Hybrid pulse width modulation-based open-winding permanent magnet synchronous motor predictive current control method

Abstract

The invention provides an open-winding permanent magnet synchronous motor predictive current control method based on hybrid pulse width modulation, and the method comprises the steps: calculating a reference voltage vector, dividing voltage vector sectors according to the reference voltage vector, and selecting a non-zero voltage vector and a zero voltage vector in each sector; according to the sector and range where the reference voltage vector is located, selecting a voltage vector which needs to be acted by the double inverters, and calculating the time when each inverter acts on the voltage vector by using a volt-second balance principle; if the action time of the selected voltage vector exceeds the control period, overmodulation processing is carried out, and the voltage vector time is redistributed; if the zero-sequence current is suppressed, the zero-sequence current and the zero-sequence voltage are suppressed by adjusting the action time of the zero-sequence voltage vector. The control performance of the open-winding permanent magnet synchronous motor can be improved, and the zero-sequence current of the open-winding permanent magnet synchronous motor can be suppressed.

Description

Technical Field

[0001] The present application relates to the technical field of open-winding permanent magnet synchronous motors, and in particular to a predictive current control method for open-winding permanent magnet synchronous motors based on hybrid pulse width modulation. Background Art

[0002] Open-winding permanent magnet synchronous motors (OW-PMSMs) are widely used in electric vehicles, electric drive systems, and high-performance motor drive control due to their simple structure, high power density, and superior efficiency. However, the control of OW-PMSMs presents a technical challenge: the generation of zero-sequence current. This zero-sequence current not only causes motor torque ripple but also affects system control accuracy and motor stability, ultimately impacting overall performance and lifespan.

[0003] Traditional OW-PMSM control methods typically rely on classic current control strategies, such as direct torque control (DTC) and field-of-control (FOC). These methods effectively control motor speed and current, but they have limitations in suppressing zero-sequence current. In dual-inverter drive systems, the generation of zero-sequence current is particularly complex, and traditional control methods struggle to effectively mitigate its negative impact. Zero-sequence current is often caused by unbalanced voltage components generated during inverter modulation. This unbalanced voltage can adversely affect motor performance, reduce control accuracy, and even cause motor overheating.

[0004] To effectively address this issue, researchers have proposed various methods to suppress zero-sequence current, such as using specialized zero-sequence current suppression algorithms, optimizing voltage vector selection, and adjusting control strategies. However, most of these methods rely on complex control algorithms or require additional hardware support, increasing system complexity and cost. Furthermore, they may not meet the actual control needs of motors with high dynamic performance requirements. Summary of the Invention

[0005] The present application aims to solve one of the technical problems in the related art at least to a certain extent.

[0006] To this end, the first objective of the present application is to propose a predictive current control method for an open-winding permanent magnet synchronous motor based on hybrid pulse width modulation.

[0007] The second objective of the present application is to propose a predictive current control device for an open-winding permanent magnet synchronous motor based on hybrid pulse width modulation.

[0008] The third objective of this application is to provide an electronic device.

[0009] The fourth object of this application is to provide a computer-readable storage medium.

[0010] A fifth object of this application is to provide a computer program product.

[0011] To achieve the above objectives, the first embodiment of the present application proposes a predictive current control method for an open-winding permanent magnet synchronous motor based on hybrid pulse width modulation, comprising:

[0012] Calculating a reference voltage vector, dividing the voltage vector into sectors according to the reference voltage vector, and selecting a non-zero voltage vector and a zero voltage vector within each sector;

[0013] Selecting the voltage vector required to be applied by the dual inverters according to the sector and range of the reference voltage vector, and calculating the time for each inverter to apply the voltage vector using the volt-second balance principle;

[0014] If the action time of the selected voltage vector exceeds the control period, overmodulation processing is performed to reallocate the voltage vector time;

[0015] If there is zero-sequence current, the zero-sequence current and voltage can be suppressed by adjusting the action time of the zero-sequence voltage vector.

[0016] Optionally, the calculating the reference voltage vector includes:

[0017] Introduce the deadbeat algorithm to calculate the components separately and The formula is:

[0018]

[0019] in, and Represents the dq0 axis reference voltage component; and Indicates the dq0 axis reference current component; L d 、L q , L0 represents the inductance component of the dq0 axis; θ e is the electrical angle; M represents the mutual inductance parameter of the three-phase winding; R represents the stator winding resistance; ω e (k) is the electrical angular velocity of the motor; ψ f With ψ f3 They represent the fundamental component and third harmonic component of the permanent magnet flux respectively;

[0020] The calculated Transform to the αβ coordinate system through inverse PARK transformation, the formula is:

[0021]

[0022] in, and Represents the reference voltage component on the αβ axis;

[0023] Calculate the reference voltage vector U ref , expressed in the αβ coordinate system as:

[0024]

[0025] Where, j is the imaginary unit;

[0026] Calculate the reference voltage vector U ref The phase angle θ1 is:

[0027]

[0028] Where θ1 is the phase angle of the reference voltage vector.

[0029] Optionally, dividing the voltage vector into sectors according to the reference voltage vector and selecting a non-zero voltage vector and a zero voltage vector in each sector includes:

[0030] The reference voltage vector U ref Mapped to the αβ coordinate system and In the figure, the voltage vector is divided into six sectors according to the phase angle θ1 of the reference voltage vector in the αβ coordinate system, and the optimal non-zero voltage vector and zero voltage vector in each sector are selected; wherein, each sector corresponds to two voltage vectors, including an odd voltage vector and an even voltage vector, and one of the three phases of the odd voltage vector is turned on and the other two phases are turned off, while one of the three phases of the even voltage vector is turned off and the other two phases are turned on.

[0031] Optionally, selecting the voltage vector required to be applied by the dual inverters according to the sector and range where the reference voltage vector is located, and calculating the time for each inverter to apply the voltage vector using the volt-second balance principle, includes:

[0032] According to the reference voltage vector U ref The phase angle θ1 and sector division in the αβ coordinate system are used to select the voltage vector required to be applied by each inverter, where each sector contains a pair of voltage vectors, the odd voltage vector is applied by inverter 1, and the odd voltage vector in the opposite direction to the even voltage vector is applied by inverter 2;

[0033] For each sector, a non-zero voltage vector and a zero voltage vector are selected to ensure that the resultant voltage vector is consistent with the reference voltage vector U ref Match;

[0034] Using the volt-second balance principle, the times t1 and t2 of the voltage vectors applied by inverter 1 and inverter 2, respectively, are calculated so that within one control cycle, the voltage vectors applied by the inverters can synthesize the required reference voltage. The calculation formulas for the times t1 and t2 are:

[0035]

[0036] Among them, U DC is the DC bus voltage, T s For the control period, u1 and u2 represent the magnitudes of the voltage vectors selected for inverter 1 and inverter 2, respectively.

[0037] Optionally, also include:

[0038] If the action time t1 and t2 are both less than the control period T s , define the open-winding permanent magnet synchronous motor system to be in state 1;

[0039] If the action time t1 is greater than the control period T s , t2 is less than the control period T s , define the open-winding permanent magnet synchronous motor system to be in state 2;

[0040] If the action time t2 is greater than the control period T s , t1 is less than the control period T s , define the open-winding permanent magnet synchronous motor system to be in state 3;

[0041] If the action time t1 and t2 are both greater than the control period T s , it is defined that the open-winding permanent magnet synchronous motor system is in state 4.

[0042] Optionally, if the action time of the selected voltage vector exceeds the control period, overmodulation processing is performed to reallocate the voltage vector time, including:

[0043] If the open-winding permanent magnet synchronous motor system is in state 2, 3 or 4, it is determined that the current system is in an overmodulation state;

[0044] When the open-winding permanent magnet synchronous motor system is in state 2, t1 is subtracted from the control period T s The difference between the two is taken as the excess duration t3, so that t′1=T s , determine the action time of the modulated voltage vector as t′1, t2 and t3;

[0045] When the open-winding permanent magnet synchronous motor system is in state 3, t2 is subtracted from the control period T s The difference between the two is taken as the excess duration t3, so that t′2=T s , determine the action time of the modulated voltage vector as t1, t′2 and t3;

[0046] When the open-winding permanent magnet synchronous motor system is in state 4, the dual inverters synthesize a voltage vector with the longest length and the same direction as the reference voltage vector. The action times t′1 and t′2 are adjusted so that the total action time meets the control cycle limit. For state 4, the overmodulated voltage vector action time is recalculated using the following formula:

[0047]

[0048] Wherein, t′1 and t′2 represent the action time of the voltage vector after the overmodulation process;

[0049] When t'1 is greater than t'2, subtract the control period T from t'1. s The difference between the two is taken as the excess duration t3, so that t′1=T s , determine the action time of the modulated voltage vector as t′1, t′2 and t3; when t′1 is less than t′2, subtract the control period T from t′2 s The difference between the two is taken as the excess duration t3, so that t′2=T s , determine the action time of the modulated voltage vector as t′1, t′2 and t3.

[0050] Optionally, if zero-sequence current exists, suppressing the zero-sequence current and voltage by adjusting the action time of the zero-sequence voltage vector includes:

[0051] Calculate the zero-sequence voltage component u of the selected voltage vectors for inverter 1 and inverter 2 0output , the formula is:

[0052] u 0output =u 0INV1 -u 0INV2

[0053] Among them, u 0INV1 with u 0INV2 Represent the zero-sequence voltage values ​​of the selected voltage vectors of inverter 1 and inverter 2 respectively;

[0054] Calculate the zero-sequence voltage output u 0output With reference zero sequence voltage The difference is:

[0055]

[0056] According to the difference Determine whether measures need to be taken to offset the zero-sequence current. If the difference is greater than 0 or less than 0, it is determined that inverter 1 or inverter 2 needs to act on the zero-sequence voltage vector U7 to offset the zero-sequence current as much as possible.

[0057] Calculate the action time t7 of the zero-sequence voltage vector U7 that needs to act, the formula is:

[0058]

[0059] in,

[0060] During the zero-sequence current suppression process, if inverter 1 or inverter 2 has excess capacity after applying the non-zero voltage vector U7, it continues to apply the zero voltage vector U0 until the control cycle ends.

[0061] To achieve the above-mentioned objectives, a second embodiment of the present application proposes a predictive current control device for an open-winding permanent magnet synchronous motor based on hybrid pulse width modulation, comprising:

[0062] a division module, configured to calculate a reference voltage vector, divide the voltage vector into sectors according to the reference voltage vector, and select a non-zero voltage vector and a zero voltage vector within each sector;

[0063] a calculation module, configured to select the voltage vector required to be applied by the dual inverters according to the sector and range where the reference voltage vector is located, and calculate the time for each inverter to apply the voltage vector using the volt-second balance principle;

[0064] An overmodulation processing module is used to perform overmodulation processing and reallocate the voltage vector time if the action time of the selected voltage vector exceeds the control period;

[0065] The suppression module is used to suppress the zero-sequence current and voltage by adjusting the action time of the zero-sequence voltage vector if there is zero-sequence current.

[0066] To achieve the above-mentioned purpose, a third embodiment of the present application provides an electronic device, comprising: a processor, and a memory communicatively connected to the processor;

[0067] The memory stores computer-executable instructions;

[0068] The processor executes the computer-executable instructions stored in the memory to implement the method as described in any one of the first aspects above.

[0069] To achieve the above-mentioned purpose, the fourth embodiment of the present application proposes a computer-readable storage medium, in which computer-executable instructions are stored. When the computer-executable instructions are executed by a processor, they are used to implement the method as described in any one of the above-mentioned first aspects.

[0070] To achieve the above-mentioned purpose, the fifth embodiment of the present application proposes a computer program product, including a computer program, which, when executed by a processor, implements the method as described in any one of the above-mentioned first aspects.

[0071] The technical solutions provided by the embodiments of this application bring at least the following beneficial effects:

[0072] By utilizing the synergistic effect of the dual inverters and accurately selecting and allocating the voltage vector, the reference voltage vector can be accurately synthesized within the control cycle, thereby effectively improving the control performance of the open-winding permanent magnet synchronous motor (OW-PMSM); by optimizing the selection of the voltage vector and the distribution of the action time, the voltage imbalance caused by the zero-sequence current can be reduced while ensuring the output power of the motor, thereby significantly reducing the negative impact of the zero-sequence current on the motor performance; the volt-second balance principle is used to calculate the voltage vector action time, making the synthesis of the voltage vector more accurate and avoiding the voltage component mismatch problem common in traditional methods. By reasonably allocating the voltage vector time of the inverter, this method can achieve more efficient power conversion and improve the overall efficiency of the motor drive system; in terms of suppressing zero-sequence current, through the precise control of the zero-sequence voltage, the system can effectively suppress the generation of zero-sequence current during dynamic changes, avoiding the motor torque pulsation and system instability caused by zero-sequence current; in addition, by adjusting the action time of the zero-sequence voltage vector, the influence of the zero-sequence current is further reduced, and the working stability and reliability of the motor are improved. In summary, this application not only solves the problem that traditional control strategies cannot effectively suppress zero-sequence current, but also optimizes the control performance of the motor, reduces the complexity of the system, and has strong practical value and application prospects.

[0073] Additional aspects and advantages of the present application will be given in part in the description below, and in part will become apparent from the description below, or will be learned through practice of the present application. BRIEF DESCRIPTION OF THE DRAWINGS

[0074] The above and / or additional aspects and advantages of the present application will become apparent and easily understood from the following description of the embodiments in conjunction with the accompanying drawings, in which:

[0075] Figure 1 A simplified flow chart of a method for predictive current control of an open-winding permanent magnet synchronous motor based on hybrid pulse width modulation provided in an embodiment of the present application;

[0076] Figure 2 A control block diagram of a method for predictive current control of an open-winding permanent magnet synchronous motor based on hybrid pulse width modulation provided in an embodiment of the present application;

[0077] Figure 3 A detailed flow chart of a method for predictive current control of an open-winding permanent magnet synchronous motor based on hybrid pulse width modulation provided in an embodiment of the present application;

[0078] Figure 4A schematic diagram of a sector plane divided according to an embodiment of the present application;

[0079] Figure 5 A schematic diagram of the reference voltage vector distribution provided in an embodiment of the present application;

[0080] Figure 6 A schematic diagram of the duty cycle synthesis range provided in an embodiment of the present application;

[0081] FIG7( a ) is a schematic diagram of a method for calculating the action time of a non-zero voltage vector in sector ① provided in an embodiment of the present application;

[0082] FIG7( b ) is an enlarged diagram of the detailed process of calculating the action time of sector ① provided in an embodiment of the present application;

[0083] Figure 8 A schematic diagram of the division of flag bits for overmodulation conditions within a sector provided by an embodiment of the present application;

[0084] Figure 9 This is a schematic diagram of the overmodulation processing in state 4 provided in an embodiment of the present application. DETAILED DESCRIPTION

[0085] The following describes in detail embodiments of the present application. Examples of the embodiments are shown in the accompanying drawings, wherein the same or similar reference numerals throughout represent the same or similar elements or elements having the same or similar functions. The embodiments described below with reference to the accompanying drawings are exemplary and are intended to be used to explain the present application, and should not be construed as limiting the present application.

[0086] Due to their high DC bus utilization and strong fault tolerance, open-winding permanent magnet synchronous motors (OW-PMSMs) have been widely used in new energy vehicles, military industry, industrial automation, and other fields. Due to their advantages mentioned above, OW-PMSMs have attracted widespread attention in the motor control field. Their design is similar to that of classic PMSMs: the neutral point of the stator winding of a conventional motor is opened and an inverter is connected at both ends.

[0087] Open-winding permanent magnet synchronous motor systems can be divided into two types: one is the isolated DC bus type, and the other is the common DC bus type. This classification is based on the different ways of providing DC power. The isolated DC bus type is powered by two voltage power supplies, while the common DC bus type is only powered by one voltage power supply. However, there is a zero-sequence loop in the common DC bus open-winding permanent magnet synchronous motor system. At this time, there is a zero-sequence current (ZSC) in the system, which makes the sum of the three-phase currents not zero. This is an inherent characteristic of the common DC bus type. For the control system, the zero-sequence current may cause current and torque ripple, resulting in problems such as motor speed loss. Therefore, it is very necessary to suppress the zero-sequence current in the system.

[0088] Many methods have been proposed to suppress zero-sequence current. For example, a new space vector pulse width modulation (SVPWM) strategy was proposed in one paper. By considering the voltage vectors on the α, β, and zero axes, it expands the modulation range of the OW-PMSM while suppressing zero-sequence current. For example, a new SVPWM scheme was proposed in another paper to select the optimal switching combination to eliminate common-mode voltage (CMV) variations and suppress zero-sequence current across the entire frequency range. Experimental verification demonstrated the results of reducing ZSC. For example, another paper proposed an SVPWM strategy based on a ZSC hysteretic controller. Basic voltage vectors with ZSC polarities opposite to the zero-sequence voltage polarity were selected for reference voltage vector synthesis. High switching frequency is also a problem that needs to be addressed. For example, another paper proposed a solution to reduce ZSC. This method has a low switching frequency. To achieve ZSC suppression, only the zero voltage vector and three non-zero voltage vectors with the same ZSV components are used. A torque controller is also used in the system to reduce torque ripple. For example, another paper proposed a method that not only suppresses ZSC but also addresses the issues caused by high switching frequency. In this scheme, the torque component generated by the q-axis current can offset the torque ripple caused by the ZSC. To improve post-fault torque control performance, a new robust fault-tolerant control strategy has been proposed. Furthermore, a new modulation method has been proposed to improve the voltage vector synthesis performance of the OW-PMSM. By keeping the low-voltage inverter in a low-potential clamped state throughout the control period, the voltage synthesis range of the high-voltage inverter is expanded, thereby covering a wider voltage synthesis range.

[0089] Model predictive control (MPC) has gained recognition in recent years for its impressive control performance. MPC offers several advantages, including its fast dynamic performance, ability to accommodate constraints, and its optimization and prediction capabilities. MPC can be categorized into two types: continuous control set MPC (CCS-MPC) and finite control set MPC (FCS-MPC). Deadbeat predictive control (DBPC) is a typical example of CCS-MPC, as used in the literature mentioned above. The aforementioned control schemes all employ SVPWM, an advanced motor drive PWM technique that optimizes the voltage vector for high efficiency and low harmonic distortion. It uses PWM to convert the desired voltage into an inverter switching sequence. However, simultaneously optimizing multiple parameters in the speed and current loop proportional-integral (PI) controllers is quite difficult, and slow dynamic performance is also a problem that needs improvement.

[0090] Compared to CCS-MPC, FCS-MPC can incorporate nonlinearities and constraints into the cost function, making it widely applicable. To date, in-depth exploration of FCS-MPC continues unabated. A paper proposed an improved model predictive current control (MPCC) method. This method first estimates the back electromotive force (EMF) and then uses it to predict the current. To improve the steady-state control performance of the system, a nonzero voltage vector and a zero voltage vector are selected to form a duty cycle modulation, and the duration of the nonzero voltage vector's action is calculated. Computational burden is also a problem faced by FCS-MPC. To address this issue, a paper proposed a control scheme that eliminates redundant trigonometric function calculations and coordinate transformations, thus eliminating redundant computations. A cost function is used to select the voltage vector that minimizes the voltage error. Experimental results demonstrate the superiority of this scheme by comparing it to another scheme applied to two-vector MPCC. A paper introduced a new MPC method based on dead-zone vectors. The impact of the dead zone on MPC was first analyzed, and voltage vectors that are beneficial and non-beneficial to the dead zone were identified. The differences between these two vectors were analyzed, and the feasibility of the method was verified through experiments.

[0091] In OW-PMSMs, dual inverters offer significant control flexibility due to their wider modulation range. To improve the control performance of the motor system, accurate voltage vector synthesis and zero-sequence current suppression are both necessary, and a cost function can be used to achieve this. A new model-predictive dual-objective current control (MPTCC) approach was proposed in a paper. This approach, based on the basic MPCC method, utilizes a new cost function to provide a more accurate voltage vector. Another paper proposes a method to improve the steady-state control performance of a single voltage vector MPCC by considering the effects of the dead zone and recalculating the dead-zone voltage vector duration. The cost function also includes the dq0 axis current to obtain the voltage vector that minimizes the cost function. Researchers have also focused on the issue of ZSC and voltage, which is caused not only by the third-harmonic back EMF but also by the dual inverter. Therefore, a paper proposes a strategy to estimate the rotor position based on a model reconstructed using the third-harmonic back EMF and ZSC to improve control effectiveness. A paper proposes a new pulse-width modulation method that eliminates ZSV and suppresses ZSC through CMV. The zero-sequence controller design in the paper effectively suppresses ZSC. Considering that high switching frequency leads to increased switching losses, a paper proposes an MPC method to suppress ZSC and optimize the switching frequency. Unlike traditional approaches, the proposed three-dimensional space vector method is performed in an ABC coordinate system. A paper also proposes a hybrid vector MPC method that uses a fast voltage vector selection strategy to select a non-zero voltage vector, which is then applied by inverter 1. A four-stage control strategy is then used to control inverter 2.

[0092] The above control schemes or methods can achieve good control effects. However, the OW-PMSM needs to suppress the zero-sequence current while maintaining good control performance.

[0093] In order to solve the above problems, the embodiment of the present application provides a method for predictive current control of an open-winding permanent magnet synchronous motor based on hybrid pulse width modulation. Control, in order to suppress the zero sequence current, Also set to 0. It can be obtained from the output of the PI controller. This method considers three parts: the voltage vector selection process of the dual inverter, the time allocation of each selected voltage vector, and the necessary part: the suppression of zero-sequence voltage and current.

[0094] For the convenience of calculation, in the embodiment of the present application, the voltage equation of the classic PMSM in the dq rotating coordinate system can be written as:

[0095]

[0096] Although the voltage equation of the common DC bus OW-PMSM in dq rotating coordinates is the same as that of the classic PMSM, the zero-sequence loop should be considered:

[0097]

[0098] Among them, u d 、u q and u0 represent the stator voltage components of the dq0 axis; i d 、i q and i0 represent the stator current components of the dq0 axis; L d 、L q , L0 represents the inductance component of the dq0 axis. Since the permanent magnet synchronous motor is a surface mount type, L d =L q ; R represents the stator winding resistance; ω e is the motor electrical angular velocity; θ e is the electrical angle; ψ f With ψ f3 They represent the fundamental component and third harmonic component of the permanent magnet flux respectively.

[0099] When predictive control is actually applied, the digital control system has a delay, which causes the voltage vector selected at the current sampling moment to not be updated to the motor in time. To solve this problem, the effect of the one-step delay should be offset by a one-step prediction process:

[0100]

[0101] Among them, i d (k), i q (k), i0(k) represents the stator current sampling of the dq0 axis at time k, e0(k)=3ω e (k)ψ f3 sin(3θ) is the third harmonic back EMF.

[0102] Taking into account the influence of the third harmonic, the electromagnetic torque equation of the OW-PMSM on the dq0 axis can be written as:

[0103]

[0104] Among them, n p Represents the number of pole pairs of the motor.

[0105] Figure 1 、 2 and Figure 3 The present invention provides a schematic diagram and block diagram of a method for predicting current control of an open-winding permanent magnet synchronous motor based on hybrid pulse width modulation. Figure 1 、 Figure 2 and Figure 3, the method comprises the following steps:

[0106] Step 101 : Calculate a reference voltage vector, divide the voltage vector into sectors according to the reference voltage vector, and select a non-zero voltage vector and a zero voltage vector in each sector.

[0107] In the embodiment of the present application, when hybrid pulse width modulation is used, in each control period, within a certain range, each inverter acts on a non-zero voltage vector and a zero voltage vector, and the reference voltage vector U is synthesized by duty cycle control. ref When the range is exceeded, the excess voltage vector will be acted upon by inverter 2 (inverter 1), and then the purpose of suppressing ZSC is achieved according to the needs of the system.

[0108] In order to accurately determine the reference voltage vector U ref The position in the coordinate system is calculated by introducing a deadbeat algorithm in this embodiment of the application, and the formula is:

[0109]

[0110] in, and Represents the dq0 axis reference voltage component; and Indicates the dq0 axis reference current component; L d 、L q , L0 represents the inductance component of the dq0 axis; θ e is the electrical angle; M represents the mutual inductance parameter of the three-phase winding; R represents the stator winding resistance; ω e (k) is the electrical angular velocity of the motor; ψ f With ψ f3 They represent the fundamental component and third harmonic component of the permanent magnet flux respectively.

[0111] Since the sector division of the reference voltage vector is carried out in the αβ coordinate system, it is necessary to convert the calculated Transform to the αβ coordinate system through inverse PARK transformation, the formula is:

[0112]

[0113] in, and Represents the reference voltage component on the αβ axis.

[0114] It can be understood that in the αβ coordinate system, the α axis represents the real axis and the β axis represents the imaginary axis, and the vector sum of the two is the reference voltage vector. ref Expressed as:

[0115]

[0116] Where j is the imaginary unit.

[0117] It should be noted that when U is obtained according to the above formula ref In order to select the voltage vector more accurately and conveniently, it is necessary to further determine its phase angle on the αβ axis, and thus the reference voltage vector U ref The calculation formula of the phase angle θ1 is:

[0118]

[0119] Where θ1 is the phase angle of the reference voltage vector.

[0120] After the reference voltage vector is calculated, the embodiment of the present application further converts the reference voltage vector U ref Mapped to the αβ coordinate system and In the figure, the voltage vector is divided into six sectors according to the phase angle θ1 of the reference voltage vector in the αβ coordinate system, and the optimal non-zero voltage vector and zero voltage vector in each sector are selected; wherein, each sector corresponds to two voltage vectors, including an odd voltage vector and an even voltage vector, and one of the three phases of the odd voltage vector is turned on and the other two phases are turned off, while one of the three phases of the even voltage vector is turned off and the other two phases are turned on.

[0121] In a possible embodiment, the αβ axis coordinate system is divided into sectors, with a total of 6 sectors, such as Figure 4 As shown. Table 1 lists the angular ranges of the six sectors, and their corresponding two adjacent voltage vectors and their switching states. Each sector has an odd voltage vector and an even voltage vector. One of the three phases of each odd voltage vector is turned on, and the other two phases are turned off. On the contrary, the even voltage vector has one of the three phases of each even voltage vector turned off, and the other two phases are turned on, as shown in Table 1. Table 2 shows the voltage components of each voltage vector in the αβ0 coordinate system. There are not only 6 non-zero voltage vectors in the table, but also 2 zero voltage vectors. Unlike U0, which has a component of 0 for the αβ0 three axes, U7 has a component of U for the 0 axis. DC .

[0122] Table 1 Sector and voltage vector

[0123] <![CDATA[Reference voltage vector angle θ1]]> Sector number <![CDATA[Odd voltage vector U x > <![CDATA[Even voltage vector U y > (0,π / 3] ① <![CDATA[U1(100)]]> <![CDATA[U2(110)]]> (π / 3,2π / 3] ② <![CDATA[U3(010)]]> <![CDATA[U2(110)]]> (2π / 3,π] ③ <![CDATA[U3(010)]]> <![CDATA[U4(011)]]> (π,4π / 3] ④ <![CDATA[U5(001)]]> <![CDATA[U4(011)]]> (4π / 3,5π / 3] ⑤ <![CDATA[U5(001)]]> <![CDATA[U6(101)]]> (5π / 3,2π] ⑥ <![CDATA[U1(100)]]> <![CDATA[U6(101)]]>

[0124] Table 2 Axis components of voltage vector

[0125]

[0126]

[0127] Since the OW-PMSM has two inverters in operation, the voltage of the motor is the voltage of inverter 1 minus the voltage of inverter 2, which is expressed as:

[0128] U ref =u inv1 -u inv2

[0129] In a specific area of ​​each sector, each inverter can accurately synthesize the reference voltage U at the current moment by selecting and applying a non-zero voltage vector and a zero voltage vector. ref Specific red areas such as Figure 6 As shown. Taking sector ① as an example, the optimal synthetic voltage for synthesizing the reference voltage is U1 and U′5, which are respectively acted by inverter 1 and inverter 2. At this time, the reference voltage U ref It is expressed as U1-U′5, which is equivalent to U1+U′2. If the voltage vectors U1 and U′5 are respectively applied by inverter 1 and inverter 2 for a whole control period T s , you can synthesize the longest diagonal of the parallelogram, such as Figure 5 and Figure 6 shown.

[0130] Step 102 : Select the voltage vector required to be applied by the dual inverters according to the sector and range where the reference voltage vector is located, and calculate the time for each inverter to apply the voltage vector using the volt-second balance principle.

[0131] In the above steps, the U ref After obtaining the non-zero voltage vectors, further calculation is required to determine the action time of these voltage vectors, so as to more accurately synthesize the reference voltage vector U ref .

[0132] In this embodiment of the present application, the action time t1 and t2 of two non-zero voltage vectors will be calculated. In order to synthesize the reference voltage vector U ref ,Each inverter needs to select a suitable voltage vector. The odd voltage vector is chosen because its zero-sequence voltage component is low, as shown in the following table:

[0133] Table 3 Voltage vector of each sector participating in the synthesis of reference voltage

[0134] Sector number Inverter 1 Inverter 2 ① <![CDATA[U1(000)]]> <![CDATA[U′5(001)]]> ② <![CDATA[U3(010)]]> <![CDATA[U′5(001)]]> ③ <![CDATA[U3(010)]]> <![CDATA[U′1(100)]]> ④ <![CDATA[U5(001)]]> <![CDATA[U′1(100)]]> ⑤ <![CDATA[U5(001)]]> <![CDATA[U′3(010)]]> ⑥ <![CDATA[U1(100)]]> <![CDATA[U′3(010)]]>

[0135] In one embodiment of the present application, sector ① is taken as an example. In order to synthesize U as shown in FIG7 (a) and (b), ref , U1 and U′5 are needed. Because the formula is U1+U′2, the triangle is based on the reference voltage U refAssume that the vertices of the triangle are O, Z, and Y (as shown in Figure 7), and draw a vertical line with line segment OZ, intersecting point T. Let the length of line segment YT be a and the length of line segment TZ be x. The length of modern OZ represents U ref The length of line segment OT is U ref -x, the angle ∠YZO is θ1, and the angle ∠YOZ is 60°-θ1.

[0136] Referring to FIG7, based on the existing conditions, the following formula can be obtained through trigonometric calculation:

[0137]

[0138] From this, the value of x can be calculated as:

[0139]

[0140] according to:

[0141]

[0142] u1 and u2 can be calculated as:

[0143]

[0144] According to the volt-second balance principle:

[0145]

[0146] The times t1 and t2 of the voltage vectors applied by inverter 1 and inverter 2, respectively, can be calculated as:

[0147]

[0148] Among them, U DC is the DC bus voltage, T s For the control period, u1 and u2 represent the magnitudes of the voltage vectors selected for inverter 1 and inverter 2, respectively.

[0149] According to the above process, the reference voltage vector U is calculated ref , the action time t1 and t2 of the voltage vector selected in each sector are obtained.

[0150] Step 103: If the action time of the selected voltage vector exceeds the control period, overmodulation processing is performed to reallocate the voltage vector time.

[0151] In actual situations, it will appear Figure 6Overmodulation outside the red area. When overmodulation occurs, it may be that the duration calculated by the above formula is longer than the control period Ts. This means that even if the entire control period is applied, the resulting voltage component still does not meet the system requirements. Therefore, the area outside the red area is divided into three parts. Including the red area, there are a total of four parts. Take sector ① as an example:

[0152] exist Figure 8 In the state 2, state 3 and state 4 are all modulation areas. Specifically, if the action time t1 and t2 are both less than the control period T s , define the open-winding permanent magnet synchronous motor system in state 1; if the action time t1 is greater than the control period T s , t2 is less than the control period T s , define the open-winding permanent magnet synchronous motor system in state 2; if the action time t2 is greater than the control period T s , t1 is less than the control period T s , define the open-winding permanent magnet synchronous motor system in state 3; if the action time t1 and t2 are both greater than the control period T s , it is defined that the open-winding permanent magnet synchronous motor system is in state 4.

[0153] Reference Figure 3 When the reference voltage is in state 2 or state 3, in this case, the redundant voltage vector that should have been applied by inverter 1 (inverter 2) will be applied by inverter 2 (inverter 1) instead. The redundant voltage vector will be applied by another inverter with the same amplitude and length but opposite direction. When the open-winding permanent magnet synchronous motor system is in state 2, t1 is subtracted from the control period T s The difference between the two is taken as the excess duration t3, so that t′1=T s , determine the modulated voltage vector action time as t′1, t2 and t3; when the open-winding permanent magnet synchronous motor system is in state 3, subtract the control period T from t2 s The difference between the two is taken as the excess duration t3, so that t′2=T s , determine the action time of the modulated voltage vector as t1, t′2 and t3.

[0154] When the situation is state 4, it means that both t1 and t2 are greater than the control period T s This situation mainly occurs in the dynamic process of the system. At this time, the dual inverter will synthesize a voltage vector with the longest length and the same direction as the reference voltage vector, intersecting with the line segment IG, as shown in Figure 1. Figure 9 shown.

[0155] From this we can deduce:

[0156]

[0157] Wherein, t′1 and t′2 represent the action time of the voltage vector after the overmodulation process;

[0158] When t'1 is greater than t'2, subtract the control period T from t'1. s The difference between the two is taken as the excess duration t3, so that t′1=T s , determine the action time of the modulated voltage vector as t′1, t′2 and t3; when t′1 is less than t′2, subtract the control period T from t′2 s The difference between the two is taken as the excess duration t3, so that t′2=T s , determine the action time of the modulated voltage vector as t′1, t′2 and t3.

[0159] Step 104 : To suppress the zero-sequence current, the zero-sequence current and voltage are suppressed by adjusting the action time of the zero-sequence voltage vector.

[0160] It's understandable that the above steps calculate the duration of the voltage vector acting on the two inverters, but don't consider the suppression of zero-sequence current. This is an unavoidable problem, as zero-sequence current can cause adverse effects such as torque ripple, leading to poor control performance of the motor control system. Therefore, in this step, the present embodiment will discuss in detail the key processes for suppressing ZSC and ZSV.

[0161] It can be concluded from Table 2 that the six non-zero voltage vectors and U7 all have zero-sequence components, while the zero-sequence component of U0 is 0. Therefore, this characteristic can be used to suppress the zero-sequence current.

[0162] In the embodiment of the present application, the zero-sequence voltage component u of the voltage vector selected by inverter 1 and inverter 2 is first calculated. 0output , the formula is:

[0163] u 0output =u 0INV1 -u 0INV2

[0164] Among them, u 0INV1 with u 0INV2 Represent the zero-sequence voltage values ​​of the selected voltage vectors of inverter 1 and inverter 2 respectively.

[0165] In addition, the following formula holds:

[0166]

[0167] The zero-sequence current is mainly generated by the zero-sequence voltage source generated by the inverter modulation, the resistance, the zero-sequence inductance and the third harmonic back electromotive force. At this time, the zero-sequence voltage generated by the modulation of the two inverters is The same is the best case. As shown in the following formula:

[0168]

[0169] However, the above formula is an ideal situation. There are three real situations:

[0170]

[0171] In the embodiment of the present application, it is necessary to Determine whether measures need to be taken to offset the zero-sequence current. If the difference is greater than 0 or less than 0, it is determined that inverter 1 or inverter 2 needs to act on the zero-sequence voltage vector U7 to offset the zero-sequence current as much as possible.

[0172] The inverter only uses the zero-sequence voltage u generated by the non-zero voltage vector 0output Subtract the reference value of the zero-sequence voltage It is expressed as u′0, which is also the value that needs to be offset by the action of U7. The action time t7 of the zero-sequence voltage vector U7 that needs to act is calculated from this formula:

[0173]

[0174] in,

[0175] It should also be noted that since u′0 can be positive or negative, U7 can be the function of inverter 1 or inverter 2. (If it is inverter 1, it is U7; if it is inverter 2, it is U7′.)

[0176] Based on the above situation, when an overmodulation condition occurs, the system outputs the following zero-sequence current:

[0177]

[0178] Obtain u through the above formula 0output , and t7 will be calculated according to the above formula. When the system is in the overmodulation state, the method of suppressing the zero-sequence current will be slightly different.

[0179] When the situation is state 2 and When the zero sequence current suppression method is the same as that in case 1 Same, at this time the action time t1 of inverter 1 is greater than the control period T s ,For inverter 2, in addition to the durations t2 and t3, there is also excess capacity to continue to act on the zero voltage vector, including U0′ and U7′, so as to achieve the purpose of suppressing zero sequence current. If the situation is state 2 and At this point, inverter 1 needs to compensate for the zero-sequence voltage, but it needs to apply a non-zero voltage vector throughout the entire cycle, leaving no excess capacity for a zero-voltage vector. If the non-zero voltage vectors applied by both inverters are changed from the initially selected odd voltage vector to an even voltage vector, inverter 2 will apply U4′ for the entire cycle, while inverter 1 will apply U1 and U2. Inverter 1 now has excess capacity for the zero-sequence voltage. After calculation, all of inverter 1's remaining capacity is applied to U7. Finally, the zero-sequence voltage obtained by the dual inverters is consistent with the odd vector case. Therefore, either odd or even voltage vectors can be applied at this time without affecting system performance.

[0180] In addition, during the zero-sequence current suppression process, if inverter 1 or inverter 2 has excess capacity after applying the non-zero voltage vector U7, it continues to apply the zero voltage vector U0 until the control cycle ends.

[0181] Furthermore, to verify the superiority and feasibility of the proposed solution, a comparative experiment was conducted on a 1.25kW OW-PMSM control platform to verify the traditional solution and the proposed solution. The experimental results show that the proposed hybrid pulse width modulation solution is superior, fully suppressing zero-sequence current while improving system control performance.

[0182] In order to implement the above embodiment, the present application further proposes a predictive current control device for an open-winding permanent magnet synchronous motor based on hybrid pulse width modulation, the device comprising:

[0183] a partitioning module, configured to calculate a reference voltage vector, partition the voltage vector into sectors according to the reference voltage vector, and select a non-zero voltage vector and a zero voltage vector within each sector;

[0184] A calculation module is used to select the voltage vector required to be applied by the dual inverters according to the sector and range where the reference voltage vector is located, and calculate the time for each inverter to apply the voltage vector using the volt-second balance principle;

[0185] An overmodulation processing module is used to perform overmodulation processing and reallocate the voltage vector time if the action time of the selected voltage vector exceeds the control period;

[0186] The suppression module is used to suppress the zero-sequence current and voltage by adjusting the action time of the zero-sequence voltage vector if there is zero-sequence current.

[0187] Regarding the apparatus in the above embodiment, the specific manner in which each module performs operations has been described in detail in the embodiment of the method, and will not be elaborated here.

[0188] In order to implement the above embodiments, the present application also proposes an electronic device, comprising: a processor, and a memory communicatively connected to the processor; the memory stores computer-executable instructions; the processor executes the computer-executable instructions stored in the memory to implement the method provided by the above embodiments.

[0189] In order to implement the above embodiments, the present application also proposes a computer-readable storage medium, in which computer-executable instructions are stored. When the computer-executable instructions are executed by a processor, they are used to implement the methods provided by the above embodiments.

[0190] In order to implement the above embodiments, the present application also proposes a computer program product, including a computer program, which implements the methods provided by the above embodiments when executed by a processor.

[0191] The collection, storage, use, processing, transmission, provision and disclosure of user personal information involved in this application are in compliance with relevant laws and regulations and do not violate public order and good morals.

[0192] It is important to note that personal information collected from users should be used for legitimate and reasonable purposes and should not be shared or sold beyond these legitimate uses. Furthermore, such collection / sharing should be conducted only after receiving the user's informed consent, including but not limited to notifying the user to read the user agreement / user notice and sign an agreement / authorization that includes the relevant user information before using the feature. Furthermore, any necessary steps must be taken to safeguard and secure access to such personal information and ensure that others with access to personal information comply with its privacy policy and procedures.

[0193] This application contemplates providing implementations that allow users to selectively block the use or access of personal information data. Specifically, this disclosure contemplates providing hardware and / or software to prevent or block access to such personal information data. Risks can be minimized by limiting data collection and deleting data once it is no longer needed. Furthermore, where applicable, such personal information can be de-identified to protect user privacy.

[0194] In the descriptions of the foregoing embodiments, the reference terms "one embodiment", "some embodiments", "example", "specific example", or "some examples" mean that the specific features, structures, materials or characteristics described in conjunction with the embodiment or example are included in at least one embodiment or example of the present application. In this specification, the schematic expressions of the above terms do not necessarily refer to the same embodiment or example. Moreover, the specific features, structures, materials or characteristics described may be combined in any one or more embodiments or examples in a suitable manner. In addition, those skilled in the art may combine and combine the different embodiments or examples described in this specification and the features of the different embodiments or examples, unless they are mutually inconsistent.

[0195] Furthermore, the terms "first" and "second" are used for descriptive purposes only and should not be construed as indicating or implying relative importance or implicitly specifying the number of the technical features being referred to. Thus, a feature defined as "first" or "second" may explicitly or implicitly include at least one of such features. Throughout the description of this application, "plurality" means at least two, for example, two, three, etc., unless otherwise specifically defined.

[0196] Any process or method description in a flowchart or otherwise described herein may be understood to represent a module, segment or portion of code comprising one or more executable instructions for implementing the steps of a custom logical function or process, and the scope of the preferred embodiments of the present application includes alternative implementations in which functions may be performed out of the order shown or discussed, including performing functions in a substantially simultaneous manner or in the reverse order depending on the functions involved, which should be understood by those skilled in the art to which the embodiments of the present application belong.

[0197] The logic and / or steps represented in the flowcharts or otherwise described herein, for example, can be considered as a sequenced list of executable instructions for implementing the logical functions, and can be embodied in any computer-readable medium for use by, or in conjunction with, an instruction execution system, apparatus, or device (e.g., a computer-based system, a system including a processor, or other system that can fetch and execute instructions from an instruction execution system, apparatus, or device). For purposes of this specification, a "computer-readable medium" can be any device that can contain, store, communicate, propagate, or transport a program for use by, or in conjunction with, an instruction execution system, apparatus, or device. More specific examples (a non-exhaustive list) of computer-readable media include the following: an electrical connection with one or more wires (electronic devices), a portable computer disk cartridge (magnetic device), random access memory (RAM), read-only memory (ROM), erasable and programmable read-only memory (EPROM or flash memory), fiber optic devices, and a portable compact disc read-only memory (CDROM). Furthermore, the computer-readable medium may even be paper or other suitable medium on which the program is printed, since the program may be obtained electronically, for example, by optically scanning the paper or other medium and then editing, interpreting or processing it in another suitable manner if necessary, and then storing it in a computer memory.

[0198] It should be understood that various parts of the present application can be implemented using hardware, software, firmware, or a combination thereof. In the above embodiments, multiple steps or methods can be implemented using software or firmware stored in a memory and executed by a suitable instruction execution system. For example, if implemented using hardware, as in another embodiment, any one of the following technologies known in the art or a combination thereof can be used to implement: a discrete logic circuit having a logic gate circuit for implementing a logic function on a data signal, an application-specific integrated circuit having a suitable combination of logic gate circuits, a programmable gate array (PGA), a field programmable gate array (FPGA), etc.

[0199] Those skilled in the art will understand that all or part of the steps in the method of the above embodiment can be completed by instructing related hardware through a program, and the program can be stored in a computer-readable storage medium. When the program is executed, it includes one or a combination of the steps of the method embodiment.

[0200] In addition, the functional units in the various embodiments of the present application may be integrated into a processing module, or each unit may exist physically separately, or two or more units may be integrated into a module. The above-mentioned integrated module may be implemented in the form of hardware or in the form of a software functional module. If the integrated module is implemented in the form of a software functional module and sold or used as an independent product, it may also be stored in a computer-readable storage medium.

[0201] The storage medium mentioned above may be a read-only memory, a magnetic disk, or an optical disk, etc. Although the embodiments of the present application have been shown and described above, it is understood that the above embodiments are exemplary and should not be construed as limiting the present application. Persons skilled in the art may make changes, modifications, substitutions, and variations to the above embodiments within the scope of the present application.

[0202] It should be understood that the various forms of the processes shown above can be used to reorder, add, or delete steps. For example, the steps described in this application can be performed in parallel, sequentially, or in a different order, as long as the desired results of the technical solution of this application can be achieved. This is not limited herein.

[0203] The above specific embodiments do not constitute a limitation on the scope of protection of this application. Those skilled in the art will appreciate that various modifications, combinations, sub-combinations, and substitutions may be made based on design requirements and other factors. Any modifications, equivalent substitutions, and improvements made within the spirit and principles of this application shall be included within the scope of protection of this application.

Claims

1. A predictive current control method for an open-winding permanent magnet synchronous motor based on hybrid pulse width modulation, characterized in that: include: Calculating a reference voltage vector, dividing the voltage vector into sectors according to the reference voltage vector, and selecting a non-zero voltage vector and a zero voltage vector within each sector; Selecting the voltage vector required to be applied by the dual inverters according to the sector and range of the reference voltage vector, and calculating the time for each inverter to apply the voltage vector using the volt-second balance principle; If the action time of the selected voltage vector exceeds the control period, overmodulation processing is performed to reallocate the voltage vector time; In order to suppress the zero-sequence current, the action time of the zero-sequence voltage vector is adjusted to suppress the zero-sequence current and voltage.

2. The method according to claim 1, characterized in that The calculating of the reference voltage vector includes: Introduce the deadbeat algorithm to calculate the components separately and The formula is: in, and Represents the dq0 axis reference voltage component; and Indicates the dq0 axis reference current component; L d , L q , L0 represents the inductance component of the dq0 axis; θ e is the electrical angle; M represents the mutual inductance parameter of the three-phase winding; R represents the stator winding resistance; ω e (k) is the electrical angular velocity of the motor; ψ f With ψ f3 They represent the fundamental component and third harmonic component of the permanent magnet flux respectively; The calculated Transform to the αβ coordinate system through inverse PARK transformation, the formula is: in, and Represents the reference voltage component on the αβ axis; Calculate the reference voltage vector U ref , expressed in the αβ coordinate system as: Where, j is the imaginary unit; Calculate the reference voltage vector U ref The phase angle θ1 is: Where θ1 is the phase angle of the reference voltage vector.

3. The method according to claim 2, characterized in that The step of dividing the voltage vector into sectors according to the reference voltage vector and selecting a non-zero voltage vector and a zero voltage vector in each sector includes: The reference voltage vector U ref Mapped to the αβ coordinate system and In the figure, the voltage vector is divided into six sectors according to the phase angle θ1 of the reference voltage vector in the αβ coordinate system, and the optimal non-zero voltage vector and zero voltage vector in each sector are selected; wherein, each sector corresponds to two voltage vectors, including an odd voltage vector and an even voltage vector, and one of the three phases of the odd voltage vector is turned on and the other two phases are turned off, while one of the three phases of the even voltage vector is turned off and the other two phases are turned on.

4. The method according to claim 3, characterized in that The step of selecting the voltage vector required to be applied by the dual inverters according to the sector and range where the reference voltage vector is located, and calculating the time for each inverter to apply the voltage vector using the volt-second balance principle, includes: According to the reference voltage vector U ref The phase angle θ1 and sector division in the αβ coordinate system are used to select the voltage vector required to be applied by each inverter, where each sector contains a pair of voltage vectors, the odd voltage vector is applied by inverter 1, and the odd voltage vector in the opposite direction to the even voltage vector is applied by inverter 2; For each sector, a non-zero voltage vector and a zero voltage vector are selected to ensure that the resultant voltage vector is consistent with the reference voltage vector U ref Match; Using the volt-second balance principle, the times t1 and t2 of the voltage vectors applied by inverter 1 and inverter 2, respectively, are calculated so that within one control cycle, the voltage vectors applied by the inverters can synthesize the required reference voltage. The calculation formulas for the times t1 and t2 are: Among them, U DC is the DC bus voltage, T s For the control period, u1 and u2 represent the magnitudes of the voltage vectors selected for inverter 1 and inverter 2, respectively.

5. The method according to claim 4, characterized in that Also includes: If the action time t1 and t2 are both less than the control period T s , define the open-winding permanent magnet synchronous motor system to be in state 1; If the action time t1 is greater than the control period T s , t2 is less than the control period T s , define the open-winding permanent magnet synchronous motor system to be in state 2; If the action time t2 is greater than the control period T s , t1 is less than the control period T s , define the open-winding permanent magnet synchronous motor system to be in state 3; If the action time t1 and t2 are both greater than the control period T s , it is defined that the open-winding permanent magnet synchronous motor system is in state 4.

6. The method according to claim 5, characterized in that If the action time of the selected voltage vector exceeds the control period, overmodulation processing is performed to reallocate the voltage vector time, including: If the open-winding permanent magnet synchronous motor system is in state 2, 3 or 4, it is determined that the current system is in an overmodulation state; When the open-winding permanent magnet synchronous motor system is in state 2, t1 is subtracted from the control period T s The difference between the two is taken as the excess duration t3, so that t′1=T s , determine the action time of the modulated voltage vector as t1′, t2 and t3; When the open-winding permanent magnet synchronous motor system is in state 3, t2 is subtracted from the control period T s The difference between the two is taken as the excess duration t3, so that t′2=T s , determine the action time of the modulated voltage vector as t1, t′2 and t3; When the open-winding permanent magnet synchronous motor system is in state 4, the dual inverters synthesize a voltage vector with the longest length and the same direction as the reference voltage vector. The action times t′1 and t′2 are adjusted so that the total action time meets the control cycle limit. For state 4, the overmodulated voltage vector action time is recalculated using the following formula: Wherein, t′1 and t′2 represent the action time of the voltage vector after the overmodulation process; When t'1 is greater than t'2, subtract the control period T from t'1. s The difference between the two is taken as the excess duration t3, so that t′1=T s , determine the action time of the modulated voltage vector as t′1, t′2 and t3; when t′1 is less than t′2, subtract the control period T from t′2 s The difference between the two is taken as the excess duration t3, so that t′2=T s , determine the action time of the modulated voltage vector as t′1, t′2 and t3.

7. The method according to claim 6, characterized in that To suppress the zero-sequence current, the zero-sequence current and voltage are suppressed by adjusting the action time of the zero-sequence voltage vector, including: Calculate the zero-sequence voltage component u of the selected voltage vectors for inverter 1 and inverter 2 0output , the formula is: in 0output =in 0INV1 -in 0INV2 Among them, u 0INV1 with u 0INV2 Represent the zero-sequence voltage values ​​of the selected voltage vectors of inverter 1 and inverter 2 respectively; Calculate the zero-sequence voltage output u 0output With reference zero sequence voltage The difference is: According to the difference Determine whether measures need to be taken to offset the zero-sequence current. If the difference is greater than 0 or less than 0, it is determined that inverter 1 or inverter 2 needs to act on the zero-sequence voltage vector U7 to offset the zero-sequence current as much as possible. Calculate the action time t7 of the zero-sequence voltage vector U7 that needs to act, the formula is: in, During the zero-sequence current suppression process, if inverter 1 or inverter 2 has excess capacity after applying the non-zero voltage vector U7, it continues to apply the zero voltage vector U0 until the control cycle ends.

8. A predictive current control device for an open-winding permanent magnet synchronous motor based on hybrid pulse width modulation, characterized in that: include: a division module, configured to calculate a reference voltage vector, divide the voltage vector into sectors according to the reference voltage vector, and select a non-zero voltage vector and a zero voltage vector within each sector; a calculation module, configured to select the voltage vector required to be applied by the dual inverters according to the sector and range where the reference voltage vector is located, and calculate the time for each inverter to apply the voltage vector using the volt-second balance principle; An overmodulation processing module is used to perform overmodulation processing and reallocate the voltage vector time if the action time of the selected voltage vector exceeds the control period; The suppression module is used to suppress the zero-sequence current and voltage by adjusting the action time of the zero-sequence voltage vector if there is zero-sequence current.

9. An electronic device, characterized in that: include: a processor, and a memory communicatively connected to the processor; The memory stores computer-executable instructions; The processor executes the computer-executable instructions stored in the memory to implement the method according to any one of claims 1 to 7.

10. A computer-readable storage medium, characterized in that The computer-readable storage medium stores computer-executable instructions, which are used to implement the method according to any one of claims 1 to 7 when executed by a processor.

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