Closed-loop control method and system for power driving and protection

Abstract

The embodiment of the invention provides a closed-loop control method for power driving and protection, and the method comprises the steps: constructing a voltage feedback type circuit through a high-speed operational amplifier, and forming a closed-loop gain structure through a feedback network of the voltage feedback type circuit; the load current of the output end of the voltage feedback type circuit is monitored in real time, and when the load current exceeds a first set threshold value, a protection mechanism is triggered to turn off the output stage; the power-on surge current is limited through the soft start control circuit; phase information of a load reflection coefficient is obtained, an inductance / capacitance combination of the variable dynamic impedance matching network is selected according to phase information table look-up, inductance / capacitance parameters are switched through a relay, and the transmission standing-wave ratio between the power amplifier and a load is optimized; monitoring the temperature of the radiating fin of the power tube, and dynamically adjusting the resistance value of the feedback network; and performing cooperative control of short-circuit protection, over-temperature protection and surge suppression according to the priority. According to the embodiment of the invention, the performance bottleneck in a load fluctuation scene is broken through.

Description

Technical Field

[0001] The present application belongs to the technical field of power electronics, and specifically relates to a closed-loop control method and system for power drive and protection. Background Art

[0002] Power amplifier systems face challenges with output distortion and stability when exposed to dynamic load impedance fluctuations (e.g., 50Ω-10kΩ). Traditional solutions employ fixed impedance matching networks coupled with discrete current-limiting protection circuits, but these solutions suffer from three major technical bottlenecks: ① The response time of mechanical matching mechanisms generally exceeds 100ms, making them incapable of adapting to high-frequency load fluctuations; ② The detection delay of the comparator-based overcurrent protection circuit is as long as 5-10µs. When a high-speed op amp in a closed-loop power amplifier shorts, the power transistors can easily exceed their safe operating area and fail; ③ The impact of temperature on the gain chain is not considered. The ±2000ppm / °C temperature coefficient of the power transistor beta causes output amplitude fluctuations, exacerbating nonlinear distortion.

[0003] In recent years, the application of wide-bandgap semiconductor devices has become widespread. While SiC / GaN power transistors possess high-frequency characteristics, they are more sensitive to reflected power caused by impedance mismatch. Experiments have shown that when the standing wave ratio (VSWR) exceeds 1.5:1, the standing wave effect can cause the junction temperature to rise by more than 30°C, significantly reducing the device lifespan.

[0004] Therefore, there is an urgent need to develop a new drive system that integrates dynamic impedance matching, fast power protection and temperature compensation to break through the performance bottleneck under load fluctuation scenarios. Summary of the Invention

[0005] The present application proposes a closed-loop control method and system for power drive and protection to address the above-mentioned defects of the prior art.

[0006] According to a first aspect of an embodiment of the present application, a closed-loop control method for power drive and protection is provided, comprising: A high-speed operational amplifier is used to construct a voltage feedback circuit, and a closed-loop gain structure is formed through a feedback network of the voltage feedback circuit; monitoring the load current at the output end of the voltage feedback circuit in real time, and triggering a protection mechanism to shut down the output stage when the load current exceeds a first set threshold value to perform short-circuit protection; The soft-start control circuit limits the power-on inrush current to perform surge suppression; Obtaining phase information of the load reflection coefficient, selecting an inductor / capacitor combination of a variable dynamic impedance matching network based on the phase information, and switching the inductor / capacitor parameters through a relay to optimize the transmission standing wave ratio between the output end of the power amplifier and the load; Monitoring the temperature of the power tube heat sink and dynamically adjusting the resistance value of the feedback network based on the temperature deviation to perform over-temperature protection; The coordinated control of the short circuit protection, the over-temperature protection and the surge suppression is performed according to priority.

[0007] In some embodiments, forming a closed-loop gain structure through the feedback network of the voltage feedback circuit includes: The feedback network having a closed-loop gain structure with deep negative feedback is constructed based on metal film resistors and precision resistors; The closed-loop gain is calculated as follows:

[0008] in, Indicates the closed-loop gain value, represents the resistance value of the metal film resistor, Indicates the resistance value of the precision resistor.

[0009] In some embodiments, the real-time monitoring of the load current at the output end of the voltage feedback circuit and triggering a protection mechanism to shut down the output stage when the load current exceeds a first set threshold to perform short-circuit protection includes: The load current is monitored in real time by a bidirectional current sensor, and the load current flows through a sampling resistor to generate a differential voltage; The differential voltage is amplified by the internal amplifier of the bidirectional current sensor and then output to the comparator; When the amplified differential voltage exceeds the threshold voltage of the comparator, a jump signal is output to trigger an interrupt of the microcontroller; After the microcontroller responds to the interrupt, the output stage of the amplifier is turned off within a preset time by a pulse width modulation controller to perform short circuit protection.

[0010] In some embodiments, the differential voltage is calculated using the following formula:

[0011] in, represents the differential voltage, represents the load current, Indicates the resistance of the sampling resistor.

[0012] In some embodiments, limiting the power-on inrush current by the soft start control circuit to perform surge suppression includes: The source current of the external field effect transistor is detected by the hot swap controller; When powered on, the internal current source of the hot-swap controller charges the current sensing pin capacitor to form a ramp current, and controls the gate voltage of the field effect transistor to gradually increase through the ramp current to generate the surge current; limiting the inrush current to a specified fraction of the rated value; When an overcurrent fault is detected, the hot-swap controller triggers a hiccup mode and restarts the power link after a preset period until the fault is resolved. When it is detected that no overcurrent fault occurs, the current operating state is maintained.

[0013] In some embodiments, the phase information of the load reflection coefficient is a phase value of the load reflection coefficient, the phase information lookup table is a preset phase interval mapping table, and obtaining the phase information of the load reflection coefficient, selecting an inductor / capacitor combination of a variable dynamic impedance matching network according to the phase information lookup table, and switching the inductor / capacitor parameters through a relay to optimize the transmission standing wave ratio between the power amplifier output end and the load includes: collecting the reflected signal between the output end of the power amplifier and the load through a dual directional coupler; extracting a phase value of a reflection coefficient of the reflected signal using a phase detector; According to the phase value, matching the corresponding inductor / capacitor combination number of the variable dynamic impedance matching network in the preset phase interval mapping table, wherein the phase interval mapping table is pre-calibrated by a vector network analyzer and is used to store the mapping relationship between the phase interval and the inductor / capacitor combination of the variable dynamic impedance matching network; The parameters of the variable dynamic impedance matching network are independently switched by at least four groups of relays, wherein the first group of relays is used to control the on and off of the main inductor, the second and third groups of relays are used to control the on and off of the two groups of parallel capacitors respectively, and the fourth group of relays is used to control the on and off of the series inductor; Within the response bandwidth, the transmission standing wave ratio between the output terminal of the power amplifier and the load is optimized to be below a specific ratio.

[0014] In some embodiments, monitoring the power tube heat sink temperature and dynamically adjusting the resistance value of the feedback network based on the temperature deviation to perform over-temperature protection includes: A platinum resistance temperature sensor is mounted on the power tube heat sink, and three sets of wires are electrically connected to the platinum resistance temperature sensor via a bridge structure to eliminate lead resistance errors. The resistance of the first set of wires is included in the high-end bias loop, the resistance of the second set of wires is isolated by the high input impedance of the instrument amplifier, and the resistance of the third set of wires is self-compensated via the bridge. Based on the linear relationship between the resistance value of the platinum resistance temperature sensor and the operating temperature, the real-time temperature is converted into a voltage signal; Dividing the operating temperature range into preset intervals, and dynamically calculating the compensation coefficient using a linear interpolation algorithm according to the interval in which the real-time temperature is located; The resistance value of the feedback network is adjusted in real time by a digital potentiometer to compensate for the gain of the temperature deviation; When the temperature deviation exceeds a second set threshold, starting the heat dissipation device and reducing the output power; When the temperature deviation exceeds the second set threshold, the current operating state is maintained.

[0015] In some embodiments, the gain for compensating for temperature deviation is calculated by the following formula:

[0016] in, Indicates real-time temperature The corresponding actual gain is, Indicates the reference temperature The corresponding initial gain is, Indicates the compensation coefficient.

[0017] In some embodiments, the priority level is such that the short circuit protection priority is higher than the over-temperature protection priority, and the over-temperature protection priority is higher than the surge suppression priority.

[0018] According to a second aspect of the present application, an adaptive power drive and protection system is provided, comprising: A feedback circuit building module is used to construct a voltage feedback circuit using a high-speed operational amplifier, and form a closed-loop gain structure through the feedback network of the voltage feedback circuit; a short-circuit protection processing module, configured to monitor the load current at the output end of the voltage feedback circuit in real time, and trigger a protection mechanism to shut down the output stage when the load current exceeds a first set threshold value, thereby performing short-circuit protection; A surge suppression processing module, used to limit the power-on surge current through a soft start control circuit to perform surge suppression; A standing wave ratio optimization module is used to obtain phase information of the load reflection coefficient, select an inductor / capacitor combination of the variable dynamic impedance matching network based on the phase information, and switch the inductor / capacitor parameters through a relay to optimize the transmission standing wave ratio between the power amplifier output terminal and the load; A temperature protection processing module is used to monitor the temperature of the power tube heat sink and dynamically adjust the resistance value of the feedback network based on the temperature deviation to perform over-temperature protection; A collaborative execution control module is used to perform collaborative control of the short-circuit protection, the over-temperature protection and the surge suppression according to priority.

[0019] The beneficial effects of the closed-loop control method and system for power drive and protection according to the embodiments of the present application include at least: The feedback network of the voltage feedback circuit constructed in the embodiment of the present application can effectively suppress the nonlinear distortion of the audio signal by stabilizing the gain of the power amplifier to form a closed-loop gain structure; combined with the current monitoring mechanism, it provides a hardware foundation for short-circuit protection and reduces the risk of power tube overload and burning; by executing the fast shutdown output stage of short-circuit protection, it can respond in milliseconds when the load current exceeds the threshold to prevent the power device from breaking down; through collaborative priority control, other operations can be interrupted at the highest priority to ensure the core safety of the system; based on the soft start control of surge suppression, the power-on current is avoided, which is caused by power supply shock and damage to input stage devices; with the mutually exclusive layout of short-circuit protection, an independent protection is built for the vulnerable link of the power supply link, which reduces the interference of false triggering; through dynamic impedance matching The phase detection and table lookup switching in the optimization can match the variable dynamic impedance matching parameters in real time based on the reflection phase, reducing the standing wave ratio to near 1:1; through direct control of relays, a mechanical delay-free switching topology is established to solve the signal reflection problem in a wide impedance range (50Ω–10kΩ); based on the dynamic adjustment of the feedback resistor in over-temperature compensation and protection, the gain deviation caused by temperature drift can be compensated to maintain the amplitude stability of the output signal; based on the hierarchical power reduction strategy, the output power can be reduced and heat dissipation can be forced when the temperature exceeds the limit, extending the service life of the power tube; based on the priority of hierarchical collaborative control, the response can be sorted according to the degree of fault hazard to maximize the effectiveness of key protection; based on the mutual exclusion and linkage of multi-directional actions, the superposition and deterioration of multiple faults are avoided. BRIEF DESCRIPTION OF THE DRAWINGS

[0020] Figure 1 A flow chart of a closed-loop control method for power drive and protection according to an embodiment of the present application; Figure 2 This is a schematic structural diagram of a closed-loop control system for power drive and protection according to an embodiment of the present application; Figure 3 A circuit diagram of an operational amplifier according to an embodiment of the present application; Figure 4 A circuit diagram of an output stage according to an embodiment of the present application; Figure 5 A circuit diagram of a power supply protection unit according to an embodiment of the present application; Figure 6 A circuit diagram of a phase detector according to an embodiment of the present application; Figure 7 A circuit diagram of a variable dynamic impedance matching network and system control related to an embodiment of the present application; Figure 8 A circuit diagram of temperature sampling and signal conditioning related to an embodiment of the present application; Figure 9 A circuit diagram of gain compensation according to an embodiment of the present application; Figure 10 A circuit diagram of a temperature-controlled fan according to an embodiment of the present application; Figure 11 This is a circuit diagram of a thermal protection relay according to an embodiment of the present application. DETAILED DESCRIPTION

[0021] In order to enable those skilled in the art to better understand the technical solution of the present application, the present application is further described in detail below with reference to the accompanying drawings and specific implementation methods.

[0022] The following detailed description of the embodiments of the present application is provided in conjunction with the accompanying drawings and examples. The following detailed description of the embodiments and the accompanying drawings are used to illustrate the principles of the present application, but are not intended to limit the scope of the present application, i.e., the present application is not limited to the described embodiments.

[0023] The embodiment of the present application provides a closed-loop control method for power drive and protection, which is executed by a closed-loop control system for power drive and protection. Figure 1 As shown, the method includes the following steps 110-160.

[0024] Step 110 : constructing a voltage feedback circuit using a high-speed operational amplifier, and forming a closed-loop gain structure through the feedback network of the voltage feedback circuit.

[0025] The embodiment of the present application preferably uses a high-speed operational amplifier to construct a voltage feedback circuit.

[0026] Among them, the voltage feedback circuit can be configured with a peak current detection unit and a power supply protection unit. The peak current detection unit can realize ±250mA threshold monitoring through a bidirectional current sensor, and the power supply protection unit can include a hot-swap controller to limit the inrush current to ≤120% of the rated value.

[0027] In some embodiments, forming a closed-loop gain structure through the feedback network of the voltage feedback circuit includes: constructing the feedback network with a closed-loop gain structure having deep negative feedback based on metal film resistors and precision resistors.

[0028] Exemplarily, the closed-loop gain is calculated by the following formula:

[0029] in, Indicates the closed-loop gain value, Indicates the resistance of the metal film resistor. Indicates the resistance value of the precision resistor.

[0030] For example, the feedback network consists of metal film resistors ( ) and precision resistors ( ), the corresponding closed-loop gain is times the depth of the negative feedback structure.

[0031] Step 120 , monitoring the load current at the output end of the voltage feedback circuit in real time, and triggering a protection mechanism to shut down the output stage when the load current exceeds a first set threshold value to perform short-circuit protection.

[0032] In some embodiments, real-time monitoring of the load current at the output end of the voltage feedback circuit and triggering a protection mechanism to shut down the output stage when the load current exceeds a first set threshold to perform short-circuit protection include: real-time monitoring of the load current through a bidirectional current sensor, the load current flowing through a sampling resistor to generate a differential voltage; the differential voltage is amplified by an internal amplifier of the bidirectional current sensor and then output to a comparator; when the amplified differential voltage exceeds a threshold voltage of the comparator, a jump signal is output to trigger a microcontroller (MCU) interrupt; after the microcontroller responds to the interrupt, the output stage of the amplifier is shut down within a preset time through a pulse width modulation (PWM) controller to perform short-circuit protection.

[0033] Exemplarily, the differential voltage is calculated by the following formula:

[0034] in, represents the differential voltage, Indicates the load current, Indicates the resistance of the sampling resistor.

[0035] For example, the output stage monitors the load current in real time through a bidirectional current sensor. Flowing through the sampling resistor (low inductance package), a differential voltage is generated , output to the comparator through the internal amplifier gain G=50 times. Comparator threshold voltage Corresponding current threshold When overcurrent is detected, the comparator outputs a jump signal that triggers a microprocessor interrupt, and the PWM controller shuts down the output stage within 800ns.

[0036] In some embodiments, the protection is triggered when the load current exceeds a ±250 mA threshold.

[0037] In one exemplary embodiment, the short-circuit protection mechanism employs a hierarchical response strategy: when an output short is detected, the current sensor triggers a protection signal within 800ns, causing the pulse-width modulation controller to immediately shut down the output stage. Once the fault is resolved, the microcontroller confirms state recovery via a 200us software delay, keeping the overall recovery time under 1ms. This embodiment integrates both hardware and software redundancy for thermal protection. For example, at the hardware level, a comparator directly drives the fan, while at the software level, the output power is dynamically adjusted based on the temperature integral. The drive amplitude is automatically reduced when the heat sink temperature exceeds 75°C. This reduces downtime during short-circuit conditions by 90% compared to conventional solutions (>10ms).

[0038] Step 130 : Limiting the power-on inrush current by the soft start control circuit to perform surge suppression.

[0039] In some embodiments, limiting the power-on inrush current through a soft-start control circuit to perform surge suppression includes: detecting the source current of an external field-effect transistor (MOSFET) through a hot-swap controller; when powered on, the internal current source of the hot-swap controller charges the current sense (CS) pin capacitor to form a ramp current, and controls the gate voltage of the field-effect transistor to gradually increase through the ramp current to generate the inrush current; limiting the inrush current to within a specific proportion of the rated value; when an overcurrent fault is detected, the hot-swap controller triggers a hiccup mode, and restarts the power link after each preset period until the fault is resolved; when it is detected that no overcurrent fault has occurred, maintaining the current operating state.

[0040] For example, a hot-swap controller implements soft-start by sensing the source current of an external field-effect transistor (FET). Upon power-up, the hot-swap controller's internal current source charges the capacitor on the current-sense pin, creating a ramp current that gradually increases the gate voltage of the external FET, limiting the inrush current to 120% of the rated value. (This means that a hot-swap controller implements soft-start, limiting the instantaneous current at power-up to ≤120% of the rated value.) If an overcurrent fault is detected, the controller triggers hiccup mode, attempting to restart every 12ms until the fault is resolved.

[0041] Step 140 , obtaining phase information of the load reflection coefficient, selecting an inductor / capacitor (L / C) combination of a variable dynamic impedance matching (LC) network based on the phase information, and switching the inductor / capacitor parameters through a relay to optimize the transmission standing wave ratio between the power amplifier output and the load.

[0042] In some implementations, the phase information of the load reflection coefficient is a phase value of the load reflection coefficient, and the phase information lookup table is a preset phase interval mapping table.

[0043] In some embodiments, obtaining phase information of a load reflection coefficient, selecting an inductor / capacitor combination of a variable dynamic impedance matching network based on the phase information through a table lookup, and switching inductor / capacitor parameters through relays to optimize the transmission standing wave ratio between the power amplifier output and the load includes: collecting a reflection signal between the power amplifier output and the load through a dual directional coupler; extracting a phase value of the reflection coefficient of the reflection signal using a phase detector; matching the corresponding number of the inductor / capacitor combination of the variable dynamic impedance matching network in a preset phase interval mapping table based on the phase value, wherein the phase interval mapping table is pre-calibrated through a vector network analyzer and is used to store a mapping relationship between phase intervals and inductor / capacitor combinations of the variable dynamic impedance matching network; independently switching the parameters of the variable dynamic impedance matching network through at least four groups of relays, wherein a first group of relays is used to control the on / off of a main inductor, a second group of relays and a third group of relays are respectively used to control the on / off of two groups of parallel capacitors, and a fourth group of relays is used to control the on / off of a series inductor; and optimizing the transmission standing wave ratio between the power amplifier output and the load to below a specific ratio within a response bandwidth.

[0044] This step of the application is related to the phase detector, variable dynamic impedance matching network and microprocessor control unit. The reflection coefficient is obtained by dual directional couplers (based on the phase difference between the incident wave and the reflected wave), and the variable dynamic impedance matching network parameters are selected by table lookup to achieve a standing wave ratio within a 10MHz bandwidth of less than 1.2:1. The accuracy of the phase detector of the application is ±0.5°, corresponding to the reflection coefficient The measurement error does not exceed 0.09.

[0045] For example, the topology of a variable dynamic impedance matching network can use a π-type network (two parallel capacitors sandwiching a series inductor), with four relays (K1-K4) switching between different inductor / capacitor combinations. Relay K1 controls the main inductor L1 (10µH), relay K2 controls the parallel capacitor C1 (100pF), relay K3 controls the parallel capacitor C2 (1nF), and relay K4 controls the series inductor L2 (1µH).

[0046] For example, the coverage of the impedance switch can be calculated based on the following formula: Verification: ; The impedance ranges corresponding to the above combinations are calculated based on the parallel / series resonance conditions and the following formula: When K1 is closed and K2-K4 are open, then: ; When K1-K4 are fully closed, the equivalent ,but: ; Based on the above combination of switch states, the range of 50Ω-10kΩ can be covered, and the matching response time is ≤15ms. This coverage range and response time meet the dynamic response requirements of the audio frequency band (20kHz), breaking through the response bottleneck of traditional mechanical matching devices (conventional solutions >100ms). Actual measurements of the embodiments of the present application show that the standing wave ratio remains stable within a 10MHz bandwidth.

[0047] For example, the standing wave ratio of this application is Calculated based on the following formula:

[0048] Based on the derivation of the above formula, the embodiment of the present application can ensure that the standing wave ratio after matching is stable below 1.2:1 (Γ<0.09).

[0049] In some embodiments, the method further includes establishing a calibration of the impedance-phase relationship, including: using a vector network analyzer (VNA) at a frequency of 10 MHz to sequentially measure the parameter S11 corresponding to each variable dynamic impedance matching combination; extracting the phase angle θ=arg(S11) of the reflection coefficient Γ, and establishing a mapping table between the variable dynamic impedance matching combination number and the phase angle θ; and inversely calculating the equivalent electrical length using the formula θ=2π×2d / λ (d is the transmission line length, λ is the wavelength).

[0050] In some embodiments, the lookup table generation method includes: dividing the phase angle θ into 10° step intervals, and the error of the corresponding reflection coefficient Γ can be controlled to be less than or equal to 0.015 (based on the Smith chart conversion relationship); storing the optimal variable dynamic impedance matching combination code corresponding to each interval in the flash memory of the microprocessor.

[0051] Step 150 : monitor the temperature of the power tube heat sink and dynamically adjust the resistance value of the feedback network based on the temperature deviation to perform over-temperature protection.

[0052] In some embodiments, monitoring the temperature of a power tube heat sink and dynamically adjusting the resistance value of the feedback network based on a temperature deviation to perform over-temperature protection includes: mounting a platinum resistance temperature sensor on the power tube heat sink, and electrically connecting three groups of wires to the platinum resistance temperature sensor through a bridge structure to eliminate lead resistance errors, wherein the resistance of a first group of wires is included in a high-end bias loop, the resistance of a second group of wires is isolated by the high input impedance of an instrumentation amplifier, and the resistance of a third group of wires is self-compensated through the bridge; converting the real-time temperature into a voltage signal based on a linear relationship between the resistance value of the platinum resistance temperature sensor and the operating temperature; dividing the operating temperature range into preset intervals, and dynamically calculating a compensation coefficient using a linear interpolation algorithm according to the interval in which the real-time temperature lies; adjusting the resistance value of the feedback network in real time through a digital potentiometer to compensate for the gain of the temperature deviation; starting a heat sink and reducing the output power when the temperature deviation exceeds a second set threshold; and maintaining the current operating state when the temperature deviation exceeds the second set threshold.

[0053] Exemplarily, the gain for compensating for temperature deviation is calculated by the following formula:

[0054] in, Indicates real-time temperature The corresponding actual gain is, Indicates the reference temperature The corresponding initial gain is, Indicates the compensation coefficient, preferably =±5ppm / ℃. The microprocessor uses the SPI interface to calculate the temperature deviation ( ) Dynamically adjust the digital potentiometer resistance and the feedback network voltage divider ratio to offset the gain drift of the power tube feedback coefficient due to temperature changes (typically -2000ppm / °C). The reference temperature is preferably 25°C.

[0055] For example, this step of the present application is related to the integrated temperature sensor and digital potentiometer, and uses a platinum resistance temperature sensor (PT100) to sample the power tube heat sink temperature, and achieves ±5ppm / ℃ gain compensation by adjusting the feedback network resistance. The temperature sampling circuit layout is a three-wire connection method, in which a bridge structure is preferably used, with a fixed balancing resistor R1 = fixed balancing resistor R2 = 1kΩ, and a sensor equivalent resistance R3 = platinum resistance temperature sensor R PT100 + Lead resistance R L , eliminate the lead resistance R through the instrument amplifier L Impact: The resistance of the first wire R L1 Taking into account the high-side bias; the second wire resistance R L2 The resistor is isolated by the high input impedance of the instrument amplifier; the third wire resistor R L3Self-compensation through bridge structure.

[0056] In some embodiments, temperature and voltage The conversion formula is as follows:

[0057] in, , ; It is the standard compensation coefficient based on platinum resistance temperature sensor.

[0058] In some embodiments, the linear interpolation algorithm is preferably a piecewise linear interpolation algorithm, and its segmentation strategy is: divide 0-50°C into 5 intervals (each 10°C), and store the endpoint temperature T of each interval. i And the corresponding compensation coefficient K i , where i is the interval number.

[0059] In some embodiments, when the real-time temperature T is between [T j , T j+1 ] interval (j is the interval number), the linear interpolation algorithm is used to dynamically calculate the compensation coefficient The formula is as follows:

[0060] In some embodiments, this step of the embodiment of the present application also includes short-circuit protection logic. When a short circuit is detected at the output end, the pulse width modulation controller shuts down the output stage within 800ns and automatically restores power within 1ms after the fault is resolved.

[0061] In some embodiments, when the temperature of the power tube heat sink exceeds 75° C., the fan is activated and the output power is reduced.

[0062] Step 160 , executing coordinated control of the short circuit protection, the over-temperature protection, and the surge suppression according to priority.

[0063] In some embodiments, the microprocessor performs a hierarchical response based on the three fault types of the short circuit protection, the over temperature protection, and the surge suppression, and the priority level is that the short circuit protection priority is higher than the over temperature protection priority, and the over temperature protection priority is higher than the surge suppression priority.

[0064] In some embodiments, the overall system performance is measured and verified by the following closed-loop error suppression formula:

[0065] in, represents the open-loop gain (=10000), Indicates the load impedance change rate (200:1), the theoretical load regulation rate The dynamic matching network's real-time correction capability allows for load regulation better than 0.1%, while harmonic distortion (THD+N) reaches 0.0043% at full power output at 20kHz (with an intrinsic value of 0.0003% and feedback network nonlinearity added). This patent application has been validated in precision testing equipment, resolving the technical bottleneck of conventional solutions with signal distortion exceeding 20% ​​within a load range of 50Ω-10kΩ, meeting the stringent harmonic distortion requirements of various audio standards.

[0066] The feedback network of the voltage feedback circuit constructed in the embodiment of the present application can effectively suppress the nonlinear distortion of the audio signal by stabilizing the gain of the power amplifier to form a closed-loop gain structure; combined with the current monitoring mechanism, it provides a hardware foundation for short-circuit protection and reduces the risk of power tube overload and burning; by executing the fast shutdown output stage of short-circuit protection, it can respond in milliseconds when the load current exceeds the threshold to prevent the power device from breaking down; through collaborative priority control, other operations can be interrupted at the highest priority to ensure the core safety of the system; based on the soft start control of surge suppression, the power-on current is avoided, which is caused by power supply shock and damage to input stage devices; with the mutually exclusive layout of short-circuit protection, an independent protection is built for the vulnerable link of the power supply link, which reduces the interference of false triggering; through dynamic impedance matching The phase detection and table lookup switching in the optimization can match the variable dynamic impedance matching parameters in real time based on the reflection phase, reducing the standing wave ratio to near 1:1; through direct control of relays, a mechanical delay-free switching topology is established to solve the signal reflection problem in a wide impedance range (50Ω–10kΩ); based on the dynamic adjustment of the feedback resistor in over-temperature compensation and protection, the gain deviation caused by temperature drift can be compensated to maintain the amplitude stability of the output signal; based on the hierarchical power reduction strategy, the output power can be reduced and heat dissipation can be forced when the temperature exceeds the limit, extending the service life of the power tube; based on the priority of hierarchical collaborative control, the response can be sorted according to the degree of fault hazard to maximize the effectiveness of key protection; based on the mutual exclusion and linkage of multi-directional actions, the superposition and deterioration of multiple faults are avoided.

[0067] Refer to the attached Figure 2 As shown, an embodiment of the present application also discloses a closed-loop control system for power drive and protection, including: a feedback circuit construction module 210, a short-circuit protection processing module 220, a surge suppression processing module 230, a standing wave ratio optimization module 240, a temperature protection processing module 250 and a collaborative execution control module 260.

[0068] The feedback circuit building module 210 is used to construct a voltage feedback circuit using a high-speed operational amplifier, and form a closed-loop gain structure through the feedback network of the voltage feedback circuit.

[0069] The short-circuit protection processing module 220 is used to monitor the load current at the output end of the voltage feedback circuit in real time, and trigger a protection mechanism to shut down the output stage when the load current exceeds a first set threshold to perform short-circuit protection.

[0070] The surge suppression processing module 230 is configured to limit the power-on surge current through a soft start control circuit to perform surge suppression.

[0071] The standing wave ratio optimization module 240 is used to obtain phase information of the load reflection coefficient, select the inductor / capacitor combination of the variable dynamic impedance matching network based on the phase information, and switch the inductor / capacitor parameters through a relay to optimize the transmission standing wave ratio between the power amplifier output end and the load.

[0072] The temperature protection processing module 250 is used to monitor the temperature of the power tube heat sink and dynamically adjust the resistance value of the feedback network based on the temperature deviation to perform over-temperature protection.

[0073] The coordinated execution control module 260 is used to perform coordinated control of the short circuit protection, the over-temperature protection and the surge suppression according to priority.

[0074] In a specific embodiment, this application is based on a proven engineering practice and is divided into four phases: hardware construction, software development, system integration, and testing and verification. The following uses a typical application scenario (audio power amplifier equipment) as an example to illustrate the specific implementation process, following phases 1 through 3.

[0075] Phase 1, hardware construction, includes the following sub-phases 1)-3).

[0076] 1) Assembly of closed-loop power amplifier unit.

[0077] Refer to the attached Figure 3 As shown in the figure, a voltage feedback circuit is constructed using a high-speed operational amplifier. First, solder the circuit to a four-layer printed circuit board (PCB), which consists of a signal layer, a power layer, a ground layer, and an auxiliary layer. The following high-frequency wiring principles should be followed during layout: the input trace should be shorter than 5mm to suppress parasitic inductance, and the feedback resistor (R f =1kΩ) and the gain resistor (R g=100Ω) uses 0.1% metal film precision resistors, and the power pins are filtered in parallel with a 10µF ceramic capacitor and a 0.1µF film capacitor. Specifically, INF1 is the BNC RF input connector; RF8 is a 49.9Ω / 1% precision resistor for input impedance matching; RF9 is a 100Ω / 0.1% precision resistor for the signal sampling and feedback path; RF10 is a 1kΩ / 0.1% precision resistor for the closed-loop feedback network; CF6 is a 10µF / 25V electrolytic capacitor for power supply decoupling; CF7 is a 0.1µF / 25V ceramic capacitor for high-frequency noise suppression; and UF2 is a high-speed operational amplifier for core signal processing. The circuit principle is as follows: the input signal is connected through the BNC connector INF1, matched by the 49.9Ω resistor RF8, and then forms a voltage divider network with the 100Ω resistor RF9, attenuating the signal and applying it to the non-inverting input of the operational amplifier UF2. The feedback network, consisting of a 1kΩ resistor RF10, connects the inverting input and output terminals, forming a closed-loop negative feedback loop, maintaining a fixed amplification factor of 11 (1 + 1kΩ / 100Ω). The power supply is decoupled in two stages, using a 10µF electrolytic capacitor CF6 and a 0.1µF ceramic capacitor CF7, to eliminate high-frequency noise. The entire circuit is referenced to ground GND, achieving stable, high-precision small-signal amplification.

[0078] Refer to the attached Figure 4 As shown, the output stage is configured with a bidirectional current sensor, and its sampling resistor R S =0.1Ω requires a low-inductance package. The output signal is connected to a comparator to detect the ±250mA threshold. Specifically, the INA138 is a three-terminal adjustable precision instrumentation amplifier; the LMV721 is a general-purpose dual-supply operational amplifier (UF3 and UF4); RF11-RF16 are precision metal film resistors (value and accuracy are marked); C1-C2 are ceramic bypass capacitors; LED_RED is a red light-emitting diode; DF1 is a fast recovery diode; and Load is the load interface. The circuit principle is as follows: This circuit implements differential voltage detection and level shifting. The input signal IN is fed into the inverting input of the INA138 via a precision voltage divider network formed by RF11-RF13. The detection threshold can be set by adjusting the resistance value of RF13. The INA138 output signal drives the voltage follower formed by UF3. Its output provides a bias reference for UF4 and also indicates the operating status via LED_RED (lit when forward conducting). UF4 forms an inverting amplifier circuit, amplifying the difference between the input signal and the reference level, ultimately outputting it through the OUT terminal. Capacitors C1 and C2 provide power decoupling and signal coupling, respectively, while diode DF1 prevents overload on the negative half-cycle signal. The entire circuit utilizes a dual-power supply architecture, ensuring linearity and stability in signal processing.

[0079] Refer to the attached Figure 5As shown, the power supply protection unit uses a controller to drive an external N-channel field-effect transistor. A 10Ω resistor is connected in series with the gate to suppress oscillation, and a 100nF snubber capacitor is connected in parallel between the source and drain. Specifically, PVR is a precision adjustable resistor; CF1-CF5 are electrolytic / ceramic filter capacitors (with withstand voltage and capacitance values ​​indicated); RF1-RF7 are precision metal film resistors (with resistance values ​​and accuracy indicated); QF1 is a fast-blow fuse; MOSFET-N is an N-channel power field-effect transistor; and UF1 is a PWM controller chip. The circuit principle is as follows: This circuit is a step-down DC-DC converter with overcurrent protection. The input power source sets a reference voltage via the PVR, which provides feedback to the LM5069 via the voltage divider network formed by RF1-RF3. UF1's internal oscillator generates a fixed-frequency PWM signal, which drives the gate of MOSFET-N and controls its conduction period. QF1 acts as an overcurrent fuse, quickly disconnecting the circuit when the load current exceeds the threshold. RF4-RF6 form a slope compensation network to improve duty cycle linearity. CF2-CF5 form a multi-stage filtering structure, with CF4-CF5 forming a π-type filter to suppress high-frequency ripple. RF7 and CF3 form a soft-start circuit to prevent inrush current from damaging the device during startup. The entire system stabilizes the output voltage (Vout) through closed-loop feedback. Undervoltage lockout (UVLO) and power status indication (PGD) ensure reliable operation over a wide range of input conditions.

[0080] 2) Implementation of dynamic impedance matching unit.

[0081] Refer to the attached Figure 6As shown, this unit consists of a phase detector, a dual directional coupler, and a variable dynamic impedance matching network switched by relays. First, a dual directional coupler with a coupling coefficient of 30dB and a frequency range of DC-50MHz is fabricated on a high-frequency printed circuit board. It is then soldered to a separate daughterboard. Its RF input is connected to a 50Ω matching resistor, and its LO terminal is connected to a reference signal (derived from the signal source synchronization signal). Specifically, INP1 / INP2 are BNC RF input connectors; RP1 / RP2 are precision metal film resistors (labeled with resistance and accuracy); CP1-CP7 are ceramic filter capacitors (labeled with capacitance and withstand voltage); RP3-RP8 are precision trimming resistors (labeled with resistance and accuracy); UP1 is an RF amplitude detection chip; and Header2×2 is a dual-channel output connector. The circuit principle is as follows: This circuit is an RF signal amplitude detection module. The input signal (INP1 / INP2) is connected via the BNC connectors. RP1 / RP2 achieves impedance matching and signal attenuation. CP1-CP4 form a π-type filter network to suppress out-of-band interference. The AD8302 chip (UP1) serves as the core detection unit. Its differential inputs, INPA / INPB, receive processed signals. An internal logarithmic detector linearly converts the RF signal amplitude into a DC voltage output. The outputs (OUTP1 / OUTP2) are gain-adjusted via the RP4-RP8 resistor network, while capacitors CP5-CP7 provide low-pass filtering and power supply decoupling. This design utilizes a fully differential architecture to suppress common-mode noise, making it suitable for high-precision amplitude monitoring in the 50MHz to 4GHz frequency range. The output voltage exhibits a logarithmic-linear relationship with the input power.

[0082] Refer to the attached Figure 7As shown, the variable dynamic impedance matching network is controlled by four relays, each corresponding to a specific impedance range (50-200Ω, 200-1kΩ, 1k-5kΩ, and 5k-10kΩ). The inductor values ​​are calibrated using a vector network analyzer (VNA), and the capacitors are made of materials that ensure temperature stability. The microprocessor drives the relays via optocouplers to control signal isolation and transmission. Specifically, IN is the signal input port; OUT is the signal output port; UP2 / UP3 / UP5 are LTV-356T-C optocouplers; DP1 / DP2 / DP4 are IN4148 high-speed switching diodes; QP1 / QP2 / QP4 are NPN power transistors; JP1 / JP2 / JP4 are relay-NO normally open modules; LP1 / LP2 / LP4 are wirewound power inductors; RP9-RP16 are precision metal film resistors; CP8-CP11 are ceramic filter capacitors; and GPIO1-GPIO4 are the microcontroller digital control interfaces. The circuit principle is as follows: This circuit is a four-channel opto-isolated relay driver module. The input signal (IN) is processed by three independent control circuits: The digital signals from GPIO1-GPIO4 sequentially trigger the UP2 / UP3 / UP5 optocouplers. Once the internal phototransistors are turned on, they provide bias current to the bases of QP1 / QP2 / QP4 via the RP9-RP12 resistor network. When the transistors are saturated and turned on, the relay coils (JP1 / JP2 / JP4) are energized, closing their normally open contacts and connecting the load at the OUT terminal. The intermediate UP4 circuit acts as an auxiliary driver, regulating the base voltage of Q3 (2SD882) via the RP13-RP16 resistor divider, enhancing the relay coil's drive capability. The LP1-LP4 inductors and the CP8-CP11 capacitors form a π-type filter network to suppress high-frequency switching noise.

[0083] 3) Temperature compensation unit integration.

[0084] Refer to the attached Figure 8As shown, the temperature sampling circuit uses a platinum resistor bonded to the power tube heat sink, with a three-wire connection to eliminate the effects of wire resistance. The signal conditioning section uses an instrumentation amplifier with a gain set to 10x, and its output is connected to the temperature sensor's analog-to-digital converter channel. Specifically, the BS-12-B3AA003 is a pressure sensor; the PT100 is a platinum temperature sensor; RB3-RB7 are precision metal film resistors (with resistance value and accuracy indicated); CB2-CB4 are ceramic filter capacitors (with capacitance and withstand voltage indicated); the UB2AD620 is a high-precision instrumentation amplifier; the MAX6611 (UB3) is a temperature-compensated voltage reference chip; BB1 is a fuse / fuse protection element; and the GPIO is a microcontroller digital control interface. The circuit principle is as follows: This circuit implements dual-parameter temperature and pressure sensing. The bridge network on the left consists of the BS-12-B3AA003 pressure sensor and the PT100 platinum resistor, forming a full-bridge connection via resistors RB3-RB6. When temperature or pressure changes, the bridge arm resistance imbalance generates a small differential voltage. This voltage is amplified by the AD620 (UB2) high-precision instrumentation amplifier and then output to the MAX6611 (UB3) for temperature compensation and reference calibration. UB3 integrates an internal bandgap reference source, and its REF pin outputs a stable reference voltage. The SHDN# pin is controlled by a GPIO for standby mode management. Capacitors CB2-CB4 provide power supply decoupling, signal coupling, and output filtering, respectively.

[0085] Refer to the attached Figure 9 As shown, the gain compensation unit uses a digital potentiometer configured in variable feedback resistor mode. Its control terminal is programmable and adjustable by a microprocessor via the I²C interface. Specifically, UB1 is the digital potentiometer chip; CB1 is a 1µF / 25V ceramic filter capacitor; RB1 / RB2 are 1kΩ metal film resistors (accuracy specified); GPIO1 / GPIO2 are the microcontroller digital interface; and the circuit interface is the power and ground connection terminals. The circuit principle is as follows: This circuit is an I²C-bus controllable digital potentiometer application module. The AD5272 chip (UB1) is the core control unit, enabling bidirectional communication with the microcontroller via the SCL (GPIO1) and SDA (GPIO2) pins. RB1 / RB2 serve as I²C bus pull-up resistors, ensuring the signal lines remain high in the idle state. Capacitor CB1 is connected between VDD and ground to suppress power supply ripple and stabilize the internal reference voltage. An optional external capacitor can be connected to the EXT_CAP pin to optimize dynamic response.

[0086] Refer to the attached Figure 10-11 As shown in the figure, the cooling system includes a temperature-controlled fan (12V / 0.3A) and a thermal protection relay. When the temperature exceeds 75°C, the microprocessor triggers the fan and reduces the output power. Figure 10As shown, GPIO is the digital output pin of the microcontroller; RB9 is a metal film current limiting resistor (47Ω); QB1 is a 2SD882NPN power transistor; DB1 is an IN4148 fast recovery diode; RB8 is a collector resistor (36Ω); the fan is a DC motor load; and Header2 is a power connector. The circuit principle is as follows: This circuit is a DC fan drive control module. The microcontroller GPIO output signal drives the QB1 base through the RB9 current limiting resistor. When the GPIO output is high, QB1 is saturated and turned on. The VCC power supply forms a loop through RB8, QB1 collector-emitter and the fan to drive the fan to rotate. The DB1 diode is connected in series at both ends of the fan to absorb the motor back electromotive force spike voltage and protect QB1 from transient overvoltage damage. In addition to the current limiting function, RB8 can also suppress high-frequency noise during motor commutation. The Header2 interface is used to connect an external fan power supply to achieve hardware isolation and flexible wiring. This design uses digital control to adjust the fan start and stop for equipment heat dissipation management. For details, refer to the attached Figure 11 As shown, GPIO is a microcontroller digital output pin; RB10 is a 200Ω metal film current-limiting resistor; QB2 is a 2SD882NPN power transistor; DB2IN4148 is a fast recovery diode; JB1Relay-NC is a normally closed relay module; and Header2 is a power connector. Circuit Principle: This circuit is a relay-controlled switch module. The microcontroller GPIO output signal drives the base of QB2 via the RB10 current-limiting resistor. When the GPIO output is high, QB2 saturates and conducts, and the +5V power supply flows through QB2's collector-emitter terminals to the relay coil (JB1). When the relay is closed, its normally closed contacts open, disconnecting Header2 from the circuit. When the GPIO output is low or unpowered, QB2 turns off, releasing the relay, restoring the closed contacts, and connecting Header2 to the circuit. Diode DB2 is connected in parallel across the relay coil to absorb the reverse induced electromotive force during power failure and prevent QB2 from overvoltage damage. The design realizes circuit on-off switching through digital control and has electrical isolation and overvoltage protection functions.

[0087] Phase 2, software development phase, includes the following sub-phases 4)-6).

[0088] 4) Impedance matching algorithm development The microprocessor's main program uses a state machine architecture, including initialization, matching detection, parameter adjustment, protection response, etc. The core algorithm is based on phase detection data and uses a table lookup method to quickly match impedance.

[0089] During the calibration process, a vector network analyzer (VNA) is used to measure the impedance-phase relationship for each variable dynamic impedance matching combination and create a lookup table. Specifically, based on Smith chart theory and the reflection coefficient model, the VNA is used to calibrate the impedance-phase characteristics of the variable dynamic impedance matching combination and create a lookup table for fast matching. The system utilizes a π-type variable dynamic impedance matching network (four relay-switched inductor / capacitor combinations) covering an impedance range of 50Ω-10kΩ. Its theoretical basis is the phase-impedance mapping relationship and the following reflection coefficient formula:

[0090] During implementation, a SOLT calibration was first performed on a high-frequency printed circuit board (PCB) with a variable dynamic impedance matching network soldered on it. The vector network analyzer parameters (10MHz bandwidth, 101 scan points) were set, and relay combinations were activated group by group. The S11 parameters were recorded, and the real impedance R and imaginary impedance X of each combination were calculated. After verifying the impedance coverage using Smith chart mapping, the 0°-360° phase was divided into 10° steps. The least squares method was used to match the optimal variable dynamic impedance matching combination according to the following formula: ; The mapping between the phase interval and the variable dynamic impedance matching code is ultimately stored in the microprocessor's flash memory. Key technologies include three-wire PT100 temperature measurement to eliminate wire errors, phase detection accuracy of ±0.5°, and a microprocessor response time of <15ms. This has resulted in an improvement in the standing wave ratio from 2.1:1 to 1.15:1, and a reduction in reflected power to 0.5%, meeting the stringent industry standards for audio equipment.

[0091] In the real-time calculation process, the following formula is used to calculate Output voltage and The relationship between the reference voltage phase at the terminal: , The microprocessor uses the phase angle relationship Look up the table to select the optimal variable dynamic impedance matching combination.

[0092] During the dynamic optimization process, when it is detected When the change exceeds 5°, a secondary matching process is initiated to avoid false triggering.

[0093] 5) Development of a temperature compensation algorithm, preferably using piecewise linear interpolation.

[0094] During the temperature acquisition process, the temperature data is read every 100ms and the current temperature T is obtained after sliding average filtering.

[0095] During the gain correction process, the compensation coefficient is calculated according to the following formula: ; in, =5×10 -6 / ℃, =25℃.

[0096] During the potentiometer control process, the compensation coefficient is converted into a digital address (0-1023) and written into the register via I2C.

[0097] 6) Design of power protection logic, preferably using interrupt priority mechanism.

[0098] During the overcurrent protection process, the detection signal is connected to the comparator, the threshold is set to 250mV (corresponding to 250mA), and the interrupt response time is <1us.

[0099] During the short-circuit processing, after the microprocessor detects the overcurrent signal, it immediately turns off the pulse width modulation output and records the fault code, and attempts to automatically recover after 1ms.

[0100] During over-temperature protection, the temperature data is updated every 500ms. When the temperature exceeds the limit (75°C), the fan is triggered and the output amplitude is reduced to 50%.

[0101] Phase 3, system integration and testing verification, includes the following sub-phases 7)-9).

[0102] 7) Hardware functional verification.

[0103] During the closed-loop amplifier test, a 1kHz sine wave (1Vpp) was input, the load was switched to 50Ω / 8Ω / 600Ω, and the output waveform was observed with an oscilloscope to confirm that the harmonic distortion (THD) was <0.005%.

[0104] During the dynamic matching network debugging process, a vector network analyzer was used to scan a 50Ω-10kΩ load to verify that the standing wave ratio was <1.2:1 (reflection coefficient Γ <0.09).

[0105] During the temperature compensation calibration process, the system was placed in a temperature chamber (0-50°C cycle) and the output amplitude changes at different temperatures were recorded to ensure that the drift was ≤0.02% / °C.

[0106] 8) Whole machine performance test.

[0107] During the load regulation test, the input voltage was 220V±10%, the load was switched from 50Ω to 10kΩ, and the output voltage fluctuation monitored by the multimeter was <0.1%.

[0108] During the short-circuit protection response process, the output end is artificially short-circuited, and the oscilloscope captures the protection action time (shutdown <800ns, recovery <1ms).

[0109] During the long-term stability test, after 72 hours of continuous operation, infrared thermal imaging showed that the temperature rise of the power tube was less than 15°C.

[0110] 9) Deployment in actual application scenarios (implementing this application in audio amplifier equipment).

[0111] During multi-way speaker switching, when the user switches between 8Ω / 4Ω / 600Ω speakers, the dynamic matching network completes parameter adjustment within 15ms, reducing the harmonic distortion (THD) from 22% in traditional solutions to 0.0045%.

[0112] During the RF signal source application, in the 10MHz carrier test, the standing wave ratio was optimized to 1.15:1, the reflection loss was increased by 9.5dB, and the junction temperature of the power tube dropped by 28°C.

[0113] Experimental verification of the embodiments of this application shows that all core performance indicators meet design requirements. Load regulation is less than 0.1% (measured at 0.08%), total harmonic distortion (THD+N) is 0.0043% at full power output at 20kHz, short-circuit protection recovery time is controlled within 1ms, a dynamic impedance matching network optimizes the voltage standing wave ratio (VSWR) to 1.15:1 (reflection coefficient Γ = 0.07), and a temperature compensation circuit reduces system gain drift to ≤ 0.02% / °C.

[0114] The embodiments of this application address the technical challenge of signal distortion >20% caused by load impedance fluctuations (50Ω-10kΩ). Measured data from the aforementioned specific embodiments demonstrates load regulation <0.1% (synergistic effect of 20MHz closed-loop gain bandwidth and 15ms dynamic matching response time), harmonic distortion <0.005% at 20kHz (open-loop gain 80dB combined with deep negative feedback calculation), and short-circuit protection recovery time <1ms (comparator response time 800ns and microprocessor reset delay 200ns). The embodiments of this application can address signal distortion and device reliability issues caused by dynamic load impedance changes in audio amplifiers, RF signal sources, and precision test equipment. By integrating closed-loop control, dynamic impedance matching, and temperature compensation technologies, they achieve high-precision power transmission over a wide load range, encompassing power amplifier design, automatic control algorithms, and thermal management techniques.

[0115] This embodiment of the present application utilizes a three-stage architecture to achieve high-precision power transmission. The first stage, based on a closed-loop power amplifier unit, uses current sensing to achieve ±250mA peak limiting and <100ns overcurrent protection. The second stage, based on a dynamic impedance matching unit, employs phase detection and relay-switched variable dynamic impedance matching networks to optimize the standing wave ratio to <1.2:1 within a 10MHz bandwidth. The third stage, based on a temperature compensation unit, uses PT100 temperature measurement and digital potentiometer adjustment to achieve ±5ppm / °C gain stability. Tests show load regulation of <0.1%, harmonic distortion of <0.005% at 20kHz (open-loop gain of 80dB + deep negative feedback suppression), and short-circuit protection recovery time of <1ms (comparator response time of 800ns + MCU reset time of 200ns). This embodiment of the present application addresses signal distortion caused by wide fluctuations in load impedance by constructing a three-stage architecture of closed-loop control, dynamic matching, and temperature compensation to achieve high-precision power transmission. Specifically, a high-speed operational amplifier (OPA) forms a closed-loop power amplifier unit with a 2000V / µs slew rate and 80dB open-loop gain. A peak current detection unit and a power supply protection unit are configured within a voltage-feedback circuit. Peak current detection uses a bidirectional current sensor to monitor the load current. When the output current exceeds the ±250mA threshold, the pulse-width modulation controller shuts down the output stage within 100ns. This response speed is slower than the maximum transient current duration allowed by the safe operating area (SOA) (typically 500ns). The power supply protection unit integrates a hot-swap controller, achieving soft-start through an external field-effect transistor, limiting inrush current to within 120% of the rated value, preventing shock to the power transistors during startup.

[0116] The embodiment of the present application has the ability to suppress load fluctuations. Through the synergistic effect of closed-loop control and dynamic matching network, the system maintains output stability within the load range of 50Ω-10kΩ. The closed-loop gain error formula shows that when the open-loop gain T=80dB (10000), even if the load impedance changes by 200 times ( =200:1), with a theoretical error of only 0.0012%. Actual measured data shows that under conditions of input voltage fluctuation of 220V±10%, the load regulation rate reaches 0.08%, which is 2 orders of magnitude higher than the traditional solution. The embodiment of the present application has improved signal fidelity. The dynamic impedance matching network optimizes the standing wave ratio from 2.1:1 of the conventional solution to 1.15:1, and the reflected power ratio is reduced from 11% to 0.5%. The combined 80dB open-loop gain and deep negative feedback (feedback coefficient β=0.1) extend the total harmonic distortion from the device's intrinsic value of 0.0003% to 0.0043% (including the nonlinear error of the feedback network). This indicator is nearly three times better than the industry standard limit (0.01%), meeting the requirements of high-fidelity audio equipment. The embodiment of the present application has established a fast power protection mechanism. The short-circuit protection uses a current sensor (response time 800ns) to work in conjunction with a pulse width modulation controller to cut off the output stage within 800us. After the fault is cleared, the MCU confirms the state recovery through a software delay of 200us. The overall recovery time is less than 1ms, which is 90% shorter than the traditional discrete component solution (5-10ms). The thermal protection function is linked to the digital potentiometer through a PT100 platinum resistor (0.1℃ resolution). When the heat sink temperature exceeds 75℃, the output power is automatically reduced, effectively preventing the device from overheating and failure. The embodiment of this application enhances temperature stability. The temperature compensation algorithm is based on the gain temperature characteristic formula. ,in =±5ppm / °C (the combination of the digital potentiometer temperature coefficient and the temperature measurement error). Experiments show that the system gain drift is ≤0.02% / °C over an ambient temperature range of 0-50°C, a 10-fold improvement compared to the uncompensated state (a power tube β temperature coefficient of -2000ppm / °C corresponds to 0.2% / °C).

[0117] It is understood that the above embodiments are merely exemplary embodiments for illustrating the principles of the present application, and the present application is not limited thereto. Those skilled in the art may make various modifications and improvements without departing from the spirit and substance of the present application, and such modifications and improvements are also considered to be within the scope of protection of the present application.

Claims

1. A closed-loop control method for power drive and protection, characterized in that: include: A high-speed operational amplifier is used to construct a voltage feedback circuit, and a closed-loop gain structure is formed through a feedback network of the voltage feedback circuit; monitoring the load current at the output end of the voltage feedback circuit in real time, and triggering a protection mechanism to shut down the output stage when the load current exceeds a first set threshold value to perform short-circuit protection; The soft-start control circuit limits the power-on inrush current to perform surge suppression; Obtaining phase information of the load reflection coefficient, selecting an inductor / capacitor combination of a variable dynamic impedance matching network based on the phase information, and switching the inductor / capacitor parameters through a relay to optimize the transmission standing wave ratio between the output end of the power amplifier and the load; Monitoring the temperature of the power tube heat sink and dynamically adjusting the resistance value of the feedback network based on the temperature deviation to perform over-temperature protection; The coordinated control of the short circuit protection, the over-temperature protection and the surge suppression is performed according to priority.

2. The closed-loop control method for power drive and protection according to claim 1, characterized in that: The forming of a closed-loop gain structure through the feedback network of the voltage feedback circuit includes: The feedback network having a closed-loop gain structure with deep negative feedback is constructed based on metal film resistors and precision resistors; The closed-loop gain is calculated as follows: in, Indicates the closed-loop gain value, represents the resistance value of the metal film resistor, Indicates the resistance value of the precision resistor.

3. The closed-loop control method for power drive and protection according to claim 1, characterized in that: The real-time monitoring of the load current at the output end of the voltage feedback circuit and triggering a protection mechanism to shut down the output stage when the load current exceeds a first set threshold to perform short-circuit protection includes: The load current is monitored in real time by a bidirectional current sensor, and the load current flows through a sampling resistor to generate a differential voltage; The differential voltage is amplified by the internal amplifier of the bidirectional current sensor and then output to the comparator; When the amplified differential voltage exceeds the threshold voltage of the comparator, a jump signal is output to trigger an interrupt of the microcontroller; After the microcontroller responds to the interrupt, the output stage of the amplifier is turned off within a preset time by a pulse width modulation controller to perform short circuit protection.

4. The closed-loop control method for power drive and protection according to claim 3, characterized in that: The differential voltage is calculated as follows: in, represents the differential voltage, represents the load current, Indicates the resistance of the sampling resistor.

5. The closed-loop control method for power drive and protection according to claim 1, characterized in that: The method of limiting the power-on surge current by using a soft start control circuit to perform surge suppression includes: The source current of the external field effect transistor is detected by the hot swap controller; When powered on, the internal current source of the hot-swap controller charges the current sensing pin capacitor to form a ramp current, and controls the gate voltage of the field effect transistor to gradually increase through the ramp current to generate the surge current; limiting the inrush current to a specified fraction of the rated value; When an overcurrent fault is detected, the hot-swap controller triggers a hiccup mode and restarts the power link after a preset period until the fault is resolved. When it is detected that no overcurrent fault occurs, the current operating state is maintained.

6. The closed-loop control method for power drive and protection according to claim 1, wherein the phase information of the load reflection coefficient is the phase value of the load reflection coefficient, and the phase information lookup table is a preset phase interval mapping table, characterized in that: The step of obtaining phase information of the load reflection coefficient, selecting an inductor / capacitor combination of a variable dynamic impedance matching network according to the phase information, and switching inductor / capacitor parameters through a relay to optimize the transmission standing wave ratio between the output end of the power amplifier and the load includes: collecting the reflected signal between the output end of the power amplifier and the load through a dual directional coupler; extracting a phase value of a reflection coefficient of the reflected signal using a phase detector; According to the phase value, matching the corresponding inductor / capacitor combination number of the variable dynamic impedance matching network in the preset phase interval mapping table, wherein the phase interval mapping table is pre-calibrated by a vector network analyzer and is used to store the mapping relationship between the phase interval and the inductor / capacitor combination of the variable dynamic impedance matching network; The parameters of the variable dynamic impedance matching network are independently switched by at least four groups of relays, wherein the first group of relays is used to control the on and off of the main inductor, the second and third groups of relays are used to control the on and off of the two groups of parallel capacitors respectively, and the fourth group of relays is used to control the on and off of the series inductor; Within the response bandwidth, the transmission standing wave ratio between the output terminal of the power amplifier and the load is optimized to be below a specific ratio.

7. The closed-loop control method for power drive and protection according to claim 1, characterized in that: The step of monitoring the heat sink temperature of the power tube and dynamically adjusting the resistance value of the feedback network based on the temperature deviation to perform over-temperature protection includes: A platinum resistance temperature sensor is mounted on the power tube heat sink, and three sets of wires are electrically connected to the platinum resistance temperature sensor via a bridge structure to eliminate lead resistance errors. The resistance of the first set of wires is included in the high-end bias loop, the resistance of the second set of wires is isolated by the high input impedance of the instrument amplifier, and the resistance of the third set of wires is self-compensated via the bridge. Based on the linear relationship between the resistance value of the platinum resistance temperature sensor and the operating temperature, the real-time temperature is converted into a voltage signal; Dividing the operating temperature range into preset intervals, and dynamically calculating the compensation coefficient using a linear interpolation algorithm according to the interval in which the real-time temperature is located; The resistance value of the feedback network is adjusted in real time by a digital potentiometer to compensate for the gain of the temperature deviation; When the temperature deviation exceeds a second set threshold, starting the heat dissipation device and reducing the output power; When the temperature deviation exceeds the second set threshold, the current operating state is maintained.

8. The closed-loop control method for power drive and protection according to claim 7, characterized in that: The gain for compensating the temperature deviation is calculated by the following formula: in, Indicates real-time temperature The corresponding actual gain is, Indicates the reference temperature The corresponding initial gain is, Indicates the compensation coefficient.

9. The closed-loop control method for power drive and protection according to claim 1, characterized in that: The priority level is that the short circuit protection priority is higher than the over-temperature protection priority, and the over-temperature protection priority is higher than the surge suppression priority.

10. A closed-loop control system for power drive and protection, characterized in that: include: A feedback circuit building module is used to construct a voltage feedback circuit using a high-speed operational amplifier, and form a closed-loop gain structure through the feedback network of the voltage feedback circuit; a short-circuit protection processing module, configured to monitor the load current at the output end of the voltage feedback circuit in real time, and trigger a protection mechanism to shut down the output stage when the load current exceeds a first set threshold value, thereby performing short-circuit protection; A surge suppression processing module, used to limit the power-on surge current through a soft start control circuit to perform surge suppression; A standing wave ratio optimization module is used to obtain phase information of the load reflection coefficient, select an inductor / capacitor combination of the variable dynamic impedance matching network based on the phase information, and switch the inductor / capacitor parameters through a relay to optimize the transmission standing wave ratio between the power amplifier output terminal and the load; A temperature protection processing module is used to monitor the temperature of the power tube heat sink and dynamically adjust the resistance value of the feedback network based on the temperature deviation to perform over-temperature protection; A collaborative execution control module is used to perform collaborative control of the short-circuit protection, the over-temperature protection and the surge suppression according to priority.

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