Upmixing harmonic energy communication system and method based on direct mixing intermodulation signal suppression

Abstract

The invention discloses an up-mixing harmonic energy communication system and method based on direct-mixing intermodulation signal suppression. The communication system includes a receive antenna and an integrated rectified transmitter. The integrated rectified transmitter includes a first branch and a second branch. The invention provides an integrated rectification transmitter with a double-branch structure, an IM2 modulation module is arranged at the tail end of a first branch, and a 90-degree phase shift network is arranged at the head end of a second branch, so that uncontrollable IMD3 signals generated in the first branch and the second branch have a phase difference of 180 degrees and are mutually counteracted; in this way, only the upmixing IMU3 signal which can be controllably and accurately modulated in the first branch is reserved in the uploaded backscattering signal, and more accurate and reliable wireless energy cooperative information transmission is achieved. According to the invention, the frequency of the IM3 signal for communication is close to that of the input dual-tone signal, so that multiplexing can be realized by only needing a small bandwidth in the design of the receiving and transmitting antenna.

Description

Technical Field

[0001] This invention belongs to the field of wireless power transmission technology, specifically relating to an upmixing harmonic energy communication system and method based on direct mixing intermodulation signal suppression. Background Technology

[0002] Radio frequency (RF)-based wireless power transfer (WPT) technology has become widespread in various applications, including portable devices, vehicles, robots, and medical devices. In these applications, limited battery life presents a significant challenge; replacing the batteries of billions of wireless sensors would lead to high environmental and maintenance costs. Reducing power consumption at both the transmitter and receiver ends of sensors is an effective solution for extending battery life and even enabling battery-free devices.

[0003] In common WPT applications such as wireless charging of devices, communication between the power transmitter and receiver is unavoidable to ensure the safety and reliability of wireless charging. There are generally two solutions for this: one is to use an additional communication system outside the WPT system, such as Bluetooth or WiFi. This method relies on the device itself for power. For devices that need wireless charging but are too weak to power on, this solution will not be able to communicate properly, potentially preventing wireless charging from taking place. The other solution is the coordinated transmission of wireless energy and data, such as the Qi protocol. In the Qi protocol, energy transmission and data transmission share the same circuitry. This operation can affect the efficiency of energy transmission and may even directly interrupt wireless energy transmission.

[0004] Third-order intermodulation (3DM) signals are parasitic signals generated when two or more signals are in a linear system. Due to nonlinear factors (typically the nonlinearity of the diode itself in a rectifier module), the second harmonic of one signal mixes with the fundamental frequency of another. 3DM signals have wide applications in various fields. In radar and other high-frequency systems, 3DM signals are widely used to improve system accuracy and stability. In radar systems, the application of 3DM signals helps enhance radar detection and anti-jamming capabilities, enabling accurate target identification and tracking in complex environments. In high-frequency systems, 3DM signals also play an important role, helping to optimize system performance and improve signal transmission efficiency. Furthermore, 3DM signals also have applications in wireless communication. Although second-order intermodulation is more common in wireless communication systems, 3DM can also play a role in certain specific situations, such as improving signal transmission efficiency and anti-jamming capabilities.

[0005] Figure 1 illustrates an existing WPT system utilizing second harmonic backscatter communication, implemented using a harmonic backscatter rectifier (HBR). When the WPT system transmits the fundamental signal to the HBR, the nonlinearity of the diodes generates both a DC signal and a second harmonic signal. Subsequently, based on the offset angle and distance between the two antennas of the WPT transmitter and the HBR, different amplitude second harmonic signals are transmitted. The DC power received by the HBR is then transmitted back to the WPT system for use in the antenna alignment system. However, the use of second harmonics for backscatter communication, coupled with the significant difference between the fundamental frequency and the second harmonic frequency, necessitates a dual-frequency antenna for the WPT system, increasing system cost. Summary of the Invention

[0006] The purpose of this invention is to provide an upmixing harmonic energy communication system and method based on direct mixing intermodulation signal suppression.

[0007] In a first aspect, the present invention provides an upmixing harmonic energy communication system based on direct-frequency intermodulation signal suppression, comprising a receiving antenna and an integrated rectifier transmitter. The integrated rectifier transmitter includes a first branch and a second branch. One end of each of the first and second branches is connected to the output terminal of the receiving antenna. The other end of the first and second branches is used to supply power to a load.

[0008] The first branch includes a first DC blocking capacitor, a first rectifier unit, a first low-pass filter, and an IM2 modulation module with controllable impedance, connected in sequence. The second branch includes a 90° phase-shift network, a second DC blocking capacitor, a second rectifier unit, and a second low-pass filter, connected in sequence. Both the first and second rectifier units include rectifier diodes.

[0009] The controller changes the uploaded signal by adjusting the impedance of the IM2 modulation module.

[0010] Preferably, the first rectifier unit is connected between the connection point of the first DC blocking capacitor and the first low-pass filter and the ground wire; the second rectifier unit is connected between the connection point of the second DC blocking capacitor and the second low-pass filter and the ground wire; the anode of the rectifier diode in the first rectifier unit is connected to the ground wire, and the cathode is connected to the first low-pass filter; the anode of the rectifier diode in the second rectifier unit is connected to the second low-pass filter, and the cathode is connected to the ground wire.

[0011] Preferably, both the first rectifier unit and the second rectifier unit include a matching unit. The matching unit includes a capacitor connected in parallel across the corresponding rectifier diode and an inductor connected in series between the corresponding rectifier diode and ground.

[0012] Preferably, both the first low-pass filter and the second low-pass filter adopt an LC filter structure.

[0013] Preferably, the IM2 modulation module includes an adjustable capacitor C. T By adjusting the adjustable capacitor C T The impedance of the IM2 modulation module is adjusted by changing the capacitance value.

[0014] Preferably, the IM2 modulation module includes an adjustable capacitor C. T Inductor L D Capacitor C D Inductance L D It is connected in series between the output of the low-pass filter and the load. Adjustable capacitor C T Series in inductor L D The end away from the load is connected to ground. Capacitor C D Series in inductor L D The end closest to the load is between the load and the ground wire.

[0015] Preferably, the 90° phase-shifting network employs a microstrip line structure to change the input dual-tone signal and the IMU3 signal generated in the second rectifier unit and reflected in the second low-pass filter by 90°.

[0016] Preferably, a transmitting antenna is also included. The transmitting antenna is used to transmit a two-tone signal and to receive an upmixed third-order intermodulation signal containing uplink information from a receiving antenna.

[0017] Preferably, an energy management unit is also included. The ports of the first and second branches away from the receiving antenna are connected to the input interface of the energy management unit. The energy management unit supplies power to the load.

[0018] Secondly, the present invention provides an upmixer harmonic energy communication method, which uses the aforementioned upmixer IMU3 harmonic energy communication system. The upmixer harmonic energy communication method includes: An external transmitting antenna is used to transmit a two-tone signal. The receiving antenna receives the two-tone signal and transmits it to the first branch and the second branch; the first rectification unit and the second rectification unit convert the two-tone signal into a DC signal to power the load.

[0019] In the first branch, the two-tone signal directly passes through the first rectifier module to generate a first derived signal comprising a second-order intermodulation component and a direct-mixed third-order intermodulation component. The direct-mixed third-order intermodulation component in the first derived signal is reflected at the first low-pass filter, and the second-order intermodulation component, after passing through the first low-pass filter, is modulated by the IM2 modulation module and then reflected. The impedance of the IM2 modulation module is controlled by the receiver's controller, ensuring that the reflected second-order intermodulation component contains the information to be uploaded during modulation. The reflected second-order intermodulation component passes through the first rectifier module again and intermodulates with the two-tone signal to generate an up-mixed third-order intermodulation signal.

[0020] In the second branch, the two-tone signal first passes through a 90° phase-shift network, undergoing a 90° phase change, and then passes through the second rectifier module to generate a second derived signal comprising a second-order intermodulation component and a direct-mixing third-order intermodulation component. The direct-mixing third-order intermodulation component of the second derived signal is reflected at the second low-pass filter, while the second-order intermodulation component is not reflected after passing through the first low-pass filter. The reflected direct-mixing third-order intermodulation component undergoes a 90° phase change when passing through the 90° phase-shift network.

[0021] The direct-mixing third-order intermodulation component reflected in the first branch cancels out the direct-mixing third-order intermodulation component reflected in the second branch due to a 180° phase difference. The up-mixing third-order intermodulation signal reflected in the first branch is transmitted through the receiving antenna and then transmitted to the transmitting antenna.

[0022] The present invention has the following beneficial effects.

[0023] 1. This invention provides an integrated rectifier transmitter with a dual-branch structure, and sets an IM2 modulation module at the tail end of the first branch and a 90° phase shift network at the head end of the second branch, so that the uncontrollable IMD3 signals generated in the first and second branches are 180° out of phase and cancel each other out, so that only the precisely controllable upmixed IMU3 signal in the first branch is retained in the backscattered signal, thereby achieving more accurate and reliable wireless energy cooperative information transmission.

[0024] 2. Both branches in this invention use rectifier modules with a single diode structure. The IMU3 signal uploaded by the rectifier module comes from the derivative signal generated by the diode itself, and no additional oscillation circuit is required. Therefore, this invention has the characteristics of compact and simple structure.

[0025] 3. In this invention, the IM3 signal used for communication has a frequency close to that of the input dual-tone signal. Therefore, this invention requires only a small bandwidth for the transceiver antenna design of the WPT system to achieve multiplexing. Simultaneously, the flexible component parameter settings reduce system construction costs. Furthermore, this invention achieves zero-energy consumption for wireless energy-coordinated information transmission, reducing energy loss and extending system lifespan. Attached Figure Description

[0026] Figure 1 This is a schematic diagram of a traditional WPT system that uses second harmonic backscattering for communication.

[0027] Figure 2 This is a block diagram of the energy and signal transmission system according to an embodiment of the present invention.

[0028] Figure 3 This is a schematic diagram of the rectifier transmitter (IRT) in an embodiment of the present invention.

[0029] Figure 4This is a schematic diagram illustrating the principle of modulating the IM2 signal into the IMU3 signal in the first branch of an embodiment of the present invention.

[0030] Figure 5 This is a superimposed schematic diagram of the two branches of the rectifier transmitter (IRT) in an embodiment of the present invention.

[0031] Figure 6 This is a schematic diagram of the overall structure of the receiving end in an embodiment of the present invention.

[0032] Figure 7 This is a schematic diagram of the equivalent circuit of the rectifier unit in an embodiment of the present invention.

[0033] Figure 8 This is a spectral distribution diagram of the circuit under dual-tone signal excitation in an embodiment of the present invention.

[0034] Figure 9 This is a schematic diagram of the mixing process of three IM3 signals in an embodiment of the present invention.

[0035] Figure 10 In this embodiment of the invention, the frequency is based on oh IM3 A schematic diagram of a 90° microstrip line phase-shifting network and its equivalent two-port network.

[0036] Figure 11 This is a signal flow diagram showing the suppression of the IMD3 signal in the direct mixing of two branches in an embodiment of the present invention.

[0037] Figure 12 This is a flow diagram of the IMU3 signal formation and modulation signal in an embodiment of the present invention.

[0038] Figure 13 In this embodiment of the invention, the IM2 signal has a large capacitance C at its frequency. D Circuit diagram.

[0039] Figure 14 This is a simplified circuit diagram corresponding to the IM2 signal in an embodiment of the present invention.

[0040] Figure 15 The diagram shows the S-parameters of the 90° phase-shifting network in this embodiment of the invention (where (a) corresponds to the Smith chart of the S(1,1) parameters and (b) corresponds to the phase of the S(2,1) parameters).

[0041] Figure 16 This is a comparison diagram of the S(1,1) parameters of the single-branch and dual-branch rectifier transmitter (IRT) in this invention (where (a) is a Smith chart and (b) is a line graph).

[0042] Figure 17 This is a schematic diagram of the DC-DC conversion efficiency of the receiving circuit in an embodiment of the present invention.

[0043] Figure 18 This is a comparison diagram of the IMD3 signal of the single-branch and dual-branch rectifier transmitter (IRT) in this invention (where part (a) corresponds to the amplitude comparison diagram of the IMD3 signal, and part (b) corresponds to the amplitude comparison diagram of the IM3 signal composed of the IMD3 signal and the IMU3 signal). Detailed Implementation

[0044] The present invention will be further described below with reference to the accompanying drawings.

[0045] Example

[0046] The abbreviations used in this embodiment are explained as follows: IM3: The Third-order Intermodulation Signal IMD3: The Direct Third-order Intermodulation (IDM) signal IMU3: The Up Third-order Intermodulation Signal WPT: Wireless Power Transfer RF: Radio Frequency PSN: Phase Shift Network LPF: Low Pass Filter HBR: Harmonic Backscatter Rectifier like Figure 2 As shown, an upmixing IMU3 harmonic energy communication system based on direct-mixing intermodulation signal IMD3 suppression includes a transmitter and a receiver. The transmitter includes a signal source and a transmitting antenna (Tx). The receiver includes a receiving antenna (Rx), an integrated rectified transmitter (IRT), a power management unit (PMU), and a controller (MCU).

[0047] The upmixing IMU3 harmonic communication system provided in this embodiment features both wireless power transfer (WPT) and uplink communication. A two-tone signal (frequency ω1, ω2) provided by a signal source is transmitted from the transmitting antenna (Tx) to the receiving antenna (Rx). The received signal output from the receiving antenna (Rx) is then converted from AC to DC by an integrated rectifier transmitter (IRT) and stored in the power management unit (PMU), which can also power the controller (MCU), completing the wireless power transfer process. Simultaneously, the controller (MCU) can load a digital signal onto the upmixing intermodulation signal IMU3 generated by the integrated rectifier transmitter (IRT). The rectifier transmitter (IRT) can suppress the direct-mixing intermodulation signal IMD3 (a signal with the same frequency as IMU3), improving signal purity. Finally, the upmixing intermodulation signal IMU3 is radiated from the receiving antenna to the transmitting antenna. Since the two-tone signal and the upmixing IMU3 signal have similar frequencies, they can be transmitted and received using the same antenna, realizing the uplink communication process.

[0048] like Figure 3 As shown, the integrated rectifier transmitter (IRT) adopts a dual-branch structure, mainly used to generate and modulate the uplink communication signal IMU3, and suppress the direct-mixer intermodulation signal IMD3. The integrated rectifier transmitter (IRT) includes a first branch and a second branch. The input terminals of both the first and second branches are connected to the output terminal of the receiving antenna (Rx).

[0049] The first branch includes a first DC blocking capacitor C block1 The system includes a first rectifier unit, a first low-pass filter (LPF), and an IM2 modulation module. The first DC blocking capacitor C... block1 The first low-pass filter and the IM2 modulation module are connected in series. The first rectifier unit is connected to the first DC blocking capacitor C. block1 Between the connection point of the first low-pass filter and the ground wire.

[0050] The second branch includes a 90° phase shift network and a second DC blocking capacitor C. Cblock2 The second rectifier unit and the second low-pass filter (LPF). The 90° phase shift network and the second DC blocking capacitor C Cblock2 The second low-pass filter is connected in series with the second rectifier unit. The second rectifier unit is connected to the second DC blocking capacitor C. Cblock2 The connection point of the second low-pass filter is between the ground wire and the ground wire.

[0051] First DC blocking capacitor C block1 Second DC blocking capacitor C Cblock2 The parameters are identical. The first and second low-pass filters are low-pass filters (LPFs) with the same structure and parameters.

[0052] The first rectifier unit includes a first rectifier diode and a first matching unit. The first matching unit includes an inductor L1 and a capacitor C1, used for impedance matching of the junction capacitance and junction resistance inside the first rectifier diode. The cathode of the first rectifier diode is connected to a first DC blocking capacitor C1. block1 The connection point of the first low-pass filter. The capacitor C1 is connected in parallel across the first rectifier diode. The inductor L1 is connected in series between the first rectifier diode and ground.

[0053] The second rectifier unit includes a second rectifier diode and a second matching unit. The parameters of the second matching unit are the same as those of the first matching unit, including inductor L2 and capacitor C2; the positive terminal of the second rectifier diode is connected to the second DC blocking capacitor C. Cblock2 The connection point of the second low-pass filter. The capacitor C2 is connected in parallel across the second rectifier diode. The inductor L2 is connected in series between the second rectifier diode and ground.

[0054] Since the first and second branches exist in parallel, in order for the DC load connected in series with the two branches in the figure to store DC energy, the rectifier diodes of the first rectifier unit and the second rectifier unit must be connected to the branches in opposite directions, so that the DC power after diode rectification flows into the DC load in only one direction.

[0055] The low-pass filter (LPF) adopts an LC filtering structure.

[0056] The IM2 modulation module includes an adjustable capacitor C. T Inductor L D Capacitor C D Inductance L D It is connected in series between the output of the low-pass filter (LPF) and the load. Adjustable capacitor C T Series in inductor L D The end away from the load is connected to ground. Capacitor C D Series in inductor L D The end closest to the load is connected to the ground wire. Meanwhile, capacitor C3 in the first low-pass filter also acts as part of the IM2 modulation module, affecting its impedance. This can be achieved by adjusting the adjustable capacitor C. T The capacitance value is used to adjust the impedance of the IM2 modulation module. The adjustable capacitor C... T The capacitance value is controlled by the upload data signal provided by the controller (MCU). In this embodiment, the load is the power management unit (PMU).

[0057] During operation, the receiving antenna (Rx) receives a dual-tone signal ( oh 1. oh2) After entering the Integrated Rectifier Transmitter (IRT), the signal is split into two paths. The first path, a two-tone signal, passes directly through the first rectifier module, while the second path first undergoes a 90° phase shift network (90° PSN) to change its phase by 90° before flowing into the second rectifier module. The two rectifier modules convert AC to DC, and simultaneously, due to the nonlinear characteristics of the diodes, an IM2 signal (frequency 10 ... oh 1 -oh (2 second-order intermodulation components), IMD3 signal (frequency of (2) oh 1- oh 2) The direct-mixing third-order intermodulation component (IMD3) and other signals. The derived signals generated by the nonlinear characteristics of the rectifier diodes are input to their respective low-pass filters (LPFs). Due to the action of the LPFs, the IMD3 signal is reflected, while the IM2 signal passes through the LPFs and propagates backward. In the first branch, the IM2 signal flows into the IM2 modulator for data modulation and is reflected. In the second branch, the IM2 signal is no longer reflected.

[0058] like Figure 4 As shown, the essence of the IM2 modulator is an embedded impedance Z n ( oh IM2 The IM2 signal generated by the rectifier diode in the first branch is a small current. I IM2 A high impedance is needed to bring it out (by increasing its voltage). I IM2 *Z n ( oh IM2 And modulation is performed; the process of loading data essentially involves changing the capacitor C. T The capacitance value is used to change the embedded impedance Z. n ( oh IM2 The amplitude and phase of the signal flowing into the modulator give it data characteristics. Subsequently, due to high impedance, the modulated IM2 signal is reflected and flows into the diode and the baseband signal. oh 1 or oh 2) Perform intermodulation to generate the modulated IMU3 signal (frequency is (2 oh 1- oh 2) The third-order intermodulation signal of the up-mixing frequency.

[0059] like Figure 5 As shown, for the first branch, the modulated IMU3 signal will be superimposed with the direct-mixer IMD3 and reflected back to the antenna; for the second branch, only the direct-mixer IMD3 is reflected, because the two-tone signal of the second branch ( oh 1. oh2) Both the reflected direct-mixed IMD3 signals and the reflected direct-mixed IMD3 signals pass through a 90° phase shifting network, so that the direct-mixed IMD3 signals reflected to the antenna end by the second branch are 180° out of phase. After superposition, they are suppressed, and only the pure IMU3 signal is used for uplink communication.

[0060] The principle behind the generation of derived signals by the rectifier diodes in the first and second branches is analyzed as follows: First, the equivalent circuit model of the Schottky diode is shown in Figure 7. This model includes the radio frequency source equivalent to the receiving antenna. V g , R g DC blocking capacitor C block Diode junction capacitance C j With junction resistance R j as well as RF choke inductors and DC loads R L Composition. When a dual-frequency signal is input, the node voltage across the diode... V j It can be expressed by equation (2-1): (twenty one) The intermodulation frequency components generated by these two signals through the rectifier diode are as follows: Figure 8 As shown, the third-order intermodulation signal IM3 has two frequencies 2 oh 1- oh 2 and 2 oh 2 - oh 1 To simplify the analysis, we will only consider frequencies of 2 in the following analysis. oh 1- oh The third-order intermodulation signal IM3 of 2.

[0061] Junction current of a typical diode From resistance and current and current capacitance The composition is expressed as equation (2-2): (twenty two) Junction current Expanding the Taylor formula to the third order at a diode bias voltage of 0, we obtain formula (2-3): (twenty three) in This is the reverse saturation current of the diode. Zero-bias junction capacitance Junction potential The gradient coefficient is a constant. , For unit charge, Emission factor Boltzmann's constant, The temperature is in Kelvin.

[0062] For the third-order intermodulation signal IM3 generated by diode mixing, considering the three mixing processes shown in Figure 9, the three components will be analyzed one by one below: A: Direct mixing process In direct mixing, the IMD3 signal is directly generated by the cubic term of equation (2-3). Substituting equation (2-1) into the cubic term of equation (2-2), we can obtain the IMD3 signal component in the direct mixing process as equation (2-4): (twenty four) Where j represents a 90° phase difference, .

[0063] B: Upmixing process As shown in Figure 6, the upmixing process has two steps. In the first step, substituting equation (2-1) into the square term of equation (2-3) yields the current at the difference frequency IM2. , formula (2-5): (2-5) in oh ∆ (IM2)= oh 1- oh 2.

[0064] In the second step, the difference frequency signal oh ∆ (IM2) and the original stimulus oh 1. When mixing is performed, the diode junction voltage is expressed as equation (2-6). Since the difference frequency current and the excitation signal have opposite directions during the mixing process, the difference frequency current in the formula is negative.

[0065] (2-6) in, This is the load impedance under differential frequency conditions.

[0066] Substituting the equation into the square term, we obtain the IM3 signal component generated during the upmixing process as equation (2-7). (2-7) C: Downmixing process The downmixing process is similar to the upmixing process, so the IM3 signal component generated during the downmixing process can be quickly obtained as Equation (2-8). (2-8) Therefore, the IM3 signal generated during the rectification process is the sum of three components, as shown in equation (2-9). (2-9) Regarding the IM3 generated by the diode's nonlinearity, calculations show that the amplitude of the down-mixing IMS3 is smaller than that of the other two mixing methods, so it can be ignored. Only the up-mixing IMU3 and the direct-mixing IMD3 are considered. Since the adjustable IM3 component comes from the up-mixing, using a 90° phase-shift network to suppress the direct-mixing IMD3 can improve signal purity, facilitate the adjustment of the amplitude and phase of the uplink signal IMU3, and benefit the identification of data at the receiving end.

[0067] The structural design of the 90° phase-shift network in this embodiment is as follows: Figure 10 As shown: This embodiment uses a microstrip line as a 90° phase-shift network (PSN). According to transmission line theory, when the input impedance ZB (ZO=ZB) of the circuit following the characteristic impedance ZO of the microstrip line passes through the microstrip line, only a phase change occurs without energy reflection. This can be analyzed from the following professional perspectives: Reflectionless analysis According to transmission line theory, the reflection coefficient at the end (load end) of a microstrip line is given by formula (2-10): (2-10) When ZO=ZB, Γ=0, indicating that no reflected wave is generated. At this time, there is only a single-direction incident wave (traveling wave) on the microstrip line, rather than a standing wave. In the traveling wave state, the voltage and current amplitudes at each point on the microstrip line are uniformly distributed, and energy propagates unidirectionally along the transmission line, with no energy loss during reflection (e.g., return loss is infinite).

[0068] B. 90° Phase Shift Analysis The 90° phase shift can be described by using an equivalent ABCD matrix for the network. This allows us to describe the transmission characteristics of a microstrip line using a transmission line ABCD matrix, ideally in a lossless condition, as shown in equation (2-11). (2-11) When ZO=ZB Substituting into the matrix, we get: This indicates that the input voltage V1 and the output voltage V2 have equal amplitudes and a phase difference of βL, meaning only a phase shift occurs without amplitude attenuation. Therefore, by choosing an appropriate length L, βL = 90° can be achieved, ensuring a 90° phase change without attenuation when the signal passes through this network. It should be noted that this microstrip line operates at a frequency of... oh IM3 Designed under conditions that are close to oh IM3 Signals at a given frequency will undergo a phase change of approximately 90° with virtually no attenuation.

[0069] The structure for suppressing IMD3 in the direct mixer in this embodiment is as follows: Figure 11 As shown: This invention employs a dual-branch microstrip line to suppress IMD3 in direct mixers, as shown in Figure 6. Figure 11 To illustrate, red represents traveling waves, and blue represents reflected waves. For ease of analysis, the phase change of the signal is shown in the diagram, relative to the input signal. oh 1. oh 2. The phase at the input terminal is used as the phase reference node, as shown in Figure 8.

[0070] Traveling wave analysis: When the receiving antenna receives a dual-tone signal oh 1. oh After step 2, the signal splits into two branches for propagation. The first branch directly inputs the two-tone signal to the rectifier diode. Due to the two-tone signal... oh 1. oh 2 and oh IM3 Since the frequencies are not significantly different, the second branch's two-tone signal undergoes a 90° phase shift after passing through the 90° phase shift network. At this point, the two-tone signals of the first and second branches are 90° out of phase, as shown in Figure 8, and then flow into the rectifier diode. After passing through the rectifier diode, the two-tone signal of the first branch, according to formula (2-3), will generate a direct mixer IMD3, with its phase consistent with the phase reference point. For the second branch, since both two-tone signals have a 90° phase shift, it is assumed that:

[0071] Substituting the cubic term into formula (2-3), we obtain the second branch direct-frequency mixer IMD3 as formula (2-12): (2-12) Comparing the direct-mixing IMD3 formula (2-4) generated by the first branch, it can be seen that the phase of the direct-mixing IMD3 of the second branch changes by 90° relative to the phase reference point, that is, the direct-mixing IMD3 signals of the first branch and the second branch are 90° out of phase.

[0072] Reflected wave analysis: To reduce rectification energy loss and achieve higher rectification efficiency, an RF choke is added after the rectifier diode. This design uses a low-pass filter (LPF) composed of LC circuitry, with a cutoff frequency of... oh 0 Much smaller than oh IM3 ( oh 1. oh 2 and oh IM3 The frequencies (which are not significantly different) will not pass through the LPF and will be reflected back. The final direct mixer IMD3 of the second branch passes through the 90° phase shift network again, resulting in a 90° phase change (i.e., a 180° phase change compared to the phase reference point). It is opposite in phase to the direct mixer IMD3 reflected from the first branch, and the two cancel each other out, achieving the suppression effect.

[0073] The upmixer IMU3 control in this embodiment is shown in Figure 12: Regarding the source of the upmixer IMU3, a simple description can be given based on Figure 12 and the above formulas (2-5)-(2-7): Step 1, dual-tone signal oh 1 and oh 2. After passing through the rectifier diode, a difference frequency signal IM2 is generated. oh ∆ ), then IM2 enters the circuit with a large capacitor C E The IM2 modulator is composed of parallel resonant circuits, where IM2 is due to C. T The capacitor is modulated to carry communication information, and due to the high impedance Z of the modulator... n ( oh IM2 This signal will be reflected back to step 2. In step 2, the modulated IM2 will interact with the two-tone signal. oh 1 or oh 2. Secondary intermodulation is performed on the rectifier diode to generate IMU3 carrying information, which is then radiated out from the antenna.

[0074] IM2 introduction Since the current IM2 generated by the diode in the first stage is small and does not meet the communication strength standard, it needs to be passed through a high-impedance embedded load Z. n ( oh IM2 This is used to draw out the voltage (adjust the voltage). Therefore, in this embodiment, a capacitor with a large capacitor C is used, as shown in Figure 12. E The parallel resonant structure serves as the high-load embedded impedance Z n ( oh IM2 The circuit diagram at the IM2 frequency is shown in Figure 13.

[0075] First, let's consider Z at the IM2 frequency. E ( oh IM2 ) is analyzed, and Z is obtained from the impedance calculation formula. E ( oh IM2 ):

[0076] Due to the use of C E It is a large capacitor in the nF range, and the IM2 frequency used in this design is 7.5MHz. Therefore, Z is calculated to be... E ( oh IM2 The impedance of ) is very small, equivalent to a large capacitor C. E Reduced load R L The impact on the circuit can be ignored, and the circuit can be simplified to the parallel resonance analysis shown in Figure 14.

[0077] In this figure, it can be seen that the small IM2 current generated by the nonlinearity of the diode in step 1 flows into the adjustable capacitor C. T In the parallel resonant circuit composed of Z and inductor L, n ( oh IM2 The calculation formula (2-13) for ) is as follows: (2-13) when oh IM2 ≈ oh 0=(LC T When )^(1 / 2), Z n ( oh IM2 It will exhibit a high impedance state, causing V to... IM2 =I IM2 ×Z n ( oh IM2 The capacitance C will increase, which is equivalent to extracting the IM2 signal for modulation. Note: The extraction of IM2 mentioned here is based on a large capacitor C. E In this case, if the large capacitor C E Remove R L The load will disrupt the high impedance state of the parallel resonance, causing it to detune, and IM2 will not be drawn out.

[0078] B. IM2 modulation generates upmixing IMU3 Since the upmixer IMU3 mentioned above is generated based on IM2, IM2 needs to be modulated so that the upmixer IMU3 also carries information. This invention uses a capacitor C connected in parallel resonance for modulation. T IM2 modulation is performed, as shown in Figure 9. From formula (2-13), it can be seen that by changing C... T The value of can change the embedding impedance Z. n ( oh IM2 The amplitude and phase of the signal are used to load data, and the modulated upmixer IMU3 is generated by secondary modulation of the modulated IM2 and the two-tone signal on the diode.

[0079] The two frequencies of the dual-tone signal described in this embodiment oh 1 and oh 2 are 910.5MHz and 918MHz respectively, resulting in a signal frequency of 7.5MHz for IM2 and IM3 for IM3. oh 1- oh 2) The frequency is 903.5MHz, and the target power of the input dual-tone signals is 0dBm (the sum of the two is 3dBm). Because the Avago HSMS286 Schottky diode has a low built-in voltage and low junction capacitance, resulting in low energy loss, it is selected as the rectifier diode. A microstrip line is used as a 90° phase shifter, made of FR4 material, with a characteristic impedance Z... O =50Ω, and other detailed parameters of the circuit design are shown in Figure 6.

[0080] The effect of the 90° phase shifter microstrip line designed in this embodiment is as follows: Figure 15 As shown in Figure 15. To verify that the designed microstrip line has the characteristics of being lossless and having a 90° phase shift, its S-parameters were analyzed using the Advance Design System (ADS, Keysight). Figure 15(a) shows that the Smith chart of the microstrip line from fL (800MHz) to fH (1200MHz) is near the origin, indicating that the characteristic impedance of the designed microstrip line is close to 50Ω, which satisfies the principle of low energy loss of the target signal passing through the microstrip line. Figure 15(b) shows that the phase change of the designed microstrip line S(2,1) is 90°, which can achieve the effect of a 90° phase change of the target signal after passing through the microstrip line.

[0081] In some embodiments, frequency oh 1 and oh The difference between 2 is relatively small, ranging from 4MHz to 10MHz, to facilitate the simultaneous transmission and reception of signals using the same transmitting antenna. This is due to the frequency of dual-tone signals. oh 1 and oh 2 and IM3 signal frequency (2 oh 1- oh 2) The values ​​are close, therefore, both the dual-tone signal and the IM3 signal can undergo a phase change of nearly 90° at the 90° phase shifter.

[0082] The following is a power transfer analysis of the circuit designed in this embodiment. First, the S-parameters and input impedance of the circuit were simulated using the large-signal S-parameter (LSSP) model in the Advance Design System (ADS, Keysight). Figure 16 (a) and (b) Figure 16 show the single and dual-branch input impedances and the S(1,1) parameter, indicating that energy can flow into the circuit. The power conversion efficiency (PCE) as a function of input power is plotted using the Pdc / Pin formula, as shown below. Figure 17 This indicates that the PCE reaches over 55% near the target power (3dBm) of the present invention.

[0083] Finally, the suppression of IMD3 in direct mixing and the control of IMU3 in upmixing were verified. For the analysis of the reflected IM3 signal components, this experiment used the CouplerSingle device from the Advance Design System (ADS, Keysight). Without IM2 control, the CouplerSingle device allowed for observation of the comparison between single and dual-branch direct mixing IMD3, as shown in Figure 18(a). This indicates that the two direct mixing IMD3 components (903MHz and 925.5MHz) using a dual-branch system with a microstrip line as a 90° phase shifter were reduced by 20dBm and 27dBm respectively compared to the single-branch direct mixing IMD3 component without the phase shifter, achieving the effect of suppressing the direct mixing IMD3 component. The verification of the upmixer IMU3 control is achieved by adding the IM2 modulation structure shown in Figure 4. The parallel resonance formed by the IM2 modulation structure, with a resonant frequency of approximately 7.5MHz, reflects the IM2 signal back and performs secondary modulation within the diode according to the formulas (2-5)-(2-7) above to generate the upmixer IMD3 signal. Figure 18 As shown in (b), after IM2 modulation, the two IM3 components (IMD3 + IMU3) increased by 30dBm and 25dBm respectively compared to before modulation, achieving the control of the upmixing IMU3. In summary, this embodiment provides a dual-branch resonant communication circuit design with microstrip lines that achieves direct-mixing IMD3 suppression, thereby improving the purity of the communication signal.

[0084] In some other embodiments, the rectifier unit can adopt various other structures of the rectifier module based on Schottky diodes, such as voltage multiplier structure or single series structure, and the required rectification function can be achieved by using these structures.

[0085] In some other embodiments, the IM2 modulation module does not employ a simple parallel resonant LC structure, but rather other structures that achieve the same purpose.

[0086] In some other embodiments, the 90° phase shift network does not use microstrip lines, but other phase shift structures, such as circularly polarized antennas, which also contain a 90° phase change and can replace microstrip lines.

Claims

1. An up-mixing harmonic energy communication system based on direct mixing intermodulation signal rejection, characterized by: The integrated rectifier transmitter comprises a first branch and a second branch, one end of the first branch and the second branch is connected with the output end of the receiving antenna, and the other end of the first branch and the second branch is used for supplying power for the load. The first branch comprises a first direct-current blocking capacitor, a first rectifier unit, a first low-pass filter and an IM2 modulation module with controllable impedance adjustment connected in sequence; the second branch comprises a 90° phase shift network, a second direct-current blocking capacitor, a second rectifier unit and a second low-pass filter connected in sequence; the first rectifier unit and the second rectifier unit each comprises a rectifier diode.

2. A summing up-mixing harmonic energy communication system based on direct mixing intermodulation signal rejection according to claim 1, characterized in that: The first rectifier unit is connected between the connection point of the first direct-current blocking capacitor and the first low-pass filter and the ground wire; the second rectifier unit is connected between the connection point of the second direct-current blocking capacitor and the second low-pass filter and the ground wire; the anode of the rectifier diode in the first rectifier unit is connected with the ground wire, and the cathode is connected with the first low-pass filter; the anode of the rectifier diode in the second rectifier unit is connected with the second low-pass filter, and the cathode is connected with the ground wire.

3. A summing up-mixing harmonic energy communication system based on direct mixing intermodulation signal rejection according to claim 2, characterized in that: The first rectifier unit and the second rectifier unit each comprises a matching unit; the matching unit comprises a capacitor connected in parallel across the corresponding rectifier diode, and an inductor connected in series between the corresponding rectifier diode and the ground wire.

4. A summing up-mixing harmonic energy communication system based on direct mixing intermodulation signal rejection according to claim 1, characterized in that: The first low-pass filter and the second low-pass filter each adopts an LC filter structure.

5. A summing up-mixing harmonic energy communication system based on direct mixing intermodulation signal rejection according to claim 1, characterized in that: The IM2 modulation module includes an adjustable capacitor C T The impedance of the IM2 modulation module is adjusted by adjusting the capacitance value of the adjustable capacitor C T .

6. A summing up-mixing harmonic energy communication system based on direct mixing intermodulation signal rejection according to claim 1, characterized in that: The IM2 modulation module includes an adjustable capacitor C T , an inductor L D , a capacitor C D ; an inductor L D connected in series between the output of the low-pass filter and the load; an adjustable capacitor C T connected in series between the inductor L D away from the load and the ground; a capacitor C D connected in series between the inductor L D close to the load and the ground.

7. A summing up-mixing harmonic energy communication system based on direct mixing intermodulation signal rejection according to claim 1, characterized in that: The 90° phase shift network adopts a microstrip line structure.

8. A summing up-mixing harmonic energy communication system based on direct mixing intermodulation signal rejection according to claim 1, characterized in that: The system further comprises a transmitting antenna; the transmitting antenna is used for transmitting a dual-tone signal and receiving an up-mixed third-order intermodulation signal containing the uploaded information from the receiving antenna.

9. A summing up-mixing harmonic energy communication system based on direct mixing intermodulation signal rejection according to claim 1, characterized in that: The system further comprises an energy management unit; the ports of the first branch and the second branch away from the receiving antenna are connected with the input interface of the energy management unit; the energy management unit supplies power for the load.

10. An up-mix harmonic energy communication method, characterized by: The system uses the up-mixed harmonic energy communication system as claimed in claim 1; The method comprises the following steps: An external transmitting antenna is used to transmit a dual-tone signal; the receiving antenna receives the dual-tone signal and transmits it to the first branch and the second branch; the first rectifier unit and the second rectifier unit convert the dual-tone signal into a direct current signal to supply power for the load; In the first branch, the dual-tone signal directly passes through the first rectifier module to generate a first derived signal comprising a second-order intermodulation component and a direct-mixed third-order intermodulation component; the direct-mixed third-order intermodulation component in the first derived signal is reflected at the first low-pass filter, and the second-order intermodulation component passes through the first low-pass filter and is reflected after being modulated in the IM2 modulation module; the reflected second-order intermodulation component passes through the first rectifier module again and is intermodulated with the dual-tone signal to generate an up-mixed third-order intermodulation signal; In the second branch, the dual-tone signal first passes through the 90° phase shift network, and then passes through the second rectifier module to generate a second derived signal comprising a second-order intermodulation component and a direct-mixed third-order intermodulation component after a 90° phase change; the direct-mixed third-order intermodulation component in the second derived signal is reflected at the second low-pass filter, and the second-order intermodulation component does not reflect after passing through the first low-pass filter; the reflected direct-mixed third-order intermodulation component is phase-shifted by 90° when passing through the 90° phase shift network; The straight-mixing third-order intermodulation component reflected in the first branch and the straight-mixing third-order intermodulation component reflected in the second branch are mutually cancelled due to the phase difference of 180°, and the up-mixing third-order intermodulation signal reflected in the first branch is emitted through the receiving antenna and transmitted to the transmitting antenna.

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