Irreversible transmission line, pseudo-traveling wave resonance device, and leakage wave antenna device

Abstract

To provide an irreversible transmission line having a two-dimensional structure and capable of uniformly distributing an electromagnetic field in a two-dimensional plane.SOLUTION: An irreversible transmission line includes first and second transmission lines that are orthogonal to each other by configuring basic cells having two transmission line portions that are orthogonal to each other and have parallel branch circuits so as to be juxtaposed and cascaded in orthogonal first and second directions. The transmission line part at a position other than an intersection point of the first and second transmission lines is formed of a material magnetized by spontaneous magnetization or the like so as to have gyro anisotropy by being magnetized in a different direction with respect to a propagation direction of a microwave. In a dispersion curve indicating a relationship between an operating frequency of a microwave signal input to each transmission line and a propagation constant of each transmission line, at least one of the parallel branch circuits of each transmission line is configured such that a wave component propagating along each of the parallel branch circuits in a longitudinal direction of each transmission line has an irreversible characteristic and is asymmetrical with respect to the transmission line portion.SELECTED DRAWING: Figure 5

Description

[Technical Field]

[0001] The present invention relates to a nonreciprocal transmission line having a two-dimensional structure in which two transmission lines are substantially orthogonal to each other, a pseudo-traveling wave resonator using the nonreciprocal transmission line, and a novel leaky wave antenna device using leaky waves from the pseudo-traveling wave resonator. In this specification, microwaves refer to, for example, microwaves in the UHF (Ultra High Frequency) band or higher, millimeter waves, quasi-millimeter waves, and terahertz waves. Hereinafter, a "nonreciprocal composite right- and left-handed transmission line" will be referred to as a "nonreciprocal composite right / left-handed transmission line" or a "nonreciprocal CRLH transmission line." [Background technology]

[0002] Recently, research into left-handed transmission (LHT) lines, which swap the inductance and capacitance of conventional distributed constant lines, has been intensifying (see, for example, Non-Patent Documents 1-3). Because left-handed transmission line circuits exhibit unique characteristics such as backward wave characteristics and lens action, expectations are high for new microwave circuit elements. However, microwave resonators and microwave circuits using them, in particular, have a problem in that the resonant frequency is determined depending on the line length, resulting in an increase in the size of the device configuration depending on the resonant frequency.

[0003] To solve this problem, the present inventors proposed a transmission-line microwave circuit in Patent Document 4, which can be significantly miniaturized compared to conventional techniques and has unique effects. However, when the transmission-line microwave circuit is applied to an antenna device, a leaky wave antenna device can be realized that can form a main beam and control the main beam direction using leaky waves from a transmission line such as a left-handed or right-handed transmission line, but there is a problem in that the radiation gain is relatively small. On the other hand, there is a problem in that a relatively long line length is required to obtain a large radiation gain.

[0004] To solve this problem, the present inventor proposed in Patent Document 5 a leaky-wave antenna device that has a larger radiation gain than conventional devices and can be made smaller in size for the same radiation gain. This leaky-wave antenna device includes a CRLH transmission line configured by cascading, between first and second ports, at least one unit cell having a series branch circuit equivalently including a capacitive element, a parallel branch circuit equivalently including an inductive element, and at least one transmission line portion that is a nonreciprocal transmission line portion or a reciprocal transmission line portion. The leaky-wave antenna device further includes a reflecting impedance circuit connected to one port of the CRLH transmission line and operating so that the impedance seen from the port toward the reflecting impedance circuit becomes substantially zero or infinite at a predetermined operating frequency. However, this leaky-wave antenna device has difficulty achieving both miniaturization and high efficiency, and suffers from the problem of beam squint, in which the beam direction changes with frequency fluctuations. Previous attempts have been made to solve this problem by using a nonreciprocal metamaterial. In addition, in a beam scanning antenna device made of a non-reciprocal metamaterial, it has become possible to construct a small, highly efficient beam scanning antenna device with reduced beam squint, but there is a problem in that it is not possible to perform both transmitting and receiving operations at the same time.

[0005] Furthermore, to solve the above problems, the present inventors proposed in Patent Document 6 a leaky wave antenna device capable of beam scanning while simultaneously transmitting and receiving to a target radio station, i.e., while maintaining a state of two-way communication. The leaky wave antenna device is configured by cascading a pair of CRLH transmission lines, each of which includes a plurality of unit cells, each of which has a series branch circuit, a parallel branch circuit, and a transmission line portion, between a pair of ports, and arranging the pair of CRLH transmission lines substantially parallel to each other and adjacent to each other so as to be electromagnetically coupled to each other. Each unit cell of each CRLH transmission line has a propagation constant with a different sign and substantially the same absolute value on its dispersion curve. At an operating frequency, when a first microwave signal is input from a first port along the first CRLH transmission line and propagates toward a second port or in the reverse direction, the microwave signal is radiated as a first leaky wave at a predetermined radiation angle, while a second microwave signal incident on the second CRLH transmission line at an incident angle equal to the radiation angle can be received. [Prior art documents] [Patent documents]

[0006] [Patent Document 1] Japanese Patent Application Publication No. 5-183329 [Patent Document 2] Japanese Patent Application Laid-Open No. 2005-124038 [Patent Document 3] Japanese Patent Application Laid-Open No. 2005-160009 [Patent Document 4] International Publication No. 2008 / 111460 [Patent Document 5] International Publication No. 2011 / 024575 [Patent Document 6] Patent application 2023-055441 [Non-patent literature]

[0007] [Non-Patent Document 1] A. Porokhnyuk et al, "Design of nonreciprocal CRLH metamaterial for non-squinting leaky-wave antenna," in IEEE MTT-S International Microwave System Digest, TH1H-1, pp. 1-3, June 2013. [Non-patent document 2] Hidefumi Yasuda et al., "Equivalent circuit of two-dimensional nonreciprocal metamaterials," IEICE Technical Report, Vol. 122, No. 256, EMT2022-62, pp. 99-104, published November 10, 2022 Summary of the Invention [Problem to be solved by the invention]

[0008] The above-mentioned nonreciprocal coupled transmission lines according to the prior art have a one-dimensional structure, and as a result, when applied to beam scanning, beam scanning is only possible in one direction, and although there have been cases in which extension to a two-dimensional structure has been explained using an equivalent circuit model (see Non-Patent Document 2), no specific structure that makes this possible has been proposed. If a conventional one-dimensional nonreciprocal coupled transmission line is simply made two-dimensional by turning it into a mesh, the directional dependence of the transmission coefficient occurs due to the nonreciprocity, as in a circulator, making it difficult to distribute the electromagnetic field uniformly within a two-dimensional plane.

[0009] Furthermore, in leaky wave antenna devices, when the antenna size is small (the line length used in the antenna's radiation section is short), the radiation efficiency decreases due to power consumption in the matching circuit inserted at the end.Regardless of whether the beam scanning method is by changing the operating frequency or by electronically changing the circuit constants, there is a problem of beam squint, in which the beam direction changes as the operating frequency changes.

[0010] The object of the present invention is to solve the above problems and to provide a non-reciprocal transmission line that has a two-dimensional structure and can uniformly distribute an electromagnetic field within a two-dimensional plane, a pseudo-traveling wave resonator using the same, and a leaky wave antenna device using the same. [Means for solving the problem]

[0011] A nonreciprocal transmission line according to one aspect of the present disclosure includes: A nonreciprocal transmission line comprising first and second nonreciprocal transmission lines (first and second transmission lines) that are orthogonal to each other, configured by arranging and cascading at least one basic cell in a first and second direction that are orthogonal to each other, each basic cell having a parallel branch circuit that equivalently includes an inductive element or a capacitive element and having two transmission line portions that are substantially orthogonal to each other, a portion of the transmission line excluding an intersection of the first and second transmission lines is made of a material that is magnetized in a direction different from the propagation direction of microwaves and is magnetized spontaneously or by an external magnetic field so as to have gyroscopic anisotropy; In a dispersion curve showing the relationship between the operating frequency of a microwave signal input to each of the transmission lines and the propagation constant of each of the transmission lines, at least any of the parallel branch circuits of each of the transmission lines is configured so that wave components propagating along each of the parallel branch circuits in the longitudinal direction of each of the transmission lines have non-reciprocal characteristics and become asymmetrical circuits with respect to the transmission line portion. [Effects of the Invention]

[0012] Therefore, the nonreciprocal transmission line according to one aspect of the present invention has a two-dimensional structure and can distribute the electromagnetic field uniformly within a two-dimensional plane. [Brief explanation of the drawings]

[0013] [Figure 1A] 1 is a perspective view showing the configuration of a basic cell of a nonreciprocal transmission line according to a first embodiment. FIG. [Figure 1B] FIG. 1B is a top view of the base cell of FIG. 1A. [Figure 1C]1B is a top view of the basic cell of FIG. 1A with the strip conductors 20 and 30 and the stub conductors 21, 22, 31, and 32 removed. [Figure 2A] FIG. 10 is an external perspective view showing the configuration of a basic cell of a nonreciprocal transmission line according to a second embodiment. [Figure 2B] FIG. 2B is a top view of the base cell of FIG. 2A. [Figure 2C] 2B is a top view of the basic cell of FIG. 2A with the strip conductors 20 and 30 and the stub conductors 21, 22, 31, and 32 removed. [Figure 3] FIG. 10 is a top view of a standard cell according to a third embodiment. [Figure 4] FIG. 1 is a top view of a standard cell according to Comparative Example 1. [Figure 5] FIG. 10 is an external perspective view showing the configuration of a nonreciprocal transmission line according to a fourth embodiment. [Figure 6] FIG. 10 is an external perspective view showing the configuration of a reciprocal transmission line according to Comparative Example 2. [Figure 7A] 6 is a graph showing the transmission characteristics of the simulation results when no DC magnetic field is applied in FIG. 5. [Figure 7B] 6 is a graph showing dispersion characteristics of the simulation results when no DC magnetic field is applied in FIG. 5. [Figure 8A] 6 is a graph showing transmission characteristics of a simulation result when a DC magnetic field is applied to the nonreciprocal transmission line of FIG. 5. [Figure 8B] 6 is a graph showing dispersion characteristics of a simulation result when a DC magnetic field is applied to the nonreciprocal transmission line of FIG. 5. [Figure 9A] 7 is a graph showing transmission characteristics of a simulation result when a DC magnetic field is applied to the reciprocal transmission line of FIG. 6. [Figure 9B] 7 is a graph showing dispersion characteristics of a simulation result when a DC magnetic field is applied to the reciprocal transmission line of FIG. 6. [Figure 10] FIG. 10 is an external perspective view showing the configuration of a pseudo traveling wave resonator according to a fifth embodiment. [Figure 11]FIG. 10 is an exploded perspective view showing the configuration of a leaky wave antenna device according to a sixth embodiment. [Figure 12A] 12 is a top view of a first layer L1 of the leaky wave antenna device of FIG. [Figure 12B] 12 is a top view of the second layer L2 of the leaky wave antenna device of FIG. [Figure 12C] 12 is a top view of the third layer L3 of the leaky wave antenna device of FIG. [Figure 12D] 12 is a top view of a fourth layer L4 of the leaky wave antenna device of FIG. [Figure 12E] 12 is a bottom view of a fifth layer L5 of the leaky wave antenna device of FIG. [Figure 13] 12 is a graph showing the return loss characteristics of the simulation results at the input port of the leaky wave antenna device of FIG. 11 in a non-magnetized state. [Figure 14] 12 is a graph showing a radiation pattern of a normalized simulation result of the leaky wave antenna device of FIG. 11 in an unmagnetized state. [Figure 15] 12 is a graph showing the return loss characteristics of the simulation results at the input port in the magnetized state of the leaky wave antenna device of FIG. 11. [Figure 16] 12 is a graph showing a radiation pattern of a normalized simulation result in the magnetization state of the leaky wave antenna device of FIG. 11. [Figure 17A] 12 is a graph showing the return loss characteristics of an experiment result at an input port of the leaky wave antenna device of FIG. 11 in a non-magnetized state. [Figure 17B] 12 is a graph showing a radiation pattern of a normalized experimental result of the leaky wave antenna device of FIG. 11 in an unmagnetized state. [Figure 18A] 12 is a graph showing the return loss characteristics of the experimental results at the input port in the magnetized state of the leaky wave antenna device of FIG. 11. [Figure 18B] 12 is a graph showing radiation patterns of normalized experimental results for the magnetization states of the leaky wave antenna device of FIG. 11. [Figure 19] FIG. 11 is an external perspective view showing the configuration of a leaky wave antenna device according to a seventh embodiment. [Figure 20] 20 is a graph showing the return loss characteristics of the simulation results of the leaky wave antenna device of FIG. 19. [Figure 21A] 20 is a graph showing a radiation pattern (radiation characteristics in a φ=90° plane) resulting from a simulation of the leaky wave antenna device of FIG. 19. [Figure 21B] 20 is a graph showing a radiation pattern (radiation characteristics in a φ=0° plane) resulting from a simulation of the leaky wave antenna device of FIG. 19. [Figure 22] FIG. 13 is a top view of a basic cell of a nonreciprocal transmission line according to an eighth embodiment. [Figure 23] FIG. 13 is a top view of a basic cell of a nonreciprocal transmission line according to a ninth embodiment. [Figure 24] 23 is an exploded perspective view showing the configuration of a leaky wave antenna device using the nonreciprocal transmission line of FIG. 22. FIG. [Figure 25A] 25 is a top view of a first layer L1 of the leaky wave antenna device of FIG. 24. FIG. [Figure 25B] 25 is a top view of the second layer L2 of the leaky wave antenna device of FIG. 24. FIG. [Figure 25C] FIG. 25 is a top view of the third layer L3 of the leaky wave antenna device of FIG. 24. [Figure 25D] FIG. 25 is a top view of a fourth layer L4 of the leaky wave antenna device of FIG. [Figure 26] FIG. 20 is a top view of a leaky wave antenna device according to a tenth embodiment. [Figure 27A] FIG. 22 is a top view of a basic cell of a nonreciprocal transmission line according to an eleventh embodiment. [Figure 27B] 27B is a top view of the basic cell of FIG. 27A with the strip conductors 20 and 30 and the stub conductors 21, 22, 32, and 34 removed. [Figure 28] FIG. 12 is an external perspective view showing a shielding conductor plate 80 (embodiment 12) placed on the top surface of the leaky wave antenna device using the nonreciprocal transmission line according to embodiments 1 to 11. [Figure 29] FIG. 13 is an external perspective view showing a shielding conductor plate 80A (embodiment 13) placed on the top surface of the leaky wave antenna device using the nonreciprocal transmission line according to embodiments 1 to 11. [Figure 30] 28 or 29 is placed on the non-reciprocal transmission line according to any one of the first to eleventh embodiments, and then a strip array antenna is placed thereon. FIG. [Figure 31] FIG. 23 is a top view of a leaky wave antenna device according to a fourteenth embodiment. [Figure 32] 32A and 32B are diagrams illustrating an AC magnetic field generated by stub conductors 21A and 22A of the leaky wave antenna device of FIG. 31. [Figure 33] 32 is a graph showing the return loss characteristics of the simulation results of the leaky wave antenna device of FIG. 31. [Figure 34] 32 is a graph showing a radiation pattern (radiation characteristics in a φ=90° plane) resulting from a simulation of the leaky wave antenna device of FIG. 31. [Figure 35] 32 is a graph showing a radiation pattern (radiation characteristics in a φ=0° plane) resulting from a simulation of the leaky wave antenna device of FIG. 31. [Figure 36] FIG. 11 is a top view of a leaky wave antenna device according to a third comparative example. [Figure 37] 37 is a diagram illustrating an AC magnetic field generated by stub conductors 21 and 22 of the leaky wave antenna device of FIG. 36. FIG. [Figure 38] 37 is a graph showing a radiation pattern (radiation characteristics in a φ=45° plane) resulting from a simulation of the leaky wave antenna device of FIG. 36. [Figure 39] FIG. 22 is a top view of a leaky wave antenna device according to a first modification of the fourteenth embodiment. [Figure 40] FIG. 20 is a top view of a leaky wave antenna device according to a second modification of the fourteenth embodiment. [Figure 41] 41A and 41B are diagrams illustrating an AC magnetic field generated by stub conductors 21C and 22C of the leaky wave antenna device of FIG. [Figure 42] 40 is a diagram illustrating an AC magnetic field generated by stub conductors 21A, 22A, 31A, and 32A of the leaky wave antenna device of FIG. 39. FIG. [Figure 43]41 is a graph showing the return loss characteristics of the simulation results of the leaky wave antenna device of FIG. 40. [Figure 44] FIG. 40 is a graph showing a radiation pattern (in a φ=45° plane) in the simulation results when no DC magnetic field is applied. [Figure 45] FIG. 40 is a graph showing a radiation pattern (in a φ=45° plane) in a simulation result when a saturation magnetization of 175 mT and an internal DC magnetic field of 90 mT are applied. [Figure 46] FIG. 20 is a top view of a leaky wave antenna device according to a third modification of the fourteenth embodiment. [Figure 47] 47 is a diagram illustrating a magnetic field generated by stub conductors 21 and 22D of the leaky wave antenna device of FIG. 46. FIG. [Figure 48] 47 is a graph showing the return loss characteristics of the simulation results of the leaky wave antenna device of FIG. 46. [Figure 49] 47 is a graph showing a radiation pattern (radiation characteristics in a φ=45° plane) resulting from a simulation of the leaky wave antenna device of FIG. 46. [Figure 50] FIG. 23 is a top view of a leaky wave antenna device according to a fifteenth embodiment. [Figure 51] FIG. 51 is a top view of one basic cell 1E in the leaky wave antenna device of FIG. [Figure 52] 51 is a graph showing the return loss characteristics of the simulation results of the leaky wave antenna device of FIG. 50. [Figure 53] 51 is a graph showing a radiation pattern (radiation characteristics in a φ=45° plane) resulting from a simulation of the leaky wave antenna device of FIG. 50. [Figure 54] 11 is a graph showing the return loss characteristics of the simulation results of the leaky wave antenna device according to Comparative Example 4. [Figure 55] 11 is a graph showing a radiation pattern (radiation characteristics in a φ=45° plane) of a simulation result of the leaky wave antenna device according to Comparative Example 4. [Figure 56] FIG. 20 is a top view of one basic cell in the leaky wave antenna device according to the sixteenth embodiment. [Figure 57] FIG. 57 is a cross-sectional view taken along line AA' in FIG. 56. [Figure 58] FIG. 20 is a cross-sectional view of one basic cell in a leaky wave antenna device according to a seventeenth embodiment. [Figure 59] FIG. 23 is a cross-sectional view of one basic cell in a leaky wave antenna device according to a modified example of the seventeenth embodiment. DETAILED DESCRIPTION OF THE INVENTION

[0014] Hereinafter, embodiments and modifications of the present invention will be described with reference to the drawings. Note that the same or similar components are denoted by the same reference numerals. Furthermore, the following numerical examples are merely examples of embodiments and may be changed to other numerical values.

[0015] (Inventor's Knowledge) According to the findings of the inventors, it is believed that by specifying the insertion locations of rotationally anisotropic magnetic materials such as ferrite, it is possible to extend the one-dimensional nonreciprocal metamaterial transmission line proposed in previous research to a two-dimensional structure by forming it into a mesh. The electromagnetic field distribution within this structure has a nearly uniform intensity distribution, and any wave vector in the leaky wave region can be uniformly distributed. Therefore, when applied to a leaky wave radiation antenna, it is possible to form a main lobe (or main beam) in the desired direction regardless of the feed position and the direction of the transmitted power. Furthermore, it becomes possible to independently control the combination of azimuth and elevation angles with only a few control variables.

[0016] In the present invention, identifying the insertion location of the ferrite material in a nonreciprocal metamaterial device is the key to expanding the dimensions of the structure of the present invention. According to the findings of the inventors, the electromagnetic field distribution within a two-dimensional nonreciprocal metamaterial structure can be uniformly distributed in intensity and with any wave number vector. Therefore, when applied to a leaky wave radiation antenna, it is possible to form a main lobe (or main beam) in a desired direction regardless of the feed position and the direction of the transmitted power. Furthermore, with only a few control variables, it is possible to dynamically control the combination of the azimuth and elevation angles of the beam with low power and low latency. Specific embodiments are described below.

[0017] (Embodiment) In the following embodiments, a method for constructing a nonreciprocal transmission line, which is a two-dimensional nonreciprocal metamaterial device, is described. First, a method for constructing a two-dimensional nonreciprocal metamaterial that allows arbitrary setting of the spatial distribution of wave vectors, i.e., the spatial phase distribution, is proposed, and the in-plane propagation characteristics and leaky-wave radiation characteristics are investigated. Next, the basic structure of the two-dimensional nonreciprocal transmission line according to the embodiment is described, followed by the transmission characteristics and dispersion characteristics of a structure with a finite number of cells in the principal axis direction. Furthermore, a two-dimensional pseudo-traveling-wave resonator is constructed using a two-dimensional nonreciprocal transmission line shielded by a metal plate, and numerical simulation results for the reflection characteristics and electromagnetic field distribution in a resonant state are presented. Furthermore, the use of leaky-wave radiation from an open-type two-dimensional pseudo-traveling-wave resonator with the metal shielding plate removed is described, and a power supply method is proposed.

[0018] (Embodiment 1) First, in the first embodiment, a basic cell, which is the basic configuration of the proposed two-dimensional nonreciprocal transmission line, is shown.

[0019] FIG. 1A is an external perspective view showing the configuration of an elementary cell of the nonreciprocal transmission line according to the first embodiment. FIG. 1B is a top view of the elementary cell of FIG. 1A, and FIG. 1C is a top view of the elementary cell of FIG. 1A with strip conductors 20 and 30 and stub conductors 21, 22, 31, and 32 removed. A two-dimensional nonreciprocal transmission line is configured by arranging a plurality of elementary cells (also referred to as "unit cells") in the x-axis direction and the y-axis direction. However, the number of elementary cells arranged in the x-axis direction or the y-axis direction may be at least one. In FIGS. 1A to 1C, the longitudinal directions (major axis directions) of strip conductors 20 and 30 (described later) are formed parallel to the x-axis and y-axis, respectively, and the thickness direction of a dielectric substrate 10 is configured parallel to the z-axis.

[0020] 1A, the basic cell has a plurality of strip conductors 20 and 30 formed substantially orthogonal to each other on a dielectric substrate 10 having a ground conductor 11 formed on the back surface thereof, thereby forming a two-dimensional square lattice structure of microstrip lines. That is, the dielectric substrate 10 is sandwiched between the strip conductors 20 and the ground conductor 11 to form a microstrip line in the x-axis direction, and the dielectric substrate 10 is sandwiched between the strip conductors 30 and the ground conductor 11 to form a microstrip line in the y-axis direction.

[0021] Ferrite blocks 40 (e.g., rectangular parallelepiped shapes) made of magnetic material are embedded in predetermined positions in the dielectric substrate 10. Here, the ferrite blocks 40 are not placed at positions on the dielectric substrate 10 near the intersections of the square lattice of the strip conductors 20, 30, but are embedded in positions in the dielectric substrate 10 in the center parts of the branches away from the intersections.

[0022] In addition, inductive stub conductors 21, 22; 31, 32 and via conductors 23, 24; 33, 34 are inserted as parallel branches on both sides of the strip conductors 20, 30 that form each lattice branch. Here, the via conductors 23, 24; 33, 34 are formed to penetrate the dielectric substrate 10 in the thickness direction, and connect each of the inductive stub conductors 21, 22; 31, 32 to the ground conductor 11. In other words, the side surfaces of the strip conductor 20 are (1) connected to the ground conductor 11 through the stub conductor 21 and the via conductor 23; (2) connected to the ground conductor 11 through the stub conductor 22 and the via conductor 24; (3) connected to the ground conductor 11 through the stub conductor 31 and the via conductor 33; (4) Connected to the ground conductor 11 through the stub conductor 32 and the via conductor 34.

[0023] The stubs of the stub conductors 21, 22; 31, 32 connected to the via conductors 23, 24; 33, 34 respectively have, for example, inductivity, but the present invention is not limited to this and they may have capacitive properties. The stub conductors 21, 22; 31, 32 are inserted asymmetrically according to the magnitude of the non-reciprocal phase shift characteristic of the wave component propagating along each branch in the direction of the main axis (x-axis or y-axis) of the strip conductors 20, 30 (i.e., the longitudinal direction). That is, s1 <l s2 ,l s3 <l s4 (Note that the length relationship may be reversed.) s1 ,l s2 ,l s3 ,l s4 are the longitudinal lengths of 31, 32, 21, and 22, respectively. However, to prevent fluctuations in the operating frequency of the nonreciprocal transmission line, the device structure is set up so that the parallel resonant frequency of the parallel branches (the frequency at which the effective dielectric constant is zero) is kept constant, that is, the combined inductance (or combined admittance) of the stubs inserted in each branch is kept constant.

[0024] When the ferrite block 40 embedded in the dielectric substrate 10 is made of a soft magnetic material, an internal DC magnetic field H0 is applied perpendicular to the dielectric substrate 10, and the saturation magnetization M s The soft magnetic ferrite block 40 may be replaced with a ferrite block having spontaneous magnetization in the vertical direction. Also, a predetermined magnetic field may be applied to the unmagnetized magnetic material from an external magnetic field generator.

[0025] The analytical model used in the numerical simulation according to this embodiment is shown below. The dielectric substrate 10 of the basic cell has a square shape with a side length p, but the present invention is not limited to this, and it may have a rectangular shape (rectangle). The dielectric substrate 10 has a dielectric constant ε d The ferrite block 40 is made of soft magnetic yttrium iron garnet (YIG) polycrystal with a relative permittivity of ε f =15, saturation magnetization μ0M s= 175mT, internal DC magnetic field μ0H0 = 90mT, magnetic loss μ0ΔH = 5mT, and its size is width w f = 2.0 mm, length l f The structural parameters of the basic cell are the width w of the central line strip conductors 20 and 30 = 2.0 mm, the insertion positions l of the inductive stub conductors 21, 22; sp = 0.5 mm, cell length p = 10 mm, width w of inductive stub conductors 21, 22; 31, 32 s = 1.0 mm, and the radius r of the via conductors 23, 24; 33, 34 is set to 0.4 mm.

[0026] Length l of the inserted inductive stub conductors 21, 22; 31, 32 s1 , l s2 , l s3 , l s4 Generally, the inductances (admittances) of the stubs inserted in each branch must be kept constant, so they are not independent of each other and are set to the length l s1 and length l s2 Between and length l s3 and length l s4 There are constraints between them. The inserted stub conductors 21, 22; 31, 32 do not need to be short-circuited with via conductors or the like, and open-ended stubs without via conductors may be used. At least one of the stub conductors 21, 22; 31, 32 may be configured as a capacitive stub, but in order to achieve parallel resonance in the parallel branches, the combined admittance of the stub conductors 21, 22; 31, 32 of each branch must exhibit inductivity. By moving the insertion positions of the stub conductors 21, 22; 31, 32 toward the boundary of the basic cell, basic cell structures such as those shown in Figures 2A to 2C can also be considered.

[0027] The nonreciprocal transmission line according to the first embodiment configured as described above comprises first and second nonreciprocal transmission lines (first and second transmission lines) that are orthogonal to each other, configured by juxtaposing and cascading at least one basic cell in first and second directions that are orthogonal to each other, each having a parallel branch circuit equivalently including an inductive element and two transmission line portions that are substantially orthogonal to each other. Each of the parallel branch circuits on both sides of each transmission line is configured so that a wave component propagating along each parallel branch circuit in the longitudinal direction of each transmission line has a nonreciprocal characteristic (to be described later) and forms an asymmetric circuit with respect to the transmission line portion.

[0028] (Embodiment 2) Fig. 2A is an external perspective view showing the configuration of a basic cell of a nonreciprocal transmission line according to embodiment 2. Fig. 2B is a top view of the basic cell of Fig. 2A, and Fig. 2C is a top view of the basic cell of Fig. 2A with strip conductors 20 and 30 and stub conductors 21, 22, 31, and 32 removed. The basic cell according to embodiment 2 differs from the basic cell according to embodiment 1 in the following respects. (1) The stub conductors 21, 22; 31, 32 and the via conductors 23, 24; 33, 34 are formed with a width half that of the first embodiment, and are formed as a single conductor together with each adjacent conductor.

[0029] The nonreciprocal transmission line according to the second embodiment configured as described above comprises first and second nonreciprocal transmission lines (first and second transmission lines) that are orthogonal to each other, configured by juxtaposing and cascading at least one basic cell in first and second directions that are orthogonal to each other, each having a parallel branch circuit equivalently including an inductive element, and each having two transmission line portions that are substantially orthogonal to each other, as in the first embodiment. Each of the parallel branch circuits on both sides of each transmission line is configured so that a wave component propagating along each parallel branch circuit in the longitudinal direction of each transmission line has a nonreciprocal characteristic, which will be described later, and forms an asymmetric circuit with respect to the transmission line portion.

[0030] In the following, we demonstrate through numerical calculations that the wave vectors of electromagnetic wave components propagating through a 2D nonreciprocal transmission line, i.e., the spatial phase gradient distribution, can be freely controlled to a desired direction and magnitude by independently controlling the presence or absence of stub conductors 21, 22; 31, 32 for each branch of the line parallel to the orthogonal x- and y-axes, or by controlling the asymmetry of the structure due to differences in stub shape. Furthermore, by opening the boundaries of the finite-cell 2D nonreciprocal transmission line structure, we construct an open-ended 2D pseudo-traveling-wave resonator and investigate the electromagnetic field distribution at resonance. In this study, we consider the case where a metallic shielding plate (shielding conductor plate) is placed parallel to and at a fixed distance from the 2D microstrip line square lattice structure to eliminate the influence of leaky wave radiation loss from the microstrip line. Furthermore, to investigate the leaky wave radiation characteristics into free space, the above-mentioned metal shielding plate (shielding conductor plate) is removed and the elevation and azimuth angle manipulation of the leaky wave radiation beam from the open-type two-dimensional pseudo-traveling wave resonator is investigated.

[0031] (Embodiment 3) 3 is a top view of the basic cell according to embodiment 3. In FIG. 3, the basic cell according to embodiment 3 differs from the basic cell according to embodiment 1 in the following points. (1) The stub conductor 21 and the via conductor 23 are connected to the side surface of the strip conductor 20 that constitutes the microstrip line, but the stub conductor 22 and the via conductor 24 are not connected. (2) The stub conductor 32 and the via conductor 34 are connected to the side surface of the strip conductor 30 that constitutes the microstrip line, but the stub conductor 31 and the via conductor 33 are not connected. The differences will be explained in detail below.

[0032] In Fig. 3, an inductive stub conductor 32 and a via conductor 34 are inserted on the +x-direction side of a microstrip line branch formed on a dielectric substrate 10 having a perpendicularly magnetized ferrite block 40 extending parallel to the y-axis, and an inductive stub conductor 21 and a via conductor 23 are inserted on the -y-direction side of a line branch extending parallel to the x-axis. The parameters of the basic cell structure are basically the same as those of the basic cell according to the first embodiment, and l s = 2.0 mm. This is because in the basic structure shown in Figs. 1A to 1C, s1 =l s3 = 0 mm (removing short-circuit via conductors 24 and 33), l s2 =l s4 = 2.0 mm.

[0033] In the basic cell according to the third embodiment shown in FIG. 3, the width of each stub conductor 21, 32 and via conductors 23, 34 may be halved as in the second embodiment to be shared with an adjacent basic cell, and a stub conductor 22 and a via conductor 24 may be added (hereinafter referred to as a modified example of the third embodiment). In the modified example of the third embodiment, the inductive stub conductors 21, 32 are inserted asymmetrically only in the +x direction with respect to the microstrip line branches parallel to the y axis, and unlike the third embodiment, the inductive stub conductors 22, 21 are inserted symmetrically in the ±y directions with respect to the branches parallel to the x axis. This structural parameter is the same as that in the structure according to the second embodiment. s1 = 0 mm (removing the short-circuit via conductor 24), length l s2 =l sx = 1.3 mm, length l s3 =l s4 =l sy = 2.6 mm.

[0034] The nonreciprocal transmission line according to the third embodiment configured as described above comprises first and second nonreciprocal transmission lines (first and second transmission lines) that are orthogonal to each other, configured by juxtaposing and cascading at least one basic cell in first and second directions that are orthogonal to each other, each having a parallel branch circuit equivalently including an inductive element, and each having two transmission line portions that are substantially orthogonal to each other, as in the first or second embodiment. Each of the parallel branch circuits on both sides of each transmission line is configured so that a wave component propagating along each parallel branch circuit in the longitudinal direction of each transmission line has a nonreciprocal characteristic described later and forms an asymmetric circuit with respect to the transmission line portion.

[0035] FIG. 4 is a top view of a basic cell according to Comparative Example 1. In this structure, inductive stub conductors 21, 22 and 31, 32 are inserted symmetrically with respect to microstrip line branches parallel to the x-axis and y-axis, respectively. The structural parameters of this basic structure shown in FIGS. 1A to 1C are the length l s1 =l s2 =l sx = 2.5 mm, length l s3 =l s4 =l sy = 2.5 mm.

[0036] (Embodiment 4) 5 is an external perspective view showing the configuration of a nonreciprocal transmission line according to embodiment 4. With reference to FIG. 5, the analysis of transmission characteristics in a principal axis direction finite cell structure will be described below.

[0037] In order to investigate the nonreciprocal phase shift characteristics of a two-dimensional nonreciprocal transmission line, an analysis is performed on a structure in which only a finite number of cells are arranged in the main axis direction. As an example, the basic cell according to the third embodiment is arranged with n cells in the y-axis direction. y = 5, and the number of cells in the x-axis direction is n x = 1 and a periodic boundary condition is applied in the x-axis direction.

[0038] 5, the nonreciprocal transmission line according to the fourth embodiment uses the structural parameters of FIGS. 1A to 1C according to the first embodiment, and has a strip conductor length l port =5.0mm, width w port Strip conductors 51 and 52 having a width of 2.2 mm are connected to both ends, and the transmission characteristics between the port P1 and the port P2 are analyzed.

[0039] The nonreciprocal transmission line according to the fourth embodiment configured as described above comprises first and second mutually orthogonal nonreciprocal transmission lines (first and second transmission lines) by arranging five basic cells in tandem in the second direction of the first and second mutually orthogonal directions, each of which has a parallel branch circuit equivalently including an inductive element, in the same manner as in the first to third embodiments. Each parallel branch circuit on both sides of each transmission line is configured so that a wave component propagating along each parallel branch circuit in the longitudinal direction of each transmission line has a nonreciprocal characteristic (to be described later) and forms an asymmetric circuit with respect to the transmission line portion.

[0040] Fig. 6 is an external perspective view showing the configuration of a reciprocal transmission line according to Comparative Example 2. Fig. 6 shows an analytical model of the transmission characteristics of the basic cell according to Comparative Example 1 (a structure in which inductive stub conductors 21, 22; 31, 32 are inserted symmetrically with respect to the line portions of the two orthogonal principal axes). The structural parameters of Comparative Example 2 are the same as those of the basic cell structure of Embodiment 1, except that the stub conductor length l s1 =l s2 =l s3 =l s4 = 2.5 mm, stub conductor width w s = 0.2 mm.

[0041] Next, we performed numerical simulations of the transmission characteristics of the finite cell structure with asymmetrically inserted stub conductors according to the third embodiment shown in Figure 5, both for the reciprocal case where no DC magnetic field is applied and for the non-reciprocal case where a DC magnetic field is applied. We also show the results of numerical simulations of the structure with stub conductors 21, 22; 31, 32 symmetrically inserted according to a modified example of the third embodiment shown in Figure 6, where a DC magnetic field is applied. For the numerical simulations, we used HFSS ver. 16.0, an electromagnetic field simulation software using the finite element method by Ansoft.

[0042] 7A is a graph showing the transmission characteristics of the simulation results when a DC magnetic field is not applied in FIG. 5. Also, FIG. 7B is a graph showing the dispersion characteristics of the simulation results when a DC magnetic field is not applied in FIG. 5. That is, for the structure according to the third embodiment of FIG. 5, assuming that a DC magnetic field is not applied to the soft magnetic ferrite, the DC magnetization is 0 μM. s The transmission characteristics and dispersion curves for the reciprocal case assuming μ = 0 mT and an internal DC magnetic field μ 0 H 0 = 0 mT are shown in Figures 7A and 7B, respectively.

[0043] As is clear from FIGS. 7A and 7B, in the reversible case, the lower cutoff frequency of the passband, which corresponds to the frequency at which the effective permittivity of the periodic structure becomes 0 and corresponds to the parallel resonance frequency of the parallel branches in the basic cell, is f c = 5.78GHz.

[0044] Fig. 8A is a graph showing the transmission characteristics of the simulation results when a DC magnetic field is applied to the nonreciprocal transmission line of Fig. 5. Fig. 8B is a graph showing the dispersion characteristics of the simulation results when a DC magnetic field is applied to the nonreciprocal transmission line of Fig. 5. That is, for the structure according to the third embodiment of Fig. 5, when a DC magnetic field is applied to soft magnetic ferrite, the DC magnetization is 0 μM. s The transmission characteristics and dispersion curves for the non-reciprocal case with μ = 175 mT and an internal DC magnetic field μ = 90 mT are shown in Figures 8A and 8B.

[0045] As is clear from FIGS. 8A and 8B, the DC magnetization μ s In the irreversible case where μ = 175 mT and the internal DC magnetic field μ = 90 mT, the cutoff frequency shifts to the high frequency side and c =6.50GHz, the irreversibility at this time is Δβ y p y The phase gradient of the magnitude of the irreversibility in the y-axis direction, which is the main axis, is given by the following equation: The leaky wave radiation angle is θ y = 19.8°.

[0046]

number

[0047] Note that β0 represents the phase constant in free space.

[0048] 9A is a graph showing the transmission characteristics of the simulation results when a DC magnetic field is applied to the reciprocal transmission line of FIG. 6. Also, FIG. 9B is a graph showing the dispersion characteristics of the simulation results when a DC magnetic field is applied to the reciprocal transmission line of FIG. 6. That is, for the structure according to Comparative Example 2 (in which stub conductors 21, 22; 31, 32 are inserted symmetrically), the DC magnetization μ0M s 9A and 9B show the transmission characteristics and dispersion curves for the reciprocal case assuming μ = 175 mT and an internal DC magnetic field μ 0 H 0 = 90 mT.

[0049] By comparing Figures 8A and 8B with Figures 9A and 9B, it can be seen that irreversibility can be controlled by the presence or absence of structural asymmetry, even when the applied DC magnetic field is the same value. Furthermore, in irreversibility control by varying the DC magnetic field, the lower cutoff frequency fc, which is the operating frequency, fluctuates by about 0.7 GHz, whereas in irreversibility control by structural asymmetry, the lower cutoff frequency f c It can be seen that irreversible operation can be performed while keeping the frequency fixed at around 6.5 GHz.

[0050] (Embodiment 5) FIG. 10 is an external perspective view showing the configuration of a pseudo traveling wave resonator according to the fifth embodiment.

[0051] In the fifth embodiment, a configuration of a two-dimensional pseudo traveling wave resonator consisting of finite cells of the two-dimensional nonreciprocal transmission line is described. Here, only the case of open termination is considered as the boundary condition of the resonator extending in the two-dimensional direction. That is, the termination of each transmission line is set to an open state. In this case, the resonant operating frequency is the frequency at which the effective permittivity becomes zero in the reciprocal case, and corresponds to the parallel resonant frequency of the parallel branches in the basic cell. In the case of a square lattice structure of two-dimensional transmission lines, by configuring the parallel resonant frequencies to coincide in the propagation characteristics in the mutually orthogonal principal axis directions, the electromagnetic field distribution becomes uniform regardless of location within the resonator. In this embodiment, to confirm the resonant operation of the two-dimensional pseudo traveling wave resonator, electromagnetic field analysis is performed using a shielding model in which the analysis space is covered with perfect electric conductor (PEC) walls, assuming no radiation loss. As an example, for the two-dimensional nonreciprocal transmission line structure according to the third embodiment, the number of cells in the x-axis direction is set to n. x = 5, and the number of cells in the y-axis direction is n y = 5, the results of numerical simulations for pseudo-traveling wave resonant operation are shown.

[0052] The structural parameters of the pseudo traveling wave resonator according to the fifth embodiment of FIG. 10 are the same as those of the basic cell according to the third embodiment. The overall size of the resonator is Lx=Ly=60 mm. The line length l port =5.0mm, width w port A microstrip line having a strip conductor 51 of 2.2 mm is connected from port P1 to the resonator via a coupling capacitance element (not shown).

[0053] In the two-dimensional pseudo traveling wave resonator shown in Figure 10, DC magnetization μ sIn the case of reversibility, assuming μ = 0mT and an internal DC magnetic field μH = 0mT, the electromagnetic field distribution in the analysis space shielded by the PEC wall was investigated. As a result of the numerical simulation, the phase distribution and electromagnetic field intensity distribution in the resonator were investigated. The results of this investigation are as follows. (1) Frequency f c At 6.15 GHz, the electromagnetic field intensity distribution is uniform across a two-dimensional plane. (2) The phase distribution is also uniform, so the zero-order resonance phenomenon that occurs when the effective refractive index is zero can be confirmed.

[0054] Next, DC magnetization μ0M s = 175mT and the internal DC magnetic field μ0H0 = 90mT, the phase distribution and electromagnetic field strength distribution were investigated and the results are as follows. (1) In the case of the basic cell according to the third embodiment, the irreversibility is Δβ along the x-axis and y-axis directions, respectively. x p x / π=0.147, Δβ y p y / π=0.147, the wave vector (Δβ x p x / π,Δβ y p y / π)=(0.147, 0.147), a phase gradient occurs in the center line section along the azimuth angle φ=45°. (2) Furthermore, since the electromagnetic field intensity is distributed almost uniformly as in the reciprocal case, it was demonstrated by numerical calculation that pseudo-traveling wave resonance appears in the two-dimensional structure.

[0055] According to the nonreciprocal transmission line of the fifth embodiment configured as described above, as in the first to fourth embodiments, a total of 25 basic cells, each having a parallel branch circuit equivalently including an inductive element and two transmission line portions substantially orthogonal to each other, are juxtaposed in a first and second direction orthogonal to each other and connected in cascade to form first and second nonreciprocal transmission lines (first and second transmission lines) that are orthogonal to each other. Each parallel branch circuit on both sides of each transmission line is configured so that wave components propagating along each parallel branch circuit in the longitudinal direction of each transmission line have nonreciprocal characteristics and form asymmetric circuits with respect to the transmission line portions. Furthermore, a pseudo-traveling wave resonator can be configured using the nonreciprocal transmission line.

[0056] Furthermore, the metal shielding plate is removed from the two-dimensional pseudo traveling wave resonator according to the fifth embodiment, and leakage wave radiation from the resonator will be described.

[0057] In the numerical simulation of the two-dimensional pseudo-traveling-wave resonator described above, we assumed a shielding structure in which the analysis space was covered in all directions with PEC walls. This resulted in a nearly uniform electromagnetic field intensity distribution within the resonator, confirming ideal pseudo-traveling-wave resonant behavior, enabling the formation of a phase gradient of any desired magnitude in any direction within the two-dimensional plane. However, to utilize the phase gradient of the electromagnetic field distribution appearing within this two-dimensional resonator as a leaky-wave antenna, it is necessary to remove the metal shielding plate to create an open structure. In this case, propagation loss occurs due to leaky-wave radiation, making it inappropriate to directly apply the electromagnetic field distribution of the shielding structure to evaluate the beam angle and directivity formed by the leaky-wave radiation. Therefore, in the following numerical simulation, we changed the boundary condition of the analysis space to an absorbing boundary and investigated the changes in the resonant characteristics and electromagnetic field distribution of the pseudo-traveling-wave resonator due to leaky-wave radiation.

[0058] First, we investigated the electromagnetic field intensity distribution in a resonant state similar to that of the pseudo-traveling wave resonator. s = 0mT and the internal DC magnetic field μ0H0 = 0mT. Next, a stub conductor is inserted asymmetrically, and the DC magnetization μ0M sThe non-reciprocal case was also investigated, with a dc magnetic field μH = 175 mT and an internal DC magnetic field μH = 90 mT. The numerical calculation results for the electromagnetic field distribution are shown at frequencies f = 6.15 GHz and 6.53 GHz, respectively, which are equal to the lower cutoff frequency mentioned above. In both the reciprocal and non-reciprocal cases, the shielded model described in the previous section, which applied PEC walls to the boundaries of the analysis space, results in a nearly uniform distribution of the electromagnetic field strength within the resonator. However, in the open-type analysis model where the shielding plates are removed, the electromagnetic field strength distribution becomes non-uniform due to the effects of leaky wave radiation, and this effect is clearly evident. This is thought to be due to the effects of propagation mode losses associated with leaky wave radiation. It can be seen that the electromagnetic field strength tends to weaken with distance from the feed position near the center on the left side. From the above, when using the electromagnetic field distribution of a two-dimensional pseudo-traveling wave resonator as a leaky wave antenna, the feed position must be taken into consideration. By setting the feed position near the center of the resonator, where the effect of attenuation is small, it is expected that the problem of uneven electromagnetic field strength within the resonator will be improved.

[0059] (Embodiment 6) Fig. 11 is an exploded perspective view showing the configuration of a leaky-wave antenna device according to a sixth embodiment. Fig. 12A is a top view of a first layer L1 of the leaky-wave antenna device of Fig. 11, Fig. 12B is a top view of a second layer L2 of the leaky-wave antenna device of Fig. 11, and Fig. 12C is a top view of a third layer L3 of the leaky-wave antenna device of Fig. 11. Fig. 12D is a top view of a fourth layer L4 of the leaky-wave antenna device of Fig. 11, and Fig. 12E is a bottom view of a fifth layer L5 of the leaky-wave antenna device of Fig. 11. In Fig. 12A, reference numeral 1 indicates the region of a basic cell. The length of the dielectric substrate 10 in the x-axis direction is denoted by Lx, and the length of the dielectric substrate 10 in the y-axis direction is denoted by Ly.

[0060] Here, the second layer L2 in Fig. 12B shows the top surface of the first layer L1 in Fig. 12A when the strip conductors 20, 30 and the stub conductors 21, 22; 31, 32 are removed. The third layer L3 in Fig. 12C shows the top surface of the second layer L2 in Fig. 12B when the dielectric substrate 10 is removed. The fourth layer L4 in Fig. 12D shows the top surface of the third layer L3 in Fig. 12C when the ground conductor 11 is removed. The fifth layer L5 in Fig. 12E shows the bottom surface of the dielectric substrate 13 of the fourth layer L4 in Fig. 12D when viewed from the backside.

[0061] 12A to 12E, the device has a multilayer structure in which two dielectric substrates 10 and 13 are combined to feed power from the center of the rear surface. The two dielectric substrates 10 and 13 have the same relative permittivity and thickness, and each has ε d = 2.53, h1 = h2 = 0.8 mm. The length and width of the dielectric substrate 10 is 60 × 60 mm, and the length and width of the dielectric substrate 13 is 60 × 65 mm (shown as 60 mm in the drawing for convenience). Power is fed from a radio signal generator 60 with an internal resistance of 50 Ω to the center of the back surface of the dielectric substrate 13 by a strip conductor 14 constituting a microstrip line. The feeder line has a width w feed = 2.0 mm, length l feed Furthermore, the distance between the end of the strip conductor 14 and the central strip conductors 20 and 30 of the two-dimensional nonreciprocal transmission line on the surface of the dielectric substrate 10 side is set to a radius r via The antenna is directly connected to a feed point 50Q by a feed line of a via conductor 50 made of a thin metal wire with a diameter of D=0.4 mm. To avoid electrical continuity with the via conductor 50, a via hole 50h with a diameter D=0.9 mm is provided in the ground conductor 11 sandwiched between the two substrates 10 and 13. In the analytical model and prototype circuit that follow, we will explain the leaky wave antenna device of Figure 11, which uses the above-mentioned multilayer pseudo traveling wave resonator fed from the center of the back surface.

[0062] The structure of the leaky wave antenna device using the two-dimensional pseudo traveling wave resonator of FIG. 11 is such that the basic cells according to the third embodiment are arranged in the x-axis direction with a cell count of n x = 5, number of cells in the y-axis direction is ny = 5. Here, the size of the entire structure is set to Lx = Ly = 60 mm. Here, the results of numerical simulations are shown for two types of two-dimensional pseudo traveling wave resonator fed by a microstrip line from the center of the back surface of the dielectric substrate 13: a reciprocal case where no DC magnetic field is applied, and a non-reciprocal case where a DC magnetic field is applied.

[0063] Fig. 13 is a graph showing the return loss characteristics of the simulation results at the input port in the unmagnetized state of the leaky wave antenna device of Fig. 11. Also, Fig. 14 is a graph showing the normalized radiation pattern of the simulation results in the unmagnetized state of the leaky wave antenna device of Fig. 11. That is, s = 0 mT and the internal DC magnetic field μ0H0 = 0 mT, the reflection loss characteristics are shown in Fig. 13, and the radiation pattern of the leaky wave radiation is shown in Fig. 14.

[0064] Fig. 15 is a graph showing the return loss characteristics of the simulation results at the input port in the magnetization state of the leaky wave antenna device of Fig. 11. Fig. 16 is a graph showing the radiation pattern of the normalized simulation results in the magnetization state of the leaky wave antenna device of Fig. 11. That is, s The reflection characteristics for the non-reciprocal case of μ0H0=175 mT and internal DC magnetic field μ0H0=90 mT are shown in FIG. 15, and the radiation pattern is shown in FIG.

[0065] DC magnetization is μ0M sIn the case of a reversible structure with a .mu.H = 0 mT and an internal DC magnetic field μH = 0 mT, as shown in Figure 13, a return loss characteristic of approximately -8.5 dB was obtained at a frequency of f = 6.15 GHz. Furthermore, at the same frequency, the inventors' simulation results showed that the electromagnetic field intensity was uniformly distributed, the phase gradient was small, and the phase was also approximately uniformly distributed, indicating that the zeroth-order resonance phenomenon, which occurs when the effective refractive index is zero, was occurring within the periodic structure. Furthermore, the radiation characteristics shown in Figure 14 reveal that the radiation beam direction in in-plane observation at an azimuth angle φ = 45° is perpendicular to the substrate surface, which corresponds to a uniform phase distribution.

[0066] On the other hand, DC magnetization μ0M s In the case of the non-reciprocal case where μ = 175 mT and the internal DC magnetic field μH = 90 mT, as is clear from Fig. 15, the resonance frequency shifts to the high frequency side, f = 6.55 GHz, but a return loss characteristic of about -8.0 dB is obtained. Furthermore, the simulation results of the inventors confirmed that the electromagnetic field strength is uniformly distributed, as in the reciprocal case, and the largest phase gradient can be observed along the azimuth angle φ = 45° direction, confirming the appearance of two-dimensional pseudo traveling wave resonance. Furthermore, as is clear from the radiation pattern in Fig. 16, a radiation angle of about 18° is obtained within the azimuth angle φ = 45° plane, which corresponds to this phase gradient. Furthermore, the absolute gain of radiation is calculated as shown in Fig. 14, when the electric field E φ component is 8.84dBi, electric field E θ In Figure 16, the electric field E φ component is 7.62dBi, electric field E θ The component was -0.84dBi.

[0067] Next, the measurement system used in the experiment will be explained below. The network analyzer used in both the reflection and radiation characteristic measurements was a ZVA67 from Rohde & Schwarz. For radiation measurements, the circuit under test connected to port P1 was placed in an anechoic chamber, and the receiving horn antenna connected to port P2 was placed opposite to it. The circuit under test was placed on a rotating stage, and by placing any face opposite to the receiving horn antenna placed at a distance, the transmission coefficient S 21 Measure the transmission power from

[0068] The radiation pattern is obtained by rotating the circuit under test together with the rotation stage. These operations were performed in four different ways: two with and without a DC magnetic field applied to the circuit under test, and two with vertical polarization and orthogonal polarization. The DC magnetic field was applied using a neodymium magnet measuring 60mm x 60mm x 10mm attached to the back of the circuit under test, and the magnetic field strength was measured using a Lakeshore Model 425 Gaussmeter. Due to the size of the ferrite, the prototype circuit is configured to protrude from the line on the opposite side of the inductive stub insertion section, but this does not have a significant effect on the transmission and radiation characteristics.

[0069] Fig. 17A is a graph showing the return loss characteristics of the experimental results at the input port in the unmagnetized state of the leaky-wave antenna device of Fig. 11. Also, Fig. 17B is a graph showing the radiation pattern of the normalized experimental results in the unmagnetized state of the leaky-wave antenna device of Fig. 11. That is, the reflection characteristics and radiation characteristics are shown in the reversible case where the externally applied magnetic field is 0 mT.

[0070] Fig. 18A is a graph showing the return loss characteristics of the experimental results at the input port in the magnetized state of the leaky-wave antenna device of Fig. 11. Also, Fig. 18B is a graph showing the radiation pattern of the normalized experimental results in the magnetized state of the leaky-wave antenna device of Fig. 11. That is, the graph shows the phase characteristics and radiation pattern in the case of a non-reciprocal externally applied magnetic field of 140 mT.

[0071] In the reversible case where no DC magnetic field is applied, as is clear from Fig. 17B, a radiation beam is obtained whose beam direction is perpendicular to the two-dimensional resonator structure (+z direction) within the plane of azimuth angle φ = 45°. On the other hand, in the non-reversible case where an externally applied DC magnetic field of 140 mT is applied by moving the neodymium magnet, a radiation pattern is obtained whose main beam direction is tilted by approximately 14° from the vertical within the plane of azimuth angle φ = 45°, as is clear from Fig. 18B.

[0072] Next, we will compare the results of the simulation and the experiment. As is clear from a comparison of Figure 13 and Figure 17A, the resonance frequency in the experiment is shifted to the higher frequency side by about 0.3 GHz compared to the resonance frequency in the numerical calculation, and it can be seen that they do not match. The reason for this mismatch is that there is an air gap between the ferrite and the substrate in the prototype circuit used in the experiment, and the inductive stub length is shorter than designed during the manufacturing process, which causes the parallel resonance frequency f sh The shape of the radiation pattern is roughly consistent between the numerical calculation and the experiment, and although there is a difference of 4° in the beam angle, it is thought that this can be adjusted because the inclination of the beam angle increases by strengthening the externally applied magnetic field.

[0073] As explained above, the numerical simulation confirmed that the beam angle tilt was directly above the structure in the reversible case where no magnetic field was applied, and 18° in the azimuth angle φ=45° plane in the non-reciprocal case where a magnetic field was applied. The experiment confirmed that the beam angle tilt was directly above the structure in the reversible case where no DC magnetic field was applied, and 14° in the azimuth angle φ=45° plane in the non-reciprocal case where a DC magnetic field was applied. In other words, both the numerical simulation and the experiment confirmed that the non-reciprocity due to the asymmetry of the structure caused by the inductive stubs makes it possible to form a radiation beam in the azimuth angle φ=45° direction, not along the main axis.

[0074] According to the nonreciprocal transmission line of the sixth embodiment configured as described above, as in the first to fifth embodiments, a total of 25 basic cells, each having a parallel branch circuit equivalently including an inductive element and two transmission line portions substantially orthogonal to each other, are juxtaposed in a first and second direction orthogonal to each other and connected in cascade to form first and second nonreciprocal transmission lines (first and second transmission lines) that are orthogonal to each other. Each parallel branch circuit on both sides of each transmission line is configured so that wave components propagating along each parallel branch circuit in the longitudinal direction of each transmission line have nonreciprocal characteristics and form asymmetric circuits with respect to the transmission line portions. Furthermore, a pseudo-traveling wave resonator can be configured using the nonreciprocal transmission line.

[0075] (Embodiment 7) Fig. 19 is an external perspective view showing the configuration of the leaky wave antenna device according to the seventh embodiment. Fig. 19 shows the results of a numerical simulation of the structure of a nonreciprocal transmission line according to a modification of the third embodiment, which has asymmetry only in the x-axis direction, which is the main axis, and a symmetrical structure in the y-axis direction. Here, when the number of cells is n x =n y = 5. The structural parameters are the same as those shown in the structure according to the modified example of embodiment 3. The size of the entire structure is Lx = Ly = 60 mm.

[0076] Fig. 20 is a graph showing the return loss characteristics of the simulation results of the leaky wave antenna device of Fig. 19. Fig. 21A is a graph showing the radiation pattern (radiation characteristics in the φ=90° plane) of the simulation results of the leaky wave antenna device of Fig. 19, and Fig. 21B is a graph showing the radiation pattern (radiation characteristics in the φ=0° plane) of the simulation results of the leaky wave antenna device of Fig. 19. That is, these are the results of a numerical simulation of a quasi-traveling wave resonator made of a two-dimensional nonreciprocal transmission line with nonreciprocity only in the y-axis direction, and are obtained by a DC magnetization of 0 μM. s FIG. 20 shows the reflection loss characteristics in the case of a non-reciprocal antenna with a dc magnetic field μ0H0 of 175 mT and an internal DC magnetic field μ0H0 of 90 mT, and the radiation patterns are shown in FIGS. 21A and 21B.

[0077] As is clear from Figure 20, the resonant frequency was approximately 6.4 GHz, and a return loss of approximately -7 dB was obtained. From the simulation results of the inventors, it was found that at the resonant frequency of 6.4 GHz, an approximately uniform electromagnetic field distribution was observed, and there was no phase change along the x-axis direction, but there was a phase gradient in the y-axis direction, confirming the phenomenon of two-dimensional pseudo traveling wave resonance. This resulted in the non-reciprocity in the x-axis direction being Δβ x = 0, it is thought that irreversibility occurs in the y-axis direction.

[0078] 21A and 21B, when the observation plane is in the plane with an azimuth angle φ=0° (xz plane), the leaky wave radiation beam direction is perpendicular to the structure. When the observation plane is in the plane with an azimuth angle φ=90° (yz plane), the radiation angle is approximately 17°. This radiation pattern also shows that the two-dimensional nonreciprocal transmission line that constitutes the pseudo traveling wave resonator has nonreciprocity only in the +y direction. In addition, the absolute gain of radiation is calculated by the electric field E in FIG. 21A. φ component is 2.28dBi, electric field E θ In Figure 21B, the electric field E φ component is -10.5dBi, electric field E θ The component was 0.67dBi.

[0079] As explained above, it was confirmed that the radiation angle can be tilted toward a single axis, the +y axis, by inserting inductive stubs asymmetrically in the x-axis direction and symmetrically in the y-axis direction in a nonreciprocal transmission line having a two-dimensional square lattice microstrip line in which the ferrite block 40 is used as part of the dielectric substrate 10. It was also confirmed that the tilt is approximately 17° in elevation angle.

[0080] According to the nonreciprocal transmission line of the seventh embodiment configured as described above, as in the first to sixth embodiments, a total of 25 basic cells, each having a parallel branch circuit equivalently including an inductive element and two transmission line portions substantially orthogonal to each other, are juxtaposed in a first and second direction orthogonal to each other and connected in cascade to form first and second nonreciprocal transmission lines (first and second transmission lines) that are orthogonal to each other. Each parallel branch circuit on both sides of each transmission line is configured so that wave components propagating along each parallel branch circuit in the longitudinal direction of each transmission line have nonreciprocal characteristics and form asymmetric circuits with respect to the transmission line portions. Furthermore, a pseudo-traveling wave resonator can be configured using the nonreciprocal transmission line, and a pseudo-traveling wave resonator can be configured using a leaky wave antenna device.

[0081] (Embodiment 8) 22 is a top view of the basic cell of the nonreciprocal transmission line according to embodiment 8. The basic cell according to embodiment 8 differs from the basic cell according to embodiment 2 (FIGS. 2A to 2C) in the following respects. (1) A series capacitor C having a fixed series capacitance between the strip conductor 30 and the stub conductor 31 S A variable capacitance diode C is inserted between the stub conductor 31 and the via conductor 33. V1 was inserted. (2) A series capacitor C having a fixed series capacitance between the strip conductor 30 and the stub conductor 32 S A variable capacitance diode C is inserted between the stub conductor 32 and the via conductor 34. V2 was inserted. (3) A series capacitor C having a fixed series capacitance between the strip conductor 20 and the stub conductor 22 S A variable capacitance diode C is inserted between the stub conductor 22 and the via conductor 24. V3 was inserted. (4) A series capacitor C having a fixed series capacitance between the strip conductor 20 and the stub conductor 21 S A variable capacitance diode C is inserted between the stub conductor 21 and the via conductor 23. V4 was inserted.

[0082] A plurality of basic cells of the nonreciprocal transmission line according to the eighth embodiment configured as described above are arranged in parallel in the x-axis direction and the y-axis direction, and each variable capacitance diode C V1 ~C V4 By varying the reverse bias voltage applied to each stub, the inductive or capacitive properties of each stub can be varied.

[0083] The nonreciprocal transmission line according to the eighth embodiment configured as described above includes at least one basic cell, each having a parallel branch circuit equivalently including an inductive element and two transmission line portions substantially orthogonal to each other, juxtaposed in a first and second direction orthogonal to each other and connected in cascade to provide first and second nonreciprocal transmission lines (first and second transmission lines) that are orthogonal to each other. At least one parallel branch circuit of each transmission line is configured so that wave components propagating along each parallel branch circuit in the longitudinal direction of each transmission line have nonreciprocal characteristics and form asymmetric circuits with respect to the transmission line portions. Furthermore, a pseudo-traveling wave resonator can be configured using the nonreciprocal transmission line, and a pseudo-traveling wave resonator can be configured using a leaky wave antenna device.

[0084] (Embodiment 9) 23 is a top view of the basic cell of the nonreciprocal transmission line according to embodiment 9. The basic cell according to embodiment 9 differs from the basic cell according to embodiment 1 (FIGS. 1A to 1C) in the following respects. (1) A series capacitor C having a fixed series capacitance between the strip conductor 30 and the stub conductor 31 S A variable capacitance diode C is inserted between the stub conductor 31 and the via conductor 33. V1 was inserted. (2) A series capacitor C having a fixed series capacitance between the strip conductor 30 and the stub conductor 32 S A variable capacitance diode C is inserted between the stub conductor 32 and the via conductor 34. V2 was inserted. (3) A series capacitor C having a fixed series capacitance between the strip conductor 20 and the stub conductor 22 S A variable capacitance diode C is inserted between the stub conductor 22 and the via conductor 24.V3 was inserted. (4) A series capacitor C having a fixed series capacitance between the strip conductor 20 and the stub conductor 21 S A variable capacitance diode C is inserted between the stub conductor 21 and the via conductor 23. V4 was inserted.

[0085] A plurality of basic cells of the nonreciprocal transmission line according to the ninth embodiment configured as described above are arranged in parallel in the x-axis direction and the y-axis direction, and each variable capacitance diode C V1 ~C V4 By varying the reverse bias voltage applied to each stub, the inductive or capacitive properties of each stub can be varied.

[0086] Fig. 24 is an exploded perspective view showing the configuration of a leaky-wave antenna device using the nonreciprocal transmission line of Fig. 22. Fig. 25A is a top view of the first layer L1 of the leaky-wave antenna device of Fig. 24, and Fig. 25B is a top view of the second layer L2 of the leaky-wave antenna device of Fig. 24. Fig. 25C is a top view of the third layer L3 of the leaky-wave antenna device of Fig. 24, and Fig. 25D is a top view of the fourth layer L4 of the leaky-wave antenna device of Fig. 24. Note that in Figs. 24 to 25D, the basic cells are arranged in a 3 x 3 lattice pattern for clarity, but the simulation results and experiments were performed on a model in which the basic cells were arranged in a 5 x 5 lattice pattern.

[0087] The leaky wave antenna device of FIG. 24 has the following features. (1) The controller 61 generates a predetermined control voltage (corresponding to the reverse bias voltage) and supplies it to the variable capacitance diode C via the control voltage line 71. V1 Apply to. (2) The controller 62 generates a predetermined control voltage (corresponding to the reverse bias voltage) and supplies it to the variable capacitance diode C via the control voltage line 72. V2 Apply to. (3) The controller 63 generates a predetermined control voltage (corresponding to the reverse bias voltage) and supplies it to the variable capacitance diode C via the control voltage line 73. V3 Apply to. (4) The controller 64 generates a predetermined control voltage (corresponding to the reverse bias voltage) and supplies it to the variable capacitance diode C via the control voltage line 74. V4 Apply to. (5) The control voltages are controlled by the controllers 61 to 64 so that the host main controller 65 controls the two-dimensional spatial phase distribution of the leaky wave antenna device to control the radiation pattern.

[0088] The control voltage lines 71 and 72 are formed on the dielectric substrate 13 , and the control voltage lines 73 and 74 are formed on the dielectric substrate 15 below the dielectric substrate 13 .

[0089] According to the nonreciprocal transmission line of the ninth embodiment configured as described above, as in the eighth embodiment, at least one basic cell having two transmission line portions, each of which has a parallel branch circuit equivalently including an inductive element and which is substantially orthogonal to each other, is juxtaposed in a first and second direction, which are orthogonal to each other, and connected in cascade to form first and second nonreciprocal transmission lines (first and second transmission lines) that are orthogonal to each other. Each parallel branch circuit of each transmission line is configured so that wave components propagating along each parallel branch circuit in the longitudinal direction of each transmission line have nonreciprocal characteristics and form asymmetric circuits with respect to the transmission line portions. Furthermore, a pseudo-traveling wave resonator can be configured using the nonreciprocal transmission line, and a pseudo-traveling wave resonator can be configured using a leaky wave antenna device.

[0090] (Embodiment 10) Fig. 26 is a top view of the leaky wave antenna device according to the embodiment 10. In Fig. 26, the leaky wave antenna device according to the embodiment 10 is characterized by having a unidirectional asymmetric stub insertion structure so as to perform non-reciprocal propagation only in the y-axis direction.

[0091] According to the nonreciprocal transmission line of the tenth embodiment configured as described above, as in the first to sixth embodiments, a total of 25 basic cells, each having a parallel branch circuit equivalently including an inductive element and two transmission line portions substantially orthogonal to each other, are juxtaposed in a first and second direction orthogonal to each other and connected in cascade to form first and second nonreciprocal transmission lines (first and second transmission lines) that are orthogonal to each other. One parallel branch circuit of each transmission line is configured so that wave components propagating along each parallel branch circuit in the longitudinal direction of the transmission line have nonreciprocal characteristics and form an asymmetric circuit with respect to the transmission line portions. Furthermore, a pseudo-traveling wave resonator can be configured using the nonreciprocal transmission line, and a pseudo-traveling wave resonator can be configured using a leaky wave antenna device.

[0092] (Embodiment 11) Fig. 27A is a top view of the basic cell of the nonreciprocal transmission line according to the embodiment 11. Fig. 27B is a top view of the basic cell of Fig. 27A with strip conductors 20 and 30 and stub conductors 21, 22, 32, and 34 removed. The basic cell of the embodiment 11 differs from the basic cell of the embodiment 1 in the following respects. (1) The strip conductors 20 and 30 are formed on the four sides of the basic cell with their widths halved. (2) In the internal region surrounded by the strip conductors 20 and 30, stub conductors 21 and 22; 31 and 32 and via conductors 23 and 24; 33 and 34 were formed as follows. A stub conductor 21 and a via conductor 23 extending in the -y-axis direction are formed on the left side surface of the strip conductor 20 on the right side of FIG. (2b) A stub conductor 22 and a via conductor 24 extending in the +y-axis direction are formed on the right side surface of the strip conductor 20 on the left side of the figure. (2c) A stub conductor 32 and a via conductor 34 extending in the +x-axis direction are formed on the lower side surface of the strip conductor 30 on the upper side of the figure. (2d) A stub conductor 31 and a via conductor 33 extending in the -x-axis direction are formed on the upper side surface of the strip conductor 30 on the lower side of the figure.

[0093] A nonreciprocal transmission line can be constructed by arranging a plurality of basic cells configured as described above in parallel in the x-axis direction and the y-axis direction. The basic cell according to the eleventh embodiment of FIG. 10 can be applied to various leaky wave antenna device embodiments or modifications described later.

[0094] According to the nonreciprocal transmission line of the tenth embodiment configured as described above, as in the first to sixth embodiments, at least one basic cell having two transmission line portions, each of which has a parallel branch circuit equivalently including an inductive element and which are substantially orthogonal to each other, is juxtaposed in a first and second direction, which are orthogonal to each other, and is connected in cascade to provide first and second nonreciprocal transmission lines (first and second transmission lines) that are orthogonal to each other. At least one parallel branch circuit of each transmission line is configured so that wave components propagating along each parallel branch circuit in the longitudinal direction of each transmission line have nonreciprocal characteristics and form asymmetric circuits with respect to the transmission line portions. Furthermore, a pseudo-traveling wave resonator can be configured using the nonreciprocal transmission line, and a pseudo-traveling wave resonator can be configured using a leaky wave antenna device.

[0095] (Embodiment 12) Fig. 28 is an external perspective view showing a shielding conductor plate 80 (embodiment 12) placed on the upper surface of a leaky-wave antenna device using a nonreciprocal transmission line according to any one of embodiments 1 to 11. In Fig. 28, the shielding conductor plate 80 is placed on the upper surface of a leaky-wave antenna device 100 using a nonreciprocal transmission line (the leaky-wave antenna device according to each of the above-described embodiments is designated by the reference numeral 100), and a plurality of coupling rectangular slots 81 having a longitudinal direction parallel to the x-axis direction or the y-axis direction are formed on the upper surface of the shielding conductor plate 80 so as to penetrate in the thickness direction. Note that the side surfaces of the dielectric substrate may or may not be covered with the conductor plate.

[0096] In the leaky wave antenna device configured as described above, the phase gradient of the electromagnetic field distribution can be manipulated in any direction within the two-dimensional plane of the two-dimensional pseudo traveling wave resonator using a nonreciprocal transmission line, and therefore, electromagnetic waves can be radiated from there through the coupling rectangular slot 81. Here, the shielding conductor plate 80 functions to suppress radiation loss.

[0097] (Embodiment 13) Fig. 29 is an external perspective view showing a shielding conductor plate 80A (embodiment 13) placed on the upper surface of a leaky-wave antenna device using a nonreciprocal transmission line according to any one of embodiments 1 to 11. In Fig. 29, the shielding conductor plate 80A is placed on the upper surface of the leaky-wave antenna device using a nonreciprocal transmission line, and a plurality of coupling cross slots 82 having longitudinal directions parallel to the x-axis direction and the y-axis direction are formed on the upper surface of the shielding conductor plate 80A so as to penetrate in the thickness direction. Note that the side surfaces of the dielectric substrate may or may not be covered with the conductor plate.

[0098] In the leaky wave antenna device configured as described above, the phase gradient of the electromagnetic field distribution can be manipulated in any direction within the two-dimensional plane of the two-dimensional pseudo traveling wave resonator using a nonreciprocal transmission line, and electromagnetic waves can be radiated therefrom via the coupling cross slots 82. Here, the shielding conductor plate 80A operates as an array antenna by utilizing the coupling cross slots 82 as radiators.

[0099] Fig. 30 is an external perspective view showing an antenna device configured by placing the shielding conductor plate 80 or 80A of Fig. 28 or 29 on the nonreciprocal transmission line according to any one of Embodiments 1 to 11, and then placing a strip array antenna on top of that. In Fig. 30, the shielding conductor plate 80 or 80A of Fig. 28 or 29 is placed on the nonreciprocal transmission line, and then a strip array antenna (also called a patch array antenna) having a plurality of rectangular patch conductors 91 arranged, for example, in a lattice pattern on a dielectric substrate 90 is placed on top of that.

[0100] In the leaky wave antenna device configured as described above, the phase gradient of the electromagnetic field distribution can be manipulated in any direction within the two-dimensional plane of the two-dimensional pseudo traveling wave resonator using a nonreciprocal transmission line, and the electromagnetic waves therefrom can excite the strip array antenna via the coupling rectangular slots 81 or the coupling cross slots 82. The slot array antenna can radiate electromagnetic waves controlled in any direction from the top surface of the antenna.

[0101] (Embodiment 14) Fig. 31 is a top view of a leaky wave antenna device according to embodiment 14. Fig. 31 shows an example of a two-dimensional structure in which, in a two-dimensional square lattice structure of microstrip lines, only electromagnetic wave components propagating along branches parallel to the y-axis exhibit non-reciprocity, while electromagnetic wave components propagating along branches parallel to the x-axis exhibit reciprocity.

[0102] In FIG. 31, reference numeral 1A denotes the area of ​​the basic cell.

[0103] Each basic cell 1A has at least first and second parallel branch circuits provided on both sides of at least one of the two transmission line portions. In the example of Fig. 31, each basic cell 1A has a first parallel branch circuit including a stub conductor 21A and a via conductor 23 provided on the -y side of the transmission line portion (i.e., strip conductor 20) along the x-axis, and a second parallel branch circuit including a stub conductor 22A and a via conductor 24 provided on the +y side.

[0104] In the xy plane, at least a portion of the first parallel branch circuit is orthogonal to at least a portion of the second parallel branch circuit. In the example of Fig. 31, stub conductors 21A and 22A, which are parts of each parallel branch circuit except for via conductors 23 and 24, are orthogonal to each other.

[0105] Furthermore, the first and second parallel branch circuits may each include at least a portion of a section that is arranged at an angle of 45 degrees relative to the transmission line portion in the xy plane. In the example of Fig. 31, the stub conductors 21A and 22A are arranged at an angle of 45 degrees relative to the strip conductor 20.

[0106] The leaky wave antenna device of FIG. 31 has stubs inserted asymmetrically with respect to the y-axis for microstrip line branches parallel to the y-axis, and has the structure of FIG. 32 for strip line branches parallel to the x-axis.

[0107] Fig. 32 is a diagram illustrating an AC magnetic field generated by the stub conductors 21A and 22A of the leaky wave antenna device of Fig. 31. Fig. 32 shows a partial region 2 of the leaky wave antenna device of Fig. 31, which includes a pair of stub conductors 21A and 22A arranged on both sides of the strip conductor 20 and close to each other. In Fig. 32, the thick dashed arrows indicate an example of the direction of the AC magnetic field observed on the stub conductors 21A and 22A at a certain moment. Fig. 32 shows the relationship between a line structure in which the pair of stub conductors inserted parallel to the y-axis on both sides of the microstrip line in the other example of Fig. 1 are rotated 45 degrees around the z-axis toward the +x direction, and the direction of the AC magnetic field component near the stubs.

[0108] Fig. 33 is a graph showing the return loss characteristics of the simulation results of the leaky wave antenna device of Fig. 31. The leaky wave antenna device has an operating frequency of 6.32 GHz.

[0109] FIG. 34 is a graph showing a radiation pattern (radiation characteristics in a φ=90° plane) obtained as a result of a simulation of the leaky wave antenna device of FIG. 31. FIG. 35 is a graph showing a radiation pattern (radiation characteristics in a φ=0° plane) obtained as a result of a simulation of the leaky wave antenna device of FIG. 31. According to FIG. 35, when the azimuth angle φ=0° plane (xz plane) is taken as the observation plane, the leaky wave antenna device has a radiation angle θ=0° perpendicular to the xy plane. According to FIG. 34, when the azimuth angle φ=90° plane (yz plane) is taken as the observation plane, the leaky wave antenna device has a radiation angle θ=15°. These radiation characteristics indicate that the two-dimensional nonreciprocal transmission line constituting the pseudo traveling wave resonator exhibits nonreciprocity only in the branches along the y-axis. The absolute gain of radiation was 7.61 dBi in the Eφ component of the electric field in FIG. 34, and 6.35 dBi in the Eθ component of the electric field in FIG. 35.

[0110] The above results show improvements of 5.33 dBi and 5.68 dBi compared to the radiation gains in Figures 21A and 21B, respectively. By arranging the stub conductors 21A and 22A so that they are orthogonal to each other, the radiation components from the stub conductors 21A and 22A are less likely to be canceled out by overlapping, which is effective in improving the radiation gain of the leaky wave.

[0111] Fig. 36 is a top view of a leaky-wave antenna device according to Comparative Example 3. The leaky-wave antenna device of Fig. 36 has a structure in which stub conductors are inserted symmetrically and perpendicularly to strip conductors parallel to the x-axis and y-axis, respectively. In other words, the leaky-wave antenna device of Fig. 36 has a configuration in which the basic cells shown in Figs. 2A to 2C are juxtaposed in the x-direction and y-direction.

[0112] FIG. 37 illustrates the AC magnetic field generated by the stub conductors 21 and 22 of the leaky wave antenna device of FIG. 36. FIG. 37 shows a partial region 2A of the leaky wave antenna device of FIG. 36, including a pair of stub conductors 21 and 22 arranged on both sides of the strip conductor 20 and located close to each other. FIG. 37 shows a top view of a line structure in which a pair of stubs are inserted symmetrically on both sides of a metal strip conductor of a microstrip line extending along the x-axis, along the y-axis, which is perpendicular to the x-axis. In FIG. 37, each triangle indicates the direction and relative magnitude of an AC magnetic field vector at a given moment, observed on a plane placed parallel to the microstrip line and 1.0 mm apart. The direction of the AC magnetic field observed on the stub conductors 21 and 22 at a given moment generally corresponds to the direction of the thick dashed arrow. Because the directions of the AC magnetic fields radiated from the stub conductors 21 and 22 are antiparallel, the radiated waves cancel each other out in the far field, equidistant from the stub conductors 21 and 22, resulting in a reduced radiation gain.

[0113] 32 and 37, it can be seen that in the leaky wave antenna device of FIG. 31, by arranging the stub conductors 21A and 22A so that they are orthogonal to each other, the radiation components from the stub conductors 21A and 22A are less likely to be cancelled out due to overlap.

[0114] Figure 38 is a graph showing the radiation pattern (radiation characteristics in the φ=45° plane) of the simulation results for the leaky wave antenna device of Figure 36. In the leaky wave antenna device of Figure 36, due to the pair of stub conductors inserted symmetrically, no nonreciprocity occurs in either the electromagnetic wave components propagating in the x-axis or y-axis directions, and the phase distribution of the electromagnetic field is uniform throughout the two-dimensional structure. In this case, leaky wave beam radiation in a direction perpendicular to the two-dimensional structure is expected. However, because the stub conductors are symmetrical, the vector directions of the current or AC magnetic field are antiparallel between the pair of stubs. Therefore, as shown in Figure 38, the radiation components directly upward in the two-dimensional structure cancel each other out, resulting in very small radiation gain. To form and radiate a beam perpendicular to the two-dimensional nonreciprocal transmission line structure, it is necessary to orthogonally orthogonalize the electromagnetic field vectors, as in Figure 32, to avoid mutual cancellation of radiation.

[0115] According to the leaky wave antenna device of Figure 31, by arranging a pair of stub conductors 21A, 22A so that they are perpendicular to each other, it is possible to prevent the leaky wave radiation components in the far field at equal distances from the stub conductors 21A, 22A from being cancelled out by superposition, regardless of the direction and magnitude of the current distribution flowing along each of the stub conductors 21A, 22A, and thus an improvement in radiation gain can be expected.

[0116] 19, the leaky wave radiation components from a pair of stub conductors 21 and 22 inserted parallel to the y-axis cancel each other out, resulting in a significant drop in radiation gain. On the other hand, in the leaky wave antenna device of Fig. 31, the stub conductors 21A and 22A are inclined at 45 degrees with respect to the strip conductor 20 and are perpendicular to each other, thereby preventing the leaky wave radiation components from overlapping and canceling each other out, and an improvement in radiation gain can be expected.

[0117] In the leaky wave antenna device of Figure 31, the stub conductors 21A and 22A have the same shape and are symmetrical about the x-axis, but similar operation can also be expected when the structures of the stub conductors are not identical but asymmetric, or when the stub conductors have different combinations of admittance.

[0118] The stub conductors 21A and 22A are not limited to being arranged at a 45-degree angle with respect to the strip conductor 20. As long as at least a portion of the first parallel branch circuit is orthogonal to at least a portion of the second parallel branch circuit in the xy plane, the first and second parallel branch circuits may be arranged at other angles with respect to the transmission line portion in the xy plane.

[0119] Fig. 39 is a top view of a leaky wave antenna device according to Modification 1 of Embodiment 14. Fig. 39 shows an example of a two-dimensional structure in which, in a two-dimensional square lattice structure of microstrip lines, both the electromagnetic wave component propagating along the branch parallel to the x-axis and the electromagnetic wave component propagating along the branch parallel to the y-axis exhibit reciprocity.

[0120] In the example of Figure 39, the leaky wave antenna device has first and second parallel branch circuits arranged on both sides of the transmission line portion along the x-axis (i.e., strip conductor 20), and also has a third parallel branch circuit including a stub conductor 31A and a via conductor 33 arranged on the -x side of the transmission line portion along the y-axis (i.e., strip conductor 30), and a fourth parallel branch circuit including a stub conductor 32A and a via conductor 34 arranged on the +x side.

[0121] In the xy plane, at least a portion of the third parallel branch circuit is orthogonal to at least a portion of the fourth parallel branch circuit. In the example of Fig. 31, stub conductors 31A and 32A, which are parts of each parallel branch circuit except for via conductors 33 and 34, are orthogonal to each other.

[0122] Furthermore, the third and fourth parallel branch circuits may each include at least a portion of a section that is arranged at an angle of 45 degrees relative to the transmission line portion in the xy plane. In the example of Fig. 31, the stub conductors 31A and 32A are arranged at an angle of 45 degrees relative to the strip conductor 30.

[0123] According to the leaky wave antenna device of Figure 39, similarly to the leaky wave antenna device of Figure 31, by arranging a pair of stub conductors 31A, 32A so that they are perpendicular to each other, it is possible to prevent the leaky wave radiation components in the far field at equal distances from the stub conductors 31A, 32A from being cancelled out by superposition, regardless of the direction and magnitude of the current distribution flowing along each of the stub conductors 31A, 32A, and thus an improvement in radiation gain can be expected.

[0124] FIG. 40 is a top view of a leaky wave antenna device according to a second modification of the fourteenth embodiment.

[0125] In FIG. 40, reference numeral 1C denotes a region of a basic cell.

[0126] Each basic cell 1C has at least first and second parallel branch circuits provided on both sides of at least one of the two transmission line portions. In the example of Fig. 40, each basic cell 1C has a first parallel branch circuit including a stub conductor 21C and a via conductor 23 provided on the -y side of the transmission line portion along the x-axis (i.e., strip conductor 20), and a second parallel branch circuit including a stub conductor 22C and a via conductor 24 provided on the +y side. Each basic cell 1C also has a third parallel branch circuit including a stub conductor 31C and a via conductor 33 provided on the -x side of the transmission line portion along the y-axis (i.e., strip conductor 30), and a fourth parallel branch circuit including a stub conductor 32C and a via conductor 34 provided on the +x side.

[0127] The first and second parallel branches are bent in the same direction along the longitudinal direction of the transmission line portion in the xy plane and include sections parallel to the longitudinal direction of the transmission line portion, while the third and fourth parallel branches are bent in the same direction along the longitudinal direction of the transmission line portion in the xy plane and include sections parallel to the longitudinal direction of the transmission line portion.

[0128] Fig. 41 is a diagram illustrating an AC magnetic field generated by the stub conductors 21C and 22C of the leaky wave antenna device of Fig. 40. Fig. 41 shows a partial region 2C of the leaky wave antenna device of Fig. 40, which includes a pair of stub conductors 21C and 22C arranged on both sides of the strip conductor 20 and located close to each other. In Fig. 41, the thick dashed arrows indicate an example of the direction of the AC magnetic field observed on the stub conductors 21C and 22C at a certain moment. Fig. 41 shows the relationship between the line structure when the pair of stub conductors 21C and 22C inserted on both sides of the microstrip line are bent into an L-shape near the connection portion and the direction of the AC magnetic field near the stub conductors 21C and 22C.

[0129] Fig. 42 is a diagram illustrating an AC magnetic field generated by stub conductors 21A, 22A, 31A, and 32A of the leaky wave antenna device of Fig. 39. Within the area shown in Fig. 42, the leaky wave antenna device includes stub conductors 21A-1 to 32A-21 and via conductors 23A-1 to 34A-2. The directions of the currents flowing through stub conductors 22A-1, 22A-2, 31A-1, and 31A-2, which are surrounded by dashed lines in the figure, or the AC magnetic fields of the leaky wave radiation components therefrom, are antiparallel to each other, and the radiation waves overlap, causing cancellation in the far field. To avoid this, a different technique from that of Fig. 31 is required, for example, the configuration of Fig. 40.

[0130] Fig. 43 is a graph showing the return loss characteristics of the simulation results of the leaky wave antenna device of Fig. 40. The leaky wave antenna device has an operating frequency of 6.31 GHz, and at this time, the reflection coefficient is -24.8 dB.

[0131] FIG. 44 is a graph showing the radiation pattern (in the φ=45° plane) of the simulation results when no DC magnetic field is applied in FIG. 40. FIG. 45 is a graph showing the radiation pattern (in the φ=45° plane) of the simulation results when a saturation magnetization of 175 mT and an internal DC magnetic field of 90 mT are applied in FIG. 40. It can be seen from FIGS. 44 and 45 that the leaky wave radiation beam direction when the observation plane is in the azimuth angle φ=45° plane is perpendicular to the two-dimensional structure in both cases. From these radiation characteristics, it can be seen that the structure in FIG. 40 does not exhibit non-reciprocity in the propagation characteristics in the x-axis and y-axis directions (i.e., is reversible) regardless of whether a DC magnetic field is applied, operates as a zeroth-order resonator utilizing a state in which the effective refractive index is zero, and the phase distribution within the structure is uniform.

[0132] The absolute radiation gain was 6.31 dBi for the electric field Eφ component in Fig. 44 and 5.64 dBi for the electric field Eφ component in Fig. 45. In the case of the radiation pattern of the structure according to the comparative example shown in Fig. 38, the radiation gain in the direction perpendicular to the structure is very small, whereas a certain degree of radiation gain is obtained in the structure of Fig. 40. From the above, it has been demonstrated that bending the stub conductor structure reduces the cancellation caused by the overlapping of electromagnetic fields, thereby improving the radiation gain.

[0133] As can be seen from the direction of the AC magnetic field shown in Figure 41, the currents flowing in the portions of the stub conductors 21C and 22C closest to the ends, or the AC magnetic field directions in the vicinity of the same structure, are parallel to each other. Therefore, in the far field at equal distances from the two stub conductors 21C and 22C, the leaky wave radiation components reinforce each other due to superposition, resulting in a large radiation gain. On the other hand, the vector directions of the AC magnetic field near the connection between the stub conductors 21C and 22C are antiparallel, and the leaky wave radiation components cancel each other out in the far field due to superposition, albeit on a small scale.

[0134] FIG. 46 is a top view of a leaky wave antenna device according to a third modification of the fourteenth embodiment.

[0135] In FIG. 46, the reference numeral 1D denotes the area of ​​the basic cell.

[0136] Each basic cell 1D has at least first and second parallel branch circuits provided on both sides of at least one of the two transmission line portions. In the example of Fig. 46, each basic cell 1D has a first parallel branch circuit including a stub conductor 21 and a via conductor 23 provided on the -y side of the transmission line portion along the x-axis (i.e., strip conductor 20), and a second parallel branch circuit including a stub conductor 22D and a via conductor 24 provided on the +y side. Each basic cell 1D also has a third parallel branch circuit including a stub conductor 31D and a via conductor 33 provided on the -x side of the transmission line portion along the y-axis (i.e., strip conductor 30), and a fourth parallel branch circuit including a stub conductor 32 and a via conductor 34 provided on the +x side.

[0137] The first parallel branch circuit includes at least a section disposed at an angle of 90 degrees relative to the transmission line portion in the xy plane. The second parallel branch circuit includes a section bent in the xy plane and parallel to the longitudinal direction of the transmission line portion. The third parallel branch circuit includes at least a section disposed at an angle of 90 degrees relative to the transmission line portion in the xy plane. The fourth parallel branch circuit includes a section bent in the xy plane and parallel to the longitudinal direction of the transmission line portion.

[0138] Fig. 47 is a diagram illustrating the magnetic field generated by the stub conductors 21 and 22D of the leaky wave antenna device of Fig. 46. Fig. 47 shows a partial region 2D of the leaky wave antenna device of Fig. 46, which includes a pair of stub conductors 21 and 22D arranged on both sides of the strip conductor 20 and located close to each other. In Fig. 47, the thick dashed arrows indicate an example of the direction of the AC magnetic field observed on the stub conductors 21 and 22D at a certain moment. Fig. 47 shows the relationship between the line structure and the direction of the AC magnetic field near the stub conductors 21 and 22D when one of the stub conductors 21 and 22D inserted on one side of the microstrip line is bent into an L-shape and the other stub conductors 21 and 22D are left unchanged.

[0139] According to the AC magnetic field distribution shown in the figure, the currents flowing through most of the left and right stub conductors are orthogonal to each other, and the AC magnetic field components of the leaky wave radiation components radiated from them are also orthogonal to each other. The AC magnetic fields of the leaky wave radiation components from parts near the connection between stub conductors 21 and 22D are antiparallel to each other, and although the scale is even smaller than in the case of Figure 41, cancellation occurs due to the overlap of the leaky wave emissions.

[0140] Fig. 48 is a graph showing the return loss characteristics of the simulation results for the leaky wave antenna device of Fig. 46. Here, a magnetic field with a saturation magnetization of 175 mT and an internal DC magnetic field of 90 mT is applied to the leaky wave antenna device. The leaky wave antenna device has an operating frequency of 6.4 GHz, and at this time, the reflection coefficient is -12.6 dB.

[0141] Figure 49 is a graph showing the radiation pattern (radiation characteristics in the φ=45° plane) of the simulation results for the leaky wave antenna device of Figure 46. As with Figure 40, Figure 49 shows that the leaky wave antenna device of Figure 46 also exhibits no non-reciprocity in the propagation characteristics in the x-axis and y-axis directions, regardless of whether a DC magnetic field is applied or not. It operates as a zero-order resonator utilizing a state in which the effective refractive index is zero, and the phase gradient within the structure is uniform. It can also be seen that the leaky wave radiation direction is perpendicular to the structure when the observation plane is within the azimuth angle φ=45° plane. The absolute radiation gain was 7.30 dBi for the electric field Eφ component. From the above, it can be confirmed that the gain is improved by bending the stub conductor.

[0142] Furthermore, since the stub structure of FIG. 47 has fewer areas where electromagnetic fields cancel each other out compared to the stub structure of FIG. 41, it can be seen that the radiation gain of FIG. 49 is slightly larger than the radiation gains of FIGS. 44 and 45.

[0143] The present invention is not limited to the case where the stub conductor 21 is arranged at an angle of 90 degrees relative to the strip conductor 20 and the stub conductor 22D includes a section parallel to the longitudinal direction of the strip conductor 20. As long as at least a portion of the section of the first parallel branch circuit is orthogonal to at least a portion of the section of the second parallel branch circuit in the xy plane, the first and second parallel branches may be arranged at other angles relative to the transmission line portion in the xy plane.

[0144] (Embodiment 15) Fig. 50 is a top view of a leaky wave antenna device according to embodiment 15. Fig. 51 is a top view of one basic cell 1E in the leaky wave antenna device of Fig. 50. The leaky wave antenna device of Fig. 50 includes strip conductors 20D and 30D instead of the strip conductors 20 and 30 of Fig. 12A. As shown in Fig. 51, the strip conductors 20D and 30D have a capacitor C0 inserted in series with the strip conductors 20D and 30D.

[0145] 51, a pattern conductor 25 is provided at the intersection of strip conductors 20D and 30D. A capacitor C0 is inserted between the strip conductors 20D and 30D and the pattern conductor 25. The capacitor C0 is inserted periodically in each of the strip conductors 20D and 30D.

[0146] The leaky wave antenna device in FIG. 50 has a feed point at the center of the left side.

[0147] Fig. 52 is a graph showing the return loss characteristics of the simulation results for the leaky wave antenna device of Fig. 50. Here, a magnetic field with a saturation magnetization of 175 mT and an internal DC magnetic field of 90 mT is applied to the leaky wave antenna device. The leaky wave antenna device has an operating frequency of 5.92 GHz, and at this time, the reflection coefficient is -7.95 dB.

[0148] Fig. 53 is a graph showing the radiation pattern (radiation characteristics in the φ=45° plane) of the simulation results for the leaky wave antenna device of Fig. 50. Here, a magnetic field with a saturation magnetization of 175 mT and an internal DC magnetic field of 90 mT is applied to the leaky wave antenna device. Fig. 53 shows that the leaky wave antenna device has a radiation angle θ=11°. The absolute gain of the radiation was 8.06 dBi for the electric field Eφ component.

[0149] Here, the simulation results of the leaky wave antenna device of FIG. 50 will be compared with the simulation results of the leaky wave antenna device according to Comparative Example 4 (see FIG. 11) that does not have the capacitor C0.

[0150] 54 is a graph showing the return loss characteristics of the simulation results for the leaky wave antenna device according to Comparative Example 4. Here, a magnetic field with a saturation magnetization of 175 mT and an internal DC magnetic field of 90 mT is applied to the leaky wave antenna device. The leaky wave antenna device has an operating frequency of 5.81 GHz, and at this frequency, the reflection coefficient is −9.3 dB.

[0151] Fig. 55 is a graph showing the radiation pattern (radiation characteristics in the φ=45° plane) of the simulation results for the leaky wave antenna device according to Comparative Example 4. Here, a magnetic field with a saturation magnetization of 175 mT and an internal DC magnetic field of 90 mT is applied to the leaky wave antenna device. Fig. 55 shows that the leaky wave antenna device has a radiation angle θ=30°. The absolute gain of the radiation was 3.97 dBi for the electric field Eφ component.

[0152] Comparing Figures 53 and 55, it can be seen that the leaky wave antenna device of Figure 50, which has a capacitor inserted in series in the transmission line portion, has an improved radiation gain compared to the leaky wave antenna device of Comparative Example 4, which does not have a capacitor inserted in series in the transmission line portion.

[0153] The stub conductors shown in FIGS. 31, 39, 40, and 46 may be configured as open-ended stubs without via conductors, instead of being short-circuited by via conductors.

[0154] The stub conductor configuration shown in FIG. 31, FIG. 39, FIG. 40, or FIG. 46 may be combined with the configuration including a capacitor shown in FIG.

[0155] (Embodiment 16) As mentioned above, in radiation from a leak with stub conductors of a symmetrical structure as shown in Figure 37, the electromagnetic fields are oriented antiparallel, and radiation can cancel each other out. Below, we will explain alternative configurations to avoid or reduce radiation cancellation.

[0156] Fig. 56 is a top view of one basic cell in the leaky wave antenna device according to embodiment 16. Fig. 57 is a cross-sectional view taken along line A-A' in Fig. 56. As shown in Fig. 57, the leaky wave antenna device according to this embodiment has a multilayer structure including a plurality of laminated dielectric layers and a plurality of laminated conductor layers.

[0157] As shown in FIG. 57, the leaky wave antenna device includes a ground conductor 11 and a shield conductor 12. The basic cell has at least first and second parallel branch circuits provided on both sides of at least one of the two transmission line sections. In the example of FIG. 57, the basic cell has a first parallel branch circuit including stub conductors 31Fa and 31Fb and via conductors 33Fa and 33Fb provided on the -x side of the transmission line section (i.e., strip conductor 30) along the y axis, and a second parallel branch circuit including stub conductor 32 and via conductor 34 provided on the +x side. The first parallel branch circuit is located on the opposite side of the ground conductor 11 from the shield conductor 12. The second parallel branch circuit includes a section (e.g., stub conductor 31Fb) located between the ground conductor 11 and the shield conductor 12.

[0158] The other stub conductors and via conductors shown in FIG. 56 are also configured in the same manner as the stub conductors and via conductors in FIG.

[0159] 57, the stub conductor 32 in one of the two parallel branch circuits is exposed on the surface of the leaky wave antenna device and contributes to the upward radiation of leaky waves. On the other hand, the stub conductor 31Fb in the other parallel branch circuit is disposed between the ground conductor 11 and the shield conductor 12 and does not contribute to external radiation. As a result, the leaky waves radiated from the stub conductors 31Fa, 31Fb, and 32 do not interfere with each other and destructively interfere with each other, resulting in a large radiation gain. The space between the stub conductor 31Fb and the shield conductor 12 may be filled with a dielectric or may be free space (i.e., air).

[0160] According to the leaky wave antenna device of this embodiment, it is possible to suppress unwanted radiation from one of a pair of stub conductors inserted as parallel branch circuits along the x-axis and y-axis directions in a two-dimensional lattice structure, and to prevent a reduction in radiation gain due to interference.

[0161] (Embodiment 17) Fig. 58 is a cross-sectional view of one basic cell in a leaky wave antenna device according to embodiment 17. In the leaky wave antenna device of Fig. 58, each parallel branch circuit of at least one of the first and second transmission lines includes a variable capacitance element. In the example of Fig. 58, the first and second parallel branch circuits each include variable capacitance diodes Cv11 and Cv12 inserted in series. The capacitances of the variable capacitance diodes Cv11 and Cv12 are controlled by a DC voltage (reverse bias voltage) applied from a controller 66.

[0162] The leaky wave antenna device of FIG. 58 may further include capacitors C1 and C2 inserted in series in the parallel branch circuits to block DC components.

[0163] According to the leaky wave antenna device of FIG. 58, by providing variable capacitance diodes Cv11 and Cv12, the phase gradient of the electromagnetic field along the ferrite substrate of the microstrip line can be dynamically changed by electronic control.

[0164] FIG. 59 is a cross-sectional view of one basic cell in a leaky-wave antenna device according to a modification of the seventeenth embodiment. Both the first and second parallel branch circuits may have sections disposed between the ground conductor 11 and the shield conductor 12. In the example of FIG. 59, the basic cell has a first parallel branch circuit including stub conductors 31Fa and 31Fb and via conductors 33Fa and 33Fb disposed on the −x side of the transmission line portion (i.e., the strip conductor 30) along the y-axis, and a second parallel branch circuit including stub conductors 32Fa and 32Fb and via conductors 34Fa and 34Fb disposed on the +x side. As described above, both the first and second parallel branch circuits have sections (e.g., stub conductors 31Fb and 32Fb) disposed between the ground conductor 11 and the shield conductor 12. The leaky-wave antenna device of FIG. 59 does not require leaky wave radiation from a pair of stub conductors and has the function of dynamically changing only the phase gradient of the electromagnetic field distribution along the microstrip line by electronic control.

[0165] In the example of Figure 59, capacitors C1 and C2 are arranged on the upper surface of the dielectric substrate 10, and variable capacitance diodes Cv11 and Cv12 are arranged on the lower surface of the dielectric substrate 10, but both may be arranged on the upper surface, or vice versa.

[0166] 58 and 59, a DC voltage is applied from controller 66 to the portion of the stub conductor between variable capacitance diode Cv11 and capacitor C1 and to the portion between variable capacitance diode Cv12 and capacitor C2. Fluctuations in the applied voltage change the capacitance of each of variable capacitance diodes Cv11 and Cv12. By changing the combination of DC voltages applied to the stub conductors while keeping the sum of the admittances of the stub conductors constant, it is possible to continuously change the phase gradient of the electromagnetic field distribution along the line while keeping the operating frequency constant.

[0167] The configuration of the stub conductor shown in FIGS. 56 to 59 may be combined with the configuration including the capacitor shown in FIG.

[0168] (Summary of the embodiment) A nonreciprocal transmission line according to an embodiment includes first and second nonreciprocal transmission lines (first and second transmission lines) that are orthogonal to each other, configured by cascading at least one basic cell in first and second directions that are orthogonal to each other, each basic cell having a parallel branch circuit equivalently including an inductive element or a capacitive element and two transmission line portions that are substantially orthogonal to each other. Here, the transmission line portions at positions other than the intersection of the first and second transmission lines are made of a material that is spontaneously magnetized or magnetized by an external magnetic field so as to have gyroscopic anisotropy by being magnetized in directions different from the propagation direction of microwaves. Furthermore, in a dispersion curve showing the relationship between the operating frequency of a microwave signal input to each transmission line and the propagation constant of each transmission line, at least any of the parallel branch circuits of each transmission line is configured so that a wave component propagating along each parallel branch circuit in the longitudinal direction of each transmission line has nonreciprocal characteristics and is asymmetric with respect to the transmission line portions.

[0169] The non-reciprocal transmission line is used to configure a pseudo traveling wave resonator, and the pseudo traveling wave resonator is used to configure a leaky wave antenna device.

[0170] In particular, by specifying the insertion locations of rotationally anisotropic materials such as ferrite, it is possible to expand the one-dimensional nonreciprocal transmission line proposed in previous research into a two-dimensional structure by forming a mesh. The electromagnetic field distribution within this structure has a nearly uniform intensity distribution, and any wave vector in the leaky wave region can be uniformly distributed. Therefore, when applied to a leaky wave radiation antenna, it is possible to form a main lobe (beam) in the desired direction regardless of the feed position (direction of transmitted power), and further, it becomes possible to independently control the combination of azimuth and elevation angles with only a few control variables.

[0171] In the nonreciprocal transmission line, specifying the insertion location of the ferrite material (ferrite block 40) is the key to expanding the dimensions of the structure according to the present invention. According to the present invention, the electromagnetic field distribution in the two-dimensional nonreciprocal transmission line can be uniformly distributed in intensity and with any wave number vector. Therefore, when applied to a leaky wave radiation antenna, it is possible to form a main lobe (beam) in the desired direction regardless of the feed position (direction of transmitted power). Furthermore, with only a few control variables, it is possible to dynamically control the combination of the azimuth and elevation angles of the beam with low power and low delay.

[0172] Simulation and experimental results of this embodiment show that in the case of a reciprocal transmission line with a uniform phase distribution, the beam direction is perpendicular to the two-dimensional structure. On the other hand, in the case of a nonreciprocal transmission line with a uniform distribution of wave vectors with non-zero magnitude, a beam is formed with an elevation angle and an azimuth angle corresponding to the phase gradient. In the nonreciprocal transmission line of this embodiment, we demonstrated that by manipulating the asymmetry of the stub structure, beam scanning can be performed by changing not only the elevation angle but also the azimuth angle. For example, beam scanning is possible not only at azimuth angles φ = 0° and 90°, which are the main axis directions of the two-dimensional lattice, but also within a plane with an azimuth angle φ = 45°.

[0173] (Variation) At least one basic cell of the nonreciprocal transmission line according to the embodiment may be juxtaposed in at least one of the x-axis direction and the y-axis direction to form the nonreciprocal transmission line.

[0174] In the nonreciprocal transmission line according to the embodiment, the number of basic cells in the first transmission line extending in the x-axis direction and the second transmission line extending in the y-axis direction may be the same or different. Furthermore, the structural parameters of the first transmission line extending in the x-axis direction and the second transmission line extending in the y-axis direction may be the same or different.

[0175] (Summary of the embodiment) A nonreciprocal transmission line according to an embodiment of the present invention provides a transmission line having a two-dimensional waveguide structure that enables electromagnetic wave propagation in a predetermined frequency band and has a "nonreciprocal phase shift characteristic" in which the magnitude of the transmission coefficient is nearly isotropic in a two-dimensional plane, while the value of the effective refractive index (determined by the phase constant or wave number) does not match for combinations of two directions in which the propagation directions are antiparallel.

[0176] In the two-dimensional periodic structure of this nonreciprocal transmission line, the Dirac point formed at the Γ point in two-dimensional wave number space can be shifted by any magnitude and in any direction within the two-dimensional plane by utilizing the nonreciprocal phase shift characteristic. Here, the minimum point of the downwardly convex dispersion surface formed at the Γ point or the maximum point (vertex) of the upwardly convex dispersion surface can be shifted by any magnitude and in any direction within the two-dimensional plane by utilizing the nonreciprocal phase shift characteristic. By selecting a Dirac point that is not at the Γ point or an extreme point of the downwardly convex or upwardly convex dispersion surface that is not at the Γ point as the operating frequency using the nonreciprocal phase shift characteristic described above, the two-dimensional waveguide structure enables electromagnetic wave propagation and can provide an electromagnetic field distribution that exhibits nonreciprocal phase shift characteristics in which the magnitude of the transmission coefficient is nearly isotropic while the direction and magnitude of the wave number vector are constant regardless of the direction of the transmitted power of the electromagnetic wave.

[0177] A two-dimensional network structure in which conventional nonreciprocal transmission lines are simply crossed does not function as the two-dimensional nonreciprocal transmission line of this embodiment. This is easily inferred from the fact that in a simple crisscross structure of two nonreciprocal lines, an input signal propagates asymmetrically with strong directional selectivity to multiple output ports. This asymmetric propagation characteristic with directional selectivity is applied to nonreciprocal circuit elements such as isolators and circulators.

[0178] By using the nonreciprocal transmission line according to this embodiment to construct a pseudo-traveling wave resonator, it is possible to construct a resonator that has a two-dimensional in-plane extent, and the resonant (operating) frequency is determined by the structural parameters of the basic cells that make up the periodic structure, regardless of its size or shape. The electromagnetic field intensity distribution within the resonator is the same (uniform distribution) throughout the resonator, and furthermore, a two-dimensional traveling wave resonator is provided in which the direction and magnitude of the phase gradient (wave vector) can be freely designed in the two-dimensional in-plane direction.

[0179] The pseudo-traveling wave resonator according to this embodiment can be said to be a resonator that extends the electromagnetic field distribution characteristics of a conventional one-dimensional pseudo-traveling wave resonator to a two-dimensional structure. Here, a mesh structure of crossed one-dimensional nonreciprocal transmission lines controls the nonreciprocal phase shift characteristics in two independent directions within a two-dimensional plane, thereby enabling the formation of a phase gradient (wave vector) in any direction within the two-dimensional plane. A two-dimensional mesh structure of simply crossed nonreciprocal transmission lines does not function as the two-dimensional pseudo-traveling wave resonator according to this embodiment. This is easily inferred from the fact that in a simple cross-cross structure of two nonreciprocal lines, an input signal propagates asymmetrically with strong directional selectivity to multiple output ports. This asymmetric directional selectivity propagation characteristic is applied to nonreciprocal circuit elements such as isolators and circulators.

[0180] As explained with reference to Figure 11, electromagnetic waves can be fed to a resonator using only a single transmission line, enabling the resonator to operate in a way that allows independent control of the two-dimensional spatial phase distribution. Normally, controlling the two-dimensional spatial phase distribution requires phase shifters in numbers comparable to the number of antenna elements, resulting in a complex control system. This technology is expected to contribute to a significant reduction in the number of phase shifters and a simplification of the feed line structure for array antennas. Here, the phase gradient is manifested by a combination of the magnitude of the DC magnetization or spontaneous magnetization induced by an externally applied DC magnetic field, and the "structural asymmetry" of the basic cells that make up the resonator.

[0181] In the case of an open-system two-dimensional pseudo-traveling wave resonator, the problem of non-uniformity of the electromagnetic field distribution within the resonator can occur due to attenuation of the electromagnetic field due to leaky wave radiation when the operating point is in the fast wave region. In this case, it is possible to cover the resonator with a metal shielding plate to reduce radiation loss. By placing a metal plate parallel to the top of the two-dimensional nonreciprocal transmission line to suppress radiation loss, and configuring a slot array antenna on that metal plate, it can also be operated as an antenna with two-dimensional beam scanning.

[0182] Furthermore, in a two-dimensional pseudo-traveling wave resonator with a metal shielding plate, by feeding power through a slot array in the metal plate to another antenna array, such as a microstrip antenna array, arranged on an outer layer, it is possible to apply this to a beam scanning antenna with two-dimensional spatial phase distribution control.

[0183] (Other variations) The nonreciprocal transmission line, which is the basic transmission line of this embodiment, has a transmission line configuration consisting of at least one basic cell, such as those shown in Figures 1A to 1C, 2A to 2C, 3, or 27A. Here, the basic cell configuration may include a transmission line portion exhibiting a nonreciprocal phase shift phenomenon in which the forward and reverse propagation constants are different, and an inductive or capacitive element may be equivalently inserted into the parallel branch circuit. Circuits or devices that can be used as the transmission line configuration include printed circuit board circuits, waveguides, and dielectric waveguides used in microwave, millimeter wave, quasi-millimeter wave, and terahertz wave applications, such as strip lines, microstrip lines, slot lines, and coplanar lines, as well as general configurations supporting guided or attenuated modes, including plasmons, polaritons, magnons, and the like, or combinations thereof, and even free space that can be described as an equivalent circuit.

[0184] The transmission line having the nonreciprocal phase shift phenomenon is a transmission line having the above-described transmission line configuration, but including a material with gyroscopic anisotropy (e.g., ferrite block 40) at a position other than the intersection of the first and second transmission lines, and being magnetized in a direction different from the propagation direction of the electromagnetic wave (preferably, a direction perpendicular to the propagation direction) and having a structure that is asymmetric with respect to the plane formed by the propagation direction and the magnetization direction. The transmission line having the nonreciprocal phase shift phenomenon also includes, in addition to the above-described transmission line, lumped-element elements that have an equivalent nonreciprocal phase shift function and are sufficiently small compared to the wavelength. The material having gyroscopic anisotropy includes all cases in which the permittivity tensor, permeability tensor, or both, which represent the properties of the material, are expressed as having gyroscopic anisotropy due to spontaneous magnetization, magnetization induced by an externally applied direct current or low-frequency magnetic field, or the circular motion of free charges. Specific examples of materials that can be targeted include ferrimagnetic materials such as ferrites used in microwaves and millimeter waves, ferromagnetic materials, solid plasmas (such as semiconductor materials), liquid and gas plasma media, and magnetic artificial media constructed by microfabrication.

[0185] The inductive element inserted into the parallel branch circuit can be a lumped-element element such as a coil used in electrical circuits, a distributed-element inductive element such as a short-circuit stub used in microwave or millimeter-wave circuits, or a circuit or element having a negative effective permittivity in the electromagnetic wave mode propagating through the transmission line. Specifically, any microwave circuit operating in a guided or attenuated mode with a negative effective permittivity, such as a spatial arrangement including at least one electric resonator such as a thin metal wire or a metal sphere, a spatial arrangement of not only metal but also dielectric resonators in an electrically resonant state, or a waveguide or parallel-plate line in which the TE mode is in the cutoff region, can be used because they can be described as a transmission line in which the parallel branch operates predominantly as an inductive element in an equivalent circuit. The inductive element inserted into the parallel branch circuit can also be a series or parallel connection of a capacitive element and an inductive element, or a combination thereof. The circuit or element to be inserted may also be inductive as a whole. [Industrial Applicability]

[0186] As described above in detail, the nonreciprocal transmission line according to the present disclosure has a two-dimensional structure and can distribute an electromagnetic field uniformly within a two-dimensional plane. Furthermore, the reciprocal transmission line can be used to configure a pseudo traveling wave resonator, and the pseudo traveling wave resonator can be used to configure a leaky wave antenna device. [Explanation of symbols]

[0187] 1, 1A, 1C~1F basic cell 10 Dielectric substrate 11 Grounding conductor 13 Dielectric substrate 14 Strip conductor 20,20D strip conductor 21, 22, 21A, 21C, 22A to 22D Stub conductors 23,24 Via conductor 25 Pattern conductor 30,30D Strip Conductor 31, 32, 31A, 31C, 31D, 31Fa, 31Fb, 32Fa, 32Fb stub conductors 33, 34, 33Fa, 33Fb, 34Fa, 34Fb Via conductor 40 Ferrite Block 50 via conductor 50h Beer Hall 50Q power supply point 51,52 Strip conductor 60 Radio Signal Generator 61~64 Controller 65 Main Controller 66 Controller 71~74 Control voltage lines 71c~74c Via conductor 71am~74pm Beer Hall 80,80A Shielding Conductor Plate 81 Rectangular slot for coupling 82 Cross slot for coupling 90 Dielectric Substrate 91 Rectangular patch conductor 100 leaky wave antenna device C0~C2 capacitors C V1 ~C V4 ,Cv11,Cv12 variable capacitance diode C S Series Capacitor L1~L5 layer P1,P2 ports

Claims

1. A nonreciprocal transmission line comprising first and second nonreciprocal transmission lines (first and second transmission lines) that are orthogonal to each other, configured by juxtaposing and cascading at least one basic cell in a first and second direction that are orthogonal to each other, each basic cell having a parallel branch circuit that equivalently includes an inductive element or a capacitive element and having two transmission line portions that are substantially orthogonal to each other, a portion of the transmission line excluding an intersection of the first and second transmission lines is made of a material that is magnetized in a direction different from the propagation direction of microwaves and is magnetized spontaneously or by an external magnetic field so as to have gyroscopic anisotropy; a non-reciprocal transmission line configured such that, in a dispersion curve showing the relationship between the operating frequency of a microwave signal input to each of the transmission lines and the propagation constant of each of the transmission lines, at least any of the parallel branch circuits of each of the transmission lines has a non-reciprocal characteristic in a wave component propagating along each of the parallel branch circuits in the longitudinal direction of each of the transmission lines, forming an asymmetric circuit with respect to the transmission line portion.

2. the first and second transmission lines of each basic cell are asymmetric circuits with respect to the transmission line portion; 2. The nonreciprocal transmission line according to claim 1.

3. the first transmission line of each of the basic cells is an asymmetric circuit with respect to the transmission line portion, and the second transmission line of each of the basic cells is a symmetric circuit with respect to the transmission line portion; 2. The nonreciprocal transmission line according to claim 1.

4. each parallel branch circuit of each of the basic cells has a width half that of the parallel branch circuit formed by the nonreciprocal transmission line, and when the basic cells are arranged in parallel and cascade-connected, the parallel branch circuits of adjacent basic cells are coupled at their longitudinal side surfaces to form the parallel branch circuits formed by the nonreciprocal transmission line; 2. The nonreciprocal transmission line according to claim 1.

5. each transmission line portion of each of the basic cells has a width half that of the transmission line portion formed by the nonreciprocal transmission line, and when the basic cells are arranged in parallel and cascade-connected, the transmission line portions of adjacent basic cells are coupled at their longitudinal side surfaces to form the respective transmission line portions formed by the nonreciprocal transmission line; 2. The nonreciprocal transmission line according to claim 1.

6. each of the basic cells has at least first and second parallel branch circuits provided on both sides of at least one of the two transmission line portions, In a plane including the first and second directions, at least a portion of the section of the first parallel branch circuit is orthogonal to at least a portion of the section of the second parallel branch circuit.

2. The nonreciprocal transmission line according to claim 1.

7. the first and second parallel branch circuits each include at least a portion of the parallel branch circuit disposed at an angle of 45 degrees with respect to the transmission line portion in a plane including the first and second directions; 7. The nonreciprocal transmission line according to claim 6.

8. the first parallel branch circuit includes at least a portion of the section disposed at an angle of 90 degrees with respect to the transmission line portion in a plane including the first and second directions; the second parallel branch circuit is bent in a plane including the first and second directions and includes a section parallel to a longitudinal direction of the transmission line portion; 7. The nonreciprocal transmission line according to claim 6.

9. each of the basic cells has at least first and second parallel branch circuits provided on both sides of at least one of the two transmission line portions, the first and second parallel branch circuits are bent in the same direction along the longitudinal direction of the transmission line portion within a plane including the first and second directions, and include sections parallel to the longitudinal direction of the transmission line portion.

2. The nonreciprocal transmission line according to claim 1.

10. the non-reciprocal transmission line includes a ground conductor and a shield conductor; each of the basic cells has at least first and second parallel branch circuits provided on both sides of at least one of the two transmission line portions, the first parallel branch circuit is disposed on an opposite side of the ground conductor from the shield conductor; the second parallel branch includes a section disposed between the ground conductor and the shield conductor; 2. The nonreciprocal transmission line according to claim 1.

11. each parallel branch circuit of at least one of the first and second transmission lines includes a variable capacitance element; 2. The nonreciprocal transmission line according to claim 1.

12. each of the basic cells has a capacitor inserted in series with the transmission line portion; The non-reciprocal transmission line according to any one of claims 1 to 11.

13. A pseudo-traveling wave resonator device comprising the nonreciprocal transmission line according to claim 1, each end of the first and second transmission lines is open; By configuring each parallel branch circuit of each basic cell of the first transmission line so that the parallel resonant frequency of the parallel branch circuit of each basic cell of the second transmission line matches the parallel resonant frequency of the parallel branch circuit of each basic cell of the first transmission line, operating the nonreciprocal transmission line as a pseudo-traveling wave resonator; Pseudo-traveling wave resonator.

14. A leaky wave antenna device comprising the pseudo traveling wave resonator device according to claim 13, A leaky wave antenna device in which a microwave signal is fed to a central portion of the pseudo traveling wave resonator where the first and second transmission lines intersect, thereby radiating a leaky wave electromagnetic wave.

15. 15. The leaky wave antenna device according to claim 14, A leaky wave antenna device comprising a conductor plate provided on a radiation surface of the leaky wave antenna device, the conductor plate having a coupling slot.

16. 16. The leaky wave antenna device according to claim 15, A leaky wave antenna device comprising a microstrip antenna provided on the radiation surface of the conductor plate.

17. The leaky wave antenna device according to any one of claims 14 to 16, A leaky wave antenna device in which circuit parameters of each parallel branch circuit of either the first or second transmission line are set so that the transmission characteristics of the transmission line become non-reciprocal transmission characteristics, thereby propagating and radiating microwaves in the first direction or the second direction.

18. The leaky wave antenna device according to any one of claims 14 to 16, each parallel branch circuit of at least one of the first and second transmission lines includes a variable capacitance element; The leaky wave antenna device changes the radiation direction of the leaky wave radiated from the leaky wave antenna device by changing the capacitance of the variable capacitance element.

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