Signal design method and device immune to phase distortion

Abstract

The present invention relates to a signal design technology immune to phase distortion and may provide a method of a transmitter, the method comprising the steps of: modulating a source bit stream to be transmitted to a receiver to generate a first symbol block; inserting at least one pilot into the first symbol block to generate a second symbol block; accumulatively linearly encoding the second symbol block to generate a third symbol block; performing real number phase transformation for the third symbol block to generate a first signal transformed into a phase value; adding a cyclic prefix (CP) to the first signal to generate a second signal; and wirelessly transmitting the second signal to the receiver.

Description

Method and device for designing a signal immune to phase distortion

[0001] The present disclosure relates to a signal design technique that is immune to phase distortion, and more particularly, to a signal design technique that is immune to phase distortion and has strong resistance to phase distortion caused by hardware damage, Doppler shift, etc.

[0002] Advances in information and communication technology (ICT) can lead to the development of various wireless communication technologies. Representative wireless communication technologies include LTE (long term evolution), NR (new radio), and 6G (6th Generation), all of which are defined by the 3rd Generation Partnership Project (3GPP) standards. LTE can be one of the 4th Generation (4G) wireless communication technologies, and NR can be one of the 5th Generation (5G) wireless communication technologies.

[0003] In order to process the rapidly increasing amount of wireless data following the commercialization of 4G communication systems (e.g., communication systems supporting LTE), 5G communication systems (e.g., communication systems supporting NR) that use a higher frequency band (e.g., a frequency band higher than 6 GHz) than the frequency band of the 4G communication system (e.g., a frequency band below 6 GHz) may be considered. 5G communication systems may support enhanced Mobile Broadband (eMBB), Ultra-Reliable and Low Latency Communication (URLLC), and massive Machine Type Communication (mMTC).

[0004] Meanwhile, most conventional wireless communication systems may be based on the cyclic prefix (CP)-orthogonal frequency division multiplexing (OFDM) transmission scheme, which can be robust to multipath fading and exhibit excellent spectral efficiency. However, the CP-OFDM transmission scheme has a high peak-to-average power ratio (PAPR), which may require high linearity of the amplifier, thereby increasing power consumption. Furthermore, the CP-OFDM transmission scheme may not be inherently robust to phase distortion caused by hardware impairments, such as carrier frequency offset (CFO) and phase noise (PhN), and phase distortion caused by high-speed mobility, such as Doppler shift or Doppler spread. Therefore, wireless communication systems may require technologies that are robust to phase distortion.

[0005] The purpose of the present disclosure to solve the above problems is to provide a signal design method and device that are immune to phase distortion and exhibit high resistance to phase distortion.

[0006] In order to achieve the above object, a method for designing a signal immune to phase distortion according to a first embodiment of the present disclosure may include, as a method of a transmitter, a step of modulating a source bit string to be transmitted to a receiver to generate a first symbol block; a step of inserting at least one pilot into the first symbol block to generate a second symbol block; a step of cumulatively linearly encoding the second symbol block to generate a third symbol block; a step of performing a real-valued phase transform on the third symbol block to generate a first signal converted into a phase value; a step of adding a cyclic prefix (CP) to the first signal to generate a second signal; and a step of wirelessly transmitting the second signal to a receiver.

[0007] Here, in the step of generating a first symbol block by modulating a source bit stream to be transmitted to the receiver, the transmitter may generate the first symbol block from the source bit stream using any one of BASK (bipolar amplitude shift keying) modulation, PSK (phase shift keying) modulation, QAM (quadrature and amplitude modulation) modulation, GMSK (Gaussian filtered minimum-shift keying) modulation, or DPSK (differential phase-shift keying) modulation.

[0008] Here, in the step of inserting at least one pilot into the first symbol block, the transmitter can insert a first pilot into a preceding time domain of the first symbol block and a second pilot into a succeeding time domain of the first symbol block.

[0009] Here, the last time domain samples of the first pilot and the first time domain samples of the second pilot may have a power value of zero.

[0010] Here, in the step of inserting at least one pilot into the first symbol block, the transmitter can insert a third pilot into the middle of the first symbol block.

[0011] Here, the first time domain samples and the last time domain samples of the third pilot may have a power value of zero.

[0012] Here, the step of generating a third symbol block by performing cumulative linear encoding on the second symbol block may include the step of dividing the second symbol block into first partial symbol blocks based on a cyclic period of a time domain sample unit; and the step of performing differential encoding on each of the first partial symbol blocks to generate the cumulative linearly encoded third symbol block.

[0013] Here, in the step of generating the cumulative linearly encoded third symbol block by performing differential encoding on each of the first partial symbol blocks, the transmitter may perform differential encoding after adding the first sample value of the previous first partial symbol block to the first sample of each of the first partial symbol blocks.

[0014] Here, the step of generating a third symbol block by performing cumulative linear encoding on the second symbol block may include the step of dividing the first symbol block within the second symbol block into second partial symbol blocks; and the step of performing differential encoding on each of the second partial symbol blocks to generate the cumulative linearly encoded third symbol block.

[0015] Here, the step of generating a third symbol block by cumulatively linearly encoding the second symbol block may include the step of differentially encoding the second symbol block to generate the cumulatively linearly encoded third symbol block.

[0016] Meanwhile, a method for designing a signal immune to phase distortion according to a second embodiment of the present disclosure for achieving the above object may include, as a method of a receiver, the steps of: converting a received signal received from a transmitter into a baseband signal; removing a CP from the baseband signal; equalizing a wireless channel with respect to the received signal from which the CP has been removed; performing a time phase transform on the received signal from which the wireless channel has been equalized; performing cumulative linear decoding on the time phase transformed signal; and demodulating the cumulative linear decoded signal to generate a source bit stream.

[0017] Here, the step of equalizing a wireless channel for the received signal from which the CP has been removed may include: a step of converting a time domain pilot signal of the received signal from which the CP has been removed into a frequency domain pilot signal; a step of converting a time domain symbol signal of the received signal from which the CP has been removed into a frequency domain symbol signal; a step of estimating a first wireless channel from the frequency domain pilot signal; a step of estimating a second wireless channel from the frequency domain symbol signal; and a step of equalizing the second wireless channel using the first wireless channel.

[0018] Here, the step of performing cumulative linear decoding on the time phase-shifted signal may include the step of dividing the time phase-shifted signal into first partial symbol blocks based on a cyclic period of a time domain sample unit; and the step of performing differential decoding on each of the first partial symbol blocks to generate the cumulative linear decoded signal.

[0019] Here, the step of performing cumulative linear decoding on the time phase-shifted signal may include the steps of: extracting a symbol block of the time phase-shifted signal; dividing the symbol block into second partial symbol blocks; and performing differential decoding on each of the second partial symbol blocks to generate the cumulative linear decoded signal.

[0020] Here, the step of performing cumulative linear decoding on the time phase-shifted signal may include the step of extracting a symbol block from the time phase-shifted signal; and the step of differentially decoding the symbol block to generate the cumulative linear decoded signal.

[0021] Here, the step of demodulating the cumulative linear decoded signal to generate a source bit stream may include the steps of demodulating the cumulative linear decoded signal to generate a first symbol block; removing a pilot from the first symbol block to generate a second symbol block; and performing symbol demapping on the second symbol block to generate the source bit stream.

[0022] Meanwhile, a signal design device immune to phase distortion according to a third embodiment of the present disclosure for achieving the above object includes a processor as a transmitter, wherein the processor can cause the transmitter to modulate a source bit string to be transmitted to a receiver to generate a first symbol block; generate a second symbol block by inserting at least one pilot into the first symbol block; generate a third symbol block by cumulatively linearly encoding the second symbol block; generate a first signal converted into a phase value by performing a real-valued phase transform on the third symbol block; generate a second signal by adding a CP (cyclic prefix) to the first signal; and wirelessly transmit the second signal to the receiver.

[0023] Here, in the step of inserting at least one pilot into the first symbol block, the processor can cause the transmitter to insert a first pilot into a preceding time domain of the first symbol block and to insert a second pilot into a succeeding time domain of the first symbol block.

[0024] Here, in the step of inserting at least one pilot into the first symbol block, the processor may cause the transmitter to insert a third pilot into the middle of the first symbol block.

[0025] Here, in the step of generating a third symbol block by cumulatively linearly encoding the second symbol block, the processor may cause the transmitter to divide the second symbol block into first partial symbol blocks based on a cyclic period of a time-domain sample unit; and perform differential encoding on each of the first partial symbol blocks to generate the cumulatively linearly encoded third symbol block.

[0026] According to the present disclosure, a wireless communication system can eliminate centrally symmetric mapping of identical data without reducing spectral efficiency. Furthermore, according to the present disclosure, the wireless communication system can effectively allocate pilots for channel estimation. Furthermore, according to the present disclosure, the wireless communication system can weight the resulting values ​​for each phase-shifted time-domain sample to a degree that quantization error does not become a problem, thereby achieving high resistance to phase distortion.

[0027] Figure 1 is a conceptual diagram illustrating a first embodiment of a communication system.

[0028] Figure 2 is a block diagram illustrating a first embodiment of a communication node constituting a communication system.

[0029] FIG. 3 is a block diagram showing a first embodiment of a signal design device immune to phase distortion.

[0030] FIG. 4 is a flowchart illustrating a first embodiment of a signal design method that is immune to phase distortion performed at a transmitter.

[0031] Figure 5a is a conceptual diagram showing the signal constellation of 2-BASK modulation.

[0032] Figure 5b is a conceptual diagram showing the signal constellation of 4-BASK modulation.

[0033] Figure 5c is a conceptual diagram showing the signal constellation of 8-BASK modulation.

[0034] Fig. 6a is a conceptual diagram showing a first embodiment of a signal constellation of 2-PSK modulation.

[0035] Figure 6b is a conceptual diagram showing a second embodiment of the signal constellation of 2-PSK modulation.

[0036] Figure 6c is a conceptual diagram showing the signal constellation of 4-PSK modulation.

[0037] Figure 6d is a conceptual diagram showing the signal constellation of 8-PSK modulation.

[0038] Figure 7a is a conceptual diagram showing the signal constellation of 4-QAM modulation.

[0039] Figure 7b is a conceptual diagram showing the signal constellation of 16-QAM modulation.

[0040] Figure 8 is a conceptual diagram showing a first embodiment of a pilot insertion process performed in a pilot insertion unit.

[0041] Figure 9 is a conceptual diagram of the CP addition process of the CP addition unit.

[0042] Fig. 10 is a flowchart illustrating a first embodiment of a signal design method that is immune to phase distortion performed at a receiving end.

[0043] Figure 11a is a conceptual diagram of the 2-BASK demapping process performed in a 2-BASK symbol demapper.

[0044] Figure 11b is a conceptual diagram of the 4-BASK demapping process performed in a 4-BASK symbol demapper.

[0045] Figure 11c is a conceptual diagram illustrating the 8-BASK demapping process performed in an 8-BASK symbol demapper.

[0046] Figure 12 is a conceptual diagram showing the 8-PSK demapping process performed in an 8-PSK symbol demapper.

[0047] Figure 13 is a conceptual diagram showing the 16-QAM demapping process performed by a 16-QAM symbol demapper.

[0048] Fig. 14 is a flowchart illustrating a second embodiment of a signal design method that is immune to phase distortion performed at a transmitter.

[0049] Figure 15 is a conceptual diagram showing a second embodiment of a pilot insertion process performed in a pilot insertion unit.

[0050] Fig. 16 is a flowchart illustrating a second embodiment of a signal design method that is immune to phase distortion performed at a receiving end.

[0051] Fig. 17 is a block diagram showing a second embodiment of a signal design device immune to phase distortion.

[0052] Fig. 18 is a flowchart illustrating a third embodiment of a signal design method that is immune to phase distortion performed at a transmitter.

[0053] Fig. 19 is a flowchart illustrating a third embodiment of a signal design method that is immune to phase distortion performed at a receiving end.

[0054] Fig. 20 is a flowchart illustrating a fourth embodiment of a signal design method that is immune to phase distortion performed at a transmitter.

[0055] Fig. 21 is a flowchart illustrating a fourth embodiment of a signal design method that is immune to phase distortion performed at a receiving end.

[0056] Fig. 22 is a block diagram showing a third embodiment of a signal design device immune to phase distortion.

[0057] This disclosure may be subject to various modifications and various embodiments. Specific embodiments are illustrated and described in detail in the drawings. However, this is not intended to limit the disclosure to specific embodiments, but rather to encompass all modifications, equivalents, and alternatives falling within the spirit and technical scope of the disclosure.

[0058] While terms such as "first" and "second" may be used to describe various components, these components should not be limited by these terms. These terms are used solely to distinguish one component from another. For example, without departing from the scope of the present disclosure, a first component could be referred to as a "second component," and similarly, a second component could also be referred to as a "first component." The term "and / or" includes a combination of multiple related items described herein or any of multiple related items described herein.

[0059] In embodiments of the present disclosure, “at least one of A and B” may mean “at least one of A or B” or “at least one of combinations of one or more of A and B.” Furthermore, in embodiments of the present disclosure, “at least one of A and B” may mean “at least one of A or B” or “at least one of combinations of one or more of A and B.”

[0060] When a component is referred to as being "connected" or "connected" to another component, it should be understood that it may be directly connected or connected to that other component, but that there may be other components intervening. Conversely, when a component is referred to as being "directly connected" or "connected" to another component, it should be understood that there are no other components intervening.

[0061] The terminology used in this disclosure is only used to describe specific embodiments and is not intended to limit the present disclosure. The singular expression includes the plural expression unless the context clearly indicates otherwise. In this disclosure, it should be understood that the terms "comprises" or "has" indicate the presence of a feature, number, step, operation, component, part, or combination thereof described in the specification, but do not preclude the presence or addition of one or more other features, numbers, steps, operations, components, parts, or combinations thereof.

[0062] Unless otherwise defined, all terms used herein, including technical or scientific terms, have the same meaning as commonly understood by a person of ordinary skill in the art to which this disclosure pertains. Terms defined in commonly used dictionaries should be interpreted as having a meaning consistent with their meaning in the context of the relevant technology, and shall not be interpreted in an idealized or overly formal sense unless explicitly defined herein.

[0063] Hereinafter, preferred embodiments of the present disclosure will be described in more detail with reference to the attached drawings. In order to facilitate an overall understanding in describing the present disclosure, identical reference numerals will be used for identical components in the drawings, and redundant descriptions of identical components will be omitted.

[0064] Figure 1 is a conceptual diagram illustrating a first embodiment of a communication system.

[0065] Referring to FIG. 1, a communication system (100) may include a plurality of communication nodes (110-1, 110-2, 110-3, 120-1, 120-2, 130-1, 130-2, 130-3, 130-4, 130-5, 130-6). Here, the communication system may be referred to as a "communication network." Each of the plurality of communication nodes may support at least one communication protocol. For example, each of the plurality of communication nodes may support a communication protocol based on CDMA (code division multiple access), a communication protocol based on WCDMA (wideband CDMA), a communication protocol based on TDMA (time division multiple access), a communication protocol based on FDMA (frequency division multiple access), a communication protocol based on OFDM (orthogonal frequency division multiplexing), a communication protocol based on OFDMA (orthogonal frequency division multiple access), a communication protocol based on SC (single carrier)-FDMA, a communication protocol based on NOMA (non-orthogonal multiple access), a communication protocol based on SDMA (space division multiple access), etc. Each of the plurality of communication nodes may have the following structure.

[0066] Figure 2 is a block diagram illustrating a first embodiment of a communication node constituting a communication system.

[0067] Referring to FIG. 2, a communication node (200) may include at least one processor (210), a memory (220), and a transceiver (230) that is connected to a network and performs communication. In addition, the communication node (200) may further include an input interface device (240), an output interface device (250), a storage device (260), etc. Each component included in the communication node (200) may be connected by a bus (270) to perform communication with each other. However, each component included in the communication node (200) may be connected through an individual interface or an individual bus centered around the processor (210), rather than a common bus (270). For example, the processor (210) may be connected to at least one of the memory (220), the transceiver (230), the input interface device (240), the output interface device (250), and the storage device (260) through a dedicated interface.

[0068] The processor (210) can execute program commands stored in at least one of the memory (220) and the storage device (260). The processor (210) may refer to a central processing unit (CPU), a graphics processing unit (GPU), or a dedicated processor in which the methods according to embodiments of the present disclosure are performed. Each of the memory (220) and the storage device (260) may be configured with at least one of a volatile storage medium and a non-volatile storage medium. For example, the memory (220) may be configured with at least one of a read-only memory (ROM) and a random access memory (RAM).

[0069] Referring again to FIG. 1, the communication system (100) may include a plurality of base stations (110-1, 110-2, 110-3, 120-1, 120-2) and a plurality of user equipment (UEs) (130-1, 130-2, 130-3, 130-4, 130-5, 130-6). Each of the first base station (110-1), the second base station (110-2), and the third base station (110-3) may form a macro cell. Each of the fourth base station (120-1) and the fifth base station (120-2) may form a small cell. The fourth base station (120-1), the third UE (130-3), and the fourth UE (130-4) may be within the coverage of the first base station (110-1). The second UE (130-2), the fourth UE (130-4), and the fifth UE (130-5) may be within the coverage of the second base station (110-2). The fifth base station (120-2), the fourth UE (130-4), the fifth UE (130-5), and the sixth UE (130-6) may be within the coverage of the third base station (110-3). The first UE (130-1) may be within the coverage of the fourth base station (120-1). The sixth UE (130-6) may be within the coverage of the fifth base station (120-2).

[0070] Here, each of the plurality of base stations (110-1, 110-2, 110-3, 120-1, 120-2) may be referred to as a NodeB, an evolved NodeB, a BTS (base transceiver station), a radio base station, a radio transceiver, an access point, an access node, a road side unit (RSU), a DU (digital unit), a CDU (cloud digital unit), a RRH (radio remote head), a RU (radio unit), a TP (transmission point), a TRP (transmission and reception point), a relay node, etc. Each of the plurality of UEs (130-1, 130-2, 130-3, 130-4, 130-5, 130-6) may be referred to as a terminal, an access terminal, a mobile terminal, a station, a subscriber station, a mobile station, a portable subscriber station, a node, a device, etc.

[0071] Each of the plurality of communication nodes (110-1, 110-2, 110-3, 120-1, 120-2, 130-1, 130-2, 130-3, 130-4, 130-5, 130-6) can support cellular communication (e.g., long term evolution (LTE), LTE-A (advanced) as defined in the 3rd generation partnership project (3GPP) standard). Each of the plurality of base stations (110-1, 110-2, 110-3, 120-1, 120-2) can operate in a different frequency band or can operate in the same frequency band. Each of the plurality of base stations (110-1, 110-2, 110-3, 120-1, 120-2) can be connected to each other via an ideal backhaul or a non-ideal backhaul, and can exchange information with each other via the ideal backhaul or the non-ideal backhaul. Each of the plurality of base stations (110-1, 110-2, 110-3, 120-1, 120-2) can be connected to a core network (not shown) via an ideal backhaul or a non-ideal backhaul. Each of the plurality of base stations (110-1, 110-2, 110-3, 120-1, 120-2) can transmit a signal received from the core network to the corresponding UE (130-1, 130-2, 130-3, 130-4, 130-5, 130-6), and can transmit a signal received from the corresponding UE (130-1, 130-2, 130-3, 130-4, 130-5, 130-6) to the core network.

[0072] Each of the plurality of base stations (110-1, 110-2, 110-3, 120-1, 120-2) can support OFDMA-based downlink transmission and SC-FDMA-based uplink transmission. In addition, each of the plurality of base stations (110-1, 110-2, 110-3, 120-1, 120-2) can support MIMO (multiple input multiple output) transmission (e.g., single user (SU)-MIMO, multi user (MU)-MIMO, massive MIMO, etc.), CoMP (coordinated multipoint) transmission, carrier aggregation transmission, transmission in an unlicensed band, device to device (D2D) communication (or, ProSe (proximity services), etc.). Here, each of the plurality of UEs (130-1, 130-2, 130-3, 130-4, 130-5, 130-6) can support the base station (110-1, 110-2, 110-3, 120-1, 120-2) and can perform operations supported by base stations (110-1, 110-2, 110-3, 120-1, 120-2).

[0073] Meanwhile, most conventional wireless communication systems are based on the cyclic prefix (CP)-orthogonal frequency division multiplexing (OFDM) transmission scheme, which can be robust to multipath fading and exhibit excellent spectral efficiency. Examples of such wireless communication systems include 4G (fourth generation) LTE (long-term evolution) and 5G (fifth generation) NR (new radio). However, the CP-OFDM transmission scheme has a high peak-to-average power ratio (PAPR), which can require high linearity from amplifiers, thereby increasing power consumption.

[0074] Therefore, wireless communication systems can utilize single-carrier (SC) transmission, which has virtually no PAPR or a very low PAPR. However, SC transmission can be vulnerable to multipath fading and has lower spectral efficiency than CP-OFDM transmission, making it a wireless transmission method rarely adopted in wireless communication standard systems.

[0075] Furthermore, CP-OFDM transmission schemes may not be inherently robust to phase distortion caused by hardware impairments, such as carrier frequency offset (CFO) and phase noise (PhN), and phase distortion caused by high-speed mobility, such as Doppler shift or Doppler spread. Therefore, wireless communication systems may require the application of phase-distortion-robust technologies.

[0076] Recently, a frequency-modulated (FM)-OFDM transmission method based on OFDM, which possesses inherent robustness against high phase-average power ratio (PAPR) and phase distortion, has been published in an academic journal. This FM-OFDM method can map data centrally symmetrically around the direct current (DC) in the frequency domain. Furthermore, FM-OFDM can transform this centrally symmetric data around the DC into the time domain, generating a signal containing only real components.

[0077] Subsequently, the FM-OFDM method can frequency-modulate signals containing only real components, convert them into phase, and transmit them with a CP. This allows the FM-OFDM method to achieve zero PAPR and effectively eliminate phase distortion caused by FM operation.

[0078] However, the FM-OFDM transmission scheme may reduce the given spectral efficiency by more than half due to the centrally symmetric mapping of the same data in the frequency domain. In addition, since the FM-OFDM transmission scheme performs phase shift after performing FM in the time domain, the estimation performance may be poor due to the effects of phase distortion and multipath wireless channels when pilots for channel estimation are allocated in the frequency domain. In addition, the FM-OFDM transmission scheme may set the weights excessively low to ensure that the time-domain sample-by-sample results of the phase-shifted FM performance are formed between [-π,π) or (-π,π).

[0079] This FM-OFDM transmission scheme may have degraded data recovery performance depending on low weight values. In addition, the FM-OFDM transmission scheme may cause quantization errors depending on low weight values, which may degrade data recovery performance. To reduce the negative effects of these low weight values, the bit width of the analog-to-digital converter (ADC) can be set to a large value when implementing the FM-OFDM transmission scheme. As a result, the FM-OFDM transmission scheme increases implementation complexity, which may increase power consumption. In addition, since the FM-OFDM transmission scheme performs phase shift after performing FM in the time domain, there may be a problem with the orthogonality of OFDM. Accordingly, the FM-OFDM transmission scheme cannot allocate data subcarriers using FM-OFDM directly adjacent to data subcarriers using CP-OFDM in the frequency domain.

[0080] In other words, the FM-OFDM transmission method may not be compatible with the CP-OFDM transmission method (Frequency Division Multiplexing, FDM), and thus may not achieve the original spectral efficiency of OFDM. Therefore, to solve this problem, a mobile communication system may need to develop a wireless transmission method that does not reduce spectral efficiency by eliminating the centrally symmetric mapping of the same data, can effectively allocate pilots for channel estimation, and has high resistance to phase distortion that minimizes quantization error in the resulting values ​​for each phase-shifted time-domain sample.

[0081] The present disclosure provides a signal design, a wireless transmission method, a new waveform, and a communication device that are highly resistant to phase distortion, can efficiently allocate pilots, and can average out errors while minimizing quantization errors, in order to solve the problems of the prior art described above.

[0082] In the present disclosure, a wireless device may be referred to as a mobile station (MS). Furthermore, a device that transmits a signal to the MS or receives a signal from the MS may be referred to as a transmission and reception point (TRP). A device that manages a TRP may be referred to as a base station (BS). The area managed by a BS may be referred to as a cell. Specific methods, procedures, and devices of the present disclosure may be described based on the configuration of the MS, TRP, BS, and cell.

[0083] Additionally, in the present disclosure, a BS may manage multiple TRPs, but may only manage a single TRP. Furthermore, multiple MSs or only one MS may exist within a cell managed by the BS. The MS may perform uplink wireless transmissions to the TRP. Furthermore, the TRP may perform downlink wireless transmissions to the MS.

[0084] FIG. 3 is a block diagram showing a first embodiment of a signal design device immune to phase distortion.

[0085] Referring to FIG. 3, the transmitter may include an M-BASK (bipolar amplitude shift keying) symbol mapper (301), a pilot insertion unit (302), a phase-shift encoding unit (303), a real phase conversion unit (304), a CP addition unit (305), and / or a baseband filter (306). The receiver may include a baseband filter (311), a CP removal unit (312), a time-frequency conversion unit (313), a frequency domain equalization unit (315), a channel estimation unit (315), a frequency-time conversion unit (316), a symbol extraction unit (317), a time-phase conversion unit (318), a phase-shift decoding unit (319), a symbol detection unit (320), and / or an M-BASK symbol demapper (321).

[0086] The transmitter may be a TRP, and the receiver may be a terminal. The transmitter may use a phase distortion-resistant single carrier (PR-SC) transmission scheme based on serrated encoding (i.e., a gear-shaped encoding). Here, the PR-SC transmission scheme based on serrated encoding can have high phase distortion resistance and a zero phase-to-noise ratio (PAPR).

[0087] FIG. 4 is a flowchart illustrating a first embodiment of a signal design method that is immune to phase distortion performed at a transmitter.

[0088] Referring to Fig. 4, the TRP can generate a source bit stream to be transmitted to the terminal (S401). Accordingly, the M-BASK symbol mapper of the TRP can perform M-BASK modulation on the generated source bit stream to generate a symbol stream or source symbol block composed of elements of n=0,1,…,N1 having only integer components (S402). Here, M and N1 can be positive integers. And, the M-BASK symbol mapper can be an example of a symbol mapper.

[0089] Figure 5a is a conceptual diagram showing the signal constellation of 2-BASK modulation.

[0090] Referring to Fig. 5a, the 2-BASK symbol mapper can receive one bit and output it by mapping, for example, the received bit 0 to -1 and the received bit 1 to 1. The absolute maximum size A2 of the output of the 2-BASK symbol mapper can be A2=1.

[0091] Figure 5b is a conceptual diagram showing the signal constellation of 4-BASK modulation.

[0092] Referring to Fig. 5b, the 4-BASK symbol mapper can receive two bits and output them by mapping, for example, the received bit 00 to -1, the received bit 01 to -2, the received bit 10 to 1, and the received bit 11 to 2. The absolute maximum size A4 of the output of the 4-BASK symbol mapper can be A4=2.

[0093] Figure 5c is a conceptual diagram showing the signal constellation of 8-BASK modulation.

[0094] Referring to Fig. 5c, the 8-BASK symbol mapper can receive three bits and output them by mapping, for example, the received bit 000 to -1, the received bit 001 to -2, the received bit 010 to -3, the received bit 011 to -4, the received bit 100 to 1, the received bit 101 to 2, the received bit 110 to 3, and the received bit 111 to 4. The absolute maximum size A8 of the output of the 8-BASK symbol mapper can be A8=4.

[0095] Since the M-BASK mapping rule can be extended in the above manner, a detailed description thereof may be omitted. Here, the integer signal constellation M-BASK mapping rules of the above bit / bits may be an example, and mapping rules of all other possible modulations may be included in the scope of the present disclosure. For example, another modulation mapping rule may be an M-PSK (phase shift keying) modulation mapping rule. An M-PSK mapper may perform M-PSK modulation on the generated source bit string to generate a symbol string or symbol block composed of elements of n=0,1,…,N1 having only a phase component.

[0096] Fig. 6a is a conceptual diagram showing a first embodiment of a signal constellation of 2-PSK modulation.

[0097] Referring to Figure 6a, the 2-PSK symbol mapper receives one bit and, for example, receives bit 0. , the received bit 1 It can be output by mapping to . The absolute maximum size A2 excluding the π value of the phase component of the output of the 2-PSK symbol mapper can be A2 = 1. Here, 2-PSK can be BPSK (binary phase shift keying).

[0098] Figure 6b is a conceptual diagram showing a second embodiment of the signal constellation of 2-PSK modulation.

[0099] Referring to Figure 6b, the 2-PSK symbol mapper receives one bit and, for example, receives bit 0. , the received bit 1 It can be output by mapping to . The absolute maximum size A2 excluding the π value of the phase component of the output of the 2-PSK symbol mapper can be A2=1 / 2.

[0100] Figure 6c is a conceptual diagram showing the signal constellation of 4-PSK modulation.

[0101] Referring to Figure 6c, the 4-PSK symbol mapper receives two bits, for example, the received bit 00. , the received bit 01 , the received bit 10 , the received bit 11 It can be output by mapping to . The absolute maximum size A4 excluding the π value of the phase component of the output of the 4-PSK symbol mapper can be A4=3 / 4. Here, 4-PSK can be QPSK (quadrature phase shift keying) or 4-QAM (quadrature and amplitude modulation).

[0102] Figure 6d is a conceptual diagram showing the signal constellation of 8-PSK modulation.

[0103] Referring to Figure 6d, the 8-PSK symbol mapper receives three bits and, as an example, receives bit 000. , the received bit 001 , the received bit 010 , the received bit 011 , the received bit is 100 , the received bit 101 , received bit 110 , the received bit 111 It can be output by mapping to . The absolute maximum size A8 excluding the π value of the phase component of the output of the 8-PSK symbol mapper can be A8=1. These mapping rules and A M The value of can be one example and all other possible mapping rules and A M The value of may be included in the scope of the present disclosure. Other modulation mapping rules may be M-QAM modulation mapping rules.

[0104] The M-QAM mapper can perform M-QAM modulation on the generated source bitstream to generate a symbol stream or symbol block consisting of elements of n=0,1,…,N1 that only have phase components. 2-QAM is identical to the BPSK described above, so its description can be omitted. 4-QAM is identical to the QPSK described above, so its description can be omitted.

[0105] Figure 7a is a conceptual diagram showing the signal constellation of 4-QAM modulation.

[0106] Referring to Figure 7a, the 4-QAM symbol mapper receives two bits, for example, the received bit 00. , the received bit 01 , the received bit 10 , the received bit 11 It can be output by mapping to . The absolute maximum size A4 excluding the π value of the phase component of the output of the 4-QAM symbol mapper can be A4 = 3 / 4. Here, 4-QAM can be QPSK or 4-PSK.

[0107] Figure 7b is a conceptual diagram showing the signal constellation of 16-QAM modulation.

[0108] Referring to Figure 7b, the 16-QAM symbol mapper receives four bits and, as an example, receives bit 0000. , the received bit 0001 , the received bit 0101 , the received bit 0100 , the received bit 0010 , the received bit 0011 , the received bit 0111 , the received bit 0110 , the received bit is 1000 , the received bit 1001 , received bit 1101 , received bit 1100 , the received bit 1010 , the received bit 1011 , the received bit 1111 , received bit 1110 It can be output by mapping to . The absolute maximum size A excluding the π value of the phase component of the output of the 16-QAM symbol mapper 16 Silver A 16 =1 may be.

[0109] The values ​​of elements of n=0,1,…,N1 having only the magnitude component above are the element values ​​after the real-to-phase converting process (in other words, The size component may be multiplied by the shape. In addition, the absolute maximum size A of the output of the mapping rule and symbol mapper above M The value of is one example and all other possible mapping rules and A M The value of may be included in the scope of the present disclosure. Here, M may be a positive integer. Other modulation mapping rules, such as GMSK (Gaussian filtered Minimum-Shift Keying) and DPSK (differential phase-shift keying), may also be included in the scope of the present disclosure. In addition, the receiver may perform demodulation and restoration by taking into account the characteristics of the modulation methods mentioned above.

[0110] Referring again to FIG. 4, the pilot insertion unit can insert pilots (e.g., pilot 1 and pilot 2) before and after the source symbol block of the signal output from the M-BASK mapper (S403). The pilots inserted in this way can easily and effectively restore or equalize the nonlinear wireless channel within each symbol of the source symbol block at the receiver.

[0111] Figure 8 is a conceptual diagram showing a first embodiment of a pilot insertion process performed in a pilot insertion unit.

[0112] Referring to FIG. 8, the pilot insertion unit can insert pilot 1 in front of the source symbol block generated by the M-BASK mapper. Then, the pilot insertion unit can insert pilot 2 in the back of the source symbol block of N1 bits generated by the M-BASK mapper. Here, pilot 1 and pilot 2 can be P bits. Here, P can be a positive integer. At this time, pilot 1 and pilot 2 can be the same. Alternatively, pilot 1 and pilot 2 can be different. Accordingly, the output x(n) of the pilot insertion unit can be composed of N-1 bits. Here, N can be a positive integer. And, N can be 2P+N1. N can be 0, 1,…, N-1.

[0113] The pilot insertion unit may assign a power value of zero to the last time-domain samples of Pilot 1 and the first time-domain samples of Pilot 2 to minimize interference caused by multipath when performing wireless channel estimation and frequency-domain equalization at the receiver. N1 may denote the length of a time-domain sample unit of a symbol block, and P may denote the length of a time-domain sample unit of Pilot 1 and Pilot 2. Pilot 1 and Pilot 2 may have the same length, but may not be limited thereto. The pilot length may be set in consideration of the length of the symbol block, frequency efficiency, etc. In addition, the pilot insertion unit may configure only Pilot 1 for reasons such as improving frequency efficiency. Alternatively, the pilot insertion unit may configure only Pilot 2 for reasons such as improving frequency efficiency. The pilot insertion process as described above may be performed after performing a floating-point encoding process.

[0114] Again, referring to FIG. 4, the phase encoding unit can perform phase encoding on the M-BASK modulated symbol block signal with the pilot inserted (S404). Here, the phase may be a sawtooth shape. At this time, the output q(n) of the phase encoding unit can be modeled as in the following mathematical expression 1 or mathematical expression 2. Here, in the case of applying the M-PSK modulation method mentioned above, the phase encoding unit can take only the value excluding π of the phase component for each constellation point and apply it to x(·) modeled in mathematical expression 1 or mathematical expression 2. In addition, in the case of applying the M-QAM modulation method mentioned above, the phase encoding unit can similarly take only the value excluding π of the phase component for each constellation point and apply the obtained value to x(·) modeled by mathematical expression 1 or mathematical expression 2.

[0115]

[0116]

[0117] In mathematical expressions 1 and 2, n is a positive real number. can be. Here, n' is a positive real number. can be, and n" is a positive real number. It can be. In mathematical equations 1 and 2, min(1,n') can be a function that compares the sizes of 1 and n' and selects the smaller one. can be a function that takes the integer part of y plus 1 when y is a positive real number with a decimal point, and takes the integer part of y as it is when y does not have a decimal point. B can be a cyclic period of the time domain sample unit. Here, B can be a positive real number.

[0118] In one embodiment, the floating-point encoding unit can perform floating-point encoding including the pilot as in Equation 1. Alternatively, in another embodiment, the floating-point encoding unit can perform floating-point encoding only on the source symbol block while leaving the pilot as in Equation 2. In this case, it can be efficient if P is constrained to be a multiple of B (e.g., P=B, 2B, 3B,…). Of course, P may not be a multiple of B. For the convenience of all descriptions of the technology of the present disclosure, P may be set to P=B.

[0119] The forward encoding unit can perform forward encoding on a source symbol block without including a pilot as in Equation 2. At this time, since the first sample to be forward encoded may be susceptible to hardware damage, the last sample(s) of the preceding pilot 1 known to both the transmitter and receiver can be taken as the first sample. In addition, the forward encoding unit can perform forward encoding in ascending order by applying x(0) or x(P) as the initial value in Equations 1 and 2. Alternatively, the forward encoding unit can perform forward encoding in descending order by applying the last sample.

[0120] Such a rolling encoding may be a method of performing a new differential encoding for every cycle B of a rolling time domain sample unit. In other words, the rolling encoding unit may divide the M-BASK modulated symbol block with the pilot inserted into it into first partial symbol blocks in N / B time domains according to mathematical expression 1. Then, the rolling encoding unit may perform differential encoding on each of the first partial symbol blocks. At this time, the rolling encoding unit may perform differential encoding after adding the first sample value of the previous first partial symbol block to the first sample of the first partial symbol blocks.

[0121] In contrast, the floating-point encoding unit can divide the source symbol block into N1 / B time domain second partial symbol blocks in the M-BASK modulated symbol block with the pilot inserted according to Equation 2. Then, the floating-point encoding unit can perform differential encoding on each of the second partial symbol blocks. At this time, the floating-point encoding unit can perform differential encoding after adding the first sample value of the previous second partial symbol block to the first sample of the second partial symbol blocks. Here, the floating-point encoding can be accumulation linear encoding.

[0122] The real-to-phase converting unit can convert the forward-encoded signal into a phase value z(n) as in Equation 3 or Equation 4 through a real-to-phase converting process (S405). In the case of applying the above-mentioned M-PSK and M-QAM modulation methods, the real-phase converting unit can follow the real-phase converting process in the case of applying the M-BASK modulation method. Here, Equation 3 may be a case where the real-phase converting process is performed on the entire forward-encoded signal. Equation 4 may be a case where the real-phase converting process is performed on a portion corresponding to a source symbol block in the forward-encoded signal.

[0123]

[0124] Here, n can be 0,1,…,N-1. Eun Yang's mistake It may be, It can be. θ is a positive real number. It can be. α is a positive real number that can converge to 0.

[0125]

[0126] Here, Eun Yang's mistake It may be, can be. θ is a positive real number. It can be. α is a positive real number that can converge to 0.

[0127] In Equations 3 and 4, the maximum value of q(n) is A due to the sawtooth encoding. M (B+1) can be A, the maximum value of q(n). M The inverse of (B+1) can be θ. q(n) of time-domain sample n is weighted by θ. This could be it. For example, θ exists in (-1,1) or [-1,1). , and if the positive α converges to 0, the ambiguity occurring in the demodulation process at the receiver can be eliminated. The above weighted value is much larger than the weighted value of the conventional FM-OFDM transmission method, and can significantly reduce the quantization error that causes performance degradation in actual implementation.

[0128] Here, if the transmitter performs a phase-shift encoding and real-time phase conversion process on the source bits, it can be robust to various phase distortions (e.g., phase distortions due to carrier frequency offset (CFO), PhN, Doppler shift, Doppler spread, etc.) while minimizing implementation-related quantization errors.

[0129] When applying the modulation method of M-BASK to the real phase-shifted signal z(n), the CP appended unit can add CP to generate a signal z'(n) to which CP has been appended (S406).

[0130] Figure 9 is a conceptual diagram of the CP addition process of the CP addition unit.

[0131] Referring to Fig. 9, the CP addition unit can generate a CP-added signal z'(n) by adding a CP when applying the M-BASK modulation method to the real phase-shifted signal z(n). Here, the CP can be Nc bits, and the real phase-shifted signal can be N bits. Accordingly, the signal with the CP added is N+N C It could be a bit. N C can be a positive real number.

[0132] Again, referring to Figure 4, TRP is a size value A corresponding to the transmission power that enables the desired communication according to the call quality of the TRP and the terminal. c,t After applying the CP to the real phase-shifted signal, digital baseband filtering can be performed by passing it through a digital baseband filter (S407). Here, A c,t can be a real number. Through such digital baseband filtering, the TRP can bandpass filter the CP-added real phase-shifted signal by a desired bandwidth. Afterwards, the TRP can wirelessly transmit the bandpass-filtered CP-added real phase-shifted signal through the antenna. Here, the CP period can be the same as the pilot 2 period embedded in the symbol period. Alternatively, the CP period can be smaller than the pilot 2 period embedded in the symbol period.

[0133] When applying the above-mentioned M-PSK modulation method and M-QAM modulation method, TRP can perform CP addition process and baseband filtering process in the same way as when applying the M-BASK modulation method. In addition, when applying the above-mentioned M-PSK modulation method, TRP can perform the size value A corresponding to the same transmission power for symbols within each symbol block in the same way as when applying the M-BASK modulation method. c,tIn contrast, when TRP applies the M-QAM modulation method, as shown in Figs. 7a and 7b, different size values ​​A are applied to each symbol within each symbol block. c,t can be applied. The signal transmitted wirelessly from the TRP can pass through the wireless fading channel. The terminal can receive the signal that has passed through the wireless fading channel from the TRP.

[0134] Fig. 10 is a flowchart illustrating a first embodiment of a signal design method that is immune to phase distortion performed at a receiving end.

[0135] Referring to FIG. 10, the terminal can perform analog bandpass filtering for a desired bandwidth by applying a bandpass filter to the signal received from the TRP. Furthermore, the terminal can convert the received signal into a baseband signal and perform digital baseband filtering using a digital baseband filter for a desired amount of bandwidth (S1001). Through this process, the terminal can add AWGN (additive white Gaussian noise) to the received signal.

[0136] Afterwards, the terminal can remove the CP to remove inter-symbol interference (ISI) or inter-carrier interference (ICI) from the complex signal (i.e., a signal having both real and complex components) that has passed through the baseband filter through the CP removal unit (S1002). The complex signal r(n) from which the CP has been removed in such a CP removal unit can be expressed as the following mathematical expression 5.

[0137]

[0138] In mathematical expression 5, φ l (n) is a mistake It could be f c can be the carrier frequency, and τ lcan be the delay value of channel path l, and β l (n) can be the phase value of the channel coefficient. A c,r can be a real number with the received power. This received power has a magnitude value A corresponding to the transmit power in TRP. c,t φ may be a size value that reflects path loss, etc. when received at the terminal. d,l (n) may be the Doppler component occurring in channel path l, and φ c (n) may be a CFO component occurring in channel path l, and φ p (n) may be a PhN component occurring in channel path l, and φ m (n) may be a phase distortion component that may occur randomly for n due to other hardware damage occurring in the channel path l. In Equation 5, w(n) may mean the AWGN added to the terminal. In Equation 5, among the phase components, φ d,l (n),φ c (n) and φ p Each of (n) can generally have very high correlations between values ​​in adjacent n φ. m (n) may have random characteristics between values ​​in adjacent n, but the impact of these components on the overall performance may not be that large.

[0139] In this disclosure, TRP can perform on-chip encoding by taking these points into consideration. In other words, to eliminate Doppler, CFO, and phase components at the receiver, the difference between values ​​in adjacent n is calculated, and φ m (n) A transmitting end may perform a forward encoding to average the components. The forward decoding at the receiving end in relation to this may be described in detail below.

[0140] The terminal can effectively estimate a wireless fading channel, convert a time-domain signal into a frequency-domain signal for recovering (or equalizing), and convert the recovered signal back into a time-domain signal by estimating and recovering it in the frequency domain.

[0141] To this end, the terminal may be equipped with a time-frequency conversion unit, a channel estimation unit, a frequency-domain equalization unit, a frequency-time conversion unit, etc. Here, the time-frequency conversion unit may perform a time-frequency conversion process for converting a time-domain signal from which a CP has been removed into a frequency-domain signal. In addition, the channel estimation unit may perform a channel estimation process for estimating a channel from a frequency-domain signal. The frequency-domain equalization unit may perform a frequency-domain equalization process for restoring a wireless channel using a frequency-domain signal and channel estimation. The frequency-time conversion unit may perform a frequency-time conversion process for converting a frequency-domain signal from which a wireless channel has been restored back into a time-domain signal.

[0142] The wireless channel environment experienced by the TRP and the terminal may only have a Line-of-Sight (LoS) path. In this case, the terminal may not require the time-frequency conversion process, channel estimation process, frequency-domain equalization process, and frequency-time conversion process. The wireless channel environment experienced by the TRP and the terminal may also have a Non-Line-of-Sight (NLoS) path. In this case, the terminal may process the received signal through the processing process of the four functional blocks mentioned above to overcome the performance degradation due to the multipath effect as shown in Equation 5. The processing process of these functional blocks may be as follows.

[0143] First, the time-frequency conversion unit can convert a signal from the time domain to the frequency domain (S1003). This time-frequency conversion unit can extract a time-domain reception signal corresponding to a pilot component from a CP-removed reception signal, as in Equation 6.

[0144]

[0145] The time-frequency transform unit performs N-point DFT (discrete Fourier transform) or FFT (fast Fourier transform) on the extracted signal of Equation 6 to obtain a signal R of the pilot component of the frequency domain as in Equation 7. p (k) can be converted to.

[0146]

[0147] W in Equation 7 p (k) can mean noise component, and H p (k) can mean frequency response, Y p (k) is a known time domain pilot signal y expressed by mathematical expression 8 shown in Fig. 7. p (n) may refer to a frequency domain signal corresponding to a pilot component converted into a frequency domain, and may be as shown in mathematical expression 9.

[0148]

[0149]

[0150] In addition, the time-frequency conversion unit can extract a time-domain reception signal corresponding to a symbol component from a reception signal from which CP has been removed, as in mathematical expression 10.

[0151]

[0152] In mathematical expression 10, the time domain reception signal r corresponding to the symbol component s The time domain transmission signal corresponding to (n) is y s(n) can be. The time frequency transform unit performs N-point DFT or FFT on the extracted signal of Equation 10 to obtain a signal R of frequency domain symbol components as in Equation 11. s (k) can be converted to.

[0153]

[0154] W in Equation 11 s (k) can mean noise component, and H s (k) can mean frequency response, Y s (k) is the time domain symbol signal y sent from the transmitter shown in Fig. 3. s (n) can mean a frequency domain signal converted into the frequency domain.

[0155] The channel estimation unit of the terminal can estimate the wireless channel in the frequency domain (S1004). This channel estimation unit is known to both the transmitter and receiver by using the mathematical expression Y of Equation 9. p (k) Using the signal R of the pilot component in the frequency domain of mathematical expression 7 p (k) Frequency response of wireless channel using least square (LS), minimum mean squared error (MMSE), discrete Fourier transform (DFT) or any other possible channel estimation method. can be obtained.

[0156] The frequency domain equalizer of the terminal can perform frequency domain equalization by compensating for the wireless channel in the frequency domain (S1005). The channel estimation unit can calculate the frequency response H reflected in the symbol component shown in Equation 11. s In order to compensate for (k), _ can be estimated through a channel estimation process. The frequency domain equalizer converts the time domain signal into a frequency domain signal as in Equations 7 and 11, thereby obtaining H in Equation 7. p (k) is H in mathematical formula 11 s(k) with a value almost similar to that of, in other words, Considered as such, it is estimated as above. Estimated value of frequency response of mathematical expression 11 It can be assumed that.

[0157] And, the frequency domain equalizer can be equalized in all possible ways for Equation 11. However, the frequency domain equalizer H p (k) is converted to H by performing a cyclic shift and performing a transformation different from Equations 10 and 11. s (k) can be made almost similar to H. In addition, the frequency domain equalizer p Rotate any phase of (k) and perform a transformation different from Equations 10 and 11 to obtain H s (k) can be made almost similar to H. Alternatively, the frequency domain equalizer can be made to H p (k) When converting from the time domain to the frequency domain, a different transformation from Equation 10 to Equation 11 is performed to obtain H s (k) can be made to be almost similar. Like this, Any possible method of doing so may be included within the scope of the present disclosure.

[0158] The frequency-time converter is a signal equalized in the frequency domain. By performing N-point IDFT (inverse discrete Fourier transform) or IFFT (inverse fast Fourier transform) on the time domain signal as in Equation 12 can be converted into (S1006).

[0159]

[0160] Here, φ may represent a phase error component that may occur during frequency-domain equalization. For convenience of explanation, the above description assumes perfect time synchronization; however, this is generally not the case, and the scope of the present disclosure can be applied without loss of generality under such imperfect time synchronization conditions.

[0161] Meanwhile, it can be assumed that the terminal has performed a TRP-based encoding on the symbol block in the signal containing the pilot as in Equation 2. In addition, the terminal can set P=B=2, N1=12, and N=16. Then, the terminal uses the symbol extraction unit to convert the signal into the time domain as in Equation 12. If symbol extraction is performed to extract only symbol components, it can be expressed as in mathematical expression 13 (S1007).

[0162]

[0163] The time phase conversion part of the terminal is the symbol extracted signal of mathematical expression 13. By performing time phase transformation on the signal, the time phase is transformed as in Equation 14. can be output (S1008).

[0164]

[0165] Here, n can be n=2,3,…,13. silver It can be. n' can be 0≤n'≤7, and n" can be 0≤n"≤2.

[0166] The terminal's on-chip decoding unit converts the signal into the phase component of mathematical expression 14. By performing the on-chip decoding in the on-chip decoded signal as in mathematical expression 15 (i.e., the restored M-BASK symbol) can be estimated (S1009).

[0167] Such a rolling decoding may be a method of performing a new differential decoding for every cycle B of a rolling time domain sample unit. In other words, the rolling decoding unit may divide a signal converted into a phase component of Equation 14 into N / B time domain partial symbol blocks. Then, the rolling decoding unit may perform differential decoding on each of the partial symbol blocks. At this time, the rolling decoding unit may perform differential decoding after subtracting the first sample value of the previous first partial symbol block from the first sample of the first partial symbol blocks. Here, the rolling decoding may be accumulation linear decoding.

[0168]

[0169] The M-BASK symbol demapper of the terminal can perform M-BASK demapping on a signal decoded in the stationary decoding unit or on a signal detected by the symbol detection unit (S1010). The terminal can generate a recovered source bit stream through this demapping process.

[0170] Figure 11a is a conceptual diagram of the 2-BASK demapping process performed in a 2-BASK symbol demapper.

[0171] Referring to Figure 11a, the 2-BASK symbol demapper receives the signal If this decision bound is greater than or equal to 0, it can be determined as bit 1. In contrast, the 2-BASK symbol demapper determines the received signal If this judgment boundary is less than 0, it can be judged as 0.

[0172] Figure 11b is a conceptual diagram of the 4-BASK demapping process performed in a 4-BASK symbol demapper.

[0173] Referring to Figure 11b, the 4-BASK symbol demapper receives the signal If this decision boundary is greater than or equal to 1.5, it can be decided as bit 11. In addition, the 4-BASK symbol demapper If this decision boundary is greater than or equal to 0 and less than 1.5, it can be decided as bit 10. In addition, the 4-BASK symbol demapper If this judgment boundary is greater than or equal to -1.5 and less than 0, it can be judged as bit 00. The 4-BASK symbol demapper receives the signal If this judgment boundary is less than -1.5, it can be judged as 01.

[0174] Figure 11c is a conceptual diagram illustrating the 8-BASK demapping process performed in an 8-BASK symbol demapper.

[0175] Referring to Figure 11c, the 8-BASK symbol demapper receives the signal If it is greater than or equal to the decision boundary of 3.5, it can be determined as bit 111, if it is greater than or equal to the decision boundary of 2.5 and less than 3.5, it can be determined as bit 110, if it is greater than or equal to the decision boundary of 1.5 and less than 2.5, it can be determined as bit 101, if it is greater than or equal to the decision boundary of 0 and less than 1.5, it can be determined as bit 100, if it is greater than or equal to the decision boundary of -1.5 and less than 0, it can be determined as bit 000, if it is greater than or equal to the decision boundary of -2.5 and less than -1.5, it can be determined as bit 001, if it is greater than or equal to the decision boundary of -3.5 and less than -2.5, it can be determined as bit 010, and if it is less than the decision boundary of -3.5, it can be determined as bit 011. Through the same process as above, the terminal can perform the M-BASK demapping process of M>4.

[0176] Figure 12 is a conceptual diagram showing the 8-PSK demapping process performed in an 8-PSK symbol demapper.

[0177] Referring to Figure 12, the 8-PSK symbol demapper receives the signal The symbol can be demapped by using the phase value between the signal constellation points as the decision boundary.

[0178] Figure 13 is a conceptual diagram showing the 16-QAM demapping process performed by a 16-QAM symbol demapper.

[0179] Referring to Figure 13, the 16-QAM symbol demapper receives the signal The symbols can be demapped by using the straight lines on the real and imaginary axes between the signal constellation points as the decision boundaries.

[0180] In the above symbol extraction process, time phase conversion process, and floating-point decoding process, if it is assumed that the terminal has performed floating-point encoding on the symbol block including the pilot of Equation 1, the symbol extraction may not be performed first. Instead, the terminal may perform time phase conversion on the frequency-time converted signal and use it as input to the floating-point decoding unit. Then, the floating-point decoding unit may perform floating-point decoding mentioned in Equation 15 on the time-phase converted signal and then perform a symbol extraction process that takes only the symbol components excluding the pilot components. Thereafter, the terminal may perform M-BASK symbol demapping on the extracted symbols.

[0181] Meanwhile, the symbol detection unit can obtain the interference component of Equation 14 by subtracting the estimated M-BASK symbol decoded temporally from the output signal of the frequency-time conversion unit. At this time, the symbol detection unit can also obtain the interference component of Equation 14 by processing the estimated M-BASK symbol decoded temporally and subtracting it from the output signal of the frequency-time conversion unit. The terminal can subtract the interference component calculated in this way from the output signal of the frequency-time conversion unit again to remove the interference, and then perform the symbol extraction process, time phase conversion process, and temporal decoding process mentioned above once or repeatedly.

[0182] In addition, even if the positions of the symbol extraction unit and the time phase conversion unit are swapped, the consistency of the technology proposed in the present disclosure can be maintained. In other words, the time phase conversion unit can process the signal first, and then the symbol extraction unit can extract the symbol components excluding the pilot component. In the PR-SC transmission method based on the overlay encoding described with reference to the above-mentioned Fig. 4, the transmitter can apply the M-PSK or M-QAM modulation method other than the M-BASK modulation method. Accordingly, the receiver can apply the M-PSK or M-QAM demapping process other than the M-BASK demapping process.

[0183] Fig. 14 is a flowchart illustrating a second embodiment of a signal design method that is immune to phase distortion performed at a transmitter.

[0184] Referring to Fig. 14, the TRP can generate a source bit stream to be transmitted to the terminal (S1401). Accordingly, the M-BASK symbol mapper of the TRP can perform M-BASK modulation on the generated source bit stream to generate a source symbol stream or source symbol block composed of elements of n=0,1,…,N1 having only integer components (S1402). Here, M and N1 can be positive integers.

[0185] Here, the integer signal constellation M-BASK mapping rules of the above bit / bits may be an example, and mapping rules of all other possible modulations may be included in the scope of the present disclosure. For example, another modulation mapping rule may be an M-PSK modulation mapping rule. An M-PSK mapper may perform M-PSK modulation on the generated source bit string to generate a source symbol string or source symbol block composed of elements of n=0,1,…,N1 having only a phase component. Another modulation mapping rule may be an M-QAM modulation mapping rule.

[0186] The M-QAM mapper can perform M-QAM modulation on the generated source bitstream to generate a source symbol stream or source symbol block composed of elements of n=0,1,…,N1 that only have phase components. Other modulation mapping rules, such as GMSK and DPSK, may also be included in the scope of the present disclosure. In addition, the receiver can perform demodulation and restoration by taking into account the characteristics of the modulation methods mentioned above.

[0187] The pilot insertion unit can insert a pilot into the middle of the source symbol block of the signal output from the M-BASK mapper (S1403). The pilot inserted in this way can easily and effectively restore or equalize a nonlinear wireless channel within each symbol of the symbol block at the receiver.

[0188] Figure 15 is a conceptual diagram showing a second embodiment of a pilot insertion process performed in a pilot insertion unit.

[0189] Referring to FIG. 15, the pilot insertion unit can insert a pilot into the middle of the source symbol blocks generated by the M-BASK mapper. In this way, the pilot insertion unit can insert a pilot into the middle of the source symbol blocks of N1 bits generated by the M-BASK mapper. Here, the pilot can be P bits. Here, P can be a positive integer. Accordingly, the output x(n) of the pilot insertion unit can be composed of N-1 bits. Here, N can be a positive integer. And, N can be P+2N1. N can be 0, 1,…, N-1.

[0190] The pilot insertion unit can assign zero power values ​​to the first and last time-domain samples of the pilot to minimize interference caused by multipath when performing wireless channel estimation and frequency-domain equalization at the receiver. N1 can represent the length of a time-domain sample unit of a symbol block, and P can represent the length of a time-domain sample unit of the pilot. The pilot length can be set by considering the length of the symbol block and frequency efficiency, etc.

[0191] Again, referring to FIG. 14, the floating-point encoding unit can perform floating-point encoding on the M-BASK modulated symbol block signal with the pilot inserted (S1404). At this time, the output q(n) of the floating-point encoding unit can be modeled as in Equation 16 or Equation 17 below. Here, in the case of applying the M-PSK modulation method mentioned above, the floating-point encoding unit can take only the value excluding π of the phase component for each constellation point and apply it to x(·) modeled in Equation 16 or Equation 17. In addition, in the case of applying the M-QAM modulation method mentioned above, the floating-point encoding unit can similarly take only the value excluding π of the phase component for each constellation point and apply the obtained value to x(·) modeled by Equation 16 or Equation 17.

[0192]

[0193] In mathematical expression 16, n is a positive real number, and n=Bn'+n"=0,1,…,N-1. Here, n' is a positive real number. can be, and n" is a positive real number 0≤n" <B일 수 있다.

[0194]

[0195] In mathematical expression 17, n is a positive real number, and can be n=Bn'+n"=0,1,…,N-1. Here, n' is a positive real number. can be, and n" is a positive real number 0≤n" <B일 수 있다.

[0196] In equations 16 and 17, min(1,n') can be a function that compares the sizes of 1 and n' and selects the smaller one. can be a function that takes the integer part of y plus 1 when y is a positive real number with a decimal point, and takes the integer part of y as it is when y does not have a decimal point. B can be a cyclic period of the time domain sample unit. Here, B can be a positive real number.

[0197] In one embodiment, the floating-point encoding unit can perform floating-point encoding including the pilot as in Equation 16. Alternatively, in another embodiment, the floating-point encoding unit can perform floating-point encoding only on the source symbol blocks while leaving the pilot as in Equation 17. In this case, it can be efficient to constrain P to be a multiple of B (e.g., P=B, 2B, 3B,…). Of course, P may not be a multiple of B. For the convenience of all descriptions of the technology of the present disclosure, P may be set to P=2B.

[0198] In addition, when the on-chip encoding unit performs on-chip encoding without including a pilot as in Equation 17, the first sample to be on-chip encoded may be susceptible to hardware damage, so the last sample(s) of the preceding pilot known to both the transmitter and receiver may be taken as the first sample.

[0199] Such a rolling encoding may be a method of performing a new differential encoding for every cycle B of a rolling time domain sample unit. In other words, the rolling encoding unit may divide the M-BASK modulated symbol block with the pilot inserted into N / B time domains into first partial symbol blocks according to Equation 16. Then, the rolling encoding unit may perform differential encoding on each of the first partial symbol blocks. At this time, the rolling encoding unit may perform differential encoding after adding the first sample value of the previous first partial symbol block to the first sample of the first partial symbol blocks.

[0200] In contrast, the floating-point encoding unit can divide the source symbol block into N1 / B time domain second partial symbol blocks in the M-BASK modulated symbol block with the pilot inserted according to Equation 17. Then, the floating-point encoding unit can perform differential encoding on each of the second partial symbol blocks. At this time, the floating-point encoding unit can perform differential encoding after adding the first sample value of the previous second partial symbol block to the first sample of the second partial symbol blocks. Here, the floating-point encoding can be cumulative linear encoding.

[0201] Meanwhile, the output q(n) of the phase encoding unit can be modeled as in the following mathematical expression 18. Here, in the case of applying the M-PSK modulation method mentioned above, the phase encoding unit can take only the value excluding π of the phase component for each constellation point and apply it to x(·) modeled in mathematical expression 18. In addition, in the case of applying the M-QAM modulation method mentioned above, the phase encoding unit can similarly take only the value excluding π of the phase component for each constellation point and apply the obtained value to x(·) modeled by mathematical expression 18.

[0202]

[0203] In mathematical expression 18, n is a positive real number, and can be n=Bn'+n"=0,1,…,N-1. Here, n' is a positive real number. can be, and n" is a positive real number, 0≤n" <B일 수 있다. 수학식 18에서 min(1,n')는 1과 n'의 크기를 비교하여 크기가 작은 것을 택하는 함수일 수 있다. can be a function that takes the integer part of y plus 1 when y is a positive real number with a decimal point, and takes the integer part of y as it is when y does not have a decimal point. B can be a cyclic period of a time-domain sample unit in the shape of a gear. Here, B can be a positive real number.

[0204] In another embodiment, the floating-point encoding unit can perform floating-point encoding only on symbol blocks while leaving the pilots as is, as in Equation 18. In this case, it can be efficient to set a constraint that p is a multiple of B (e.g., P=B,2B,3B,…). Of course, P may not be a multiple of B. For the convenience of all descriptions of the technology of the present disclosure, P may be set to P=2B.

[0205] In contrast, the floating-point encoding unit can divide the source symbol block into N1 / B time domain second partial symbol blocks in the M-BASK modulated symbol block with the pilot inserted according to Equation 18. Then, the floating-point encoding unit can perform differential encoding on each of the second partial symbol blocks. At this time, the floating-point encoding unit can perform differential encoding after adding the first sample value of the previous second partial symbol block to the first sample of the second partial symbol blocks. Here, the floating-point encoding can be cumulative linear encoding. Equations 17 and 18 can have differences in the starting values ​​when performing the second differential encoding.

[0206] The real-to-phase conversion unit can convert the signal encoded in the real-phase into a phase value z(n) as in Equation 19 or Equation 20 through a real-to-phase converting process (S1405). In the case of applying the above-mentioned M-PSK and M-QAM modulation methods, the real-to-phase conversion unit can follow the real-phase conversion process when applying the M-BASK modulation method.

[0207]

[0208] Here, n can be 0,1,…,N-1. Eun Yang's mistake It may be, It can be. θ is a positive real number. It can be. α is a positive real number that can converge to 0.

[0209]

[0210] Here, Eun Yang's mistake It may be, It can be. θ is a positive real number. It can be. α is a positive real number that can converge to 0.

[0211] In Equations 19 and 20, the maximum value of q(n) is A due to the gear-shaped encoding. M (B+1) can be A, the maximum value of q(n). M The inverse of (B+1) can be . q(n) of time-domain sample n is weighted by θ. This could be it. For example, θ exists in (-1,1) or [-1,1). If it is possible to do so, and if the positive α converges to 0, the ambiguity occurring in the demodulation process at the receiver can be eliminated. The above weighted value is much larger than the weighted value of the conventional FM-OFDM transmission method, and can significantly reduce the quantization error that causes performance degradation in actual implementation.

[0212] Here, if the transmitter performs a phase-shift encoding and real-phase conversion process on the source bits, it can be robust to various phase distortions (e.g., phase distortions due to CFO, PhN, Doppler shift, Doppler spread, etc.) while minimizing implementation-side quantization errors.

[0213] When applying the modulation method of M-BASK to the real phase-shifted signal z(n), the CP appended unit can generate a signal z'(n) with CP appended (S1406). TRP is a size value A corresponding to the transmission power that enables the desired communication according to the call quality of the TRP and the terminal. c,t After applying the CP to the real phase-shifted signal, digital baseband filtering can be performed by passing it through a digital baseband filter (S1407). Here, A c,t can be a mistake. Through such digital baseband filtering, the TRP can bandpass filter the CP-added real phase-shifted signal by a desired bandwidth. Afterwards, the TRP can wirelessly transmit the bandpass-filtered CP-added real phase-shifted signal through the antenna. The signal wirelessly transmitted from the TRP can pass through the wireless fading channel. The terminal can receive the signal that has passed through the wireless fading channel from the TRP.

[0214] Fig. 16 is a flowchart illustrating a second embodiment of a signal design method that is immune to phase distortion performed at a receiving end.

[0215] Referring to Figure 16, the terminal can perform analog bandpass filtering for a desired bandwidth by applying a bandpass filter to the signal received from the TRP. Furthermore, the terminal can convert the received signal into a baseband signal and perform digital baseband filtering using a digital baseband filter for a desired amount of bandwidth (S1601). Through this process, the terminal can add AWGN to the received signal.

[0216] Afterwards, the terminal can remove the CP to remove ISI or ICI from the complex signal (i.e., a signal having both real and complex components) that has passed through the baseband filter through the CP removal unit (S1602). The complex signal r(n) from which the CP has been removed in such a CP removal unit can be expressed as in mathematical expression 5.

[0217] In mathematical expression 5, φ l (n) is a mistake _ may be f c can be the carrier frequency, and τ l can be the delay value of channel path l, and β l (n) can be the phase value of the channel coefficient. A c,r can be a real number with the received power. This received power has a magnitude value A corresponding to the transmit power in TRP. c,t φ may be a size value that reflects path loss, etc. when received at the terminal. d,l (n) may be the Doppler component occurring in channel path l, and φ c (n) may be a CFO component occurring in channel path l, and φ p (n) may be a PhN component occurring in channel path l, and φ m(n) may be a phase distortion component that may occur randomly for n due to other hardware damage occurring in the channel path l. In Equation 5, w(n) may mean the AWGN added to the terminal. In Equation 5, among the phase components, φ d,l (n), φ c (n) and φ p Each of (n) can generally have very high correlations between values ​​in adjacent n φ. m (n) may have random characteristics between values ​​in adjacent n, but the impact of these components on the overall performance may not be that large.

[0218] In this disclosure, TRP can perform on-chip encoding by taking these points into consideration. In other words, to eliminate Doppler, CFO, and phase components at the receiver, the difference between values ​​in adjacent n is calculated, and φ m (n) A transmitting end may perform a forward encoding to average the components. The forward decoding at the receiving end in relation to this may be described in detail below.

[0219] The terminal can effectively estimate a wireless fading channel, convert a time-domain signal into a frequency-domain signal for recovery (or equalization), estimate and recover in the frequency domain, and then convert the recovered signal back into a time-domain signal.

[0220] To this end, the terminal may be equipped with a time-frequency conversion unit, a channel estimation unit, a frequency-domain equalization unit, a frequency-time conversion unit, etc. Here, the time-frequency conversion unit may perform a time-frequency conversion process for converting a time-domain signal from which a CP has been removed into a frequency-domain signal. In addition, the channel estimation unit may perform a channel estimation process for estimating a channel from a frequency-domain signal. The frequency-domain equalization unit may perform a frequency-domain equalization process for restoring a wireless channel using the frequency-domain signal and channel estimation. The frequency-time conversion unit may perform a frequency-time conversion process for converting the channel-restored frequency-domain signal back into a time-domain signal. The wireless channel environment experienced by the TRP and the terminal may only have an LoS path. Then, the time-frequency conversion process, the channel estimation process, the frequency-domain equalization process, and the frequency-time conversion process may not be required in the terminal.

[0221] The wireless channel environment experienced by the TRP and the terminal may also include NLoS paths. Therefore, to overcome the performance degradation caused by the multipath effect described in Equation 5, the terminal can process the received signal through the four functional blocks mentioned above. The processing steps of these functional blocks can be as follows.

[0222] First, the time-frequency conversion unit can convert a signal from the time domain to the frequency domain (S1603). This time-frequency conversion unit can extract a time-domain reception signal corresponding to a pilot component from a CP-removed reception signal, as in Equation 21.

[0223]

[0224] The time-frequency transform unit performs N-point DFT or FFT on the extracted signal of Equation 21 to obtain a signal R of the pilot component of the frequency domain as in Equation 22. p (k) can be converted to.

[0225]

[0226] W in equation 22 p (k) can mean noise component, and H p (k) can mean frequency response, Y p (k) is a known time domain pilot signal y expressed by mathematical expression 23 shown in Fig. 15. p (n) may mean a frequency domain pilot signal corresponding to a pilot component converted into a frequency domain, and may be as shown in mathematical expression 24.

[0227]

[0228]

[0229] In addition, the time-frequency conversion unit can extract a time-domain reception signal corresponding to a symbol component from a reception signal from which CP has been removed, as in mathematical expression 25.

[0230]

[0231] In mathematical expression 25, the time domain reception signal r corresponding to the symbol component s The time domain transmission signal corresponding to (n) is y s (n) can be. The time frequency transform unit performs N-point DFT or FFT on the extracted signal of Equation 25 to obtain a signal R of frequency domain symbol components as in Equation 26. s (k) can be converted to.

[0232]

[0233] W in Equation 26 s (k) can mean noise component, and H s (k) can mean frequency response, Y s (k) is the time domain symbol signal y sent from the transmitter. s (n) can mean a frequency domain signal converted into the frequency domain.

[0234] The channel estimation unit of the terminal can estimate the wireless channel in the frequency domain (S1604). This channel estimation unit is known to both the transmitter and receiver by using the mathematical expression Y of Equation 24. p (k) Using the signal R of the pilot component in the frequency domain of mathematical expression 22 p (k) Frequency response of the wireless channel using LS, MMSE, DFT or any other possible channel estimation method. can be obtained.

[0235] The frequency domain equalizer of the terminal can perform frequency domain equalization by compensating for the wireless channel in the frequency domain (S1605). The channel estimation unit can calculate the frequency response H reflected in the symbol component shown in Equation 26. s To compensate for (k), we go through a channel estimation process. can be estimated. The frequency domain equalizer converts the time domain signal into a frequency domain signal as in Equations 22 and 26, thereby obtaining H in Equation 22. p (k) is H in mathematical expression 26 s (k) with a value almost similar to that of, in other words, Considered as such, it is estimated as above. Estimated value of frequency response of mathematical expression 26 It can be assumed that the frequency domain equalizer is can be equalized in all possible ways for Equation 26. However, the frequency domain equalizer H p (k) is converted to H by performing a cyclic shift and performing a transformation different from Equations 25 and 26. s (k) can be made almost similar to H. In addition, the frequency domain equalizer p Rotate any phase of (k) and perform a transformation different from Equations 25 and 26 to obtain H s(k) can be made almost similar to H. Alternatively, the frequency domain equalizer can be made to H p (k) When converting from the time domain to the frequency domain, a different transformation from Equations 25 and 26 is performed to obtain H s (k) can be made to be almost similar. Like this, Any possible method of doing so may be included within the scope of the present disclosure.

[0236] The frequency-time converter is a signal equalized in the frequency domain. Perform N-point IDFT or IFFT on the time domain signal as in Equation 27. can be converted to (S1606).

[0237]

[0238] Here, φ may represent a phase error component that may occur during frequency-domain equalization. For convenience of explanation, the above description assumes perfect time synchronization; however, this is generally not the case, and the scope of the present disclosure can be applied without loss of generality under such imperfect time synchronization conditions.

[0239] Meanwhile, it can be assumed that the terminal has performed a TRP-based encoding on the symbol block in the signal containing the pilot as in Equations 17 and 18. In addition, the terminal can set B=2, P=2B, N1=6, and N=16. Then, the terminal uses the symbol extraction unit to convert the signal into the time domain as in Equation 27. If symbol extraction is performed to extract only symbol components, it can be expressed as in mathematical expression 28 (S1607).

[0240]

[0241] The time phase conversion part of the terminal is the symbol extracted signal of mathematical expression 28. By performing time phase transformation on the signal, the time phase is transformed as in Equation 29. can be output (S1608). Alternatively, the time phase conversion unit of the terminal can output the symbol extracted signal of mathematical expression 28. By performing time phase transformation on the signal, the time phase is transformed as in mathematical expression 30. can be printed.

[0242]

[0243]

[0244] Here, n can be n=2n'+n", and n=0,1,…,5. silver It can be. The terminal's on-board decoding unit converts the signal into the phase component of mathematical expression 29. By performing the on-chip decoding in the above, the on-chip decoded signal is obtained as in Equation 31. (i.e., the restored M-BASK symbol) can be estimated (S1609). Alternatively, the terminal's on-chip decoding unit can estimate the signal converted into the phase component of mathematical expression 30. By performing the on-chip decoding in the above, the on-chip decoded signal is obtained as in mathematical expression 32. (i.e., the restored M-BASK symbol) can be estimated.

[0245]

[0246]

[0247] The M-BASK symbol demapper of the terminal can perform M-BASK demapping on a signal decoded in the stationary decoding unit or on a signal detected by the symbol detection unit (S1610). The terminal can generate a recovered bit string through this demapping process.

[0248] In the above symbol extraction process, time phase conversion process, and floating-point decoding process, if it is assumed that the terminal has performed floating-point encoding on the symbol block including the pilot of Equation 16, the symbol extraction may not be performed first. Instead, the terminal may perform time phase conversion on the frequency-time converted signal and use it as input to the floating-point decoding unit. Then, the floating-point decoding unit may perform floating-point decoding mentioned in Equation 29 on the time-phase converted signal and then perform a symbol extraction process that takes only the symbol components excluding the pilot components. Thereafter, the terminal may perform M-BASK symbol demapping on the extracted symbols.

[0249] The symbol detection unit can obtain the interference components of Equations 29 and 30 by subtracting the estimated M-BASK symbol decoded temporally from the output signal of the frequency-time conversion unit. At this time, the symbol detection unit can also obtain the interference components of Equations 29 and 30 by processing the estimated M-BASK symbol decoded temporally and subtracting it from the output signal of the frequency-time conversion unit. The terminal can subtract the interference component calculated in this way from the output signal of the frequency-time conversion unit again to remove the interference, and then perform the symbol extraction process, time phase conversion process, and temporal decoding process mentioned above once or repeatedly.

[0250] In addition, even if the positions of the symbol extraction unit and the time phase conversion unit are swapped, the consistency of the technology proposed in the present disclosure can be maintained. In other words, the time phase conversion unit can process the signal first, and then the symbol extraction unit can extract the symbol components excluding the pilot component. In the PR-SC transmission method based on the overlay encoding described with reference to the above-mentioned Fig. 14, the transmitter can apply the M-PSK or M-QAM modulation method other than the M-BASK modulation method. Accordingly, the receiver can apply the M-PSK or M-QAM demapping process other than the M-BASK demapping process.

[0251] Fig. 17 is a block diagram showing a second embodiment of a signal design device immune to phase distortion.

[0252] Referring to FIG. 17, the transmitter may include an M-BASK symbol mapper (1701), a pilot insertion unit (1702), a differential encoder (1703), a real phase converter (1704), a CP addition unit (1705), and / or a baseband filter (1706). The receiver may include a baseband filter (1711), a CP removal unit (1712), a time-frequency converter (1713), a frequency-domain equalizer (1715), a channel estimation unit (1715), a frequency-time converter (1716), a symbol extractor (1717), a time-phase converter (1718), a differential decoder (1719), a symbol detector (1720), and / or an M-BASK symbol demapper (1721).

[0253] The transmitter may be a TRP, and the receiver may be a terminal. The transmitter may use a differential encoding-based phase distortion-resistant single carrier (PR-SC) transmission scheme. Here, the differential encoding-based PR-SC transmission scheme can have high phase distortion resistance and a zero phase-to-noise ratio (PAPR).

[0254] Fig. 18 is a flowchart illustrating a third embodiment of a signal design method that is immune to phase distortion performed at a transmitter.

[0255] Referring to FIG. 18, the TRP can generate a source bit stream to be transmitted to the terminal (S1801). Accordingly, the M-BASK symbol mapper of the TRP can perform M-BASK modulation on the generated source bit stream to generate a source symbol stream or source symbol block composed of elements of n=0,1,…,N1 having only integer components (S1802). Here, M and N1 can be positive integers. Here, the above-described bit / bit integer signal constellation M-BASK mapping rules can be one example, and mapping rules of all other possible modulations can be included in the scope of the present disclosure. As an example, another modulation mapping rule can be an M-PSK modulation mapping rule. The M-PSK mapper can perform M-PSK modulation on the generated source bit stream to generate a symbol stream or symbol block composed of elements of n=0,1,…,N1 having only phase components. Another modulation mapping rule can be an M-QAM modulation mapping rule. The M-QAM mapper can generate a symbol sequence or symbol block consisting of elements of n=0,1,…,N1 that have only phase components by performing M-QAM modulation on the generated source bit sequence.

[0256] The pilot insertion unit can insert pilots (e.g., pilot 1 and pilot 2) before and after symbol blocks of signals output from the M-BASK mapper (S1803). These inserted pilots can easily and effectively restore or equalize nonlinear wireless channels within each symbol of the symbol block at the receiver.

[0257] The pilot insertion unit may assign zeros to the last time-domain samples of Pilot 1 and the first time-domain samples of Pilot 2 to minimize interference caused by multipath when performing wireless channel estimation and frequency-domain equalization at the receiver. N1 may denote the length of a time-domain sample unit of a symbol block, and P may denote the length of a time-domain sample unit of Pilot 1 and Pilot 2. Pilot 1 and Pilot 2 may have the same length, but may not be limited thereto. The pilot length may be set in consideration of the length of the symbol block, frequency efficiency, etc. In addition, the pilot insertion unit may configure only Pilot 1 for reasons such as improving frequency efficiency. Alternatively, the pilot insertion unit may configure only Pilot 2 for reasons such as improving frequency efficiency. The pilot insertion process as described above may be performed after performing a floating-point encoding process.

[0258] The differential encoding unit can perform differential encoding on an M-BASK modulated symbol block signal with a pilot inserted (S1804). At this time, the output q(n) of the differential encoding unit can be modeled as in the following mathematical expression 33 or mathematical expression 34.

[0259]

[0260] In mathematical expression 33, n is a positive real number, and n=0,1,…,N-1.

[0261]

[0262] In Equation 34, n can be a positive real number. Here, n' can be a positive real number. The differential encoding unit may perform differential encoding including a pilot as in Equation 33 in one embodiment. Alternatively, the differential encoding unit may perform differential encoding only on the symbol block while leaving the pilot as in Equation 43 in another embodiment. In addition, when the differential encoding unit performs differential encoding without including a pilot as in Equation 34, the first sample to be differentially encoded may be less susceptible to hardware damage, so the last sample(s) of the preceding pilot 1 known to both the transmitter and receiver may be used as the first sample. As described above, the differential encoding unit may not perform differential encoding on all samples N1 of the corresponding symbol block, and may perform differential encoding by resetting to a unit N1 smaller than N1 (for example, N1=4N2) in order to reduce the quantization error mentioned above. However, if the differential encoding unit performs differential encoding in this way, there may be a burden of having to allocate known symbols, as each symbol that is reset is not differentially encoded. To reduce this burden and improve channel estimation performance, the differential encoding unit can allocate known symbols as pilot symbols, and these distributed pilot symbols can also be utilized.

[0263] The real phase conversion unit can convert the differentially encoded signal into a phase value z(n) as in Equation 35 or Equation 36 through a real phase conversion process (S1805). In the case of applying the above-mentioned M-PSK and M-QAM modulation methods, the real phase conversion unit can follow the real phase conversion process when applying the M-BASK modulation method.

[0264]

[0265] Here, n can be 0,1,…,N-1. Eun Yang's mistake It may be, _ can be.θ can be a positive real number.

[0266]

[0267] Here, Eun Yang's mistake It may be, It can be. θ can be a positive real number. If the bits exist in (-1,1) or [-1,1), the ambiguity that occurs during the demodulation process at the receiver can be eliminated. Here, if the transmitter performs differential encoding and real-valued phase transformation on the source bits, it can be robust to various phase distortions (e.g., phase distortions due to CFO, PhN, Doppler shift, Doppler spread, etc.) while minimizing implementation-related quantization errors.

[0268] The CP appender can generate a CP-added signal z'(n) by adding CP when applying the M-BASK modulation method to the real-phase-shifted signal z(n) (S1806). TRP is a size value A corresponding to the transmission power that enables the desired communication according to the call quality of the TRP and the terminal. c,t After applying the CP to the real phase-shifted signal, digital baseband filtering can be performed by passing it through a digital baseband filter (S1807). Here, A c,t may be a real number. Through such digital baseband filtering, the TRP can bandpass filter the CP-added real phase-shifted signal by a desired bandwidth. Afterwards, the TRP can wirelessly transmit the bandpass-filtered CP-added real phase-shifted signal through the antenna. Here, the CP period may be the same as the pilot 2 period embedded in the symbol period. Alternatively, the CP period may be smaller than the pilot 2 period embedded in the symbol period. The signal wirelessly transmitted from the TRP can pass through a wireless fading channel. The terminal can receive the signal that has passed through the wireless fading channel from the TRP.

[0269] Fig. 19 is a flowchart illustrating a third embodiment of a signal design method that is immune to phase distortion performed at a receiving end.

[0270] Referring to Figure 19, the terminal can perform analog bandpass filtering for a desired bandwidth by applying a bandpass filter to the signal received from the TRP. Furthermore, the terminal can convert the received signal into a baseband signal and perform digital baseband filtering using a digital baseband filter for a desired amount of bandwidth (S1901). Through this process, the terminal can add AWGN to the received signal.

[0271] Afterwards, the terminal can remove the CP to remove ISI or ICI from the complex signal (i.e., a signal having both real and complex components) that has passed through the baseband filter via the CP removal unit (S1902). The complex signal r(n) from which the CP has been removed by the CP removal unit can be expressed as the following mathematical expression 37.

[0272]

[0273] In equation 37, φ l (n) is a mistake It could be f c can be the carrier frequency, and τ l can be the delay value of channel path l, and β l (n) can be the phase value of the channel coefficient. A c,r can be a real number with the received power. This received power has a magnitude value A corresponding to the transmit power in TRP. c,t φ may be a size value that reflects path loss, etc. when received at the terminal. d,l (n) may be the Doppler component occurring in channel path l, and φ c (n) may be a CFO component occurring in channel path l, and φ p(n) may be a PhN component occurring in channel path l, and φ m (n) may be a phase distortion component that may occur randomly for n due to other hardware damage occurring in channel path l. In Equation 37, w(n) may represent the AWGN added to the terminal. In Equation 37, among the phase components, φ d,l (n), φ c (n) and φ p Each of (n) can generally have very high correlations between values ​​in adjacent n φ. m (n) may have random characteristics between values ​​in adjacent n, but the impact of these components on the overall performance may not be that large.

[0274] In this disclosure, TRP can perform differential encoding by taking these points into consideration. In other words, to eliminate Doppler, CFO, and phase components at the receiver, the difference between values ​​in adjacent n is calculated, and φ m Differential encoding can be performed at the transmitter to average the (n) components. Differential decoding at the receiver in relation to this can be described in detail below.

[0275] The terminal can effectively estimate a wireless fading channel, convert a time-domain signal into a frequency-domain signal for recovery (or equalization), estimate and recover in the frequency domain, and then convert the recovered signal back into a time-domain signal.

[0276] To this end, the terminal may be equipped with a time-frequency conversion unit, a channel estimation unit, a frequency-domain equalization unit, a frequency-time conversion unit, etc. Here, the time-frequency conversion unit may perform a time-frequency conversion process for converting a time-domain signal from which a CP has been removed into a frequency-domain signal. In addition, the channel estimation unit may perform a channel estimation process for estimating a channel from a frequency-domain signal. The frequency-domain equalization unit may perform a frequency-domain equalization process for restoring a wireless channel using the frequency-domain signal and channel estimation. The frequency-time conversion unit may perform a frequency-time conversion process for converting the channel-restored frequency-domain signal back into a time-domain signal. The wireless channel environment experienced by the TRP and the terminal may only have an LoS path. Then, the time-frequency conversion process, the channel estimation process, the frequency-domain equalization process, and the frequency-time conversion process may not be required in the terminal.

[0277] The wireless channel environment experienced by the TRP and the terminal may also include NLoS paths. Therefore, to overcome the performance degradation caused by the multipath effect described in Equation 37, the terminal can process the received signal through the processing steps of the four functional blocks mentioned above. The processing steps of these functional blocks can be as follows.

[0278] First, the time-frequency conversion unit can convert a signal from the time domain to the frequency domain (S1903). This time-frequency conversion unit can extract a time-domain reception signal corresponding to the pilot component, as in Equation 38, from the signal from which the CP has been removed in Equation 37.

[0279]

[0280] The time-frequency transform unit performs N-point DFT or FFT on the extracted signal of Equation 38 to obtain a signal R of the pilot component of the frequency domain as in Equation 39. p(k) can be converted to.

[0281]

[0282] W in equation 39 p (k) can mean noise component, and H p (k) can mean frequency response, Y p (k) is a known time domain pilot signal y expressed by mathematical expression 40. p (k) may refer to a frequency domain signal corresponding to a pilot component converted into a frequency domain, and may be as shown in mathematical expression 41.

[0283]

[0284]

[0285] In addition, the time-frequency conversion unit can extract a time-domain reception signal corresponding to a symbol component from a reception signal from which CP has been removed, as in mathematical expression 42.

[0286]

[0287] In mathematical expression 42, the time domain reception signal r corresponding to the symbol component s The time domain transmission signal corresponding to (n) is y s (n) can be. The time frequency transform unit performs N-point DFT or FFT on the extracted signal of Equation 42 to obtain a signal R of frequency domain symbol components as in Equation 43. s (k) can be converted to.

[0288]

[0289] W in equation 43 s (k) can mean noise component, and H s (k) can mean frequency response, Y s (k) is the time domain symbol signal y sent from the transmitter. s (n) can mean a frequency domain signal converted into the frequency domain.

[0290] The channel estimation unit of the terminal can estimate the wireless channel in the frequency domain (S1904). This channel estimation unit is known to both the transmitter and receiver by using the mathematical expression Y of Equation 41. p (k) Using the signal R of the pilot component in the frequency domain of mathematical expression 39 p (k) Frequency response of the wireless channel using LS, MMSE, DFT or any other possible channel estimation method. can be obtained.

[0291] The frequency domain equalizer of the terminal can perform frequency domain equalization by compensating for the wireless channel in the frequency domain (S1905). The channel estimation unit can calculate the frequency response H reflected in the symbol component shown in Equation 43. s To compensate for (k), we go through a channel estimation process. can be estimated. The frequency domain equalizer converts the time domain signal into a frequency domain signal as in Equations 39 and 43, thereby obtaining H in Equation 43. p (k) is H in mathematical expression 43 s (k) with a value almost similar to that of, in other words, Considered as such, it is estimated as above. Estimated value of frequency response of mathematical expression 43 It can be assumed that the frequency domain equalizer is can be equalized in all possible ways for Equation 43. However, the frequency domain equalizer H p (k) is transformed by performing a cyclic shift on H, which is different from equations 42 and 43. s (k) can be made almost similar to H. In addition, the frequency domain equalizer p Rotate any phase of (k) and perform a transformation different from Equations 42 and 43 to obtain H s (k) can be made almost similar to H. Alternatively, the frequency domain equalizer can be made to Hp (k) When converting from the time domain to the frequency domain, a different transformation from Equations 42 and 43 is performed to obtain H s (k) can be made to be almost similar. Like this, Any possible method of doing so may be included within the scope of the present disclosure.

[0292] The frequency-time converter is a signal equalized in the frequency domain. Perform N-point IDFT or IFFT on the time domain signal as in Equation 44. can be converted to (S1906).

[0293]

[0294] Here, φ may represent a phase error component that may occur during frequency-domain equalization. For convenience of explanation, the above description assumes perfect time synchronization; however, this is generally not the case, and the scope of the present disclosure can be applied without loss of generality under such imperfect time synchronization conditions.

[0295] Meanwhile, it can be assumed that the terminal has performed differential encoding on the symbol block in the signal containing the pilot as in Equation 35 in TRP. In addition, the terminal can set P=B=2, N1=12, and N=16. Then, the terminal uses the symbol extraction unit to convert the signal into the time domain as in Equation 44. If symbol extraction is performed to extract only symbol components, it can be expressed as in mathematical expression 45 (S1907).

[0296]

[0297] The time phase conversion part of the terminal is the symbol extracted signal of mathematical expression 45. By performing time phase transformation on the signal, the time phase is transformed as in mathematical expression 46. can be printed (S1908).

[0298]

[0299] Here, n can be n=2,3,…,13. The differential decoding unit of the terminal converts the signal into the phase component of mathematical expression 46. By performing differential decoding in , the differentially decoded signal is obtained as in Equation 47. (i.e., the restored M-BASK symbol) can be estimated (S1909).

[0300]

[0301] The M-BASK symbol demapper of the terminal can perform M-BASK demapping on a signal differentially decoded by the differential decoding unit or on a signal symbol-detected by the symbol detection unit (S1910). The terminal can generate a recovered bit string through this demapping process.

[0302] In the above symbol extraction process, time phase conversion process, and differential decoding process, if it is assumed that the terminal has performed differential encoding on the symbol block including the pilot of Equation 33, the symbol extraction may not be performed first. Instead, the terminal may perform time phase conversion on the frequency-time converted signal and use it as an input to the differential decoding unit. Then, the differential decoding unit may perform the differential decoding mentioned in Equation 47 on the time-phase converted signal and then perform a symbol extraction process that takes only the symbol components excluding the pilot components. Thereafter, the terminal may perform M-BASK symbol demapping on the extracted symbols.

[0303] Meanwhile, the symbol detection unit can obtain the interference component of Equation 46 by subtracting the differentially decoded estimated M-BASK symbol from the output signal of the frequency-time conversion unit. At this time, the symbol detection unit can also obtain the interference component of Equation 46 by processing the differentially decoded estimated M-BASK symbol and subtracting it from the output signal of the frequency-time conversion unit. The terminal can subtract the interference component thus obtained from the output signal of the frequency-time conversion unit again to remove the interference, and then perform the symbol extraction process, time phase conversion process, and differential decoding process mentioned above once or repeatedly.

[0304] In addition, even if the positions of the symbol extraction unit and the time phase conversion unit are swapped, the consistency of the technology proposed in the present disclosure can be maintained. In other words, the time phase conversion unit can process the signal first, and then the symbol extraction unit can extract the symbol components excluding the pilot component. In the differential encoding-based PR-SC transmission method described with reference to FIG. 18 above, the transmitter can apply an M-PSK or M-QAM modulation scheme other than the M-BASK modulation scheme. Accordingly, the receiver can apply an M-PSK or M-QAM demapping process other than the M-BASK demapping process.

[0305] Fig. 20 is a flowchart illustrating a fourth embodiment of a signal design method that is immune to phase distortion performed at a transmitter.

[0306] Referring to Fig. 20, the TRP can generate a source bitstream to be transmitted to the terminal (S2001). Accordingly, the M-BASK symbol mapper of the TRP can perform M-BASK modulation on the generated source bitstream to generate a source symbol stream or source symbol block composed of elements of n=0,1,…,N1 having only integer components (S2002). Here, M and N1 can be positive integers.

[0307] Here, the integer signal constellation M-BASK mapping rules of the above bit / bits may be an example, and mapping rules of all other possible modulations may be included in the scope of the present disclosure. For example, another modulation mapping rule may be an M-PSK modulation mapping rule. An M-PSK mapper may perform M-PSK modulation on the generated source bit string to generate a symbol string or symbol block composed of elements of n=0,1,…,N1 having only a phase component. Another modulation mapping rule may be an M-QAM modulation mapping rule.

[0308] The M-QAM mapper can perform M-QAM modulation on the generated source bitstream to generate a symbol stream or symbol block composed of elements of n=0,1,…,N1 that only have phase components. Other modulation mapping rules, such as GMSK and DPSK, may also be included in the scope of the present disclosure. In addition, the receiver can perform demodulation and restoration by taking into account the characteristics of the modulation methods mentioned above.

[0309] The pilot insertion unit can insert a pilot into the middle of a symbol block of a signal output from the M-BASK mapper (S2003). The pilot inserted in this way can simply and effectively restore or equalize a nonlinear wireless channel within each symbol of the symbol block at the receiver. The pilot insertion unit can insert a pilot into the middle of the symbol blocks generated from the M-BASK mapper. In this way, the pilot insertion unit can insert a pilot into the middle of the symbol blocks of N1 bits generated from the M-BASK mapper. Here, the pilot can be P bits. Here, P can be a positive integer. Accordingly, the output x(n) of the pilot insertion unit can be composed of N-1 bits. Here, N can be a positive integer. And, N can be P+2N1. N can be 0, 1,…, N-1.

[0310] The pilot insertion unit can assign zero power values ​​to the first and last time-domain samples of the pilot to minimize interference caused by multipath when performing wireless channel estimation and frequency-domain equalization at the receiver. N1 can represent the length of a time-domain sample unit of a symbol block, and P can represent the length of a time-domain sample unit of the pilot. The pilot length can be set by considering the length of the symbol block and frequency efficiency, etc.

[0311] The differential encoding unit can perform differential encoding on an M-BASK modulated symbol block signal with a pilot inserted (S2004). At this time, the output q(n) of the differential encoding unit can be modeled as in the following mathematical expression 48 or mathematical expression 49.

[0312]

[0313] In mathematical expression 48, n is a positive real number, and n=0,1,…,N-1. Here, n' is a positive real number, and n'=N1+P.

[0314]

[0315] In Equation 49, n can be a positive real number. Here, n' can be a positive real number such that n'=N1+P. In one embodiment, the differential encoding unit can perform differential encoding by including the pilot as in Equation 48. Alternatively, in another embodiment, the differential encoding unit can perform differential encoding only on symbol blocks while leaving the pilot as in Equation 49. In addition, when the differential encoding unit performs differential encoding without including the pilot as in Equation 48, the first sample to be differentially encoded may be vulnerable to hardware damage, so the last sample(s) of the preceding pilot known to both the transmitter and receiver may be taken as the first sample. In addition, when the differential encoding unit performs differential encoding without including the pilot as in Equation 49, the first sample to be differentially encoded may be vulnerable to hardware damage, so the last sample(s) of the preceding pilot 1 known to both the transmitter and receiver may be taken as the first sample. As described above, the differential encoding unit may not perform differential encoding on all samples N1 of the corresponding symbol block, and may perform differential encoding by resetting to a unit N1 smaller than N1 (for example, N1 = 4N2) to reduce the quantization error mentioned above. However, if the differential encoding unit performs differential encoding in this way, there may be a burden of having to allocate known symbols because not every symbol that is reset is differentially encoded. To reduce this burden and improve channel estimation performance, the differential encoding unit may allocate known symbols as pilot symbols, and may also utilize such distributed pilot symbols.

[0316] The real phase conversion unit can convert the differentially encoded signal into a phase value z(n) as in Equation 50 or Equation 51 through a real phase conversion process (S2005). In the case of applying the above-mentioned M-PSK and M-QAM modulation methods, the real-phase conversion unit can follow the real-phase conversion process when applying the M-BASK modulation method.

[0317]

[0318] Here, n can be 0,1,…,N-1. Eun Yang's mistake It may be, It can be. θ is a positive real number. It can be. α is a positive real number that can converge to 0.

[0319]

[0320] Here, Eun Yang's mistake It may be, It can be. θ is a positive real number. _ can be. α is a positive real number that can converge to 0.

[0321] In equations 50 and 51 If it exists in (-1,1) or [-1,1), the ambiguity that occurs during the demodulation process at the receiver can be eliminated. The above weighted value is much larger than the weighted value of the conventional FM-OFDM transmission method, and can significantly reduce the quantization error that causes performance degradation in actual implementation.

[0322] Here, if the transmitter performs differential encoding and real phase conversion on the source bits, it can be robust to various phase distortions (e.g., phase distortions due to CFO, PhN, Doppler shift, Doppler spread, etc.) while minimizing implementation-related quantization errors.

[0323] The CP appender can generate a CP-added signal z'(n) by adding CP when applying the M-BASK modulation method to the real-phase-shifted signal z(n) (S2006). TRP is a size value A corresponding to the transmission power that enables the desired communication according to the call quality of the TRP and the terminal. c,t After applying the CP to the real phase-shifted signal, digital baseband filtering can be performed by passing it through a digital baseband filter (S2007). Here, A c,t can be a mistake. Through such digital baseband filtering, the TRP can bandpass filter the CP-added real phase-shifted signal by a desired bandwidth. Afterwards, the TRP can wirelessly transmit the bandpass-filtered CP-added real phase-shifted signal through the antenna. The signal wirelessly transmitted from the TRP can pass through the wireless fading channel. The terminal can receive the signal that has passed through the wireless fading channel from the TRP.

[0324] Fig. 21 is a flowchart illustrating a fourth embodiment of a signal design method that is immune to phase distortion performed at a receiving end.

[0325] Referring to Figure 21, the terminal can perform analog bandpass filtering for a desired bandwidth by applying a bandpass filter to the signal received from the TRP. Furthermore, the terminal can convert the received signal into a baseband signal and perform digital baseband filtering using a digital baseband filter for a desired amount of bandwidth (S2101). Through this process, the terminal can add AWGN to the received signal.

[0326] Afterwards, the terminal can remove the CP to remove ISI or ICI from the complex signal (i.e., a signal having both real and complex components) that has passed through the baseband filter via the CP removal unit (S2102). The complex signal r(n) from which the CP has been removed in such a CP removal unit can be expressed as in mathematical expression 37.

[0327] In equation 37, φ l (n) is a mistake It could be f c can be the carrier frequency, and τ l can be the delay value of channel path l, and β l (n) can be the phase value of the channel coefficient. A c,r can be a real number with the received power. This received power has a magnitude value A corresponding to the transmit power in TRP. c,t φ may be a size value that reflects path loss, etc. when received at the terminal. d.l (n) may be the Doppler component occurring in channel path l, and φ c (n) may be a CFO component occurring in channel path l, and φ p (n) may be a PhN component occurring in the channel path, and φ m (n) may be a phase distortion component that may occur randomly for n due to other hardware damage occurring in channel path l. In Equation 37, w(n) may represent the AWGN added to the terminal. In Equation 37, among the phase components, φ d,l (n), φ c (n) and φ p Each of (n) can generally have very high correlations between values ​​in adjacent n φ. m (n) may have random characteristics between values ​​in adjacent n, but the impact of these components on the overall performance may not be that large.

[0328] In this disclosure, TRP can perform differential encoding by taking these points into consideration. In other words, to eliminate Doppler, CFO, and phase components at the receiver, the difference between values ​​in adjacent n is calculated, and φ m Differential encoding can be performed at the transmitter to average the (n) components. Differential decoding at the receiver in relation to this can be described in detail below.

[0329] The terminal can effectively estimate a wireless fading channel, convert a time-domain signal into a frequency-domain signal for recovery (or equalization), estimate and recover in the frequency domain, and then convert the recovered signal back into a time-domain signal.

[0330] To this end, the terminal may be equipped with a time-frequency conversion unit, a channel estimation unit, a frequency-domain equalization unit, a frequency-time conversion unit, etc. Here, the time-frequency conversion unit may perform a time-frequency conversion process for converting a time-domain signal from which a CP has been removed into a frequency-domain signal. In addition, the channel estimation unit may perform a channel estimation process for estimating a channel from a frequency-domain signal. The frequency-domain equalization unit may perform a frequency-domain equalization process for restoring a wireless channel using the frequency-domain signal and channel estimation. The frequency-time conversion unit may perform a frequency-time conversion process for converting the channel-restored frequency-domain signal back into a time-domain signal. The wireless channel environment experienced by the TRP and the terminal may only have an LoS path. Then, the time-frequency conversion process, the channel estimation process, the frequency-domain equalization process, and the frequency-time conversion process may not be required in the terminal.

[0331] The wireless channel environment experienced by the TRP and the terminal may also include NLoS paths. Therefore, to overcome the performance degradation caused by the multipath effect described in Equation 37, the terminal can process the received signal through the processing steps of the four functional blocks mentioned above. The processing steps of these functional blocks can be as follows.

[0332] First, the time-frequency conversion unit can convert a signal from the time domain to the frequency domain (S2103). This time-frequency conversion unit can extract a time-domain reception signal corresponding to a pilot component from a CP-removed reception signal, as in Equation 52.

[0333]

[0334] The time-frequency conversion unit can perform N-point DFT or FFT on the extracted signal of Equation 52 to convert it into a signal Rp(k) of a pilot component in the frequency domain as in Equation 53.

[0335]

[0336] W in equation 53 p (k) can mean noise component, and H p (k) can mean frequency response, Y p (k) is a known time domain pilot signal y expressed by mathematical expression 54. p (n) may refer to a frequency domain signal corresponding to a pilot component converted into a frequency domain, and may be as in mathematical expression 55.

[0337]

[0338]

[0339] In addition, the time-frequency conversion unit can extract a time-domain reception signal corresponding to a symbol component from a reception signal from which CP has been removed, as in mathematical expression 56.

[0340]

[0341] In mathematical expression 56, the time domain reception signal r corresponding to the symbol component s The time domain transmission signal corresponding to (n) is y s (n) can be. The time frequency transform unit performs N-point DFT or FFT on the extracted signal of Equation 56 to obtain a signal R of frequency domain symbol components as in Equation 57. s (k) can be converted to.

[0342]

[0343] W in equation 57 s (k) can mean noise component, and H s (k) can mean frequency response, Y s (k) is the time domain symbol signal y sent from the transmitter. s (n) can mean a frequency domain signal converted into the frequency domain.

[0344] The channel estimation unit of the terminal can estimate the wireless channel in the frequency domain (S2104). This channel estimation unit is known to both the transmitter and receiver by using the mathematical expression Y of Equation 55. p (k) Using the signal R of the pilot component in the frequency domain of mathematical expression 53 p (k) Frequency response of the wireless channel using LS, MMSE, DFT or any other possible channel estimation method. can be obtained.

[0345] The frequency domain equalizer of the terminal can perform frequency domain equalization by compensating for the wireless channel in the frequency domain (S2105). The channel estimation unit can calculate the frequency response H reflected in the symbol component shown in Equation 57. s To compensate for (k), we go through a channel estimation process. can be estimated. The frequency domain equalizer converts the time domain signal into a frequency domain signal as in Equations 53 and 55, thereby obtaining H in Equation 53.p (k) is H in mathematical expression 55 s (k) with a value almost similar to that of, in other words, Considered as such, it is estimated as above. Estimated value of frequency response of mathematical expression 55 It can be assumed that the frequency domain equalizer is can be equalized in all possible ways for Equation 55. However, the frequency domain equalizer H p (k) is transformed by performing a cyclic shift on H, which is different from equations 54 and 55. s (k) can be made almost similar to H. In addition, the frequency domain equalizer p Rotate any phase of (k) and perform a transformation different from Equations 54 and 55 to obtain H s (k) can be made almost similar to H. Alternatively, the frequency domain equalizer can be made to H p (k) When converting from the time domain to the frequency domain, a different transformation from Equations 54 and 55 is performed to obtain H s (k) can be made to be almost similar. Like this, Any possible method of doing so may be included within the scope of the present disclosure.

[0346] The frequency-time converter is a signal equalized in the frequency domain. Perform N-point IDFT or IFFT on the time domain signal as in Equation 58 can be converted to (S2106).

[0347]

[0348] Here, φ may represent a phase error component that may occur during frequency-domain equalization. For convenience of explanation, the above description assumes perfect time synchronization; however, this is generally not the case, and the scope of the present disclosure can be applied without loss of generality under such imperfect time synchronization conditions.

[0349] Meanwhile, it can be assumed that the terminal has performed a TRP-based encoding on the symbol block in the signal containing the pilot as in Equation 49. In addition, the terminal can set B=2, P=2B, N1=6, and N=16. Then, the terminal uses the symbol extraction unit to convert the signal into the time domain as in Equation 58. If symbol extraction is performed to extract only symbol components, it can be expressed as in mathematical expression 59 (S2107).

[0350]

[0351] The time phase conversion part of the terminal is the symbol extracted signal of mathematical expression 59. By performing time phase transformation on the signal, the time phase is transformed as in mathematical expression 60. can be output (S2108). Alternatively, the time phase conversion unit of the terminal can output the symbol extracted signal of mathematical expression 59. By performing time phase transformation on the signal, the time phase is transformed as in mathematical expression 61. can be printed.

[0352]

[0353]

[0354] The differential decoding unit of the terminal converts the signal into the phase component of mathematical expression 60. Differential decoding is performed in the differentially decoded signal as in mathematical expression 62. (i.e., the restored M-BASK symbol) can be estimated (S2109). Alternatively, the differential decoding unit of the terminal can estimate the signal converted into the phase component of mathematical expression 61. By performing differential decoding in , the differentially decoded signal is obtained as in mathematical expression 63. (i.e., the restored M-BASK symbol) can be estimated.

[0355]

[0356]

[0357] The M-BASK symbol demapper of the terminal can perform M-BASK demapping on a signal differentially decoded by the differential decoding unit or on a signal symbol-detected by the symbol detection unit (S2110). The terminal can generate a recovered bit string through this demapping process.

[0358] In the above symbol extraction process, time phase conversion process, and differential decoding process, if it is assumed that the terminal has performed differential encoding on the symbol block including the pilot as in Equation 48, the symbol extraction may not be performed first. Instead, the terminal may perform time phase conversion on the frequency-time converted signal and use it as input to the differential decoding unit. Then, the differential decoding unit may perform differential decoding as mentioned in Equation 59 on the time-phase converted signal and then perform a symbol extraction process that takes only the symbol components excluding the pilot components. Thereafter, the terminal may perform M-BASK symbol demapping on the extracted symbols.

[0359] The symbol detection unit can obtain the interference components of Equations 60 and 61 by subtracting the differentially decoded estimated M-BASK symbol from the output signal of the frequency-time conversion unit. At this time, the symbol detection unit can also obtain the interference components of Equations 60 and 61 by processing the differentially decoded estimated M-BASK symbol and subtracting it from the output signal of the frequency-time conversion unit. The terminal can subtract the interference component thus obtained from the output signal of the frequency-time conversion unit again to remove the interference, and then perform the symbol extraction process, time phase conversion process, and differential decoding process mentioned above once or repeatedly.

[0360] In addition, even if the positions of the symbol extraction unit and the time phase conversion unit are swapped, the consistency of the technology proposed in the present disclosure can be maintained. In other words, the time phase conversion unit can process the signal first, and then the symbol extraction unit can extract the symbol components excluding the pilot component. In the differential encoding-based PR-SC transmission method described with reference to FIG. 20 above, the transmitter can apply an M-PSK or M-QAM modulation scheme other than the M-BASK modulation scheme. Accordingly, the receiver can apply an M-PSK or M-QAM demapping process other than the M-BASK demapping process.

[0361] Fig. 22 is a block diagram showing a third embodiment of a signal design device immune to phase distortion.

[0362] Referring to FIG. 22, the transmitter may include a QAM symbol mapper (2201), a spreading part (2202), a centrally symmetric symbol mapper (2203), a subcarrier indexing part (2204), a time domain transform part (2205), a cumulative linear encoding part (2206), a real phase transform part (2207), a CP adding part (2208), and / or a baseband filter (2209). The receiver may include a baseband filter (2211), a CP removing part (2212), a time phase transform part (2213), a cumulative linear decoding part (2214), a frequency domain transform part (2215), a subcarrier deindexing part (2216), a centrally symmetric symbol demapper (2217), a despreading part (2218), a symbol detector (2219), and / or a QAM symbol demapper (2220).

[0363] The transmitter may be a TRP, and the receiver may be a terminal. The transmitter and receiver may use a phase distortion-resistant orthogonal frequency division multiplexing (PR-OFDM) wireless transmission method. Here, the PR-OFDM transmission method can have high phase distortion resistance and a zero phase-to-noise ratio (PAPR).

[0364] A transmitter can generate a source bitstream to be transmitted to a receiver. A QAM symbol mapper can perform symbol mapping using the Quadrature Amplitude Modulation (QAM) modulation scheme on the generated source bitstream to generate a complex signal block. Of course, the transmitter can perform symbol mapping using any possible modulation scheme other than QAM modulation to generate a complex signal block.

[0365] FM-OFDM may have problems such as Out-Of-Band (OOB) problem, high PAPR problem, etc. To solve these problems of FM-OFDM, the spreading unit is a symbol-mapped signal sequence vector d (each element d k , k=0,1,…,M-1) by encoding with all possible types of spreading matrices S (M×M diagonal matrices and / or M×M off-diagonal matrices) and / or performing spreading as in Equation 63 to generate a symbol sequence vector y. In this way, the spreading unit can pass the symbol sequence vector as an input to the centrally symmetric symbol mapper.

[0366]

[0367] A centrally symmetric symbol mapper can perform centrally symmetric symbol mapping in the frequency domain. To this end, the centrally symmetric symbol mapper can allocate complex signals of signal blocks generated in ascending order of subcarriers from DC (direct current) to upper subcarrier groups of available subcarriers, excluding DC. In addition, the centrally symmetric symbol mapper can allocate conjugate signals of complex signals of signal blocks generated in ascending order of subcarriers from DC to lower subcarrier groups of available subcarriers, centered on DC.

[0368] The subcarrier indexing unit can perform subcarrier indexing to transmit signal block information to be sent to a desired frequency band in the RF in the frequency domain. To this end, the subcarrier indexing unit can directly map signals assigned to a lower subcarrier group including DC to the first part of the input terminal of the time domain transform unit. In addition, the subcarrier indexing unit can directly map signals assigned to an upper subcarrier group to the last part of the input terminal of the time domain transform unit. The subcarrier indexing unit can map zero values ​​to other elements of the input terminal of the time domain transform unit.

[0369] The time domain transform unit can generate a signal in the inter-domain using a subcarrier-indexed N-point IDFT or IFFT. At this time, the generated signal can have only real components according to the characteristics of the centrally symmetric symbol mapping.

[0370] Next, the cumulative linear encoding unit can perform cumulative linear encoding on the generated time-domain signal block. At this time, the cumulative linear encoding can be differential encoding or stepwise encoding. The real phase transform unit can transform the cumulative linear encoded signal into the phase domain. And, the CP adding unit can add a CP to the signal transformed into the phase domain. The transmitter can use a baseband filter to upscale the baseband signal and transmit it to the receiver. At this time, the transmitter can perform RF bandpass filtering.

[0371] A signal transmitted from a transmitter can be received by a receiver via a wireless channel. AWGN can be added to such a signal. The receiver can perform bandpass filtering on the received signal. Furthermore, the receiver can perform baseband filtering using a baseband filter. Subsequently, the receiver can perform a CP removal process using a CP remover. The receiver can perform time phase transformation using a time phase shifter. The receiver can perform cumulative linear decoding using a cumulative linear decoding unit. Here, the cumulative linear decoding can be differential decoding or stepwise decoding. The receiver can convert a time domain signal into a frequency domain signal using a frequency domain transform unit. The frequency domain transform unit can be an N-point DFT or FFT.

[0372] Next, the frequency domain transform unit can transmit the signal block that has undergone the N-point DFT or FFT process to the subcarrier deindexing unit. The subcarrier deindexing unit can perform a process corresponding to the reverse order of the process performed by the subcarrier indexing unit. For example, the subcarrier deindexing unit can move the signals from element 0 to element N / 2-1 of the output terminal of the N-point DFT or FFT to behind the signals from element N / 2 to element N-1. In other words, the subcarrier deindexing unit can cyclically shift the received signal to the left by N / 2. The subcarrier deindexing unit can transmit the subcarrier-deindexed signal to a centrally symmetric symbol demapper.

[0373] A centrally symmetric symbol demapper can perform centrally symmetric symbol demapping by receiving a subcarrier-deindexed signal from a subcarrier deindexing unit. In other words, the centrally symmetric symbol mapper can perform conjugation on signals of a lower subcarrier group to which a signal mentioned at a transmitter is assigned, starting from element N / 2+1 of an input terminal. In addition, the centrally symmetric symbol mapper can extract signals corresponding to a symbol block transmitted by combining signals of an upper subcarrier group to which a signal mentioned at a transmitter is assigned, starting from element N / 2-1 of an input terminal, and conjugated signals corresponding to these signals.

[0374] The despreading part extracts the signal sequence vector r (each element d k ,k=0,1,…,M-1), in order to solve the OOB problem and high PAPR problem of FM-OFDM, the symbol sequence vector obtained by performing despreading as in Equation 65, for example can be passed to the QAM symbol demapper.

[0375]

[0376] Here, stands for the Hermitian matrix of S. can mean the inverse matrix of S. The symbol detector can detect a symbol in a signal containing a pilot. Then, the QAM symbol demapper despreads the symbol sequence vector The source bitstream can be restored by performing QAM symbol demapping.

[0377] The operations of the method according to the embodiments of the present disclosure can be implemented as a computer-readable program or code on a computer-readable recording medium. A computer-readable recording medium includes any type of recording device that stores information readable by a computer system. Furthermore, a computer-readable recording medium can be distributed across network-connected computer systems, allowing the computer-readable program or code to be stored and executed in a distributed manner.

[0378] Additionally, the computer-readable recording medium may include hardware devices specifically configured to store and execute program instructions, such as ROM, RAM, flash memory, etc. The program instructions may include not only machine language codes produced by a compiler, but also high-level language codes that can be executed by a computer using an interpreter, etc.

[0379] While some aspects of the present disclosure have been described in the context of a device, they may also represent a description of a corresponding method, wherein a block or device corresponds to a method step or a feature of a method step. Similarly, aspects described in the context of a method may also be described as a corresponding block or item or a feature of a corresponding device. Some or all of the method steps may be performed by (or using) a hardware device, such as, for example, a microprocessor, a programmable computer, or an electronic circuit. In some embodiments, at least one or more of the most important method steps may be performed by such a device.

[0380] In embodiments, a programmable logic device (e.g., a field-programmable gate array) may be used to perform some or all of the functions of the methods described herein. In embodiments, the field-programmable gate array may operate in conjunction with a microprocessor to perform one of the methods described herein. In general, the methods are preferably performed by some hardware device.

[0381] Although the present disclosure has been described with reference to preferred embodiments thereof, it will be understood by those skilled in the art that various modifications and changes may be made to the present disclosure without departing from the spirit and scope of the present disclosure as set forth in the claims below.

Claims

1. As a method of transmitter, A step of generating a first symbol block by modulating a source bit string to be transmitted to a receiver; A step of generating a second symbol block by inserting at least one pilot into the first symbol block; A step of generating a third symbol block by cumulatively linearly encoding the second symbol block; A step of performing a real phase transformation on the third symbol block to generate a first signal converted into a phase value; A step of generating a second signal by adding a CP (cyclic prefix) to the first signal; and Comprising the step of wirelessly transmitting the second signal to the receiver, Method of transmitter.

2. In claim 1, In the step of generating a first symbol block by modulating a source bit stream to be transmitted to the receiver, the transmitter generates the first symbol block from the source bit stream using any one of BASK (bipolar amplitude shift keying) modulation, PSK (phase shift keying) modulation, QAM (quadrature and amplitude modulation) modulation, GMSK (Gaussian filtered minimum-shift keying) modulation, or DPSK (differential phase-shift keying) modulation. Method of transmitter.

3. In claim 1, In the step of inserting at least one pilot into the first symbol block, the transmitter inserts a first pilot into a preceding time domain of the first symbol block and inserts a second pilot into a succeeding time domain of the first symbol block. Method of transmitter.

4. In claim 3, The last time domain samples of the first pilot and the first time domain samples of the second pilot have a power value of zero. Method of transmitter.

5. In claim 1, In the step of inserting at least one pilot into the first symbol block, the transmitter inserts a third pilot in the middle of the first symbol block. Method of transmitter.

6. In claim 5, The first and last time domain samples of the third pilot have a power value of zero. Method of transmitter.

7. In claim 1, The step of generating a third symbol block by cumulatively linearly encoding the second symbol block is: A step of dividing the second symbol block into first partial symbol blocks based on the cyclic period of the time domain sample unit; and A step of generating the third symbol block that is cumulatively linearly encoded by performing differential encoding on each of the first partial symbol blocks, Method of transmitter.

8. In claim 7, In the step of generating the third symbol block that is cumulatively linearly encoded by performing differential encoding on each of the first partial symbol blocks, The transmitter performs differential encoding after adding the first sample value of the previous first partial symbol block to the first sample of each of the first partial symbol blocks. Method of transmitter.

9. In claim 1, The step of generating a third symbol block by cumulatively linearly encoding the second symbol block is: A step of dividing the first symbol block within the second symbol block into second partial symbol blocks; and A step of generating the third symbol block that is cumulatively linearly encoded by performing differential encoding on each of the second partial symbol blocks, Method of transmitter.

10. In claim 1, The step of generating a third symbol block by cumulatively linearly encoding the second symbol block is: A step of generating the third symbol block by differentially encoding the second symbol block and generating the cumulative linearly encoded third symbol block, Method of transmitter.

11. As a method of receiver, A step of converting a received signal from a transmitter into a baseband signal; A step of removing CP from the above baseband signal; A step of equalizing a wireless channel for a received signal from which the CP has been removed; A step of performing time phase conversion on a received signal for which the wireless channel is equalized; A step of performing cumulative linear decoding on the time phase-shifted signal; and A step of demodulating the above-described accumulated linear decoded signal to generate a source bit string, Method of receiver.

12. In claim 11, The step of equalizing the wireless channel for the received signal from which the CP has been removed is as follows: A step of converting a time domain pilot signal of a received signal from which the CP has been removed into a frequency domain pilot signal; A step of converting a time domain symbol signal of a received signal from which the CP has been removed into a frequency domain symbol signal; A step of estimating a first wireless channel from the frequency domain pilot signal; A step of estimating a second wireless channel from the frequency domain symbol signal; and including a step of equalizing the second wireless channel using the first wireless channel; Method of receiver.

13. In claim 11, The step of performing cumulative linear decoding on the above time phase transformed signal is: A step of dividing the time phase-converted signal into first partial symbol blocks based on the cyclic period of the time domain sample unit; and A step of generating the cumulative linear decoded signal by performing differential decoding on each of the first partial symbol blocks, Method of receiver.

14. In claim 12, The step of performing cumulative linear decoding on the above time phase transformed signal is: A step of extracting a symbol block of the time phase transformed signal; a step of dividing the symbol block into second partial symbol blocks; and A step of generating the cumulative linear decoded signal by performing differential decoding on each of the second partial symbol blocks, Method of receiver.

15. In claim 12, The step of performing cumulative linear decoding on the above time phase transformed signal is: A step of extracting a symbol block from the time phase transformed signal; comprising a step of differentially decoding the symbol block to generate the cumulative linearly decoded signal; Method of receiver.

16. In claim 11, The step of demodulating the above-mentioned accumulated linear decoded signal to generate a source bit string is: A step of demodulating the above-described accumulated linear decoded signal to generate a first symbol block; A step of generating a second symbol block by removing a pilot from the first symbol block; and A step of generating the source bit string by performing symbol demapping on the second symbol block, Method of receiver.

17. As a transmitter, Includes a processor, The above processor is the transmitter, Modulating a source bit string to be transmitted to a receiver to generate a first symbol block; Generating a second symbol block by inserting at least one pilot into the first symbol block; Generating a third symbol block by cumulatively linearly encoding the second symbol block; Performing a real phase transformation on the third symbol block to generate a first signal converted to a phase value; Generating a second signal by adding a CP (cyclic prefix) to the first signal; and causing the second signal to be wirelessly transmitted to the receiver; transmitter.

18. In claim 17, In the step of inserting at least one pilot into the first symbol block, the processor causes the transmitter to insert a first pilot into a preceding time domain of the first symbol block and to insert a second pilot into a succeeding time domain of the first symbol block. transmitter.

19. In claim 17, In the step of inserting at least one pilot into the first symbol block, the processor causes the transmitter to insert a third pilot in the middle of the first symbol block. transmitter.

20. In claim 17, In the step of generating a third symbol block by cumulatively linearly encoding the second symbol block, the processor causes the transmitter to: Dividing the second symbol block into first partial symbol blocks based on the cyclic period of the time domain sample unit; and Causing to generate the third symbol block that is cumulatively linearly encoded by performing differential encoding on each of the first partial symbol blocks, transmitter.

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