Hybrid coupler with second harmonic termination

Abstract

A 90 degree hybrid coupler with harmonic termination is presented. According to one aspect, the harmonic is a second harmonic that is connected between the direct port and the coupled port of the hybrid coupler. The harmonic termination includes a resonant circuit that includes at least one capacitor and one inductor in series connection. A resonant frequency of the resonant circuit is selected to be between the operating frequency and the second harmonic. According to one aspect, the resonant frequency is selected to be at a distance of about one third of the operating frequency from the second harmonic. According to yet another aspect, the inductor includes a coupled-line structure that provides an inductance of the inductor. The direct port and coupled port of the coupled-line structure are shorted to ground and the inductance is an inductively coupled inductance between the input port and isolated port of the coupled-line structure.

Description

HYBRID COUPLER WITH SECOND HARMONIC TERMINATIONCROSS-REFERENCE TO RELATED APPLICATIONS

[0001] The present application claims priority to U.S. Provisional Patent Application No. 63 / 597,814 filed on November 10, 2023, the contents of all of which is incorporated herein by reference in their entirety.TECHNICAL FIELD

[0002] The present teachings relate to radio frequency (RF) circuits. More particularly, the present teachings relate to methods and apparatuses for suppression of second harmonics in a hybrid coupler.BACKGROUND

[0003] Operation of active electronic circuits at the limit for linearity may engender harmonics of the operating (center) frequency of such circuits. This is the case, for example, of a radio frequency (RF) amplifier operating at the IdB compression point, PldB. In some cases, increased linearity may be provided via use of a well-known in the art balanced RF amplifier that includes two amplifiers operating in quadrature, where inputs to the two amplifiers are provided via an input (90 degrees) hybrid coupler, and quadrature outputs of the two amplifiers are combined via an output (90 degrees) hybrid coupler.

[0004] Hybrid couplers may be designed according to a target operating frequency, for example, a center frequency of an RF signal to be processed. Although such hybrid couplers may provide consistent (90 degrees) phase balance over an extended frequency range that may include the operating frequency and a second harmonic thereof, amplitude balance may only be provided over a small bandwidth around the operating frequency. Such imbalance of the amplitude at higher frequencies, including at the second harmonic of the operating frequency, may in turn adversely affect performance of, for example, a balanced RF amplifier, including an AM-AM distortion (undesired amplitude modulation) performance and / or an AM-PM performance (undesired phase modulation).

[0005] The amplitude balance performance of a hybrid coupler may be considered as inherent to the design of the hybrid coupler, with some designs including improvement in the amplitude balance performance at the cost of a degradation of the phase balance performance. For example, a twisted hybrid coupler design may include a wider bandwidth amplitude balance when compared to a coupled-line hybrid coupler but at a cost of a narrower bandwidth phase balance. Furthermore, the twisted hybrid coupler may require a substantial larger physical area when compared to the coupled-line hybrid coupler.

[0006] A prior art solution for reducing the effect of the second harmonics on the performance of an RF circuit, such as an RF amplifier, including a balanced RF amplifier, may include coupling of second harmonic traps (e.g., notch filter, band-rejection filter) to the RF circuit. Although such prior art solution may effectively reduce the amplitude of the second harmonics, and therefore their effect on the linearity related performance of the RF circuit (e.g., AM-AM, AM-PM, etc.), the quality-factor of the second harmonic traps may degrade other performance metrics of the RF circuit, such as, for example, gain and output power.

[0007] Teachings according to the present disclosure aim at overcoming the above shortcomings via use of a novel 90 degree hybrid coupler with second harmonic termination that may be based on a coupled-line design.SUMMARY

[0008] According to a first aspect of the present disclosure, a circuit is presented, the circuit comprising: a 90 degree hybrid coupler comprising an input port, a direct port, a coupled port and an isolated port, the isolated port coupled to ground through a termination resistor; and a resonant circuit including a first capacitor, a second capacitor, and an inductor in series connection, the inductor connected between the first capacitor and the second capacitor, the resonant circuit connected between the direct port and the coupled port, wherein the 90 degree hybrid coupler is configured for operation at a center frequency, and respective first and second capacitances of the first and second capacitor and inductance of the inductor are selected so that an impedance of the resonant circuit resonates at a resonant frequency that isabout one third of the center frequency smaller than a second harmonic of the center frequency.

[0009] According to a second aspect of the present disclosure, a balanced amplifier configured for operation at a center frequency is presented, the balanced amplifier comprising: an input 90 degree hybrid coupler comprising respective input port, direct port, coupled port and isolated port, the isolated port coupled to ground through a termination resistor; an output 90 degree hybrid coupler comprising respective input port, direct port, coupled port and isolated port, the isolated port coupled to ground through a termination resistor; first and second amplifiers comprising: respective first and second inputs respectively connected to the direct port and coupled port of the input 90 degree hybrid coupler; and respective first and second outputs respectively connected to the coupled port and the direct port of the output 90 degree hybrid coupler; and a resonant circuit including a first capacitor, a second capacitor, and an inductor in series connection, the inductor connected between the first capacitor and the second capacitor, the resonant circuit connected between the direct port and the coupled port of the output 90 degree hybrid coupler, wherein respective first and second capacitances of the first and second capacitor and inductance of the inductor are selected so that an impedance of the resonant circuit resonates at a resonant frequency that is greater than the center frequency and smaller than a second harmonic of the center frequency.

[0010] According to a third aspect of the present disclosure, a circuit is presented, the circuit comprising: at least one active or passive device; and a shorted coupled-line inductor coupled to the at least one active or passive device at a first or second terminal of the shorted coupled- line inductor, wherein the shorted coupled-line inductor comprises a coupled-line structure that includes two same lengths conductor lines arranged at close proximity in parallel along their respective lengths, a direct port and a coupled port of the coupled-line structure of the shorted coupled-line inductor are respectively shorted to ground, and an input port and an isolated port of the coupled-line structure of the shorted coupled-line inductor respectively provide the first and second terminals of the inductor.

[0011] Further aspects of the disclosure are provided in the description, drawings and claims of the present application.BRIEF DESCRIPTION OF DRAWINGS

[0012] The accompanying drawings, which are incorporated into and constitute a part of this specification, illustrate one or more embodiments of the present disclosure and, together with the description of example embodiments, serve to explain the principles and implementations of the disclosure.

[0013] FIG. 1 A shows a circuital representation of a prior art 90 degree hybrid coupler.

[0014] FIG. IB shows a coupled-line configuration that may be used in the implementation of the 90 degree hybrid coupler of FIG. 1A.

[0015] FIG. 2A shows a circuital representation of a 90 degree hybrid coupler with second harmonic termination according to the present disclosure.

[0016] FIG. 2B shows graphs representing amplitude and phase response of the hybrid couplers of FIG. 1A and FIG. 2A.

[0017] FIG. 3A shows details of a second harmonic termination according to an embodiment of the present disclosure, including an inductance.

[0018] FIG. 3B shows various embodiments according to the present disclosure of the inductance shown in FIG. 3A, including an embodiment based on a coupled-line inductance according to the present disclosure.

[0019] FIG. 4A shows a circuital representation of the coupled-line inductance.

[0020] FIG. 4B shows a circuital representation of the 90 degree hybrid coupler of FIG. 2A with the second harmonic termination of FIG. 4A.

[0021] FIG. 5A shows a circuital representation of a balanced amplifier according to an embodiment of the present disclosure.

[0022] FIG. 5B shows graphs representative of performance of the balanced amplifier of FIG. 5 A, further contrasted to a prior art balanced amplifier.

[0023] FIG. 6 shows circuital representations of various circuits, including LC stages, according to the present disclosure.

[0024] Like reference numbers and designations in the various drawings indicate like elements.DETAILED DESCRIPTION

[0025] Throughout this description, embodiments and variations are described for the purpose of illustrating uses and implementations of the inventive concept. The illustrative description should be understood as presenting examples of the inventive concept, rather than as limiting the scope of the concept as disclosed herein.

[0026] FIG. 1A shows a circuital representation of a prior art 90 degree hybrid coupler (100A). The hybrid coupler (100A) may include a passive device (110) that includes four ports: a coupled port, CPL, a direct port, DIR, an input / common port, IN, and an isolated port, ISO, with paths between any two ports providing either zero degrees, 0°, phase delay or ninety degrees, 90°, phase delay as shown in FIG. 1A. The hybrid coupler (100 A) can be used as: i) a power splitter, where an input signal (e.g., RFIN) provided at the input port (IN) is split into two signals (e.g., RFCPL, RFDIR) of equal power (e.g., -3dB relative to the input signal) but ninety degrees, 90°, out of phase (e.g., quadrature, A(|> = 90°) that are provided at the coupled port, CPL, and the direct port, DIR, or ii) as a power combiner where two equal power quadrature signals (e.g., RFCPL, RFDIR) provided at the coupled port, CPL, and thedirect port, DIR, are combined at the input / common port, IN. As shown in FIG. 1 A, operation of the prior art 90 degree hybrid coupler (100A) precludes terminating the isolated port, ISO, of the passive device (110) with a termination impedance (e.g., ZT) that is equal to a characteristic system impedance (e.g., 50 ohms, 50Q).

[0027] Teachings according to the present disclosure may allow implementation of the passive device (110) of a 90 degree hybrid coupler via a (prior art) coupled-line configuration (110) shown in FIG. IB. As shown in FIG. IB, the coupled-line configuration (110) may include two equal length (e.g., LCL) transmission lines (e.g., microstrips, conductors), CL1 and CL2, that are arranged in close proximity and parallel to one another such as a signal being conducted through a first line, CL1, may inductively couple to a second line, CL2, e.g., via a mutual inductance between CL1 and CL2. In particular, the two transmission lines, CL1 and CL2, may be considered as face-coupled, or in other words, respective (metal) widths of the lines CL1 and CL2 may overlap along the length, LCL, of the lines, so to provide increased (inductive) coupling between the two lines. In turn, such increased coupling may allow for a more compact implementation of the 90 degree hybrid coupler when compared to other implementations, such as for example, a twisted 90 degree hybrid coupler or a side- coupled 90 degrees hybrid implementation. It should be noted that the length, LCL, of the two transmission lines CL1 and CL2 may be based on a (target) frequency of operation of the corresponding 90 degree hybrid coupler and typically equal to about half of quarter wavelength, 1 / 8 X, of the frequency of operation (e.g., center frequency, fundamental frequency) compared to about 1 / 4 X of the frequency of operation for a side-coupled 90 degree hybrid coupler. In other words, the two transmission lines CL1 and CL2 may be quarter wavelength transmission lines. It should be noted that teachings according to the present disclosure may cover operating RF frequencies in the range of few GHz (e.g., as small as 1 GHz) and up to (including) the millimeter band (e.g., 30-300 GHz).

[0028] FIG. 2A shows a circuital representation (200A) of a 90 degree hybrid coupler (110) with second harmonic termination (200A) according to the present disclosure. According to an embodiment of the present disclosure, the 90 degree hybrid coupler (110) may be based on the coupled-line configuration (110) of FIG. IB with ports CPL and DIR coupled to respective terminals of the second harmonic termination (220). Accordingly, the RF signalsRFTCPL and RFTDIR respectively coupled to the ports CPL and DIR may be considered as second harmonic terminated signals, or in other words, having amplitudes and / or phases modified at about the second harmonic. According to an embodiment of the present disclosure, a bandwidth of operation of the second harmonic termination (220) is away from a bandwidth of operation of the RF signal conducted through the coupler (110). In other words, as shown in the graphs of FIG. 2B, the second harmonic termination (220) may not (substantially) affect in-band amplitude and phase of the RF signal at about the center frequency of operation, Hl.

[0029] FIG. 2B shows graphs representing amplitude (top graphs labelled Mag for RFTCPL and RFTDIR) and phase (bottom graph labelled \<) for RFT) response of the hybrid coupler with second harmonic termination shown in FIG. 2A contrasted to the amplitude (top graphs labelled Mag for RFCPL and RFDIR) and phase (bottom graph labelled \<) for RF) response of the prior art hybrid coupler of FIG. 1 A. It should be noted that the top amplitude response graphs of FIG. 2B show the amplitudes at each of the ports CPL and DIR for each of the present case (e.g., RFTCPL and RFTDIR) that includes a coupled second harmonic termination (220) across such ports and the prior art case (e.g., RFCPL and RFDIR) where no second harmonic termination is provided. On the other hand, the phase response graphs at the bottom of FIG. 2B show the phase difference between the two ports CPL and DIR for each of the two cases (RFT and RF). It should be noted that in the exemplary graphs of FIG. 2B, the operating frequency, Hl, is assumed to be at about 28 GHz, and therefore the second harmonic, H2, may be at about 56 GHz.

[0030] For the case where no second harmonic termination is provided, as shown in the top amplitude response graphs of FIG. 2B, at Hl and about Hl, the amplitudes of RFCPL and RFDIR may be balanced (i.e., equal or substantially equal). However, as the frequency increases, the two amplitudes diverge with RFCPL amplitude increasing and RFDIR amplitude decreasing. Accordingly, for the case where no second harmonic termination is provided, there may be an imbalance in the amplitudes of RFCPL and RFDIR (e.g., of about 4 dB) at the second harmonic, H2. Furthermore, for the same case, as shown in the bottom phase response graph of FIG. 2B labelled RF, the phase difference between the two ports CPL and DIR may remain substantially constant at about 90 degrees as the frequency varies from Hl to H2. Asdiscussed earlier in the present disclosure, the amplitude imbalance at the second harmonic, H2, may negatively affect performance of a circuit (e.g., balanced amplifier) using the prior art configuration whether or not the two RF signals RFCPL and RFDIR at the ports CPL and DIR are to be combined.

[0031] For the case where the second harmonic termination (220) is provided, as shown in the top amplitude response graphs (labelled RFTCPL and RFTDIR) and the bottom phase response graph (labelled RFT) of FIG. 2B, at Hl and about Hl, the amplitudes of RFTCPL and RFTDIR may be balanced (i.e., equal or substantially equal) and the phases of RFTCPL and RFTDIR may be balanced (i.e., phase difference equal or substantially equal to 90 degrees). Furthermore, at the second harmonic, H2, the amplitudes of RFTCPL and RFTDIR may be equal or substantially equal. Furthermore, as shown in the bottom phase response graph of FIG. 2B labelled RFT, at the second harmonic, H2, the phases of RFTCPL and RFTDIR may be balanced (i.e., phase difference equal or substantially equal to 90 degrees). It should be noted that although the equal or substantially equal amplitudes of the RFTCPL and RFTDIR at the H2 frequency shown in FIG. 2B may be slightly different from -3dB (e.g., about -3.5 dB), optimization of values of component (e.g., capacitance, inductance) that make the second harmonic termination (220) may allow obtaining amplitudes closer to -3dB if desired.

[0032] Accordingly, the second harmonic termination (220) according to the present disclosure may provide balancing of the otherwise unbalanced split RF power at the CPL and DIR ports of a traditional 90 degree hybrid coupler. In other words, inherent H2 related deficiencies of a traditional 90 degree hybrid coupler, including one based on the (compact) coupled-line configuration of FIG. IB, may be compensated by coupling the second harmonic termination (220) according to the present teachings between the CPL and DIR ports. Because the resultant RF signals, RFTCPL and RFTDIR respectively coupled to the ports CPL and DIR may have, at the second harmonic frequency, substantially same amplitudes and a 90 degrees phase difference, combining such RF signals via, for example, another 90 degree hybrid coupler, may result in a combined RF signal with a reduced H2 harmonic and derivatives / products thereof. It should be noted that the expression “balanced” or “balancing” as used in the present disclosure may be contextual and refer to: a) equal or substantially equal in the context of amplitudes; or b) a difference that is equal to or substantially equal to90 degrees in the context of phases. It should be noted that the amplitude of the RFTCPL signal at the second harmonic, H2, may generally be much larger than the amplitude of RFTDIR signal at the second harmonic, H2. In turn, this may allow a significant level of the second harmonic to exit. Adding a second harmonic trap between CPL and DIR ports (e.g., at the frequency FR shown in FIG. 2B) may force the amplitudes of the RFTCPL and RFTDIR signals to balance at both frequencies, FR and H2, as shown in FIG 2B. Amplitude balance at the second harmonic, H2, may reduce the second harmonic significantly with insignificant effect on amplitudes at the operating frequency, Hl.

[0033] According to an embodiment of the present disclosure, and as shown in the top amplitude response graphs of FIG. 2B, the second harmonic termination (220) may be configured to generate (realize, provide, create, etc.) a first (amplitude) notch at the DIR port having a first notch (minimum amplitude) point, MINDIR, at a first notch frequency, FNI, and a second (amplitude) notch at the CPL port having a second notch (minimum amplitude) point, MINCPL at a second notch frequency, FN2, wherein FNI is smaller than FN2. AS shown in FIG. 2B, the first and second notches may be defined by respective falling and rising (amplitude) portions that generate the respective first and second notch points, MINDIR and MINCPL. Because the notches “ride” on the native / inherent amplitude response curves (e.g., RFCPL, RFDIR of FIG. 2B) at the respective DIR and CPL ports, and because at higher frequencies (e.g., substantially above Hl and around H2) the amplitude at the DIR port is smaller than the amplitude at the CPL port, then as shown in FIG. 2B, the amplitude at the first notch point, MINDIR, is smaller than the amplitude at the second notch point, MINCPL. In turn, and due to the relatively narrow (e.g., 3dB) bandwidth of the first and second notches in combination with the distance between the first and second notch frequencies, FNI and FN2, a rising portion of the first notch intersects with a falling portion of the second notch at a first amplitude intersection frequency, FR. Furthermore, as shown in FIG. 2B, when the two notches gradually vanish at frequencies away from (greater than) the respective first and second notch frequencies, FNI and FN2, the inherent amplitude responses at the respective DIR port (e.g., decreasing amplitude) and CPL port (e.g., increasing amplitude) cause a second amplitude intersection frequency that can be made to be equal to the second harmonic, H2.

[0034] According to an embodiment of the present disclosure, the above described first and second notches may be realized via a resonant circuit that includes reactive elements, including at least one inductor and at least one capacitor. It follows that according to an embodiment of the present disclosure, the second harmonic termination (220) comprises a resonant circuit that includes at least one inductor and at least one capacitor. FIG. 3 A shows details of a second harmonic termination (220) according to an embodiment of the present disclosure, including an inductance, L220.

[0035] In particular, as shown on the right side of FIG. 3 A, the second harmonic termination (220) may include a first capacitor, Cl 220, the inductance, L220, and a second capacitor, Cl 220, that are arranged in series connection. The first capacitor, CI220, may be connected (e.g., directly connected) at a first terminal to the CPL port and at a second terminal to a first terminal of the inductance, L220, via a first node, Nl. On the other hand, the second capacitor, C2220, (e.g., directly connected) may be connected at a first terminal to the DIR port and at a second terminal to a second terminal of the inductance, L220, via a second node, N2.

[0036] According to an embodiment of the present disclosure, values of the capacitors CI220, and C2220, of the LC network (CI220, L220, C2220) shown in FIG. 3 A may be selected so to present respective high impedances at and about the center frequency of operation, Hl. In other words, an impedance of the LC network (Cl 220, L220, C2220) and therefore of the second harmonic termination (220) according to the present teachings, is an open circuit (e.g., decoupled via the capacitors CI220, and C2220) at the center frequency of operation, Hl, and accordingly may not (substantially) affect in-band signals at the CPL and DIR ports. Accordingly, as shown in the graphs of FIG. 2B, the second harmonic termination (220) may not (substantially) affect in-band amplitude and phase of the RF signal at about the center frequency of operation, Hl.

[0037] According to an embodiment of the present disclosure, values of the reactive elements CI220, L220, and C2220, of the LC network (CI220, L220, C2220) shown in FIG. 3 A may be selected to generate (a series) resonance at the frequency, FR, shown in FIG. 2B. In other words, the LC network (Cl 220, L220, C2220) resonates at the resonant frequency, FR. Becauseat the resonant frequency, FR, the impedance of the LC network (CI220, L220, C2220) is equal (e.g., substantially equal) to a short (e.g., real part and imaginary part of the impedance is equal to zero) then as shown in the top graphs of FIG. 2B, the amplitude at the CPL port (RFCPL) is equal to the amplitude at the DIR port (e.g., RFDIR) at the resonant frequency, FR. In addition, because at the resonant frequency, FR, the impedance of the LC network (CI220, L220, C2220) is equal to a short, then as shown in the bottom RFT graph of FIG. 2B, the phase difference between the CPL and DIR ports at the resonant frequency, FR. is zero. On the other hand, at the second harmonic, H2, the (purely reactive) impedance of the LC network may cause balancing of the amplitudes and phases between the CPL and DIR ports. Accordingly, the short at the resonant frequency, FR, may force the amplitude at the CPL port to decrease and the amplitude at the DIR port to increase versus frequency, so to create, as shown in FIG. 2B, another amplitude balance point at the second harmonic, H2.

[0038] According to an embodiment of the present disclosure, the resonant frequency, FR, may be selected to be about one third (i.e., 1 / 3) of the frequency Hl smaller than the frequency H2. In other words, a nominal value, FRN, of the resonant frequency, FR, may satisfy the equation: FRN = H2 - (1 / 3) x HL It should be noted that some latitude in the selection of the value of the resonant frequency, FR, may be possible in view of specific circuit implementations and optimization of the frequency response graphs shown in FIG. 2B. Accordingly, an optimized value, FRO, of the resonant frequency may be within a range of [0.9 x FRN, LI x FRN], or in other words, FRO = FRN + / - 0.1 x FRN. Simulation and optimization routines via methods and techniques known in the art may further be used to tweak respective shapes of the first and second notches, including respective positions of the first and second notch points, MINDIR and MINCPL, as well as corresponding notch bandwidths. According to an embodiment of the present disclosure, capacitance values of CI220 and C2220 are selected to be equal and such that insignificant effect on amplitudes at the operating frequency, Hl, is provided.

[0039] The inductance, L220, according to the present teachings may be realized by way of different inductive structures. Three such inductive structures are shown in FIG. 3B. These include an inductance, L220A, provided by a coil structure, TC; an inductance, L220B, provided by a transmission line (microstrip, delay line, conductor) DL having a length, LDL; and aninductance, L220C, provided by a coupled-line structure that similarly to the coupled-line structure described above with reference to FIG. IB, includes two equal length (e.g., LSCL) transmission lines (e.g., microstrips, conductors), CL3 and CL4, that are arranged in close proximity and parallel to one another.

[0040] Because the inductance, L220, according to the present teachings should provide the desired inductance at a broadband frequency range that encompasses the frequencies Hl and H2, different challenges may arise in the realization of the structures that provide such inductance. Such challenges may include, for example, realization of a relatively large inductance (e.g., few nanohenries or more) that may be substantially devoid of parasitic / stray components at the broadband frequency range, a sufficient small resistance so to minimize signal loss, and for some cases, a reduced form factor (physical area / footprint).

[0041] In the case of the inductance, L220A, of FIG. 3B realized by the coil structure, TC, by methods and techniques known in the art, the relatively large physical size, possible parasitic / stray components at the higher frequencies, as well as an inherent resistance of the coil structure, TC, may render such solution less desirable in some implementations.

[0042] In the case of the inductance, L220B, of FIG. 3B realized by the transmission line (microstrip, delay line, single conductor), DL, by methods and techniques known in the art, issues with the parasitic / stray components and inherent resistance related to the coil structure, TC, may be absent, overcome, or otherwise substantially reduced. However, the length, LDL, for provision of a desired (relatively large) inductance may become sufficiently large to render such solution less desirable in some cases. For example, to implement the required inductance for operation according to the exemplary frequencies (e.g., Hl = 28 GHz; H2 = 56 GHz) shown in the graphs of FIG. 2B, the length, LDL, of the transmission line, DL, may be substantially larger (e.g., by a factor of at last two, 1 / 4 versus 1 / 8 X the length, LCL, of the transmission lines, CL1 and CL2, used to realize the 90 degree hybrid coupler (e.g., 110) of FIG. 2A.

[0043] The above shortcomings for realization of the inductance, L220, may be overcome by the inductance, L220C, shown in FIG. 3B. According to an embodiment of the present disclosure, the inductance, L220C, is realized by a coupled-line structure, similar in structure to one described above with reference to FIG. IB, that includes two equal length (e.g., LSCL) transmission lines (e.g., microstrips, conductors), CL3 and CL4, that are arranged in close proximity and parallel to one another.

[0044] According to an embodiment of the present disclosure, functionality according to an inductance (e.g., of an inductor) of the coupled-line structure shown in FIG. 3B may be provided by shorting the respective CPL and DIR ports of the coupled-line structure (CL3, CL4) to obtain the inductance, L220C, between the respective IN and ISO ports. Shorting of the port DIR may cause the port IN to see a high impedance (open) at the port DIR and accordingly an RF signal coupled to the port IN may be (totally) reflected at the port DIR. Likewise, shorting of the port CPL may cause the port ISO to see a high impedance (open) at the port CPL and accordingly an RF signal coupled to the port ISO may be (totally) reflected at the port CPL. Accordingly, the inductive coupling provided by the relative arrangement of the two transmission lines, CL3 and CL4, of the coupled-line structure (CL3, CL4) couples the ports IN to the port ISO. In other words, the port IN is coupled to the port ISO via a mutual inductance of the coupled-line structure (CL3, CL4). It should be noted that for consistency reasons, nomenclatures for the ports of the coupled-line structure (CL3, CL4) are maintained according to the known in the art nomenclatures in spite of different functionalities provided by the couple-line structure (CL3, CL4) via shorting of the DIR and CPL ports. Accordingly, shorting the coupled and direct ports, CPL and DIR, may allow realization of a large inductor using a compact / short face-coupled line.

[0045] A circuital representation (arbitrary) of the inductance, L220C, is shown at the bottom of FIG. 4 A, including respective connections of the ports IN and ISO to the nodes N1 and N2. In the present disclosure, such circuit may be referred to as a “shorted coupled-line inductance” circuit or a “shorted coupled-line inductor” that by definition includes a coupled- line structure (CL3, CL4) with shorts at ports DIR and CPL, and a coupled inductance between ports IN and ISO. It should be noted that due to symmetry of operation of the shorted coupled-line inductance circuit, L220C, connections to nodes N1 and N2 may bereversed such that port IN may be connected to node N2 and port ISO may be connected to node Nl. It should be further noted that in contrast to the inductances L220A and L220B described above with reference to FIG. 3B, the shorted coupled-line inductance, L220C, does not provide a signal conduction path between nodes Nl and N2, rather such nodes are coupled to one another via inductive coupling between the lines CL3 and CL4.

[0046] With further reference to FIG. 4A, the coupled-line inductance circuit, L220C, according to the present disclosure may provide a relatively large (mutual) inductance between the ports IN and ISO by way of the relative arrangement of the face-coupled transmission lines CL3 and CL4 and their respective (metal) widths that may overlap along the length, LSCL. In turn, such increased coupling may allow for a more compact implementation of the inductance (e.g., L220 of FIG. 2A) for use in the second harmonic termination (e.g., 220 of FIG. 2A) when compared to other implementations described above with reference to FIG. 3B. It should be noted the length, LSCL, in combination with the relative distance between the transmission lines CL3 and CL4, and the width of the transmission lines CL3 and CL4, may provide a value of the inductance between the ports IN and ISO. For example, to implement the required inductance for operation according to the exemplary frequencies (e.g., Hl = 28 GHz; H2 = 56 GHz) shown in the graphs of FIG. 2B, the length, LSCL, of the transmission lines, CL3 and CL4, may be substantially smaller (e.g., by a factor of at least 2.5, 180 pm versus 480 pm) the length, LCL, of the transmission lines, CL1 and CL2, used to realize the 90 degrees hybris coupler (e.g., 110) of FIG. 2A. Accordingly, the compact coupled line configuration shown in FIG.4A that includes coupled and direct ports, CPL and DIR, that are shorted (to ground), functions as an inductor. The length, LSCL, is about 1 / 16 X.

[0047] FIG. 4B shows a circuital representation of the 90 degree hybrid coupler of FIG. 2A with the second harmonic termination (220) of FIG. 4A. As shown in FIG. 4B, the second harmonic termination (220) may include an LC network (CI220, L220C, C2220) that includes the above-described shorted coupled-line inductance circuit, L220C, to generate the abovedescribed resonant frequency, FR. Such configuration may be used, for example, to reduce / terminate second harmonic components that may be present in, for example, a balanced RF amplifier, as shown in FIG. 5 A.

[0048] FIG. 5A shows a circuital representation of a balanced amplifier (500A) according to an embodiment of the present disclosure. As shown in FIG. 5A, the balanced amplifier (500A) may include an input 90 degree hybrid coupler (110a) that splits an input RF signal, RFIN, at the respective IN port, to two equal amplitude / power quadrature signals at the respective DIR and CLP ports that are fed as respective inputs to (single-ended) amplifiers (530a) and (530b), wherein each such amplifier may include one or more cascaded stages (e.g., 530al, ..., 530bl, ...). Respective outputs of the amplifiers (530a) and (530b) in turn coupled / connected to respective CPL and DIR ports of an output 90 degree hybrid coupler (100b) that combines the signals art the respective CPL and DIR ports to the respective IN port, thereby resulting in an output RF signal, RFOUT, that is an amplified version of RFIN. Because the combination of the input 90 degree hybrid coupler (110a) and the amplifiers (530a, 530b) may generate and / or amplify second harmonic components of the operating frequency of the input RF signal, RFIN, the balanced amplifier (500) may include a second harmonic termination (220) coupled between the respective CPL and DIR ports of the output 90 degree hybrid coupler (110b). Although the second harmonic termination (220) of FIG. 5A is shown to include the shorted coupled-line inductance circuit, L220C, according to, e.g., FIG. 4B, other inductances, included inductances L220A and L220B described above with reference to FIG. 3B, may be used. Effects of the coupling of the second harmonic termination (220) to the output 90 degree hybrid coupler (110b) on a performance of the balanced amplifier (500A) is shown in the graphs of FIG. 5B.

[0049] FIG. 5B shows graphs representative of performance of the balanced amplifier (500A, graphs labelled H2T) of FIG. 5 A that includes a second harmonic termination (e.g., 220) according to the present teachings, further contrasted to a prior art balanced amplifier (graphs labelled / H2T) that excludes a second harmonic termination. As shown in the top AM-PM distortion graph and the bottom AM-AM distortion graph, inclusion of the second harmonic termination (e.g., 220) increases linearity performance of the balanced amplifier (500A) when contrasted to a traditional balanced amplifier that does not include the second harmonic termination.

[0050] It should be noted that teachings according to the present disclosure may not be limited to reducing effects of a second harmonic as other frequencies higher than the centerfrequency of operation (e.g., Hl), including higher order harmonics, may be targeted. In other words, component values of the second harmonic termination according to the present disclosure may be selected / optimized for operation (e.g., according to graphs of FIG. 2B) at other frequencies. According to some embodiment of the present disclosure, more than one second harmonic termination according to the present disclosure tuned at different frequencies (e.g., different resonant frequencies) may be applied concurrently across same two nodes (e.g., CPL and DIR ports of FIG. 2 A) or across different nodes of a circuit to terminate / reduce different targeted frequencies.

[0051] Furthermore, it should be noted that the shorted coupled-line inductance circuit (e.g., L220C) according to the present disclosure described above with reference to, e.g., FIG. 4A, may be used as a standalone inductance in any passive or active circuit. Such circuits may include, for example, active or passive filter circuits that may include one or more LC T- stages (e.g., stage (a) of FIG. 6) or Pi-stages (e.g., stage (b) of FIG. 6), or resonant circuits that may include one or more LC parallel stages (e.g., stage (c) of FIG. 6) or LC series stages (e.g., stage (d) of FIG. 6). Generally, such circuits can be any passive or active circuit (e.g., 680 of FIG. 6) that may include at least one active or passive device coupled / connected to the shorted coupled-line inductance circuit.

[0052] Applications that may include the novel apparatus and systems of various embodiments include electronic circuitry used in high-speed computers, communication and signal processing circuitry, modems, single or multi-processor modules, single or multiple embedded processors, data switches, and application-specific modules, including multilayer, multi-chip modules. Such apparatus and systems may further be included as sub-components within a variety of electronic systems, such as televisions, cellular telephones, personal computers (e.g., laptop computers, desktop computers, handheld computers, tablet computers, etc.), workstations, radios, video players, audio players (e.g., mp3 players), vehicles, medical devices (e.g., heart monitor, blood pressure monitor, etc.) and others. Some embodiments may include a number of methods.

[0053] The term “MOSFET” technically refers to metal-oxide-semiconductor-field-effect- transistors; another synonym for MOSFET is “MISFET”, for metal-insulator-semiconductorFET. However, “MOSFET” has become a common label for most types of insulated-gate FETs (“IGFETs”). Despite that, it is well known that the term “metal” in the names MOSFET and MISFET is now often a misnomer because the previously metal gate material is now often a layer of polysilicon (polycrystalline silicon). Similarly, the “oxide” in the name MOSFET can be a misnomer, as different dielectric materials are used with the aim of obtaining strong channels with smaller applied voltages. Accordingly, the term “MOSFET” as used herein is not to be read as literally limited to metal-oxide-semiconductor FETs, but instead includes IGFETs in general.

[0054] As should be readily apparent to one of ordinary skill in the art, various embodiments of the invention can be implemented to meet a wide variety of specifications. Unless otherwise noted above, selection of suitable component values is a matter of design choice and various embodiments of the invention may be implemented in any suitable IC technology (including but not limited to MOSFET and IGFET structures), or in hybrid or discrete circuit forms. Integrated circuit embodiments may be fabricated using any suitable substrates and processes, including but not limited to standard bulk silicon, silicon-on-insulator (SOI), silicon-on-sapphire (SOS), GaN HEMT, GaAs pHEMT, and MESFET technologies. However, the inventive concepts described above are particularly useful with an SOI-based fabrication process (including SOS), and with fabrication processes having similar characteristics. Fabrication in CMOS on SOI or SOS enables low power consumption, the ability to withstand high power signals during operation due to FET stacking, good linearity, and high frequency operation (in excess of about 10 GHz, and particularly above about 20 GHz). Monolithic IC implementation is particularly useful since parasitic capacitances generally can be kept low (or at a minimum, kept uniform across all units, permitting them to be compensated) by careful design.

[0055] Voltage levels may be adjusted or voltage and / or logic signal polarities reversed depending on a particular specification and / or implementing technology (e.g., NMOS, PMOS, or CMOS, and enhancement mode or depletion mode transistor devices). Component voltage, current, and power handling capabilities may be adapted as needed, for example, by adjusting device sizes, serially “stacking” components (particularly FETs) to withstand greater voltages, and / or using multiple components in parallel to handle greater currents.Additional circuit components may be added to enhance the capabilities of the disclosed circuits and / or to provide additional functions without significantly altering the functionality of the disclosed circuits.

[0056] The examples set forth above are provided to give those of ordinary skill in the art a complete disclosure and description of how to make and use the embodiments of the gate drivers for stacked transistor amplifiers of the disclosure and are not intended to limit the scope of what the applicant considers to be the invention. Such embodiments may be, for example, used within mobile handsets for current communication systems (e.g., WCDMA, LTE, WiFi, etc.) wherein amplification of signals with frequency content of above 100 MHz and at power levels of above 50 mW may be required. The skilled person may find other suitable implementations of the presented embodiments.

[0057] Modifications of the above-described modes for carrying out the methods and systems herein disclosed that are obvious to persons of skill in the art are intended to be within the scope of the following claims. All patents and publications mentioned in the specification are indicative of the levels of skill of those skilled in the art to which the disclosure pertains. All references cited in this disclosure are incorporated by reference to the same extent as if each reference had been incorporated by reference in its entirety individually.

[0058] It is to be understood that the disclosure is not limited to particular methods or systems, which can, of course, vary. It is also to be understood that the terminology used herein is for the purpose of describing particular embodiments only and is not intended to be limiting. As used in this specification and the appended claims, the singular forms “a”, “an”, and “the” include plural referents unless the content clearly dictates otherwise. The term "plurality" includes two or more referents unless the content clearly dictates otherwise. Unless defined otherwise, all technical and scientific terms used herein have the same meaning as commonly understood by one of ordinary skill in the art to which the disclosure pertains.

[0059] A number of embodiments of the disclosure have been described. Nevertheless, it will be understood that various modifications can be made without departing from the spirit and scope of the present disclosure. Accordingly, other embodiments are within the scope of the following claims.

Claims

CLAIMS1. A circuit comprising: a 90 degree hybrid coupler comprising an input port, a direct port, a coupled port and an isolated port, the isolated port coupled to ground through a termination resistor; and a resonant circuit including a first capacitor, a second capacitor, and an inductor in series connection, the inductor connected between the first capacitor and the second capacitor, the resonant circuit connected between the direct port and the coupled port, wherein the 90 degree hybrid coupler is configured for operation at a center frequency, and respective first and second capacitances of the first and second capacitor and inductance of the inductor are selected so that an impedance of the resonant circuit resonates at a resonant frequency that is about one third of the center frequency smaller than a second harmonic of the center frequency.

2. The circuit according to claim 1, wherein: the first capacitor is connected at a first terminal of the first capacitor to the direct port; the inductor is connected at a first terminal of the inductor to a second terminal of the first capacitor; and the second capacitor is connected, at a first terminal of the second capacitor to the coupled port, and at a second terminal of the second capacitor to a second terminal of the inductor.

3. The circuit according to claim 1 or claim 2, wherein: at the resonant frequency, the impedance of the resonant circuit is a short, and at the second harmonic, the impedance of the resonant circuit is purely reactive.

4. The circuit according to any one of claims 1-3, wherein: at the second harmonic, the impedance of the resonant circuit in combination with a frequency response characteristic of the 90 degree hybrid coupler provide balanced amplitudes between the direct port and coupled port.

5. The circuit according to any one of claims 1-3, wherein: at the second harmonic, the impedance of the resonant circuit in combination with a frequency response characteristic of the 90 degree hybrid coupler provide balanced amplitudes and phases between the direct port and coupled port.

6. The circuit according to any one of claims 1-5, wherein: the 90 degree hybrid coupler comprises a coupled-line structure that includes a first conductor line between the input port and the direct port, and a second conductor line between the isolated port and the coupled port.

7. The circuit according to claim 6, wherein: the first conductor line and the second conductor line have a same length that is based on a wavelength corresponding to the center frequency, and the first conductor line and the second conductor line are arranged at close proximity in parallel along respective lengths.

8. The circuit according to any one of claims 1-7, wherein: the inductor is a coil inductor.

9. The circuit according to any one of claims 1-7, wherein: the inductor is a conductor line inductor.

10. The circuit according to any one of claims 1-7, wherein: the inductor is a shorted coupled-line inductor whose inductance is provided via inductive coupling between respective first and second terminals of the inductor.

11. The circuit according to claim 10, wherein: the shorted coupled-line inductor comprises a coupled-line structure that includes two same lengths conductor lines arranged at close proximity in parallel along their respective lengths,a direct port and a coupled port of the coupled-line structure of the shorted coupled- line inductor are respectively shorted to ground, and an input port and an isolated port of the coupled-line structure of the shorted coupled- line inductor respectively provide the first and second terminals of the inductor.

12. The circuit according to claim 10, wherein: the inductance of the shorted coupled-line inductor is a mutual inductance between the two same lengths conductor lines.

13. A balanced amplifier configured for operation at a center frequency, comprising: an input 90 degree hybrid coupler comprising respective input port, direct port, coupled port and isolated port, the isolated port coupled to ground through a termination resistor; an output 90 degree hybrid coupler comprising respective input port, direct port, coupled port and isolated port, the isolated port coupled to ground through a termination resistor; first and second amplifiers comprising: respective first and second inputs respectively connected to the direct port and coupled port of the input 90 degree hybrid coupler; and respective first and second outputs respectively connected to the coupled port and the direct port of the output 90 degree hybrid coupler; and a resonant circuit including a first capacitor, a second capacitor, and an inductor in series connection, the inductor connected between the first capacitor and the second capacitor, the resonant circuit connected between the direct port and the coupled port of the output 90 degree hybrid coupler, wherein respective first and second capacitances of the first and second capacitor and inductance of the inductor are selected so that an impedance of the resonant circuit resonates at a resonant frequency that is greater than the center frequency and smaller than a second harmonic of the center frequency.

14. The balanced amplifier according to claim 13, wherein:the resonant frequency is about one third of the center frequency smaller than the second harmonic of the center frequency.

15. The balanced amplifier according to claim 13 or claim 14, wherein: the center frequency is in a range from about few GHz to about 300 GHz.

16. The balanced amplifier according to any one of claims 13-15, wherein: the inductor is a shorted coupled-line inductor whose inductance is provided via inductive coupling between respective first and second terminals of the inductor.

17. The circuit according to claim 16, wherein: the shorted coupled-line inductor comprises a coupled-line structure that includes two same lengths conductor lines arranged at close proximity in parallel along their respective lengths, a direct port and a coupled port of the coupled-line structure of the shorted coupled- line inductor are respectively shorted to ground, and an input port and an isolated port of the coupled-line structure of the shorted coupled- line inductor respectively provide the first and second terminals of the inductor.

18. The circuit according to claim 16, wherein: the inductance of the shorted coupled-line inductor is a mutual inductance between the two same lengths conductor lines.

19. A circuit, comprising: at least one active or passive device; and a shorted coupled-line inductor coupled to the at least one active or passive device at a first or second terminal of the shorted coupled-line inductor, wherein the shorted coupled-line inductor comprises a coupled-line structure that includes two same lengths conductor lines arranged at close proximity in parallel along their respective lengths,a direct port and a coupled port of the coupled-line structure of the shorted coupled-line inductor are respectively shorted to ground, and an input port and an isolated port of the coupled-line structure of the shorted coupled-line inductor respectively provide the first and second terminals of the inductor.

20. The circuit according to claim 19, wherein: an inductance of the shorted coupled-line inductor is a mutual inductance provided via inductive coupling between the two same lengths conductor lines.

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