A two-phase DC bias vernier motor controller and control method

Abstract

The present invention discloses a two-phase DC biased sinusoidal current motor controller and control method, comprising a first subtractor, a second subtractor, a third subtractor, a fourth subtractor, a speed regulator, a d-axis current regulator, a q-axis current regulator, a 0-axis current regulator, a pulse width modulator, a current transformer, a motor position sensor, an A-phase and a B-phase current sensor, and a current converter. After topology optimization, the present invention can reduce the drive resources to half, and the control signal that the control chip needs to output is also reduced to half of the original.

Description

Technical Field

[0001] The invention belongs to an AC motor drive control device, and specifically relates to a two-phase DC bias vernier motor controller and a control method, which can be used for the drive control of a novel two-phase DC bias vernier reluctance and vernier permanent magnet motor. Background Art

[0002] Permanent magnet motors have the excellent characteristics of low speed and high torque, but the high price of permanent magnet materials leads to high costs of permanent magnet motors, which seriously restricts their application in industrial occasions. Although the traditional switched reluctance motor has a simple structure and low cost, its unique operating mode determines that the motor has large noise and vibration, and the torque pulsation is also large. These defects affect the application of switched reluctance motors. In order to combine the advantages of the two motors, some scholars have proposed a three-phase DC biased vernier reluctance motor in recent years. However, the three-phase DC biased vernier motor requires too many power switching devices, resulting in high costs for the matching drive and control units. The structure of a two-phase motor is as follows: Figure 1 As shown, it includes 01 stator, 02 rotor, 03 winding, and other common structural parts such as shaft, housing, end cover, position sensor, etc. This type of motor can use single-layer fractional slot non-overlapping concentrated winding, and its winding contains two phases A and B. Each phase winding consists of two small phases. The connection method of the two small phases is as follows Figure 2 As shown. The current of each phase winding is shown in Figure 3(a), Figure 3(b) and Figure 3(c). The current of each phase winding consists of a DC component and an AC component. The DC component is used to construct an excitation magnetic field circuit, and the AC component is used to generate electromagnetic torque. When the DC biased sinusoidal current motor adopts the maximum torque current ratio control method, the amplitude of the DC excitation component must be kept equal to the amplitude of the AC component.

[0003] For this type of DC biased motor where each phase current contains both AC and DC components, the existing controller generally uses a single-phase full-bridge converter to control each phase current. The wiring method of one phase winding of a DC biased motor controller using a single-phase full-bridge converter is as follows: Figure 4 As shown, there are four turn-off power devices S 11 , S 12 , S 13 , S 14 The converter used in the controller occupies too many power electronic devices and controller resources. Summary of the invention

[0004] The purpose of the present invention is to provide a two-phase DC bias vernier motor controller and control method to solve the problem of excessive number of power electronic devices and controller resources occupied by the converter used in the existing controller. The present invention has a small size and weight, and in principle avoids the risk of bridge arm direct-through in the traditional structure, improves system reliability, and effectively reduces the number of power electronic devices used, especially the number of power devices used, while ensuring the performance of the controller, which can greatly reduce the cost of the controller and is suitable for the drive control of a new two-phase DC bias vernier reluctance motor.

[0005] In order to achieve the above object, the present invention adopts the following technical scheme:

[0006] The two-phase DC biased sinusoidal current motor controller provided by the present invention comprises a first subtractor, a second subtractor, a third subtractor, a fourth subtractor, a speed regulator, a d-axis current regulator, a q-axis current regulator, a 0-axis current regulator, a pulse width modulator, a converter, a motor position sensor, A-phase and B-phase current sensors, and a current converter.

[0007] Motor speed given value n r and the motor speed measurement value n are sent to the first subtractor, n r The difference between -n is sent to the speed regulator, and the speed regulator outputs the q-axis current given value i * q , by adjusting i q , so that n r The difference of -n is always zero, that is, n always follows n r The DC biased sinusoidal current motor adopts the strategy of setting the d-axis current to 0, and the 0-axis current setting is equal to the q-axis current setting, so the d-axis current setting i can be obtained. * d , q-axis current given value i * q , 0 axis current given value i * 0 ;

[0008] The motor speed measurement value is the rotor position signal θ measured by the motor position sensor. r After processing,

[0009] D-axis current given value i * d and the d-axis current feedback value i d Sent to the second subtractor, i * d -i d The difference is sent to the d-axis current regulator, and the d-axis current regulator outputs the d-axis voltage given value V * d, by adjusting V * d , so that i * d -i d The difference is always zero, that is, for i * d Conduct error-free tracking;

[0010] Q-axis current given value i * q and the q-axis current feedback value i q Sent to the third subtractor, i * q -i q The difference is sent to the q-axis current regulator, and the q-axis current regulator outputs the q-axis voltage given value V * q , by adjusting V * q , so that the difference is always zero, that is, for i * q Conduct error-free tracking;

[0011] 0 axis current given value i * 0 and 0-axis current feedback value i 0 Sent to the fourth subtractor, i * 0 -i 0 The difference is sent to the 0-axis current regulator, and the 0-axis current regulator outputs the 0-axis voltage given value V * 0 , by adjusting V * 0 , so that i * 0 -i 0 The difference is always zero, that is, for i * 0 Conduct error-free tracking;

[0012] V * d 、V * q 、V * 0 The pulse width modulator is input to perform space vector modulation to generate an A-phase PWM signal and a B-phase PWM signal respectively;

[0013] The converter is composed of a DC power supply, four MOS field effect transistors and five freewheeling diodes, wherein the drains of the first, second, third and fourth MOS field effect transistors are connected to the positive electrode of the DC power supply; the source of the first MOS field effect transistor is connected to the negative electrode of the first freewheeling diode and connected to one end of the A phase of the two-phase motor; the source of the third MOS field effect transistor is connected to the negative electrode of the third freewheeling diode and connected to the other end of the A phase of the two-phase motor; the source of the second MOS field effect transistor is connected to the negative electrode of the second freewheeling diode and connected to one end of the B phase of the two-phase motor; the source of the fourth MOS field effect transistor is connected to the negative electrode of the fourth freewheeling diode and connected to the other end of the B phase of the two-phase motor; the positive electrodes of the first, second, third, fourth and fifth freewheeling diodes are connected to the negative electrode of the DC power supply; the midpoint of the A and B phase windings of the motor is connected to the positive electrode of the fifth freewheeling diode; the A phase PWM signal is used as the control signal of the gate of the first and third MOS field effect transistors of the inverter, and the B phase PWM signal is used as the control signal of the gate of the second and fourth MOS field effect transistors of the inverter;

[0014] The output voltage of the converter acts on the DC biased sinusoidal current motor phase winding, controls the current of the A and B phases of the DC biased sinusoidal current motor winding, and generates a sinusoidal current signal with DC bias corresponding to the input PWM signal; finally, the error-free tracking of the d, q, and 0 axis currents is achieved;

[0015] The A-phase and B-phase current sensors respectively measure the two phase currents i of the A-phase A+ 、i A- , the two split-phase currents i of phase B B+ 、i B- , the motor rotor position signal θ detected by the motor position sensor r and A+ 、i A 、i B+ 、i B- The current is input into the current converter, and the current converter performs a stationary-rotating coordinate transformation to obtain the d-axis, q-axis, and 0-axis current signals, which are respectively used as the d-axis current feedback value i d , q-axis current i q , 0 axis current feedback value i 0 ;

[0016] The speed regulator outputs the q-axis current given value i* according to the following process q ;

[0017] The speed regulator judges its input, if n r -n difference is zero, then keep i* q is a fixed value and is output;

[0018] If r-n is positive, the q-axis current given value i* is continuously increased. q , until the motor speed feedback value n r Increase so that n r -n is zero, keep i* q is a constant value to maintain n r The difference of -n is always zero;

[0019] If r -n is negative, the q-axis current given value i* is continuously reduced. q , until the motor speed feedback value n r Reduce, so that n r -n is zero, keep i* q is a constant value to maintain n r The difference of -n is always zero;

[0020] The d-axis current regulator outputs the d-axis voltage given value V according to the following process * d ;

[0021] The d-axis current regulator judges its input. If i * d -i d The difference is zero, then keep V * d is a fixed value and is output;

[0022] If i * d -i d If the difference is positive, the d-axis voltage set value V is continuously increased. * d , until the d-axis current feedback value i d Increase, so that i * d -i d is zero, keep V * d is a constant value to maintain i * d -i d The difference is always zero;

[0023] If i * d -i d If the difference is negative, the d-axis voltage set value V is continuously reduced. * d , until the d-axis current feedback value i d Reduce, so that i * d -i d is zero, keep V *d is a constant value to maintain i * d -i d The difference is always zero;

[0024] The q-axis current regulator outputs a q-axis voltage given value V * q , 0 axis current regulator outputs 0 axis voltage given value V * 0 The process is the same as above, only the corresponding parameters need to be changed.

[0025] The pulse width modulator generates A and B phase PWM signals including the following process:

[0026] (A) The pulse width modulator is used to adjust the motor rotor position signal θ r and V * d 、V * q Perform rotation and static transformation to obtain V * a 、V * b ;

[0027]

[0028] Where V * a is the given value of the A-phase AC voltage in the stationary coordinate system, V * b is the given value of the B-phase AC voltage in the stationary coordinate system, and p is the number of pole pairs of the motor;

[0029] (B) The A-phase AC voltage duty cycle signal T generated by modulation a =V * a / V dc , B phase AC voltage duty cycle signal T b =V * b / V dc ; For V * 0 Output a DC voltage duty cycle signal T 0 =V * 0 / V dc , where V dc is the DC bus power supply voltage of the converter;

[0030] (C) Finally, the duty cycle of the A-phase PWM signal is obtained. A :T A =T a +T0 ;B phase PWM signal duty cycle T B :T B =T b +T 0 .

[0031] The current converter is used to convert the motor rotor position signal θ r and A+ 、i A- 、i B+ 、i B- Perform the stationary-rotating coordinate transformation as follows to obtain the d-axis current feedback value i d , q-axis current feedback value i q , 0-axis current feedback value i 0 is the phase current amplitude of phase A or phase B;

[0032]

[0033]

[0034] Where RME(i A+ ) is the RMS value of the A+ phase current.

[0035] The present invention can realize the optimization control strategy of the novel two-phase DC bias vernier reluctance motor, according to the DC bias sinusoidal motor electromagnetic torque formula: Where T e is the electromagnetic torque of the DC biased sinusoidal current motor, p is the number of motor poles, L dc is the motor 0-axis equivalent inductance, i 0 is the motor 0 axis current, i q is the motor q-axis current, and the effective value of the motor phase current is defined as: That is i d 、i q 、i 0 Resulting vector currents in three-dimensional space.

[0036] Compared with the prior art, the present invention has the following beneficial technical effects:

[0037] The converter of the present invention optimizes the traditional single-phase full-bridge converter topology structure. According to the single-phase current characteristics of the load motor, a turn-off power device of each half-bridge is replaced by a freewheeling diode. The freewheeling diode used needs to have a fast recovery characteristic. When the turn-off device performs a turn-off action, the freewheeling diode can quickly enter a conducting state to form a single-phase current control loop of the motor. Since each bridge arm of the motor unidirectional current control loop only includes one turn-off power device and one freewheeling diode, the danger of bridge arm direct conduction is physically avoided, and the bridge arm direct conduction problem caused by the traditional full-bridge structure converter is completely solved, thereby enhancing the reliability and stability of the system.

[0038] Compared with the traditional structure, in the power electronic devices constituting the converter of the present invention, the number of turn-off devices and diodes is reduced by half. Since the gate drive circuit driving the turn-off devices is correspondingly reduced by half, and the pulse width modulation output demand for digital signal processing used to generate PWM drive signals is also reduced synchronously, the overall number of devices required is less, the volume and weight of the system are reduced, and the power density of the system is improved.

[0039] In summary, the present invention has a small size and weight, and in principle avoids the risk of bridge arm direct-through in traditional structures, improves system reliability, and effectively reduces the number of power electronic devices used, especially the number of power devices used, while ensuring the performance of the controller. It can greatly reduce the cost of the controller and is suitable for the drive control of a new two-phase DC biased vernier reluctance motor. BRIEF DESCRIPTION OF THE DRAWINGS

[0040] Figure 1 is a schematic cross-sectional view of a two-phase DC biased vernier reluctance motor;

[0041] Figure 2 This is the wiring diagram of the A-phase winding of the two-phase DC bias vernier reluctance motor;

[0042] FIG3( a ) is a schematic diagram of the DC component of the phase current of a DC biased vernier reluctance motor;

[0043] FIG3( b ) is a schematic diagram of the AC component of the phase current of a DC biased vernier reluctance motor;

[0044] FIG3( c ) is a schematic diagram of the phase current of one phase of a DC biased vernier reluctance motor;

[0045] Figure 4 A schematic diagram of the connection mode of one-phase winding of an existing DC bias motor controller (single-phase full-bridge inverter topology structure);

[0046] Figure 5 It is a structural schematic diagram of the present invention;

[0047] Figure 6 It is a schematic diagram of the converter structure of the present invention.

[0048] Among them, 1, the first subtractor; 2, the second subtractor; 3, the third subtractor; 4, the fourth subtractor; 5, the speed regulator; 6d, the d-axis current regulator; 6q, the q-axis current regulator; 60, the 0-axis current regulator; 7, the pulse width modulator; 8, the converter; 9, the motor; 10, the motor position sensor; 11, the A-phase and B-phase current sensors; 12, the current converter. DETAILED DESCRIPTION

[0049] The present invention will be further described below in conjunction with the accompanying drawings:

[0050] The present invention makes the DC component in the DC bias circuit slightly larger than the AC component amplitude. At this time, the output torque decreases slightly, but it can ensure that the current direction of the motor remains unchanged during operation, and only the current magnitude changes. Figure 4 Device S 12 and S 13 It only plays the role of conducting and continuing current, and all four devices use pulse width modulation to output the command voltage. 12 and S 13 The winding current is not controlled and there is no need for a switch action, so it can be simplified to a freewheeling diode.

[0051] Through the above analysis, it can be found that the number of turn-off power devices of the motor controller is reduced from 4 to 2 per phase, and only one freewheeling diode is added per phase. The number of turn-off power devices required for the entire system is reduced from 12 to 4. Since the price of turn-off power devices is higher than that of freewheeling diodes, the system cost is greatly reduced.

[0052] The relevant issues are explained as follows: In the field of motor control, the natural coordinate system ABC (i.e., the three-phase stationary coordinate system) is often transformed into a two-phase stationary coordinate system (α-β coordinate system) and a synchronous rotating coordinate system dq0 (rotating coordinate system). The ABC axes are 120° electrical angle apart in space; the d, q, and 0 axes are perpendicular to each other in the three-dimensional dq0 space and rotate around the 0 axis at pω. r The speed of the motor rotates synchronously, where p is the number of motor poles, ω r is the motor mechanical angular velocity. In the present invention, since there are only two-phase windings, the transformation between the two-phase stationary coordinate system and the rotating coordinate system can be performed directly.

[0053] In a traditional topology converter, each phase winding needs to be configured with two PWM signals, which requires two sets of drive units and two control signals. After the topology optimization of the present invention, the drive resources can be reduced by half, and the control signal that the control chip needs to output is also reduced to half of the original.

[0054] like Figure 5 As shown, the present invention includes a first subtractor 1, a second subtractor 2, a third subtractor 3, a fourth subtractor 4, a speed regulator 5, a d-axis current regulator 6d, a q-axis current regulator 6q, a 0-axis current regulator 60, a pulse width modulator 7, a converter 8, a motor 9, a motor position sensor 10, an A-phase and a B-phase current sensor 11, and a current converter 12;

[0055] Motor speed given value n r and the motor speed measurement value n are sent to the first subtractor 1, n rThe difference between -n is sent to the speed regulator 5, and the speed regulator 5 outputs the q-axis current given value i * q , by adjusting i q , so that n r The difference of -n is always zero, that is, n always follows n r When the DC biased sinusoidal current motor is in the maximum torque current ratio working state, the d-axis current given value is 0, and the q-axis and 0-axis current given values ​​are equal, so the d-axis current given value i can be obtained. * d , q-axis current given value i * q , 0 axis current given value i * 0 ;

[0056] The motor speed measurement value is a rotor position signal θ measured by the motor position sensor 10. r After processing, it is obtained by

[0057] get;

[0058] D-axis current given value i * d and the d-axis current feedback value i d Sent to the second subtractor 2, i * d -i d The difference is sent to the d-axis current regulator 6d, and the d-axis current regulator 6d outputs the d-axis voltage given value V * d , by adjusting V * d , so that i * d -i d The difference is always zero, that is, for i * d Conduct error-free tracking;

[0059] Q-axis current given value i * q and the q-axis current feedback value i q Sent to the third subtractor 3, i * q -i q The difference is sent to the q-axis current regulator 6q, and the q-axis current regulator 6q outputs the q-axis voltage given value V * q , by adjusting V * q , so that i * q -i qThe difference is always zero, that is, for i * q Conduct error-free tracking;

[0060] 0 axis current given value i * 0 and 0-axis current feedback value i 0 Sent to the fourth subtractor 4, i * 0 -i 0 The difference is sent to the 0-axis current regulator 60, and the 0-axis current regulator 60 outputs the 0-axis voltage given value V * 0 , by adjusting V * 0 , so that i * 0 -i 0 The difference is always zero, that is, for i * 0 Conduct error-free tracking;

[0061] V * d 、V * q 、V * 0 The pulse width modulator 7 is input to perform space vector modulation to generate an A-phase PWM signal and a B-phase PWM signal respectively;

[0062] like Figure 6 As shown, the converter 8 is composed of a DC power supply, four MOS field effect transistors and five freewheeling diodes, wherein the drains of the first MOS field effect transistor S1, the second MOS field effect transistor S2, the third MOS field effect transistor S3 and the fourth MOS field effect transistor S4 are connected to the positive electrode of the DC power supply; the source of the first MOS field effect transistor S1 is connected to the cathode of the first freewheeling diode D1 and connected to one end A of the A phase of the two-phase motor; the source of the third MOS field effect transistor S3 is connected to the cathode of the third freewheeling diode D3 and connected to the other end A of the A phase of the two-phase motor The source of the second MOS field effect transistor S2 is connected to the cathode of the second freewheeling diode D2, and is connected to the other end B of the B phase of the two-phase motor; the source of the fourth MOS field effect transistor S4 is connected to the cathode of the third freewheeling diode D4, and is connected to the other end Y of the B phase of the two-phase motor; the positive electrodes of the first freewheeling diode D1, the second freewheeling diode D2, the third freewheeling diode D3, and the fourth freewheeling diode D4 are connected to the cathode of the DC power supply; the cathode of the fifth freewheeling diode D5 is connected to the cathode of the DC power supply, and the positive electrode is connected to the midpoint A of the A and B phase windings of the two-phase motor N , B NThe A-phase PWM signal is used as a control signal for the gates of the first MOS field effect transistor S1 and the third MOS field effect transistor S3 of the inverter, and the B-phase PWM signal is used as a control signal for the gates of the second MOS field effect transistor S2 and the fourth MOS field effect transistor S4 of the inverter;

[0063] like Figure 5 As shown, the output voltage of the converter 8 acts on the DC bias sinusoidal current motor phase winding, controls the current of the A and B phases of the DC bias vernier reluctance motor 9 winding, and generates a sinusoidal current signal with DC bias corresponding to the input PWM signal; finally, the d, q, and 0 axis currents are tracked without difference;

[0064] The A-phase and B-phase current sensors 11 respectively measure two phase current signals i of the A-phase. A+ 、i A- , B phase two phase current signal i B+ 、i B- , the motor rotor position signal θ detected by the motor position sensor 10 r and A+ 、i A- 、i B+ 、i B- The current is input into the current converter, and the current converter 12 performs a stationary-rotating coordinate transformation to obtain d, q, and 0 axis current signals, which are respectively used as d axis current feedback values ​​i d , q-axis current i q , 0 axis current feedback value i 0 ;

[0065] The speed regulator 5 outputs the q-axis current given value i* according to the following process q ;

[0066] The speed regulator 5 judges its input. If n r -n difference is zero, then keep i* q is a fixed value and is output;

[0067] If r -n is positive, the q-axis current given value i* is continuously increased. q , until the motor speed feedback value n r Increase so that n r -n zero, keep i* at this time q is a constant value to maintain n r The difference of -n is always zero;

[0068] If r -n is negative, the q-axis current given value i* is continuously reduced. q , until the motor speed feedback value n rReduce, so that n r -n is zero, keep i* q is a constant value to maintain n r The difference of -n is always zero;

[0069] The d-axis current regulator outputs the d-axis voltage given value V according to the following process * d ;

[0070] The d-axis current regulator 6d judges its input. If i * d -i d The difference is zero, then keep V * d is a fixed value and is output;

[0071] If i * d -i d If the difference is positive, the d-axis voltage set value V is continuously increased. * d , until the d-axis current feedback value i d Increase, so that i * d -i d is zero, keep V * d is a constant value to maintain i * d -i d The difference is always zero;

[0072] If i * d -i d If the difference is negative, the d-axis voltage set value V is continuously reduced. * d , until the d-axis current feedback value i d Reduce, so that i * d -i d is zero, keep V * d is a constant value to maintain i * d -i d The difference is always zero;

[0073] The q-axis current regulator 6q outputs a q-axis voltage given value V * q , 0 axis current regulator 60 outputs 0 axis voltage given value V * 0 The process is the same as above, only the corresponding parameters need to be changed.

[0074] The pulse width modulator 7 generates the A and B phase PWM signals including the following process:

[0075] (A) The pulse width modulator 7 is used to control the motor rotor position signal θ r and V * d 、V * q 、V * d0 Rotation and static transformation, get V * a 、V * b ;

[0076]

[0077] in is the given value of the A-phase AC voltage in the stationary coordinate system, V * b is the given value of the B-phase AC voltage in the stationary coordinate system, and p is the number of pole pairs of the motor;

[0078] (B) The A-phase AC voltage duty cycle signal T generated by modulation a =V * a / V dc , B phase AC voltage duty cycle signal T b =V * b / V dc ; For V * 0 Output a DC voltage duty cycle signal T 0 =V * 0 / V dc , where V dc is the DC bus power supply voltage of the converter;

[0079] (C) Finally, the duty cycle of the A-phase PWM signal is obtained. A :T A =T a +T 0 ;B phase PWM signal duty cycle T B :T B =T b +T 0 .

[0080] The current converter 12 is used to convert the motor rotor position signal θ r and A+ 、i A- 、i B+ 、i B-Perform the stationary-rotating coordinate transformation as follows to obtain the d-axis current feedback value i d ,q axial flow feedback value i q , 0-axis current feedback value i 0 is the current amplitude of phase A or phase B;

[0081]

[0082]

[0083] The present invention is suitable for motor drive applications that are sensitive to cost control, including but not limited to consumer-grade washing machines, air conditioners, vacuum cleaners, refrigerators and other household appliances drive motors, lawn mowers and low-speed electric vehicle drive motors. These applications often do not have particularly high requirements for motor performance and control accuracy, but considering commercial production, they are extremely sensitive to cost. A low-cost, high-reliability DC bias vernier motor drive system is a good choice.

Claims

1. A two-phase DC bias vernier motor controller, characterized in that: The invention comprises a first subtractor (1), a second subtractor (2), a third subtractor (3), a fourth subtractor (4), a speed regulator (5), a d-axis current regulator (6d), a q-axis current regulator (6q), a 0-axis current regulator (60), a pulse width modulator (7), a current converter (8), a motor (9), a motor position sensor (10), A-phase and B-phase current sensors (11) and a current converter (12); The first subtractor (1) is used to receive the motor speed given value n r and the motor speed measurement value n, and the difference n r -n is sent to the speed regulator (5), the output end of the speed regulator (5) is connected to the d-axis current regulator (6d), the q-axis current regulator (6q) and the 0-axis current regulator (60) respectively through the second subtractor (2), the third subtractor (3) and the fourth subtractor (4), the output ends of the d-axis current regulator (6d), the q-axis current regulator (6q) and the 0-axis current regulator (60) are all connected to the pulse width modulator (7), the output end of the pulse width modulator (7) is connected to the converter (8), the converter (8) The output end of the motor (8) is connected to the motor (9), the motor (9) is connected to a motor position sensor (10), and A-phase and B-phase current sensors (11) are also connected between the converter (8) and the motor (9), the output ends of the A-phase and B-phase current sensors (11) are respectively connected to the second subtractor (2), the third subtractor (3) and the fourth subtractor (4) through the current converter (12), and the output end of the motor position sensor (10) is respectively connected to the first subtractor (1), the pulse width modulator (7) and the current converter (12); The converter (8) is composed of a DC power supply, a first MOS field effect transistor S1, a second MOS field effect transistor S2, a third MOS field effect transistor S3, a fourth MOS field effect transistor S4, a first freewheeling diode D1, a second freewheeling diode D2, a third freewheeling diode D3, a fourth freewheeling diode D4 and a fifth freewheeling diode D5; The drains of the first MOS field effect transistor S1, the second MOS field effect transistor S2, the third MOS field effect transistor S3, and the fourth MOS field effect transistor S4 are connected to the positive electrode of the DC power supply; the source of the first MOS field effect transistor S1 is connected to the negative electrode of the first freewheeling diode D1, and is connected to the A end of the A phase winding of the motor (9); the source of the second MOS field effect transistor S2 is connected to the negative electrode of the second freewheeling diode D2, and is connected to the B end of the B phase winding of the motor (9); the source of the third MOS field effect transistor S3 is connected to the negative electrode of the third freewheeling diode D1, and is connected to the B end of the B phase winding of the motor (9); The cathode of the first MOS field effect transistor S4 is connected to the cathode of the fourth freewheeling diode D4 and connected to the X end of the A phase winding of the motor (9); the source of the fourth MOS field effect transistor S4 is connected to the cathode of the fourth freewheeling diode D4 and connected to the Y end of the B phase winding of the motor (9); the anodes of the first freewheeling diode D1, the second freewheeling diode D2, the third freewheeling diode D3 and the fourth freewheeling diode D4 are connected to the cathode of the DC power supply, and the cathode of the fifth freewheeling diode D5 is connected to the cathode of the DC power supply; the anode of the fifth freewheeling diode D5 is connected to the midpoint A of the A and B phase windings of the motor (9); N , B N The A-phase PWM signal generated by the pulse width modulator (7) is used as a control signal for the gates of the first MOS field effect transistor S1 and the third MOS field effect transistor S3, and the B-phase PWM signal generated by the pulse width modulator (7) is used as a control signal for the gates of the second MOS field effect transistor S2 and the fourth MOS field effect transistor S4.

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