Methods and Systems for Continuous Glucose Monitoring

Abstract

A method for optionally externally calibrating a calibration-free glucose sensor uses the value of the measured working electrode current (Isig) and EIS data to calculate a final sensor glucose (SG) value. The counter electrode voltage (Vcntr) can also be used as an input. The raw Isig and Vcntr values can be preprocessed, and low-pass filtering, averaging, and / or feature generation can be applied. One or more models for predicting SG calculations can be used to generate the SG value. Complex redundancy can be employed to take advantage of the operational benefits of the different characteristics of two or more different or dissimilar sensors, the characteristics including, for example, those related to the hydration, stability, and durability of such sensors. A fusion algorithm, EIS, and an application-specific integrated circuit (ASIC) can be used to implement the use of such redundant glucose sensors, devices, and sensor systems in a manner that can bridge the gap between fast startup, sensor lifetime, and calibration-free algorithm accuracy.

Description

Technical Field

[0001] Embodiments of the present invention generally relate to sensor technology, including sensors and sensor devices for sensing various physiological parameters such as glucose concentration. More specifically, embodiments of the present invention relate to optional calibration in calibration-free systems, devices, and methods, and the use of complex redundancy in glucose sensors, devices, and sensor systems including closed-loop insulin infusion systems, as well as fusion algorithms, electrochemical impedance spectroscopy (EIS), and application-specific integrated circuits (ASICs) for implementing the use of such redundant glucose sensors, devices, and sensor systems. Background Art

[0002] Subjects (e.g., patients) and medical personnel desire to monitor readings of the physiological condition within a subject. Illustratively, a subject desires to continuously monitor the blood glucose level within the subject. Currently, a patient can use a BG measurement device (i.e., a blood glucose meter) such as a test strip meter, a continuous glucose measurement system (or continuous glucose monitor), or a hospital hemacue to measure his / her blood glucose (BG). BG measurement devices use various methods to measure the patient's BG level, such as using a blood sample from the patient, a sensor in contact with body fluid, an optical sensor, an enzyme sensor, or a fluorescence / fluorescence quenching sensor. When a BG measurement device has generated a BG measurement value, the measurement value is displayed on the BG measurement device.

[0003] Infusion pump devices and systems are relatively well known in the medical field for delivering or dispensing prescription drugs such as insulin to a patient. In one form, such devices include a relatively compact pump housing adapted to receive a syringe or reservoir carrying a prescription drug for administration to the patient through an infusion tube and an associated catheter or infusion set. A programmable controller can operate the infusion pump continuously or at periodic intervals to obtain tightly controlled and accurate drug delivery over an extended period of time. Such infusion pumps are used for administering insulin and other drugs, and exemplary pump configurations are shown and described in U.S. Pat. Nos. 4,562,751; 4,678,408; 4,685,903; 5,080,653; and 5,097,122, which are incorporated herein by reference.

[0004] Everyone has a basal insulin requirement, which in diabetic individuals can typically be maintained by continuously or intermittently administering a basal amount of insulin to the patient using an infusion pump. However, when additional glucose (i.e., above basal levels) is present in a diabetic individual, such as when the individual consumes a meal, the amount and timing of insulin to be administered must be determined in order to adequately account for the additional glucose while avoiding over-infusion of insulin. Typically, a large dose of insulin is administered to compensate for the meal (i.e., a meal bolus). Diabetic patients typically determine the amount of insulin they may need to cover an anticipated meal based on the carbohydrate content of the meal.

[0005] In recent years, a variety of electrochemical glucose sensors have been developed for obtaining an indication of the blood glucose level of a diabetic patient. Such readings are useful for monitoring and / or adjusting a treatment regimen that typically includes the periodic administration of insulin to the patient. Typically, small and flexible electrochemical sensors can be used to obtain periodic readings over an extended period of time. In one form, a flexible subcutaneous sensor is constructed according to thin-film shielding technology. Typical thin-film sensors are described in co-owned U.S. Patents Nos. 5,390,671; 5,391,250; 5,482,473; and 5,586,553, which are incorporated herein by reference.

[0006] These electrochemical sensors have been applied to a telemetry-based characteristic monitoring system. As described, for example, in co-owned U.S. Patent No. 6,809,653 (the " '653 patent"), which is incorporated herein by reference in its entirety, the telemetry system includes a remotely located data receiving device, a sensor for generating a signal indicative of a user characteristic, and a transmitter device for processing the signal received from the sensor and for wirelessly transmitting the processed signal to the remotely located data receiving device. The data receiving device can be a characteristic monitor, a data receiver that provides data to another device, an RF programmer, a drug delivery device (such as an infusion pump), etc.

[0007] Current continuous glucose measurement systems include subcutaneous (or short-term) sensors and implantable (or long-term) sensors. For each of the short-term and long-term sensors, the patient must wait a certain amount of time for the continuous glucose sensor to stabilize and provide an accurate reading. In many continuous glucose sensors, the subject must wait three hours for the continuous glucose sensor to stabilize before any glucose measurements can be made. This is inconvenient for the patient and, in some cases, may cause the patient not to utilize the continuous glucose measurement system.

[0008] In addition, when a glucose sensor is first inserted into a patient's skin or subcutaneous layer, the glucose sensor does not operate in a steady state. The sensor electrical readings representing the patient's glucose level vary over a wide range of readings. In the past, sensor stabilization often took several hours. For example, the techniques for sensor stabilization are described in detail in the '653 patent, where the initialization process for sensor stabilization can be reduced to about one hour. A high voltage (e.g., 1.0 to 1.2 volts) can be applied for 1 to 2 minutes to stabilize the sensor, and then a low voltage (e.g., 0.5 to 0.6 volts) can be applied for the remaining time of the initialization process (e.g., about 58 minutes).

[0009] It is also necessary to fully "wet" or hydrate the electrodes of the sensor before using the electrodes of the sensor. If the electrodes of the sensor are not fully hydrated, the result may be inaccurate readings of the patient's physiological condition. Current users of blood glucose sensors can be instructed not to power on the sensors immediately. If these sensors are utilized too early, such blood glucose sensors may not operate in an optimal or efficient manner.

[0010] Many of the prior arts of continuous glucose monitoring (CGM) are mainly auxiliary, which means that the readings provided by CGM devices (including, for example, implantable or subcutaneous sensors) cannot be used to make clinical decisions without a reference value. The reference value must in turn be obtained from a finger stick using, for example, a BG meter. The reason for the need for a reference value is that the amount of information that can be obtained from the sensor / sensing assembly is limited. Specifically, the sensing assembly may only provide the raw sensor value (i.e., the sensor current or Isig) and a voltage for processing. Therefore, during analysis, if the raw sensor signal seems abnormal (e.g., if the signal is decreasing), the only way to distinguish between a sensor failure and a physiological change (i.e., a change in glucose level) in the user / patient's body may be to obtain a reference glucose value by finger stick. It is well known that reference finger sticks are also used to calibrate the sensor.

[0011] The art has sought ways to eliminate or at least minimize the number of finger sticks required to calibrate and assess the health of the sensor. However, given the number and complexity level of multiple sensor failure modes, no satisfactory solution has been found. Only diagnostic methods based on a direct assessment of Isig or a comparison of two Isigs have been developed. In either case, since Isig tracks the glucose level in the human body, Isig is clearly not analyte-independent. Therefore, Isig itself is not a reliable source of information for sensor diagnosis nor a reliable predictor for the sensor to continue to perform.

[0012] Another limitation that has existed in the art to date is the lack of sensor electronics that can not only operate the sensors, but also perform real-time sensor and electrode diagnostics, and can diagnose redundant electrodes, redundant sensors, complementary sensors, and redundant and complementary sensors, while managing the power supply of the sensors. To be sure, the concept of electrode redundancy has been around for some time. However, in the past, using electrode redundancy (and / or complementary and redundant electrodes) has made it nearly impossible to obtain more than one reading at a time, and has made it nearly impossible to assess the relative health of the redundant electrodes, the overall reliability of the sensors, and how often, if at all, a calibration reference value is needed.

[0013] In addition, even when redundant sensing electrodes have been used, the number has typically been limited to two. Again, this partly comes down to the lack of advanced electronics that can operate, assess, and manage multiple independently operating electrodes (e.g., up to 5 or more) in real time. However, another reason is the following limited view: using redundant electrodes to obtain "independent" sensor signals, and for this purpose, two redundant electrodes are sufficient. As noted, while this is one function of using redundant electrodes, it is not the only function. SUMMARY OF THE INVENTION

[0014] According to an embodiment of the present invention, a method for optionally externally calibrating a calibration-free glucose sensor for measuring glucose levels in a user's body, wherein the glucose sensor includes physical sensor electronics, a microcontroller, and working electrodes, the method comprising: periodically measuring, by the physical sensor electronics, an electrode current (Isig) signal of the working electrodes; performing, by the microcontroller, an electrochemical impedance spectroscopy (EIS) procedure to generate EIS-related data of the working electrodes; calculating, by the microcontroller, a corresponding sensor glucose (SG) value for each of the SG prediction models based on the Isig signal, the EIS-related data, and a plurality of calibration-free SG prediction models; calculating, by the microcontroller, an SG variance estimate value for each of the corresponding SG values; determining, by the microcontroller, whether an external blood glucose (BG) value is available, and when available, incorporating the BG value into the calculation of the SG values; fusing, by the microcontroller, the corresponding SG values from the plurality of SG prediction models to obtain a single fused SG value; applying, by the microcontroller, an unscented Kalman filter to any of the SG values; and calculating, by the microcontroller, a calibrated SG value to be displayed to the user.

[0015] According to other embodiments of the present invention, a glucose monitoring system includes a glucose sensor device for determining the glucose concentration in a user's body during a total sensor device wear time, where the total sensor device wear time includes a first time window, a subsequent second time window, and a transition period between the first time window and the second time window, and the glucose sensor device includes a first glucose sensor and a second glucose sensor, where the first glucose sensor and the second glucose sensor have different characteristics in at least one of hydration, stability, and durability. The glucose sensor device further includes sensor electronics, where the sensor electronics includes at least one physical microprocessor configured to: (a) periodically receive a corresponding first output signal from the first glucose sensor indicating the glucose concentration level in the user's body; (b) calculate the glucose concentration level in the user's body during the first time window based entirely on the first output signal; (c) periodically receive a corresponding second output signal from the second glucose sensor indicating the glucose concentration level in the user's body; (d) calculate the glucose concentration level during the transition period based on both the first output signal and the second output signal; and (e) calculate the glucose concentration level in the user's body during the second time window based entirely on the second output signal. BRIEF DESCRIPTION OF THE DRAWINGS

[0016] Embodiments of the present invention will be described in detail with reference to the drawings, in which like numbers represent corresponding parts in the figures.

[0017] Figure 1 is a perspective view of a subcutaneous sensor insertion device and a block diagram of a sensor electronics device according to an embodiment of the present invention.

[0018] Figure 2A shows a substrate having two sides, a first side containing an electrode configuration and a second side containing electronic circuitry.

[0019] Figure 2B shows an overall block diagram of an electronic circuit for sensing the output of a sensor.

[0020] Figure 3 shows a block diagram of a sensor electronics device and a sensor including a plurality of electrodes according to an embodiment of the present invention.

[0021] Figure 4 shows an alternative embodiment of the present invention including a sensor and a sensor electronics device according to an embodiment of the present invention.

[0022] Figure 5 shows an electronic block diagram of a sensor electrode according to an embodiment of the present invention and the voltage applied to the sensor electrode.

[0023] Figure 6A Shows a method of applying a pulse during a stabilization time frame to reduce the stabilization time frame according to an embodiment of the present invention.

[0024] Figure 6B Shows a method of stabilizing a sensor according to an embodiment of the present invention.

[0025] Figure 6C Shows the utilization of feedback when stabilizing a sensor according to an embodiment of the present invention.

[0026] Figure 7 Shows the effect of stabilizing a sensor according to an embodiment of the present invention.

[0027] Figure 8A Shows a block diagram of a sensor electronic device and a sensor including a voltage generating device according to an embodiment of the present invention.

[0028] Figure 8B Shows a voltage generating device for implementing this embodiment of the present invention.

[0029] Figure 8C Shows a voltage generating device for generating two voltage values according to an embodiment of the present invention.

[0030] Figure 8D Shows a voltage generating device having three voltage generating systems according to an embodiment of the present invention.

[0031] Figure 9A Shows a sensor electronic device including a microcontroller for generating voltage pulses according to an embodiment of the present invention.

[0032] Figure 9B Shows a sensor electronic device including an analysis module according to an embodiment of the present invention.

[0033] Figure 10 Shows a block diagram of a sensor system including a hydrated electronic device according to an embodiment of the present invention.

[0034] Figure 11 Shows an embodiment of the present invention including a mechanical switch for assisting in determining the hydration time.

[0035] Figure 12 Shows a method of detecting hydration according to an embodiment of the present invention.

[0036] Figure 13A Shows a method of hydrating a sensor according to an embodiment of the present invention.

[0037] Figure 13B Shows an additional method for verifying the hydration of a sensor according to an embodiment of the present invention.

[0038] Figure 14A , Figure 14B and Figure 14C illustrate a method of combining hydration of a sensor with stabilization of the sensor, according to an embodiment of the present invention.

[0039] Figure 15A illustrate an EIS-based analysis of the system response to the application of a periodic AC signal, according to an embodiment of the present invention.

[0040] Figure 15B illustrate a known circuit model for electrochemical impedance spectroscopy.

[0041] Figure 16A illustrate an example of a Nyquist plot, according to an embodiment of the present invention, where an AC voltage plus a DC voltage (DC bias) is applied to a working electrode for a selected frequency spectrum from 0.1 Hz to 1000 MHz.

[0042] Figure 16B illustrate another example of a Nyquist plot having a linear fit for relatively low frequencies and an intercept approaching the real impedance value at relatively high frequencies.

[0043] Figure 16C and Figure 16D illustrate the infinite and finite glucose sensor responses to a sinusoidal working potential, respectively.

[0044] Figure 16E illustrate a Bode plot of magnitude, according to an embodiment of the present invention.

[0045] Figure 16F illustrate a Bode plot of phase, according to an embodiment of the present invention.

[0046] Figure 17 illustrate a Nyquist plot of sensor impedance that changes as the sensor ages, according to an embodiment of the present invention.

[0047] Figure 18 illustrate a method of applying EIS technology in stabilizing a sensor and detecting the sensor lifetime, according to an embodiment of the present invention.

[0048] Figure 19 illustrate a schedule for performing an EIS procedure, according to an embodiment of the present invention.

[0049] Figure 20 illustrate a method of using an EIS procedure in combination with a remedial measure to detect and repair a sensor, according to an embodiment of the present invention.

[0050] Figure 21A and Figure 21BAn example of a sensor remedy according to an embodiment of the present invention is shown.

[0051] Figure 22 A Nyquist plot of a normally operating sensor is shown, where the Nyquist slope gradually increases and the intercept gradually decreases as the sensor wear time progresses.

[0052] Figure 23A An example of the raw current signal (Isig) from two redundant working electrodes and the corresponding real impedance of the electrodes at 1 kHz according to an embodiment of the present invention is shown.

[0053] Figure 23B Shows Figure 23A The Nyquist plot of the first working electrode (WE1).

[0054] Figure 23C Shows Figure 23A The Nyquist plot of the second working electrode (WE2).

[0055] Figure 24 An example of a signal drop of two redundant working electrodes and the corresponding real impedance of the electrodes at 1 kHz according to an embodiment of the present invention is shown.

[0056] Figure 25A An example of the substantially glucose-independent real impedance, imaginary impedance, and phase of a normally operating glucose sensor at a relatively high frequency according to an embodiment of the present invention is shown.

[0057] Figure 25B An illustrative example of the level of change in glucose correlation of the real impedance at a relatively low frequency according to an embodiment of the present invention is shown.

[0058] Figure 25C An illustrative example of the level of change in glucose correlation of the phase at a relatively low frequency according to an embodiment of the present invention is shown.

[0059] Figure 26 An example of the trends of the 1 kHz real impedance, 1 kHz imaginary impedance, and relatively high frequency phase when a glucose sensor loses sensitivity due to hypoxia at the sensor insertion site according to an embodiment of the present invention is shown.

[0060] Figure 27 An example of Isig and phase of in vitro simulated hypoxia at different glucose concentrations according to an embodiment of the present invention is shown.

[0061] Figures 28A to 28C An example of the sensitivity loss caused by hypoxia and the EIS-based parameters of the electrodes in the case of redundant working electrodes WE1 and WE2 according to an embodiment of the present invention is shown.

[0062] Figure 28D shows the EIS-induced spikes in the raw Isig for an example of Figures 28A to 28C .

[0063] Figure 29 shows an example of the sensitivity loss caused by hypoxia due to occlusion according to an embodiment of the present invention.

[0064] Figures 30A to 30C shows an example of the sensitivity loss caused by bio-fouling, as well as the redundant working electrodes WE1 and WE2 and the EIS-based parameters of the electrodes, according to an embodiment of the present invention.

[0065] Figure 30D shows the EIS-induced spikes in the raw Isig for an example of Figures 30A to 30C .

[0066] Figure 31 shows a diagnostic program for sensor fault detection according to an embodiment of the present invention.

[0067] Figure 32A and Figure 32B shows another diagnostic program for sensor fault detection according to an embodiment of the present invention.

[0068] Figure 33A shows a top-level flowchart of a fusion algorithm based on current (Isig) according to an embodiment of the present invention.

[0069] Figure 33B shows a top-level flowchart of a fusion algorithm based on sensor glucose (SG) according to an embodiment of the present invention.

[0070] Figure 34 shows the details of the sensor glucose (SG)-based fusion algorithm of Figure 33B according to an embodiment of the present invention.

[0071] Figure 35 shows the details of the current (Isig)-based fusion algorithm of Figure 33A according to an embodiment of the present invention.

[0072] Figure 36 is an illustration of the calibration of a sensor in a steady state according to an embodiment of the present invention.

[0073] Figure 37 is an illustration of the calibration of a sensor in transition according to an embodiment of the present invention.

[0074] Figure 38AIllustration of EIS-based dynamic slope (and slope adjustment) according to an embodiment of the present invention for sensor calibration.

[0075] Figure 38B Shows an EIS-assisted sensor calibration flowchart according to an embodiment of the present invention involving low-start detection.

[0076] Figure 39 Shows sensor current (Isig) and 1 kHz impedance magnitude for in vitro simulation of an interferent very close to the sensor according to an embodiment of the present invention.

[0077] Figure 40A and Figure 40B respectively show the Bode plots for Figure 39 the phase and impedance of the simulation shown.

[0078] Figure 40C Shows the Nyquist plot for Figure 39 the simulation shown.

[0079] Figure 41 Shows another in vitro simulation of an interferent according to an embodiment of the present invention.

[0080] Figure 42A and Figure 42B Shows an ASIC block diagram according to an embodiment of the present invention.

[0081] Figure 43 Shows the potentiostat configuration of a sensor with redundant working electrodes according to an embodiment of the present invention.

[0082] Figure 44 Shows the equivalent AC inter-electrode circuit of a sensor with the Figure 43 potentiostat configuration shown.

[0083] Figure 45 Shows some main blocks of the EIS circuitry in the analog front-end IC of a glucose sensor according to an embodiment of the present invention.

[0084] Figures 46A to 46F Shows the Figure 45 simulation of the current for which the signal of the EIS circuitry shown is multiplied by 0 degrees phase with 0 degrees phase.

[0085] Figures 47A to 47F Shows the Figure 45 simulation of the current for which the signal of the EIS circuitry shown is multiplied by 0 degrees phase with 90 degrees phase.

[0086] Figure 48 Shows a circuit model according to an embodiment of the present invention.

[0087] Figures 49A to 49C Shows a diagram of a circuit model according to an alternative embodiment of the present invention.

[0088] Figure 50A Is a Nyquist plot of a superposition equivalent circuit simulation according to an embodiment of the present invention.

[0089] Figure 50B Is Figure 50A An enlarged view of the high-frequency part of

[0090] Figure 51 Shows a Nyquist plot in which Cdl increases in the direction of arrow A according to an embodiment of the present invention.

[0091] Figure 52 Shows a Nyquist plot in which α increases in the direction of arrow A according to an embodiment of the present invention.

[0092] Figure 53 Shows a Nyquist plot in which Rp increases in the direction of arrow A according to an embodiment of the present invention.

[0093] Figure 54 Shows a Nyquist plot in which the Warburg admittance increases in the direction of arrow A according to an embodiment of the present invention.

[0094] Figure 55 Shows a Nyquist plot in which λ increases in the direction of arrow A according to an embodiment of the present invention.

[0095] Figure 56 Shows the influence of the membrane capacitance on the Nyquist plot according to an embodiment of the present invention.

[0096] Figure 57 Shows a Nyquist plot in which the membrane resistance increases in the direction of arrow A according to an embodiment of the present invention.

[0097] Figure 58 Shows a Nyquist plot in which Rsol increases in the direction of arrow A according to an embodiment of the present invention.

[0098] Figures 59A to 59C Shows the variation of EIS parameters related to circuit elements during startup and calibration according to an embodiment of the present invention.

[0099] Figures 60A to 60C Shows the variation of a different set of EIS parameters related to circuit elements during startup and calibration according to an embodiment of the present invention.

[0100] Figures 61A to 61C Shows the variation of another different set of EIS parameters related to circuit elements during startup and calibration according to an embodiment of the present invention.

[0101] Figure 62 Shows the EIS response for multiple electrodes according to an embodiment of the present invention.

[0102] Figure 63 Is a Nyquist plot showing the effect of Isig calibration by increasing glucose according to an embodiment of the present invention.

[0103] Figure 64 Shows the effect of the oxygen (Vcntr) response on the Nyquist plot according to an embodiment of the present invention.

[0104] Figure 65 Shows the shift in the Nyquist plot due to temperature change according to an embodiment of the present invention.

[0105] Figure 66 Shows the relationship between Isig and blood glucose according to an embodiment of the present invention.

[0106] Figures 67A to 67B Shows sensor drift according to an embodiment of the present invention.

[0107] Figure 68 Shows the increase in membrane resistance during sensitivity loss according to an embodiment of the present invention.

[0108] Figure 69 Shows the decrease in Weber admittance during sensitivity loss according to an embodiment of the present invention.

[0109] Figure 70 Shows a calibration curve according to an embodiment of the present invention.

[0110] Figure 71 Shows the high-frequency semi-circle that becomes visible on the Nyquist plot according to an embodiment of the present invention.

[0111] Figure 72A and Figure 72B Shows the Vcntr drop (rail) and the decrease in Cdl according to an embodiment of the present invention.

[0112] Figure 73 Shows the changing slope of the calibration curve according to an embodiment of the present invention.

[0113] Figure 74 Shows the changing length of the Nyquist plot according to an embodiment of the present invention.

[0114] Figure 75 Shows Figure 74 An enlarged view of the low-frequency and high-frequency regions of the Nyquist plot of

[0115] Figure 76A and Figure 76BShows the combined effect of increased film resistance, decreased Cdl, and decreased Vcntr according to an embodiment of the present invention.

[0116] Figure 77 Shows the relative Cdl values of two working electrodes according to an embodiment of the present invention.

[0117] Figure 78 Shows the relative Rp values of two working electrodes according to an embodiment of the present invention.

[0118] Figure 79 Shows the combined effect of varying EIS parameters on a calibration curve according to an embodiment of the present invention.

[0119] Figure 80 Shows that, according to an embodiment of the present invention, the length of the Nyquist plot in the low-frequency region is longer than in the region where there is a sensitivity loss.

[0120] Figure 81 Is a flowchart of sensor self-calibration based on the detection of sensitivity changes according to an embodiment of the present invention.

[0121] Figure 82 Shows the horizontal shift in the Nyquist plot due to sensitivity loss according to an embodiment of the present invention.

[0122] Figure 83 Shows a method for developing a heuristic EIS metric based on the Nyquist plot according to an embodiment of the present invention.

[0123] Figure 84 Shows the relationship between Rm and the calibration factor according to an embodiment of the present invention.

[0124] Figure 85 Shows the relationship between Rm and the normalized Isig according to an embodiment of the present invention.

[0125] Figure 86 Shows the Isig curves of various glucose levels changing over time according to an embodiment of the present invention.

[0126] Figure 87 Shows the Cdl curves of various glucose levels changing over time according to an embodiment of the present invention.

[0127] Figure 88 Shows according to an embodiment of the present invention Figure 86 of the second inflection point of the graph.

[0128] Figure 89 Shows according to an embodiment of the present invention corresponding to Figure 88 the second inflection point of Rm for the peak in.

[0129] Figure 90 A graph showing the relationship between the calibration factor (CF) and Rmem+Rsol according to an embodiment of the present invention.

[0130] Figure 91A A graph showing the in vivo results of MARD for all valid BGs within approximately the first 8 hours of sensor life according to an embodiment of the present invention.

[0131] Figure 91B A graph showing the median ARD count for all valid BGs within approximately the first 8 hours of sensor life according to an embodiment of the present invention.

[0132] Figures 92A to 92C Shows the calibration factor adjustment according to an embodiment of the present invention.

[0133] Figures 93A to 93C Shows the calibration factor adjustment according to an embodiment of the present invention.

[0134] Figures 94A to 94C Shows the calibration factor adjustment according to an embodiment of the present invention.

[0135] Figure 95 Shows an illustrative example of the initial decay of Cdl according to an embodiment of the present invention.

[0136] Figure 96 Shows the elimination of the influence of the illegal Faraday current on Isig according to an embodiment of the present invention.

[0137] Figure 97A Shows the calibration factor before removing the illegal Faraday current of two working electrodes according to an embodiment of the present invention.

[0138] Figure 97B Shows the calibration factor after removing the illegal Faraday current of two working electrodes according to an embodiment of the present invention.

[0139] Figure 98A and Figure 98B Shows the elimination of the influence of the illegal Faraday current on MARD according to an embodiment of the present invention.

[0140] Figure 99 A graph of the double-layer capacitance over time according to an embodiment of the present invention.

[0141] Figure 100 Shows the shift of Rmem+Rsol and the appearance of a high-frequency semi-circle during sensitivity loss according to an embodiment of the present invention.

[0142] Figure 101A Shows a flowchart for detecting sensitivity loss using combinational logic according to an embodiment of the present invention.

[0143] Figure 101B Shows a flowchart for detecting sensitivity loss using combinational logic according to another embodiment of the present invention.

[0144] Figure 102 Shows an illustrative method for using the Nyquist slope as a marker to distinguish between a new sensor and an old sensor according to an embodiment of the present invention.

[0145] Figures 103A to 103C Shows an illustrative example of Nyquist plots with different lengths for different sensor configurations according to an embodiment of the present invention.

[0146] Figure 104 Shows for Figures 103A to 103C the sensor, the Nyquist plot length varying with time.

[0147] Figure 105 Shows a flowchart for blanking sensor data or terminating a sensor according to an embodiment of the present invention.

[0148] Figure 106 Shows a flowchart for sensor termination according to an embodiment of the present invention.

[0149] Figure 107 Shows a flowchart for signal dip detection according to an embodiment of the present invention.

[0150] Figure 108A Shows Isig and Vcntr varying with time, and Figure 108B Shows glucose varying with time according to an embodiment of the present invention.

[0151] Figure 109A is the calibration ratio varying with time, and Figure 109B Shows glucose varying with time according to an embodiment of the present invention.

[0152] Figure 110A and Figure 110B Shows the calibration factor trend varying with time according to an embodiment of the present invention.

[0153] Figure 111 Shows a flowchart for first day calibration (FDC) according to an embodiment of the present invention.

[0154] Figure 112 Shows a flowchart for EIS-based calibration according to an embodiment of the present invention.

[0155] Figure 113 Shows a flowchart of an existing calibration method.

[0156] Figure 114Shows a calibration flow chart according to an embodiment of the present invention.

[0157] Figure 115 Shows a calibration flow chart according to other embodiments of the present invention.

[0158] Figure 116 Shows a calibration flow chart according to still other embodiments of the present invention.

[0159] Figure 117 Shows a calibration flow chart according to other embodiments of the present invention.

[0160] Figure 118 Shows a table of comparative MARD values calculated based on embodiments of the present invention.

[0161] Figure 119 Shows a flow chart for calculating the original fusion weight according to an embodiment of the present invention.

[0162] Figure 120 Shows a sensor glucose (SG) fusion logic diagram according to an embodiment of the present invention.

[0163] Figure 121 Shows a flow chart of a calibration-free backtracking algorithm according to an embodiment of the present invention.

[0164] Figure 122 Shows a decision tree model according to an embodiment of the present invention.

[0165] Figure 123 Shows a decision tree model for blanking data according to an embodiment of the present invention.

[0166] Figure 124 Is a table showing examples of parameters for a blanking algorithm according to an embodiment of the present invention.

[0167] Figure 125 Shows the fusion, filtering, and blanking results according to an embodiment of the present invention.

[0168] Figure 126 Shows a flow chart of optional calibration logic according to an embodiment of the present invention.

[0169] Figure 127 Shows a comparison table between two different glucose sensor designs.

[0170] Figure 128 Shows an example of complex redundancy according to an embodiment of the present invention.

[0171] Figure 129 Shows a block diagram including a calibrated model and an uncalibrated model according to an embodiment of the present invention.

[0172] Figure 130 A diagram showing the fusion logic according to an embodiment of the present invention.

[0173] Figure 131 A diagram showing the fusion logic with one calibrated model and one uncalibrated model according to an embodiment of the present invention.

[0174] Figure 132 A diagram showing the fusion logic with two uncalibrated models according to an embodiment of the present invention.

[0175] Figure 133 A diagram showing the fusion logic with two calibrated models according to an embodiment of the present invention.

[0176] Figure 134 A diagram showing the fusion logic with multiple calibrated models and / or multiple uncalibrated models according to an embodiment of the present invention. Detailed Description

[0177] In the following description, reference is made to the accompanying drawings which form a part hereof and show several embodiments of the present invention. It should be understood that other embodiments may be utilized and structural and operational changes may be made without departing from the scope of the present invention.

[0178] The present invention will be described below with reference to flowcharts of methods, systems, apparatuses, devices, and programming and computer program products. It should be understood that each block of the flowchart illustrations and combinations of blocks in the flowchart illustrations can be implemented by programming instructions, which include computer program instructions (such as any menu screen described in the figures). These computer program instructions can be loaded onto a computer or other programmable data processing device (such as a controller, microcontroller, or processor in a sensor electronic device) to produce a machine such that the instructions executed on the computer or other programmable data processing device produce instructions for implementing the functions specified in one or more flowchart blocks. These computer program instructions can also be stored in a computer-readable memory, which can direct the computer or other programmable data processing device to act in a particular manner such that the instructions stored in the computer-readable memory produce an article of manufacture that includes instructions for implementing the functions specified in one or more flowchart blocks. The computer program instructions can also be loaded onto a computer or other programmable data processing device such that a series of operational steps are performed on the computer or other programmable device to produce a computer-implemented process such that the instructions executed on the computer or other programmable device provide steps for implementing the functions specified in one or more flowchart blocks, and / or the menus presented herein. The programming instructions can also be stored in and / or implemented by an electronic circuit system that includes integrated circuits (ICs) and application-specific integrated circuits (ASICs) used in conjunction with sensor devices, equipment, and systems.

[0179] Figure 1 is a perspective view of a subcutaneous sensor insertion device and a block diagram of a sensor electronic device according to an embodiment of the present invention. As Figure 1 shown, a subcutaneous sensor set 10 is provided for subcutaneously placing an active portion of a flexible sensor 12 (e.g., see FIG. 2) and the like at a selected site within a user's body. The subcutaneous or transcutaneous portion of the sensor set 10 includes a hollow slotted insertion needle 14 and a cannula 16. The needle 14 is used to facilitate the quick and easy subcutaneous placement of the cannula 16 at a subcutaneous insertion site. The sensing portion 18 of the sensor 12 is within the cannula 16, and the sensing portion is used to expose one or more sensor electrodes 20 to a user's body fluid through a window 22 formed in the cannula 16. In an embodiment of the present invention, one or more sensor electrodes 20 can include a counter electrode, a reference electrode, and one or more working electrodes. After insertion, the insertion needle 14 is withdrawn to leave the cannula 16, and the sensing portion 18 and the sensor electrodes 20 remain in place at the selected insertion site.

[0180] In a particular embodiment, subcutaneous sensor set 10 facilitates precise placement of flexible thin-film electrochemical sensor 12, which is of the type for monitoring specific blood parameters indicative of a user's condition. Sensor 12 monitors glucose levels in the body and can be used in conjunction with an external or implantable type of automatic or semi-automatic drug infusion pump, as described in U.S. Patent Nos. 4,562,751; 4,678,408; 4,685,903 or 4,573,994, to control insulin delivery to a diabetic patient.

[0181] A particular embodiment of flexible electrochemical sensor 12 is constructed in accordance with thin-film shielding technology to include an elongated thin-film conductor embedded or encapsulated between a selected insulating material (such as a polyimide film or polyimide sheet) and a membrane. When the sensing portion 18 (or active portion) of sensor 12 is placed subcutaneously at the insertion site, the sensor electrode 20 at the tip of the sensing portion 18 is exposed through one of the insulating layers to make direct contact with the patient's blood or other body fluid. The sensing portion 18 is joined to a connecting portion 24, which terminates in a conductive contact pad or the like, which is also exposed through one of the insulating layers. In alternative embodiments, other types of implantable sensors may be used, such as chemically-based sensors, optically-based sensors, and the like.

[0182] As is known in the art, connecting portion 24 and the contact pads are generally adapted for direct wired electrical connection to a suitable monitor or sensor electronics 100 to monitor the user's condition in response to signals obtained from sensor electrode 20. Further description of this general type of flexible thin-film sensor can be found, for example, in U.S. Patent No. 5,391,250, which is incorporated herein by reference. Connecting portion 24 can be conveniently electrically connected to monitor or sensor electronics 100, or connected through a connector block 28 (or the like) as shown and described in U.S. Patent No. 5,482,473, which is also incorporated herein by reference. Thus, according to an embodiment of the present invention, subcutaneous sensor set 10 can be configured or shaped to work with a wired or wireless characteristic monitoring system.

[0183] Sensor electrode 20 can be used in a variety of sensing applications and can be configured in a variety of ways. For example, sensor electrode 20 can be used in physiological parameter sensing applications where some type of biomolecule serves as a catalyst. For example, sensor electrode 20 can be used in a glucose and oxygen sensor having glucose oxidase (GOx) that catalyzes a reaction with sensor electrode 20. The reaction produces gluconic acid (C6H 12 O7) and hydrogen peroxide (H2O2) that is proportional to the amount of glucose present.

[0184] The sensor electrode 20 and biomolecules or some other catalyst can be placed within a human body in a vascular or non-vascular environment. For example, the sensor electrode 20 and the biomolecules can be placed in a vein and subjected to blood flow, or can be placed in a subcutaneous or peritoneal region of the human body.

[0185] The monitor 100 can also be referred to as the sensor electronic device 100. The monitor 100 can include a power supply 110, a sensor interface 122, processing electronics 124, and data formatting electronics 128. The monitor 100 can be coupled to the sensor set 10 via a cable 102 through a connector of a connector block 28 that is electrically coupled to the connection portion 24. In an alternative embodiment, the cable can be omitted. In this embodiment of the present invention, the monitor 100 can include a connector adapted to be directly connected to the connection portion 104 of the sensor set 10. The sensor set 10 can be modified to locate the connector portion 104 at a different location (e.g., located above the sensor set) to facilitate placement of the monitor 100 above the sensor set.

[0186] In an embodiment of the present invention, the sensor interface 122, the processing electronics 124, and the data formatting electronics 128 are formed as separate semiconductor chips. However, alternative embodiments can combine the various semiconductor chips into a single or multiple custom semiconductor chips. The sensor interface 122 is connected to the cable 102, which is connected to the sensor set 10.

[0187] The power supply 110 can be a battery. The battery can include three silver oxide 357 batteries in series. In alternative embodiments, different battery chemistries can be used, such as lithium-based chemistries, alkaline batteries, nickel metal hydride, etc., and different numbers of batteries can be used. The monitor 100 provides power to the sensor set via the power supply 110 through the cable 102 and the cable connector 104. In an embodiment of the present invention, the power is a voltage provided to the sensor set 10. In an embodiment of the present invention, the power is a current provided to the sensor set 10. In an embodiment of the present invention, the power is a voltage provided to the sensor set 10 at a specific voltage.

[0188] Figure 2A and 2B An implantable sensor and electronics for driving the implantable sensor according to an embodiment of the present invention are shown. Figure 2A A substrate 220 having two sides is shown, with a first side 222 containing an electrode configuration and a second side 224 containing an electronic circuitry. As in Figure 2AAs can be seen, the first side 222 of the substrate includes two counter - working electrode pairs 240, 242, 244, 246 on opposite sides of the reference electrode 248. The second side 224 of the substrate includes an electronic circuitry. As shown, the electronic circuitry can be enclosed in a hermetically sealed housing 226, thereby providing a protective housing for the electronic circuitry. This allows the sensor substrate 220 to be inserted into a vascular environment or other environments where the electronic circuitry may be exposed to fluids. By sealing the electronic circuitry in the hermetically sealed housing 226, the electronic circuitry can operate without the risk of being short - circuited by the surrounding fluid. Figure 2A Also shown are pads 228 to which the input and output lines of the electronic circuitry can be connected. The electronic circuitry itself can be fabricated in a variety of ways. According to an embodiment of the present invention, the electronic circuitry can be fabricated as an integrated circuit using techniques commonly used in the industry.

[0189] Figure 2B A general block diagram of an electronic circuit for sensing the output of a sensor according to an embodiment of the present invention is shown. At least one pair of sensor electrodes 310 can interface with a data converter 312, and the output of the data converter can interface with a counter 314. The counter 314 can be controlled by control logic 316. The output of the counter 314 can be connected to a line interface 318. The line interface 318 can be connected to input and output lines 320 and can also be connected to the control logic 316. The input and output lines 320 can also be connected to a power rectifier 322.

[0190] The sensor electrodes 310 can be used in a variety of sensing applications and can be configured in a variety of ways. For example, the sensor electrodes 310 can be used in physiological parameter sensing applications where some types of biomolecules act as catalysts. For example, the sensor electrodes 310 can be used in a glucose and oxygen sensor having glucose oxidase (GOx) that catalyzes a reaction with the sensor electrodes 310. The sensor electrodes 310 along with the biomolecule or some other catalyst can be placed in the human body in a vascular or non - vascular environment. For example, the sensor electrodes 310 and the biomolecule can be placed in a vein and subjected to blood flow.

[0191] Figure 3A block diagram of a sensor electronic device and a sensor including a plurality of electrodes according to an embodiment of the present invention is shown. The sensor group or system 350 includes a sensor 355 and a sensor electronic device 360. The sensor 355 includes a counter electrode 365, a reference electrode 370, and a working electrode 375. The sensor electronic device 360 includes a power supply 380, a regulator 385, a signal processor 390, a measurement processor 395, and a display / transmission module 397. The power supply 380 supplies power (in the form of voltage, current, or voltage including current) to the regulator 385. The regulator 385 transmits the regulated voltage to the sensor 355. In an embodiment of the present invention, the regulator 385 transmits the voltage to the counter electrode 365 of the sensor 355.

[0192] The sensor 355 generates a sensor signal indicative of the concentration of the physiological characteristic being measured. For example, the sensor signal may indicate a blood glucose reading. In an embodiment of the present invention using a subcutaneous sensor, the sensor signal may represent the hydrogen peroxide level in a subject's body. In an embodiment of the present invention using a blood or cranial sensor, the amount of oxygen is measured by the sensor and represented by the sensor signal. In an embodiment of the present invention using an implantable or long-term sensor, the sensor signal may represent the oxygen level in a subject's body. The sensor signal may be measured at the working electrode 375. In an embodiment of the present invention, the sensor signal may be a current measured at the working electrode. In an embodiment of the present invention, the sensor signal may be a voltage measured at the working electrode.

[0193] After the sensor signal (e.g., a measured current or voltage) is measured at the sensor 355 (e.g., the working electrode), the signal processor 390 receives the sensor signal. The signal processor 390 processes the sensor signal and generates a processed sensor signal. The measurement processor 395 receives the processed sensor signal and calibrates the processed sensor signal using a reference value. In an embodiment of the present invention, the reference value is stored in a reference memory and provided to the measurement processor 395. The measurement processor 395 generates a sensor measurement value. The sensor measurement value may be stored in a measurement memory (not shown). The sensor measurement value may be sent to a display / transmission device to be displayed on a display in a housing together with the sensor electronics, or transmitted to an external device.

[0194] The sensor electronic device 360 can be a monitor that includes a display for presenting readings of physiological characteristics. The sensor electronic device 360 can also be installed in a desktop computer, a pager, a television with communication capabilities, a laptop computer, a server, a network computer, a personal digital assistant (PDA), a portable telephone with computer capabilities, an infusion pump with a display, a glucose sensor with a display, and / or a combined infusion pump / glucose sensor. The sensor electronic device 360 can be housed in a cellular telephone, a smart phone, a network device, a home network device, and / or other devices connected to a home network.

[0195] Figure 4 An alternative embodiment that includes a sensor and a sensor electronic device is shown. The sensor group or sensor system 400 includes the sensor electronic device 360 and the sensor 355. The sensor includes a counter electrode 365, a reference electrode 370, and a working electrode 375. The sensor electronic device 360 includes a microcontroller 410 and a digital-to-analog converter (DAC) 420. The sensor electronic device 360 can also include a current-to-frequency converter (I / F converter) 430.

[0196] The microcontroller 410 includes software program code or programmable logic that, when executed, causes the microcontroller 410 to transmit a signal to the DAC 420, where the signal represents a voltage level or voltage value to be applied to the sensor 355. The DAC 420 receives the signal and generates a voltage value at the level indicated by the microcontroller 410. In an embodiment of the invention, the microcontroller 410 can change the representation of the voltage level in the signal frequently or infrequently. Illustratively, the signal from the microcontroller 410 can indicate that the DAC 420 applies a first voltage value for one second and a second voltage value for two seconds.

[0197] The sensor 355 can receive a voltage level or a voltage value. In an embodiment of the present invention, the counter electrode 365 can receive the output of an operational amplifier that has a reference voltage and a voltage value from the DAC 420 as inputs. The application of the voltage level causes the sensor 355 to generate a sensor signal indicative of the concentration of the physiological characteristic being measured. In an embodiment of the present invention, the microcontroller 410 can measure the sensor signal (e.g., a current value) from the working electrode. Illustratively, the sensor signal measurement circuit 431 can measure the sensor signal. In an embodiment of the present invention, the sensor signal measurement circuit 431 can include a resistor, and current can flow through the resistor to measure the value of the sensor signal. In an embodiment of the present invention, the sensor signal can be a current level signal, and the sensor signal measurement circuit 431 can be a current-to-frequency (I / F) converter 430. The current-to-frequency converter 430 can measure the sensor signal based on the current reading, convert the sensor signal into a frequency-based sensor signal, and transmit the frequency-based sensor signal to the microcontroller 410. In an embodiment of the present invention, the microcontroller 410 can more easily receive the frequency-based sensor signal compared to a non-frequency-based sensor signal. The microcontroller 410 receives the frequency-based or non-frequency-based sensor signal and determines the value of the physiological characteristic of the subject, such as the blood glucose level. The microcontroller 410 can include program code that, when executed or run, is capable of receiving the sensor signal and converting the sensor signal into a physiological characteristic value. In one embodiment, the microcontroller 410 can convert the sensor signal into a blood glucose level. In some embodiments, the microcontroller 410 can utilize the measured values stored in the internal memory to determine the blood glucose level of the subject. In some embodiments, the microcontroller 410 can utilize the measured values stored in a memory external to the microcontroller 410 to assist in determining the blood glucose level of the subject.

[0198] After the microcontroller 410 determines the physiological characteristic value, the microcontroller 410 can store the measured values of the physiological characteristic value over several time periods. For example, the blood glucose value can be sent from the sensor to the microcontroller 410 every second or every five seconds, and the microcontroller can save the sensor measurements for five or ten minutes of BG readings. The microcontroller 410 can transmit the measured values of the physiological characteristic value to a display on the sensor electronics 360. For example, the sensor electronics 360 can be a monitor that includes a display providing a blood glucose reading of the subject. In one embodiment, the microcontroller 410 can transmit the measured values of the physiological characteristic value to the output interface of the microcontroller 410. The output interface of the microcontroller 410 can transmit the measured values of the physiological characteristic value, such as the blood glucose value, to an external device, such as an infusion pump, a combined infusion pump / blood glucose meter, a computer, a personal digital assistant, a pager, a network device, a server, a cellular phone, or any computing device.

[0199] Figure 5 An electronic block diagram shows a sensor electrode and a voltage applied to the sensor electrode according to one embodiment. In Figure 5 the embodiment shown, an operational amplifier 530 or other servo control device may be connected to the sensor electrode 510 through a circuit / electrode interface 538. The operational amplifier 530 utilizes feedback through the sensor electrode and attempts to maintain a specified voltage between the reference electrode 532 and the working electrode 534 by adjusting the voltage at the counter electrode 536 (the DAC may desire that the applied voltage be the specified voltage). Current may then flow from the counter electrode 536 to the working electrode 534. This current can be measured to determine the electrochemical reaction between the sensor electrode 510 and a biomolecule of a sensor placed near the sensor electrode 510 and used as a catalyst. Figure 5 The circuit system disclosed herein can be used in long-term or implantable sensors, or can be used in short-term or subcutaneous sensors.

[0200] In an embodiment of a long-term sensor, in the case where the glucose oxidase (GOx) enzyme is used as a catalyst in the sensor, current can flow from the counter electrode 536 to the working electrode 534 only when oxygen is present near the enzyme and the sensor electrode 510. Illustratively, if the voltage set at the reference electrode 532 is maintained at approximately 0.5 volts, then the amount of current flowing from the counter electrode 536 to the working electrode 534 has a fairly linear relationship with the unit slope of the amount of oxygen present in the region surrounding the enzyme and the electrode. Therefore, by maintaining the reference electrode 532 at approximately 0.5 volts and using this region of the current-voltage curve to vary the blood oxygen level, the accuracy of determining the amount of oxygen in the blood can be improved. Different embodiments may utilize different sensors with biomolecules other than glucose oxidase, and thus a voltage other than 0.5 volts may be set at the reference electrode.

[0201] As discussed above, during the initial implantation or insertion of the sensor 510, the sensor 510 may provide inaccurate readings due to the subject's adjustment of the sensor and electrochemical by-products generated by the catalyst used in the sensor. Many sensors require a stabilization period in order for the sensor 510 to provide accurate readings of the subject's physiological parameters. During the stabilization period, the sensor 510 does not provide accurate blood glucose measurements. The users and manufacturers of the sensor may desire to improve the stabilization time frame of the sensor so that the sensor can be quickly utilized after being inserted into the subject's body or subcutaneous layer.

[0202] In prior sensor electrode systems, the stabilization period or time frame could be in the range of one hour to three hours. To reduce the stabilization period or time frame and improve the timeliness of the sensor's accuracy, the sensor (or the electrodes of the sensor) can be subjected to several pulses instead of applying one pulse and then applying another voltage. Figure 6A A method of applying pulses during the stabilization time frame to reduce the stabilization time frame is shown. In this embodiment, the voltage application device applies a first voltage of 600 to the electrode at a first time or time period. In one embodiment, the first voltage can be a DC constant voltage. This causes an anodic current to be generated. In an alternative embodiment, a digital-to-analog converter or another voltage source can supply voltage to the electrode during the first time period. An anodic current means that electrons are driven towards the electrode to which the voltage is applied. In some embodiments, the application device can apply current instead of voltage. In embodiments where voltage is applied to the sensor, after applying the first voltage to the electrode, the voltage regulator can wait (i.e., not apply voltage) for a second time, time frame, or time period 605. In other words, the voltage application device waits until the second time period has passed. Not applying voltage causes a cathodic current, which causes the electrode to which no voltage is applied to acquire electrons. Repeat 610 applying the first voltage to the electrode during the first time period and then not applying voltage during the second time period for several iterations. This can be referred to as anodic and cathodic cycling. In one embodiment, the total number of iterations of the stabilization method is three, i.e., the voltage is applied three times during the first time period and no voltage is applied during the second time period after each application. In an embodiment, the first voltage can be 1.07 volts. In additional embodiments, the first voltage can be 0.535 volts, or can be approximately 0.7 volts.

[0203] Repeatedly applying voltage and not applying voltage causes the sensor (and thus the electrodes) to be subjected to anodic-cathodic cycling. Anodic-cathodic cycling reduces the electrochemical by-products generated due to the patient's body's reaction to the insertion or implantation of the sensor. Electrochemical by-products cause the generation of background current, which leads to inaccurate measurements of the subject's physiological parameters. Under certain operating conditions, the electrochemical by-products can be removed. Under other operating conditions, the electrochemical by-products can be reduced or significantly reduced. A successful stabilization method will bring the anodic-cathodic cycling to equilibrium, significantly reduce the electrochemical by-products, and minimize the background current.

[0204] In one embodiment, the first voltage applied to the electrodes of the sensor can be a positive voltage. In an alternative embodiment, the applied first voltage can be a negative voltage. Additionally, the first voltage can be applied to the working electrode. In some embodiments, the first voltage can be applied to the counter electrode or the reference electrode.

[0205] In some embodiments, the duration of the voltage pulse and the duration without voltage application can be equal, for example, three minutes each. In other embodiments, the duration of the voltage application or the voltage pulse can be different values. For example, the first time and the second time can be different. In one embodiment, the first time period can be five minutes, and the waiting time period can be two minutes. In one variant, the first time period can be two minutes, and the waiting time period (or the second time frame) can be five minutes. In other words, the duration of applying the first voltage can be two minutes, and no voltage may be applied within five minutes. This time frame is illustrative only and should not be restrictive. For example, the first time frame can be two minutes, three minutes, five minutes, or ten minutes, and the second time frame can be five minutes, ten minutes, twenty minutes, etc. The time frames (e.g., the first time and the second time) can depend on the unique characteristics of different electrodes, the sensor, and / or the physiological characteristics of the patient.

[0206] In combination with the foregoing, more or fewer than three pulses can be utilized to stabilize the glucose sensor. In other words, the number of iterations can be greater than 3 or less than 3. For example, four voltage pulses (e.g., high voltage then no voltage application) can be applied to one of the electrodes, or six voltage pulses can be applied to one of the electrodes.

[0207] Illustratively, three consecutive 1.07-volt pulses (followed by the corresponding waiting time) can be sufficient for a subcutaneously implanted sensor. In one embodiment, three consecutive 0.7-volt voltage pulses can be utilized. For a sensor implanted in blood or cranial fluid, such as a long-term or permanent sensor, the three consecutive pulses can have higher or lower voltage values: negative voltage or positive voltage. Additionally, more than three pulses (e.g., five, eight, twelve) can be utilized to create an anodic-cathodic cycle between the anodic current and the cathodic current in any of the subcutaneous, blood, or cranial fluid sensors.

[0208] Figure 6B A method for stabilizing a sensor according to an embodiment of the present invention is shown. In Figure 6B the embodiment shown, the voltage application device can apply a first voltage to the sensor for a first time 630 to initiate an anodic cycle at the electrodes of the sensor. The voltage application device can be a DC power supply, a digital-to-analog converter, or a voltage regulator. After the first time period has elapsed, a second voltage is applied 635 to the sensor for a second time to initiate a cathodic cycle at the electrodes of the sensor. Illustratively, as Figure 6AAs shown in the method, a different voltage (different from the first voltage) is applied to the sensor during the second time frame, rather than no voltage being applied. In an embodiment of the present invention, applying the first voltage during the first time and the second voltage during the second time is repeated for several iterations. In some embodiments, applying the first voltage during the first time and the second voltage during the second time can each be applied for a stabilization time frame, such as 10 minutes, 15 minutes, or 20 minutes, rather than several iterations. This stabilization time frame is the entire time frame of the stabilization sequence, such as until the sensor (and electrodes) stabilizes. The benefits of this stabilization method are faster break - in of the sensor, less background current (in other words, some background current is suppressed), and better glucose response.

[0209] In one embodiment, the first voltage can be 0.535 volts applied for five minutes, and the second voltage can be 1.070 volts applied for two minutes. The first voltage of 0.535 volts can be applied for five minutes, the second voltage of 1.070 volts can be applied for two minutes, the first voltage of 0.535 volts can be applied for five minutes, and the second voltage of 1.070 volts can be applied for two minutes. In other words, in this embodiment, there are three iterations of the voltage pulse scheme. The pulse method can be changed because the second time frame, such as the time frame for applying the second voltage, can be extended from two minutes to five minutes, ten minutes, fifteen minutes, or twenty minutes. Additionally, in this embodiment of the present invention, after three iterations are applied, a nominal operating voltage of 0.535 volts can be applied.

[0210] 1.070 and 0.535 volts are illustrative values. Other voltage values can be selected based on various factors. These factors can include the type of enzyme utilized in the sensor, the membrane utilized in the sensor, the operating period of the sensor, the length of the pulse, and / or the magnitude of the pulse. Under certain operating conditions, the first voltage can be in the range of 1.00 to 1.09 volts, and the second voltage can be in the range of 0.510 to 0.565 volts. In other operating embodiments, depending on the voltage sensitivity of the electrodes in the sensor, the range including the first voltage and the second voltage can have a higher range, for example, 0.3 volts, 0.6 volts, 0.9 volts. Under other operating conditions, the voltage can be in the range of 0.8 volts to 1.34 volts, while another voltage can be in the range of 0.335 to 0.735. Under other operating conditions, the range of the higher voltage can be less than the range of the lower voltage. Illustratively, the higher voltage can be in the range of 0.9 to 1.09 volts, and the lower voltage can be in the range of 0.235 to 0.835 volts.

[0211] In an embodiment, the first voltage and the second voltage can be positive voltages, or in other embodiments, they can be negative voltages. In another embodiment, the first voltage can be positive and the second voltage can be negative, or the first voltage can be negative and the second voltage can be positive. For each iteration, the first voltage can be a different voltage level. Additionally, the first voltage can be a DC constant voltage. Further, the first voltage can be a ramp voltage, a sinusoidal voltage, a step voltage, or other common voltage waveforms. In an embodiment, the second voltage can be a DC constant voltage, a ramp voltage, a sinusoidal voltage, a step voltage, or other common voltage waveforms. In an alternative embodiment, the first voltage or the second voltage can be an AC signal on a DC waveform. Generally, the first voltage can be one type of voltage, such as a ramp voltage, and the second voltage can be a second type of voltage, such as a sinusoidal voltage, and the first voltage (or the second voltage) can have a different waveform shape for each iteration. For example, if there are three cycles in the stabilization method, then in the first cycle, the first voltage can be a ramp voltage, in the second cycle, the first voltage can be a constant voltage, and in the third cycle, the first voltage can be a sinusoidal voltage.

[0212] In an embodiment, the duration of the first time frame and the duration of the second time frame can have the same value, or the durations of the first time frame and the second time frame can have different values. For example, the duration of the first time frame can be two minutes, and the duration of the second time frame can be five minutes, and the number of iterations can be three. As discussed above, the stabilization method can include several iterations. In various embodiments, during different iterations of the stabilization method, the duration of each of the first time frames can change, and the duration of each of the second time frames can change. Illustratively, during the first iteration of the anode-cathode cycle, the first time frame can be 2 minutes, and the second time frame can be 5 minutes. During the second iteration, the first time frame can be 1 minute, and the second time frame can be 3 minutes. During the third iteration, the first time frame can be 3 minutes, and the second time frame can be 10 minutes.

[0213] In one embodiment, a first voltage of 0.535 volts is applied to the electrodes in the sensor for two minutes to start the anode cycle, and then a second voltage of 1.07 volts is applied to the electrodes for five minutes to start the cathode cycle. Then, the first voltage of 0.535 volts is applied again for two minutes to start the anode cycle, and the second voltage of 1.07 volts is applied to the sensor for five minutes. In the third iteration, 0.535 volts is applied for two minutes to start the anode cycle, and then 1.07 volts is applied for five minutes. Then, during the actual operating time frame of the sensor, for example, when the sensor provides a reading of the subject's physiological characteristics, the voltage applied to the sensor is 0.535.

[0214] In Figure 6A and Figure 6B embodiments, voltage pulses of shorter duration can be utilized. The voltage pulses of shorter duration can be used to apply a first voltage, a second voltage, or both. In one embodiment, the magnitude of the voltage pulse of shorter duration for the first voltage is -1.07 volts, and the magnitude of the voltage pulse of shorter duration for the second voltage is approximately half of the higher magnitude, e.g., -0.535 volts. Alternatively, the magnitude of the pulse of shorter duration for the first voltage can be 0.535 volts, and the magnitude of the pulse of shorter duration for the second voltage is 1.07 volts.

[0215] In embodiments utilizing pulses of short duration, the voltage can be applied discontinuously throughout a first time period. Alternatively, the voltage application device can transmit several pulses of short duration during the first time period. In other words, several voltage pulses of small width or short duration can be applied to the electrodes of the sensor during the first time period. Each small width or short duration pulse can have a width of several milliseconds. Illustratively, this pulse width can be 30 milliseconds, 50 milliseconds, 70 milliseconds, or 200 milliseconds. These values are intended to be illustrative rather than restrictive. In one embodiment, for example, in the embodiment shown in Figure 6A , these short duration pulses are applied to the sensor (electrodes) during the first time period and then no voltage is applied during a second time period.

[0216] Each short duration pulse can have the same duration during the first time period. For example, each short duration voltage pulse can have a time width of 50 milliseconds, and each pulse delay between the pulses can be 950 milliseconds. In this example, if the measured time of the first time frame is two minutes, then 120 short duration voltage pulses can be applied to the sensor. Alternatively, each of the short duration voltage pulses can have a different duration. In various embodiments, each of the short duration voltage pulses can have the same amplitude value, or can have different amplitude values. By utilizing voltage pulses of short duration instead of applying voltage to the sensor continuously, the same anodic and cathodic cycles can be performed, and over time, the total energy or charge of the sensor (e.g., the electrodes) is reduced. Using voltage pulses of short duration utilizes less power compared to applying a continuous voltage to the electrodes because less energy is applied to the sensor (and the electrodes).

[0217] Figure 6CIllustrates the utilization of feedback in stabilizing a sensor according to one embodiment. A sensor system may include a feedback mechanism for determining whether additional pulses are needed to stabilize the sensor. In one embodiment, a sensor signal generated by an electrode (e.g., a working electrode) may be analyzed to determine whether the sensor signal is stable. A voltage of 630 is applied to a first electrode in a first time frame to initiate an anodic cycle. A voltage of 635 is applied to the electrode in a second time frame to initiate a cathodic cycle. In an embodiment of the present invention, an analysis module may analyze the sensor signal (e.g., the current emitted by the sensor signal, the resistance at a specific point in the sensor, the impedance at a specific node in the sensor) and determine whether a threshold measurement value 637 has been reached (e.g., by comparing with the threshold measurement value to determine whether the sensor is providing an accurate reading). If the sensor reading is determined to be accurate, which indicates that the electrode (and thus the sensor) has stabilized 642, then the first voltage and / or the second voltage may not be applied additionally. If stability is not achieved, then additional anodic / cathodic cycles may be initiated by applying the first voltage 630 to the electrode for a first time period and then applying the second voltage 635 to the electrode for a second time period.

[0218] In some embodiments, the analysis module may be employed after three anodic / cathodic cycles of applying the first voltage and the second voltage to the electrodes of the sensor. However, the analysis module may be employed after applying the first voltage and the second voltage once, as Figure 6C shown.

[0219] The analysis module may be utilized to measure the voltage emitted after current has been introduced across one electrode or across two electrodes. The analysis module may monitor the voltage level at the electrode or at the receiving level. In one embodiment, if the voltage level is higher than a certain threshold, then this may mean that the sensor has stabilized. In one embodiment, if the voltage level drops below the threshold level, then this may indicate that the sensor has stabilized and is ready to provide a reading. In one embodiment, current may be introduced into the electrode or across a pair of electrodes. The analysis module may monitor the current level emitted from the electrode. In this embodiment, if the current differs from the sensor signal current by an order of magnitude, then the analysis module is capable of monitoring the current. If the current is higher than or lower than the current threshold, then this may indicate that the sensor has stabilized.

[0220] In an embodiment of the present invention, an analysis module may measure the impedance between two electrodes of a sensor. The analysis module may compare the impedance to a threshold or target impedance value, and if the measured impedance is below the target or threshold impedance, then the sensor (and thus the sensor signal) may stabilize. In one embodiment, the analysis module may measure the resistance between two electrodes of the sensor. In this embodiment of the present invention, if the analysis module compares the resistance to a threshold or target resistance value and the measured resistance value is less than the threshold or target resistance value, then the analysis module may determine that the sensor has stabilized and that the sensor signal may be utilized.

[0221] Figure 7 Shows the effect of stabilizing a sensor according to an embodiment of the present invention. Line 705 represents the blood glucose sensor readings of a glucose sensor using a prior single-pulse stabilization method. Line 710 represents the blood glucose readings of a glucose sensor to which three voltage pulses are applied (e.g., 3 voltage pulses with a duration of 2 minutes, with no voltage applied within 5 minutes after each voltage pulse). The x-axis 715 represents the amount of time. Points 720, 725, 730, and 735 represent the measured glucose readings, which are obtained using finger pricks and then input into a blood glucose meter. As shown in the graph, the prior single-pulse stabilization method takes approximately 1 hour 30 minutes to stabilize to a desired glucose reading, such as 100 units. In contrast, the three-pulse stabilization method takes only approximately 15 minutes to stabilize the glucose sensor and significantly improves the stabilization time frame.

[0222] Figure 8A Shows a block diagram of a sensor electronic device and a sensor including a voltage generating device. The voltage generating or applying device 810 includes electronics, logic, or circuitry for generating voltage pulses. The sensor electronic device 360 may also include an input device 820 for receiving reference values and other useful data. In one embodiment, the sensor electronic device may include a measurement memory 830 for storing sensor measurements. In this embodiment, a power supply 380 may supply power to the sensor electronic device. The power supply 380 may supply power to a regulator 385, which supplies a regulated voltage to the voltage generating or applying device 810. The connection terminal 811 represents that in the illustrated embodiment, the connection terminal couples or connects the sensor 355 to the sensor electronic device 360.

[0223] In Figure 8AIn the illustrated embodiment, a voltage generation or application device 810 supplies a voltage, such as a first voltage or a second voltage, to an input terminal of an operational amplifier 840. The voltage generation or application device 810 may also supply a voltage to a working electrode 375 of a sensor 355. Another input terminal of the operational amplifier 840 is coupled to a reference electrode 370 of the sensor. Applying a voltage from the voltage generation or application device 810 to the operational amplifier 840 drives the voltage measured at the counter electrode 365 to be close to or equal to the voltage applied at the working electrode 375. In an embodiment, the voltage generation or application device 810 may be used to apply a desired voltage between the counter electrode and the working electrode. This may be done by directly applying a fixed voltage to the counter electrode.

[0224] In an embodiment such as Figure 6A and Figure 6B shown, a voltage generation device 810 generates a first voltage to be applied to a sensor during a first time frame. The voltage generation device 810 transmits this first voltage to the operational amplifier 840, which drives the voltage at the counter electrode 365 of the sensor 355 to the first voltage. In some embodiments, the voltage generation device 810 may also directly transmit the first voltage to the counter electrode 365 of the sensor 355. In Figure 6A the illustrated embodiment, the voltage generation device 810 then does not transmit the first voltage to the sensor 355 during a second time frame. In other words, the voltage generation device 810 is turned off or disconnected. The voltage generation device 810 may be programmed to continue cycling between applying the first voltage and not applying a voltage over several iterations or stable time frames, such as over twenty minutes. Figure 8B A voltage generation device for implementing this embodiment of the present invention is shown. A voltage regulator 385 transmits a regulated voltage to the voltage generation device 810. A control circuit 860 controls the closing and opening of a switch 850. If the switch 850 is closed, then a voltage is applied. If the switch 850 is open, then no voltage is applied. A timer 865 provides a signal to the control circuit 860 to indicate to the control circuit 860 to turn the switch 850 on and off. The control circuit 860 includes logic that can indicate the circuit to close and open the switch 850 several times (to match the necessary iterations). In one embodiment, the timer 865 may also transmit a stabilization signal to identify that a stabilization sequence has been completed, i.e., a stable time frame has passed.

[0225] In one embodiment, the voltage generation device generates a first voltage during a first time frame and a second voltage during a second time frame. Figure 8CA voltage generation device for generating two voltage values to implement this embodiment is shown. In this embodiment, a two-position switch 870 is utilized. Illustratively, if the control circuit 860 is instructed by the timer 865 to turn on or off the first switch position 871, then the voltage generation device 810 generates a first voltage within the first time frame. After the first voltage has been applied within the first time frame, the timer sends a signal indicating that the first time frame has elapsed to the control circuit 860, and the control circuit 860 guides the switch 870 to move to the second position 872. When the switch 870 is in the second position 872, the regulated voltage is directed to the voltage step-down or buck converter 880 to reduce the regulated voltage to a smaller value. Then, the smaller value is delivered to the operational amplifier 840 within the second time frame. After the timer 865 has sent a signal to the control circuit 860 indicating that the second time period has elapsed, the control circuit 860 moves the switch 870 back to the first position. This continues until the required number of iterations has been completed or the stabilization time frame has elapsed. In an embodiment of the present invention, after the sensor stabilization time frame has elapsed, the sensor transmits the sensor signal 350 to the signal processor 390.

[0226] Figure 8D A voltage application device 810 for performing a more complex application of voltage to a sensor is shown. The voltage application device 810 may include a control device 860, a switch 890, a sinusoidal voltage generation device 891, a ramp voltage generation device 892, and a constant voltage generation device 893. In other embodiments, the voltage application may generate an AC wave on top of a DC signal or other various voltage pulse waveforms. In Figure 8DIn the illustrated embodiment, the control device 860 can move the switch to one of three voltage generation systems 891 (sinusoidal), 892 (ramp), 893 (constant DC). This causes each of the voltage generation systems to generate the identified voltage waveform. Under certain operating conditions, for example, in the case where sinusoidal pulses are to be applied for three pulses, the control device 860 can cause the switch 890 to connect the voltage from the voltage regulator 385 to the sinusoidal voltage generator 891 so that the voltage application device 810 generates a sinusoidal voltage. Under other operating conditions, for example, when a ramp voltage is applied as the first voltage of the first pulse of three pulses to the sensor, a sinusoidal voltage is applied as the first voltage of the second pulse of three pulses to the sensor, and a constant DC voltage is applied as the first voltage of the third pulse of three pulses to the sensor, the control device 860 can cause the switch 890 to move between the following during the first time frame in the anode / cathode cycle: connecting the voltage from the voltage generation or application device 810 to the ramp voltage generation system 892, then to the sinusoidal voltage generation system 891, and then to the constant DC voltage generation system 893. In this embodiment, the control device 860 can also direct or control the switch to connect certain voltage generation subsystems to the voltage from the regulator 385 during the second time frame, for example, during the application of the second voltage.

[0227] Figure 9A A sensor electronic device including a microcontroller for generating voltage pulses is shown. An advanced sensor electronic device can include a microcontroller 410 (see Figure 4 ), a digital-to-analog converter (DAC) 420, an operational amplifier 840, and a sensor signal measurement circuit 431. In one embodiment, the sensor signal measurement circuit can be a current-to-frequency (I / F) converter 430. In Figure 9A the illustrated embodiment, software or programmable logic in the microcontroller 410 provides instructions for transmitting a signal to the DAC 420, which in turn instructs the DAC 420 to output a specific voltage to the operational amplifier 840. As Figure 9AAs shown by line 911 therein, it can also instruct the microcontroller 410 to output a specific voltage to the working electrode 375. As discussed above, applying a specific voltage to the operational amplifier 840 and the working electrode 375 can drive the voltage measured at the counter electrode to a specific voltage magnitude. In other words, the microcontroller 410 outputs a signal indicating the voltage or voltage waveform to be applied to the sensor 355 (e.g., the operational amplifier 840 connected to the sensor 355). In an alternative embodiment, a fixed voltage can be set by directly applying a voltage from the DAC 420 between the reference electrode and the working electrode 375. Similar results can also be obtained by applying voltages to each of the electrodes, with the voltage difference equal to the fixed voltage applied between the reference electrode and the working electrode. Additionally, a fixed voltage can be set by applying a voltage between the reference electrode and the counter electrode. Under certain operating conditions, the microcontroller 410 can generate pulses of a specific magnitude, and the DAC 420 interprets the pulses as indicating that a specific magnitude of voltage is to be applied to the sensor. After the first time frame, the microcontroller 410 (through a program or programmable logic) outputs a second signal that indicates that the DAC 420 does not output a voltage (for the sensor electronic device 360 operating according to the method described in Figure 6A the sensor electronic device 360 described in Figure 6B or outputs a second voltage (for the sensor electronic device 360 operating according to the method described in

[0228] After the second time frame has passed, the microcontroller 410 then repeats the following cycle: sending a signal indicating the first voltage to be applied (within the first time frame), and then sending the signal to indicate no voltage application or the second voltage to be applied (within the second time frame).

[0229] The microcontroller 410 may include programmable logic or a program for continuing this loop within a stable time frame or for several iterations. Illustratively, the microcontroller 410 may include counting logic for identifying when the first time frame or the second time frame has passed. Additionally, the microcontroller 410 may include counting logic for identifying when the stable time frame has passed. After any of the foregoing time frames has passed, the counting logic may instruct the microcontroller to send a new signal or to stop transmitting signals to the DAC 420.

[0230] The use of the microcontroller 410 allows for the application of multiple voltage magnitudes in several sequences over several durations. In an embodiment of the present invention, the microcontroller 410 may include control logic or a program for instructing the digital-to-analog converter 420 to transmit voltage pulses having a magnitude of approximately 1.0 volts during a first time period of 1 minute, and then to transmit voltage pulses having a magnitude of approximately 0.5 volts during a second time period of 4 minutes, and to repeat this loop for four iterations. In one embodiment, the microcontroller 420 may be programmed to transmit signals to cause the DAC 420 to apply voltage pulses of the same magnitude for each first voltage during each iteration. The microcontroller 410 may be programmed to transmit signals to cause the DAC to apply voltage pulses of different magnitudes for each first voltage during each iteration. In this embodiment, the microcontroller 410 may also be programmed to transmit signals to cause the DAC 420 to apply voltage pulses of different magnitudes for each second voltage during each iteration. Illustratively, the microcontroller 410 may be programmed to transmit signals to cause the DAC 420 to apply a first voltage pulse of approximately 1.0 volts during the first iteration, a second voltage pulse of approximately 0.5 volts during the first iteration, a first voltage of 0.7 volts and a second voltage of 0.4 volts during the second iteration, and a first voltage of 1.2 volts and a second voltage of 0.8 volts during the third iteration.

[0231] The microcontroller 410 can also be programmed to instruct the DAC 420 to provide a number of voltage pulses of short duration within a first time frame. In this embodiment of the present invention, instead of applying one voltage throughout the first time frame (e.g., two minutes), a number of shorter duration pulses can be applied to the sensor. In this embodiment, the microcontroller 410 can also be programmed to instruct the DAC 420 to provide a number of voltage pulses of short duration to the sensor within a second time frame. Illustratively, the microcontroller 410 can send a signal to cause the DAC to apply a number of voltage pulses of short duration, where the short duration is 50 milliseconds or 100 milliseconds. Between these short duration pulses, the DAC may not apply a voltage, or the DAC may apply a minimum voltage. The microcontroller can cause the DAC 420 to apply voltage pulses of short duration within a first time frame (e.g., two minutes). Then, the microcontroller 410 can send a signal to cause the DAC not to apply any voltage or to apply voltage pulses of short duration to the sensor in an amount of a second voltage within the second time frame. For example, the second voltage can be 0.75 volts, and the second time frame can be five minutes. In one embodiment, the microcontroller 410 can send a signal to the DAC 420 to cause the DAC 420 to apply different magnitude voltages for each short duration pulse in the first time frame and / or the second time frame. In an embodiment, the microcontroller 410 can send a signal to the DAC 420 to cause the DAC 420 to apply a pattern of voltage magnitudes to the voltage pulses of short duration within the first time frame or the second time frame. For example, the microcontroller can transmit a signal or pulse indicating that the DAC 420 applies thirty 20-millisecond pulses to the sensor during the first time frame. Each of the thirty 20-millisecond pulses can have the same magnitude or can have different magnitudes. In this embodiment, the microcontroller 410 can instruct the DAC 420 to apply short duration pulses during the second time frame, or can instruct the DAC 420 to apply another voltage waveform during the second time frame.

[0232] Although the disclosure in FIGS. 6-8 discloses the application of voltage, current can also be applied to the sensor to initiate the stabilization process. Illustratively, in Figure 6BIn the illustrated embodiment, a first current can be applied during a first time frame to initiate an anodic or cathodic response, and a second current can be applied during a second time frame to initiate an opposite anodic or cathodic response. The application of the first and second currents can continue for several iterations or can continue within a stabilization time frame. In one embodiment, a first current can be applied during a first time frame, and a first voltage can be applied during a second time frame. In other words, one of the anodic or cathodic cycles can be triggered by applying a current to the sensor, and one of the anodic or cathodic cycles can be triggered by applying a voltage to the sensor. As described above, the applied current can be a constant current, a ramp current, a step pulse current, or a sinusoidal current. Under certain operating conditions, the current can be applied as a series of short-duration pulses during the first time frame.

[0233] Figure 9B A sensor and sensor electronics according to an embodiment of the present invention are shown, which utilize an analysis module to provide feedback during a stabilization period. Figure 9B The analysis module 950 is incorporated into the sensor electronics 360. The analysis module 950 utilizes feedback from the sensor to determine whether the sensor is stable. In one embodiment, the microcontroller 410 can include instructions or commands for controlling the DAC 420 to cause the DAC 420 to apply a voltage or current to a portion of the sensor 355. Figure 9B It is shown that a voltage or current can be applied between the reference electrode 370 and the working electrode 375. However, the voltage or current can be applied between the electrodes or directly to one of the electrodes, and the present invention should not be limited Figure 9B by the illustrated embodiment. The application of the voltage or current is shown by the dashed line 955. The analysis module 950 can measure the voltage, current, resistance, or impedance in the sensor 355. Figure 9BMeasurements are shown being made at the working electrode 375, but this should not limit the invention, as other embodiments may measure voltage, current, resistance, or impedance between electrodes of the sensor or directly at the reference electrode 370 or counter electrode 365. The analysis module 950 may receive the measured voltage, current, resistance, or impedance and may compare the measured values to stored values (e.g., thresholds). The dashed line 956 represents the analysis module 950 reading or making a measurement of voltage, current, resistance, or impedance. Under certain operating conditions, if the measured voltage, current, resistance, or impedance is above the threshold, then the sensor stabilizes and the sensor signal provides an accurate reading of the patient's physiological condition. Under other operating conditions, if the measured voltage, current, resistance, or impedance is below the threshold, then the sensor stabilizes. Under other operating conditions, the analysis module 950 may verify that the measured voltage, current, resistance, or impedance is stable within a particular time frame, e.g., one or two minutes. This may indicate that the sensor 355 has stabilized and that the sensor signal is transmitting an accurate measurement of the subject's physiological parameter, such as a blood glucose level. After the analysis module 950 has determined that the sensor has stabilized and that the sensor signal is providing an accurate measurement, the analysis module 950 may transmit a signal (e.g., a sensor stabilization signal) to the microcontroller 410, which indicates that the sensor has stabilized and the microcontroller 410 may begin to use or receive the sensor signal from the sensor 355. This is represented by the dashed line 957.

[0234] Figure 10 A block diagram of a sensor system including a hydrated electronic device is shown. The sensor system includes a connector 1010, a sensor 1012, and a monitor or sensor electronics 1025. The sensor 1012 includes electrodes 1020 and a connection portion 1024. In one embodiment, the sensor 1012 may be connected to the sensor electronics 1025 via the connector 1010 and a cable. In other embodiments, the sensor 1012 may be directly connected to the sensor electronics 1025. In some embodiments, the sensor 1012 may be incorporated into the same physical device as the sensor electronics 1025. The monitor or sensor electronics 1025 may include a power supply 1030, a regulator 1035, a signal processor 1040, a measurement processor 1045, and a processor 1050. The monitor or sensor electronics 1025 may also include a hydration detection circuit 1060. The hydration detection circuit 1060 interfaces with the sensor 1012 to determine whether the electrodes 1020 of the sensor 1012 are sufficiently hydrated. If the electrodes 1020 are not sufficiently hydrated, then the electrodes 1020 do not provide an accurate glucose reading, so it is important to know when the electrodes 1020 are sufficiently hydrated. Once the electrodes 1020 are sufficiently hydrated, an accurate glucose reading can be obtained.

[0235] InFigure 10 In the illustrated embodiment, the hydration detection circuit 1060 may include a delay or timer module 1065 and a connection detection module 1070. In embodiments utilizing short-term or subcutaneous sensors, after the sensor 1012 is inserted into the subcutaneous tissue, the sensor electronics or monitor 1025 is connected to the sensor 1012. The connection detection module 1070 identifies that the sensor electronics 1025 has been connected to the sensor 1012 and sends a signal to the timer module 1065. This is illustrated by arrow 1084 in Figure 10 which indicates that the detector 1083 detects the connection and sends a signal to the connection detection module 1070 indicating that the sensor 1012 has been connected to the sensor electronics 1025. In embodiments utilizing implantable or long-term sensors, the connection detection module 1070 identifies that an implantable sensor has been inserted into the body. The timer module 1065 receives the connection signal and waits for a set or established hydration time. Illustratively, the hydration time may be two minutes, five minutes, ten minutes, or 20 minutes. These examples are illustrative and not restrictive. The time frame need not be a set number of minutes and may include any number of seconds. In one embodiment, after the timer module 1065 has waited for the set hydration time, the timer module 1065 may notify the processor 1050 that the sensor 1012 is hydrated by sending a hydration signal, as shown by line 1086.

[0236] In this embodiment, the processor 1050 may receive the hydration signal and only start utilizing the sensor signal (e.g., sensor measurements) after the hydration signal has been received. In another embodiment, the hydration detection circuit 1060 may be coupled between the sensor (sensor electrode 1020) and the signal processor 1040. In this embodiment, the hydration detection circuit 1060 may prevent the sensor signal from being sent to the signal processor 1040 until the timer module 1065 has notified the hydration detection circuit 1060 that the set hydration time has elapsed. This is illustrated by the dashed lines labeled 1080 and 1081. Illustratively, the timer module 1065 may transmit the connection signal to a switch (or transistor) to turn on the switch and allow the sensor signal to proceed to the signal processor 1040. In an alternative embodiment, the timer module 1065 may transmit the connection signal to turn on (or close) the switch 1088 in the hydration detection circuit 1060 to allow the voltage from the regulator 1035 to be applied to the sensor 1012 after the hydration time has elapsed. In other words, in this embodiment, the voltage from the regulator 1035 is not applied to the sensor 1012 until after the hydration time has elapsed.

[0237] Figure 11An embodiment including a mechanical switch for assisting in determining a hydration time is shown. In one embodiment, a single housing may contain the sensor assembly 1120 and the sensor electronics 1125. In another embodiment, the sensor assembly 1120 may be in one housing and the sensor electronics 1125 may be in a separate housing, but the sensor assembly 1120 and the sensor electronics 1125 may be connected together. In this embodiment, the connection detection mechanism 1160 may be a mechanical switch. The mechanical switch may detect that the sensor 1120 is physically connected to the sensor electronics 1125. When the mechanical switch 1160 detects that the sensor 1120 is connected to the sensor electronics 1125, the timer circuit 1135 may also be activated. In other words, the mechanical switch may close and a signal may be transmitted to the timer circuit 1135. Once the hydration time has passed, the timer circuit 1135 transmits a signal to the switch 1140 to allow the regulator 1035 to apply a voltage to the sensor 1120. In other words, the voltage is not applied until the hydration time has passed. In one embodiment, once the hydration time has passed, current may be applied instead of voltage to the sensor. In an alternative embodiment, when the mechanical switch 1160 recognizes that the sensor 1120 has been physically connected to the sensor electronics 1125, power may be applied to the sensor 1120 first. Sending power to the sensor 1120 causes a sensor signal to be output from the working electrode in the sensor 1120. The sensor signal may be measured and sent to the processor 1175. The processor 1175 may include a counter input. Under certain operating conditions, after a set hydration time has elapsed since the sensor signal was input into the processor 1175, the processor 1175 may begin processing the sensor signal as an accurate measurement of glucose in a subject's body. In other words, the processor 1170 has received the sensor signal from the potentiostat circuit 1170 for a certain amount of time, but does not process the signal until it receives an instruction from the counter input of the processor identifying that the hydration time has passed. In an embodiment, the potentiostat circuit 1170 may include a current-to-frequency converter 1180. In this embodiment, the current-to-frequency converter 1180 may receive the sensor signal as a current value and may convert the current value to a frequency value, which is easier for the processor 1175 to process.

[0238] When the sensor 1120 has been disconnected from the sensor electronics 1125, the mechanical switch 1160 may also notify the processor 1175. This is shown by Figure 11is represented by the dashed line 1176 therein. This may cause the processor 1170 to power down or reduce the power of several components, chips, and / or circuits of the sensor electronics 1125. If the sensor 1120 is not connected, the battery or power source may be depleted if the components or circuits of the sensor electronics 1125 are powered on. Therefore, if the mechanical switch 1160 detects that the sensor 1120 has been disconnected from the sensor electronics 1125, the mechanical switch can indicate this to the processor 1175, and the processor 1175 can power down or reduce the power of one or more of the electronic circuits, chips, or components of the sensor electronics 1125.

[0239] Figure 12 An electrical method for detecting hydration according to an embodiment of the present invention is shown. In one embodiment, an electrical detection mechanism for detecting the connection of a sensor can be utilized. In this embodiment, the hydration detection electronics 1250 may include an AC source 1255 and a detection circuit 1260. The hydration detection electronics 1250 may be located in the sensor electronics 1225. The sensor 1220 may include a counter electrode 1221, a reference electrode 1222, and a working electrode 1223. As Figure 12 shown, the AC source 1255 is connected to a voltage setting device 1275, the reference electrode 1222, and the detection circuit 1260. In this embodiment, the AC signal from the AC source is applied to the reference electrode connection, as Figure 12 represented by the dashed line 1291 therein. The AC signal can be impedance-connected to the sensor 1220, and if the sensor 1220 is connected to the sensor electronics 1225, the connected signal will be significantly attenuated. Therefore, a low-level AC signal exists at the input of the detection circuit 1260. This can also be referred to as a highly attenuated signal or a signal with a high attenuation level. Under certain operating conditions, the voltage level of the AC signal can be Vapplied*(Ccoupling) / (Ccoupling+Csensor). If the detection circuit 1260 detects a high-level AC signal (low-attenuation signal) at the input terminals of the detection circuit 1260, no interrupt is sent to the microcontroller 410 because the sensor 1220 is not sufficiently hydrated or activated. For example, the input of the detection circuit 1260 can be a comparator. If the sensor 1220 is sufficiently hydrated (or wetted), an effective capacitance (e.g., Figure 12 the capacitance C in r-c ) is formed between the counter electrode and the reference electrode, and an effective capacitance (e.g., Figure 12 the capacitance C in w-rThat is to say, the effective capacitance is related to the capacitance formed between two nodes and does not represent placing an actual capacitor in the circuit between two electrodes. In one embodiment, the AC signal from the AC source 1255 is capacitively coupled by capacitors C r-c and C w-r sufficiently attenuated, and the detection circuit 1260 detects the presence of a low-level or highly attenuated AC signal from the AC source 1255 at the input terminal of the detection circuit 1260. This embodiment is important because the existing connection between the sensor 1120 and the sensor electronics 1125 is utilized to reduce the number of connections to the sensor. That is to say, Figure 11 the mechanical switches disclosed in [reference] require switches and associated connections between the sensor 1120 and the sensor electronics 1125. Removing the mechanical switch is advantageous because the size of the sensor 1120 is continuously shrinking, and removing components helps to achieve this size reduction. In an alternative embodiment, the AC signal can be applied to different electrodes (e.g., counter electrode or working electrode), and the present invention can operate in a similar manner.

[0240] As described above, after the detection circuit 1260 has detected the presence of a low-level AC signal at the input terminal of the detection circuit 1260, the detection circuit 1260 may later detect the presence of a high-level AC signal with low attenuation at the input terminal. This indicates that the sensor 1220 has been disconnected from the sensor electronics 1225, or the sensor is not operating properly. If the sensor has been disconnected from the sensor electronics 1225, then the AC source can be coupled to the input of the detection circuit 1260 with little or low attenuation. As described above, the detection circuit 1260 can generate an interrupt to the microcontroller. This interrupt can be received by the microcontroller, and the microcontroller can reduce or remove the power of one or more components or circuits in the sensor electronics 1225. This can be referred to as a second interrupt. Similarly, this helps to reduce the power consumption of the sensor electronics 1225, especially when the sensor 1220 is not connected to the sensor electronics 1225.

[0241] In an alternative embodiment, the AC signal can be applied to the reference electrode 1222, as shown by reference numeral 1291, and the impedance measurement device 1277 can measure the impedance of a region in the sensor 1220. Illustratively, the region can be the region between the reference electrode and the working electrode, as Figure 12as shown by the dashed line 1292 in. Under certain operating conditions, if the measured impedance has decreased below an impedance threshold or other set criterion, the impedance measurement device 1277 can transmit a signal to the detection circuit 1260. This indicates that the sensor is sufficiently hydrated. Under other operating conditions, once the impedance is above the impedance threshold, the impedance measurement device 1277 can transmit a signal to the detection circuit 1260. The detection circuit 1260 then transmits an interrupt to the microcontroller 410. In another embodiment, the impedance measurement device 1277 can transmit an interrupt or a signal directly to the microcontroller.

[0242] In an alternative embodiment, the AC source 1255 can be replaced by a DC source. If a DC source is used, a resistance measurement element can be used instead of the impedance measurement element 1277. In an embodiment using a resistance measurement element, once the resistance drops below a resistance threshold or set criterion, the resistance measurement element can transmit a signal to the detection circuit 1260 (represented by the dashed line 1293) or directly to the microcontroller, the signal indicating that the sensor is sufficiently hydrated and power can be applied to the sensor.

[0243] In Figure 12 the embodiment shown, if the detection circuit 1260 detects a low-level or highly attenuated AC signal from the AC source, an interrupt is generated to the microcontroller 410. This interrupt indicates that the sensor is sufficiently hydrated. In this embodiment, in response to the interrupt, the microcontroller 410 generates a signal that is transmitted to the digital-to-analog converter 420 to indicate or cause the digital-to-analog converter 420 to apply a voltage or current to the sensor 1220. As described above in Figure 6A 、 Figure 6B or Figure 6CAny of the pulses of the different sequences or pulses of short duration described in the associated text describing the pulse application can be applied to the sensor 1220. Illustratively, the voltage from the DAC 420 can be applied to the operational amplifier 1275, and the output of the operational amplifier is applied to the counter electrode 1221 of the sensor 1220. This causes the sensor, such as the working electrode 1223 of the sensor, to generate a sensor signal. Since the sensor is fully hydrated, as identified by the interruption, the sensor signal generated at the working electrode 1223 is accurately measuring glucose. The sensor signal is measured by the sensor signal measuring device 431, and the sensor signal measuring device 431 transmits the sensor signal to the microcontroller 410, where the parameters of the physiological condition of the subject are measured. The generation of the interruption indicates that the sensor is fully hydrated, and the sensor 1220 is now providing accurate glucose measurements. In this embodiment, the hydration period can depend on the type and / or manufacturer of the sensor, and on the response of the sensor to being inserted or implanted into the subject. Illustratively, one sensor 1220 can have a hydration time of five minutes, and one sensor 1220 can have a hydration time of one minute, two minutes, three minutes, six minutes, or 20 minutes. Similarly, any amount of time can be an acceptable hydration time amount for the sensor, but a lesser amount of time is preferred.

[0244] If the sensor 1220 is connected but not fully hydrated or wetted, then the effective capacitance C r-c and C w-r may not attenuate the AC signal from the AC source 1255. The electrodes in the sensor 1120 are dry before insertion, and since the electrodes are dry, there is no good circuit path (or conductive path) between the two electrodes. Therefore, the detection circuit 1260 can still detect a high-level AC signal or a low-attenuation AC signal and may not generate an interruption. Once the sensor is inserted, the electrodes are immersed in the conductive body fluid. This creates a leakage path with a lower DC resistance. Also, a boundary layer capacitor is formed at the metal / fluid interface. In other words, a significant capacitance is formed between the metal / fluid interfaces, and this large capacitance appears as two capacitors connected in series between the electrodes of the sensor. This can be referred to as the effective capacitance. In fact, the conductivity of the electrolyte above the electrodes is being measured. In some embodiments of the present invention, the glucose limiting membrane (GLM) also exhibits an impedance that hinders electrical efficiency. The unhydrated GLM produces a high impedance, while the highly hydrated GLM produces a low impedance. Accurate sensor measurements require a low impedance.

[0245] Figure 13AA method of hydrating a sensor according to an embodiment of the present invention is shown. In one embodiment, the sensor can be physically connected 1310 to the sensor electronics. After the connection, in one embodiment, a timer or counter can be started to count 1320 the hydration time. After the hydration time has elapsed, a signal can be transmitted 1330 to a subsystem in the sensor electronics to start applying a voltage to the sensor. As discussed above, in one embodiment, the microcontroller can receive the signal and instruct the DAC to apply a voltage to the sensor, or in another embodiment of the present invention, a switch can receive the signal, and the signal allows the regulator to apply a voltage to the sensor. The hydration time can be five minutes, two minutes, ten minutes, and can vary depending on the subject and the type of sensor.

[0246] In an alternative embodiment, after connecting the sensor to the sensor electronics, an AC signal (e.g., a low voltage AC signal) can be applied 1340 to the sensor, such as the reference electrode of the sensor. The AC signal can be applied because the connection of the sensor to the sensor electronics allows the AC signal to be applied to the sensor. After applying the AC signal, an effective capacitance is formed 1350 between the electrode to which the applied voltage is applied and the other two electrodes in the sensor. The detection circuit determines 1360 what level of AC signal is present at the input of the detection circuit. If a low level AC signal (or a highly attenuated AC signal) is present at the input of the detection circuit, then since the effective capacitance forms a good electrical conduit between the electrodes and the AC signal is attenuated, the detection circuit generates 1370 an interrupt and the interrupt is sent to the microcontroller.

[0247] The microcontroller receives the interrupt generated by the detection circuit and transmits 1380 a signal to the digital-to-analog converter, and the signal indicates or causes the digital-to-analog converter to apply a voltage to the electrodes of the sensor, such as to the counter electrode. Applying a voltage to the electrodes of the sensor causes the sensor to generate or produce a sensor signal 1390. The sensor signal measuring device 431 measures the generated sensor signal and transmits the sensor signal to the microcontroller. The microcontroller receives 1395 the sensor signal from the sensor signal measuring device connected to the working electrode and processes the sensor signal to extract a measurement of the physiological characteristics of the subject or patient.

[0248] Figure 13B Another method for verifying the hydration of a sensor according to an embodiment of the present invention is shown. In Figure 13BIn the illustrated embodiment, the sensor is physically connected 1310 to the sensor electronics. An AC signal is applied 1341 to an electrode in the sensor, such as a reference electrode. Alternatively, in another embodiment, a DC signal is applied 1341 to an electrode in the sensor. If an AC signal is applied, then an impedance measurement element measures 1351 the impedance at a point within the sensor. Alternatively, if a DC signal is applied, then a resistance measurement element measures 1351 the resistance at a point within the sensor. If the resistance or impedance is below a resistance threshold or impedance threshold, respectively (or other set criteria), then the impedance (or resistance) measurement element transmits 1361 (or allows transmission of) a signal to the detection circuit, and the detection circuit transmits an interrupt indicating that the sensor has hydrated to the microcontroller. Reference numerals 1380, 1390, and 1395 are the same in Figure 13A and Figure 13B because they represent the same action.

[0249] The microcontroller receives the interrupt and transmits 1380 a signal to the digital-to-analog converter to apply a voltage to the sensor. In an alternative embodiment, the digital-to-analog converter may apply a current to the sensor, as discussed above. The sensor, such as a working electrode, generates 1390 a sensor signal that represents a physiological parameter of the patient. The microcontroller receives 1395 the sensor signal from a sensor signal measuring device that measures the sensor signal at an electrode (e.g., a working electrode) in the sensor. The microcontroller processes the sensor signal to extract a measurement of a physiological characteristic of the subject or patient, such as the blood glucose level of the patient.

[0250] Figure 14A and Figure 14B illustrates a method of combining hydration of a sensor with stabilization of the sensor according to an embodiment of the present invention. In Figure 14AIn the illustrated embodiment of the present invention, a sensor is connected 1405 to a sensor electronic device. An AC signal is applied 1410 to the electrodes of the sensor. A detection circuit determines 1420 what level of AC signal is present at the input of the detection circuit. If the detection circuit determines that a low-level AC signal is present at the input (indicating a high level of attenuation of the AC signal), then an interrupt is sent 1430 to the microcontroller. Once the interrupt is sent to the microcontroller, the microcontroller knows to begin or initiate 1440 a stabilization sequence, i.e., to apply several voltage pulses to the electrodes of the sensor, as described above. For example, the microcontroller may cause a digital-to-analog converter to apply three voltage pulses (magnitude of +0.535 volts) to the sensor, where each of the three voltage pulses is followed by a period of three voltage pulses (voltage magnitude to be applied is 1.07 volts). This may be referred to as a stabilization sequence of transmitted voltages. The microcontroller may cause this to happen by executing a software program in a read-only memory (ROM) or a random access memory. After the stabilization sequence has been executed, the sensor may generate 1450 a sensor signal, which is measured and transmitted to the microcontroller.

[0251] The detection circuit may determine 1432 that a high-level AC signal continues to be present at the input of the detection circuit (e.g., the input of a comparator), even after a hydration time threshold has passed. For example, the hydration time threshold may be 10 minutes. After 10 minutes have elapsed, the detection circuit may still detect that a high-level AC signal is present. At this point in time, the detection circuit may transmit 1434 a hydration assist signal to the microcontroller. If the microcontroller receives the hydration assist signal, then the microcontroller may transmit 1436 a signal to cause the DAC to apply a voltage pulse or a series of voltage pulses to assist in hydrating the sensor. In one embodiment, the microcontroller may transmit a signal to cause the DAC to apply a portion of the stabilization sequence or other voltage pulses to assist in hydrating the sensor. In this embodiment, the application of the voltage pulses may cause a low-level AC signal (or a highly attenuated signal) to be detected 1438 at the detection circuit. At this point, the detection circuit may transmit an interrupt, as disclosed in step 1430, and the microcontroller may begin the stabilization sequence.

[0252] Figure 14BShows a second embodiment of the combination of a hydration method and a stabilization method, where feedback is utilized during the stabilization process. A sensor connection 1405 is made to sensor electronics. An AC signal (or DC signal) is applied 1411 to the sensor. In an embodiment, the AC signal (or DC signal) is applied to an electrode of the sensor, such as a reference electrode. An impedance measurement device (or resistance measurement device) measures 1416 the impedance (or resistance) within a specified region of the sensor, such as between the reference electrode and the working electrode. The measured impedance (or resistance) can be compared 1421 with an impedance or resistance value to see if the impedance (or resistance) in the sensor is low enough, which indicates that the sensor is hydrated. If the impedance (or resistance) is below the impedance (or resistance) value or other set criteria (which can be a threshold), then the transmission 1431 to the microcontroller will be interrupted. After receiving the interruption, the microcontroller transmits 1440 a signal to the DAC, which signal instructs the DAC to apply a stabilization sequence of voltage (or current) to the sensor. After the stabilization sequence is applied to the sensor, a sensor signal is generated in the sensor (e.g., at the working electrode), measured by a sensor signal measurement device, transmitted by the sensor signal measurement device, and received by the microcontroller 1450. Since the sensor is hydrated and a stabilization sequence of voltage has been applied to the sensor, the sensor signal is accurately measuring the physiological parameter (i.e., blood glucose).

[0253] Figure 14CA third embodiment is shown where the stabilization method and the hydration method are combined. In this embodiment, the sensor connection 1500 is connected to the sensor electronics. After the sensor is physically connected to the sensor electronics, an AC signal (or a DC signal) is applied 1510 to the electrodes of the sensor (e.g., the reference electrode). At the same time or approximately at the same time, the microcontroller transmits a signal to cause the DAC to apply 1520 a sequence of stabilization voltages to the sensor. In an alternative embodiment, a sequence of stabilization currents rather than a sequence of stabilization voltages can be applied to the sensor. The detection circuit determines 1530 what level of AC signal (or DC signal) is present at the input terminals of the detection circuit. If a low-level AC signal (or DC signal) representing a highly attenuated AC signal (or DC signal) is present at the input terminals of the detection circuit, then the transmission 1540 to the microcontroller will be interrupted. Since the microcontroller has started the stabilization sequence, the microcontroller receives the interruption and sets 1550 the first indicator that the sensor is sufficiently hydrated. After the stabilization sequence is completed, the microcontroller sets 1555 the second indicator that indicates the completion of the stabilization sequence. The application of the stabilization sequence voltage causes the sensor, e.g., the working electrode, to generate 1560 a sensor signal that is measured by the sensor signal measurement circuit and sent to the microcontroller. If the second indicator indicating the completion of the stabilization sequence is set and the first indicator indicating the completion of hydration is set, then the microcontroller can utilize 1570 the sensor signal. If one or both of the indicators are not set, then the microcontroller may not be able to utilize the sensor signal because the sensor signal may not represent an accurate measurement of the physiological measurement of the subject.

[0254] Generally, the above hydration and stabilization processes can be used as part of a larger continuous glucose monitoring (CGM) method. The current state of the art in continuous glucose monitoring is largely adjunctive, meaning that the readings provided by CGM devices (including, for example, implantable or subcutaneous sensors) cannot be used to make clinical decisions without a reference value. The reference value typically must be obtained from a fingerstick using, for example, a BG meter. The reason for the need for a reference value is the limited amount of information that can be obtained from the sensor / sensing assembly. Specifically, the sensing assembly may only provide the raw sensor value (i.e., the sensor current or Isig) and the counter voltage as the voltage between the counter electrode and the reference electrode (e.g., see Figure 5 ) for processing. Thus, during analysis, if the raw sensor signal appears abnormal (e.g., if the signal is decreasing), the only way to distinguish between a sensor failure and a physiological change (i.e., a change in glucose level) in the user / patient's body may be to obtain a reference glucose value via a fingerstick. It is well known that reference fingersticks are also used to calibrate the sensor.

[0255] Embodiments of the invention described herein are directed to advancements and improvements in continuous glucose monitoring, resulting in more autonomous systems and related devices and methods, where the need for reference finger sticks can be minimized or eliminated, and clinical decisions can thus be made very reliably based on information obtained only from sensor signals. From a sensor design perspective, according to embodiments of the invention, such autonomy can be achieved through electrode redundancy, sensor redundancy (including, for example, complex redundancy between two or more sensors), sensor diagnostics, and Isig and / or sensor glucose (SG) fusion.

[0256] In the discussion herein, and for the purposes of the present invention, "redundancy" refers to the presence / use of two or more electrodes, whether contained on / within a single probe (or "flexible portion"), or on / within two or more flexible portions, and "complex redundancy" refers to the presence / use of two (or more) sensors, where at least two of the sensors are different. Thus, "redundant" electrodes can be contained on / within a single flexible portion, two or more identical flexible portions, or two or more different flexible portions. As will be further explored below, redundancy can be achieved, for example, by using multiple working electrodes to generate multiple signals indicative of a patient's blood glucose (BG) level. The multiple signals can in turn be used to generate a fused glucose value, as well as to assess the relative health of the (working) electrodes, the overall reliability of the sensor, and the frequency at which a calibration reference value (if any) is needed.

[0257] For example, it is known that obtaining signals from multiple electrochemical sensors can provide improved performance in the form of simple redundancy, which is achieved either through multiple electrodes on the same probe (or flexible portion), or by taking advantage of spatial separation and two separate probes. For example, Medtronic sells a hospital glucose sensor that includes two probes, each with two working electrodes, thus generating four independent glucose signals.

[0258] In contrast to simple redundancy, orthogonal redundancy can be defined as two devices that use two different technologies to achieve the same goal, where the failure modes of the two devices are completely unique and non - overlapping. Thus, orthogonal redundancy can be generated by combining, for example, optical sensing and electrochemical sensing technologies. Clearly, the advantage of orthogonal redundancy is that two types of sensors, such as optical and electrochemical (or "echem") sensors, are subject to different types of interference, failure modes, and body responses. On the other hand, the use of two completely different technologies introduces an additional level of design and computational complexity to the measurement and analysis of glucose levels in a patient's body.

[0259] On the other hand, pseudo-orthogonal redundancy can be achieved by leveraging the same techniques, but with variations to produce complementary glucose measurements while minimizing additional design and / or computational complexity. For example, two or more electrochemical sensors can be employed, where one (or more) of the sensors can be a conventional peroxide-based sensor, and one (or more) of the sensors can measure glucose by calculating the oxygen difference between two working electrodes (usually on the same sensor).

[0260] In yet another specific type of redundancy, as will be explored in more detail below, in embodiments of the present invention, a complex redundant sensor system can include two (or more) sensors, where at least two of the sensors are different from each other in design (and can also employ different chemistries and / or sizes). Here, one (or more) of the sensors can be designed to have, for example, significantly better hydration and / or stability characteristics, but may not last for 2 or 3 days. On the other hand, other sensors can have durable endurance, but slower initial hydration and / or stability. In this case, an algorithm can be designed in which the first sensor is used to generate glucose data during early wear, after which the first sensor can be used to calibrate the second sensor, and then the second sensor can be switched to to generate glucose data during the remaining life of the glucose sensor system.

[0261] Sensor diagnostics involves using additional (diagnostic) information that can provide a real-time understanding of the sensor health. In this regard, it has been found that electrochemical impedance spectroscopy (EIS) provides such additional information in the form of sensor impedance and impedance-related parameters at different frequencies. Additionally, advantageously, it has further been found that for certain frequency ranges, the impedance and / or impedance-related data is substantially glucose-independent. This glucose-independence not only enables a variety of EIS-based markers or indicators to generate robust, highly reliable sensor glucose values (through a fusion method, which will be described in more detail below), but also enables the assessment of the condition, health, lifespan, and efficiency of individual electrodes and the overall sensor to be substantially independent of the glucose-related Isig.

[0262] For example, the analysis of impedance data unrelated to glucose uses values such as 1 kHz real impedance, 1 kHz imaginary impedance, and Nyquist slope to provide information about the efficiency of the sensor in terms of its hydration and readiness for data acquisition (which will be described in more detail below). Additionally, impedance data unrelated to glucose provides information about potential occlusions that may be present on the surface of the sensor membrane, which may temporarily prevent glucose from entering the sensor and thus cause a signal drop (using, for example, the value of 1 kHz real impedance). Further, impedance data unrelated to glucose uses values such as phase angle and / or imaginary impedance at 1 kHz and higher frequencies to provide information about loss of sensor sensitivity during long-term wear - potentially due to local hypoxia at the insertion site.

[0263] In the context of (electrode) redundancy and EIS, a fusion algorithm can be used to obtain the diagnostic information provided by EIS for each redundant electrode and independently assess the reliability of each electrode. Then weights can be added to each independent signal as a measure of reliability, and a single fused signal can be calculated that can be used to generate the sensor glucose value seen by the patient / subject. From the foregoing, it can be seen that the combined use of redundancy, sensor diagnostics using EIS, and an EIS-based fusion algorithm results in a more reliable overall CGM system. Additionally, EIS diagnostics can automatically check the health of each electrode without reference to glucose values (finger sticks), thus reducing the number of reference values required.

[0264] The EIS or AC impedance method studies the response of a system to a periodically applied small-amplitude AC signal. This is schematically shown in Figure 15A where E is the applied potential, I is the current, and impedance (Z) is defined as ΔE / ΔI. However, although impedance can itself be simply defined mathematically as ΔE / ΔI, to date, there has been no commercial success in applying EIS technology to continuous glucose monitoring. This is partly because glucose sensors are very complex systems, and so far, no mathematical model has been developed that can fully explain the complexity of the EIS output of glucose sensors.

[0265] Figure 15B A simplified circuit model that has been used to describe electrochemical impedance spectroscopy is shown in. In this illustration, IHP represents the inner Helmholtz plane, OHP represents the outer Helmholtz plane, CE is the counter electrode, WE is the working electrode, C d is the double-layer capacitance, R p is the polarization resistance, Z w is the Warburg impedance, and R s is the solution resistance. The latter four components - double-layer capacitance (C d ), Warburg impedance (Z w)、Polarization resistance (R p ) and solution resistance (R s ) - each of which can play an important role in sensor performance and can be measured separately by applying a low - frequency or high - frequency alternating operating potential. For example, the Warburg impedance is closely related to the diffusion impedance of an electrochemical system - mainly the low - frequency impedance - and thus exists in all diffusion - limited electrochemical sensors. Therefore, by associating one or more of these components with one or more components and / or layers of a glucose sensor, EIS technology can be used as a sensor diagnostic tool.

[0266] It is well known that impedance can be defined in terms of its magnitude and phase, where the magnitude (|Z|) is the ratio of the voltage - difference amplitude to the current amplitude, and the phase (θ) is the phase shift of the current ahead of the voltage. When a circuit is driven only with direct current (DC), the impedance is the same as the resistance, i.e., resistance is a special case of impedance with a zero phase angle. However, as a complex number, impedance can also be represented by its real and imaginary parts. In this regard, the following equations can be used to derive the real and imaginary impedances from the impedance magnitude and phase:

[0267] Real impedance (ω) = magnitude (ω) x cos(phase (ω) / 180xπ)

[0268] Imaginary impedance (ω) = magnitude (ω) x sin(phase (ω) / 180xπ)

[0269] where ω represents the input frequency at which the magnitude (in ohms) and phase (in degrees) are measured. The relationship between impedance on the one hand and current and voltage on the other hand - including how impedance can be calculated from measured values of current and voltage - will be explored more fully below in connection with sensor electronics that includes an application - specific integrated circuit (ASIC), which has been developed for embodiments of the invention described herein.

[0270] Continue Figure 15B For the circuit model shown, the total system impedance can be simplified to:

[0271]

[0272] where Z w (ω) is the Warburg impedance, ω is the angular velocity, j is the imaginary unit (used instead of the traditional "i" to avoid confusion with current), and C d 、R p and R s are the double - layer capacitance, polarization resistance, and solution resistance (as previously defined), respectively. The Warburg impedance can be calculated as

[0273]

[0274]

[0275]

[0276] where D is the diffusion rate, L is the sensor membrane thickness, C is the peroxide concentration, and m:1 / 2 corresponds to a 45° Nyquist slope.

[0277] A Nyquist plot is a graphical representation where the real part of the impedance (Real Z) is plotted against the imaginary part of the impedance (Img Z) over the entire frequency spectrum. Figure 16A A general example of a Nyquist plot is shown, where the X value is the real part of the impedance and the Y value is the imaginary part of the impedance. The phase angle is the angle between the impedance point (X,Y) - which defines a vector with magnitude |Z| - and the X-axis.

[0278] Figure 16A The Nyquist plot of... is generated by applying an AC voltage plus a DC voltage (DC bias) between the working electrode and the counter electrode at a selected frequency (i.e., a swept frequency) from 0.1 Hz to 1000 MHz. Starting from the right, the frequency increases from 0.1 Hz. For each frequency, the real and imaginary impedances can be calculated and plotted. As shown, a typical Nyquist plot of an electrochemical system may look like a semicircle connected to a straight line at an inflection point, where the semicircle and the straight line represent the plotted impedances. In some embodiments, the impedance at the inflection point is of particular interest because it is the most easily identifiable in the Nyquist plot and can define the intercept. Typically, the inflection point is close to the X-axis, and the X value of the inflection point is approximately the sum of the polarization resistance and the solution resistance (R p +R s ).

[0279] Reference Figure 16B , the Nyquist plot can generally be described with respect to a low-frequency region 1610 and a high-frequency region 1620, where the labels "high frequency" and "low frequency" are used in a relative sense and are not meant to be limiting. Thus, for example, the low-frequency region 1610 can exemplarily contain data points obtained for a frequency range between approximately 0.1 Hz and approximately 100 Hz (or higher), and the high-frequency region 1620 can exemplarily contain data points obtained for a frequency range between approximately 1 kHz (or lower) and approximately 8 kHz (or higher). In the low-frequency region 1610, the Nyquist slope represents the gradient of the linear fit 1630 of the low-frequency data points in the Nyquist plot. As shown, in the high-frequency region 1620, the value of the imaginary impedance is minimal and can become negligible. Thus, the intercept 1600 is essentially the value of the real impedance at higher frequencies (e.g., in this case, approximately in the range of 1 kHz to 8 kHz). In Figure 16B , the intercept 1600 is at approximately 25 kiloohms.

[0280] Figure 16C and Figure 16D illustrate how the glucose sensor responds to a sinusoidal (i.e., alternating) working potential. In these figures, GLM is the glucose-limiting membrane of the sensor, AP is the adhesive promoter, HSA is human serum albumin, GOX is the glucose oxidase (layer), E dc is the DC potential, E ac is the AC potential, and C′ 过氧化物 is the peroxide concentration during AC application. As Figure 16C shown, if the sensor diffusion length, which varies with the AC potential frequency, molecular diffusivity, and membrane thickness, is smaller than the membrane (GOX) length, then the system gives a relatively linear response (i.e., infinite) with a constant phase angle. Conversely, if the diffusion length equals the membrane (GOX) length, then the system response becomes finite, resulting in a semicircular Nyquist plot, as Figure 16D shown. The latter typically applies to low-frequency EIS, where non-Faradaic processes are negligible.

[0281] When performing EIS analysis, AC voltages and DC biases of various frequencies can be applied, for example, between a working electrode and a reference electrode. In this regard, EIS is an improvement over previous methods that may have limited the application to simple DC currents or single-frequency AC voltages. Although EIS can typically be performed at frequencies in the range of μHz to MHz, in the embodiments of the present invention described herein, a narrower frequency range (e.g., between about 0.1 Hz and about 8 kHz) may be sufficient. Thus, in some embodiments, an AC potential can be applied in the frequency range between about 0.1 Hz and about 8 kHz, where the programmable amplitude is up to at least 100 mV, and preferably about 50 mV.

[0282] In the above frequency range, relatively high frequencies, i.e., typically between about 1 kHz and about 8 kHz, are used to examine the capacitive properties of the sensor. Depending on the membrane thickness and permeability, at relatively high frequencies, the typical range of impedance can be, for example, between about 500 ohms and 25 kΩ, and the typical range of phase can be, for example, between 0 degrees and -40 degrees. On the other hand, relatively low frequencies, i.e., typically between about 0.1 Hz and about 100 Hz, are used to examine the resistive properties of the sensor. Here, depending on the electrode design and degree of metallization, the typical operating range for the output real impedance can be, for example, between about 50 kΩ and 300 kΩ, and the typical range for the phase can be between about -50 degrees and about -90 degrees. The above illustrative ranges are shown, for example, in the Bode plots of Figure 16E and Figure 16F .

[0283] As previously mentioned, the phrases "high frequency" and "low frequency" are meant to be used relative to each other, not in an absolute sense, and they, along with the typical impedance and phase ranges mentioned above, are intended to be illustrative rather than restrictive. However, the basic principle remains the same: the capacitive and resistive characteristics of a sensor can be examined by analyzing impedance data over the entire spectrum, where, generally, lower frequencies provide information about more resistive components (such as electrodes, etc.), while higher frequencies provide information about capacitive components (such as membranes). However, the actual frequency range in each case depends on the overall design, including, for example, the type of electrode, the surface area of the electrode, the membrane thickness, the permeability of the membrane, and so on. Also see regarding the general correspondence between high-frequency circuit components and the sensor membrane, and between low-frequency circuit components and Faradaic processes involving, for example, electrodes Figure 15B .

[0284] EIS can be used in sensor systems where the sensor includes a single working electrode, as well as in sensor systems where the sensor includes multiple (redundant) working electrodes. In one embodiment, EIS provides valuable information about the lifespan (or aging) of the sensor. Specifically, at different frequencies, the magnitude and phase angle of the impedance change. As seen in Figure 17 , the sensor impedance, particularly the sum of Rp and Rs, reflects the sensor lifespan as well as the operating conditions of the sensor. Thus, as seen from the different graphs in Figure 17 , new sensors typically have a higher impedance than used sensors. In this way, by considering the X value of the sum of Rp and Rs, a threshold can be used to determine when the lifespan of the sensor has exceeded the specified operating lifespan of the sensor. It should be noted that although for the illustrative examples shown in Figure 17 through Figure 21 and discussed below, the value of the real impedance at the inflection point (i.e., Rp + Rs) is used to determine the aging, status, stability, and hydration of the sensor, alternative embodiments may use other EIS-based parameters in addition to or instead of the real impedance, such as the imaginary impedance, phase angle, Nyquist slope, etc.

[0285] Figure 17An example of a Nyquist plot over the useful life of a sensor is shown. The points indicated by the arrows are the corresponding inflection points for each scan in the spectrum. For example, before initialization (at time t = 0), Rs+Rp is higher than 8.5 kΩ, and after initialization (at time t = 0.5 hr), the value of Rs+Rp drops below 8 kΩ. Over the next six days, Rs+Rp continues to decrease, such that at the end of the specified sensor life, Rs+Rp drops below 6.5 kΩ. Based on such examples, a threshold can be set to specify when an Rs+Rp value indicates the end of the specified operating life of the sensor. Thus, EIS technology allows for closing the loop that permits reuse of the sensor after a specified operating time. In other words, if a patient attempts to reuse the sensor after it has reached its specified operating time by disconnecting and then reconnecting the sensor again, then EIS will measure an abnormally low impedance, enabling the system to reject the sensor and prompt the patient to replace it with a new sensor.

[0286] Additionally, EIS can detect sensor failure by detecting when the impedance of the sensor drops below a low impedance threshold level, indicating that the sensor may be overly worn and unable to operate properly. The system can then terminate the sensor before its specified operating life. As will be explored in more detail below, sensor impedance can also be used to detect other sensor failure (modes). For example, when the sensor enters a low current state (i.e., sensor failure) for various reasons, the sensor impedance may also increase above a certain high impedance threshold. If the impedance becomes abnormally high during sensor operation, for example due to protein or polypeptide fouling, macrophage attachment, or any other factor, the system can also terminate the sensor before its specified operating life.

[0287] Figure 18 Shows how EIS technology can be applied during sensor stabilization and when detecting the life of a sensor, according to certain embodiments. Figure 18 The logic at 1800 begins after the aforementioned hydration procedure and sensor initialization procedure have been completed. In other words, the sensor has been considered fully hydrated, and the first initialization procedure has been applied to initialize the sensor. The initialization procedure can preferably take the form of a voltage pulse, as previously described in the detailed description. However, in alternative embodiments, different waveforms can be used for the initialization procedure. For example, a sine wave can be used instead of a pulse to accelerate the wetting or conditioning of the sensor. Additionally, it may be necessary for a portion of the waveform to be greater than the normal operating voltage of the sensor, i.e., 0.535 volts.

[0288] At block 1810, an EIS procedure is applied and the impedance is compared with both a first high threshold and a first low threshold. Examples of the first high threshold and the first low threshold are 7 kiloohms and 8.5 kiloohms, respectively, but they can be set higher or lower as needed. If the impedance (e.g., Rp+Rs) is higher than the first high threshold, then the sensor undergoes additional initialization procedures (e.g., applying one or more additional pulses) at block 1820. Ideally, the total number of initialization procedures applied to initialize the sensor will be optimized to limit the impact on the sensor battery life and the total amount of time required to stabilize the sensor. Thus, by applying EIS, fewer initializations can be performed first, and the number of initializations can be incrementally added to provide only the correct number of initializations to prepare the sensor for use. Similarly, in an alternative embodiment, EIS can be applied to the hydration procedure to minimize the number of initializations required to assist the hydration process, as shown in FIGS. 13 to 14.

[0289] On the other hand, if the impedance (e.g., Rp+Rs) is lower than the first low threshold, then the sensor will be determined to be faulty and will be immediately terminated at block 1860. A message will be given to the user to replace the sensor and start the hydration process again. If the impedance is within the high and low thresholds, then the sensor will start normal operation at block 1830. Then, the logic proceeds to block 1840, where additional EIS is performed to check the life of the sensor. When the logic first reaches block 1840, the microcontroller will perform EIS to measure the life of the sensor, thus filling the gap that the user can insert and remove the same sensor. In future iterations of the EIS procedure, when the logic returns to block 1840, the microprocessor will perform EIS at fixed intervals during the specified life of the sensor. In a preferred embodiment, the fixed interval is set to every 2 hours, however, longer or shorter time periods can be easily used.

[0290] At block 1850, the impedance is compared to a second set of high and low thresholds. An example of such a second high threshold and low threshold can be 5.5 kilo-ohms and 8.5 kilo-ohms respectively, but the values can be set higher or lower as needed. As long as the impedance value remains within the second high and low thresholds, the logic proceeds to block 1830, where the sensor operates normally until a specified sensor lifetime, such as 5 days, is reached. Of course, as described with respect to block 1840, the EIS will be performed at regularly scheduled intervals throughout the specified sensor lifetime. However, if after performing the EIS, it is determined at block 1850 that the impedance has dropped below the second lower threshold or risen above the second higher threshold, then the sensor is terminated at block 1860. In other alternative embodiments, a secondary check can be implemented for faulty sensor readings. For example, if the EIS indicates that the impedance is outside the range of the second high and low thresholds, then the logic can perform a second EIS to confirm that the second set of thresholds is indeed not met (and that the first EIS was performed correctly) before ending the sensor at block 1860.

[0291] Figure 19 Depending on the above description and details the possible schedules for performing diagnostic EIS procedures. Each diagnostic EIS procedure is optional and no diagnostic EIS procedures can be scheduled or any combination of one or more diagnostic EIS procedures can be made as needed. Figure 19 The schedule begins at the insertion of the sensor at point 1900. After the sensor is inserted, the sensor undergoes a hydration period 1910. This hydration period is important because an inadequately hydrated sensor may give the user inaccurate readings, as previously mentioned. During this hydration period 1910, a first optional diagnostic EIS procedure at point 1920 is scheduled to ensure that the sensor is adequately hydrated. The first diagnostic EIS procedure 1920 measures the sensor impedance value to determine if the sensor has been adequately hydrated. If the first diagnostic EIS procedure 1920 determines that the impedance is within the set high and low thresholds, which indicates adequate hydration, then the sensor controller will allow the start of the sensor at point 1930. Conversely, if the first diagnostic EIS procedure 1920 determines that the impedance is outside the set high and low thresholds, which indicates inadequate hydration, then the sensor hydration period 1910 can be extended. After the extended hydration, once a certain capacitance is reached between the electrodes of the sensor, which means the sensor is adequately hydrated, the start at point 1930 can be made.

[0292] After the sensor is activated at point 1930 but before sensor initialization begins at point 1950, a second optional diagnostic EIS procedure 1940 is scheduled. This scheduled second diagnostic EIS procedure 1940 can detect whether the sensor is being reused before initialization begins at 1950. The test to determine whether the sensor is being reused is described in detail in Figure 18 . However, different from the previous description regarding Figure 18 , in the case where the aging test is performed after initialization is completed, the aging test is shown in Figure 19 to be performed before initialization. It is important to recognize that the timeline of the EIS procedure described in Figure 19 can be rearranged without affecting the overall teachings of the application, and the order of some steps can be interchanged. As previously mentioned, the second diagnostic EIS procedure 1940 detects a reused sensor by determining the impedance value of the sensor and then comparing the impedance value with a set high threshold and a set low threshold. If the impedance exceeds the set threshold, which indicates that the sensor is being reused, the sensor may be rejected and the user is prompted to replace it with a new sensor. This can avoid complications caused by reusing an old sensor. Conversely, if the impedance is within the set threshold, then sensor initialization 1950 can begin with the confidence that a new sensor is being used.

[0293] After initialization begins at point 1950, a third optional diagnostic EIS procedure 1960 is scheduled. The third diagnostic EIS procedure 1960 tests the impedance value of the sensor to determine whether the sensor is fully initialized. The third diagnostic EIS procedure 1960 should be performed for the minimum amount of time required to fully initialize any sensor. When performed at this time, the sensor life is maximized by limiting the time that a fully initialized sensor is not in use, and over-initialization is avoided by confirming the full initialization of the sensor before performing too much initialization. Preventing over-initialization is important because over-initialization can cause current suppression, which can lead to inaccurate readings. However, under-initialization is also a problem, so if the third diagnostic EIS procedure 1960 indicates under-initialization of the sensor, then optional initialization at point 1970 can be performed to fully initialize the sensor. Under-initialization is disadvantageous because it can produce an excessive current that is not related to the actual glucose concentration. Due to the risks of under-initialization and over-initialization, the third diagnostic EIS procedure plays an important role in ensuring proper operation of the sensor when in use.

[0294] Alternatively, a periodic diagnostic EIS procedure 1980 can be scheduled at an optional time after the sensor is fully initialized. The EIS procedure 1980 can be scheduled at any set interval. As will be discussed in more detail below, the EIS procedure 1980 can also be triggered by other sensor signals, such as an abnormal current or an abnormal counter electrode voltage. Additionally, fewer or more EIS procedures 1980 can be scheduled as needed. In a preferred embodiment, the EIS procedure used during the hydration process, sensor life check, initialization process, or periodic diagnostic test is the same procedure. In an alternative embodiment, depending on the need to focus on a particular impedance range, the EIS procedure can be shortened or lengthened for various EIS procedures (i.e., fewer or more ranges of inspection frequencies). The periodic diagnostic EIS procedure 1980 monitors impedance values to ensure that the sensor continues to operate at an optimal level.

[0295] If the sensor current drops due to contaminant species, sensor life, or a combination of contaminant species and sensor life, then the sensor may not be able to operate at an optimal level. A sensor that has aged beyond a certain duration is no longer useful, but a sensor hindered by contaminant species may potentially be repaired. Contaminant species can reduce the surface area of the electrode or the diffusion path of the analyte and reaction by-products, resulting in a drop in the sensor current. These contaminant species are charged and gradually accumulate on the electrode or membrane surface at a certain voltage. Previously, contaminant species would undermine the effectiveness of the sensor. Now, if the periodic diagnostic EIS procedure 1980 detects impedance values indicating the presence of contaminant species, then remedial measures can be taken. Regarding Figure 20 describing when remedial measures are taken. Thus, the periodic diagnostic EIS procedures 1980 become very useful because they can trigger sensor remedial measures, which can restore the sensor current to a normal level and extend the life of the sensor. Two possible embodiments of sensor remedial measures are described in the descriptions below in Figure 21A and Figure 21B .

[0296] Additionally, when certain events are determined to be imminent, any scheduled diagnostic EIS procedure 1980 can be suspended or rescheduled. Such events can include any situation where the patient is required to check the sensor readings, including, for example, when the patient uses a test strip meter to measure his or her BG level to calibrate the sensor, when the patient is warned of a calibration error and needs to use the test strip meter again to measure his or her BG level, or when a hyperglycemic or hypoglycemic warning has been issued but not confirmed.

[0297] Figure 20 shows a method of combining a diagnostic EIS procedure with sensor remedial measures. The block 2000 diagnostic procedure can be as Figure 19Any periodic diagnostic EIS procedure 1980 described in detail. The logic of this method begins when the diagnostic EIS procedure is executed at block 2000 to detect the impedance value of the sensor. As noted, in a particular embodiment, the EIS procedure applies a combination of a DC bias and an AC voltage of variable frequency, where the impedance detected by executing the EIS procedure is mapped on a Nyquist plot, and the inflection point in the Nyquist plot approximates the sum of the polarization resistance and the solution resistance (i.e., the real impedance value). After the diagnostic EIS procedure at block 2000 detects the impedance value of the sensor, the logic moves to block 2010.

[0298] At block 2010, the impedance value is compared with a set high threshold and a low threshold to determine whether the impedance value is normal. If at block 2010 the impedance is within the set bounds of the high threshold and the low threshold, then normal sensor operation is resumed at block 2020, and Figure 20 the logic will end until the time for another diagnostic EIS procedure is scheduled. Conversely, if at block 2010 it is determined that the impedance is abnormal (i.e., outside the set bounds of the high threshold and the low threshold), then a remedial action is triggered at block 2030. Examples of the high threshold and the low threshold that can be acceptable during the sensor lifetime are 5.5 kiloohms and 8.5 kiloohms respectively, but they can be set higher or lower as needed.

[0299] The remedial action of block 2030 is performed to remove any contaminant species that may have caused the abnormal impedance value. In a preferred embodiment, the remedial action is performed by applying a reverse current or a reverse voltage between the working electrode and the reference electrode. The details of the remedial action will be described in more detail with respect to FIG. 21. After the remedial action is performed at block 2030, the impedance value is retested at block 2040 by the diagnostic EIS procedure. Then, when the impedance value from the diagnostic EIS procedure at block 2040 is compared with the set high threshold or low threshold, it is determined at block 2050 whether the remedial action was successful. As in block 2010, if the impedance is within the set threshold, then this is considered normal, and if the impedance is outside the set threshold, then this is considered abnormal.

[0300] If it is determined at block 2050 that the impedance value of the sensor has returned to normal, then normal sensor operation at block 2020 will proceed. If the impedance remains abnormal, which indicates that sensor aging is the cause of the impedance anomaly or that the remedial measures have failed to successfully remove the contaminant species, then the sensor is terminated at block 2060. In an alternative embodiment, instead of immediately terminating the sensor, the sensor can generate a sensor message that first requests the user to wait and then performs further remedial measures after a set period of time. This alternative step can be combined with separate logic to determine whether, after performing the initial remedial measures, the impedance value is getting closer to within the bounds of the high and low thresholds. For example, if no change in the sensor impedance value is detected, then the sensor can decide to terminate. However, if the sensor impedance value is getting closer to the preset bounds but still outside the bounds after the initial remedial measures, then additional remedial measures can be performed. In yet another alternative embodiment, the sensor can generate a message requesting the user to calibrate the sensor by performing a finger stick meter measurement to further confirm whether the sensor has indeed failed. All of the above embodiments are intended to prevent the user from using a faulty sensor that produces inaccurate readings.

[0301] Figure 21A An embodiment of the foregoing sensor remedial measures is shown. In this embodiment, the blockage created by contaminant species is removed by reversing the voltage applied to the sensor between the working electrode and the reference electrode. The reversed DC voltage lifts the charged contaminant species from the electrode or membrane surface, thus clearing the diffusion path. Through the cleared path, the current of the sensor returns to normal levels and the sensor can provide accurate readings. Thus, the remedial measures save the user time and money associated with replacing an otherwise functional sensor.

[0302] Figure 21B An alternative embodiment of the previously mentioned sensor remedial measures is shown. In this embodiment, the reversed DC voltage applied between the working electrode and the reference electrode is coupled with an AC voltage. By adding the AC voltage, certain tightly adsorbed species or species on the surface layer can be removed because the AC voltage can further extend the force of the AC voltage from the electrode and penetrate all layers of the sensor. The AC voltage can take any number of different waveforms. Some examples of waveforms that can be used include square waves, triangular waves, sine waves, or pulses. As in the previous embodiment, once the contaminant species are cleared, the sensor can return to normal operation and both sensor life and accuracy are improved.

[0303] Although the above examples mainly illustrate the use of real impedance data in sensor diagnostics, embodiments of the invention described herein also contemplate the use of other EIS-based and substantially analyte-independent parameters (other than real impedance) in sensor diagnostic procedures. For example, as previously described, analysis of impedance data (substantially) independent of glucose provides information about the efficiency of the sensor in terms of its hydration and rate of readiness for data acquisition, such impedance data being, for example, the values of the 1 kHz real impedance and 1 kHz imaginary impedance, as well as the Nyquist slope. Additionally, impedance data (substantially) independent of glucose, such as the value of the 1 kHz real impedance, provides information about potential occlusions that may be present on the surface of the sensor membrane, which occlusions may temporarily prevent glucose from entering the sensor and thus cause a signal drop.

[0304] Furthermore, impedance data (substantially) independent of glucose, such as the values of the high-frequency phase angle and / or imaginary impedance at 1 kHz and higher frequencies, provides information about loss of sensor sensitivity during long-term wear, which loss of sensitivity may be due to local hypoxia at the insertion site. In this regard, the basic mechanism of sensitivity loss due to hypoxia can be described as follows: when there is local hypoxia, the sensor output (i.e., Isig and SG) will depend on oxygen rather than glucose, and thus, the sensor loses sensitivity to glucose. Other markers, including the 0.1 Hz real impedance, the counter electrode voltage (Vcntr), and EIS-induced spikes in Isig, can also be used to detect sensitivity loss due to hypoxia. Additionally, in a sensor system with redundant electrodes, the relative differences in the 1 kHz real impedance, 1 kHz imaginary impedance, and 0.1 Hz real impedance between two or more working electrodes can be used to detect sensitivity loss due to biofouling.

[0305] According to embodiments of the invention described herein, EIS-based sensor diagnostics requires consideration and analysis of EIS data related to one or more of at least three main factors, namely potential sensor failure modes: (1) signal initiation; (2) signal drop; and (3) sensitivity loss. Importantly, it has been found that most impedance-related parameters used in such diagnostic analyses and procedures can be studied at a certain frequency or within a certain frequency range, where the parameters are substantially independent of the analyte, and this finding allows the implementation of sensor diagnostic procedures independent of the analyte level in the patient. Thus, while EIS-based sensor diagnostics can be triggered by, for example, large fluctuations in Isig related to the analyte, the impedance-related parameters used in such sensor diagnostic procedures are themselves substantially independent of the analyte level. As will be explored in more detail below, it has also been found that in most cases where glucose can be seen to have an effect on the magnitude (or other characteristics) of EIS-based parameters, this effect is typically small enough that, for example, there is at least an order of magnitude difference between the EIS-based measurement and the effect of glucose on the measurement, and thus glucose can be screened out of the measurement, for example, by software in the IC.

[0306] By definition, "initiation" refers to the integrity of the sensor signal during the first few hours after insertion (e.g., t = 0 to 6 hours). For example, in many current devices, the signal during the first 2 hours after insertion is considered unreliable and thus the patient / user does not see the sensor glucose value. In cases where the sensor takes a longer amount of time to hydrate, the sensor signal is low for several hours after insertion. By using EIS, additional impedance information can be obtained immediately after inserting the sensor (by running an EIS program). In this regard, the total impedance equation can be used to explain the principle of using 1 kHz real impedance for low initiation detection. At relatively high frequencies (in this case, 1 kHz and above), the imaginary impedance is very small (confirmed by in vivo data), and thus the total impedance reduces to:

[0307]

[0308] As the wetting of the sensor gradually completes, the double-layer capacitance (C d ) increases. As a result, the total impedance will decrease because, as shown in the above equation, the total impedance is inversely proportional to C d . This is shown in the form of an intercept of 1600 on the real impedance axis as, for example, Figure 16B shown. Importantly, 1 kHz imaginary impedance can also be used for the same purpose because it also contains a capacitance component and is inversely proportional to the capacitance component.

[0309] Another marker for low start-up detection is the Nyquist slope, which depends only on the impedance at relatively low frequencies, which in turn corresponds to the Warburg impedance component of the total impedance (see, for example, Figure 15B ). Figure 22 Figure Figure 22 shows the Nyquist plot of a sensor operating normally, where arrow A indicates the passage of time starting at t = 0, i.e., the sensor wear time. Thus, immediately after the sensor is inserted (time t = 0), EIS is performed at a relatively low frequency, which generates real and imaginary impedance data that are plotted with a first (Nyquist) slope by a first linear fit 2200. A second (lower) sweep frequency is run at time intervals after t = 0, and the second sweep frequency generates a second linear fit 2210 that has a second (Nyquist) slope greater than the first Nyquist slope, and so on. As the sensor becomes more hydrated, the Nyquist slope increases and the intercept decreases, as reflected by lines 2200, 2210, etc., which become steeper and move closer to the Y-axis. In combination with low start-up detection, clinical data indicate that the Nyquist slope typically increases significantly after sensor insertion and initialization and then stabilizes to a certain level. One explanation for this is that as the sensor gradually becomes wet, both the species diffusion rate and concentration undergo significant changes, which are reflected in the Warburg impedance.

[0310] In Figure 23A , the Isig 2230 for the first working electrode WE1 starts lower than expected (at about 10 nA) and takes some time to catch up with the Isig 2240 of the second working electrode WE2. Thus, in this particular instance, WE1 is represented as having a low start-up. The EIS data reflects this low start-up in two ways. First, as Figure 23A shows, the real impedance of WE1 at 1 kHz (2235) is much higher than the 1 kHz real impedance 2245 of WE2. Second, when compared to the Nyquist slope of WE2 ( Figure 23C ), the Nyquist slope of WE1 ( Figure 23B ) starts out lower, has a larger intercept 2237, and takes more time to stabilize. As will be discussed later, these two characteristics, the 1 kHz real impedance and the Nyquist slope, can be used as diagnostic inputs in a fusion algorithm to decide which of the two electrodes can carry a higher weight when calculating the fused signal. Additionally, one or both of these markers can be used in a diagnostic procedure to determine whether the sensor as a whole is acceptable or whether it should be terminated and replaced.

[0311] By definition, a signal (or Isig) dip is an instance of a low sensor signal that is mostly transient in nature, e.g., on the order of a few hours. Such a low signal can be caused, for example, by some form of biological occlusion on the sensor surface or by pressure applied at the insertion site (e.g., while sleeping on the side). During this period, the sensor data is considered unreliable; however, the signal does eventually recover. In EIS data, this type of signal dip, as opposed to those caused by changes in blood glucose within the patient, is reflected in the 1 kHz real impedance data, as Figure 24 shown.

[0312] Specifically, in Figure 24 , the Isig 2250 of the first working electrode WE1 and the Isig 2260 of the second working electrode WE2 both start at approximately 25 nA at the far left (i.e., 6 PM). As time progresses, both Isigs fluctuate, which reflects glucose fluctuations near the sensor. For approximately the first 12 hours or so (i.e., until approximately 6 AM), both Isigs are fairly stable, as are their corresponding 1 kHz real impedances 2255, 2265. However, between approximately 12 and 18 hours, i.e., between 6 AM and noon, the Isig 2260 of WE2 starts to dip and continues to trend downward over the next few hours until approximately 9 PM. During this period, the Isig 2250 of WE1 also exhibits a certain dip, but is much more stable and has a much smaller dip magnitude compared to the Isig 2260 of WE2. The characteristics of the Isigs of WE1 and WE2 are also reflected in their corresponding 1 kHz real impedance data. Thus, as Figure 24 shown, during the above time period, while the 1 kHz real impedance (2255) of WE1 remains fairly stable, the 1 kHz real impedance (2265) of WE2 increases significantly.

[0313] By definition, sensitivity loss is a situation where the sensor signal (Isig) becomes low and unresponsive for an extended period of time and generally does not recover. Sensitivity loss can occur for various reasons. For example, electrode poisoning can greatly reduce the effective surface area of the working electrode, thus severely limiting the current amplitude. Sensitivity loss can also occur at the insertion site due to low or lack of oxygen. Additionally, sensitivity loss can occur due to some forms of extreme surface occlusion (i.e., a more permanent form of signal dip caused by biological or other factors) that limit the passage of glucose and oxygen through the sensor membrane, thereby reducing the number / frequency of chemical reactions that generate current in the electrode and ultimately the sensor signal (Isig). It should be noted that the various causes of sensitivity loss described above apply to both short-term (7 to 10 days of wear) and long-term (6 months of wear) sensors.

[0314] In EIS data, sensitivity loss typically occurs early by increasing the absolute value of the phase (|phase|) and the imaginary impedance (|imaginary impedance|) at relatively high frequency ranges (e.g., 128 Hz and above, and 1 kHz and above, respectively). Figure 25A An example of a normally operating glucose sensor is shown, where the sensor current 2500 responds to glucose, i.e., Isig 2500 tracks glucose fluctuations, but all relevant impedance outputs, such as the 1 kHz real impedance 2510, the imaginary 1 kHz impedance 2530, and the phase (2520) at a frequency equal to or higher than approximately 128 Hz, remain stable because they are essentially glucose-independent.

[0315] Specifically, Figure 25A The top curve plot in shows that after the first few hours, the 1 kHz real impedance 2510 remains fairly stable at approximately 5 kiloohms (and the 1 kHz imaginary impedance 2530 remains fairly stable at approximately -400 ohms). In other words, at 1 kHz, the real impedance data 2510 and the imaginary impedance data 2530 are essentially glucose-independent, and thus they can be used as characteristics or independent indicators of the health, condition, and ultimately the reliability of a particular sensor in an analysis. However, as previously mentioned, different impedance-related parameters can exhibit glucose-independence at different frequency ranges, and the range can depend on the overall sensor design in each case, such as electrode type, electrode surface area, membrane thickness, membrane permeability, etc.

[0316] Thus, in the example Figure 25B for a 90% short tubeless electrode design, the top curve plot also shows that the sensor current 2501 responds to glucose, and after the first few hours, the 1 kHz real impedance 2511 remains fairly stable at approximately 7.5 kiloohms. Figure 25B The bottom curve plot in shows the real impedance data at frequencies between 0.1 Hz (2518) and 1 kHz (2511). It can be seen that the real impedance data at 0.1 Hz (2518) is highly glucose-related. However, as indicated by the reference numerals 2516, 2514, and 2512, as the frequency increases from 0.1 Hz to 1 kHz, i.e., for impedance data measured at frequencies close to 1 kHz, the real impedance becomes increasingly glucose-independent.

[0317] Return Figure 25A, the middle curve shows that at relatively high frequencies, the phase 2520 is substantially independent of glucose. However, it should be noted that the "relatively high frequency" associated with this parameter (phase) of the sensor in the analysis means a frequency of 128 Hz and above. In this regard, the curve shows that over the entire period shown, the phase at all frequencies between 128 Hz and 8 kHz is stable. On the other hand, as can be seen from the bottom curve of Figure 25C , although the phase 2522 at 128 Hz (and above) is stable, the phase 2524 fluctuates at frequencies increasingly less than 128 Hz, i.e., the phase becomes increasingly glucose-related and varies to different degrees. It should be noted that the electrode design for the Figure 25C example is the same as the electrode design for the Figure 25B , and the top curve of the former is the same as the top curve of the latter.

[0318] Figure 26 shows an example of sensitivity loss due to hypoxia at the insertion site. In this case, the insertion site becomes hypoxic after day 4 (indicated by the dark vertical line in Figure 26 ), which makes the sensor current 2600 low and unresponsive. The 1 kHz real impedance 2610 remains stable, indicating no physical blockage on the sensor. However, as indicated by the corresponding downward arrows, the changes in the phase 2622 at relatively high frequencies and the 1 kHz imaginary impedance 2632 are consistent with the loss of sensitivity, indicating that this loss is due to hypoxia at the insertion site. Specifically, Figure 26 shows that the phase at higher frequencies (2620) and the 1 kHz imaginary impedance (2630) become more negative (indicated by the dark vertical line) before the sensor loses sensitivity, and continue to decline as the sensor sensitivity continues to be lost. Thus, as described above, this sensitivity loss is anticipated or predicted by an increase in the absolute value of the phase (|phase|) and the imaginary impedance (|imaginary impedance|) in a relatively high frequency range (e.g., 128 Hz and above and 1 kHz and above, respectively).

[0319] The above features can be verified by in vitro testing, an example of which is shown in Figure 27 . Figure 27Shows the results of in vitro testing of the sensor, where hypoxia was simulated at different glucose concentrations. In the top graph, as the glucose concentration increased from 100 mg / dl (2710) to 200 mg / dl (2720), 300 mg / dl (2730), and 400 mg / dl (2740) and then decreased to 200 mg / dl (2750), Isig fluctuated with the glucose concentration. In the bottom graph, the phase at relatively high frequencies was generally stable, indicating that the phase is independent of glucose. However, at extremely low oxygen concentrations, such as at 0.1% O2, the phase at relatively high frequencies fluctuated, as indicated by the circled area and arrows 2760, 2770. It should be noted that the magnitude and / or direction of the fluctuation (i.e., positive or negative) depends on various factors. For example, the higher the ratio of glucose concentration to oxygen concentration, the greater the magnitude of the phase change. Additionally, the specific sensor design and sensor lifetime (i.e., measured by the time after implantation) can affect this fluctuation. Thus, for example, the older the sensor, the more susceptible it is to interference.

[0320] Figures 28A to 28D Shows another example of the sensitivity loss caused by hypoxia for the redundant working electrodes WE1 and WE2. As Figure 28A shown, even though the sensor current 2800 fluctuated and eventually became unresponsive, the 1 kHz real impedance 2810 was stable. And, as previously described, the change in the 1 kHz imaginary impedance 2820 was consistent with the sensitivity loss of the sensor. However, additionally, Figure 28B shows the real impedance data and imaginary impedance data (2830 and 2840, respectively) at 0.105 Hz. The imaginary impedance data can more commonly be referred to as "0.1 Hz data", which indicates that although the imaginary impedance at 0.1 Hz appears to be rather stable, the 0.1 Hz real impedance 2830 increased significantly as the sensor lost sensitivity. Additionally, as Figure 28C shown, in the case of sensitivity loss caused by hypoxia, V cntr 2850 dropped to 1.2 volts.

[0321] In summary, the figure shows the following findings: The sensitivity loss caused by hypoxia is accompanied by a lower 1 kHz imaginary impedance (i.e., the latter becomes more negative), a higher 0.105 Hz real impedance (i.e., the latter becomes more positive), and a V cntr drop. Additionally, the hypoxia process and the V cntr drop are generally accompanied by an increase in the capacitive components in the electrochemical circuit. It should be noted that in some of the diagnostic procedures to be described later, the 0.105 Hz real impedance may not be used because the real impedance data at this relatively low frequency may seem to be analyte-related.

[0322] Finally, in combination with Figures 28A to 28BFor examples, it should be noted that impedance measurements at 1 kHz or higher frequencies typically cause spikes in Isig due to EIS. This is shown in Figure 28D where the raw Isig of WE2 is plotted against time. Due to the double-layer capacitance charge, the sharp increase in Isig at the start of the spike is a non-Faradaic process. Therefore, the sensitivity loss caused by hypoxia may also be accompanied by higher EIS-induced spikes, as well as lower 1 kHz imaginary impedance, higher 0.105 Hz real impedance, and V cntr drop, as discussed above.

[0323] Figure 29 Another example of sensitivity loss is shown. This situation can be considered an extreme version of the Isig sudden drop described above in connection with Figure 24 Here, it is observed that the sensor current 2910 is low from the insertion time, indicating a problem with the insertion procedure, which results in electrode occlusion. Compared with the same parameter values of a normally operating sensor shown in Figure 25A , the 1 kHz real impedance 2920 is significantly higher, while the phase 2930 at relatively high frequencies and the 1 kHz imaginary impedance 2940 are both shifted to significantly more negative values. The shift of the phase 2930 at relatively high frequencies and the 1 kHz imaginary impedance 2940 indicates that the sensitivity loss may be due to hypoxia, which in turn may be caused by occlusion of the sensor surface.

[0324] Figures 30A to 30D Data of another redundant sensor is shown, where the relative differences in the 1 kHz real impedance, 1 kHz imaginary impedance, and 0.1 Hz real impedance between two or more working electrodes can be used to detect sensitivity loss due to biofouling. In this example, WE1 shows a greater sensitivity loss than WE2, which can be seen from the higher 1 kHz real impedance 3010, lower 1 kHz imaginary impedance 3020, and much higher real impedance of WE2 at 0.105 Hz (3030). However, additionally, in this example, V cntr 3050 does not drop. Furthermore, as shown in Figure 30D , over time, the height of the spikes in the raw Isig data does not change much. This indicates that for sensitivity loss due to biofouling, the V cntr drop and the increase in spike height are related. Additionally, the fact that the height of the spikes in the raw Isig data does not change significantly over time indicates that the capacitive component of the circuit does not change significantly over time. Therefore, the sensitivity loss due to biofouling is related to the resistive component (i.e., diffusion) of the circuit.

[0325] The various impedance-related parameters described above can be used alone or in combination as inputs to the following to generate a more reliable sensor glucose value: (1) an EIS-based sensor diagnostic procedure; and / or (2) a fusion algorithm. Regarding the former, Figure 31 illustrates how impedance-related parameters or characteristics, i.e., EIS-based data, can be used in a diagnostic procedure to determine in real time whether a sensor is operating properly or whether the sensor should be replaced.

[0326] In Figure 31 the diagnostic procedure illustrated in the flowchart of is based on periodically collecting EIS data, e.g., every hour, every half hour, every 10 minutes, or at any other interval, including continuously, which may be applicable to a particular sensor under analysis. At each such interval, the EIS can operate over the entire spectrum (i.e., "full scan"), or can operate within a selected frequency range or even at a single frequency. Thus, for example, for an hourly data collection scheme, the EIS can be performed at frequencies in the range from μHz to MHz, or can operate within a narrower frequency range, e.g., between about 0.1 Hz and about 8 kHz, as discussed above. In various embodiments, EIS data acquisition can be implemented alternately between a full scan and a narrower range spectrum, or according to other schemes.

[0327] The temporal frequency of EIS implementation and data collection can be determined by various factors. For example, each implementation of the EIS consumes a certain amount of power, which is typically provided by the sensor's battery, i.e., the battery that powers the sensor electronics, including the ASIC described later. Thus, the battery capacity and the remaining sensor life may contribute to determining the number of times the EIS is run, and the frequency breadth sampled for each such run. Additionally, particular circumstances may require monitoring of EIS parameters (e.g., real impedance at 1 kHz) at a particular frequency based on a first schedule (e.g., every few seconds or minutes), while other parameters and / or the same parameter at other frequencies can be monitored based on a second schedule (e.g., at a lower frequency). In these cases, the diagnostic procedure can be customized for a particular sensor and requirements such that battery power can be conserved and unnecessary and / or redundant EIS data acquisition can be avoided.

[0328] It should be noted that in some embodiments, e.g., Figure 31The diagnostic procedure shown requires a series of individual "tests" that are implemented to perform real-time monitoring of the sensor. Multiple tests or markers, also referred to as "multiple markers", are implemented because each time an EIS is run, i.e., each time the EIS program is executed, data may be collected on multiple impedance-based parameters or characteristics that can be used to detect the sensor status or quality, including, for example, whether the sensor has failed or is failing. When performing sensor diagnostics, there may sometimes be diagnostic tests that may indicate a fault, while other diagnostics may indicate no fault. Therefore, the availability of multiple impedance-related parameters and the implementation scheme of the multi-test program are advantageous because some of the multiple tests can act as a validity check for certain other tests. Thus, real-time monitoring using a multi-marker program can include a certain degree of built-in redundancy.

[0329] In view of the above, Figure 31 The logic of the diagnostic procedure shown starts at 3100 after the sensor has been inserted / implanted and an EIS run has been performed to provide the EIS data as input. At 3100, using the EIS data as input, it is first determined whether the sensor is still in place. Thus, if it is found that the |Z| slope is constant within the test frequency band (or range), and / or the phase angle is approximately -90°, then it is determined that the sensor is no longer in place, and a warning indicating that the sensor has been pulled out is sent to the patient / user, for example. The specific parameters (and their corresponding values) described herein for detecting sensor pull-out are based on the finding that once the sensor leaves the body and the membrane is no longer hydrated, the impedance spectral response is like that of a capacitor.

[0330] If it is determined that the sensor is still in place, then the logic moves to step 3110 to determine whether the sensor is properly initialized. As shown, the "Init. check" is performed by determining the following conditions: (i) whether |(Z n - Z1) / Z1| > 30% at 1 kHz, where Z1 is the real impedance of the first measurement and Z n is the impedance measured at the next interval, as discussed above; and (2) whether the phase angle change is greater than 10° at 0.1 Hz. If the answer to either of the questions is "yes", then the test is satisfactory, i.e., test 1 is passed. Otherwise, test 1 is marked as failed.

[0331] At step 3120, test 2 asks whether the frequency difference (f2 - f1) between two consecutive EIS runs is greater than 10 Hz at a phase angle of -45°. Again, an answer of "no" is marked as failed; otherwise, test 2 is fully satisfied.

[0332] Test 3 at step 3130 is a hydration test. Here, what is asked is the current impedance Z nIs it less than the impedance Z after initialization at 1 kHz? pi If so, then this test is satisfied; otherwise, Test 3 is marked as failed. Test 4 in Step 3140 is also a hydration test, but this time at a lower frequency. Thus, this test asks whether Z n is less than 300 kΩ at 0.1 Hz during sensor operation after initialization. Again, a "no" answer indicates that the sensor has failed Test 4.

[0333] At Step 3150, Test 5 asks whether the low-frequency Nyquist slope increases overall from 0.1 Hz to 1 Hz. As previously discussed, for a sensor operating normally, the Nyquist slope at relatively low frequencies should increase over time. Thus, if the answer to the question is "yes", then this test is satisfied; otherwise, the test will be marked as failed.

[0334] Step 3160 is the last test of this embodiment of the diagnostic program. Here, it is asked whether the real impedance decreases overall. Here, as previously discussed, in a sensor operating normally, it is expected that the real impedance should decrease over time. Thus, answering "yes" here means that the sensor is operating normally; otherwise, the sensor has failed Test 6.

[0335] Once all 6 tests have been performed, a decision is made at 3170 as to whether the sensor is operating normally or whether a failure has occurred. In this embodiment, if the sensor passes at least 3 of the 6 tests, then the sensor is determined to be operating normally (3172). In other words, in order to be determined as failed (3174), the sensor must fail at least 4 of the 6 tests. In alternative embodiments, different rules can be used to evaluate normal operation and sensor failure. Additionally, in some embodiments, when determining overall sensor operation (normal vs. failed), each test can be weighted such that the assigned weights reflect, for example, the importance of the test or the importance of the specific parameter being asked about by the test. For example, the weight of one test might be twice that of another test, but only half that of a third test, and so on.

[0336] In other alternative embodiments, different numbers of tests and / or different sets of EIS-based parameters can be used for each test. Figure 32A and Figure 32B illustrates an example of a diagnostic program for real-time monitoring that includes 7 tests. Refer to Figure 32A, the logic starts at 3200 after the sensor has been inserted / implanted and the EIS procedure has been executed to provide the EIS data as input. At 3200, using the EIS data as input, it is first determined whether the sensor is still in the proper position. Thus, if it is found that the |Z| slope is constant within the tested frequency band (or range), and / or the phase angle is approximately -90°, then it is determined that the sensor is no longer in the proper position, and a warning indicating that sensor pull-out has occurred is sent to the patient / user, for example. On the other hand, if it is determined that the sensor is in the proper position, then the logic moves to start the diagnostic check (3202).

[0337] At 3205, Test 1 is similar to Test 1 of the diagnostic procedure discussed above in connection with Figure 31 but this Test 1 specifies that a follow-up measurement of Z is made 2 hours after the first measurement n . Thus, in this example, Z n = Z 2hr . More specifically, Test 1 compares the real impedance 2 hours after (sensor implantation and) initialization with the value before initialization. Similarly, the second part of Test 1 asks whether the difference between the phase 2 hours after initialization and the phase before initialization is greater than 10° at 0.1 Hz. As previously mentioned, if the answer to either question is affirmative, then it is determined that the sensor has been properly hydrated and initialized and has passed Test 1; otherwise, the sensor has failed this test. It should be noted that although this test asks about impedance and phase changes 2 hours after initialization, depending on various factors, the time interval between any two consecutive EIS runs may be shorter or longer, such factors including, for example, sensor design, electrode redundancy level, the degree to which the diagnostic procedure includes redundant tests, battery power, etc.

[0338] Moving to 3210, the logic next performs a sensitivity loss check by asking whether the percentage change in the impedance magnitude at 1 kHz and the percentage change in the impedance magnitude in Isig after a 2-hour interval (n + 2) are greater than 30%. If the answer to both questions is "yes", then it is determined that the sensor is losing sensitivity, and thus, it is determined that Test 2 has failed. It should be noted that although Test 2 is shown herein based on a preferred percentage difference of 30%, in other embodiments, for the purpose of performing this test, the percentage difference in the impedance magnitude at 1 kHz and in Isig may be in the range of 10% to 50%.

[0339] Test 3 (at 3220) is similar to Figure 31Test 5 of the algorithm shown. Here, as previously mentioned, the question is whether the low-frequency Nyquist slope increases overall from 0.1 Hz to 1 Hz. If so, then this test passes; otherwise, the test fails. As shown in 3220, this test is also capable of setting a threshold or acceptable range for the percentage change in the low-frequency Nyquist slope, beyond which the sensor may be considered faulty or at least trigger additional diagnostic tests. In an embodiment of the present invention, such a threshold / acceptable range for the percentage change in the low-frequency Nyquist slope can be in the range of about 2% to about 20%. In some preferred embodiments, the threshold can be about 5%.

[0340] The logic then moves to 3230, which is another low-frequency test, this time involving phase and impedance magnitude. More specifically, the phase test asks whether the phase at 0.1 Hz increases continuously over time. If so, then the test fails. Like other tests that monitor the trend of a parameter, the low-frequency phase test of Test 4 is also capable of setting a threshold or acceptable range for the percentage change in the low-frequency phase, beyond which the sensor can be considered faulty or at least cause problems. In some preferred embodiments, such a threshold / acceptable range for the percentage change in the low-frequency phase can be in the range of about 5% to about 30%. In some preferred embodiments, the threshold can be about 10%.

[0341] As noted, Test 4 also includes a low-frequency impedance magnitude test, which asks whether the impedance magnitude at 0.1 Hz increases continuously over time. If so, then the test fails. It should be noted that if the phase test or the impedance magnitude test fails, then Test 4 is considered "failed". The low-frequency impedance magnitude test of Test 4 is also suitable for setting a threshold or acceptable range for the percentage change in the low-frequency impedance magnitude, beyond which the sensor can be considered faulty or at least cause problems. In some preferred embodiments, such a threshold / acceptable range for the percentage change in the low-frequency impedance magnitude can be in the range of about 5% to about 30%. In some preferred embodiments, the threshold can be about 10%, where the impedance magnitude of a typical sensor usually ranges between about 100 kiloohms and about 200 kiloohms.

[0342] Test 5 (at 3240) is another sensitivity loss check and can be considered complementary to Test 2. Here, if both the percentage change in Isig and the percentage change in the impedance magnitude at 1 kHz are greater than 30%, then it is determined that the sensor is recovering from a sensitivity loss. In other words, even if Test 2 did not detect a sensitivity loss for some reason, it is determined that the sensor has previously experienced a certain amount of sensitivity loss. As with Test 2, although Test 5 is shown based on a preferred percentage difference of 30%, in other embodiments, for the purpose of conducting this test, the percentage difference in Isig and the impedance magnitude at 1 kHz can be in the range of 10% to 50%.

[0343] Moving to 3250, Test 6 provides specific failure criteria for sensor functionality tests, which have been determined based on the observed data and a specific sensor design. Specifically, in one embodiment, if at least two of the following three criteria are met, then the sensor can be determined to have failed and thus be unable to respond to glucose: (1) Isig is less than 10 nA; (2) the imaginary impedance at 1 kHz is less than -1500 ohms; and (3) the phase at 1 kHz is less than -15°. Thus, if any two of (1) through (3) are not met, then it is determined that Test 6 has passed. It should be noted that in other embodiments, if Isig is less than approximately 5 nA to approximately 20 nA, then the Isig prong of this test may fail. Similarly, if the imaginary impedance at 1 kHz is less than approximately -1000 ohms to approximately -2000 ohms, then the second prong may fail. Finally, if the phase at 1 kHz is less than approximately -10° to approximately -20°, then the phase prong may fail.

[0344] Finally, step 3260 provides another sensitivity check where the parameter is evaluated at a low frequency. Thus, Test 7 asks whether the magnitude of the difference between the ratio (n + 2) of the imaginary impedance to Isig on the one hand and the permeable value of the ratio on the other hand at 0.1 Hz is greater than 30% of the magnitude of the previous value of the ratio. If so, then the test fails; otherwise, the test passes. Here, although Test 7 is shown based on a preferred percentage difference of 30%, in other embodiments, for the purpose of conducting this test, the percentage difference can be in the range of 10% to 50%.

[0345] Once all 7 tests have been implemented, a decision is made at 3270 as to whether the sensor is operating properly or whether a warning should be issued indicating that the sensor has failed (or may be failing). As shown, in this embodiment, if the sensor passes at least 4 of the 7 tests, then it is determined that the sensor is operating properly (3272). In other words, in order to be determined to have failed or at least caused a problem (3274), the sensor must fail at least 4 of the 7 tests. If it is determined that the sensor is "bad" (3274), then a warning of the effect can be sent to the patient / user, for example. As previously mentioned, in alternative embodiments, different rules can be used to assess the relationship between normal operation and sensor failure / problems. Additionally, in some embodiments, when determining overall sensor operation (normal vs. failed), each test can be weighted such that the assigned weights reflect, for example, the importance of the test or the importance of the particular parameter being interrogated by the test.

[0346] As previously mentioned, in the embodiments of the invention described herein, the various impedance-related parameters described above can be used, either individually or in combination, as inputs to one or more fusion algorithms to produce a more reliable sensor glucose value. Specifically, it is well known that, unlike a single-sensor (i.e., single working electrode) system, multiple sensing electrodes provide a more reliable glucose reading because multiple signals obtained from two or more working electrodes can be fused to provide a single sensor glucose value. This signal fusion utilizes the quantitative inputs provided by EIS to calculate the most reliable output sensor glucose value from the redundant working electrodes. It should be noted that although the following discussion may describe various fusion algorithms with respect to a first working electrode (WE1) and a second working electrode (WE2) as redundant electrodes, this is for illustration only and not a limitation, because the algorithms and their basic principles described herein are applicable to and can be used in redundant sensor systems having more than 2 working electrodes. Additionally, the redundant electrodes can be included in the (same) sensor on or within a single flexible portion or multiple flexible portions, or the redundant electrodes can be included in different sensors on or within a single flexible portion or multiple flexible portions (e.g., in a complex redundant sensor system having two or more sensors, where at least two of the sensors have different designs from each other).

[0347] Figure 33A and Figure 33B Top-level flowcharts of two alternative methods are shown, each of which includes a fusion algorithm. Specifically, Figure 33A is a flowchart of a fusion algorithm involving current (Isig), and Figure 33B is a flowchart for sensor glucose (SG) fusion. As can be seen from the figure, the main difference between the two methods is the calibration time. Thus, Figure 33AIt is shown that for Isig fusion, calibration 3590 is performed after fusion 3540 is completed. That is, the redundant Isigs from WE1 to WEn are fused into a single Isig 3589, and then the Isig 3589 is calibrated to produce a single sensor glucose value 3598. On the other hand, for SG fusion, calibration 3435 is completed for each individual Isig from WE1 to WEn to produce a calibrated SG value (such as 3436, 3438) for each working electrode. Thus, the SG fusion algorithm achieves independent calibration of each of the multiple Isigs, which may be preferred in some embodiments of the present invention described herein. After calibration, the multiple calibrated SG values are fused into a single SG value 3498.

[0348] Importantly, it should be noted that Figure 33A and Figure 33B each of the flowcharts shown includes a spike filtering process (3520, 3420). As discussed above regarding sensitivity loss, impedance measurements at 1 kHz or higher frequencies typically cause spikes in the Isig due to EIS. Thus, once the EIS procedure has been performed for each of the electrodes WE1 to WEn, it is preferred for both SG fusion and Isig fusion to first filter the Isigs 3410, 3412, etc. and 3510, 3512, etc. to obtain the corresponding filtered Isigs 3422, 3424, etc. and 3522, 3524, etc. The filtered Isigs are then used for Isig fusion, or first calibrated and then used for SG fusion, as described below. It will become apparent in the subsequent discussion that both of these fusion algorithms require calculating and assigning weights based on various factors.

[0349] Figure 34 Details of the fusion algorithm 3440 for SG fusion are shown. In essence, four factors need to be checked before determining the fusion weights. First, the integrity check 3450 involves determining whether each of the following parameters is within the specified range for normal sensor operation (e.g., a predetermined lower threshold and upper threshold): (i) Isig; (ii) 1 kHz real and imaginary impedances; (iii) 0.105 Hz real and imaginary impedances; and (iv) Nyquist slope. As shown, the integrity check 3450 includes a boundary check 3452 and a noise check 3456, where for each check, the above parameters are used as input parameters. It should be noted that for brevity, the real and / or imaginary impedances at one or more frequencies are presented Figures 33A to 35 above, and the real and imaginary impedances are referred to simply as "Imp" to denote impedance. Additionally, the magnitude and phase of the impedance (also shown as input Figure 33A and 33B above) can be used to calculate both the real and imaginary impedances.

[0350] The output from each of the boundary check 3452 and the noise check 3458 is the corresponding reliability index (RI) of each of the redundant working electrodes. Thus, the output from the boundary check includes, for example, RI_bound_We1(3543) and RI_bound_We2(3454). Similarly, for the noise check, the output includes, for example, RI_noise_We1(3457) and RI_noise_We2(3458). Based on the consistency with the above ranges of normal sensor operation, the boundary and noise reliability indices of each working electrode are calculated. Thus, if any parameter is outside the specified range of a particular electrode, the reliability index of the particular electrode is reduced.

[0351] It should be noted that the thresholds or ranges of the above parameters can depend on various factors, including the specific sensor and / or electrode design. However, in a preferred embodiment, the typical ranges of some of the above parameters can be, for example, as follows: the boundary threshold of the 1 kHz real impedance = [0.3e+4 2e+4]; the boundary threshold of the 1 kHz imaginary impedance = [-2e+3,0]; the boundary threshold of the 0.105 Hz real impedance = [2e+4 7e+4]; the boundary threshold of the 0.105 Hz imaginary impedance = [-2e+5 -0.25e+5]; and the boundary threshold of the Nyquist slope = [2 5]. The noise can be calculated, for example, using the second-order central difference method, where if the noise is higher than a certain percentage (e.g., 30%) of the median of each variable buffer, then the noise is considered to be outside the noise range.

[0352] Secondly, the sensor current (Isig) and the 1 kHz real impedance can be used to detect a sensor dropout. Thus, as Figure 34 shown, Isig and "Imp" are used as inputs to the dropout detection 3460. Here, the first step is to determine whether there is any divergence between the Isigs, and whether any such divergence is reflected in the 1 kHz real impedance data. This can be achieved by using the mapping 3465 between the Isig similarity index (RI_sim_isig12) 3463 and the 1 kHz real impedance similarity index (RI_sim_imp12) 3464. This mapping is crucial because it helps to avoid false positives in cases where the dropout is not true. When the Isig divergence is true, the algorithm will select the sensor with the higher Isig.

[0353] According to one embodiment, the divergence / convergence of two signals (e.g., two Isigs, or two 1 kHz real impedance data points) can be calculated as follows:

[0354] diff_va1 = abs(va1 - (va1 + va2) / 2);

[0355] diff_va2 = abs(va2 - (va1 + va2) / 2);

[0356] RI_sim = 1 - (diff_va1 + diff_va2) / (mean(abs(va1 + va2)) / 4)

[0357] Where va1 and va2 are two variables, and RI_sim (similarity index) is an index for measuring the convergence or divergence of a signal. In this embodiment, RI_sim must be restricted between 0 and 1. Thus, if RI_sim calculated as above is less than 0, then it is set to 0, and if RI_sim is greater than 1, then it is set to 1.

[0358] Mapping 3465 is performed by using ordinary linear regression (OLR). However, when OLR does not work well, robust median slope linear regression (RMSLR) can be used. For example, for the Isig similarity index and the 1kHz real impedance index, two mapping programs are required: (i) mapping the Isig similarity index to the 1kHz real impedance similarity index; and (ii) mapping the 1kHz real impedance similarity index to the Isig similarity index. These two mapping programs will generate two residuals: res12 and res21. Each of the dip reliability indices 3467, 3468 can then be calculated as follows:

[0359] RI_dip = 1 - (res12 + res21) / (RI_sim_isig + RI_sim_1K_real_impedance).

[0360] The third factor is the sensitivity loss 3470, which can be detected using, for example, the trend of the 1kHz imaginary impedance over the past 8 hours. If the trend of one sensor becomes negative, then the algorithm will rely on another sensor. If both sensors lose sensitivity, then the simple average will be taken. The 1kHz imaginary impedance is smoothed by using a powerful low-pass filter (which tends to be noisy), and the trend can be calculated by using, for example, the correlation coefficient or the linear regression with respect to time over the past 8-hour period to determine whether the correlation coefficient is negative or the slope is negative. Then, each of the sensitivity loss reliability indices 3473, 3474 is assigned a binary value of 1 or 0.

[0361] The total reliability index (RI) for each of we1, we2,..., wen is calculated as follows:

[0362] RI_we1 = RI_dip_we1 × RI_sensitivity_loss_we1 × RI_bound_we1 × RI_noise_we1

[0363] RI_we2 = RI_dip_we2 × RI_sensitivity_loss_we2 × RI_bound_we2 × RI_noise_we2

[0364] RI_we3 = RI_dip_we3 × RI_sensitivity_loss_we3 × RI_bound_we3 × RI_noise_we3

[0365] RI_we4 = RI_dip_we4 × RI_sensitivity_loss_we4 × RI_bound_we4 × RI_noise_we4 . . .

[0369] RI_we n = RI_dip_we n × RI_sensitivity_loss_we n × RI_bound_we n × RI_noise_we n

[0370] After calculating the corresponding reliability indices of individual working electrodes, the weights of each electrode can be calculated as follows:

[0371] weight_we1 = RI_we1 / (RI_we1 + RI_we2 + RI_we3 + RI_we4 + … + RI_we n )

[0372] weight_we2 = RI_we2 / (RI_we1 + RI_we2 + RI_we3 + RI_we4 + … + RI_we n )

[0373] weight_we3 = RI_we3 / (RI_we1 + RI_we2 + RI_we3 + RI_we4 + … + RI_we n )

[0374] weight_we4 = RI_we4 / (RI_we1 + RI_we2 + RI_we3 + RI_we4 + … + RI_we n ) . . .

[0378] weight_we n = RI_wen / (RI_we1 + RI_we2 + RI_we3 + RI_we4 + … + RI_we n )

[0379] Based on the above, the fused SG 3498 is calculated as follows:

[0380] SG = weight_we1 × SG_we1 + weight_we2 × SG_we2 + weight_we3 × SG_we3 + weight_we4 × SG_we4 +... + weight_we n × SG_we n

[0381] The last factor is related to artifacts in the final sensor readings, which may be caused, for example, by the instantaneous weight changes in sensor fusion. This can be avoided by applying the low-pass filter 3480 to smooth the RI of each electrode, or by applying the low-pass filter to the final SG. When using the former method, the filtered reliability indices, such as RI_We1* and RI_We2* (3482, 3484), are used to calculate the weights of each electrode and thus to calculate the fused SG 3498.

[0382] Figure 35 Details of the fusion algorithm 3540 for Isig fusion are shown. It can be seen that this algorithm is basically similar to Figure 34 the algorithm shown for SG fusion, but there are two differences. First, as previously mentioned, for Isig fusion, calibration constitutes the final step of the process, in which the single fused Isig 3589 is calibrated to produce a single sensor glucose value 3598. See also Figure 33B . Second, although SG fusion uses the SG values of multiple electrodes to calculate the final SG value 3498, the fused Isig value 3589 is calculated using the filtered Isig of multiple electrodes (3522, 3524, etc.).

[0383] In a closed-loop study involving non-diabetic subjects, it has been found that the above fusion algorithm achieves a significant improvement in mean absolute relative difference (MARD) on day 1 (when the low-startup problem is most severe and may thus have a significant impact on the accuracy and thus reliability of the sensor) and overall (i.e., over the 7-day lifespan of the sensor). The study evaluated data from an 88% distributed layout electroplated with high current density (nominal) using three different methods: (1) calculating a sensor glucose value (SG) by fusion using the Ferrari algorithm 1.0 from Medtronic Minimed (which is the SG fusion algorithm discussed above); (2) calculating an SG by using 1 kHz EIS data to identify a better ISIG value (by the Isig fusion algorithm discussed above); and (3) calculating an SG by using a higher ISIG value (i.e., without using EIS). Details of the study data are as follows:

[0384] (1) SG for an 88% distributed layout electroplated with high current density (nominal), based on Ferrari 1.0 Alg

[0385]

[0386]

[0387]

[0388]

[0389] (2) SG for an 88% distributed layout electroplated with high current density (nominal), based on better ISIG, using 1 kHz EIS

[0390]

[0391]

[0392]

[0393]

[0394] (3) SG for an 88% distributed layout electroplated with high current density (nominal), based on higher ISIG

[0395]

[0396]

[0397]

[0398]

[0399] Based on the above data, it has been found that for the first method, the MARD (%) on day 1 was 19.52%, and the total MARD was 12.28%. For the second method, the MARD on day 1 was 15.96%, and the total MARD was 11.83%. Finally, for the third method, the MARD on day 1 was 17.44%, and overall it was 12.26%. Thus, for this design with redundant electrodes, it seems that calculating SG based on better ISIG and using 1 kHz EIS (i.e., the second method) provides the greatest advantage. Specifically, the lower MARD on day 1 may be due to, for example, better low-start detection using EIS. Additionally, in this study, for WE1 and WE2, the total MARD percentage was more than 1% lower than the total average MARD of 13.5%. It should be noted that in the above methods, data transitions can be processed, for example, by filtering methods that minimize the severity of the transitions, such as by using the low-pass filter 3480 discussed above in connection with Figures 33A to 35 as described.

[0400] It is worth emphasizing that sensor diagnostics, including, for example, the assessment of low-start, sensitivity loss, and signal dropout events, depends on various factors, including sensor design, the number of electrodes (i.e., redundancy), electrode distribution / configuration, etc. Thus, the EIS-based parameters can be substantially independent of the actual frequency or frequency range of glucose, and thus one or more of the above fault modes' independent markers or predictors can also depend on the specific sensor design. For example, although it has been found that, as described above, sensitivity loss can be predicted using the imaginary impedance at relatively high frequencies - where the imaginary impedance is substantially independent of glucose - the glucose level correlation and thus the specific frequency range for using the imaginary impedance as a marker for sensitivity loss may shift (higher or lower), depending on the actual sensor design.

[0401] More specifically, as sensor designs increasingly tend to use redundant working electrodes, the size of the redundant working electrodes must become smaller and smaller to maintain the overall size of the sensor. The size of the electrodes, in turn, affects the frequency at which specific diagnostics can be queried. In this regard, it is important to note that the fusion algorithms described herein and shown in FIGS. 33A to Figure 35 are considered illustrative rather than restrictive, since each algorithm can be modified as needed based on the type of sensor in the analysis to use EIS-based parameters at frequencies that exhibit minimal glucose correlation.

[0402] Additionally, the experimental data indicates that the human body tissue structure may also affect glucose correlation at different frequencies. For example, for children, it has been found that the real impedance at 0.105 Hz is an indicator that is substantially independent of glucose for low startup detection. It is believed that this is due to changes in the tissue structure of children, such as changes in the Weber impedance, which is mainly related to the resistance component. See also the subsequent discussion regarding interferent detection.

[0403] Embodiments of the invention described herein also relate to the use of EIS in optimizing sensor calibration. As background, in current methods, the slope of the BG vs. Isig curve that can be used to calibrate subsequent Isig values is calculated as follows:

[0404]

[0405] where α is an exponential function of the time constant, β is a function of the blood glucose variance, and the offset is a constant. For a sensor under stable conditions, this method provides reasonably accurate results. As shown, for example, Figure 36 BG and Isig follow a fairly linear relationship, and the offset can be considered a constant.

[0406] However, there are cases where the above linear relationship does not hold, such as during periods when the sensor is undergoing a transition. As shown, Figure 37 it is clear that Isig-BG pairs 1 and 2 are significantly different from pairs 3 and 4 in terms of the Isig vs. BG relationship. For these types of conditions, using a constant offset tends to produce inaccurate results.

[0407] To address this issue, one embodiment relates to the use of an EIS-based dynamic offset, where EIS measurements are used to define a sensor state vector as follows:

[0408] V = {real_imp_1K, img_imp_1K, Nyquist_slope, Nyquist_R_square}

[0409] where all elements in the vector are substantially independent of BG. It should be noted that Nyquist_R_square is the R-squared of the linear regression used to calculate the Nyquist slope, i.e., the square of the correlation coefficient between the real impedance and the imaginary impedance at relatively low frequencies, and a low R-squared indicates abnormal sensor performance. For each Isig-BG pair, a state vector is assigned. If a significant difference in the state vectors is detected, such as Figure 37 |V2 - V3| in the example shown, then different offset values are assigned to 3 and 4 when compared to 1 and 2. Thus, by using this dynamic offset method, it is possible to maintain the linear relationship between Isig and BG.

[0410] In the second embodiment, an EIS-based segmentation method can be used for calibration. Using Figure 37 the instances and the vector V, it can be determined that the sensor state during 1 and 2 is significantly different from the sensor state during 3 and 4. Therefore, the calibration buffer can be divided into two segments as follows:

[0411] Isig_buffer1 = [Isig1, Isig2]; BG_buffer1 = [BG1, BG2]

[0412] Isig_buffer2 = [Isig3, Isig4]; BG_buffer2 = [BG3, BG4]

[0413] Therefore, when the sensor operates during 1 and 2, Isig_buffer1 and BG_buffer1 will be used for calibration. However, when the sensor operates during 3 and 4, i.e., during the transition period, Isig_buffer2 and BG_buffer2 will be used for calibration.

[0414] In yet another embodiment, an EIS-based dynamic slope method can be used for calibration purposes, in which EIS is used to adjust the slope. Figure 38A An example of how this method can be used to improve sensor accuracy is shown. In this figure, data points 1 to 4 are discrete blood glucose values. From Figure 38A it can be seen that there is a sensor drop 3810 between data points 1 and 3, and this drop can be detected using the above-mentioned sensor state vector V. During the drop, the slope can be adjusted upward to reduce underreading, as shown by reference numeral 3820 in Figure 38A .

[0415] In another embodiment, EIS diagnostics can be used to determine the timing of sensor calibration, which is very useful for, for example, low-start events, sensitivity loss events, and other similar situations. As is well known, most current methods require periodic calibration based on a preset schedule, such as 4 times a day. However, in the case of using EIS diagnostics, calibration becomes event-driven, and thus can only be performed when necessary and at the most effective time. Here, the state vector V can also be used to determine when the sensor state has changed and to request calibration when it has indeed changed.

[0416] More specifically, in the illustrative example, Figure 38B a flowchart of EIS-assisted sensor calibration involving low-start detection is shown. By using the Nyquist slope, the 1 kHz real impedance, and the boundary check 3850 (e.g., in combination with Figures 33A to 35For the fusion algorithm, refer to the aforementioned boundary check of the EIS-based parameters and the associated thresholds), a reliability index 3853 can be developed for startup such that when the 1 kHz real impedance 3851 and the Nyquist slope 3852 are below their corresponding upper bounds, RI_startup = 1, and the sensor is ready for calibration. In other words, the reliability index 3853 is "high" (3854), and the logic can proceed to calibration at 3860.

[0417] On the other hand, when the 1 kHz real impedance and the Nyquist slope are higher than their corresponding upper bounds (or thresholds), RI_startup = 0 (i.e., "low"), and the sensor is not ready for calibration (3856), that is, there may be a low startup problem. Here, the trends of the 1 kHz real impedance and the Nyquist slope can be used to predict when both parameters are within the range (3870). If it is estimated that this will only take a very short time (e.g., less than one hour), then the algorithm will wait until the sensor is ready, that is, until the aforementioned EIS-based parameters are within the bounds (3874), at which point the algorithm proceeds to calibration. However, if the waiting time is relatively long (3876), then the sensor can be calibrated now, and then the slope or offset can be gradually adjusted based on the trends of the 1 kHz real impedance and the Nyquist slope (3880). It should be noted that by performing the adjustment, serious over-reading or under-reading due to low startup can be avoided. As previously mentioned, the EIS-based parameters and related information used in the instant calibration algorithm are basically independent of glucose.

[0418] It should be noted that although the above description in combination with Figure 38B shows the calculation of the reliability index for starting a single working electrode, this is for illustration only and not a limitation. Therefore, in a redundant sensor containing two or more working electrodes, the boundary check can be performed for each of the multiple (redundant) working electrodes, and the startup reliability index can be calculated. Then, based on the corresponding reliability indices, at least one working electrode that can proceed to obtain a glucose measurement value can be identified. In other words, in a sensor with a single working electrode, if the working electrode exhibits low startup, then the actual use of the sensor (for measuring glucose) may have to be delayed until the low startup period ends. This period can typically be about one hour or longer, which is clearly disadvantageous. In contrast, in a redundant sensor, using the method described herein allows for adaptive or "intelligent" startup, where the electrodes that can collect data can be identified in a relatively short order, e.g., within about a few minutes. This in turn reduces the MARD because low startup typically increases the MARD by about 1 / 2%.

[0419] In yet another embodiment, the EIS can assist in adjusting the calibration buffer. For existing calibration algorithms, the buffer size is always 4, i.e., 4 Isig-BG pairs, and the weights are based on α and β, as previously described, where α is an exponential function of the time constant and β is an exponential function of the blood glucose variance. Here, the EIS can help determine when to refresh the buffer, how to adjust the buffer weights, and determine an appropriate buffer size.

[0420] In some embodiments, the EIS can also be used for interferent detection. Specifically, it may be desirable to provide a drug infusion device that includes a combined sensor and a drug infusion catheter, where the sensor is placed within the infusion catheter. In such a system, the physical location of the infusion catheter relative to the sensor may cause some problems, mainly due to the potential effects (i.e., interference) on the sensor signal, which may be caused by the drug being infused and / or its inactive components.

[0421] For example, the diluent used with insulin contains m-cresol as a preservative. In in vitro studies, it has been found that if insulin (and thus m-cresol) is infused very close to the sensor, then m-cresol will have an adverse effect on the glucose sensor. Therefore, a system in which the sensor and the infusion catheter are combined in a single needle must be able to detect and adjust the effect of m-cresol on the sensor signal. Since m-cresol affects the sensor signal, a way to detect such an interferent independently of the sensor signal itself would be preferred.

[0422] Experiments have shown that the effect of m-cresol on the sensor signal is temporary and thus reversible. However, when insulin is infused too close to the sensor, m-cresol tends to "poison" the electrode, such that the electrode is no longer able to detect glucose until the insulin (and m-cresol) has been absorbed into the patient's tissue. In this regard, it has been found that there is typically a period of about 40 minutes between the start of insulin infusion and the sensor regaining the ability to detect glucose. Advantageously, however, it has also been found that within the same time period, the 1 kHz impedance magnitude increases significantly and is completely independent of the glucose concentration.

[0423] Specifically, Figure 39 The Isig and impedance data of an in vitro experiment are shown, where the sensor is placed in a 100 mg / dL glucose solution, and the 1 kHz impedance is measured every 10 minutes, as shown by the circled data point 3920. Then m-cresol is added to bring the solution to 0.35% m-cresol (3930). It can be seen that after adding m-cresol, the Isig 3940 initially increases significantly and then starts to drift downward. Then, by adding an additional 100 mg / dL glucose, the concentration of glucose in the solution is doubled. However, this has no effect on the Isig 3940 because the electrode is unable to detect glucose.

[0424] On the other hand, m-cresol has a significant impact on both the impedance magnitude and the phase. Figure 40A A Bode plot of the phase is shown, and Figure 40B a Bode plot of the impedance magnitude before and after the addition of m-cresol is shown. It can be seen that after the addition of m-cresol, the impedance magnitude 4010 increases by at least one order of magnitude spectrally from its post-initialization value 4020. At the same time, the phase 4030 is completely changed compared to its post-initialization value 4040. In Figure 40C the Nyquist plot. Here, the pre-initialization curve 4050 and the post-initialization curve 4060 appear as expected for a normally operating sensor. However, after the addition of m-cresol, the curve 4070 becomes completely different.

[0425] The above experiment identifies an important practical flaw that continues to rely on Isig after the addition of m-cresol. Return reference Figure 39 , the patient / user monitoring the sensor signal may be mistaken that his glucose level has just reached the peak and he should administer a bolus. Then, the user administers a bolus, at which time Isig has already started to drift downward again. In other words, everything seems normal to the patient / user. However, in reality, what actually happens is that the patient has just administered an unnecessary dose of insulin, which depends on the patient's glucose level before the bolus administration, and this may put the patient at risk of experiencing a hypoglycemic event. This situation makes it even more necessary to have a method for detecting interferents that is as glucose-independent as possible.

[0426] Figure 41 Another experiment is shown, where the sensor is initialized to a 100 mg / dL glucose solution, after which the glucose is increased to 400 mg / dL over one hour and then restored to 100 mg / dL. Then m-cresol is added to increase the concentration to 0.35%, and the sensor is kept in this solution for 20 minutes. Finally, the sensor is placed in a 100 mg / dL glucose solution to allow Isig to recover after exposure to m-cresol. It can be seen that after initialization, the 1 kHz impedance magnitude 4110 is approximately 2 kiloohms. When m-cresol is added, Isig 4120 reaches a spike, as does the impedance magnitude 4110. In addition, when the sensor is restored to the 100 md / dL glucose solution, the impedance magnitude 4110 also returns to near the normal level.

[0427] As can be seen from the above experiments, EIS can be used to detect the presence of an interferent - in this case, the interferent is m-cresol. Specifically, since the interferent affects the sensor in a way that increases the impedance magnitude across the entire spectrum, the impedance magnitude can be used to detect the interference. Once the interference is detected, the sensor operating voltage can be changed to a point where no interferent is measured, or data reporting can be paused, and the sensor will indicate to the patient / user that due to the administered drug, the sensor cannot report data (until the measured impedance returns to the pre-infusion level). It should be noted that since the effect of the interferent is caused by the preservative contained in the insulin, the impedance magnitude will exhibit the same behavior as described above regardless of whether the insulin infusion is rapid or slow.

[0428] Importantly, as described above, the impedance magnitude, and of course the magnitude at 1 kHz, is essentially independent of glucose. Referring Figure 41 , it can be seen that as the glucose concentration increases from 100 mg / dL to 400 mg / dL - a four-fold increase - the 1 kHz impedance magnitude increases from approximately 2000 ohms to approximately 2200 ohms, or approximately 10%. In other words, the effect of glucose on the impedance magnitude measurement appears to be approximately one order of magnitude less than the measured impedance. This "signal-to-noise ratio" level is typically small enough to allow filtering out the noise (i.e., the glucose effect), such that the resulting impedance magnitude is essentially independent of glucose. Additionally, it should be emphasized that the impedance magnitude exhibits a higher glucose independence in actual human tissue compared to the buffer solution used in the in vitro experiments described above.

[0429] Embodiments of the invention described herein also relate to an analog front-end integrated circuit (AFE IC), which is a custom application-specific integrated circuit (ASIC) that provides the necessary analog electronics to, in particular, provide the following: (i) support multiple potentiostats and interface with an oxygen- or peroxide-based multi-terminal glucose sensor; (ii) interface with a microcontroller to form a micropower sensor system; and (iii) implement EIS diagnostics, fusion algorithms, and other EIS-based processes based on the measurement of EIS-based parameters. More specifically, the ASIC incorporates diagnostic capabilities to measure the real and imaginary impedance parameters of the sensor over a wide frequency range, and also incorporates a digital interface circuitry for two-way communication with a microprocessor chip. In addition, the ASIC includes power control circuitry, a real-time clock, and a crystal oscillator that operate at extremely low standby and operating power, such that the power to the external microprocessor can be turned off.

[0430] Figure 42A and Figure 42B shows a block diagram of the ASIC, and Table 1 below provides pad signal descriptions (in Figure 42A and Figure 42Bshown on the left side of), where some signals are multiplexed onto a single pad.

[0431]

[0432] Reference will now be made to Figure 42A and Figure 42B and Table 1 to describe the ASIC.

[0433] Power plane

[0434] The ASIC has a power plane powered by the supply pad VBAT(4210), and the power plane has an operating input range from 2.0 volts to 4.5 volts. This power plane has a regulator for reducing the voltage of some circuits in this plane. The power supply is called VDDBU(4212) and has output pads for testing and bypassing. The circuits on the VBAT power supply include an RC oscillator, a real-time clock (RC osc) 4214, a battery protection circuit, a regulator control, a power-on reset circuit (POR), and various input / outputs. The pads on the VBAT power plane are configured to draw less than 75 nA at 40 °C and VBAT = 3.50 V.

[0435] The ASIC also has a VDD power supply to supply logic. The VDD power supply voltage range can be programmed from at least 1.6 volts to 2.4 volts. The circuits on the VDD power plane include most of the digital logic, a timer (32 kHz), and a real-time clock (32 kHz). The VDD power plane includes level shifters connected to another voltage plane as needed. The level shifters in turn have interfaces that are adjusted such that if another power plane is not powered, the current increase in any powered power plane will not exceed 10 nA.

[0436] The ASIC includes an on-board regulator (with shutdown control) and the option of an external VDD source. The regulator input is a separate pad REG_VDD_IN(4216), which has electrostatic discharge (ESD) protection like other I / Os on VBAT. The on-board regulator has an output pad REG_VDD_OUT(4217). The ASIC also has an input pad for VDD, which is separate from the REG_VDD_OUT pad.

[0437] The ASIC includes an analog power plane, called VDDA(4218), which is powered by the VDD on-board regulator or an external source and is typically supplied through filtering the VDD. The circuits supplied by VDDA are configured to operate within 0.1 volts of VDD, thus eliminating the need for a level shift between the VDDA and VDD power planes. The VDDA power supplies the sensor analog circuits, analog measurement circuits, and any other noise-sensitive circuit systems.

[0438] The ASIC includes pad power VPAD for specifying digital interface signals. The pad power has an operating voltage range of at least 1.8V to 3.3V. These pads have separate power pads and are powered by an external source. The pads also incorporate level shifters into other on-board circuits to allow a flexible pad power range independent of the VDD logic power voltage. The ASIC can regulate the VPAD pad ring signal such that when the VPAD power is not enabled, the other supply currents do not increase by more than 10 nA.

[0439] Bias generator

[0440] The ASIC has a bias generator circuit BIAS_GEN(4220), which is supplied from the VBAT power supply and generates a bias current that is stable when supplying voltage to the system. The output current has the following specifications: (i) power supply sensitivity: the power supply voltage of 1.6V to 4.5V < ±2.5%; and (ii) current accuracy: < ±3% after trimming.

[0441] The BIAS_GEN circuit generates switched and unswitched output currents to supply circuits that require a bias current to operate. The operating current consumption of the BIAS_GEN circuit is less than 0.3 uA at 25 °C, where VBAT is 2.5V to 4.5V (excluding any bias output current). Finally, the temperature coefficient of the bias current typically ranges between 4,000 ppm / °C and 6,000 ppm / °C.

[0442] Voltage reference

[0443] As described herein, the ASIC is configured to have a low-power voltage reference, which is powered by the VBAT power supply. The voltage reference has an enable input, which can accept a signal from logic powered by VBAT or VDDBU. The ASIC is designed such that when VBAT is powered, the enable signal does not cause the current of any power supply from this signal interface to increase by more than 10 nA.

[0444] The reference voltage has the following specifications: (i) Output voltage: 1.220 ± 3 mV after trimming; (ii) Power supply sensitivity: < ±6 mV for 1.6 V to 4.5 V input; (iii) Temperature sensitivity: < ±5 mV for 0 °C to 60 °C; and (iv) Output voltage default accuracy (without trimming): 1.220 V ± 50 mV. Additionally, at 4.5 V and 40 °C, the power supply current should be less than 800 nA. In this embodiment, when the reference is disabled, the reference output will be forced to VSSA to prevent the VDD voltage regulator from overshooting to a level exceeding the logic breakdown voltage.

[0445] 32 kHz oscillator

[0446] The ASIC includes a low-power 32.768 kHz crystal oscillator 4222, which is powered by a power supply derived from the VDDA power supply, and the capacitance of the crystal oscillator pads (XTALI, XTALO) can be trimmed by software. Specifically, the frequency trimming range is at least -50 ppm to +100 ppm, with a maximum step size of 2 ppm throughout the trimming range. Here, it can be assumed that the load capacitance of the crystal at each crystal terminal is 7 pF, Ls = 6.9512 kH, Cs = 3.3952 fF, Rs = 70 k, the parallel capacitance = 1 pF, and the PC board parasitic capacitance is 2 pF.

[0447] The ASIC has a VPAD level output available at the pad CLK_32kHZ, where the output can be disabled under software and logic control. The default value disables the 32 kHz oscillator. The input pin OSC32K_BYPASS (4224) can disable the 32 kHz oscillator (without power consumption) and allow digital input to the XTALI pad. Configure the circuit associated with this function so that when OSC32K_BYPASS is low, the ASIC current does not increase by more than 10 nA in either state of the OSC32K_BYPASS signal outside the oscillator current.

[0448] When the VDDA plane is powered, except for the bypass condition, the 32 kHz oscillator is required to be always operable. If OSC32K_BYPASS is true, then the 32 kHz oscillator analog circuit system is put into a low-power state, and the XTALI pad is configured to accept digital input with a level from 0 to VDDA. It should be noted that the duty cycle of the 32 kHz oscillator output is between 40% and 60%.

[0449] Timer

[0450] The ASIC includes a timer 4226, which is clocked from a 32 kHz oscillator divided by 2. The timer is presettable and has two programmable timeouts. The timer has 24 programmable bits for a total time of 17 minutes and 4 seconds. The timer also has a programmable delay for disabling the clocking of the CLK_32KHz pad and setting the microprocessor (uP) interface signals on the VPAD plane to a predefined state (see the section on microprocessor wake-up control signals below). This will enable the microprocessor to enter the suspend mode without an external clock. However, this function may be disabled by software with programmable bits.

[0451] The timer also includes a programmable delay for waking up the microprocessor by enabling the CLK_32KHZ clock output and setting UP_WAKEUP high. The transition of POR2 (VDD POR) from the power low state to the power good state will enable the 32 kHz oscillator, the CLK_32KHZ clock output, and set UP_WAKEUP high. Power-off and power-on are configured to be controlled by programmable control bits.

[0452] Real Time Clock (RTC)

[0453] The ASIC also has a 48-bit readable / writable binary counter, which is operated by a non-gated, free-running 32 kHz oscillator. Writing to the real time clock 4228 requires writing the address with a key before the clock can be written. The write access to the clock is configured to terminate between 1 millisecond and 20 milliseconds after writing to the key address.

[0454] The real time clock 4228 is configured to be reset to half count (MSB = 1, all other bits = 0) by power-on reset from POR1_IN (VBAT POR) or POR2_IN (VDD_POR). In an embodiment of the present invention, the real time clock has programmable interrupt capabilities and is designed to be robust against single event upsets (SEUs), which can be achieved by layout techniques or by adding capacitance to appropriate nodes as needed.

[0455] RC oscillator

[0456] The ASIC further includes an RC clock powered by the VBAT power supply or a power supply derived from VBAT. The RC oscillator is always running, but the oscillator can be bypassed by writing to a register bit in the analog test mode (see the chapter on digital testing) and applying a signal from 0 to the VBAT level to GPIO_VBAT. The RC oscillator is not trimable and has the following specifications: (i) a frequency between 750 Hz and 1500 Hz; (ii) a duty cycle between 50% ± 10%; (iii) a current consumption less than 200 nA at 25 °C; (iv) a frequency variation of less than ±2% for a 1 V to 4.5 V VBAT power supply, and better than 1% for a 1.8 V to 4.5 V VBAT power supply; and (v) a frequency variation of less than +2, -2% for a temperature from 15 °C to 40 °C, where VBAT = 3.5 V. A 32 kHz crystal oscillator or an external frequency source can be used to measure the RC frequency (see the oscillator calibration circuit).

[0457] Real-time RC clock (based on RC oscillator)

[0458] The ASIC includes a 48-bit readable / writable binary ripple counter based on the RC oscillator. Writing to the RC real-time clock requires writing the address with a key before the clock can be written. After writing to the key address, write access to the clock terminates between 1 millisecond and 20 milliseconds, where the time of the protection window is configured to be generated with the RC clock.

[0459] If the crystal oscillator is turned off, the real-time RC clock allows relative timestamps and is configured to reset to half count (MSB = 1, all others = 0) on POR1_IN (BAT POR). The real-time RC clock is designed to be robust against single event upsets (SEUs) either through layout techniques or by adding capacitance to appropriate nodes when needed. At the falling edge of POR2_IN, or if the ASIC enters the low battery state, the RT real-time clock value that can be read through the SPI port can be captured into a register. This register and the associated logic are on the VBAT or VDDBU power plane.

[0460] Battery protection circuit

[0461] The ASIC includes a battery protection circuit 4230 that uses a comparator to monitor the battery voltage and is powered by a power supply derived from the VBAT power plane. The battery protection circuit is configured to always be running when powering the VBAT power supply. The battery protection circuit can use the RC oscillator for the clock signal and has an average current consumption of less than 30 nA, including a 3 MΩ total resistance external voltage divider.

[0462] The battery protection circuit uses an external switched divider with a ratio of.421 for a 2.90V battery threshold. The ASIC also has an internal divider with a ratio of.421 ± 0.5%. This divider is connected between BATT_DIV_EN(4232) and VSSA(4234), and the divider output is a pin called BATT_DIV_INT(4236). To conserve pins in the packaged component, BATT_DIV_INT in this embodiment is internally connected to BATT_DIV inside the package. Also in this configuration, BATT_DIV_EN does not need to be taken out of the package, saving two package pins.

[0463] The battery protection circuit is configured to sample the voltage on input pin BATT_DIV(4238) at a rate of approximately 2 times per second, where the sampling time is generated by an RC oscillator. The ASIC is able to adjust the divider of the RC oscillator to adjust the sampling time interval to.500 seconds ± 5 milliseconds, where the RC oscillator operates within its operating tolerance. In a preferred embodiment, the ASIC has a test mode that allows for more frequent sampling intervals during testing.

[0464] The comparator input is configured to accept inputs from 0 to VBAT volts. For inputs from 0 to VBAT volts, the input current to comparator input BATT_DIV is less than 10nA. The comparator sampling circuit outputs a positive pulse to pad BATT_DIV_EN, which can be used by an external circuitry to enable an off-chip resistor divider only during the sampling time to save power. The high voltage logic level is the VBAT voltage, and the low level is the VSS level.

[0465] At VBAT = 3.0V, the output resistance of the BATT_DIV_EN pad should be less than 2 kiloohms. This allows the divider to be driven directly from this output. After a programmable number of consecutive samples indicate a low battery condition, the comparator control circuitry triggers an interrupt to interrupt output pad UP_INT. The default number of samples is 4, but the number of consecutive samples can be programmed from 4 to 120.

[0466] After a programmable number of consecutive samples indicate a low battery after the above UP_INT is generated, the comparator control circuitry is configured to generate a signal that will put the ASIC into a low power mode: the VDD regulator will be deactivated, and a low signal will be asserted to pad VPAD_EN. This will be referred to as the low battery state. Also, the number of consecutive samples can be programmed from 4 to 120 samples, with a default value of 4 samples.

[0467] The comparator has separate programmable thresholds for the falling and rising voltages on BATT_DIV. This is implemented in digital logic to multiplex two values into the circuit depending on the state of the low battery state. Thus, if the low battery state is low, then the falling threshold applies; if the low battery state is high, then the rising threshold applies. Specifically, the comparator has 16 programmable thresholds from 1.22 to 1.645 ± 3%, where the DNL of the programmable threshold is set to less than 0.2 LSB.

[0468] The comparator threshold varies less than + / - 1% from 20°C to 40°C. The default threshold for the falling voltage is 1.44V (for the nominal divider, the VBAT threshold is 3.41V), and the default threshold for the rising voltage is 1.53V (for the nominal divider, the VBAT threshold is 3.63V). After the ASIC enters the low battery state, if the comparator senses 4 consecutive indications of a good battery, then the ASIC will start the microprocessor startup sequence.

[0469] Battery power plane power-on reset

[0470] If the input VBAT swings more than 1.2 volts within a 50 microsecond period, or the VBAT voltage is below 1.6 ±.3 volts, then a power-on reset (POR) output is generated on the pad nPOR1_OUT(4240). This POR extends to a minimum pulse width of 5 milliseconds. The output of the POR circuit is configured to be active low and goes to the pad nPOR1_OUT on the VBAT power plane.

[0471] The IC has an input pad nPOR1_IN(4242) for the battery power plane POR. This input pad has RC filtering such that pulses shorter than 50 nanoseconds do not cause a logic reset. In this embodiment, nPOR1_OUT is externally connected to nPOR1_IN during normal operation, thus separating the analog circuitry from the digital circuitry for testing. nPOR1_IN causes a reset of all logic on any power plane and initializes all registers to their default values. Thus, the reset status register POR bit is set and all other bits of the reset status register are cleared. The POR reset circuitry is configured to consume no more than 0.1uA from the VBAT power within 5 seconds after power-on.

[0472] VDD power-on reset (POR)

[0473] The ASIC also has a voltage comparator circuit that generates a VDD voltage plane reset signal after power-up or when VDD drops below a programmable threshold. The range can be programmed with several voltage thresholds. The default value is 1.8V - 15% (1.53V). POR2 has a programmable threshold for the rising voltage, and the threshold implements hysteresis. The rising threshold is also programmable, with a default value of 1.60V ± 3%.

[0474] The POR signal is active low and has an output pad nPOR2_OUT(4244) on the VDD power plane. The ASIC also has an active low POR open-drain output nPOR2_OUT_OD(4246) on the VBAT power plane. This can be used to apply POR to other system components.

[0475] The VDD-powered logic has a POR that originates from the input pad nPOR2_IN(4248). The nPOR2_IN pad is on the VDD power plane and has RC filtering such that pulses shorter than 50 nanoseconds do not cause a logic reset. In normal use, nPOR2_OUT is configured to be externally connected to the nPOR2_IN input pad, thus separating the analog circuit system from the digital circuit system.

[0476] After VDD exceeds the programmable threshold, the generated reset extends for an active time of at least 700 milliseconds to ensure the crystal oscillator is stable. The POR reset circuit system consumes no more than 0.1uA from the VDD power supply for more than 5 seconds after power-up and no more than 0.1uA from the VBAT power supply for more than 5 seconds after power-up. The register storing the POR threshold is powered by the VDD power plane.

[0477] Sensor interface electronics

[0478] In the embodiments of the present invention described herein, the sensor circuit system supports up to five sensor WORK electrodes (4310) in any combination of peroxide or oxygen sensors, but in other embodiments, a greater number of such electrodes can also be accommodated. When the peroxide sensor WORK electrode provides current, the oxygen sensor WORK electrode absorbs current. For this embodiment, the sensor can be configured in a potentiostat configuration as Figure 43 shown.

[0479] The sensor electronics has programmable power control for each electrode interface circuit to minimize current consumption by turning off the current to unused sensor electronics. The sensor electronics also includes electronics for driving the COUNTER electrode 4320, which uses feedback from the RE (reference) electrode 4330. When this circuitry is not in use, its current can be programmed off to save power. The interface electronics includes a multiplexer 4250 such that the COUNTER and RE electrodes can be connected to any (redundant) WORK electrode.

[0480] The ASIC is configured to provide the following sensor interfaces: (i) RE: a reference electrode that can establish a reference potential for the electronics to set the WORK voltage; (ii) WORK1 to WORK5: working sensor electrodes where the required reduction / oxidation (redox) reactions occur; and (iii) COUNTER: the output from this pad maintains a known voltage on the RE electrode with respect to system VSS. In this embodiment, the ASIC is configured to be able to individually set the WORK voltage of up to 5 WORK electrodes, with a resolution and accuracy better than or equal to 5 mV.

[0481] In the oxygen mode, the WORK voltage can be programmed between at least 0 and 1.22 V with respect to VSSA. In the peroxide mode, the WORK voltage can be programmed between at least 0.6 volts and 2.054 volts with respect to VSSA. If VDDA is less than 2.15 V, then the WORK voltage can operate up to VDDA - 0.1 V. The ASIC includes a current measurement circuit to measure the WORK electrode current in the peroxide sensor mode. For example, this can be implemented by a current-voltage or current-frequency converter, which can have the following specifications: (i) current range: 0 to 300 nA; (ii) voltage output range: the same as the WORK electrode in the peroxide / oxygen mode; (iii) output offset voltage: maximum ±5 mV; and (iv) uncalibrated resolution: ±.25 nA.

[0482] After applying the calibration factor to the gain and assuming an acquisition time of 10 seconds or less, the current measurement accuracy is:

[0483] 5 pA - 1 nA: ±3% ±20 pA

[0484] 1 nA - 10 nA: ±3% ±20 pA

[0485] 10 nA - 300 nA: ±3% ±.2 nA

[0486] For the current-to-frequency converter (ItoF) only, the frequency range can be between 0 Hz and 50 kHz. In peroxide mode, the current converter must operate within the specified voltage range with respect to VSS of the WORK electrode. Here, the current consumption of the 2.5 V power supply is less than 2 μA, where the WORK electrode current of each converter is less than 10 nA, including the digital-to-analog (DAC) current.

[0487] The current converter can be enabled or disabled through software control. When disabled, the WORK electrode will exhibit an extremely high impedance value, i.e., greater than 100 megohms. Additionally, for the ItoF only, the output of the I-F converter will go into a 32-bit counter, which can be read, written to, and cleared by the microprocessor and test logic. During counter reading, the clocking of the counter is paused to ensure accurate reading.

[0488] In the embodiments of the present invention described herein, the ASIC also includes a current measurement circuit to measure the WORK electrode current in oxygen sensor mode. The circuit can be implemented as a current-to-voltage or current-to-frequency converter, and programmable bits can be used to configure the current converter to operate in oxygen mode. As previously mentioned, in oxygen mode, the current converter must operate within the specified voltage range of the WORK electrode with respect to VSS. Here, again, the current range is 3.7 pA to 300 nA, the voltage output range is the same as that of the WORK electrode in oxygen mode, the output offset voltage is at most ±5 mV, and the uncalibrated resolution is 3.7 pA ± 2 pA.

[0489] After applying the calibration factor to the gain and assuming an acquisition time of 10 seconds or less, the current measurement accuracy is:

[0490] 5 pA - 1 nA: ±3% ± 20 pA

[0491] 1 nA - 10 nA: ±3% ± 20 pA

[0492] 10 nA - 300 nA: ±3% ± 0.2 nA

[0493] For the current-to-frequency converter (ItoF) only, the frequency range can be between 0 Hz and 50 kHz, and the current consumption of the 2.5 V power supply is less than 2 μA, where the WORK electrode current of each converter is less than 10 nA, including the DAC current. The current converter can be enabled or disabled through software control. When disabled, the WORK electrode will exhibit an extremely high impedance value, i.e., greater than 100 megohms. Also, for the ItoF only, the output of the I-F converter will go into a 32-bit counter, which can be read, written to, and cleared by the microprocessor and test logic. During counter reading, the clocking of the counter is paused to ensure accurate reading.

[0494] In an embodiment of the invention described herein, the reference electrode (RE) 4330 has an input bias current of less than.05 nA at 40°C. The COUNTER electrode adjusts its output to maintain a desired voltage on the RE electrode. This is achieved by amplifier 4340, the output of which to the COUNTER electrode 4320 attempts to minimize the difference between the actual RE electrode voltage and the target RE voltage, the target RE voltage being set by the DAC.

[0495] The RE set voltage can be programmed between at least 0 and 1.80 V, and the common-mode input range of the COUNTER amplifier encompasses at least.20 to (VDD-.20) V. When necessary, register bits can be used to select the common-mode input range and to program the operating mode of the COUNTER. The WORK voltage is set to a resolution and accuracy that are better than or equal to 5 mV. It should be noted that in the normal mode, the COUNTER voltage seeks to maintain the RE voltage at the level of the programmed RE target value. However, in the force counter mode, the COUNTER voltage is forced to the programmed RE target voltage.

[0496] All electrode drive circuits are configured to be able to drive the electrodes to the electrode load and not oscillate under any usage conditions. Figure 44 An equivalent ac inter-electrode circuit with a potentiostat configuration as shown in Figure 43 is shown. Figure 44 The equivalent circuit shown can be located between any electrodes, namely WORK1 to WORK5, COUNTER, and RE, where the value ranges of the corresponding circuit components are as follows:

[0497] Ru = [200 - 5k] ohms

[0498] Cc = [10 - 2000] pF

[0499] Rpo = [1 - 20] kilo-ohms

[0500] Rf = [200 - 2000] kilo-ohms

[0501] Cf = [2 - 30] uF

[0502] During initialization, the drive currents for the WORK and COUNTER electrodes need to supply a higher current than in the aforementioned normal potentiostat operation. Therefore, if additional drive is required, programmable register bits can be used to program the electrode drive circuits to a higher power state. It is important to achieve low-power operation in the normal potentiostat mode, in which the electrode current is typically less than 300 nA.

[0503] In a preferred embodiment, during initialization, the WORK1 to WORK5 electrodes can be programmed in steps equal to or less than 5 mV between 0 and VDD volts, and the drive or sink current output capability of the electrodes is a minimum of 20 μA from 0.20 V to (VDD - 0.20 V). Also during initialization, the ASIC is typically configured to be able to measure the current of up to 20 μA of one WORK electrode with an accuracy of measured value ±2% ± 40 nA. In addition, during initialization, the RE set voltage is programmable as described above, the COUNTER DRIVE CIRCUIT output must be able to source or sink 50 μA minimum using the COUNTER electrodes from 0.20 V to (VDD - 0.20 V), and the supply current (VDDA and VDDA) to the initialization circuitry is required to exceed any output current provided by less than 50 μA.

[0504] Current calibrator

[0505] In an embodiment of the present invention, the ASIC has a current reference that can be directed to any WORK electrode for calibration. In this regard, the calibrator includes programmable bits to sink or source current output. Assuming a 0 tolerance external precision resistor, the programmable currents include at least 10 nA, 100 nA, and 300 nA with an accuracy better than ±1% ± 1 nA. For the reference resistor, the calibrator uses a 1 megohm precision resistor connected to pad TP_RES(4260). Additionally, for initialization and / or sensor state purposes, the current reference can be directed to the COUNTER or RE electrodes. A constant current can be applied to the COUNTER or RE electrodes, and the electrode voltage can be measured using the ADC.

[0506] High-speed RC oscillator

[0507] Returning to reference FIG. 42, the ASIC further includes a high-speed RC oscillator 4262 that supplies the analog-to-digital converter (ADC) 4264, the ADC sequencer 4266, and other digital functions that require a clock speed higher than 32 kHz. The high-speed RC oscillator is phase-locked to a 32 kHz clock (32.768 kHz) to provide an output frequency that can be programmed between 524.3 kHz and 1048 kHz. Additionally, the duty cycle of the high-speed RC oscillator is 50% ± 10%, the phase jitter is less than 0.5% rms, the current is less than 10 μA, and the frequency is stable within the VDD operating range (voltage range of 1.6 to 2.5 V). The default value of the high-speed RC oscillator is "off" (i.e., disabled), in which case the current consumption is less than 10 nA. However, the ASIC has programmable bits to enable the high-speed RC oscillator.

[0508] Analog-to-digital converter

[0509] The ASIC includes a 12-bit ADC (4264) with the following characteristics: (i) capable of performing a conversion in less than 1.5 milliseconds when operating according to a 32 kHz clock; (ii) capable of performing a faster conversion when clocked from a high-speed RC oscillator; (iii) an accuracy of at least 10 bits (12 bits ± 4 counts); (iv) a reference voltage input of 1.220 V, and between 20 °C and 40 °C, the temperature sensitivity is less than 0.2 mV / °C; (v) a full-scale input range of 0 to 1.22 V, 0 to 1.774 V, 0 to 2.44 V, and 0 to VDDA, where the 1.774 and 2.44 V ranges have programmable bits to reduce the conversion range to a lower value to accommodate a lower VDDA voltage; (vi) the current consumed from the power supply is less than 50 uA; (vi) has a converter capable of operating according to a 32 kHz clock or a high-speed RC clock; (vii) DNL is less than 1 LSB; and (viii) issues an interrupt at the end of the conversion.

[0510] As Figure 42A and 42B shown, the ASIC has an analog multiplexer 4268 at the input of the ADC 4264, and both the ADC and the multiplexer can be controlled by software. In a preferred embodiment, at least the following signals are connected to the multiplexer:

[0511] (i) VDD - core voltage and regulator output

[0512] (ii) VBAT - battery power supply

[0513] (iii) VDDA - analog power supply

[0514] (iv) RE - reference electrode of the sensor

[0515] (v) COUNTER - counter electrode of the sensor

[0516] (vi) WORK1 to WORK5 - working electrodes of the sensor

[0517] (vii) temperature sensor

[0518] (viii) at least two external pin analog signal inputs

[0519] (ix) EIS integrator output

[0520] (x) ItoV current converter output.

[0521] The ASIC is configured such that for the input COUNTER, RE, WORK1 to WORK5, the temperature sensor, and any other input that may be adversely affected by the load, the load on the ADC does not exceed ±0.01 nA. The multiplexer includes a voltage divider for any input with a voltage higher than the ADC input voltage range, and a buffer amplifier that reduces the input resistance of the voltage-divided input to less than 1 nA for load-sensitive inputs. The buffer amplifier in turn has a common-mode input range of at least 0.8V to VDDA voltage, and an offset of less than 3 mV between an input range of 0.8V to VDDA - 0.1V.

[0522] In a preferred embodiment, the ASIC has a mode for performing ADC measurements in a programmed sequence. Thus, the ASIC includes a programmable sequencer 4266 that supervises the measurement of up to 8 input sources for ADC measurements with the following programmable parameters:

[0523] (i) ADC MUX input

[0524] (ii) ADC range

[0525] (iii) Delay time before measurement, where the delay can be programmed from 0 to 62 milliseconds in steps of 0.488 milliseconds

[0526] (iv) Number of measurements per input from 0 to 255

[0527] (v) Number of measurement cycles: 0 to 255, where a measurement cycle refers to multiple repetitions of a sequence of up to 8 input measurements (e.g., as an outer loop in a program)

[0528] (vi) Delay between measurement cycles, where the delay can be programmed from 0 to 62 milliseconds in steps of 0.488 milliseconds.

[0529] The sequencer 4266 is configured to start upon receiving an automatic measurement start command, and the measured values can be stored in the ASIC for retrieval via the SPI interface. It should be noted that the sequencer time base can be programmed between a 32 kHz clock and a high-speed RC oscillator 4262.

[0530] Sensor diagnostics

[0531] As described in detail previously, embodiments of the invention described herein are directed to using impedance and impedance-related parameters, for example, in sensor diagnostic procedures and Isig / SG fusion algorithms. To this end, in a preferred embodiment, when in potentiostat configuration, the ASIC described herein is capable of measuring the magnitude and phase angle of the impedance of any WORK sensor electrode relative to the RE and COUNTER electrodes. This is done, for example, by measuring the amplitude and phase of the current waveform in response to a sine wave form superimposed on the WORK electrode voltage. See, for example Figure 42B diagnostic circuitry 4255 in

[0532] The ASIC is capable of measuring the resistance and capacitance components from any electrode to any electrode, for example, via electrode multiplexer 4250. It should be noted that such measurements may disturb the sensor balance and may require a stabilization time or sensor initialization to record a stable electrode current. As previously mentioned, although the ASIC can be used for impedance measurements over a wide frequency spectrum, for the purposes of the embodiments of the invention, a relatively narrow frequency range can be used. Specifically, the sine wave measurement capabilities of the ASIC can include test frequencies from about 0.10 Hz to about 8192 Hz. When making such measurements, the minimum frequency resolution according to embodiments of the invention may be limited as shown in Table 2 below:

[0533] Table 2

[0534]

[0535] The sine wave amplitude can be programmed in steps of at least 5 mV from at least 10 mVp-p to 50 mVp-p and in steps of at least 10 mV from 60 mVp-p to 100 mVp-p. In a preferred embodiment, the amplitude accuracy is better than ±5% or ±5 mV, whichever is greater. Additionally, the ASIC can measure the electrode impedance with the accuracy specified in Table 3 below:

[0536]

[0537] In embodiments of the invention, the ASIC can measure the phase of the input waveform relative to a time base, and the input waveform phase can be used for impedance calculations to improve accuracy. The ASIC can also have on-chip resistors to calibrate the above-described electrode impedance circuitry. The on-chip resistors, in turn, can be calibrated by comparison with a known 1-megohm off-chip precision resistor.

[0538] The data sampling of the waveform can also be used to determine impedance. The data can be transmitted to an external microprocessor using a Serial Peripheral Interface (SPI) for calculation and processing. The converted current data is sufficiently buffered to be able to transfer 2000 ADC data conversions to an external device through the SPI interface without losing data. This assumes a maximum latency time of 8 milliseconds for servicing data transfer request interrupts.

[0539] In an embodiment of the present invention, instead of or in addition to measuring the electrode impedance using a sine wave, the ASIC can also use a stepped input to measure the electrode current. Here, the ASIC can provide a programmable amplitude step of 10 to 200 mV for the electrode with a resolution better than 5 mV and sample (measure) the resulting current waveform. The sampling duration can be programmed in.25 second steps to at least 2 seconds, and the sampling interval for measuring the current can include at least five programmable binary weighted steps of approximately.5 milliseconds to 8 milliseconds.

[0540] The resolution of the electrode voltage samples is less than 1 mV and the range is up to ±.25 volts. This measurement can be made with respect to a suitable stable voltage to reduce the required data conversion dynamic range. Similarly, the resolution of the electrode current samples is less than.04 uA and the range is up to 20 uA. If the measurement polarity is programmable, then the current measurement can be unipolar.

[0541] In an embodiment of the present invention, the current measurement can use an I-V converter. In addition, the ASIC can have on-chip resistors to calibrate the current measurement. The on-chip resistors can in turn be calibrated by comparison with a known 1 megohm off-chip precision resistor. The current measurement sample accuracy is better than ±3% or ±10 nA, whichever is greater. As previously mentioned, the converted current data is sufficiently buffered to be able to transfer 2000 ADC data conversions to an external device through the SPI interface without losing data. This assumes a maximum latency time of 8 milliseconds for servicing data transfer request interrupts.

[0542] Calibration voltage

[0543] The ASIC includes a precision voltage reference for calibrating the ADC. The output voltage is 1.000V ±3%, the production variation is less than ±1.5%, and within the temperature range of 20°C to 40°C, the stability is better than ±3 mV. This precision calibration voltage can be calibrated by the on-chip ADC by comparing it with an external precision voltage during manufacturing. In manufacturing, the calibration factor can be in the system non-volatile memory (not on this ASIC) to achieve higher accuracy.

[0544] The current consumption of the calibration voltage circuit is preferably less than 25 μA. Additionally, the calibration voltage circuit can power down to less than 10 nA when not in use to save battery power.

[0545] Temperature sensor

[0546] The ASIC has a temperature transducer with a sensitivity between 9 and 11 mV per degree Celsius between -10 °C and 60 °C. The output voltage of the temperature sensor enables the ADC to measure the temperature-related voltage using a 0 to 1.22 V ADC input range. The current consumption of the temperature sensor is preferably less than 25 μA, and the temperature sensor can power down to less than 10 nA when not in use to save battery power.

[0547] VDD voltage regulator

[0548] The ASIC has a VDD voltage regulator with the following characteristics:

[0549] (i) Minimum input voltage range: 2.0 V to 4.5 V.

[0550] (ii) Minimum output voltage: 1.6 to 2.5 V ± 5%, default value is 2.0 V.

[0551] (iii) Dropout voltage: At Iload = 100 μA, Vin = 2.0 V, Vin - Vout <.15 V.

[0552] (iv) The output voltage is programmable with an accuracy within 2% of the values shown in Table 4 below:

[0553] Table 4

[0554]

[0555] (v) The regulator can supply a 1 mA output at 2.5 V with an input voltage of 2.8 V.

[0556] (vi) The regulator also has input and output pads that can be open if an external regulator is used. In this non-operational mode, the current consumption of the regulator circuit is preferably less than 100 nA.

[0557] (vii) The change in output voltage from a 10 μA load to a 1 mA load is preferably less than 25 mV.

[0558] (viii) The current consumption of the source excluding the output current at a 1 mA load is less than 100 μA.

[0559] (ix) The current consumption of the source excluding the output current at a 0.1 mA load is less than 10 μA.

[0560] (x) The current consumption of the source without the output current under a 10 μA load is less than 1 μA.

[0561] General comparator

[0562] The ASIC includes at least two comparators 4270, 4271 powered by VDDA. The comparators use 1.22 V as a reference to generate a threshold. The outputs of the comparators can be read by the processor and will generate a maskable interrupt on the rising or falling edge determined by the configuration register.

[0563] The comparators have power control to reduce power when not in use, and the current supply for each comparator is less than 50 nA. For a 20 mV overdrive signal, the response time of the comparator is preferably less than 50 microseconds, and the offset voltage is less than ±8 mV.

[0564] The comparators also have programmable hysteresis, where the hysteresis options include: threshold on the rising input = 1.22 V + Vhyst, threshold on the falling input = 1.22 - Vhyst, or no hysteresis (Vhyst = 25 ± 10 mV). The output from either comparator can be used for any GPIO on any power plane. (See the "GPIO" section).

[0565] Sensor connection sensing circuitry on RE

[0566] The analog switched capacitor circuit monitors the impedance of the RE connection to determine if a sensor is connected. Specifically, a capacitor of approximately 20 pF is switched at a frequency of 16 Hz by an inverter with an output swing from VSS to VDD. The comparator will sense the voltage swing on the RE pad, and if the swing is less than the threshold, the comparator output will indicate a connection. The above comparison is made on both transitions of the pulse. A swing below the threshold on both transitions is required to indicate a connection, and a comparison indicating a high swing on either phase will indicate a disconnection. The connection signal / disconnection signal is debounced so that a transition of its state requires a stable indication of the new state for at least 1 / 2 second.

[0567] The circuit has six thresholds defined by the following resistors in parallel with the 20 pF capacitor: 500 kΩ, 1 MΩ, 2 MΩ, 4 MΩ, 8 MΩ, and 16 MΩ. This parallel equivalent circuit is between the RE pad and a virtual ground, which can be at any voltage between the power rails. The threshold accuracy is better than ±30%.

[0568] If a sensor is connected or disconnected, the output of the sensor connection sensing circuitry can generate an interrupt or processor startup in a programmable manner. This circuitry is active whenever nPOR2_IN is high and VDD and VDDA are present. The current consumption of this circuitry is on average less than 100 nA.

[0569] WAKEUP pad

[0570] The WAKEUP circuitry is powered by the VDD supply, where the input range is from 0 V to VBAT. The WAKEUP pad 4272 has a weak pull - down of 80 ± 40 nA. This current can be sourced from the output of the BIAS_GEN 4220. In the case of a 0 v input, the average current consumed by the circuitry is less than 50 nA.

[0571] The rising input voltage threshold Vih of the WAKEUP input is 1.22 ± 0.1 V, and the falling input threshold is - 25 mV ± 12 mV of the rising threshold. In a preferred embodiment, the circuitry associated with the WAKEUP input draws no more than 100 nA of current (this current does not include the input pull - down current) for any input with a value between -.2 and the VBAT voltage. The WAKEUP pad is debounced for at least 1 / 2 second.

[0572] If the WAKEUP pad changes state, the output of the WAKEUP circuitry can generate an interrupt or processor startup in a programmable manner. (See the event handler section). Importantly, it should be noted that if the battery protection circuitry indicates a low - battery state, the WAKEUP pad circuitry is configured to assume a low current of <1 nA.

[0573] UART WAKEUP

[0574] The ASIC is configured to monitor the nRX_EXT pad 4274. If the nRX_EXT level remains high (UART BREAK) for more than 1 / 2 second, then a UART WAKEUP event is generated. Due to sampling, the UART WAKEUP event may be generated with a continuous high level as short as 1 / 4 second. The UART WAKEUP event can generate an interrupt, a WAKEUP, and / or a microprocessor reset (nRESET_OD) in a programmable manner. (See the event handler section).

[0575] In a preferred embodiment, the circuit associated with the UART WAKEUP input draws no more than 100 nA, and if the battery protection circuitry indicates a low battery state, then the UART WAKEUP pad circuitry is configured to assume a low current of <1 nA. The UART wakeup input has a rising input voltage threshold Vih of 1.22 ± 0.1 V. The falling input threshold is -25 mV ± 12 mV of the rising threshold.

[0576] Microprocessor wake-up control signal

[0577] The ASIC is capable of generating signals to assist in controlling the power management of the microprocessor. Specifically, the ASIC can generate the following signals:

[0578] (i) nSHUTDN - nSHUTDN can control the power enable of the off-chip VDD regulator. The nSHUTDN pad is on the VBAT power rail. If the battery protection circuitry indicates a low battery state, then nSHUTDN is low, otherwise nSHUTDN is high.

[0579] (ii) VPAD_EN - VPAD_EN can control the power enable of the external regulator supplying the VPAD power. The internal signal corresponding to this external signal ensures that when the VPAD power is disabled, the input from the VPAD pad does not generate additional current due to a floating input. The VPAD_EN pad is an output on the VBAT power rail. If the battery protection signal indicates a low battery, then the VPAD_EN signal is low. The VPAD_EN signal can be set low by a software command that starts a timer; the terminal count of the timer forces VPAD_EN to be low. If the battery protection signal indicates a good battery, then the following events may cause the VPAD_EN signal to go high (see the event handler for more details): nPOR2_IN transitions from low to high; SW / timer (programmable); WAKEUP transition; low to high and / or high to low (programmable); sensor connection transition; low to high and / or high to low (programmable); UART interrupt; and RTC time event (programmable).

[0580] (iii) UP_WAKEUP - UP_WAKEUP can be connected to the microprocessor wake-up pad. It is intended to wake up the microprocessor from the sleep mode or a similar power-down mode. The UP_WAKEUP pad is an output on the VPAD power rail. The UP_WAKEUP signal can be programmed to be low-active, high-active, or pulsed. The UP_WAKEUP signal can be set low by a software command that starts a timer; the terminal count of the timer forces UP_WAKEUP to be low. If the battery protection signal indicates a good battery charge, then the following events may cause the UP_WAKEUP signal to go high (see the event handler for more details): nPOR2_IN transitions from low to high; SW / Timer (programmable); WAKEUP transition; low to high and / or high to low (programmable); sensor connection transition; low to high and / or high to low (programmable); UART interrupt; and RTC time event (programmable). The WAKEUP signal may be delayed by a programmable amount. If WAKEUP is programmed as a pulse, then the pulse width can be programmed.

[0581] (iv) CLK_32KHZ - The CLK_32KHZ pad can be connected to the microprocessor to supply a low-speed clock. The clock is on-off programmable and is turned on in a programmable manner for wake-up events. The CLK_32KHZ pad is an output on the VPAD power rail. If the battery protection signal indicates a low battery charge, then the CLK_32KHZ signal is low. The CLK_32KHZ output can be programmed to be off by a programmable bit. The default is on. The CLK_32KHZ signal can be deactivated by a software command that starts a timer; the terminal count of the timer forces CLK_32KHZ to be low. If the battery protection signal indicates a good battery charge, then the following events may cause the CLK_32KHZ signal to be enabled (see the event handler for more details): nPOR2_IN transitions from low to high; SW / Timer (programmable); WAKEUP transition; low to high and / or high to low (programmable); sensor connection transition; low to high and / or high to low (programmable); UART interrupt; RTC time event (programmable); and detection of low battery charge by the battery protection circuit.

[0582] (v) nRESET_OD - nRESET_OD can be connected to the microprocessor to cause a microprocessor reset. nRESET_OD is programmable for wake-up events. The nRESET_OD pad is an output on the VPAD power rail. This pad is open-drain (nfet output). If the battery protection signal indicates low battery, then the nRESET_OD signal is low. The nRESET_OD active time can be programmed from 1 to 200 milliseconds. The default value is 200 ms. The following events may assert the nRESET_OD signal low (see the event handler for more details): nPOR2_IN; SW / Timer (programmable); WAKEUP transition; from low to high and / or from high to low (programmable); sensor connection transition; low to high and / or high to low (programmable); UART interrupt; and RTC time event (programmable).

[0583] (vi) UP_INT - UP_INT can be connected to the microprocessor to convey an interrupt. UP_INT is programmable for wake-up events. The UP_INT pad is an output on the VPAD power rail. If the battery protection signal indicates low battery, then the UP_INT signal is low. The UP_INT signal can be set high by a software command that starts a timer; the terminal count of the timer forces UP_INT high. If the battery protection signal indicates good battery, then the following events may assert the UP_INT signal high (see the event handler for more details): SW / Timer (programmable); WAKEUP transition; low to high and / or high to low (programmable); sensor connection transition; low to high and / or high to low (programmable); UART interrupt; RTC time event (programmable); detection of low battery by the battery protection circuit; and any ASIC interrupt when unmasked.

[0584] The ASIC has GPIO1 and GPIO0 pads that can be used as start-up mode control for the microprocessor. The POR2 event will reset a 2-bit counter, and the bits of the counter are mapped to GPIO1 and GPIO0 (MSB and LSB respectively). The rising edge of a UART interrupt increments the counter by one, where the counter counts modulo 4 and goes to zero in the case of an increment in state 11. The start-up mode counter can be preset via SPI.

[0585] Event handler / watchdog

[0586] The ASIC incorporates an event handler to define the response to events, which include changes in system state and input signals. Events include all interrupt sources (e.g., UART_BRK, WAKE_UP, sensor connection, etc.). Software can program the response of the event handler to stimuli via the SPI interface. However, some responses may be hard-wired (not programmable).

[0587] The actions of the event handler include enabling / disabling VPAD_EN, enabling / disabling CLK_32KHZ, asserting nRESET_OD, asserting UP_WAKEUP, and asserting UP_INT. Event watchdog timers 1 to 5 can be individually programmed from 250 milliseconds to 16,384 seconds in 250-millisecond increments. The timeouts of event watchdog timers 6 to 8 are hard-coded. The timeouts of Timer6 and Timer7 are 1 minute; the timeout of Timer8 is 5 minutes.

[0588] The ASIC also has a watchdog function to monitor the response of the microprocessor when an event is triggered. When the microprocessor fails to acknowledge the activity caused by the event, the event watchdog is activated. Once activated, the event watchdog performs a series of programmable actions, namely event watchdog timers 1 to 5, and then a series of hard-wired actions, namely event watchdog timers 6 to 8, to regain the response of the microprocessor. The series of actions includes interrupt, reset, wake-up, asserting the 32KHz clock, powering down and powering up the processor.

[0589] During the series of actions, if the microprocessor regains its ability to acknowledge the recorded activity, then the event watchdog is reset. If the ASIC fails to obtain an acknowledgment from the microprocessor, then the event watchdog powers down the microprocessor under conditions that will allow UART_BRK to restart the microprocessor and will activate an alarm. When activated, the alarm condition generates a square wave in a programmable repeat pattern on pad ALARM, with a frequency of approximately 1 kHz. The programmable pattern has two programmable sequences that have programmable burst on and off times. The alarm has another programmable pattern that can be programmed through the SPI port. The pattern will have two programmable sequences that have programmable burst on and off times.

[0590] Digital-to-analog (D / A)

[0591] In a preferred embodiment, the ASIC has two 8-bit D / A converters 4276, 4278, which have the following characteristics:

[0592] (i) The D / A stabilizes with a load of less than 50 pF in less than 1 millisecond.

[0593] (ii) The D / A has at least 8-bit accuracy.

[0594] (iii) The output range is programmable from 0 to 1.22V or 0 to VDDA.

[0595] (iv) The temperature sensitivity of the D / A voltage reference is less than 1 mV / °C

[0596] (v) The DNL is less than 1 LSB.

[0597] (vi) The current consumed by the D / A from the VDDA power supply is less than 2 μA.

[0598] (vii) Each D / A has an output l to the pad.

[0599] (viii) The D / A output is of high impedance. The load current must be less than 1 nA.

[0600] (ix) The D / A pad can be programmed to output a digital signal from the register. The output swing is from VSSA to VDDA.

[0601] Charger / data downloader interface

[0602] TX_EXT_OD 4280 is an open-drain output whose input is the signal on the TX_UP input pad. This will allow the TX_EXT_OD pad to be turned on in the UART idle condition. The TX_EXT_OD pad has a comparator that monitors its voltage. If the voltage is higher than the comparator threshold voltage within the debounce period (1 / 4 second), then the output nBAT_CHRG_EN (4281) will go low. This comparator and other associated circuitry with this function are on the VBAT and / or VDDBU planes.

[0603] The circuitry associated with this function must allow a low level on the TX_EXT_OD pad, which is due to the assertion of nBAT_CHRG_EN not being disabled during normal communication with an external device. If POR1 is active, then nBAT_CHRG_EN will be high (not asserted). The threshold voltage of the comparator is between 0.50 V and 1.2 V. The comparator will have hysteresis; the falling threshold is approximately 25 mV lower than the rising threshold.

[0604] The nRX_EXT pad inverts the signal on this pad and outputs it to RX_UP. Thus, the nRX_EXT signal will be idle low. nRX_EXT must accept inputs up to the VBAT voltage. The nRX_EXT threshold is 1.22 V ± 3%. The output of this comparator will be obtained via the SPI bus for the microprocessor to read.

[0605] The nRX_EXT pad is also incorporated with a programmable current supply method, and the current will be 80±30 nA with a maximum voltage of VBAT. The ASIC layout has a shielding programmable option to adjust this current from 30 nA to 200 nA in steps of less than 50 nA with a minimum number of shield layer replacements. The programmable bit will be available to block UART interrupt detection and force RX_UP to be high. In normal operation, this bit will be set high before enabling the current source for nRX_EXT and then set low after disabling the current source to ensure no glitch is generated on RX_UP or to ensure a UART interrupt event is generated. Attention should be paid to implementing a wet connector detector. When the current source entering nRX_EXT is active, the RX comparator output indicating a low input voltage will indicate the leakage current. The ASIC includes a pull-down resistor of approximately 100 kΩ on the nRX_EXT pad. This pull-down will be disconnected when the current source is active.

[0606] Sensor connection switch

[0607] The ASIC should have a pad SEN_CONN_SW(4282) that can detect low tolerance to VSS(4284). SEN_CONN_SW provides a current of 5 to 25 μA when SEN_CONN_SW = 0V, and the maximum open-circuit voltage is 0.4V. The ASIC layout has a shielding programmable option to adjust this current from 1 μA to 20 μA in steps of less than 5 μA with a minimum number of shield layer replacements. SEN_CONN_SW has an associated circuit system that can detect the presence of a resistor between SEN_CONN_SW and VSSA(4234), and the threshold of this resistor is between 2 and 15 kΩ. The average current consumption of this circuit is at most 50 nA. Sampling must be used to achieve this low current.

[0608] Oscillator calibration circuit

[0609] The ASIC has counters whose inputs can be routed to an internal or external clock source. One counter generates a programmable gating interval for the other counter. The gating interval includes 1 to 15 seconds from the 32 kHz oscillator. The clocks that can be routed to either counter are the 32 kHz RC oscillator, the high-speed RC oscillator, and the input from any GPIO pad.

[0610] Oscillator bypass

[0611] The ASIC can use an external clock to replace each output of the oscillator. The ASIC has a register that can only be written when a specific TEST_MODE is asserted. This register has bits to enable the external input of the RC oscillator and can be shared with other analog test control signals. However, if TEST_MODE is not active, then this register will not allow any oscillator bypass bits to be active.

[0612] The ASIC also has an input pad for the external clock to bypass the RC oscillator. The pad GPIO_VBAT is on the VBAT power plane. The ASIC further includes a bypass enable pad OSC32K_BYPASS for the 32KHZ oscillator. When high, the 32KHZ oscillator output is supplied by driving the OSC32KHZ_IN pad. It should be noted that typically, the OSC32KHZ_IN pad is connected to a crystal.

[0613] The ASIC has an input for the external clock to bypass the HS_RC_OSC. The bypass is enabled by programmable register bits. The HS_RC_OSC can be programmably supplied through GPIO on the VDD plane or GPIO on the VPAD plane.

[0614] SPI slave port

[0615] The SPI slave port contains an interface that consists of a chip select input (SPI_nCS) 4289, a clock input (SPI_CK) 4286, a serial data input (SPI_MOSI) 4287, and a serial data output (SPI_MISO) 4288. The chip select input (SPI_nCS) is an active-low input that is asserted by an off-chip SPI master to initiate and delimit an SPI transaction. When SPI_nCS is asserted low, the SPI slave port configures itself as an SPI slave device and performs data transactions based on the clock input (SPI_CK). When SPI_nCS is inactive, the SPI slave port resets itself and remains in the reset mode. Since this SPI interface supports block transfers, the master device should keep SPI_nCS low until the transfer is complete.

[0616] The SPI clock input (SPI_CK) will always be asserted by the SPI master. The SPI slave port uses the rising edge of SPI_CK to latch the incoming data on the SPI_MOSI input and uses the falling edge of SPI_CK to drive the outgoing data on the SPI_MISO output. The serial data input (SPI_MOSI) is used to transfer data from the SPI master to the SPI slave. All data bits are asserted after the falling edge of SPI_CK. The serial data output (SPI_MISO) is used to transfer data from the SPI slave to the SPI master. All data bits are asserted after the falling edge of SPI_CK.

[0617] SPI_nCS, SPI_CK, and SPI_MOSI are always driven by the SPI master, unless the SPI master is powered down. If VPAD_EN is low, then these inputs are adjusted such that the current consumption associated with these inputs is less than 10 nA and the SPI circuitry remains reset or inactive. When SPI_nCS is active, SPI_MISO is driven only by the SPI slave port; otherwise, SPI_MISO is tri-state.

[0618] The chip select (SPI_nCS) defines and forms the data transfer packet for the SPI data transaction. The data transfer packet consists of three parts. There is a 4-bit command part, followed by a 12-bit address part, and then any number of 8-bit data bytes. Command bit 3 is used as the direction bit. A '1' indicates a write operation, and a '0' indicates a read operation. The combination of command bits 2, 1, 0 has the following definitions. Unused combinations are undefined.

[0619] (i) 0000: Read data and increment the address.

[0620] (ii) 0001: Read data, address unchanged

[0621] (iii) 0010: Read data and decrement the address

[0622] (iv) 1000: Write data and increment the address

[0623] (v) 1001: Write data, address unchanged

[0624] (vi) 1010: Write data and decrement the address

[0625] (vii) x011: Test port addressing

[0626] The 12-bit address part defines the starting byte address. If SPI_nCS remains active after the first data byte to indicate a multi-byte transfer, the address will increment by one after each byte is transferred. Bit <11> of the address (address<11:0>) indicates the highest address bit. The address wraps around after reaching the limit.

[0627] Data is in byte format, and block transfer can be performed by extending SPI_nCS to allow all bytes to be transferred in one data packet.

[0628] Microprocessor interrupt

[0629] The ASIC has an output at the VPAD logic level UP_INT for sending an interrupt to the host microprocessor. The microprocessor interrupt module consists of an interrupt status register, an interrupt mask register, and a function for logically ORing all interrupt statuses into a microprocessor interrupt. The interrupt is implemented to support edge-sensitive and level-sensitive styles. The polarity of the interrupt is programmable. The default interrupt polarity is TBD.

[0630] In a preferred embodiment, all interrupt sources on the AFE ASIC will be recorded in the interrupt status register. Writing a '1' to the corresponding interrupt status bit clears the corresponding pending interrupt. All interrupt sources on the AFE ASIC can be masked through the interrupt mask register. Writing a '1' to the corresponding interrupt mask bit masks the corresponding pending interrupt. Writing a '0' to the corresponding interrupt mask bit deactivates the masking of the corresponding interrupt. The default state of the interrupt mask register is TBD.

[0631] General-purpose input / output (GPIO) / parallel test port

[0632] In an embodiment, the ASIC can have eight GPIOs operating on the VPAD level signal. The ASIC has one GPIO operating on the VBAT level signal and one GPIO operating on the VDD level signal. All GPIOs have at least the following characteristics:

[0633] (i) Register bits control the selection and direction of each GPIO.

[0634] (ii) The ASIC is capable of configuring the GPIO as an input that can be read through the SPI interface.

[0635] (iii) The ASIC is capable of configuring the GPIO as an input for generating an interrupt.

[0636] (iv) The ASIC is capable of configuring each GPIO as an output controlled by register bits and writable through the SPI interface.

[0637] (v) The ASIC can output the input signal applied to GPIO_VBAT or GPIO_VDD to the GPIO (on the VPAD power plane) in a programmable manner. (Level shifting function).

[0638] (vi) The ASIC can configure each GPIO as an input to the oscillator calibration circuit.

[0639] (vii) The ASIC can configure each general comparator output as at least one GPIO on each power plane. The polarity of the comparator output can be programmed by programmable bits.

[0640] (viii) The GPIO has a microprocessor interrupt generation function.

[0641] (ix) The GPIO can be programmed as an open-drain output.

[0642] (x) The GPIO on the VPAD power plane can be configured to implement the startup control of the microprocessor.

[0643] The parallel test port shares 8-bit GPIO on the VPAD voltage plane. The test port will be used to observe the register contents and various internal signals. In normal mode, the output of this port is controlled by the port configuration register. Writing 8'hFF to the GPIO_O1S_REG and GPIO_O2S_REG registers will direct the test port data on the GPIO output, while writing 8'h00 to the GPIO_ON_REG register will disable the test port data and enable the GPIO data to the GPIO output.

[0644] By addressing the target register via SPI from the port, the registers and pre-grouped internal signals can be observed through this test port. The command bits of the SPI data packet are set to 4'b0011, followed by a 12-bit target register address. The parallel test port continues to display the contents of the addressed register until the next test port addressing command is received.

[0645] Analog test port

[0646] The IC has a multiplexer that feeds the pad TP_ANAMUX (4290), and the multiplexer will provide visibility for internal analog circuit nodes for testing. The IC also has a multiplexer that feeds the pad TP_RES (4260), and the multiplexer will provide visibility for internal analog circuit nodes for testing. In common applications, this pad will also accommodate a resistor with a precision of 1 meg to perform various system calibrations.

[0647] Chip ID

[0648] The ASIC contains a 32-bit masked programmable ID. A microprocessor using the SPI interface will be able to read this ID. This ID will be placed in the analog electronic block so that changing the ID does not require chip re-layout. The design should be such that changing the ID only requires one metal or one contact mask.

[0649] Spare test output

[0650] The ASIC has 16 spare digital output signals that can be multiplexed to 8-bit GPIO according to commands sent through the SPI interface. These signals will be organized into two 8-bit bytes and will be connected to VSS when not in use.

[0651] Digital test

[0652] The ASIC has a test mode controller that uses two input pins, TEST_CTL0(4291) and TEST_CTL1(4292). The test controller generates signals based on the combination of test control signals with the following functionality (TEST_CTL<1:0>):

[0653] (i) 0 is the normal operation mode;

[0654] (ii) 1 is the analog test mode;

[0655] (iii) 2 is the scan mode;

[0656] (iv) 3 is the analog test mode where VDD_EN is controlled by the input to GPIO_VBAT.

[0657] The test controller logic is separated between the VDD and VDDBU power planes. During the scan mode, the test LT_VBAT should be asserted high to regulate the analog output into digital logic. The ASIC has scan chains implemented in as much digital logic as possible for fast digital testing.

[0658] Leakage test pin

[0659] The ASIC has a pin called LT_VBAT that, when high, puts all analog blocks into an inactive mode so that only leakage current is drawn from the power supply. LT_VBAT makes all digital outputs from the analog blocks in a stable high or low state to avoid affecting the interface logic current consumption. The LT_VBAT pad is on the VBAT plane and has a pull-down resistance between 10 kΩ and 40 kΩ.

[0660] Power requirement

[0661] In an embodiment of the present invention, the ASIC includes a low-power mode, in which, minimally, the microprocessor clock is turned off, the 32 kHz real-time clock runs, and the circuitry is active to detect sensor connection, a change in the level of the WAKE_UP pin, or a BREAK on the nRX_EXT input. This mode has a total current consumption of VBAT (VDDBU), VDD, and VDDA to a maximum of 4.0 uA. When the battery protection circuit detects low battery level (see battery protection circuit description), the ASIC enters a mode in which only the VBAT and VDDBU power planes are active. This is called the low battery state. The VBAT current in this mode is less than.3 uA.

[0662] In the case of programming the ASIC into a potentiostat configuration and assuming that the WORK electrode current is measured once per minute, the average current consumption of all power supplies is less than 7 uA. In the potentiostat configuration, any one of the WORK electrodes is active in the H2O2 (peroxide) mode and its voltage is set to 1.535 V, the COUNTER amplifier is turned on and VSET_RE is set to 1.00 V, a 20 MEG load resistor is connected between WORK and COUNTER, and COUNTER and RE are connected together. The current measured after calibration should be 26.75 nA ± 3%. In the case where the WORK electrode current is 25 nA, adding an additional WORK electrode may increase the combined current consumption by less than 2 uA.

[0663] In the case of programming the ASIC into a potentiostat configuration enabling diagnostic functions to measure the impedance of one of the WORK electrodes relative to the COUNTER electrode, the ASIC is configured to meet the following requirements:

[0664] (i) Test frequencies: 0.1, 0.2, 0.3, 0.5 Hz, 1.0, 2.0, 5.0, 10, 100, 1000, and 4000 Hz.

[0665] (ii) The measurement at the above ...

Claims

1. A method for optionally externally calibrating a calibration-free glucose sensor for measuring glucose levels in a user's body, the glucose sensor comprising physical sensor electronics, a microcontroller, and a working electrode, the method comprising: Periodically measuring, by the physical sensor electronics, an electrode current Isig signal of the working electrode; Executing, by the microcontroller, an electrochemical impedance spectroscopy EIS program to generate EIS-related data of the working electrode; Calculating, by the microcontroller, a corresponding SG value for each of the SG prediction models based on the Isig signal, the EIS-related data, and a plurality of calibration-free sensor glucose SG prediction models; Calculating, by the microcontroller, an SG variance estimate value for each corresponding SG value; Fusing, by the microcontroller, the corresponding SG values from a plurality of the SG prediction models to obtain a single fused SG value; Determining, by the microcontroller, whether an external blood glucose BG value is available; In response to determining that the BG value is available, comparing the BG value with the fused SG value to determine whether the difference between the fused SG value and the BG value exceeds a threshold; In response to determining that the difference between the fused SG value and the BG value exceeds the threshold, modulating the fused SG value for a period of time by scaling the corresponding SG value according to a modulation factor, the modulation factor being proportional to the ratio of the BG value to the corresponding SG value; Applying, by the microcontroller, an unscented Kalman filter to the fused SG value; And Calculating, by the microcontroller, a calibrated SG value to be displayed to the user.

2. The method according to claim 1, wherein the physical sensor electronics further measures a counter electrode voltage Vcntr value of the glucose sensor.

3. The method according to claim 2, wherein the microcontroller further preprocesses the Isig signal and the Vcntr value before calculating the corresponding SG value.

4. The method according to claim 3, which further comprises applying a low-pass filter to the Isig signal.

5. The method according to claim 3, wherein the preprocessing includes downsampling the Isig signals that are close in time.

6. The method according to claim 1, wherein the plurality of SG prediction models are machine learning models.

7. The method according to claim 6, wherein the machine learning model comprises at least one of a genetic programming algorithm, a regression decision tree, and a bagging decision tree.

8. The method according to claim 1, wherein the plurality of SG prediction models are analytical models.

9. The method according to claim 1, wherein each SG variance estimate value for each corresponding SG value is calculated empirically based on training data.

10. The method according to claim 1, wherein the Kalman filter contains a set of measurement functions when BG calibration is available based on a comparison between the BG value and each corresponding SG value, and a set of measurement functions when BG calibration is not available based on a comparison between the BG value and each corresponding SG value.

11. The method according to claim 1, wherein the sensor comprises a plurality of working electrodes.

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