Switching converter and method for operating a switching converter

Abstract

A switching converter and a method for operating the switching converter are described herein. According to one example, the switching converter includes a transformer including a primary winding, a secondary winding, and at least one auxiliary winding. The switching converter also includes a first electronic switch coupled to the primary winding and configured to turn on and off a primary current through the primary winding according to a first drive signal. A synchronous rectifier circuit is coupled between the secondary winding and an output node of the switching converter. In addition, the switching converter includes a pre-magnetization circuit including a capacitor circuit and a second electronic switch connected between the auxiliary winding and the capacitor circuit. The capacitor circuit is configured to apply a first voltage to the auxiliary winding during a pre-magnetization period when the second electronic switch is closed and to be (re)charged with a second voltage when the second electronic switch is disconnected; the second voltage is higher than the first voltage.

Description

Technical Field

[0001] Embodiments of the present invention relate to voltage conversion circuits, such as flyback converters. Background Art

[0002] Switched mode voltage converters (switch mode power supplies, SMPS) are widely used for power conversion in automotive, industrial or consumer electronic applications. A flyback converter is a specific type of switched mode voltage converter, which includes a transformer having a primary winding and a secondary winding. An electronic switch is connected in series with the primary winding, and the transformer is magnetized when the electronic switch is closed and demagnetized when the electronic switch is open. Magnetizing the transformer includes storing energy in the transformer, and demagnetizing the transformer includes transferring the stored energy to the secondary winding and a load coupled to the secondary winding.

[0003] The flyback converter can be operated in a discontinuous conduction mode (DCM). In this operating mode, there is a delay time between the time when the transformer has been completely demagnetized and the time when the electronic switch is turned on again. During this delay time, parasitic oscillations of the voltage across the electronic switch may occur. In DCM, the flyback converter can be operated in a quasi-resonant operating mode, where the electronic switch is closed at those times when the voltage across the electronic switch reaches a minimum value. When the flyback converter is operated in a quasi-resonant mode, the switching frequency of the electronic switch changes, and operating the electronic switch in the flyback converter at a fixed frequency may result in increased switching losses. However, there may be scenarios where it is desired to operate the flyback converter at a fixed frequency in DCM. Although the switching frequency is substantially constant (fixed frequency resonant operation), there is a concept that allows (quasi) resonant operation to be maintained. However, in this operating mode, problems may arise when synchronous rectifiers are used at the secondary side of the flyback converter. Summary of the invention

[0004] A switching converter is described herein. According to one example, the switching converter includes a transformer including a primary winding, a secondary winding, and at least one auxiliary winding. The switching converter also includes a first electronic switch coupled to the primary winding and configured to turn on and off a primary current through the primary winding according to a first drive signal. A synchronous rectifier circuit is coupled between the secondary winding and an output node of the switching converter. In addition, the switching converter includes a pre-magnetization circuit including a capacitor circuit and a second electronic switch connected between the auxiliary winding and the capacitor circuit. The capacitor circuit is configured to apply a first voltage to the auxiliary winding during a pre-magnetization period when the second electronic switch is closed and to be (re)charged with a second voltage when the second electronic switch is disconnected; the second voltage is higher than the first voltage.

[0005] In addition, a method for operating a switching converter is described herein, wherein the switching converter includes: a transformer including a primary winding, a secondary winding, and at least one auxiliary winding; and a first electronic switch coupled to the primary winding; a synchronous rectifier circuit coupled between the secondary winding and an output node of the switching converter; and a pre-magnetization circuit including a capacitor circuit and a second electronic switch connected between the auxiliary winding and the capacitor circuit. According to one example, the method includes: using the first electronic switch to turn on and off a primary current through the primary winding according to a first drive signal; closing the second electronic switch during a pre-magnetization period to apply a first voltage provided by the capacitor circuit to the auxiliary winding; and charging the capacitor circuit using a second voltage provided by the auxiliary winding when the second electronic switch is open; wherein the second voltage is higher than the first voltage. BRIEF DESCRIPTION OF THE DRAWINGS

[0006] The embodiments described herein may be better understood with reference to the following description and accompanying drawings. The components in the accompanying drawings are not necessarily drawn to scale, but emphasis is placed on illustrating the principles of the embodiments. In addition, in the accompanying drawings, like reference numerals represent corresponding components. In the accompanying drawings:

[0007] Figure 1 An exemplary implementation of a flyback converter is shown.

[0008] Figure 2 A timing diagram showing quasi-resonant operation of a flyback converter is included.

[0009] Figure 3 A timing diagram showing fixed frequency operation of a flyback converter is included.

[0010] Figure 4 An exemplary implementation of a flyback converter configured to perform fixed frequency (quasi) resonant operation is shown.

[0011] Figure 5 One embodiment of a flyback converter using synchronous rectifiers is shown.

[0012] Figure 6 Included is shown in Figure 5 Timing diagram of the operation of the synchronous rectifier used in a flyback converter.

[0013] Figure 7 Included is a timing diagram illustrating undesirable early turn-on of synchronous rectifiers that may occur during fixed frequency operation of a flyback converter.

[0014] Figure 8 It shows that it can be Figure 5An example of an improved pre-charging circuit for avoiding premature turn-on of the mentioned synchronous rectifier used in the embodiments.

[0015] Fig. 9 It shows that it can be Figure 5 Another example of an improved pre-charging circuit for avoiding premature turn-on of the mentioned synchronous rectifier used in the embodiments of FIG. DETAILED DESCRIPTION

[0016] In the following detailed description, reference is made to the accompanying drawings. The accompanying drawings form a part of the specification and show by way of illustration specific embodiments in which the invention can be practiced. It should be understood that the features of the various embodiments described herein may be combined with each other unless otherwise specifically stated.

[0017] Figure 1 An exemplary implementation of a voltage converter (switch mode power supply, SMPS) is shown. Figure 1 The voltage converter shown has a flyback converter topology and will be referred to as a flyback converter hereinafter. The flyback converter includes: a circuit configured to receive an input voltage V IN The first input node and the second input node are input; and having an output voltage V OUT The first output node and the second output node are output. A load coupled to the output node ( Figure 1 (not shown) can respectively receive the output voltage V obtained at the output OUT and the output current i OUT The flyback converter includes a transformer 20 having a primary winding L P and the primary winding L P The magnetically coupled secondary winding L S . Primary winding L P and the secondary winding L S The electronic switch T1 (eg, MOS transistor) is connected to the primary winding L P Series connection, where the primary winding L P A series circuit of an electronic switch T1 and an electronic switch T2 is connected between the first input node and the second input node to receive an input voltage V IN .

[0018] Optionally, capacitor C IN —hereinafter referred to as the input capacitor—is connected between the input nodes. The input capacitor C IN Can help filter the input voltage V IN The ripple is particularly when the input voltage V is generated by a rectifier circuit (not shown) according to the alternating current voltage (AC voltage). INTherefore, the input voltage is a DC voltage and is referenced to the first ground node GND1, while the output voltage is referenced to the second ground node GND2.

[0019] Figure 1 The flyback converter also includes a secondary winding L connected S A rectifier circuit 30 is provided between the output and the Figure 1 In the embodiment shown, the rectifier circuit 30 includes a rectifier element D such as a diode. R and capacitor C R The series circuit and the secondary winding L S connected in parallel, where the output voltage V OUT The capacitor C R However, this is only a simplified exemplary implementation of the rectifier circuit 30. Other implementations of the rectifier circuit 30 may also be used and examples will be discussed later.

[0020] The control circuit 10 is configured to receive a feedback signal S from a feedback circuit (control loop) based on the feedback signal S FB To drive the electronic switch T1. The feedback circuit may include receiving the output voltage V OUT The filter 40 (including, for example, a PID regulator) and the transmitter circuit OC. Figure 1 In the embodiment shown, the filter 40 is on the secondary side of the transformer 20, and the transmitter OC transmits the output signal of the filter 40 from the secondary side to the primary side through the galvanic isolation, and the output signal of the transmitter OC is the feedback signal S received by the control circuit 10. FB The filter 40 is configured to generate a voltage according to the output voltage and the reference signal ( Figure 1 (not shown) generates an error signal and generates a feedback signal S based on the error signal FB This is generally known so that no further detailed explanation is required in this regard.

[0021] According to one embodiment, the filter 40 has one of a proportional (P) characteristic, a proportional integral (PI) characteristic, and a proportional integral derivative (PID) characteristic. According to another embodiment (not shown), the positions of the filter 40 and the transmitter OC in the feedback loop are interchanged, so that the transmitter transmits a signal representing the output voltage V from the secondary side to the primary side through the galvanic isolation. OUT The filter receives the signal sent by the transmitter and generates a feedback signal S FB In the illustrated embodiment, the transmitter OC comprises an optocoupler. However, this is merely an example. Other transmitters suitable for transmitting signals via galvanic isolation (also referred to as a barrier) may also be used. Examples of such transmitters include transmitters having a transformer such as a coreless transformer.

[0022] The control circuit 10 is configured to operate the electronic switch T1 in a pulse width modulation (PWM) manner. According to one embodiment, the electronic switch T1 is a transistor. Figure 1 In the embodiment shown in , the transistor is a MOSFET (metal oxide semiconductor field effect transistor), in particular an n-type MOSFET. However, this is only an example. Other types of transistors such as IGBT (insulated gate bipolar transistor), JFET (junction field effect transistor), BJT (bipolar junction transistor) or p-type MOSFET may also be used.

[0023] Refer to the following Figure 2 Instructions Figure 1 One approach to a flyback converter is shown in FIG. Figure 2 shows the load path voltage V across the load path of the electronic switch T1 DS , the auxiliary winding L of the transformer 20 AUX The auxiliary voltage V AUX , the drive signal S1 received by the electronic switch T1 from the control circuit 10, the load current i through the electronic switch T1 P (the primary current of the transformer 20) and the magnetization intensity M of the transformer 20 TR Timing diagram of Figure 1 In the case of MOSFET T1 shown, the load path voltage V DS is the drain-source voltage and the load current i P is the drain-source current. The drive signal S1 is received by the gate node of the MOSFET T1. The drive signal S1 may have one of a first signal level that turns on the electronic switch T1 and a second signal level that turns off the electronic switch T1. The first level is referred to as the on-level and the second signal level is referred to as the off-level hereinafter. For illustration purposes only, Figure 2 In the figure, the on-level of the driving signal S1 is plotted as a high signal level, and the off-level is plotted as a low level.

[0024] Reference Figure 2 The operation of the flyback converter includes a plurality of consecutive driving cycles, wherein in each driving cycle, the control circuit 10 turns on the electronic switch T1 for a conduction period T ON1 , and during the conduction period T ON1 After that, the electronic switch T1 is turned off for the off period T OFF .exist Figure 2 In the embodiment shown in , one of these drive cycles starts at time t1 and ends at time t4, which is the time when the next drive cycle starts. ON1 During this period, the input voltage V IN Make the load current iP Flowing through the primary winding L P and electronic switch T1, so that the primary current i P The current level during the conduction period T ON1 This increases the primary current i P By increasing the magnetization intensity M of the transformer 20 TR This magnetization is associated with magnetically storing energy in the transformer 20 (more precisely, in the air gap of the transformer 20), and the stored energy increases with the primary current i P During the conduction period T ON1 During this period, the load path voltage V of the electronic switch T1 DS is essentially zero, and the primary winding L P The voltage across the terminals is essentially equal to the input voltage V IN (Ignore the voltage drop across the electronic switch T1 when the electronic switch T1 is turned on). Figure 1 In the embodiment shown in FIG. 1 , the auxiliary winding L AUX and the primary winding L P have opposite winding directions. In this case, the auxiliary voltage V AUX The voltage level is given by:

[0025] V AUX = -(N AUX / N P )·V P , (1)

[0026] Among them, N AUX Indicates the auxiliary winding L AUX The number of windings, N P Indicates the primary winding L P The number of windings, and V P is the primary winding L P Therefore, during the conduction period T ON1 During this period, the auxiliary voltage V AUX The voltage level is -N AUX / N P ·V IN (When the switch is on, V P ≈V IN ).

[0027] When the electronic switch T1 is turned off, the energy stored in the transformer 20 is transferred to the secondary winding L S , rectifier circuit 30 and finally delivered to the load. This demagnetizes transformer 20. Figure 2 In, T DEMAG represents the time period during which the transformer 20 is demagnetized. DEMAGDuring this period, energy is transferred to the secondary side of the transformer 20. DEMAG - hereinafter also referred to as the demagnetization period - during which the load path voltage V DS Basically equal to the input voltage V IN Plus the reflected voltage V REFLECT . Reflected voltage V REFLECT Basically it is given by:

[0028] V REFLECT =N P / N S ·(V OUT +V R ), (2)

[0029] Among them, N P is the primary winding L P The number of windings, N S is the secondary winding L S The number of windings, and V R is the voltage across the rectifier circuit 30. The voltage V across the rectifier circuit 30 R Depends on the secondary winding L S The current i s The current level of the secondary current. s During the demagnetization period T DEMAG period decreases, making the reflected voltage V REFLECT Decrease and during the demagnetization period T DEMAG At the end, N P / N S ·V OUT .

[0030] In the so-called quasi-resonant (QR) mode, there is a delay time between time t3 when the transformer 20 has been completely demagnetized and time t4 when the next drive cycle starts, i.e., when the electronic switch T1 is turned on again. During this period, the load path voltage V DS Oscillation. This is caused by a parasitic resonant circuit, which includes the primary winding L P and the parasitic capacitance of the electronic switch T1. Such parasitic capacitance can be represented by capacitor C DS (See Figure 1 ) indicates that the capacitor C DS In the QR mode, after the transformer 20 has been demagnetized, the control circuit 10 controls the load path voltage V DS Reach the minimum value V DSmin1 However, the minimum voltage V DSmin1The voltage level of may be relatively high so that relatively high switching losses may occur. These switching losses include, for example, losses associated with discharging the parasitic capacitance of the electronic switch T1. The higher these losses are, the higher the voltage V at the time t1 when the electronic switch T1 is turned on. DS In addition, the switching frequency in the quasi-resonant mode changes and depends in particular on the on-time T ON1 However, there are applications where such variations in the switching frequency are undesirable and it is desirable to operate the flyback converter at a predefined frequency. This predefined frequency may be fixed or may depend on the power consumption of the load.

[0031] Figure 3 A timing diagram is shown that provides a method of switching the electronic switch T1 at a predefined frequency and providing low switching losses. Figure 3 shows the load path voltage V DS , auxiliary voltage V AUX , the drive signal S1 of the electronic switch T1, the load current i of the electronic switch T1 P and the magnetization intensity M of transformer 20 TR The magnetization intensity represents the magnetic flux and the magnetic flux density in the core (not shown) of the transformer 20, respectively.

[0032] exist Figure 3 In the example shown in FIG. 1 , when the electronic switch T1 is turned on for the conduction period T ON1 (See Figure 3 , before the drive signal S1), the transformer 20 is pre-magnetized for a pre-magnetization period T in each drive cycle. ON2 (See Figure 3 , drive signal S2). After pre-magnetizing the transformer 20, the first delay time T is waited before turning on the electronic switch T1 again. DEL1 The pre-magnetization transformer 20 may include another electronic switch that is turned on differently from the electronic switch T1. Examples of such another electronic switch are described in more detail below.

[0033] Below, with the primary winding L P The electronic switch T1 connected in series will be referred to as a first electronic switch, and the other electronic switch for pre-magnetizing the transformer 20 will be referred to as a second electronic switch. The drive signal S2 for driving the second electronic switch is also Figure 3 As shown in (see Figure 3 , the third timing diagram from the top). According to Figure 3 In the example of , the high level represents the on-level of turning on the second electronic switch to pre-magnetize the transformer 20, and the low level represents the off-level of turning off the second switch. Pre-magnetizing the transformer 20 includes magnetizing the transformer 20 so that after turning on the first electronic switch T1, the primary current i Pimmediately becomes negative (the direction of the positive primary current is determined by Figure 1 The primary current i P remains negative until the transformer 20 is demagnetized (i.e., the pre-magnetization is compensated). Therefore, the magnetization M TR During the on-time T ON1 During this time, it also changes its polarity (see Figure 3 , the sixth timing diagram). Figure 3 In, t 11 Represents the magnetization intensity M TR is zero and changes its polarity. ON2 The magnetization intensity (magnetic flux) M obtained in TR will be called negative magnetization and at time t 11 The magnetization thereafter will be referred to as positive magnetization.

[0034] Negative pre-magnetization of the transformer 20 includes magnetically storing energy in the transformer 20. During the pre-magnetization period T ON2 After that, that is, at the first delay time T DEL1 (During the pre-magnetization period T ON2 With the conduction period T ON1 During the period between 20 and 100 Hz, the energy stored in the transformer 20 and the parasitic capacitance C of the electronic switch T1 DS (See Figure 1 ) causes the voltage V DS Oscillation (see Figure 3 , the first timing diagram). Figure 2 The demagnetization period T DEMAG The amplitude of the oscillation at the end is higher than that at the end of the oscillation, because the transformer 20 is negatively magnetized. DS At the first delay time T DEL1 The minimum voltage V reached during DSmin2 Specific voltage V DS During the demagnetization period T DEMAG The minimum value V reached after the end DSmin1 According to one embodiment, the first delay time T DEL1 is selected so that the first electronic switch T1 is DS Reach the minimum value V DSmin2 The first delay time T DEL1 Controlled by the control circuit 10.

[0035] During the on-time T ON1 At the beginning of the pre-magnetization period T ON2 The energy stored in the transformer 20 is transferred to the input and output capacitors C INThe time period during which the energy is transferred to the input is determined by the primary current i P is negative (see Figure 3 , the fifth timing diagram) is shown in the time period. When the primary current i P At time t 11 reaches zero (and the magnetization intensity M TR When the transformer 20 reaches zero, the transformer 20 is demagnetized. 11 After that, the transformer 20 is positively magnetized until the first electronic switch T1 is turned off at time t2. ON1 Afterwards, during the demagnetization period T DEMAG During this period, the energy stored in the transformer 20 is transferred to the secondary winding L S , rectifier circuit 30 and load, as previously referred to Figure 1 and Figure 2 As described.

[0036] When Figure 3 When operating the flyback converter as shown, the switching losses are lower than those in QR mode (see Figure 2 ) is low, because the first electronic switch T1 is at a lower level voltage V DS conduction, that is, with the same V DSmin1 Relative V DSmin2 Pre-magnetizing the transformer 20 is not associated with significant losses because the energy used to pre-magnetize the transformer 20 is fed back to the input and is thus fed back to the input capacitor C IN The voltage level V DSmin2 In particular, it depends on the pre-magnetization period T ON2 , where the voltage level V DSmin2 With the pre-magnetization period T ON2 According to one embodiment, the pre-magnetization period T ON2 is adjusted so that the second minimum value V DSmin2 The voltage level of is above zero. The electronic switch T1 may have a parasitic capacitance C DS (See Figure 1 ), parasitic capacitance C DS As the load path voltage V DS Therefore, the desired voltage level V DSmin2 The lower the pre-magnetization period T ON2 The energy to be stored in the transformer 20 increases disproportionately during the first delay time T DEL1 During this period, it is necessary to discharge the parasitic capacitance. According to one embodiment, the pre-magnetization period T ON2 is set so that the voltage level V DSmin2 Between 5 volts and 50 volts.

[0037] In each driving cycle, the pre-magnetization period T ON2 During the demagnetization period T DEMAG That is, it is not necessary to start at any time during the demagnetization period T DEMAG Then the voltage V DS The pre-magnetization period T starts at a specific phase of oscillation ON2 Therefore, the transformer 20 can be pre-magnetized at a predefined (fixed) frequency. That is, the pre-magnetization period T in one driving cycle ON2 The beginning of the continuous drive cycle is the pre-magnetization period T ON2 The time period T between the start of the pre-magnetization period T can be constant. For example, if the pre-magnetization period T ON2 The first delay time T is substantially the same in each driving cycle and DEL1 is substantially the same in each driving cycle, the switching frequency of the first electronic switch T1 is equal to the predefined (fixed) frequency f=1 / T at which the transformer is pre-magnetized. Therefore, Figure 3 The illustrated concept for operating a flyback converter may provide a fixed frequency resonant operation of the electronic switch T1 in the flyback converter. "Fixed frequency resonant operation" means that the switching frequency is substantially constant during the operation of the flyback converter, independent of the power consumption of the load, while maintaining the advantages of quasi-resonant operation. According to another embodiment, the controller 10 is configured to change the switching frequency based on the power consumption of the load connected to the flyback converter, wherein the switching frequency may decrease as the power consumption decreases. The power consumption of the load is determined by the feedback signal S FB express.

[0038] Depends on the feedback signal S FB And thus depending on the power consumption of the load connected to the output of the flyback converter, the conduction period T of the first electronic switch T1 can be controlled in a conventional manner. ON When the power consumption of the load increases, the conduction period T ON1 Therefore, the demagnetization period T DEMAG becomes longer and the second delay time T DEL2 Become shorter (see Figure 3 , the sixth timing diagram). On the contrary, if the power consumption of the load decreases, the conduction period T ON1 Therefore, the demagnetization period T DEMAG becomes shorter and the second delay time T DEL2 Become longer.

[0039] exist Figure 3 In, T OSC Indicates the demagnetization period T DEMAG Then the voltage V DS The period of parasitic oscillation. OSC Basically equal to the voltage VDS Reduced first delay time T DEL1 The first delay time T DEL1 It can be basically one quarter of the oscillation period, that is, T OSC / 4(See Figure 3 , the first and sixth figures). The oscillation period T OSC Depends on the input voltage V IN Therefore, according to one embodiment, according to the input voltage V IN The voltage level adjusts the first delay time T DEL1 . Input voltage V IN The voltage level of V is usually constant or changes only slowly during the operation of the flyback converter. Therefore, the voltage level can be assumed to be substantially constant in a number of subsequent drive cycles and varies according to the input voltage V IN Adjust the first delay time T DEL1 The operation of the first electronic switch T1 at the predefined frequency f=1 / T mentioned above is not affected. The same applies to the operation of the first electronic switch T1 at the predefined frequency f=1 / T mentioned above. IN Adjust the pre-magnetization period T ON2 According to one embodiment, the pre-magnetization period T ON2 is adjusted so that it follows the input voltage V IN becomes longer as the voltage level increases.

[0040] The second delay time T in a driving cycle can be DEL2 The oscillation period is measured during T OSC , and based on this measurement, the first delay time T may be adjusted in one or more subsequent drive cycles DEL1 As mentioned above, the oscillation period T OSC Depends on the input voltage V IN The voltage level is constant or changes only slowly, so the oscillation period T is not measured in each drive cycle. OSC May be sufficient (but only measured occasionally).

[0041] The auxiliary voltage V AUX (See Figure 1 ) to measure the oscillation period T OSC . Measure the oscillation period T OSC It can include: detecting the auxiliary voltage V AUX becomes zero; and measure the auxiliary voltage V AUX The first time it reaches zero is t 31 With auxiliary voltage V AUX The third time it reaches zero is t 32 According to another embodiment, the auxiliary voltage VAUX The time difference between two subsequent moments of reaching zero. This time corresponds to the oscillation period T OSC half of T OSC / 2. Based on this, the oscillation period T can be determined OSC .

[0042] Figure 3 The concept shown can be implemented in different ways. According to one embodiment, the pre-magnetization period T ON2 Start with a predefined frequency f=1 / T. Referring to above, the predefined frequency may be fixed or may depend on the power consumption of the load. The clock signal S CLK It can be used to define these pre-magnetization periods T ON2 The beginning of Figure 3 In, t 01 and t4 represent the times when the pre-magnetization period starts. ON2 , the first delay time T DEL1 and conduction period T ON1 It can be adjusted as described above. DEMAG and the second delay time T DEL2 Depends on the conduction period T ON1 And automatically adjust as follows:

[0043] T DEL2 +T DEMAG =T–(T ON1 +T DEL1 +T ON2 ). (3)

[0044] According to another embodiment, the conduction period T ON1 The control circuit 10 starts with a predefined frequency f=1 / T, which may be fixed or depend on the power consumption of the load. That is, the control circuit 10 turns on the first electronic switch T1 at the predefined frequency. The clock signal S CLK Can be used to limit the conduction period T ON1 Those times at the beginning, that is, those times when the electronic switch T1 is turned on. Figure 3 In the figure, t1 and t5 represent the conduction period T ON1 Again, the conduction period T ON1 , pre-magnetization period T ON2 and the first delay time T DEL1 The adjustment can be made as described above. Based on these parameters and based on the period T of one driving cycle, the pre-magnetization period T can be calculated. ON2 Start time. Figure 3 In the example, a conduction period T ON1 The beginning of the next pre-magnetization period TON2 The time period between the start and end of the conduction period is equal to the conduction period T ON1 Plus the demagnetization period T DEMAG Add the second delay time T DEL2 (T ON1 +T DEMAG +T DEL2 ). As follows, according to the period T of a driving cycle, the pre-magnetization period T ON2 and the first delay time T DEL1 This time period can be easily calculated.

[0045] T ON1 +T DEMAG +T DEL2 =T-(T ON2 +T DEL1 ). (4)

[0046] According to another embodiment, by adjusting (calculating) the second delay time T DEL2 To adjust the period T and switching frequency (which is 1 / T) of a driving cycle respectively. Figure 3 , the period T of one driving cycle is given by:

[0047] T=T ON1 +T DEMAG +T DEL2 +T ON2 +T DEL1 , (5a)

[0048] The conduction period T ON1 , pre-magnetization period T ON2 and the first delay time T DEL1 The adjustment (calculation) can be performed as mentioned above. Demagnetization period T DEMAG According to the conduction period T ON1 Therefore, considering the parameter T ON1 、T ON2 and T DEL1 (by the controller 10) is adjusted and T DEMAG Automatically adjust the second delay time T DEL2 To adjust the desired period T of a driving cycle. In this case, there is no need to define the conduction period T separately. ON1 Or pre-magnetization period T ON2 Based on equation (5a), the second delay time T DEL2 can be calculated as follows:

[0049] T DEL2 =T-(T ON1 +T DEMAG +TON2 +T DEL1 ). (5b)

[0050] The period T of the desired drive cycle may be fixed or may depend on the power consumption of the load. In the latter case, the period T of the drive cycle may increase as the power consumption decreases, so that the switching frequency decreases as the power consumption decreases.

[0051] Adjust the second delay time T based on equation (5a) and equation (5b) DEL2 And therefore adjusting the period T of the drive cycle (switching frequency) requires measuring the demagnetization period T DEMAG Measuring the demagnetization period may include measuring the conduction period T at time t2 ON1 The time period between the end of the pre-magnetization period T and the time t3 when the transformer has been demagnetized. Time t2 is the time when the drive signal S1 switches to the off level. In addition, time t2 is determined by the pre-magnetization period T ON2 The time at the start plus the pre-magnetization period T ON2 , the first delay time T DEL1 and conduction period T ON1 The latter is adjusted (calculated) as described above. DEL2 As defined above, one driving cycle is in the pre-magnetization period T ON2 Start from the beginning of .

[0052] According to one embodiment, by adjusting the third delay time T DEL2 'To adjust the period T of the driving cycle. Figure 3 , the third delay time T DEL2 ' is equal to the second delay time T DEL2 Subtract the oscillation period T OSC That is,

[0053] T DEL2 '=T DEL2 -T OSC / 4. (6)

[0054] Based on equations (5b) and (6), the third delay time T DEL2 ' can be calculated as follows:

[0055] T DEL2 '=T-(T ON1 +T DEMAG +T OSC / 4+T ON2 +T DEL1 ). (7)

[0056] In other words, considering T ON1 , T ON2 and TDEL1 Adjust as described above and T DEMAG and T OSC / 4 automatically adjusts, and can be adjusted by adjusting the third delay time T DEL2 'To adjust the period T of the driving cycle. Adjust the second delay time T DEL2 And therefore, adjusting the period T (switching frequency) of the driving cycle based on equation (7) requires measuring the demagnetization period T DEMAG Plus an oscillation period T OSC Measure the demagnetization period T DEMAG Plus an oscillation period T OSC A quarter of the time period T may include: detecting the conduction period T at time t2 ON1 The end of the conduction period T ON1 After the auxiliary voltage V AUX When does it first cross zero? Figure 3 In, t 31 Indicates that during the conduction period T ON1 After the auxiliary voltage V AUX The first time it crossed zero.

[0057] Figure 4 shows the use of Figure 3 An example of a flyback converter for the discussed constant frequency switching concept. Figure 4 The flyback converter shown with Figure 1 The flyback converter is essentially the same. However, Figure 4 The flyback converter shown has an additional supply circuit 50, which is coupled to the auxiliary winding L AUX and is configured to generate a supply voltage V for the control circuit 10 CC The supply circuit 50 includes a rectifier element D such as a diode. S and the auxiliary winding L AUX The capacitor C connected in parallel S In this example, the above Figure 3 The second electronic switch mentioned in the discussion is a diode D S Electronic switch T2 connected in parallel. Supply voltage V CC The capacitor C S The second electronic switch T2 receives the second drive signal S2 provided by the control circuit 10. As described above, the drive signal S2 defines the pre-magnetization period of the transformer 20 (see Figure 3 ).

[0058] References have been made above including Figure 3 The timing diagram in Figure 4In the demagnetization period T DEMAG During this period, energy is not only transferred to the secondary winding L S , rectifier circuit 30 and load, and through the auxiliary winding L AUX and diode D S is delivered to the capacitor C of the supply circuit 50 S During the pre-magnetization period T ON2 During this period, the control circuit 10 closes the second electronic switch T2. This causes the transformer 20 to be pre-magnetized, and during the pre-magnetization period T ON2 The energy stored in the transformer 20 is supplied from the capacitor C of the circuit 50 during S Obtained.

[0059] Optionally, the control circuit 10 may be connected via a resistor R IN Receive input voltage V IN The input voltage V received by the control circuit 10 IN It can be used to supply the control circuit 10 before the electronic switch T1 is turned on for the first time, that is, before the flyback converter starts. Alternatively, the control circuit 10 can use the input voltage V IN The level information is used to control the pre-magnetization period T ON2 and the first delay time T DEL1 Alternatively, the oscillation period T may be measured as discussed further above. OSC To adjust the first delay time T DEL1 .

[0060] The control circuit 10 may be implemented using dedicated analog circuits or using hardware and software. According to one embodiment, the control circuit comprises a microprocessor or a microcontroller on which a program configured to perform a reference operation is executed. Figure 3 Software for the described method.

[0061] The control circuit 10 can operate in current mode (CM). In this case, the control circuit 10 receives a current from a current sensing resistor R connected in series with the first electronic switch T1. CS Receive voltage V CS The voltage V CS With the primary current i P In this embodiment, the control circuit 10 is configured to be proportional to the voltage V CS and feedback signal S FB For the conduction period T ON1 The control circuit 10 is configured to adjust the output voltage based only on the feedback signal S FB To calculate the conduction period TON1 .

[0062] As discussed above, the transformer 20 is pre-magnetized during the pre-magnetization period T of each driving cycle. ON2 During (see Figure 3 ) is pre-magnetized by Figure 4 The electronic switch T2 in the supply circuit 50 in the example of FIG. 5 is triggered. Therefore, the supply circuit 50 has two functions / purposes, namely, first, to generate a supply voltage V for the control circuit 10. CC , and second, for using the auxiliary winding L AUX The transformer 20 is pre-magnetized. Figure 5 (Its Figure 5 In this example, which is very similar to the previous example of , the two functions are separated. Therefore, in Figure 5 In the example of FIG. 1 , the supply circuit 50 is only configured to generate a supply voltage V for the control circuit 10. CC , and the pre-magnetization function is coupled to another auxiliary winding L AUX Therefore, the supply circuit 50 basically includes a diode D coupled in series. S and capacitor C S , where the series circuit is connected in parallel to the auxiliary winding L AUX . Supply voltage V CC is the capacitor C S The pre-magnetization circuit 60 includes a capacitor C PM and a second electronic switch T2 which is implemented as a MOS transistor in this example. The electronic switch T2 is coupled to the capacitor C PM With another auxiliary winding L AUX ', so that when the electronic switch T2 is closed, the capacitor C PM and another auxiliary winding L AUX ' parallel connection. When the electronic switch T2 is turned off, the intrinsic reverse diode of MOSFETT2 allows the capacitor C to supply the circuit. S The charging is carried out in a similar manner to the capacitor C PM to charge.

[0063] Figure 5 Examples also include connecting in parallel to the auxiliary winding L AUX The resistor R1 and the resistor R2 form a voltage divider. Figure 5 Unlike the previous examples, the control circuit receives a scaled voltage V AUX '≈V AUX · R2 / (R1 + R2). Optionally, a small capacitor C2 may be connected in parallel to the resistor R2.

[0064] In addition to the aspects described above (separation of the pre-magnetization circuit 60 from the supply circuit 50, the voltage divider R1, and the voltage divider R2), Figure 5 The primary side of the flyback converter is Figure 5 The primary side of the flyback converter in the previous example is substantially the same and is described above with reference to the above. On the secondary side, the rectifier circuit 30 is implemented in a different manner than in the previous example. Figure 4 In the example, a simple diode is used as a rectifying element (see Figure 4 , diode D R ),but Figure 5 The rectifier circuit 30 basically comprises a transistor T SR The so-called synchronous rectifier is composed of a MOSFET (for example) and a controller circuit 31. The synchronous rectifier is used to replace the ordinary silicon diode because the MOSFET T SR When turned on, this results in a significantly lower forward voltage than a silicon diode and thus improves power dissipation. However, synchronous rectifiers require a controller circuit ( Figure 5 , the controller circuit 31 is labeled "SR controller"), which turns the MOSFET on and off at the right moment. Controller circuits for synchronous rectifiers are known and available from various manufacturers (e.g., synchronous rectifier controller NCP4304 from ON Semiconductor). The controller circuit 31 can be configured to monitor the transistor T SR The voltage V across the load current path SR (i.e., the drain-source voltage in the case of a MOSFET) and at a voltage V SR When the MOSFET T SR In other words, MOSFET T SR At voltage V SR The controller circuit 31 turns on when the falling edge of the voltage V SR Shortly before θ becomes positive, the MOSFET is turned off again.

[0065] SR controller circuit 31 How to switch MOSFET T SR An example is included in Figure 6 The top diagram shows the timing diagram of the MOSFET T SR The voltage V across the load current path SR , while the bottom graph shows the waveform generated by the SR controller circuit 31 and fed to the MOSFET T SR The gate supply of the corresponding gate signal V GSR .like Figure 6 As shown, MOSFET T SRThe turn-on level is lower than the threshold level V SRTH ( Figure 6 The time t ON ) voltage V SR Various concepts can be used to detect the MOSFET T SR The moment of being turned off again is t OFF It should be noted that the threshold level V SRTH Can be close to zero (positive or negative) or zero. Different thresholds can be used to detect the MOSFET T SR For example, the threshold for determining the on-time may be negative, such as -0.2V, and the threshold for determining the off-time may be zero volts.

[0066] Refer again Figure 5 Obviously, the transistor T SR The voltage V across the load current path SR Equal to the output voltage V OUT With the secondary winding L S The voltage drop across the terminals is V LS Assuming the output voltage is essentially constant or changes only slowly, the voltage V SR The falling edge (see Figure 6 , top figure) also exists in the secondary winding L S The voltage across the two ends is V LS middle.

[0067] When the conventional QR mode in which the pre-magnetization circuit 60 is not required (see, for example Figure 2 ) when operating, Figure 5 The circuit shown and in particular the rectifier circuit 30 (synchronous rectifier) ​​operates as discussed above. However, when operating in a fixed frequency resonant mode, i.e., in the above referenced Figure 3 During the fixed frequency operation discussed, problems may occur. The negative pre-magnetization of the transformer 20 produces a negative magnetization gradient (see Figure 3 , t 01 With t 02 The magnetization intensity M TR ). This negative magnetization gradient causes a voltage V LS During the pre-magnetization period T ON2 During the demagnetization period T DEMAG In a similar way, during the demagnetization period, T DEMAG The magnetization intensity gradient is also negative during the pre-magnetization period T ON2 During the voltage V LS The decrease in the current may trigger the electronic switch T included in the rectifier circuit 30. SRThis undesired turn-on can cause an undesired short circuit (shoot-through), which can damage the MOSFET T of the synchronous rectifier. SR In addition, depending on the implementation, the MOSFETT SR The premature turn-on of the first electronic switch T1 may also damage or destroy the first electronic switch T1. SR And make MOSFET T SR The on-state is maintained for a minimum on-time in order to avoid undesired switching. During this minimum on-time, the first electronic switch T1 is also turned on and then a shoot-through (short circuit) occurs, which may damage the MOSFET T SR and the first electronic switch T1.

[0068] The above-mentioned undesirable transistor T is performed by the synchronous rectifier controller circuit 31. SR The conduction is also included in Figure 7 The timing diagram is shown in FIG. Figure 7 The first three timing diagrams (from the top) in Figure 3 These figures show exemplary waveforms of the drive signals S1 and S2 for the first electronic switch T1 and the second electronic switch T2, respectively. The fourth timing diagram shows the transistor T1 of the synchronous rectifier circuit 30. SR The voltage across the two ends is V SR The fifth figure is a corresponding waveform of the drive signal (gate signal) V generated by the synchronous rectifier controller circuit 31. GSR .like Figure 7 It can be seen that the pre-magnetization period T ON2 The pre-magnetization period results in a voltage V SR The corresponding falling edge is detected by the synchronous rectifier controller circuit 31, so the synchronous rectifier controller circuit 31 is used for the transistor T SR The driving signal V GSR As mentioned above, the SR controller can be (or needs to be) designed so that the MOSFET T SR (i.e., the SR switch) remains in the on state for a minimum period of time T MIN In order to avoid undesired switching. It may happen that the minimum time period T MIN The conduction period T of the first electronic switch T1 ON1 Overlapping situation. In this case, when the two switches T1 and T SR When both switches T1 and T SR may be damaged or destroyed. Figure 7 In the case shown, the dashed line shows the minimum time period TMIN And a short circuit occurs at time t1 when the first electronic switch T1 is turned on.

[0069] As mentioned above, the transistor T SR The voltage V across the load current path SR Equal to the output voltage V OUT With the secondary winding L S The voltage across the two ends is V LS The difference between (V SR =V OUT -V LS ). In the pre-magnetization period T ON2 During this period, the secondary winding L S The voltage across the two ends is V LS Basically, with another auxiliary winding L AUX The voltage across the two ends is V PM Therefore,

[0070] V SR =V OUT -V PM ·N S / N AUX ', (8)

[0071] Among them, N S Indicates the secondary winding L S The number of windings and N AUX ' represents another auxiliary winding L AUX 'The number of windings.

[0072] As can be seen from equation (8), during the pre-magnetization period T ON2 During this period, the auxiliary winding L can be applied to reduce AUX 'Voltage V PM To reduce the voltage signal V SR To avoid the transistor T included in the synchronous rectifier circuit 30 SR To prevent the undesired early turn-on of the MOSFET, the embodiments described below include an improved pre-magnetization circuit 60. These pre-magnetization circuits are configured to ON2 During this period, the voltage V PM Applied to another auxiliary winding L AUX ', voltage V PM This low makes the generated transistor T SR The voltage across the two ends is V SR The drop of can not trigger the transistor T SR conduction.

[0073] exist Figure 8 In the embodiment of the present invention, the pre-magnetization circuit 60 includes not only a single capacitor (such as Figure 5 ), and includes a first capacitor C PM1 , the second capacitor C PM2 and three diodes D P1 , D P2 and D P3 A more complex capacitor circuit. Capacitor C PM1 , C PM2 Through the diode D P1 , D P2 and D P3 coupled, so that when the capacitor circuit passes through the intrinsic reverse diode D of the transistor R When charging, the capacitor C PM1 , C PM2 effectively connected in series (via diode D P3 ). Therefore, the capacitor C PM1 , C PM2 A capacitive voltage divider is formed, where the voltage across each capacitor is approximately V PM / 2 (when the voltage drop across the diode is neglected). Diode D P1 , D P2 reverse bias and therefore the capacitor C PM1 , C PM2 Blocked during charging. During the pre-magnetization period T ON2 During this period, transistor T2 is turned on, and in this case, diode D P3 is reverse biased and blocked, and thus capacitor C PM1 , C PM2 are effectively connected in parallel. Therefore, during the pre-magnetization period T ON2 During this period, the voltage V PM / 2 is applied to another auxiliary winding L AUX ', the voltage V PM / 2 is low enough to avoid undesired premature triggering of the synchronous rectifiers.

[0074] exist Fig. 9 In the embodiment, the pre-magnetization circuit 60 includes a capacitor C PM , capacitor C PM The second electronic switch T2 can be turned off through the diode D P3 Charging. Due to the diode D P3 The voltage drop across the terminals (forward voltage V F ≈0.7V), the capacitor is charged to a value equal to V PM -V F Voltage. In the pre-magnetization period T ON2 During this period, transistor T2 is turned on, and in this case, diode D P3 is reverse biased. Therefore, the voltage V PM-V F Through diode D P1 is applied to another auxiliary winding L AUX ', the diode D P1 effectively connected in series with capacitor C PM When the diode D P1 Cause V F The other auxiliary winding “sees” V PM -2V F If this level is not low enough to avoid (or reduce) the voltage level of the transistor T in the synchronous rectifier circuit 30, SR The voltage across the two ends is V SR The diode D P1 It can be replaced by a series circuit of two or more diodes.

[0075] In both implementations ( Figure 8 and Fig. 9 ), the pre-magnetization circuit 60 includes a capacitor circuit, which is charged to a maximum voltage level V PM (The second electronic switch T2 is turned off while the capacitor circuit is charged.) However, when the energy stored in the capacitor circuit is released to the other auxiliary winding L AUX ' (ie, pre-magnetization period T ON2 During this period, see Figure 3 ), applied to another auxiliary winding L AUX The maximum voltage of ' is lower than the maximum voltage level V in the charging stage PM .

[0076] Although various embodiments have been shown and described with respect to one or more specific embodiments, changes and / or modifications may be made to the illustrated examples without departing from the spirit and scope of the features and structures described herein. In particular, with respect to the various functions performed by the above-described components or structures (units, components, devices, circuits, systems, etc.), unless otherwise indicated, terms used to describe such components (including references to "means") are intended to correspond to any component or structure that performs the specified function of the described component (e.g., functionally equivalent), even if structurally not equivalent to the disclosed structure that performs the functions in the exemplary implementation of the present disclosure shown herein.

Claims

1. A switching converter, comprising: transformer, which includes a primary winding (L P ), secondary winding (L S ) and at least one auxiliary winding (L AUX '); A first electronic switch (T1) is coupled to the primary winding (L P ) and is configured to make the primary current (i P ) turn on and off; A synchronous rectifier circuit (30) is coupled to the secondary winding (L S ) between the output node of the switching converter and including a transistor (T SR ) and is configured to turn on and off the transistor (T SR ) of the controller (31); The pre-magnetization circuit (60) comprises a capacitor circuit and a second electronic switch (T2), wherein the second electronic switch (T2) is connected to the auxiliary winding (L AUX ') and the capacitive circuit; The capacitor circuit is configured to: when the second electronic switch (T2) is closed, during the pre-magnetization period (T ON2 ) during which a first voltage is applied to the auxiliary winding (L AUX ') and when the second electronic switch (T2) is turned off, it is charged to a second voltage (V PM ), the first voltage is lower than the second voltage, so that in the pre-magnetization period (T ON2 ) to prevent the controller (31) from turning on the transistor (T SR ).

2. The switching converter according to claim 1, in, The capacitive circuit includes at least one capacitor coupled to the secondary winding via a diode, the diode causing a voltage drop (V F ), so that during the pre-magnetization period (T ON2 ) is applied to the auxiliary winding (L AUX ') is greater than the second voltage (V PM ) is at least the voltage drop (V F ).

3. The switching converter according to claim 2, in, The capacitive circuit includes a first capacitor (C PM1 ) and connected to the first capacitor (C PM1 ) of the first diode (D P1 ), and further comprising a second capacitor (C PM2 ) and connected to the second capacitor (C PM2 ) of the second diode (D P2 ), wherein the first capacitor (C PM1 ) and the first diode (D P1 ) between the common circuit node and the second capacitor (C PM2 ) and the second diode (D P2 ) through a third diode (D P3 )connect.

4. The switching converter according to any one of claims 1 to 3, in, The synchronous rectifier circuit (30) includes a transistor (T SR ) and the output capacitor (C R ), the output capacitor (C R ) through the transistor (T SR ) is coupled to the secondary winding (L S ),and The synchronous rectifier circuit (30) comprises a controller circuit (31), wherein the controller circuit (31) is configured to sense the transistor (T SR ) of the load current path (V SR ) and when the transistor (T SR ) of the load current path (V SR ) drops to a predefined threshold level (V SRTH ) below, the transistor (T SR ) is turned on.

5. The switching converter according to any one of claims 1 to 3, further comprising: A control circuit (10) configured to generate the first drive signal (S1) for the first electronic switch (T1) and the second drive signal (S2) for the second electronic switch (T2), wherein in each switching cycle, the first drive signal (S1) turns on the first electronic switch (T1) for a conduction period (T ON1 ).

6. The switching converter according to claim 5, in, The first driving signal (S1) or the second driving signal (S2) is synchronized with a clock signal.

7. The switching converter according to any one of claims 1 to 3, in, During operation, the auxiliary winding (L AUX ') in the conduction period (T ON1 ) demagnetization period (T DEMAG ) to provide the second voltage (V PM ).

8. The switching converter according to claim 5, in, In each switching cycle, the second drive signal (S2) turns on the second electronic switch (T2) for the pre-magnetization period (T ON2 ), the pre-magnetization period (T ON2 ) occurs during the previous demagnetization period (T DEMAG )after.

9. The switching converter according to any one of claims 1 to 3, further comprising: Another auxiliary winding (L AUX ); as well as A supply circuit (50) coupled to the other auxiliary winding (L AUX ) and is configured to generate a supply voltage (V CC ).

10. The switching converter according to any one of claims 1 to 3, in, During the pre-magnetization period (T ON2 ) is applied to the auxiliary winding (L AUX The maximum level of the first voltage during the charging phase in which the capacitor circuit is charged is lower than the second voltage (V PM )’s maximum level.

11. A method for operating a switching converter, the switching converter comprising: transformer, which includes a primary winding (L P ), secondary winding (L S ) and at least one auxiliary winding (L AUX '); a first electronic switch (T1) coupled to the primary winding (L P ); a synchronous rectifier circuit (30) coupled to the secondary winding (L S ) between the output node of the switching converter and including a transistor (T SR ) and is configured to turn on and off the transistor (T SR ) controller (31); and a pre-magnetization circuit (60), which includes a capacitor circuit and a second electronic switch (T2), wherein the second electronic switch (T2) is connected to the auxiliary winding (L AUX ') between the capacitor circuit; the method comprising: The first electronic switch (T1) is used to make the primary current (i P ) turn on and off; During the pre-magnetization period (T ON2 ) during the period of closing the second electronic switch (T2) to apply the first voltage provided by the capacitor circuit to the auxiliary winding (L AUX ');as well as Before closing the second electronic switch (T2), when the second electronic switch (T2) is turned off, the capacitor circuit is charged to a second voltage (V PM ), wherein the first voltage is lower than the second voltage, so that during the pre-magnetization period (T ON2 ) to prevent the controller (31) from turning on the transistor (T SR ).

12. The method according to claim 11, in, The difference between the second voltage and the first voltage is caused at least in part by a voltage drop across a diode included in the capacitive circuit.

13. The method according to claim 11 or 12, in, During the pre-magnetization period (T ON2 ) is applied to the auxiliary winding (L AUX The maximum level of the first voltage during the charging phase in which the capacitor circuit is charged is lower than the second voltage (V PM )’s maximum level.

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