Methods, systems and devices for continuous glucose monitoring

Abstract

A continuous glucose monitoring system can utilize electrode current (Isig) signals, electrochemical impedance spectroscopy (EIS), and Vcntr values ​​to optimize sensor glucose (SG) calculations in a manner that can reduce the need for blood glucose (BG) calibration requests from a user.

Description

[0001] Related application information

[0002] This application claims the benefit of the filing date of U.S. Provisional Application Serial No. 62 / 800,297, filed February 1, 2019, which is incorporated herein by reference in its entirety. Technical Field

[0003] The present technology generally relates to sensor technology, including sensors for sensing various physiological parameters, such as glucose concentration, and to achieving reduced need for blood glucose (BG) calibration requests from users in glucose sensing systems, devices, and methods. Background Art

[0004] Over the years, a variety of sensors have been developed for detecting and / or quantifying specific agents or compositions in a patient's blood, enabling patients and medical personnel to monitor physiological conditions within the patient's body. Illustratively, a subject may wish to continuously monitor the blood glucose level within the subject's body. Consequently, glucose sensors have been developed to obtain an indication of a diabetic patient's blood glucose level. Such readings are useful for monitoring and / or adjusting treatment regimens, which typically include the regular administration of insulin to the patient.

[0005] Currently, patients can measure their blood glucose (BG) using a blood glucose measuring device (i.e., a glucometer) such as a test strip meter, a continuous glucose measurement system (or continuous glucose monitor), or a hospital hemacue. BG measuring devices use various methods to measure a patient's BG level, such as a blood sample from the patient, a sensor that contacts bodily fluids, an optical sensor, an enzymatic sensor, or a fluorescent sensor. When a BG measuring device has generated a BG measurement value, the measurement value is displayed on the BG measuring device.

[0006] Current continuous glucose measurement systems include subcutaneous (or short-term) sensors and implantable (or long-term) sensors. The sensors have been used in telemetry characteristic monitoring systems. As described, for example, in commonly assigned U.S. Patent No. 6,809,653, the entire contents of which are incorporated herein by reference, a telemetry system using an electrochemical sensor includes a remotely located data receiving device, a sensor for generating a signal indicating a user characteristic, and a transmitter device for processing the signal received from the sensor and for wirelessly transmitting the processed signal to the remotely located data receiving device. The data receiving device can be a characteristic monitor, a data receiver providing data to another device, an RF programmer, a drug delivery device (e.g., an infusion pump), etc.

[0007] Regardless of whether the data receiving device (e.g., glucose monitor), the transmitter device, and the sensor (e.g., glucose sensor) communicate wirelessly or by connecting a wire, the characteristic monitoring system of the above type only has practical use after being calibrated based on the unique characteristics of an individual user. According to the prior art, the user needs to perform external calibration on the sensor. More specifically, and in conjunction with the illustrative example of a diabetic patient, the latter is required to take finger-stick blood glucose meter readings two to four times a day on average for the duration used by the continuous characteristic monitoring system. Each time blood is drawn from the user's finger and analyzed by the blood glucose meter to provide the user's real-time blood glucose level. The user then inputs this data into the glucose monitor as the user's current blood glucose level, and the data is used to calibrate the glucose monitoring system.

[0008] However, such external calibration is disadvantageous for various reasons. For example, blood glucose meters are not completely accurate and contain inherent margins of error. Furthermore, even if completely accurate, blood glucose meters are susceptible to misuse; for example, if the user handles sugar or other sugary substances immediately before performing a fingerstick, and some of the sugar gets on the user's finger, the blood glucose analysis will produce an inaccurate indication of the blood glucose level. Furthermore, there are costs associated with each fingerstick application, not to mention the pain and discomfort.

[0009] The current state of the art in continuous glucose monitoring (CGM) is largely adjunctive, meaning that the readings provided by a CGM device (comprising, for example, an implantable or subcutaneous sensor) cannot be used to make clinical decisions without a reference value. Reference values, in turn, must be obtained from a fingerstick using, for example, a BG meter. The need for a reference value stems from the limited amount of information available from the sensor / sensing assembly. Specifically, the only pieces of information currently provided by the sensing assembly for processing are the raw sensor value (i.e., sensor current or Isig) and the reference voltage. Therefore, during analysis, if the raw sensor signal appears abnormal (e.g., if the signal is decreasing), the only way to distinguish between a sensor malfunction and a physiological change in the user / patient's body (i.e., a change in glucose levels in the body) is by obtaining a reference glucose value via a fingerstick. As is well known, reference fingersticks are also used to calibrate sensors.

[0010] The art has sought ways to eliminate or at least minimize the number of fingersticks required to calibrate and assess sensor health. However, given the number and complexity of multiple sensor failure modes, a satisfactory solution has yet to be found. At best, diagnostics based on direct evaluation of Isig or comparison of multiple Isigs, such as from redundant and / or orthogonally redundant sensors and / or electrodes, have been developed. In either case, because Isig tracks glucose levels in the human body, Isig is clearly not analyte-independent. Therefore, Isig itself is not a reliable source of information for sensor diagnostics, nor is it a reliable predictor of continued sensor performance.

[0011] Another limitation that has existed in the art to date has been the lack of sensor electronics that can not only operate the sensor, but also perform real-time sensor and electrode diagnostics, as well as diagnostics for redundant electrodes, all while managing the sensor's power supply. To be sure, the concept of electrode redundancy has been around for some time. However, until now, using electrode redundancy to obtain more than one reading at a time and assess the relative health of the redundant electrodes, the overall reliability of the sensor, and the required frequency and, if possible, calibration reference values ​​has been largely unsuccessful.

[0012] The art has also sought to provide self-calibrating sensors and more accurate and reliable methods for performing sensor diagnostics by developing various circuit models. These models typically attempt to associate circuit elements with parameters that can be used for intelligent diagnostics, comprehensive fault analysis, and real-time self-calibration. However, to date, most of these models have met with limited success. Summary of the Invention

[0013] In one aspect, the present disclosure provides a method for optimizing sensor glucose calculation of a glucose sensor for measuring glucose levels in a user's body, the sensor comprising a physical sensor electronics device, a microcontroller, a counter electrode, and a working electrode, the method comprising periodically measuring, by the physical sensor electronics device, an electrode current (Isig) signal of the working electrode; periodically executing, by the microcontroller, an electrochemical impedance spectroscopy (EIS) procedure to generate EIS-related data of the working electrode; periodically measuring, by the physical sensor electronics device, a potential difference (Vcntr) between the counter electrode and the working electrode; periodically obtaining a blood glucose (BG) value of the user; determining whether the BG is valid, and if the BG is valid, pairing the BG value with the Isig, Vcntr, and EIS-related data; determining, by the microcontroller, whether there is a calibration error; and when there is no calibration error, calculating, by the microcontroller, a sensor glucose (SG) value based on the paired BG, Isig, Vcntr, and EIS-related data.

[0014] The details of one or more aspects of the disclosure are set forth in the accompanying drawings and description below. Other features, objects, and advantages of the techniques described in this disclosure will be apparent from the description and drawings, and from the claims. BRIEF DESCRIPTION OF THE DRAWINGS

[0015] Embodiments of the present invention will be described in detail with reference to the accompanying drawings, wherein like numerals designate corresponding parts throughout the drawings.

[0016] Figure 1 is a perspective view of the subcutaneous sensor insertion group and a block diagram of the sensor electronics.

[0017] Figure 2A A substrate with two sides is shown, a first side containing an electrode arrangement and a second side containing electronic circuitry.

[0018] Figure 2B Shown is a general block diagram of the electronic circuit used to sense the output of the sensor.

[0019] Figure 3 Shown is a block diagram of sensor electronics and a sensor comprising multiple electrodes.

[0020] Figure 4 An alternative embodiment of the present invention comprising a sensor and sensor electronics is shown.

[0021] Figure 5 An electronic block diagram showing sensor electrodes and voltages applied to the sensor electrodes.

[0022] Figure 6A A method of applying pulses during the stabilization time frame to reduce the stabilization time frame is shown.

[0023] Figure 6B A method of stabilizing the sensor is shown.

[0024] Figure 6C The use of feedback in stabilizing the sensor is shown.

[0025] Figure 7 The effect of stabilizing the sensor is shown.

[0026] Figure 8A A block diagram of the sensor electronics and the sensor including the voltage generating device is shown.

[0027] Figure 8B A voltage generating device is shown.

[0028] Figure 8C A voltage generating device for generating two voltage values ​​is shown.

[0029] Figure 8DA voltage generating device having three voltage generating systems is shown.

[0030] Figure 9A The sensor electronics including a microcontroller for generating the voltage pulses are shown.

[0031] Figure 9B The sensor electronics including the analysis module are shown.

[0032] Figure 10 A block diagram of a sensor system incorporating hydrated electronics is shown.

[0033] Figure 11 An embodiment is shown that includes a mechanical switch to assist in determining hydration time.

[0034] Figure 12 A method for detecting hydration is shown.

[0035] Figure 13A A method of hydrating the sensor is shown.

[0036] Figure 13B An additional method for verifying hydration of the sensor is shown.

[0037] Figure 14A 、 14B 14C and 14C illustrate a method for combining sensor hydration with sensor stabilization.

[0038] Figure 15A An EIS-based analysis of the system response to an applied periodic AC signal is shown.

[0039] Figure 15B A known circuit model for electrochemical impedance spectroscopy is shown.

[0040] Figure 16A An example of a Nyquist plot is shown where an AC voltage plus a DC voltage (DC bias) is applied to the working electrode for a selected spectrum from 0.1 Hz to 1000 MHz.

[0041] Figure 16B Another example of a Nyquist plot is shown having a linear fit for relatively low frequencies and an intercept approaching the real impedance value at relatively high frequencies.

[0042] Figure 16C and Figure 16D Infinite and finite glucose sensor responses to a sinusoidal working potential are shown, respectively.

[0043] Figure 16E A Bode plot of the magnitudes is shown.

[0044] Figure 16F A Bode plot of the phase is shown.

[0045] Figure 17 Nyquist plot showing the variation of sensor impedance as a function of sensor usage.

[0046] Figure 18 A method of applying EIS technology in stabilizing a sensor and detecting the age of the sensor is shown.

[0047] Figure 19 A timeline for performing the EIS procedure is shown.

[0048] Figure 20 A method for detecting and repairing a sensor using an EIS procedure in conjunction with remedial measures is shown.

[0049] Figure 21A and Figure 21B An example of sensor remediation is shown.

[0050] Figure 22 The Nyquist plot of a properly functioning sensor is shown, where the Nyquist slope gradually increases and the intercept gradually decreases as the sensor wearing time progresses.

[0051] Figure 23A Shown are the raw current signals (Isig) from two redundant working electrodes and the corresponding real impedances of the electrodes at 1 kHz.

[0052] Figure 23B Shown Figure 23A Nyquist plot of the first working electrode (WE1).

[0053] Figure 23C Shown Figure 23A Nyquist plot of the second working electrode (WE2).

[0054] Figure 24 Examples of signal dips for two redundant working electrodes and the corresponding real impedances of the electrodes at 1 kHz are shown.

[0055] Figure 25A The substantial glucose independence of the real impedance, imaginary impedance, and phase of a properly operating glucose sensor at relatively high frequencies is shown.

[0056] Figure 25B Illustrative examples of varying levels of glucose dependence of real impedance at relatively low frequencies are shown.

[0057] Figure 25C Illustrative examples of varying levels of glucose correlation of phase at relatively low frequencies are shown.

[0058] Figure 26 Shown are the trends of 1 kHz real impedance, 1 kHz imaginary impedance, and relatively high frequency phase as the glucose sensor loses sensitivity due to oxygen starvation at the sensor insertion site.

[0059] Figure 27 Isig and phase of in vitro simulated hypoxia at different glucose concentrations are shown.

[0060] Figures 28A-28C An example of sensitivity loss due to oxygen starvation and EIS-based parameters of the electrodes is shown in the case of redundant working electrodes WE1 and WE2.

[0061] Figure 28D Shown for Figures 28A to 28C The original Isig of an example of an EIS-induced spike.

[0062] Figure 29 An example of sensitivity loss caused by oxygen deficiency due to occlusion is shown.

[0063] Figures 30A-30C An example of sensitivity loss due to biofouling in the case of redundant working electrodes WE1 and WE2 is shown, along with EIS-based parameters of the electrodes.

[0064] Figure 30D Shown for Figures 30A to 30C The original Isig of an example of an EIS-induced spike.

[0065] Figure 31 A diagnostic routine for sensor fault detection is shown.

[0066] Figure 32A and 32B Another diagnostic routine for sensor fault detection is shown.

[0067] Figure 33A A top-level flow chart is shown involving the current (Isig) based fusion algorithm.

[0068] Figure 33B A top-level flow chart is shown involving the sensor glucose (SG) based fusion algorithm.

[0069] Figure 34 Shown Figure 33B Details of the sensor glucose (SG) based fusion algorithm.

[0070] Figure 35 Shown Figure 33A Details of the current (Isig) based fusion algorithm.

[0071] Figure 36 is a graphical representation of the calibration of the sensor in steady state.

[0072] Figure 37 is a graphical representation of the calibration of the sensor in transition.

[0073] Figure 38A is a graphical representation of the dynamic slope (with slope adjustment) based on EIS.

[0074] Figure 38B A flow chart of EIS-assisted sensor calibration involving low-start detection is shown.

[0075] Figure 39 Shown are sensor current (Isig) and 1 kHz impedance magnitudes for in vitro simulations of interferents in close proximity to the sensor.

[0076] Figure 40A and Figure 40B Shown are the Figure 39 Bode plots of the simulated phase and impedance are shown.

[0077] Figure 40C Shown for Figure 39 The Nyquist plot of the simulation is shown.

[0078] Figure 41 Another in vitro simulation of an interferent is shown.

[0079] Figure 42A and 42B An ASIC block diagram is shown.

[0080] Figure 43 Shown is the potentiostat configuration of the sensor with redundant working electrodes.

[0081] Figure 44 Shown with Figure 43 Equivalent AC inter-electrode circuit of the sensor in the potentiostat configuration shown.

[0082] Figure 45 Some of the main blocks of the EIS circuitry in the analog front end IC of the glucose sensor are shown.

[0083] Figures 46A to 46F Shown Figure 45 The signal of the EIS circuit system shown is a simulation of the current multiplied by the 0-degree phase for 0 degrees.

[0084] Figures 47A to 47F Shown Figure 45 The signal of the EIS circuit system shown is a simulation of the current with a phase of 0 degrees multiplied by a phase of 90 degrees.

[0085] Figure 48 The circuit model is shown.

[0086] Figures 49A-49C A diagram of the circuit model is shown.

[0087] Figure 50A It is the Nyquist plot of the superimposed equivalent circuit simulation.

[0088] Figure 50B yes Figure 50A A magnified view of the high-frequency part.

[0089] Figure 51 The Nyquist plot shows that Cd1 increases in the direction of arrow A.

[0090] Figure 52 The Nyquist plot is shown as α increases in the direction of arrow A.

[0091] Figure 53 The Nyquist plot shows that Rp increases in the direction of arrow A.

[0092] Figure 54 The Nyquist diagram shows that the Weber admittance increases in the direction of arrow A.

[0093] Figure 55 The Nyquist plot is shown as λ increases in the direction of arrow A.

[0094] Figure 56 The effect of membrane capacitance on the Nyquist plot is shown.

[0095] Figure 57 The Nyquist plot shows that the membrane resistance increases in the direction of arrow A.

[0096] Figure 58 The Nyquist plot shows Rsol increasing in the direction of arrow A.

[0097] Figures 59A-59C The changes in EIS parameters associated with circuit elements during startup and calibration are shown.

[0098] Figures 60A-60C The changes in different sets of EIS parameters related to circuit elements during startup and calibration are shown.

[0099] Figures 61A-61C The changes in yet a different set of EIS parameters associated with circuit elements during startup and calibration are shown.

[0100] Figure 62 The EIS responses for multiple electrodes are shown.

[0101] Figure 63is a Nyquist plot showing the effect of calibration by adding glucose Isig.

[0102] Figure 64 The effect of oxygen (Vcntr) response on the Nyquist plot is shown.

[0103] Figure 65 The shift in the Nyquist plot due to temperature changes is shown.

[0104] Figure 66 The relationship between Isig and blood sugar is shown.

[0105] 67A-67B illustrate sensor drift.

[0106] Figure 68 The increase in membrane resistance during sensitivity loss is shown.

[0107] Figure 69 The drop in Weber admittance during sensitivity loss is shown.

[0108] Figure 70 The calibration curve is shown.

[0109] Figure 71 The high frequency semicircle that becomes visible on the Nyquist plot is shown.

[0110] 72A and 72B illustrate Vcntr track movement and Cdl reduction.

[0111] Figure 73 The changing slopes of the calibration curves are shown.

[0112] Figure 74 The varying lengths of the Nyquist plot are shown.

[0113] Figure 75 Shown Figure 74 A magnified view of the low-frequency and high-frequency regions of the Nyquist plot.

[0114] Figures 76A and 76B show the combined effect of increased membrane resistance, reduced Cdl, and Vcntr track shift.

[0115] Figure 77 The relative Cdl values ​​of the two working electrodes are shown.

[0116] Figure 78 The relative Rp values ​​of the two working electrodes are shown.

[0117] Figure 79 The combined effect of varying EIS parameters on the calibration curve is shown.

[0118] Figure 80It is shown that, according to an embodiment of the present invention, the length of the Nyquist diagram in the low-frequency region is longer than in the region where there is a sensitivity loss.

[0119] Figure 81 is a flow chart of sensor self-calibration based on detection of sensitivity changes.

[0120] Figure 82 The horizontal shift in the Nyquist plot due to sensitivity loss is shown.

[0121] Figure 83 A method for developing a heuristic EIS metric based on the Nyquist plot is shown.

[0122] Figure 84 The relationship between Rm and the calibration factor is shown.

[0123] Figure 85 The relationship between Rm and normalized Isig is shown.

[0124] Figure 86 Isig plots showing various glucose levels over time.

[0125] Figure 87 Cdl curves are shown for various glucose levels over time.

[0126] Figure 88 Shown Figure 86 The second inflection point of the graph.

[0127] Figure 89 Shown is the corresponding Figure 88 The second inflection point of the peak in Rm.

[0128] Figure 90 A diagram showing the relationship between the calibration factor (CF) and Rmem+Rsol.

[0129] Figure 91A is a graph showing in vivo results of MARD on all active BGs within approximately the first 8 hours of sensor life.

[0130] Figure 91B is a graph showing the median number of ARDs across all active BGs within approximately the first 8 hours of sensor life.

[0131] Figures 92A-92C The calibration factor adjustment is shown.

[0132] Figures 93A-93C The calibration factor adjustment is shown.

[0133] Figures 94A-94C The calibration factor adjustment is shown.

[0134] Figure 95 An illustrative example of the initial decay of Cd1 is shown.

[0135] Figure 96 The effect of removing the non-Faradaic current on Isig is shown.

[0136] Figure 97A The calibration factors before removing the non-Faradaic currents of the two working electrodes are shown.

[0137] Figure 97B The calibration factors after removing the non-Faradaic currents of the two working electrodes are shown.

[0138] Figure 98A and 98B The effect of removing the non-Faraday current on MARD is shown.

[0139] Figure 99 is a graph of the double layer capacitance changing with time.

[0140] Figure 100 The shift of Rmem+Rsol and the appearance of a high-frequency semicircle during sensitivity loss are shown.

[0141] Figure 101A A flow chart showing the use of combinational logic to detect sensitivity loss is shown.

[0142] Figure 101B A flow chart showing the use of combinational logic to detect sensitivity loss is shown.

[0143] Figure 102 An illustrative method for using the Nyquist slope as a marker to distinguish between new and used sensors is shown.

[0144] 103A-103C show illustrative examples of Nyquist plots for different sensor configurations with different lengths.

[0145] Figure 104 103A to Figure 103C The Nyquist plot length of the sensor over time.

[0146] Figure 105 A flow chart for blanking sensor data or terminating a sensor is shown.

[0147] Figure 106 A flow chart showing sensor termination is shown.

[0148] Figure 107 A flow chart of signal sag detection is shown.

[0149] FIG. 108A shows Isig and Vcntr over time, and FIG. 108B shows glucose over time.

[0150] FIG. 109A is the calibration ratio over time, and FIG. 109B shows glucose over time.

[0151] Figure 110A and 110B The trend of the calibration factor over time is shown.

[0152] Figure 111 A flow chart of the first day calibration (FDC) is shown.

[0153] Figure 112 A flow chart of EIS-based calibration is shown.

[0154] Figure 113 A flow chart of the calibration-free backtracking algorithm is shown.

[0155] Figure 114 A decision tree model is shown.

[0156] Figure 115 A decision tree model for elided data is shown.

[0157] Figure 116 is a table showing an example of parameters of the blanking algorithm.

[0158] Figure 117 The fusion, filtering and blanking results are shown.

[0159] Figure 118 A flow chart and schematic diagram showing the modules of an algorithm for correcting time-varying errors between sensor glucose (SG) and blood glucose (BG) calculations.

[0160] Figure 119 A flow chart and schematic diagram showing the modules of an algorithm for correcting time-varying errors between sensor glucose (SG) and blood glucose (BG) calculations.

[0161] Figure 120 is a table of various models that may be used in accordance with various embodiments.

[0162] Figure 121 is a graph of the model performance.

[0163] Figure 122 A flow chart for operationalizing the optional external calibration algorithm is shown.

[0164] Figure 123A and 123B A flow chart for generating physiological calibration factors is shown.

[0165] Figure 124A and 124B A flow chart for generating environmental calibration factors is shown.

[0166] Figure 125 An idealized elevation conversion scale for physiological and environmental factors is shown.

[0167] Figure 126 An idealized inhibitory conversion scale for physiological and environmental factors is shown.

[0168] Figures 127A-127D Shown are variance estimates calculated for eight different sensor glucose calculation units for 1821 sensors worn by 537 subjects on days 1, 3, 5, and 7 of sensor wear across a range of blood glucose.

[0169] Figure 128 A flow chart comprising modules according to an embodiment of the present invention is shown.

[0170] Figure 129 The embodiment according to the present invention is shown Figure 128 Schematic diagram of the sensor feature generation module.

[0171] Figure 130 Shown Figure 129 Logic diagram of the sensor feature generation module.

[0172] Figure 131 The embodiment according to the present invention is shown Figure 128 Schematic diagram of the blood glucose calibration module.

[0173] Figure 132 Show Figure 131 Logic diagram of the blood glucose calibration module.

[0174] Figure 133 The embodiment according to the present invention is shown Figure 128 Decision logic description of the sensor glucose modeling module shown in .

[0175] Figures 134A-134E A partition matrix including feature names and coefficient values ​​for each partition used for sensor glucose (SG) value calculation according to an embodiment of the present invention is shown. DETAILED DESCRIPTION

[0176] In the following description, reference is made to the accompanying drawings which form a part hereof and which illustrate several embodiments of the present invention. It is understood that other embodiments may be utilized and structural and operational changes may be made without departing from the scope of the present invention.

[0177] The present invention is described below with reference to flowchart illustrations of methods, systems, devices, apparatuses, and programming products and computer program products. It should be understood that each block of the flowchart illustrations and the combination of blocks in the flowchart illustrations can be implemented by programming instructions, which include computer program instructions (such as any menu screens described in the diagrams). These computer program instructions can be loaded onto a computer or other programmable data processing device (e.g., a controller, microcontroller, or processor in a sensor electronic device) to produce a machine, such that the instructions executed on the computer or other programmable data processing device produce instructions for implementing the functions specified in one or more flowchart blocks. These computer program instructions can also be stored in a computer-readable memory, which can direct the computer or other programmable data processing device to function in a specific manner, such that the instructions stored in the computer-readable memory produce an article of manufacture, which includes instructions for implementing the functions specified in one or more flowchart blocks. The computer program instructions can also be loaded onto a computer or other programmable data processing device to cause a series of operating steps to be performed on the computer or other programmable device to produce a computer-implemented process, such that the instructions executed on the computer or other programmable device provide steps for implementing the functions specified in one or more flowchart blocks, and / or the menus presented herein. The programming instructions may also be stored in and / or implemented by electronic circuit systems, including integrated circuits (ICs) and application-specific integrated circuits (ASICs) used in conjunction with sensor devices, apparatuses, and systems. The following terms and definitions may also be used herein:

[0178]

[0179]

[0180] Figure 1 Figure 1 is a perspective view of the subcutaneous sensor insertion assembly and a block diagram of the sensor electronics. Figure 1 As shown, a subcutaneous sensor group 10 is provided for subcutaneously placing an active portion of a flexible sensor 12 (e.g., see FIG. 2 ), etc., within a selected location within the body of a user. The subcutaneous or percutaneous portion of the sensor group 10 comprises a hollow slotted insertion needle 14 and a cannula 16. The needle 14 is used to facilitate quick and easy subcutaneous placement of the cannula 16 at the subcutaneous insertion site. The sensing portion 18 of the sensor 12 is inside the cannula 16, and the sensing portion is used to expose one or more sensor electrodes 20 to the user's bodily fluids through a window 22 formed in the cannula 16. In one embodiment, the one or more sensor electrodes 20 may include a counter electrode, a reference electrode, and one or more working electrodes. After insertion, the insertion needle 14 is withdrawn to leave the cannula 16 with the sensing portion 18 and the sensor electrodes 20 in place at the selected insertion site.

[0181] In certain embodiments, the subcutaneous sensor assembly 10 facilitates the precise placement of a flexible, thin-film electrochemical sensor 12 of a type used to monitor a specific blood parameter indicative of a user's condition. The sensor 12 monitors glucose levels within the body and can be used in conjunction with an external or implantable automatic or semi-automatic drug infusion pump of the type described in, for example, U.S. Patent Nos. 4,562,751; 4,678,408; 4,685,903; or 4,573,994 to control insulin delivery to a diabetic patient.

[0182] Certain embodiments of the flexible electrochemical sensor 12 are constructed according to thin film shielding technology to include an elongated thin film conductor embedded or coated between a selected insulating material (e.g., a polyimide film or sheet) and the membrane. When the sensing portion 18 (or active portion) of the sensor 12 is placed subcutaneously at the insertion site, the sensor electrode 20 at the tip of the sensing portion 18 is exposed through one of the insulating layers to come into direct contact with the patient's blood or other bodily fluids. The sensing portion 18 is joined to a connecting portion 24, which terminates in a conductive contact pad or the like, which is also exposed through one of the insulating layers. In alternative embodiments, other types of implantable sensors may be used, such as chemical-based sensors, optical-based sensors, etc.

[0183] As is known in the art, the connecting portion 24 and the contact pads are generally adapted for direct wired electrical connection to a suitable monitor or sensor electronics 100 to monitor a condition of the user in response to signals derived from the sensor electrodes 20. Further description of this general type of flexible thin film sensor is found in U.S. Patent No. 5,391,250, entitled "METHOD OFFABRICATING THIN FILM SENSORS," which is incorporated herein by reference. The connecting portion 24 can be conveniently electrically connected to the monitor or sensor electronics 100, or connected via a connector block 28 (or the like) as shown and described in U.S. Patent No. 5,482,473, entitled "FLEX CIRCUIT CONNECTOR," which is also incorporated herein by reference. Thus, according to embodiments of the present invention, the subcutaneous sensor group 10 can be configured or formed to operate with either a wired or wireless property monitoring system.

[0184] The sensor electrode 20 can be used in a variety of sensing applications and can be configured in a variety of ways. For example, the sensor electrode 20 can be used in physiological parameter sensing applications in which some types of biomolecules serve as catalysts. For example, the sensor electrode 20 can be used in glucose and oxygen sensors having glucose oxidase (GOx) that catalyzes reactions with the sensor electrode 20. The sensor electrode 20 and the biomolecule or some other catalyst can be placed in the human body in a vascular or non-vascular environment. For example, the sensor electrode 20 and the biomolecule can be placed in a vein and subjected to blood flow, or can be placed in the subcutaneous or peritoneal area of ​​the human body.

[0185] The monitor 100 may also be referred to as sensor electronics 100. The monitor 100 may include a power supply 110, a sensor interface 122, processing electronics 124, and data formatting electronics 128. The monitor 100 may be coupled to the sensor stack 10 using a cable 102 through a connector electrically coupled to the connector block 28 of the connection portion 24. In an alternative embodiment, the cable may be omitted. In this embodiment, the monitor 100 may include a suitable connector for connecting directly to the connection portion 104 of the sensor stack 10. The sensor stack 10 may be modified to position the connector portion 104 in a different location (e.g., above the sensor stack) to facilitate placement of the monitor 100 above the sensor stack.

[0186] In one embodiment, sensor interface 122, processing electronics 124, and data formatting electronics 128 are formed as separate semiconductor chips, however, alternative embodiments may combine the various semiconductor chips into a single or multiple custom semiconductor chips. Sensor interface 122 is connected to cable 102, which is connected to sensor group 10.

[0187] The power source 110 can be a battery. The battery can include three silver oxide 357 cells connected in series. In alternative embodiments, different battery chemistries can be utilized, such as lithium-based chemistries, alkaline batteries, nickel metal hydride, etc., and different numbers of batteries can be used. The monitor 100 provides power to the sensor group 10 via the power source 110 via the cable 102 and the cable connector 104. In one embodiment, the power is a voltage provided to the sensor group 10. In another embodiment, the power is a current provided to the sensor group 10. In one embodiment, the power is a voltage provided to the sensor group 10 at a specific voltage.

[0188] Figure 2A and 2B An implantable sensor and electronic devices for driving the implantable sensor according to one embodiment are shown. Figure 2A A substrate 220 is shown having two sides, a first side 222 containing an electrode configuration and a second side 224 containing electronic circuitry. Figure 2A As can be seen in the figure, the first side 222 of the substrate includes two counter electrode-working electrode pairs 240, 242, 244, 246 on opposite sides of the reference electrode 248. The second side 224 of the substrate includes the electronic circuit system. As shown, the electronic circuit system can be enclosed in a hermetically sealed housing 226, thereby providing a protective housing for the electronic circuit. This allows the sensor substrate 220 to be inserted into a vascular environment or other environment where the electronic circuit system may be subjected to fluids. By sealing the electronic circuit system in the hermetically sealed housing 226, the electronic circuit system can operate without the risk of being shorted by the surrounding fluid. Figure 2A Also shown are pads 228 to which the input and output lines of the electronic circuitry can be connected. The electronic circuitry itself can be manufactured in a variety of ways. According to an embodiment, the electronic circuitry can be manufactured as an integrated circuit using techniques common in the industry.

[0189] Figure 2B A general block diagram of an electronic circuit for sensing the output of a sensor according to one embodiment is shown. At least one pair of sensor electrodes 310 can interface with a data converter 312, the output of which can interface with a counter 314. Counter 314 can be controlled by control logic 316. The output of counter 314 can be connected to a line interface 318. Line interface 318 can be connected to input and output lines 320 and can also be connected to control logic 316. Input and output lines 320 can also be connected to a power rectifier 322.

[0190] The sensor electrode 310 can be used in a variety of sensing applications and can be configured in a variety of ways. For example, the sensor electrode 310 can be used in physiological parameter sensing applications in which some types of biomolecules serve as catalysts. For example, the sensor electrode 310 can be used in glucose and oxygen sensors having glucose oxidase (GOx) that catalyzes reactions with the sensor electrode 310. The sensor electrode 310 and the biomolecule or some other catalyst can be placed in the human body in a vascular or non-vascular environment. For example, the sensor electrode 310 and the biomolecule can be placed in a vein and subjected to blood flow.

[0191] Figure 3A block diagram of a sensor electronics device and a sensor comprising multiple electrodes according to an embodiment of the present invention is shown. A sensor group or system 350 includes a sensor 355 and a sensor electronics device 360. The sensor 355 includes a counter electrode 365, a reference electrode 370, and a working electrode 375. The sensor electronics device 360 ​​includes a power supply 380, a regulator 385, a signal processor 390, a measurement processor 395, and a display / transmission module 397. The power supply 380 provides power (in the form of voltage, current, or voltage containing current) to the regulator 385. The regulator 385 transmits the regulated voltage to the sensor 355. In an embodiment of the present invention, the regulator 385 transmits the voltage to the counter electrode 365 of the sensor 355.

[0192] The sensor 355 generates a sensor signal that indicates the concentration of the physiological property being measured. For example, the sensor signal may indicate a blood glucose reading. In embodiments utilizing a subcutaneous sensor, the sensor signal may represent the level of hydrogen peroxide in the subject. In embodiments utilizing a blood or cranial sensor, the amount of oxygen is measured by the sensor and represented by the sensor signal. In embodiments utilizing an implantable or long-term sensor, the sensor signal may represent the level of oxygen in the subject. The sensor signal is measured at the working electrode 375. In one embodiment, the sensor signal may be a current measured at the working electrode. In one embodiment, the sensor signal may be a voltage measured at the working electrode.

[0193] After the sensor signal is measured at the sensor 355 (e.g., the working electrode), the signal processor 390 receives the sensor signal (e.g., the measured current or voltage). The signal processor 390 processes the sensor signal and generates a processed sensor signal. The measurement processor 395 receives the processed sensor signal and calibrates the processed sensor signal using a reference value. In one embodiment, the reference value is stored in a reference memory and provided to the measurement processor 395. The measurement processor 395 generates the sensor measurement value. The sensor measurement value can be stored in a measurement memory (not shown). The sensor measurement value can be sent to a display / transmission device to be displayed on a display in the housing together with the sensor electronics, or transmitted to an external device.

[0194] Sensor electronics 360 can be a monitor that includes a display to show physiological characteristic readings. Sensor electronics 360 can also be installed in a desktop computer, a pager, a television with communication capabilities, a laptop computer, a server, a network computer, a personal digital assistant (PDA), a portable phone with computer functionality, an infusion pump with a display, a glucose sensor with a display, and / or a combination infusion pump / glucose sensor. Sensor electronics 360 can be housed in a Blackberry, a network device, a home network device, or an appliance connected to a home network.

[0195] Figure 4 An alternative embodiment including a sensor and sensor electronics is shown. A sensor set or sensor system 400 includes sensor electronics 360 and a sensor 355. The sensor includes a counter electrode 365, a reference electrode 370, and a working electrode 375. The sensor electronics 360 includes a microcontroller 410 and a digital-to-analog converter (DAC) 420. The sensor electronics 360 may also include a current-to-frequency converter (I / F converter) 430.

[0196] The microcontroller 410 includes software program code or programmable logic that, when executed, causes the microcontroller 410 to transmit a signal to the DAC 420, wherein the signal represents a voltage level or voltage value to be applied to the sensor 355. The DAC 420 receives the signal and generates a voltage value at the level indicated by the microcontroller 410. In one embodiment, the microcontroller 410 may change the representation of the voltage level in the signal frequently or infrequently. Illustratively, the signal from the microcontroller 410 may instruct the DAC 420 to apply a first voltage value for one second and a second voltage value for two seconds.

[0197] The sensor 355 can receive a voltage level or voltage value. In one embodiment, the counter electrode 365 can receive the output of an operational amplifier having a reference voltage and a voltage value from the DAC 420 as inputs. The application of the voltage level causes the sensor 355 to generate a sensor signal indicative of the concentration of the physiological characteristic being measured. In an embodiment, the microcontroller 410 can measure the sensor signal (e.g., a current value) from the working electrode. Illustratively, the sensor signal measurement circuit 431 can measure the sensor signal. In one embodiment, the sensor signal measurement circuit 431 can include a resistor, and a current can flow through the resistor to measure the value of the sensor signal. In one embodiment, the sensor signal can be a current level signal, and the sensor signal measurement circuit 431 can be a current-to-frequency (I / F) converter 430. The I / F converter 430 can measure the sensor signal based on the current reading, convert the sensor signal into a frequency-based sensor signal, and transmit the frequency-based sensor signal to the microcontroller 410. In some embodiments, the microcontroller 410 can more easily receive a frequency-based sensor signal than a non-frequency-based sensor signal. The microcontroller 410 receives the sensor signal, whether frequency-based or non-frequency-based, and determines a value of a physiological characteristic of the subject, such as a blood glucose level. The microcontroller 410 may include program code that, when executed or run, is capable of receiving the sensor signal and converting the sensor signal into a physiological characteristic value. In one embodiment, the microcontroller 410 may convert the sensor signal into a blood glucose level. In some embodiments, the microcontroller 410 may determine the blood glucose level of the subject using measurement results stored in an internal memory. In some embodiments, the microcontroller 410 may determine the blood glucose level of the subject using measurement results stored in a memory external to the microcontroller 410.

[0198] After the microcontroller 410 determines the physiological characteristic value, the microcontroller 410 may store the measured value of the physiological characteristic value for a number of time periods. For example, the blood glucose value may be sent from the sensor to the microcontroller 410 every second or every five seconds, and the microcontroller may store the sensor measurement value for five or ten minutes of the blood glucose reading. The microcontroller 410 may transmit the measured value of the physiological characteristic value to a display on the sensor electronics 360. For example, the sensor electronics 360 may be a monitor that includes a display that provides a blood glucose reading of the subject. In one embodiment, the microcontroller 410 may transmit the measured value of the physiological characteristic value to an output interface of the microcontroller 410. The output interface of the microcontroller 410 may transmit the measured value of the physiological characteristic value, such as the blood glucose value, to an external device, such as an infusion pump, a combination infusion pump / blood glucose meter, a computer, a personal digital assistant, a pager, a network device, a server, a cell phone, or any computing device.

[0199] Figure 5 An electronic block diagram of sensor electrodes and voltages applied to the sensor electrodes according to an embodiment is shown. Figure 5 In the illustrated embodiment, an operational amplifier 530 or other servo control device can be connected to the sensor electrode 510 via a circuit / electrode interface 538. The operational amplifier 530 uses feedback from the sensor electrode to attempt to maintain a specified voltage between the reference electrode 532 and the working electrode 534 (the DAC may expect the applied voltage to be the specified voltage) by adjusting the voltage at the counter electrode 536. Current can then flow from the counter electrode 536 to the working electrode 534. This current can be measured to determine the electrochemical reaction between the sensor electrode 510 and a biomolecule of the sensor that has been placed near the sensor electrode 510 and acts as a catalyst. Figure 5 The circuit system disclosed in can be used in a long-term or implantable sensor, or can be used in a short-term or subcutaneous sensor.

[0200] In an embodiment of a long-term sensor, where the glucose oxidase (GOx) enzyme is used as the catalyst in the sensor, current can flow from the counter electrode 536 to the working electrode 534 only when oxygen is present near the enzyme and the sensor electrode 510. Illustratively, if the voltage set at the reference electrode 532 is maintained at approximately 0.5 volts, the amount of current flowing from the counter electrode 536 to the working electrode 534 has a fairly linear relationship with the unit slope of the amount of oxygen present in the area surrounding the enzyme and the electrode. Thus, by maintaining the reference electrode 532 at approximately 0.5 volts and utilizing this region of the current-voltage curve to vary the blood oxygen level, the accuracy of determining the amount of oxygen in the blood can be improved. Different embodiments may utilize different sensors with biomolecules other than glucose oxidase and, therefore, may set a voltage other than 0.5 volts at the reference electrode.

[0201] As discussed above, during the initial implantation or insertion of sensor 510, sensor 510 may provide inaccurate readings due to the subject's adjustment to the sensor and electrochemical byproducts produced by the catalyst used in the sensor. Many sensors require a stabilization period in order for sensor 510 to provide accurate readings of the subject's physiological parameters. During the stabilization period, sensor 510 does not provide accurate blood glucose measurements. Sensor users and manufacturers may wish to improve the sensor's stabilization timeframe so that the sensor can be quickly utilized after insertion into a subject's body or into the subcutaneous layer of a subject.

[0202] In previous sensor electrode systems, the stabilization period or time frame was one to three hours. To reduce the stabilization period or time frame and improve the timeliness of the sensor's accuracy, the sensor (or the sensor's electrodes) can be subjected to several pulses rather than applying one pulse and then applying another voltage. Figure 6A A method for applying a pulse during a stabilization time frame to reduce the stabilization time frame according to an embodiment is presented. In this embodiment, a voltage application device applies 600 a first voltage to an electrode for a first time or time period. In one embodiment, the first voltage can be a DC constant voltage. This generates an anodic current. In alternative embodiments, a digital-to-analog converter or another voltage source can supply voltage to the electrode during the first time period. Anodic current means that electrons are driven toward the electrode to which the voltage is applied. In one embodiment, the application device can apply a current instead of a voltage. In embodiments in which a voltage is applied to a sensor, after applying the first voltage to the electrode, a voltage regulator can wait (i.e., not apply the voltage) for a second time, time frame, or time period 605. In other words, the voltage application device waits until the second time period has elapsed. Not applying the voltage generates a cathodic current, which causes the electrode to which the voltage is not applied to gain electrons. The process of applying the first voltage to the electrode for the first time period, followed by not applying the voltage for the second time period, is repeated 610 for several iterations. This can be referred to as an anodic and cathodic cycle. In one embodiment, the total number of iterations of the stabilization method is three, i.e., applying the voltage three times during the first time period, followed by not applying the voltage during the second time period after each application. In an embodiment, the first voltage may be 1.07 volts. In an embodiment, the first voltage may be 0.535 volts. In an embodiment, the first voltage may be 0.7 volts.

[0203] Repeated application of voltage and non-application of voltage causes the sensor (and therefore the electrode) to undergo an anode-cathode cycle. The anode-cathode cycle reduces the electrochemical byproducts produced by the patient's body in response to the insertion of the sensor or the implantation of the sensor. The electrochemical byproducts cause the generation of background current, which results in inaccurate measurements of the subject's physiological parameters. In one embodiment, the electrochemical byproducts can be eliminated. Under other operating conditions, the electrochemical byproducts can be reduced or significantly reduced. A successful stabilization method will allow the anode-cathode cycle to reach equilibrium, significantly reduce electrochemical byproducts, and minimize background current.

[0204] In one embodiment, the first voltage applied to the electrode of the sensor can be a positive voltage. In an alternative embodiment, the first voltage applied can be a negative voltage. In one embodiment, the first voltage can be applied to the working electrode. In another embodiment, the first voltage can be applied to the counter electrode or the reference electrode.

[0205] In the embodiments herein, the duration of the voltage pulse and the duration of not applying the voltage can be equal, for example, three minutes each. In other embodiments, the duration of voltage application or voltage pulse can be different values, for example, the first time and the second time can be different. In one embodiment, the first time period can be five minutes, and the waiting period can be two minutes. In one embodiment, the first time period can be two minutes, and the waiting period (or second time frame) can be five minutes. In other words, the duration of applying the first voltage can be two minutes, and the voltage may not be applied within five minutes. This time frame is only illustrative and should not be restrictive. For example, the first time frame can be two minutes, three minutes, five minutes or ten minutes, and the second time frame can be five minutes, ten minutes, twenty minutes, etc. The time frame (for example, the first time and the second time) can depend on the unique characteristics of different electrodes, sensors and / or the physiological characteristics of the patient.

[0206] In embodiments, more or fewer than three pulses may be used to stabilize the glucose sensor. In other words, the number of iterations may be greater than three or less than three. For example, four voltage pulses (e.g., high voltage followed by no voltage) may be applied to one of the electrodes, or six voltage pulses may be applied to one of the electrodes.

[0207] Illustratively, three consecutive 1.07 volt pulses (followed by a corresponding wait time) may be sufficient for a subcutaneously implanted sensor. In one embodiment, three consecutive 0.7 volt voltage pulses may be utilized. For sensors implanted in blood or cranial fluid, such as long-term or permanent sensors, the three consecutive pulses may have higher or lower voltage values: negative or positive. Additionally, more than three pulses (e.g., five, eight, twelve) may be utilized to generate an anodic-cathodic cycle between the anodic current and the cathodic current in any of the subcutaneous, blood, or cranial fluid sensors.

[0208] Figure 6B A method for stabilizing the sensor is shown. Figure 6B In the illustrated embodiment, a voltage applying device may apply 630 a first voltage to the sensor for a first time period to initiate an anodic cycle at the sensor's electrode. The voltage applying device may be a DC power supply, a digital-to-analog converter, or a voltage regulator. After the first time period has elapsed, a second voltage is applied 635 to the sensor for a second time period to initiate a cathodic cycle at the sensor's electrode. Illustratively, as Figure 6AAs shown in the method of , a different voltage (different from the first voltage) is applied to the sensor during the second time frame instead of applying no voltage. In one embodiment, applying the first voltage for the first time and the second voltage for the second time is repeated for more than 640 iterations. In one embodiment, a stabilization time frame, for example, 10 minutes, 15 minutes, or 20 minutes, is applied to the first voltage for the first time and the second voltage for the second time, instead of multiple iterations. This stabilization time frame is the entire time frame of the stabilization sequence, for example, until the sensor (and electrode) stabilizes. The benefits of this stabilization method are faster run-in of the sensor, less background current (in other words, some background current is suppressed), and better glucose response.

[0209] In one embodiment, the first voltage may be 0.535 volts for five minutes, the second voltage may be 1.070 volts for two minutes, the first voltage of 0.535 volts may be applied for five minutes, the second voltage of 1.070 volts may be applied for two minutes, the first voltage of 0.535 volts may be applied for five minutes, and the second voltage of 1.070 volts may be applied for two minutes. In other words, in this embodiment, there are three iterations of the voltage pulse scheme. The pulse method can be varied, such that the second time frame, for example, the time frame for applying the second voltage, can be extended from two minutes to five minutes, ten minutes, fifteen minutes, or twenty minutes. In addition, after applying three iterations, a nominal operating voltage of 0.535 volts may be applied.

[0210] 1.070 and 0.535 volts are illustrative values. Other voltage values ​​can be selected based on a variety of factors. These factors can include the type of enzyme utilized in the sensor, the membrane utilized in the sensor, the operating period of the sensor, the length of the pulse and / or the magnitude of the pulse. Under certain operating conditions, the first voltage can be in the range of 1.00 to 1.09 volts, and the second voltage can be in the range of 0.510 to 0.565 volts. In other operating embodiments, depending on the voltage sensitivity of the electrode in the sensor, the range comprising the first voltage and the second voltage can have a higher range, for example, 0.3 volt, 0.6 volt, 0.9 volt. Under other operating conditions, the voltage can be in the range of 0.8 volt to 1.34 volts, and another voltage can be in the range of 0.335 to 0.735. Under other operating conditions, the range of the higher voltage can be smaller than the range of the lower voltage. Illustratively, the higher voltage can be in the range of 0.9 to 1.09 volts, and the lower voltage can be in the range of 0.235 to 0.835 volts.

[0211] In embodiments, the first and second voltages may be positive voltages, or in other embodiments, negative voltages. In one embodiment, the first voltage may be positive and the second voltage may be negative, or alternatively, the first voltage may be negative and the second voltage may be positive. The first voltage may be a different voltage level for each iteration. In one embodiment, the first voltage may be a constant DC voltage. In other embodiments, the first voltage may be a ramp voltage, a sinusoidal voltage, a stepped voltage, or other commonly used voltage waveforms. In one embodiment, the second voltage may be a constant DC voltage, a ramp voltage, a sinusoidal voltage, a stepped voltage, or other commonly used voltage waveforms. In one embodiment, the first or second voltage may be an AC signal riding on a DC waveform. In one embodiment, the first voltage may be one type of voltage, such as a ramp voltage, and the second voltage may be a second type of voltage, such as a sinusoidal voltage. In one embodiment, the first voltage (or second voltage) may have a different waveform shape for each iteration. For example, if there are three cycles in the stabilization method, in the first cycle, the first voltage may be a ramp voltage, in the second cycle, the first voltage may be a constant voltage, and in the third cycle, the first voltage may be a sinusoidal voltage.

[0212] In an embodiment, the duration of the first time frame and the duration of the second time frame may have the same value, or alternatively, the duration of the first time frame and the second time frame may have different values. For example, the duration of the first time frame may be two minutes, and the duration of the second time frame may be five minutes, and the number of iterations may be three. As discussed above, the stabilization method may include several iterations. In some embodiments, the duration of each of the first time frames may change, and the duration of each of the second time frames may change during different iterations of the stabilization method. Illustratively, during the first iteration of the anode-cathode cycle, the first time frame may be 2 minutes, and the second time frame may be 5 minutes. During the second iteration, the first time frame may be 1 minute, and the second time frame may be 3 minutes. During the third iteration, the first time frame may be 3 minutes, and the second time frame may be 10 minutes.

[0213] In one embodiment, a first voltage of 0.535 volts is applied to the electrodes in the sensor for two minutes to initiate the anodic cycle, followed by a second voltage of 1.07 volts applied to the electrodes for five minutes to initiate the cathodic cycle. The first voltage of 0.535 volts is then applied again for two minutes to initiate the anodic cycle, and a second voltage of 1.07 volts is applied to the sensor for five minutes. In a third iteration, 0.535 volts is applied for two minutes to initiate the anodic cycle, followed by 1.07 volts for five minutes. Subsequently, during the actual operating time frame of the sensor, e.g., when the sensor is providing readings of a subject's physiological characteristics, the voltage applied to the sensor is 0.535 volts.

[0214] exist Figure 6A and Figure 6B In some embodiments, a shorter duration voltage pulse may be utilized. The shorter duration voltage pulse may be used to apply the first voltage, the second voltage, or both. In one embodiment, the magnitude of the shorter duration voltage pulse of the first voltage is -1.07 volts, and the magnitude of the shorter duration voltage pulse of the second voltage is approximately half of the higher magnitude, for example, -0.535 volts. Alternatively, the magnitude of the shorter duration pulse of the first voltage may be 0.535 volts, and the magnitude of the shorter duration pulse of the second voltage may be 1.07 volts.

[0215] In embodiments utilizing short duration pulses, the voltage may be applied discontinuously throughout the first time period. Alternatively, the voltage applying device may transmit several short duration pulses during the first time period. In other words, several small width or short duration voltage pulses may be applied to the electrodes of the sensor during the first time period. Each small width or short duration pulse may have a width of several milliseconds. Illustratively, this pulse width may be 30 milliseconds, 50 milliseconds, 70 milliseconds, or 200 milliseconds. These values ​​are intended to be illustrative and not restrictive. In one embodiment, for example, Figure 6A In the embodiment shown in , these short duration pulses are applied to the sensor (electrodes) for a first time period, and then no voltage is applied for a second time period.

[0216] In one embodiment, each short duration pulse can have the same duration in the first time period. For example, each short duration voltage pulse can have a time width of 50 milliseconds, and each pulse delay between pulses can be 950 milliseconds. In this example, if the measured time of the first time frame is two minutes, 120 short duration voltage pulses can be applied to the sensor. In one embodiment, each of the short duration voltage pulses can have different durations. In one embodiment, each of the short duration voltage pulses can have the same amplitude value. In one embodiment, each of the short duration voltage pulses can have different amplitude values. By utilizing a short duration voltage pulse rather than continuously applying voltage to the sensor, the same anode and cathode cycle can be performed, and over time, the total energy or charge of the sensor (such as an electrode) decreases. Compared with applying a continuous voltage to the electrode, the power utilized using a short duration voltage pulse is less, because the energy applied to the sensor (and electrode) is less.

[0217] Figure 6C The use of feedback in stabilizing a sensor according to an embodiment is illustrated. The sensor system may include a feedback mechanism for determining whether additional pulses are needed to stabilize the sensor. In one embodiment, a sensor signal generated by an electrode (e.g., a working electrode) may be analyzed to determine whether the sensor signal is stable. A first voltage is applied 630 to the electrode during a first time frame to initiate an anodic cycle. A second voltage is applied 635 to the electrode during a second time frame to initiate a cathodic cycle. In one embodiment, an analysis module may analyze the sensor signal (e.g., the current emitted by the sensor signal, the resistance at a specific point in the sensor, the impedance at a specific node in the sensor) and determine 637 whether a threshold measurement has been reached (e.g., by comparing with a threshold measurement to determine whether the sensor is providing an accurate reading). If the sensor reading is determined to be accurate, indicating that the electrode (and therefore the sensor) has stabilized 642, then the first voltage and / or the second voltage may not be applied additionally. In one embodiment, if stability is not achieved, additional anodic / cathodic cycles may be initiated by applying 630 the first voltage to the electrode during a first time period and then applying 635 the second voltage to the electrode during a second time period.

[0218] In some embodiments, the analysis module may be employed after applying the first voltage and the second voltage to the electrodes of the sensor for three anodic / cathodic cycles. In one embodiment, the analysis module may be employed after applying the first voltage and the second voltage once. Figure 6C shown.

[0219] In one embodiment, an analysis module can be used to measure the voltage emitted after a current has been introduced across one electrode or across two electrodes. The analysis module can monitor the voltage level at the electrodes or receiving level. In one embodiment, if the voltage level is above a certain threshold, then this may mean that the sensor has stabilized. In one embodiment, if the voltage level drops below a certain threshold, then this may mean that the sensor has stabilized and is ready to provide a reading. In one embodiment, a current can be introduced into the electrodes or across the electrode couple. The analysis module can monitor the current level emitted from the electrodes. In this embodiment, the analysis module can monitor the current if it differs by an order of magnitude from the sensor signal current. If the current is above or below a current threshold, then this may indicate that the sensor has stabilized.

[0220] In one embodiment, the analysis module can measure the impedance between two electrodes of the sensor. The analysis module can compare the impedance to a threshold or target impedance value, and if the measured impedance is lower than the target or threshold impedance, then the sensor (and therefore the sensor signal) can be stabilized. In one embodiment, the analysis module can measure the resistance between two electrodes of the sensor. In this embodiment, if the analysis module compares the resistance to a threshold or target resistance value and the measured resistance value is lower than the threshold or target resistance value, then the analysis module can determine that the sensor is stabilized and that the sensor signal can be utilized.

[0221] Figure 7 : The effect of stabilizing the sensor according to one embodiment is shown. Line 705 represents the blood glucose sensor reading of a glucose sensor utilizing a previous single pulse stabilization method. Line 710 represents the blood glucose reading of a glucose sensor with three voltage pulses applied (e.g., 3 voltage pulses of 2 minutes duration, with no voltage applied for 5 minutes after each voltage pulse). The x-axis 715 represents the amount of time. Points 720, 725, 730, and 735 represent measured glucose readings, which are obtained using a finger stick and then input into the blood glucose meter. As shown in the graph, the previous single pulse stabilization method took approximately 1 hour and 30 minutes to stabilize to the desired glucose reading, for example, 100 units. In contrast, the three-pulse stabilization method took only approximately 15 minutes to stabilize the glucose sensor and significantly improved the stabilization time frame.

[0222] Figure 8AA block diagram of sensor electronics and a sensor including a voltage generating device is shown according to one embodiment. Voltage generating or applying device 810 includes electronics, logic, or circuitry for generating voltage pulses. Sensor electronics 360 may also include input devices 820 for receiving reference values ​​and other useful data. In one embodiment, sensor electronics may include a measurement memory 830 for storing sensor measurements. In this embodiment, power supply 380 may provide power to the sensor electronics. Power supply 380 may provide power to regulator 385, which supplies a regulated voltage to voltage generating or applying device 810. Connector terminals 811 are shown, which, in the illustrated embodiment, couple or connect sensor 355 to sensor electronics 360.

[0223] exist Figure 8A In the illustrated embodiment, a voltage generating or applying device 810 supplies a voltage, such as a first voltage or a second voltage, to an input terminal of an operational amplifier 840. The voltage generating or applying device 810 can also supply a voltage to the working electrode 375 of the sensor 355. Another input terminal of the operational amplifier 840 is coupled to the reference electrode 370 of the sensor. Applying a voltage from the voltage generating or applying device 810 to the operational amplifier 840 drives the voltage measured at the counter electrode 365 to be close to or equal to the voltage applied at the working electrode 375. In an embodiment, the voltage generating or applying device 810 can be used to apply a desired voltage between the counter electrode and the working electrode. This can be done by applying a fixed voltage directly to the counter electrode.

[0224] exist Figure 6A and Figure 6B In one embodiment shown, the voltage generating device 810 generates a first voltage to be applied to the sensor during a first time frame. The voltage generating device 810 transmits this first voltage to the operational amplifier 840, which drives the voltage at the counter electrode 365 of the sensor 355 to the first voltage. In one embodiment, the voltage generating device 810 can also transmit the first voltage directly to the counter electrode 365 of the sensor 355. Figure 6A In the illustrated embodiment, the voltage generating device 810 then does not transmit the first voltage to the sensor 355 for a second time frame. In other words, the voltage generating device 810 is turned off or disconnected. The voltage generating device 810 can be programmed to continue cycling between applying the first voltage and not applying the voltage for a number of iterations or stabilization time frames, for example, for twenty minutes. Figure 8BA voltage generating device for implementing this embodiment is shown. Voltage regulator 385 transmits the regulated voltage to voltage generating device 810. Control circuit 860 controls the closing and opening of switch 850. If switch 850 is closed, voltage is applied. If switch 850 is open, no voltage is applied. Timer 865 provides a signal to control circuit 860 to instruct control circuit 860 to turn switch 850 on and off. Control circuit 860 includes logic that can instruct the circuit to close and open switch 850 a number of times (to match the necessary iterations). In one embodiment, timer 865 can also transmit a stabilization signal to identify the completion of the stabilization sequence, that is, the expiration of the stabilization time frame.

[0225] In one embodiment, the voltage generating device generates a first voltage within a first time frame and generates a second voltage within a second time frame. Figure 8C A voltage generating device is shown for generating two voltage values ​​to implement this embodiment. In this embodiment, a two-position switch 870 is utilized. Illustratively, if the control circuit 860 is instructed by timer 865 to open or close a first switch position 871, the voltage generating device 810 generates a first voltage within a first time frame. After applying the first voltage within the first time frame, the timer sends a signal to the control circuit 860 indicating that the first time frame has elapsed, and the control circuit 860 directs the switch 870 to move to a second position 872. When the switch 870 is in the second position 872, the regulated voltage is directed to a voltage step-down or buck converter 880 to reduce the regulated voltage to a smaller value. The smaller value is then supplied to the operational amplifier 840 within a second time frame. After the timer 865 signals the control circuit 860 that the second time period has elapsed, the control circuit 860 returns the switch 870 to the first position. This process continues until the desired number of iterations has been completed or the stabilization time frame has elapsed. In one embodiment, the sensor transmits the sensor signal 350 to the signal processor 390 after the sensor stabilization time frame has elapsed.

[0226] Figure 8D A voltage applying device 810 is shown for performing a more complex application of voltage to the sensor. The voltage applying device 810 may include a control device 860, a switch 890, a sinusoidal voltage generating device 891, a ramp voltage generating device 892, and a constant voltage generating device 893. In other embodiments, the voltage application may generate an AC wave on top of a DC signal or other various voltage pulse waveforms. Figure 8DIn the illustrated embodiment, the control device 860 can cause a switch to move to one of three voltage generating systems 891 (sinusoidal), 892 (ramp), or 893 (constant DC). This causes each of the voltage generating systems to generate a recognized voltage waveform. Under certain operating conditions, for example, when a sinusoidal pulse is to be applied for three pulses, the control device 860 can cause the switch 890 to connect the voltage from the voltage regulator 385 to the sinusoidal voltage generator 891 so that the voltage applying device 810 generates a sinusoidal voltage. Under other operating conditions, for example, when a ramp voltage is to be applied to the sensor as the first voltage of the first of the three pulses, a sinusoidal voltage is to be applied to the sensor as the first voltage of the second of the three pulses, and a constant DC voltage is to be applied to the sensor as the first voltage of the third of the three pulses, the control device 860 can cause the switch 890 to move, during the first time frame of the anode / cathode cycle, between connecting the voltage from the voltage generating or applying device 810 to the ramp voltage generating system 892, then to the sinusoidal voltage generating system 891, and then to the constant DC voltage generating system 893. In this embodiment, the control device 860 may also direct or control the switches to connect certain voltage generating subsystems to the voltage from the regulator 385 during the second time frame, such as during application of the second voltage.

[0227] Figure 9A FIG. 4 shows a sensor electronics device including a microcontroller for generating voltage pulses according to one embodiment. An advanced sensor electronics device may include a microcontroller 410 (see FIG. Figure 4 ), digital-to-analog converter (DAC) 420, operational amplifier 840 and sensor signal measurement circuit 431. In one embodiment, the sensor signal measurement circuit may be a current-to-frequency (I / F) converter 430. Figure 9A In the embodiment shown, software or programmable logic in the microcontroller 410 provides instructions for transmitting a signal to the DAC 420, which in turn instructs the DAC 420 to output a specific voltage to the operational amplifier 840. Figure 9AAs shown in line 911 in , the microcontroller 410 can also be instructed to output a specific voltage to the working electrode 375. As discussed above, applying a specific voltage to the operational amplifier 840 and the working electrode 375 can drive the voltage measured at the counter electrode to a specific voltage magnitude. In other words, the microcontroller 410 outputs a signal indicating the voltage or voltage waveform to be applied to the sensor 355 (e.g., the operational amplifier 840 connected to the sensor 355). In an alternative embodiment, a fixed voltage can be set by applying a voltage from the DAC 420 directly between the reference electrode and the working electrode 375. In the case where the voltage difference is equal to the fixed voltage applied between the reference electrode and the working electrode, a similar result can be obtained by applying a voltage to each of the electrodes. In addition, a fixed voltage can be set by applying a voltage between the reference electrode and the counter electrode. Under certain operating conditions, the microcontroller 410 can generate a pulse of a specific magnitude, which the DAC 420 interprets as indicating that a voltage of a specific magnitude is to be applied to the sensor. After the first time frame, the microcontroller 410 (via program or programmable logic) outputs a second signal that instructs the DAC 420 to output no voltage (for Figure 6A sensor electronics 360) or outputting a second voltage (for use in accordance with Figure 6B ). After the second time frame has elapsed, the microcontroller 410 then repeats the cycle of sending a signal indicating that the first voltage is to be applied (within the first time frame), and then sending a signal indicating that no voltage is to be applied or that a second voltage is to be applied (within the second time frame).

[0228] Under other operating conditions, the microcontroller 410 can generate a signal to the DAC 420 that instructs the DAC to output a ramp voltage. Under other operating conditions, the microcontroller 410 can generate a signal to the DAC 420 that instructs the DAC 420 to output a voltage that simulates a sinusoidal voltage. These signals can be incorporated into any of the pulse methods discussed above in the previous paragraphs or previously in the application. In one embodiment, the microcontroller 410 can generate instructions and / or pulse sequences that the DAC 420 receives and interprets as meaning that a certain pulse sequence is to be applied. For example, the microcontroller 410 can transmit a series of instructions (via signals and / or pulses) that instruct the DAC 420 to generate a constant voltage for a first iteration of a first time frame, a ramp voltage for a first iteration of a second time frame, a sinusoidal voltage for a second iteration of the first time frame, and a square wave with two values ​​for a second iteration of the second time frame.

[0229] Microcontroller 410 may include programmable logic or a program for continuing this cycle for a stable time frame or several iterations. Specifically, microcontroller 410 may include counting logic for identifying when the first time frame or the second time frame has elapsed. Additionally, microcontroller 410 may include counting logic for identifying when a stable time frame has elapsed. After any of the aforementioned time frames has elapsed, the counting logic may instruct the microcontroller to send a new signal or to stop transmitting a signal to DAC 420.

[0230] The use of microcontroller 410 allows for the application of multiple voltage magnitudes over multiple durations and in multiple sequences. In one embodiment, microcontroller 410 may include control logic or programming for instructing digital-to-analog converter 420 to transmit a voltage pulse having a magnitude of approximately 1.0 volt for a first time period of 1 minute, followed by a voltage pulse having a magnitude of approximately 0.5 volt for a second time period of 4 minutes, and repeating this cycle for four iterations. In one embodiment, microcontroller 420 may be programmed to transmit a signal causing DAC 420 to apply a voltage pulse of the same magnitude for each first voltage in each iteration. In one embodiment, microcontroller 410 may be programmed to transmit a signal causing DAC 420 to apply a voltage pulse of a different magnitude for each first voltage in each iteration. In this embodiment, microcontroller 410 may also be programmed to transmit a signal causing DAC 420 to apply a voltage pulse of a different magnitude for each second voltage in each iteration. Illustratively, the microcontroller 410 may be programmed to transmit signals to cause the DAC 420 to apply a first voltage pulse of approximately 1.0 volts in a first iteration, a second voltage pulse of approximately 0.5 volts in a first iteration, a first voltage of 0.7 volts and a second voltage of 0.4 volts in a second iteration, and a first voltage of 1.2 volts and a second voltage of 0.8 volts in a third iteration.

[0231] The microcontroller 410 can also be programmed to instruct the DAC 420 to provide several short-duration voltage pulses within a first time frame. In this embodiment, rather than applying a single voltage throughout the entire first time frame (e.g., two minutes), multiple pulses of shorter duration can be applied to the sensor. In this embodiment, the microcontroller 410 can also be programmed to instruct the DAC 420 to provide several short-duration voltage pulses to the sensor within a second time frame. Illustratively, the microcontroller 410 can send a signal to cause the DAC to apply several short-duration voltage pulses, where the short duration is 50 milliseconds or 100 milliseconds. Between these short-duration pulses, the DAC can apply no voltage, or the DAC can apply a minimum voltage. The microcontroller can cause the DAC 420 to apply short-duration voltage pulses within a first time frame (e.g., two minutes). Then, the microcontroller 410 can send a signal to cause the DAC to apply no voltage or to apply short-duration voltage pulses to the sensor at a second voltage level within a second time frame. For example, the second voltage can be 0.75 volts, and the second time frame can be five minutes. In one embodiment, microcontroller 410 may send a signal to DAC 420 causing DAC 420 to apply a different voltage magnitude to each of the short-duration pulses within a first time frame and / or a second time frame. In one embodiment, microcontroller 410 may send a signal to DAC 420 causing DAC 420 to apply a pattern of voltage magnitudes to the short-duration voltage pulses within the first time frame or the second time frame. For example, the microcontroller may transmit a signal or pulse instructing DAC 420 to apply thirty 20-millisecond pulses to the sensor during a first time frame. Each of the thirty 20-millisecond pulses may have the same magnitude or may have a different magnitude. In this embodiment, microcontroller 410 may instruct DAC 420 to apply the short-duration pulses during a second time frame, or may instruct DAC 420 to apply a different voltage waveform during the second time frame.

[0232] Although the disclosure in Figures 6 to 8 discloses the application of a voltage, a current may also be applied to the sensor to initiate the stabilization process. Figure 6BIn the illustrated embodiment, a first current may be applied during a first time frame to initiate an anodic or cathodic response, and a second current may be applied during a second time frame to initiate an opposite anodic or cathodic response. The application of the first and second currents may continue for several iterations or may continue within a stable time frame. In one embodiment, the first current may be applied during the first time frame, and the first voltage may be applied during the second time frame. In other words, one of the anodic or cathodic cycles may be triggered by applying a current to the sensor, and one of the anodic or cathodic cycles may be triggered by applying a voltage to the sensor. As described above, the applied current may be a constant current, a ramp current, a step pulse current, or a sinusoidal current. Under certain operating conditions, the current may be applied as a series of short duration pulses during the first time frame.

[0233] Figure 9B A sensor and sensor electronics utilizing an analysis module for feedback during a stabilization period is shown according to one embodiment of the present invention. Figure 9B An analysis module 950 is incorporated into the sensor electronics 360. The analysis module 950 utilizes feedback from the sensor to determine whether the sensor is stable or unstable. In one embodiment, the microcontroller 410 may include instructions or commands for controlling the DAC 420 such that the DAC 420 applies a voltage or current to a portion of the sensor 355. Figure 9B It is shown that a voltage or current can be applied between the reference electrode 370 and the working electrode 375. However, the voltage or current can be applied between the electrodes or directly to one of the electrodes, and the present invention should not be limited to Figure 9B The application of voltage or current is shown by dashed line 955. Analysis module 950 can measure the voltage, current, resistance, or impedance in sensor 355. Figure 9BMeasurements are shown to be taken at the working electrode 375, but this should not limit the present invention, as other embodiments may measure voltage, current, resistance, or impedance between the electrodes of the sensor or directly at the reference electrode 370 or the counter electrode 365. Analysis module 950 may receive the measured voltage, current, resistance, or impedance and may compare the measured value to a stored value (e.g., a threshold value). Dashed line 956 indicates that analysis module 950 reads or performs a voltage, current, resistance, or impedance measurement. Under certain operating conditions, if the measured voltage, current, resistance, or impedance is above the threshold value, the sensor is stable and the sensor signal provides an accurate reading of the patient's physiological condition. Under other operating conditions, if the measured voltage, current, resistance, or impedance is below the threshold value, the sensor is stable. Under other operating conditions, analysis module 950 may verify that the measured voltage, current, resistance, or impedance is stable within a specific time frame, such as one or two minutes. This may indicate that sensor 355 is stable and the sensor signal is transmitting an accurate measurement of the subject's physiological parameter, such as blood glucose level. After the analysis module 950 has determined that the sensor has stabilized and the sensor signal is providing accurate measurements, the analysis module 950 can transmit a signal (e.g., a sensor stabilization signal) to the microcontroller 410 indicating that the sensor has stabilized and the microcontroller 410 can begin using or receiving the sensor signal from the sensor 355. This is represented by the dashed line 957.

[0234] Figure 10 A block diagram of a sensor system including hydration electronics, according to one embodiment, is shown. The sensor system includes a connector 1010, a sensor 1012, and monitor or sensor electronics 1025. Sensor 1012 includes electrodes 1020 and a connection portion 1024. In one embodiment, sensor 1012 can be connected to sensor electronics 1025 via connector 1010 and a cable. In other embodiments, sensor 1012 can be directly connected to sensor electronics 1025. In other embodiments, sensor 1012 and sensor electronics 1025 can be incorporated into the same physical device. Monitor or sensor electronics 1025 can include a power supply 1030, a regulator 1035, a signal processor 1040, a measurement processor 1045, and a processor 1050. Monitor or sensor electronics 1025 can also include hydration detection circuitry 1060. Hydration detection circuitry 1060 interfaces with sensor 1012 to determine whether electrodes 1020 of sensor 1012 are adequately hydrated. If the electrodes 1020 are not adequately hydrated, then the electrodes 1020 do not provide accurate glucose readings, so it is important to know when the electrodes 1020 are adequately hydrated. Once the electrodes 1020 are adequately hydrated, accurate glucose readings can be obtained.

[0235] exist Figure 10 In the embodiment shown, the hydration detection circuit 1060 may include a delay or timer module 1065 and a connection detection module 1070. In embodiments utilizing a short-term sensor or subcutaneous sensor, after the sensor 1012 is inserted into the subcutaneous tissue, the sensor electronics or monitor 1025 is connected to the sensor 1012. The connection detection module 1070 recognizes that the sensor electronics 1025 has been connected to the sensor 1012 and sends a signal to the timer module 1065. This Figure 10 1085. The hydration timer module 1065 may be configured to detect the hydration timer 1012 and to detect the hydration timer 1012. The hydration timer module 1065 may be configured to detect the hydration timer 1012 and to detect the hydration timer 1012. The hydration timer module 1065 may be configured to detect the hydration timer 1012 and to detect the hydration timer 1012. The hydration timer module 1065 may be configured to detect the hydration timer 1012 and to detect the hydration timer 1012. The hydration timer module 1065 may be configured to detect the hydration timer 1012 and to detect the hydration timer 1012. The hydration timer module 1065 may be configured to detect the hydration timer 1012 and to detect the hydration timer 1012.

[0236] In this embodiment, the processor 1050 can receive a hydration signal and begin utilizing the sensor signal (e.g., sensor measurement) only after receiving the hydration signal. In another embodiment, the hydration detection circuit 1060 can be coupled between the sensor (sensor electrode 1020) and the signal processor 1040. In this embodiment, the hydration detection circuit 1060 can prevent the sensor signal from being sent to the signal processor 1040 until the timer module 1065 has notified the hydration detection circuit 1060 that a set hydration time has elapsed. This is illustrated by the dashed lines labeled with reference numerals 1080 and 1081. Illustratively, the timer module 1065 can transmit a connection signal to a switch (or transistor) to turn on the switch and allow the sensor signal to proceed to the signal processor 1040. In an alternative embodiment, the timer module 1065 can transmit a connection signal to turn on (or close) switch 1088 in the hydration detection circuit 1060 to allow the voltage from the regulator 1035 to be applied to the sensor 1012 after the hydration time has elapsed. In other words, in this embodiment, the voltage from regulator 1035 is not applied to sensor 1012 until after the hydration time has elapsed.

[0237] Figure 11An embodiment is shown that includes a mechanical switch to assist in determining hydration time. In one embodiment, a single housing can contain sensor assembly 1120 and sensor electronics 1125. In one embodiment, sensor assembly 1120 can be in one housing, and sensor electronics 1125 can be in a separate housing, but sensor assembly 1120 and sensor electronics 1125 can be connected together. In this embodiment, connection detection mechanism 1160 can be a mechanical switch. The mechanical switch can detect that sensor 1120 is physically connected to sensor electronics 1125. In one embodiment, timer circuit 1135 can also be activated when mechanical switch 1160 detects that sensor 1120 is connected to sensor electronics 1125. In other words, the mechanical switch can close and transmit a signal to timer circuit 1135. Once the hydration time has elapsed, timer circuit 1135 transmits a signal to switch 1140, allowing regulator 1035 to apply voltage to sensor 1120. In other words, voltage is not applied until the hydration time has elapsed. In one embodiment, once the hydration time has elapsed, current can replace the voltage applied to the sensor. In an alternative embodiment, when the mechanical switch 1160 recognizes that the sensor 1120 has been physically connected to the sensor electronics 1125, power may be initially applied to the sensor 1120. This power delivery to the sensor 1120 causes a sensor signal to be output from the working electrode in the sensor 1120. The sensor signal may be measured and transmitted to the processor 1175. The processor 1175 may include a counter input. Under certain operating conditions, after a set hydration time has elapsed since the sensor signal was input into the processor 1175, the processor 1175 may begin processing the sensor signal as an accurate measurement of glucose in the subject. In other words, the processor 1175 may have received the sensor signal from the potentiostat circuit 1170 for a certain amount of time, but may not process the signal until it receives an indication from the processor's counter input that the hydration time has elapsed. In an embodiment, the potentiostat circuit 1170 may include a current-to-frequency converter 1180. In this embodiment, the current-to-frequency converter 1180 may receive the sensor signal as a current value and convert it into a frequency value, which is easier for the processor 1175 to process.

[0238] In one embodiment, the mechanical switch 1160 can also notify the processor 1175 when the sensor 1120 has been disconnected from the sensor electronics 1125. Figure 111176 in FIG. This may cause the processor 1170 to power down or reduce the power to several components, chips, and / or circuits of the sensor electronics 1125. If the sensor 1120 is not connected, the battery or power supply may be depleted while the components or circuits of the sensor electronics 1125 are powered on. Therefore, if the mechanical switch 1160 detects that the sensor 1120 has been disconnected from the sensor electronics 1125, the mechanical switch may indicate this to the processor 1175, and the processor 1175 may power down or reduce the power to one or more of the electronic circuits, chips, or components of the sensor electronics 1125.

[0239] Figure 12 An electrical method of detecting hydration according to one embodiment is shown. In one embodiment, an electrical detection mechanism for detecting the connection of the sensor can be utilized. In this embodiment, the hydration detection electronics 1250 can include an AC source 1255 and a detection circuit 1260. The hydration detection electronics 1250 can be located in the sensor electronics 1225. The sensor 1220 can include a counter electrode 1221, a reference electrode 1222, and a working electrode 1223. Figure 12 As shown, the AC source 1255 is coupled to a voltage setting device 1275, a reference electrode 1222, and a detection circuit 1260. In this embodiment, the AC signal from the AC source is applied to the reference electrode connection, as shown in FIG. Figure 12 . In one embodiment, the AC signal is coupled to the sensor 1220 through an impedance, and if the sensor 1220 is connected to the sensor electronics 1225, the coupled signal is significantly attenuated. Therefore, there is a low level AC signal at the input of the detection circuit 1260. This may also be referred to as a highly attenuated signal or a signal with a high attenuation level. Under certain operating conditions, the voltage level of the AC signal may be Vapplied*(Ccoupling) / (Ccoupling+Csensor). If the detection circuit 1260 detects the presence of a high level AC signal (a low attenuation signal) at the input terminal of the detection circuit 1260, then no interrupt is sent to the microcontroller 410 because the sensor 1220 is not fully hydrated or activated. For example, the input of the detection circuit 1260 may be a comparator. If the sensor 1220 is fully hydrated (or wetted), then an effective capacitance is formed between the counter electrode and the reference electrode (e.g., Figure 12 The capacitor C r-c ), and an effective capacitance is formed between the reference electrode and the working electrode (e.g., Figure 12 The capacitor C w-rIn other words, effective capacitance relates to the capacitance formed between the two nodes and does not represent the placement of an actual capacitor in the circuit between the two electrodes. In one embodiment, the AC signal from the AC source 1255 is passed through the capacitor C r-c and C w-r The AC signal from the AC source 1255 is sufficiently attenuated and the detection circuit 1260 detects the presence of a low level or highly attenuated AC signal from the AC source 1255 at the input terminals of the detection circuit 1260. This embodiment is important because it reduces the number of connections to the sensor by utilizing the existing connections between the sensor 1120 and the sensor electronics 1125. In other words, Figure 11 The mechanical switch disclosed in requires a switch and associated connections between the sensor 1120 and the sensor electronics 1125. Removing the mechanical switch is advantageous because the size of the sensor 1120 continues to decrease, and removing components facilitates this size reduction. In alternative embodiments, the AC signal can be applied to a different electrode (e.g., a counter electrode or a working electrode), and the present invention can operate in a similar manner.

[0240] As described above, after detection circuit 1260 has detected the presence of a low-level AC signal at its input terminal, detection circuit 1260 may later detect the presence of a high-level AC signal with low attenuation at its input terminal. This indicates that sensor 1220 has been disconnected from sensor electronics 1225 or that the sensor is not operating properly. If the sensor has been disconnected from sensor electronics 1225, an AC source with minimal or low attenuation may be connected to the input of detection circuit 1260. As described above, detection circuit 1260 may generate an interrupt to the microcontroller. This interrupt may be received by the microcontroller, which may reduce or remove power to one or more components or circuits in sensor electronics 1225. This may be referred to as a second interrupt. Again, this helps reduce the power consumption of sensor electronics 1225, particularly when sensor 1220 is not connected to sensor electronics 1225.

[0241] exist Figure 12 In the illustrated alternative embodiment, an AC signal may be applied to the reference electrode 1222, as shown at 1291, and the impedance measurement device 1277 may measure the impedance of a region in the sensor 1220. Illustratively, the region may be the region between the reference electrode and the working electrode, as shown at 1291. Figure 121292 in FIG. Under certain operating conditions, if the measured impedance has decreased below an impedance threshold or other set criteria, the impedance measurement device 1277 can transmit a signal to the detection circuit 1260. This indicates that the sensor is sufficiently hydrated. Under other operating conditions, once the impedance is above the impedance threshold, the impedance measurement device 1277 can transmit a signal to the detection circuit 1260. The detection circuit 1260 then transmits an interrupt to the microcontroller 410. In another embodiment, the impedance measurement device 1277 can transmit an interrupt or signal directly to the microcontroller.

[0242] In an alternative embodiment, the AC source 1255 can be replaced by a DC source. If a DC source is utilized, a resistance measuring element can be utilized in place of the impedance measuring element 1277. In embodiments utilizing a resistance measuring element, once the resistance drops below a resistance threshold or set standard, the resistance measuring element can transmit a signal to the detection circuit 1260 (represented by dashed line 1293) or directly to the microcontroller indicating that the sensor is sufficiently hydrated and power can be applied to the sensor.

[0243] exist Figure 12 In the embodiment shown, if the detection circuit 1260 detects a low level or highly attenuated AC signal from the AC source, an interrupt is generated to the microcontroller 410. This interrupt indicates that the sensor is sufficiently hydrated. In this embodiment, in response to the interrupt, the microcontroller 410 generates a signal that is transmitted to the digital-to-analog converter 420 to instruct or cause the digital-to-analog converter 420 to apply a voltage or current to the sensor 1220. Figure 6A 、 Figure 6B or Figure 6CAny of the various sequences of pulses or short-duration pulses described in the associated text describing pulse application can be applied to sensor 1220. Illustratively, the voltage from DAC 420 can be applied to operational amplifier 1275, the output of which is applied to counter electrode 1221 of sensor 1220. This causes the sensor, for example, working electrode 1223 of the sensor, to generate a sensor signal. Because the sensor is fully hydrated, as identified by the interrupt, the sensor signal generated at working electrode 1223 accurately measures glucose. The sensor signal is measured by sensor signal measurement device 431, which transmits the sensor signal to microcontroller 410, where a parameter of the subject's physiological condition is measured. The generation of the interrupt indicates that the sensor is fully hydrated and that sensor 1220 is now providing accurate glucose measurements. In this embodiment, the hydration period may depend on the type and / or manufacturer of the sensor, as well as on the sensor's response to insertion or implantation into the subject. Illustratively, one sensor 1220 may have a hydration time of five minutes, and one sensor 1220 may have a hydration time of one minute, two minutes, three minutes, six minutes, or 20 minutes. Again, any amount of time may be an acceptable hydration time for a sensor, but a lesser amount of time is preferred.

[0244] If the sensor 1220 is connected but not fully hydrated or wet, the effective capacitance C r-c and C w-r The AC signal from the AC source 1255 may not be attenuated. The electrodes in the sensor 1120 are dry before insertion, and because the electrodes are dry, there is no good electrical path (or conductive path) between the two electrodes. Therefore, the detection circuit 1260 can still detect a high-level AC signal or a low-attenuation AC signal and will not generate an interrupt. Once the sensor is inserted, the electrodes are immersed in the conductive body fluid. This creates a leakage path with a lower DC resistance. In addition, a boundary layer capacitor is formed at the metal / fluid interface. In other words, a considerable capacitance is formed between the metal / fluid interface, and this large capacitance looks like two capacitors connected in series between the electrodes of the sensor. This can be called effective capacitance. In fact, the conductivity of the electrolyte above the electrodes is being measured. In some embodiments, the glucose limiting membrane (GLM) also shows an impedance that hinders electrical efficiency. Unhydrated GLM produces high impedance, while high-water content GLM produces low impedance. Accurate sensor measurements require low impedance.

[0245] Figure 13AA method of hydrating a sensor according to an embodiment of the present invention is shown. In one embodiment, the sensor can be physically connected 1310 to the sensor electronics. After connection, in one embodiment, a timer or counter can be started to count the hydration time 1320. After the hydration time has elapsed, a signal can be transmitted 1330 to a subsystem in the sensor electronics to begin applying voltage to the sensor. As discussed above, in one embodiment, a microcontroller can receive a signal and instruct a DAC to apply voltage to the sensor, or in another embodiment, a switch can receive a signal allowing a regulator to apply voltage to the sensor. The hydration time can be five minutes, two minutes, ten minutes, and can vary depending on the subject and the type of sensor.

[0246] In an alternative embodiment, after connecting the sensor to the sensor electronics, an AC signal (e.g., a low voltage AC signal) can be applied 1340 to the sensor, such as a reference electrode of the sensor. The AC signal can be applied because the connection of the sensor to the sensor electronics allows the AC signal to be applied to the sensor. After the AC signal is applied, an effective capacitance is formed 1350 between the electrode to which the applied voltage is applied and the other two electrodes in the sensor. The detection circuit determines 1360 what level of AC signal is present at the input of the detection circuit. If a low level AC signal (or a highly attenuated AC signal) is present at the input of the detection circuit, an interrupt is generated 1370 by the detection circuit and sent to the microcontroller because the effective capacitance forms a good electrical conduit between the electrodes and the attenuation of the generated AC signal.

[0247] The microcontroller receives the interrupt generated by the detection circuit and transmits a signal 1380 to the digital-to-analog converter, which instructs or causes the digital-to-analog converter to apply a voltage to an electrode of the sensor, such as the counter electrode. Applying the voltage to the sensor electrode causes the sensor to generate or produce a sensor signal 1390. The sensor signal measurement device 431 measures the generated sensor signal and transmits the sensor signal to the microcontroller. The microcontroller receives 1395 the sensor signal from the sensor signal measurement device coupled to the working electrode and processes the sensor signal to extract a measurement of a physiological characteristic of the subject or patient.

[0248] Figure 13B An additional method for verifying the hydration of a sensor according to an embodiment of the present invention is shown. Figure 13BIn the embodiment shown, the sensor is physically connected 1310 to the sensor electronics. In one embodiment, an AC signal is applied 1341 to an electrode in the sensor, such as a reference electrode. Alternatively, in one embodiment, a DC signal is applied 1341 to an electrode in the sensor. If an AC signal is applied, the impedance measuring element measures 1351 the impedance at a point within the sensor. Alternatively, if a DC signal is applied, the resistance measuring element measures 1351 the resistance at a point within the sensor. If the resistance or impedance is below a resistance threshold or impedance threshold, respectively (or other set criteria), the impedance (or resistance) measuring element transmits 1361 a signal (or allows the signal to be transmitted) to the detection circuit, and the detection circuit transmits an interrupt to the microcontroller identifying that the sensor is hydrated. Reference numerals 1380, 1390, and 1395 are shown in FIG. Figure 13A and Figure 13B are the same in , because they represent the same action.

[0249] The microcontroller receives the interrupt and transmits 1380 a signal to a digital-to-analog converter to apply a voltage to the sensor. In an alternative embodiment, the digital-to-analog converter can apply a current to the sensor, as discussed above. The sensor, such as a working electrode, generates 1390 a sensor signal that represents a physiological parameter of the patient. The microcontroller receives 1395 the sensor signal from a sensor signal measuring device that measures the sensor signal at an electrode in the sensor (e.g., a working electrode). The microcontroller processes the sensor signal to extract a measurement of a physiological characteristic of the subject or patient, such as the patient's blood glucose level.

[0250] Figure 14A and 14B A method for combining sensor hydration with sensor stabilization according to an embodiment of the present invention is shown. Figure 14AIn the illustrated embodiment, a sensor is connected 1405 to sensor electronics. An AC signal is applied 1410 to the sensor's electrodes. A detection circuit determines 1420 what level of AC signal is present at the detection circuit's input. If the detection circuit determines that a low-level AC signal is present at the input (indicating a high level of attenuation of the AC signal), an interrupt is sent 1430 to the microcontroller. Once the interrupt is sent to the microcontroller, the microcontroller knows to begin or initiate 1440 a stabilization sequence, i.e., to apply several voltage pulses to the sensor's electrodes, as described above. For example, the microcontroller can cause a digital-to-analog converter to apply three voltage pulses (with a magnitude of +0.535 volts) to the sensor, where each of the three voltage pulses is followed by a period of three voltage pulses (with a magnitude of 1.07 volts). This can be referred to as a stabilization sequence for the transmission voltage. The microcontroller can initiate this by executing a software program in read-only memory (ROM) or random access memory. After the stabilization sequence completes, the sensor can generate 1450 a sensor signal, which is measured and transmitted to the microcontroller.

[0251] In one embodiment, the detection circuit may determine 1432 that a high-level AC signal continues to be present at the input of the detection circuit (e.g., the input of a comparator) even after a hydration time threshold has passed. For example, the hydration time threshold may be 10 minutes. After 10 minutes have passed, the detection circuit may still detect the presence of a high-level AC signal. At this point, the detection circuit may transmit 1434 a hydration assist signal to the microcontroller. If the microcontroller receives the hydration assist signal, the microcontroller may transmit 1436 a signal to cause the DAC to apply a voltage pulse or series of voltage pulses to assist in hydrating the sensor. In one embodiment, the microcontroller may transmit a signal to cause the DAC to apply a portion of a stabilization sequence or other voltage pulses to assist in hydrating the sensor. In this embodiment, the application of the voltage pulses may cause a low-level AC signal (or a highly attenuated signal) to be detected 1438 at the detection circuit. At this point, the detection circuit may transmit an interrupt, as disclosed in step 1430, and the microcontroller may begin a stabilization sequence.

[0252] Figure 14BA second embodiment of a combination of a hydration method and a stabilization method is shown, wherein feedback is utilized during the stabilization process. The sensor is connected 1405 to the sensor electronics. An AC signal (or DC signal) is applied 1411 to the sensor. In an embodiment, the AC signal (or DC signal) is applied to an electrode of the sensor, such as a reference electrode. An impedance measuring device (or resistance measuring device) measures 1416 the impedance (or resistance) within a specified area of ​​the sensor. In one embodiment, the impedance (or resistance) between the reference electrode and the working electrode can be measured. The measured impedance (or resistance) can be compared 1421 to an impedance or resistance value to see if the impedance (or resistance) in the sensor is low enough, indicating that the sensor is hydrated. If the impedance (or resistance) is lower than the impedance (or resistance) value or other set criteria (which can be a threshold), an interrupt is transmitted 1431 to the microcontroller. After receiving the interrupt, the microcontroller transmits 1440 a signal to the DAC, which instructs the DAC to apply a stable sequence of voltage (or current) to the sensor. After the stabilization sequence is applied to the sensor, a sensor signal is generated in the sensor (e.g., at the working electrode), measured by the sensor signal measurement device, transmitted by the sensor signal measurement device, and received by the microcontroller 1450. Because the sensor is hydrated and the stabilization sequence of voltages has been applied to the sensor, the sensor signal accurately measures the physiological parameter (i.e., blood glucose).

[0253] Figure 14CA third embodiment combining a stabilization method and a hydration method is shown. In this embodiment, a sensor is connected 1500 to sensor electronics. After the sensor is physically connected to the sensor electronics, an AC signal (or DC signal) is applied 1510 to an electrode (e.g., a reference electrode) of the sensor. At or approximately the same time, a microcontroller transmits a signal to cause a DAC to apply 1520 a stabilization voltage sequence to the sensor. In alternative embodiments, a stabilization current sequence may be applied to the sensor instead of a stabilization voltage sequence. A detection circuit determines 1530 what level of AC signal (or DC signal) is present at the input terminals of the detection circuit. If a low-level AC signal (or DC signal) is present at the input terminals of the detection circuit, indicating a highly attenuated AC signal (or DC signal), an interrupt is transmitted 1540 to the microcontroller. Because the microcontroller has already begun the stabilization sequence, the microcontroller receives the interrupt and sets 1550 a first indicator indicating that the sensor is fully hydrated. After the stabilization sequence is complete, the microcontroller sets 1555 a second indicator indicating that the stabilization sequence is complete. Application of the stabilization sequence voltage causes a sensor, such as a working electrode, to generate a sensor signal 1560, which is measured by sensor signal measurement circuitry and sent to a microcontroller. If the second indicator of stabilization sequence completion is set and the first indicator of hydration completion is set, the microcontroller can utilize the sensor signal 1570. If one or both of the indicators are not set, the microcontroller may not utilize the sensor signal because the sensor signal may not represent an accurate measurement of the subject's physiological measurement.

[0254] Typically, the hydration and stabilization process described above can be used as part of a larger continuous glucose monitoring (CGM) approach. The current state of the art in continuous glucose monitoring is largely adjunctive, meaning that the readings provided by a CGM device (comprising, for example, an implantable or subcutaneous sensor) cannot be used to make clinical decisions without a reference value. The reference value, in turn, must be obtained from a finger stick using, for example, a BG meter. The reason for the need for a reference value is that the amount of information available from the sensor / sensing assembly is limited. Specifically, the only pieces of information currently provided for processing by the sensing assembly are the raw sensor value (i.e., the sensor current or Isig) and the counter voltage, which is the voltage between the counter electrode and a reference electrode (see, e.g., Figure 5 ). Therefore, during analysis, if the raw sensor signal appears abnormal (e.g., if the signal is decreasing), the only way to distinguish between a sensor malfunction and a physiological change in the user / patient (i.e., a change in glucose level in the body) is by obtaining a reference glucose value via a fingerstick. As is well known, a reference fingerstick is also used to calibrate the sensor.

[0255] Some embodiments described herein are directed to advancing and improving continuous glucose monitoring, resulting in more autonomous systems and related devices and methods, in which the need for reference fingersticks can be minimized or eliminated, and clinical decisions can be made reliably based on information derived from sensor signals alone. From a sensor design perspective, according to embodiments herein, this autonomy can be achieved through electrode redundancy, sensor diagnostics, and Isig and / or sensor glucose (SG) fusion.

[0256] As will be explored further below, redundancy can be achieved by using multiple working electrodes (e.g., plus a counter electrode and a reference electrode) to generate multiple signals indicative of a patient's blood glucose (BG) level. Furthermore, the multiple signals can be used to assess the relative health of the (working) electrodes, the overall reliability of the sensor, and the frequency (if any) required to calibrate the reference value.

[0257] Sensor diagnostics involves the use of additional (diagnostic) information that can provide real-time insights into the health of the sensor. In this regard, it has been discovered that electrochemical impedance spectroscopy (EIS) provides this additional information in the form of sensor impedance and impedance-related parameters at different frequencies. Furthermore, advantageously, it has been further discovered that for certain frequency ranges, the impedance and / or impedance-related data is substantially glucose-independent. This glucose independence enables the use of multiple EIS-based markers or indicators to generate robust, highly reliable sensor glucose values ​​(via fusion methods), and also to assess the condition, health, age, and efficiency of individual electrodes and the overall sensor substantially independently of glucose-related Isig.

[0258] For example, analysis of the glucose-independent impedance data using values ​​such as 1 kHz real impedance, 1 kHz imaginary impedance, and Nyquist slope provides information about the efficiency of the sensor in terms of how quickly it hydrates and becomes ready for data acquisition (described in greater detail below). Furthermore, the glucose-independent impedance data provides information about potential occlusions that may exist on the sensor membrane surface, which may temporarily prevent glucose from entering the sensor and thus cause a signal dip (using values ​​such as 1 kHz real impedance). Furthermore, the glucose-independent impedance data provides information about loss of sensor sensitivity during extended wear, possibly due to localized hypoxia at the insertion site, using values ​​such as phase angle and / or imaginary impedance at frequencies of 1 kHz and higher.

[0259] In the context of electrode redundancy and EIS, as well as other contexts, as described in further detail below, a fusion algorithm can be used to extract the diagnostic information provided by the EIS for each redundant electrode and independently assess the reliability of each electrode. A weight can then be added to each independent signal as a measure of reliability, and a single fused signal can be calculated that can be used to generate the sensor glucose value seen by the patient / subject.

[0260] As can be seen from the above, the combined use of redundancy, sensor diagnostics using EIS, and EIS-based fusion algorithms allows for an overall CGM system that is more reliable than currently available systems. Redundancy is advantageous in at least two respects. First, redundancy eliminates the risk of single point failure by providing multiple signals. Second, providing multiple (working) electrodes where a single electrode may be sufficient allows the output of the redundant electrode to be used as a check on the primary electrode, thereby reducing and potentially eliminating the need for frequent calibration. Additionally, EIS diagnostics can automatically check the health of each electrode without the need for a reference glucose value (fingerstick), thereby reducing the number of reference values ​​required. However, the use of EIS technology and EIS diagnostic methods is not limited to redundant systems, i.e., those systems having more than one working electrode. Instead, as discussed below in conjunction with the embodiments herein, EIS can be advantageously used in conjunction with single electrode and / or multi-electrode sensors.

[0261] EIS or AC impedance method studies the response of a system to the application of a periodic small amplitude AC signal. Figure 15A This is schematically shown in Figure 1, where E is the applied potential, I is the current, and impedance (Z) is defined as ΔE / ΔI. However, although impedance itself can be mathematically simply defined as ΔE / ΔI, to date, there has been no commercial success in applying EIS technology to continuous glucose monitoring. This is partly because glucose sensors are very complex systems, and to date, no mathematical model has been developed that can fully explain the complexity of the EIS output of glucose sensors.

[0262] Figure 15B A simplified circuit model that has been used to describe electrochemical impedance spectroscopy is shown in Figure 1. In this diagram, IHP stands for inner Helmholtz plane, OHP stands for outer Helmholtz plane, CE is the counter electrode, WE is the working electrode, and C d is the double layer capacitance, R p is the polarization resistance, Z w is the Weber impedance, and R s is the solution resistance. The last four components are the double layer capacitance (C d ), Warburg impedance (Z w ), polarization resistance (R p ) and solution resistance (R sEach of these components can play an important role in sensor performance and can be measured individually by applying a low-frequency or high-frequency AC working potential. For example, Weber impedance is closely related to the diffusion impedance of the electrochemical system—primarily the low-frequency impedance—and is therefore present in all diffusion-limited electrochemical sensors. Therefore, by correlating one or more of these components with one or more components and / or layers of a glucose sensor, EIS technology can be used as a sensor diagnostic tool.

[0263] As is well known, impedance can be defined in terms of its magnitude and phase, where the magnitude (|Z|) is the ratio of the voltage difference amplitude to the current amplitude, and the phase (θ) is the phase shift of the current ahead of the voltage. When a circuit is driven solely by direct current (DC), the impedance is the same as the resistance, meaning that resistance is a special case of impedance with zero phase angle. However, as a complex number, impedance can also be represented by its real and imaginary parts. In this regard, the following equations can be used to derive the real and imaginary impedances from the impedance magnitude and phase:

[0264] Real impedance (ω) = magnitude (ω) x cos(phase (ω) / 180 x π)

[0265] Imaginary impedance (ω) = magnitude (ω) x sin(phase (ω) / 180 x π)

[0266] Where ω represents the input frequency of the measured magnitude (in ohms) and phase (in degrees). The relationship between impedance, on the one hand, and current and voltage, on the other hand, including how the former can be calculated based on measurements of the latter, will be discussed more fully below in conjunction with sensor electronics including an application-specific integrated circuit (ASIC) developed for use in the embodiments herein.

[0267] continue Figure 15B For the circuit model shown, the total system impedance can be simplified to:

[0268]

[0269] where Z w (ω) is the Weber impedance, ω is the angular velocity, j is an imaginary unit (instead of the traditional "i" to avoid confusion with current), and C d 、R p and R s are the double layer capacitance, polarization resistance, and solution resistance (as defined previously). The Weber impedance can be calculated as

[0270]

[0271]

[0272]

[0273] where D is the diffusivity, L is the sensor film thickness, C is the peroxide concentration, and m:1 / 2 corresponds to a 45° Nyquist slope.

[0274] A Nyquist plot is a graphical representation where the real part of the impedance (Real Z) is plotted against the imaginary part of the impedance (Img Z) over the frequency spectrum. Figure 16A A general example of a Nyquist plot is shown, where the X value is the real part of the impedance and the Y value is the imaginary part of the impedance. The phase angle is the angle between the impedance point (X, Y) and the X axis, defining a vector of magnitude |Z|.

[0275] Figure 16A The Nyquist plot of an electrochemical system is generated by applying an AC voltage plus a DC voltage (DC bias) between the working electrode and the counter electrode at a selected frequency (i.e., a frequency sweep) from 0.1 Hz to 1000 MHz. Starting from the right, the frequency increases from 0.1 Hz. For each frequency, the real impedance and the imaginary impedance can be calculated and plotted. As shown, a typical Nyquist plot for an electrochemical system may look like a semicircle connected to a straight line at an inflection point, where the semicircle and the straight line represent the plotted impedance. In certain embodiments, the impedance at the inflection point is particularly interesting because it is the easiest to identify in the Nyquist plot and the intercept can be defined. Typically, the inflection point is close to the X-axis, and the X value of the inflection point is approximately the sum of the polarization resistance and the solution resistance (R p +R s ).

[0276] refer to Figure 16B , the Nyquist plot can generally be described in terms of a lower frequency region 1610 and a higher frequency region 1620, where the labels "higher frequency" and "lower frequency" are used in a relative sense and are not intended to be limiting. Thus, for example, the low frequency region 1610 may illustratively include data points obtained for a frequency range between about 0.1 Hz and about 100 Hz (or higher), and the high frequency region 1620 may illustratively include data points obtained for a frequency range between about 1 kHz (or lower) and about 8 kHz (or higher). In the low frequency region 1610, the Nyquist slope represents the gradient of the linear fit 1630 of the low frequency data points in the Nyquist plot. As shown, in the high frequency region 1620, the value of the imaginary impedance is smallest and can become negligible. Thus, the intercept 1600 is essentially the value of the real impedance at higher frequencies (e.g., in this case, approximately in the range of 1 kHz to 8 kHz). Figure 16B In the example, the intercept 1600 is at about 25 kilo-ohms.

[0277] Figure 16C and Figure 16DIn these figures, GLM is the glucose limiting membrane of the sensor, AP is the adhesion promoter, HSA is human serum albumin, GOX is the glucose oxidase (layer), E is the glucose oxidase (layer), and GLM is the glucose limiting membrane of the sensor. dc is the DC potential, E ac is the AC potential, and C′ 过氧化物 is the peroxide concentration during AC application. Figure 16C As shown in , if the sensor diffusion length, which varies with the AC potential frequency, molecular diffusion rate, and membrane thickness, is smaller than the membrane (GOX) length, the system gives a relatively linear response with a constant phase angle (i.e., infinite). Conversely, if the diffusion length is equal to the membrane (GOX) length, the system response will become finite, resulting in a semicircular Nyquist plot, as shown in Figure 16D The latter is generally applicable to low-frequency EIS, where non-Faraday processes are negligible.

[0278] When performing an EIS analysis, AC voltages and DC biases of various frequencies can be applied, for example, between a working electrode and a reference electrode. In this respect, EIS is an improvement over previous methods, which may limit their application to simple DC currents or AC voltages of a single frequency. Although, in general, EIS can be performed at frequencies in the μHz to MHz range, in embodiments, a narrower frequency range (e.g., between about 0.1 Hz and about 8 kHz) may be sufficient. Thus, in some embodiments, an AC potential can be applied that falls within a frequency range between about 0.1 Hz and about 8 kHz, with a programmable amplitude of up to at least 100 mV, and preferably about 50 mV.

[0279] Within the above-mentioned frequency range, the capacitive properties of the sensor are checked using relatively high frequencies, i.e., typically frequencies between about 1 kHz and about 8 kHz. Depending on the thickness and permeability of the membrane, at relatively high frequencies, a typical range of impedance can be, for example, between about 500 ohms and 25 kilo-ohms, and a typical range of phase can be, for example, between 0 degrees and -40 degrees. On the other hand, the resistive properties of the sensor are checked using relatively low frequencies, i.e., typically frequencies between about 0.1 Hz and about 100 Hz. Here, depending on the electrode design and the degree of metallization, a typical operating range for the output real impedance can be, for example, between about 50 kilo-ohms and 300 kilo-ohms, and a typical range for the phase can be between about -50 degrees and about -90 degrees. The above-mentioned illustrative ranges are, for example, between Figure 16E and Figure 16F The Bode plot is shown in FIG.

[0280] As previously mentioned, the phrases "high frequency" and "low frequency" are meant to be used relative to each other, not in an absolute sense, and they, as well as the typical impedance and phase ranges mentioned above, are intended to be illustrative, not restrictive. However, the basic principle remains the same: the capacitive and resistive characteristics of the sensor can be examined by analyzing the impedance data over the entire frequency spectrum, where, in general, lower frequencies provide information about the more resistive components (e.g., electrodes, etc.), while higher frequencies provide information about the capacitive components (e.g., membranes). However, the actual frequency range in each case depends on the overall design, including, for example, the type of electrodes, the surface area of ​​the electrodes, the thickness of the membrane, the permeability of the membrane, etc. See also for a general correspondence between high frequency circuit components and the sensor membrane, and between low frequency circuit components and the Faraday process including, for example, electrodes. Figure 15B .

[0281] EIS can be used in sensor systems where the sensor contains a single working electrode, as well as in sensor systems where the sensor contains multiple (redundant) working electrodes. In one embodiment, EIS provides valuable information about the life (or aging) of the sensor. Specifically, at different frequencies, the magnitude and phase angle of the impedance change. As in Figure 17 As seen in Figure 2, the sensor impedance, specifically the sum of Rp and Rs, reflects the sensor lifetime as well as the sensor operating conditions. Figure 17 As can be seen in the different graphs in Figure 1, new sensors generally have higher impedance than used sensors. In this way, by considering the value of X, which is the sum of Rp and Rs, a threshold value can be used to determine when the life of the sensor exceeds the specified operating life of the sensor. It should be noted that although for Figure 17 21 and discussed below, the value of the real impedance at the inflection point (i.e., Rp+Rs) is used to determine the aging, state, stability, and hydration of the sensor, but alternative embodiments may use other EIS-based parameters in addition to or instead of the real impedance, such as imaginary impedance, phase angle, Nyquist slope, etc.

[0282] Figure 17An example of a Nyquist plot over the life of a sensor is shown. The points indicated by the arrows are the corresponding inflection points for each scan in the spectrum. For example, before initialization (at time t=0), Rs+Rp is above 8.5 kiloohms, and after initialization (at time t=0.5 hr), the value of Rs+Rp drops below 8 kiloohms. Over the next six days, Rs+Rp continues to decrease, so that at the end of the specified sensor life, Rs+Rp drops below 6.5 kiloohms. Based on such examples, a threshold can be set to specify when the Rs+Rp value indicates the end of the specified operating life of the sensor. Thus, EIS technology allows for closing the loophole that allows reuse of sensors after a specified operating time. In other words, if a patient attempts to reuse a sensor after it has reached its specified operating time by disconnecting and then reconnecting it, EIS will measure an abnormally low impedance, enabling the system to reject the sensor and prompt the patient to replace it with a new one.

[0283] Additionally, EIS can detect sensor failure by detecting when the sensor's impedance drops below a low-impedance threshold level, indicating that the sensor may be excessively worn and unable to operate properly. The system can then terminate the sensor before its designated operating life. As will be discussed in more detail below, sensor impedance can also be used to detect other sensor failure modes. For example, when a sensor enters a low-current state (i.e., sensor failure) for various reasons, the sensor impedance may also increase to exceed a certain high-impedance threshold. The system can also terminate the sensor before its designated operating life if the impedance becomes abnormally high during sensor operation, for example, due to protein or peptide fouling, macrophage attachment, or any other factor.

[0284] Figure 18 It demonstrates how EIS techniques can be applied during sensor stabilization and when detecting sensor age according to certain embodiments. Figure 18 The logic begins at 1800 after the aforementioned hydration and sensor initialization procedures have completed. In other words, the sensor is deemed fully hydrated, and the first initialization procedure has been applied to initialize the sensor. The initialization procedure preferably takes the form of a voltage pulse, as previously described in the detailed description. However, in alternative embodiments, different waveforms may be used for the initialization procedure. For example, a sine wave may be used instead of a pulse to accelerate sensor wetting or conditioning. Furthermore, it may be necessary to have a portion of the waveform be greater than the sensor's normal operating voltage, which is 0.535 volts.

[0285] At box 1810, the EIS program is applied and the impedance is compared with the first high threshold and the first low threshold. Examples of the first high threshold and the first low threshold are 7 kilo-ohms and 8.5 kilo-ohms respectively, but they can be set to higher or lower as needed. If the impedance (e.g., Rp+Rs) is higher than the first high threshold, the sensor undergoes another initialization program (e.g., applying one or more other pulses) at box 1820. Ideally, the number of total initialization programs applied to initialize the sensor will be optimized to limit the impact on the sensor battery life and the total amount of time required to stabilize the sensor. Therefore, by applying EIS, less initialization can be performed first, and the initialization number can be incrementally added to provide only the correct number of initializations to prepare for use of the sensor. Similarly, in an alternative embodiment, EIS can be applied to a hydration program to minimize the number of initializations required for the auxiliary hydration process, as shown in Figures 13 to 14.

[0286] On the other hand, if the impedance (e.g., Rp+Rs) is below the first low threshold, the sensor will be determined to be faulty and will terminate immediately at box 1860. The user will be prompted to replace the sensor and begin the hydration process again. If the impedance is within the high and low thresholds, the sensor will begin normal operation at box 1830. The logic then proceeds to box 1840 where another EIS is performed to check the age of the sensor. The first time the logic reaches box 1840, the microcontroller will perform an EIS to measure the life of the sensor, thereby compensating for the user being able to plug and unplug the same sensor. In future iterations of the EIS procedure, when the logic returns to box 1840, the microprocessor will perform an EIS at fixed intervals during the specified life of the sensor. In a preferred embodiment, the fixed interval is set to every 2 hours, however, longer or shorter time periods can easily be used.

[0287] At block 1850, the impedance is compared to a second set of high and low thresholds. Examples of such second high and low thresholds are 5.5 kilo-ohms and 8.5 kilo-ohms, respectively, but these values ​​can be set higher or lower as needed. As long as the impedance value remains within the second high and low thresholds, the logic proceeds to block 1830, where the sensor operates normally until a specified sensor lifespan, such as 5 days, is reached. Of course, as described with respect to block 1840, EIS will be performed at regularly scheduled intervals throughout the specified sensor lifespan. However, if, after performing the EIS, it is determined at block 1850 that the impedance has fallen below the second lower threshold or risen above the second higher threshold, the sensor is terminated at block 1860. In other alternative embodiments, a secondary check can be implemented for erroneous sensor readings. For example, if the EIS indicates that the impedance is outside the range of the second high and low thresholds, the logic can perform a second EIS to confirm that the second set of thresholds has indeed not been met (and to confirm that the first EIS was performed correctly) before terminating the sensor at block 1860.

[0288] Figure 19 Building upon the above description, and detailing a possible schedule for executing diagnostic EIS procedures according to a preferred embodiment of the present invention, each diagnostic EIS procedure is optional, and any diagnostic EIS procedure may be omitted or any combination of one or more diagnostic EIS procedures may be performed as desired. Figure 19 The timeline begins at sensor insertion at point 1900. After sensor insertion, the sensor undergoes a hydration period 1910. This hydration period is important because an insufficiently hydrated sensor may provide the user with inaccurate readings, as previously described. During this hydration period 1910, a first optional diagnostic EIS procedure at point 1920 is scheduled to ensure that the sensor is adequately hydrated. The first diagnostic EIS procedure 1920 measures the sensor impedance value to determine whether the sensor is adequately hydrated. If the first diagnostic EIS procedure 1920 determines that the impedance is within the set high and low thresholds, indicating adequate hydration, the sensor controller will allow the sensor to be activated at point 1930. Conversely, if the first diagnostic EIS procedure 1920 determines that the impedance exceeds the set high and low thresholds, indicating insufficient hydration, the sensor hydration period 1910 may be extended. After the extended hydration, once a certain capacitance is reached between the sensor's electrodes, indicating that the sensor is adequately hydrated, activation at point 1930 may be performed.

[0289] After sensor activation at point 1930 but before sensor initialization begins at point 1950, a second optional diagnostic EIS procedure 1940 is scheduled. The second diagnostic EIS procedure 1940, as scheduled here, can detect whether the sensor is being reused before initialization begins at 1950. The test to determine whether the sensor is being reused is performed at Figure 18 However, unlike the description of Figure 18 The difference from the previous description is that in the case where the aging test is performed after the initialization is completed, the aging test is performed after Figure 19 is shown as being executed before initialization. It is important to realize that Figure 19 The timeline of the EIS procedure described in

[1940] can be rearranged without affecting the overall teaching of the application, and the order of some steps can be interchanged. As previously described, the second diagnostic EIS procedure 1940 detects a reused sensor by determining the sensor's impedance value and then comparing it to set high and low thresholds. If the impedance falls outside the set thresholds, indicating that the sensor is being reused, the sensor may be rejected and the user may be prompted to replace it with a new sensor. This avoids complications caused by reusing an old sensor. Conversely, if the impedance is within the set thresholds, sensor initialization 1950 can begin with confidence that a new sensor is being used.

[0290] After initialization begins at point 1950, a third optional diagnostic EIS procedure 1960 is arranged. The third diagnostic EIS procedure 1960 tests the impedance value of the sensor to determine whether the sensor is fully initialized. The third diagnostic EIS procedure 1960 should be executed with the minimum amount of time required to fully initialize any sensor. When executed at this time, the sensor life is maximized by limiting the time that the fully initialized sensor is not used, and over-initialization is avoided by confirming the complete initialization of the sensor before performing excessive initialization. Preventing over-initialization is important because over-initialization can cause current suppression, which can cause inaccurate readings. However, insufficient initialization is also a problem. Therefore, if the third diagnostic EIS procedure 1960 indicates that the sensor is not initialized enough, the optional initialization at point 1970 can be executed to fully initialize the sensor. Insufficient initialization is disadvantageous because it will generate excessive current that is unrelated to the actual glucose concentration. Due to the dangers of insufficient initialization and over-initialization, the third diagnostic EIS procedure plays an important role in ensuring that the sensor operates properly when in use.

[0291] In addition, an optional periodic diagnostic EIS procedure 1980 can be scheduled at a time after the sensor is fully initialized. The EIS procedure 1980 can be scheduled at any set interval. As will be discussed in more detail below, the EIS procedure 1980 can also be triggered by other sensor signals, such as abnormal current or abnormal counter electrode voltage. In addition, fewer or more EIS procedures 1980 can be scheduled as needed. In a preferred embodiment, the EIS procedure used during the hydration process, sensor life check, initialization process, or periodic diagnostic test is the same procedure. In an alternative embodiment, the EIS procedure can be shortened or extended for various EIS procedures (i.e., fewer or more ranges of checking frequency), depending on the need to focus on a specific impedance range. The periodic diagnostic EIS procedure 1980 monitors the impedance value to ensure that the sensor continues to operate at an optimal level.

[0292] If the sensor current drops due to a contaminant species, sensor age, or a combination of contaminant species and sensor age, the sensor may not be operating at its optimum level. A sensor that has aged beyond a certain time is no longer useful, but a sensor that is hampered by a contaminant species may be repairable. Contaminant species can cause the sensor current to drop by reducing the surface area of ​​the electrode or the diffusion path for the analyte and reaction byproducts. These contaminant species are electrically charged and gradually accumulate on the electrode or membrane surface at a certain voltage. Previously, contaminant species would destroy the effectiveness of the sensor. Now, if the periodic diagnostic EIS procedure 1980 detects an impedance value that indicates the presence of a contaminant species, remedial action can be taken. About Figure 20 Describes when to take remedial action. Therefore, periodic diagnostic EIS procedures 1980 become very useful because they can trigger sensor remedial actions, which can restore sensor current to normal levels and extend sensor life. Figure 21A and Figure 21B Two possible embodiments of sensor remediation are described in the description of .

[0293] Additionally, any scheduled diagnostic EIS procedure 1980 may be paused or rescheduled when it is determined that certain events are imminent. Such events may include any situation where the patient is required to check a sensor reading, including, for example, when the patient is measuring his or her BG level using a test strip meter to calibrate a sensor, when the patient is warned of a calibration error and the need to measure his or her BG level again using a test strip meter, or when a hyperglycemia or hypoglycemia warning has been issued but not acknowledged.

[0294] Figure 20 A method of combining a diagnostic EIS procedure with sensor remediation measures according to an embodiment of the present invention is shown. The block 2000 diagnostic procedure may be as follows Figure 19The method logic begins by executing any periodic diagnostic EIS procedure 1980 as described in detail in block 2000. The method logic begins when a diagnostic EIS procedure is executed at block 2000 to detect the impedance value of the sensor. As noted, in certain embodiments, the EIS procedure applies a combination of a DC bias voltage and a variable frequency AC voltage, wherein the impedance detected by executing the EIS procedure is mapped onto a Nyquist plot, and the inflection point in the Nyquist plot approximates the sum of the polarization resistance and the solution resistance (i.e., the real impedance value). After the diagnostic EIS procedure at block 2000 detects the impedance value of the sensor, the logic moves to block 2010.

[0295] At block 2010, the impedance value is compared to the set high and low thresholds to determine if the impedance value is normal. If the impedance is within the set limits of the high and low thresholds at block 2010, normal sensor operation is resumed at block 2020, and Figure 20 The logic will end until another diagnostic EIS procedure is scheduled. Conversely, if the impedance is determined to be abnormal (i.e., outside the set limits of the high and low thresholds) at block 2010, then remedial action is triggered at block 2030. Examples of acceptable high and low thresholds over the life of the sensor are 5.5 kilo-ohms and 8.5 kilo-ohms, respectively, but they can be set higher or lower as needed.

[0296] The remedial measures of box 2030 are performed to remove any contaminant species that may have caused the abnormal impedance value. In a preferred embodiment, the remedial measures are performed by applying a reverse current or reverse voltage between the working electrode and the reference electrode. The details of the remedial measures will be described in more detail with respect to Figure 21. After the remedial measures are performed at box 2030, the impedance value is tested again by a diagnostic EIS procedure at box 2040. The remedial measures are then determined to be successful at box 2050 when the impedance value from the diagnostic EIS procedure of box 2040 is compared to a set high threshold or low threshold. As in box 2010, if the impedance is within the set threshold, then this is considered normal, and if the impedance exceeds the set threshold, then this is considered abnormal.

[0297] If it is determined at block 2050 that the sensor's impedance value has returned to normal, normal sensor operation at block 2020 will proceed. If the impedance remains abnormal, indicating that sensor aging is the cause of the impedance anomaly or that the remedial measures have failed to successfully remove the contaminant species, the sensor is terminated at block 2060. In an alternative embodiment, instead of immediately terminating the sensor, the sensor may generate a sensor message that first requests the user to wait and then, after a set period of time, initiates further remedial measures. This alternative step may be combined with separate logic to determine whether the impedance value is increasingly within the bounds of upper and lower thresholds after the initial remedial measures have been taken. For example, if no change in the sensor's impedance value is detected, the sensor may decide to terminate. However, if the sensor's impedance value is increasingly within the preset limits but remains outside the limits after the initial remedial measures, additional remedial measures may be taken. In yet another alternative embodiment, the sensor may generate a message requesting the user to calibrate the sensor by performing a fingerstick meter measurement to further confirm whether the sensor is indeed faulty. All of the above embodiments are intended to prevent users from using faulty sensors that produce inaccurate readings.

[0298] Figure 21A An example of the aforementioned sensor remediation procedure is shown. In this example, blockage caused by contaminant species is removed by reversing the voltage applied to the sensor between the working and reference electrodes. The reversed DC voltage lifts the charged contaminant species from the electrode or membrane surface, thereby clearing the diffusion path. Through the cleared path, the sensor current returns to normal levels, and the sensor can provide accurate readings. Thus, this remediation procedure saves users the time and money associated with replacing a previously valid sensor.

[0299] Figure 21B An alternative embodiment to the previously mentioned sensor remediation measures is shown. In this embodiment, a reversed DC voltage applied between the working electrode and the reference electrode is coupled with an AC voltage. By adding the AC voltage, certain tightly adsorbed species or species on the surface layer can be removed because the AC voltage can extend the force of the AC voltage further from the electrode and penetrate all layers of the sensor. The AC voltage can take any number of different waveforms. Some examples of waveforms that can be used include square waves, triangle waves, sine waves, or pulses. As with the previous embodiments, once the contaminant species are cleaned, the sensor can return to normal operation with improved sensor life and accuracy.

[0300] While the above examples primarily illustrate the use of real impedance data in sensor diagnostics, embodiments also relate to the use of other EIS-based, substantially analyte-independent parameters (besides real impedance) in sensor diagnostic procedures. For example, as previously described, analysis of (substantially) glucose-independent impedance data (e.g., the values ​​of real impedance at 1 kHz and imaginary impedance at 1 kHz) and the Nyquist slope provides information about the effectiveness of the sensor in rapidly hydrating and preparing the sensor for data acquisition. Furthermore, (substantially) glucose-independent impedance data, such as the value of real impedance at 1 kHz, provides information about potential occlusions that may be present on the sensor membrane surface, which may temporarily prevent glucose from entering the sensor and thus cause a signal dip.

[0301] Additionally, (substantially) glucose-independent impedance data, such as the values ​​of high-frequency phase angle and / or imaginary impedance at 1 kHz and higher, provide information about sensor sensitivity loss during extended wear, which may be due to localized hypoxia at the insertion site. In this regard, the basic mechanism of sensitivity loss due to hypoxia can be described as follows: when there is localized hypoxia, the sensor output (i.e., Isig and SG) will depend on oxygen rather than glucose, and, as a result, the sensor loses sensitivity to glucose. Other markers including 0.1 Hz real impedance, counter electrode voltage (Vcntr), and EIS-induced spikes in Isig can also be used to detect hypoxia-induced sensitivity loss. Furthermore, in redundant sensor systems, the relative differences in 1 kHz real impedance, 1 kHz imaginary impedance, and 0.1 Hz real impedance between two or more working electrodes can be used to detect sensitivity loss due to biofouling.

[0302] According to an embodiment, EIS-based sensor diagnostics require consideration and analysis of EIS data related to at least three primary factors, i.e., one or more of the following potential sensor failure modes: (1) signal onset; (2) signal dip; and (3) sensitivity loss. Importantly, it was discovered herein that most impedance-related parameters used in such diagnostic analyses and procedures can be studied at a frequency or frequency range where the parameters are substantially independent of the analyte, which allows the sensor diagnostic procedures to be implemented independently of the analyte levels in the patient. Thus, while EIS-based sensor diagnostics can be triggered by, for example, large fluctuations in Isig associated with the analyte, the impedance-related parameters used in such sensor diagnostic procedures are themselves substantially independent of the analyte levels. As will be discussed in more detail below, it was also discovered that in most cases where glucose can be seen to have an effect on the magnitude (or other characteristics) of an EIS-based parameter, such an effect is typically sufficiently small, e.g., there is at least an order of magnitude difference between the EIS-based measurement and the effect of glucose on the measurement, so that glucose can be screened out of the measurement, e.g., by software in the IC.

[0303] By definition, "start-up" refers to the integrity of the sensor signal during the first few hours after insertion (e.g., t=0 to 6 hours). For example, in current devices, the signal within the first 2 hours after insertion is considered unreliable, and therefore, the sensor glucose value is not visible to the patient / user. In cases where the sensor takes a longer amount of time to hydrate, the sensor signal is low for a few hours after insertion. By using EIS, additional impedance information can be obtained immediately after inserting the sensor (by running the EIS program). In this regard, the total impedance equation can be used to explain the principle of low startup detection using 1kHz real impedance. At relatively high frequencies (in this case, 1kHz and above), the imaginary impedance is very small (confirmed by in vivo data), so the total impedance is reduced to:

[0304]

[0305] As the wetting of the sensor is gradually completed, the double layer capacitance (C d ) increases. As a result, the total impedance will decrease because, as shown in the above equation, the total impedance is related to C d This is inversely proportional to Figure 16B This is shown as an intercept 1600 on the real impedance axis. Importantly, the 1 kHz imaginary impedance can also be used for the same purpose since it also contains and is inversely proportional to the capacitive component.

[0306] Another marker for low start detection is the Nyquist slope, which depends only on the relatively low frequency impedance, which in turn corresponds to the Weber impedance component of the total impedance (see, for example, Figure 15B ). Figure 22 A Nyquist plot for a properly operating sensor is shown, with arrow A indicating the passage of time starting at t=0, i.e., the time the sensor is worn. Thus, EIS is performed at a relatively low frequency immediately after sensor insertion (time t=0), generating real and imaginary impedance data, which are plotted using a first linear fit 2200 having a first (Nyquist) slope. At a time interval after t=0, a second (lower) frequency sweep is run, generating a second linear fit 2210 having a second (Nyquist) slope greater than the first, and so on. As the sensor becomes more hydrated, the Nyquist slope increases and the intercept decreases, as reflected by lines 2200, 2210, etc. becoming steeper and moving closer to the Y-axis. Combined with low-start detection, clinical data indicates that the Nyquist slope typically increases significantly after sensor insertion and initialization, then stabilizes to a certain level. One explanation for this is that as the sensor becomes increasingly wetted, species diffusivity and concentration undergo significant changes, which are reflected in the Weber impedance.

[0307] exist Figure 23A In the example, Isig 2230 of the first working electrode WE1 starts out lower than expected (about 10 nA) and takes some time to catch up to Isig 2240 of the second working electrode WE2. Therefore, in this particular example, WE1 is shown as having a low start. The EIS data reflects this low start in two ways. First, as Figure 23A As shown, the real impedance of WE1 at 1kHz (2235) is much higher than the real impedance of WE2 at 1kHz (2245). Secondly, when compared with the Nyquist slope of WE2 ( Figure 23C ) compared to the Nyquist slope of WE1 ( Figure 23B ) starts out lower, has a larger intercept 2237, and takes more time to stabilize. As will be discussed later, both the 1kHz real impedance and the Nyquist slope can be used as diagnostic inputs in the fusion algorithm to determine which of the two electrodes should carry more weight when calculating the fused signal. Additionally, one or both of these markers can be used in a diagnostic routine to determine whether the sensor as a whole is acceptable or should be terminated and replaced.

[0308] By definition, a signal (or Isig) dip is an instance of low sensor signal that is mostly temporary in nature, e.g., on the order of a few hours. Such a low signal may be due to, for example, some form of biological occlusion on the sensor surface or due to pressure applied at the insertion site (e.g., when sleeping on one's side). During this period, the sensor data is considered unreliable; however, the signal does eventually recover. In the EIS data, such a signal dip, as opposed to the signal dip caused by changes in blood glucose in the patient, is reflected in the 1 kHz real impedance data, e.g., Figure 24 shown.

[0309] Specifically, in Figure 24 , Isig 2250 of the first working electrode WE1 and Isig 2260 of the second working electrode WE2 both start at about 25nA at the far left end (i.e., 6 p.m.). Over time, both Isigs fluctuate, reflecting the glucose fluctuations near the sensor. For about the first 12 hours or so (i.e., until about 6 a.m.), both Isigs are quite stable, as are their corresponding 1kHz real impedances 2255, 2265. However, between about 12 and 18 hours (i.e., between 6 a.m. and noon), Isig 2260 of WE2 begins to plummet and continues to trend downward over the next few hours until about 9 p.m. During this period, Isig 2250 of WE1 also shows some plummeting, but compared to Isig 2260 of WE2, Isig 2250 is much more stable and the amount of the plummet is much smaller. The characteristics of Isig of WE1 and WE2 are also reflected in their corresponding 1kHz real impedance data. Therefore, as shown in FIG. Figure 24 As shown, during the above time period, while the 1 kHz real impedance (2255) of WE1 remains fairly stable, the 1 kHz real impedance (2265) of WE2 increases significantly.

[0310] By definition, sensitivity loss refers to a situation where the sensor signal (Isig) becomes low and unresponsive for an extended period of time, and is generally unrecoverable. Sensitivity loss may occur for a variety of reasons. For example, electrode poisoning can greatly reduce the effective surface area of ​​the working electrode, severely limiting the current amplitude. Sensitivity loss may also occur at the insertion site due to hypoxia or anoxia. In addition, sensitivity loss may occur due to certain forms of extreme surface occlusion (i.e., a more permanent form of signal drop caused by biological or other factors), which restricts the passage of glucose and oxygen through the sensor membrane, thereby reducing the number / frequency of chemical reactions that generate current in the electrode and ultimately generate the sensor signal (Isig). Note that the various causes of sensitivity loss described above apply to both short-term (7 to 10 days of wear) and long-term (6 months of wear) sensors.

[0311] In EIS data, sensitivity loss is typically preceded by an increase in the absolute values ​​of the phase (|phase|) and the imaginary impedance (|imaginary impedance|) in relatively higher frequency ranges (eg, 128 Hz and above and 1 kHz and above, respectively). Figure 25A An example of a properly operating glucose sensor is shown where the sensor current 2500 responds to glucose, i.e., Isig 2500 tracks glucose fluctuations, but all relevant impedance outputs, such as the 1 kHz real impedance 2510, the imaginary 1 kHz impedance 2530, and the phase (2520) at frequencies equal to or above about 128 Hz, remain stable because they are essentially independent of glucose.

[0312] Specifically, Figure 25A The top graph in FIG shows that, after the first few hours, the 1 kHz real impedance 2510 remains fairly stable at approximately 5 kiloohms (and the 1 kHz imaginary impedance 2530 remains fairly stable at approximately -400 ohms). In other words, at 1 kHz, the real impedance data 2510 and the imaginary impedance data 2530 are essentially independent of glucose, and thus can be used as characteristics or independent indicators of the health, condition, and ultimately reliability of the particular sensor under analysis. However, as previously discussed, different impedance-related parameters may exhibit glucose independence over different frequency ranges, and the ranges in each case may depend on the overall sensor design, such as electrode type, electrode surface area, membrane thickness, membrane permeability, etc.

[0313] Therefore, in the example Figure 25B , for the 90% short tubeless electrode design, the top graph also shows that the sensor current 2501 responds to glucose and that after the first few hours the 1 kHz real impedance 2511 remains fairly stable at approximately 7.5 kiloohms. Figure 25B The bottom graph in FIG shows real impedance data at frequencies between 0.1 Hz (2518) and 1 kHz (2511). As can be seen, the real impedance data at 0.1 Hz (2518) is highly correlated with glucose. However, as shown by reference numerals 2516, 2514, and 2512, as the frequency increases from 0.1 Hz to 1 kHz, i.e., for impedance data measured at frequencies close to 1 kHz, the real impedance becomes increasingly independent of glucose.

[0314] return Figure 25A, the middle graph shows that the phase 2520 at relatively high frequencies is essentially independent of glucose. However, it should be noted that "relatively high frequencies" with respect to this parameter (phase) of the sensor under analysis refers to frequencies of 128 Hz and above. In this regard, the graph shows that the phase is stable for all frequencies between 128 Hz and 8 kHz throughout the period shown. On the other hand, Figure 25C As can be seen in the bottom graph of , while the phase 2522 is stable at 128 Hz (and above), the phase 2524 fluctuates at frequencies increasingly less than 128 Hz, i.e., the phase becomes increasingly glucose-dependent and varies to varying degrees. Figure 25C Examples of electrode designs and applications Figure 25B The electrode designs in are the same, and the top graph of the former is the same as the top graph of the latter.

[0315] Figure 26 An example of sensitivity loss due to hypoxia at the insertion site is shown. In this case, the insertion site became hypoxic after day 4 ( Figure 26 26), which causes the sensor current 2600 to become low and unresponsive. The 1kHz real impedance 2610 remains stable, indicating that there is no physical occlusion on the sensor. However, as shown by the corresponding downward arrows, the changes in the phase 2622 at relatively higher frequencies and the 1kHz imaginary impedance 2632 are consistent with a loss of sensitivity, indicating that this loss is caused by a lack of oxygen at the insertion site. Specifically, Figure 26 As shown, the phase at higher frequencies (2620) and the imaginary impedance (2630) at 1 kHz becomes more negative (indicated by the dark vertical line) before the sensor loses sensitivity, and continues to trend downward as the sensor continues to lose sensitivity. Thus, as described above, this sensitivity loss is anticipated or predicted by the increasing absolute values ​​of the phase (|phase|) and the imaginary impedance (|imaginary impedance|) at relatively high frequency ranges (e.g., 128 Hz and above and 1 kHz and above, respectively).

[0316] The above characteristics can be verified by in vitro tests, examples of which are Figure 27 Shown in. Figure 27Results from in vitro testing of the sensor are shown, simulating hypoxia at various glucose concentrations. In the top graph, Isig fluctuates with glucose concentration as glucose concentration increases from 100 mg / dl (2710) to 200 mg / dl (2720), 300 mg / dl (2730), and 400 mg / dl (2740), and then decreases to 200 mg / dl (2750). In the bottom graph, the phase at relatively high frequencies is generally stable, indicating that it is independent of glucose. However, at very low oxygen concentrations, such as 0.1% O2, the phase at relatively high frequencies fluctuates, as indicated by the circled area and arrows 2760 and 2770. It should be noted that the magnitude and / or direction (i.e., positive or negative) of the fluctuation depends on various factors. For example, the higher the ratio of glucose concentration to oxygen concentration, the greater the magnitude of the phase change. Additionally, the specific sensor design and sensor lifetime (i.e., measured by time since implantation) can affect this fluctuation. So, for example, the older the sensor, the more susceptible it is to interference.

[0317] 28A to 28D Another example of sensitivity loss caused by oxygen deficiency of the redundant working electrodes WE1 and WE2 is shown. Figure 28A As shown, the 1kHz real impedance 2810 is stable even though the sensor current 2800 fluctuates and eventually becomes unresponsive. And, as previously mentioned, the change in the 1kHz imaginary impedance 2820 is consistent with the loss of sensitivity of the sensor. However, in addition, Figure 28B The real impedance data and imaginary impedance data at 0.105 Hz are shown (2830 and 2840, respectively). The imaginary impedance data may be more generally referred to as "0.1 Hz data," indicating that although the imaginary impedance at 0.1 Hz appears fairly stable, the 0.1 Hz real impedance 2830 increases significantly as the sensor loses sensitivity. Figure 28C As shown, in the case of sensitivity loss due to oxygen deficiency, V cntr The 2850 drops to 1.2 volts.

[0318] In summary, the figure shows the following findings: the hypoxia-induced sensitivity loss is accompanied by a lower 1 kHz imaginary impedance (i.e., the latter becomes more negative), a higher 0.105 Hz real impedance (i.e., the latter becomes more positive), and a V cntr In addition, the hypoxic process and V cntr The decrease is typically accompanied by an increase in the capacitive component of the electrochemical circuit. Note that in some diagnostic procedures described later, the 0.105 Hz real impedance may not be used because this relatively low frequency real impedance data appears likely to be analyte-related.

[0319] Finally, combine FIG. 28A to FIG. 28BFor example, it should be noted that impedance measurements at frequencies of 1kHz or higher will often cause EIS-induced spikes in Isig. Figure 28D , where the raw Isig of WE2 is plotted against time. The sharp increase in Isig at the beginning of the spike is a non-Faradaic process due to the double layer capacitance charge. Therefore, the sensitivity loss caused by hypoxia may also be accompanied by a higher EIS-induced spike, as well as a lower 1kHz imaginary impedance, a higher 0.105Hz real impedance, and a V cntr Decline, as discussed above.

[0320] Figure 29 Another example of sensitivity loss is shown. This situation can be considered as the above combined with Figure 24 An extreme version of the Isig dip described. Here, the sensor current 2910 is observed to be low starting at the time of insertion, indicating a problem with the insertion procedure, resulting in electrode occlusion. Figure 25A Compared to the same parameter values ​​for a properly functioning sensor, the 1 kHz real impedance 2920 is significantly higher, while the relatively high frequency phase 2930 and the 1 kHz imaginary impedance 2940 are both shifted to significantly more negative values. The shift in the relatively high frequency phase 2930 and the 1 kHz imaginary impedance 2940 indicates that the loss of sensitivity may be due to hypoxia, which may be caused by occlusion of the sensor surface.

[0321] 30A to 30D Data from another redundant sensor is shown, where the relative differences in 1 kHz real impedance, 1 kHz imaginary impedance, and 0.1 Hz real impedance between two or more working electrodes can be used to detect sensitivity loss due to biofouling. In this example, WE1 exhibits a greater sensitivity loss than WE2, as evident from WE2's higher 1 kHz real impedance 3010, lower 1 kHz imaginary impedance 3020, and higher real impedance at 0.105 kHz (3030). However, in addition, in this example, V cntr 3050 does not decrease. In addition, if Figure 30D As shown, the height of the peak in the raw Isig data does not change much over time. This indicates that for sensitivity loss due to biofouling, V cntr The decrease and increase in peak height are related. In addition, the fact that the height of the peak in the raw Isig data does not change much over time indicates that the capacitive component of the circuit does not change significantly over time, and therefore, the sensitivity loss due to biofouling is related to the resistive component of the circuit (i.e., diffusion).

[0322] The various impedance-related parameters described above can be used individually or in combination as input to: (1) EIS-based sensor diagnostics; and / or (2) fusion algorithms to generate more reliable sensor glucose values. Figure 31 It shows how EIS-based data, ie, impedance-related parameters or characteristics, can be used in a diagnostic routine to determine in real time whether a sensor is functioning properly or should be replaced.

[0323] exist Figure 31 The diagnostic procedure shown in the flowchart is based on periodically collecting EIS data, such as every hour, every half hour, every 10 minutes, or at any other interval, including continuously, as may be appropriate for the particular sensor under analysis. At each such interval, the EIS can be run over the entire spectrum (i.e., a "full scan"), or can be run over a selected frequency range or even at a single frequency. Thus, for example, for an hourly data collection scheme, the EIS can be performed at frequencies in the μHz to MHz range, or can be run over a narrower frequency range, such as between about 0.1 Hz and about 8 kHz, as discussed above. In embodiments, EIS data acquisition can be performed alternately between a full scan and a narrower range of spectra, or according to other schemes.

[0324] The frequency with which EIS is performed and data is collected can be determined by various factors. For example, each execution of EIS consumes a certain amount of power, which is typically provided by the sensor's battery, which is the battery that runs the sensor electronics, including the ASIC described later. Therefore, the battery capacity and the remaining sensor life may help determine the number of times EIS is run, and the breadth of frequencies sampled for each such run. In addition, the embodiments herein contemplate situations where it may be desirable to monitor an EIS parameter (e.g., real impedance at 1 kHz) at a specific frequency based on a first schedule (e.g., once every few seconds or minutes), while monitoring other parameters and / or the same parameters at other frequencies based on a second schedule (e.g., less frequently). In these situations, diagnostic routines can be customized for specific sensors and requirements so that battery power can be conserved and unnecessary and / or redundant EIS data acquisition can be avoided.

[0325] Note that in embodiments, the diagnostic program (e.g. Figure 31The procedure shown in FIG. 1 ) requires a series of separate "tests" that are implemented in order to perform real-time monitoring of a sensor. Multiple tests or markers (also referred to as "multiple markers") are implemented because each time an EIS is run (i.e., each time an EIS procedure is performed), data or characteristics regarding multiple impedance-based parameters can be collected that can be used to detect the state or quality of the sensor (including, for example, whether the sensor has failed or is failing). When performing sensor diagnostics, there may sometimes be diagnostic tests that may indicate a fault, while other diagnostics may indicate no fault. Therefore, the availability of multiple impedance-related parameters and the implementation of a multi-test procedure are advantageous because some of the multiple tests can serve as validity checks for certain other tests. Therefore, real-time monitoring using a multi-marker procedure can include a degree of built-in redundancy.

[0326] In view of the above, Figure 31 The logic of the illustrated diagnostic procedure begins at 3100 after the sensor has been inserted / implanted and an EIS run has been performed to provide EIS data as input. At 3100, using the EIS data as input, it is first determined whether the sensor is still in place. Thus, if the |Z| slope is found to be constant within the tested frequency band (or range), and / or the phase angle is approximately -90°, then the sensor is determined to be no longer in place, and, for example, a warning is sent to the patient / user indicating that sensor removal has occurred. The specific parameters (and their corresponding values) described herein for detecting sensor removal are based on the observation that once the sensor is removed from the body and the membrane is no longer hydrated, the impedance spectrum responds like a capacitor.

[0327] If it is determined that the sensor is still in place, then the logic moves to step 3110 to determine if the sensor is properly initialized. As shown, an "Init. Check" is performed by determining the following conditions: (i) Is |(Z n -Z1) / Z1|>30%, where Z1 is the real impedance of the first measurement, and Z n is the impedance measured at the next interval, as discussed above; and (2) whether the phase angle changes by more than 10° at 0.1 Hz. If the answer to either question is "yes," then the test is satisfactory, i.e., Test 1 is passed. Otherwise, Test 1 is marked as failed.

[0328] At step 3120, Test 2 asks whether the frequency difference (f2-f1) between two consecutive EIS runs is greater than 10 Hz at a phase angle of -45°. Again, a "no" answer is marked as a failure; otherwise, Test 2 is fully satisfied.

[0329] Test 3 at step 3130 is a hydration test. Here, the current impedance Z is askedn Is it less than the initialization impedance Z at 1kHz? pi If yes, then this test is satisfied; otherwise, test 3 is marked as failed. Test 4 in step 3140 is also a hydration test, but this time at a lower frequency. Therefore, this test asks whether Z n Is it less than 300 kOhms at 0.1 Hz? Again, a "no" answer indicates the sensor has failed Test 4.

[0330] At step 3150, Test 5 asks whether the low-frequency Nyquist slope increases overall from 0.1 Hz to 1 Hz. As previously discussed, for a properly operating sensor, the relatively low-frequency Nyquist slope should increase over time. Therefore, if the answer to this query is "yes," then this test is satisfied; otherwise, the test is marked as failed.

[0331] Step 3160 is the final test of this embodiment of the diagnostic procedure. Here, the question is whether the real impedance has decreased overall. As previously discussed, in a properly operating sensor, the real impedance should be expected to decrease over time. Therefore, a "yes" answer here means the sensor is operating normally; otherwise, the sensor fails Test 6.

[0332] Once all six tests have been performed, a decision is made at 3170 as to whether the sensor is operating normally or has failed. In this embodiment, if the sensor passes at least three of the six tests, then the sensor is determined to be operating normally (3172). In other words, in order to be determined to be a failure (3174), the sensor must fail at least four of the six tests. In alternative embodiments, different rules may be used to assess normal operation versus sensor failure. Additionally, in some embodiments, each of the tests may be weighted so that the assigned weight reflects, for example, the importance of the test, or the importance of a particular parameter queried for the test, in determining overall sensor operation (normal versus failure). For example, one test may be weighted twice as much as another test, but only half as much as a third test, and so on.

[0333] In other alternative embodiments, a different number of tests and / or a different set of EIS-based parameters may be used for each test. Figure 32A and Figure 32B An example of a diagnostic program for real-time monitoring consisting of 7 tests is shown. Figure 32AThe logic begins at 3200 after the sensor has been inserted / implanted and an EIS procedure has been performed, providing EIS data as input. Using the EIS data as input, at 3200, it is first determined whether the sensor is still in place. Thus, if the |Z| slope is found to be constant within the tested frequency band (or range), and / or the phase angle is approximately -90°, then the sensor is determined to be no longer in place, and, for example, a warning is sent to the patient / user indicating that sensor removal has occurred. On the other hand, if the sensor is determined to be in place, then the logic moves to starting a diagnostic check (3202).

[0334] At 3205, Test 1 is similar to the above combined Figure 31 Test 1 of the diagnostic procedure in question, but this Test 1 specifies that a follow-up measurement Z be performed 2 hours after the first measurement n Therefore, in this example, Z n =Z 2hr . More specifically, Test 1 compares the actual impedance 2 hours after (sensor implantation and) initialization with the pre-initialization value. Similarly, the second part of Test 1 asks whether the difference between the phase 2 hours after initialization and the phase before initialization is greater than 10° at 0.1 Hz. As previously described, if the answer to either query is yes, then the sensor is determined to have been properly hydrated and initialized and meets Test 1; otherwise, the sensor fails this test. It should be noted that even though this test asks about the impedance and phase changes 2 hours after initialization, the time interval between any two consecutive EIS runs may be shorter or longer depending on a variety of factors, including, for example, sensor design, electrode redundancy level, the extent to which the diagnostic program includes redundant testing, battery charge, etc.

[0335] Moving to 3210, the logic next performs a sensitivity loss check by querying whether the percentage change in the impedance magnitude at 1 kHz and the percentage change in the impedance magnitude in Isig after the 2-hour interval (n+2) are greater than 30%. If the answer to both queries is "yes," then it is determined that the sensor is losing sensitivity and, therefore, that Test 2 has failed. It should be noted that while Test 2 is shown herein based on a preferred percentage difference of 30%, in other embodiments, the percentage difference in the impedance magnitude at 1 kHz and in Isig can be in the range of 10% to 50% for the purposes of performing this test.

[0336] Test 3 (at 3220) is similar to Figure 31Test 5 of the algorithm shown. Here, as previously described, the question is whether the low-frequency Nyquist slope increases from 0.1 Hz to 1 Hz overall. If so, then this test passes; otherwise, the test fails. As shown in 3220, this test can also set a threshold or acceptable range for the percentage change in the low-frequency Nyquist slope, above which the sensor may be considered faulty or at least trigger additional diagnostic testing. In embodiments, this threshold / acceptable range for the percentage change in the low-frequency Nyquist slope may fall within a range of approximately 2% to approximately 20%. In some preferred embodiments, the threshold may be approximately 5%.

[0337] The logic next moves to 3230, another low-frequency test, this time involving phase and impedance magnitudes. More specifically, the phase test asks whether the phase at 0.1 Hz is continuously increasing over time. If so, the test fails. As with other tests that monitor parameter trends, the low-frequency phase test of Test 4 also enables the setting of a threshold or acceptable range for the percentage change in low-frequency phase, beyond which the sensor can be considered faulty, or at least causing problems. In embodiments, this threshold / acceptable range for the percentage change in low-frequency phase can fall within a range of approximately 5% to approximately 30%. In some preferred embodiments, the threshold can be approximately 10%.

[0338] As noted, Test 4 also includes a low-frequency impedance magnitude test, which inquires whether the impedance magnitude at 0.1 Hz increases continuously over time. If so, the test fails. It should be noted that if either the phase test or the impedance magnitude test fails, then Test 4 is considered a "failure." The low-frequency impedance magnitude test of Test 4 is also suitable for setting a threshold or acceptable range for the percentage change in the low-frequency impedance magnitude, above which the sensor can be considered to have failed, or at least caused a problem. In embodiments, this threshold / acceptable range for the percentage change in the low-frequency impedance magnitude can fall within a range of approximately 5% to approximately 30%. In some preferred embodiments, the threshold can be approximately 10%, with the impedance magnitude of common sensors typically ranging from approximately 100 kilo-ohms to approximately 200 kilo-ohms.

[0339] Test 5 (at 3240) is another sensitivity loss check that can be considered complementary to Test 2. Here, if both the percentage change in Isig and the percentage change in the impedance magnitude at 1 kHz are greater than 30%, then the sensor is determined to be recovering from sensitivity loss. In other words, even if Test 2 does not detect sensitivity loss for some reason, it is determined that the sensor has previously experienced some sensitivity loss. As with Test 2, while Test 5 is shown based on a preferred percentage difference of 30%, in other embodiments, the percentage difference in Isig and the impedance magnitude at 1 kHz can be in the range of 10% to 50% for the purposes of this test.

[0340] Moving to 3250, Test 6 provides specific failure criteria for the sensor functionality test, which have been determined based on observed data and the specific sensor design. Specifically, in one embodiment, if at least two of the following three criteria are met, then the sensor can be determined to have failed and therefore cannot respond to glucose: (1) Isig is less than 10nA; (2) imaginary impedance at 1kHz is less than -1500 ohms; and (3) phase at 1kHz is less than -15°. Therefore, if any two conditions in (1) to (3) are not met, then Test 6 is determined to have passed. It should be noted that in other embodiments, if Isig is less than about 5nA to about 20nA, then the Isig prong of this test may fail. Similarly, if the imaginary impedance at 1kHz is less than about -1000 ohms to about -2000 ohms, then the second prong may fail. Finally, if the phase at 1kHz is less than about -10° to about -20°, then the phase prong may fail.

[0341] Finally, step 3260 provides another sensitivity check, in which parameters are evaluated at low frequencies. Thus, Test 7 asks whether, at 0.1 Hz, the magnitude of the difference between the ratio (n+2) of imaginary impedance to Isig, on the one hand, and the permeable value of that ratio, on the other hand, is greater than 30% of the magnitude of the previous value of that ratio. If so, the test fails; otherwise, the test passes. While Test 7 is shown here based on a preferred percentage difference of 30%, in other embodiments, the percentage difference may be in the range of 10% to 50% for the purposes of this test.

[0342] Once all seven tests have been performed, a decision is made at 3270 as to whether the sensor is operating normally or whether a warning should be issued indicating that the sensor has failed (or may be failing). As shown, in this embodiment, if the sensor passes at least four of the seven tests, then the sensor is determined to be operating normally (3272). In other words, in order to be determined to have failed or at least caused a problem (3274), the sensor must fail at least four of the seven tests. If the sensor is determined to be "bad" (3274), then a warning to the effect can be sent to the patient / user, for example. As previously described, in alternative embodiments, different rules can be used to assess the relationship between normal operation and sensor failure / problems. In addition, in some embodiments, each of the tests can be weighted so that the assigned weight reflects, for example, the importance of the test, or the importance of a specific parameter queried for the test, when determining overall sensor operation (normal vs. failed).

[0343] As previously described, in an embodiment, the various above-mentioned impedance-related parameters can be used alone or in combination as inputs to one or more fusion algorithms in the fusion algorithm to generate a more reliable sensor glucose value. Specifically, it is well known that, unlike a single sensor (i.e., a single working electrode) system, multiple sensing electrodes provide glucose readings with higher reliability because the multiple signals obtained from two or more working electrodes can be fused to provide a single sensor glucose value. This signal fusion utilizes the quantitative input provided by EIS to calculate the most reliable output sensor glucose value from the redundant working electrodes. It should be noted that although the subsequent discussion may describe various fusion algorithms with respect to the first working electrode (WE1) and the second working electrode (WE2) as redundant electrodes, this is for illustration only and not for limitation because the algorithms and their underlying principles described herein are applicable to and can be used in redundant sensor systems with more than 2 working electrodes.

[0344] Figure 33A and Figure 33B A top-level flow chart of two alternative approaches is shown, each of which incorporates a fusion algorithm. Specifically, Figure 33A is a flow chart involving a fusion algorithm based on current (Isig), and Figure 33B This is a flow chart for sensor glucose (SG) fusion. As can be seen from the figure, the main difference between the two methods is the calibration time. Therefore, Figure 33A As shown, for Isig fusion, calibration 3590 is performed after fusion 3540 is completed. That is, the redundant Isigs from WE1 to WEn are fused into a single Isig 3589, which is then calibrated to produce a single sensor glucose value 3598. On the other hand, for SG fusion, calibration 3435 is completed for each individual Isig from WE1 to WEn to produce a calibrated SG value (e.g., 3436, 3438) for each working electrode. Thus, the SG fusion algorithm provides independent calibration of each of the multiple Isigs, which may be preferred in some embodiments. After calibration, the multiple calibrated SG values ​​are fused into a single SG value 3498.

[0345] It is important to note that Figure 33A and Figure 33BEach of the flowcharts shown includes a spike filtering process (3520, 3420). As discussed above in the discussion of sensitivity loss, impedance measurements at frequencies of 1 kHz or higher typically cause EIS-induced spikes in Isig. Therefore, once the EIS procedure has been performed on each of the electrodes WE1 to WEn, it is preferred that Isig 3410, 3412, etc. and 3510, 3512, etc. be first filtered to obtain corresponding filtered Isig 3422, 3424, etc. and 3522, 3524, etc. for both SG fusion and Isig fusion. The filtered Isig is then used for Isig fusion, or is first calibrated and then used for SG fusion, as described below. As will become apparent in the subsequent discussion, both fusion algorithms require calculation and assignment of weights based on various factors.

[0346] Figure 34 Details of the fusion algorithm 3440 for SG fusion are shown. Essentially, four factors need to be checked before the fusion weights are determined. First, a sanity check 3450 involves determining whether each of the following parameters is within a specified range for normal sensor operation (e.g., predetermined lower and upper thresholds): (i) Isig; (ii) 1kHz real and imaginary impedance; (iii) 0.105Hz real and imaginary impedance; and (iv) Nyquist slope. As shown, the sanity check 3450 includes a bounds check 3452 and a noise check 3456, where the above parameters are used as input parameters for each check. It should be noted that for brevity, the sig values ​​are not shown in the Nyquist slope. Figures 33A to 35 The real impedance and / or imaginary impedance presented at one or more frequencies, referred to as "Imp" for short, can be used to refer to impedance. In addition, the impedance magnitude and phase (in Figure 33A and 33B , which is also shown above as an input) to calculate both real and imaginary impedances.

[0347] The output from each of the boundary check 3452 and the noise check 3458 is a corresponding reliability index (RI) for each of the redundant working electrodes. Thus, the output from the boundary check includes, for example, RI_bound_We1 (3543) and RI_bound_We2 (3454). Similarly, for the noise check, the output includes, for example, RI_noise_We1 (3457) and RI_noise_We2 (3458). Based on consistency with the above ranges for normal sensor operation, the boundary and noise reliability indices for each working electrode are calculated. Thus, if any parameter exceeds the specified range for a particular electrode, the reliability index for that particular electrode is reduced.

[0348] It should be noted that the thresholds or ranges for the above parameters may depend on various factors, including the specific sensor and / or electrode design. However, in a preferred embodiment, typical ranges for some of the above parameters may be as follows: boundary threshold for 1 kHz real impedance = [0.3e+4 2e+4]; boundary threshold for 1 kHz imaginary impedance = [-2e+3, 0]; boundary threshold for 0.105 Hz real impedance = [2e+4 7e+4]; boundary threshold for 0.105 Hz imaginary impedance = [-2e+5 -0.25e+5]; and boundary threshold for Nyquist slope = [2 5]. For example, the noise may be calculated using a second-order central difference method, where the noise is considered to have exceeded the noise boundary if it is above a certain percentage (e.g., 30%) of the median value of each variable buffer.

[0349] Secondly, the sensor current (Isig) and 1kHz real impedance can be used to detect sensor dips. Figure 34 As shown, Isig and Imp are used as inputs for sag detection 3460. Here, the first step is to determine if there is any divergence between Isigs and whether any such divergence is reflected in the 1kHz real impedance data. This can be achieved by using a mapping 3465 between the Isig similarity index (RI_sim_isig12) 3463 and the 1kHz real impedance similarity index (RI_sim_imp12) 3464. This mapping is crucial because it helps avoid false positives when a sag is not true. When Isig divergence is true, the algorithm will select the sensor with the higher Isig.

[0350] According to some embodiments, the divergence / convergence of two signals (e.g., two Isig or two 1kHz real impedance data points) can be calculated as follows:

[0351] diff_va1=abs(va1-(va1+va2) / 2);

[0352] diff_va2=abs(va2-(va1+va2) / 2);

[0353] RI_sim=1-(diff_va1+diff_va2) / (mean(abs(va1+va2)) / 4)

[0354] Where va1 and va2 are two variables, and RI_sim (Similarity Index) is an index that measures the convergence or divergence of the signal. In this embodiment, RI_sim must be limited between 0 and 1. Therefore, if RI_sim calculated as above is less than 0, it is set to 0, and if RI_sim is greater than 1, it is set to 1.

[0355] The mapping 3465 is performed by using ordinary linear regression (OLR). However, when OLR does not work well, robust median slope linear regression (RMSLR) can be used. For example, for the Isig similarity index and the 1kHz real impedance index, two mapping procedures are required: (i) mapping the Isig similarity index to the 1kHz real impedance similarity index; and (ii) mapping the 1kHz real impedance similarity index to the Isig similarity index. These two mapping procedures will produce two residuals: res12 and res21. Each of the sag reliability indicators 3467, 3468 can then be calculated as follows:

[0356] RI_dip=1-(res12+res21) / (RI_sim_isig+RI_sim_1K_real_impedance).

[0357] The third factor is sensitivity loss 3470, which can be detected using, for example, the trend of the 1kHz imaginary impedance over the past 8 hours. If the trend of one sensor becomes negative, the algorithm will rely on the other sensor. If both sensors lose sensitivity, a simple average will be taken. The 1kHz imaginary impedance is smoothed using a strong low-pass filter (which tends to be noisy) and the trend can be calculated by using, for example, the correlation coefficient over the past 8 hours or a linear regression with respect to time to determine whether the correlation coefficient is negative or the slope is negative. Each of the sensitivity loss reliability indicators 3473, 3474 is then assigned a binary value of 1 or 0.

[0358] The total reliability index (RI) of each of we1, we2, ..., wen is calculated as follows:

[0359] RI_we1=RI_dip_we1×RI_sensitivity_loss_we1×RI_bound_we1×RI_noise_we1

[0360] RI_we2=RI_dip_we2×RI_sensitivity_loss_we2×RI_bound_we2×RI_noise_we2

[0361] RI_we3=RI_dip_we3×RI_sensitivity_loss_we3×RI_bound_we3×RI_noise_we3

[0362] RI_we4=RI_dip_we4×RI_sensitivity_loss_we4×RI_bound_we4×RI_noise_we4 . . .

[0366] RI_we n =RI_dip_we n ×RI_sensitivity_loss_we n ×RI_bound_we n ×RI_noise_we n

[0367] After calculating the corresponding reliability index of the individual working electrodes, the weight of each electrode can be calculated as follows:

[0368] weight_we1=RI_we1 / (RI_we1+RI_we2+RI_we3+RI_we4+…+RI_we n )

[0369] weight_we2=RI_we2 / (RI_we1+RI_we2+RI_we3+RI_we4+…+RI_we n )

[0370] weight_we3=RI_we3 / (RI_we1+RI_we2+RI_we3+RI_we4+…+RI_we n )

[0371] weight_we4=RI_we4 / (RI_we1+RI_we2+RI_we3+RI_we4+…+RI_we n ) . . .

[0375] weight_we n =RI_we n / (RI_we1+RI_we2+RI_we3+RI_we4+…+RI_we n )

[0376] Based on the above, the fused SG 3498 is then calculated as follows:

[0377] SG=weight_we1×SG_we1+weight_we2×SG_we2+weight_we3×SG_we3+

[0378] weight_we4×SG_we4+……+weight_we n ×SG_we n

[0379] The last factor is related to artifacts in the final sensor readings, such as those caused by instantaneous weight changes in sensor fusion. This can be avoided by applying a low-pass filter 3480 to smooth the RI of each electrode, or by applying a low-pass filter to the final SG. In the former case, the filtered reliability indices, such as RI_We1* and RI_We2* (3482, 3484), are used to calculate the weights of each electrode and, therefore, the fused SG 3498.

[0380] Figure 35 The details of the fusion algorithm 3540 for Isig fusion are shown. It can be seen that this algorithm is basically similar to Figure 34 The algorithm for SG fusion is shown in , but with two differences. First, as previously described, for Isig fusion, calibration constitutes the final step of the process in which the individual fused Isigs 3589 are calibrated to produce a single sensor glucose value 3598. See also Figure 33B Secondly, although SG fusion uses the SG values ​​of multiple electrodes to calculate the final SG value 3498, the fused Isig value 3589 is calculated using the filtered Isig (3522, 3524, etc.) of the multiple electrodes.

[0381] In a closed-loop study involving a non-diabetic population, it was found that the fusion algorithm described above achieved significant improvements in mean absolute relative difference (MARD) both on day 1 (when the low start problem is most severe and therefore may have a significant impact on sensor accuracy and therefore reliability) and overall (i.e., over the 7-day life of the sensor). Data from an 88% distributed layout design using high current density (nominal) plating was evaluated using three different methods: (1) calculating a sensor glucose value (SG) by fusion using Medtronic Minimed's Ferrari algorithm 1.0 (which is the SG fusion algorithm discussed above); (2) calculating a SG by identifying better ISIG values ​​using 1 kHz EIS data (via the Isig fusion algorithm discussed above); and (3) calculating a SG by using higher ISIG values ​​(i.e., without using EIS). Details of the study data are as follows:

[0382] (1) Based on Ferrari 1.0Alg, for 88% distributed layout SG using high current density (nominal) electroplating

[0383]

[0384]

[0385]

[0386]

[0387] (2) Based on better ISIG, 1kHz EIS for 88% distributed layout using high current density (nominal) electroplating

[0388]

[0389]

[0390]

[0391]

[0392] (3) SG based on higher ISIG for 88% distributed layout using high current density (nominal) electroplating

[0393]

[0394]

[0395]

[0396]

[0397]

[0398] Based on the above data, it has been found that for the first method, the MARD (%) on day 1 was 19.52%, and the total MARD was 12.28%. For the second method, the MARD on day 1 was 15.96%, and the total MARD was 11.83%. Finally, for the third method, the MARD on day 1 was 17.44%, and the overall MARD was 12.26%. Therefore, for this design with redundant electrodes, it seems that calculating the SG based on the better ISIG using 1kHz EIS (i.e., the second method) provides the greatest advantage. Specifically, the lower MARD on day 1 may be due to, for example, better low-start detection using EIS. In addition, in this study, for WE1 and WE2, the total MARD percentage was more than 1% lower than the overall average MARD of 13.5%. Note that in the above methods, data transitions can be processed, for example, by filtering methods to minimize the severity of the transitions, such as by using methods such as those described above in combination with Figures 33A-35 The low-pass filter 3480 in question.

[0399] It is worth emphasizing that sensor diagnostics, including, for example, assessment of low startup, sensitivity loss, and signal dip events, depend on various factors, including sensor design, number of electrodes (i.e., redundancy), electrode distribution / configuration, etc. Thus, the actual frequency or frequency range at which an EIS-based parameter may be substantially independent of glucose, and thus an independent marker or predictor of one or more of the above-described failure modes, may also depend on the specific sensor design. For example, while it has been found, as described above, that sensitivity loss can be predicted using imaginary impedance at relatively high frequencies—where imaginary impedance is substantially independent of glucose—the glucose level correlation, and therefore the specific frequency range for using imaginary impedance as a marker for sensitivity loss, may shift (higher or lower) depending on the actual sensor design.

[0400] More specifically, as sensor designs increasingly trend toward the use of redundant working electrodes, the size of the redundant working electrodes must be reduced to maintain the overall size of the sensor. The size of the electrodes, in turn, affects the frequency with which a particular diagnostic can be queried. In this regard, it is important to note that the sensor designs described herein and FIG. 33A to FIG. Figure 35 The fusion algorithms shown are to be considered illustrative rather than limiting, as each algorithm may be modified as needed based on the type of sensor under analysis to use EIS-based parameters at frequencies that exhibit minimal glucose correlation.

[0401] Additionally, experimental data indicates that human tissue structure may also affect glucose correlation at different frequencies. For example, in children, real impedance at 0.105 Hz has been found to be a substantially glucose-independent indicator for low-start detection. This is believed to be due to variations in children's tissue structure, such as Weber impedance variations, which are primarily related to the resistive component. See also the subsequent discussion on interferent detection.

[0402] Embodiments herein also relate to the use of EIS in optimizing sensor calibration. As background, in the current method, the slope of the BG vs. Isig curve, which can be used to calibrate subsequent Isig values, is calculated as follows:

[0403]

[0404] Where α is an exponential function of the time constant, β is a function of the blood glucose variance, and the offset is a constant. For sensors under stable conditions, this method provides fairly accurate results. For example, Figure 36 As shown, BG and Isig follow a fairly linear relationship, and the offset can be considered a constant.

[0405] However, there are situations where the above linear relationship does not hold true, such as during periods when the sensor is experiencing a transition. Figure 37As shown, it is clear that Isig-BG pairs 1 and 2 differ significantly in the Isig to BG relationship from pairs 3 and 4. For these types of conditions, using a constant offset will often produce inaccurate results.

[0406] To address this issue, one embodiment is directed to the use of EIS-based dynamic offset, where EIS measurements are used to define the sensor state vector as follows:

[0407] V={real_imp_1K, img_imp_1K, Nyquist_slope, Nyquist_R_square}

[0408] All elements in the vector are essentially independent of BG. It should be noted that Nyquist_R_square is the R square of the linear regression used to calculate the Nyquist slope, that is, the square of the correlation coefficient between the real impedance and the imaginary impedance at relatively low frequencies, and a low R square indicates abnormal sensor performance. For each Isig-BG pair, a state vector is assigned. If a significant difference in the state vector is detected, such as Figure 37 For the example shown, |V2-V3|, then 3 and 4 are assigned different offset values ​​when compared to 1 and 2. Therefore, by using this dynamic offset method, it is possible to maintain a linear relationship between Isig and BG.

[0409] In a second embodiment, a segmentation method based on EIS can be used for calibration. Figure 37 Instances of and vector V, it can be determined that the sensor state during 1 and 2 is significantly different from the sensor state during 3 and 4. Therefore, the calibration buffer can be divided into two segments as follows:

[0410] Isig_buffer1=[Isig1,Isig2]; BG_buffer1=[BG1,BG2]

[0411] Isig_buffer2=[Isig3,Isig3]; BG_buffer2=[BG3,BG3]

[0412] Therefore, when the sensor operates during 1 and 2, Isig_buffer1 and BG_buffer1 will be used for calibration. However, when the sensor operates during 3 and 4, ie, during the transition period, Isig_buffer2 and BG_buffer2 will be used for calibration.

[0413] In yet another embodiment, an EIS-based dynamic slope method may be used for calibration purposes, in which the slope is adjusted using EIS. Figure 38AAn example of how this method can be used to improve sensor accuracy is shown in Figure 1. In this figure, data points 1 to 4 are discrete blood glucose values. Figure 38A It can be seen that there is a sensor dip 3810 between data points 1 and 3, which can be detected using the above sensor state vector V. During the dip, the slope can be adjusted upward to reduce underreading, as shown in FIG. Figure 38A As shown in the figure mark 3820.

[0414] In another embodiment, EIS diagnostics can be used to determine the timing of sensor calibration, which is useful for low-start events, sensitivity loss events, and other similar situations. As is well known, most current methods require regular calibration based on a preset schedule, such as four times per day. However, when using EIS diagnostics, calibration becomes event-driven, so it can only be performed when necessary and most efficient. Here, the state vector V can also be used to determine when the sensor state has changed and, if so, request calibration.

[0415] More specifically, in the illustrative example, Figure 38B A flow chart for EIS-assisted sensor calibration involving low start detection is shown. By using Nyquist slope, 1kHz real impedance and boundary checking 3850 (e.g., in combination with Figures 33A to 35 fusion algorithm (see the aforementioned bounds check and associated thresholds for EIS-based parameters), a reliability indicator 3853 can be developed for startup such that when the 1 kHz real impedance 3851 and the Nyquist slope 3852 are below their respective upper bounds, RI_startup = 1 and the sensor is ready for calibration. In other words, the reliability indicator 3853 is "high" (3854), and the logic can proceed to calibration at 3860.

[0416] On the other hand, when the 1kHz real impedance and the Nyquist slope are above their corresponding upper bounds (or thresholds), RI_startup = 0 (i.e., "low"), and the sensor is not yet ready for calibration (3856), i.e., there may be a low startup problem. Here, the trends of the 1kHz real impedance and the Nyquist slope can be used to predict when both parameters are within range (3870). If it is estimated that this will only take a very short time (e.g., less than an hour), then the algorithm will wait until the sensor is ready, i.e., until the above-mentioned EIS-based parameters are within the bounds (3874), at which point the algorithm proceeds to calibration. However, if the wait time is relatively long (3876), then the sensor can be calibrated now, and then the slope or offset can be gradually adjusted based on the 1kHz real impedance and Nyquist slope trends (3880). It should be noted that by performing adjustments, severe over-reading or under-reading due to low startup can be avoided. As previously mentioned, the EIS-based parameters and related information used in the instant calibration algorithm are essentially independent of glucose.

[0417] It should be noted that although the combination Figure 38B The above description shows a single working electrode and the calculation of a reliability index for starting the working electrode, but this is for illustration only and not limitation. Therefore, in a redundant sensor comprising two or more working electrodes, a boundary check can be performed for each of the multiple (redundant) working electrodes and a startup reliability index calculated. Then, based on the corresponding reliability index, at least one working electrode that can proceed to obtain a glucose measurement value can be identified. In other words, in a sensor with a single working electrode, if the working electrode exhibits low startup, then the actual use of the sensor (for measuring glucose) may have to be delayed until the low startup period ends. This period can typically be about one hour or longer, which is obviously disadvantageous. In contrast, in a redundant sensor, the method described herein allows for adaptive or "smart" startup, where electrodes that can perform data collection can be identified in a relatively short order, for example in about a few minutes. This in turn reduces MARD, because low startup typically increases MARD by about 1 / 2%.

[0418] In yet another embodiment, EIS can assist in adjusting the calibration buffer. For existing calibration algorithms, the buffer size is always 4, representing 4 Isig-BG pairs, and the weights are based on α and β. As previously mentioned, α is an exponential function of the time constant, and β is an exponential function of the blood glucose variance. Here, EIS can help determine when to flush the buffer, how to adjust the buffer weights, and determine the appropriate buffer size.

[0419] Embodiments also relate to using EIS for interference detection. Specifically, it may be desirable to provide a drug infusion device comprising a combined sensor and a drug infusion catheter, wherein the sensor is positioned within the infusion catheter. In such a system, the physical location of the infusion catheter relative to the sensor may cause problems due to potential effects (i.e., interference) on the sensor signal that may be caused by the infused drug and / or its inactive components.

[0420] For example, the diluent used with insulin contains m-cresol as a preservative. In vitro studies have shown that m-cresol can adversely affect glucose sensors if insulin (and therefore m-cresol) is infused in close proximity to the sensor. Therefore, systems combining the sensor and infusion catheter in a single needle must be able to detect and adjust for the effects of m-cresol on the sensor signal. Because m-cresol affects the sensor signal, a method for detecting this interferent independently of the sensor signal itself would be preferred.

[0421] Experiments have shown that the effect of m-cresol on the sensor signal is temporary and therefore reversible. However, when insulin is infused too close to the sensor, m-cresol tends to "poison" the electrode, rendering it no longer able to detect glucose until the insulin (and m-cresol) have been absorbed into the patient's tissues. In this regard, it has been found that there is typically a period of approximately 40 minutes between the start of insulin infusion and the sensor regaining the ability to detect glucose. However, advantageously, it has also been found that, during the same period, the 1 kHz impedance magnitude increases significantly, completely independent of glucose concentration.

[0422] Specifically, Figure 39 Isig and impedance data for an in vitro experiment are shown, wherein the sensor was placed in a 100 mg / dL glucose solution and the 1 kHz impedance was measured every 10 minutes, as shown by the circled data points 3920. Meta-cresol was then added to bring the solution to 0.35% meta-cresol (3930). As can be seen, after the addition of meta-cresol, Isig 3940 initially increased significantly and then began to drift downward. The concentration of glucose in the solution was then doubled by adding 100 mg / dL glucose. However, this had no effect on Isig 3940, as the electrode was unable to detect glucose.

[0423] On the other hand, m-cresol had a significant effect on both impedance magnitude and phase. Figure 40A shows the Bode plot of the phase, and Figure 40BThe Bode plots of the impedance magnitude before and after the addition of m-cresol are shown. It can be seen that after the addition of m-cresol, the impedance magnitude 4010 increases spectrally by at least one order of magnitude from its initialization value 4020. At the same time, the phase 4030 changes completely from its initialization value 4040. Figure 40C Here, the pre-initialization curve 4050 and the post-initialization curve 4060 appear as expected for a properly functioning sensor. However, after adding m-cresol, the curve 4070 becomes completely different.

[0424] The above experiments identify important practical drawbacks to continued reliance on Isig after m-cresol has been added. Figure 39 , a patient / user monitoring the sensor signal may be misled into believing that their glucose level has just peaked and that they should administer a bolus. The user then administers the bolus, at which point Isig has already begun to drift downward again. In other words, everything looks normal to the patient / user. However, in reality, what has actually happened is that the patient has simply administered an unnecessary dose of insulin, which, depending on the patient's glucose level before the bolus was administered, may put the patient at risk of experiencing a hypoglycemic event. This situation makes even more desirable methods for detecting interferents that are as glucose-independent as possible.

[0425] Figure 41 Another experiment is shown, in which the sensor was initialized with a 100 mg / dL glucose solution. The glucose was then raised to 400 mg / dL over one hour and then returned to 100 mg / dL. m-Cresol was then added to raise the concentration to 0.35%, and the sensor was left in this solution for 20 minutes. Finally, the sensor was placed in a 100 mg / dL glucose solution to allow the Isig to recover after exposure to the m-Cresol. As can be seen, after initialization, the 1 kHz impedance magnitude 4110 was approximately 2 kiloohms. When m-Cresol was added, the Isig 4120 peaked, as did the impedance magnitude 4110. Furthermore, when the sensor was returned to the 100 mg / dL glucose solution, the impedance magnitude 4110 also returned to near-normal levels.

[0426] As can be seen from the above experiment, EIS can be used to detect the existence of interfering agents-in this case, interfering agents are meta-cresol. Specifically, because interfering substances affect the sensor in a manner that increases the impedance magnitude on the whole frequency spectrum, impedance magnitude can be used to detect interference. Once interference is detected, the sensor operating voltage can be changed to a point where the interfering substance is not measured, or data reporting can be suspended, and the sensor can indicate to the patient / user that the sensor cannot report data (until the measured impedance returns to the level before the infusion) due to the administration of medication. It should be noted that because the impact of the interfering substance is because the preservative contained in the insulin causes, no matter infusion of insulin is fast or slow, impedance magnitude all can show the same behavior as above.

[0427] Importantly, as mentioned above, the impedance magnitude, and certainly the magnitude at 1 kHz, is essentially independent of glucose. Figure 41 , it can be seen that as the glucose concentration increases from 100 mg / dL to 400 mg / dL—a fourfold increase—the 1 kHz impedance magnitude increases from approximately 2000 ohms to approximately 2200 ohms, or approximately a 10% increase. In other words, the effect of glucose on the impedance magnitude measurement appears to be approximately an order of magnitude smaller than the measured impedance. This "signal-to-noise ratio" level is generally small enough to allow the noise (i.e., glucose effect) to be filtered out, resulting in an impedance magnitude that is essentially glucose-independent. Furthermore, it should be emphasized that the impedance magnitude exhibits a higher degree of glucose independence in actual human tissue compared to the buffer solutions used in the in vitro experiments described above.

[0428] Embodiments herein also relate to an analog front-end integrated circuit (AFE IC), which is a custom application-specific integrated circuit (ASIC) that provides the necessary analog electronics to: (i) support multiple potentiostats and interface with oxygen- or peroxide-based multi-terminal glucose sensors; (ii) interface with a microcontroller to form a micropower sensor system; and (iii) implement EIS diagnostics, fusion algorithms, and other EIS-based processes based on the measurement of EIS-based parameters. More specifically, the ASIC combines diagnostic capabilities to measure the real and imaginary impedance parameters of the sensor over a wide frequency range, as well as digital interface circuitry to enable bidirectional communication with a microprocessor chip. In addition, the ASIC includes power control circuitry and a real-time clock and crystal oscillator to enable operation at very low standby and active power, so that the power to the external microprocessor can be shut down.

[0429] Figure 42A and Figure 42B A block diagram of the ASIC is shown, and Table 1 below provides pad signal descriptions (in Figure 42A and Figure 42B ), where some of the signals are multiplexed onto a single pad.

[0430]

[0431]

[0432] Now refer to Figure 42A and Figure 42B And Table 1 describes the ASIC.

[0433] Power plane

[0434] The ASIC has a power plane powered by power pad VBAT (4210) with an operating input range of 2.0 volts to 4.5 volts. This power plane has a regulator for reducing the voltage of some circuits in this plane. The power supply is called VDDBU (4212) and has output pads for testing and bypassing. The circuits on the VBAT supply include an RC oscillator, a real-time clock (RC osc) 4214, a battery protection circuit, regulator controls, a power-on reset circuit (POR), and various inputs / outputs. The pads on the VBAT power plane are configured to draw less than 75 nA at 40°C and VBAT = 3.50 V.

[0435] The ASIC also has a VDD power supply for powering the logic. The VDD supply voltage range is programmable from at least 1.6 volts to 2.4 volts. Circuitry on the VDD power plane includes most of the digital logic, timers (32 kHz), and a real-time clock (32 kHz). The VDD power plane includes level shifters that connect to other voltage planes as needed. The level shifters, in turn, have interfaces that are regulated so that if the other power plane is unpowered, the current of any powered power plane will not increase by more than 10 nA.

[0436] The ASIC includes an onboard regulator (with shutdown control) and the option of an external VDD power supply. The regulator input is a separate pad, REG_VDD_IN (4216), which has electrostatic discharge (ESD) protection like the other I / O on VBAT. The onboard regulator has an output pad, REG_VDD_OUT (4217). The ASIC also has an input pad for VDD, which is separate from the REG_VDD_OUT pad.

[0437] The ASIC includes an analog power plane called VDDA (4218) that is powered by the VDD onboard regulator or an external power supply, and is typically supplied by a filtered VDD supply. The circuits supplied by VDDA are configured to operate within 0.1 volts of VDD, eliminating the need for level shifting between the VDDA and VDD power planes. The VDDA supply powers the sensor analog circuitry, analog measurement circuitry, and any other noise-sensitive circuitry.

[0438] The ASIC includes pad power supplies VPAD for designated digital interface signals. The pad power supplies have an operating voltage range of at least 1.8V to 3.3V. These pads have separate power supply pads and are powered by an external source. The pads also incorporate level shifters for other onboard circuitry, allowing for a flexible pad power supply range independent of the VDD logic supply voltage. The ASIC regulates the VPAD pad ring signal so that when the VPAD power supply is disabled, the current added to other supplies does not exceed 10nA.

[0439] Bias generator

[0440] The ASIC has a bias generator circuit BIAS_GEN (4220) that is supplied from the VBAT power supply and generates a bias current that is stable when the system is supplied with voltage. The output current has the following specifications: (i) power supply sensitivity: <±2.5% with a power supply voltage of 1.6V to 4.5V; and (ii) current accuracy: <±3% after trimming.

[0441] The BIAS_GEN circuit generates switched and unswitched output currents to supply circuits that require bias current for operation. The BIAS_GEN circuit's operating current consumption is less than 0.3uA at 25°C, with VBAT between 2.5V and 4.5V (excluding any bias output current). Finally, the bias current's temperature coefficient is typically between 4,000ppm / °C and 6,000ppm / °C.

[0442] Voltage Reference

[0443] As described herein, the ASIC is configured with a low-power voltage reference that is powered by the VBAT supply. The voltage reference has an enable input that can accept a signal from logic powered by either VBAT or VDDBU. The ASIC is designed so that when VBAT is powered, the enable signal does not cause the current from any power supply interfaced with this signal to increase by more than 10 nA.

[0444] The reference voltage has the following specifications: (i) output voltage: 1.220 ± 3 mV after trimming; (ii) supply sensitivity: < ± 6 mV from 1.6 V to 4.5 V input; (iii) temperature sensitivity: < ± 5 mV from 0°C to 60°C; and (iv) output voltage default accuracy (no trimming): 1.220 V ± 50 mV. Additionally, the supply current should be less than 800 nA at 4.5 V and 40°C. In this embodiment, when the reference is disabled, the reference output is forced to VSSA to prevent the VDD voltage regulator from overshooting to a level exceeding the logic breakdown voltage.

[0445] 32kHz oscillator

[0446] The ASIC includes a low-power 32.768kHz crystal oscillator 4222, which is powered by a power supply derived from the VDDA supply. The capacitance of the crystal oscillator pads (XTALI, XTALO) can be trimmed via software. Specifically, the frequency trim range is from at least -50ppm to +100ppm, with a maximum step size of 2ppm across the entire trim range. It is assumed that the crystal has a load capacitance of 7pF at each crystal terminal, Ls = 6.9512kH, Cs = 3.3952fF, Rs = 70k, a shunt capacitance of 1pF, and a PC board parasitic capacitance of 2pF.

[0447] The ASIC has a VPAD-level output available on the CLK_32kHZ pad, which can be disabled under software and logic control. The default value disables the 32kHz oscillator. The input pin OSC32K_BYPASS (4224) disables the 32kHz oscillator (no power consumption) and allows digital input to the XTALI pad. The circuitry associated with this function is configured so that, in either state of the OSC32K_BYPASS signal, instead of the oscillator current, it does not increase the ASIC current by more than 10nA when OSC32K_BYPASS is low.

[0448] When the VDDA plane is powered, the 32kHz oscillator is required to be operational at all times, except in bypass conditions. If OSC32K_BYPASS is true, the 32kHz oscillator analog circuitry is put into a low-power state and the XTALI pad is configured to accept a digital input with a level of 0 to VDDA. Note that the duty cycle of the 32kHz oscillator output is between 40% and 60%.

[0449] Timer

[0450] The ASIC includes a timer 4226 that is clocked from a 32kHz oscillator divided by 2. The timer is presettable and has two programmable timeouts. The timer has 24 programmable bits for a total time of 17 minutes and 4 seconds. The timer also has a programmable delay for deactivating clocking of the CLK_32KHz pad and setting the microprocessor (uP) interface signals on the VPAD plane to a predetermined state (see the section below on microprocessor wake-up control signals). This allows the microprocessor to enter suspend mode without the need for an external clock. However, this feature may be disabled by software with a programmable bit.

[0451] The timer also includes a programmable delay for waking up the microprocessor by enabling the CLK_32KHZ clock output and setting UP_WAKEUP high. The transition of POR2 (VDD POR) from the power low state to the power good state enables the 32kHz oscillator, the CLK_32KHZ clock output, and sets UP_WAKEUP high. Power down and power up are configured to be controlled by programmable control bits.

[0452] Real-time clock (RTC)

[0453] The ASIC also has a 48-bit readable / writable binary counter that is operated by an ungated, free-running 32kHz oscillator. Writing to the real-time clock 4228 requires writing to the address with the key before the clock can be written. Write access to the clock is configured to terminate between 1 millisecond and 20 milliseconds after writing to the key address.

[0454] The real-time clock 4228 is configured to be reset to half count (MSB = 1, all other bits are 0) by POR1_IN (VBAT POR) or POR2_IN (VDD_POR) via a power-on reset. In an embodiment, the real-time clock has programmable interrupt capability and is designed to be robust against single event upsets (SEUs), which can be achieved through layout techniques or by adding capacitance to appropriate nodes as needed.

[0455] RC oscillator

[0456] The ASIC further includes an RC clock powered by the VBAT supply or a supply derived from VBAT. The RC oscillator is always running, but can be bypassed by writing to a register bit in analog test mode (see the section on digital testing) and applying a signal to GPIO_VBAT at a level between 0 and VBAT. The RC oscillator is not trimmable and includes the following specifications: (i) frequency between 750 Hz and 1500 Hz; (ii) duty cycle between 50% ± 10%; (iii) current consumption less than 200 nA at 25°C; (iv) frequency variation less than ± 2% for a 1V to 4.5V VBAT supply and better than 1% for a 1.8V to 4.5V VBAT supply; and (v) frequency variation less than +2, -2% over a temperature range of 15°C to 40°C, with VBAT = 3.5V. The RC frequency can be measured using a 32kHz crystal oscillator or an external frequency source (see oscillator calibration circuit).

[0457] Real-time RC clock (based on RC oscillator)

[0458] The ASIC includes a 48-bit readable / writable binary ripple counter based on an RC oscillator. Writing to the RC real-time clock requires writing to an address using a key before the clock can be written. After writing to the key address, write access to the clock terminates between 1 millisecond and 20 milliseconds, where the protection window is configured to be generated using the RC clock.

[0459] The real-time RC clock allows relative timestamping even when the crystal oscillator is off and is configured to reset to half count (MSB = 1, all other values ​​0) on POR1_IN (BAT POR). The real-time RC clock is designed to be robust to single-event upsets (SEUs) through layout techniques or by adding capacitance to appropriate nodes when needed. On the falling edge of POR2_IN, or if the ASIC enters a low-battery state, the RT real-time clock value can be captured into a register that can be read through the SPI port. This register and associated logic are on the VBAT or VDDBU power plane.

[0460] Battery protection circuit

[0461] The ASIC includes a battery protection circuit 4230 that uses a comparator to monitor the battery voltage and is powered by a power supply derived from the VBAT power plane. The battery protection circuit is configured to always operate while power is supplied to the VBAT power supply. The battery protection circuit uses an RC oscillator for its clock signal and has an average current consumption of less than 30nA, including a 3MΩ total resistance external voltage divider.

[0462] The battery protection circuit uses an external switched voltage divider with a ratio of .421 for a 2.90V battery threshold. The ASIC also has an internal voltage divider with a ratio of .421 ± 0.5%. This voltage divider is connected between BATT_DIV_EN (4232) and VSSA (4234), and the voltage divider output is a pin called BATT_DIV_INT (4236). To conserve pins within the packaged component, BATT_DIV_INT in this embodiment is internally connected to BATT_DIV within the package. Also in this configuration, BATT_DIV_EN does not need to come out of the package, saving two package pins.

[0463] The battery protection circuit is configured to sample the voltage on input pin BATT_DIV (4238) approximately twice per second, where the sampling time is generated by an RC oscillator. The ASIC is capable of adjusting the voltage divider of the RC oscillator to adjust the sampling time interval to 0.500 seconds ± 5 milliseconds, where the RC oscillator operates within its operating tolerances. In a preferred embodiment, the ASIC has a test mode that enables more frequent sampling intervals during testing.

[0464] The comparator input is configured to accept inputs from 0 to VBAT volts. For inputs from 0 to VBAT volts, the input current to the comparator input BATT_DIV is less than 10 nA. The comparator sampling circuit outputs a positive pulse to the BATT_DIV_EN pad, which can be used by external circuitry to enable the off-chip resistor divider only during the sampling time to save power. The voltage high logic level is the VBAT voltage, and the low level is the VSS level.

[0465] At VBAT = 3.0V, the output resistance of the BATT_DIV_EN pad should be less than 2 kiloohms. This allows the voltage divider to be driven directly from this output. After a programmable number of consecutive samples indicating a low-battery condition, the comparator control circuitry triggers an interrupt on the interrupt output pad UP_INT. The default number of samples is 4, but the number of consecutive samples can be programmed from 4 to 120.

[0466] After a programmable number of consecutive samples indicates a low battery condition following the generation of the aforementioned UP_INT, the comparator control circuitry is configured to generate a signal that will cause the ASIC to enter a low-power mode: the VDD regulator will be disabled, and a low signal will be asserted on the VPAD_EN pad. This is referred to as a low-battery condition. Again, the number of consecutive samples is programmable from 4 to 120 samples, with a default value of 4 samples.

[0467] The comparator features separate programmable thresholds for falling and rising voltages on BATT_DIV. This is implemented in digital logic to multiplex two values ​​into the circuit depending on the state of the low-battery condition. Therefore, if the low-battery condition is low, the falling threshold applies; if the low-battery condition is high, the rising threshold applies. Specifically, the comparator features 16 programmable thresholds ranging from 1.22 to 1.645 ± 3%, with the DNL of the programmable thresholds set to less than 0.2 LSB.

[0468] The comparator threshold varies by less than + / - 1% from 20°C to 40°C. The default threshold for falling voltage is 1.44V (for a nominal voltage divider, the VBAT threshold is 3.41V), and the default threshold for rising voltage is 1.53V (for a nominal voltage divider, the VBAT threshold is 3.63V). After the ASIC enters the low battery state, if the comparator senses four consecutive indications of a good battery, the ASIC will begin the microprocessor startup sequence.

[0469] Battery power plane power-on reset

[0470] If the input VBAT swings more than 1.2 volts within a 50 microsecond period, or the VBAT voltage falls below 1.6 ± 0.3 volts, a power-on reset (POR) output is generated on pad nPOR1_OUT (4240). This POR is extended to a minimum pulse width of 5 milliseconds. The output of the POR circuit is configured as active low and is directed to pad nPOR1_OUT on the VBAT power plane.

[0471] The IC has an input pad nPOR1_IN (4242) for the battery power plane POR. This input pad has RC filtering so that pulses shorter than 50 nanoseconds do not cause a logic reset. In this embodiment, nPOR1_OUT is externally connected to nPOR1_IN during normal operation, thereby separating the analog circuitry from the digital circuitry for testing. nPOR1_IN resets all logic on any of the power planes and initializes all registers to their default values. As a result, the reset status register POR bit is set and all other reset status register bits are cleared. The POR reset circuitry is configured to consume no more than 0.1uA from the VBAT power supply for more than 5 seconds after power-up.

[0472] VDD Power-On Reset (POR)

[0473] The ASIC also features a voltage comparator circuit that generates a VDD voltage level reset signal after power-up or when VDD drops below a programmable threshold. The range is programmable across several voltage thresholds. The default value is 1.8V-15% (1.53V). POR2 has a programmable threshold for rising voltages, which implements hysteresis. The rising threshold is also programmable, with a default value of 1.60V ±3%.

[0474] The POR signal is active low and has an output pad nPOR2_OUT (4244) on the VDD power plane. The ASIC also has an active low POR open-drain output nPOR2_OUT_OD (4246) on the VBAT power plane. This can be used to apply the POR to other system components.

[0475] The VDD powered logic has a POR sourced from the input pad nPOR2_IN (4248). The nPOR2_IN pad is located on the VDD power plane and has RC filtering so that pulses shorter than 50 nanoseconds do not cause a logic reset. In normal use, nPOR2_OUT is configured to be connected externally to the nPOR2_IN input pad, thereby separating the analog circuitry from the digital circuitry.

[0476] After VDD exceeds the programmable threshold, the reset generated is extended to at least 700 milliseconds of active time to ensure that the crystal oscillator is stable. The POR reset circuitry consumes no more than 0.1uA from the VDD supply for more than 5 seconds after power-up and no more than 0.1uA from the VBAT supply for more than 5 seconds after power-up. The registers storing the POR thresholds are powered by the VDD power plane.

[0477] Sensor interface electronics

[0478] In an embodiment, the sensor circuit system supports up to five sensor WORK electrodes (4310) in any combination of peroxide or oxygen sensors, but in other embodiments, a greater number of such electrodes may be accommodated. When the peroxide sensor WORK electrode sources current, the oxygen sensor WORK electrode sinks current. For this embodiment, the sensor may be configured as follows: Figure 43 in the potentiostat configuration shown.

[0479] The sensor electronics have programmable power control for each electrode interface circuit to minimize current consumption by shutting off the current to unused sensor electronics. The sensor electronics also contain electronics for driving the COUNTER electrode 4320, which uses feedback from the RE (reference) electrode 4330. When this circuitry is not in use, its current can be programmed to shut down to save power. The interface electronics contain a multiplexer 4250 so that the COUNTER and RE electrodes can be connected to any (redundant) WORK electrode.

[0480] The ASIC is configured to provide the following sensor interfaces: (i) RE: Reference electrode, which establishes the reference potential of the solution for the electronics to set the WORK voltage; (ii) WORK1 to WORK5: Working sensor electrodes, where the desired reduction / oxidation (redox) reactions occur; and (iii) COUNTER: The output from this pad maintains a known voltage at the RE electrode relative to system VSS. In this embodiment, the ASIC is configured to be able to individually set the WORK voltages of up to five WORK electrodes with a resolution and accuracy better than or equal to 5 mV.

[0481] In oxygen mode, the WORK voltage is programmable between at least 0 and 1.22V relative to VSSA. In peroxide mode, the WORK voltage is programmable between at least 0.6V and 2.054V relative to VSSA. If VDDA is less than 2.15V, the WORK voltage can be operated to VDDA-0.1V. The ASIC includes current measurement circuitry to measure the WORK electrode current in peroxide sensor mode. For example, this can be implemented using a current-to-voltage or current-to-frequency converter, which can have the following specifications: (i) current range: 0 to 300nA; (ii) voltage output range: same as the WORK electrode in peroxide / oxygen mode; (iii) output offset voltage: ±5mV maximum; and (iv) uncalibrated resolution: ±0.25nA.

[0482] After applying the calibration factor to the gain and assuming an acquisition time of 10 seconds or less, the current measurement accuracy is:

[0483] 5pA-1nA:±3%±20pA

[0484] 1nA-10nA:±3%±20pA

[0485] 10nA-300nA: ±3% ±.2nA

[0486] For the current-to-frequency converter (ItoF) only, the frequency range can be between 0 Hz and 50 kHz. In peroxide mode, the current converter must operate within the specified voltage range relative to the work electrode's VSS. Here, the current consumption is less than 2 μA from a 2.5 V supply, with the work electrode current per converter being less than 10 nA, including the digital-to-analog (DAC) current.

[0487] The current converter can be enabled or disabled by software control. When disabled, the WORK electrode will exhibit extremely high impedance values, i.e., greater than 100 megohms. Again, for ItoF only, the output of the IF converter will enter a 32-bit counter, which can be read, written, and cleared by the microprocessor and test logic. During the counter read, the clock control to the counter is suspended to ensure accurate reading.

[0488] In one embodiment, the ASIC also includes a current measurement circuit for measuring the work electrode current in oxygen sensor mode. This circuit can be implemented as a current-to-voltage or current-to-frequency converter, and programmable bits can be used to configure the current converter for operation in oxygen mode. As previously mentioned, in oxygen mode, the current converter must operate within the specified voltage range of the work electrode relative to VSS. Here, the current range is similarly 3.7pA to 300nA, and the voltage output range is the same as for the work electrode in oxygen mode, with a maximum output offset voltage of ±5mV and an uncalibrated resolution of 3.7pA ±2pA.

[0489] After applying the calibration factor to the gain and assuming an acquisition time of 10 seconds or less, the current measurement accuracy is:

[0490] 5pA-1nA:±3%±20pA

[0491] 1nA-10nA:±3%±20pA

[0492] 10nA-300nA: ±3% ±.2nA

[0493] For the current-to-frequency converter (ItoF) only, the frequency range can be between 0Hz and 50kHz, and the current consumption of the 2.5V power supply is less than 2uA, where the WORK electrode current of each converter is less than 10nA, including the DAC current. The current converter can be enabled or disabled by software control. When disabled, the WORK electrode will exhibit an extremely high impedance value, i.e., greater than 100 megohms. Similarly, for ItoF only, the output of the IF converter will enter a 32-bit counter, which can be read, written, and cleared by a microprocessor and test logic. During the counter read, the clock control of the counter is suspended to ensure accurate reading.

[0494] In an embodiment, the reference electrode (RE) 4330 has an input bias current of less than 0.05 nA at 40°C. The COUNTER electrode adjusts its output to maintain a desired voltage on the RE electrode. This is achieved by an amplifier 4340, whose output to the COUNTER electrode 4320 attempts to minimize the difference between the actual RE electrode voltage and the target RE voltage, which is set by the DAC.

[0495] The RE set voltage can be programmed between at least 0 and 1.80V, and the common mode input range of the COUNTER amplifier includes at least 0.20 to (VDD-0.20)V. Register bits can be used to select the common mode input range, if necessary, and provide programming of the COUNTER's operating mode. The WORK voltage is set to a resolution and accuracy better than or equal to 5mV. It should be noted that in normal mode, the COUNTER voltage seeks to maintain the RE voltage at the programmed RE target value. However, in force counter mode, the COUNTER voltage is forced to the programmed RE target voltage.

[0496] All electrode drive circuits are configured to drive the electrodes to the electrode load without oscillation under any usage conditions. Figure 44 Shown according to Figure 43 Equivalent ac inter-electrode circuit for an embodiment of the potentiostat configuration shown. Figure 44 The equivalent circuit shown can be located between any of the electrodes, namely WORK1 to WORK5, COUNTER, and RE, where the value ranges of the corresponding circuit components are as follows:

[0497] Ru=[200-5k]ohm

[0498] Cc=[10-2000]pF

[0499] Rpo=[1-20] kiloohms

[0500] Rf=[200-2000] kiloohms

[0501] Cf=[2-30]uF

[0502] During initialization, the drive current for the work and counter electrodes requires a higher current than that required for normal potentiostat operation. Therefore, if additional drive is required, the electrode drive circuit can be programmed to a higher power state using programmable register bits. Low power operation is crucial in normal potentiostat mode, where electrode current is typically less than 300 nA.

[0503] In a preferred embodiment, during initialization, the WORK1 through WORK5 electrodes can be programmed between 0 and VDD volts in steps of 5 mV or less, and the electrodes are capable of driving or sinking a current output of 20 uA minimum from 0.20 V to (VDD - 0.20 V). Also during initialization, the ASIC is typically configured to measure the current of a WORK electrode up to 20 uA with an accuracy of ±2% of the measured value ±40 nA. Additionally, during initialization, the RE set voltage is programmable as previously described, the COUNTER DRIVE CIRCUIT output must be capable of sourcing or sinking 50 uA minimum using a COUNTER electrode from 0.20 V to (VDD - 0.20 V), and the supply current to the initialization circuitry (VDDA and VDDA) is required to exceed any output current provided by less than 50 uA.

[0504] Current Calibrator

[0505] In an embodiment, the ASIC has a current reference that can be manipulated to any WORK electrode for calibration purposes. In this regard, the calibrator includes a programmable bit that causes the current output to sink current or provide current. The programmable current includes at least 10nA, 100nA, and 300nA, assuming the tolerance of the external precision resistor is 0, its accuracy is better than ±1% ±1nA. For the reference resistor, the calibrator uses a 1 megohm precision resistor connected to pad TP_RES (4260). In addition, for the purpose of initialization and / or sensor status, the current reference can be directed to the COUNTER or RE electrode. A constant current can be applied to the COUNTER or RE electrode, and the electrode voltage can be measured using an ADC.

[0506] High-speed RC oscillator

[0507] Referring back to FIG. 42 , the ASIC further includes a high-speed RC oscillator 4262 that supplies an analog-to-digital converter (ADC) 4264, an ADC sequencer 4266, and other digital functions that require a clock speed higher than 32 kHz. The high-speed RC oscillator is phase-locked to a 32 kHz clock (32.768 kHz) to provide an output frequency programmable from 524.3 kHz to 1048 kHz. In addition, the high-speed RC oscillator has a duty cycle of 50% ± 10%, a phase jitter of less than 0.5% rms, a current of less than 10 uA, and a frequency stability over the entire VDD operating range (voltage range of 1.6 to 2.5 V). The high-speed RC oscillator defaults to “off” (i.e., disabled), in which case the current consumption is less than 10 nA. However, the ASIC has a programmable bit for enabling the high-speed RC oscillator.

[0508] analog-to-digital converters

[0509] The ASIC includes a 12-bit ADC (4264) with the following features: (i) capable of converting in less than 1.5 milliseconds when operating from a 32kHz clock; (ii) capable of performing faster conversions when clocked from a high-speed RC oscillator; (iii) accuracy of at least 10 bits (12 bits ± 4 counts); (iv) a reference voltage input of 1.220V with a temperature sensitivity of less than 0.2mV / °C between 20°C and 40°C; (v) full-scale input ranges of 0 to 1.22V, 0 to 1.774V, 0 to 2.44V, and 0 to VDDA, with the 1.774 and 2.44V ranges having programmable bits to reduce the conversion range to lower values ​​to accommodate lower VDDA voltages; (vi) consuming less than 50uA of current from the power supply; (vii) having a converter capable of operating from either a 32kHz clock or a high-speed RC clock; (vii) a DNL of less than 1LSB; and (viii) issuing an interrupt at the end of a conversion.

[0510] like Figure 42A and 42B As shown, the ASIC has an analog multiplexer 4268 at the input of the ADC 4264, both the ADC and the multiplexer being controllable by software. In a preferred embodiment, at least the following signals are connected to the multiplexer:

[0511] (i) VDD - core voltage and regulator output

[0512] (ii) VBAT - battery power

[0513] (iii) VDDA - analog power supply

[0514] (iv) Reference electrode of RE-sensor

[0515] (v) COUNTER - Counter electrode of the sensor

[0516] (vi) WORK1 to WORK5 - working electrodes of the sensor

[0517] (vii) Temperature sensor

[0518] (viii) At least two external pins for analog signal input

[0519] (ix) EIS integrator output

[0520] (x)ItoV current converter output.

[0521] The ASIC is configured so that the ADC load does not exceed ±0.01 nA for the COUNTER, RE, WORK1 to WORK5 inputs, the temperature sensor, and any other inputs that may be adversely affected by the load. The multiplexer includes a voltage divider for any input with a voltage above the ADC input voltage range, and a buffer amplifier that reduces the input resistance of the divided input to less than 1 nA for load-sensitive inputs. The buffer amplifier has a common-mode input range of at least 0.8 V to the VDDA voltage and an offset of less than 3 mV between inputs of 0.8 V and VDDA-0.1 V.

[0522] In a preferred embodiment, the ASIC has a mode for performing ADC measurements in a programmed sequence. Therefore, the ASIC includes a programmable sequencer 4266 that supervises the measurement of up to 8 input sources for ADC measurements with the following programmable parameters:

[0523] (i) ADC MUX input

[0524] (ii) ADC range

[0525] (iii) Delay time before measurement, where the delay is programmable from 0 to 62 milliseconds in steps of 0.488 milliseconds

[0526] (iv) The number of measurements for each input is 0 to 255

[0527] (v) Number of measurement cycles: 0-255, where a measurement cycle is a series of up to 8 input measurements repeated multiple times (eg, as an outer loop in a program).

[0528] (vi) Delay between measurement cycles, where the delay is programmable from 0 to 62 milliseconds in steps of 0.488 milliseconds.

[0529] The sequencer 4266 is configured to start upon receipt of an automatic measurement start command and the measured values ​​can be stored in the ASIC for retrieval via the SPI interface. Note that the sequencer time base is programmable between the 32kHz clock and the high speed RC oscillator 4262.

[0530] Sensor diagnostics

[0531] As previously described in detail, embodiments relate to the use of impedance and impedance-related parameters in, for example, sensor diagnostics and Isig / SG fusion algorithms. To this end, in a preferred embodiment, the ASIC described herein is capable of measuring the impedance magnitude and phase angle of any WORK sensor electrode relative to the RE and COUNTER electrodes when in a potentiostat configuration. This is done, for example, by measuring the amplitude and phase of the current waveform in response to a sinusoidal waveform superimposed on the WORK electrode voltage. See, for example, Figure 42B Diagnostic circuit system 4255 in.

[0532] The ASIC is capable of measuring the resistance and capacitance components from any electrode to any electrode, for example, via the electrode multiplexer 4250. It should be noted that such measurements may disturb the sensor balance and may require stabilization time or sensor initialization to record stable electrode currents. As previously described, although the ASIC can be used to perform impedance measurements across a wide frequency spectrum, for the purposes of the embodiment, a relatively narrow frequency range can be used. Specifically, the ASIC's sine wave measurement capability can include test frequencies from about 0.10 Hz to about 8192 Hz. When performing this measurement, the minimum frequency resolution according to one embodiment can be limited as shown in Table 2 below:

[0533] Table 2

[0534]

[0535] The sine wave amplitude is programmable from at least 10 mVp-p to 50 mVp-p in steps of at least 5 mV, and from 60 mVp-p to 100 mVp-p in steps of at least 10 mV. In a preferred embodiment, the amplitude accuracy is better than ±5% or ±5 mV, whichever is greater. In addition, the ASIC can measure the electrode impedance with the accuracy specified in Table 3 below:

[0536]

[0537] In one embodiment, the ASIC can measure the input waveform phase relative to the time base, which can be used in the impedance calculation to improve accuracy. The ASIC can also have on-chip resistors to calibrate the electrode impedance circuit described above. The on-chip resistors can be calibrated by comparing them to a known 1 megohm off-chip precision resistor.

[0538] The waveform data samples can also be used to determine the impedance. The data can be transferred to an external microprocessor using a serial peripheral interface (SPI) for calculation and processing. The converted current data is fully buffered to enable 2000 ADC data conversions to be transferred to an external device via the SPI interface without data loss. This assumes a maximum latency of 8 milliseconds to service the data transfer request interrupt.

[0539] In an embodiment, rather than measuring electrode impedance with a sine wave (or in addition to measuring electrode impedance with a sine wave), the ASIC can measure electrode current with a step input. Here, the ASIC can provide programmable amplitude steps of 10 to 200 mV to the electrode with a resolution better than 5 mV and sample (measure) the resulting current waveform. The duration of the sampling can be programmable to at least 2 seconds in 0.25 second steps, and the sampling interval for measuring current can contain at least five programmable binary-weighted steps of approximately 0.5 milliseconds to 8 milliseconds.

[0540] The electrode voltage sample has a resolution of less than 1 mV and a range of at most ±0.25 volts. This measurement can be made with respect to a suitable stable voltage to reduce the required dynamic range for data conversion. Similarly, the electrode current sample has a resolution of less than 0.04 uA and a range of at most 20 uA. If the measurement polarity is programmable, the current measurement can be unipolar.

[0541] In one embodiment, current measurement can use an IV converter. Additionally, the ASIC can include on-chip resistors to calibrate current measurements. These on-chip resistors can be calibrated by comparison to a known 1-megohm off-chip precision resistor. Current measurement sample accuracy is better than ±3% or ±10nA, whichever is greater. As previously mentioned, the converted current data is fully buffered to enable transfer of 2000 ADC data conversions to an external device via the SPI interface without data loss. This assumes a maximum latency of 8 milliseconds to service the data transfer request interrupt.

[0542] Calibration voltage

[0543] The ASIC includes a precision voltage reference for calibrating the ADC. The output voltage is 1.000V ±3%, with a production variation of less than ±1.5%, and stability better than ±3mV over the temperature range of 20°C to 40°C. This precision calibration voltage can be calibrated by the on-chip ADC by comparing it to an external precision voltage during manufacturing. During manufacturing, the calibration factor can be stored in system non-volatile memory (not on this ASIC) for even higher accuracy.

[0544] The current consumption of the calibration voltage circuit is preferably less than 25 uA. In addition, the calibration voltage circuit can be powered down to less than 10 nA when not in use to save battery power.

[0545] Temperature sensor

[0546] The ASIC has a temperature transducer with a sensitivity between 9 and 11 mV per degree Celsius between -10°C and 60°C. The temperature sensor's output voltage allows the ADC to measure temperature-related voltages using a 0 to 1.22V ADC input range. The temperature sensor's current consumption is preferably less than 25uA, and the temperature sensor can be powered down to less than 10nA when not in use to conserve battery power.

[0547] VDD voltage regulator

[0548] The ASIC has a VDD voltage regulator with the following characteristics:

[0549] (i) Minimum input voltage range: 2.0V to 4.5V.

[0550] (ii) Minimum output voltage: 1.6 to 2.5 V ± 5%, default value is 2.0 V.

[0551] (iii) Dropout voltage: When Iload = 100uA, Vin = 2.0V, Vin-Vout < 0.15V.

[0552] (iv) The output voltage is programmable with an accuracy within 2% of the values ​​shown in Table 4 below:

[0553] Table 4

[0554]

[0555] (v) The regulator can supply 1mA output at 2.5V where the input voltage is 2.8V.

[0556] (vi) The regulator also has input and output pads which may be open circuited if an external regulator is used. In this inoperative mode the current consumption of the regulator circuit is preferably less than 100 nA.

[0557] (vii) The change in output voltage from a 10 uA load to a 1 mA load is preferably less than 25 mV.

[0558] (viii) The current consumption of the source, excluding the output current under a 1 mA load, is less than 100 uA.

[0559] (ix) The current consumption of the source excluding the output current under a 0.1 mA load is less than 10 uA.

[0560] (x) The current consumption of the source excluding the output current under a 10uA load is less than 1uA.

[0561] General purpose comparator

[0562] The ASIC contains at least two comparators 4270, 4271 powered by VDDA. The comparators use 1.22V as a reference to generate thresholds. The output of the comparators can be read by the processor and will generate a maskable interrupt on a rising or falling edge determined by a configuration register.

[0563] The comparators have power control to reduce power when not in use, and each comparator draws less than 50 nA of current. For a 20 mV overdrive signal, the comparator response time is preferably less than 50 microseconds, and the offset voltage is less than ±8 mV.

[0564] The comparators also have programmable hysteresis, with hysteresis options including: threshold on rising input = 1.22V + Vhyst, threshold on falling input = 1.22 - Vhyst, or no hysteresis (Vhyst = 25 ± 10mV). The output from any comparator can be used on any GPIO on any power plane. (See the "GPIO" section.)

[0565] Sensor connection sensing circuit system on RE

[0566] The analog switched capacitor circuit monitors the impedance of the RE connection to determine if the sensor is connected. Specifically, a capacitor of approximately 20 pF is switched at a frequency of 16 Hz, driven by an inverter whose output swings from VSS to VDD. A comparator senses the voltage swing on the RE pad, and if the swing is less than a threshold, the comparator output indicates a connection. This comparison is performed on both transitions of the pulse. A swing below the threshold on both transitions is required to indicate a connection, and a comparison indicating a high swing on either phase indicates a disconnect. The connect / disconnect signals are debounced so that the transition of their state requires at least 1 / 2 second for a stable indication of the new state.

[0567] The circuit has six thresholds defined by the following resistors in parallel with a 20pF capacitor: 500 kΩ, 1 MΩ, 2 MΩ, 4 MΩ, 8 MΩ, and 16 MΩ. This parallel equivalent circuit is between the RE pad and a virtual ground, which can be at any voltage between the power rails. Threshold accuracy is better than ±30%.

[0568] The output of the sensor connection sensing circuitry can programmably generate an interrupt or a processor start if a sensor is connected or disconnected. This circuit is active whenever nPOR2_IN is high and VDD and VDDA are present. The circuit consumes less than 100nA of current on average.

[0569] Wakeup Pad

[0570] The wakeup circuitry is powered by the VDD supply, with an input range of 0V to VBAT. The wakeup pad 4272 has a weak pull-down current of 80±40nA. This current can be sourced from the output of BIAS_GEN 4220. With a 0V input, the circuit consumes an average current of less than 50nA.

[0571] The rising input voltage threshold, Vih, of the WAKEUP input is 1.22 ± 0.1 V, and the falling input threshold is -25 mV ± 12 mV of the rising threshold. In a preferred embodiment, the circuitry associated with the WAKEUP input draws no more than 100 nA of current for any input voltage between -.2 and VBAT (this current does not include the input pull-down current). The WAKEUP pad is debounced for at least 1 / 2 second.

[0572] The output of the WAKEUP circuit can programmatically generate an interrupt or processor start if the WAKEUP pad changes state (see the Event Handlers section). It is important to note that the WAKEUP pad circuitry is configured to assume a low current of <1 nA if the battery protection circuit indicates a low battery condition.

[0573] UART WAKEUP

[0574] The ASIC is configured to monitor the nRX_EXT pad 4274. If nRX_EXT remains high (UART BREAK) for more than 1 / 2 second, a UART WAKEUP event is generated. Due to sampling, a UART WAKEUP event can occur with a continuous high level for as little as 1 / 4 second. The UART WAKEUP event can programmatically generate an interrupt, a wakeup, and / or a microprocessor reset (nRESET_OD). (See the Event Handler section.)

[0575] In a preferred embodiment, the circuitry associated with the UART WAKEUP input draws no more than 100 nA, and if the battery protection circuitry indicates a low battery condition, the UART WAKEUP pad circuitry is configured to assume a low current of < 1 nA. The UART wakeup input has a rising input voltage threshold, Vih, of 1.22 ± 0.1 V. The falling input threshold is -25 mV ± 12 mV of the rising threshold.

[0576] Microprocessor wake-up control signal

[0577] ASICs are able to generate signals to help control the power management of the microprocessor. Specifically, ASICs can generate the following signals:

[0578] (i) nSHUTDN - nSHUTDN controls the power enable of the off-chip VDD regulator. nSHUTDN is connected to the VBAT power rail. If the battery protection circuitry indicates a low battery condition, nSHUTDN is low; otherwise, nSHUTDN is high.

[0579] (ii) VPAD_EN - VPAD_EN controls the power enable of the external regulator that supplies the VPAD power. An internal signal corresponding to this external signal ensures that when the VPAD power is disabled, the input from the VPAD pad does not draw additional current due to a floating input. The VPAD_EN pad is an output on the VBAT power rail. If the battery protection signal indicates a low battery level, the VPAD_EN signal is low. The VPAD_EN signal can be set low by a software command that starts a timer; the timer's terminal count forces VPAD_EN low. The following events can cause the VPAD_EN signal to go high if the battery protection signal indicates a good battery (see event handler for more details): nPOR2_IN transition from low to high; SW / timer (programmable); WAKEUP transition; from low to high and / or from high to low (programmable); sensor connection transition; from low to high and / or from high to low (programmable); UART interrupt; and RTC time event (programmable).

[0580] (iii) UP_WAKEUP - UP_WAKEUP can be connected to a microprocessor wake-up pad. It is intended to wake the microprocessor from sleep mode or similar power-down modes. The UP_WAKEUP pad is an output on the VPAD power rail. The UP_WAKEUP signal can be programmed to be active low, active high, or pulsed. The UP_WAKEUP signal can be set low by a software command that starts a timer; the terminal count of the timer forces UP_WAKEUP low. If the battery protection signal indicates good battery power, the following events may cause the UP_WAKEUP signal to go high (see event handler for more details): nPOR2_IN transitions from low to high; SW / timer (programmable); WAKEUP transitions; low to high and / or high to low (programmable); sensor connection transitions; low to high and / or high to low (programmable); UART interrupts; and RTC time events (programmable). The WAKEUP signal may be delayed by a programmable amount. If WAKEUP is programmed to be pulsed, the pulse width can be programmed.

[0581] (iv) CLK_32KHZ - The CLK_32KHZ pad can be connected to a microprocessor to supply a low-speed clock. The clock is on-off programmable and is programmably turned on in response to a wake-up event. The CLK_32KHZ pad is an output on the VPAD power rail. If the battery protection signal indicates low battery, the CLK_32KHZ signal is low. The CLK_32KHZ output can be programmed to be off via a programmable bit. It is on by default. The CLK_32KHZ signal can be disabled by a software command that starts a timer; the timer's terminal count forces CLK_32KHZ low. If the battery protection signal indicates good battery level, the following events may enable the CLK_32KHZ signal (see event handler for more details): nPOR2_IN transition from low to high; SW / timer (programmable); WAKEUP transition; low to high and / or high to low (programmable); sensor connection transition; low to high and / or high to low (programmable); UART interrupt; RTC time event (programmable); and detection of low battery level by the battery protection circuit.

[0582] (v) nRESET_OD - nRESET_OD can be connected to a microprocessor to cause a microprocessor reset. nRESET_OD is programmable for wake-up events. The nRESET_OD pad is an output on the VPAD power rail. This pad is open-drain (NFET output). If the battery protection signal indicates low battery, the nRESET_OD signal is low. The nRESET_OD active time is programmable from 1 to 200 milliseconds. The default value is 200ms. The following events may cause the nRESET_OD signal to be asserted low (see Event Handler for more details): nPOR2_IN; SW / timer (programmable); WAKEUP transition; from low to high and / or from high to low (programmable); sensor connection transition; from low to high and / or high to low (programmable); UART interrupt; and RTC time event (programmable).

[0583] (vi) UP_INT - UP_INT can be connected to the microprocessor to communicate interrupts. UP_INT is programmable for wake-up events. The UP_INT pad is an output on the VPAD power rail. If the battery protection signal indicates low battery, the UP_INT signal is low. The UP_INT signal can be set high by a software command that starts a timer; the timer's terminal count forces UP_INT high. If the battery protection signal indicates good battery, the following events may cause the UP_INT signal to be asserted high (see event handler for more details): SW / timer (programmable); WAKEUP transition; low to high and / or high to low (programmable); sensor connection transition; low to high and / or high to low (programmable); UART interrupt; RTC time event (programmable); detection of low battery by the battery protection circuit; and any ASIC interrupt when unmasked.

[0584] The ASIC has GPIO1 and GPIO0 pads that can be used as boot mode control for a microprocessor. A POR2 event resets a 2-bit counter whose bits are mapped to GPIO1 and GPIO0 (MSB and LSB, respectively). A rising edge on a UART interrupt increments the counter by one, where the counter counts modulo 4 and reaches zero if incremented in state 11. The boot mode counter can be preset via the SPI.

[0585] Event Handler / Watchdog

[0586] The ASIC incorporates event handlers to define responses to events, including changes in system state and input signals. Events include all interrupt sources (e.g., UART_BRK, WAKE_UP, sensor connection, etc.). Software can program the event handler's responses to stimuli via the SPI interface. However, some responses may be hardwired (non-programmable).

[0587] Actions in the event handler include enabling / disabling VPAD_EN, enabling / disabling CLK_32KHZ, asserting nRESET_OD, asserting UP_WAKEUP, and asserting UP_INT. Event watchdog timers 1 through 5 are individually programmable from 250 ms to 16,384 seconds in 250 ms increments. The timeouts for event watchdog timers 6 through 8 are hard-coded. Timer 6 and Timer 7 have a timeout of 1 minute; Timer 8 has a timeout of 5 minutes.

[0588] The ASIC also includes a watchdog function to monitor the microprocessor's response when an event is triggered. If the microprocessor fails to acknowledge event-triggered activity, the event watchdog is activated. Once activated, the event watchdog executes a series of programmable actions (Event Watchdog Timers 1 through 5), followed by a series of hardwired actions (Event Watchdog Timers 6 through 8) to regain microprocessor responsiveness. This series of actions includes interrupting, resetting, waking up, asserting the 32 kHz clock, and powering down and powering up the processor.

[0589] During this sequence of events, if the microprocessor regains its ability to acknowledge recorded activity, the event watchdog is reset. If the ASIC fails to obtain acknowledgement from the microprocessor, the event watchdog powers down the microprocessor under conditions that will allow UART_BRK to restart the microprocessor and activate the alarm. When activated, the alarm condition generates a square wave on the ALARM pad in a programmable repetitive pattern with a frequency of approximately 1 kHz. The programmable pattern has two programmable sequences with programmable burst on and off times. The alarm has another programmable mode that can be programmed via the SPI port. This mode has two programmable sequences with programmable burst on and off times.

[0590] Digital to Analog (D / A)

[0591] In the preferred embodiment, the ASIC has two 8-bit D / A converters 4276, 4278 with the following characteristics:

[0592] (i) The D / A settles in less than 1 millisecond with a load less than 50 pF.

[0593] (ii) The D / A has at least 8-bit accuracy.

[0594] (iii) The output range can be programmed from 0 to 1.22V or 0 to VDDA.

[0595] (iv) The temperature sensitivity of the D / A voltage reference is less than 1mV / °C

[0596] (v)DNL is less than 1LSB.

[0597] (vi) The D / A consumes less than 2uA from the VDDA power supply.

[0598] (vii) Each D / A has an output 1 to a pad.

[0599] (viii) The D / A output is high impedance. The load current must be less than 1nA.

[0600] (ix) The D / A pad can be programmed to output digital signals from the register. The output swing is from VSSA to VDDA.

[0601] Charger / data downloader interface

[0602] TX_EXT_OD 4280 is an open-drain output whose input is the signal on the TX_UP input pad. This allows the TX_EXT_OD pad to be turned on during the UART idle condition. The TX_EXT_OD pad has a comparator that monitors its voltage. If the voltage is above the comparator threshold voltage within the debounce period (1 / 4 second), the output nBAT_CHRG_EN (4281) will go low. This comparator and other associated circuitry with this function are on the VBAT and / or VDDBU planes.

[0603] The circuitry associated with this function must allow a low level on the TX_EXT_OD pad due to normal communication with an external device without deasserting nBAT_CHRG_EN. If POR1 is active, nBAT_CHRG_EN will be high (not asserted). The comparator threshold voltage is between 0.50V and 1.2V. The comparator will have hysteresis; the falling threshold is approximately 25mV lower than the rising threshold.

[0604] The nRX_EXT pad inverts the signal on this pad and outputs it to RX_UP. This way, the nRX_EXT signal will idle low. nRX_EXT must accept inputs up to the VBAT voltage. The nRX_EXT threshold is 1.22V ± 3%. The output of this comparator is available via the SPI bus for reading by the microprocessor.

[0605] The nRX_EXT pad also incorporates a programmable current source of 80 ± 30 nA, with a maximum voltage of VBAT. The ASIC layout features a shield programmable option to adjust this current from 30 nA to 200 nA in steps of less than 50 nA, with minimal shield changes. A programmable bit can be used to prevent UART interrupt detection and force RX_UP high. In normal operation, this bit is set high before enabling the current source to nRX_EXT, and then set low after disabling the current source to ensure that no glitches are generated on RX_UP or a UART interrupt event is generated. Care should be taken to implement a wet connector detector; when the current source into nRX_EXT is active, the RX comparator output, indicating a low input voltage, will indicate leakage current. The ASIC includes a pull-down resistor of approximately 100 kiloohms on the nRX_EXT pad. This pull-down is disconnected when the current source is active.

[0606] Sensor connection switch

[0607] The ASIC should have a pad SEN_CONN_SW (4282) that can detect low tolerance to VSS (4284). SEN_CONN_SW provides a current of 5 to 25uA with SEN_CONN_SW = 0V and a maximum open circuit voltage of 0.4V. The ASIC layout has a shield programmable option to adjust this current from 1uA to 20uA in steps of less than 5uA with minimal shield changes. SEN_CONN_SW has associated circuitry that can detect the presence of a resistor between SEN_CONN_SW and VSSA (4234) with a threshold between 2 and 15 kiloohms. The average current consumption of this circuit is a maximum of 50nA. Sampling must be used to achieve this low current.

[0608] Oscillator calibration circuit

[0609] The ASIC has counters whose inputs can be directed to internal or external clock sources. One counter generates a programmable gating interval for the other counter. The gating interval ranges from 1 to 15 seconds from the 32kHz oscillator. The clocks that can be directed to either counter are the 32kHz RC oscillator, the high-speed RC oscillator, and the input from any GPIO pad.

[0610] Oscillator Bypass

[0611] The ASIC can replace each oscillator output with an external clock. The ASIC has a register that can only be written when a specific TEST_MODE is asserted. This register has bits that enable the external input of the RC oscillator and can be shared with other analog test control signals. However, if TEST_MODE is not active, this register will not allow any oscillator bypass bits to be active.

[0612] The ASIC also has an input pad for an external clock, bypassing the RC oscillator. The GPIO_VBAT pad is on the VBAT power plane. The ASIC further includes a bypass enable pad, OSC32K_BYPASS, for the 32kHz oscillator. When driven high, the 32kHz oscillator output is supplied by driving the OSC32KHZ_IN pad. Note that the OSC32KHZ_IN pad is typically connected to a crystal.

[0613] The ASIC has an input for an external clock to bypass the HS_RC_OSC. The bypass is enabled via a programmable register bit. The HS_RC_OSC can be supplied programmably via a GPIO on the VDD plane or a GPIO on the VPAD plane.

[0614] SPI slave port

[0615] The SPI slave port includes an interface consisting of a chip select input (SPI_nCS) 4289, a clock input (SPI_CK) 4286, a serial data input (SPI_MOSI) 4287, and a serial data output (SPI_MISO) 4288. The chip select input (SPI_nCS) is an active-low input asserted by an off-chip SPI master to initiate and qualify an SPI transaction. When SPI_nCS is asserted low, the SPI slave port configures itself as an SPI slave and executes data transactions based on the clock input (SPI_CK). When SPI_nCS is inactive, the SPI slave port resets itself and remains in reset mode. Because this SPI interface supports block transfers, the master should hold SPI_nCS low until the transfer is complete.

[0616] The SPI clock input (SPI_CK) is always asserted by the SPI master. The SPI slave port uses the rising edge of SPI_CK to latch incoming data on the SPI_MOSI input and the falling edge of SPI_CK to latch outgoing data on the SPI_MISO output. The serial data input (SPI_MOSI) is used to transfer data from the SPI master to the SPI slave. All data bits are asserted after the falling edge of SPI_CK. The serial data output (SPI_MISO) is used to transfer data from the SPI slave to the SPI master. All data bits are asserted after the falling edge of SPI_CK.

[0617] SPI_nCS, SPI_CK, and SPI_MOSI are always driven by the SPI master unless the SPI master is powered down. If VPAD_EN is low, these inputs are regulated so that the current consumption associated with them is less than 10nA and the SPI circuitry remains in reset or inactive. SPI_MISO is only driven by the SPI slave port when SPI_nCS is active; otherwise, SPI_MISO is tri-stated.

[0618] The chip select (SPI_nCS) defines and forms the data packet for an SPI data transaction. A data packet consists of three parts: a 4-bit command segment, a 12-bit address segment, and then any number of 8-bit data bytes. Command bit 3 serves as the direction bit. A "1" indicates a write operation, and a "0" indicates a read operation. The combination of command bits 2, 1, and 0 has the following definitions. Unused combinations are undefined.

[0619] (i)0000: Read data and increment address.

[0620] (ii) 0001: Read data, the address remains unchanged

[0621] (iii)0010: read data, decrement address

[0622] (iv) 1000: Write data and increment address

[0623] (v)1001: Write data, address remains unchanged

[0624] (vi)1010: write data, decrement address

[0625] (vii) x011: Test port addressing

[0626] The 12-bit address portion defines the starting byte address. If SPI_nCS remains active after the first data byte to indicate a multi-byte transfer, the address is incremented by one after each byte is transferred. The bits of the address (Address<11:0>) <11> Indicates the highest address bit. The address wraps around after reaching the limit.

[0627] The data is in byte format and block transfers can be performed by extending SPI_nCS to allow all bytes to be transferred in one packet.

[0628] Microprocessor interrupts

[0629] The ASIC has an output at the VPAD logic level UP_INT for sending an interrupt to the host microprocessor. The microprocessor interrupt module consists of an interrupt status register, an interrupt mask register, and functionality for logically ORing all interrupt states into a microprocessor interrupt. The interrupt is implemented to support both edge-sensitive and level-sensitive interrupts. The polarity of the interrupt is programmable. The default interrupt polarity is TBD.

[0630] In a preferred embodiment, all interrupt sources on the AFE ASIC are recorded in the interrupt status register. Writing a "1" to the corresponding interrupt status bit clears the corresponding pending interrupt. All interrupt sources on the AFE ASIC can be masked using the interrupt mask register. Writing a "1" to the corresponding interrupt mask bit masks the corresponding pending interrupt. Writing a "0" to the corresponding interrupt mask bit disables the masking of the corresponding interrupt. The default state of the interrupt mask register is TBD.

[0631] General-purpose input / output (GPIO) / parallel test port

[0632] In an embodiment, the ASIC may have eight GPIOs that operate on VPAD level signals. The ASIC may have one GPIO that operates on VBAT level signals and one GPIO that operates on VDD level signals. All GPIOs may have at least the following characteristics:

[0633] (i) Register bits control the selection and direction of each GPIO.

[0634] (ii) The ASIC is able to configure the GPIOs as inputs that can be read via the SPI interface.

[0635] (iii) The ASIC can configure the GPIO as an input for generating an interrupt.

[0636] (iv) The ASIC is able to configure each GPIO as an output controlled by a register bit that can be written to via the SPI interface.

[0637] (v) The ASIC can programmably output the input signal applied to GPIO_VBAT or GPIO_VDD to the GPIO (on the VPAD power plane) (level shifting function).

[0638] (vi) The ASIC can configure each GPIO as an input to the oscillator calibration circuit.

[0639] (vii) The ASIC has means to configure each general purpose comparator output to at least one GPIO on each power plane. The polarity of the comparator output can be programmed via a programmable bit.

[0640] (viii)GPIO has the function of generating microprocessor interrupts.

[0641] (ix)GPIO can be programmed as open-drain output.

[0642] (x) GPIOs on the VPAD power plane can be configured to implement microprocessor startup control.

[0643] The parallel test port shares an 8-bit GPIO on the VPAD voltage plane. This test port can be used to observe register contents and various internal signals. In normal mode, the output of this port is controlled by the port configuration register. Writing 8'hFF' to the GPIO_O1S_REG and GPIO_O2S_REG registers directs test port data to the GPIO outputs, while writing 8'h00' to the GPIO_ON_REG register disables test port data and enables GPIO data to the GPIO outputs.

[0644] By addressing the target register via the SPI slave port, registers and pre-grouped internal signals can be observed through this test port. The command bits of the SPI packet are set to 4'b0011, followed by the 12-bit target register address. The parallel test port continues to display the contents of the addressed register until the next test port addressing command is received.

[0645] Analog test port

[0646] The IC has a multiplexer feeding pad TP_ANAMUX (4290) that will provide visibility to internal analog circuit nodes for testing. The IC also has a multiplexer feeding pad TP_RES (4260) that will provide visibility to internal analog circuit nodes for testing. In a common application, this pad will also accommodate a precision 1 meg resistor to perform various system calibrations.

[0647] Chip ID

[0648] The ASIC contains a 32-bit shielded programmable ID. A microprocessor using the SPI interface will be able to read this ID. This ID will be located in the analog electronics block, allowing changes without rewiring the chip. The design should ensure that changing the ID only requires a metal or contact mask.

[0649] Spare test output

[0650] The ASIC has 16 spare digital output signals that can be multiplexed to 8-bit GPIOs under command sent through the SPI interface. These signals will be organized into two 8-bit bytes and will be connected to VSS when not in use.

[0651] Digital Test

[0652] The ASIC has a test mode controller using two input pins TEST_CTL0 (4291) and TEST_CTL1 (4292). The test controller generates signals based on the combination of test control signals with the following functions (TEST_CTL<1:0>):

[0653] (i) 0 is the normal operation mode;

[0654] (ii) 1 is the simulation test mode;

[0655] (iii)2 is scanning mode;

[0656] (iv)3 is the analog test mode, where VDD_EN is controlled by the input to GPIO_VBAT.

[0657] The test controller logic is split between the VDD and VDDBU power planes. During scan mode, TEST_VBAT should be asserted high to condition the analog outputs to the digital logic. The ASIC has scan chains implemented in as much of the digital logic as possible for fast digital testing.

[0658] Leakage test pin

[0659] The ASIC has a pin called LT_VBAT, which, when high, puts all analog blocks into an inactive mode, drawing only leakage current from the power supply. LT_VBAT keeps all digital outputs from the analog blocks in a stable high or low state to avoid affecting the interface logic current consumption. The LT_VBAT pad is located on the VBAT plane and has a pull-down resistor between 10 kOhm and 40 kOhm.

[0660] Power requirements

[0661] In one embodiment, the ASIC includes a low-power mode in which at least the microprocessor clock is shut down, the 32kHz real-time clock is running, and circuitry is active to detect sensor connections, changes in the WAKE_UP pin, or a BREAK on the nRX_EXT input. This mode has a total current draw of VBAT (VDDBU), VDD, and VDDA of 4.0uA maximum. When the battery protection circuit detects low battery (see Battery Protection Circuit Description), the ASIC enters a mode in which only the VBAT and VDDBU power planes are active. This is called a low-battery state. The VBAT current in this mode is less than 0.3uA.

[0662] With the ASIC programmed for potentiostat configuration, and assuming the WORK electrode current is measured once per minute, the average current consumption of all power supplies is less than 7uA. In the potentiostat configuration, any one WORK electrode is active in H2O2 (peroxide) mode and its voltage is set to 1.535V, the COUNTER amplifier is on and VSET_RE is set to 1.00V, a 20MEG load resistor is connected between WORK and COUNTER, and COUNTER and RE are connected together. The measured current after calibration should be 26.75nA ± 3%. With a WORK electrode current of 25nA, adding an additional WORK electrode may increase the combined current consumption by less than 2uA.

[0663] In the case where the ASIC is programmed to a potentiostat configuration with the diagnostic function enabled to measure the impedance of one of the WORK electrodes relative to the COUNTER electrode, the ASIC is configured to satisfy the following:

[0664] (i) Test frequencies: 0.1 Hz, 0.2 Hz, 0.3 Hz, 0.5 Hz, 1.0 Hz, 2.0 Hz, 5.0 Hz, 10 Hz, 100 Hz, 1000 Hz and 4000 Hz.

[0665] (ii) The measurement of the above frequencies should not exceed 50 seconds.

[0666] (iii) The total charge supplied to the ASIC is less than 8 millicoultry.

[0667] environment

[0668] In a preferred embodiment, the ASIC:

[0669] (i) Operate within the commercial temperature range of 0 to 70°C and meet all requirements.

[0670] (ii) Functionally operates between -20°C and 80°C but may also do so with reduced accuracy.

[0671] (iii) It is expected to be handled after storage in the temperature range of -30 to 80°C.

[0672] (iv) Expected to operate in a relative humidity range of 1% to 95%.

[0673] (v) Unless otherwise specified, ESD protection is greater than ±2KV when using the Human Body Model on all pins in the TBD package.

[0674] (vi) Configured so that the WORK1 to WORK5, COUNTER, RE, TX_EXT_OD, and nRX_EXT pads withstand greater than ±4KV Human Body Model.

[0675] (vii) Configured so that the leakage current of WORK1 to WORK5 and RE pads is less than 0.05 nA at 40°C.

[0676] In an embodiment, the ASIC may be fabricated in a .25 micron CMOS process, and the backup data for the ASIC is on DVD disk 916-TBD.

[0677] As described in detail above, the ASIC provides the necessary analog electronics to: (i) support multiple potentiostats and interface with oxygen- or peroxide-based multi-terminal glucose sensors; (ii) interface with a microcontroller to form a micropower sensor system; and (iii) implement EIS diagnostics based on measurements of EIS-based parameters. The measurement and calculation of EIS-based parameters will now be described according to the embodiments herein.

[0678] As previously mentioned, impedance at frequencies ranging from 0.1 Hz to 8 kHz can provide information about the state of the sensor electrodes. The AFE IC circuitry incorporates circuitry for generating the measurement forcing signal and for performing the measurements used to calculate impedance. Design considerations for this circuitry include current consumption, accuracy, measurement speed, required processing power, and the on-time required by the controlling microprocessor.

[0679] In a preferred embodiment, the technique used by the AFE IC to measure electrode impedance is to superimpose a sinusoidal voltage on the DC voltage driving the electrode and measure the phase and amplitude of the resulting AC current. To generate the sine wave, the AFE IC incorporates a digitally synthesized sine wave current. This digital technique is used because the frequency and phase can be precisely controlled using a crystal-derived timebase, and frequencies from DC to 8 kHz can be easily generated. The sinusoidal current is applied to a resistor in series with a voltage source to add an AC component to the electrode voltage. This voltage is the AC forcing voltage. It is then buffered by the amplifier that drives the selected sensor electrode.

[0680] The current driving the electrode contains an AC current component derived from the excitation sine wave and is converted to a voltage. This voltage is then processed by multiplying it by a square wave with a fixed phase relative to the resulting sine wave. This multiplied voltage is then integrated. After a programmable number of integration intervals—an integer number of 1 / 2 cycles of the driving sine wave—the voltage is measured by the ADC. Calculations involving the integrated voltage value provide the real and imaginary components of the impedance.

[0681] The advantage of using an integrator for impedance measurement is that the noise bandwidth of the measurement is significantly reduced compared to simply sampling the waveform. In addition, the sampling time requirement is significantly reduced, thus relaxing the speed requirements of the ADC.

[0682] Figure 45 The main blocks of the EIS circuitry in the AFE IC are shown (in Figure 42B (Indicated by reference numeral 4255 in the figure). IDAC 4510 generates a step-by-step sine wave synchronously with the system clock. The higher frequency of this system clock causes the IDAC to step through a lookup table containing a digital code. This code drives the IDAC, which produces an output current that approximates a sine wave. This sine wave current is forced through a resistor to provide an AC component Vin_ac and a DC offset VSET8 (4520). When the IDAC circuit is disabled, the DC output voltage returns to VSET8, thereby minimizing the disturbance to the electrode balance. This voltage is then buffered by amplifier 4530, which drives the electrode through a series resistor Rsense. The differential voltage across Rsense is proportional to the current. This voltage is presented to multiplier 4540, which multiplies the voltage by +1 or -1. This is done through switches and a differential amplifier (instrumentation amplifier). The system clock is divided to produce phase clock 4550, which controls the multiplication function and can be set to 0, 90, 180, or 270 degrees relative to the sine wave.

[0683] Figures 46A-46F and 47A-47F show the graphs Figure 45The signal of the circuit shown is a simulation of the current with a 0-degree phase shift representing a real resistor. For these example simulations, the simulated input values ​​were chosen to provide a current sensing voltage equal to 0.150V. In order to obtain enough information to derive the impedance and phase, two integrations are required: one for the 0-degree phase multiplication ( Figures 46A-46F ), a 90-degree phase multiplication is performed ( Figures 47A to 47F ).

[0684] Impedance calculation

[0685] The equation that describes the integrator output is given below. For simplicity, only 1 / 2 of the sine wave period is considered. From Figures 46A to 46F and Figures 47A to 47F As can be seen from the curve, the total integrator output is approximately the integrated value of 1 / 2 sine wave cycle multiplied by the number of 1 / 2 cycles being integrated. It should be noted that the multiplication switch, which is related to the integration time, performs a "gating" function on the integrator signal; this can be thought of as setting the integration limits. The multiplication signal has a fixed phase with the generated sine wave. This phase can be set to 0, 90, 180, or 270 degrees using software. If the sine wave is in phase with the multiplied square wave (0-degree shift), the limits of integration will be π (180°) and 0 (0°). If the sine wave is shifted 90 degrees, the limits of integration can be thought of as 3 / 4π (270°) and 1 / 4π (90°).

[0686] The following formula shows a multiplied square wave that is in phase (0°) with the driving sine wave. This produces a voltage proportional to the real component of the current. Note that Φ is the phase shift of the sine wave relative to the multiplied square wave; Vout is the integrator output, and Aampl is the current sine wave amplitude. Furthermore, the period of the sine wave is 1 / f, and RC is the integrator time constant.

[0687]

[0688]

[0689]

[0690]

[0691] If Φ = 0, then This corresponds to the real part of the current.

[0692] For a multiplied square wave in quadrature phase (90°) with respect to the driving sine wave, to produce an output proportional to the imaginary component of the current:

[0693]

[0694]

[0695]

[0696]

[0697] If Φ = 0, then This corresponds to the imaginary part of the current.

[0698] exist Figures 46A to 46F In the first example curve shown, A ampl =0.150V, the frequency is 1kHz, Φ = 0, and the RC of the integrator is 20 megohms and 25pF, which gives RC = 0.5 milliseconds. Substituting these numbers into the equation gives 0.09549V, which compares favorably with the integrator output of the curve in Figure 46. It should be noted that the integrator output during integration is the incremental voltage from the start of integration to the measurement.

[0699] For 90° square wave multiplication, since sin(0) = 0, the result should be 0. The simulation results are close to this value.

[0700] To calculate the phase:

[0701] because From this we can conclude that: Where V out90 is the integrator output with a 90° phase shift of multiplication, and V out0 is the integrator output with a 0° phase shift. V out90 and V out0 The output must be integrated over the same number of 1 / 2 cycles or normalized by the number of cycles. It is important to note that in a practical software (e.g., ASIC) implementation, only an integer number of cycles (360°) is allowed, because an integer number of cycles compensates for any offset in the circuitry before the multiplier.

[0702] The magnitude of the current can be obtained from and or or This current has the phase angle calculated above.

[0703] The above analysis shows that the current amplitude and its phase relative to the multiplication signal can be determined. The excitation voltage is generated with a fixed phase (0, 90, 180 or 270 degrees) relative to the multiplication signal - this is done digitally, so it can be precisely controlled. However, there is at least one amplifier in the path before the excitation sine wave is applied to the electrode; this introduces undesirable phase shift and amplitude errors. This can be compensated by integrating the excitation sine wave signal obtained electronically near the electrode. Therefore, the amplitude and any phase shift of the excitation voltage can be determined. Since the paths of the current and voltage waveforms will be processed by the same circuit, any analog circuit gain and phase errors will be eliminated.

[0704] Since the variable of interest is impedance, it may not be necessary to actually calculate A ampl Because the current waveform and the voltage waveform are integrated over the same path, a simple relationship exists between the ratio of current to voltage. The integrated current sense voltage V is represented by an additional subscript. I_out and the integrated electrode voltage is called V V_out To describe the phase of the multiplication function:

[0705]

[0706]

[0707] Impedance will be voltage divided by current. Therefore,

[0708]

[0709] The magnitude of the voltage and current can also be obtained from the square root of the square of the integrated voltage at 0 and 90 degrees phase. Therefore, the following can also be used:

[0710]

[0711] For relatively high frequencies, such as those above about 256 Hz, the integration of the waveform can be performed with a hardware integrator. High frequencies require four measurement cycles: (i) a cycle for the in-phase sensor current; (ii) a cycle for the 90-degree out-of-phase sensor current; (iii) a cycle for the in-phase excitation voltage; and (iv) a cycle for the 90-degree out-of-phase excitation voltage.

[0712] Two integrators can be used for relatively low frequencies, such as frequencies below about 256 Hz, where the integrated value is formed by numerically combining the integrator results in the system microprocessor. Knowing how many integrations there are per cycle enables the microprocessor to properly calculate the 0 and 90 degree components.

[0713] Synchronizing the integration with the excitation AC waveform and dividing the integration into at least four parts at lower frequencies eliminates the need for a hardware multiplier, since the combination of the integration parts in the microprocessor can implement the multiplication function. Thus, only one integration pass is required to obtain both real and imaginary current information. For lower frequencies, the amplifier phase error becomes smaller, so below a frequency, for example, between 1 Hz and 50 Hz, and preferably below about 1 Hz, it is not necessary to determine the excitation voltage phase. Furthermore, for lower frequencies, the amplitude can be assumed to be constant, so that only one measurement cycle is required to determine the impedance after stabilization.

[0714] As mentioned above, although one hardware integrator is used for relatively high frequencies, two integrators can be used for relatively low frequencies. Figure 45 The schematic in Figure 1 shows the EIS circuitry in an AFE IC for relatively high EIS frequencies. At these frequencies, the integrator does not saturate when integrating within a single cycle. In practice, multiple cycles are integrated for the highest frequencies because this provides a larger output signal, resulting in a greater signal-to-noise ratio.

[0715] For relatively low frequencies, such as those below about 500 Hz, the integrator outputs can saturate with the common parameter. Therefore, for these frequencies, two integrators that are switched alternately are used. That is, while the first integrator is integrating, the second integrator is read by the ADC and then reset (cleared) at the end of the integration time of the first integrator to prepare it for integration. In this way, the signal can be integrated without gaps in the integration. This will contribute to Figure 45 The EIS circuitry shown adds a second integrator and associated timing control.

[0716] Stable cycle considerations

[0717] The above analysis applies to steady-state conditions, where the current waveform does not change between cycles. Due to the initial state of the capacitor, this condition is not met immediately after applying a sine wave to the resistor-capacitor (RC) network. The current phase starts at 0 degrees and then develops to a steady-state value. However, it is desirable to minimize measurement time to reduce current consumption and allow sufficient time for the DC sensor measurement (Isigs). Therefore, it is necessary to determine the number of cycles necessary to obtain a sufficiently accurate measurement.

[0718] The equation for a simple RC circuit consisting of a resistor in series with a capacitor is

[0719]

[0720] Solving the above equation for I(t) yields:

[0721]

[0722] Where V c0 is the initial value of the capacitor voltage, V m is the magnitude of the driving sine wave, and ω is the frequency in radians (2πf).

[0723] The first term contains the term that defines the unsteady-state condition. One way to speed up the stabilization of the system is to make the first term equal to 0, which can be done, for example, by setting

[0724] or

[0725] Although this may not be necessary in practice, it is possible to set the initial phase of the excitation sine wave to jump immediately from the DC steady-state point to V cinit This technique can be evaluated for specific frequencies and expected phase angles to find possible time reductions.

[0726] The unsteady-state term is multiplied by an exponential function of time. This will determine how quickly steady-state conditions are reached. The RC value can be determined as a first-order approximation based on the impedance calculation information. Given the following equation:

[0727] and R=Zcosφ, from which we can get

[0728] For a sensor with a 5-degree phase angle at 100 Hz, this means a time constant of 18.2 milliseconds. For a settling of less than 1%, this means a settling time of approximately 85 milliseconds, or 8.5 cycles. On the other hand, for a sensor with a 65-degree phase angle at 0.10 Hz, this would mean a time constant of 0.75 seconds. For a settling of less than 1%, this would mean a settling time of approximately 3.4 seconds.

[0729] Thus, in the embodiment detailed above, the ASIC includes (at least) 7 electrode pads, 5 of which are designated as WORK electrodes (i.e., sensing electrodes or working electrodes, or WE), one of which is labeled COUNTER (i.e., counter electrode, or CE), and one of which is labeled REFERENCE (i.e., reference electrode, or RE). Counter amplifier 4321 (see Figure 42B ) can be connected to any of the COUNTER, REFERENCE, and / or assigned WORK pads in a programmable manner, as well as any combination thereof. As already mentioned, some embodiments may include, for example, more than five WEs. In this regard, embodiments may also relate to an ASIC that interfaces with more than five working electrodes.

[0730] It is important to note that for the ASIC described herein, each of the five working electrodes, counter electrode, and reference electrode can be individually and independently addressed. Thus, any one of the five working electrodes can be turned on and Isig (electrode current) can be measured, and any one can be turned off. In addition, any one of the five working electrodes can be operably connected / coupled to an EIS circuit system for measuring EIS-related parameters, such as impedance and phase. In other words, EIS can be selectively run on any one or more working electrodes. In addition, the corresponding voltage level of each of the five working electrodes can be independently programmed in amplitude and sign relative to the reference electrode. This has many applications, for example, changing the voltage on one or more electrodes to make the electrodes less sensitive to interference.

[0731] In embodiments where two or more working electrodes are used as redundant electrodes, the EIS techniques described herein can be used, for example, to determine which of the multiple redundant electrodes is operating optimally (...

Claims

1. A method for optimizing sensor glucose SG calculation of a glucose sensor, the glucose sensor being used to measure the glucose level in a user's body, the glucose sensor comprising a physical sensor electronics, a microcontroller, a counter electrode, and a working electrode, the method comprising: (a) periodically measuring the electrode current Isig signal of the working electrode by the physical sensor electronic device; (b) periodically executing an electrochemical impedance spectroscopy (EIS) program by the microcontroller to generate EIS-related data of the working electrode; (c) periodically measuring, by the physical sensor electronics, a potential difference Vcntr between the counter electrode and the working electrode; (d) periodically obtaining the blood glucose (BG) value of the user; (e) determining whether the BG is valid, and if the BG is a valid BG, pairing the BG value with the Isig, the Vcntr, and the EIS related data; (f) determining, by the microcontroller, whether there is a calibration error; as well as (g) When there is no calibration error, the microcontroller calculates the SG value based on the paired BG, Isig, Vcntr and EIS related data.

2. The method of claim 1, further comprising displaying the SG value to the user.

3. The method according to claim 1, wherein: In step (e), the BG is paired with the first Isig measured after obtaining the BG value.

4. The method of claim 3, wherein the BG is further paired with a second Isig measured after the first Isig.

5. The method of claim 1, further comprising storing calibration information in a calibration buffer, wherein the calibration information comprises at most 15 calibration data sets, each of the data sets comprising a paired Isig-BG value and a timestamp of the BG value. The method of claim 5 , further comprising calculating a calibration ratio CR based on the information stored in the calibration buffer.

7. The method of claim 6, wherein the calculated CR is weighted based on the most recent calibration time and the time of each of the BGs in the calibration buffer.

8. The method of claim 1, further comprising determining, by the microcontroller, whether the calculated SG value is valid, and displaying the SG value to the user if the SG value is valid.

9. The method of claim 8, wherein if the SG value is below 10 mg / dL, it is ineffective.

10. The method according to claim 1, wherein: If there is a calibration error, the SG value is not displayed to the user.

11. The method of claim 1, further comprising: before step (g), if it is determined that a calibration error exists, repeating steps (d)-(f), and terminating the glucose sensor when two consecutive calibration errors have occurred.

12. The method of claim 1, further comprising calculating a noise signal level based on the Isig signal, and terminating the glucose sensor when the noise signal level exceeds a threshold within a predetermined time window.

13. The method according to claim 12, wherein the predetermined time window comprises three consecutive time periods, each of the time periods having a duration of two hours.

14. The method of claim 13, wherein the noise signal level calculation starts when the glucose sensor has been used for at least 24 hours.

15. The method of claim 12, wherein the glucose sensor is terminated without calibration.

16. The method of claim 1, further comprising monitoring a value of 1 kHz real impedance, and terminating the glucose sensor when the value of the 1 kHz real impedance exceeds a threshold.

17. The method of claim 16, wherein the glucose sensor is terminated when the 1 kHz real impedance exceeds 12,000 ohms.

18. The method of claim 16, wherein the glucose sensor is terminated when the 1 kHz real impedance exceeds 10,500 ohms for two hours or more.

19. The method of claim 1, further comprising calculating, by the microcontroller, a rate of change of SG over a period of time, and invalidating the calculated SG when the rate of change of SG exceeds a threshold.

20. The method of claim 1, further comprising calculating, by the microcontroller, a rate of change of Isig within a certain period of time, and invalidating the calculated SG when the rate of change of Isig exceeds a threshold value.

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