A method for designing inductor parameters and controlling switching frequency of a bidirectional DC converter based on a switched quasi-Z source

Abstract

The present invention relates to an inductance parameter design and a variable switching frequency control method based on a switch quasi-Z source bidirectional DC converter, and belongs to the field of parameter design and control technology of circuit topology in power electronics technology. In the prior art, there are problems of turn-on loss of the main power switch tube and reverse recovery loss of the synchronous rectifier tube in the switch quasi-Z source bidirectional DC converter. The method provided by the present invention is based on the switch quasi-Z source bidirectional DC converter, and designs the inductance parameters in the circuit according to the rated operating range and basic switching frequency of the converter, and calculates and adjusts the switching frequency according to the real-time operating conditions of the converter based on the traditional voltage closed-loop control method. By adopting the method provided by the present invention, the zero voltage turn-on of all power switch tubes in the converter and the quasi-zero current turn-off of the synchronous rectifier tube can be realized, the turn-on loss of all power switch tubes in the converter and the reverse recovery loss of the synchronous rectifier tube are eliminated, and the operating efficiency and power density of the converter are further improved.

Description

Technical Field

[0001] The present invention belongs to the technical field of parameter design and control technology of circuit topology in power electronics technology, and specifically relates to an inductance parameter design and a variable switching frequency control method based on a switching quasi-Z source bidirectional DC converter. Background Art

[0002] Compared with traditional fuel vehicles that use non-renewable fossil energy as a power source, electric vehicles use electricity as a power source, which is not only clean and environmentally friendly, but also has high energy conversion efficiency. In recent years, electric vehicles have gradually replaced traditional fuel vehicles and occupied an increasingly wide range of the automobile market. Among the related technologies of electric vehicles, the research on its energy source system has received widespread attention. The early electric vehicle energy source system only used power batteries as a single energy source. When the vehicle frequently switched working conditions, the sudden power was provided by the power battery, which greatly reduced the service life of the power battery. In the electric vehicle multi-source system composed of power batteries and supercapacitors, the power battery with high energy density only needs to provide low-frequency power to maintain a stable DC bus voltage, while the high-frequency power during load transients is absorbed and released by the supercapacitor with high power density. The voltage fluctuation range of the supercapacitor is large, so the bidirectional DC converter that interfaces the supercapacitor and the DC bus needs to have a wide voltage gain range to achieve voltage matching between the supercapacitor and the DC bus. In addition, the converter also needs to have a high operating efficiency to fully utilize the energy of the multi-source system. The switching quasi-Z-source bidirectional DC converter with a common ground between the low-voltage side and the high-voltage side has a simple topology structure, can achieve a wide voltage gain range, and has a low voltage stress on the power switch tube, so it is suitable for the power interface between the supercapacitor and the DC bus. However, under the traditional inductor design and control method, the power switch of the converter is hard-on, and there is a reverse recovery process of the diode inside the synchronous rectifier tube. This will not only produce higher switching losses, but also limit the increase in switching frequency. In order to further improve the operating efficiency and power density of the switching quasi-Z-source bidirectional DC converter, the hard switching problem of the power switch tube must be solved.

[0003] References to related patent applications:

[0004] Existing soft switching technologies include adding auxiliary circuits and coupling inductors. [1] realizes zero voltage turn-on of all power switches based on the traditional BUCK converter, but adds two additional power switch tubes, which will generate additional conduction losses. [2] invented a non-isolated soft switching DC converter with low input current ripple and high output voltage gain. The converter uses a soft switching auxiliary circuit to achieve zero voltage turn-on of the power switch tube and zero current turn-off of the diode. However, the auxiliary switch tube cannot achieve zero voltage turn-on, which may limit the further increase of the converter switching frequency. [3] adds an auxiliary circuit containing inductors, MOSFETs, and diodes to achieve soft switching of all power switch tubes in the continuous mode of the BUCK converter, but the added magnetic components will bring additional losses.

[0005] [1] Teng Hui, Fan Xiaobo, Ren Yuanhang. A soft-switching bidirectional DC / DC conversion circuit and converter [P]. Guangdong Province: CN108566092B, 2020-07-07.

[0006] [2] Wang Ping, Li Bo. Non-isolated low current ripple high voltage gain soft switching DC-DC converter [P]. Tianjin: CN111464028A, 2020-07-28.

[0007] [3] Lu Pengfei. A soft-switching Buck converter and its control method [P]. Guangdong Province: CN111224545A, 2020-06-02. Summary of the invention

[0008] In view of the defects existing in the prior art, the purpose of the present invention is to realize soft switching of a switching quasi-Z-source bidirectional DC converter within the full gain range and load range, improve its operating efficiency and power density, and thus improve the overall performance of an electric vehicle multi-source system.

[0009] To achieve the above objectives, the technical solution adopted by the present invention is: an inductance parameter design and variable switching frequency control method, comprising the following steps:

[0010] (1) According to the rated operating range of the converter, calculate the total equivalent parallel inductance that satisfies the zero voltage turn-on condition of all power switches in the full operating range at the basic switching frequency.

[0011] (2) According to the real-time operating conditions of the converter and the determined total equivalent parallel inductance, the switching frequency that satisfies the quasi-zero current shutdown of all synchronous rectifiers is calculated, and the switching frequency is adjusted based on the traditional voltage closed loop.

[0012] (3) Based on the variable switching frequency control, the specific values ​​of the inductance on the low-voltage side and the high-voltage side are calculated to make the inductance current on the low-voltage side continuous at the rated power level.

[0013] Further, in step (1), according to the basic switching frequency f s0 , Maximum boost gain M within the operating range max , Maximum high voltage side current I high_max , high voltage side voltage U high Design the inductor parameters. At the basic switching frequency, in order to make all power switches of the converter open at zero voltage in the full operating range, calculate the low-voltage side inductor L of the converter according to the following formula 1 With the high-voltage side inductor L 2 The equivalent parallel inductance L.

[0014]

[0015] Further, in step (2), based on the traditional voltage closed-loop control, according to the converter boost gain M, the high-voltage side current I high , the remaining current margin δ (δ>0) is used to calculate and adjust the switching frequency in real time. In order to make all the switch tubes in the converter turn on with zero voltage and the synchronous rectifier tube turn off with zero current, the switching frequency f is determined according to the following formula: s .

[0016]

[0017] Further, in step (3), according to the determined total equivalent parallel inductance L, current margin δ, and minimum boost gain M within the operating range, min , Maximum high voltage side current I high_max Calculate the low voltage side inductance L 1 With the high-voltage side inductor L 2 In order to achieve soft switching operation of the converter and keep the low-voltage side inductor current continuous at the rated power level, the low-voltage side inductor L is selected according to the following formula: 1 With the high-voltage side inductor L 2 The inductance value.

[0018]

[0019] The effects of the present invention are:

[0020] 1. By designing the inductance parameters, the converter can achieve zero voltage turn-on of all switches in the full operating range at a given switching frequency, thus avoiding the turn-on loss of the switches during the operation of the converter and improving the operating efficiency of the converter;

[0021] 2. By controlling the switching frequency, the switch tube in the converter can always operate in a triangle conduction mode when the operating conditions change, thereby achieving quasi-zero current shutdown of the synchronous rectifier tube, avoiding reverse recovery loss in the synchronous rectifier tube in the converter, and further improving the operating efficiency of the converter within the operating conditions;

[0022] 3. It retains the characteristics of low voltage stress and wide gain range of the switching quasi-Z-source bidirectional DC converter, does not require additional components and auxiliary circuits, reduces the switching loss of the converter in each switching cycle, and further improves the operating efficiency and power density of the converter. BRIEF DESCRIPTION OF THE DRAWINGS

[0023] Figure 1 Electric vehicle multi-source system architecture

[0024] Figure 2 Switching Quasi-Z-Source Bidirectional DC Converter Topology

[0025] Figure 3 Schematic diagram of the steps of the present invention

[0026] Figure 4 Control system block diagram

[0027] Figure 5 Switching Quasi-Z Source Bidirectional DC Converter Boost Mode Operation Mode (a) S 1 S 2 S 3 =000; (b) S 1 S 2 S 3 =100; (c) S 1 S 2 S 3 =000; (d) S 1 S 2 S 3 =011;

[0028] Figure 6 Boost mode power switch voltage and current experimental waveform (a) Main power tube Q 1 ; (b) Synchronous rectifier Q 2 ; (c) Synchronous rectifier Q 3 ;

[0029] Figure 7 Switching Quasi-Z Source Bidirectional DC Converter Buck Mode Operation Mode (a) S 1 S 2 S 3 =000; (b) S 1 S 2 S 3 =011; (c) S 1 S 2S 3 =000; (d) S 1 S 2 S 3 =100;

[0030] Figure 8 Buck mode power switch voltage and current experimental waveforms (a) Synchronous rectifier tube Q 1 ; (b) Main power tube Q 2 ; (c) Main power tube Q 3 ;

[0031] The main symbol names in the above figure are: P sc , P bat , P load Respectively represent the power of supercapacitor energy storage unit, power battery energy storage unit and bus load; U low , U high are the low-voltage side and high-voltage side voltages of the converter respectively; Q 1 , Q 2 , Q 3 is the power switch tube in the converter, S 1 , S 2 , S 3 Q 1 , Q 2 , Q 3 The gate signal, C s1 , C s2 , C s3 Q 1 , Q 2 , Q 3 The parasitic capacitance, C low , C high They are the low-voltage side and high-voltage side capacitors of the converter respectively; L 1 , L 2 are the inductances of the low-voltage side and high-voltage side of the converter respectively; i L1 、i L2 L 1 , L 2 The inductor current in the bidirectional working mode is Figure 5 , Figure 6 As shown in; U ref ,I high are the given values ​​of the high-voltage side voltage and the high-voltage side current of the converter respectively; D, f s are the duty cycle and switching frequency of the main power switch tube, u ds1 -u ds3 、i ds1 -i ds3 They are power switch Q 1 -Q 3 The voltage and current on. DETAILED DESCRIPTION

[0032] A specific embodiment of the present invention is described below in conjunction with the accompanying drawings, and the technical solution principle and effect of the present invention are further explained.

[0033] Multi-source systems for electric vehicles such as Figure 1 As shown, the switch quasi-Z source interfaces the supercapacitor with the DC bus to achieve bidirectional power flow and match the dynamically fluctuating supercapacitor voltage U low With stable DC bus voltage U high The topology of the switching quasi-Z source bidirectional DC converter is as follows: Figure 2 As shown, Q 2 , Q 3 The gate signal is the same as Q 1 Complementary and retain dead zone. In boost mode, Q 1 Working as the main power tube, Q 2 , Q 3 It is a synchronous rectifier tube; in buck mode, Q 2 , Q 3 Working as the main power tube, Q 1 It is a synchronous rectifier tube.

[0034] Combine the following Figure 3 The steps shown in the flowchart are Figure 4 The switching quasi-Z-source bidirectional DC converter control system shown in the figure specifically describes the steps of the embodiment of the present invention for the operating condition range shown in the following table.

[0035] Table 1 Operating conditions

[0036]

[0037] According to step (1), the maximum boost gain within the operating condition is first calculated from the low-voltage side voltage and the high-voltage side voltage range.

[0038] M max =6 (8)

[0039] From the rated operating power P n With high voltage side voltage U high Calculate the maximum high voltage side current

[0040]

[0041] In order to make all power switches of the converter open at zero voltage in the full range of operating conditions, the maximum boost gain, maximum high-voltage side current, basic switching frequency, and high-voltage side voltage are used to calculate f s0 =The value range of the total equivalent parallel inductance L of all switches turned on at zero voltage within the rated operating range at 36kHz

[0042]

[0043] According to the above formula, select L = 57.7μH, that is

[0044]

[0045] The principle of step (1) is as follows:

[0046] The zero voltage turn-on condition of the power switch is to provide a reverse discharge current for the parasitic capacitance of the switch before the driving signal. After the parasitic capacitance is completely discharged, the terminal voltage drops to zero, achieving zero voltage turn-on and avoiding turn-on loss. The working mode of the switch quasi-Z source bidirectional DC converter is analyzed. The synchronous rectifier is always turned on at zero voltage due to reverse conduction. The reference directions of the inductor current in the boost and buck operation modes are as follows: Figure 3 , Figure 4 As shown, in the boost mode, the main power switch tube Q 1 The conduction current i S1_boost and the minimum inductor current i L1_min 、i L2_min The relationship is as follows

[0047] i S1_boost =i L1_min +i L2_min (12)

[0048] In the buck mode, the main power switch tube Q 2 , Q 3 The conduction current i S2_buck 、i S3_buck They are

[0049]

[0050] Therefore, the zero voltage turn-on condition for all switches in the bidirectional working mode of the converter is that the sum of the minimum values ​​of the two inductor currents in the circuit is less than zero, that is,

[0051] i L1_min +i L2_min <0 (14)

[0052] The value range of the total equivalent parallel inductance L can be derived from the above formula:

[0053]

[0054] According to step (2), in order to make all the switch tubes in the converter turn on with zero voltage and the synchronous rectifier tubes turn off with zero current, the design is as follows: Figure 6The control system shown in the figure. The voltage closed-loop control stabilizes the high-voltage side voltage at 300V. By obtaining the input and output voltages and currents of the converter under different operating conditions, the switching frequency is calculated, where the current margin δ = 2A is retained.

[0055]

[0056] The switching frequency calculated by the above formula is input into the pulse width modulation module through a low-pass filter and step limiter to adjust the switching frequency of the power switch tube, so as to realize zero voltage turn-on of all switch tubes and quasi-zero current turn-off of all synchronous rectifier tubes within the operating range shown in Table 1. The adjustment range of the switching frequency is about 30kHz-180kHz.

[0057] The principle of step (2) is as follows:

[0058] Synchronous rectifier Q in boost mode 2 , Q 3 The shutdown current i S2_boost 、i S3_boost And the synchronous rectifier Q in buck mode 1 The shutdown current i S1_buck The relationship between the inductor current and

[0059]

[0060]

[0061] In order to ensure that the parasitic capacitance of the main power tube is completely discharged and ZVS is achieved, a certain current margin δ (δ>0) is retained. Then the following equation is established, and the synchronous rectifier tube is quasi-zero current turned off.

[0062] i L1_min +i L2_min +δ=0 (19)

[0063] The calculation expression of switching frequency can be derived from the above formula.

[0064] According to step (3), in order to ensure that when the converter works under variable switching frequency control, all switches are always turned on at zero voltage within the rated operating range, the synchronous rectifier is turned off at quasi-zero current, and the low-voltage side inductor current is continuous under the rated power level. According to the total equivalent parallel inductance L, the current margin δ, and the minimum boost gain M within the operating range, min , Maximum high voltage side current I high_max Calculate the inductance L 1 , L 2 The value range of

[0065]

[0066] Determine the inductance L from the above formula 1 =147μH,L 2 =95μH.

[0067] The principle of step (2) is as follows:

[0068] Under variable switching frequency control, the ripple θ of the inductor current on the low-voltage side of the converter is calculated. 1

[0069]

[0070] From the above formula, it can be seen that the ripple of the low-voltage side inductor current is inversely correlated with the voltage gain when the power level remains unchanged. In order to ensure the continuity of the low-voltage side current within the full gain range at the rated power level, the inductance value calculated by (16) must satisfy

[0071]

[0072] Since the total equivalent parallel inductance L has been determined, the inductance L can be obtained by combining 1 , L 2 The value range of .

[0073] The following is an explanation of the implementation effects of the embodiments of the present invention. According to the energy flow direction, the working modes of the converter in the step-up and step-down operation modes are explained respectively.

[0074] 1. Boost Mode

[0075] When the switching quasi-Z-source bidirectional DC converter operates in boost mode, the converter experiences four operating modes in one switching cycle. The corresponding topological current flow paths are as follows: Figure 5 shown. Figure 6 is the voltage and current waveform of the power switch. Before the following mode starts, the power switch Q 1 Shutdown, Q 2 , Q 3 Conductivity.

[0076] Mode 1: When S 1 S 2 S 3 =000, the power switch Q 2 , Q 3 Quasi-zero current shutdown, parasitic capacitance C s2 , C s3 Charging, while C s1 Discharge, the current flow path of the topology is as follows Figure 5 (a) shows the energy storage capacitor C in this mode. 1 , C 2 , C high The voltage of C s2 , Cs3 Charging is completed, i.e. Q 2 , Q 3 Completely shut down, C s1 Fully discharged, Q 1 The internal freewheeling diode is turned on.

[0077] Mode 2: When S 1 S 2 S 3 =100, the power switch Q 1 Zero voltage turn-on, inductance L 1 The energy storage is charged by the low-voltage DC source, and the current flow path of the topology is as follows Figure 5 (b) is shown. Inductance L 2 and capacitor C 2 By C 1 Charging, the high-voltage side load is charged by the high-voltage side capacitor C high Energy supply.

[0078] Mode 3: When S 1 S 2 S 3 =000, the power switch Q 1 Shutdown. Parasitic capacitance C s1 Charging, while C s2 , C s3 Discharge. When C s1 Charging is completed, C s2 , C s3 Fully discharged, Q 2 , Q 3 The internal freewheeling diode is turned on. The current flow path of the topology is as follows Figure 5 (c) as shown.

[0079] Mode 4: When S 1 S 2 S 3 =001, the power switch Q 2 , Q 3 Reverse zero voltage conduction, the current flow path of the topology is as follows Figure 5 (d) The DC source on the low voltage side is L 1 , C 1 Charging energy storage and C 2 The series connection provides energy for the high voltage side. 2 Discharge, Q 2 , Q 3 The conduction current of the inductor decreases gradually with the change of the inductor current. Before the end of this mode, Q 2 , Q 3 The freewheeling diode is turned off with zero current.

[0080] 2. Buck Mode

[0081] When the switching quasi-Z-source bidirectional DC converter operates in the buck mode, the converter experiences four operating modes in one switching cycle. The corresponding topological current flow paths are as follows: Figure 7 shown. Figure 8 is the voltage and current waveform of the power switch. Before the following mode starts, the power switch Q 1 On, Q 2 , Q 3 Shut down.

[0082] Mode 1: When S 1 S 2 S 3 =000, the power switch Q 1 Quasi-zero current shutdown. Parasitic capacitance C s1 Charging, while C s2 , C s3 Discharge. The current flow path of the topology is as follows Figure 7 (a) shows the energy storage capacitor C in this mode. 1 , C 2 , C high The voltage of C s1 Charging is completed, i.e. Q 1 Completely shut down, C s2 , C s3 Fully discharged, Q 2 , Q 3 The internal freewheeling diode is turned on.

[0083] Mode 2: When S 1 S 2 S 3 =011, the power switch Q 2 , Q 3 Zero voltage turn-on, the current flow path of the topology is as follows Figure 7 (b) shows the capacitor C 1 is the inductance L 1 Charge and store energy and supply energy to the low voltage side load, capacitor C 2 and inductor L 2 Charged by a high voltage side DC source.

[0084] Mode 3: When S 1 S 2 S 3 =000, the power switch Q 2 , Q 3 The current flow path of the topology is as follows: Figure 7 (c) shows the parasitic capacitance C s2 , C s3 Charging, C s1 At the same time, discharge is performed. s2 , C s3 Charging is completed, i.e. Q2 , Q 3 Completely shut down, C s1 Fully discharged, Q 1 The internal freewheeling diode is turned on.

[0085] Mode 4: When S 1 S 2 S 3 =100, the power switch Q 1 Reverse zero voltage conduction, the current flow path of the topology is as follows Figure 7 (d) is shown. Inductance L 2 With capacitor C 2 Series connection C 1 Charging. Inductor L 1 Discharge, Q 1 The conduction current of the inductor decreases gradually with the change of the inductor current. Before the end of this mode, Q 1 The freewheeling diode is turned off with zero current.

[0086] The experimental results obtained in this embodiment can illustrate

[0087] 1. All switches in the converter are turned on at zero voltage in both boost and buck modes, thus avoiding the turn-on loss of the switches during converter operation;

[0088] 2. When the converter is in boost mode, the synchronous rectifier Q 2 , Q 3 The freewheeling diode is turned off at zero current; in the buck working mode, the synchronous rectifier Q 1 The freewheeling diode is turned off at zero current. Therefore, the synchronous rectifier achieves quasi-zero current turn-off, avoiding reverse recovery loss;

[0089] 3. No additional components or auxiliary circuits are added to the converter, and the steady-state characteristics of the converter are not changed, but the turn-on loss of the power switch and the reverse recovery loss during shutdown are avoided, thereby improving the operating efficiency and power density of the converter.

[0090] The present invention may be implemented in other specific forms without departing from its spirit or essential features. The described embodiments are considered to be illustrative and non-restrictive in all aspects, for example:

[0091] 1) Application topologies include various converters containing switched quasi-Z source networks;

[0092] 2) The duty cycle control link in the control system is not limited to the voltage closed loop;

[0093] 3) Current ripple index when designing inductor;

[0094] 4) Selection of various parameters, etc.

[0095] The scope of the invention is therefore indicated by the appended claims rather than the foregoing description, and all changes that come within the meaning and range of equivalent technical solutions of the claims are embraced within their scope.

Claims

1. An inductor parameter design and variable switching frequency control method based on a switching quasi-Z source bidirectional DC converter. The following steps are involved: (1) According to the rated operating range of the converter, calculate the total equivalent parallel inductance that satisfies the zero voltage turn-on condition of all power switches within the full operating range at the basic switching frequency; (2) According to the real-time operating conditions of the converter and the determined total equivalent parallel inductance, the switching frequency that satisfies the quasi-zero current shutdown of all synchronous rectifiers is calculated, and the switching frequency is adjusted based on the voltage closed loop; (3) Based on the variable switching frequency control, the specific values ​​of the inductance on the low-voltage side and the high-voltage side are calculated to make the inductance current on the low-voltage side continuous at the rated power level.

2. The inductor parameter design and variable switching frequency control method based on the switching quasi-Z source bidirectional DC converter according to claim 1, Features: In step (1), according to the basic switching frequency f s0 , Maximum boost gain M within the operating range max , Maximum high voltage side current I high_max , high voltage side voltage U high To design the inductor parameters, at the basic switching frequency, in order to make all power switches of the converter open at zero voltage in the full operating range, the low-voltage side inductor L of the converter is calculated according to the following formula: 1 With the high-voltage side inductor L 2 The equivalent parallel inductance L.

3. The inductor parameter design and variable switching frequency control method based on the switching quasi-Z source bidirectional DC converter according to claim 1, Features: In step (2), based on the voltage closed-loop control, according to the converter boost gain M, the high-voltage side current I high , the current margin δ is retained, and the switching frequency is calculated and adjusted in real time. In order to make all the switches in the converter turn on with zero voltage and the synchronous rectifier turn off with zero current, the switching frequency f is determined according to the following formula s .

4. The inductor parameter design and variable switching frequency control method based on the switching quasi-Z source bidirectional DC converter according to claim 1, Features: In step (3), according to the determined total equivalent parallel inductance L, current margin δ, and minimum boost gain M within the operating range, min , Maximum high voltage side current I high_max Calculate the low voltage side inductance L 1 With the high-voltage side inductor L 2 In order to achieve soft switching operation of the converter and keep the low-voltage side inductor current continuous at the rated power level, the low-voltage side inductor L is selected according to the following formula: 1 With the high-voltage side inductor L 2 The inductance value.

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