Sensor fusion method, system and device

Abstract

A single, optimal, fused sensor glucose value may be calculated based on respective sensor glucose values ​​of a plurality of redundant working electrodes (WE) of a glucose sensor. For each of the WEs, a respective electrochemical impedance spectroscopy (EIS) procedure may be performed to obtain a membrane resistance (Rmem) value for each WE. A noise value and a calibration factor (CF) value may be calculated for each WE, and respective fusion weights of Rmem, noise, and CF may be calculated for each WE. Thereafter, an overall fusion weight may be calculated based on the Rmem fusion weight, the noise fusion weight, and the CF fusion weight of the WE, so that a single, optimal, fused sensor glucose value may be calculated based on the respective overall fusion weight and the sensor glucose value for each of the plurality of redundant working electrodes.

Description

[0001] This application is a divisional application of the Chinese invention patent application with application number 201680082782.5, application date July 22, 2016, and invention name “Method, system and device for sensor fusion”. Technical Field

[0002] Embodiments of the present invention generally relate to subcutaneous and implantable sensor devices and, in particular embodiments, to methods, systems, and devices for providing a single, optimal, fused sensor value to a user. Background Art

[0003] In recent years, a variety of sensors have been developed for detecting and / or quantifying specific pharmaceutical agents or compositions in a patient's blood. These sensors enable patients and medical personnel to monitor physiological conditions within the patient's body. For example, a patient may wish to continuously monitor their blood sugar levels. Consequently, glucose sensors have been developed to obtain blood sugar readings in diabetic patients. These readings are very useful for monitoring and / or adjusting treatment regimens, which often include regular insulin administration to the patient.

[0004] Currently, a patient can measure their blood glucose (BG) using a BG measurement device (i.e., a blood glucose meter), such as a test strip, a continuous glucose measurement system (or continuous glucose monitor), or a hospital Hemacue. BG measurement devices measure a patient's BG level using various methods, such as a blood sample from the patient, a sensor that contacts bodily fluids, an optical sensor, an enzymatic sensor, or a fluorescent sensor. Once the BG measurement device has generated a BG measurement value, the measurement value is displayed on the BG measurement device.

[0005] Current continuous glucose measurement system includes subcutaneous (or short-term) sensor and implantable (or long-term) sensor.Sensor has been applied to telemetry characteristic monitor system.For example, described in the United States Patent (USP) the 6th,809,653 of common assignee (its entire content is incorporated herein by reference), the telemetry system using electrochemical sensor includes remote data receiving device, the sensor for generating the signal indicating user characteristic and the transmitting device for processing the signal received from the sensor and wirelessly transmitting the processed signal to the remote data receiving device.Data receiving device can be characteristic monitor, provides data to the data receiver, RF program device, drug delivery device (for example, insulin pump) etc. of another device.

[0006] No matter data receiving device (such as glucose monitor), sending device and sensor (such as glucose sensor) are wireless communication or are connected to communicate by wire, the characteristic monitoring system of the above type is only put into practical application after it has been calibrated based on the unique characteristics of individual users. According to the current state of the art, users need to carry out external calibration to the sensor. More specifically, taking diabetic patients as example, the latter need to use finger puncture blood glucose meter to read two to four times a day on average within the time period of using the characteristic monitor system. Each time blood needs to be sampled from the user's fingertip and analyzed by the blood glucose meter to provide the user's real-time blood glucose level. The user then inputs this data into the glucose monitor, as the user's current blood glucose level for calibrating the glucose monitoring system.

[0007] However, this external calibration has disadvantages for a variety of reasons. For example, blood glucose meters are not completely accurate and include inherent margins of error. Furthermore, even if a blood glucose meter were completely accurate, it could be susceptible to misuse. For example, if a user has handled candy or another sugary substance immediately prior to performing a finger prick, and some sugar is stuck to the user's finger, the blood glucose analysis will produce an inaccurate indication of blood glucose levels. Furthermore, blood glucose meters incur a cost, not to mention the pain and discomfort associated with each finger prick.

[0008] The current state of the continuous glucose monitoring (CGM) field is largely auxiliary, that is, the readings provided by CGM devices (including, for example, implantable or subcutaneous sensors) cannot be used to make clinical decisions without a reference value. In turn, the reference value must be obtained by finger pricking using, for example, a BG meter. The reference value is needed because the amount of information available from the sensor / detection element is very limited. Specifically, the only few pieces of information currently provided by the detection element for processing are the raw sensor value (i.e., sensor current or Isig) and the voltage. Therefore, during analysis, if the raw sensor signal looks abnormal (e.g., if the signal decreases), then the only way to distinguish between a sensor failure and a physiological change in the user / patient's body (i.e., a change in glucose level in the body) is to obtain a reference glucose value by finger pricking. As is known, reference finger pricking is also used to calibrate the sensor.

[0009] The art has explored multiple approaches to eliminate or at least minimize the number of fingertip punctures required to calibrate and evaluate sensor status. However, considering the number and complexity of most sensor failure modes, no satisfactory solution has yet been found. At most, diagnostic methods based on direct evaluation of Isig or comparison of multiple Isigs (e.g., from redundant and / or orthogonal redundancy, sensors and / or electrodes) have been developed. In either case, since Isig follows the level of glucose in the body, it is, by definition, not independent of the analyte. Thus, Isig itself is not a reliable source of information for sensor diagnostics, nor is it a reliable predictor of continuous sensor performance.

[0010] Another limitation that has existed in the prior art to date has been the lack of sensor electronics that can not only operate the sensor but also perform real-time sensor and electrode diagnostics, including for redundant electrodes, while managing the sensor's power supply. Granted, the concept of electrode redundancy has been around for some time. However, until now, there has been little success in using electrode redundancy to not only obtain more than one reading at a time but also to assess the relative condition of the redundant electrodes, the overall reliability of the sensor, and how often a calibration reference value is needed (if necessary), and to deliver a single, optimal, fused sensor value to the user. Summary of the Invention

[0011] According to an embodiment of the present invention, a method for calculating a single, fused sensor glucose value based on the respective sensor glucose values ​​of multiple redundant working electrodes of a glucose sensor includes: performing a respective electrochemical impedance spectroscopy (EIS) procedure on each of the multiple redundant working electrodes to obtain a membrane resistance (Rmem) value of each of the multiple working electrodes; calculating the corresponding Rmem fusion weight of each of the multiple working electrodes based on the corresponding Rmem value of each of the multiple working electrodes; measuring the noise value of each of the multiple working electrodes; calculating the corresponding noise fusion weight of each of the multiple working electrodes based on the corresponding noise value of each of the multiple working electrodes; measuring the calibration factor (CF) value of each of the multiple working electrodes; calculating the corresponding CF fusion weight of each of the multiple working electrodes based on the corresponding CF value of each of the multiple working electrodes; for each of the multiple working electrodes, calculating the overall fusion weight based on the Rmem fusion weight, noise fusion weight and CF fusion weight of the electrode; and calculating a single, fused sensor glucose value based on the corresponding overall fusion weight and the respective sensor glucose value of each of the multiple redundant working electrodes.

[0012] According to another embodiment of the present invention, a program code storage device includes: a computer-readable medium and a computer-readable program code stored on the computer-readable medium, the computer-readable program code having instructions that, when executed, cause a physical microcontroller to perform a method for calculating a single, fused sensor glucose value based on respective sensor glucose values ​​of a plurality of redundant working electrodes of a glucose sensor by: performing a respective electrochemical impedance spectroscopy (EIS) procedure on each of the plurality of redundant working electrodes to obtain a membrane resistance (Rmem) value of each of the working electrodes; calculating a corresponding Rmem fusion weight for each of the plurality of working electrodes based on the corresponding Rmem value of each of the plurality of working electrodes; Obtain a noise value for each of the plurality of working electrodes; calculate a corresponding noise fusion weight for each of the plurality of working electrodes based on the corresponding noise value for each of the plurality of working electrodes; obtain a calibration factor (CF) value for each of the plurality of working electrodes; calculate a corresponding CF fusion weight for each of the plurality of working electrodes based on the corresponding CF value for each of the plurality of working electrodes; for each of the plurality of working electrodes, calculate an overall fusion weight based on the Rmem fusion weight, noise fusion weight and CF fusion weight of the electrode; calculate a single, fused sensor glucose value based on the corresponding overall fusion weight for each of the plurality of working electrodes and the respective sensor glucose values. BRIEF DESCRIPTION OF THE DRAWINGS

[0013] Embodiments of the present invention will be described in detail with reference to the accompanying drawings, wherein like reference numerals denote like components throughout the figures.

[0014] Figure 1 is a perspective view of a subcutaneous sensor insertion assembly and a block diagram of sensing electronics according to an embodiment of the present invention.

[0015] Figure 2A A substrate is shown having two sides, a first of which contains electrode structures and a second of which contains electronic circuitry.

[0016] Figure 2B The figure shows the overall block diagram of the circuit used to detect the sensor output.

[0017] Figure 3 A block diagram of a sensing electronics device and a sensor including a plurality of electrodes according to an embodiment of the present invention is illustrated.

[0018] Figure 4 An alternative embodiment of the present invention is illustrated that includes a sensor and sensor electronics according to an embodiment of the present invention.

[0019] Figure 5An electronic block diagram of sensor electrodes and voltages applied to the sensor electrodes according to an embodiment of the present invention is illustrated.

[0020] Figure 6A A method of applying a pulse during a stabilization time range to reduce the stabilization time range according to an embodiment of the present invention is illustrated.

[0021] Figure 6B A method of stabilizing a sensor according to an embodiment of the present invention is illustrated.

[0022] Figure 6C The diagram illustrates the utilization of feedback information in stabilizing a sensor according to an embodiment of the present invention.

[0023] Figure 7 The function of a stabilized sensor according to an embodiment of the present invention is illustrated.

[0024] Figure 8A A block diagram of a sensor electronics device and a sensor according to one embodiment of the present invention is shown, wherein the sensor electronics device includes a voltage generating device.

[0025] Figure 8B A voltage generating device embodying this embodiment of the present invention is shown in the figure.

[0026] Figure 8C The voltage generating device is shown to generate two voltage values ​​to implement the embodiment of the present invention.

[0027] Figure 8D A voltage generating apparatus having three voltage generating systems according to an embodiment of the present invention is illustrated.

[0028] Figure 9A A sensing electronics device including a microcontroller for generating voltage pulses according to an embodiment of the present invention is illustrated.

[0029] Figure 9B A sensor electronics device including an analysis module according to an embodiment of the present invention is illustrated.

[0030] Figure 10 A block diagram of a sensor system including hydrated electronics according to an embodiment of the present invention is illustrated.

[0031] Figure 11 An embodiment of the present invention is illustrated that includes a mechanical switch to help determine hydration time.

[0032] Figure 12 A method for detecting hydration according to an embodiment of the present invention is illustrated.

[0033] Figure 13A A method of hydrating a sensor according to an embodiment of the present invention is illustrated.

[0034] Figure 13B Other methods for verifying sensor hydration according to embodiments of the present invention are illustrated.

[0035] Figure 14A 、 Figure 14B and Figure 14C A combined approach to sensor hydration and sensor stabilization according to an embodiment of the present invention is illustrated.

[0036] Figure 15A Illustrated is an EIS-based analysis of a system responding to the application of a periodic AC signal in accordance with an embodiment of the present invention.

[0037] Figure 15B A known circuit model for electrochemical impedance spectroscopy is illustrated.

[0038] Figure 16A An example of a Nyquist plot according to an embodiment of the present invention is illustrated, wherein an AC voltage plus a DC voltage (DC bias) is applied to a working electrode at a selected frequency spectrum of 0.1 Hz to 1000 MHz.

[0039] Figure 16B Another example of a Nyquist plot is shown with a straight line fit for relatively low frequencies and an intercept close to the real impedance value at relatively high frequencies.

[0040] Figure 16C and Figure 16D An infinite glucose sensor and a finite glucose sensor are shown, respectively, in response to a sinusoidal working potential.

[0041] Figure 16E Bode plots of amplitudes according to embodiments of the present invention are shown.

[0042] Figure 16F Bode plots of phase according to an embodiment of the present invention are shown.

[0043] Figure 17 Graphed is a Nyquist plot of sensor impedance as it changes with sensor age, according to an embodiment of the present invention.

[0044] Figure 18 A method for applying EIS technology in sensor stabilization and detecting sensor age according to an embodiment of the present invention is illustrated.

[0045] Figure 19 A timeline for performing an EIS procedure according to an embodiment of the present invention is illustrated.

[0046] Figure 20A method for detecting and repairing a sensor using an EIS procedure in conjunction with remedial measures according to an embodiment of the present invention is illustrated.

[0047] Figure 21A and Figure 21B An example of sensor remediation according to an embodiment of the present invention is illustrated.

[0048] Figure 22 The Nyquist curve of a normally functioning sensor is shown, wherein the Nyquist slope gradually increases and the intercept gradually decreases as the sensor wearing time elapses.

[0049] Figure 23A Shown are the raw current signals (Isig) from two redundant working electrodes and the real impedance of each electrode at 1 kHz, according to an embodiment of the present invention.

[0050] Figure 23B Shown Figure 23A Nyquist plot of the first working electrode (WE1).

[0051] Figure 23C Shown Figure 23A Nyquist plot of the second working electrode (WE2).

[0052] Figure 24 An example of signal dropout for two redundant working electrodes and the respective real impedances of the electrodes at 1 kHz is illustrated in accordance with an embodiment of the present invention.

[0053] Figure 25A The graph illustrates the substantial glucose independence of the real impedance, imaginary impedance, and phase at relatively high frequencies for a properly functioning glucose sensor according to an embodiment of the present invention.

[0054] Figure 25B Illustrative examples of different glucose dependency levels of real impedance at relatively low frequencies according to embodiments of the present invention are shown.

[0055] Figure 25C Illustrative examples of different glucose dependency levels of phase at relatively low frequencies according to embodiments of the present invention are shown.

[0056] Figure 26 Shown are the trends of 1 kHz real impedance, 1 kHz imaginary impedance, and relatively high frequency phase as glucose sensor sensitivity is lost due to hypoxia at the sensor insertion site, according to an embodiment of the present invention.

[0057] Figure 27 Isig and phase are shown for simulating hypoxia under different glucose concentration conditions in vitro according to an embodiment of the present invention.

[0058] Figures 28A to 28C An example of EIS-based parameters with redundant working electrodes WE1 and WE2 and electrodes with loss of sensitivity due to oxygen deficiency according to an embodiment of the present invention is shown.

[0059] Figure 28D Shown Figures 28A to 28C Example of EIS-induced spike signal in the original Isig.

[0060] Figure 29 An example of sensitivity loss due to occlusion-induced hypoxia according to an embodiment of the present invention is shown.

[0061] Figures 30A to 30C An example of redundant working electrodes WE1 and WE2 with loss of sensitivity due to biofouling and EIS-based parameters of the electrodes is shown in accordance with an embodiment of the present invention.

[0062] Figure 30D Shown Figures 30A to 30C Example of the original Isig spike signal induced by EIS.

[0063] Figure 31 A diagnostic procedure for sensor fault detection according to an embodiment of the present invention is shown.

[0064] Figure 32A and Figure 32B Another diagnostic routine for sensor fault detection according to an embodiment of the present invention is shown.

[0065] Figure 33A A top-level flow chart is shown involving a current (Isig) based fusion algorithm according to an embodiment of the present invention.

[0066] Figure 33B A top-level flow chart is shown involving a sensor glucose (SG) based fusion algorithm according to an embodiment of the present invention.

[0067] Figure 34 The embodiment of the present invention is shown Figure 33B Details of the sensor glucose (SG) based fusion algorithm.

[0068] Figure 35 The embodiment of the present invention is shown Figure 33A Details of the current (Isig) based fusion algorithm.

[0069] Figure 36 is an illustration of a calibrated steady-state sensor according to an embodiment of the present invention.

[0070] Figure 37is an illustration of a sensor in a calibration transformation according to an embodiment of the present invention.

[0071] Figure 38A is an illustration of EIS-based dynamic slope (with slope adjustment) for sensor calibration according to an embodiment of the present invention.

[0072] Figure 38B A flow chart of EIS-assisted sensor calibration including low start-up detection according to an embodiment of the present invention is shown.

[0073] Figure 39 Shown are the sensor current (Isig) and 1 kHz impedance amplitude of a simulated interferent in close proximity to the sensor in vitro according to an embodiment of the present invention.

[0074] Figure 40A and Figure 40B Shown respectively Figure 39 Bode plots of phase and impedance in the simulation shown.

[0075] Figure 40C Shown Figure 39 The simulated Nyquist plot is shown.

[0076] Figure 41 Another in vitro simulation with interferents according to an embodiment of the present invention is shown.

[0077] Figure 42A and Figure 42B A block diagram of an ASIC according to an embodiment of the present invention is shown.

[0078] Figure 43 A potentiostat configuration for a sensor with redundant working electrodes is shown according to an embodiment of the present invention.

[0079] Figure 44 Shown with Figure 43 Equivalent AC circuit between the electrodes of the sensor in the potentiostat configuration is shown.

[0080] Figure 45 FIG. 4 is a main block diagram of an EIS circuit in an analog front-end IC of a glucose sensor according to an embodiment of the present invention.

[0081] Figures 46A to 46F Shows the Figure 45 The signal of the EIS circuit shown is simulated as a current of 0 degree phase with 0 degree phase multiplication.

[0082] Figures 47A to 47F Shows the Figure 45 The signal of the EIS circuit shown is simulated as a current with a 0 degree phase and a 90 degree phase multiplication.

[0083] Figure 48 A circuit model according to an embodiment of the present invention is shown.

[0084] Figures 49A to 49C An example of a circuit model according to an alternative embodiment of the present invention is shown.

[0085] Figure 50A is a Nyquist curve of an overlay equivalent circuit simulation according to an embodiment of the present invention.

[0086] Figure 50B yes Figure 50A A magnified view of the high-frequency part.

[0087] Figure 51 The Nyquist plot shows that Cdl increases in the direction of arrow A according to an embodiment of the present invention.

[0088] Figure 52 The Nyquist curve is shown as α increases in the direction of arrow A according to an embodiment of the present invention.

[0089] Figure 53 shows an embodiment of the present invention. p Nyquist curve that increases in the direction of arrow A.

[0090] Figure 54 The Nyquist curve shows that the Warburg admittance increases in the direction of arrow A according to an embodiment of the present invention.

[0091] Figure 55 The Nyquist curve is shown as λ increases in the direction of arrow A according to an embodiment of the present invention.

[0092] Figure 56 The effect of the membrane capacitance on the Nyquist curve according to an embodiment of the present invention is shown.

[0093] Figure 57 The Nyquist plot shows that the membrane resistance increases in the direction of arrow A according to an embodiment of the present invention.

[0094] Figure 58 The Nyquist plot shows that Rsol increases in the direction of arrow A according to an embodiment of the present invention.

[0095] Figures 59A to 59C Shown are changes in EIS parameters associated with circuit elements during startup and calibration according to an embodiment of the present invention.

[0096] Figures 60A to 60C Shown are changes in a set of different EIS parameters associated with circuit elements during startup and initialization, according to an embodiment of the present invention.

[0097] Figures 61A to 61C Another different set of EIS parameter changes associated with circuit elements during startup and initialization according to an embodiment of the present invention are shown.

[0098] Figure 62 Shown are the EIS responses of multiple electrodes according to embodiments of the present invention.

[0099] Figure 63 The Isig calibration of the Nyquist curve affected by adding glucose is shown according to an embodiment of the present invention.

[0100] Figure 64 The effect of oxygen (Vcntr) response on the Nyquist plot is shown according to an embodiment of the present invention.

[0101] Figure 65 The shift in the Nyquist plot due to temperature change according to an embodiment of the present invention is shown.

[0102] Figure 66 The relationship between Isig and blood glucose according to an embodiment of the present invention is shown.

[0103] Figures 67A to 67B The sensor is shown moved down according to an embodiment of the present invention.

[0104] Figure 68 The increase in membrane resistance during sensitivity loss according to an embodiment of the present invention is shown.

[0105] Figure 69 The drop in Warburg admittance during sensitivity loss according to an embodiment of the present invention is shown.

[0106] Figure 70 A calibration curve according to an embodiment of the present invention is shown.

[0107] Figure 71 It is shown that a higher frequency semicircle becomes apparent on the Nyquist plot according to an embodiment of the present invention.

[0108] Figure 72A and Figure 72B Vcntr rail and Cdl reduction are shown according to an embodiment of the present invention.

[0109] Figure 73 The change in the slope of the calibration curve according to an embodiment of the present invention is shown.

[0110] Figure 74 The change in the length of the Nyquist curve according to an embodiment of the present invention is shown.

[0111] Figure 75 Shown Figure 74 Zoomed-in views of the lower and higher frequency regions of the Nyquist plot.

[0112] Figure 76A and Figure 76B The combined effect of increased membrane resistance, reduced Cdl, and the Vcnr rail is shown according to an embodiment of the present invention.

[0113] Figure 77 Shown are the Cdl values ​​of two working electrodes according to an embodiment of the present invention.

[0114] Figure 78 Rp values ​​for two working electrodes according to an embodiment of the present invention are shown.

[0115] Figure 79 The combined effects of varying EIS parameters on the calibration curve according to an embodiment of the present invention are shown.

[0116] Figure 80 It is shown that according to the embodiment of the present invention, in the lower frequency region, the length of the Nyquist curve is longer in the presence of sensitivity loss.

[0117] Figure 81 is a flow chart of sensor self-calibration based on detection of sensitivity changes according to an embodiment of the present invention.

[0118] Figure 82 Graph illustrating a horizontal shift of the Nyquist curve due to sensitivity loss according to an embodiment of the present invention.

[0119] Figure 83 A method for developing an exploratory EIS indicator based on a Nyquist curve according to an embodiment of the present invention is shown.

[0120] Figure 84 The relationship between Rm and the calibration factor according to an embodiment of the present invention is shown.

[0121] Figure 85 The relationship between Rm and normalized Isig according to an embodiment of the present invention is shown.

[0122] Figure 86 Isig curves as a function of time are shown for various glucose levels according to an embodiment of the present invention.

[0123] Figure 87 Shown are Cdl curves as a function of time for various glucose levels according to an embodiment of the present invention.

[0124] Figure 88The embodiment of the present invention is shown Figure 86 The second inflection point of the curve.

[0125] Figure 89 The embodiment of the present invention is shown Figure 88 The peak value in corresponds to the second inflection point of Rm.

[0126] Figure 90 An example of the relationship between the calibration factor (CF) and Rmem+Rsol according to an embodiment of the present invention is shown.

[0127] Figure 91A is a graph showing in vivo results based on MARD of all active BGs within approximately the first 8 hours of sensor life, according to an embodiment of the present invention.

[0128] Figure 91B is a graph showing the median ARD number based on all valid BGs within approximately the first 8 hours of sensor life, in accordance with an embodiment of the present invention.

[0129] Figures 92A to 92C A calibration factor adjustment according to an embodiment of the present invention is shown.

[0130] Figures 93A to 93C A calibration factor adjustment according to an embodiment of the present invention is shown.

[0131] Figures 94A to 94C A calibration factor adjustment according to an embodiment of the present invention is shown.

[0132] Figure 95 An illustrative example of initial decay in Cd1 according to an embodiment of the present invention is shown.

[0133] Figure 96 The effect of removing the non-Faraday current on Isig according to an embodiment of the present invention is shown.

[0134] Figure 97A The calibration factors for two working electrodes before removal of non-Faradaic current are shown, in accordance with an embodiment of the present invention.

[0135] Figure 97B The calibration factors for two working electrodes after removal of non-Faradaic current are shown, in accordance with an embodiment of the present invention.

[0136] Figure 98A and Figure 98B The effect of removing the non-Faradaic current on the MARD according to an embodiment of the present invention is shown.

[0137] Figure 99 is a graph showing the change in double layer capacitance over time according to an embodiment of the present invention.

[0138] Figure 100 The shift of Rmem+Rsol and the appearance of a higher frequency semicircle during sensitivity loss according to an embodiment of the present invention are shown.

[0139] Figure 101A A flow chart is shown for detecting sensitivity loss using combinational logic according to an embodiment of the present invention.

[0140] Figure 101B A flow chart is shown for detecting sensitivity loss using combinational logic according to an embodiment of the present invention.

[0141] Figure 102 An exemplary method of using the Nyquist slope as a marker for distinguishing new sensors from old sensors according to an embodiment of the present invention is shown.

[0142] Figures 103A to 103C Illustrative examples of Nyquist plots with different lengths for different sensor configurations according to embodiments of the present invention are shown.

[0143] Figure 104 Shown Figures 103A to 103C The Nyquist curve length of the sensor as a function of time.

[0144] Figure 105 A flowchart of clearing sensor data or stopping a sensor according to an embodiment of the present invention is shown.

[0145] Figure 106 A flow chart of sensor stopping according to an embodiment of the present invention is shown.

[0146] Figure 107 A flow chart of signal drop detection according to an embodiment of the present invention is shown.

[0147] Figure 108A shows Isig and Vcntr as a function of time according to an embodiment of the present invention; Figure 108B Shown is glucose as a function of time according to an embodiment of the present invention.

[0148] Figure 109A shows the calibration ratio as a function of time according to an embodiment of the present invention; Figure 109B Shown is glucose as a function of time according to an embodiment of the present invention.

[0149] Figure 110A and Figure 110B Shown is the trend of the calibration factor as a function of time according to an embodiment of the present invention.

[0150] Figure 111A flow chart of first day calibration (FDC) according to an embodiment of the present invention is shown.

[0151] Figure 112 A flow chart of EIS-based calibration according to an embodiment of the present invention is shown.

[0152] Figure 113 A flow chart of an existing calibration method is shown.

[0153] Figure 114 A calibration flow chart according to an embodiment of the present invention is shown.

[0154] Figure 115 FIG. 2 shows a calibration flow chart according to another embodiment of the present invention.

[0155] Figure 116 FIG. 4 shows a calibration flow chart according to yet another embodiment of the present invention.

[0156] Figure 117 FIG. 2 shows a calibration flow chart according to another embodiment of the present invention.

[0157] Figure 118 A comparison table of MARD values ​​calculated based on the embodiments of the present invention is shown.

[0158] Figure 119 A flowchart for calculating original fusion weights according to an embodiment of the present invention is shown.

[0159] Figure 120 A sensor glucose (SG) fusion logic diagram according to an embodiment of the present invention is shown. DETAILED DESCRIPTION

[0160] In the following description, reference is made to the accompanying drawings which form a part of the present invention and which illustrate several embodiments of the present invention. It should be understood that other embodiments may be used and changes may be made in structure and operation without departing from the scope of the present invention.

[0161] The present invention is described below with reference to example flow charts of methods, systems, devices, apparatuses, and programming and computer program products. It should be understood that each block in the example flow charts and combinations of blocks in the example flow charts can be implemented by programming instructions, which include computer program instructions (as can be described as any menu screen in the figure). These computer program instructions can be loaded into a computer or other programmable data processing device (for example, a controller, microcontroller or processor in a sensor electronic device) to produce a machine, so that the instructions executed on the computer or other programmable data processing device produce instructions for implementing the functions specified in the flow chart or multiple flow charts. These computer program instructions can also be stored in a computer-readable memory and can instruct the computer or other programmable data processing device to operate in a specific manner, so that the instructions stored in the computer-readable memory produce an article of manufacture including instructions for implementing the functions specified in the flow chart or multiple flow charts. The computer program instructions can also be loaded into a computer or other programmable data processing device to generate a series of operating steps that are executed in the computer or other programmable device to generate a computer-executable program, so that the instructions executed on the computer or other programmable device provide steps for implementing the functions specified in the flow chart or multiple flow charts and / or in the menus present herein. Programming instructions may also be stored in and / or implemented by electronic circuitry, including integrated circuits (ICs) and application specific integrated circuits (ASICs) used with sensor devices, apparatuses, and systems.

[0162] Figure 1 FIG is a perspective view of a subcutaneous sensor insertion assembly and a block diagram of the sensing electronics according to an embodiment of the present invention. Figure 1 As shown, the subcutaneous sensor assembly 10 is configured to subcutaneously place an active portion or the like of a flexible sensor 12 (see, for example, FIG. 2 ) at a selected site within the body of a user. The subcutaneous or percutaneous portion of the sensor assembly 10 includes a hollow, slotted insertion needle 14 and a cannula 16. The needle 14 is used to facilitate quick and easy subcutaneous placement of the cannula 16 at the subcutaneous insertion site. Inside the cannula 16 is a sensing portion 18 of the sensor 12, and one or more sensor electrodes 20 are exposed to the user's bodily fluids through a window 22 formed in the cannula 16. In an embodiment of the present invention, the one or more sensor electrodes 20 may include a counter electrode, a reference electrode, and one or more working electrodes. After insertion, the insertion needle 14 is withdrawn to leave the cannula 16 with the sensing portion 18 and sensor electrodes 20 in place at the selected insertion site.

[0163] In certain embodiments, the subcutaneous sensor assembly 10 facilitates the precise placement of a flexible, thin-film electrochemical sensor 12 of the type used to monitor specific blood parameters representative of the user's condition. The sensor 12 monitors glucose levels in the body and can be used in conjunction with an external or implantable automatic or semi-automatic drug infusion pump, such as those described in U.S. Pat. Nos. 4,562,751, 4,678,408, 4,685,903, or 4,573,994, to control insulin delivery to a diabetic patient.

[0164] In certain embodiments, the flexible electrochemical sensor 12 can be constructed using thin film masking technology to include an elongated thin film conductor embedded or encased between layers and films of a selected insulating material, such as a polyimide film or sheet. When the sensing portion 18 (or active portion) of the sensor 12 is placed subcutaneously at the insertion site, the sensor electrode 20 at the distal end of the sensing portion 18 is exposed through one of the insulating layers and is in direct contact with the patient's blood or other body fluids. The sensing portion 18 is connected to a connecting portion 24, which terminates in a conductive contact or similar element that is also exposed through one of the insulating layers. In alternative embodiments, other types of implantable sensors may also be used, such as chemical-based implantable sensors, optical-based implantable sensors, and the like.

[0165] As is known in the art, the connecting portion 24 and the contacts are generally adapted for direct wire electrical connection to a suitable monitor or sensing electronics device 100 for monitoring a condition of a user in response to signals from the sensor electrodes 20. Further description of this general type of flexible thin film sensor can be found in U.S. Patent No. 5,391,250, entitled "METHOD OFFABRICATING THIN FILM SENSORS," which is incorporated herein by reference. The connecting portion 24 can be conveniently electrically connected to the monitor or sensing electronics device 100, or can be conveniently electrically connected to the monitor or sensing electronics device 100 via a connecting block 28 (or the like), which is shown and described in U.S. Patent No. 5,482,473, entitled "FLEX CIRCUIT CONNECTOR," which is incorporated herein by reference. Thus, according to embodiments of the present invention, the subcutaneous sensor assembly 10 can be configured or formed to work with a wired or wireless characteristic monitoring system.

[0166] The sensor electrode 20 can be used in a variety of detection applications and can be configured in a variety of ways. For example, the sensor electrode 20 can be used in physiological parameter detection applications in which some type of biomolecule acts as a catalyst. For example, the sensor electrode 20 can be used in a glucose and oxygen sensor having glucose oxidase (GOx) that catalyzes a reaction with the sensor electrode 20. The sensor electrode 20, along with the biomolecule or some other catalyst, can be placed in a vascular or non-vascular environment within the human body. For example, the sensor electrode 20 and biomolecule can be placed in a vein and exposed to blood flow, or can be placed in the subcutaneous or peritoneal area of ​​the human body.

[0167] The monitor 100 may also be referred to as a sensor electronics device 100. The monitor 100 may include a power supply 110, a sensor interface 122, processing electronics 124, and data formatting electronics 128. The monitor 100 may be connected to the sensor assembly 10 by a cable 102 via a connector that is electrically connected to a connection block 28 of the connection portion 24. In alternative embodiments, the cable may be omitted. In this embodiment of the invention, the monitor 100 may include a suitable connector that connects directly to the connection portion 104 of the sensor assembly 10. The sensor assembly 10 may be modified to place the connector portion 104 in a different location, such as on top of the sensor assembly, to facilitate placement of the monitor 100 on the sensor assembly.

[0168] In one embodiment of the present invention, the sensor interface 122, processing electronics 124, and data formatting electronics 128 are formed as multiple separate semiconductor chips. However, alternative embodiments may combine the various different semiconductor chips into a single custom semiconductor chip or multiple custom semiconductor chips. The sensor interface 122 is connected to the cable 102 connected to the sensor assembly 10.

[0169] The power source 110 may be a battery. The battery may include three groups of silver oxide 357 cells. In alternative embodiments, different battery chemistries may be used, such as lithium-based chemistries, alkaline batteries, nickel metal hydride, and the like, and different numbers of cells may be used. The monitor 100 provides power to the sensor assembly via the power source 110 via the cable 102 and the cable connector 104. In an embodiment of the present invention, the power is a voltage provided to the sensor assembly 10. In an embodiment of the present invention, the power is a current provided to the sensor assembly 10. In an embodiment of the present invention, the power is a voltage provided to the sensor assembly 10 at a specific number of volts.

[0170] Figure 2A and Figure 2B An implantable sensor and electronic components for driving the implantable sensor according to an embodiment of the present invention are illustrated. Figure 2AA substrate 220 is shown having two sides, a first side 222 of the two sides containing electrode structures and a second side 224 of the two sides containing electronic circuitry. Figure 2A As can be seen in the figure, the first side 222 of the substrate includes two counter-electrode-working electrode pairs 240, 242, 244, and 246 located on opposite sides of a reference electrode 248. The second side 224 of the substrate includes circuitry. As shown, the electronic circuitry can be housed within a sealed housing 226, which provides a protective enclosure for the electronic circuitry. This allows the sensor substrate 220 to be inserted into a vascular environment or other environment where the electronic circuitry could be exposed to fluids. By sealing the electronic circuitry within the sealed housing 226, the electronic circuitry can operate without the risk of short circuiting due to the surrounding fluids. Figure 2A Also shown are contacts 228 to which the input and output lines of the electronic circuit can be connected. The electronic circuit itself can be manufactured in a variety of ways. According to an embodiment of the present invention, the electronic circuit can be manufactured into an integrated circuit using common industry techniques.

[0171] Figure 2B The following illustrates a general block diagram of a circuit for detecting sensor output according to an embodiment of the present invention. At least one pair of sensor electrodes 310 may be connected to a data converter 312, and the output of the data converter 312 may be connected to a counter 314. Counter 314 may be controlled by control logic 316. The output of counter 314 may be connected to a line interface 318. Line interface 318 may be connected to input / output lines 320 and may also be connected to control logic 316. Input / output lines 320 may also be connected to a power rectifier 322.

[0172] Sensor electrode 310 can be used in a variety of detection applications and can be configured in a variety of ways. For example, sensor electrode 310 can be used in physiological parameter detection applications in which some type of biomolecule acts as a catalyst. For example, sensor electrode 310 can be used in glucose and oxygen sensors having glucose oxidase (GOx) that catalyzes a reaction with sensor electrode 310. Sensor electrode 310, along with biomolecules or some other type of catalyst, can be placed in a vascular or non-vascular environment within the human body. For example, sensor electrode 310 and biomolecules can be placed in a vein and exposed to blood flow.

[0173] Figure 3A block diagram of a sensor electronic device and a sensor including multiple electrodes according to an embodiment of the present invention is illustrated. The sensor assembly or system 350 includes a sensor 355 and a sensor electronic device 360. The sensor 355 includes a counter electrode 365, a reference electrode 370, and a working electrode 375. The sensor electronic device 360 ​​includes a power supply 380, a regulator 385, a signal processor 390, a measurement processor 395, and a display / transmission module 397. The power supply 380 provides power (in the form of voltage, current, or voltage including current) to the regulator 385. The regulator 385 transmits the regulated voltage to the sensor 355. In one embodiment of the present invention, the regulator 385 transmits the voltage to the counter electrode 365 of the sensor 355.

[0174] The sensor 355 generates a sensor signal that represents the concentration of the physiological characteristic being measured. For example, the sensor signal may represent a blood glucose reading. In an embodiment of the present invention using a subcutaneous sensor, the sensor signal may represent the level of hydrogen peroxide in the subject. In an embodiment of the present invention using a blood sensor or a cranial sensor, the sensor signal represents the amount of oxygen being measured by the sensor. In an embodiment of the present invention using an implantable sensor or a long-term sensor, the sensor signal may represent the level of oxygen in the subject. The sensor signal is measured at the working electrode 375. In an embodiment of the present invention, the sensor signal may be a current measured at the working electrode. In an embodiment of the present invention, the sensor signal may be a voltage measured at the working electrode.

[0175] After the sensor signal is measured at the sensor 355 (e.g., working electrode), the signal processor 390 receives the sensor signal (e.g., measured current or voltage). The signal processor 390 processes the sensor signal and generates a processed sensor signal. The measurement processor 395 receives the processed sensor signal and calibrates the processed sensor signal using a reference value. In an embodiment of the present invention, the reference value is stored in a reference memory and provided to the measurement processor 395. The measurement processor 395 generates sensor measurements. The sensor measurements may be stored in a measurement memory (not shown). The sensor measurements may be sent to a display / transmission device to be displayed on a display in a housing with a sensing electronic device or transmitted to an external device.

[0176] The sensing electronic device 360 ​​can be a monitor including a display that displays physiological characteristic readings. The sensing electronic device 360 ​​can also be installed in a desktop computer, a pager, a television with communication capabilities, a laptop computer, a server, a network computer, a personal digital assistant (PDA), a portable phone with computer functions, an infusion pump with a display, a glucose sensor with a display, and / or a combination of an infusion pump / glucose sensor. The sensing electronic device 360 ​​can be installed in a BlackBerry device, a network device, a home network device, or a device connected to a home network.

[0177] Figure 4 An alternative embodiment of the present invention is illustrated that includes a sensor and sensor electronics according to an embodiment of the present invention. Sensor assembly or sensor system 400 includes sensor electronics 360 and sensor 355. The sensor includes a counter electrode 365, a reference electrode 370, and a working electrode 375. Sensor electronics 360 includes a microcontroller 410 and a digital-to-analog converter (DAC) 420. Sensor electronics 360 may also include a current-to-frequency converter (I / F converter) 430.

[0178] Microcontroller 410 includes software program code or programmable logic that, when executed, causes microcontroller 410 to transmit a signal to DAC 420, or the programmable logic to transmit a signal to DAC 420, wherein the signal represents a voltage level or voltage value to be applied to sensor 355. DAC 420 receives the signal and generates a voltage value at the level indicated by microcontroller 410. In embodiments of the present invention, microcontroller 410 may change the representation of the voltage level in the signal frequently or infrequently. For example, a signal from microcontroller 410 may instruct DAC 420 to apply a first voltage value for one second and a second voltage value for two seconds.

[0179] The sensor 355 may receive a voltage level or voltage value. In embodiments of the present invention, the counter electrode 365 may receive the output of an operational amplifier having a reference voltage and a voltage value from the DAC 420 as inputs. Application of the voltage level causes the sensor 355 to generate a sensor signal representing the concentration of the physiological characteristic being measured. In embodiments of the present invention, the microcontroller 410 may measure the sensor signal (e.g., a current value) from the working electrode. For example, the sensor signal measurement circuit 431 may measure the sensor signal. In embodiments of the present invention, the sensor signal measurement circuit 431 may include a resistor, and a current may flow through the resistor to measure the value of the sensor signal. In embodiments of the present invention, the sensor signal may be a current level signal, and the sensor signal measurement circuit 431 may be a current-to-frequency (I / F) converter 430. The I / F converter 430 may measure the sensor signal based on the current reading, convert it into a frequency-based sensor signal, and transmit the frequency-based sensor signal to the microcontroller 410. In embodiments of the present invention, the microcontroller 410 may more easily receive a frequency-based sensor signal than a non-frequency-based sensor signal. The microcontroller 410 receives the sensor signal, whether frequency-based or non-frequency-based, and determines a physiological characteristic value of the subject, such as a blood glucose level. The microcontroller 410 may include program code that, when executed or run, is capable of receiving the sensor signal and converting the sensor signal into a physiological characteristic value. In an embodiment of the present invention, the microcontroller 410 may convert the sensor signal into a blood glucose level. In an embodiment of the present invention, the microcontroller 410 may use measured values ​​stored in an internal memory to determine the blood glucose level of the subject. In an embodiment of the present invention, the microcontroller 410 may use measured values ​​stored in a memory external to the microcontroller 410 to help determine the blood glucose level of the subject.

[0180] After the physiological characteristic value is determined by the microcontroller 410, the microcontroller 410 may store the measured value of the physiological characteristic value for a period of time. For example, the blood glucose value may be sent from the sensor to the microcontroller 410 every one second or every five seconds, and the microcontroller may store the sensor measured value of the blood glucose reading for 5 minutes or 10 minutes. The microcontroller 410 may transmit the measured value of the physiological characteristic value to a display on the sensing electronic device 360. For example, the sensing electronic device 360 ​​may be a monitor including a display that provides the subject with a blood glucose reading. In an embodiment of the present invention, the microcontroller 410 may transmit the measured value of the physiological characteristic value to an output interface of the microcontroller 410. The output interface of the microcontroller 410 may transmit the measured value of the physiological characteristic value (e.g., blood glucose value) to an external device, such as an infusion pump, a combination infusion pump / blood glucose meter, a computer, a personal digital assistant, a pager, a network device, a server, a mobile phone, or any computing device.

[0181] Figure 5 An electronic block diagram illustrating sensor electrodes and voltages applied to the sensor electrodes according to an embodiment of the present invention is illustrated. Figure 5 In the illustrated embodiment of the present invention, an operational amplifier 530 or other servo control device can be connected to the sensor electrode 510 via a circuit / electrode interface 538. The operational amplifier 530, using feedback from the sensor electrode, attempts to maintain a specified voltage between the reference electrode 532 and the working electrode 534 (the specified voltage is the voltage that the DAC can expect to be applied) by adjusting the voltage at the counter electrode 536. Current can then flow from the counter electrode 536 to the working electrode 534. This current can be measured to detect the electrochemical reaction between the sensor electrode 510 and the sensor biomolecule, which is placed in the vicinity of the sensor electrode 510 and acts as a catalyst. Figure 5 The disclosed circuits may be used in long-term or implantable sensors, or in short-term or subcutaneous sensors.

[0182] In a long-term sensor embodiment, where glucose oxidase (GOx) is used as the catalyst in the sensor, current can flow from the counter electrode 536 to the working electrode 534 as long as oxygen is present in the vicinity of the enzyme and the sensor electrode 510. For example, if the voltage setpoint at the reference electrode 532 is maintained at approximately 0.5 volts, the amount of current flowing from the counter electrode 536 to the working electrode 534 is linearly related to the amount of oxygen present in the enzyme and the area surrounding the electrodes, with a uniform slope. Thus, improved accuracy in determining the oxygen content in the blood can be achieved by maintaining the reference electrode 532 at approximately 0.5 volts and using the current-voltage curve in this area to change the blood oxygen level. Different embodiments may use different sensors with biomolecules other than glucose oxidase and, thus, may have a voltage setpoint other than 0.5 volts at the reference electrode.

[0183] As discussed above, during the initial implantation or insertion of sensor 510, sensor 510 may produce inaccurate readings due to the subject adjusting the sensor and electrochemical byproducts produced by the catalyst used in the sensor. A stabilization period is required for many sensors to allow sensor 510 to provide accurate readings of the subject's physiological parameters. During this stabilization period, sensor 510 may not provide accurate blood glucose measurements. Sensor users and manufacturers would like to improve the stabilization period of sensors so that the sensors can be used quickly after insertion into the subject's body or subcutaneously.

[0184] In previous sensor electrode systems, the stabilization time or time range was one to three hours. To shorten the stabilization time or time range and improve the timeliness of sensor accuracy, the sensor (or the sensor's electrodes) can be subjected to multiple pulses rather than applying a pulse once and then applying another voltage. Figure 6AA method for applying multiple pulses during a stable time range to shorten the stable time range according to an embodiment of the present invention is illustrated. In an embodiment of the present invention, a voltage applying device applies a first voltage to an electrode for a first time or a first time period (600). In an embodiment of the present invention, the first voltage can be a DC constant voltage. This causes an anodic current to be generated. In an optional embodiment of the present invention, a digital-to-analog converter or another voltage source can apply a voltage to the electrode for a first time period. Anodic current means driving electrons toward the electrode to which the voltage is applied. In an embodiment of the present invention, instead of a voltage, the applying device can apply a current. In an embodiment of the present invention in which a voltage is applied to a sensor, after applying the first voltage to the electrode, the voltage regulator can wait (i.e., not apply the voltage) for a second time, time range, or time period (605). In other words, the voltage applying device waits until the second time period elapses. Not applying the voltage generates a cathode current, causing the electrode to which the voltage is not applied to acquire electrons. After applying the first voltage to the electrode for the first time period, not applying the voltage for the second time period, and the process is repeated several times (610). This can be referred to as an anodic and cathode cycle. In an embodiment of the present invention, the total number of repetitions in the stabilization method is three, i.e., the voltage is applied three times for a first time period, each time followed by no voltage for a second time period. In an embodiment of the present invention, the first voltage may be 1.07 volts. In an embodiment of the present invention, the first voltage may be 0.535 volts. In an embodiment of the present invention, the first voltage may be approximately 0.7 volts.

[0185] Repeated application of voltage and non-application of voltage causes the sensor (and electrode) to undergo an anode-cathode cycle. The anode-cathode cycle reduces the electrochemical byproducts produced by the patient's body reacting with the insertion of the sensor or the implantation of the sensor. In an embodiment of the present invention, the electrochemical byproducts lead to the generation of background current, which causes inaccurate measurements of the subject's physiological parameters. In an embodiment of the present invention, the electrochemical byproducts can be eliminated. Under other operating conditions, the electrochemical byproducts can be reduced or significantly reduced. A successful stabilization method allows the anode-cathode cycle to reach a balance, with electrochemical byproducts significantly reduced and background current minimized.

[0186] In an embodiment of the present invention, the first voltage applied to the electrode of the sensor may be a positive voltage. In an embodiment of the present invention, the first voltage applied may be a negative voltage. In an embodiment of the present invention, the first voltage may be applied to the working electrode. In an embodiment of the present invention, the first voltage may be applied to the counter electrode or the reference electrode.

[0187] In an embodiment of the present invention, the duration of the voltage pulse and the duration of no voltage application may be equal, for example, both are three minutes. In an embodiment of the present invention, the duration of voltage application or the duration of the voltage pulse may be different values, for example, the first time and the second time may be different. In an embodiment of the present invention, the first time period may be five minutes and the waiting time may be two minutes. In an embodiment of the present invention, the first time period may be two minutes and the waiting time (or second time range) may be five minutes. In other words, the duration period of applying the first voltage may be two minutes and not applying voltage lasts for five minutes. This time range is intended to be illustrative only and is not a limitation of the present invention. For example, the first time range may be two minutes, three minutes, five minutes or ten minutes, and the second time range may be five minutes, ten minutes, twenty minutes, and so on. The time ranges (for example, the first time and the second time) may depend on the unique characteristics of different electrodes, sensors and / or patient's physiological characteristics.

[0188] In embodiments of the present invention, more or fewer than three pulses may be used to stabilize the glucose sensor. In other words, the number of repetitions may be greater than three or less than three. For example, four voltage pulses may be applied to one of the electrodes (e.g., no voltage followed by a high voltage) or six voltage pulses may be applied to one of the electrodes.

[0189] For example, three consecutive pulses of 1.07 volts (followed by respective waiting times) may be sufficient for a sensor implanted subcutaneously. In embodiments of the present invention, three consecutive voltage pulses of 0.7 volts may be used. For sensors implanted in blood or cranial fluid (e.g., long-term or permanent sensors), the three consecutive pulses may have higher or lower voltage values ​​and may be negative or positive. Furthermore, more than three (e.g., five, eight, twelve) pulses may be used to generate an anodic-cathodic cycle between the anodic current and the cathodic current in any of the subcutaneous, blood, or cranial fluid sensors.

[0190] Figure 6B The method for stabilizing a sensor according to an embodiment of the present invention is illustrated. Figure 6B In the embodiment of the present invention shown, a voltage applying device may apply a first voltage to the sensor for a first time to initiate an anodic cycle (630) at the sensor electrode. The voltage applying device may be a DC power supply, a digital-to-analog converter, or a voltage regulator. After the first time period has elapsed, a second voltage is applied to the sensor for a second time to initiate a cathodic cycle (635) at the sensor electrode. For example, instead of applying no voltage, as in Figure 6AAs exemplified in the method of , a different voltage (different from the first voltage) is applied to the sensor for a second time frame. In an embodiment of the present invention, applying the first voltage for a first time and applying the second voltage for a second time are repeated several times (640). In an embodiment of the present invention, applying the first voltage for a first time and applying the second voltage for a second time can each be applied for a stable time frame, for example, 10 minutes, 15 minutes or 20 minutes, rather than being repeated several times. The stable time frame is the entire time frame of the stabilization process, for example, until the sensor is stabilized (and thus the electrode is stabilized). The benefits of this stabilization method are faster sensor operation, lower background current (in other words, suppressing some background current) and better glucose response.

[0191] In an embodiment of the present invention, the first voltage may be 0.535 volts and applied for five minutes, and the second voltage may be 1.070 volts and applied for two minutes. The first voltage of 0.535 volts may be applied for five minutes, and the second voltage of 1.070 volts may be applied for two minutes. The first voltage of 0.535 volts may be applied for five minutes, and the second voltage of 1.070 volts may be applied for two minutes. In other words, in this embodiment, the voltage pulse scheme is repeated three times. The pulse method can be varied so that the second time frame, for example, the time frame for applying the second voltage, can be extended from two minutes to five minutes, ten minutes, fifteen minutes, or twenty minutes. Furthermore, in this embodiment, after the three repeated applications, the rated operating voltage of 0.535 volts may be applied.

[0192] 1.070 and 0.535 volts are exemplary voltage values. Other voltage values ​​may be selected based on a variety of factors. These factors may include the type of enzyme used in the sensor, the membrane used in the sensor, the operating time of the sensor, the pulse length, and / or the pulse amplitude. Under some operating conditions, the first voltage may range from 1.00 to 1.09 volts, and the second voltage may range from 0.510 to 0.565 volts. In other operating embodiments, the range encompassing the first and second voltages may have a higher range, for example, 0.3 volts, 0.6 volts, or 0.9 volts, depending on the voltage sensitivity of the sensor electrode. Under other operating conditions, the voltage range may range from 0.8 volts to 1.34 volts, and other voltage ranges may range from 0.335 to 0.735 volts. Under other operating conditions, the range of the higher voltage may be narrower than the range of the lower voltage. For example, the range of the higher voltage may range from 0.9 volts to 1.09 volts, while the range of the lower voltage may range from 0.235 volts to 0.835 volts.

[0193] In an embodiment of the present invention, the first voltage and the second voltage may be positive voltages, alternatively, in other embodiments of the present invention, they may be negative voltages. In an embodiment of the present invention, the first voltage may be a positive voltage, and the second voltage may be a negative voltage, alternatively, the first voltage may be a negative voltage, and the second voltage may be a positive voltage. The first voltage may have a different voltage level at each repetition. In an embodiment of the present invention, the first voltage may be a DC constant voltage. In other embodiments of the present invention, the first voltage may be a ramp power supply, a sinusoidal voltage, a step voltage, or other commonly used voltage waveforms. In an embodiment of the present invention, the second voltage may be a DC constant voltage, a ramp voltage, a sinusoidal voltage, a step voltage, or other commonly used voltage waveforms. In an embodiment of the present invention, the first voltage or the second voltage may be an AC signal superimposed on a DC waveform. In an embodiment of the present invention, the first voltage may be one type of voltage, such as a ramp voltage, and the second voltage may be another type of voltage, such as a sinusoidal voltage. In an embodiment of the present invention, the first voltage (or the second voltage) may have a different waveform at each repetition. For example, if there are three cycles in the stabilization method, in the first cycle, the first voltage may be a ramp voltage, in the second cycle, the first voltage may be a constant voltage, and in the third cycle, the first voltage may be a sine wave voltage.

[0194] In an embodiment of the present invention, the duration of the first time range and the duration of the second time range may have the same value, alternatively, the duration value of the first time range and the duration value of the second time range may be different. For example, the duration of the first time range may be two minutes, while the duration of the second time range may be five minutes, and the number of repetitions may be three. As discussed above, the stabilization method may include multiple repetitions. In an embodiment of the present invention, during different repetitions of the stabilization method, the duration of the first time range in each repetition may change and the duration of the second time range in each repetition may change. For example, during the first repetition of the anode-cathode cycle, the first time range may be two minutes and the second time range may be five minutes. During the second repetition, the first time range may be one minute and the second time range may be three minutes. During the third repetition, the first time range may be three minutes and the second time range may be ten minutes.

[0195] In an embodiment of the present invention, a first voltage of 0.535 volts is applied to the electrodes of the sensor for two minutes to initiate the anodic cycle, followed by a second voltage of 1.07 volts applied to the electrodes for five minutes to initiate the cathodic cycle. The first voltage of 0.535 volts is then applied again for two minutes to initiate the anodic cycle, and a second voltage of 1.07 volts is applied to the sensor for five minutes. During the third repetition, 0.535 volts is applied for two minutes to initiate the anodic cycle, followed by 1.07 volts applied for five minutes. A voltage of 0.535 volts is then applied to the sensor for the duration of the sensor's actual operating time, e.g., while the sensor is providing readings of a subject's physiological characteristics.

[0196] Shorter duration voltage pulses can be used to Figure 6A and Figure 6B In an embodiment of the present invention, a shorter duration voltage pulse can be used to apply the first voltage, the second voltage, or both. In an embodiment of the present invention, the amplitude of the shorter duration voltage pulse for the first voltage is -1.07 volts, and the amplitude of the shorter duration voltage pulse for the second voltage is approximately half the higher amplitude, for example, -0.535 volts. Alternatively, the amplitude of the shorter duration pulse for the first voltage can be 0.535 volts, and the amplitude of the shorter duration pulse for the second voltage can be 1.07 volts.

[0197] In embodiments of the present invention using short pulses, the voltage may not be applied continuously throughout the first time period. Instead, the voltage applying device may transmit multiple short pulses during the first time period. In other words, multiple small width or short voltage pulses may be applied to the sensor electrode during the first time period. Each small width or short pulse may be several milliseconds wide. For example, the pulse width may be 30 milliseconds, 50 milliseconds, 70 milliseconds, or 200 milliseconds. These values ​​are intended to illustrate and not to limit the present invention. In embodiments of the present invention, for example, Figure 6A In the illustrated embodiment, these short pulses are applied to the sensor (electrode) for a first period of time, followed by no voltage being applied for a second period of time.

[0198] In an embodiment of the present invention, within the first time period, each short pulse may have the same duration. For example, each short voltage pulse may have a time width of 50 milliseconds and the pulse delay between each pulse may be 950 milliseconds. In this embodiment, if the measurement time of the first time range is two minutes, 120 short voltage pulses may be applied to the sensor. In an embodiment of the present invention, each short voltage pulse may have a different duration. In an embodiment of the present invention, each short voltage pulse may have the same amplitude. In an embodiment of the present invention, each short voltage pulse may have a different amplitude. By using short voltage pulses, rather than continuously applying voltage to the sensor, the same anode and cathode cycles can be produced and the sensor (e.g., electrode) consumes less total energy or charge over time. Relative to applying a continuous voltage to the electrode, the application of short voltage pulses uses less electricity because less energy is applied to the sensor (and thereby the electrode).

[0199] Figure 6C The use of feedback in stabilizing a sensor according to an embodiment of the present invention is illustrated. The sensor system may include a feedback mechanism to determine whether additional pulses are needed to stabilize the sensor. In an embodiment of the present invention, a sensor signal generated by an electrode (e.g., a working electrode) may be analyzed to determine whether the sensor signal is stable. A first voltage is applied to the electrode for a first time range to begin an anodic cycle (630). A second voltage is applied to the electrode for a second time range to begin a cathodic cycle (635). In an embodiment of the present invention, an analysis module may analyze the sensor signal (e.g., a current emitted by the sensor signal, a resistance at a specific point in the sensor, an impedance at a specific node in the sensor) and determine whether a threshold measurement value has been reached (e.g., by comparing with the threshold measurement value to determine whether the sensor is providing an accurate reading) (637). If the sensor reading is determined to be accurate, indicating that the electrode is stable (and therefore the sensor is stable) (642), additional first voltages and / or second voltages may not be applied. In an embodiment of the present invention, if stability is not achieved, then additional anodic / cathodic cycles are initiated by applying the first voltage to the electrode for a first time period (630) and then applying the second voltage to the electrode for a second time period (635).

[0200] In an embodiment of the present invention, the analysis module may be used after applying three anodic / cathodic cycles of the first voltage and the second voltage to the electrodes of the sensor. Figure 6C For example, the analysis module may be used after applying the first voltage and the second voltage once.

[0201] In an embodiment of the present invention, the analysis module can be used to measure the voltage generated after the current has flowed through the electrode or through both electrodes. The analysis module can monitor the voltage level at the electrode or the voltage level at the receiving electrode. In an embodiment of the present invention, if the voltage level is above a certain threshold, this may mean that the sensor is stable. In an embodiment of the present invention, if the voltage level drops below the threshold level, this may indicate that the sensor is stable and ready to provide a reading. In an embodiment of the present invention, current can be introduced to the electrode or flow through both electrodes. The analysis module can monitor the current level sent by the electrode. In this embodiment of the present invention, the analysis module is able to monitor the current if the current differs by an order of magnitude from the sensor signal current. If the current is above or below the current threshold, then this may indicate that the sensor is stable.

[0202] In an embodiment of the present invention, the analysis module may measure the impedance between two electrodes of the sensor. The analysis module may compare the impedance to a threshold impedance value or a target impedance value, and if the measured impedance is lower than the target impedance or the threshold impedance, then the sensor may be stable (and thus the sensor signal is stable). In an embodiment of the present invention, the analyte module may measure the resistance between two electrodes of the sensor. In this embodiment of the present invention, if the analysis module compares the resistance to a threshold resistance value or a target resistance value, and the measured resistance value is less than the threshold resistance value or the target resistance value, then the analysis module may determine that the sensor is stable and the sensor signal is usable.

[0203] Figure 7 The effect of the stable sensor according to an embodiment of the present invention is illustrated. Line 705 represents the blood glucose sensor reading of the glucose sensor when using the previous single pulse stabilization method. Line 710 represents the blood glucose reading of the glucose sensor when applying three voltage pulses (for example, three voltage pulses last for two minutes, and no voltage is applied for 5 minutes after each voltage pulse). X-axis 715 represents the amount of time. Points 720, 725, 730 and 735 represent the glucose readings measured using the finger puncture method and are subsequently input to the blood glucose meter. As shown in the figure, the previous single pulse stabilization method takes about 1 hour and 30 minutes to stabilize to the desired glucose reading, for example 100 units. In contrast, the three pulse stabilization method only takes about 15 minutes to stabilize the glucose sensor and produces a significantly improved stabilization time range.

[0204] Figure 8AA block diagram of a sensing electronic device and a sensor according to an embodiment of the present invention is illustrated, which includes a voltage generating device. The voltage generating or applying device 810 includes electronic components, logic or circuitry that generates voltage pulses. The sensing electronic device 360 ​​may also include an input device 820 for receiving reference values ​​and other useful data. In an embodiment of the present invention, the sensing electronic device may include a measurement memory 830 for storing sensor measurements. In this embodiment of the present invention, a power supply 380 may provide power to the sensing electronic device. The power supply 380 may supply power to a regulator 385, which provides a regulated voltage to the voltage generating or applying device 810. In the illustrated embodiment of the present invention, the connection terminal 811 represents a connection terminal that couples or connects the sensor 355 to the sensing electronic device 360.

[0205] exist Figure 8A In the embodiment of the present invention shown, voltage generation or application device 810 provides voltage (for example, first voltage or second voltage) to the input of operational amplifier 840. Voltage generation or application device 810 can also provide voltage to the working electrode 375 of sensor 355. The other input of operational amplifier 840 is connected to the reference electrode 370 of sensor. Voltage generation or application device 810 applies voltage to operational amplifier 840 to drive the voltage measured at counter electrode 365 to be close to or equal to the voltage applied to working electrode 375. In an embodiment of the present invention, voltage generation or application device 810 can be used for applying desired voltage between counter electrode and working electrode. This can occur by applying fixed voltage directly to counter electrode.

[0206] exist Figure 6A and Figure 6B In the embodiment of the present invention shown, the voltage generating device 810 generates a first voltage to be applied to the sensor in a first time range. The voltage generating device 810 sends the first voltage to the operational amplifier 840, which drives the voltage at the counter electrode 365 of the sensor 355 to reach the first voltage. In an embodiment of the present invention, the voltage generating device 810 can also transmit the first voltage directly to the counter electrode 365 of the sensor 355. Figure 6A In the illustrated embodiment of the present invention, the voltage generating device 810 then does not transmit the first voltage to the sensor 355 for a second time period. In other words, the voltage generating device 810 is turned off or switched off. The voltage generating device 810 can be programmed to continuously cycle between applying the first voltage and not applying the voltage a number of times or continuously cycle for a stable time period (e.g., for twenty minutes). Figure 8BThe voltage generating device of this embodiment of the present invention is illustrated. Voltage regulator 385 transmits the regulated voltage to voltage generating device 810. Control circuit 860 controls the closing and opening of switch 850. If switch 850 is closed, voltage is applied. If switch 850 is open, no voltage is applied. Timer 865 provides a signal to control circuit 860 to instruct control circuit 860 to turn switch 850 on or off. Control circuit 860 includes logic that can instruct the circuit to open and close switch 850 multiple times (to match the required number of repetitions). In an embodiment of the present invention, timer 865 can also send a stabilization signal to identify the end of the stabilization sequence, that is, the stabilization time range has expired.

[0207] In an embodiment of the present invention, the voltage generating device generates a first voltage for a first time range and generates a second voltage for a second time range. Figure 8C To illustrate an embodiment of the present invention, a voltage generating device generates two voltage values. In this embodiment, a two-position switch 870 is used. For example, if the first switch position 871 is turned on or closed by timer 865 commanding control circuit 860, voltage generating device 810 generates a first voltage for a first time frame. After applying the first voltage for the first time frame, the timer sends a signal to control circuit 860 indicating that the first time frame has elapsed, and control circuit 860 instructs switch 870 to move to a second position 872. When switch 870 is in second position 872, the regulated voltage enters a voltage step-down (buck) converter 880, which reduces the regulated voltage to a lower value. This lower value is then passed to operational amplifier 840 for a second time frame. After timer 865 signals control circuit 860 that the second time frame has elapsed, control circuit 860 returns switch 870 to the first position. This process continues until the desired number of repetitions is completed or the stabilization time frame has elapsed. In an embodiment of the present invention, the sensor sends the sensor signal 350 to the signal processor 390 after the sensor stabilization time frame has elapsed.

[0208] Figure 8D The following illustrates a voltage applying device 810 for applying voltage to the sensor in a more complex manner. The voltage applying device 810 may include a control device 860, a switch 890, a sine wave voltage generating device 891, a ramp voltage generating device 892, and a constant voltage generating device 893. In other embodiments of the present invention, the voltage applying device may generate an AC wave based on a DC signal or other various voltage pulse waveforms. Figure 8DIn the illustrated embodiment of the present invention, the control device 860 can cause the switch to move to one of three voltage generating systems: 891 (sine wave), 892 (ramp), and 893 (constant DC). This causes each of the voltage generating systems to generate a recognized voltage waveform. Under certain operating conditions, for example, when applying a sine wave pulse for three pulses, the control device 860 can cause the switch 890 to connect the voltage from the voltage regulator 385 to the sine wave voltage generator 891, thereby causing the voltage applying device 810 to generate a sine wave voltage. Under other operating conditions, for example, when a ramp voltage is applied to the sensor as the first voltage of the first of three pulses, a sinusoidal voltage is applied to the sensor as the first voltage of the second of the three pulses, and a constant DC voltage is applied to the sensor as the first voltage of the third of the three pulses, the control device 860 may cause the switch 890 to move, during a first time frame in the anode / cathode cycle, between connecting the voltage from the voltage generating or applying device 810 to the ramp voltage generating system 892, then to the sinusoidal voltage generating system 891, and then to the constant DC voltage generating system 893. In this embodiment of the invention, the control device 860 may also instruct or control the switch to connect one of the voltage generating subsystems to the voltage from the regulator 385 during a second time frame (e.g., during application of the second voltage).

[0209] Figure 9A An example of a sensor electronics device including a microcontroller for generating voltage pulses according to an embodiment of the present invention is shown. The advanced sensor electronics device may include a microcontroller 410 (see Figure 4 ), digital-to-analog converter (DAC) 420, operational amplifier 840, and sensor signal measurement circuit 431. In an embodiment of the present invention, the sensor signal measurement circuit may be a current-to-frequency (I / F) converter 430. Figure 9A In the embodiment of the present invention shown, software or programmable logic in the microcontroller 410 provides instructions to send a signal to the DAC 420, which in turn instructs the DAC 420 to output a specific voltage to the operational amplifier 840. Figure 9AAs shown by line 911 in FIG, the microcontroller 410 can also be instructed to output a specific voltage to the working electrode 375. As discussed above, applying a specific voltage to the operational amplifier 840 and the working electrode 375 can drive the voltage measured at the counter electrode to a specific voltage amplitude. In other words, the microcontroller 410 outputs a signal representing a voltage or voltage waveform to be applied to the sensor 355 (e.g., the operational amplifier 840 connected to the sensor 355). In an alternative embodiment of the present invention, a fixed voltage can be set by applying a voltage directly from the DAC 420 between the reference electrode and the working electrode 375. A similar result can also be achieved by applying a voltage to each electrode, where the difference between the voltages of each electrode is equal to the fixed voltage applied between the reference electrode and the working electrode. In addition, the fixed voltage can be set by applying a voltage between the reference electrode and the counter electrode. Under certain operating conditions, the microcontroller 410 can generate pulses of a specific amplitude, which the DAC 420 interprets as indicating that a voltage of a specific amplitude is to be applied to the sensor. After the first time frame, the microcontroller 410 (via a program or programmable logic) outputs a second signal that instructs the DAC 420 to output no voltage (for Figure 6A The sensor electronic device 360 ​​operates according to the method) or outputs a second voltage (for Figure 6B (For the sensing electronic device 360 ​​operating as described above), the microcontroller 410 repeats the following cycle after the second time frame has elapsed: sending a signal indicating that the first voltage is to be applied (for the first time frame) and then sending a signal indicating that no voltage is to be applied or that a second voltage is to be applied (for the second time frame).

[0210] Under other operating conditions, the microcontroller 410 may generate a signal to the DAC 420 that instructs the DAC 420 to output a ramp voltage. Under other operating conditions, the microcontroller 410 may generate a signal to the DAC 420 that instructs the DAC 420 to output a voltage that simulates a sine wave voltage. These signals may be incorporated into any of the pulse modulation methods discussed in the preceding paragraphs or earlier in this application. In an embodiment of the present invention, the microcontroller 410 may generate a series of instructions and / or pulses that the DAC 420 receives and interprets as indicating that a certain pulse sequence is to be applied. For example, the microcontroller 410 may send a series of instructions (via signals and / or pulses) instructing the DAC 420 to generate a constant voltage for the first repetition of a first time frame, a ramp voltage for the first repetition of a second time frame, a sine wave voltage for the second repetition of the first time frame, and a square wave having two values ​​for the second repetition of the second time frame.

[0211] Microcontroller 410 may include programmable logic or a program to continue this cycle for a stable time frame or a number of times. For example, microcontroller 410 may include counting logic to identify when the first time frame or the second time frame has elapsed. Furthermore, microcontroller 410 may include counting logic to identify when the stable time frame has elapsed. If any of the aforementioned time frames have elapsed, the counting logic may instruct the microcontroller to send a new signal or to stop sending a signal to DAC 420.

[0212] The use of the microcontroller 410 allows for multiple voltage amplitudes to be applied in multiple sequences and for multiple durations. In an embodiment of the present invention, the microcontroller 410 may include control logic or programming to instruct the digital-to-analog converter 420 to transmit a voltage pulse having an amplitude of approximately 1.0 volts for a first period of 1 minute, followed by a voltage pulse having an amplitude of approximately 0.5 volts for a second period of 4 minutes, and repeat this cycle four times. In an embodiment of the present invention, the microcontroller 420 may be programmed to signal the DAC 420 to apply voltage pulses of the same amplitude to each first voltage during each repetition. In an embodiment of the present invention, the microcontroller 410 may be programmed to signal the DAC 420 to apply voltage pulses of different amplitudes to each first voltage during each repetition. In this embodiment of the present invention, the microcontroller 410 may also be programmed to signal the DAC 420 to apply voltage pulses of different amplitudes to each second voltage during each repetition. For example, the microcontroller 410 may be programmed to send a signal to cause the DAC 420 to apply a first voltage pulse of approximately 1.0 volts in a first repetition, a second voltage pulse of approximately 0.5 volts in a second repetition, a first voltage of 0.7 volts and a second voltage of 0.4 volts in a second repetition, and a first voltage of 1.2 volts and a second voltage of 0.8 volts in a third repetition.

[0213] The microcontroller 410 can also be programmed to instruct the DAC 420 to apply multiple short voltage pulses for a first timeframe. In this embodiment of the invention, rather than applying a single voltage for the entire first timeframe (e.g., two minutes), multiple shorter pulses can be applied to the sensor. In this embodiment, the microcontroller 410 can also be programmed to instruct the DAC 420 to apply multiple short voltage pulses to the sensor for a second timeframe. For example, the microcontroller 410 can send a signal to cause the DAC to apply multiple short voltage pulses, where the short duration is 50 milliseconds or 100 milliseconds. Between these short pulses, the DAC can apply no voltage or a minimum voltage. The microcontroller can cause the DAC 420 to apply the short voltage pulses for a first timeframe, e.g., two minutes. The microcontroller 410 can then send a signal to cause the DAC to apply no voltage or to apply short voltage pulses to the sensor at a second voltage level for a second timeframe. For example, the second voltage can be 0.75 volts and the second timeframe can be five minutes. In an embodiment of the present invention, microcontroller 410 may send a signal to DAC 420 to cause DAC 420 to apply a voltage of different amplitude to each short pulse within a first time range and / or a second time range. In this embodiment of the present invention, microcontroller 410 may send a signal to DAC 420 to cause DAC 420 to apply a voltage amplitude with an undulating pattern to the short voltage pulses for the first time range or the second time range. For example, the microcontroller may send a signal or pulse instructing DAC 420 to apply thirty 20-millisecond pulses to the sensor within a first time range. Each of the thirty 20-millisecond pulses may have the same amplitude or a different amplitude. In this embodiment of the present invention, microcontroller 410 may instruct DAC 420 to apply the short pulses within a second time range or may instruct DAC 420 to apply a different voltage waveform within the second time range.

[0214] Although the disclosure of Figures 6 to 8 discloses the application of voltage, current can also be applied to the sensor to initiate the stabilization process. Figure 6BIn the illustrated embodiment of the invention, a first current may be applied within a first timeframe to initiate an anodic or cathodic response, and a second current may be applied within a second timeframe to initiate an opposing anodic or cathodic response. The application of the first and second currents may be repeated multiple times in succession or may last for a stable timeframe. In an embodiment of the invention, a first current may be applied within a first timeframe and a first voltage may be applied within a second timeframe. In other words, one of the anodic or cathodic cycles may be triggered by a current applied to the sensor and the other of the anodic or cathodic cycles may be triggered by a voltage applied to the sensor. As described above, the applied current may be a constant current, a ramped current, a step pulse current, or a sine wave current. In some operating conditions, the current is applied within the first timeframe as a series of short pulses.

[0215] According to an embodiment of the present invention, Figure 9B An example is given of sensors and sensor electronics that process feedback information using an analysis module during a stable period. Figure 9B The analysis module 950 is introduced into the sensor electronics 360. The analysis module 950 uses the feedback information from the sensor to determine whether the sensor is stable. In an embodiment of the present invention, the microcontroller 410 may include instructions or commands to control the DAC 420 so that the DAC 420 applies a voltage or current to a portion of the sensor 355. Figure 9B It is illustrated that the voltage or current can be applied between the reference electrode 370 and the working electrode 375. However, the voltage or current can be applied between the electrodes or directly to one of the electrodes, and the present invention is not limited to Figure 9B The application of voltage or current is shown by dashed line 955. Analysis module 950 can measure the voltage, current, resistance or impedance in sensor 355. Figure 9BWhile the measurement is illustrated as occurring at the working electrode 375, this should not be considered limiting, as other embodiments of the present invention may measure voltage, current, resistance, or impedance between the electrodes of the sensor or directly at the reference electrode 370 or the counter electrode 365. Analysis module 950 may receive the measured voltage, current, resistance, or impedance and compare the measured value to a stored value (e.g., a threshold value). Dashed line 956 indicates that analysis module 950 reads or acquires the measured value of the voltage, current, resistance, or impedance. Under some operating conditions, if the measured voltage, current, resistance, or impedance is above the threshold value, the sensor is stable and the sensor signal provides an accurate reading of the patient's physiological condition. Under other operating conditions, if the measured voltage, current, resistance, or impedance is below the threshold value, the sensor is stable. Under other operating conditions, analysis module 950 may verify that the measured voltage, current, resistance, or impedance is stable within a specified time frame (e.g., one or two minutes). This may indicate that sensor 355 is stable and that the sensor signal is transmitting an accurate measurement of the subject's physiological parameter (e.g., blood glucose level). After the analysis module 950 has determined that the sensor is stable and the sensor signal provides accurate measurements, the analysis module 950 may transmit a signal (e.g., a sensor stability signal) to the microcontroller 410 indicating that the sensor is stable and that the microcontroller 410 may begin using or receiving signals from the sensor 355. This is represented by the dashed line 957.

[0216] According to an embodiment of the present invention, Figure 10 A block diagram of a sensor system including hydration electronics is illustrated. The sensor system includes a connector 1010, a sensor 1012, and a monitor or sensing electronics device 1025. The sensor 1012 includes an electrode 1020 and a connection portion 1024. In embodiments of the present invention, the sensor 1012 may be connected to the sensing electronics device 1025 via the connector 1010 and a cable. In other embodiments of the present invention, the sensor 1012 may be directly connected to the sensing electronics device 1025. In other embodiments of the present invention, the sensor 1012 may be incorporated into the same physical device as the sensing electronics device 1025. The monitor or sensing electronics device 1025 may include a power supply 1030, a regulator 1035, a signal processor 1040, a measurement processor 1045, and a processor 1050. The monitor or sensing electronics device 1025 may also include a hydration detection circuit 1060. The hydration detection circuit 1060 is connected to the sensor 1012 to determine whether the electrode 1020 of the sensor 1012 is sufficiently hydrated. If the electrodes 1020 are not fully hydrated, then the electrodes 1020 do not provide accurate glucose readings, therefore, it is very important to know when the electrodes 1020 are fully hydrated. Once the electrodes 1020 are fully hydrated, then accurate glucose readings can be obtained.

[0217] exist Figure 10 In the embodiment of the invention shown, the hydration detection circuit 1060 may include a delay or timer module 1065 and a connection detection module 1070. In the embodiment of the invention using a short-term sensor or subcutaneous sensor, after the sensor 1012 has been inserted into the subcutaneous tissue, the sensing electronics or monitor 1025 is connected to the sensor 1012. The connection detection module 1070 recognizes that the sensing electronics 1025 is connected to the sensor 1012 and sends a signal to the timer module 1065. This Figure 10 1086. In the embodiment of the present invention, the timer module 1065 receives the connection signal and waits for a set or determined hydration time. For example, the hydration time can be two minutes, five minutes, ten minutes, or twenty minutes. These embodiments are intended to illustrate and not to limit the present invention. The time range does not have to be a set number of minutes and can include any number of seconds. In an embodiment of the present invention, after the timer module 1065 has waited the set hydration time, the timer module 1065 can notify the processor 1050 by sending a hydration signal that the sensor 1012 is hydrated, which is represented by line 1086.

[0218] In this embodiment of the invention, the processor 1050 may receive a hydration signal and begin using the sensor signal (e.g., sensor measurement) only after receiving the hydration signal. In another embodiment of the invention, the hydration detection circuit 1060 may be connected between the sensor (sensor electrode 1020) and the signal processor 1040. In this embodiment of the invention, the hydration detection circuit 1060 may prevent the sensor signal from being sent to the signal processor 1040 until the timer module 1065 notifies the hydration detection circuit 1060 that a set hydration time has elapsed. This is represented by the dashed lines labeled 1080 and 1081. For example, the timer module 1065 may send a connection signal to a switch (or transistor) to turn on the switch and allow the sensor signal to be transmitted to the signal processor 1040. In an alternative embodiment of the invention, in the hydration detection circuit 1060, the timer module 1065 may send a connection signal to turn on (or close) the switch 1088 after the hydration time has elapsed, thereby applying the voltage from the regulator 1035 to the sensor 1012. In other words, in embodiments of the present invention, the voltage from the regulator 1035 may not be applied to the sensor 1012 until the hydration time has elapsed.

[0219] Figure 11An embodiment of the present invention that includes a mechanical switch that helps determine hydration time is described. In this embodiment of the present invention, a single housing may include sensor assembly 1120 and sensor electronics 1125. In this embodiment of the present invention, sensor assembly 1120 may be located in one housing and sensor electronics 1125 may be located in another housing, but sensor assembly 1120 and sensor electronics 1125 may be connected together. In this embodiment of the present invention, connection detection mechanism 1160 may be a mechanical switch. The mechanical switch may detect that sensor 1120 is physically connected to sensor electronics 1125. In this embodiment of the present invention, timer circuit 1135 may also be activated when mechanical switch 1160 detects that sensor 1120 is connected to sensor electronics 1125. In other words, the mechanical switch may close and a signal may be transmitted to timer circuit 1135. Once the hydration time has elapsed, timer circuit 1135 sends a signal to switch 1140 to cause regulator 1035 to apply voltage to sensor 1120. In other words, no voltage is applied until the hydration time has elapsed. In embodiments of the present invention, once the hydration time has elapsed, current may be applied to the sensor instead of voltage. In an alternative embodiment of the present invention, when the mechanical switch 1160 recognizes that the sensor 1120 is physically connected to the sensing electronics 1125, power may be initially applied to the sensor 1120. The power supplied to the sensor 1120 causes a sensor signal to be output from the working electrode of the sensor 1120. The sensor signal may be measured and transmitted to the processor 1175. The processor 1175 may include a counter input. Under certain operating conditions, after a set hydration time has elapsed from the time the sensor signal was input to the processor 1175, the processor 1175 may begin processing the sensor signal as an accurate measurement of glucose in the subject. In other words, the processor 1175 may have been receiving the sensor signal from the constant potential circuit 1170 for a period of time but not processing the signal until it receives an instruction from the processor's counter input recognizing that the hydration time has elapsed. In embodiments of the present invention, the constant potential circuit 1170 may include a current-to-frequency converter 1180. In this embodiment of the invention, the current-to-frequency converter 1180 may receive the sensor signal as a current value and may convert the current value into a frequency value, which is easier for the processor 1175 to process.

[0220] In an embodiment of the present invention, the mechanical switch 1160 may also notify the processor 1175 when the sensor 1120 is disconnected from the sensor electronics 1125. Figure 111176. This can cause the processor 1170 to shut down or reduce power to several components, chips, and / or circuits of the sensing electronics 1125. If the sensor 1120 is not connected, and if components or circuits of the sensing electronics 1125 are in an on state, the battery or power source may be depleted. Therefore, if the mechanical switch 1160 detects that the sensor 1120 has been disconnected from the sensing electronics 1125, the mechanical switch can indicate this to the processor 1175, which can shut down or reduce power to one or more of the circuits, chips, or components of the sensing electronics 1125.

[0221] Figure 12 An electrical method for detecting hydration according to an embodiment of the present invention is illustrated. In an embodiment of the present invention, an electronic detection mechanism can be used to detect sensor connection. In this embodiment of the present invention, the hydration detection electronic component 1250 can include an AC source 1255 and a detection circuit 1260. The hydration detection electronic component 1250 can be located in the sensing electronic device 1225. The sensor 1220 can include a counter electrode 1221, a reference electrode 1222, and a working electrode 1223. Figure 12 As shown, AC source 1255 is connected to voltage setting device 1275, reference electrode 1222, and detection circuit 1260. In this embodiment of the invention, the AC signal from the AC source is applied to the reference electrode connection, as shown in FIG. Figure 12 As shown by the dashed line 1291 in FIG. In an embodiment of the present invention, the AC signal is coupled to the sensor 1220 through the impedance and if the sensor 1220 is connected to the sensing electronic device 1225, the coupled signal is significantly attenuated. Therefore, a low-level AC signal appears at the input of the detection circuit 1260. This is also called a high-attenuation signal or a signal with a high attenuation level. Under some operating conditions, the voltage level of the AC signal can be V 施加 *(C 耦合 ) / (C 耦合 +C 传感器 If the detection circuit 1260 detects that a high level AC signal (low attenuation signal) is present at the input of the detection circuit 1260, no interrupt signal is sent to the microcontroller 410 because the sensor 1220 is not yet fully hydrated or activated. For example, the input of the detection circuit 1260 can be a comparator. If the sensor 1220 is fully hydrated (or wetted), an effective capacitance is formed between the counter electrode and the reference electrode (e.g., Figure 12 The capacitor C r-c ), and an effective capacitance is formed between the reference electrode and the working electrode (e.g., Figure 12 The capacitor C w-rIn other words, effective capacitance refers to the capacitance formed between two nodes and does not mean that an actual capacitor is placed between two electrodes in the circuit. In an embodiment of the present invention, the AC signal from the AC source 1255 is connected to the capacitor C r-c and C w-r The low level or highly attenuated AC signal from the AC source 1255 is sufficiently attenuated and the detection circuit 1260 detects the presence of the low level or highly attenuated AC signal at the input of the detection circuit 1260. This embodiment of the present invention is important because the use of the existing connections between the sensor 1120 and the sensing electronics 1125 reduces the number of connections to the sensor. In other words, Figure 11 The disclosed mechanical switch requires the switch and associated connections between the sensor 1120 and the sensing electronics 1125. Eliminating the mechanical switch is advantageous because the sensor 1120 continues to shrink in size and the elimination of components helps achieve this size reduction. In alternative embodiments of the present invention, the AC signal can be applied to a different electrode (e.g., a counter electrode or a working electrode) and the present invention can operate in a similar manner.

[0222] As described above, after detection circuit 1260 has detected the presence of a low-level AC signal at the input of detection circuit 1260, detection circuit 1260 may later detect the presence of a high-level, low-attenuation AC signal at the input. This indicates that sensor 1220 has been separated from sensing electronics 1225 or that the sensor is not functioning properly. If the sensor has been separated from sensing electronics 1225, the AC source may be coupled to the input of detection circuit 1260 with little or no attenuation. As described above, detection circuit 1260 may generate an interrupt signal that is sent to the microcontroller. This interrupt signal may be received by the microcontroller, and the microcontroller may reduce or remove power to one or more components or circuits in sensing electronics 1225. This may be referred to as a second interrupt signal. Furthermore, this helps reduce power consumption of sensing electronics 1225, particularly when sensor 1220 is not connected to sensing electronics 1225.

[0223] exist Figure 12 In an alternative embodiment of the present invention, an AC signal may be applied to the reference electrode 1222, as shown by reference numeral 1291, and the impedance measurement device 1277 may measure the impedance of a region in the sensor 1220. For example, the region may be the region between the reference electrode and the working electrode, as shown in FIG. Figure 121292 in FIG. Under some operating conditions, impedance measurement device 1277 may send a signal to detection circuit 1260 if the measured impedance drops below an impedance threshold or other set criteria. This indicates that the sensor is sufficiently hydrated. Under other operating conditions, impedance measurement device 1277 may send a signal to detection circuit 1260 once the impedance rises above the impedance threshold. Detection circuit 1260 then sends an interrupt signal to microcontroller 410. In another embodiment of the present invention, impedance measurement device 1277 may send an interrupt signal or other signal directly to the microcontroller.

[0224] In an alternative embodiment of the present invention, the AC source 1255 may be replaced by a DC source. If a DC source is used, a resistance measuring element may be used in place of the impedance measuring element 1277. In embodiments of the present invention using a resistance measuring element, once the resistance drops below a resistance threshold or set standard, the resistance measuring element may send a signal (represented by dashed line 1293) to the detection circuit 1260 or directly to the microcontroller indicating that the sensor is sufficiently hydrated and power may be applied to the sensor.

[0225] exist Figure 12 In the embodiment of the present invention shown, if the detection circuit 1260 detects a low level or high attenuation AC signal from the AC source, an interrupt signal is generated to the microcontroller 410. The interrupt signal indicates that the sensor is fully hydrated. In this embodiment of the present invention, in response to the interrupt signal, the microcontroller 410 generates a signal that is transmitted to the digital-to-analog converter 420 to instruct or cause the digital-to-analog converter 420 to apply a voltage or current to the sensor 1220. Figure 6A 、 Figure 6B or Figure 6CAny of the various pulses or short pulse sequences described in the related text describing pulse application can be applied to sensor 1220. For example, the voltage from DAC 420 can be applied to operational amplifier 1275, the output of which is applied to counter electrode 1221 of sensor 1220. This causes the sensor (e.g., working electrode 1223 of the sensor) to generate a sensor signal. Because the sensor is fully hydrated, as identified by the interrupt signal, the sensor signal generated at working electrode 1223 is an accurate measurement of glucose. The sensor signal is measured by sensor signal measurement device 431, and sensor signal measurement device 431 transmits the sensor signal to microcontroller 410, where a physiological condition parameter of the subject is measured. The generation of the interrupt signal indicates that the sensor is fully hydrated and that sensor 1220 is providing accurate glucose measurements. In this embodiment of the invention, the hydration time may depend on the sensor type and / or manufacturer, as well as the sensor's response to insertion or implantation into the subject. For example, one sensor 1220 may have a hydration time of five minutes and one sensor 1220 may have a hydration time of one minute, two minutes, three minutes, six minutes, or 20 minutes. Again, any amount of time may be an acceptable amount of hydration time for a sensor, but smaller amounts of time are preferred.

[0226] If the sensor 1220 is connected but not fully hydrated or wet, then the effective capacitance C r-c and C w-c The AC signal from the AC source 1255 may not be attenuated. The electrodes in the sensor 1120 are dry before insertion, and because the electrodes are dry, there is no good electrical path (or conductive path) between the two electrodes. Therefore, a high-level AC signal or a low-attenuation AC signal can still be detected by the detection circuit 1260 and no interrupt signal will be generated. Once the sensor is inserted, the electrodes are immersed in the conductive body fluid. This creates a leakage path with a lower DC resistance. Moreover, a boundary layer capacitor is formed at the metal / fluid interface. In other words, a considerable capacitance is formed between the metal / fluid interface and this large capacitance looks like two capacitors connected in series between the sensor electrodes. This can be called effective capacitance. In practice, the conductivity of the electrolyte on the electrode is measured. In some embodiments of the present invention, a glucose limiting membrane (GLM) also exemplifies the impedance that blocks electrical efficiency. Non-hydrated GLM produces high impedance, while high-water content GLM produces low impedance. Low impedance is desirable for accurate sensor measurements.

[0227] Figure 13AA method of hydrating a sensor according to an embodiment of the present invention is illustrated. In an embodiment of the present invention, the sensor may be physically connected to a sensing electronic device (1310). After connection, a timer or counter is started to calculate the hydration time (1320). After the hydration time has elapsed, a signal may be transmitted to a subsystem in the sensing electronic device to begin applying voltage to the sensor (1330). As discussed above, in an embodiment of the present invention, a microcontroller may receive the signal and instruct a DAC to apply voltage to the sensor, or in another embodiment of the present invention, a switch may receive the signal, which causes a regulator to apply voltage to the sensor. The hydration time may be five minutes, two minutes, ten minutes and may vary depending on the patient and the type of sensor.

[0228] In an optional embodiment of the present invention, after the sensor is connected to the sensing electronic device, an AC signal (e.g., a low voltage AC signal) may be applied to the sensor, for example, the reference electrode (1340) of the sensor. Because the connection of the sensor to the sensing electronic device allows the AC signal to be applied to the sensor, the AC signal may be applied. After the AC signal is applied, an effective capacitance (1350) is formed between the electrode of the sensor to which the voltage is applied and the other two electrodes. The detection circuit determines the level of the AC signal present at the input of the detection circuit (1360). If a low level AC signal (or a highly attenuated AC signal) is present at the input of the detection circuit, an interrupt signal is generated by the detection circuit and sent to the microcontroller (1370) due to the effective capacitance forming a good conductive path between the electrodes and the resulting attenuation of the AC signal.

[0229] The microcontroller receives the interrupt signal generated by the detection circuit and sends a signal to the digital-to-analog converter, instructing or causing the digital-to-analog converter to apply a voltage to an electrode of the sensor (e.g., a counter electrode) (1380). Applying the voltage to the electrode of the sensor causes the sensor to generate or generate a sensor signal (1390). The sensor signal measurement device 431 measures the generated sensor signal and transmits the sensor signal to the microcontroller. The microcontroller receives the sensor signal from the sensor signal measurement device connected to the working electrode and processes the sensor signal to extract a measurement value of a physiological characteristic of the subject or patient (1395).

[0230] Figure 13B Other methods for verifying sensor hydration according to embodiments of the present invention are illustrated. Figure 13BIn the embodiment of the invention shown, the sensor is physically connected to the sensing electronics (1310). In an embodiment of the invention, an AC signal (1341) is applied to an electrode in the sensor (e.g., a reference electrode). Alternatively, in an embodiment of the invention, a DC signal (1341) is applied to an electrode in the sensor. If an AC signal is applied, the impedance measuring element measures the impedance of a point in the sensor (1351). Alternatively, if a DC signal is applied, the resistance measuring element measures the resistance of a point in the sensor (1351). If the resistance or impedance is below a resistance threshold or impedance threshold (or other set standard), the impedance (or resistance) measuring element sends a signal (or causes a signal to be transmitted) to a detection circuit (1361), and the detection circuit sends an interrupt signal to the microcontroller, identifying that the sensor is hydrated. Reference numerals 1380, 1390, and 1395 are shown in FIG. Figure 13A and Figure 13B are the same in , because they represent the same operation.

[0231] The microcontroller receives the interrupt signal and sends a signal to the digital-to-analog converter to apply a voltage to the sensor (1380). In an optional embodiment of the present invention, as discussed above, the digital-to-analog converter can apply a current to the sensor. The sensor, such as the working electrode, generates a sensor signal (1390) representing a physiological parameter of the patient. The microcontroller receives a sensor signal (1395) from a sensor signal measuring device that measures the sensor signal at an electrode in the sensor (e.g., the working electrode). The microcontroller processes the sensor signal to extract a measurement value of a physiological characteristic of the subject or patient, such as the patient's blood glucose level.

[0232] Figure 14A and Figure 14B The combined method of sensor hydration and sensor stabilization according to an embodiment of the present invention is illustrated. Figure 14AIn the illustrated embodiment of the present invention, a sensor is connected to sensing electronics (1405). An AC signal is applied to the sensor electrodes (1410). A detection circuit determines the level of the AC signal present at the input of the detection circuit (1420). If the detection circuit determines that a low-level AC signal is present at the input (indicating that the AC signal is attenuated at a high level), an interrupt signal is sent to a microcontroller (1430). Once the interrupt signal is sent to the microcontroller, the microcontroller knows to begin or initiate a stabilization sequence (1440), i.e., applying a plurality of voltage pulses to the sensor electrodes, as described above. For example, the microcontroller may cause a digital-to-analog converter to apply three voltage pulses (with an amplitude of +0.535 volts) to the sensor, each of which is followed by a duration of three voltage pulses (with an amplitude of 1.07 volts to be applied). This may be referred to as a voltage stabilization sequence. The microcontroller may initiate the voltage stabilization sequence by executing a software program in read-only memory (ROM) or random access memory. After the stabilization sequence is complete, the sensor may generate a sensor signal, which is measured and transmitted to the microcontroller (1450).

[0233] In an embodiment of the present invention, the detection circuit may determine that a high-level AC signal has been continuously present at the input of the detection circuit (e.g., the input of a comparator) even after a hydration time threshold has passed (1432). For example, the hydration time threshold may be 10 minutes. After 10 minutes have passed, the detection circuit may still detect the presence of a high-level AC signal. At this point, the detection circuit may send a hydration assist signal to the microcontroller (1434). If the microcontroller receives the hydration assist signal, the microcontroller may send a signal to cause the DAC to apply a voltage pulse or series of voltage pulses to assist in hydrating the sensor (1436). In an embodiment of the present invention, the microcontroller may send a signal to cause the DAC to apply a portion of a stabilization sequence or other voltage pulses to assist in hydrating the sensor. In this embodiment of the present invention, the application of the voltage pulses may generate a low-level AC signal (or a highly attenuated signal), which is detected in the detection circuit (1438). At this point, the detection circuit may send an interrupt signal, and the microcontroller may begin a stabilization sequence, as disclosed in step 1430.

[0234] Figure 14BA second embodiment of a combination of a hydration method and a stabilization method is illustrated, wherein feedback information is used during the stabilization process. The sensor is connected to a sensing electronic device (1405). An AC signal (or DC signal) is applied to the sensor (1411). In an embodiment of the present invention, the AC signal (or DC signal) is applied to an electrode of the sensor, such as a reference electrode. An impedance measuring device (or resistance measuring device) measures the impedance (or resistance) within a specified area of ​​the sensor (1416). In an embodiment of the present invention, the impedance (or resistance) between the reference electrode and the working electrode can be measured. The measured impedance (or resistance) can be compared with a certain impedance value or resistance value (1421) to observe whether the impedance (or resistance) within the sensor is low enough, which indicates that the sensor is hydrated. If the impedance (or resistance) is lower than the impedance value (resistance value) or other set standard (which can be a threshold), an interrupt signal is transmitted to the microcontroller (1431). After receiving the interrupt signal, the microcontroller sends a signal to the DAC, instructing the DAC to apply a stabilization sequence of voltage (or current) to the sensor (1440). After the stabilization sequence is applied to the sensor, a sensor signal is generated in the sensor (e.g., at the working electrode), measured by the sensor signal measuring device, transmitted through the sensor signal measuring device, and received by the microcontroller (1450). Because the sensor is hydrated and the voltage stabilization sequence has been applied to the sensor, the sensor signal is an accurately measured physiological parameter (i.e., blood glucose).

[0235] Figure 14CA third embodiment of the present invention is illustrated, combining a stabilization method and a hydration method. In this embodiment of the invention, a sensor is connected to a sensing electronics device (1500). After the sensor is physically connected to the sensing electronics device, an AC signal (or DC signal) is applied to an electrode (e.g., a reference electrode) of the sensor (1510). Simultaneously or substantially simultaneously, a microcontroller sends a signal to a DAC to apply a stabilization voltage sequence to the sensor (1520). In an alternative embodiment of the invention, a stabilization current sequence may be applied to the sensor instead of a stabilization voltage sequence. A detection circuit determines the level of the AC signal (or DC signal) present at the input of the detection circuit (1530). If a low-level AC signal (or DC signal) is present at the input of the detection circuit, indicating that the AC signal (or DC signal) is highly attenuated, an interrupt signal is sent to the microcontroller (1540). Because the microcontroller has already initiated the stabilization sequence, the microcontroller receives the interrupt signal and sets a first indicator that the sensor is sufficiently hydrated (1550). After the stabilization sequence ends, the microcontroller sets a second indicator indicating that the stabilization sequence is complete (1555). Application of the stabilization voltage sequence causes the sensor (e.g., the working electrode) to generate a sensor signal (1560), which is measured by the sensor signal measurement circuit and sent to the microcontroller. If the second indicator indicating completion of the stabilization sequence is set and the first indicator indicating completion of hydration is set, the microcontroller can use the sensor signal (1570). If one or both of the indicators are not set, the microcontroller cannot use the sensor signal because the sensor signal does not represent an accurate measurement of the subject's physiological measurement.

[0236] In general, the hydration and stabilization processes described above can be used as part of a larger continuous glucose monitoring (CGM) approach. The current state of continuous glucose monitoring in the prior art is primarily adjunctive, meaning that the readings provided by a CGM device (including, for example, an implantable sensor or a subcutaneous sensor) cannot be used to make clinical decisions without a reference value. The reference value must be obtained by, for example, a finger prick using a BG meter. A reference value is necessary because the available information from the sensor / detection element is very limited. Specifically, the only pieces of information currently provided by the detection element for processing are the raw sensor value (i.e., the sensor current or Isig) and the counter voltage, which is the voltage between the counter electrode and the reference electrode (see, e.g., Figure 5 ). Therefore, during analysis, if the raw sensor signal appears abnormal (e.g., if the signal decreases), the only way for one skilled in the art to distinguish between a sensor malfunction and a physiological change in the user / patient (i.e., a change in glucose levels in the body) is to obtain a reference glucose value via a finger prick. As is known in the art, finger prick reference values ​​are also used to calibrate sensors.

[0237] Embodiments of the invention described herein relate to advancements and improvements in continuous glucose monitoring, resulting in more independent systems and related devices and methods, wherein the need for finger-prick reference values ​​can be minimized or eliminated, and thus clinical decisions with a high level of confidence can be made based solely on information from sensor signals. From a sensor design perspective, this independence can be achieved through electrode redundancy, sensor diagnostics, and Isig and / or sensor glucose (SG) fusion, according to embodiments of the invention.

[0238] As will be explored further below, redundancy can be achieved by using multiple working electrodes (e.g., in addition to a counter electrode and a reference electrode) to generate multiple signals representing a patient's blood glucose (BG) level. These multiple signals can then be used to assess the relative health of the electrodes (working electrodes), the overall reliability of the sensor, and the frequency with which the reference value needs to be calibrated (if necessary).

[0239] Sensor diagnostics include the use of additional (diagnostic) information that can provide real-time insights into the health of the sensor. In this regard, electrochemical impedance spectroscopy (EIS) has been found to provide this additional information in the form of sensor impedance and impedance-related parameters at different frequencies. Moreover, advantageously, it has been further discovered that for some frequency ranges, the impedance and / or impedance-related data are substantially independent of glucose. This glucose independence enables the use of multiple EIS-based markers or signatures to not only generate robust, highly reliable sensor glucose values ​​(via fusion methods), but also to assess the condition, health, age, and effectiveness of individual electrodes and the sensor as a whole, substantially independent of glucose-dependent Isig.

[0240] For example, analysis of the glucose-independent impedance data provides information about sensor efficiency, such as how quickly the sensor hydrates and becomes ready for data acquisition, using, for example, 1 kHz real impedance values, 1 kHz imaginary impedance values, and Nyquist slope values ​​(described in detail below). Furthermore, the glucose-independent impedance data provides information about potential occlusions (e.g., using 1 kHz real impedance values) that may exist on the sensor membrane surface, which may temporarily block the passage of glucose into the sensor and thereby cause a signal drop. Furthermore, the glucose-independent impedance data provides information about sensor sensitivity loss during extended wear, potentially due to localized hypoxia at the insertion site, using, for example, phase angle and / or imaginary impedance values ​​at 1 kHz and higher frequencies.

[0241] In the description of electrode redundancy and EIS, and elsewhere described in detail below, a fusion algorithm can be used to obtain the diagnostic information provided by EIS about each redundant electrode and independently assess the reliability of each electrode. A weight, a measure of reliability, can then be added to each individual signal, and a single fused signal can be calculated that can be used to generate the sensor glucose value observed by the patient / subject.

[0242] As can be seen from the above, the combined use of redundancy, sensor diagnostics using EIS, and EIS-based fusion algorithms make the entire CGM system more reliable than currently available CGM systems. Redundancy is advantageous in at least two respects. First, redundancy eliminates the risk of single point failure by providing multiple signals. Second, providing multiple electrodes (working electrodes) where a single electrode may be sufficient allows the output of the redundant electrode to be used as a check on the primary electrode, thereby reducing and possibly eliminating the need for frequent calibration. In addition, the EIS diagnostic method independently and in detail checks the health of each electrode without the need for a reference glucose value (finger prick), thereby reducing the number of reference values ​​required. However, the use of EIS technology and EIS diagnostics is not limited to redundant systems (i.e., those with more than one working electrode). In particular, as discussed below in conjunction with embodiments of the present invention, EIS can be advantageously used in conjunction with single electrode and / or multi-electrode sensors.

[0243] EIS or AC impedance method studies the system's response to the application of a periodic small amplitude AC signal. Figure 15A As shown in Figure 1, where E is the applied potential, I is the current, and impedance (Z) is defined as ΔE / ΔI. However, while impedance itself can be mathematically defined simply as ΔE / ΔI, commercial success has yet to be achieved in applying EIS technology to continuous glucose monitoring. This is partly due to the fact that glucose sensors are very complex systems and, to date, no mathematical model has been developed in the art that fully accounts for the complexity of the EIS output of glucose sensors.

[0244] Figure 15B A simplified circuit model for describing electrochemical impedance spectroscopy is shown. In this example, IHP stands for inner Helmholtz plane, OHP stands for outer Helmholtz plane, CE is the counter electrode, WE is the working electrode, and C d is the double layer capacitance, R p is the polarization resistance, Z w is the Warburg impedance, and R s is the solution resistance. The last four components—the double layer capacitance (C d )、Warburg impedance (Z w ), polarization resistance (Rp ) and solution resistance (R s )—each of which can play an important role in sensor performance and can be measured individually by applying a low-frequency or high-frequency alternating working potential. For example, Warburg impedance is closely related to the diffusion impedance of the electrochemical system—which is primarily a low-frequency impedance—and, as such, is present in all diffusion-limited electrochemical sensors. Therefore, by correlating one or more of these components with one or more components and / or layers in a glucose sensor, one skilled in the art can use EIS technology as a sensor diagnostic tool.

[0245] As is known in the art, impedance can be defined in terms of magnitude and phase, where the magnitude (|Z|) is the ratio of the voltage difference amplitude to the current amplitude, and the phase (θ) is the phase shift of the current over the voltage. When a circuit is driven solely by direct current (DC), impedance is the same as resistance, i.e., resistance is a special case of impedance at zero phase angle. However, as a complex number, impedance can also be represented by its real and imaginary parts. In this regard, the real and imaginary impedances can be derived from the impedance magnitude and phase using the following equations:

[0246] Real Impedance(ω)=Magnitude(ω)x cos(Phase(ω) / 180xπ)

[0247] Imaginary Impedance(ω)=Magnitude(ω)x sin(Phase(ω) / 180xπ)

[0248] Where ω represents the input frequency at which the amplitude (in ohms) and phase (in degrees) are measured. The relationship between impedance and current, on the one hand, and between impedance and voltage, on the other hand, including how to calculate the former based on measurements of the latter, will be examined in detail below in conjunction with a sensor electronics device comprising an application-specific integrated circuit (ASIC) developed for use in embodiments of the present invention.

[0249] then Figure 15B Using the circuit model shown, the total system impedance can be simplified to:

[0250]

[0251] Among them, Z w (ω) is the Warburg impedance, ω is the angular velocity, j is an imaginary unit (used instead of the traditional "i" so as not to be confused with current), and C d、R p and R s are the double layer capacitance, polarization resistance, and solution resistance (as defined above). The Warburg impedance can be calculated as follows:

[0252]

[0253]

[0254]

[0255] where D is the diffusivity, L is the sensor film thickness, C is the peroxide concentration, and m:1 / 2 corresponds to a 45° Nyquist slope.

[0256] The Nyquist plot is a graphical technique where the real part of the impedance (Real Z) is plotted against its imaginary part (Img Z) across the frequency spectrum. Figure 16A Shows a generalized example of a Nyquist plot, where the X value is the real part of the impedance and the Y value is the imaginary part of the impedance. The phase angle is the angle between the impedance point (X,Y)—defined as a vector with magnitude |Z|—and the X-axis.

[0257] Figure 16A The Nyquist curve of an electrochemical system is generated by applying an AC voltage plus a DC voltage (DC bias) between a working electrode and a counter electrode at a selected frequency of 0.1 Hz to 1000 MHz (i.e., a frequency sweep). Starting from the right, the frequency increases from 0.1 Hz. The real impedance and imaginary impedance under each frequency condition can be calculated and plotted. As shown in the figure, a typical Nyquist curve of an electrochemical system may look like a semicircle connected to a straight line at an inflection point, wherein the semicircle and the straight line represent the impedance being plotted. In some embodiments, the impedance at the inflection point is very important because it is most easily identified in the Nyquist curve and the intercept can be defined. Typically, the inflection point is close to the X-axis, and the X value of the inflection point is close to the sum of the polarization resistance and the solution resistance (R p +R s ).

[0258] refer to Figure 16B, the Nyquist curve can be generally described in terms of a lower frequency region 1610 and a higher frequency region 1620, where the labels "higher frequency" and "lower frequency" are used in a relative sense, but are not limiting of the present invention. Thus, for example, the lower frequency region 1610 may illustratively include data points obtained in the frequency range of approximately 0.1 Hz to approximately 100 Hz (or higher), and the higher frequency region 1620 may illustratively include data points obtained in the frequency range of approximately 1 kHz (or lower) to approximately 8 kHz (and higher). In the lower frequency region 1610, the Nyquist slope represents the slope of the linear fit 1630 of the lower frequency data points in the Nyquist curve. As shown, in the higher frequency region 1620, the imaginary impedance value is minimal and negligible. Therefore, the intercept 1600 is essentially the value of the real impedance at higher frequencies (e.g., in this case, approximately in the range of 1 kHz to 8 kHz). In Figure 16B , the intercept 1600 is approximately 25k ohms.

[0259] Figure 16C and Figure 16D Shows how a glucose sensor responds to a sinusoidal (i.e., alternating) operating potential. In these figures, GLM is the glucose limiting membrane of the sensor. AP is an adhesion promoter, HSA is human serum albumin, GOX is glucose oxidase (layer), and E dc is the DC potential, E ac is the AC potential, and C' 过氧化物 is the concentration of peroxide during the application of AC. Figure 16C As shown in FIG, if the sensor diffusion length is smaller than the membrane (GOX) length, which is a function of the AC potential frequency, molecular diffusion rate, and membrane thickness, then the system provides a relatively linear response (i.e., infinite) with a constant phase angle. Conversely, if the diffusion length is equal to the membrane (GOX) length, then the system response becomes finite, resulting in a Figure 16D The semicircular Nyquist plot is shown in Figure 2. The latter is generally applicable for low-frequency EIS, where non-Faraday processes are negligible.

[0260] In performing an EIS analysis, AC voltages and DC biases of various frequencies may be applied, for example, between a working electrode and a reference electrode. In this regard, EIS is an improvement over previous methods, which may have been limited to applying a simple DC current or a single-frequency AC voltage. Although, in general, EIS can be performed at frequencies ranging from μHz to MHz, in embodiments of the present invention, a narrower frequency range (e.g., about 0.1 Hz to about 8 kHz) may be sufficient to perform EIS. Thus, in embodiments of the present invention, an AC potential in the frequency range of about 0.1 Hz to about 8 kHz may be applied, with a programmable amplitude of up to at least 100 mV, and preferably about 50 mV.

[0261] Within the above frequency range, relatively higher frequencies - i.e., those frequencies typically in the range of about 1 kHz to about 8 kHz - are used to examine in detail the capacitive properties of the sensor. Based on the thickness and permeability of the membrane, a typical impedance range at relatively higher frequencies can be, for example, about 500 ohms to 25k ohms, and a typical phase range can be, for example, 0 degrees to -40 degrees. On the other hand, relatively lower frequencies - i.e., those frequencies typically in the range of about 0.1 Hz to about 100 Hz - are used to examine in detail the resistive properties of the sensor. Thus, based on the electrode design and the degree of metallization, a typical functional range of the output real impedance can be, for example, about 50k ohms to 300k ohms, and a typical range of phase can be about -50 degrees to about -90 degrees. The above exemplary ranges are, for example, Figure 16E and Figure 16F The Bode plots are shown.

[0262] As mentioned above, the phrases "higher frequency" and "lower frequency" are intended to be used relative to each other, not in an absolute sense, and the "higher frequency" and "lower frequency" and the typical impedance and phase ranges described above are exemplary and not limiting of the invention. Nevertheless, the basic principle remains the same: the capacitive and resistive behavior of a sensor can be examined in detail by analyzing the impedance data over the entire frequency spectrum, where lower frequencies generally provide information about components with greater resistance (e.g., electrodes, etc.) and higher frequencies provide information about capacitive components (e.g., membranes). However, the actual frequency range in each case depends on the overall design, including, for example, the type of electrodes, the surface area of ​​the electrodes, the membrane thickness, the permeability of the membrane, etc. See also Figure 15B , regarding the general correspondence between high-frequency circuit elements and sensor membranes and the general correspondence between low-frequency circuit elements and Faraday processes (including, for example, electrodes).

[0263] EIS can be used in sensor systems where the sensor includes a single working electrode, or in sensor systems where the sensor includes multiple (redundant) working electrodes. In one embodiment, EIS provides useful information about the age (or aging) of the sensor. Specifically, the magnitude and phase angle of the impedance change under different frequency conditions. Figure 17 As shown in Figure 2, the sensor impedance—specifically, the sum of Rp and Rs—reflects the sensor age and sensor operation. Figure 17 As can be seen from the different curves, new sensors generally have higher impedance than used sensors. Thus, considering the value of X, which is the sum of Rp and Rs, a threshold value can be used to determine when the age of the sensor exceeds the specified sensor operating life. It should be noted that although Figure 17 21 shows an exemplary embodiment and is discussed below, however, the real impedance value at the inflection point (i.e., Rp+Rs) is used to determine the aging, state, stability, and hydration of the sensor, and alternative embodiments may use other EIS-based parameters such as imaginary impedance, phase angle, Nyquist slope, etc. in addition to the real impedance, or, alternative embodiments may use other EIS-based parameters such as imaginary impedance, phase angle, Nyquist slope, etc. in place of the real impedance.

[0264] Figure 17 An example of a Nyquist plot over sensor life is illustrated. The points indicated by arrows are the respective inflection points for each frequency sweep within the spectrum. For example, before startup (time t=0), Rs+Rp is above 8.5 kOhms, and after startup (time t=0.5 hours), the value of Rs+Rp drops below 8 kOhms. Over the course of the next six days, Rs+Rp continues to decrease, such that at the end of the specified sensor life, Rs+Rp drops below 6.5 kOhms. Based on these examples, a threshold can be set to specify when the Rs+Rp value indicates the end of the specified sensor operating life. Thus, EIS technology eliminates the vulnerability of reusing a sensor beyond its specified operating time. In other words, if a patient attempts to re-use a sensor by disconnecting and reconnecting it after it has reached its specified operating time, EIS will measure an abnormally low impedance, causing the system to reject the sensor and prompting the patient to use a new sensor.

[0265] Furthermore, EIS can detect sensor failure by detecting when the sensor impedance drops below a low impedance threshold level, which indicates that the sensor may be overused and not functioning properly. The system can then stop the sensor before the specified operating life. As discussed in more detail below, sensor impedance can also be used to detect other sensor failure modes. For example, when the sensor enters a low current state (i.e., sensor failure) for any reason, the sensor impedance may also increase above a high impedance threshold. If the impedance becomes abnormally high during sensor operation due to, for example, protein or peptide contamination, macrophage attachment, or any other factor, the system can also stop the sensor before the specified sensor operating life.

[0266] Figure 18 The invention illustrates how EIS technology can be used in sensor stabilization and sensor age detection according to an embodiment of the invention. Figure 18 The logic begins after the hydration process and the aforementioned sensor initialization process has completed (1800). In other words, the sensor is considered fully hydrated and the first initialization process has been used to initialize the sensor. Preferably, the initialization process can be in the form of a voltage pulse as described in detail above. However, in alternative embodiments, different waveforms can be used for the initialization process. For example, a sine wave can be used instead of a pulse to accelerate sensor wetting or conditioning. In addition, some portions of the waveform may need to be greater than the normal operating voltage of the sensor, i.e., 0.535 volts.

[0267] In box 1810, the EIS program is applied and the impedance is compared with a first high threshold and a first low threshold. Examples of the first high threshold and the first low threshold can be 7k ohms and 8.5k ohms, respectively, although the value can be set higher or lower as needed. If the impedance, such as Rp+Rs, is higher than the first high threshold, then, in box 1820, the sensor undergoes an additional initialization process (e.g., one or more additional pulses are applied). Ideally, the number of total initialization processes for initializing the sensor can be optimized to limit the impact on the battery life of the sensor and the total time required to stabilize the sensor. Therefore, by using EIS, a smaller initialization process can be performed initially, and the number of initializations can be gradually increased to obtain the right number of initializations to prepare the sensor for use. Similarly, in an optional embodiment, EIS can be used for the hydration process to minimize the number of initializations required for the auxiliary hydration process shown in Figures 13 to 14.

[0268] On the other hand, if the impedance, e.g., Rp+Rs, is below a first low threshold, then, at block 1860, the sensor may be determined to be faulty and immediately stopped. A message will be sent to the user to replace the sensor and restart the hydration process. If the impedance is within the high and low thresholds, then, at block 1830, the sensor may begin normal operation. The logic then proceeds to block 1840, where an additional EIS is performed to check the sensor's age. The first time the logic reaches block 1840, the microcontroller performs an EIS to assess the sensor's age, thereby eliminating the vulnerability of the user being able to plug and unplug the same sensor. During subsequent iterations of the EIS procedure, when the logic returns to block 1840, the microcontroller may perform an EIS at fixed intervals over the specified lifespan of the sensor. In a preferred embodiment, the fixed interval is set to every two hours, however, longer or shorter time periods may simply be used.

[0269] At block 1850, the impedance is compared to a second set of upper and lower thresholds. Examples of the second upper and lower thresholds are 5.5 kOhms and 8.5 kOhms, respectively, although these values ​​can be set higher or lower as desired. As long as the impedance value is within the range of the second upper and lower thresholds, the logic proceeds to block 1830, where the sensor operates normally until the specified sensor lifespan, e.g., five days, has been reached. Of course, as described with reference to block 1840, EIS can be performed at regularly scheduled intervals throughout the specified sensor lifespan. However, if, after performing an EIS, at block 1850, the impedance is determined to have fallen below the second lower threshold or risen above the second higher threshold, the sensor is stopped at block 1860. In further alternative embodiments, a secondary check can be performed upon a sensor fault reading. For example, if the EIS indicates that the impedance is outside the second upper and lower thresholds, the logic can perform a second EIS to confirm that the second set of thresholds has indeed been violated (and that the first EIS was performed correctly) before declaring the sensor dead (block 1860).

[0270] Figure 19 Building upon the above detailed description and detailing a possible schedule for performing a diagnostic EIS procedure according to a preferred embodiment of the present invention, each diagnostic EIS procedure is optional, and it is possible to schedule no diagnostic EIS procedure or to have any combination of one or more diagnostic EIS procedures as desired. Figure 19The timeline begins with sensor insertion at time 1900. After sensor insertion, the sensor undergoes a hydration period 1910. This hydration period is important because, as mentioned above, an inadequately hydrated sensor may provide inaccurate readings to the user. During hydration period 1910, at time 1920, a first optional diagnostic EIS procedure is performed to ensure that the sensor is adequately hydrated. First diagnostic EIS procedure 1920 measures the sensor impedance to determine whether the sensor is adequately hydrated. If first diagnostic EIS procedure 1920 determines that the impedance is within a set upper and lower threshold range, indicating adequate hydration, then at time 1930, the sensor controller may allow the sensor to be powered on. Conversely, if first diagnostic EIS procedure 1920 determines that the impedance is outside the set upper and lower threshold ranges, indicating insufficient hydration, then the sensor hydration period 1910 may be extended. After the extended hydration period, once a certain capacitance is reached between the sensor electrodes, indicating that the sensor is adequately hydrated, power-on at time 1930 may occur.

[0271] The second optional diagnostic EIS procedure 1940 may be scheduled after the sensor is powered on at time 1930, but before the sensor is initialized at time 1950. In this arrangement, the second diagnostic EIS procedure 1940 may detect whether the sensor is reused before the initialization at time 1950 begins. The test to determine whether the sensor is reused is performed at Figure 18 However, unlike the previous Figure 18 As described, wherein an aging test is performed after initialization is completed, Figure 19 In the example, the burn-in test is shown as being performed before initialization. It is important to understand that Figure 19 The timeline of the EIS procedure can be rearranged without affecting the overall teaching of the present application, and the order of some steps can be interchanged. As previously explained, the second diagnostic EIS procedure 1940 detects a reused sensor by determining the sensor impedance value and then comparing the value with a set high threshold and a set low threshold. If the impedance is outside the range of the set threshold, this indicates that the sensor has been reused, so the sensor can then be rejected and the user can be prompted to replace it with a new sensor. This prevents complications that may arise from reusing an old sensor. Conversely, if the impedance falls within the set threshold, then initialization 1950 of the sensor can begin and it is certain that a new sensor is being used.

[0272] The third optional diagnostic EIS procedure 1960 can be arranged after the initialization at time point 1950 begins. The third diagnostic EIS procedure 1960 tests the impedance value of the sensor to determine whether the sensor is fully initialized. The third diagnostic EIS procedure 1960 should be performed in the minimum amount of time required for any sensor to be fully initialized. When the third diagnostic EIS procedure is performed at this time, the sensor life is maximized by limiting the time that the fully initialized sensor is not used, and over-initialization is avoided by determining that the sensor is fully initialized before over-initialization occurs. Preventing over-initialization is very important because over-initialization generates undercurrent (suppressed current), which can produce inaccurate readings. However, insufficient initialization is also a problem. Therefore, if the third diagnostic EIS procedure 1960 indicates that the sensor is insufficiently initialized, then an optional initialization can be performed at time point 1970 to fully initialize the sensor. Insufficient initialization is disadvantageous because excessive current (excessive current) causes it to be unrelated to the actual glucose concentration. Due to the dangers of insufficient initialization and over-initialization, the third diagnostic EIS procedure plays an important role in ensuring that the sensor operates normally when it is used.

[0273] In addition, optional periodic diagnostic EIS program 1980 can be arranged according to time after the sensor is fully initialized. EIS program 1980 can be arranged at any set interval. As discussed in detail below, EIS program 1980 can also be triggered by other sensor signals, for example, abnormal current or abnormal counter electrode voltage. In addition, as needed, as few or as many EIS programs as possible can be arranged. In a preferred embodiment, the EIS program used in the hydration process, the EIS program used in the sensor life check process, the EIS program used in the initialization process or the EIS program used in the periodic diagnostic test process are the same program. In an optional embodiment, for different EIS programs, the EIS program can be shortened or extended (that is, checking fewer frequency ranges or more frequency ranges), depending on the needs for a specific impedance range. Periodic diagnostic EIS program 1980 monitors impedance values ​​to ensure that the sensor continues to operate at an optimal level.

[0274] If the sensor current has dropped due to contaminants, sensor age, or a combination of contaminants and sensor age, then the sensor may not be operating at an optimal level. Sensors that have aged beyond a certain point are no longer useful, but sensors that are hampered by contaminants may be repaired. Contaminants cause the sensor current to drop by reducing the surface area of ​​the electrode or the diffusion pathways for the analyte and reaction byproducts. These contaminants are charged and gradually accumulate on the electrode or membrane surface under certain voltage conditions. Previously, contaminants would destroy the effectiveness of the sensor. Now, if the periodic diagnostic EIS procedure 1980 detects impedance values ​​that indicate the presence of contaminants, remedial action can be taken. Reference Figure 20 Describes when to take remedial action. Therefore, regular diagnostic EIS procedures 1980 become essential because this process can trigger sensor remedial actions that can restore sensor current to normal levels and extend sensor life. Two possible implementations of sensor remedial actions are described below. Figure 21A and Figure 21B described in .

[0275] In addition, any scheduled diagnostic EIS procedure 1980 may be aborted or rescheduled when it is determined that certain events are imminent. These events may include any situation where the patient needs to check the sensor reading, including, for example, when the patient measures his or her BG level using a test strip in order to calibrate the sensor, when the patient is warned of a calibration error and needs to measure his or her BG level again using a test strip, or when a hyperglycemic or hypoglycemic warning has been issued but not received.

[0276] Figure 20 The method of combining a diagnostic EIS procedure with sensor remediation measures according to an embodiment of the present invention is illustrated. The diagnostic procedure of block 2000 may be Figure 19 Any periodic diagnostic EIS procedure 1980 described in detail. The logic of the method begins as follows: At block 2000, a diagnostic EIS procedure is performed to detect the impedance value of the sensor. As described herein, in certain embodiments, the EIS procedure uses a combination of a DC bias and an AC voltage of varying frequencies, wherein the impedance detected by performing the EIS procedure is plotted on a Nyquist plot, and the inflection point of the Nyquist plot is approximately the sum of the polarization resistance and the solution resistance (i.e., the actual impedance value). After the diagnostic EIS procedure detects the impedance value of the sensor at block 2000, the logic moves to block 2010.

[0277] At block 2010, the impedance value is compared to the set high and low thresholds to determine if the impedance value is normal. If the impedance value is within the set high and low thresholds at block 2010, then, at block 2020, normal sensor operation is resumed, and Figure 20The logic will stop until another diagnostic EIS procedure is scheduled. Conversely, if the impedance is determined to be abnormal (i.e., outside the set high and low threshold ranges) in block 2010, then remedial action is triggered in block 2030. Examples of acceptable high and low thresholds over the life of the sensor are 5.5 kOhms and 8.5 kOhms, respectively, although these values ​​can be set higher or lower as needed.

[0278] The remedial measures of box 2030 are performed to remove any contaminants that may have caused the abnormal impedance value. In a preferred embodiment, the remedial measures are performed by applying a reverse current or reverse voltage between the working electrode and the reference electrode. The specific description of the remedial measures will be described in detail with reference to Figure 21. After the remedial measures are performed at box 2030, the impedance value is again detected by a diagnostic EIS procedure at box 2040. The success of the remedial measures is then determined at box 2050 when the impedance value from the diagnostic EIS procedure of box 2040 is compared with a set high threshold or low threshold. As in box 2010, if the impedance is within the set threshold range, then the impedance is considered normal, and if the impedance is outside the set threshold range, then the impedance is considered abnormal.

[0279] If the sensor's impedance value is determined to have returned to normal at block 2050, then normal sensor operation proceeds at block 2020. If the impedance remains abnormal, indicating that sensor age is the cause of the abnormal impedance or that remedial measures to remove contaminants have been unsuccessful, then the sensor is subsequently deactivated at block 2060. In an alternative embodiment, instead of immediately deactivating the sensor, the sensor may initially generate a sensor message requesting the user to wait and then perform further remedial measures after a set period of time has elapsed. This optional step may be combined with separate logic to determine whether the impedance value is near or within a range of upper and lower thresholds after the initial remedial measures have been performed. For example, if the sensor's impedance value has not changed, then the sensor may subsequently decide to deactivate. However, if the sensor's impedance value is near a pre-set range after the initial remedial measures, but remains outside of that range, then additional remedial measures may be performed. In yet another alternative embodiment, the sensor may generate a message requesting the user to calibrate the sensor using a finger prick measurement to further confirm whether the sensor has truly failed. All of the above embodiments are intended to prevent users from using a faulty sensor that produces inaccurate readings.

[0280] Figure 21AAn example of one embodiment of the aforementioned sensor remediation measure is described. In this embodiment, the blockage caused by the contaminant is removed by reversing the voltage applied between the working and reference electrodes of the sensor. The reversed DC voltage dissipates the charged contaminant from the electrode or membrane surface, clearing the diffusion pathway. Through this cleared pathway, the sensor current returns to normal levels, and the sensor can produce accurate readings. Thus, the remediation measure saves the user the time and expense associated with replacing an otherwise valid sensor.

[0281] Figure 21B An alternative embodiment of the above-mentioned sensor remediation measure is illustrated. In this embodiment, a reversed DC voltage applied between the working electrode and the reference electrode is combined with an AC voltage. By adding the AC voltage, some substances that are tightly adsorbed on the surface layer or on the surface layer can be removed because the AC voltage can extend its driving force further from the electrode and penetrate all layers of the sensor. The AC voltage can appear in any number of different waveforms. Some examples of waveforms that can be used include square waves, triangle waves, sine waves, or pulses. As with the previous embodiment, once the contaminants are removed, the sensor can return to normal operation, and both sensor life and accuracy are improved.

[0282] While the above embodiments primarily illustrate the use of real impedance data in sensor diagnostics, embodiments of the present invention also contemplate the use of other EIS-based, substantially analyte-independent parameters (besides real impedance) in sensor diagnostics. For example, as described above, analysis of (substantially) glucose-independent impedance data (e.g., 1 kHz real and imaginary impedance values, and the Nyquist slope) can provide information about sensor efficiency, such as how quickly the sensor hydrates and is ready to acquire data. Furthermore, (substantially) glucose-independent impedance data, such as the 1 kHz real impedance value, provides information about potential occlusions that may be present on the sensor membrane surface, which could temporarily block glucose from entering the sensor and thereby cause a signal drop.

[0283] In addition, (substantially) glucose-independent impedance data, such as high-frequency phase angle values ​​and / or imaginary impedance values ​​at 1 kHz and higher frequencies, provide information about sensor sensitivity loss during extended wear, which may potentially be caused by local hypoxia at the insertion site. In this regard, the basic mechanism of hypoxia-induced sensitivity loss can be described as follows: when there is a local lack of oxygen, the sensor output (i.e., Isig and SG) becomes dependent on oxygen rather than glucose, and thus, the sensor loses sensitivity to glucose. Other markers including 0.1 Hz real impedance, counter electrode voltage (Vcntr), and EIS-induced Isig spikes can also be used to detect hypoxia-induced sensitivity loss. Moreover, in redundant sensor systems, the relative differences in 1 kHz real impedance, 1 kHz imaginary impedance, and 0.1 Hz real impedance between two or more working electrodes can be used to detect sensitivity loss due to biocontamination.

[0284] According to embodiments of the present invention, EIS-based sensor diagnostics require consideration and analysis of EIS data related to one or more of at least three primary factors, namely, potential sensor failure modes: (1) signal start-up; (2) signal drop; and (3) sensitivity loss. Significantly, it has been discovered herein that most impedance-related parameters used in these diagnostic analyses and diagnostic procedures can be investigated at a frequency or across a range of frequencies where the parameters are substantially analyte-independent, allowing sensor-diagnostic procedures to be performed independently of analyte levels in a patient. Thus, while EIS-based sensor diagnostics can be triggered by, for example, large fluctuations in Isig that are analyte-dependent, the impedance-related parameters used in these sensor diagnostic procedures are themselves substantially independent of analyte levels. As discussed in greater detail below, it has been discovered that in most cases where glucose is observed to affect the amplitude (or other characteristics) of an EIS-based parameter, the effect is typically sufficiently small—e.g., the difference between the EIS-based measurement and the glucose that affects it is at least an order of magnitude—that the effect can be filtered out of the measurement by software in the IC.

[0285] By definition, "start-up" refers to the integrity of the sensor signal during the first few hours after insertion (e.g., t=0-6 hours). For example, in current devices, the sensor signal is considered unreliable during the first two hours after insertion, and as such, the sensor glucose values ​​are shielded from the patient / user. In cases where the sensor takes a longer time to hydrate, the sensor signal is lower for several hours after insertion. By using EIS, additional impedance information can be provided (by performing the EIS procedure) after the sensor is inserted. In this regard, the total impedance equation can be used to explain the principle behind low startup detection using 1 kHz real impedance. At relatively high frequencies - 1 kHz and above - the imaginary impedance is very small (as confirmed by in vivo data), so the total impedance is summarized as:

[0286]

[0287] As the sensor is gradually wetted, the double layer capacitance (C d ) increases. Therefore, the total impedance will decrease because, as shown in the above equation, the total impedance is proportional to C d This is inversely proportional to Figure 16B This is illustrated in the form of an intercept 1600 on the real impedance axis. Importantly, the 1 kHz imaginary impedance can also be used for the same purpose since it also includes and is inversely proportional to the capacitive component.

[0288] Another indicator for low start detection is the Nyquist slope, which is entirely dependent on the impedance at relatively low frequencies, which in turn corresponds to the Warburg impedance component of the total impedance (see, e.g. Figure 15B ). Figure 22The Nyquist slope of a properly functioning sensor is shown, where arrow A represents the progression of time, i.e., the sensor's wear time, starting at t=0. Thus, an EIS at a relatively low frequency is performed immediately after sensor insertion (t=0), generating real and imaginary impedance data that are plotted using a first linear fit 2200 having a first (Nyquist) slope. At a time interval after t=0, a second (lower frequency) frequency sweep is run, generating a second linear fit 2210 having a second (Nyquist) slope greater than the first Nyquist slope, and so on. As the sensor becomes more hydrated, the Nyquist slope increases and the intercept decreases, as reflected by lines 2200, 2210, etc., becoming steeper and closer to the Y-axis. Regarding low-start detection, clinical data show that the Nyquist slope typically increases sharply after sensor insertion and start-up, then stabilizes to a certain level. One explanation for this is that as the sensor becomes increasingly wetted, the substance's diffusivity and concentration undergo a dramatic change, which is reflected in the Warburg impedance.

[0289] exist Figure 23A In the example, the Isig 2230 of the first working electrode WE1 starts out lower than expected (about 10 nA) and takes some time to catch up to the Isig 2240 of the second working electrode WE2. Therefore, in this particular example, WE1 is designated as having a low start. The EIS data reflects this low start in two ways. First, as Figure 23A As shown, the real impedance of WE1 at 1kHz (2235) is much higher than the real impedance of WE2 at 1kHz (2245). Second, when compared with the Nyquist slope of WE2 ( Figure 23C ), Nyquist slope of WE1 ( Figure 23B ) starts lower, has a larger intercept 2237, and takes more time to stabilize. As discussed later, these two markers—1 kHz real impedance and Nyquist slope—can be used as diagnostic inputs in the fusion algorithm to determine which of the two electrodes should carry a higher weight when calculating the fused signal. In addition, one or both of these two markers can be used in a diagnostic procedure to determine whether the sensor as a whole is acceptable or whether the sensor should be decommissioned and replaced.

[0290] By definition, a signal (or Isig) dip is an instance of low sensor signal that is mostly temporary in nature, e.g., a few hours. Such a low signal could be caused by, for example, some form of biological occlusion on the sensor surface, or by pressure exerted on the insertion site (e.g., when sleeping on one's side). During this period, the sensor data is considered unreliable, however, the signal does eventually recover. In EIS data, this type of signal dip—as opposed to a signal dip caused by changes in blood glucose within the patient—is reflected in the 1 kHz real impedance data, e.g., Figure 24 shown.

[0291] Specifically, in Figure 24 , both Isig 2250 of the first working electrode WE1 and Isig 2260 of the second working electrode WE2 start at about 25nA at the far left end (i.e., 6pm). Over time, these two Isigs fluctuate, reflecting the glucose fluctuations near the sensor. For about the first 12 hours (i.e., until about 6am), both Isigs are quite stable, as are their respective 1kHz real impedances 2255, 2265. However, between about 12 and 18 hours—i.e., 6am to noon—Isig 2260 of WE2 begins to decline and continues on a downward trend for the next few hours until about 9pm. During this period, Isig 2250 of WE1 also shows some decline, but Isig 2250 is much more stable than Isig 2260 of WE2 and declines less than Isig 2260 of WE2. The changes in Isig of WE1 and WE2 are also reflected in their respective 1kHz real impedance data. Therefore, as shown in Figure 24 As shown, during the above time period, although the 1 kHz real impedance of WE1 (2255) remains quite stable, the 1 kHz real impedance of WE2 (2265) increases significantly.

[0292] By definition, sensitivity loss is a condition where the sensor signal (Isig) becomes low and unresponsive for an extended period of time and generally cannot be recovered. Sensitivity loss can occur for a variety of reasons. For example, electrode poisoning significantly reduces the active surface area of ​​the working electrode, severely limiting the current amplitude. Sensitivity loss can also occur due to hypoxia or oxygen deficiency at the insertion site. In addition, sensitivity loss can occur due to some forms of extreme surface occlusion (i.e., a more permanent form of signal drop caused by biological or other factors) that restricts the passage of glucose and oxygen through the sensor membrane, thereby reducing the number / frequency of chemical reactions that generate current in the electrode and ultimately the sensor signal (Isig). It should be noted that the various causes of sensitivity loss described above apply to both short-term (7 to 10 days of wear) and long-term (6 months of wear) sensors.

[0293] In EIS data, the absolute value of phase (|phase|) and the absolute value of imaginary impedance (|imaginary impedance|) generally increase in a relatively high frequency range (eg, 128 Hz and above, and 1 kHz and above) before sensitivity is lost. Figure 25A An example of a properly functioning glucose sensor is shown in which the sensor current 2500 responds to glucose—i.e., Isig 2500 follows glucose fluctuations—but the phase of all relevant impedance outputs, such as the 1 kHz real impedance 2510, the 1 kHz imaginary impedance 2530, and frequencies around 128 Hz or greater (2520), remain constant because they are essentially independent of glucose.

[0294] Specifically, Figure 25A The top graph in FIG shows that after the first few hours, the 1 kHz real impedance 2510 remains fairly stable at approximately 5 kOhms (and the 1 kHz imaginary impedance 2530 remains fairly stable at approximately -400 Ohms). In other words, at 1 kHz, the real impedance data 2510 and the imaginary impedance data 2530 are essentially independent of glucose, and thus, depending on the analysis, can be used as a marker or independent indicator of the health status and ultimately the reliability of a particular sensor. However, as previously mentioned, different impedance-related parameters may exhibit glucose independence over different frequency ranges, and in each case, the range may depend on the overall sensor design, e.g., electrode type, electrode surface area, membrane thickness, membrane permeability, etc.

[0295] Therefore, in the embodiment Figure 25B In the -90% short tubeless electrode design, the top graph again shows that the sensor current 2501 responds to glucose, and after the first few hours, the 1 kHz real impedance 2511 remains fairly stable at about 7.5 kOhms. Figure 25B The bottom graph in FIG shows real impedance data at frequencies between 0.1 Hz (2518) and 1 kHz (2511). As can be seen from the figure, the real impedance data at 0.1 Hz (2518) is substantially dependent on glucose. However, as shown by reference numerals 2516, 2514, and 2512, the real impedance becomes increasingly independent of glucose as the frequency increases from 0.1 Hz to 1 kHz (i.e., the impedance data is measured closer to 1 kHz).

[0296] Back to Figure 25A, the middle graph shows that the phase 2520 at relatively high frequencies is essentially independent of glucose. However, it should be noted that "relatively high frequencies" in relation to this parameter (phase) of the sensor during analysis means frequencies of 128 Hz and above. In this regard, the graph shows that the phase of all frequencies between 128 Hz and 8 kHz is stable over the entire time period shown. On the other hand, as Figure 25C As shown in the bottom graph of FIG, while the phase 2522 is stable at 128 Hz (and above), the phase 2524 fluctuates at frequencies progressively less than 128 Hz—that is, the phase becomes increasingly glucose-dependent and the phase degree changes. Figure 25C The electrode design of the embodiment is Figure 25B The electrodes used are of the same design and Figure 25C The top picture and Figure 25B Top Figure 1 To.

[0297] Figure 26 An example of sensitivity loss due to hypoxia at the insertion site is shown. In this case, the insertion site was Figure 26 ) immediately after the insertion site, which becomes hypoxic, causing the sensor current 2600 to become low and unresponsive. The 1 kHz real impedance 2610 remains stable, indicating that there is no physical occlusion at the sensor. However, as indicated by the respective downward arrows, the changes in the relatively high frequency phase 2622 and the 1 kHz imaginary impedance 2632 are consistent with a loss of sensitivity, indicating that this type of loss is due to hypoxia at the insertion site. Specifically, Figure 26 It is shown that the phase at higher frequencies (2620) and the 1 kHz imaginary impedance (2630) become more negative before the sensor loses sensitivity—represented by the black vertical line—and continue to trend downward as the sensor sensitivity continues to lose. Thus, as described above, such sensitivity loss is preceded by an increase in the absolute value of the phase (|phase|) and the absolute value of the imaginary impedance (|imaginary impedance|) within the relatively higher frequency ranges (e.g., 128 Hz and above, and 1 kHz and above), or in other words, such sensitivity loss is predicted by an increase in the absolute value of the phase (|phase|) and the absolute value of the imaginary impedance (|imaginary impedance|) within the relatively higher frequency ranges (e.g., 128 Hz and above, and 1 kHz and above).

[0298] The above mentioned markers can be confirmed by in vitro tests. Figure 27 Examples of in vitro tests are shown. Figure 27Results from in vitro testing of the sensor, simulating hypoxia under conditions of varying glucose concentrations, are shown. In the top graph, Isig fluctuates as glucose concentration increases from 100 mg / dl (2710) to 200 mg / dl (2720), 300 mg / dl (2730), 400 mg / dl (2740), and then back down to 200 mg / dl (2750). In the bottom graph, the phase at relatively high frequencies is generally stable, indicating that the phase is independent of glucose. However, at very low oxygen concentrations, e.g., 0.1% O2, the phase fluctuates at relatively high frequencies, as indicated by the circled area and arrows 2760 and 2770. It should be noted that the magnitude and / or direction (i.e., positive or negative) of the fluctuations depends on various factors. For example, the higher the ratio of glucose to oxygen concentration, the higher the magnitude of the phase fluctuations. Furthermore, the specific sensor design and sensor age (i.e., measured over time since implantation) influence these fluctuations. So, for example, the older the sensor, the more prone it is to fluctuations.

[0299] 28A to 28D Another example of sensitivity loss due to oxygen deficiency using redundant working electrodes WE1 and WE2 is shown. Figure 28A As shown, the 1kHz real impedance 2810 is stable even as the sensor current 2800 fluctuates and eventually becomes unresponsive. Moreover, as previously mentioned, the changes in the 1kHz imaginary impedance 2820 are consistent with the loss of sensor sensitivity. However, in addition, Figure 28B The real and imaginary impedance data at 0.105 Hz are shown (represented as 2830 and 2840, respectively). The latter, more commonly referred to as the "0.1 kHz data," shows that although the imaginary impedance at 0.1 kHz appears fairly stable, the real impedance 2830 at 0.1 kHz increases significantly as the sensor loses sensitivity. Furthermore, as Figure 28C As shown, under the condition of sensitivity loss due to oxygen deficiency, V cntr The 2850 reaches the voltage rail at 1.2 volts.

[0300] Briefly, the graph illustrates the finding that the sensitivity loss due to hypoxia is associated with a lower 1kHz imaginary impedance (i.e., the latter becomes more negative), a higher 0.105Hz real impedance (i.e., the latter becomes more positive), and V cntr Moreover, the hypoxia process is related to the Vcntr-rail (the counter electrode voltage rail: V cntr Rail) is generally associated with an increase in the capacitive component of the electrochemical circuit. It should be noted that in some diagnostic procedures to be described later, 0.105 Hz real impedance may not be used because this relatively low frequency real impedance data appears to be analyte-dependent.

[0301] Finally, about 28A to 28D For example, it should be noted that 1kHz or higher frequency impedance measurements usually cause EIS-induced spikes in Isig. Figure 28D , where the raw Isig of WE2 is plotted against time. The significant increase in Isig at the start of the spike is a non-Faradaic process due to the double layer capacitance charging. Therefore, in addition to the lower 1kHz imaginary impedance, the higher 0.105Hz real impedance, and V discussed above cntr Besides the sensitivity loss due to hypoxia, the sensitivity loss can also be associated with higher EIS-induced spike signals.

[0302] Figure 29 Another example of sensitivity loss is given. This example can be considered as Figure 24 Here, the sensor current 2910 is observed to decrease from the time of insertion, indicating that a problem occurred during the insertion process that caused the electrode to occlude. Figure 25A For the same parameter values ​​shown for a properly functioning sensor, the 1 kHz real impedance 2920 is significantly higher, while the relatively high frequency phase 2930 and the 1 kHz imaginary impedance 2940 both change to more negative values. The changes in the relatively high frequency phase 2930 and the 1 kHz imaginary impedance 2940 indicate a loss of sensitivity, possibly due to hypoxia, which in turn may be caused by occlusion of the sensor surface.

[0303] 30A to 30D Data for another redundant sensor is shown, in which the relative differences in the 1 kHz real impedance and the 1 kHz imaginary impedance, as well as the 0.1 Hz real impedance, between two or more working electrodes can be used to detect sensitivity loss due to biofouling. In this example, WE1 exhibits more sensitivity loss than WE2, as evident by the higher 1 kHz real impedance 3010 of WE2, the lower 1 kHz imaginary impedance 3020 of WE2, and the much higher real impedance of WE2 at 0.105 Hz (3030). However, in addition, in this example, V cntr The 3050 does not reach the voltage rail. Moreover, as Figure 30D As shown in Figure 2, the height of the spike signal in the raw Isig data does not change much over time. This indicates that for the sensitivity loss caused by biofouling, V cntr - rail and the increase in spike height are correlated. In addition, the fact that the spike height in the raw Isig data does not change much over time suggests that the capacitive component of the circuit does not change significantly over time, so the sensitivity loss caused by biofouling is related to the resistive component of the circuit (i.e., diffusion).

[0304] The various impedance-related parameters described above can be used individually or in combination as input to: (1) EIS-based sensor diagnostics; and / or (2) fusion algorithms for generating more reliable sensor glucose values. Figure 31 It illustrates how EIS-based data—ie, impedance-related parameters or signatures—can be used in diagnostic procedures to determine in real time whether a sensor is functioning properly or should be replaced.

[0305] Figure 31 The diagnostic procedure shown in the flow chart of is based on the periodic acquisition of EIS data, for example, every hour, every half hour, every ten minutes, or at any other interval—including continuous acquisition—as may be appropriate for the particular sensor under analysis. In each such interval, the EIS may be run across the entire frequency spectrum (i.e., a "full sweep"), or may be run over a selected frequency range, or may even be run at a single frequency. Thus, for example, for a scheme where data is collected every hour, the EIS may be performed over a frequency range from μHz to MHz, or may be run over a narrower frequency range, such as about 0.1 Hz to about 8 kHz, as described above. In embodiments of the invention, EIS data acquisition may alternate between a full sweep and a narrower frequency spectrum, or may be performed according to other schemes.

[0306] The temporal frequency of EIS execution and data acquisition can be dictated by various factors. For example, each execution of EIS consumes a certain amount of power, which is typically provided by the sensor's battery, i.e., the battery that runs the sensor electronics, including the ASIC described below. Thus, the battery capacity and the remaining sensor life can help determine the number of EIS runs and the frequency width of the samples taken during each run. Furthermore, embodiments of the present invention contemplate situations in which it may be desired to monitor an EIS parameter (e.g., real impedance at 1 kHz) at a particular frequency based on a first schedule (e.g., every few seconds or minutes), while the same parameter may be monitored at other frequencies based on a second schedule (e.g., at a lower frequency). In these situations, the diagnostic routine can be tailored to the specific sensor and requirements so that battery power can be conserved and unnecessary and / or redundant EIS data acquisition can be avoided.

[0307] It should be noted that in embodiments of the present invention, diagnostic procedures, e.g. Figure 31The diagnostic procedure shown performs a series of individual "tests" that are performed to monitor the sensor in real time. Multiple tests or signatures—also referred to as "multiple signatures"—are used because each time an EIS is run (i.e., each time an EIS procedure is performed), data about multiple impedance-based parameters or characteristics can be collected, which can be used to detect sensor condition or quality, including whether the sensor has failed or malfunctioned. In performing sensor diagnostics, sometimes a diagnostic test may indicate a sensor malfunction, while other diagnostics may indicate no malfunction. Therefore, the availability of multiple impedance-related parameters and the implementation of a multi-test procedure are advantageous because some of the multiple tests can serve as validity checks for some of the other tests. Therefore, real-time monitoring using a multi-signature approach can include a degree of built-in redundancy.

[0308] From the above content, we can see that Figure 31 The logic of the diagnostic procedure shown begins at step 3100 after the sensor has been inserted / implanted and EIS is run, providing EIS data as input. At step 3100, using the EIS data as input, it is first determined whether the sensor is still in the proper position. Thus, if the |Z| slope is found to be constant across the entire test frequency band (or range), and / or the phase angle is approximately -90°, then the sensor is determined to be no longer in the proper position and an alert is sent, for example, to the patient / user, indicating that the sensor has been removed. The specific parameters (and their respective values) described herein for detecting sensor removal are based on the discovery that once the sensor is outside the body and the membrane is no longer hydrated, the impedance spectrum response looks like a capacitor.

[0309] If it is determined that the sensor is still in the proper position, then the logic moves to step 3110 to determine if the sensor has been properly initialized. As shown, "Init.Check" is performed by determining the following: (1) Is |(Z n -Z1) / Z1|>30%, where, as discussed above, Z1 is the real impedance measured at the first time, Z n is the impedance measured at the next interval, and (2) whether the change in phase angle is greater than 10° at 0.1 Hz. If the answer to either of the above two questions is "yes", then the test is satisfied, that is, Test 1 passes. Otherwise, Test 1 is marked as failed.

[0310] In step 3120, Test 2 asks whether the frequency difference (f2-f1) between two consecutive EIS runs is greater than 10 Hz at a phase angle of -45°. Again, a "no" answer is marked as a failure, otherwise, Test 2 is satisfactory.

[0311] Test 3 in step 3130 is a hydration test. In this step, the current impedance Z is queried at 1kHz. n Is it less than the initial impedance Z? pi If yes, then the test is satisfied, otherwise, test 3 is marked as failed. Test 4 in step 3140 is also a hydration test, but this time performed at a lower frequency. Thus, this test asks if Z is 0.1 Hz during operation of the sensor after initialization. n Is it less than 300k ohms? Again, a "no" answer means the sensor has failed test 4.

[0312] In step 3150, Test 5 asks whether the low-frequency Nyquist slope increases over the entire range from 0.1 Hz to 1 Hz. As discussed above, for a properly functioning sensor, the relatively low-frequency Nyquist slope should increase over time. Therefore, if the answer to this query is "yes," then the test is satisfied; otherwise, the test is marked as failed.

[0313] Step 3160 is the final test of this diagnostic procedure implementation. In this step, the question is asked whether the real impedance is decreasing over the entire range. Again, as discussed above, in a properly functioning sensor, it is expected that the real impedance will decrease over time. Therefore, a "yes" answer in this step indicates that the sensor is functioning properly; otherwise, the sensor fails Test 6.

[0314] Once all six tests have been performed, a determination is made in 3170 as to whether the sensor is operating normally or is faulty. In this embodiment, if the sensor passes at least three of the six tests, then the sensor is determined to be operating normally (3172). In other words, in order to determine that the sensor has faulted (3174), the sensor must fail at least four of the six tests. In alternative embodiments, different rules may be used to evaluate normal operation and sensor faults. Furthermore, in embodiments of the present invention, each test may be weighted such that the assigned weight reflects, for example, the importance of the test or the importance of the particular parameter questioned by the test in determining overall sensor operation (normal vs. faulty). For example, one test may be weighted twice as much as another test, but only half as much as a third test, and so on.

[0315] In other alternative embodiments, a different number of tests and / or a different set of EIS-based parameters for each test may be used. Figure 32A and Figure 32B An example of a diagnostic program for real-time monitoring consisting of seven tests is shown. Figure 32AAfter the sensor has been inserted / implanted, the logic begins at step 3200 and an EIS procedure is executed, providing EIS data as input. At 3200, using the EIS data as input, a determination is first made as to whether the sensor is still properly positioned. Thus, if the |Z| slope is found to be constant within the detection frequency bandwidth (or range), and / or the phase angle is approximately -90°, then the sensor is determined to no longer be properly positioned, and an alert is sent, e.g., to the patient / user, indicating that the sensor has been removed. On the other hand, if the sensor is determined to be properly positioned, then the logic moves to starting a diagnostic check (3202).

[0316] In step 3205, Test 1 is similar to the above with respect to Figure 31 Test 1 of the diagnostic procedure discussed, except that Test 1 of this embodiment specifies that Z be measured two hours after the first measurement. n Thus, in this embodiment, Z n =Z 2hr . More specifically, Test 1 compares the actual impedance two hours after (sensor implantation and) initialization to the value before initialization. Similarly, another part of Test 1 asks whether the difference between the phase two hours after initialization and the phase before initialization is greater than 10° under 0.1 Hz conditions. As previously described, if the answer to any of the inquiries is yes, then the sensor is determined to be normally hydrated and initialized, and Test 1 is satisfied, otherwise, the sensor fails the test. It should be noted that even though the test of this embodiment asks about the impedance and phase changes two hours after initialization, the time interval between any two consecutive EIS runs can be shorter or longer, depending on a variety of factors, including, for example, sensor design, electrode redundancy level, the extent to which the diagnostic program includes redundant testing, battery power, etc.

[0317] Moving to step 3210, the logic then performs a sensitivity loss check by asking whether the percentage change in impedance magnitude at 1 kHz and the percentage change in Isig after the two-hour interval (n+2) are greater than 30%. If the answer to both of these queries is "yes," then the sensor is determined to have lost sensitivity, and thus, Test 2 is determined to have failed. It should be noted that while Test 2 is illustrated herein based on a preferred 30% difference percentage, in other embodiments, the percentage difference in impedance magnitude at 1 kHz and the percentage difference in Isig may fall within the range of 10% to 50% for the purposes of performing this test.

[0318] Test 3 (in 3220) is similar to Figure 311 . Example Test 5 algorithm. In this example, as previously described, the question is whether the low frequency Nyquist slope increases over the entire range from 0.1 Hz to 1 Hz. If so, then the test passes, otherwise, the test fails. As shown in 3220, the test can also be modified to set a threshold or acceptable range for the percentage change in the low frequency Nyquist slope, above which the sensor is considered to have failed the test, or at the very least, further diagnostic testing can be triggered. In embodiments of the invention, such a threshold / acceptable range for the percentage change in the low frequency Nyquist slope can fall within the range of about 2% to about 20%. In some preferred embodiments, the threshold can be about 5%.

[0319] The logic then moves to step 3230, which is another low frequency test, this time involving phase and impedance magnitude. More specifically, the phase test asks whether the phase at 0.1 Hz is increasing consistently over time. If so, the test fails. As with the other tests involving monitoring parameter trends, the low frequency phase test of Test 4 can also be modified to set a threshold or acceptable range for the percentage change in low frequency phase, beyond which the sensor can be considered to have failed the test, or at the very least, to be a cause for concern. In embodiments of the present invention, such a threshold / acceptable range for the percentage change in low frequency phase can fall within the range of about 5% to about 30%. In some preferred embodiments, the threshold can be about 10%.

[0320] As mentioned above, Test 4 also includes a low-frequency impedance magnitude test, which inquires whether the impedance magnitude at 0.1 Hz continues to increase over time. If so, the test fails. It should be noted that Test 4 is considered "failed" if either the phase test or the impedance magnitude test fails. The low-frequency impedance magnitude test of Test 4 can also be modified to set a threshold or acceptable range for the percentage change in low-frequency impedance magnitude, beyond which the sensor can be considered to have failed the test, or at the very least, to be a cause of concern. In embodiments of the present invention, such a threshold / acceptable range for the percentage change in low-frequency impedance magnitude can fall within a range of approximately 5% to approximately 30%. In some preferred embodiments, the threshold can be approximately 10%, where the impedance magnitude of a normal sensor typically ranges from approximately 100K ohms to approximately 200K ohms.

[0321] Test 5 (in step 3240) is another sensitivity loss check that can be considered complementary to Test 2. In this test, if the percentage change in Isig and the percentage change in impedance magnitude at 1 kHz are both greater than 30%, then the sensor is determined to be recovering from sensitivity loss. In other words, the sensor is determined to have previously experienced some sensitivity loss, even if, for some reason, sensitivity loss was not detected by Test 2. As with Test 2, while Test 5 is illustrated based on a preferred 30% difference percentage, in other embodiments, for the purposes of this test, the percentage difference in Isig and the percentage difference in impedance magnitude at 1 kHz may fall within the range of 10% to 50%.

[0322] Moving to step 3250, Test 6 provides a functional test of the sensor using specific failure criteria, which are determined based on the observed data and the specific sensor design. Specifically, in one embodiment, if at least two of the following three criteria are met: (1) Isig is less than 10nA; and (2) the imaginary impedance at 1kHz is less than -1500 ohms; and (3) the phase at 1kHz is less than -15°, then the sensor may be determined to have failed the test, such that the sensor is unlikely to be responsive to glucose. Therefore, if any two of (1) to (3) are not met, then Test 6 is determined to have passed. It should be noted that in other embodiments, if Isig is less than about 5nA to about 20nA, then the Isig test in this test may fail. Similarly, if the imaginary impedance at 1kHz is less than about -1000 ohms to about -2000 ohms, then the second test may fail. Finally, if the phase at 1kHz is less than about -10° to about -20°, then the phase test may fail.

[0323] Finally, step 3260 provides another sensitivity check, in which the parameters are evaluated under low-frequency conditions. Thus, Test 7 asks whether the difference between the ratio of the imaginary impedance to Isig(n+2) at 0.1 Hz and the previous ratio is greater than 30% of the magnitude of the previous ratio. If so, the test fails; otherwise, the test passes. In this embodiment, while Test 7 is illustrated based on a preferred 30% difference percentage, in other embodiments, the difference percentage may be within the range of 10% to 50% for the purpose of performing this test.

[0324] Once all seven tests have been performed, a decision is made in 3270 as to whether the sensor is operating properly or whether an alarm should be issued to indicate a sensor failure (or possible failure). As shown, in this embodiment, if the sensor passes at least four of the seven tests, then the sensor is determined to be operating properly (3272). In other words, in order to determine that the sensor has failed or at least caused concern (3274), the sensor must fail at least four of the seven tests. If the sensor is determined to be "bad" (3274), then an alarm indicating this condition can be sent to, for example, the patient / user. As previously described, in alternative embodiments, different rules can be used to evaluate normal operation and sensor failure / concern. In addition, in embodiments of the present invention, each test can be weighted so that the assigned weight reflects, for example, the importance of the test, or the importance of the specific parameter interrogated by the test, in determining the overall operation of the sensor (normal vs. failure).

[0325] As described above, in embodiments of the present invention, the various impedance-related parameters described above may be used individually or in combination as inputs to one or more fusion algorithms for generating more reliable sensor glucose values. Specifically, it is known that, unlike a single sensor (i.e., a single working electrode) system, multiple detection electrodes provide glucose readings with higher reliability, as multiple signals, the glucose readings are obtained from two or more working electrodes, which can be fused to provide a single sensor glucose value. This signal fusion uses the quantitative input provided by EIS to calculate the most reliable output sensor glucose value for the redundant working electrodes. It should be noted that although the following discussion may describe various different fusion algorithms based on the first working electrode (WE1) and the second working electrode (WE2) as redundant electrodes, this is by way of example and not limitation, as the algorithms and their underlying principles described herein are applicable to and can be used in redundant sensor systems having more than two working electrodes.

[0326] Figure 33A and Figure 33B A top-level flow chart for two alternative methods is shown, each of which includes a fusion algorithm. Specifically, Figure 33A This is a flow chart of the fusion algorithm based on current (Isig), Figure 33B Figure 1 is a flow chart involving sensor glucose (SG) fusion. As can be seen from the figure, the main difference between the two methods is the calibration time. Therefore, Figure 33AAs shown, for Isig fusion, calibration 3590 is performed after fusion 3540 is completed. That is, the redundant Isigs from WE1 to WEn are fused into a single Isig 3589, which is then calibrated to produce a single sensor glucose value 3598. On the other hand, for SG fusion, calibration 3435 is completed for each individual Isig from WE1 to WEn to produce a calibrated SG value (e.g., 3436, 3438) for each working electrode. Thus, the SG fusion algorithm provides independent calibration of each of the multiple Isigs, which may be preferred in embodiments of the present invention. Once calibrated, the multiple calibrated SG values ​​are fused into a single SG value 3498.

[0327] It is important to note that Figure 33A and Figure 33B Each of the flowcharts shown in includes a spike filtering process (3520, 3420). As described above in the discussion of sensitivity loss, frequency impedance measurements of 1 kHz or higher typically cause EIS-induced spikes in Isig. Therefore, once the EIS procedure is performed on each of the electrodes WE1 to WEn, it is preferred that Isig 3410, 3412, etc. and 3510, 3512, etc. are first filtered for both SG fusion and Isig fusion to obtain respective filtered Isig 3422, 3424, etc. and 3522, 3524, etc. As discussed in detail below, the filtered Isigs are then used for Isig fusion, or are first calibrated and then used for SG fusion. As will become apparent in the subsequent discussion, the two fusion algorithms calculate and assign weights based on a variety of different factors.

[0328] Figure 34 Details of the fusion algorithm 3440 for SG fusion are shown. Basically, four factors need to be checked before determining the fusion weights. First, the integrity check 3450 includes determining whether each of the following parameters falls within a specified range (e.g., predetermined lower and upper thresholds) for normal sensor operation: (i) Isig; (ii) 1kHz real impedance and imaginary impedance; (iii) 0.105Hz real impedance and imaginary impedance; and (iv) Nyquist slope. As shown, the integrity check 3450 includes a bound check 3452 and a noise check 3456, where the above parameters are used as input parameters for each check. It should be noted that, in short, the real impedance and / or imaginary impedance under the conditions of one frequency or more than one frequency appear in Figures 33A to 35 Impedance is abbreviated as "Imp" in the text. In addition, both real and imaginary impedances can be expressed using impedance magnitude and phase (in Figure 33A and Figure 33B) is calculated.

[0329] The output of each of the bound check 3452 and the noise check 3458 is a respective reliability index (RI) for each of the redundant working electrodes. Thus, the output of the bound check includes, for example, RI_bound_We1 (3543) and RI_bound_We2 (3454). Similarly, for the noise check, the output includes, for example, RI_noise_We1 (3457) and RI_noise_We2 (3458). The bound and noise reliability indices for each working electrode can be calculated based on compliance with the above ranges for normal sensor operation. Thus, if any of the parameters is outside the specified range for a particular electrode, the reliability index for that particular electrode is reduced.

[0330] It should be noted that the thresholds or ranges for the above parameters may depend on a variety of factors, including the specific sensor and / or electrode design. Nevertheless, in a preferred embodiment, typical ranges for some of the above parameters may be as follows: limit threshold for 1 kHz real impedance = [0.3e+4 2e+4]; limit threshold for 1 kHz imaginary impedance = [-2e+3, 0]; limit threshold for 0.105 Hz real impedance = [2e+47e+4]; limit threshold for 0.105 imaginary impedance = [-2e+5-0.25e+5]; limit threshold for Nyquist slope = [2 5]. Noise can be calculated, for example, using a second-order central difference method, where the noise is considered to exceed the noise limit if it is above a certain percentage (e.g., 30%) of the median value of each variable buffer.

[0331] Second, the sensor current (Isig) and 1kHz real impedance can be used to detect sensor drops. Figure 34 As shown, Isig and Imp are used as inputs for dip detection 3460. Here, the first step is to determine if there is any divergence between Isigs and whether any such divergence is reflected in the 1kHz real impedance data. This is achieved by using a mapping 3465 between the Isig similarity index (RI_sim_isig12) 3463 and the 1kHz real impedance similarity index (RI_sim_imp12) 3464. This mapping is very important because it helps avoid false positives in cases where dips are not real. In the case where Isig deviation is real, the algorithm will choose the sensor with the higher Isig.

[0332] According to an embodiment of the present invention, the divergence / convergence of two signals (e.g., two Isig, or two 1 kHz real impedance data points) can be calculated as follows:

[0333] diff_va1=abs(va1-(va1+va2) / 2);

[0334] diff_va2=abs(va2-(va1+va2) / 2);

[0335] RI_sim=1-(diff_va1+diff_va2) / (mean(abs(va1+va2)) / 4)

[0336] Where va1 and va2 are two variables, and RI_sim (similarity index) is an index that measures the convergence or divergence of the signal. In this embodiment, RI_sim must be between 0 and 1. Therefore, if RI_sim calculated as above is less than 0, RI_sim is set to 0, and if it is greater than 1, RI_sim is set to 1.

[0337] The mapping 3465 is performed by using ordinary linear regression (OLR). However, when OLR does not work well, a robust median slope linear regression (RMSLR) can be used. For example, for the Isig similarity index and the 1kHz real impedance index, two mapping processes are required: (i) mapping the Isig similarity index to the 1kHz real impedance similarity index; and (ii) mapping the 1kHz real impedance similarity index to the Isig similarity index. These two mapping processes will produce two residuals: res12 and res21. Each of the descent reliability indices 3467, 3468 can then be calculated as follows:

[0338] RI_dip=1–(res12+res21) / (RI_sim_isig+RI_sim_1K_real_impedance).

[0339] The third factor is sensitivity loss 3470, which can be detected using, for example, the trend of the 1 kHz imaginary impedance over the past eight hours. If the trend of one sensor becomes negative, the algorithm will rely on the other sensor. If both sensors lose sensitivity, a simple average is taken. The trend can be calculated by using a strong low-pass filter to eliminate the 1 kHz imaginary impedance, which becomes noise, and the trend can be calculated by using a correlation coefficient or linear regression related to time (for example, the past eight hours) to determine whether the correlation coefficient is negative or the slope is negative. Each of the sensitivity loss reliability indices 3473, 3474 is then assigned a binary value of 1 or 0.

[0340] The overall reliability index (RI) of each of we1, we2, ... wen is calculated as follows:

[0341] RI_we1=RI_dip_we1×RI_sensitivity_loss_we1×RI_bound_we1×RI_noise_we1

[0342] RI_we2=RI_dip_we2×RI_sensitivity_loss_we2×RI_bound_we2×RI_noise_we2

[0343] RI_we3=RI_dip_we3×RI_sensitivity_loss_we3×RI_bound_we3×RI_noise_we3

[0344] RI_we4=RI_dip_we4×RI_sensitivity_loss_we4×RI_bound_we4×RI_noise_we4 . . .

[0348] RI_we n =RI_dip_we n ×RI_sensitivity_loss_we n ×RI_bound_we n ×RI_noise_we n

[0349] After calculating the reliability index of each working electrode, the weight of each electrode is calculated as follows:

[0350] weight_we1=RI_we1 / (RI_we1+RI_we2+RI_we3+RI_we4+…+RI_we n )

[0351] weight_we2=RI_we2 / (RI_we1+RI_we2+RI_we3+RI_we4+…+RI_we n )

[0352] weight_we3=RI_we3 / (RI_we1+RI_we2+RI_we3+RI_we4+…+RI_we n )

[0353] weight_we4=RI_we4 / (RI_we1+RI_we2+RI_we3+RI_we4+…+RI_we n ) . . .

[0357] weight_we n =RI_we n / (RI_we1+RI_we2+RI_we3+RI_we4+…+RI_we n )

[0358] Based on the above, the fused SG 3498 is then calculated as follows:

[0359] SG=weight_we1×SG_we1+weight_we2×SG_we2+weight_we3×SG_we3+weight_we4×SG_we4+...+weight_we n ×SG_we n

[0360] The last factor involves false information in the final sensor readout, which may be caused, for example, by transient weight changes in sensor fusion. This can be avoided by using a low-pass filter 3480 to smooth the RI of each electrode or by applying a low-pass filter to the final SG. When the former is used, the filtered reliability indices—for example, RI_We1* and RI_We2* (3482, 3484)—are used to calculate the weights of each electrode and, from there, the fused SG 3498.

[0361] Figure 35 The details of the Isig fusion algorithm 3540 are shown. As can be seen from the figure, the algorithm is the same as Figure 34 The fusion algorithm shown for SG fusion is essentially similar with only two differences. First, as previously mentioned, for Isig fusion, calibration constitutes the final step of the method, where the single fused Isig 3589 is calibrated to produce a single sensor glucose value 3598. See also Figure 33B Second, although SG fusion uses SG values ​​of multiple electrodes to calculate the final SG value 3498, the fused Isig value 3589 is calculated by using filtered Isig (3522, 3524, etc.) of the multiple electrodes.

[0362] In a closed-loop study involving non-diabetic subjects, the fusion algorithm described above was found to provide significant improvements in mean absolute relative difference (MARD) on day one and over time (i.e., over the seven-day life of the sensor), where the low start-up problem is most pronounced on day one and can have a significant impact on sensor accuracy and reliability. The study evaluated an 88% distributed array design with high current density (rated) plating using three different methods: (1) calculating a sensor glucose value (SG) by fusion using Medtronic Minimed's Ferrari Algorithm 1.0 (this is the SG fusion algorithm discussed previously); (2) calculating a SG by identifying better ISIG values ​​using 1 kHz EIS data (using the Isig fusion algorithm described above); and (3) calculating a SG using higher ISIG values ​​(i.e., without using EIS). Details of the study data are as follows:

[0363] (1) For 88% distributed arrangement with high current density (rated) coating, SG based on Ferrari 1.0Alg

[0364]

[0365]

[0366]

[0367]

[0368]

[0369]

[0370] (2) For 88% distributed arrangement with high current density (rated) coating, SG based on better ISIG using 1kHz EIS

[0371]

[0372]

[0373]

[0374]

[0375]

[0376] (3) For 88% distributed arrangement with high current density (rated) coating, SG based on higher ISIG

[0377]

[0378]

[0379]

[0380]

[0381]

[0382] It was found from the above data that, by the first method, the MARD (%) on the first day was 19.52%, and the MARD for the entire process was 12.28%. For the second method, the MARD on the first day was 15.96%, and the MARD for the entire process was 11.83%. Finally, for the third method, the MARD on the first day was 17.44%, and the MARD for the entire process was 12.26%. Therefore, for this design using redundant electrodes, it can be seen that the SG calculated based on the better ISIG using 1kHz EIS (i.e., the second method) provides the greatest advantage. Specifically, the lower first-day MARD can be attributed to, for example, better detection of low starts using EIS. In addition, the percentage of MARD for the entire process in this study was lower than the average MARD of 13.5% for WE1 and WE2, which is greater than 1%. It should be noted that in the above method, data conversion can be controlled by, for example, filtering methods to minimize the degree of conversion, such as by using the above-mentioned Figures 33A to 35 The low-pass filter 3480 in question.

[0383] It is worth repeating that sensor diagnostics, including, for example, low start-up assessments, sensitivity loss assessments, and signal drop event assessments, depend on a variety of factors, including sensor design, number of electrodes (i.e., redundancy), electrode distribution / configuration, and the like. Thus, the actual frequency or frequency range based on the ESI parameter may be substantially independent of glucose, and thus, the individual signatures or indicators for one or more of the aforementioned failure modes may also depend on the specific sensor design. For example, as described above, while it has been found that sensitivity loss can be predicted using imaginary impedance at relatively high frequencies—wherein the imaginary impedance is substantially independent of glucose—the glucose-dependent level, therefore, the specific frequency range of the imaginary impedance that is indicative of sensitivity loss may vary (become higher or lower) depending on the actual sensor design.

[0384] More specifically, as sensor designs increasingly move toward redundant working electrodes, the latter must become increasingly smaller in order to maintain the overall size of the sensor. The size of the electrodes, in turn, affects the frequency required for a particular diagnosis. In this regard, it is important to note that the Figures 33A to 35The fusion algorithms shown are to be considered exemplary and not limiting, and based on the type of sensor under analysis, each algorithm may be modified as needed to utilize EIS-based parameters at frequencies that exhibit the least amount of glucose dependence.

[0385] Furthermore, experimental data suggest that human tissue structure can influence glucose dependence at different frequencies. For example, in children, real impedance at 0.105 Hz was found to be a low-start detection indicator that is largely independent of glucose. This is believed to be due to changes in children's tissue structure, such as the Warburg impedance, which primarily involves a resistive component. See also the discussion below on interfering substance detection.

[0386] Embodiments of the invention herein also relate to the use of EIS in optimizing sensor calibration. As background, in the current method, the slope of the BG-Isig curve, which can be used to calibrate subsequent Isig values, is calculated as follows:

[0387]

[0388] Where α is an exponential function of the time constant, β is a function of the blood glucose variation, and offset is a constant. For sensors in stable conditions, this method provides fairly accurate results. For example Figure 36 As shown, BG and Isig follow a fairly linear relationship, and offset can be obtained as a constant.

[0389] However, there are situations where the above linear relationship does not apply, such as during the conversion process of the sensor. Figure 37 It is clear that the Isig-BG pairs 1 and 2 are significantly different from the Isig-BG pairs 3 and 4, based on the relationship between Isig and BG. For these types of cases, the use of a constant offset is prone to producing inaccurate results.

[0390] To address this issue, one embodiment of the present invention involves using an EIS-based dynamic offset, where the EIS measurements are used to define the sensor state vector as follows:

[0391] V={real_imp_1K,img_imp_1K,Nyquist_slope,Nyquist_R_square}

[0392] where all elements in the vector are substantially independent of the BG. It should be noted that Nyquist_R_square is the R square of the linear regression used to calculate the Nyquist slope, i.e., the square of the correlation coefficient between the real and imaginary impedances at relatively low frequencies, and that a low R square indicates an anomaly in sensor performance. A state vector is assigned to each Isig-BG pair. If a significant difference in the state vector is detected—e.g. Figure 37 |V2-V3| of the example shown—different offset values ​​are assigned to 3 and 4 when they are compared to 1 and 2. Therefore, by using this motion compensation method, the linear relationship between Isig and BG can be maintained.

[0393] In a second embodiment, an EIS-based segmentation method can be used for calibration. Figure 37 According to the embodiment and vector V, it can be determined that the sensor states in cases 1 and 2 are significantly different from the sensor states in cases 3 and 4. Therefore, the calibration buffer memory can be divided into the following two parts:

[0394] Isig_buffer1=[Isig1,Isig2]; BG_buffer1=[BG1,BG2]

[0395] Isig_buffer2=[Isig3,Isig3]; BG_buffer2=[BG3,BG3]

[0396] Therefore, Isig_buffer1 and BG_buffer1 can be used for calibration when the sensor operates in situations 1 and 2. However, when the sensor operates in situations 3 and 4, that is, during conversion, Isig_buffer2 and BG_buffer2 can be used for calibration.

[0397] In yet another embodiment, an EIS-based dynamic slope method can be used for calibration purposes, where EIS is used to adjust the slope. Figure 38A An example of how this approach can be used to improve sensor accuracy is shown in Figure 1. In this figure, data points 1 to 4 are discrete blood glucose values. Figure 38A As can be seen in FIG, a sensor dip 3810 occurs between data points 1 and 3, which can be detected using the sensor state vector V described above. During the dip, the slope can be adjusted upward to reduce insufficient readings, such as Figure 38A As shown in the figure marked 3820.

[0398] In further embodiments, EIS diagnostics can be used to determine when to calibrate the sensor, which is useful for events such as low-start events, sensitivity loss events, and other similar situations. Most current methods known in the art require regular calibration based on a preset schedule, such as four times per day. However, using EIS diagnostics, calibration becomes event-driven, allowing calibration to be performed only as needed and when it is most effective. Similarly, the state vector V can be used to determine when the sensor state has changed and to request calibration if the sensor state has indeed changed.

[0399] More specifically, in an exemplary embodiment, Figure 38B An EIS-assisted sensor calibration procedure including low start detection is shown. Using Nyquist slope, 1kHz real impedance and limit check 3850 (see, e.g., the aforementioned Figures 33A to 35 fusion algorithm related limit checks and related thresholds based on EIS parameters), reliability index 3853 can be developed for startup, such that when the 1 kHz real impedance 3851 and the Nyquist slope 3852 are lower than their respective upper limits, RI_startup = 1 and the sensor is ready for calibration. In other words, reliability index 3853 is "high" (3854), and the logic can proceed to calibration at 3860.

[0400] On the other hand, when the 1 kHz real impedance and the Nyquist slope are above their respective upper limits (or thresholds), RI_startup = 0 (i.e., "low"), and the sensor is not ready for calibration (3856), i.e., there may be a low startup problem. Here, the trends of the 1 kHz real impedance and the Nyquist slope can be used to predict when these two parameters are within range (3870). If it is estimated that only a very short time is required (e.g., less than 1 hour), then the algorithm waits until the sensor is ready, i.e., until the above-mentioned EIS-based parameters are within range (3874), at which time the algorithm calibrates. However, if the wait time is relatively long (3876), then the sensor can be calibrated now and then the slope or offset can be gradually adjusted based on the 1 kHz real impedance and Nyquist slope trends (3880). It should be noted that by making such adjustments, severe over-readings or under-readings caused by low startup can be avoided. As previously described, the EIS-based parameters and related information used for the present calibration algorithm are essentially independent of glucose.

[0401] It should be noted that although the above Figure 38BThe description shows a single working electrode and the calculation of the startup reliability index for that working electrode, but this is exemplary and not limiting. Therefore, in a redundant sensor comprising two or more working electrodes, each of the multiple (redundant) working electrodes can be limit checked and a startup reliability index can be calculated. Then, based on the respective reliability index, at least one working electrode can be identified as being able to continue to operate to obtain glucose measurements. In other words, in a sensor with a single working electrode, if a working electrode exhibits low startup, then actual use of the sensor (for measuring glucose) may be postponed until the low startup process has passed. This time period is typically on the order of one hour or more, which is clearly disadvantageous. In contrast, in a redundant sensor, using the methods described herein allows for adaptive startup or "smart" startup, wherein electrodes that can perform data acquisition can be identified in a relatively short period of time, for example, on the order of minutes. This in turn reduces MARD, as low startup typically provides an improvement in MARD of approximately 1 / 2%.

[0402] In yet another embodiment, EIS can help adjust the calibration cache memory. For existing calibration algorithms, the cache size is typically 4, i.e., 4 Isig-BG pairs, and the weights are based on α and β. As previously mentioned, α is an exponential function of the time constant, and β is a function of the blood glucose variation. Here, EIS can help determine when to flush the cache, how to adjust the cache weights, and what the appropriate cache size is.

[0403] Embodiments of the present invention also relate to the application of EIS for interfering substance detection. Specifically, it is desirable to provide a drug infusion assembly comprising a sensor and a drug infusion catheter, wherein the sensor is positioned within the infusion catheter. In such a system, the physical position of the infusion catheter relative to the sensor needs to be considered, primarily due to the potential impact on the sensor signal (i.e., interference with the sensor signal) caused by the infused drug and / or its inactive components.

[0404] For example, the diluent used with insulin contains m-cresol as a preservative. In in vitro studies, m-cresol was found to adversely affect glucose sensors if insulin (and therefore, m-cresol) was injected in close proximity to the sensor. Therefore, systems in which the sensor and infusion catheter are combined in a single needle must be able to detect or adjust for the effects of m-cresol on the sensor signal. Because m-cresol affects the sensor signal, it would be advantageous to provide a method for detecting this interfering substance independently of the sensor signal itself.

[0405] Experiments have shown that the effect of m-cresol on the sensor signal is temporary and reversible. Nevertheless, when insulin is infused too close to the sensor, m-cresol tends to "poison" the electrode, so that the latter no longer detects glucose until the insulin (and m-cresol) are absorbed into the patient's tissue. In this regard, it has been found that the time period between the start of insulin infusion and the sensor regaining the ability to detect glucose is typically about 40 minutes. However, it has been found that, over the same time period, there is a significant increase in the 1 kHz impedance amplitude, which is completely independent of glucose concentration.

[0406] Specifically, Figure 39 Isig and impedance data from an in vitro experiment are shown, wherein the sensor was placed in a 100 mg / dL glucose solution and the 1 kHz impedance was measured every 10 minutes, represented by the circled data points 3920. Meta-cresol was then added to bring the solution to 0.35% meta-cresol (3930). As can be seen, once meta-cresol was added, Isig 3940 began to increase significantly and then began to gradually decrease. The glucose concentration in the solution was doubled by adding another 100 mg / dL glucose. However, this had no effect on Isig 3940 because the electrode could not detect glucose.

[0407] On the other hand, m-cresol has a significant effect on the impedance amplitude and phase. Figure 40A Bode plots showing the phase before and after addition of m-cresol, Figure 40B The Bode plots of the impedance magnitude before and after the addition of m-cresol are shown. As can be seen from the figure, after the addition of m-cresol, the impedance magnitude 4010 increases by at least one order of magnitude from its initial value 4020 across the entire spectrum. At the same time, the phase 4030 completely changes relative to its initial value 4040. Figure 40C In the Nyquist plot of FIG. 40, the pre-initialization curve 4050 and the post-initialization curve 4060 indicate a properly functioning sensor as expected. However, after the addition of m-cresol, the curve 4070 is significantly different.

[0408] The above experiments identified a significant practical drawback of relying on Isig even after the addition of m-cresol. Figure 39, a patient / user monitoring the sensor signal may be given the false impression that his glucose level has just suddenly increased and that he should take a bolus. The user then takes a bolus, while Isig has already begun to gradually decrease. In other words, everything may appear normal to the patient / user. However, in reality, what actually happened is that the patient was given an unnecessary dose of insulin based on the patient's glucose level before the bolus, which may put the patient at risk of experiencing a hypoglycemic event. This assumption emphasizes the need for methods that are as independent of glucose detection interferents as possible.

[0409] Figure 41 Another experiment is shown, in which the sensor was initialized in a 100 mg / dL glucose solution. The glucose level was then raised to 400 mg / dL for one hour, and then returned to 100 mg / dL. Meta-cresol was then added to raise its concentration to 0.35%, and the sensor remained in this solution for 20 minutes. Finally, the sensor was placed in a 100 mg / dL glucose solution to allow Isig to recover after exposure to meta-cresol. As can be seen from the figure, after initialization, the 1 kHz impedance amplitude 4110 was approximately 2 k ohms. When meta-cresol was added, Isig 4120 suddenly increased, and the impedance amplitude 4110 also suddenly increased. Moreover, when the sensor was returned to the 100 mg / dL glucose solution, the impedance amplitude 4110 also returned to near normal levels.

[0410] As can be seen from the above experiment, EIS can be used for detecting the existence of interfering agent---be meta-cresol in the present embodiment. Specifically, because interfering substance affects the sensor in the mode that impedance amplitude increases within the whole frequency spectrum, so impedance amplitude can be used for detecting described interference. Once interference is detected, the sensor operating voltage can be changed to the point that can't measure the interfering substance so, or data reporting can be temporarily suspended, and sensor can point out to patient / user that due to drug administration, sensor can't report data (level before infusion until the impedance measured). It should be noted that, because the impact of interfering substance is owing to being included in the preservative in insulin, so impedance amplitude can show above-mentioned identical variation, no matter insulin is fast injection or slow injection.

[0411] Importantly, as mentioned above, the impedance magnitude is essentially independent of glucose, and the magnitude at 1 kHz is certainly also essentially independent of glucose. Figure 41, it can be seen that as the glucose concentration increases from 100 mg / dL to 400 mg / dL—a fourfold increase—the 1 kHz impedance amplitude increases from approximately 2000 ohms to approximately 2200 ohms, or about 10%. In other words, the effect of glucose on the impedance amplitude measurement appears to be approximately an order of magnitude smaller than the measured amplitude. This level of "signal-to-noise ratio" is generally small enough to filter out the noise (i.e., the glucose effect), such that the remaining impedance amplitude is essentially independent of glucose. Furthermore, it should be emphasized that the impedance amplitude exhibits an even greater degree of glucose independence in actual human tissue compared to the buffer solution used in the in vitro experiments described above.

[0412] Embodiments of the present invention also relate to an Analog Front End Integrated Circuit (AFEIC), a custom application-specific integrated circuit (ASIC) that provides the necessary analog electronics to: (i) support multiple potentiostats and an oxygen- or peroxide-based multi-terminal glucose sensor interface; (ii) interface with a microcontroller to form a micropower sensor system; and (iii) perform EIS diagnostics, fusion algorithms, and other EIS-based methods based on EIS parameter measurements. More specifically, the ASIC combines the sensor's diagnostic capabilities for measuring both real and imaginary impedance parameters over a wide frequency range, along with digital interface circuitry for bidirectional communication with a microprocessor chip. Furthermore, the ASIC includes power control circuitry, a real-time clock, and a crystal oscillator capable of operating at very low standby and operating power levels, allowing the external microprocessor power supply to be turned off.

[0413] Figure 42A and 42B The block diagram of the ASIC is shown, and Table 1 below provides a description of the pad signals (in Figure 42A and 42B shown on the left), where some signals are multiplexed onto a single pad.

[0414]

[0415]

[0416] Reference Figure 42A and 42B And Table 1 describes the ASIC.

[0417] Power Plane

[0418] The ASIC has a power supply board that is powered via the supply pad VBAT (4210), which has an operating input range of 2.0 volts to 4.5 volts. The power supply board has a regulator to reduce the voltage for some circuits on the board. The power supply (supply) is called VDDBU (4212) and has output pads for testing and shunting. The circuits on the VBAT supply include an RC oscillator, a real-time clock (RC osc) 4214, a battery protection circuit, a regulator control, a power-on reset circuit (POR), and various inputs / outputs. The pads on the VBAT supply board are configured to draw less than 75nA at 40°C and VBAT = 3.50V.

[0419] The ASIC also has a VDD power supply that provides logic. The VDD power supply voltage range is programmable from at least 1.6 volts to 2.4 volts. The circuitry on the VDD power plane includes most of the digital logic, a timer (32 kHz), and a real-time clock (32 kHz). The VDD power plane includes a level shifter that connects to other voltage planes as needed. The level shifter further has an interface that is regulated so that any powered power plane does not increase current by more than 10 nA if the other power plane is not powered.

[0420] The ASIC includes an onboard regulator (with shutdown control) and external VDD source selection. The regulator input is a separate pad, REG_VDD_IN (4216), which has electrostatic discharge (ESD) protection along with the other I / O on VBAT. The onboard regulator has an output pad, REG_VDD_OUT (4217). The ASIC also has an input pad for VDD that is independent of the REG_VDD_OUT pad.

[0421] The ASIC includes an analog power plane, called VDDA (4218), which is powered by the VDD onboard regulator or an external source, and is typically powered by filtered VDD. The VDDA supply circuit is configured to operate within a 0.1 volt range of VDD, eliminating the need for level conversion between the VDDA and VDD power planes. The VDDA supply provides power to the sensor analog circuitry, analog measurement circuitry, and any other noise-sensitive circuitry.

[0422] The ASIC includes a pad power supply, VPAD, for specifying digital interface signals. The pad power supply operates over a voltage range of at least 1.8V to 3.3V. These pads have separate power pads and are powered from an external source. The pads also incorporate level shifters into other onboard circuitry to enable flexible pad power supply ranges independent of the VDD logic supply voltage. The ASIC conditions the VPAD pad ring signal so that when the VPAD supply is not enabled, the current drawn by other supplies does not increase by more than 10nA.

[0423] Bias generator

[0424] The ASIC has a bias generator circuit, BIAS_GEN (4220), which is powered by the VBAT power supply and generates a bias current that is stable when the system is supplied with voltage. The output current has the following specific characteristics: (i) supply sensitivity: <±2.5% under the condition of supply voltage of 1.6V to 4.5V; and (ii) current accuracy after regulation: <±3%.

[0425] The BIAS_GEN circuit generates switched and unswitched output currents to power circuits requiring bias current for operation. The operating current consumption of the BIAS_GEN circuit is less than 0.3uA (excluding any bias output current) at 25°C with a VBAT range of 2.5V to 4.5V. Finally, the temperature coefficient of the bias current is typically 4,000ppm / °C to 6,000ppm / °C.

[0426] Reference voltage

[0427] As described herein, the ASIC is configured with a low reference supply voltage, powered by the VBAT supply. The reference voltage has an enable input that accepts a signal from logic powered by either VBAT or VDDBU. The ASIC is designed so that when VBAT is powered, the enable signal does not increase any current by more than 10 nA from any supply from the signal interface.

[0428] The reference voltage has the following specific characteristics: (i) output voltage: 1.220 ± 3 mV after regulation; (ii) supply sensitivity: < ± 6 mV from 1.6 V to 4.5 V input; (iii) temperature sensitivity: < ± 5 mV from 0°C to 60°C; and (iv) output voltage default accuracy (unregulated): 1.220 V ± 50 mV. In addition, the source current is less than 800 nA at 4.5 V and 40°C. In this embodiment, when the reference is disabled, the reference output drives to VSSA, thereby preventing the VDD voltage regulator from overshooting to a level exceeding the logic breakdown voltage.

[0429] 32kHz oscillator

[0430] The ASIC includes a low-power 32.768kHz crystal oscillator 4222, which is powered by power from the VDDA supply and has software-adjustable capacitance across the crystal oscillator pads (XTALI, XTALO). Specifically, the frequency adjustment range is from at least -50ppm to +100ppm, using a maximum 2ppm step size across the entire adjustment range. The crystal is assumed to have a load capacitance of 7pF, Ls = 6.9512kH, Cs = 3.3952fF, Rs = 70k, a shunt capacitance of 1pF, and 2pF of PC board parasitic capacitance at each crystal terminal.

[0431] The ASIC has a VPAD level output available on the CLK_32kHZ pad, which can be disabled under software and logic control. The default value drives the 32kHz oscillator. The input pin OSC32K_BYPASS (4224) allows the 32kHz oscillator to be disabled (without power consumption) and allows the XTALI pad digital input. The circuitry associated with this function is configured so that when OSC32K_BYPASS is low, no additional ASIC current is drawn in excess of 10nA in any state of the OSC32K_BYPASS signal, other than the oscillator current.

[0432] The 32kHz oscillator is required to always be running when the VDDA board is powered, except in shunt conditions. If OSC32K_BYPASS is true, the 32kHz oscillator analog circuitry enters a low-power state and the XTALI pad is configured to receive a digital input with a level between 0 and VDDA. It should be noted that the 32kHz oscillator output has a duty cycle of 40% to 60%.

[0433] Timer

[0434] The ASIC includes a timer 4226 that is timed by dividing the 32kHz oscillator by 2. This timer is presettable and has two programmable timeout settings. It has 24 programmable bits, providing a total timeout of 17 minutes and 4 seconds. The timer also has a programmable delay to disable the clock to the CLK_32KHz pad and set the microprocessor (uP) interface signal to a predetermined state on the VPAD board (see the Microprocessor Wake-up Control Signal section below). This allows the microprocessor to enter a suspend mode without an external clock. However, this feature can be disabled by software using a programmable bit.

[0435] The timer also includes a programmable delay to wake up the microprocessor by enabling the CLK_32kHz clock output and setting UP_WAKEUP high. The transition of POR2 (VDD POR) from the low power state to the OK power state enables the 32kHz oscillator, the CLK_32kHz clock output, and sets UP_WAKEUP high. Power down and power up are configured under programmable control bits.

[0436] Real-time clock (RTC)

[0437] The ASIC also has a 48-bit readable / writable binary counter that runs from a non-gated free-running 32kHz oscillator. Write operations to the real-time clock 4228 require writing a key address before the clock can be written. The clock's write path is configured to stop 1 to 20 milliseconds after writing the key address.

[0438] The real-time clock 4228 is configured to be reset to half count (MSB = 1, all other bits are 0) by a power-on reset via POR1_IN (VBATPOR) or POR2_IN (VDD_POR). In an embodiment of the present invention, the real-time clock has programmable interrupt capability and is designed to be robust against single-event upsets (SEUs), which can be achieved through layout techniques or by adding capacitance to appropriate nodes, if necessary.

[0439] RC oscillator

[0440] The ASIC also includes an RC clock powered by the VBAT supply or a supply derived from VBAT. The RC oscillator is always running, except that it can be bypassed by writing to a register bit in analog test mode (see the Digital Test section) and applying a signal to GPIO_VBAT with a level between 0 and VBAT. The RC oscillator is not adjustable and includes the following specific characteristics: (i) frequency of 750Hz to 1500Hz; (ii) duty cycle of 50% ±10%; (iii) current consumption of less than 200nA at 25°C; (iv) frequency variation of less than ±2% with a VBAT supply of 1V to 4.5V and greater than 1% with a VBAT supply of 1.8V to 4.5V; and (v) frequency variation of less than +2, -2% with VBAT = 3.5V over a temperature range of 15°C to 40°C. The RC frequency can be measured using a 32kHz crystal oscillator or an external frequency source (see the Oscillator Calibration Circuit).

[0441] Real-time RC clock (based on RC oscillator)

[0442] The ASIC includes a 48-bit readable / writable binary ripple counter based on an RC oscillator. Writing to the RC real-time clock requires writing a key address before writing to the clock. The clock write path is stopped 1 to 20 milliseconds after writing the key address, with the protection window time configured to be generated by the RC clock.

[0443] If the crystal oscillator is shut down, the real-time RC clock allows for relative timestamp generation and is configured to be reset to half count (MSB = 1, all others 0) via POR1_IN (BAT POR). The real-time RC clock is designed to be robust against single-event upsets (SEUs), which can be achieved through routing techniques or by adding capacitors to appropriate nodes (if necessary). On the falling edge of POR2_IN, or if the ASIC enters a low-battery state, the RC real-time clock value can be captured into a register that can be read via the SPI port. This register and associated logic are located on the VBAT or VDDBU power plane.

[0444] Battery protection circuit

[0445] The ASIC includes a battery protection circuit 4230 that monitors the battery voltage using a comparator and is powered by power from the VBAT power supply. The battery protection circuit is configured to operate continuously using power applied to the VBAT power supply. The battery protection circuit can use an RC oscillator for timing signals and can have an average current consumption of less than 30nA. The battery protection circuit includes a 3M ohm total resistance external voltage divider.

[0446] The battery protection circuit uses an external conversion voltage divider with a 0.421 ratio for a 2.90V battery threshold. The ASIC also has an internal voltage divider with a ratio of 0.421 ± 0.5%. This divider is connected between BATT_DIV_EN (4232) and VSSA (4234), and the output of the divider is a pin called BATT_DIV_INT (4236). To save pins on the packaged component, BATT_DIV_INT in this embodiment is internally connected to BATT_DIV in the package. Moreover, in this configuration, BATT_DIV_EN does not need to extend from the package, saving two package pins.

[0447] The battery protection circuit is configured to sample the voltage on the BATT_DIV (4238) input pin approximately twice per second, where the sampling time is generated by an RC oscillator. The ASIC is capable of adjusting the voltage divider of the RC oscillator to adjust the sampling interval to 0.500 seconds ± 5 milliseconds when the RC oscillator operates within its operating tolerance. In a preferred embodiment, the ASIC has a test mode that allows for more frequent sampling intervals during testing.

[0448] The comparator inputs are configured to receive inputs between 0 volts and VBAT volts. For inputs between 0 volts and VBAT volts, the comparator input current, BATT_DIV, is less than 10 nA. The comparator sampling circuit outputs a positive-going pulse to pad BATT_DIV_EN, which can be used by external circuitry to enable the off-chip resistor divider only during sampling time to conserve power. The logic high voltage level is the VBAT voltage, and the logic low voltage level is the VSS level.

[0449] Under VBAT = 3.0V, the output resistance of the BATT_DIV_EN pin can be less than 2k ohms. This allows the voltage divider to be driven directly from this output. After a programmable number of consecutive samples indicating a low-battery condition, the comparator control circuit triggers an interrupt signal to the interrupt output pin UP_INT. The default number of samples is 4, although the number of consecutive samples is programmable from 4 to 120.

[0450] After a programmable number of consecutive samples indicates low battery and after the aforementioned UP_INT is generated, the comparator control circuit is configured to generate a signal that causes the ASIC to enter low-power mode: the VDD regulator is disabled and a low signal is sent to the VPAD_EN pad. This is referred to as a low-battery state. Furthermore, the number of consecutive samples is programmable from 4 to 120 samples, with a default of 4 samples.

[0451] The comparator has separate programmable thresholds for lowering and raising the voltage on BATT_DIV. This is implemented in the digital logic to multiplex two values ​​within the circuit based on the low-battery state. Therefore, if the low-battery state is low, the lower threshold is used, and if the low-battery state is high, the higher threshold is used. Specifically, the comparator has 16 programmable thresholds, ranging from 1.22 to 1.645 ± 3%, with the DNL of the programmable thresholds set to less than 0.2 LSB.

[0452] The comparator threshold varies by less than + / - 1% from 20°C to 40°C. The default threshold for decreasing the voltage is 1.44V (the nominal VBAT threshold for the voltage divider is 3.41V) and the default threshold for increasing the voltage is 1.53V (the nominal VBAT threshold for the voltage divider is 3.63V). After the ASIC enters the low battery state, if the comparator detects four consecutive battery OK flags, the ASIC can initiate the microprocessor boot sequence.

[0453] Battery power board power reset

[0454] If the input VBAT swings greater than 1.2 volts within a 50 microsecond period or if the VBAT voltage falls below 1.6 ± 0.3 volts, a power-on reset (POR) output is generated on pad nPOR1_OUT (4240). This POR is extended to a minimum pulse width of 5 milliseconds. The output of the POR circuit is set to active low and is routed to pad nPOR1_OUT on the VBAT power plane.

[0455] The IC has an input pad, nPOR1_IN (4242), for the battery power board POR. This input pad has RC filtering so that pulses shorter than 50 nanoseconds do not cause a logic reset. In this embodiment, nPOR1_OUT is externally connected to nPOR1_IN in normal operation, thereby separating the analog circuitry from the digital circuitry for testing purposes. nPOR1_IN causes all logic on any power board to be reset and initializes all registers to their default values. Therefore, the reset status register POR bit is set and all other reset status register bits are cleared. The POR reset circuit is configured to consume no more than 0.1uA of the VBAT power supply for a period greater than 5 seconds after power is applied.

[0456] VDD Power-On Reset (POR)

[0457] The ASIC also features a voltage comparator circuit that generates a VDD voltage board reset signal after power-up or if VDD drops below a programmable threshold. The threshold range is programmable using several voltage thresholds. The default value is 1.8V-15% (1.53V). POR2 has a programmable threshold for rising voltages, which incorporates hysteresis. The rising threshold is also programmable, with a default value of 1.60V ±3%.

[0458] The POR signal is active low and has an output pad nPOR2_OUT (4244) on the VDD power plane. The ASIC also has an active low POR open-drain output, nPOR2_OUT_OD (4246) on the VBAT power plane. This can be used to apply the POR to other system components.

[0459] The VDD powered logic has a POR sourced from the input pad nPOR2_IN (4248). The nPOR2_IN pad is located on the VDD power plane and has RC filtering so that pulses shorter than 50 nanoseconds do not cause a logic reset. nPOR2_OUT is configured to be connected externally to the nPOR2_IN input pad under normal use conditions, thereby separating the analog circuitry from the digital circuitry.

[0460] The reset generated is stretched to at least 700 milliseconds after VDD rises above a programmable threshold to ensure crystal oscillator stability. The POR reset circuit consumes no more than 0.1uA from the VDD supply for more than 5 seconds after power-up, and no more than 0.1uA from the VBAT supply for more than 5 seconds after power-up. The registers storing the POR thresholds are powered by the VDD power plane.

[0461] Sensor interface electronics

[0462] In an embodiment of the present invention, in any peroxide or oxygen sensor combination, the sensor circuit supports up to five sensor working (WORK) electrodes (4310), although, in other embodiments, a greater number of such electrodes may be used. The peroxide sensor WORK electrodes source current, while the oxygen sensor WORK electrodes sink current. For this embodiment, the sensors may be configured as follows: Figure 43 The potentiostatic configuration is shown.

[0463] The sensing electronics have programmable power control for each electrode interface circuit to minimize current consumption by shutting off current to unused sensing electronics. The sensor electronics also include electronics to drive the COUNTER electrode 4320, which uses feedback information from the RE (reference) electrode 4330. The current to this circuit can be programmably shut off when not in use to save power. The interface electronics include a multiplexer 4250 so that the COUNTER electrode and the RE electrode can be connected to either of the (redundant) WORK electrodes.

[0464] The ASIC is configured to provide the following sensor interfaces: (i) RE: Reference electrode, which establishes the reference potential of the solution for the electronics to use in setting the WORK voltage; (ii) WORK1-WORK5: Sensor working electrodes, where the desired reduction / oxidation (redox) reaction occurs; and (iii) COUNTER: The output from this pad maintains a known voltage on the RE electrode relative to the system VSS. In this embodiment of the invention, the ASIC is configured to be able to individually set the WORK voltages of up to five WORK electrodes with a resolution and accuracy greater than or equal to 5 mV.

[0465] In oxygen mode, the WORK voltage is programmable between at least 0 and 1.22V relative to VSSA. In peroxide mode, the WORK voltage is programmable between at least 0.6V and 2.054V relative to VSSA. If VDDA is less than 2.15V, the WORK voltage can be set to VDDA-0.1V. The ASIC includes current measurement circuitry to measure the WORK electrode current in peroxide sensor mode. This can be achieved, for example, using a current-to-voltage or current-to-frequency converter, which may have the following specific features: (i) current range: 0-300nA; (ii) voltage output range: same as the WORK electrode in peroxide / oxygen mode; (iii) output offset voltage: ±5mV maximum; and (iv) uncalibrated resolution: ±0.25nA.

[0466] After applying the calibration factor to the gain and assuming an acquisition time of 10 seconds or less, the current measurement accuracy is:

[0467] 5pA-1nA:±3%±20pA

[0468] 1nA-10nA:±3%±20pA

[0469] 10nA-300nA: ±3% ±.2nA

[0470] For the current-to-frequency converter (IF) only, the frequency range can be 0 Hz to 50 kHz. In peroxide mode, the current converter must operate within a specific voltage range relative to the work electrode's VSS. Here, for each converter, including the digital-to-analog (DAC) current, the current consumption from the 2.5V supply must be less than 2uA, and the work electrode current must be less than 10nA.

[0471] The current converter can be enabled or disabled via software control. When disabled, the WORK electrode will exhibit very high impedance values, i.e., greater than 100M ohms. Furthermore, for IF only, the IF converter outputs a 32-bit counter that can be read, written, and cleared by the microprocessor and test logic. During counter readings, the counter clock is paused to ensure accurate readings.

[0472] In an embodiment of the present invention, the ASIC also includes current measurement circuitry for measuring the WORK electrode current in oxygen sensor mode. This circuitry can be implemented as a current-to-voltage or current-to-frequency converter, and programmable bits can be used to configure the current converter to operate in oxygen mode. As previously mentioned, in oxygen mode, the current converter must operate within a specific voltage range of the WORK electrode relative to VSS. Here, too, the current range is 3.7 pA to 300 nA, and the voltage output range is the same as for the WORK electrode in oxygen mode, with a maximum output offset voltage of ±5 mV and an uncalibrated resolution of 3.7 pA ±2 pA.

[0473] After applying the calibration factor to the gain and assuming an acquisition time of 10 seconds or less, the current measurement accuracy is:

[0474] 5pA–1nA: ±3% ±20pA

[0475] 1nA–10nA: ±3% ±20pA

[0476] 10nA–300nA: ±3% ±.2nA

[0477] For the current-to-frequency converter (IF) only, the frequency range can be 0 Hz to 50 kHz, and, for each converter, including the DAC current, the current consumption from the 2.5 V power supply is less than 2 uA and the work electrode current is less than 10 nA. The current converter can be enabled or disabled by software control. When disabled, the work electrode can exhibit very high impedance values, i.e., greater than 100 M ohms. Furthermore, for the IF only, the IF converter outputs a 32-bit counter that can be read, written, and cleared by the microprocessor and test logic. During the counter reading, the counter clock is paused to ensure accurate readings.

[0478] In an embodiment of the present invention, the input bias current of the reference electrode (RE) 4330 is less than 0.05 nA at 40°C. The COUNTER electrode adjusts its output to maintain the desired voltage on the RE electrode. This is accomplished by an amplifier 4340, whose output to the COUNTER electrode 4320 attempts to minimize the difference between the actual RE electrode voltage and the target RE voltage, which is set by the DAC.

[0479] The RE set voltage is programmable between at least 0 and 1.80V, and the common mode input range of the COUNTER amplifier includes at least 0.20 to (VDD-0.20)V. Register bits can be used to select the common mode input range (if desired) and to program the operating mode of the COUNTER. The WORK voltage is set to a resolution and accuracy greater than or equal to 5mV. It should be noted that in normal mode, the COUNTER voltage seeks to maintain the RE voltage to the level of the programmed RE target value. However, in drive relative mode, the COUNTER electrode voltage is driven to the programmed RE target voltage.

[0480] All electrode drive circuits are configured to be able to drive the electrodes to the electrode loads without oscillation for any application scenario. Figure 44 The embodiment according to the present invention is shown Figure 43 Equivalent AC circuit between electrodes in the potentiostatic configuration shown. Figure 44 The equivalent circuit shown can be between any of the electrodes (i.e., WORK1 to WORK5, COUNTER, and RE electrodes), and for each circuit element, the respective value ranges are as follows:

[0481] Ru=[200-5k]ohm

[0482] Cc=[10-2000]pF

[0483] Rpo=[1-20]k ohm

[0484] Rf=[200-2000]k ohms

[0485] Cf=[2-30]uF.

[0486] During initialization, the drive currents for the work and counter electrodes may need to be higher than those used in normal potentiostatic operation. Therefore, programmable register bits can be used to program the electrode drive circuitry to a higher power state if additional drive is required. It is important to achieve low-power operation in normal potentiostatic mode, where electrode currents are typically less than 300nA.

[0487] In a preferred embodiment, during initialization, the WORK1 through WORK5 electrodes are programmable from 0 to VDD volts in steps of 5 mV or less, and their drive or sink output capability is a minimum of 20 uA over the range of 0.20 V to (VDD - 0.20 V). Furthermore, during initialization, the ASIC is typically configured to measure up to 20 uA of current from a single WORK electrode with an accuracy of ±2% ±40 nA. Furthermore, during initialization, the RE set voltage is programmable as described above, and the COUNTER driver circuit output must be capable of sourcing or sinking a minimum of 50 uA of current using the COUNTER electrode over the range of 0.20 V to (VDD - 0.20 V). Furthermore, the source current (VDD and VDDA) flowing to the initialization circuit is required to be less than 50 uA in excess of any provided output current.

[0488] Current Calibrator

[0489] In an embodiment of the present invention, the ASIC has a reference current that can be applied to any WORK electrode for calibration purposes. In this regard, the calibrator includes a programmable bit that causes the current output to sink or source current. Assuming a 0 tolerance external precision resistor, the programmable currents include at least 10nA, 100nA, and 300nA, with an accuracy better than ±1% ±1nA. The calibrator uses a 1 megohm precision resistor connected to the contact TP_RES (4260) for the reference resistor. In addition, for the purpose of initialization and / or sensor status, the reference current can be used for the COUNTER electrode or RE electrode. A constant current can be applied to the COUNTER electrode or RE electrode and the electrode voltage can be measured by the ADC.

[0490] High-speed RC oscillator

[0491] Referring to FIG. 42 , the ASIC also includes a high-speed RC oscillator 4262 that provides power to an analog-to-digital converter (ADC) 4264, an ADC sequencer 4266, and other digital functions that require a high-speed clock greater than 32 kHz. The high-speed RC oscillator is phase-locked to a 32 kHz clock (32.768 kHz) to provide a programmable output frequency between 524.3 kHz and 1048 kHz. Furthermore, the high-speed RC oscillator has a duty cycle of 50% ± 10%, phase jitter of less than 0.5% rms, a current of less than 10 uA, and a frequency that is stable within the VDD operating range (voltage range of 1.6 V to 2.5 V). The high-speed RC oscillator is “off” (i.e., disabled) by default, in which case the current consumption is less than 10 nA. However, the ASIC has a programmable bit to enable the high-speed RC oscillator.

[0492] Analog-to-digital converter

[0493] The ASIC includes a 12-bit ADC (4264) having the following features: (i) the ability to effect conversions in less than 1.5 milliseconds by operating from a 32kHz clock; (ii) the ability to perform fast conversions when clocked by a high-speed RC oscillator; (iii) having at least 10 bits of accuracy (12 bits ± 4 counts); (iv) having a 1.220V reference voltage input with a temperature sensitivity of less than 0.2mV / °C at 20°C to 40°C; (v) full-scale input ranges of 0 to 1.22V, 0 to 1.774V, 0 to 2.44V, and 0 to VDDA, wherein the 1.774V and 2.44V ranges have programmable bits to reduce the conversion range to lower values ​​to accommodate lower VDDA voltages; (vi) consuming less than 50uA of current from the power supply; (vii) having a converter capable of operating from a 32kHz clock or a high-speed RC clock; (vii) having a DNL of less than 1LSB; and (viii) issuing an interrupt signal at the end of a conversion.

[0494] like Figure 42A and 42B As shown, the ASIC has an analog multiplexer 4268 at the input of ADC4264, both of which are controlled by software. In a preferred embodiment, at least the following signals are connected to the multiplexer:

[0495] (i) VDD – core voltage and regulator output

[0496] (ii) VBAT – Battery Power

[0497] (iii) VDDA – analog power supply

[0498] (iv) RE – reference electrode of the sensor

[0499] (v) COUNTER – the counter electrode of the sensor

[0500] (vi)WORK1-WORK5—Sensor working electrodes

[0501] (vii) Temperature sensor

[0502] (viii) At least two external pins for analog signal input

[0503] (ix) EIS integrated output

[0504] (x)IV current converter output.

[0505] For inputs COUNTER, RE, WORK1-WORK5, the temperature sensor, and any other inputs that could be adversely affected by load, the ASIC is configured to load the ADC no more than ±0.01 nA. The multiplexer includes a voltage divider for any input with a voltage higher than the ADC's input voltage range, and a buffer amplifier that reduces the input resistance of the divided input to less than 1 nA for load-sensitive inputs. The buffer amplifier further has a common-mode input range of at least 0.8 V to VDDA and an offset of less than 3 mV compared to an input range of 0.8 V to VDDA - 0.1 V.

[0506] In a preferred embodiment, the ASIC has a mode for acquiring ADC measurements in a programmed sequence. Therefore, the ASIC includes a programmable sequencer 4266 that monitors measurements on up to eight ADC measurement input sources, with the following programmable parameters:

[0507] (i) ADC MUX input

[0508] (ii) ADC range

[0509] (iii) Delay before measurement, where the delay can be programmed from 0 to 62 ms in steps of 0.488 ms

[0510] (iv) The number of measurement values ​​for each input is 0 to 255

[0511] (v) Number of measurement cycles: 0-255, where a measurement cycle is a sequence of up to 8 input measurements repeated multiple times (e.g. as an outer loop in a program)

[0512] (vi) Delay between measurement cycles, where the delay is programmable from 0 to 62 milliseconds in 0.488 millisecond steps

[0513] The sequencer 4266 is configured to start after receiving an automatic measurement start command, and the measured values ​​can be stored in the ASIC for retrieval on the SPI interface. It should be noted that the sequencer time base can be designed between the 32kHz clock and the high speed RC oscillator 4262.

[0514] Sensor diagnostic methods

[0515] As described in detail above, embodiments of the present invention relate to the application of impedance and impedance-related parameters in, for example, sensor diagnostics and Isig / SG fusion algorithms. To this end, in a preferred embodiment, the ASIC described herein has the ability to measure the impedance magnitude and phase angle between any WORK sensor electrode and the RE and COUNTER electrodes in a constant potential configuration. This is achieved, for example, by measuring the amplitude and phase of a current waveform that responds to a sinusoidal-like waveform superimposed on the WORK electrode voltage. See, for example, Figure 42B Diagnostic circuit 4255 in.

[0516] The ASIC has the ability to measure the resistance component and capacitance component between any electrodes through, for example, the electrode multiplexer 4250. It should be noted that this measurement may be disturbed by the sensor balance and may require a set time or sensor initialization to record a stable electrode current. As discussed above, although the ASIC can be used for impedance measurement within a wide frequency spectrum, for the purposes of embodiments of the present invention, a relatively narrow frequency range can be used. Specifically, the sine wave measurement capability of the ASIC can include test frequencies of approximately 0.10 Hz to approximately 8192 Hz. When performing these measurements, the minimum frequency resolution according to embodiments of the present invention can be defined as shown in Table 2 below:

[0517] Table 2

[0518]

[0519]

[0520] The sine wave amplitude can be programmed between at least 10 mVp-p and 50 mVp-p in 5 mV steps and between 60 mVp-p and 100 mVp-p in 10 mV steps. In a preferred embodiment, the amplitude accuracy is better than ±5% or ±5 mV, whichever is greater. In addition, the ASIC can measure electrode impedance with the accuracy specified in Table 3 below.

[0521]

[0522] In embodiments of the present invention, the ASIC can measure the phase of the input waveform relative to a time baseline, which can be used to calculate the impedance to provide accuracy. The ASIC can also include an on-chip resistor for calibrating the electrode impedance circuit. The on-chip resistor can be calibrated by comparing it to a known 1 megohm off-chip precision resistor.

[0523] The waveform data samples can also be used to determine the impedance. The data can be transferred to an external microprocessor with a serial external interface (SPI) for calibration and processing. The converted current data is fully buffered to enable 2000 ADC conversion data to be transferred to an external device via the SPI interface without data loss. This assumes a maximum latency of 8 milliseconds to service the data transfer request interrupt.

[0524] In an embodiment of the present invention, as an alternative to, or in addition to, measuring the electrode impedance of a sinusoidal wave, the ASIC can measure the electrode current of a step input. Here, the ASIC can provide a programmable amplitude step of 10 to 200 mV with a resolution better than 5 mV to the electrode and sample (measure) the resulting current waveform. The sampling time can be programmed to at least 2 seconds in 0.25 second steps, and the sampling interval for measuring the current can include at least five programmable binary-weighted steps of approximately 0.5 milliseconds to 8 milliseconds.

[0525] The electrode voltage is sampled with a resolution of less than 1mV and a range of up to ±0.25 volts. This measurement can be associated with a suitable stabilizing voltage to reduce the required dynamic range for data conversion. Similarly, the electrode current is sampled with a resolution of less than 0.04uA and a range of up to 20uA. If the measurement polarity is programmable, the current measurement can be unipolar.

[0526] In embodiments of the present invention, current measurements can be made using an IV converter. Furthermore, the ASIC may include an on-chip resistor for calibrating current measurements. This on-chip resistor can then be calibrated by comparing it to a known 1 megohm off-chip precision resistor. Current measurement sampling accuracy is better than ±3% or ±10nA, whichever is greater. As previously mentioned, the converted current data can be sufficiently buffered to enable transmission of the 2000 ADC conversion data to an external device via the SPI interface without data loss. This assumes a maximum latency of 8 milliseconds for the data transfer request interrupt.

[0527] Calibration voltage

[0528] The ASIC includes a precision voltage reference for calibrating the ADC. The output voltage is 1.000V ±3%, with less than ±1.5% variation during production and stability better than ±3mV over the 20°C to 40°C temperature range. This precision calibration voltage can be calibrated during production by the ADC by comparing it to an external precision voltage. During production, the calibration factor can be stored in system non-volatile memory (not on the ASIC) for even higher accuracy.

[0529] The current consumption of the calibration voltage circuit is preferably less than 25uA. Moreover, when not in use, the calibration voltage circuit can reduce power consumption to less than 10nA to conserve battery power.

[0530] Temperature sensor

[0531] The ASIC features a temperature sensor with a sensitivity of 9mV / °C to 11mV / °C over the -10°C to 60°C range. The temperature sensor's output voltage allows the ADC to measure temperature-dependent voltages within the 0 to 1.22V ADC input range. The temperature sensor's current consumption is preferably less than 25uA, and when not in use, the temperature sensor can be powered down to less than 10nA to conserve battery power.

[0532] VDD voltage regulator

[0533] The ASIC has a VDD voltage regulator with the following features:

[0534] (i) Minimum input voltage range: 2.0V–4.5V.

[0535] (ii) Minimum output voltage: 1.6-2.5V±5%, default 2.0V.

[0536] (iii) Voltage drop: Vin–Vout<.15V at Iload (load current) = 100uA, Vin = 2.0V.

[0537] (iv) The output voltage is programmable with an accuracy within 2% of the values ​​shown in Table 4 below:

[0538] Table 4

[0539]

[0540] (v) The regulator can provide a 2.5V, 1mA output from a 2.8V input.

[0541] (vi) The regulator also has input and output contacts which can open the circuit in the event of use of an external regulator. The current consumption of the regulator circuit in this non-operating mode is preferably less than 100 nA.

[0542] (vii) The variation in output voltage from a load of 10 uA to 1 mA is preferably less than 25 mV.

[0543] (viii) In addition to the output current under 1mA load, the current consumption of the power supply is less than 100uA.

[0544] (ix) In addition to the output current under 0.1mA load, the current consumption of the power supply is less than 10uA.

[0545] (x) In addition to the output current under a 10uA load, the current consumption of the power supply is less than 1uA.

[0546] General purpose comparator

[0547] The ASIC includes at least two comparators 4270 and 4271 powered by VDDA. The comparators use 1.22V as a reference to generate thresholds. The comparator outputs can be read by the processor and can generate maskable interrupt signals on rising or falling edges determined by configuration registers.

[0548] The comparator has power control to reduce power when not in use, and the current supply is less than 50nA per comparator.The response time of the comparator is preferably less than 50 microseconds for a 20mV overdrive signal, and the offset voltage is less than ±8mV.

[0549] The comparators also have programmable hysteresis, where hysteresis options include threshold = 1.22V + Vhyst on rising input, threshold = 1.22 - Vhyst on falling input, or no hysteresis (Vhyst = 25 ± 10mV). The output from either comparator can be provided to any GPIO on any power board (see the GPIO section).

[0550] Sensor connection detection circuit on RE

[0551] The analog conversion capacitor circuit monitors the impedance of the RE connection to determine whether the sensor is connected. Specifically, a capacitor of approximately 20pF is switched at a frequency of 16Hz under the drive of the inverter, and the output swings between VSS and VDD. The comparator can detect the voltage swing on the RE contact, and if the voltage swing is less than the threshold, the comparator output can indicate a connection. The above comparison can be performed on two transitions of the pulse. A swing below the threshold on both transitions is required to indicate a connection, and a comparison showing a high swing on either phase can indicate a disconnection. The connected signal / unconnected signal is debounced for at least 1 / 2 second, so that its state transition is required to stably show the new state.

[0552] The circuit has six thresholds defined by the following resistors in parallel with a 20pF capacitor: 500k ohms, 1 megohm, 2 megohm, 4 megohm, 8 megohm, and 16 megohm. This parallel equivalent circuit is between the RE contact and a virtual ground, which can be any voltage between the power rails. Threshold accuracy is better than ±30%.

[0553] The output of the sensor connection detection circuit can be programmed to generate an interrupt signal (if a sensor is not connected) or a processor start signal (if a sensor is connected). This circuit is active regardless of whether nPOR2_IN is high or VDD and VDDA are present. The circuit consumes an average current of less than 100nA.

[0554] WAKEUP (wake-up) touch pad

[0555] The wakeup circuit is powered by the VDD supply and has an input range of 0V to VBAT. The wakeup pad 4272 has a weak pull-down current of 80±40nA. This current can be derived from the output of the BIAS_GEN 4220. The average current consumed by the circuit is less than 50nA at 0V input.

[0556] The WAKEUP input has a rising input voltage threshold, Vih, of 1.22 ± 0.1V, and a falling input threshold of -25 mV ± 12 mV relative to the rising threshold. In a preferred embodiment, the circuitry associated with the WAKEUP input consumes no more than 100 nA for any input value between -0.2 and VBAT volts (this current does not include the input pull-down current). The WAKEUP pad is debounced for at least 1 / 2 second.

[0557] The output of the wakeup circuit can be programmed to generate an interrupt or a processor start signal if the wakeup pad changes state (see the Event Handler section). It is important to note that the wakeup pad circuit is configured to draw a low current of <1nA if the battery protection circuit indicates a low battery condition.

[0558] UART WAKEUP

[0559] The ASIC is configured to monitor the nRX_EXT pin 4274. If the nRX_EXT level remains high (UART interrupt) for longer than 1 / 2 second, a UARTWAKEUP event is generated. This is because the UARTWAKEUP event can be generated for a duration as short as 1 / 4 second. The UARTWAKEUP event can be programmable to generate an interrupt signal, a WAKEUP signal, and / or a microprocessor reset signal (nRESET_OD). (See the Event Handler section.)

[0560] In a preferred embodiment, the circuitry associated with the UARTWAKEUP input consumes no more than 100 nA, and the UARTWAKEUP pad circuit is configured to draw a low current of < 1 nA if the battery protection circuit indicates a low battery condition. The UARTWAKEUP input has a rising input voltage threshold, Vih, of 1.22 ± 0.1 V. The falling input threshold is -25 mV ± 12 mV relative to the rising threshold.

[0561] Microprocessor WAKEUP control signal

[0562] The ASIC can generate signals to help control the power management of the microprocessor. Specifically, the ASIC can generate the following signals:

[0563] (i) nSHUTDN - nSHUTDN controls the power to enable the off-chip VDD regulator. The nSHUTDN pad is located on the VBAT power rail. If the battery protection circuit indicates a low battery condition, nSHUTDN is low; otherwise, nSHUTDN is high.

[0564] (ii) VPAD_EN - VPAD_EN controls the power enabled to the external regulator that provides the VPAD power supply. An internal signal responsive to this external signal ensures that the input from the VPAD pad does not draw additional current due to the floating input when the VPAD power supply is disabled. The VPAD_EN pad is an output on the VBAT power rail. If the battery protection signal indicates a low battery condition, the VPAD_EN signal is low. The VPAD_EN signal can be set low by a software instruction that starts a timer; the timer stops counting, driving VPAD_EN low. If the battery protection signal indicates a good battery condition, the following events can cause the VPAD_EN signal to rise (see the Event Handler section for a more detailed description): nPOR2_IN transitioning from low to high; SW / timer (programmable); WAKEUP transitions; low to high, and / or high to low (programmable); sensor connection transitions; low to high, and / or high to low (programmable); UART interrupts; and RTC time events (programmable).

[0565] (iii) UP_WAKEUP - UP_WAKEUP can be connected to the microprocessor wakeup pin. It is intended to wake the microprocessor from sleep mode or similar low power modes. The UP_WAKEUP pin is an output on the VPAD power rail. The UP_WAKEUP signal can be programmed to be active low, active high, or pulsed. The UP_WAKEUP signal can be set low by a software instruction that starts a timer; the timer stops counting and drives UP_WAKEUP low. If the battery protection signal indicates that the battery is good, then the following events can cause the UP_WAKEUP signal to rise (see the Event Handler section for more detailed description): nPOR2_IN transitions from low to high, SW / timer (programmable); WAKEUP transitions; low to high, and / or high to low, (programmable); sensor connection transitions; low to high, and / or high to low, (programmable); UART interrupt; and RTC time event (programmable). The WAKEUP signal can be delayed by a programmable amount. If WAKEUP is programmed to be pulsed, then the pulse width is programmed.

[0566] (iv) CLK_32KHZ - The CLK_32KHZ pad can be connected to a microprocessor to provide a low-speed clock. The clock is programmable on and off and can be programmably turned on to a wake-up event. The CLK_32KHZ pad is an output on the VPAD power rail. If the battery protection signal indicates that the battery is low, the CLK_32KHZ signal is low. The CLK_32KHZ output can be programmed to off via a programmable bit. The default is ON. The CLK_32KHZ signal can be disabled by a software instruction that starts a timer. Stopping the timer count can drive CLK_32KHZ down. If the battery protection signal shows that the battery is good, the following events can enable the CLK_32KHZ signal (see the event handler section for a more detailed description): nPOR2_IN transitions from low to high; SW / timer (programmable); WAKEUP transitions; low to high, and / or high to low, (programmable); sensor connection transitions; low to high, and / or high to low, (programmable); UART interrupt; RTC time event (programmable); and battery protection circuit detection of battery power.

[0567] (v) nRESET_OD - nRESET_OD can be connected to a microprocessor to reset the microprocessor. nRESET_OD is programmable as a wake-up event. The nRESET_OD pin is an output on the VPAD power rail. This pin is open drain (nfet output). If the battery protection signal indicates that the battery is low, the nRESET_OD signal is low. The nRESET_OD active time is programmable from 1 millisecond to 200 milliseconds. The default is 200ms. The following events can cause the nRESET_OD signal to be activated low (see the event controller section for a more detailed description): nPOR2_IN; SW / timer (programmable); WAKEUP transition; low to high, and / or high to low, (programmable); sensor connection transition; low to high, and / or high to low, (programmable); UART interrupt; and RTC time event (programmable).

[0568] (vi) UP_INT - UP_INT can be connected to the microprocessor to transmit an interrupt signal. UP_INT can be programmed as a wake-up event. The UP_INT pin is an output on the VPAD power rail. If the battery protection signal indicates that the battery is low, the UP_INT signal is low. The UP_INT signal can be set high by a software instruction that starts a timer; the timer's stop count drives UP_INT high. If the battery protection signal indicates that the battery is good, the following events can cause the UP_INT signal to be activated high (see the Event Controller section for a more detailed description): SW / Timer (programmable); WAKEUP transition; low to high, and / or high to low, (programmable); sensor connection transition; low to high and / or high to low, (programmable); UART interrupt; RTC time event (programmable); detection of low battery by the battery protection circuit; and any ASIC interrupt signal when not masked.

[0569] The ASIC features GPIO1 and GPIO0 pins that can act as boot mode controls for the microprocessor. A POR2 event resets a 2-bit counter whose bits are mapped to GPIO1 and GPIO0 (MSB, LSB, respectively). A rising edge on the UART interrupt signal increments the counter modulo 4 and resets to zero if the counter is incremented in state 11. The boot mode counter is presettable via the SPI.

[0570] Event Handlers / Monitors

[0571] The ASIC incorporates event handlers to define responses to events, including changes in system state and input signal changes. Events include all interrupt sources (e.g., UART_BRK, WAKE_UP, sensor connection, etc.). The event handler responses to stimuli are software-programmable via the SPI interface. However, some responses may be fixed (non-programmable).

[0572] Event handler functions include enabling / disabling VPAD_EN, enabling / disabling CLK_32KHZ, activating nRESET_OD, activating UP_WAKEUP, and activating UP_INT. Event monitor timers 1 through 5 are independently programmable from 250 milliseconds to 16,384 seconds in 250 millisecond increments. The timeout settings for event monitor timers 6 through 8 are hard-coded. The timeout settings for timers 6 and 7 are 1 minute, and the timeout setting for timer 8 is 5 minutes.

[0573] The ASIC also includes a monitor function to monitor the microprocessor's response to event triggers. The event monitor can be activated when the microprocessor fails to respond to event-induced activity. Once activated, the event monitor executes a programmable sequence of actions for event monitor timers 1 through 5, followed by a fixed sequence of actions for event monitor timers 6 through 8 to obtain a response from the microprocessor again. This sequence of actions includes interrupting, resetting, waking up, sending a 32 kHz clock, and shutting down and restarting the microprocessor.

[0574] During this sequence of events, if the microprocessor regains the ability to respond to recorded activity, the event monitor is reset. If the ASIC fails to receive a response from the microprocessor, the event monitor shuts down the microprocessor while allowing UART_BRK to restart the microprocessor and activates the alarm. When the alarm is activated, the alarm condition generates a square wave with a frequency of approximately 1 kHz on the ALARM pad in a programmable repeating pattern. The programmable pattern has two programmable sequences with programmable burst on and off times. The alarm has another programmable pattern that can be programmed via the SPI port. This other programmable pattern has two programmable sequences with programmable burst on and off times.

[0575] Digital-Analog (D / A)

[0576] In a preferred embodiment, the ASIC has two 8-bit D / A converters 4276, 4278 with the following features:

[0577] (i) The D / A adjusts in less than 1 millisecond with a load less than 50 pF.

[0578] (ii) The D / A has an accuracy of at least 8 bits.

[0579] (iii) The programmable output range is 0 to 1.22V or 0 to VDDA.

[0580] (iv) The temperature sensitivity of the D / A reference voltage is less than 1mV / °C.

[0581] (v)DNL is less than 1LSB.

[0582] (vi) The current consumption of D / A from VDDA power supply is less than 2uA.

[0583] (vii) Each D / A has an output 1 to a pad.

[0584] (viii) The D / A output has high impedance. The load current must be less than 1nA.

[0585] (ix) The D / A pads can be programmed to output digital signals from the registers. The output swing is from VSSA to VDDA.

[0586] Charger / data download interface

[0587] TX_EXT_OD 4280 is an open-drain output whose input is the signal on the TX_UP input pad. This causes the TX_EXT_OD pad to open when the UART is idle. The TX_EXT_OD pad has a comparator that monitors its voltage. If the voltage exceeds the comparator threshold voltage for the debounce period (1 / 4 second), the output nBAT_CHRG_EN (4281) goes low. This comparator and other related circuitry for this functionality is located on the VBAT and / or VDDBU boards.

[0588] The circuitry associated with this function must generate a low level on the TX_EXT_OD pin without disabling the assertion of nBAT_CHRG_EN, due to normal communication with an external device. If POR1 is active, nBAT_CHRG_EN will be high (inactive). The comparator threshold voltage is 0.50V to 1.2V. The comparator may have hysteresis. The falling threshold is approximately 25mV lower than the rising threshold.

[0589] The nRX_EXT pad inverts the signal on it and outputs it to RX_UP. This allows the nRX_EXT signal to fall slowly. nRX_EXT must accept inputs up to VBAT voltage. The nRX_EXT threshold is 1.22V ± 3%. The output of this comparator can be accessed via the SPI bus for reading by a microprocessor.

[0590] The nRX_EXT pad also incorporates a programmable current source that can be 80 ± 30 nA, with a maximum voltage of VBAT. The ASIC arrangement has a shielded programmable option to adjust the current from 30 nA to 200 nA in steps of less than 50 nA, with minimal shield layer changes. A programmable bit can be used to block UART interrupt detection and cause RX_UP to rise. In normal operation, this bit can be set high before enabling current sourcing into nRX_EXT and set low after current sourcing is disabled to ensure that no glitches are generated on RX_UP or UART interrupt events are generated. It should be noted that with a wet connection detector, although current sourcing into nRX_EXT is active, the RX comparator output indicating a low input voltage may indicate leakage current. The ASIC includes a pull-down resistor of approximately 100 k ohms on the nRX_EXT pad. This pull-down resistor is not connected when current sourcing is active.

[0591] Sensor connection switch

[0592] The ASIC may have a contact SEN_CONN_SW (4282) that is capable of detecting a low resistance to VSS (4284). SEN_CONN_SW draws a current of 5 to 25uA at SEN_CONN_SW = 0V and has a maximum open circuit voltage of 0.4V. The ASIC arrangement has a programmable option for the shield to adjust this current from 1uA to 20uA in steps of less than 5uA with minimal shield variation. SEN_CONN_SW has associated circuitry to detect the presence of a resistance between SEN_CONN_SW and VSSA (4234) with a threshold of 2k ohms to 15k ohms. The average current consumption of this circuit is a maximum of 50nA. Sampling must be used to achieve this low current.

[0593] Oscillator calibration circuit

[0594] The ASIC features counters whose inputs can be directed to either an internal or external clock source. One counter generates a programmable gating interval for the other counters. Depending on the 32kHz oscillator, the gating interval ranges from 1 second to 15 seconds. The clocks that can be directed to any counter include 32kHz, an RC oscillator, a high-speed RC oscillator, and input from any GPIO pad.

[0595] Oscillator Bypass

[0596] The ASIC can replace each of the oscillator outputs with an external clock. The ASIC has a register that can be written only when a specific TEST_MODE is active. This register has a bit that enables the external input of the RC oscillator and can be shared with other analog test control signals. However, if TEST_MODE is not active, this register does not enable any oscillator bypass bits.

[0597] The ASIC also features an input pin for an external clock, bypassing the RC oscillator. The GPIO_VBAT pin is located on the VBAT power plane. The ASIC also includes a bypass enable pin, OSC32K_BYPASS, for the 32kHz oscillator. When OSC32K_BYPASS is high, driving the OSC32KHZ_IN pin provides a 32kHz oscillator output. Note that the OSC32KHZ_IN pin is typically connected to a crystal.

[0598] The ASIC has an input for an external clock to bypass the HS_RC_OSC. The bypass is enabled via a programmable register bit. The HS_RC_OSC can be powered programmably via a GPIO on the VDD board or via a GPIO on the VPAD board.

[0599] SPI slave port

[0600] The SPI slave port includes an interface consisting of a chip select input (SPI_nCS) 4289, a clock input (SPI_CK) 4286, a serial data input (SPI_MOSI) 4287, and a serial data output (SPI_MISO) 4288. The chip select input (SPI_nCS) is an active-low input that is activated by an off-chip SPI master to initiate and terminate an SPI transaction. When SPI_nCS is activated low, the SPI slave port configures itself as an SPI slave and processes data based on the clock input (SPI_CK). When SPI_nCS is deactivated, the SPI slave port resets itself and remains in reset mode. Because this SPI interface supports block transfers, the master should maintain SPI_nCS at a level until the transfer is complete.

[0601] The SPI clock input (SPI_CK) can always be activated by the SPI master. The SPI slave port uses the rising edge of SPI_CK to latch input data on the SPI_MOSI input and uses the falling edge of SPI_CK to drive output data on the SPI_MISO output. The serial data input (SPI_MOSI) is used to transfer data from the SPI master to the SPI slave. All data bits become active after the falling edge of SPI_CK. The serial data output (SPI_MISO) is used to transfer data from the SPI slave to the SPI master. All data bits become active after the falling edge of SPI_CK.

[0602] SPI_nCS, SPI_CK, and SPI_MOSI are normally driven by the SPI master unless the SPI master is disabled. If VPAD_EN is low, these inputs are regulated so that the current consumption associated with these inputs is less than 10nA and the SPI circuitry remains in reset or disabled. SPI_MISO is driven by the SPI slave port only when SPI_nCS is active; otherwise, SPI_MISO is tri-stated.

[0603] The chip select (SPI_nCS) defines and structures the data transfer packet for SPI data transactions. A data transfer packet consists of three parts: a 4-bit command portion, followed by a 12-bit address portion, and then any number of 8-bit data bytes. Command bit 3 serves as an indicator. A "1" indicates a write operation, and a "0" indicates a read operation. The combinations of command bits 2, 1, and 0 have the following definitions. Unused combinations are undefined.

[0604] (i)0000: read data and increase address

[0605] (ii) 0001: Read data without changing the address

[0606] (iii)0010: Read data, decrease address

[0607] (iv) 1000: Write data and increase address

[0608] (v)1001: Write data without changing the address

[0609] (vi) 1010: Write data without increasing the address

[0610] (vii) x011: Test port addressing

[0611] The 12-bit address portion defines the starting byte address. If SPI_nCS remains valid after the first data byte to indicate a multi-byte transfer, the address is incremented by one after each byte is transferred. <11> (Address<11:0>) represents the most significant address bit. The address wraps around after reaching the address boundary.

[0612] Data is in bytes and can be transferred in blocks by extending SPI_nCS so that all bytes are transferred in one packet.

[0613] Microprocessor interrupts

[0614] The ASIC features an output, UP_INT, at the VPAD logic level, for sending interrupt signals to the host microprocessor. The microprocessor interrupt module consists of an interrupt status register, an interrupt mask register, and functionality that logically ORs all interrupt states into a single microprocessor interrupt. Interrupts are implemented to support both edge-sensitive and level-sensitive interrupts. The interrupt signal polarity is programmable. The default interrupt polarity is TBD.

[0615] In a preferred embodiment, all interrupt signal sources on the AFE ASIC are recorded in an interrupt status register. Writing a "1" to the corresponding interrupt status bit clears the corresponding pending interrupt signal. All interrupt signal sources on the AFE ASIC are maskable using the interrupt mask register. Writing a "1" to the corresponding interrupt signal mask bit enables masking of the corresponding pending interrupt signal. Writing a "0" to the corresponding interrupt signal mask bit disables masking of the corresponding interrupt signal. The default state of the interrupt signal mask register is TBD.

[0616] General-purpose input / output (GPIO) / parallel test port

[0617] In an embodiment of the present invention, the ASIC may have eight GPIOs that operate with VPAD level signals. The ASIC may have one GPIO that operates with VBAT level signals and one GPIO that operates with VDD level signals. All GPIOs have at least the following characteristics:

[0618] (i) Register bits control the selection and indication of each GPIO.

[0619] (ii) The ASIC has a way to configure the GPIOs as inputs that can be read via the SPI interface.

[0620] (iii) The ASIC has a way to configure the GPIO as an input that generates an interrupt signal.

[0621] (iv) The ASIC has a way to configure each GPIO as an output controlled by a register bit that can be written through the SPI interface.

[0622] (v) The ASIC can programmably output the input signal applied to GPIO_VBAT or GPIO_VDD to GPIO (VPAD power supply board). (Level shift function).

[0623] (vi) The ASIC has a way to configure each GIPO as an input to the oscillator calibration circuit.

[0624] (vii) The ASIC has a way to configure the output of each general purpose comparator to at least one GIPO on each power board. The polarity of the comparator output is programmable via a programmable bit.

[0625] (viii)GPIO has the ability to generate microprocessor interrupt signals.

[0626] (ix)GPIO can be programmed to open-drain output.

[0627] (x) The GPIO on the VPAD power board can be configured to implement microprocessor boot control.

[0628] The parallel test port shares the same 8-bit GPIO as the VPAD voltage board. This test port can be used to observe register contents and various internal signals. In normal mode, the output of this port is controlled by the port configuration register. Writing 8'hFF to the GPIO_O1S_REG and GPIO_O2S_REG registers directs test port data to the GPIO outputs, while writing 8'h00 to the GPIO_ON_REG register disables test port data and enables GPIO data to the GPIO outputs.

[0629] Registers and pre-grouped internal signals can be observed through this test port by addressing the target register via the SPI slave port. The SPI packet has the command bit set to 4'b0011, followed by the 12-bit target register address. The parallel test port continuously displays the contents of the addressed register until the next test port addressing command is received.

[0630] Analog test port

[0631] The IC has a multiplexer feed pad TP_ANAMUX (4290) that provides visibility to internal analog circuit nodes for testing. The IC also has a multiplexer feed pad TP_RES (4260) that provides visibility to internal analog circuit nodes for testing. The pads can also accommodate precision 1M resistors in conventional applications for various system calibrations.

[0632] Chip ID

[0633] The ASIC includes a 32-bit mask-programmable ID. This ID can be read by a microprocessor using an SPI interface. This ID is to be placed in the analog electronics block so that ID changes do not require chip rerouting. The design should require only a metal or contact mask change to change the ID.

[0634] Spare test output

[0635] The ASIC has 16 spare digital output signals that can be multiplexed into 8-bit GPIOs under instructions sent through the SPI interface. These signals can be organized into two 8-bit bytes and can be connected to VSS if not used.

[0636] Digital Test

[0637] The ASIC has a test mode controller that uses two input pins TEST_CTL0 (4291) and TEST_CTL1 (4292). The test controller generates signals by combining test control signals with the following functionality (TEST_CTL<1:0>):

[0638] (i) 0 is the normal operating mode;

[0639] (ii) 1 is the simulation test mode;

[0640] (iii) 2 is scanning mode;

[0641] (iv) 3 is the analog test mode with VDD_EN controlled by the input of GPIO_VBAT.

[0642] The test controller logic is split between the VDD and VDDBU power planes. During scan mode, TEST_VBAT should be active high to condition the analog outputs for the digital logic. The ASIC has scan chains that are implemented in as much of the digital logic as reasonably possible for fast digital testing.

[0643] Leakage test pin

[0644] The ASIC has a pin called LT_VBAT, which, when high, puts all analog blocks into a disabled mode, drawing only leakage current from the power supply. LT_VBAT causes all digital outputs from the analog blocks to be in a stable high or low state, without affecting the interface logic current consumption. The LT_VBAT pad is located on the VBAT pad and has a pull-down resistor between 10k ohms and 40k ohms.

[0645] Power demand

[0646] In an embodiment of the present invention, the ASIC includes a low-power mode in which, at a minimum, the microprocessor clock is shut down, the 32kHz real-time clock is running, and circuitry is activated to detect sensor connection, a level change on the WAKE_UP pin, or a break (interrupt) on the nRX_EXT input. This mode consumes a maximum of 4.0uA of total current from VBAT (VDDBU), VDD, and VDDA. When the battery protection circuit detects low battery (see the description of the battery protection circuit), the ASIC enters a mode in which only the VBAT and VDDBU power rails are active. This is referred to as a low-battery state. The VBAT current in this mode is less than 0.3uA.

[0647] The average current consumption of all power supplies is less than 7uA when the ASIC is programmed for the following potentiostat configuration: any one work electrode is active in H2O2 (peroxide) mode with its voltage set to 1.535V; the counter amplifier is powered on with VSET_RE set to 1.00V; a 20MEG load resistor is connected between the work and counter electrodes; the counter and RE are connected together; and, assuming one working electrode current is measured every minute. After calibration, the measured current should be 26.75nA ± 3%. Enabling additional working electrodes increases the combined current consumption by less than 2uA, with the working electrode current being 25nA.

[0648] Insofar as the ASIC is programmed in a potentiostatic configuration, wherein the diagnostic function is enabled to measure the impedance of one of the WORK electrodes relative to the COUNTER electrode, the ASIC is configured to conform to the following characteristics:

[0649] (i) Test frequencies: 0.1, 0.2, 0.3, 0.5 Hz, 1.0, 2.0, 5.0, 10, 100, 1000 and 4000 Hz.

[0650] (ii) The above frequency measurements shall not exceed 50 seconds.

[0651] (iii) The total charge provided to the ASIC is less than 8 millicoulombs.

[0652] environment

[0653] In a preferred embodiment of the present invention, the ASIC:

[0654] (i) Operate and meet all specifications over the commercial operating temperature range of 0°C to 70°C.

[0655] (ii) Functional operation from -20°C to 80°C, but with reduced accuracy.

[0656] (iii) Expected to operate after storage in the temperature range of -30°C to 80°C.

[0657] (iv) Expected to operate in a relative humidity range of 1% to 95%.

[0658] (v)ESD protection is better than ±2kV, Human Body Model on all pins unless otherwise specified when packaged in TBD package.

[0659] (vi) Configure the WORK1-WORK5, COUNTER, RE, TX_EXT_OD, and nRX_EXT contacts to withstand more than ±4KV Human Body Model.

[0660] (vii) The leakage current of WORK1-WORK5 and RE contacts configured at 40°C is less than 0.50nA.

[0661] In an embodiment of the present invention, the ASIC may be manufactured using a .25 micron CMOS process and backup data for the ASIC is located on a DVD disc 916-TBD.

[0662] As described in detail above, the ASIC provides the necessary analog electronics to: (i) support a variety of potentiostats and interface with oxygen- or peroxide-based multi-terminal glucose sensors; (ii) interface with a microprocessor to form a micropower sensor system; and (iii) perform EIS diagnostics based on measurements of EIS-based parameters. Measurement and calibration of EIS-based parameters are described herein according to embodiments of the present invention.

[0663] As previously mentioned, impedance at frequencies ranging from 0.1 Hz to 8 kHz has previously provided information about the sensor electrode's state. The AFE IC incorporates circuitry to generate the measurement drive signal and to perform the measurement used to calculate the impedance. Design considerations for this circuitry include current consumption, accuracy, measurement speed, the amount of processing required, and the amount of time required for the control microprocessor.

[0664] In a preferred embodiment of the present invention, the technique used by the AFE IC to measure electrode impedance is to superimpose a sinusoidal voltage on the DC voltage driving the electrode and measure the phase and amplitude of the resulting AC current. To generate the sine wave, a digitally synthesized sinusoidal current is incorporated into the AFE IC. This digital technique is used because the frequency and phase can be precisely controlled using a crystal-generated time base, and it can easily generate frequencies from DC to 8 kHz. The sinusoidal current is applied through a resistor in series with the voltage source, thereby adding an AC component to the electrode voltage. This voltage is the AC drive voltage. This voltage is then buffered by the amplifier that drives the selected sensor electrode.

[0665] The current driving the electrodes contains an AC current component derived from the driving sine wave and is converted to a voltage. This voltage is then processed by multiplying it with a square wave that has a fixed phase relative to the resulting sine wave. This multiplied voltage is then integrated. After a programmable number of integration intervals—one equal to 1 / 2 cycle of the driving sine wave—the voltage is measured by the ADC. The real and imaginary components of the impedance are obtained by calculations involving the integrated voltage.

[0666] The advantage of using an integrator for impedance measurement is that the measured noise bandwidth is significantly reduced compared to simply sampling the waveform. Furthermore, the sampling time...

Claims

1. A method for calculating a single, fused sensor glucose value based on individual sensor glucose values ​​of a plurality of redundant working electrodes of a glucose sensor, the method comprising: Performing a respective electrochemical impedance spectroscopy (EIS) procedure on each of the plurality of redundant working electrodes to obtain a membrane resistance Rmem value of each of the working electrodes; Calculate the Rmem fusion weight corresponding to each of the working electrodes based on the Rmem value corresponding to each of the multiple working electrodes; measuring a noise value of each of the plurality of working electrodes; Calculate the noise fusion weight corresponding to each working electrode based on the noise value corresponding to each working electrode of the plurality of working electrodes; Measuring a calibration factor CF value of each of the plurality of working electrodes; Calculate the CF fusion weight corresponding to each working electrode based on the CF value corresponding to each working electrode in the plurality of working electrodes; For each of the plurality of working electrodes, calculating an overall fusion weight based on the Rmem fusion weight, the noise fusion weight and the CF fusion weight of the electrode; calculating a single, fused sensor glucose value based on the corresponding overall fusion weight and the respective sensor glucose value of each of the plurality of redundant working electrodes, Wherein, between sensor activation and expiration of the calculated time period, the value of the Rmem fusion weight is modified by a first variable to generate a modified Rmem fusion weight, wherein the first variable is a function of the variance of the respective Rmem fusion weight values ​​of the plurality of working electrodes.

2. The method of claim 1, further comprising: A validity check is performed on the respective sensor glucose value.

3. The method of claim 2, wherein: If all sensor glucose values ​​are invalid, then the fused sensor glucose value is determined to be invalid.

4. The method of claim 2, wherein: The glucose sensor comprises two redundant working electrodes, and wherein if the sensor glucose value of one of the two working electrodes is valid and the sensor glucose value of the other of the two working electrodes is invalid, the fused sensor glucose value is equal to the valid sensor glucose value.

5. The method of claim 1, wherein: The Rmem values ​​are obtained based on the respective 1 kHz real impedance values.

6. The method of claim 1, further comprising: For each working electrode, the respective noise fusion weights and CF fusion weights are combined to calculate the combined noise and CF fusion weights.

7. The method of claim 6, wherein: The combined noise and CF fusion weights are calculated by weighting and averaging the respective noise fusion weights and CF fusion weights according to predefined parameters.

8. The method of claim 1, wherein: After a predetermined period of time from sensor activation, the value of the Rmem fusion weight is equal to 0.

9. The method of claim 8, wherein: The predetermined period of time is approximately 25 hours.

10. The method of claim 6, further comprising: The combined noise and CF fusion weight is modified by a second variable to calculate a modified combined noise and CF fusion weight.

11. The method of claim 10, wherein: The overall fusion weight is calculated based on the modified Rmem fusion weight and the modified combined noise and CF fusion weight.

12. The method of claim 10, wherein: The sum of the values ​​of the first variable and the second variable is equal to 1.

13. The method of claim 1, wherein each of said respective EIS procedures is performed over a range of frequencies.

14. A program code storage device comprising: computer readable medium; as well as A computer readable program code stored on the computer readable medium, the computer readable program code having instructions that, when executed, cause the physical microcontroller to perform a method for calculating a single, fused sensor glucose value based on respective sensor glucose values ​​of a plurality of redundant working electrodes of a glucose sensor by: Performing a respective electrochemical impedance spectroscopy (EIS) procedure on each of the plurality of redundant working electrodes to obtain a membrane resistance Rmem value of each of the working electrodes; Calculate the Rmem fusion weight corresponding to each of the working electrodes based on the Rmem value corresponding to each of the multiple working electrodes; Obtaining a noise value of each of the plurality of working electrodes; Calculate the noise fusion weight corresponding to each working electrode based on the noise value corresponding to each working electrode of the plurality of working electrodes; Obtaining a calibration factor CF value of each working electrode of the plurality of working electrodes; Calculate the CF fusion weight corresponding to each working electrode based on the CF value corresponding to each working electrode in the plurality of working electrodes; For each of the plurality of working electrodes, calculating an overall fusion weight based on the Rmem fusion weight, the noise fusion weight and the CF fusion weight of the electrode; calculating a single, fused sensor glucose value based on the corresponding overall fusion weight and the respective sensor glucose value of each of the plurality of redundant working electrodes, In which, the computer-readable program code has the following instructions, which, when executed, also cause the microcontroller to modify the value of the Rmem fusion weight of each working electrode through a first variable to generate a modified Rmem fusion weight, wherein the first variable is a function of the variance of the respective Rmem fusion weight values ​​of the multiple working electrodes.

15. The apparatus of claim 14, wherein: The computer readable program code has instructions which, when executed, further cause the microcontroller to perform a validity check on the respective sensor glucose value.

16. The apparatus of claim 14, wherein: The computer readable program code has the following instructions, which, when executed, also cause the microcontroller to combine the respective noise fusion weights and CF fusion weights of each working electrode to calculate the combined noise and CF fusion weights, and the combined noise and CF fusion weights are calculated by weighting and averaging the respective noise fusion weights and CF fusion weights according to predefined parameters.

17. The apparatus of claim 16, wherein: The computer readable program code has instructions which, when executed, further cause the microcontroller to modify the combined noise and CF fusion weights by a second variable to calculate modified combined noise and CF fusion weights.

18. The apparatus of claim 17, wherein: The computer readable program code has instructions that, when executed, further cause the microcontroller to calculate the overall fusion weight based on the modified Rmem fusion weight and the modified combined noise and CF fusion weight.

19. A method for calculating a single, fused sensor glucose value based on individual sensor glucose values ​​of a plurality of redundant working electrodes (WE1, WE2) of a glucose sensor, the method comprising: Performing a respective electrochemical impedance spectroscopy (EIS) procedure on each of the plurality of redundant working electrodes to obtain a membrane resistance Rmem value of each of the working electrodes; Calculate the Rmem fusion weight corresponding to each of the working electrodes based on the Rmem value corresponding to each of the multiple working electrodes; measuring a noise value of each of the plurality of working electrodes; Calculate the noise fusion weight corresponding to each working electrode based on the noise value corresponding to each working electrode of the plurality of working electrodes; Measuring a calibration factor CF value of each of the plurality of working electrodes; Calculate the CF fusion weight corresponding to each working electrode based on the CF value corresponding to each working electrode in the plurality of working electrodes; For each of the plurality of working electrodes, calculating an overall fusion weight based on the Rmem fusion weight, the noise fusion weight and the CF fusion weight of the electrode; A single, fused sensor glucose value is calculated based on the corresponding overall fusion weight and the respective sensor glucose value for each of the plurality of redundant working electrodes.

20. The method of claim 19, further comprising: performing a validity check on said respective sensor glucose value, Wherein, if all sensor glucose values ​​are invalid, the fused sensor glucose value is determined to be invalid, or Wherein the glucose sensor comprises two redundant working electrodes, and wherein if the sensor glucose value of one of the two working electrodes is valid and the sensor glucose value of the other of the two working electrodes is invalid, the fused sensor glucose value is equal to the valid sensor glucose value.

21. The method of claim 19, wherein: The Rmem values ​​are obtained based on the respective 1 kHz real impedance values.

22. The method of claim 19, further comprising: For each working electrode, the respective noise fusion weights and CF fusion weights are combined to calculate a combined noise and CF fusion weight, wherein the combined noise and CF fusion weight is calculated by weighting and averaging the respective noise fusion weights and CF fusion weights according to predefined parameters.

23. The method of claim 19, wherein: After a predetermined period of time from sensor activation, the value of the Rmem fusion weight is equal to 0.

24. The method of claim 23, wherein: Between sensor activation and expiration of the calculated time period, the value of the Rmem fusion weight is modified by a first variable to generate a modified Rmem fusion weight, the first variable being a function of a variance of the respective Rmem fusion weight values ​​of the plurality of working electrodes.

25. The method of claim 24, further comprising: For each working electrode, the respective noise fusion weights and CF fusion weights are combined to calculate a combined noise and CF fusion weight, wherein the fused noise and CF fusion weight is calculated by weighting and averaging the respective noise fusion weights and CF fusion weights according to predefined parameters.

26. The method of claim 25, further comprising: The combined noise and CF fusion weight is modified by a second variable to calculate a modified combined noise and CF fusion weight.

27. The method of claim 26, wherein: The overall fusion weight is calculated based on the modified Rmem fusion weight and the modified combined noise and CF fusion weight.

28. The method of claim 26, wherein: The sum of the values ​​of the first variable and the second variable is equal to 1.

29. The method of claim 19, wherein each of said respective EIS procedures is performed over a range of frequencies.

30. A program code storage device comprising: computer readable medium; as well as A computer readable program code stored on the computer readable medium, the computer readable program code having instructions that, when executed, cause the physical microcontroller to perform a method for calculating a single, fused sensor glucose value based on respective sensor glucose values ​​of a plurality of redundant working electrodes of a glucose sensor by: Performing a respective electrochemical impedance spectroscopy (EIS) procedure on each of the plurality of redundant working electrodes to obtain a membrane resistance Rmem value of each of the working electrodes; Calculate the Rmem fusion weight corresponding to each of the working electrodes based on the Rmem value corresponding to each of the multiple working electrodes; Obtaining a noise value of each of the plurality of working electrodes; Calculate the noise fusion weight corresponding to each working electrode based on the noise value corresponding to each working electrode of the plurality of working electrodes; Obtaining a calibration factor CF value of each working electrode of the plurality of working electrodes; Calculate the CF fusion weight corresponding to each working electrode based on the CF value corresponding to each working electrode in the plurality of working electrodes; For each of the plurality of working electrodes, calculating an overall fusion weight based on the Rmem fusion weight, the noise fusion weight and the CF fusion weight of the electrode; A single, fused sensor glucose value is calculated based on the corresponding overall fusion weight and the respective sensor glucose value for each of the plurality of redundant working electrodes.

31. The apparatus of claim 30, wherein: The computer readable program code has instructions which, when executed, further cause the microcontroller to perform a validity check on the respective sensor glucose value.

32. The apparatus of claim 30, wherein: The computer readable program code has the following instructions, which, when executed, also cause the microcontroller to combine the respective noise fusion weights and CF fusion weights of each working electrode to calculate the combined noise and CF fusion weights, and the combined noise and CF fusion weights are calculated by weighting and averaging the respective noise fusion weights and CF fusion weights according to predefined parameters.

33. The apparatus of claim 32, wherein: The computer-readable program code has the following instructions, which, when executed, also cause the microcontroller to modify the value of the Rmem fusion weight of each working electrode through a first variable to generate a modified Rmem fusion weight, wherein the first variable is a function of the variance of the respective Rmem fusion weight values ​​of the multiple working electrodes.

34. The apparatus of claim 33, wherein: The computer readable program code has the following instructions, which, when executed, also cause the microcontroller to modify the combined noise and CF fusion weight by a second variable to calculate a modified combined noise and CF fusion weight, and calculate the overall fusion weight based on the modified Rmem fusion weight and the modified combined noise and CF fusion weight.

Images