A Soft-Switching High-Frequency Link Four-Leg Matrix Inverter Topology and Pulse Width Modulation Method

Abstract

The invention discloses a soft-switching high-frequency link four-leg matrix inverter topology and a pulse width modulation method, belonging to the technical field of power electronic power conversion. The topology includes a push-pull forward high-frequency inverter structure with a clamping capacitor, a high-frequency transformer, a post-stage four-leg matrix converter, and an output LC filter connected in sequence; the topology suppresses overvoltage spikes of all switching tubes by introducing a clamping capacitor; the pulse width modulation method includes an SPWM signal generation method, a deconstruction and reconstruction method, and a fourth-leg independent control method, which can achieve zero-current-switching (ZCS) turn-on and zero-voltage-switching (ZVS) turn-off of the front-stage switching tubes in the topology, and ZVS turn-on and ZVS turn-off of the post-stage switching tubes, and can reduce the switching losses of the switching tubes when operating in a high-frequency state; at the same time, under unbalanced working conditions, the topology can output three-phase balanced voltages. The invention can achieve two-stage power conversion and has the advantages of high circuit efficiency, simple control method, etc.

Description

Technical Field

[0001] The present invention relates to the technical field of power electronic power conversion, and in particular to a soft-switching high-frequency link four-leg matrix inverter topology and a pulse width modulation method. Background Art

[0002] An inverter is a topological device that converts direct current electrical energy into alternating current electrical energy. The high-frequency link inverter uses a high-frequency transformer to replace the power-frequency transformer, overcoming the disadvantages of the traditional transformer such as large volume, high noise, and high cost. The conversion process of the high-frequency link matrix inverter has three power characteristics: DC / HFAC / LFAC, where HFAC: high-frequency alternating current, LFAC: low-frequency alternating current. It can be seen that a DC / AC, i.e., direct current / alternating current inversion link, appears in such an inverter, and this link is located on the primary side of the transformer. An AC / AC, i.e., alternating current / alternating current conversion link, also appears, and this link is often referred to as a cycloconverter or matrix converter link and is located on the secondary side of the transformer. Compared with the traditional converter, the matrix converter has no intermediate energy storage link, uses bidirectional switches, can realize bidirectional energy flow, has a compact structure, small volume, high efficiency, and the output voltage amplitude and frequency can be independently controlled.

[0003] The invention patent "A Modulation Method for a Single-Phase Push-Pull Forward High-Frequency Link Matrix Inverter Topology" (CN201710821717.6) proposes a single-phase push-pull forward high-frequency link matrix converter topology structure and a matching modulation scheme. The front stage combines a forward converter and a push-pull converter through a clamping capacitor, integrating the advantages of both and overcoming their disadvantages at the same time. However, due to the limitation of the topology structure, it can only handle single-phase loads. The three-phase four-leg matrix converter is adopted in the rear stage of the soft-switching high-frequency link four-leg matrix inverter topology proposed in this paper, so it can handle three-phase unbalanced loads.

[0004] To meet the unbalanced working conditions that often occur in real life, a push-pull forward three-phase four-leg high-frequency link matrix inverter is proposed. By adding a fourth leg, when the system is connected to a three-phase unbalanced load, the switching tube of the fourth leg can be controlled to provide a current path for the zero-sequence current, so as to output three-phase balanced voltage.

[0005] Due to the existence of the leakage inductance of the high-frequency transformer, when the high-frequency link matrix inverter commutates, a large voltage overshoot is generated on the power tubes of the matrix converter on the secondary side of the high-frequency transformer. Therefore, the safe commutation of the matrix / cycloconverter on the secondary side of the high-frequency transformer has always been a technical difficulty restricting the wide promotion of high-frequency link inverters. Currently, there are mainly the following several safe commutation strategies: ① By adding an active clamp to suppress the voltage overshoot, soft switching can be achieved, but the introduced clamp circuit increases the cost, and the additional controllable power tubes also make the control more complex; ② The unipolar and bipolar phase-shifted control strategies achieve the natural commutation of the inductor current by means of the commutation overlap of the cycloconverter, and achieve ZVS of the power tubes, but there are problems such as the commutation overlap time being difficult to control; ③ Introduce a series resonance circuit in the front-stage inverter to achieve soft commutation of the power tubes. At this time, it is required that the power tube switching occurs at the zero-current moment, and the control of the output energy requires judging the resonant working state of the resonance circuit, making the control method complex.

[0006] However, although the above strategies can achieve safe commutation, they make the modulation and control of the inverter more complex, resulting in a decrease in system reliability, thus affecting the popularization and use of this type of converter. Summary of the Invention

[0007] The technical problem to be solved by the present invention is to provide a soft-switching high-frequency link four-leg matrix inverter topology and a pulse width modulation method, which can achieve soft switching of all switching tubes in the topology, reduce the common-mode voltage output by the inverter, reduce the switching tube loss, improve the efficiency of the converter, achieve the balance of the three-phase output voltage under unbalanced working conditions through the fourth leg, and make the modulation process simpler, making the high-frequency transformer easier to promote and use.

[0008] To solve the above technical problems, the technical solution adopted by the present invention is:

[0009] A soft-switching high-frequency link four-leg matrix inverter topology, including a push-pull forward high-frequency inverter structure with a clamping capacitor, a high-frequency transformer, a three-phase four-leg matrix converter, an output LC filter, and a three-phase unbalanced load connected in sequence;

[0010] The push-pull forward high-frequency inverter structure with a clamping capacitor includes a DC input voltage Vdc, a controllable switch tube S1, a controllable switch tube S2, and a capacitor Cs;

[0011] The high-frequency transformer T includes a primary side and a secondary side. The primary side includes a coil N11 and a coil N12, and the secondary side includes a coil N2;

[0012] The three-phase four-leg matrix converter includes a first leg, a second leg, and a third leg, and a fourth leg for dealing with unbalanced working conditions;

[0013] The first bridge arm includes controllable switching tubes Sa1, Sa4, Sa2, and Sa3; the second bridge arm includes controllable switching tubes Sb1, Sb4, Sb2, and Sb3; the third bridge arm includes controllable switching tubes Sc1, Sc4, Sc2, and Sc3; and the fourth bridge arm includes controllable switching tubes Sd1, Sd4, Sd2, and Sd3.

[0014] The LC filter includes a first inductor La, a second inductor Lb, a third inductor Lc, a fourth inductor Ld, a first capacitor Ca, a second capacitor Cb, and a third capacitor Cc.

[0015] The three-phase unbalanced load includes loads Ra, Rb, and Rc with different load parameters.

[0016] A further improvement of the technical solution of the present invention is that the positive pole of the DC input voltage Vdc is respectively connected to the collector of the controllable switching tube S1 and one end of the coil N12, and the negative pole of the DC input voltage Vdc is respectively connected to the emitter of the controllable switching tube S2 and one end of the coil N11; the emitter of the controllable switching tube S1 is respectively connected to one end of the clamping capacitor Cs and the other end of the coil N11; the collector of the controllable switching tube S2 is respectively connected to the other end of the clamping capacitor Cs and the other end of the coil N12.

[0017] One end of the secondary side N2 of the high-frequency transformer T is respectively connected to the collectors of the controllable switching tubes Sa1, Sb1, Sc1, and Sd1, and the other end of the secondary side N2 of the high-frequency transformer T is respectively connected to the collectors of the controllable switching tubes Sa3, Sb3, Sc3, and Sd3; the emitter of the controllable switching tube Sa1 is connected to the emitter of the controllable switching tube Sa4, the emitter of the controllable switching tube Sb1 is connected to the emitter of the controllable switching tube Sb4, the emitter of the controllable switching tube Sc1 is connected to the emitter of the controllable switching tube Sc4, and the emitter of the controllable switching tube Sd1 is connected to the emitter of the controllable switching tube Sd4.

[0018] The emitter of the controllable switching tube Sa2 is connected to the emitter of the controllable switching tube Sa3, the emitter of the controllable switching tube Sb2 is connected to the emitter of the controllable switching tube Sb3, the emitter of the controllable switching tube Sc2 is connected to the emitter of the controllable switching tube Sc3, and the emitter of the controllable switching tube Sd2 is connected to the emitter of the controllable switching tube Sd3.

[0019] The collector of the controllable switch tube Sa4 is connected to the collector of the controllable switch tube Sa2 and then connected to one end of the first inductor La. The other end of the first inductor La is connected to one end of the first capacitor Ca and one end of the load Ra. The other end of the load Ra is respectively connected to the other ends of the load Rb and the load Rc. The other end of the first capacitor Ca is respectively connected to the other ends of the second capacitor Cb and the third capacitor Cc. The neutral points of the load Ra, the load Rb, and the load Rc are connected to the neutral points of the first capacitor Ca, the second capacitor Cb, and the third capacitor Cc.

[0020] The collector of the controllable switch tube Sb4 is connected to the collector of the controllable switch tube Sb2 and then connected to one end of the second inductor Lb. The other end of the second inductor Lb is connected to one end of the second capacitor Cb and one end of the load Rb.

[0021] The collector of the controllable switch tube Sc4 is connected to the collector of the controllable switch tube Sc2 and then connected to one end of the third inductor Lc. The other end of the third inductor Lc is connected to one end of the third capacitor Cc and one end of the load Rc.

[0022] The collector of the controllable switch tube Sd4 is connected to the collector of the controllable switch tube Sd2 and then connected to one end of the fourth inductor Ld. The other end of the fourth inductor Ld is connected to the other ends of the first capacitor Ca, the second capacitor Cb, the third capacitor Cc, the load Ra, the load Rb, and the load Rc.

[0023] A further improvement of the technical solution of the present invention lies in: including an SPWM signal generation method, a deconstruction and reconstruction method, and a fourth bridge arm independent control method.

[0024] A further improvement of the technical solution of the present invention lies in: the SPWM signal generation method generates six pulse width modulation signals SPWM1, SPWM2, SPWM3, SPWM4, SPWM5, and SPWM6, which specifically include the following steps:

[0025] A1. Given a balanced three-phase sine signal of the desired output;

[0026] A2. The three-phase sine signal is compared with a sawtooth carrier wave to generate complementary SPWM1 and SPWM2, complementary SPWM3 and SPWM4, and complementary SPWM5 and SPWM6.

[0027] A further improvement of the technical solution of the present invention lies in: the deconstruction and reconstruction method specifically includes the following steps:

[0028] B1. SPWM2, SPWM4, and SPWM6 generate signal A through "AND" logic; SPWM2, SPWM4, and SPWM6 generate signal B through "OR" logic;

[0029] B2. Signal A is frequency-divided by two to obtain V 1 and inverted to generate Meanwhile, signal B is also divided by two and inverted to generate V 2 ,

[0030] B3, V 1 and perform a NAND logic operation to obtain an intermediate signal V m , V 2 and perform a NAND logic operation to obtain an intermediate signal V p ;

[0031] B4, invert V m to obtain the drive signal of the pre-stage controllable switch tube S 1 and invert V p to obtain the drive signal of the pre-stage controllable switch tube S 2 ;

[0032] B5, the intermediate signal V p and 6-channel SPWM modulation signals perform an OR logic operation to generate the drive signals of the first three bridge arm switch tubes S a1 , S b1 , S c1 , S a2 , S b2 , S c2 , and the intermediate signal V m and 6-channel SPWM modulation signals perform an OR logic operation to generate the drive signals of the first three bridge arm switch tubes S a3 , S b3 , S c3 , S a4 , S b4 , S c4 ;

[0033] A further improvement of the technical solution of the present invention lies in that: the control of the fourth bridge arm of the three-phase four-bridge arm matrix converter is independent of the first three bridge arms, and the specific control method of the fourth bridge arm includes the following steps:

[0034] C1. Sample the phase voltages of the three-phase unbalanced load, sum the three-phase voltages and multiply by the amplification factor to obtain the modulation signal of the fourth bridge arm;

[0035] C2. Compare the modulation signal of the fourth bridge arm with the carrier wave to generate complementary modulation signals PWM7 and PWM8;

[0036] C3. The intermediate signal Vp and the PWM7 and PWM8 signals perform an OR logic operation respectively to generate the drive signals of the fourth bridge arm switch tubes Sd1 and Sd2, and the intermediate signal Vm and the PWM7 and PWM8 signals perform an OR logic operation respectively to generate the drive signals of the fourth bridge arm switch tubes Sd3 and Sd4.

[0037] A further improvement of the technical solution of the present invention lies in that: under the control of the pulse width modulation method, the controllable switch tubes S1 and S2 are alternately turned on, and there is a built-in dead zone between the drives of the controllable switch tubes S1 and S2; when the controllable switch tube S1 is turned on and the controllable switch tube S2 is turned off, the front-stage circuit outputs a positive voltage to the rear-stage; when the controllable switch tube S2 is turned on and the controllable switch tube S1 is turned off, the front-stage circuit outputs a negative voltage to the rear-stage; when both the controllable switch tubes S1 and S2 are turned off, the front-stage circuit outputs a zero voltage to the rear-stage.

[0038] A further improvement of the technical solution of the present invention lies in that: the driving signal characteristics of the front-stage switch tube are different from those of the front-stage switch tube under the modulation method of the single-phase push-pull forward high-frequency link matrix converter.

[0039] A further improvement of the technical solution of the present invention lies in that: when the front-stage circuit outputs a positive voltage to the rear-stage, the positive-group switch tubes Sa1, Sb1, Sc1, Sd1, Sa2, Sb2, Sc2, Sd2 are in the high-frequency modulation state, and the negative-group switch tubes Sa3, Sb3, Sc3, Sd3, Sa4, Sb4, Sc4, Sd4 are all turned on; when the front-stage circuit outputs a negative voltage to the rear-stage, the negative-group switch tubes Sa3, Sb3, Sc3, Sd3, Sa4, Sb4, Sc4, Sd4 are in the high-frequency modulation state, and the positive-group switch tubes Sa1, Sb1, Sc1, Sd1, Sa2, Sb2, Sc2, Sd2 are all turned on; when the front-stage circuit transmits a zero voltage to the rear-stage, all the switch tubes of the rear-stage matrix converter are turned on to enter the freewheeling state.

[0040] A further improvement of the technical solution of the present invention lies in that: under unbalanced operating conditions, the clamping capacitor can be used to absorb the energy of the leakage inductance of the transformer, realizing the ZCS turn-on and ZVS turn-off of the front-stage switch tube, and making the voltage stress across the switch tube be in [0, 2Vdc]. The pulse width modulation method generates zero voltage moments on the primary and secondary sides of the transformer, enabling the ZVS turn-on and ZVS turn-off of the rear-stage switch tube; and because the driving signals of the front-stage switch tubes S1 and S2 are driven with equal width, the primary and secondary side currents of the transformer and the voltage and current across the clamping capacitor are continuous and smooth; while in the single-phase push-pull forward high-frequency link matrix inverter, due to the driving signals of the front-stage switch tubes S1 and S2 changing according to the sine law, the primary and secondary side currents of the transformer and the voltage and current across the clamping capacitor also pulsate sinusoidally at twice the power frequency.

[0041] Due to the adoption of the above technical solution, the technical progress obtained by the present invention is:

[0042] 1. The present invention suppresses the overvoltage stress across all the switch tubes of the front and rear stages, improving the reliability of the circuit.

[0043] 2. The ZVS soft commutation of the matrix converter on the secondary side of the high-frequency transformer can be achieved without relying on the overlapping commutation of the auxiliary circuit and the matrix converter on the secondary side of the high-frequency transformer, and the soft commutation implementation scheme is not limited by the load.

[0044] 3. At any time, only one power transistor in the same arm of the matrix converter on the secondary side of the high-frequency transformer is in the off state, and the power switch transistor is in the on state for half a cycle and in the square-wave modulation state for the other half cycle. Therefore, the present invention can reduce the switching frequency of the power transistor while reducing the switching loss of the power transistor.

[0045] 4. The added fourth arm is only related to the unbalanced current, independent of and interfering with the control of the first three arms, improving the compatibility of the control system. And this control method is simple and easy to implement, can compensate for the influence of the unbalanced load on the circuit, and enables the three-phase four-arm inverter to output three-phase balanced voltage under unbalanced conditions.

[0046] 5. The modulation method of the present invention is simple to control and easy to implement in the process, can effectively improve the overall efficiency and system reliability of the machine, helps the wide promotion of the converter, and is especially suitable for use in fields such as new energy power generation and motor control.

[0047] 6. The present invention can well achieve the functions of electrical isolation between the DC side and the AC side and adjusting the voltage ratio. It not only has a small volume and light weight, but also has bidirectional symmetric magnetization of the high-frequency transformer core, a duty cycle adjustment greater than 0.5, and small input current ripple. Therefore, it is widely used in the occasions of low voltage and large current. BRIEF DESCRIPTION OF THE DRAWINGS

[0048] Figure 1 is the topology diagram of the soft-switching high-frequency link three-phase four-arm matrix inverter in the present invention;

[0049] Figure 2 is the principle waveform diagram of the working state of the inverter in the present invention within a high-frequency period;

[0050] Figure 3 is the voltage-type decoupling schematic diagram of the matrix converter on the secondary side of the high-frequency transformer;

[0051] Figure 4 is the modal circuit diagram of the inverter in the present invention within a high-frequency period. DETAILED DESCRIPTION OF THE INVENTION

[0052] The present invention will be further described in detail below with reference to the drawings and embodiments:

[0053] Such as Figure 1As shown in the figure, a soft-switching high-frequency-link four-leg matrix inverter topology includes a push-pull forward high-frequency inverter structure with a clamping capacitor, a high-frequency transformer, a three-phase four-leg matrix converter (referred to as the matrix converter for short), an LC filter, and a three-phase unbalanced load, which are connected in sequence.

[0054] The push-pull forward high-frequency inverter structure with a clamping capacitor includes a DC input voltage V dc , a controllable switch tube S 1 , a controllable switch tube S 2 , and a capacitor C s .

[0055] The high-frequency transformer T includes a primary side and a secondary side. The primary side includes a coil N 11 and a coil N 12 . The secondary side includes a coil N 2 .

[0056] The three-phase four-leg matrix converter includes controllable switch tubes S a1 , S a2 , S a3 , S a4 , S b1 , S b2 , S b3 , S b4 , S c1 , S c2 , S c3 , S c4 , S d1 , S d2 , S d3 , and S d4 .

[0057] The LC filter includes a first inductor L a , a second inductor L b , a third inductor L c , a fourth inductor L d , a first capacitor C a , a second capacitor C b , a third capacitor C c , a load R a , a load R b , and a load R c .

[0058] The positive pole of the DC input voltage V dc is respectively connected to the collector of the controllable switch tube S 1 , one end of the coil N 12 . The DC input voltage Vdc The negative electrodes of 2 are respectively connected to the emitter of the controllable switch tube S and one end of the coil N 11 ; the emitter of the controllable switch tube S 1 is respectively connected to one end of the clamping capacitor C and the other end of the coil N s ; the collector of the controllable switch tube S 11 is respectively connected to the other end of the clamping capacitor C and the other end of the coil N 2 ; s 12

[0059] One end of the secondary side N of the high-frequency transformer T 2 is respectively connected to the collectors of the controllable switch tubes S a1 , S b1 , S c1 , S d1 , S 2 ; the other end of the secondary side N of the high-frequency transformer T a3 is respectively connected to the collectors of the controllable switch tubes S b3 , S c3 , S d3 , S a1 ; the emitter of the controllable switch tube S a4 is connected to the emitter of the controllable switch tube S b1 , the emitter of the controllable switch tube S b4 is connected to the emitter of the controllable switch tube S c1 , the emitter of the controllable switch tube S c4 is connected to the emitter of the controllable switch tube S d1 , the emitter of the controllable switch tube S d4 is connected to the emitter of the controllable switch tube S a2 ; the emitter of the controllable switch tube S a3 is connected to the emitter of the controllable switch tube S b2 , the emitter of the controllable switch tube S b3 is connected to the emitter of the controllable switch tube S c2 , the emitter of the controllable switch tube S c3 is connected to the emitter of the controllable switch tube S d2 , the emitter of the controllable switch tube S d3 is connected to the emitter of the controllable switch tube S

[0060] The collector of the controllable switch tube S a4 is connected to the collector of the controllable switch tube S a2 and then connected to one end of the first inductor L a ; one end of the first inductor L a ​​The other end is connected to the first capacitor C a One end, the load R a One end is connected, and the load R a The other end is respectively connected to the load R b 、the load R c The other end is connected; the first capacitor C a The other end is respectively connected to the second capacitor C b 、the third capacitor C c The other end is connected; the load R a 、the load R b 、the load R c The neutral point and the first capacitor C a 、the second capacitor C b 、the third capacitor C c The neutral points are connected.

[0061] The collector of the controllable switch tube S b4 is connected to the collector of the controllable switch tube S b2 and then connected to one end of the second inductor L b The other end of the second inductor L b is connected to the second capacitor C b 、the load R b One end is connected.

[0062] The collector of the controllable switch tube S c4 is connected to the collector of the controllable switch tube S c2 and then connected to one end of the third inductor L c The other end of the third inductor L c is connected to the third capacitor C c 、the load R c One end is connected.

[0063] The collector of the controllable switch tube S d4 is connected to the collector of the controllable switch tube S d2 and then connected to one end of the fourth inductor L d The other end of the fourth inductor L d is connected to the first capacitor C a 、the second capacitor C b 、the third capacitor C c 、the load R a 、the load R b 、the load R c The other end is connected.

[0064] A pulse width modulation method for a soft-switching high-frequency link four-leg matrix inverter topology includes an SPWM signal generation method, a decomposition and reconstruction method, and a fourth-leg independent control method.

[0065] The SPWM signal generation method generates six pulse-width modulation signals SPWM1, SPWM2, SPWM3, SPWM4, SPWM5, and SPWM6, which specifically include the following steps:

[0066] A1. Given a balanced three-phase sinusoidal signal of the desired output;

[0067] A2. Comparing the three-phase sinusoidal signal with a sawtooth carrier wave to generate complementary SPWM1 and SPWM2, complementary SPWM3 and SPWM4, and complementary SPWM5 and SPWM6.

[0068] The deconstruction and reconstruction method specifically includes the following steps:

[0069] B1. SPWM2, SPWM4, and SPWM6 generate signal A through "AND" logic; SPWM2, SPWM4, and SPWM6 generate signal B through "OR" logic;

[0070] B2. Signal A is divided by two to obtain V 1 and inverted to generate At the same time, signal B is also divided by two and inverted to generate V 2 、

[0071] B3. V 1 and perform "NAND" logic to obtain the intermediate signal V m , V 2 and perform "NAND" logic to obtain the intermediate signal V p ;

[0072] B4. Inverting V m to obtain the drive signal of the front-stage controllable switch tube S 1 and inverting V p to obtain the drive signal of the front-stage controllable switch tube S 2 ;

[0073] B5. The intermediate signal V p and the six SPWM modulation signals perform "OR" logic to generate the drive signals of the first three-bridge arm switch tubes S a1 , S b1 , S c1 , S a2 , S b2 , S c2 , and the intermediate signal V m and the six SPWM modulation signals perform "OR" logic to generate the drive signals of the first three-bridge arm switch tubes S a3 , S b3 , S c3 , S a4 , S b4 , Sc4 Drive signal

[0074] The control of the fourth bridge arm of the three-phase four-leg matrix converter is independent of the first three bridge arms. The specific control method of the fourth bridge arm includes the following steps:

[0075] C1. Sample the phase voltages of the three-phase unbalanced load, sum the three-phase voltages and multiply by the amplification factor to obtain the modulation signal of the fourth bridge arm;

[0076] C2. Compare the modulation signal of the fourth bridge arm with the carrier wave to generate complementary modulation signals PWM7 and PWM8;

[0077] C3. The intermediate signal V p and the PWM7 and PWM8 signals are respectively subjected to "OR" logic to generate the drive signals of the switching tubes S d1 and S d2 of the fourth bridge arm. The intermediate signal V m and the PWM7 and PWM8 signals are respectively subjected to "OR" logic to generate the drive signals of the switching tubes S d3 and S d4 of the fourth bridge arm.

[0078] The paper "High-frequency Link Three-phase Four-leg Matrix Inverter and Its Arm Control" proposes a topology of a high-frequency link three-phase four-leg matrix inverter. The front stage adopts an H-bridge high-frequency inverter structure, while the topology of the soft-switching high-frequency link four-leg matrix inverter proposed in the present invention adopts a push-pull forward high-frequency inverter structure. Compared with the H-bridge high-frequency inverter structure, the number of switching tubes used in the push-pull forward high-frequency inverter structure is reduced by half. And in the fourth bridge arm control method proposed in the comparative document, the sampling and control of the inductor current of the fourth bridge arm are added, which slightly increases the complexity of the control, while in this paper, only the output three-phase phase voltages are sampled and controlled, and the control method is relatively simple to implement.

[0079] The pulse modulation method can make the bridge arm composed of the controllable switching tubes S 1 and S 2 conduct alternately, and there is a dead zone between the drives of the controllable switching tubes S 1 and S 2 ; when the controllable switching tube S 1 conducts and the controllable switching tube S 2 turns off, the front-stage circuit outputs a positive voltage to the rear stage; when the controllable switching tube S 2 conducts and the controllable switching tube S 1 turns off, the front-stage circuit outputs a negative voltage to the rear stage; when the controllable switching tubes S 1 and S 2When all are turned off, the front-stage circuit outputs zero voltage to the rear stage. The control of the fourth bridge arm of the matrix converter is independent of the first three bridge arms, providing a current path when the entire system is connected to an unbalanced load, causing unbalanced current, so as to output balanced three-phase voltage.

[0080] The driving signals of the front-stage switching tubes are driven with approximately equal widths. In the invention patent "A Modulation Method for a Single-Phase Push-Pull Forward High-Frequency Link Matrix Inverter Topology" (CN201710821717.6), the driving signals of the front-stage switching tubes change according to a sine law. On the one hand, this can reduce the high-frequency transformer bias magnetic problem and the loss of the clamping capacitor caused by inconsistent conduction times of the switching tubes in adjacent cycles. On the other hand, the overall efficiency can be improved by optimizing the dead time to reduce the charging process of the clamping capacitor; and this modulation method combined with the topology can achieve soft switching of the switching tubes and suppress the voltage stress across the switching tubes between 0 and 2V dc The three-phase four-bridge-arm matrix converter is decomposed into positive and negative two groups of voltage-source inverters. When the front-stage circuit outputs a positive voltage to the rear stage, the positive-group switching tubes S a1 、S b1 、S c1 、S d1 、S a2 、S b2 、S c2 、S d2 are in the high-frequency modulation state, and the negative-group switching tubes S a3 、S b3 、S c3 、S d3 、S a4 、S b4 、S c4 、S d4 are all turned on; when the front-stage circuit outputs a negative voltage to the rear stage, the negative-group switching tubes S a3 、S b3 、S c3 、S d3 、S a4 、S b4 、S c4 、S d4 are in the high-frequency modulation state, and the positive-group switching tubes S a1 、S b1 、S c1 、S d1 、S a2 、S b2 、S c2 、S d2 are all turned on; when the front-stage circuit transmits zero voltage to the rear stage, all the switching tubes of the rear-stage matrix converter are turned on and enter the freewheeling state, and ZVS of the rear-stage switching tubes can be achieved.

[0081] Working process:

[0082] The primary push - pull forward high - frequency inverter of the high - frequency transformer uses two complementary signals with built - in dead - time to modulate the input DC voltage into a bipolar three - state high - frequency AC voltage wave. The matrix converter on the secondary side of the high - frequency transformer is equivalently decomposed, so as to convert the high - frequency AC voltage wave transmitted by the high - frequency transformer into a unipolar SPWM wave. The on - off of the switching tubes in the matrix converter on the secondary side of the high - frequency transformer are all completed during the period when the voltage of the high - frequency transformer is zero. Therefore, zero - voltage switching (ZVS) of the switching tubes can be achieved, and natural commutation of the leakage inductance current of the high - frequency transformer and the filter inductance current can be realized.

[0083] Figure 2 This is the waveform diagram of the working state principle within a high - frequency period of the present invention. In the figure, S 1 and S 2 are the drive signals of the switching tubes of the high - frequency inverter at the front stage of the high - frequency transformer, and S a1 ~S a4 、S b1 ~S b4 、S c1 ~S c4 、S d1 ~S d4 are the drive signals of the switching tubes of the matrix converter at the rear stage of the high - frequency transformer. u N11 、u N12 are the voltage waveforms at both ends of the primary windings N 11 and N 12 of the high - frequency transformer, and i N11 、i N12 are the currents flowing through the primary windings N 11 and N 12 of the high - frequency transformer. u CS is the voltage across the clamping capacitor C S , and i Lf is the current flowing through the filter inductance L a 、L b or L c . It can be seen from Figure 2 that the front - stage switching tubes S 1 and S 2 conduct alternately, and the modulation method has a built - in dead - time; at the same time, within the dead - time of the front - stage, the primary voltage of the high - frequency transformer is zero, providing the switching time for the positive and negative groups of the switching tubes at the rear - stage, which can avoid the over - voltage spikes caused by interrupting the flow path of the leakage current of the high - frequency transformer.

[0084] Figure 3 is the circuit equivalent decomposition schematic diagram of the matrix converter at the rear stage of the high - frequency transformer. This modulation method decomposes the matrix converter into two ordinary voltage - type inverters. When the input voltage of the high - frequency transformer is positive, S a1 、Sa2 , S b1 , S b2 , S c1 , S c2 , S d1 , S d2 is in the high-frequency modulation state, and S of the negative-group inverter a3 , S a4 , S b3 , S b4 , S c3 , S c4 , S d3 , S d4 is in the conducting state; when the input current signal of the high-frequency transformer is negative, S of the negative-group inverter a3 , S a4 , S b3 , S b4 , S c3 , S c4 , S d3 , S d4 is in the high-frequency modulation state, and S of the positive-group inverter a1 , S a2 , S b1 , S b2 , S c1 , S c2 , S d1 , S d2 is in the conducting state.

[0085] Figure 4 This is the modal circuit diagram of the soft-switching high-frequency-link four-leg matrix inverter of the present invention within a high-frequency cycle. Figures (a)-(j) are the following operating modes 1-10 respectively. Assume that all components in the topology are ideal components, and the inverter operates under unbalanced conditions (A-phase load > B-phase load = C-phase load). According to the working principle, there are 10 operating states within a high-frequency cycle. The specific modal analysis is as follows:

[0086] (1) Operating mode 1 [t 0 -t 1 , t 0 Before the moment, S 1 , S 2 are not conducting, and the leakage inductance of the primary side of the high-frequency transformer and the clamping capacitor resonate. The input voltage V dc charges the clamping capacitor through the loop V dc -N 12 -C S -N 11 -V dc . The filter inductors L a , L b and L c freewheel through the matrix converter. At t0 At this moment, the pre-stage switch tube S 1 conducts, and the current flows as shown in Figure 4 (a). Since no current flowed through the switch tube S before time t 0 , the switch tube S is turned on with zero current. V 1 is applied across the coil N through the loop V 1 -S dc -N dc -V 1 -N 11 -V dc is applied across the coil N 11 , and the current in the coil N 11 increases rapidly. The high-frequency transformer starts to transfer energy to the post-stage; V CS is applied across the coil N through the loop C S -N 12 -S 1 -C S is applied across the coil N 12 , C S is charged and resonates with the leakage inductance of the coil N 12 . The current in the coil N 12 decreases rapidly. The positive-group inverter in the matrix converter operates, and the switch tubes in the negative-group inverter are all in the conducting state. The currents in the filter inductors L a , L b and L c start to rise linearly.

[0087] (2) Operating mode 2 [t 1 -t 2 , at time t 1 , the converter starts to operate stably, and the current in the coil N 12 drops to 0. At steady state, the current flows as shown in Figure 4 (b). Both currents are input at the same-named ends. V dc is connected in parallel across the coil N 11 , and V CS is connected in parallel across the coil N 12 , which is equivalent to the parallel connection of two single-ended forward circuits. In this stage, the voltage stress on S 2 is the largest, which is V dc +V CS ≈2V dc . The operating state of the matrix converter remains unchanged.

[0088] (3) Operating mode 3 [t 2 -t 3 , at time t 2 , the pre-stage still maintains the previous operating state, and the switch tubes of the post-stage matrix converter act. S c1 changes from the conducting state to the off state, S c2From the off state to the on state, S d1 From the on state to the off state, S d2 From the off state to the on state, due to the existence of the parasitic capacitance of the switching tube, S c1 、S d1 Is zero-voltage turn-off. The current flow direction of phase C changes from input to output, as Figure 4 (c) shown.

[0089] (4) Working mode 4 [t 3 -t 4 , t 3 At this moment, all the front-stage power switching tubes are turned off and enter the dead time. Due to the existence of the parasitic capacitance of the switching tube, the switching tube S 1 Is zero-voltage turn-off. As Figure 4 (d) shown, since V dc Is directly applied to N 12 On, so the current flowing through the coil N 12 Rapidly decreases through zero and reverses and increases. At this time, the current flowing through the coil N 11 Is greater than the current flowing through the coil N 12 . According to Kirchhoff's current law, the anti-parallel diode of the power switch S 2 Naturally conducts. The anti-parallel diodes of N 11 、S 2 And C S Form a freewheeling circuit. The leakage inductance on N 11 Resonates with C S . C S Is in the charging state. At the same time, since the voltage of S 1 Is clamped at V dc +V CS , thus reducing the overvoltage stress on S 1 .

[0090] Similarly, since the current flowing through the coil N 12 Is less than the current flowing through the coil N 11 , and the current flowing through the coil N 11 Flows in from the same-name terminal all the way, and the current flowing through the coil N 12 Flows in from the opposite-name terminal. The high-frequency transformer still has energy transferred to the rear stage, but the transferred energy gradually decreases until t 4 At this moment, there is no energy transfer. All the rear-stage switching tubes are turned on during this process and enter the all-“1” state. The filter inductor current starts to linearly decrease and enters the freewheeling state. Since all the switching tubes in the rear-stage matrix converter are in the on state during this stage, the current path of the leakage inductance is not interrupted. Therefore, the voltage overshoot problem existing in the decoupling strategy is solved.

[0091] (5) Working mode 5 [t 4-t 5 , t 4 At this moment, a circulating current is formed in the loop composed of the primary side current of the high-frequency transformer in V dc -N 12 -C S -N 11 -V dc as shown in Figure 4 (e). At t 4 At this moment, since the currents flowing through the two coils on the primary side flow in from the same-named ends and the other flows in from the different-named ends, and the two are equal, no energy is transferred to the secondary side, and the voltages of the primary and secondary sides of the high-frequency transformer are zero, and the output filter inductor remains in the freewheeling state.

[0092] (6) Operating mode 6 [t 5 -t 6 , t 5 At this moment, the pre-stage switch tube S 2 turns on, and the current flow is as shown in Figure 4 (f). V dc is applied to the coil N dc -N 12 -S 2 -V dc and the current on the coil N 12 increases rapidly. The high-frequency transformer starts to transfer energy to the post-stage; V 12 is applied to the coil N CS through the loop C S -N 11 -S 2 -C S and C 11 is charged and resonates with the leakage inductance on the coil N S . The current on the coil N 11 decreases rapidly. The negative-group inverter in the matrix converter operates, and the switch tubes in the positive-group inverter are all in the on state. The filter inductor currents L 11 and L a and L b and L c start to rise linearly.

[0093] (7) Operating mode 7 [t 6 -t 7 , t 6 At this moment, the converter starts to work stably, and the current on the coil N 11 drops to 0. The current flow at steady state is as shown in Figure 4 (g). The two currents are both output from the same-named ends. V dc is connected in parallel across the coil N 12 ends, and V CS is connected in parallel across the coil N 11At both ends, it is equivalent to the parallel connection of two single-ended forward circuits. During this stage, S 1 undergoes the maximum voltage stress, which is V dc +V CS ≈2V dc . The operating state of the matrix converter remains unchanged.

[0094] (8) Operating mode 8 [t 7 -t 8 , at time t 7 , the front stage still maintains the previous operating state, and the switching tubes of the rear-stage matrix converter act. S c4 changes from the off state to the on state, S c3 changes from the on state to the off state, S d4 changes from the off state to the on state, S d3 changes from the on state to the off state. Due to the existence of the parasitic capacitance of the switching tube, S c3 and S d3 are turned off under zero voltage. The current flow direction of phase C changes from input to output, as shown in Figure 4 (h).

[0095] (9) Operating mode 9 [t 8 -t 9 , at time t 8 , all the front-stage power switching tubes are turned off, and the dead time is entered. As shown in Figure 4 (i), at this time, the current flowing through coil N 11 rapidly decreases to zero and then increases in the reverse direction. The current flowing through coil N 12 is greater than the current flowing through coil N 11 . According to Kirchhoff's current law, the anti-parallel diode of power switch S 1 conducts naturally. The anti-parallel diodes of N 12 and S 1 and C S form a freewheeling circuit. The leakage inductance on N 12 resonates with C S , and C S is in the charging state. At the same time, since the voltage of S 2 is clamped at V dc +V CS , the overvoltage stress on S 1 is thus reduced. Similarly, all the rear-stage switching tubes are turned on during this process and enter the all-“1” state. The filter inductor current starts to decrease linearly and enters the freewheeling state.

[0096] (10) Operating mode 10 [t 9 -t 10 , at time t 9 , the primary side current of the high-frequency transformer is at V dc-N 12 -C S -N 11 -V dc A circulating current is formed in the circuit composed of, as Figure 4 (j) shown. At time t 9 , since the currents flowing through the two primary-side coils flow into the same-named terminals in one path and into the opposite-named terminals in the other path, and the two are equal, no energy is transferred to the secondary side, and the voltages of the primary and secondary sides of the high-frequency transformer are zero. The output filter inductor remains in the freewheeling state.

[0097] It can be seen from the above working process that the overvoltage stress on the switching tubes in the push-pull forward-type high-frequency link four-leg matrix inverter is significantly reduced, the circuit operates stably, and a three-phase balanced output voltage can be provided when the three-phase load is unbalanced.

[0098] In summary, the present invention provides a soft-switching push-pull forward-type high-frequency link four-leg matrix inverter topology scheme with few power conversion levels and simple modulation, and a deconstruction and reconstruction type pulse width modulation method matched therewith, which can achieve soft switching of all switching tubes in the topology, reduce the common-mode voltage of the inverter output, reduce the switching tube loss, improve the efficiency of the converter, balance the three-phase output voltages under unbalanced operating conditions by adding a fourth leg, and make the modulation process simpler, making the transformer easier to promote and use.

Claims

1. A pulse width modulation method for a soft-switching high-frequency link four-leg matrix inverter topology, characterized in that: The soft-switching high-frequency link four-leg matrix inverter topology includes a push-pull forward high-frequency inverter structure with a clamping capacitor, a high-frequency transformer, a three-phase four-leg matrix converter, an output LC filter, and a three-phase unbalanced load connected in sequence; The push-pull forward high-frequency inverter structure with a clamping capacitor includes a DC input voltage V dc , a controllable switch tube S 1 , a controllable switch tube S 2 and a capacitor C s ; The high-frequency transformer T includes a primary side and a secondary side. The primary side includes coil N 11 and coil N 12 , and the secondary side includes coil N 2 ; The three-phase four-leg matrix converter includes a first leg, a second leg, a third leg, and a fourth leg for dealing with unbalanced conditions; The first bridge arm includes controllable switching transistors S a1 , controllable switching transistors S a4 , controllable switching transistors S a2 and controllable switching transistors S a3 . The second bridge arm includes controllable switching transistors S b1 , controllable switching transistors S b4 , controllable switching transistors S b2 and controllable switching transistors S b3 . The third bridge arm includes controllable switching transistors S c1 , controllable switching transistors S c4 , controllable switching transistors S c2 and controllable switching transistors S c3 . The fourth bridge arm includes controllable switching transistors S d1 , controllable switching transistors S d4 , controllable switching transistors S d2 and controllable switching transistors S d3 ; The LC filter includes a first inductor L a , a second inductor L b , a third inductor L c , a fourth inductor L d , a first capacitor C a , a second capacitor C b and a third capacitor C c ; The three-phase unbalanced load includes load R a , load R b and load R c , and the load parameters are different; The pulse width modulation method for the soft-switching high-frequency link four-leg matrix inverter topology includes an SPWM signal generation method, a decomposition and reconstruction method, and a fourth leg independent control method; The SPWM signal generation method generates 6 pulse width modulation signals SPWM1, SPWM2, SPWM3, SPWM4, SPWM5, SPWM6, and specifically includes the following steps: A1. Given a balanced three-phase sinusoidal signal of the desired output; A2. Comparing the three-phase sinusoidal signal with a sawtooth carrier to generate complementary SPWM1 and SPWM2, complementary SPWM3 and SPWM4, and complementary SPWM5 and SPWM6; The decomposition and reconstruction method specifically includes the following steps: B1. SPWM2, SPWM4, SPWM6 generate signal A through "AND" logic; SPWM2, SPWM4, SPWM6 generate signal B through "OR" logic; The signal A is divided by two to obtain V 1 and inverted to generate At the same time, the signal B is also divided by two and inverted to generate V 2 , B3, V 1 and perform a NAND logic operation to obtain an intermediate signal V m , V 2 and perform a NAND logic operation to obtain an intermediate signal V p ; B4. Invert V m to obtain the drive signal of the pre-stage controllable switch tube S 1 and invert V p to obtain the drive signal of the pre-stage controllable switch tube S 2 ; B5, Intermediate signal V p and 6-channel SPWM modulation signals perform an "OR" logic operation to generate the drive signals for the first three bridge arm switching transistors S a1 , S b1 , S c1 , S a2 , S b2 , S c2 . The intermediate signal V m and 6-channel SPWM modulation signals perform an "OR" logic operation to generate the drive signals for the first three bridge arm switching transistors S a3 , S b3 , S c3 , S a4 , S b4 , S c4 .

2. The pulse width modulation method for a soft-switching high-frequency link four-leg matrix inverter topology according to claim 1, characterized in that: DC input voltage V dc The positive poles are respectively connected to the collector of the controllable switch tube S 1 and one end of the coil N 12 ; the negative poles of the DC input voltage V dc are respectively connected to the emitter of the controllable switch tube S 2 and one end of the coil N 11 ; the emitter of the controllable switch tube S 1 is respectively connected to one end of the clamping capacitor C s and the other end of the coil N 11 ; the collector of the controllable switch tube S 2 is respectively connected to the other end of the clamping capacitor C s and the other end of the coil N 12 ; The secondary side N of the high-frequency transformer T 2 One end of each is connected to the collector of the controllable switch tube S a1 The collector of the controllable switch tube S b1 The collector of the controllable switch tube S c1 The collector of the controllable switch tube S d1 The collector is connected. The other end of the secondary side N of the high-frequency transformer T 2 One end of each is connected to the collector of the controllable switch tube S a3 The collector of the controllable switch tube S b3 The collector of the controllable switch tube S c3 The collector of the controllable switch tube S d3 The collector is connected. The emitter of the controllable switch tube S a1 The emitter is connected to the emitter of the controllable switch tube S a4 The emitter of the controllable switch tube S b1 The emitter is connected to the emitter of the controllable switch tube S b4 The emitter of the controllable switch tube S c1 The emitter is connected to the emitter of the controllable switch tube S c4 The emitter of the controllable switch tube S d1 The emitter is connected to the emitter of the controllable switch tube S d4 The emitter is connected; Controllable switch tube S a2 The emitter of which is connected to the emitter of the controllable switch tube S a3 The emitter of the controllable switch tube S b2 The emitter of which is connected to the emitter of the controllable switch tube S b3 The emitter of the controllable switch tube S c2 The emitter of which is connected to the emitter of the controllable switch tube S c3 The emitter of the controllable switch tube S d2 The emitter of which is connected to the emitter of the controllable switch tube S d3 The emitter of which is connected to the emitter of the controllable switch tube S; Controllable switch tube S a4 The collector of the controllable switch tube S a2 is connected to the collector of the controllable switch tube S a and then connected to one end of the first inductor L a The other end of the first inductor L a is connected to one end of the first capacitor C a and one end of the load R a The other end of the load R b is respectively connected to the load R c and the other end of the load R a The other end of the first capacitor C b is respectively connected to the other end of the second capacitor C c and the other end of the third capacitor C a The load R b and the load R c neutral points and the first capacitor C a and the second capacitor C b and the third capacitor C c neutral points are connected; Controllable switch tube S b4 The collector of the controllable switch tube S b2 is connected to the collector of the controllable switch tube S b and then connected to one end of the second inductor L b The other end of the second inductor L b is connected to one end of the second capacitor C b and the load R Controllable switch tube S c4 The collector of the controllable switch tube S c2 is connected to the collector of the controllable switch tube S c and then connected to one end of the third inductor L c The other end of the third inductor L c is connected to one end of the third capacitor C c and the load R Controllable switch tube S d4 The collector of which is connected to the collector of the controllable switch tube S d2 and then connected to one end of the fourth inductor L d The other end of the fourth inductor L d is connected to the other ends of the first capacitor C a , the second capacitor C b , the third capacitor C c , the load R a , the load R b , the load R c and the other ends are connected together.

3. The pulse width modulation method for a soft-switching high-frequency link four-leg matrix inverter topology according to claim 1, characterized in that: The control of the fourth leg of the three-phase four-leg matrix converter is independent of the first three legs. The fourth leg independent control method specifically includes the following steps: C1. Sampling the phase voltages of the three-phase unbalanced load, summing the three-phase phase voltages and multiplying by an amplification factor to obtain the modulation signal of the fourth leg; C2. Comparing the modulation signal of the fourth leg with the carrier to generate complementary modulation signals PWM7, PWM8; C3, Intermediate signal V p respectively perform "OR" logic with PWM7 and PWM8 signals to generate the drive signals for the fourth leg switching transistors S d1 , S d2 . The intermediate signal V m respectively perform "OR" logic with PWM7 and PWM8 signals to generate the drive signals for the fourth leg switching transistors S d3 , S d4 .

4. The pulse width modulation method for a soft-switching high-frequency link four-leg matrix inverter topology according to claim 1, characterized in that: Under the control of the pulse width modulation method, the controllable switch tube S 1 and the controllable switch tube S 2 conduct alternately, and there is a built-in dead zone between the drives of the controllable switch tube S 1 and the controllable switch tube S 2 ; when the controllable switch tube S 1 conducts and the controllable switch tube S 2 turns off, the front-stage circuit outputs a positive voltage to the rear-stage; when the controllable switch tube S 2 conducts and the controllable switch tube S 1 turns off, the front-stage circuit outputs a negative voltage to the rear-stage; when the controllable switch tube S 1 and the controllable switch tube S 2 are both turned off, the front-stage circuit outputs a zero voltage to the rear-stage.

5. The pulse width modulation method for a soft-switching high-frequency link four-leg matrix inverter topology according to claim 1, characterized in that: The driving signal characteristics of the front-stage switching tubes are different from those of the front-stage switching tubes under the modulation method of a single-phase push-pull forward high-frequency link matrix converter.

6. The pulse width modulation method for a soft-switching high-frequency link four-leg matrix inverter topology according to claim 1, characterized in that: When the current - stage circuit outputs a positive voltage to the subsequent stage, the positive - group switching transistors S a1 、S b1 、S c1 、S d1 、S a2 、S b2 、S c2 、S d2 are in the high - frequency modulation state, and the negative - group switching transistors S a3 、S b3 、S c3 、S d3 、S a4 、S b4 、S c4 、S d4 are all turned on; when the current - stage circuit outputs a negative voltage to the subsequent stage, the negative - group switching transistors S a3 、S b3 、S c3 、S d3 、S a4 、S b4 、S c4 、S d4 are in the high - frequency modulation state, and the positive - group switching transistors S a1 、S b1 、S c1 、S d1 、S a2 、S b2 、S c2 、S d2 are all turned on; when the current - stage circuit transmits a zero voltage to the subsequent stage, all the switching transistors of the subsequent - stage matrix converter are turned on to enter the free - wheeling state.

7. The pulse width modulation method for a soft-switching high-frequency link four-leg matrix inverter topology according to claim 1, characterized in that: Under unbalanced operating conditions, the clamping capacitor can be used to absorb the energy of the transformer leakage inductance, realizing ZCS turn-on and ZVS turn-off of the front-stage switch tube, and keeping the voltage stress across the switch tube within [0, 2Vdc]. The pulse-width modulation method generates zero-voltage moments on the primary and secondary sides of the transformer, enabling ZVS turn-on and ZVS turn-off of the rear-stage switch tube. Moreover, due to the front-stage switch tubes S 1 and S 2 being driven with equal-width driving signals, the primary and secondary currents of the transformer and the voltage and current across the clamping capacitor are continuous and smooth. However, in a single-phase push-pull forward high-frequency link matrix inverter, since the driving signals of the front-stage switch tubes S 1 and S 2 vary according to a sine law, the primary and secondary currents of the transformer and the voltage and current across the clamping capacitor also pulsate sinusoidally at twice the power frequency.

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