A motor driving method, apparatus, motor system, and storage medium

Abstract

The application discloses a motor driving method and device, a motor system and a storage medium. The method comprises the following steps: obtaining three-phase current values, three-phase voltage values and a rotating speed of a motor after starting the motor; performing Clark transformation on the three-phase voltage values of the motor to obtain measured values of alpha and beta axis stator voltage components; determining command values of the alpha and beta axis stator voltage components of the motor according to the three-phase current values, the rotating speed and the measured values of the alpha and beta axis stator voltage components; determining alpha and beta axis voltage values of the motor by using a preset quasi-proportional resonant controller based on the measured values of the alpha and beta axis stator voltage components and the command values of the alpha and beta axis stator voltage components; and driving the motor to operate after the alpha and beta axis voltage values of the motor are subjected to space vector pulse width modulation and then are subjected to inversion by an inverter. According to the scheme, the current harmonic existing at present is suppressed by using the quasi-proportional resonant controller, and the operation performance of the motor is improved.

Description

TECHNICAL FIELD

[0001] The present application belongs to the technical field of electric machines, and particularly relates to an electric machine driving method, device, electric machine system and storage medium, in particular to an electric machine driving method, device, electric machine system and storage medium with current harmonic suppression function. BACKGROUND

[0002] Due to the manufacturing process of the electric machine itself, air gap magnetic field distortion and dead zone problem of the power electronic device itself, the non-linear problem exists in the magnetic potential of the electric machine itself, the magnetic circuit, and the power supply and load connected with the electric machine, thereby causing the harmonic torque ripple of the electric machine in the running process. Due to the harmonic torque ripple of the output torque of the electric machine in the running process, the smoothness of the output torque of the electric machine is affected, and the operation performance of the electric machine required by many high-precision industries cannot be met. For example, permanent magnet synchronous motors have been used in various industries in current industrial production, and the torque ripple problem existing in the operation process brings many adverse effects to many high-precision production equipment, and cannot meet the requirements of these industries on the control performance of the electric machine.

[0003] The above content is only used to assist in understanding the technical solutions of the present application, and does not represent the acknowledgement of the above content as prior art. SUMMARY

[0004] The purpose of the present application is to provide an electric machine driving method, device, electric machine system and storage medium, to solve the problem that the electric machine (such as a permanent magnet synchronous motor) will generate harmonic torque ripple in the running process, affect the operation performance of the electric machine, and achieve the effect of suppressing the current harmonic existing at present by using a proportional resonant controller, which is beneficial to improve the operation performance of the electric machine.

[0005] The present application provides an electric machine driving method, comprising: obtaining three-phase current values of the electric machine, obtaining three-phase voltage values of the electric machine, and obtaining a rotating speed of the electric machine after starting the electric machine; performing Clark transformation on the three-phase voltage values of the electric machine to obtain measurement values of alpha and beta axis stator voltage components of the electric machine; determining command values of the alpha and beta axis stator voltage components of the electric machine according to the three-phase current values of the electric machine, the rotating speed of the electric machine, and the measurement values of the alpha and beta axis stator voltage components of the electric machine; determining alpha and beta axis voltage values of the electric machine by using a preset proportional resonant controller based on the measurement values of the alpha and beta axis stator voltage components of the electric machine and the command values of the alpha and beta axis stator voltage components of the electric machine; and driving the electric machine to run after the alpha and beta axis voltage values of the electric machine are subjected to space vector pulse width modulation and then are inverted by an inverter.

[0006] In some embodiments, determining the command value of the stator voltage component of the motor in the α-axis and the β-axis of the motor according to the three-phase current value of the motor, the rotating speed of the motor, and the measured value of the stator voltage component of the motor in the α-axis and the β-axis comprises: performing Clark transformation on the three-phase current value of the motor to obtain the measured value of the stator current component of the motor in the α-axis and the β-axis; observing the stator flux of the motor and the angle between the d-axis and the α-axis of the motor in the synchronous rotating coordinate system by a flux observer based on the measured value of the stator voltage component of the motor in the α-axis and the β-axis and the measured value of the stator current component of the motor in the α-axis and the β-axis; performing Park transformation based on the measured value of the stator current component of the motor in the α-axis and the β-axis and the angle between the d-axis and the α-axis of the motor in the synchronous rotating coordinate system to obtain the measured value of the stator current component of the motor in the d-axis and the q-axis; and performing Park inverse transformation after PI operation in the speed loop and the flux outer loop based on the rotating speed of the motor, the measured value of the stator current component of the motor in the d-axis, the stator flux of the motor, and the angle between the d-axis and the α-axis of the motor in the synchronous rotating coordinate system to obtain the command value of the stator voltage component of the motor in the α-axis and the β-axis.

[0007] In some embodiments, the method further comprises: analyzing the amplitude variation and the resonance frequency of the q-axis stator current of the motor based on the measured value of the q-axis stator current component of the motor, determining the current harmonic condition of the motor, and displaying the determined current harmonic condition of the motor.

[0008] In some embodiments, determining the voltage value of the motor in the α-axis and the β-axis based on the measured value of the stator voltage component of the motor in the α-axis and the β-axis and the command value of the stator voltage component of the motor in the α-axis and the β-axis comprises: inputting the difference between the command value of the stator voltage component of the motor in the α-axis and the measured value of the stator voltage component of the motor in the α-axis into a preset PR controller to obtain the voltage value of the motor in the α-axis; inputting the difference between the command value of the stator voltage component of the motor in the β-axis and the measured value of the stator voltage component of the motor in the β-axis into another preset PR controller to obtain the voltage value of the motor in the β-axis; and obtaining the voltage value of the motor in the α-axis and the β-axis based on the voltage value of the motor in the α-axis and the voltage value of the motor in the β-axis.

[0009] In some embodiments, the preset one PR controller and the preset another PR controller comprise a PR main controller, an odd harmonic suppressor and an even PR controller; the transfer function of the PR main controller is:

[0010]

[0011] wherein k p is a proportional link coefficient, k r is a resonance link coefficient, ω0 is an angular frequency of a sine signal, i.e., a resonance angular frequency, and ω c is a cut-off angular frequency.

[0012] The transfer function of the odd harmonic suppressor is:

[0013]

[0014] wherein k r is an odd resonance link coefficient, ω0 is an angular frequency of an odd sine signal, i.e., a resonance angular frequency, and ω c is an odd cut-off angular frequency.

[0015] The transfer function of the even PR controller is:

[0016]

[0017] wherein k r6 and k r12 are 6th and 12th resonance link coefficients, ω0 is an angular frequency of an even sine signal, i.e., a resonance angular frequency, and ω c is an even cut-off angular frequency.

[0018] To match the above method, the application further provides a motor driving device, comprising: an acquisition unit configured to acquire three-phase current values of the motor after the motor is started, acquire three-phase voltage values of the motor, and acquire a rotating speed of the motor; an operation unit configured to perform Clark transformation on the three-phase voltage values of the motor to obtain measured values of α and β axis stator voltage components of the motor; the operation unit is further configured to calculate the three-phase current values of the motor, the rotating speed of the motor, and the measured values of the α and β axis stator voltage components of the motor u sα and u sβdetermining an instruction value of the stator voltage component of the motor in the α-axis and the β-axis; the operation unit is further configured to determine the voltage value of the motor in the α-axis and the β-axis by using a preset PR controller based on the measured value of the stator voltage component of the motor in the α-axis and the β-axis and the instruction value of the stator voltage component of the motor in the α-axis and the β-axis; and the operation unit is further configured to drive the motor to operate after the voltage value of the motor in the α-axis and the β-axis is subjected to space vector pulse width modulation and then subjected to inversion by an inverter.

[0019] In some embodiments, the operation unit determines the instruction value of the stator voltage component of the motor in the α-axis and the β-axis according to the three-phase current value of the motor, the rotating speed of the motor, and the measured value of the stator voltage component of the motor in the α-axis and the β-axis, including: performing Clark transformation on the three-phase current value of the motor to obtain the measured value of the stator current component of the motor in the α-axis and the β-axis; observing the stator flux of the motor and the angle between the d-axis and the α-axis of the motor in the synchronous rotating coordinate system by a flux observer based on the measured value of the stator voltage component of the motor in the α-axis and the β-axis and the measured value of the stator current component of the motor in the α-axis and the β-axis; performing Park transformation based on the measured value of the stator current component of the motor in the α-axis and the β-axis and the angle between the d-axis and the α-axis of the motor in the synchronous rotating coordinate system to obtain the measured value of the stator current component of the motor in the d-axis and the q-axis; and performing PI operation in the speed loop and the flux outer loop respectively and then performing Park inverse transformation based on the rotating speed of the motor, the measured value of the stator current component of the motor in the d-axis, the stator flux of the motor, and the angle between the d-axis and the α-axis of the motor in the synchronous rotating coordinate system to obtain the instruction value of the stator voltage component of the motor in the α-axis and the β-axis.

[0020] In some embodiments, the operation unit is further configured to analyze the amplitude variation and the resonance frequency of the q-axis stator current of the motor based on the measured value of the q-axis stator current component of the motor, determine the current harmonic condition of the motor, and display the determined current harmonic condition of the motor.

[0021] In some embodiments, the operation unit determines the alpha and beta axis voltage values of the motor based on the measured values of the alpha and beta axis stator voltage components of the motor and the command values of the alpha and beta axis stator voltage components of the motor using a preset PR controller, including: inputting the difference between the command value of the alpha axis stator voltage component in the command values of the alpha and beta axis stator voltage components of the motor and the measured value of the alpha axis stator voltage component of the motor in the measured values of the alpha and beta axis stator voltage components of the motor into a preset PR controller to obtain the alpha axis voltage value of the motor; inputting the difference between the command value of the beta axis stator voltage component in the command values of the alpha and beta axis stator voltage components of the motor and the measured value of the beta axis stator voltage component of the motor in the measured values of the alpha and beta axis stator voltage components of the motor into another preset PR controller to obtain the beta axis voltage value of the motor; and obtaining the alpha and beta axis voltage values of the motor based on the alpha axis voltage value of the motor and the beta axis voltage value of the motor.

[0022] In some embodiments, the PR controller in the preset PR controller and the other preset PR controller includes a PR main controller, an odd harmonic suppressor, and an even PR controller; the transfer function of the PR main controller is:

[0023]

[0024] wherein k p is a proportional link coefficient, k r is a resonance link coefficient, ω0 is the angular frequency of a sine signal, i.e., the resonance angular frequency, and ω c is the cutoff angular frequency.

[0025] The transfer function of the odd harmonic suppressor is:

[0026]

[0027] wherein k r is an odd resonance link coefficient, ω0 is the angular frequency of an odd sine signal, i.e., the resonance angular frequency, and ω c is the odd cutoff angular frequency.

[0028] The transfer function of the even PR controller is:

[0029]

[0030] wherein k r6 and k r12 are the 6th and 12th resonance link coefficients, respectively, ω0 is the angular frequency of an even sine signal, i.e., the resonance angular frequency, and ω ceven order cutoff angular frequency.

[0031] In another aspect of the present application, there is provided a motor system, which is matched with the above motor drive device, and which comprises the above motor drive device.

[0032] In another aspect of the present application, there is provided a storage medium, which is matched with the above method, and which comprises a stored program, wherein the program, when executed, controls a device in which the storage medium is located to perform the above motor drive method.

[0033] Therefore, by analyzing the amplitude variation and the resonance frequency of the quadrature-axis current, the current harmonic characteristics currently existing are obtained, and the current harmonics currently existing are suppressed by using the quasi-proportional-resonant controller, so that the current harmonics can be quickly suppressed, and the oscillation of the motor system can be reduced, which is beneficial to improving the operation performance of the motor.

[0034] Other features and advantages of the present application will be set forth in the following description, and in part will become apparent to those skilled in the art from the description, or can be learned by practice of the application.

[0035] The technical solutions of the present application will be described in further detail below with reference to the accompanying drawings and examples. BRIEF DESCRIPTION OF DRAWINGS

[0036] Figure 1 FIG. 1 is a flowchart of an embodiment of the motor drive method of the present application;

[0037] Figure 2 FIG. 2 is a flowchart of an embodiment of the method of the present application for determining the command values of the alpha-axis and beta-axis stator voltage components of the motor;

[0038] Figure 3 FIG. 3 is a flowchart of an embodiment of the method of the present application for determining the alpha-axis and beta-axis voltage values of the motor by using the preset quasi-proportional-resonant controller;

[0039] Figure 4 FIG. 4 is a structural schematic diagram of an embodiment of the motor drive device of the present application;

[0040] Figure 5 FIG. 5 is a structural schematic diagram of an embodiment of the current harmonic suppression control system;

[0041] Figure 6 FIG. 6 is a structural schematic diagram of an embodiment of the quasi-proportional-resonant controller;

[0042] Figure 7 FIG. 7 is a schematic diagram of the overall working process of an embodiment of the driving method with the current harmonic suppression function;

[0043] Figure 8 Fig. 1 is a schematic diagram of a harmonic operation interface of an embodiment of a current harmonic suppression control system.

[0044] In the embodiments of the present application, the reference signs are as follows in combination with the drawings:

[0045] 102 - acquisition unit; 104 - operation unit. DETAILED DESCRIPTION

[0046] In order to make the objectives, technical solutions and advantages of the present application clearer, the technical solutions of the present application will be described clearly and completely below in combination with specific embodiments of the present application and corresponding drawings. Obviously, the described embodiments are only some of the embodiments of the present application, rather than all the embodiments. Based on the embodiments in the present application, all other embodiments obtained by those skilled in the art without creative work fall within the protection scope of the present application.

[0047] In some solutions, the predicted current harmonics are calculated in a prediction model through current motor parameters and states, and are brought into a cost function in advance for suppression. The prediction model method is comparative, but for real-time changes in industrial sites, the prediction model method cannot predict and process occasional current harmonics.

[0048] In some other solutions, a filter and a wave trap are cooperated to eliminate current harmonics. However, the current resonance suppression method is aimed at high-frequency harmonics and only processes one kind of harmonics, and it is difficult to process if multiple current harmonics are superimposed.

[0049] The solution of the present application proposes a driving solution with a current harmonic suppression function, which is used to solve the torque ripple problem of a permanent magnet synchronous motor, improve the operation performance of the permanent magnet synchronous motor, and improve the motor speed accuracy.

[0050] According to the embodiments of the present application, a motor driving method is provided, as shown in Figure 1 Fig. 1 is a flowchart of an embodiment of the method of the present application. The motor driving method can include steps S110 to S150.

[0051] At step S110, after the motor is started, the three-phase current values of the motor are acquired, the three-phase voltage values of the motor are acquired, and the rotating speed of the motor is acquired. The three-phase current values of the motor are current values i a , i b , i c of U, V and W phases of the motor, the three-phase voltage values of the motor are voltage values u a , u b , u c of U, V and W phases of the motor, and the rotating speed of the motor is rotating speed ω.

[0052] At step S120, a Clark transformation is performed on the three-phase voltage values of the motor to obtain measured values of the α-axis and β-axis stator voltage components of the motor. The measured values of the α-axis and β-axis stator voltage components of the motor are u sα and u sβ .

[0053] At step S130, based on the three-phase current values of the motor, the rotational speed of the motor, and the measured values of the α-axis and β-axis stator voltage components of the motor u sα and u sβ , command values of the α-axis and β-axis stator voltage components of the motor are determined. The command values of the α-axis and β-axis stator voltage components of the motor are u sa * and u sβ * .

[0054] In some embodiments, the specific process of determining the command values of the α-axis and β-axis stator voltage components of the motor based on the three-phase current values of the motor, the rotational speed of the motor, and the measured values of the α-axis and β-axis stator voltage components of the motor in step S130 is described in the following exemplary description.

[0055] The specific process of determining the command values of the α-axis and β-axis stator voltage components of the motor in step S130 is further described below in conjunction with an embodiment flowchart of determining the command values of the α-axis and β-axis stator voltage components of the motor in the method of the present application shown in Figure 2 .

[0056] At step S210, a Clark transformation is performed on the three-phase current values of the motor to obtain measured values of the α-axis and β-axis stator current components of the motor. The measured values of the α-axis and β-axis stator current components of the motor are i α and i β .

[0057] At step S220, based on the measured values of the α-axis and β-axis stator voltage components of the motor and the measured values of the α-axis and β-axis stator current components of the motor, the stator flux of the motor and the angle between the d-axis and the α-axis of the motor in the synchronous rotating coordinate system are observed by a flux observer. The stator flux of the motor is ψ s , and the angle between the d-axis and the α-axis of the motor in the synchronous rotating coordinate system is θ.

[0058] Step S230, based on the measured value of the motor's alpha, beta axis stator current component, and the angle between the d-axis and the alpha-axis of the motor in the synchronous rotating coordinate system, Park transformation is performed to obtain the measured value of the motor's d, q-axis stator current component. The measured value of the motor's d, q-axis stator current component is i d and i q .

[0059] Step S240, based on the motor speed, the measured value of the motor's d-axis stator current component, the motor stator flux linkage, and the angle between the d-axis and the alpha-axis of the motor in the synchronous rotating coordinate system, PI operation is performed in the speed loop and the flux linkage outer loop respectively, and then Park inverse transformation is performed to obtain the command value of the motor's alpha, beta axis stator voltage component.

[0060] Figure 5 It is a structural schematic diagram of an embodiment of the current harmonic suppression control system. As shown in Figure 5 , in the scheme of the present application, the current harmonic suppression control system (i.e. the current harmonic suppression driving device) is composed of a Clark transformation module, a Park inverse transformation module, a flux linkage observer, and a quasi-proportional resonant controller. The quasi-proportional resonant control system based on the stator flux linkage is composed of three loops, which are the speed loop, the flux linkage outer loop, and the stator voltage inner loop. In the example shown in Figure 5 , the current loop adopts a closed-loop control scheme, and needs to obtain accurate current feedback value, voltage feedback value, and flux linkage feedback value, and its current harmonic suppression effect is in the stator voltage inner loop.

[0061] Figure 6 It is a structural schematic diagram of an embodiment of the quasi-proportional resonant controller. Figure 5 In the example shown in Figure 6 , the specific structure of the quasi-proportional resonant controller can be referred to the example shown in

[0062] Figure 7 It is a whole working flow schematic diagram of an embodiment of a driving method with current harmonic suppression function. Referring to the examples shown in Figure 5 , Figure 6 and Figure 7 , the driving method with current harmonic suppression function provided by the scheme of the present application comprises:

[0063] Step 1, after the motor is started, when there is current harmonic, the U, V, W phases of the motor are sampled to obtain the corresponding current values i a , i b , i c and voltage values u a , u b , u cAfter sampling the current, frequency analysis is performed. When there is a large peak at a certain frequency in the frequency spectrum, there is a current harmonic.

[0064] Step 2, based on the sampled motor U, V, W phase current value i a b c The measured value of the stator current component is obtained by Clark transformation i α and i β Based on the sampled motor U, V, W phase voltage value u a b c The measured value of the stator voltage component is obtained by Clark transformation u sα and u sβ Based on the measured value of the stator current component i α and i β , and the measured value of the stator voltage component u sα and u sβ , the flux linkage ψ s and angle θ are obtained by flux observer. At the same time, the q-axis torque command T e * .

[0065] Based on the measured value of the stator current component i α and i β , and the angle θ, Park transformation is performed to obtain the measured value of the d, q-axis stator current component i d and i q Frequency analysis is performed on the measured value of the q-axis stator current component i q to obtain the harmonic frequency, amplitude, etc.

[0066] Based on the flux linkage ψ s and angle θ, the measured value of the d-axis stator current component i d , and the q-axis torque command T e * , PI operation and Park inverse transformation are performed to obtain the command value of the stator voltage component u sa * and u sβ * .

[0067] Where, the measured value is in the feedback link, the command value is in the forward link, and in Figure 5 It can be seen that both are obtained through a series of transformations. The measured value is obtained in practice, indicating the current; the command value is the target value of the next stage.

[0068] ​​​​Wherein, the Clark transformation and Park inverse transformation formula are as follows:

[0069]

[0070]

[0071] Wherein: θ is the included angle between the d-axis and the a-axis in the synchronous rotation coordinate system, I a , I b , I c is the a, b, c phase current measured by the current sensor. α , I β is the axis current value after the Clark transformation. d , U q is the d, q axis voltage value. α , U β is the α, β axis voltage value after the Park inverse transformation. In formula (1) and formula (2), the meaning of the different symbols in the upper and lower cases is the same.

[0072] At step S140, based on the measured value of the α, β axis stator voltage component of the motor and the instruction value of the α, β axis stator voltage component of the motor, the preset quasi-proportional resonant controller is used to determine the α, β axis voltage value of the motor. The α, β axis voltage value of the motor is u α * and u β * .

[0073] In some embodiments, the specific process of determining the α, β axis voltage value of the motor based on the measured value of the α, β axis stator voltage component of the motor and the instruction value of the α, β axis stator voltage component of the motor in step S140 using the preset quasi-proportional resonant controller is described in the following exemplary description.

[0074] The following describes an embodiment flowchart for determining the α, β axis voltage value of the motor using the preset quasi-proportional resonant controller in the method of the application shown in Figure 3 FIG. 1, which further describes the specific process of determining the α, β axis voltage value of the motor using the preset quasi-proportional resonant controller in step S140, including steps S310 to S330.

[0075] At step S310, the difference between the instruction value of the α-axis stator voltage component in the instruction value of the α, β axis stator voltage component of the motor and the measured value of the α-axis stator voltage component of the motor in the measured value of the α, β axis stator voltage component of the motor is input into a preset quasi-proportional resonant controller to obtain the α-axis voltage value of the motor.

[0076] Step S320, inputting the difference between the command value of the β-axis stator voltage component in the command values of the α, β-axis stator voltage components of the motor and the measured value of the β-axis stator voltage component of the motor in the measured values of the α, β-axis stator voltage components of the motor into another preset PR controller to obtain the β-axis voltage value of the motor.

[0077] Step S330, obtaining the α, β-axis voltage values of the motor based on the α-axis voltage value of the motor and the β-axis voltage value of the motor.

[0078] Referring to the example shown in Figure 5 、 Figure 6 and Figure 7 , the driving method with current harmonic suppression function provided by the scheme of the application further comprises:

[0079] Step 3, inputting the difference between the measured value of the stator voltage component u sa * and the measured value of the stator voltage component u sβ * fed back into a PR controller. sα sβ respectively.

[0080] Step 4, inputting the output u α * and u β * of the two PR controllers into a vector space control link and driving the motor to operate normally through an inverter.

[0081] Among them, the vector space control link is a space vector pulse width modulation (SVPWM).

[0082] Among them, the PR controller in the preset PR controller and the other preset PR controller comprises a PR main controller, an odd harmonic suppressor and an even PR controller.

[0083]

[0084] Among them, k p is a proportional link coefficient, k r is a resonance link coefficient, ω0 is the angular frequency of a sine signal, i.e. the resonance angular frequency, and ω c is the cut-off angular frequency.

[0085] The transfer function of the odd harmonic suppressor is:

[0086] ​

[0087] where k r is the odd order resonant pole coefficient, ω0 is the angular frequency of the odd order sinusoidal signal, i.e., the resonant angular frequency, and ω c is the odd order cutoff angular frequency.

[0088] The transfer function of the even order quasi-resonant controller is:

[0089]

[0090] where k r6 and k r12 are the 6th and 12th order resonant pole coefficients, respectively, ω0 is the angular frequency of the even order sinusoidal signal, i.e., the resonant angular frequency, and ω c is the even order cutoff angular frequency.

[0091] Referring to the examples shown in Figure 5 , Figure 6 and Figure 7 , the driving method with current harmonic suppression function provided by the scheme of the present application further comprises:

[0092] The quasi-proportional resonant controller is composed of the following parts:

[0093] The first part, the transfer function of the quasi-proportional resonant main controller is:

[0094]

[0095] where k p is the proportional pole coefficient, k r is the resonant pole coefficient, ω0 is the angular frequency of the sinusoidal signal, i.e., the resonant angular frequency, and ω c is the cutoff angular frequency.

[0096] The second part, the odd order harmonic suppressor:

[0097]

[0098] where k r is the odd order resonant pole coefficient, ω0 is the angular frequency of the odd order sinusoidal signal, i.e., the resonant angular frequency, and ω c is the odd order cutoff angular frequency.

[0099] The third part, the 6th and 12th order harmonics have greater impact on the smoothness of the output torque during the operation of the motor, and therefore the even order quasi-resonant controller is set. The even order quasi-resonant controller:

[0100]

[0101] where k r6 and k r126th and 12th harmonic resonant element coefficients, ω0 is the angular frequency of the even order sinusoidal signal, i.e. the resonance angular frequency, ω c is the even order cutoff angular frequency.

[0102] The above controllers are connected in parallel to obtain a proportional resonant controller as shown in formula (5). Figure 6 The proportional resonant controller. And according to the needs of current resonance, the corresponding odd or even resonant controller can be selected. Through frequency analysis, the amplitude, frequency and composition of each order of resonance are obtained, so as to adjust the coefficients k p , k r , ω0 and ω c to eliminate the resonance effect.

[0103] When the current harmonic current is measured by frequency analysis at a certain time, a large peak exists at 150hz, and generally, the frequency of the fundamental current is 50hz, so 150 / 50=3; the odd order resonant controller is selected. After suppression, the current is analyzed again, and when the frequency spectrum does not exist or has a small peak, the suppression is successful. In the case where the resonance is not suppressed, kr and kp are usually increased.

[0104] At step S150, based on the alpha and beta axis voltage values of the motor, after space vector pulse width modulation and inverter inversion, the motor is driven to operate.

[0105] The scheme of the application realizes zero steady-state error control by detecting the resonant frequency of the current and realizing the sinusoidal signal with the same frequency as the resonant frequency, and controls the harmonic in the motor system by selecting appropriate resonant frequency and other parameters, so as to achieve the suppression effect of the output voltage, and further suppress the shaft current, suppress the motor harmonic torque ripple, reduce the motor harmonic content, realize the smooth operation of the motor, and improve the motor working performance.

[0106] In some embodiments, the motor driving method further comprises the process of determining the current harmonic frequency of the motor.

[0107] Based on the measured value of the q-axis stator current component of the motor, the amplitude change and resonant frequency of the q-axis stator current of the motor are analyzed, the current harmonic condition of the motor is determined, and the determined current harmonic condition of the motor is displayed.

[0108] Referring to Figure 5 , Figure 6 and Figure 7The example shows that the driving method with the current harmonic suppression function provided by the scheme of the application further includes: when the current harmonic exists in the entire driving device, the harmonic state information is uploaded to the display panel of the driving device, and the display interface is entered through the "switch" key to check which harmonic exists and the frequency of the harmonic. Figure 8 The harmonic operation interface diagram of an embodiment of the current harmonic suppression control system is shown in the figure. Figure 8 Figure 8 The "3" in the figure represents the display content, for example, if the sampling detects a 3rd harmonic of 150 Hz, the display is "3rd harmonic 150 Hz". In this way, the scheme of the application modularizes the current harmonic suppression, as an operationally simple driving device, can perform harmonic detection and information display, and can target the harmonic for processing, suppress the current harmonic, and improve the speed and accuracy.

[0109] For some occasions with low mechanical and electrical interference factors, the even-order quasi-resonant controller can also achieve the effect of the scheme of the application. For example, when the field wiring is good and the shielding layer of each device or wire is good, the 6th and 12th even-order quasi-resonant controllers can be removed.

[0110] The related scheme needs to know the prediction mathematical model of the current system in advance, and for the case where the prediction mathematical model of the controlled system is difficult to obtain, the calculation and classification are undoubtedly blind, and the adaptability of the system is reduced. And the situation in the industrial field is often different from the theory, it is difficult to master through model prediction, and it is difficult to suppress unexpected harmonics. The harmonic current detection method provided by the scheme of the application detects the harmonic current in actual application, has pertinence, can improve the adaptability of the motor system, and can use a more targeted quasi-proportional resonant controller to control the current harmonic, improve the anti-interference ability of the motor system, and the scheme is more reasonable.

[0111] The related scheme is also based on the motor mathematical model for the detection of the current harmonic, and the motor is connected with the mechanical member in the industrial field, and there are external factors. The model prediction often deviates from the actual working condition. And its current resonance suppression method is aimed at high-frequency harmonics and only processes one kind of harmonic, which is not easy to handle if multiple current harmonics are superimposed. The harmonic current detection method provided by the scheme of the application detects the harmonic current in actual application, and can handle the case where multiple resonant currents are superimposed, reduces the cost, and ensures the normal operation of the motor.

[0112] ​The scheme of the present application analyzes the amplitude variation and resonance frequency of the quadrature-axis current to obtain the current harmonic characteristics currently existing, and further suppresses the current harmonic currently existing by using a quasi-proportional resonant controller, so that the current is suppressed by a current harmonic suppression scheme, the system oscillation is reduced, and the speed oscillation fluctuation in the steady state and transient state is reduced. The amplitude variation and resonance frequency of the quadrature-axis current are detected and analyzed, the suppression of the current harmonic is targeted, the suppression speed is fast, the scheme is simple and easy to implement, the scheme is modularized and beneficial to engineering application, and the scheme has the functions of detection, display and suppression. It can be seen that the scheme of the present application can realize high-precision and low-oscillation current harmonic suppression control, and reduce the speed fluctuation. The design is simple, the suppression is fast, the observation is easy, and the practicability is high.

[0113] By analyzing the amplitude variation and resonance frequency of the quadrature-axis current, the current harmonic characteristics currently existing are obtained. Further, the current harmonic currently existing is suppressed by using a quasi-proportional resonant controller, so that the current harmonic is suppressed by using the quasi-proportional resonant controller, the current harmonic can be quickly suppressed, and the oscillation of the motor system can be reduced, which is beneficial to improving the operation performance of the motor.

[0114] According to the embodiment of the present application, a motor driving device corresponding to the motor driving method is also provided. Referring to Figure 4 , a structure schematic diagram of an embodiment of the device of the present application is shown. The motor driving device can include an acquisition unit 102 and an operation unit 104.

[0115] The acquisition unit 102 is configured to acquire three-phase current values of the motor, acquire three-phase voltage values of the motor, and acquire a rotating speed of the motor after the motor is started. The three-phase current values of the motor are current values i a , i b , i c of U, V, and W phases of the motor, the three-phase voltage values of the motor are voltage values u a , u b , u c of U, V, and W phases of the motor, and the rotating speed of the motor is a rotating speed ω. The specific functions and processes of the acquisition unit 102 are described with reference to step S110.

[0116] The operation unit 104 is configured to perform Clark transformation on the three-phase voltage values of the motor to obtain measured values of α and β axis stator voltage components of the motor. The measured values of the α and β axis stator voltage components of the motor are u sα and u sβ . The specific functions and processes of the operation unit 104 are described with reference to step S120.

[0117] The operation unit 104 is further configured to determine the command values of the stator voltage components of the motor in the α-axis and β-axis based on the three-phase current values of the motor, the rotational speed of the motor, and the measured values of the stator voltage components of the motor in the α-axis and β-axis. The command values of the stator voltage components of the motor in the α-axis and β-axis are u sα and u sβ . The specific functions and processes of the operation unit 104 will also be described in step S130. sa * and u sβ * . The specific functions and processes of the operation unit 104 will also be described in step S130.

[0118] In some embodiments, the operation unit 104 determines the command values of the stator voltage components of the motor in the α-axis and β-axis based on the three-phase current values of the motor, the rotational speed of the motor, and the measured values of the stator voltage components of the motor in the α-axis and β-axis, including:

[0119] The operation unit 104 is further configured to perform Clark transformation on the three-phase current values of the motor to obtain the measured values of the stator current components of the motor in the α-axis and β-axis. The measured values of the stator current components of the motor in the α-axis and β-axis are i α and i β . The specific functions and processes of the operation unit 104 will also be described in step S210.

[0120] The operation unit 104 is further configured to observe the stator flux of the motor and the angle between the d-axis and the α-axis of the motor in the synchronous rotating coordinate system by a flux observer based on the measured values of the stator voltage components of the motor in the α-axis and β-axis and the measured values of the stator current components of the motor in the α-axis and β-axis. The stator flux of the motor is ψ s , and the angle between the d-axis and the α-axis of the motor in the synchronous rotating coordinate system is θ. The specific functions and processes of the operation unit 104 will also be described in step S220.

[0121] The operation unit 104 is further configured to perform Park transformation based on the measured values of the stator current components of the motor in the α-axis and β-axis and the angle between the d-axis and the α-axis of the motor in the synchronous rotating coordinate system to obtain the measured values of the stator current components of the motor in the d-axis and q-axis. The measured values of the stator current components of the motor in the d-axis and q-axis are i d and i q . The specific functions and processes of the operation unit 104 will also be described in step S230.

[0122] The calculation unit 104 is further configured to perform PI calculations on the speed loop and the outer loop of the flux linkage based on the motor's rotational speed, the measured value of the motor's d-axis stator current component, the motor's stator flux linkage, and the angle between the d-axis and α-axis of the motor in the synchronous rotating coordinate system, and then perform an inverse Park transformation to obtain the command values ​​of the motor's α- and β-axis stator voltage components. The specific functions and processing of this calculation unit 104 are further described in step S240.

[0123] Figure 5 This is a schematic diagram of one embodiment of a current harmonic suppression control system. Figure 5 As shown, in the scheme of this invention, the current harmonic suppression control system (i.e., the driving device for current harmonic suppression) consists of a Clark transform module, a Park inverse transform module, a flux linkage observer, and a quasi-proportional resonant controller. The quasi-proportional resonant control system based on stator flux linkage consists of three loops: a velocity loop, an outer flux linkage loop, and an inner stator voltage loop. Figure 5 In the example shown, the current loop adopts a closed-loop control scheme, which requires accurate current feedback values, voltage feedback values, and flux linkage feedback values. Its current harmonic suppression effect is in the inner loop of the stator voltage.

[0124] Figure 6 A schematic diagram of one embodiment of a quasi-proportional resonant controller. Figure 5 For the specific structure of the quasi-proportional resonant controller in the example shown, please refer to [link / reference needed]. Figure 6 The example shown.

[0125] Figure 7 This is a schematic diagram illustrating the overall workflow of an embodiment of a drive device with current harmonic suppression function. See also... Figure 5 , Figure 6 and Figure 7 As shown in the example, the driving device with current harmonic suppression function provided by the present invention includes:

[0126] Step 1: After the motor starts, when current harmonics are present, sample the U, V, and W phases of the motor to obtain the corresponding current values ​​i. a i b i c and voltage value u a u b u c .

[0127] Step 2: Based on the sampled current values ​​i of the U, V, and W phases of the motor a i b i c The measured values ​​of the α and β axis stator current components i are obtained through Clark transformation. α and i βThe voltage values u a , u b , u c of the U, V, and W phases of the motor are sampled sα , u sβ The measured values u α , u β of the stator voltage components are obtained by Clark transformation based on the measured values i sα , i sβ of the α and β axis stator current components s , θ are obtained by a flux observer based on the measured values u e , u * of the stator voltage components and the angle θ. Meanwhile, the torque command T

[0128] The measured values i α , i β of the d and q axis stator current components are obtained by Park transformation based on the measured values i d , i q of the α and β axis stator current components and the angle θ. The measured value i q of the q axis stator current component is subjected to frequency analysis to obtain the frequency, amplitude, and other components of the existing harmonics.

[0129] The α and β axis stator voltage component command values u s , u d are obtained by PI operation and Park inverse transformation based on the flux ψ e , T * , and the measured value i sa , u sβ * of the q axis torque command T

[0130] The Clark transformation and Park inverse transformation formulas are as follows:

[0131]

[0132]

[0133] In the formulas, θ is the included angle between the d axis and the α axis in the synchronous rotating coordinate system, I a , I b , I c are the a, b, and c phase currents measured by the current sensor. I α , I β are the axis current values after Clark transformation. U d , U q are the d and q axis voltage values. U α , U βPark inverse transformation voltage values of the α-axis and the β-axis.

[0134] The operation unit 104 is further configured to determine the α-axis voltage value and the β-axis voltage value of the motor by using a preset PR controller based on the measured value of the α-axis stator voltage component and the β-axis stator voltage component of the motor and the instruction value of the α-axis stator voltage component and the β-axis stator voltage component of the motor. The α-axis voltage value and the β-axis voltage value of the motor are u α * and u β * The specific functions and processes of the operation unit 104 are also described in step S140.

[0135] In some embodiments, the operation unit 104 determines the α-axis voltage value and the β-axis voltage value of the motor by using a preset PR controller based on the measured value of the α-axis stator voltage component and the β-axis stator voltage component of the motor and the instruction value of the α-axis stator voltage component and the β-axis stator voltage component of the motor, including:

[0136] The operation unit 104 is further configured to input the difference between the instruction value of the α-axis stator voltage component in the instruction value of the α-axis stator voltage component and the β-axis stator voltage component of the motor and the measured value of the α-axis stator voltage component in the measured value of the α-axis stator voltage component and the β-axis stator voltage component of the motor into a preset PR controller to obtain the α-axis voltage value of the motor. The specific functions and processes of the operation unit 104 are also described in step S310.

[0137] The operation unit 104 is further configured to input the difference between the instruction value of the β-axis stator voltage component in the instruction value of the α-axis stator voltage component and the β-axis stator voltage component of the motor and the measured value of the β-axis stator voltage component in the measured value of the α-axis stator voltage component and the β-axis stator voltage component of the motor into another preset PR controller to obtain the β-axis voltage value of the motor. The specific functions and processes of the operation unit 104 are also described in step S320.

[0138] The operation unit 104 is further configured to obtain the α-axis voltage value and the β-axis voltage value of the motor based on the α-axis voltage value and the β-axis voltage value of the motor. The specific functions and processes of the operation unit 104 are also described in step S330.

[0139] Referring to the examples shown in Figure 5 , Figure 6 and Figure 7 The driving device with current harmonic suppression function provided by the scheme of the present application further comprises:

[0140] Step 3, u sa * and u sβ* measured stator voltage component u sα and u sβ are respectively subtracted and input into the quasi-proportional resonant controllers.

[0141] Step 4, two quasi-proportional resonant controller outputs u α * and u β * enter the vector space control link, and drive the motor to run normally through the inverter. Among them, the vector space control link is the space vector pulse width modulation (SVPWM).

[0142] Among them, the preset one quasi-proportional resonant controller and the quasi-proportional resonant controller in the preset another quasi-proportional resonant controller include a quasi-proportional resonant main controller, an odd harmonic wave suppressor and an even quasi-resonant controller.

[0143] The transfer function of the quasi-proportional resonant main controller is:

[0144]

[0145] Among them, k p is a proportional link coefficient, k r is a resonant link coefficient, ω0 is the angular frequency of a sine signal, i.e. the resonant angular frequency, and ω c is the cutoff angular frequency.

[0146] The transfer function of the odd harmonic wave suppressor is:

[0147]

[0148] Among them, k r is an odd resonant link coefficient, ω0 is the angular frequency of an odd sine signal, i.e. the resonant angular frequency, and ω c is the odd cutoff angular frequency.

[0149] The transfer function of the even quasi-resonant controller is:

[0150]

[0151] Among them, k r6 and k r12 are 6th and 12th resonant link coefficients, ω0 is the angular frequency of an even sine signal, i.e. the resonant angular frequency, and ω c is the even cutoff angular frequency.

[0152] Referring to Figure 5 , Figure 6 and Figure 7The example shown, the scheme of the application provides a driving device with current harmonic suppression function, further comprising:

[0153] The quasi-proportional resonant controller consists of the following parts:

[0154] The first part, the quasi-proportional resonant main controller transfer function:

[0155]

[0156] Wherein, k p is the proportional link coefficient, k r is the resonant link coefficient, ω0 is the angular frequency of the sinusoidal signal, i.e. the resonant angular frequency, ω c is the cutoff angular frequency.

[0157] The second part, the odd harmonic suppressor:

[0158]

[0159] Wherein, k r is the odd resonant link coefficient, ω0 is the angular frequency of the odd sinusoidal signal, i.e. the resonant angular frequency, ω c is the odd cutoff angular frequency.

[0160] The third part, the even quasi-resonant controller. The even quasi-resonant controller:

[0161]

[0162] Wherein, k r6 and k r12 are the 6th and 12th resonant link coefficients, ω0 is the angular frequency of the even sinusoidal signal, i.e. the resonant angular frequency, ω c is the even cutoff angular frequency.

[0163] The above controllers are connected in parallel to obtain a quasi-proportional resonant controller. Figure 6 The odd or even resonant controller can be selected according to the current resonance needs. Through frequency analysis, the amplitude, frequency and composition of each resonant are obtained, so as to adjust the coefficients k p , k r , ω0, ω c to achieve the effect of eliminating resonance.

[0164] The operation unit 104 is further configured to drive the motor to operate based on the alpha and beta axis voltage values of the motor after space vector pulse width modulation and inverter inversion. The specific functions and processing of the operation unit 104 are also referred to step S150.

[0165] The scheme of the present application realizes the sine signal with the same frequency as the resonant frequency of the current and further realizes zero steady-state error control by detecting the resonant frequency of the current, and the output voltage can be suppressed by selecting appropriate resonant frequency and other parameters to control the harmonic in the motor system, thereby suppressing the shaft current, suppressing the motor harmonic torque ripple, reducing the motor harmonic content, achieving smooth operation of the motor, and improving the motor performance.

[0166] In some embodiments, the motor drive device further comprises a process of determining the current harmonic frequency of the motor.

[0167] The operation unit 104 is further configured to analyze the amplitude variation and resonant frequency of the q-axis stator current of the motor based on the measured value of the q-axis stator current component of the motor, determine the current harmonic condition of the motor, and display the determined current harmonic condition of the motor.

[0168] Referring to the examples shown in Figure 5 , Figure 6 and Figure 7 The scheme of the present application provides a drive device with current harmonic suppression function, which further comprises the following steps: when there is current harmonic in the entire drive device, the harmonic state information is uploaded to the display panel of the drive device, and the "switch" key is used to enter the display interface to view which harmonics exist at present and the frequency of the harmonics. Figure 8 The harmonic operation interface diagram of an embodiment of the current harmonic suppression control system is shown in Figure 8 . Figure 8 The "?" in the above formula represents the display content, for example, if the sampling detects a 3rd harmonic of 150 Hz, the display is "3rd harmonic 150 Hz". In this way, the scheme of the present application modularizes the current harmonic suppression module as a drive device with simple operation, which can detect and display harmonic information and can process the harmonic in a targeted manner to suppress the current harmonic while improving the speed and accuracy.

[0169] For some occasions with low mechanical and electrical interference factors, the even-order quasi-resonant controller can also achieve the effect of the scheme of the present application.

[0170] Existing solutions require prior knowledge of the current system's predictive mathematical model. When the predictive mathematical model for the controlled system is difficult to obtain, this undoubtedly increases the randomness of calculations and classifications, reducing the system's adaptability. Furthermore, the conditions encountered in industrial settings often differ from theoretical predictions, making it difficult to grasp the situation through model predictions and suppress unexpected harmonics. In contrast, the harmonic current detection device proposed in this invention detects harmonic currents in practical applications, providing targeted detection and improving the adaptability of the motor system. Moreover, it allows for the use of a more targeted quasi-proportional resonant controller to control current harmonics, enhancing the motor system's immunity to disturbances. Therefore, this solution is more rational.

[0171] Existing solutions rely on mathematical models of motors for current harmonic detection. However, in industrial settings, motors are connected to mechanical components, increasing the influence of external factors and causing model predictions to deviate from actual operating conditions. Furthermore, their current resonance suppression devices target high-frequency harmonics and only handle one type of harmonic; they are ineffective when multiple current harmonics are superimposed. The harmonic current detection device proposed in this invention detects harmonic currents in practical applications and can handle situations where multiple resonant currents are superimposed, reducing costs and ensuring normal motor operation.

[0172] The present invention analyzes the amplitude change and resonant frequency of the quadrature-axis current to obtain the characteristics of the existing current harmonics. Then, a quasi-proportional resonant controller is used to suppress the existing current harmonics. Thus, a current harmonic suppression scheme reduces system oscillations while simultaneously reducing velocity oscillation fluctuations in both steady-state and transient processes. The detection and analysis of the amplitude change and resonant frequency of the quadrature-axis current allows for targeted, fast, and simple implementation of current harmonic suppression. Furthermore, its modular design facilitates engineering applications, providing detection, display, and suppression functions. Therefore, the present invention achieves high-precision, low-oscillation current harmonic suppression control while reducing velocity fluctuations. It is simple in design, provides rapid suppression, is easy to observe, and has strong practicality.

[0173] Since the processing and functions implemented by the device in this embodiment are basically the same as the embodiments, principles and examples of the aforementioned methods, any details not covered in the description of this embodiment can be found in the relevant descriptions in the aforementioned embodiments, and will not be repeated here.

[0174] By employing the technical solution of this invention, the characteristics of the current harmonics are obtained by analyzing the amplitude change and resonant frequency of the quadrature-axis current; then, the current harmonics are suppressed by a quasi-proportional resonant controller, which can specifically handle the harmonics, suppressing the current harmonics while improving speed accuracy.

[0175] According to an embodiment of the present application, a motor system corresponding to the motor driving device is also provided.

[0176] Since the processing and functions realized by the motor system of the present embodiment are basically corresponding to the above-mentioned embodiments, principles and examples of the device, the descriptions of the present embodiment not elaborated can be referred to the relevant descriptions in the above-mentioned embodiments, which will not be repeated here.

[0177] By analyzing the amplitude variation and resonance frequency of the quadrature-axis current, the current harmonic characteristics currently existing are obtained, and the current harmonics currently existing are suppressed by using the quasi-proportional resonant controller, so that the smooth operation of the motor is realized, and the working performance of the motor is improved.

[0178] According to an embodiment of the present application, a storage medium corresponding to the motor driving method is also provided, which comprises a stored program, wherein when the program is running, the device where the storage medium is located is controlled to perform the above-mentioned motor driving method.

[0179] Since the processing and functions realized by the storage medium of the present embodiment are basically corresponding to the above-mentioned embodiments, principles and examples of the method, the descriptions of the present embodiment not elaborated can be referred to the relevant descriptions in the above-mentioned embodiments, which will not be repeated here.

[0180] By analyzing the amplitude variation and resonance frequency of the quadrature-axis current, the current harmonic characteristics currently existing are obtained, and the current harmonics currently existing are suppressed by using the quasi-proportional resonant controller, so that the smooth operation of the motor is realized, and the working performance of the motor is improved.

[0181] In summary, those skilled in the art can easily understand that the above-mentioned advantageous modes can be freely combined and superimposed without conflict.

[0182] The above-mentioned only is the embodiment of the present application, and is not used to limit the present application, and for those skilled in the art, the present application can have various changes and variations. Any modification, equivalent replacement, improvement, etc. made within the spirit and principles of the present application shall be included in the scope of claims of the present application.

Claims

1. A motor driving method, characterized in that, include: After the motor starts, the three-phase current value of the motor, the three-phase voltage value of the motor, and the speed of the motor are obtained. The three-phase voltage values ​​of the motor are subjected to Clark transformation to obtain the measured values ​​of the α and β axis stator voltage components of the motor. The three-phase current values ​​of the motor are subjected to Clark transformation to obtain the measured values ​​of the α and β axis stator current components of the motor. Based on the measured values ​​of the α and β axis stator voltage components and the measured values ​​of the α and β axis stator current components of the motor, the stator flux linkage of the motor and the angle between the d-axis and α-axis of the motor in the synchronous rotating coordinate system are observed by a flux linkage observer. Based on the measured values ​​of the α and β axis stator current components and the angle between the d-axis and α-axis of the motor in the synchronous rotating coordinate system, a Park transformation is performed to obtain the measured values ​​of the d and q axis stator current components of the motor. Based on the motor speed, the measured values ​​of the d-axis stator current components, the stator flux linkage of the motor, and the angle between the d-axis and α-axis of the motor in the synchronous rotating coordinate system, PI calculations are performed in the speed loop and the outer flux linkage loop respectively, followed by an inverse Park transformation to obtain the command values ​​of the α and β axis stator voltage components of the motor. Based on the measured values ​​of the α and β axis stator voltage components of the motor and the command values ​​of the α and β axis stator voltage components of the motor, the α and β axis voltage values ​​of the motor are determined using a preset quasi-proportional resonant controller. Based on the α and β axis voltage values ​​of the motor, the motor is driven to run after being modulated by space vector pulse width modulation and then inverted by an inverter.

2. The motor driving method according to claim 1, characterized in that, Also includes: Based on the measured values ​​of the q-axis stator current component of the motor, the amplitude variation and resonant frequency of the q-axis stator current of the motor are analyzed to determine the current harmonic situation of the motor, and the determined current harmonic situation of the motor is displayed.

3. The motor driving method according to claim 1 or 2, characterized in that, Based on the measured values ​​of the α and β axis stator voltage components of the motor, and the command values ​​of the α and β axis stator voltage components of the motor, the α and β axis voltage values ​​of the motor are determined using a preset quasi-proportional resonant controller, including: The difference between the command value of the α-axis stator voltage component and the measured value of the α-axis stator voltage component of the motor is input into a preset quasi-proportional resonant controller to obtain the α-axis voltage value of the motor. The difference between the command value of the β-axis stator voltage component in the command value of the α and β-axis stator voltage components of the motor and the measured value of the β-axis stator voltage component in the measured value of the α and β-axis stator voltage components of the motor is input to another preset quasi-proportional resonant controller to obtain the β-axis voltage value of the motor. The α-axis voltage value and the β-axis voltage value of the motor are used to obtain the α-axis and β-axis voltage values ​​of the motor.

4. The motor driving method according to claim 3, characterized in that, in, The preset quasi-proportional resonant controller, and another preset quasi-proportional resonant controller, include: a quasi-proportional resonant main controller, an odd-order harmonic suppressor, and an even-order quasi-resonant controller; wherein, The transfer function of the quasi-proportional resonant main controller is: ； in, This is the coefficient for the proportional element. The coefficient of the resonant element. This is the angular frequency of the sinusoidal signal, i.e., the resonant angular frequency. The cutoff angular frequency; The transfer function of the odd harmonic suppressor is: ； in, The coefficients of the odd-order resonant elements, This refers to the angular frequency of an odd-order sinusoidal signal, i.e., the resonant angular frequency. The odd-order cutoff angular frequency; The transfer function of the even-order quasi-resonant controller is: ； in, and The coefficients of the 6th and 12th resonant elements are respectively. The angular frequency of an even-order sinusoidal signal, i.e., the resonant angular frequency. It is an even-order cutoff angular frequency.

5. A motor drive device that implements motor drive using the motor drive method as described in claim 1, characterized in that, include: The acquisition unit is configured to acquire the three-phase current value of the motor, acquire the three-phase voltage value of the motor, and acquire the speed of the motor after the motor is started. The arithmetic unit is configured to perform Clark transformation on the three-phase voltage values ​​of the motor to obtain the measured values ​​of the α and β axis stator voltage components of the motor. The computing unit is further configured to calculate the three-phase current value of the motor, the speed of the motor, and the measured values ​​of the α and β axis stator voltage components of the motor. and The command values ​​for the α and β axis stator voltage components of the motor are determined. The computing unit is also configured to determine the α and β axis voltage values ​​of the motor using a preset quasi-proportional resonant controller based on the measured values ​​of the α and β axis stator voltage components of the motor and the command values ​​of the α and β axis stator voltage components of the motor. The computing unit is also configured to drive the motor by performing space vector pulse width modulation based on the α and β axis voltage values ​​of the motor, followed by inverter inversion.

6. The motor drive device according to claim 5, characterized in that, The arithmetic unit determines the command values ​​of the α and β axis stator voltage components of the motor based on the three-phase current values ​​of the motor, the motor speed, and the measured values ​​of the α and β axis stator voltage components of the motor, including: The three-phase current values ​​of the motor are subjected to Clark transformation to obtain the measured values ​​of the α and β axis stator current components of the motor. Based on the measured values ​​of the α and β axis stator voltage components of the motor and the measured values ​​of the α and β axis stator current components of the motor, the stator flux of the motor and the angle between the d-axis and the α-axis of the motor in the synchronous rotating coordinate system are observed by the flux linkage observer. Based on the measured values ​​of the α and β axis stator current components of the motor, and the angle between the d axis and the α axis of the motor in the synchronous rotating coordinate system, Park transformation is performed to obtain the measured values ​​of the d and q axis stator current components of the motor. Based on the motor's rotational speed, the measured value of the motor's d-axis stator current component, the motor's stator flux linkage, and the angle between the motor's d-axis and α-axis in the synchronous rotating coordinate system, PI calculations are performed in the speed loop and the flux linkage outer loop, followed by Park inverse transformation to obtain the command values ​​of the motor's α and β-axis stator voltage components.

7. The motor drive device according to claim 6, characterized in that, Also includes: The computing unit is also configured to analyze the amplitude variation and resonant frequency of the q-axis stator current of the motor based on the measured value of the q-axis stator current component, determine the current harmonic situation of the motor, and display the determined current harmonic situation of the motor.

8. The motor drive device according to any one of claims 5 to 7, characterized in that, The arithmetic unit, based on the measured values ​​of the α and β axis stator voltage components of the motor and the command values ​​of the α and β axis stator voltage components of the motor, determines the α and β axis voltage values ​​of the motor using a preset quasi-proportional resonant controller, including: The difference between the command value of the α-axis stator voltage component and the measured value of the α-axis stator voltage component of the motor is input into a preset quasi-proportional resonant controller to obtain the α-axis voltage value of the motor. The difference between the command value of the β-axis stator voltage component in the command value of the α and β-axis stator voltage components of the motor and the measured value of the β-axis stator voltage component in the measured value of the α and β-axis stator voltage components of the motor is input to another preset quasi-proportional resonant controller to obtain the β-axis voltage value of the motor. The α-axis voltage value and the β-axis voltage value of the motor are used to obtain the α-axis and β-axis voltage values ​​of the motor.

9. The motor drive device according to claim 8, characterized in that, in, The preset quasi-proportional resonant controller, and another preset quasi-proportional resonant controller, include: a quasi-proportional resonant main controller, an odd-order harmonic suppressor, and an even-order quasi-resonant controller; wherein, The transfer function of the quasi-proportional resonant main controller is: ； in, This is the coefficient for the proportional element. The coefficient of the resonant element. This is the angular frequency of the sinusoidal signal, i.e., the resonant angular frequency. The cutoff angular frequency; The transfer function of the odd harmonic suppressor is: ； in, The coefficients of the odd-order resonant elements, This refers to the angular frequency of an odd-order sinusoidal signal, i.e., the resonant angular frequency. The odd-order cutoff angular frequency; The transfer function of the even-order quasi-resonant controller is: ； in, and The coefficients of the 6th and 12th resonant elements are respectively. The angular frequency of an even-order sinusoidal signal, i.e., the resonant angular frequency. It is an even-order cutoff angular frequency.

10. A motor system, characterized in that, include: The motor drive device as described in any one of claims 5 to 9.

11. A storage medium, characterized in that, The storage medium includes a stored program, wherein, when the program is executed, it controls the device containing the storage medium to perform the motor drive method according to any one of claims 1 to 4.

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