Switched capacitor converter and switching circuit and switch thereof

Abstract

A switched capacitor converter and a switching circuit and a switch thereof. The switched capacitor converter includes a plurality of switch units for switching a coupling relationship between a capacitor and a first power supply and a second power supply, wherein at least one of the plurality of switch units includes a switching circuit. The switching circuit includes a first switch, a second switch and a switching driving circuit, wherein an on-resistance value of the first switch is greater than an on-resistance value of the second switch, and a parasitic capacitance value of the first switch is less than a parasitic capacitance value of the second switch. The switching driving circuit controls the first switch to be turned on earlier than the second switch and / or controls the first switch to be turned off later than the second switch, so that a switching loss of the switching circuit is less than a preset target value.

Description

Technical Field

[0001] This invention relates to a switching capacitor converter, and more particularly to a switching capacitor converter with low switching loss and low conduction loss. The invention also relates to a switching circuit suitable for a switching capacitor converter and having low switching power loss and low conduction power loss. Background Technology

[0002] Please refer to Figure 1 , Figure 1 This is a schematic diagram of a prior art switching capacitor converter 100. (See diagram below.) Figure 1 As shown, the switching capacitor converter 100 is a circuit employing charge pump technology (charge pump technology is well known to those skilled in the art and will not be described in detail here) to convert the power from the power supply (POW) into charging power for charging the battery. The switching capacitor converter 100 is commonly used in various types of chargers (e.g., mobile phone chargers, laptop chargers, etc.). Due to the inclusion of multiple switches Q1-Q4, the existing switching capacitor converter 100 suffers from significant switching loss and excessively high temperature during fast charging due to the parasitic capacitance of the switches Q1-Q4.

[0003] Furthermore, since most existing switching capacitor converters 100 use MOSFETs as their internal switches, the charging efficiency of these converters is also compromised by the manufacturing process of the MOSFETs themselves. Please also refer to... Figure 2 and Figure 3 , Figure 2 This is a schematic diagram of the layout of a metal-oxide-semiconductor (MOS) transistor in the prior art. Figure 3 This is a characteristic diagram of the safe operating region of a prior art metal-oxide-semiconductor transistor. For example... Figure 2 and Figure 3 As shown, the prior art metal-oxide-semiconductor (MOSFET) has a ring-shaped body contact point B, which provides a large safe operating area (SOA) to prevent damage (as shown by solid line L1). However, due to the large area of ​​the ring-shaped body contact point B, the prior art MOSFET has a large on-resistance value, which in turn results in a large conduction loss for the prior art switching capacitor converter 100. Summary of the Invention

[0004] This invention provides a switching circuit, comprising: a first switch, including a first control terminal, a first inverting output terminal, and a first non-inverting output terminal; a second switch, including a second control terminal, a second inverting output terminal, and a second non-inverting output terminal, wherein the second inverting output terminal is coupled to the first inverting output terminal, and the second non-inverting output terminal is coupled to the first non-inverting output terminal; the on-resistance of the first switch is greater than the on-resistance of the second switch, and the parasitic capacitance of the first control terminal is less than the parasitic capacitance of the second control terminal; and a switching drive. A circuit is used to generate a first control signal and a second control signal, thereby controlling the first switch and the second switch respectively; wherein the switching drive circuit controls the first switch to turn on before the second switch turns on and / or controls the first switch to turn off after the second switch turns off, when the second switch turns on and / or turns off, the voltage difference between the second inverting output terminal and the second non-inverting output terminal is less than a preset voltage difference target value, so that all switching losses of the switching circuit are less than a preset target value.

[0005] In some embodiments, the layout area of ​​the first switch is smaller than the layout area of ​​the second switch.

[0006] In some embodiments, the ratio of the layout area of ​​the first switch to the layout area of ​​the second switch is between 1 / 5 and 1 / 3.

[0007] In some embodiments, the second switch is a MOSFET, wherein the second control terminal is a gate, the second inverting output terminal is a drain, and the second non-inverting output terminal is a source.

[0008] In some embodiments, the open-circuit tolerance voltage of the second switch is less than that of the first switch.

[0009] In some embodiments, the area of ​​the safe operating area (SOA) of the second switch is smaller than the area of ​​the safe operating area of ​​the first switch.

[0010] In some embodiments, the sheet resistance of the second switch is less than that of the first switch.

[0011] In some embodiments, the area ratio of the body electrode contact point of the second switch is lower than that of the body electrode contact point of the first switch, so that the plate resistance value of the second switch is less than that of the first switch.

[0012] In some embodiments, the safe operating area of ​​the first switch is non-rectangular.

[0013] In some embodiments, the safe operating area of ​​the second switch is non-rectangular.

[0014] In some embodiments, the safe operating area of ​​the second switch has an inner bend, wherein the withstand voltage value of the inner bend is less than the open-circuit withstand voltage of the second switch.

[0015] In some embodiments, the withstand voltage value corresponding to the aforementioned inner inflection point is 1 / 10 to 1 / 5 of the open-circuit withstand voltage of the second switch.

[0016] In some embodiments, the voltage at the control terminal of the inflection point corresponding to the aforementioned inflection point is greater than the conduction threshold of the second switch.

[0017] In some embodiments, the switching drive circuit includes a clamp coupled between the first inverting output terminal and the first control terminal to clamp the voltage difference between the first inverting output terminal and the first control terminal below a clamping limit, such that the voltage difference between the first inverting output terminal and the first non-inverting output terminal does not exceed the absolute maximum rated value of the first switch, wherein the clamping limit is less than the absolute maximum rated value of the first switch.

[0018] In some embodiments, the switching drive circuit further includes a current source coupled between the first control terminal and the first in-phase output terminal to bias the clamp.

[0019] In another embodiment, the switching drive circuit includes a clamp coupled between the second inverting output terminal and the second control terminal to clamp the voltage difference between the second inverting output terminal and the second control terminal below a clamping limit, such that the voltage difference between the second inverting output terminal and the second non-inverting output terminal does not exceed the absolute maximum rated value of the second switch, wherein the clamping limit is less than the absolute maximum rated value of the second switch.

[0020] In another embodiment, the switching drive circuit further includes a current source coupled between the second control terminal and the second in-phase output terminal to bias the clamp.

[0021] In some embodiments, the switching circuit described above is applicable to a switching capacitor converter, which is used to switch the coupling relationship between a capacitor and a first power supply and a second power supply, thereby converting the first power supply to the second power supply, or converting the second power supply to the first power supply, wherein the voltage of the first power supply is higher than the voltage of the second power supply.

[0022] In some embodiments, the aforementioned switching capacitor converter includes a first switching unit, a second switching unit, a third switching unit, and a fourth switching unit, wherein at least one of the first switching unit, the second switching unit, the third switching unit, or the fourth switching unit includes the switching circuit. The first switching unit is coupled between the first power supply and the first terminal of the capacitor, the second switching unit is coupled between the first terminal of the capacitor and the second power supply, the third switching unit is coupled between the second power supply and the second terminal of the capacitor, and the fourth switching unit is coupled between the second terminal of the capacitor and a ground potential. The first switching unit, the second switching unit, the third switching unit, and the fourth switching unit are used to periodically switch the first terminal of the capacitor between the first power supply and the second power supply, and correspondingly switch the second terminal of the capacitor between the second power supply and the ground potential, thereby converting the first power supply to the second power supply or converting the second power supply to the first power supply, wherein the voltage of the first power supply is twice the voltage of the second power supply, and the current of the first power supply is half the current of the second power supply.

[0023] The present invention further proposes a switching capacitor converter, comprising: a plurality of switching units for switching the coupling relationship between a capacitor and a first power supply and a second power supply, thereby converting the first power supply to the second power supply, or converting the second power supply to the first power supply, wherein the voltage of the first power supply is higher than the voltage of the second power supply; wherein at least one of the switching units includes a switching circuit, the switching circuit comprising: a first switch including a first control terminal, a first inverting output terminal, and a first non-inverting output terminal; and a second switch including a second control terminal, a second inverting output terminal, and a second non-inverting output terminal, the second inverting output terminal being coupled to the first inverting output terminal, and the second non-inverting output terminal being coupled to the first non-inverting output terminal. The first switch has a greater on-resistance than the second switch, and the parasitic capacitance of the first control terminal is less than the parasitic capacitance of the second control terminal. A switching drive circuit is also included to generate a first control signal and a second control signal, which in turn control the first switch and the second switch, respectively. The switching drive circuit controls the first switch to turn on before the second switch turns on and / or controls the first switch to turn off after the second switch turns off. When the second switch turns on and / or turns off, the voltage difference between the second inverting output terminal and the second non-inverting output terminal is less than a preset voltage difference target value, so that all switching losses in the switching circuit are less than a preset target value.

[0024] In some embodiments, the aforementioned switching unit includes a first switching unit, a second switching unit, a third switching unit, and a fourth switching unit, wherein at least one of the first switching unit, the second switching unit, the third switching unit, or the fourth switching unit includes the switching circuit. The first switching unit is coupled between a first power supply and a first terminal of a capacitor, the second switching unit is coupled between the first terminal of the capacitor and a second power supply, the third switching unit is coupled between the second power supply and a second terminal of the capacitor, and the fourth switching unit is coupled between the second terminal of the capacitor and a ground potential. The first switching unit, the second switching unit, the third switching unit, and the fourth switching unit are used to periodically switch the first terminal of the capacitor between the first power supply and the second power supply, and correspondingly switch the second terminal of the capacitor between the second power supply and the ground potential, thereby converting the first power supply to the second power supply or converting the second power supply to the first power supply, wherein the voltage of the first power supply is twice the voltage of the second power supply, and the current of the first power supply is half the current of the second power supply.

[0025] In some embodiments, the open-circuit withstand voltage of the second switch is less than that of the first switch.

[0026] In some embodiments, the safe operating area of ​​the second switch is smaller than the safe operating area of ​​the first switch.

[0027] In some embodiments, the plate resistance value of the second switch is less than the plate resistance value of the first switch.

[0028] In some embodiments, the area ratio of the body electrode contact point of the second switch is lower than that of the body electrode contact point of the first switch, so that the plate resistance value of the second switch is less than that of the first switch.

[0029] In some embodiments, the safe operating area of ​​the first switch is non-rectangular.

[0030] In some embodiments, the safe operating area of ​​the second switch is non-rectangular.

[0031] In some embodiments, the safe operating area of ​​the second switch has an inner bend, wherein the withstand voltage value of the inner bend is less than the open-circuit withstand voltage of the second switch.

[0032] In some embodiments, the withstand voltage value corresponding to the aforementioned inner inflection point is 1 / 10 to 1 / 5 times the open-circuit withstand voltage of the second switch.

[0033] In some embodiments, the voltage at the control terminal of the inflection point corresponding to the aforementioned inflection point is greater than the conduction threshold of the second switch.

[0034] In some embodiments, the voltage at the control terminal corresponding to the inner inflection point is less than the conduction threshold of the second switch.

[0035] In some embodiments, the switching circuit further includes a clamper coupled between the first inverting output terminal and the first control terminal to clamp the voltage difference between the first inverting output terminal and the first control terminal below a clamping limit, such that the voltage difference between the first inverting output terminal and the first non-inverting output terminal does not exceed the absolute maximum rated value of the first switch, wherein the clamping limit is less than the absolute maximum rated value of the first switch.

[0036] In some embodiments, the switching circuit further includes a current source coupled between the first control terminal and the first in-phase output terminal to bias the clamp.

[0037] In another embodiment, the switching circuit further includes a clamp coupled between the second inverting output terminal and the second control terminal to clamp the voltage difference between the second inverting output terminal and the second control terminal below a clamping limit, such that the voltage difference between the second inverting output terminal and the second non-inverting output terminal does not exceed the absolute maximum rated value of the second switch, wherein the clamping limit is less than the absolute maximum rated value of the second switch.

[0038] In another embodiment, the switching circuit further includes a current source coupled between the second control terminal and the second in-phase output terminal to bias the clamp.

[0039] The present invention also provides a switch for switching the coupling relationship between a capacitor and a first power supply and a second power supply, thereby converting the first power supply to the second power supply or the second power supply to the first power supply, wherein the voltage of the first power supply is higher than the voltage of the second power supply. The switch includes a control terminal, an inverting output terminal, and a non-inverting output terminal, wherein the switch has a non-rectangle safe operating area (SOA), a control voltage is provided between the control terminal and the non-inverting output terminal, and a switching voltage is provided between the inverting output terminal and the non-inverting output terminal, wherein when the switch is used to switch the capacitor for power conversion, an operating trajectory (locus) corresponding to the control voltage and the switching voltage is restricted within the non-rectangle safe operating area.

[0040] In some embodiments, the switch is a metal-oxide-semiconductor transistor, the control terminal is a gate, the inverting output terminal is a drain, and the non-inverting output terminal is a source.

[0041] In some embodiments, the aforementioned non-rectangular safety operating area has an inner fold point.

[0042] In some embodiments, the withstand voltage value of the inward bend corresponding to the aforementioned bend is less than the open-circuit withstand voltage of the switch.

[0043] In some embodiments, the withstand voltage value corresponding to the aforementioned inner inflection point is 1 / 10 to 1 / 5 times the open-circuit withstand voltage of the switch.

[0044] In some embodiments, the voltage at the control terminal corresponding to the inner inflection point is greater than the conduction threshold of the switch.

[0045] In some embodiments, the voltage at the control terminal corresponding to the inner inflection point is less than the conduction threshold of the switch.

[0046] This invention proposes a switching capacitor converter and its switching circuit, which can not only reduce the overall loss of the switching capacitor converter to improve its charging efficiency under different loads, but also reduce the temperature of the switching capacitor converter during fast charging.

[0047] The following detailed description through specific embodiments will make it easier to understand the purpose, technical content, features and effects achieved by the present invention. Attached Figure Description

[0048] Figure 1 This is a schematic diagram of a switching capacitor converter using existing technology.

[0049] Figure 2 This is a schematic diagram of the layout of a metal-oxide-semiconductor transistor (MOS) in the prior art.

[0050] Figure 3 This is a characteristic diagram of the safe operating region of a prior art metal-oxide-semiconductor transistor.

[0051] Figure 4 This is a schematic diagram of a switching circuit according to an embodiment of the present invention.

[0052] Figure 5 This is a schematic diagram of a switching capacitor converter according to an embodiment of the present invention.

[0053] Figure 6A This is a flowchart (a) illustrating the operation of a switching capacitor converter according to an embodiment of the present invention.

[0054] Figure 6B This is a flowchart (II) of the operation of switching a capacitor converter according to an embodiment of the present invention.

[0055] Figure 7 This is an operational waveform diagram of switching a capacitor converter according to an embodiment of the present invention.

[0056] Figure 8A This is a schematic diagram of the layout of a metal-oxide-semiconductor transistor according to an embodiment of the present invention.

[0057] Figure 8B This is a schematic diagram of the layout of a metal-oxide-semiconductor transistor according to another embodiment of the present invention.

[0058] Figure 9 These are characteristic diagrams of the safe operating region of a metal-oxide-semiconductor transistor according to different embodiments of the present invention.

[0059] Figure 10A This is an operational trajectory diagram of the control voltage of the first switch and the switch voltage across the circuit, and a characteristic diagram of its safe operating range, according to an embodiment of the present invention.

[0060] Figure 10B This is an operational trajectory diagram of the control voltage of the second switch and the switching voltage across the switch, and a characteristic diagram of its safe operating range, according to an embodiment of the present invention.

[0061] Figure 10C This is an operational trajectory diagram of the control voltage of the second switch and the switching voltage across the switch, and a characteristic diagram of its safe operating range, according to another embodiment of the present invention.

[0062] Figure 11 This is a schematic diagram of a switching capacitor converter with a clamp according to an embodiment of the present invention.

[0063] Figure 12 This is a schematic diagram of a switching capacitor converter with a clamp according to another embodiment of the present invention.

[0064] Figure 13 This is an efficiency curve corresponding to the output current in a switching capacitor converter according to a partial embodiment of the present invention.

[0065] Explanation of symbols in the diagram

[0066] 100: Switching to a capacitive converter

[0067] 200: Switching circuit

[0068] 210: Switching drive circuit

[0069] 300: Switching Capacitor Converter

[0070] 310: First Switching Unit

[0071] 320: Second switching unit

[0072] 330: Third Switching Unit

[0073] 340: Fourth Switching Unit

[0074] 350: Switching drive circuit

[0075] 360: Clamping Device

[0076] B: Body contact point

[0077] BAT: Battery

[0078] CFLY: Capacitor

[0079] CH: First control terminal

[0080] CL: Second control terminal

[0081] CN: The second terminal of the capacitor

[0082] CP: The first terminal of the capacitor

[0083] CV1: Curve

[0084] CV2: Curve

[0085] D: Drain

[0086] G: Gate

[0087] G0: Control signal

[0088] G1: Control signal

[0089] G1H: Control signal

[0090] G1L: Control signal

[0091] G2: Control signal

[0092] G2H: Control signal

[0093] G2L: Control signal

[0094] G3: Control signal

[0095] G3H: Control signal

[0096] G3L: Control Signal

[0097] G4: Control signal

[0098] G4H: Control signal

[0099] G4L: Control Signal

[0100] GH: First control signal

[0101] GL: Second control signal

[0102] Iout: Output current

[0103] Is: Current source

[0104] IBUS: Current of the power supply

[0105] L1: Solid line

[0106] L2: Solid line

[0107] L3: Dashed line

[0108] L4: Dashed line

[0109] L5: Solid line

[0110] L6: Dashed line

[0111] L7: One-point chain

[0112] L8: Two-point chain

[0113] NDD: Voltage Input Terminal

[0114] NOH: First non-inverting output terminal

[0115] NOL: Second non-inverting output terminal

[0116] NSS: Voltage Output Terminal

[0117] POW: Power Supply

[0118] Q0: Path switch

[0119] Q1: Switch

[0120] Q1H: First switch

[0121] Q1L: Second Switch

[0122] Q2: Switch

[0123] Q2H: First switch

[0124] Q2L: Second Switch

[0125] Q3: Switch

[0126] Q3H: First switch

[0127] Q3L: Second Switch

[0128] Q4: Switch

[0129] Q4H: First switch

[0130] Q4L: Second switch

[0131] QH: First switch

[0132] QL: Second Switch

[0133] R1: Solid box

[0134] R2: Solid box

[0135] R3: Solid box

[0136] R4: Solid box

[0137] Rcv1: Turning Point

[0138] Rcv2: Turning Point

[0139] ROH: First inverting output terminal

[0140] ROL: Second inverting output terminal

[0141] S: Source

[0142] S10-S80: Steps

[0143] t0-t8: Time points

[0144] Td1, Td2: Lag time

[0145] VBAT: Secondary power supply

[0146] VBUS: Power Supply

[0147] Vds, Vds1H, Vds1L: Switching voltage across the circuit

[0148] Vgs, Vgs1H, Vgs1L: Control voltage

[0149] VMID: First Power Supply Detailed Implementation

[0150] The accompanying drawings in this invention are schematic and are primarily intended to illustrate the coupling relationships between circuits and the relationships between signal waveforms. The circuits, signal waveforms, and frequencies are not drawn to scale. For clarity, many practical details will be described in the following description, but this is not intended to limit the scope of the patent application.

[0151] Please refer to Figure 4 , Figure 4 This is a schematic diagram of a switching circuit 200 according to an embodiment of the present invention. Figure 4As shown, the switching circuit 200 includes a first switch QH, a second switch QL, and a switching drive circuit 210. The first switch QH includes a first control terminal CH, a first inverting output terminal ROH, and a first non-inverting output terminal NOH. The second switch QL includes a second control terminal CL, a second inverting output terminal ROL, and a second non-inverting output terminal NOL. The second inverting output terminal ROL and the first inverting output terminal ROH are coupled to a voltage input terminal NDD, and the second non-inverting output terminal NOL and the first non-inverting output terminal NOH are coupled to a voltage output terminal NSS. In other words, the current conduction paths of the first switch QH and the second switch QL (i.e., between the corresponding inverting and non-inverting output terminals) are connected in parallel. The switching drive circuit 210 is coupled to the first control terminal CH and the second control terminal CL. The switching drive circuit 210 generates a first control signal GH and a second control signal GL, thereby controlling the conduction of the first switch QH and the second switch QL, respectively.

[0152] In some embodiments, the layout area of ​​the first switch QH is smaller than the layout area of ​​the second switch QL, wherein the ratio of the layout area of ​​the first switch QH to the layout area of ​​the second switch QL is between 1 / 5 and 1 / 3, that is, the layout area of ​​the second switch QL is 3 to 5 times the layout area of ​​the first switch QH, with preferred embodiments being 3 times, 4 times, and 5 times. The layout area of ​​the switches affects their on-capacitance and parasitic capacitance values. Therefore, when the layout area of ​​the first switch QH is smaller than the layout area of ​​the second switch QL, the on-resistance of the first switch QH is greater than the on-resistance of the second switch QL, and the parasitic capacitance of the first control terminal CH is less than the parasitic capacitance of the second control terminal CL. In some embodiments, the first switch QH and the second switch QL are components manufactured using the same process, for example, both the first switch QH and the second switch QL are MOSFETs or bipolar junction transistors (BJTs), and they have the same process parameters. In some embodiments, when the second switch QL is a metal-oxide-semiconductor (MOSFET), the second control terminal CL corresponds to the gate of the MOSFET, the second inverting output terminal ROL corresponds to the drain of the MOSFET, and the second non-inverting output terminal NOL corresponds to the source of the MOSFET. In some embodiments, the first switch QH and / or the second switch QL of the switching circuit 200 are MOSFETs, wherein the first control terminal CH and / or the second control terminal CL are the gate, the first inverting output terminal ROH and / or the second inverting output terminal ROL are the drain, and the first non-inverting output terminal NOH and / or the second non-inverting output terminal NOL are the source.

[0153] In some embodiments, the architecture of the switching circuit 200 can be applied to switching the capacitor converter 300. Please refer to... Figure 5 , Figure 5This is a schematic diagram of a switching capacitor converter 300 according to an embodiment of the present invention. The switching capacitor converter 300 includes a plurality of switching units, which are used to switch the coupling relationship between a capacitor CFLY and a first power supply VMID and a second power supply VBAT, thereby converting the first power supply VMID (e.g., the intermediate power supply of a USB system) to the second power supply VBAT (e.g., battery voltage), or converting the second power supply VBAT to the first power supply VMID, wherein the voltage of the first power supply VMID is higher than the voltage of the second power supply VBAT.

[0154] In some embodiments, the switching capacitor converter 300 can be used, for example, in a USB PD charging system and / or a USB PD power supply system (i.e., OTG, On-The-Go). In some embodiments, the switching capacitor converter 300 further includes a path switch Q0 for controlling the conduction between a first power supply VMID and a power supply VBUS. In constant current charging mode, the switching capacitor converter 300 can be used to multiply the power supply current IBUS to generate an output current Iout, thereby charging the battery BAT with a higher output current. The switching capacitor converter 300 can also be used in constant voltage charging mode, or in OTG mode, to multiply the battery voltage VBAT to generate a first power supply VMID, thereby outputting the power supply VBUS.

[0155] In some embodiments, the switching capacitor converter 300 includes four switching units: a first switching unit 310, a second switching unit 320, a third switching unit 330, and a fourth switching unit 340, wherein at least one of the first switching unit 310, the second switching unit 320, the third switching unit 330, or the fourth switching unit 340 includes a switching circuit 200. In this embodiment (e.g.) Figure 5 As shown, the first switch unit 310, the second switch unit 320, the third switch unit 330, and the fourth switch unit 340 all include switch circuits 200. The first switch unit 310 includes a first switch Q1H and a second switch Q1L; the second switch unit 320 includes a first switch Q2H and a second switch Q2L; the third switch unit 330 includes a first switch Q3H and a second switch Q3L; and the fourth switch unit 340 includes a first switch Q4H and a second switch Q4L. In this embodiment, the switching drive circuits 210 in each switch circuit 200 are integrated into a single switching drive circuit 350 to generate control signals G0, G1L-G4L, and G1H-G4H to control the corresponding switches. It should be noted that in some embodiments, not all switch units include switch circuits 200 (for example, only one switch unit among multiple switch units may include a switch circuit 200). Users can adjust the number of switch circuits 200 as switch units according to their needs.

[0156] In some embodiments, the switching circuit 200 employs split switches to mitigate the negative impact of the Miller effect caused by the parasitic capacitances of the first switch QH and the second switch QL (the Miller effect is well known to those skilled in the art and will not be elaborated upon), thereby reducing the switching loss of the first switch QH and the second switch QL, and thus reducing the overall loss of the switching capacitor converter 300. The principle of the split switch technology and the operation flow of the switching capacitor converter 300 will be explained in detail below.

[0157] Please refer to further details. Figure 6A , Figure 6B and Figure 7 , Figure 6A and Figure 6B This is a flowchart illustrating the operation of a switching capacitor converter 300 according to an embodiment of the present invention. Figure 7 This is an operational waveform diagram of the switching capacitor converter 300 according to an embodiment of the present invention. Figure 6A and Figure 7 As shown, when the switching capacitor converter 300 is enabled, the switching drive circuit 350 of the switching capacitor converter 300 controls the path switch Q0 to be in the on state via the control signal G0. Figure 6A Step S10, Figure 7 At time point t0, the voltage of the first power supply VMID is equal to the voltage of the power supply VBUS. Next, the switching drive circuit 350 controls the first switch Q1H of the first switching unit 310 and the first switch Q3H of the third switching unit 330 to be in the conducting state through control signals G1H and G3H, respectively, and controls the first switch Q2H of the second switching unit 320 and the first switch Q4H of the fourth switching unit 340 to be in the non-conducting state through control signals G2H and G4H, respectively (step S20, time point t1). After a delay time Td1 (in this embodiment, the delay time Td1 is, for example, less than 30 nanoseconds), the switching drive circuit 350 controls the second switch Q1L of the first switching unit 310 and the second switch Q3L of the third switching unit 330 to be in a conducting state through control signals G1L and G3L, respectively, and controls the second switch Q2L of the second switching unit 320 and the second switch Q4L of the fourth switching unit 340 to be in a non-conducting state through control signals G2L and G4L, respectively (step S30, time point t2).

[0158] When the first switch unit 310 and the third switch unit 330 are in the on state and the second switch unit 320 and the fourth switch unit 340 are in the off state (as shown at time points t1 to t4), the capacitor CFLY and the battery BAT form a series circuit, wherein the capacitor CFLY and the battery BAT are connected in series between the first power supply VMID and the ground potential. At this time, the first power supply VMID will charge the capacitor CFLY and the battery BAT simultaneously (step S40, hereinafter referred to as the first state).

[0159] Continuing step S40 (e.g.) Figure 6B and Figure 7 As shown, the switching drive circuit 350 controls the second switch Q2L of the second switching unit 320 and the second switch Q4L of the fourth switching unit 340 to be in a conducting state via control signals G2L and G4L, respectively, and controls the second switch Q1L of the first switching unit 310 and the second switch Q3L of the third switching unit 330 to be in a non-conducting state via control signals G1L and G3L, respectively (step S50, time point t3). After a lag time Td2 (in this embodiment, the lag time Td2 is, for example, less than 30 nanoseconds), the switching drive circuit 350 controls the first switch Q2H of the second switching unit 320 and the first switch Q4H of the fourth switching unit 340 to be in a conducting state via control signals G2H and G4H, respectively, and controls the first switch Q1H of the first switching unit 310 and the first switch Q3H of the third switching unit 330 to be in a non-conducting state via control signals G1H and G3H, respectively (step S60, time point t4). When the second switch unit 320 and the fourth switch unit 340 are in the on state and the first switch unit 310 and the third switch unit 330 are in the off state (as shown at time point t5 to time point t8), the capacitor CFLY and the battery BAT form a parallel circuit. At this time, the capacitor CFLY will discharge to charge the battery BAT (step S70, hereinafter referred to as the second state).

[0160] In some embodiments, due to the periodic switching of the plurality of switching units, the coupling of capacitor CFLY and battery BAT periodically switches between a first state and a second state. In steady state, according to the principle of capacitive voltage division, the voltage of the first power supply VMID is twice the voltage of the second power supply VBAT. For example, assuming the voltage of the power supply VBUS is 20 volts (V), then the voltage of the first power supply VMID is 20 volts and the voltage of the second power supply VBAT is 10 volts. Furthermore, ignoring losses, the input and output power are approximately equal; therefore, the current of the first power supply VMID is half the current of the second power supply VBAT. For example, assuming the current of the power supply IBUS is 5 amps and the current of the second power supply VBAT is 10 amps. Finally, the switching drive circuit 350 periodically repeats steps S10 to S70 until the switching drive circuit 350 generates a control signal G0 to control the path switch Q0 to be in a non-conducting state, at which point the switching drive circuit 350 stops operating (step S80).

[0161] by Figure 7 Taking the switching process of the first switching unit 310 (corresponding to the aforementioned switching circuit 200) as an example, the principle of the split-switch technology has two key points: First, the switching drive circuit 210 first controls the first switch QH (such as the first switch Q1H of the first switching unit 310) with a small parasitic capacitance value to be in the conducting state (as shown at time point t1) to mitigate the losses caused by the Miller effect. In addition, since the current conduction paths of the first switch QH and the second switch QL (such as the second switch Q1L of the first switching unit 310) are connected in parallel, when the second switch QL then turns on, the voltage difference between the second inverting output terminal ROL and the second non-inverting output terminal NOL (i.e., the switching voltage Vds1L) is less than a preset voltage difference target value. For example, when the second switch Q1L of the first switching unit 310 turns on at time t2, since the first switch Q1H of the first switching unit 310 has already turned on, the switching voltage Vds1H of the first switch Q1H and the switching voltage Vds1L of the second switch Q1L are both 0 volts. Therefore, when the second switch Q1L turns on at time t2, its parasitic capacitance (e.g., the gate-drain capacitance of the second switch Q1L) will not exhibit the Miller effect, thereby making the switching loss of the switching circuit 200 (as corresponding to the first switching unit 310) less than a preset target value. In other words, the Miller effect loss caused by the large parasitic capacitance value of the second switch QL of the switching circuit 200 will be reduced due to the influence of the first switch QH of the switching circuit 200 turning on first. In addition, since the parasitic capacitance value of the first switch QH is small, the loss caused by the Miller effect when the first switch QH turns on first can also be significantly reduced.

[0162] On the other hand, when the second switch Q1L of the first switching unit 310 turns off at time t3, since the first switch Q1H of the first switching unit 310 is still in the on state, the switching voltage Vds1H of the first switch Q1H and the switching voltage Vds1L of the second switch Q1L are both 0 volts. Therefore, when the second switch Q1L turns off at time t3, its parasitic capacitance (e.g., the gate-drain capacitance of the second switch Q1L) will not exhibit the Miller effect. As a result, the switching loss of the switching circuit (as corresponding to the first switching unit 310) is less than a preset target value.

[0163] Furthermore, during the main conduction period (as shown at time points t2 to t3), the split-switch technology uses the switching drive circuit 210 to control the second switch QL, which has a smaller on-resistance value, to be in the conducting state, thus allowing the switching circuit 200 to maintain the advantage of a smaller on-resistance value (resistance parallel principle). In other words, the split-switch technology not only mitigates the negative impact of the Miller effect caused by the parasitic capacitance of the first switch QH and the second switch QL in the switching circuit 200, but also allows the switching circuit 200 to maintain the advantage of a smaller on-resistance value.

[0164] The aforementioned disconnect switch technology allows the second switch QL to avoid the high voltage caused by ringing during switching. Therefore, the second switch QL can employ a transistor with a smaller open-circuit tolerance voltage. In some embodiments, the open-circuit tolerance voltage of the second switch QL is less than that of the first switch QH, where the open-circuit tolerance voltage is defined as the voltage between the inverting and non-inverting output terminals of the switch when the control terminal voltage is zero (i.e., the switch is controlled to be non-conducting). In other words, an open circuit means that when the switch is controlled to be non-conducting, there is an open circuit between its inverting and non-inverting output terminals. In one embodiment, a transistor with a smaller open-circuit tolerance voltage has a smaller on-resistance value per unit area. Therefore, when the second switch QL is implemented with a transistor with a smaller open-circuit tolerance voltage, the switching circuit can have a smaller on-resistance value, further reducing the overall loss of the switching circuit 200.

[0165] In some embodiments, the switching circuit 200 employs non-rectangle safe operating area (SOA) technology of metal-oxide-semiconductor transistors to reduce the on-resistance of the first switch QH and / or the second switch QL, compared to Figure 3 Compared with the prior art with a rectangular safe operating area, the non-rectangular safe operating area technology can reduce the conduction loss of the first switch QH and / or the second switch QL, thereby reducing the overall loss of the switching capacitor converter 300.

[0166] In some embodiments, a non-rectangular safety operating area can be achieved by reducing the area ratio of the body electrode contact point B. Please also refer to... Figure 8A , Figure 8B , Figure 8A and Figure 8B These are schematic diagrams of the layout of metal-oxide-semiconductor transistors according to different embodiments of the present invention, wherein... Figure 8A A gold-oxide-semiconductor transistor with a dual-gate (2-finger) process was revealed. Figure 8B This paper discloses a metal-oxide-semiconductor (MOSFET) with a 10-finger process. Both dual-gate and 10-finger MOSFETs include a gate (G), drain (D), source (S), and a body contact (B). The difference between dual-gate and 10-finger MOSFETs lies in the area ratio of the body contact (B). This area ratio affects the on-resistance of the switch. A larger area ratio of the body contact (B) results in a smaller area ratio of the transistor channel per unit area, thus a larger on-resistance. Conversely, a smaller area ratio of the body contact (B) results in a larger area ratio of the transistor channel per unit area, thus a smaller on-resistance. In this embodiment (e.g.) Figure 8A and Figure 8B As shown in the diagram, in a dual-gate MOSFET, one body contact point B is provided for every two gates; while in a deca-gate MOSFET, one body contact point B is provided for every ten gates. In other words, the area ratio of the body contact point B in a deca-gate MOSFET is smaller than that in a dual-gate MOSFET, meaning the transistor channel area of ​​a deca-gate MOSFET is larger. Therefore, under the same unit area, the on-resistance of a deca-gate MOSFET is smaller than that of a dual-gate MOSFET. Furthermore, for the same reason, the on-resistance values ​​of both the deca-gate and dual-gate MOSFETs are relatively... Figure 2 The on-resistance of the metal-oxide-semiconductor transistor with an annular body electrode contact point B shown is relatively small.

[0167] Please refer to further details. Figure 9 , Figure 9 This is a characteristic diagram of the safe operating area (SOA) of a metal-oxide-semiconductor (MOSFET) according to different embodiments of the present invention, where the solid line L2 is the waveform of the safe operating area of ​​a dual-gate MOSFET, and the dashed line L3 is the waveform of the safe operating area of ​​a deca-gate MOSFET. Figure 9In the characteristic diagram, the horizontal axis represents the switching voltage of the MOSFET (such as the gate-drain voltage Vds), in volts; the vertical axis represents the control voltage of the MOSFET (such as the gate-source voltage Vgs), in volts. Figure 9 As shown, the safe operating regions of both the dual-gate MOSFET and the deca-gate MOSFET are non-rectangular. The safe operating region of the deca-gate MOSFET has an inward bend (as shown in the solid box R1). The safe operating region is used to ensure that the MOSFET can be operated under safe conditions. Taking solid line L2 as an example, when the control voltage of the dual-gate MOSFET is 3 volts, the switching voltage of the dual-gate MOSFET must not exceed approximately 8 volts to avoid the dual-gate MOSFET from collapsing or being damaged. Taking dashed line L3 as an example, when the control voltage of the deca-gate MOSFET is 1 volt, the switching voltage of the deca-gate MOSFET must not exceed approximately 9 volts to avoid the deca-gate MOSFET from collapsing or being damaged.

[0168] In some embodiments, the safe operating area of ​​the first switch QH and / or the second switch QL of the switching circuit 200 is non-rectangular, wherein the first switch QH and / or the second switch QL are deca-gate metal-oxide-semiconductor transistors to reduce the on-resistance value, thereby reducing the conduction loss of the first switch QH and / or the second switch QL, and thus reducing the overall loss of the switching capacitor converter 300.

[0169] In some embodiments, within the same unit area, the on-resistance of a switch having a non-rectangular safe operating area (e.g., a dual-gate metal-oxide-semiconductor or a deca-gate metal-oxide-semiconductor) is less than the on-resistance of another switch having a rectangular safe operating area (e.g., ...). Figure 2 The metal-oxide-semiconductor (MOSFET) having an annular body electrode contact point B, wherein in one embodiment, the above-mentioned on-resistance value relationship is provided that the area of ​​the non-safe operating region is smaller than the area of ​​the rectangular safe operating region.

[0170] Please refer to the following at the same time Figure 10A and Figure 10B , Figure 10A According to an embodiment of the present invention, the control voltage Vgs of the first switch QH (for example, corresponding to...) Figure 5 The control voltage Vgs1H of the first switch Q1H in the first switching unit 310 and the switching voltage Vds (for example, corresponding to...) Figure 5 The operating trajectory diagram (locus) between the first switch Q1H of the first switch unit 310 and the switching voltage Vds1H is shown in the characteristic diagram of its safe operating region (L3). Figure 10BAccording to an embodiment of the present invention, the control voltage Vgs of the second switch QL (for example, corresponding to...) Figure 5 The control voltage Vgs1L of the first switch Q1H in the first switching unit 310 and the switching voltage Vds (for example, corresponding to...) Figure 5 The operating trajectory diagram CV2 between the switching voltage Vds1L of the first switch Q1H in the first switching unit 310 and its characteristic diagram of the safe operating region (L3) is shown. It should be noted that the operating trajectory diagram CV2 is derived from... Figure 5 When the switching capacitor converter 300 performs periodic switching, the characteristic curves are obtained by recording the control voltage and switching voltage of the first switch Q1H and the second switch Q1L of the first switching unit 310 at all time points.

[0171] like Figure 10A As shown, the operating trajectory diagram CV1 between the control voltage and the switching voltage of the first switch QH falls within the safe operating region of the deca-gate metal-oxide-semiconductor transistor; as... Figure 10B As shown, the operating trajectory diagram CV2 between the control voltage and the switching voltage of the second switch QL also falls within the safe operating range of the deca-gate metal-oxide-semiconductor transistor. In other words, when both the first switch QH and the second switch QL use deca-gate metal-oxide-semiconductor transistors, both the first switch QH and the second switch QL can operate safely without any problems of collapse or damage.

[0172] Please refer to further details. Figure 10C , Figure 10C This is a characteristic diagram of the operating trajectory CV2 between the control voltage and the switching voltage of the second switch QL according to another embodiment of the present invention, and its safe operating region (L4). Figure 10C As shown, in some embodiments, since the operating trajectory diagram CV2 between the control voltage and the switching voltage of the second switch QL is very close to the horizontal axis (Vds) and the vertical axis (Vgs), the area of ​​the safe operating region (L4) of the second switch QL can be smaller than the area of ​​the safe operating region (L3) of the first switch QL. Specifically, as Figure 10C As shown, the second switch QL can, for example, be a metal-oxide-semiconductor (e.g., a 15-gate metal-oxide-semiconductor) with a smaller area ratio of the body electrode contact point B, so that the second switch QL has a smaller safe operating area (L4) and an inward fold point closer to 0 (as shown by the dashed line L4 and the solid line box R2), thereby enabling the second switch QL to obtain a lower on-resistance value to further reduce conduction losses.

[0173] In some embodiments, since the second switch QL can be a metal-oxide-semiconductor transistor with a lower area ratio of the body electrode contact point B, or a transistor with a lower open-circuit withstand voltage, the sheet resistance of the second switch QL is less than that of the first switch QH, wherein the sheet resistance is defined as the resistance of the switch per unit area.

[0174] In such Figure 2 The rectangular safe operating region of the present invention does not have an inward fold point. In contrast, in some embodiments, the safe operating region of the transistor (such as the second switch QL) of the present invention has an inward fold point (such as...). Figure 9 (As shown in the solid box R1), the withstand voltage value corresponding to the inner inflection point is less than the open-circuit withstand voltage of the second switch QL itself. As explained above, the smaller the safe operating area, the smaller its plate resistance; therefore, the lower the inflection point withstand voltage value, the smaller the plate resistance. In some embodiments, the inflection point withstand voltage value is 1 / 10 to 1 / 5 times the open-circuit withstand voltage of the second switch QL, that is, the open-circuit withstand voltage of the second switch QL is 5 to 10 times the inflection point withstand voltage value, with preferred embodiments being 5, 6, 7, 8, 9, and 10 times, thereby allowing the second switch QL to have a smaller plate resistance value.

[0175] In some embodiments, the transistor of the present invention having an inward breakpoint has an inward breakpoint control terminal voltage greater than the transistor's turn-on threshold. In some embodiments, since the first switch QH of the switching circuit 200 needs to turn on when its switching voltage is high, or turn off when its switching voltage is low, therefore, as Figure 10A As shown, the control terminal voltage (Vgs) corresponding to the inflection point Rcv1 of its operating trajectory CV1 is higher than the conduction threshold of the first switch QH. In some embodiments, the first switch QH (e.g., Figure 5 The voltage at the breakpoint control terminal of the first switch Q1H in the first switch unit 310 (i.e., Vgs corresponding to the breakpoint shown in solid box R1) in the safe operation area is greater than the conduction threshold of the first switch Q1H.

[0176] Please continue reading. Figure 10B and Figure 10C In some embodiments, because the switching voltage across the second switch QL of the switching circuit 200 is low (e.g., 0 volts) when it turns on, or high when it turns off, the control terminal voltage (Vgs) corresponding to the inflection point Rcv2 of the operating trajectory CV2 is almost equal to 0 volts. Therefore, the second switch QL (e.g., Figure 5 The second switch Q1L of the first switch unit 310 can have the same safe operating area as the first switch QH (e.g., Figure 10BThe L3 transistor, or, in one embodiment, such as Figure 10C As shown, the second switch QL (for example, Figure 5 The second switch Q1L of the first switch unit 310 can have a smaller safe operating range than the first switch QH (e.g., Figure 10C The transistor is L4, therefore, in some embodiments, the second switch QL (e.g., Figure 5 The voltage at the breakpoint control terminal of the safe operation area of ​​the second switch Q1L in the first switching unit 310 (i.e., Vgs corresponding to the breakpoint shown in solid box R2) is less than the conduction threshold of the second switch QL.

[0177] In one embodiment, the second switch QL (e.g., Figure 5 The voltage at the breakpoint control terminal of the second switch Q1L in the first switching unit 310 (i.e., Vgs corresponding to the breakpoint shown in the solid box R2) in the safe operating region is less than that of the first switch QH (e.g., Figure 5 The voltage at the breakpoint control terminal of the safe operating area of ​​the first switch Q1H of the first switch unit 310 (i.e., Vgs corresponding to the breakpoint shown in the solid box R1).

[0178] In some embodiments, the switching circuit 200 employs clamping circuit technology to limit the voltage difference between the first inverting output terminal ROH and the first non-inverting output terminal NOH, and / or between the second inverting output terminal ROL and the second non-inverting output terminal NOL. This allows for a reduction in the process dimensions of the first switch QH and / or the second switch QL, thereby reducing the on-resistance of the first switch QH and / or the second switch QL. In other words, in some embodiments, the switching circuit 200 further includes a clamp, which can indirectly reduce the conduction losses of the first switch QH and / or the second switch QL, thereby reducing the overall loss of the switching capacitor converter 300. In some embodiments, the clamp may include, for example, a Zener diode, or a diode-coupled transistor to achieve the clamping effect. Clamps are well known to those skilled in the art and will not be described in detail here.

[0179] Please refer to Figure 11 , Figure 11 This is a schematic diagram of a switching capacitor converter 300 with a clamp according to an embodiment of the present invention. Figure 11As shown, the switching capacitor converter 300 also includes a clamp 260, which is coupled between the first inverting output terminal and the first control terminal of the first switch Q1H in the first switching unit 310. It should be noted that in other embodiments, the clamp 260 may also be coupled between the first inverting output terminal and the first control terminal of the first switch in other switching units (for example, coupling the clamp 260 between the first inverting output terminal and the first control terminal of the first switch Q2H in the second switching unit 320). The user can adjust the position of the clamp 260 as needed.

[0180] In some embodiments, clamp 260 is used to clamp the voltage difference between the first inverting output terminal and the first control terminal of the first switch Q1H in the first switching unit 310 below a clamping limit, such that the voltage difference between the first inverting output terminal and the first non-inverting output terminal of the first switch Q1H does not exceed the absolute maximum rated value of the first switch Q1H, wherein the clamping limit is less than the absolute maximum rated value of the first switch Q1H. In other words, because clamp 260 limits the voltage value of the first switch Q1H during operation, the process size of the first switch Q1H in the first switching unit 310 can be reduced, thereby reducing the on-resistance value of the first switch Q1H. Therefore, clamp technology can effectively reduce the overall loss of the switching capacitor converter 300.

[0181] Please refer to Figure 12 , Figure 12 This is a schematic diagram of a switching capacitor converter 300 with a clamp 360 according to another embodiment of the present invention. Figure 12 As shown, the switching capacitor converter 300 further includes a clamp 360, which is coupled between the second inverting output terminal and the second control terminal of the second switch Q1L in the first switching unit 310. In some embodiments, the switching circuit 200 further includes a current source Is, which is used to bias the clamp. Similarly, in other embodiments, the clamp 360 may also be coupled between the second inverting output terminal and the second control terminal of the second switch in other switching units (for example, coupling the clamp 360 between the second inverting output terminal and the second control terminal of the second switch Q2L in the second switching unit 320), and the user can adjust the position of the clamp 360 as needed.

[0182] In some embodiments, clamp 360 is used to clamp the voltage difference between the first inverting output terminal and the first control terminal of the second switch Q1L in the first switching unit 310 below a clamping limit, such that the voltage difference between the first inverting output terminal and the first non-inverting output terminal of the second switch Q1L does not exceed the absolute maximum rated value of the second switch Q1L, wherein the clamping limit is less than the absolute maximum rated value of the second switch Q1L. In other words, because clamp 360 limits the voltage value of the second switch Q1L during operation, the manufacturing size of the second switch Q1L in the first switching unit 310 can be reduced, thereby reducing the on-resistance value of the second switch Q1L. Therefore, clamp technology can effectively reduce the overall loss of the switching capacitor converter 300.

[0183] Please refer to Figure 13 , Figure 13 This is an efficiency curve diagram corresponding to the output current Iout in the switching capacitor converters 100 and 300 according to some embodiments of the present invention. The solid line L5 is the efficiency curve diagram of the prior art switching capacitor converter 100; the dashed line L6 is the efficiency curve diagram of the switching capacitor converter 300 using only split-switch technology; the single-point link line L7 is the efficiency curve diagram of the switching capacitor converter 300 using both split-switch technology and non-rectangular safe operating area technology; and the two-point link line L8 is the efficiency curve diagram of the switching capacitor converter 300 using both split-switch technology, non-rectangular safe operating area technology, and clamping technology. Figure 13 In the efficiency curve graph, the horizontal axis represents the output current Iout of the switching capacitor converter 300, in amperes; the vertical axis represents the power conversion efficiency of the switching capacitor converter 300, in percentage (%). Figure 13 As shown in the solid-line box R3, the split-switch technology effectively improves the charging efficiency of the switching capacitor converter 300 under light loads by reducing switching losses (in this embodiment, the output current Iout under light load is, for example, 1 ampere); and so on. Figure 13 As shown in the solid box R4, the non-rectangular safe operating area technology and clamp technology effectively improve the charging efficiency of the switching capacitor converter 300 under heavy loads by reducing conduction losses (in this embodiment, the output current Iout of the heavy load is, for example, 5 amps). When the switching capacitor converter 300 simultaneously employs the split-switch technology, the non-rectangular safe operating area technology, and the clamp technology (as shown by the two-point link L8), the charging efficiency of the switching capacitor converter 300 is improved under any load. Furthermore, in some embodiments, the non-rectangular safe operating area technology or clamp technology can also effectively reduce the temperature of the switching capacitor converter 300 when operating under heavy loads by reducing conduction losses.

[0184] In summary, when the switching circuit 200 and / or the switching capacitor converter 300 in some embodiments of the present invention are applied to various types of chargers, they can not only effectively improve the charging efficiency of the charger under different loads, but also effectively reduce the temperature of the charger during fast charging, so as to avoid the charger overheating and causing danger.

[0185] The present invention has been described above with reference to preferred embodiments. However, the above description is only intended to facilitate understanding of the invention by those skilled in the art and is not intended to limit the broadest scope of the invention. The described embodiments are not limited to individual application and can also be used in combination. For example, two or more embodiments can be used in combination, and some components of one embodiment can be used to replace corresponding components in another embodiment. Furthermore, within the same spirit of the invention, those skilled in the art can conceive of various equivalent changes and combinations. For example, the phrase "processing or calculating based on a signal or generating an output result" in the present invention is not limited to the signal itself, but also includes, when necessary, performing voltage-to-current conversion, current-to-voltage conversion, and / or proportional conversion on the signal, and then processing or calculating based on the converted signal to generate an output result. Therefore, within the same spirit of the invention, those skilled in the art can conceive of various equivalent changes and combinations, and there are many combinations, which will not be listed here. Therefore, the scope of the present invention should cover the above and all other equivalent changes.

Claims

1. A switching circuit, characterized by, The switching circuit comprises: a first switch comprising a first control terminal, a first inverting output terminal, and a first non-inverting output terminal; a second switch comprising a second control terminal, a second inverting output terminal, and a second non-inverting output terminal, the second inverting output terminal being coupled to the first inverting output terminal, and the second non-inverting output terminal being coupled to the first non-inverting output terminal, the first switch having a conduction resistance value greater than a conduction resistance value of the second switch, and the first control terminal having a parasitic capacitance value less than a parasitic capacitance value of the second control terminal; a switching driving circuit configured to generate a first control signal and a second control signal to control the first switch and the second switch, respectively; wherein the switching driving circuit controls the first switch to turn on before the second switch turns on and / or controls the first switch to turn off after the second switch turns off, and when the second switch turns on and / or turns off, a voltage difference between the second inverting output terminal and the second non-inverting output terminal is less than a preset voltage difference target value, so that a switching loss of the switching circuit is less than a preset target value; wherein the second switch has an open circuit voltage resistance less than an open circuit voltage resistance of the first switch. The first switch has a layout area less than a layout area of the second switch.

2. The switching circuit of claim 1, wherein, The first switch has a layout area to layout area ratio of the second switch in a range of 1 / 5 to 1 / 3.

3. The switching circuit of claim 2, wherein, The second switch is a metal-oxide-semiconductor transistor, wherein the second control terminal is a gate, the second inverting output terminal is a drain, and the second non-inverting output terminal is a source.

4. The switching circuit of claim 1, wherein, The second switch has a safe operating area range less than a safe operating area range of the first switch.

5. The switching circuit of claim 1, wherein, The second switch has a sheet resistance value less than a sheet resistance value of the first switch.

6. The switching circuit of claim 5, wherein, The second switch has a body contact area ratio less than a body contact area ratio of the first switch, so that the second switch has a sheet resistance value less than a sheet resistance value of the first switch.

7. The switching circuit of claim 6, wherein, The safe operating area of the first switch is non-rectangular.

8. The switching circuit of claim 5, wherein, The safe operating area of the second switch is non-rectangular.

9. The switching circuit of claim 5, wherein, The safe operating area of the second switch has an inflection point, wherein an inflection point voltage corresponding to the inflection point is less than the open circuit voltage resistance of the second switch.

10. The switching circuit of claim 9, wherein, The inflection point voltage corresponding to the inflection point is 1 / 10 to 1 / 5 times the open circuit voltage resistance of the second switch.

11. The switching circuit of claim 10, wherein, A voltage of the second control terminal corresponding to the inflection point of the second switch is greater than a conduction threshold value of the second switch.

12. The switching circuit of claim 10, wherein, The safe operating area of the first switch has an inflection point, wherein a voltage of the second control terminal corresponding to the inflection point of the safe operating area of the second switch is less than a voltage of the first control terminal corresponding to the inflection point of the safe operating area of the first switch.

13. The switching circuit of claim 10, wherein, The switching circuit further comprises a clamp coupled between the first inverting output terminal and the first control terminal to clamp a voltage difference between the first inverting output terminal and the first control terminal to be less than a clamping limit value, so that a voltage difference between the first inverting output terminal and the first non-inverting output terminal does not exceed an absolute maximum rating of the first switch, wherein the clamping limit value is less than the absolute maximum rating of the first switch.

14. The switching circuit of claim 1, wherein, ​ 15. The switching circuit of claim 14, wherein, The switching circuit further comprises a current source coupled between the first control terminal and the first non-inverting output terminal for biasing the clamp.

16. The switching circuit of claim 1, wherein, The switching circuit further comprises a clamp coupled between the second non-inverting output terminal and the second control terminal for clamping a voltage difference between the second non-inverting output terminal and the second control terminal below a clamp limit, such that a voltage difference between the second non-inverting output terminal and the second non-inverting output terminal does not exceed an absolute maximum rating of the second switch, wherein the clamp limit is less than the absolute maximum rating of the second switch.

17. The switching circuit of claim 16, wherein, The switching circuit further comprises a current source coupled between the second control terminal and the second non-inverting output terminal for biasing the clamp.

18. The switching circuit of any one of claims 1 to 17, wherein, The switching circuit is suitable for a switched-capacitor converter for switching a coupling relationship between a capacitor and a first power source and a second power source, thereby converting the first power source to the second power source or converting the second power source to the first power source, wherein a voltage of the first power source is higher than a voltage of the second power source.

19. The switching circuit of claim 18, wherein, The switched-capacitor converter comprises a first switch unit, a second switch unit, a third switch unit, and a fourth switch unit, wherein at least one of the first switch unit, the second switch unit, the third switch unit, or the fourth switch unit comprises the switching circuit, the first switch unit is coupled between the first power source and a first terminal of the capacitor, the second switch unit is coupled between the first terminal of the capacitor and the second power source, the third switch unit is coupled between the second power source and a second terminal of the capacitor, and the fourth switch unit is coupled between the second terminal of the capacitor and a ground potential, the first switch unit, the second switch unit, the third switch unit, and the fourth switch unit are used to periodically switch the first terminal of the capacitor between the first power source and the second power source, and correspondingly switch the second terminal of the capacitor between the second power source and the ground potential, thereby converting the first power source to the second power source or converting the second power source to the first power source, wherein a voltage of the first power source is twice a voltage of the second power source, and a current of the first power source is half a current of the second power source.

20. A switched-capacitor converter, comprising: The switching circuit further comprises a current source coupled between the first control terminal and the first non-inverting output terminal for biasing the clamp. The switching circuit is suitable for a switched-capacitor converter for switching a coupling relationship between a capacitor and a first power source and a second power source, thereby converting the first power source to the second power source or converting the second power source to the first power source, wherein a voltage of the first power source is higher than a voltage of the second power source. The switching circuit further comprises a current source coupled between the first control terminal and the first non-inverting output terminal for biasing the clamp. The switching circuit is suitable for a switched-capacitor converter for switching a coupling relationship between a capacitor and a first power source and a second power source, thereby converting the first power source to the second power source or converting the second power source to the first power source, wherein a voltage of the first power source is higher than a voltage of the second power source. The switching circuit further comprises a current source coupled between the first control terminal and the first non-inverting output terminal for biasing the clamp. The switching circuit is suitable for a switched-capacitor converter for switching a coupling relationship between a capacitor and a first power source and a second power source, thereby converting the first power source to the second power source or converting the second power source to the first power source, wherein a voltage of the first power source is higher than a voltage of the second power source. The switching circuit further comprises a current source coupled between the first control terminal and the first non-inverting output terminal for biasing the clamp. A switching driving circuit is configured to generate a first control signal and a second control signal to control the first switch and the second switch, respectively. The switching driving circuit controls the first switch to be turned on before the second switch and / or controls the first switch to be turned off after the second switch, and a voltage difference between the second inverting output terminal and the second non-inverting output terminal is less than a preset voltage difference target value when the second switch is turned on and / or turned off, so that a switching loss of the switching circuit is less than a preset target value.

21. The switched-capacitor converter of claim 20, wherein, The switching units include a first switching unit, a second switching unit, a third switching unit, and a fourth switching unit. At least one of the first switching unit, the second switching unit, the third switching unit, or the fourth switching unit includes the switching circuit. The first switching unit is coupled between the first power supply and a first terminal of the capacitor. The second switching unit is coupled between the first terminal of the capacitor and the second power supply. The third switching unit is coupled between the second power supply and a second terminal of the capacitor. The fourth switching unit is coupled between the second terminal of the capacitor and a ground potential. The first switching unit, the second switching unit, the third switching unit, and the fourth switching unit periodically switch the first terminal of the capacitor between the first power supply and the second power supply, and correspondingly switch the second terminal of the capacitor between the second power supply and the ground potential, thereby converting the first power supply to the second power supply or converting the second power supply to the first power supply. The voltage of the first power supply is twice the voltage of the second power supply, and the current of the first power supply is half the current of the second power supply.

22. The switched-capacitor converter of claim 21, wherein, The open circuit voltage resistance of the second switch is less than the open circuit voltage resistance of the first switch.

23. The switched-capacitor converter of claim 22, wherein, The safe operating area of the second switch is less than the safe operating area of the first switch.

24. The switched-capacitor converter of claim 23, wherein, The sheet resistance of the second switch is less than the sheet resistance of the first switch.

25. The switched-capacitor converter of claim 24, wherein, The area ratio of the body contact point of the second switch is lower than the body contact point of the first switch, so that the sheet resistance of the second switch is less than the sheet resistance of the first switch.

26. The switched-capacitor converter of claim 23, wherein, The safe operating area of the first switch is non-rectangular.

27. The switched-capacitor converter of claim 23, wherein, The safe operating area of the second switch is non-rectangular.

28. The switched-capacitor converter of claim 27, wherein, The safe operating area of the second switch has an inflection point, wherein a fold point voltage resistance value corresponding to the inflection point is less than the open circuit voltage resistance of the second switch.

29. The switched-capacitor converter of claim 28, wherein, The fold point voltage resistance value corresponding to the inflection point is 1 / 10 to 1 / 5 times the open circuit voltage resistance of the second switch.

30. The switched-capacitor converter of claim 28, wherein, The voltage of the second control terminal corresponding to the inflection point of the second switch is greater than the turn-on threshold of the second switch.

31. The switched-capacitor converter of claim 28, wherein, The safe operating area of the first switch has an inflection point, wherein the voltage of the second control terminal corresponding to the inflection point of the safe operating area of the second switch is less than the voltage of the first control terminal corresponding to the inflection point of the safe operating area of the first switch.

32. The switched-capacitor converter of any of claims 20-31, wherein, The clamp is coupled between the first inverting output and the first control terminal to clamp a voltage difference between the first inverting output and the first control terminal below a clamp limit, such that a voltage difference between the first inverting output and the first non-inverting output does not exceed an absolute maximum rating of the first switch, wherein the clamp limit is less than the absolute maximum rating of the first switch.

33. The switched-capacitor converter of claim 32, wherein, The clamp is coupled between the second inverting output and the second control terminal to clamp a voltage difference between the second inverting output and the second control terminal below a clamp limit, such that a voltage difference between the second inverting output and the second non-inverting output does not exceed an absolute maximum rating of the second switch, wherein the clamp limit is less than the absolute maximum rating of the second switch.

34. The switched-capacitor converter of any one of claims 20-31, wherein, The clamp is coupled between the second inverting output and the second control terminal to clamp a voltage difference between the second inverting output and the second control terminal below a clamp limit, such that a voltage difference between the second inverting output and the second non-inverting output does not exceed an absolute maximum rating of the second switch, wherein the clamp limit is less than the absolute maximum rating of the second switch.

35. The switched-capacitor converter of claim 34, wherein, The clamp is coupled between the second inverting output and the second control terminal to clamp a voltage difference between the second inverting output and the second control terminal below a clamp limit, such that a voltage difference between the second inverting output and the second non-inverting output does not exceed an absolute maximum rating of the second switch, wherein the clamp limit is less than the absolute maximum rating of the second switch.

36. A switch for switching a coupling relationship between a capacitor and a first power source and a second power source, thereby converting the first power source to the second power source or converting the second power source to the first power source, characterized by, The switch has a non-rectangular safe operating area, a control voltage between the control terminal and the non-inverting output, and a switch cross voltage between the inverting output and the non-inverting output, wherein a corresponding operating locus between the control voltage and the switch cross voltage when the switch is used to switch the capacitor for power conversion is limited in the non-rectangular safe operating area.

37. The switch of claim 36, wherein, The switch is a metal-oxide-semiconductor transistor, the control terminal is a gate, the inverting output is a drain, and the non-inverting output is a source.

38. The switch of claim 36, wherein, The non-rectangular safe operating area has an inflection point.

39. The switch of claim 38, wherein, A breakdown voltage value corresponding to the inflection point is less than an open circuit breakdown voltage of the switch.

40. The switch of claim 39, wherein, The breakdown voltage value corresponding to the inflection point is 1 / 10 to 1 / 5 times the open circuit breakdown voltage of the switch.

41. The switch of claim 38, wherein, A breakdown control terminal voltage corresponding to the inflection point is greater than a turn-on threshold of the switch.

42. The switch of claim 38, wherein, A breakdown control terminal voltage corresponding to the inflection point is less than a turn-on threshold of the switch.

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