A virtual capacitor control method for a bidirectional DC-DC converter

Abstract

The application relates to a virtual capacitor control method of a bidirectional DC-DC converter, mainly comprising four parts of I-U droop control, virtual capacitor control, voltage control and current control; the droop control can realize voltage-current droop function, is beneficial to current distribution when multiple machines of the bidirectional DC-DC converter are interconnected; the virtual capacitor control can make the bidirectional DC-DC converter have greater inertia; the voltage control comprises a leading element and proportional-integral control, the leading element can provide greater damping, reduce the influence of sudden change of a DC side load on output voltage, improve the dynamic performance of the converter, and the proportional-integral control can follow the voltage instruction value without static error; the current loop adopts proportional-integral control, and can follow the current instruction value without static error. The application makes the bidirectional DC-DC converter have stronger inertia and damping, can suppress output voltage fluctuation caused by load change, and improves the dynamic performance of the converter.

Description

TECHNICAL FIELD

[0001] The present application relates to the field of bidirectional DC-DC converter control, in particular to a virtual capacitor control method of bidirectional DC-DC converter. BACKGROUND

[0002] With the development of photovoltaic, energy storage, wind power and other distributed generation technologies, and the increasing proportion of DC loads in terminal power consumption, the development of DC microgrid is promoted. However, in the DC microgrid, all distributed power sources, energy storage elements and DC loads are connected through power electronic devices, which are different from traditional motors and are non-rotating static devices, so they do not have the rotational inertia and damping characteristics of traditional motors, thus showing low inertia characteristics. Since the output power of the distributed power source has intermittency and volatility with the change of its own environment, the load is also fluctuating, especially the load fluctuation has obvious randomness. Since the DC microgrid does not have rotating machines and other inertia devices, the droop control adopted does not provide inertia support for the droop characteristics, and cannot effectively respond to the voltage fluctuation caused by power deviation. The temporary power difference between the power source and the load can easily cause significant fluctuations in the DC bus voltage. Especially when the DC load power fluctuates, the instantaneous voltage impact and fluctuation have a great impact on the DC output voltage, which is not conducive to the stable operation of the system.

[0003] To solve the above problems, domestic and foreign researchers have proposed some improvement methods, such as: literature 1(Y. Wang, C. Wang, L. Xu, J. Meng and Y. Hei, "Adjustable Inertial Response From the Converter With Adaptive Droop Control in DC Grids," in IEEE Transactions on Smart Grid, vol. 10, no. 3, pp. 3198-3209, May 2019, doi: 10.1109 / TSG.2018.2820160.) proposed a variable droop coefficient control method, which links the DC bus voltage change rate to the droop coefficient based on the DC droop control, so that the droop curve changes in response to voltage changes; Literature 2(Shi N, Zhang H, Xiao X. Improved droop control strategy to improve the dynamic characteristics of DC microgrid [J]. Transactions of Electrical Engineering, 2016, 31(03): 31-39. DOI: 10.19595 / j.cnki.1000-6753.tces.2016.03.005.) added damping characteristics and improved the dynamic performance of the bus voltage by connecting a high-pass filter in parallel with the voltage droop control loop. However, variable droop coefficient control and parallel high-pass filter control will produce a certain degree of voltage mutation at the moment of disturbance, so the control method needs to be further improved. Literature 3(S. Sheng, H. Liu, Z. Zeng, Z. Lv, Q. Tan, Q. Duan, L. Ran, "An energy router based on virtual motor control," Proceedings of the CSEE, vol. 35, no. 14, pp. 3541-3550, 2015, doi: 10.13334 / j.0258-8013.pcsee.2015.14.008.) proposed a virtual DC motor control method that can simulate the inertia and damping characteristics of traditional motors in DC systems; Literature 4(Cui J, Lv Z, Sheng W, Wu M, Wang J, Zhang W, Fan S. A new virtual DC motor control technology [J]. Proceedings of the CSEE, 2019, 39(10): 3029-3038. DOI: 10.13334 / j.0258-8013.pcsee.180782.) estimated the load disturbance through an extended state observer and introduced it as a compensation term into the virtual DC motor control. The derivation principle and control method of the above two methods are complex and difficult to apply in practice.Reference 5 (N. Zhi, X. Ming, Y. Ding, L. Du and H. Zhang, "Power-Loop-Free Virtual DC Machine Control With Differential Compensation," in IEEE Transactions on Industry Applications, vol. 58, no. 1, pp. 413-422, Jan.-Feb. 2022, doi: 10.1109 / TIA.2021.3119512.) adds differential compensation to virtual DC motor control. Although this can improve the dynamic performance of the DC bus voltage, the implementation of an ideal differential compensation circuit is difficult, and the differential circuit can lead to a decrease in noise immunity and amplification of high-frequency noise.

[0004] In summary, although the existing technologies mentioned above can improve the dynamic performance of the output voltage of DC microgrids, they still have their own shortcomings. The present invention improves the dynamic performance of bidirectional DC-DC converters based on the virtual capacitor control method, which is simple and easy to implement. Summary of the Invention

[0005] This invention improves upon the traditional IU droop control method by simulating inertia characteristics through virtual capacitor control after the bidirectional DC-DC converter, and by adding a lead element in series at the outer voltage loop to increase damping, thereby enhancing the dynamic performance of output voltage changes when the load changes.

[0006] To achieve the above objectives, the technical solution adopted by the present invention is as follows:

[0007] A virtual capacitor control method for a bidirectional DC-DC converter includes IU droop control, virtual capacitor control, voltage control, and current control;

[0008] Construct a bidirectional DC-DC converter topology;

[0009] Collect the inductor current i on the power supply side of the converter respectively b Load side current i dc DC bus voltage on the load side u dc ;

[0010] The input current i is calculated using IU droop control. in , will current i in Input the virtual capacitor control section and calculate the load-side bus voltage command value u. dc * ;

[0011] Voltage control includes a lead element and proportional-integral control, which will control u dc* The input voltage control section calculates the load-side current command value i. dc * ;

[0012] The current control uses proportional-integral control, which controls i dc * The input current control section outputs to the PWM pulse signal generator, and the generated pulse signal controls the switching on and off of the switching transistors in the bidirectional DC-DC converter.

[0013] Furthermore, the current i in The expression is as follows:

[0014] i in =(u ref -u dc )K d (1)

[0015] In the above formula, u ref K is the reference value for the DC bus voltage. d This is the voltage-current droop factor;

[0016] The u dc * The expression is as follows:

[0017]

[0018] In the above formula, C v is the virtual capacitance value connected in parallel on the load side, and s is the Laplace operator;

[0019] The i dc * The expression is as follows:

[0020]

[0021] In the above formula, T1 and T2 are the time constants of the lead element, and satisfy T1≥T2, k vp k is the proportional gain of the voltage loop PI controller. vi The integral coefficient of the voltage loop PI controller;

[0022] The expression for the current control part is as follows:

[0023]

[0024] In the above formula, D pwm For the modulated signal, U dc U is the steady-state voltage value of the load-side bus. b k is the steady-state voltage value of the power supply. ip k is the proportional gain of the current loop PI controller.ii is the integral coefficient of the current loop PI controller.

[0025] Furthermore, the small-signal model for the virtual capacitor control of the bidirectional DC-DC converter is as follows:

[0026]

[0027]

[0028] In the above formula, Δu b , Δi b Δd and Δd are the power supply voltage u, respectively. b Inductor current i b The steady-state values ​​corresponding to the disturbances with duty cycle d are U and d, respectively. b I b and D; Δu dc and Δi dc The DC bus voltage u on the load side is respectively dc and load-side current i dc The disturbances, and their corresponding steady-state values ​​are U dc and I dc L is the inductance value on the power supply side, R is the parasitic resistance value of the inductance on the power supply side, and C is the capacitance value on the load side.

[0029] Furthermore, the small-signal model for the virtual capacitor control of the bidirectional DC-DC converter is obtained by linearizing the average value model of the bidirectional DC-DC converter. The average value model of the bidirectional DC-DC converter is expressed as follows:

[0030]

[0031]

[0032] Linearizing equations (7) and (8) respectively yields equations (5) and (6).

[0033] Furthermore, the time constant of the lead element in the voltage control is determined based on the stability criterion and impedance ratio criterion of the power supply subsystem. Let the small-signal closed-loop transfer function of the small-signal model of the virtual capacitor control of the bidirectional DC-DC converter be G. If G has no poles in the right half-plane, it indicates that the power supply subsystem is stable. The impedance ratio T0 is used to judge the stability of the cascaded system. If the Nyquist curve of the impedance ratio does not enclose (-1,0), the cascaded system is stable; otherwise, the cascaded system is in an unstable state.

[0034] Furthermore, the method for obtaining the small-signal closed-loop transfer function G is as follows:

[0035] Applying the Laplace transform to equations (5) and (6) respectively, we obtain the following equation:

[0036]

[0037]

[0038]

[0039]

[0040] Ignoring energy losses, the power balance relationship between the power supply side and the load side of the bidirectional DC-DC converter is as follows:

[0041] u b i b =u dc i dc (5)

[0042] Linearizing equation (13) and ignoring higher-order perturbation terms, we get:

[0043]

[0044] The transfer function of the voltage control section is:

[0045]

[0046] The transfer function of the current control section is:

[0047]

[0048] Therefore, the small-signal closed-loop transfer function G is derived as follows:

[0049]

[0050] The impedance ratio T0 is obtained as follows:

[0051] Output impedance Z on the power supply side bat =G;

[0052] Assuming the load side is connected to the bidirectional DC-DC converter via a series inductor and a parallel capacitor, then the load side input impedance Z L for:

[0053]

[0054] In the above formula, R L L is the parasitic resistance value of the line inductance. load C is the line inductance value. load R is the line capacitance value. load This is the load resistance value;

[0055] Through loop gain T0 = Z bat / Z L To determine the stability of a cascaded system.

[0056] Furthermore, the specific method for determining the time constant of the lead element in the voltage control is as follows:

[0057] Step 1: Select a time constant T2 and input it into the system;

[0058] Step 2: Preset a time constant T1, satisfying T1≥T2;

[0059] Step 3: Solve for the distribution of poles of the small-signal closed-loop transfer function G, and plot the Nyquist curve of T0;

[0060] Step 4: If G has no poles in the right half-plane and the Nyquist curve of T0 does not enclose (-1,0), then proceed to step 5. If the above conditions are not met, then proceed to step 6.

[0061] Step 5: Let T1 = T1 + ΔT, where ΔT is the iteration step size of T1, and then return to step 3;

[0062] Step 6: Output the critical value T1.

[0063] Furthermore, in step 2, during the initial stage of converter startup, T1 = T2 is preset.

[0064] This invention utilizes the ability of capacitors to prevent sudden changes in DC voltage to simulate the effect of a virtual capacitor in the control of a bidirectional DC-DC converter. Furthermore, an advance element is added to the voltage control to further enhance the system's damping. This invention enables the bidirectional DC-DC converter to have strong inertia and damping, effectively suppressing large fluctuations in output voltage caused by load changes, improving the converter's dynamic performance, and is easy to implement. Attached Figure Description

[0065] Figure 1 This is a schematic diagram of the virtual capacitor control principle of the bidirectional DC-DC converter in the embodiment;

[0066] Figure 2 for Figure 1 Virtual capacitor control block diagram of the circuit;

[0067] Figure 3 The diagram shows a comparison of the simulated DC bus voltage output waveforms under load fluctuations using the virtual capacitor control method of this invention and the traditional droop control method. Detailed Implementation

[0068] The technical solutions of the present invention will be clearly and completely described below with reference to the accompanying drawings in the embodiments of the present invention.

[0069] This embodiment discloses a virtual capacitor control method for a bidirectional DC-DC converter. The bidirectional DC-DC converter mentioned here refers to... Figure 1 The bidirectional Buck / Boost converter shown is Figure 1 in, u b Where L is the power supply voltage, R is the power supply inductance, C is the load-side capacitance, and 1 and 2 are two switching transistors. b i is the inductor current on the power supply side. in i is the input current before the parallel virtual capacitor. dc For the load-side current, C v The virtual capacitor is connected in parallel on the load side, Δi is the current disturbance flowing through the virtual capacitor, and u dc This is the DC bus voltage on the load side.

[0070] This invention is aimed at Figure 1 The topology shown adopts Figure 2 The control method shown employs virtual capacitor control, which can be divided into four parts: IU droop control, virtual capacitor control, voltage control, and current control. IU droop control enables voltage-current droop, facilitating current distribution in multi-machine interconnection of bidirectional DC-DC converters. Virtual capacitor control provides the bidirectional DC-DC converter with greater inertia. More importantly, in voltage control, in addition to proportional-integral (PI) control, this invention introduces a lead element. The lead element provides significant damping, reducing the impact of sudden load changes on the output voltage and improving the converter's dynamic performance. PI control can follow the voltage command value without static error. Current control also employs PI control, enabling it to follow the current command value without static error.

[0071] Specifically, such as Figure 2 As shown, the virtual control method of the present invention is as follows:

[0072] (1) First, the inductor current i on the power supply side of the converter is collected by the sensor. b Load side current i dc DC bus voltage on the load side u dc ;

[0073] (2) Then the input current i is calculated through the IU droop control section. in ;

[0074] (3) The current i in Input the virtual capacitor control section and calculate the load-side bus voltage command value u. dc* ;

[0075] (4) will u dc * The input voltage control section calculates the load-side current command value i. dc * ;

[0076] (5) Change i dc * The input current control section outputs to the PWM pulse signal generator, and the generated pulse signal controls the switching on and off of the switching transistors in the bidirectional DC-DC converter.

[0077] The current-voltage relationship in traditional IU droop control is as follows:

[0078] i in =(u ref -u dc )K d (1)

[0079] In the above formula, u ref K is the reference value for the DC bus voltage. d This is the voltage-current droop coefficient.

[0080] This invention simulates connecting a capacitor in parallel on the load side after the bidirectional DC-DC converter (e.g., Figure 1 (As shown in the dashed box), the inertia of the bidirectional DC-DC converter is improved by utilizing the characteristic of capacitors to prevent sudden changes in DC voltage. Based on the relationship between the voltage and current of the virtual capacitor, the following equation can be obtained in the virtual capacitor control section:

[0081]

[0082] In the above formula, C v Let be the virtual capacitance value connected in parallel on the load side, and s be the Laplace operator.

[0083] The voltage loop expression for an existing bidirectional DC-DC converter using IU droop control is as follows:

[0084]

[0085] This invention simulates inertia characteristics through virtual capacitor control. To ensure the bidirectional DC-DC converter also possesses damping characteristics, according to automatic control principles, a lead element can increase damping and improve the system's dynamic characteristics. However, the voltage outer loop of traditional IU droop control provides a negative damping effect. Therefore, this invention connects a lead element in series at the voltage outer loop of the voltage control section to improve dynamic performance. Thus, the expression for the voltage control section of this invention is as follows:

[0086]

[0087] In the above formula, T1 and T2 are the time constants of the lead element, and satisfy T1≥T2, k vp k is the proportional gain of the voltage loop PI controller. vi This represents the integral coefficient of the voltage loop PI controller.

[0088] The expression for the current control part of this invention is as follows:

[0089]

[0090] In the above formula, D pwm For the modulated signal, U dc U is the steady-state voltage value of the load-side bus. b k is the steady-state voltage value of the power supply. ip k is the proportional gain of the current loop PI controller. ii is the integral coefficient of the current loop PI controller.

[0091] The small-signal model established in the virtual capacitor control-based method described above is as follows:

[0092]

[0093]

[0094] In the above formula, Δu b , Δi b Δd and Δd are the power supply voltage u, respectively. b Inductor current i b The steady-state values ​​corresponding to the disturbances with duty cycle d are U and d, respectively. b I b and D; Δu dc and Δi dc The DC bus voltage u on the load side is respectively dc and load-side current i dc The disturbances, and their corresponding steady-state values ​​are U dc and I dc L is the inductance value on the power supply side, R is the parasitic resistance value of the inductance on the power supply side, and C is the capacitance value on the load side.

[0095] The small-signal model given above can be obtained by linearizing the average value model of the bidirectional DC-DC converter represented by equations (7) and (8):

[0096]

[0097]

[0098] To further explain, the time constants of the voltage control lead element in this invention, namely T1 and T2 in formula (3), are determined based on the stability criterion and impedance ratio criterion of the power supply subsystem. The impedance ratio criterion presupposes that each subsystem in the cascaded system is stable; therefore, the time constant design of the lead element should be based on the stability judgment of the power supply subsystem and the impedance ratio criterion. The determination of whether the power supply subsystem is stable mainly depends on whether the small-signal closed-loop transfer function G of the small-signal model has a pole in the right half-plane. If no pole in the right half-plane exists, it indicates that the power supply subsystem is stable. From the impedance ratio criterion, it can be seen that the loop gain T0 = Z... bat / Z L The stability of a cascaded system can be determined by the following: if the Nyquist curve of T0 does not enclose (-1,0), the cascaded system is stable; otherwise, the cascaded system is in an unstable state.

[0099] Specifically, the small-signal closed-loop transfer function G is obtained as follows:

[0100] Applying the Laplace transform to equations (5) and (6) respectively, we obtain the following equation:

[0101]

[0102]

[0103]

[0104]

[0105] Ignoring energy losses, the power balance relationship between the power supply side and the load side of the bidirectional DC-DC converter is as follows:

[0106] u b i b =u dc i dc (14)

[0107] Linearizing equation (13) and ignoring higher-order perturbation terms, we get:

[0108]

[0109] The transfer function of the voltage control section is:

[0110]

[0111] The transfer function of the current control section is:

[0112]

[0113] Therefore, the small-signal closed-loop transfer function G is derived as follows:

[0114]

[0115] The relationship between the output impedance on the power supply side and the small-signal closed-loop transfer coefficient is: Z bat =G.

[0116] Assuming the load side is connected to the bidirectional DC-DC converter via a series inductor and a parallel capacitor, then the load side input impedance Z L for:

[0117]

[0118] In the above formula, R L L is the parasitic resistance value of the line inductance. load C is the line inductance value. load R is the line capacitance value. load This represents the load resistance value.

[0119] The time constant design for the lead element can be carried out using the following method:

[0120] Step 1: First, select a value for a time constant T2 and input it into the system, for example, T2 = 0.05;

[0121] Step 2: Then, preset a time constant T1, which needs to satisfy T1≥T2;

[0122] Step 3: Solve for the distribution of poles of the small-signal closed-loop transfer function G, and plot the Nyquist curve of T0;

[0123] Step 4: If G has no poles in the right half-plane and the Nyquist curve of T0 does not enclose (-1,0), then proceed to step 5. If the above conditions are not met, then proceed to step 6.

[0124] Step 5: Let T1 = T1 + ΔT, where ΔT is the iteration step size of T1, and then return to step 3;

[0125] Step 6: Output the critical value T1. At this point, selecting a T1 value less than the critical value can ensure stability.

[0126] Regarding the preset T1 in step 2 above, it should be noted that if a larger value is directly selected as T1, it will affect the startup characteristics of the converter. Therefore, in practical applications, a smaller T1 value can be used first to ensure that the converter can start normally. For example, in step 2, T1 = T2 is preset first, and a larger T1 value is selected after the system reaches steady state to improve the dynamic characteristics of the converter.

[0127] To verify the effectiveness of the control method of this invention in improving dynamic performance, the output waveforms were compared according to the virtual capacitor control provided in this invention and the traditional IU droop control, as shown below. Figure 3 As shown, when the load fluctuates, the output DC bus voltage fluctuates significantly when using the traditional IU droop control method, resulting in a large instantaneous change in the DC bus voltage. However, the fluctuation is very small when using the virtual capacitor control method of this invention. It can be seen that the control method given by this invention can significantly reduce the impact of sudden changes in DC load on the output voltage.

[0128] Although embodiments of the invention have been shown and described, it will be understood by those skilled in the art that various changes, modifications, substitutions and alterations can be made to these embodiments without departing from the principles and spirit of the invention, the scope of which is defined by the appended claims and their equivalents.

Claims

1. A virtual capacitor control method for a bidirectional DC-DC converter, characterized in that: This includes IU droop control, virtual capacitor control, voltage control, and current control; Construct a bidirectional DC-DC converter topology; Collect the inductor current i on the power supply side of the converter respectively b Load side current i dc DC bus voltage on the load side u dc ; The input current i is calculated using IU droop control. in , will current i in Input the virtual capacitor control section and calculate the load-side bus voltage command value u. dc * ; Voltage control includes a lead element and proportional-integral control, which will control u dc * The input voltage control section calculates the load-side current command value i. dc * ; The current control uses proportional-integral control, which controls i dc * The input current control section outputs to the PWM pulse signal generator, and the generated pulse signal controls the switching transistors in the bidirectional DC-DC converter. The current i in The expression is as follows: and in =(in ref -in dc )K d (1) In the above formula, u ref K is the reference value for the DC bus voltage. d This is the voltage-current droop factor; The u dc * The expression is as follows: In the above formula, C v is the virtual capacitance value connected in parallel on the load side, and s is the Laplace operator; The i dc * The expression is as follows: In the above formula, T1 and T2 are the time constants of the lead element, and satisfy T1≥T2, k vp k is the proportional gain of the voltage loop PI controller. vi The integral coefficient of the voltage loop PI controller; The expression for the current control part is as follows: In the above formula, D pwm For the modulated signal, U dc U is the steady-state voltage value of the load-side bus. b k is the steady-state voltage value of the power supply. ip k is the proportional gain of the current loop PI controller. ii is the integral coefficient of the current loop PI controller.

2. The virtual capacitor control method for a bidirectional DC-DC converter according to claim 1, characterized in that: The small-signal model for the virtual capacitor control of the bidirectional DC-DC converter is as follows: In the above formula, Δu b , Δi b Δd and Δd are the power supply voltage u, respectively. b Inductor current i b The steady-state values ​​corresponding to the disturbances with duty cycle d are U and d, respectively. b I b and D; Δu dc and Δi dc The DC bus voltage u on the load side is respectively dc and load-side current i dc The disturbances, and their corresponding steady-state values ​​are U dc and I dc L is the inductance value on the power supply side, R is the parasitic resistance value of the inductance on the power supply side, and C is the capacitance value on the load side.

3. The virtual capacitor control method for a bidirectional DC-DC converter according to claim 2, characterized in that: The small-signal model for the virtual capacitor control of the bidirectional DC-DC converter is obtained by linearizing the average value model of the bidirectional DC-DC converter. The average value model of the bidirectional DC-DC converter is expressed as follows: Linearizing equations (7) and (8) respectively yields equations (5) and (6).

4. A virtual capacitor control method for a bidirectional DC-DC converter according to claim 1 or 2, characterized in that: The time constant of the lead element in the voltage control is determined based on the stability criteria and impedance ratio criteria of the power supply subsystem. Let the small-signal closed-loop transfer function of the small-signal model of the virtual capacitor control of the bidirectional DC-DC converter be G. If G has no poles in the right half-plane, it indicates that the power supply subsystem is stable. The impedance ratio T0 is used to judge the stability of the cascaded system. If the Nyquist curve of the impedance ratio does not enclose (-1,0), the cascaded system is stable; otherwise, the cascaded system is in an unstable state.

5. The virtual capacitor control method for a bidirectional DC-DC converter according to claim 4, characterized in that: The method for obtaining the small-signal closed-loop transfer function G is as follows: Applying the Laplace transform to equations (5) and (6) respectively, we obtain the following equation: Ignoring energy losses, the power balance relationship between the power supply side and the load side of the bidirectional DC-DC converter is as follows: in b and b =in dc and dc (5) Linearizing equation (13) and ignoring higher-order perturbation terms, we get: The transfer function of the voltage control section is: The transfer function of the current control section is: Therefore, the small-signal closed-loop transfer function G is derived as follows: The impedance ratio T0 is obtained as follows: Output impedance Z on the power supply side bat =G; Assuming the load side is connected to the bidirectional DC-DC converter via a series inductor and a parallel capacitor, then the load side input impedance Z L for: In the above formula, R L L is the parasitic resistance value of the line inductance. load C is the line inductance value. load R is the line capacitance value. load This is the load resistance value; Through loop gain T0 = Z bat / Z L To determine the stability of a cascaded system.

6. The virtual capacitor control method for a bidirectional DC-DC converter according to claim 4, characterized in that: The specific method for determining the time constant of the lead element in the voltage control is as follows: Step 1: Select a time constant T2 and input it into the system; Step 2: Preset a time constant T1, satisfying T1≥T2; Step 3: Solve for the distribution of poles of the small-signal closed-loop transfer function G, and plot the Nyquist curve of T0; Step 4: If G has no poles in the right half-plane and the Nyquist curve of T0 does not enclose (-1,0), then proceed to step 5. If the above conditions are not met, then proceed to step 6. Step 5: Let T1 = T1 + ΔT, where ΔT is the iteration step size of T1, and then return to step 3; Step 6: Output the critical value T1.

7. The virtual capacitor control method for a bidirectional DC-DC converter according to claim 6, characterized in that: In step 2, during the initial stage of converter startup, T1 = T2 is preset.

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