Three-phase high-frequency PWM AC chopper and control method thereof

Abstract

The present invention discloses a three-phase high-frequency PWM AC chopper and a control method thereof. The three-phase high-frequency PWM AC chopper includes six transistors, three filter inductors, three filter resistors, and three filter capacitors. The collectors of the first, third, and fifth transistors are respectively connected to a three-phase power supply. The emitters of the first, third, and fifth transistors are respectively connected to corresponding loads via corresponding filter inductors and filter resistors. One end of the three filter capacitors is respectively connected to the connection node between the filter resistors and the corresponding loads, and the other ends of the three filter capacitors are interconnected. The emitters of the first, third, and fifth transistors are respectively connected to the collectors of the second, fourth, and sixth transistors, and the emitters of the second, fourth, and sixth transistors are interconnected. The present invention has the advantages of strong anti-interference ability, simple wiring, and simple overall structure. The control method can reduce harmonics and enable the response signal to quickly and smoothly achieve accurate and error-free tracking of a given reference voltage.

Description

Technical Field

[0001] The present invention relates to the technical field of power electronic voltage regulators, and in particular to a three-phase high-frequency PWM AC chopper and a control method thereof. Background Art

[0002] Power electronic voltage regulators are generally controlled using a phase-controlled method. While this method has achieved some application in engineering practice, it generates a large number of harmonics during circuit operation, resulting in significant energy consumption during the transmission and transformation stages, which is detrimental to both the grid and the load. Furthermore, the power factor of phase-controlled voltage regulation is low, and the response signal cannot quickly and smoothly achieve accurate and error-free tracking of a given reference voltage. Furthermore, phase-controlled power electronic voltage regulators suffer from poor anti-interference capabilities, complex wiring, and complex structures, which conflict with the trend toward miniaturization and lightweighting of modern power electronic devices. To overcome the inherent drawbacks of phase-controlled power electronic voltage regulators, such as complex wiring and high harmonic content, indirect AC / AC converters can be used to achieve AC / AC conversion. This technology can regulate both voltage and frequency, but the multi-stage conversion results in low conversion efficiency and has numerous drawbacks, such as complex control circuits. Summary of the Invention

[0003] The present invention aims to at least partially address one of the technical problems in the related art. To this end, a first object of the present invention is to provide a three-phase high-frequency PWM AC chopper having the advantages of strong anti-interference capability, simple wiring, a simple overall structure, high power factor variation, and high efficiency. The corresponding chopper control method can reduce harmonics and enable the response signal to quickly and smoothly achieve accurate and error-free tracking of a given reference voltage.

[0004] A second object of the present invention is to provide a three-phase high-frequency PWM AC chopper control method.

[0005] To achieve the above object, the present invention is implemented through the following technical solutions:

[0006] A three-phase high-frequency PWM AC chopper comprises: first to sixth triodes, three filter inductors, three filter resistors, and three filter capacitors; the collectors of the first, third, and fifth triodes are respectively connected to a three-phase power supply, and the emitters of the first, third, and fifth triodes are respectively connected to corresponding loads via corresponding filter inductors and filter resistors, wherein the connection between the filter resistor of each phase and the load of each phase has a first node to a third node, one end of the three filter capacitors is respectively connected to the first node to the third node, and the other ends of the three filter capacitors are interconnected; the emitters of the first, third, and fifth triodes are also respectively connected to the collectors of the second, fourth, and sixth triodes, and the emitters of the second, fourth, and sixth triodes are interconnected.

[0007] Preferably, a diode is connected in anti-parallel between the emitter and the collector of each transistor.

[0008] Preferably, the first triode and the second triode are complementary switching tubes, the third triode and the fourth triode are complementary switching tubes, and the fifth triode and the sixth triode are complementary switching tubes.

[0009] Preferably, the inductance value, resistance value and capacitance value of the filter inductor, filter resistor and filter capacitor of each phase are the same.

[0010] To achieve the above object, a second aspect of the present invention provides a three-phase high-frequency PWM AC chopper control method for controlling the above-mentioned three-phase high-frequency PWM AC chopper, comprising:

[0011] Obtaining a three-phase output voltage of a three-phase high-frequency PWM AC chopper, wherein the three-phase output voltage is processed by a phase-locked loop to obtain a phase-locked angle;

[0012] Based on the phase-locked angle, performing Clark transformation and Park transformation on the three-phase output voltage and the three-phase filter inductor current to obtain dq-axis components of the three-phase output voltage and the dq-axis components of the three-phase filter inductor current;

[0013] Based on the dq-axis components of the three-phase output voltage and the three-phase filter inductor current, a modulation wave is obtained through double closed-loop control in a dq two-phase rotating coordinate system, and the modulation wave is compared with a triangular carrier to obtain six PWM pulse signals, so that six transistors can be driven by the six PWM pulse signals to realize output voltage control of the three-phase high-frequency PWM AC chopper.

[0014] Preferably, before performing Clark transformation, the mathematical model of the three-phase high-frequency PWM AC chopper in the three-phase stationary coordinate system is:

[0015]

[0016] Among them, i La 、i Lb 、i Lc They are the three-phase filter inductor current, u oa 、u ob 、u oc are the three-phase output voltages, t is the time, r, L, and C are the resistance of the filter resistor, the inductance of the filter inductor, and the capacitance of the filter capacitor, respectively. oa 、i ob 、i oc They are the three-phase output current, u ra 、u rb 、u rc are the midpoint voltages of the three-phase bridge arms respectively.

[0017] Preferably, after Clark transformation, the mathematical model of the three-phase high-frequency PWM AC chopper in the αβ two-phase stationary coordinate system is:

[0018]

[0019] Among them, i Lα 、i Lβ They are the α-axis component and β-axis component of the three-phase filter inductor current in the αβ two-phase stationary coordinate system, u oα 、u oβ are the α-axis component and β-axis component of the three-phase output voltage in the αβ two-phase stationary coordinate system, i oα 、i oβ They are the α-axis component and β-axis component of the three-phase output current in the αβ two-phase stationary coordinate system, u α 、u β They are the α-axis component and β-axis component of the three-phase bridge arm midpoint voltage in the αβ two-phase stationary coordinate system.

[0020] Preferably, after Park transformation, the mathematical model of the three-phase high-frequency PWM AC chopper in the dq two-phase rotating coordinate system is:

[0021]

[0022] Among them, u od 、u oq are the d-axis component and q-axis component of the three-phase output voltage in the dq two-phase rotating coordinate system, i Ld 、i Lq They are the d-axis component and q-axis component of the three-phase filter inductor current in the dq two-phase rotating coordinate system, i od 、i oq They are the d-axis component and q-axis component of the three-phase output current in the dq two-phase rotating coordinate system, u rd 、u rqare the d-axis component and q-axis component of the three-phase bridge arm midpoint voltage in the dq two-phase rotating coordinate system, and ω is the angular frequency.

[0023] Preferably, based on the dq axis components of the three-phase output voltage and the three-phase filter inductor current, a modulation wave is obtained by double closed-loop control in a dq two-phase rotating coordinate system, including:

[0024] After comparing the d-axis component of the three-phase output voltage with the reference voltage, the first parameter value is obtained by processing through a proportional-integral link. The q-axis component of the three-phase output voltage is processed through a first compensation link to obtain a second parameter value. The d-axis component of the three-phase output voltage is processed through a proportional link to obtain a third parameter value. The first parameter value is added to the third parameter value and the second parameter value is subtracted from the first parameter value to obtain a fourth parameter value. After comparing the fourth parameter value with the d-axis component of the three-phase filter inductor current, the fifth parameter value is obtained by processing through a proportional-integral link. The q-axis component of the three-phase filter inductor current is processed through a second compensation link to obtain a sixth parameter value. The d-axis component of the three-phase output voltage is added to the fifth parameter value and the sixth parameter value is subtracted from the sixth parameter value to obtain a d-axis voltage component used to generate the modulated wave.

[0025] After comparing the q-axis component of the three-phase output voltage with 0, the component is processed by a proportional-integral link to obtain a seventh parameter value. The d-axis component of the three-phase output voltage is processed by a first compensation link to obtain an eighth parameter value. The q-axis component of the three-phase output voltage is processed by a proportional link to obtain a ninth parameter value. The seventh parameter value is added to the ninth parameter value and subtracted from the eighth parameter value to obtain a tenth parameter value. The tenth parameter value is compared with the q-axis component of the three-phase filter inductor current and processed by a proportional-integral link to obtain an eleventh parameter value. The d-axis component of the three-phase filter inductor current is processed by a second compensation link to obtain a twelfth parameter value. The q-axis component of the three-phase output voltage is added to the eleventh parameter value and subtracted from the twelfth parameter value to obtain a q-axis voltage component used to generate the modulated wave.

[0026] The modulated wave is generated based on the d-axis voltage component and the q-axis voltage component.

[0027] Preferably, the modulation wave is compared with a triangular carrier wave to obtain six PWM pulse signals, including:

[0028] Inputting the modulated wave and the triangular carrier wave into a comparator to compare the voltages thereof;

[0029] When the voltage of the modulated wave is greater than the voltage of the triangular carrier, the comparator outputs a high level;

[0030] When the voltage of the modulated wave is less than the voltage of the triangular carrier, the comparator outputs a low level;

[0031] The first PWM pulse signal is obtained according to the output result of the comparator, the second and third PWM pulse signals are obtained by performing phase shift processing on the first PWM pulse signal, and the six PWM pulse signals are obtained by inverting the three PWM pulse signals.

[0032] The present invention has at least the following technical effects:

[0033] The present invention provides a three-phase high-frequency PWM AC chopper and a control method thereof. The three-phase high-frequency PWM AC chopper includes six transistors, three filter inductors, three filter resistors, and three filter capacitors. The three transistors are arranged on a three-phase circuit and connected to a load via the three filter inductors, filter resistors, and filter capacitors, respectively. The three transistors and three complementary transistors also form a bridge circuit. The three filter capacitors are respectively connected to the nodes where the corresponding filter resistors connect to the load. The three-phase high-frequency PWM AC chopper has the advantages of simple wiring, a simple overall structure, high power factor variation efficiency, and strong anti-interference capability. The control method of a three-phase high-frequency PWM AC chopper includes obtaining the three-phase output voltage of the three-phase high-frequency PWM AC chopper, processing the three-phase output voltage through a phase-locked loop to obtain a phase-locked angle, and then based on the phase-locked angle, performing Clark transformation and Park transformation on the three-phase output voltage and the three-phase filter inductor current to obtain the dq-axis components of the three-phase output voltage and the dq-axis components of the three-phase filter inductor current. Then, based on the dq-axis components of the three-phase output voltage and the three-phase filter inductor current, a modulation wave is obtained through double closed-loop control in a dq two-phase rotating coordinate system, and the modulation wave is compared with a triangular carrier to obtain six PWM pulse signals. Then, six transistors are driven by the six PWM pulse signals to realize output voltage control of the three-phase high-frequency PWM AC chopper. The present invention adopts a dual closed-loop control method based on dq decoupling to enable the chopper to have good volt-ampere characteristics, thereby being able to quickly and accurately track a given reference voltage, and enabling the response signal to quickly and smoothly achieve zero-error tracking of the given reference voltage, making the chopper suitable for use in different environments. In addition, the output voltage curve can be smooth, with fewer harmonics and a small overshoot, which facilitates the realization of zero-error control. The control method has the advantage of strong robustness, enabling the chopper to cope with different extreme situations, such as sudden load changes and grid voltage disturbances.

[0034] Additional aspects and advantages of the present invention will be set forth in part in the description which follows and, in part, will be obvious from the description which follows, or may be learned through practice of the present invention. BRIEF DESCRIPTION OF THE DRAWINGS

[0035] Figure 1 Schematic diagram of the structure of a three-phase high-frequency PWM AC chopper according to an embodiment of the present invention.

[0036] Figure 2Flowchart of a three-phase high-frequency PWM AC chopper control method according to an embodiment of the present invention.

[0037] Figure 3 Schematic diagram of the voltage and current dual closed-loop control structure of an embodiment of the present invention. DETAILED DESCRIPTION

[0038] The present embodiment is described in detail below. Examples of the embodiment are shown in the accompanying drawings, wherein the same or similar reference numerals throughout represent the same or similar elements or elements having the same or similar functions. The embodiments described below with reference to the accompanying drawings are exemplary and are intended to explain the present invention, but are not to be construed as limiting the present invention.

[0039] To solve the problems in the background technology, this embodiment uses a bridge circuit and an LC filter to form a three-phase high-frequency PWM (pulse width modulation) AC chopper. Its control method is to use Clark (Clark coordinate transformation method) transformation and Park (Park coordinate transformation method) transformation to obtain a mathematical model in the dq two-phase rotating coordinate system, and then use decoupling control to determine the dual closed-loop control method in the dq two-phase rotating coordinate system. The dual closed-loop control method in the dq two-phase rotating coordinate system is used to obtain a modulation wave. The modulation wave is compared with the triangular carrier to obtain a switch tube PWM pulse, and the on and off of each switch tube is determined, thereby completing the chopping control of the three-phase high-frequency PWM AC chopper. Among them, the three-phase high-frequency PWM AC chopper has the advantages of simple wiring, simple overall structure, high power factor change efficiency and strong anti-interference ability. In addition, by adopting the dual closed-loop control strategy in the dq two-phase rotating coordinate system, the response signal can be quickly and smoothly achieved error-free tracking, with small overshoot, good response curve, and strong robustness.

[0040] The three-phase high-frequency PWM AC chopper and the control method thereof of this embodiment will be described below with reference to the accompanying drawings.

[0041] Figure 1 FIG is a schematic structural diagram of a three-phase high-frequency PWM AC chopper according to an embodiment of the present invention. Figure 1 As shown, the three-phase high-frequency PWM AC chopper includes the first to sixth transistors VT1-VT6, three filter inductors L1-L3, three filter resistors r1-r3 and three filter capacitors C1-C3; the collectors of the first transistor VT1, the third transistor VT3 and the fifth transistor VT5 are respectively connected to the three-phase power supply u sa 、u sb and u scThe emitters of the first transistor VT1, the third transistor VT3, and the fifth transistor VT5 are connected to corresponding loads R1-R3 via corresponding filter inductors L1-L3 and filter resistors r1-r3, respectively. The filter resistors of each phase are connected to the loads of each phase from first to third nodes, respectively. One end of the three filter capacitors C1-C3 is connected to the first to third nodes, respectively. The other ends of the three filter capacitors C1-C3 are interconnected and connected to a common point O. Furthermore, the emitters of the first transistor VT1, the third transistor VT3, and the fifth transistor VT5 are also connected to the collectors of the second transistor VT2, the fourth transistor VT4, and the sixth transistor VT6, respectively. The emitters of the second transistor VT2, the fourth transistor VT4, and the sixth transistor VT6 are interconnected. A diode is connected in antiparallel between the emitter and collector of each transistor.

[0042] In this embodiment, the first transistor VT1 and the second transistor VT2 are complementary switching transistors, the third transistor VT3 and the fourth transistor VT4 are complementary switching transistors, and the fifth transistor VT5 and the sixth transistor VT6 are complementary switching transistors.

[0043] It should be noted that the inductance value of the three-phase filter inductor is L, the resistance value of the three-phase filter resistor is r, and the capacitance value of the three-phase filter capacitor is C.

[0044] Figure 2 Flowchart of a three-phase high-frequency PWM AC chopper control method according to an embodiment of the present invention. The method includes:

[0045] Step S1: obtaining the three-phase output voltage of the three-phase high-frequency PWM AC chopper, and processing the three-phase output voltage through a phase-locked loop to obtain a phase-locked angle.

[0046] like Figure 3 As shown, the three-phase output voltage u of the three-phase high-frequency PWM AC chopper can be obtained oa 、u ob 、u oc , then the three-phase output voltage u oa 、u ob 、u oc The phase-locked loop (PLL) processes the phase-locked angle (ωt), where ω is the angular frequency and t is the time.

[0047] Step S2: Based on the phase-locked angle, the three-phase output voltage and the three-phase filter inductor current are subjected to Clark transformation and Park transformation to obtain the dq-axis components of the three-phase output voltage and the dq-axis components of the three-phase filter inductor current.

[0048] Specifically, the three-phase filter inductor current and the three-phase filter capacitor voltage can be used as state variables, and the state equations for the filter inductor and filter capacitor can be written respectively. According to Kirchhoff's voltage-current law, the expressions of the three-phase filter inductor current equation and the three-phase filter capacitor equation can be obtained. Then, by sorting out the mathematical model of the three-phase high-frequency PWM AC chopper in the abc three-phase stationary coordinate system before Clark transformation, it can be obtained as follows:

[0049]

[0050] Among them, i La 、i Lb 、i Lc They are the three-phase filter inductor current, u oa 、u ob 、u oc are the three-phase output voltages, r, L, and C are the resistance of the filter resistor, the inductance of the filter inductor, and the capacitance of the filter capacitor, respectively. oa 、i ob 、i oc They are the three-phase output current, u ra 、u rb 、u rc are the midpoint voltages of the three-phase bridge arms respectively.

[0051] To reduce the number of variables and simplify the analysis, this embodiment introduces the Clark transform, originally applied to motor control, into the analysis of a three-phase high-frequency PWM AC chopper. The Clark transform in this embodiment converts three variables in the abc three-phase stationary coordinate system into two variables in the αβ two-phase stationary coordinate system. The mathematical model of the three-phase high-frequency PWM AC chopper in the αβ two-phase stationary coordinate system obtained through the Clark transform is:

[0052]

[0053] Among them, i Lα 、i Lβ They are the α-axis component and β-axis component of the three-phase filter inductor current in the αβ two-phase stationary coordinate system, u oα 、u oβ are the α-axis component and β-axis component of the three-phase output voltage in the αβ two-phase stationary coordinate system, i oα 、i oβ They are the α-axis component and β-axis component of the three-phase output current in the αβ two-phase stationary coordinate system, u α 、u β They are the α-axis component and β-axis component of the three-phase bridge arm midpoint voltage in the αβ two-phase stationary coordinate system.

[0054] The Clark transformation described above transforms the three static, interrelated variables into two static, independent variables, reducing the number of variables. However, the mapping of each variable on the coordinate axis is still an AC quantity, which is clearly detrimental to control. When using traditional PI (proportional-integral) control, the control of the AC quantity always has a static error. To achieve a control effect without static error, the AC quantity in the αβ two-phase stationary coordinate system must be transformed into a DC quantity in the dq two-phase rotating coordinate system based on the phase-locked angle through the Park transformation. After calculation and organization, the mathematical model of the three-phase high-frequency PWM AC chopper in the dq two-phase rotating coordinate system is:

[0055]

[0056] Among them, u od 、u oq are the d-axis component and q-axis component of the three-phase output voltage in the dq two-phase rotating coordinate system, i Ld 、i Lq They are the d-axis component and q-axis component of the three-phase filter inductor current in the dq two-phase rotating coordinate system, i od 、i oq They are the d-axis component and q-axis component of the three-phase output current in the dq two-phase rotating coordinate system, u rd 、u rq They are the d-axis component and q-axis component of the three-phase bridge arm midpoint voltage in the dq two-phase rotating coordinate system.

[0057] Thus, the d-axis component and q-axis component u of the three-phase output voltage in the dq two-phase rotating coordinate system can be obtained. od 、u oq , and the d-axis component and q-axis component i of the three-phase filter inductor current in the dq two-phase rotating coordinate system Ld 、i Lq After obtaining the d-axis component and q-axis component of the three-phase output voltage in the dq two-phase rotating coordinate system and the d-axis component and q-axis component of the three-phase filter inductor current in the dq two-phase rotating coordinate system, the d-axis component and the q-axis component are decoupled and controlled. Based on the decoupling control, the voltage and current dual closed-loop control structure in the dq two-phase rotating coordinate system is designed. The voltage and current dual closed-loop control structure in the dq two-phase rotating coordinate system is shown in FIG. Figure 3 shown.

[0058] Step S3: Based on the dq-axis components of the three-phase output voltage and the three-phase filter inductor current, a modulation wave is obtained through double closed-loop control in the dq two-phase rotating coordinate system, and the modulation wave is compared with the triangular carrier to obtain six PWM pulse signals, so as to drive six transistors through the six PWM pulse signals to realize the output voltage control of the three-phase high-frequency PWM AC chopper.

[0059] like Figure 3 As shown in the figure, the double closed-loop control method in the dq two-phase rotating coordinate system is as follows: the d-axis component u of the three-phase output voltage od With reference voltage v ref After comparison, the first parameter value is obtained through proportional integral link PI processing, the q-axis component u of the three-phase output voltage oq The second parameter value is obtained by the first compensation link ωcf, the d-axis component u of the three-phase output voltage od The third parameter value is obtained by multiplying the proportional link by the coefficient k; the fourth parameter value is obtained by adding the third parameter value to the first parameter value and subtracting the second parameter value; the fourth parameter value is added to the d-axis component i of the three-phase filter inductor current. Ld After comparison, the fifth parameter value is obtained through proportional integral link PI processing, the q-axis component i of the three-phase filter inductor current Lq The sixth parameter value is obtained by processing the second compensation link ωLf; the d-axis component u of the three-phase output voltage is od Add the fifth parameter value and subtract the sixth parameter value to obtain the d-axis voltage component v used to generate the modulation wave d .

[0060] The q-axis component u of the three-phase output voltage oq After comparing with 0, the seventh parameter value is obtained through the proportional integral link PI processing, the d-axis component u of the three-phase output voltage od The eighth parameter value is obtained by the first compensation link ωcf, the q-axis component u of the three-phase output voltage oq The ninth parameter value is obtained by multiplying the coefficient k by the proportional link; the tenth parameter value is obtained by adding the ninth parameter value to the seventh parameter value and subtracting the eighth parameter value; the tenth parameter value is added to the q-axis component i of the three-phase filter inductor current. Lq After comparison, the eleventh parameter value is obtained through proportional integral link PI processing, the d-axis component i of the three-phase filter inductor current Ld The twelfth parameter value is obtained by processing the second compensation link ωLf; the q-axis component u of the three-phase output voltage is oq Adding the eleventh parameter value and subtracting the twelfth parameter value yields the q-axis voltage component v used to generate the modulation wave. q Then, the d-axis voltage component v d and the q-axis voltage component v q Find the sum of the squares and the square root to get the modulated wave.

[0061] In one embodiment of the present invention, a modulation wave is compared with a triangular carrier to obtain six PWM pulse signals, including: inputting the modulation wave and the triangular carrier into a comparator to compare the voltages thereof; when the voltage of the modulation wave is greater than the voltage of the triangular carrier, the comparator outputs a high level; when the voltage of the modulation wave is less than the voltage of the triangular carrier, the comparator outputs a low level; obtaining a first PWM pulse signal based on the comparator output result, obtaining second and third PWM pulse signals by performing phase shifting processing on the first PWM pulse signal, and obtaining six PWM pulse signals by inverting the three PWM pulse signals.

[0062] The comparator output generates the first PWM pulse signal. Since the three-phase PWM pulse signals differ by 120°, phase shifting yields three PWM pulse signals. Furthermore, since the switches on the bridge arms are complementary, six PWM pulse signals are obtained by inverting the three PWM pulse signals. Furthermore, these six PWM pulse signals drive the six switching transistors (VT1-VT6), achieving output voltage control of the three-phase high-frequency PWM AC chopper.

[0063] In summary, the three-phase high-frequency PWM AC chopper provided by this embodiment has a simple topology and a small number of switching elements, resulting in low cost. The three-phase high-frequency PWM AC chopper has a good response curve. The three-phase high-frequency PWM AC chopper control method of this embodiment can make the chopper output voltage curve smooth, with fewer harmonics and a small overshoot, facilitating the implementation of zero-error control. The three-phase high-frequency PWM AC chopper has high tracking accuracy. Due to the use of the three-phase high-frequency PWM AC chopper control method based on dq decoupling control, the chopper can have good volt-ampere characteristics, thereby enabling rapid, accurate, and stable zero-error tracking of a given reference voltage. The three-phase high-frequency PWM AC chopper has strong robustness. The three-phase high-frequency PWM AC chopper control method of this embodiment can make the chopper have strong anti-interference capabilities and can cope with various extreme situations, such as sudden load changes and grid voltage disturbances.

[0064] It should be noted that, in this document, relational terms such as first and second, etc., are used only to distinguish one entity or operation from another entity or operation, and do not necessarily require or imply the existence of any such actual relationship or order between these entities or operations. Moreover, the terms "comprises," "comprising," or any other variants thereof are intended to cover non-exclusive inclusion, so that a process, method, article, or device comprising a series of elements includes not only those elements, but also other elements not explicitly listed, or elements inherent to such process, method, article, or device. In the absence of further limitations, an element defined by the phrase "comprising a ..." does not exclude the presence of other identical elements in the process, method, article, or device comprising the element.

[0065] Although the present invention has been described in detail through the above preferred embodiments, it should be understood that the above description is not intended to limit the present invention. After reading the above description, various modifications and substitutions of the present invention will become apparent to those skilled in the art. Therefore, the scope of protection of the present invention should be defined by the appended claims.

Claims

1. A three-phase high-frequency PWM AC chopper control method for controlling a three-phase high-frequency PWM AC chopper, the three-phase high-frequency PWM AC chopper comprising: The first to sixth transistors, three filter inductors, three filter resistors and three filter capacitors; the collectors of the first, third and fifth transistors are respectively connected to the three-phase power supply, and the emitters of the first, third and fifth transistors are respectively connected to the corresponding loads through the corresponding filter inductors and filter resistors, wherein the connection points between the filter resistors of each phase and the loads of each phase are respectively the first node to the third node, one end of the three filter capacitors is respectively connected to the first node, the second node and the third node, and the other ends of the three filter capacitors are interconnected; the emitters of the first, third and fifth transistors are also respectively connected to the collectors of the second, fourth and sixth transistors, and the emitters of the second, fourth and sixth transistors are interconnected, characterized in that the method comprises: Obtaining a three-phase output voltage of a three-phase high-frequency PWM AC chopper, wherein the three-phase output voltage is processed by a phase-locked loop to obtain a phase-locked angle; Based on the phase-locked angle, performing Clark transformation and Park transformation on the three-phase output voltage and the three-phase filter inductor current to obtain dq-axis components of the three-phase output voltage and the dq-axis components of the three-phase filter inductor current; Based on the dq-axis components of the three-phase output voltage and the three-phase filter inductor current, dual closed-loop control is performed in a dq two-phase rotating coordinate system to obtain a d-axis voltage component and a q-axis voltage component for generating a modulation wave, and Euclidean norm calculation is performed on the d-axis voltage component and the q-axis voltage component to obtain a modulation wave. The modulation wave is then compared with a triangular carrier to obtain six PWM pulse signals, so that six transistors are driven by the six PWM pulse signals to realize output voltage control of a three-phase high-frequency PWM AC chopper.

2. The three-phase high-frequency PWM AC chopper control method according to claim 1, characterized in that: A diode is connected in anti-parallel between the emitter and collector of each transistor.

3. The three-phase high-frequency PWM AC chopper control method according to claim 1, wherein: The first triode and the second triode are complementary switching tubes, the third triode and the fourth triode are complementary switching tubes, and the fifth triode and the sixth triode are complementary switching tubes.

4. The three-phase high-frequency PWM AC chopper control method according to claim 1, wherein: The inductance, resistance and capacitance values ​​of the filter inductors, filter resistors and filter capacitors of each phase are the same.

5. The three-phase high-frequency PWM AC chopper control method according to claim 1, wherein: Before Clark transformation, the mathematical model of the three-phase high-frequency PWM AC chopper in the three-phase stationary coordinate system is: Among them, i La 、i Lb 、i Lc They are the three-phase filter inductor current, u oa 、u ob 、u oc are the three-phase output voltages, t is the time, r, L, and C are the resistance of the filter resistor, the inductance of the filter inductor, and the capacitance of the filter capacitor, respectively. oa 、i ob 、i oc They are the three-phase output current, u ra 、u rb 、u rc are the midpoint voltages of the three-phase bridge arms respectively.

6. The three-phase high-frequency PWM AC chopper control method according to claim 5, characterized in that: After Clark transformation, the mathematical model of the three-phase high-frequency PWM AC chopper in the αβ two-phase stationary coordinate system is: Among them, i Lα 、i Lβ They are the α-axis component and β-axis component of the three-phase filter inductor current in the αβ two-phase stationary coordinate system, u oα 、u oβ are the α-axis component and β-axis component of the three-phase output voltage in the αβ two-phase stationary coordinate system, i oα 、i oβ They are the α-axis component and β-axis component of the three-phase output current in the αβ two-phase stationary coordinate system, u α 、u β They are the α-axis component and β-axis component of the three-phase bridge arm midpoint voltage in the αβ two-phase stationary coordinate system.

7. The three-phase high-frequency PWM AC chopper control method according to claim 6, characterized in that: After Park transformation, the mathematical model of the three-phase high-frequency PWM AC chopper in the dq two-phase rotating coordinate system is: Among them, u od 、u oq are the d-axis component and q-axis component of the three-phase output voltage in the dq two-phase rotating coordinate system, i Ld 、i Lq They are the d-axis component and q-axis component of the three-phase filter inductor current in the dq two-phase rotating coordinate system, i od 、i oq They are the d-axis component and q-axis component of the three-phase output current in the dq two-phase rotating coordinate system, u rd 、u rq are the d-axis component and q-axis component of the three-phase bridge arm midpoint voltage in the dq two-phase rotating coordinate system, and ω is the angular frequency.

8. The three-phase high-frequency PWM AC chopper control method according to claim 1, wherein: Based on the dq axis components of the three-phase output voltage and the three-phase filter inductor current, a modulation wave is obtained through double closed-loop control in a dq two-phase rotating coordinate system, including: After comparing the d-axis component of the three-phase output voltage with the reference voltage, the first parameter value is obtained by processing through a proportional-integral link. The q-axis component of the three-phase output voltage is processed through a first compensation link to obtain a second parameter value. The d-axis component of the three-phase output voltage is processed through a proportional link to obtain a third parameter value. The first parameter value is added to the third parameter value and the second parameter value is subtracted from the first parameter value to obtain a fourth parameter value. After comparing the fourth parameter value with the d-axis component of the three-phase filter inductor current, the fifth parameter value is obtained by processing through a proportional-integral link. The q-axis component of the three-phase filter inductor current is processed through a second compensation link to obtain a sixth parameter value. The d-axis component of the three-phase output voltage is added to the fifth parameter value and the sixth parameter value is subtracted from the sixth parameter value to obtain a d-axis voltage component used to generate the modulated wave. After comparing the q-axis component of the three-phase output voltage with 0, the component is processed by a proportional-integral link to obtain a seventh parameter value. The d-axis component of the three-phase output voltage is processed by a first compensation link to obtain an eighth parameter value. The q-axis component of the three-phase output voltage is processed by a proportional link to obtain a ninth parameter value. The seventh parameter value is added to the ninth parameter value and subtracted from the eighth parameter value to obtain a tenth parameter value. The tenth parameter value is compared with the q-axis component of the three-phase filter inductor current and processed by a proportional-integral link to obtain an eleventh parameter value. The d-axis component of the three-phase filter inductor current is processed by a second compensation link to obtain a twelfth parameter value. The q-axis component of the three-phase output voltage is added to the eleventh parameter value and subtracted from the twelfth parameter value to obtain a q-axis voltage component used to generate the modulated wave. The modulated wave is generated based on the d-axis voltage component and the q-axis voltage component.

9. The three-phase high-frequency PWM AC chopper control method according to claim 1, wherein: Comparing the modulated wave with the triangular carrier wave to obtain six PWM pulse signals, including: Inputting the modulated wave and the triangular carrier wave into a comparator to compare the voltages thereof; When the voltage of the modulated wave is greater than the voltage of the triangular carrier, the comparator outputs a high level; When the voltage of the modulated wave is less than the voltage of the triangular carrier, the comparator outputs a low level; The first PWM pulse signal is obtained according to the output result of the comparator, the second and third PWM pulse signals are obtained by performing phase shift processing on the first PWM pulse signal, and the six PWM pulse signals are obtained by inverting the three PWM pulse signals.

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