Target detection method, electronic equipment and storage medium

Abstract

The embodiment of the invention relates to the technical field of signal processing, and discloses a target detection method, electronic equipment and a storage medium. The target detection method can be applied to a DDM radar sensor with at least one transceiving channel, and comprises the following steps: performing distance dimension Fourier transform processing on an echo signal received by a receiving channel to obtain distance dimension FFT data, and performing phase shift compensation on the distance dimension FFT data by adopting a phase shift error coefficient of a transmitting channel to obtain distance dimension FFT compensation data; doppler dimension FFT processing is carried out on the distance dimension FFT compensation data, and Doppler dimension FFT data are obtained; and determining speed information of the target object based on the Doppler dimension FFT data. According to the target detection method provided by the embodiment of the invention, the error of the phase shifter can be compensated, and the accuracy of angle calculation of the radar system is improved, so that the accuracy of the detected target is improved.

Description

Technical Field

[0001] The embodiments of the present application relate to the field of signal processing technology, and in particular to a target detection method, an electronic device, and a storage medium. Background Art

[0002] In a radar system, a series of continuous linear frequency modulated pulse signals are emitted by the transmitting antenna, and the echo of the pulse signal is received by the receiving antenna. After processing, the detected target information can be output. This process is generally called a frame in time. The radar system detects the surrounding moving or stationary objects in real time through continuous frame processing.

[0003] In millimeter-wave radar systems, in order to synthesize a large-aperture receiving antenna array and improve the angular resolution performance of the radar system, multiple transmitting antennas are usually used to virtualize more receiving arrays to increase the equivalent receiving aperture, which is called a multiple input multiple output (MIMO) radar system. The Doppler Division Multiplexing (DDM) scheme can separate the signals transmitted by different transmitting antennas, but the DDM scheme requires each transmitting antenna to modulate an initial phase that steps with the chirp (a linear frequency modulated continuous wave), which is generally implemented by a phase shifter in the RF circuit. The phase shifter generally has a phase shift error, and the phase shifter also affects the amplitude of the pulse signal. Therefore, when different phase shift values ​​are set for the transmitting antenna, the amplitude of the pulse signal will have a slight fluctuation. When the phase shift error and amplitude fluctuation are large, the target object cannot be accurately detected.

[0004] Therefore, the calibration of the phase shifter plays an important role in the performance of the DDM MIMO radar system. The current calibration method is to find the appropriate phase shifter setting under different operating parameters and temperatures so that the phase shifter output is closest to the ideal phase shift value, thereby achieving the purpose of phase shifter calibration. However, there is still a residual error between the phase shift value of the calibrated phase shifter and the ideal phase shift value, and the performance of the radar system cannot be well improved. Summary of the invention

[0005] The purpose of the embodiments of the present application is to provide a target detection method, an electronic device and a storage medium, which can compensate for the error of the phase shifter, improve the accuracy of the angle solution of the radar system, and thus improve the accuracy of the detected target.

[0006] To solve the above technical problems, an embodiment of the present application provides a target detection method, which is applied to a DDM radar sensor having at least one transceiver channel, comprising:

[0007] Performing range-dimensional Fourier transform processing on the echo signal received by the receiving channel to obtain range-dimensional FFT data, and performing phase shift compensation on the range-dimensional FFT data using the phase shift error coefficient of the transmitting channel to obtain range-dimensional FFT compensation data;

[0008] Performing Doppler dimension Fourier transform processing on the distance dimension FFT compensation data to obtain Doppler dimension FFT data;

[0009] Based on the Doppler FFT data, velocity information of the target object is determined.

[0010] The target detection method provided in the present application can obtain accurate range-dimensional FFT data by performing phase shift compensation on the range-dimensional FFT data using the compensation coefficients of each transmitting channel after obtaining the range-dimensional FFT data; then, the Doppler-dimensional FFT data is obtained through the compensated range-dimensional FFT data, and the speed information of the target object determined based on the Doppler-dimensional FFT data is the accurate speed information, thereby achieving accurate detection of the target object.

[0011] In some embodiments, the phase shift compensation of the range dimension FFT data using the phase shift error coefficient of the transmission channel includes:

[0012] For the data corresponding to each chirp sequence in the range-dimensional FFT data, phase shift compensation is performed using the phase shift error coefficients of the transmission channels corresponding to each chirp sequence.

[0013] In some embodiments, the method further comprises:

[0014] Based on the Doppler FFT data, angle information of the target object is determined.

[0015] In some embodiments, determining the angle information of the target object based on the Doppler FFT data includes:

[0016] The complex data at the peak of the Doppler dimension FFT data generated by the corresponding transmission channel is extracted and compensated from the Doppler dimension FFT data, and a receiving vector is synthesized to determine the angle information of the target object.

[0017] In some embodiments, the range-dimensional FFT data is composed of a plurality of range-gate FFT data, and the phase shift compensation is performed using the phase shift error coefficients of the transmission channels corresponding to the chirp sequences in the range-dimensional FFT data, respectively, including:

[0018] For each of the range gate FFT data, phase shift compensation is performed using the phase shift error coefficients of the transmission channels corresponding to each of the chirp sequences to obtain range gate FFT compensation data corresponding to each of the range gate FFT data.

[0019] In some embodiments, performing Doppler dimension Fourier transform processing on the distance dimension FFT compensation data to obtain Doppler dimension FFT data includes:

[0020] The Doppler dimension Fourier transform processing is performed on each of the range gate FFT compensation data to obtain Doppler gate FFT data corresponding to each of the range gate FFT compensation data.

[0021] In some embodiments, the transceiver channel is composed of at least two transmitting channels and one receiving channel, and the determining the speed information of the target object based on the Doppler dimension FFT data includes:

[0022] Based on all the Doppler gate FFT data, the velocity information of the target object is determined.

[0023] In some embodiments, the transceiver channel is composed of at least two transmitting channels and at least two receiving channels, and the determining the speed information of the target object based on the Doppler dimension FFT data includes:

[0024] Extracting Doppler gate FFT data in each receiving channel respectively, wherein the Doppler gate FFT data in each receiving channel includes Doppler gate FFT data compensated by phase shift error coefficients of different transmitting channels;

[0025] Combining the Doppler gate FFT data in all the receiving channels that have been compensated by the phase shift error coefficient of the same transmitting channel to obtain multiple groups of Doppler gate FFT data corresponding to each of the transmitting channels;

[0026] For each group of the Doppler gate FFT data, extract target Doppler gate FFT data therefrom, wherein the target Doppler gate FFT data is the echo signal of the transmission channel corresponding to the group of the Doppler gate FFT data;

[0027] The target Doppler gate FFT data extracted from each set of the Doppler gate FFT data are synthesized, and the velocity information of the target object is determined based on the target Doppler gate FFT data extracted from each set of the Doppler gate FFT data.

[0028] In some embodiments, the phase shift error coefficient of the transmitting channel is obtained by the following steps:

[0029] The chirp sequence corresponding to the transmission channel is compared with a standard chirp sequence generated by a standard phase shifter to obtain a phase shift error coefficient of the transmission channel.

[0030] In some embodiments, the using the phase error coefficient of the transmitting channel to perform phase compensation on the range-dimensional FFT data to obtain the range-dimensional FFT compensation data includes:

[0031] The phase shift error coefficient of the transmitting channel is multiplied or added to the range-dimensional FFT data to obtain range-dimensional FFT compensation data.

[0032] In some embodiments, the phase shift error coefficient includes a phase shift phase error and / or a phase shift amplitude error.

[0033] In some embodiments, the angle information of the target object includes a pitch angle and / or an azimuth angle of the target object.

[0034] In some embodiments, the transmission link corresponding to the transmission channel includes an analog signal source and a phase shifter, the analog signal source can be configured to provide an initial analog signal, the phase shifter can be configured to generate a phase-shifted signal, and shift the phase of the initial analog signal based on the phase-shifted signal to perform a preset phase shift operation on the initial analog signal; wherein the phase shifter is a delay line phase shifter or a digital phase shifter.

[0035] In some embodiments, when the phase shifter is a digital phase shifter, the digital phase shifter includes a digital phase shift signal source, a digital-to-analog converter and a mixer. The digital phase shifter can be configured to generate a digital phase shift signal. The digital-to-analog converter can be configured to convert the received digital phase shift signal into an analog phase shift signal. The mixer can be configured to perform a mixing operation on the received initial analog signal using the received analog phase shift signal so as to perform a preset phase shift operation on the initial analog signal.

[0036] The embodiment of the present application provides a signal transceiver link, including a transmitting link for transmitting an electromagnetic wave signal and a receiving link for receiving an echo signal formed based on the electromagnetic wave signal;

[0037] The transmission link includes an analog signal source and a digital phase shifter, wherein the analog signal source may be configured to provide an initial analog signal, and the digital phase shifter may be configured to generate a digital phase-shifted signal, and perform a phase shift on the initial analog signal based on the digital phase-shifted signal, so as to perform a preset phase shift operation on the initial analog signal;

[0038] The receiving link includes an analog-to-digital converter and a digital baseband processing module, wherein the analog-to-digital converter may be configured to perform analog-to-digital conversion on the received echo signal to obtain a digital baseband signal, and the digital baseband signal processing module may be configured to perform distance-dimensional Fourier transform and velocity-dimensional Fourier transform on the digital baseband signal in sequence;

[0039] The digital baseband processing module includes a phase compensation unit, and the phase compensation unit can be configured to adopt a phase error coefficient of a digital phase shifter in the transmission link to obtain a phase compensation for the distance-dimensional FFT data by Fourier transforming the distance-dimensional FFT data; and

[0040] The digital baseband processing module may be configured to perform the speed dimension Fourier transform according to the compensated distance dimension FFT data.

[0041] An embodiment of the present application further provides a computer-readable storage medium storing a computer program, which implements the target detection method as described above when executed by a processor.

[0042] An embodiment of the present application also provides an integrated circuit, including: a signal transceiver channel for transmitting radio signals and receiving echo signals formed by the radio signals being reflected by a target object; and a target detection module for detecting the target object according to the echo signals based on the method described above.

[0043] In some embodiments, the target detection module includes: a first processing unit, used to perform range-dimensional Fourier transform processing on the echo signal received by the receiving channel to obtain range-dimensional FFT data, and use the phase shift error coefficient of the transmitting channel to perform phase shift compensation on the range-dimensional FFT data to obtain range-dimensional FFT compensated data; a second processing unit, used to perform Doppler-dimensional Fourier transform processing on the range-dimensional FFT compensated data to obtain Doppler-dimensional FFT data; a detection unit, used to determine the velocity information of the target object based on the Doppler-dimensional FFT data.

[0044] An embodiment of the present application further provides a radio device, comprising: a carrier; the integrated circuit as described above, arranged on the carrier; and an antenna, arranged on the carrier, for transmitting and receiving radio signals.

[0045] An embodiment of the present application further provides a terminal device, comprising: a device body; and a radio device as described above disposed on the device body, wherein the radio device is used for target detection and / or communication. BRIEF DESCRIPTION OF THE DRAWINGS

[0046] One or more embodiments are exemplarily described by pictures in the corresponding drawings, and these exemplified descriptions do not constitute limitations on the embodiments. Elements with the same reference numerals in the drawings represent similar elements, and unless otherwise stated, the figures in the drawings do not constitute proportional limitations.

[0047] Figure 1 is a flow chart of a target detection method provided according to an embodiment of the present application;

[0048] Figure 2a is a schematic diagram of chirp stepped phase modulation of a transmitting antenna according to an embodiment of the present application;

[0049] Figure 2b yes Figure 2a The corresponding 2D FFT plane;

[0050] Figure 3a is a schematic diagram of a process of performing range-dimensional FFT processing on a single receiving channel according to an embodiment of the present application;

[0051] Figure 3b It is Figure 3a 1D FFT data plane diagram obtained during the processing;

[0052] Figure 4a 1D FFT data of the Mth range gate is subjected to phase shift compensation according to an embodiment of the present application. Figure 1 ;

[0053] Figure 4b is a second schematic diagram of a process of performing phase shift compensation on 1D FFT data of an Mth range gate according to an embodiment of the present application;

[0054] Figure 5 is a schematic diagram of a process of combining detection of multiple receiving channels and solving a synthetic angle of a virtual array according to an embodiment of the present application;

[0055] Figure 6 A simplified schematic diagram of a signal transmission link of a digital phase shifter architecture in an embodiment of the present application;

[0056] Figure 7 Based on Figure 6 A schematic diagram of a digital phase shifter architecture in a signal transmission link is shown;

[0057] Figure 8 A schematic diagram of a transmission link including a specific compensation module in an embodiment of the present application;

[0058] Fig. 9 A schematic diagram of a transceiver link in an embodiment of the present application;

[0059] Fig.10 It is a schematic diagram of the structure of an integrated circuit provided according to an embodiment of the present application. DETAILED DESCRIPTION

[0060] To make the purpose, technical scheme and advantages of the embodiments of the present application clearer, the embodiments of the present application will be described in detail below in conjunction with the accompanying drawings. However, it will be appreciated by those skilled in the art that in the present application, many technical details are proposed in order to enable the reader to better understand the present application. However, even without these technical details and various changes and modifications based on the following embodiments, the technical scheme claimed in the present application can also be implemented. The division of the following embodiments is for the convenience of description, and the specific implementation of the present application should not be construed as any limitation, and the various embodiments can be combined and referenced with each other under the premise of no contradiction.

[0061] An embodiment of the present application relates to a target detection method, which is applied to a DDM radar sensor having at least one transceiver channel. The implementation details of the target detection method of this embodiment are described in detail below. The following content is only for the convenience of understanding the implementation details, and is not necessary for implementing this solution.

[0062] The specific process of the target detection method of this embodiment can be as follows Figure 1 As shown, including:

[0063] Step 101: Perform range-dimensional Fourier transform processing on the echo signal received by the receiving channel to obtain range-dimensional FFT data, and use the phase shift error coefficient of the transmitting channel to perform phase shift compensation on the range-dimensional FFT data to obtain range-dimensional FFT compensation data.

[0064] In existing DDM MIMO radar systems, a pre-calibration method is usually used, that is, to find a suitable phase shifter under different operating parameters and temperatures so that the phase shifter output is closest to the ideal phase shift value to perform phase shift processing on the chirp emitted by the transmission channel to reduce the error caused by the phase shifter. However, in fact, there is still a residual error between the calibrated phase shift value and the ideal phase shift value.

[0065] Each transmitting antenna has a phase shifter, and the phase shift errors of different transmitting antennas are different. This error has no effect on the range-dimensional FFT data, but only affects the accuracy of the Doppler-dimensional FFT data. Therefore, the present application proposes that after performing a range-dimensional Fourier transform (also called "1D FFT") on the echo signals received by each receiving channel to obtain the range-dimensional FFT data (also called "1DFFT data"), the compensation coefficients of each transmitting channel are used to perform phase shift compensation on the range-dimensional FFT data to effectively eliminate the influence of the phase shifter error, thereby obtaining accurate range-dimensional FFT data.

[0066] The core improvement of this embodiment is that the phase shift error compensation of the phase shifter of the transmitting antenna is realized by performing phase shift compensation processing on the distance dimension FFT data, and the specific operation process of how to use the phase shift error coefficient to perform phase shift compensation processing, the specific coefficients included in the phase shift error coefficient, and the method steps of how to obtain the phase shift error coefficient are not limited. In the subsequent embodiments, these contents will be explained through multiple embodiments and will not be repeated here.

[0067] Step 102: Perform Doppler dimension Fourier transform processing on the range dimension FFT compensation data to obtain Doppler dimension FFT data.

[0068] After obtaining the distance dimension FFT compensation data, Doppler dimension Fourier transform (also called "2DFFT") processing may be further performed on the data to obtain Doppler dimension FFT data (also called "2D FFT data" or "velocity dimension FFT data").

[0069] Step 103: Determine the velocity information of the target object based on the Doppler FFT data.

[0070] By performing constant false alarm rate (CFAR) detection on the Doppler dimension FFT data, false targets can be removed from the energy dimension. At this time, the target information contains information in the distance and speed dimensions.

[0071] The target detection method of this embodiment can obtain accurate range-dimensional FFT data by using the compensation coefficients of each transmission channel to perform phase shift compensation on the range-dimensional FFT data after obtaining the range-dimensional FFT data; then, the Doppler-dimensional FFT data is obtained through the compensated range-dimensional FFT data, and the speed information of the target object determined based on the Doppler-dimensional FFT data is the accurate speed information, thereby achieving accurate detection of the target object.

[0072] The phase shift compensation process in the target detection method in the above embodiment will be described below through multiple embodiments.

[0073] In some embodiments, phase shift compensation is performed on range-dimensional FFT data using a phase shift error coefficient of a transmission channel, including: for data corresponding to each chirp sequence in the range-dimensional FFT data, phase shift compensation is performed using a phase shift error coefficient of a transmission channel corresponding to each chirp sequence.

[0074] Take the DDM radar sensor containing two transmitting channels (corresponding to two transmitting antennas TX0 and TX1) as an example. Figure 2aAs shown in the figure, it is a schematic diagram of chirp step phase modulation of 2 transmitting antennas. Among them, in the chirp sequence (chirp sequence) sent by the transmitting antenna TX0, each chirp has not been modulated by the phase shifter (it can also be considered that the phase sequence with a step of 0 degrees is modulated), so its echo signal has no speed offset; in the chirp sequence (chirp sequence) sent by the transmitting antenna TX1, the modulation phase step is the initial phase of θ=π / 2. When the modulated phase error is 0, the receiver processes the received echo signal through a two-dimensional Fourier transform (2D FFT) to obtain a 2D FFT plane as shown Figure 2b As shown in FIG. 1 , the two peaks on the 2D FFT plane are 2D FFT peaks formed by the echo signals emitted by TX0 and TX1 and reflected by a target with a certain distance and speed. According to the pre-modulated step phase of TX0 and TX1, the distances of these two peaks in the Doppler dimension are known, so they can correspond to TX0 and TX1, achieving the purpose of multiplexing and smooth separation in the Doppler dimension.

[0075] From this, it can be seen that, in principle, as long as the echo signals of TX0 and TX1 are obtained, the phase shift error coefficient of the transmitting channel corresponding to TX0 can be used to perform phase shift compensation for the data corresponding to the chirp sequence sent by TX0 in the range gate FFT data obtained by performing 1DFFT processing on the echo signals, and the phase shift error coefficient of the transmitting channel corresponding to TX1 can be used to perform phase shift compensation for the data corresponding to the chirp sequence sent by TX0.

[0076] However, in the range gate FFT data, the data corresponding to the chirp sequences belonging to TX0 and TX1 can only be separated by the Doppler dimension obtained after 2DFFT processing. Therefore, when compensating the range gate FFT data, the data corresponding to the two chirp sequences can be distinguished first, that is, the phase shift error coefficients corresponding to the two transmission channels (TX0, TX1) are respectively used for the entire obtained range gate FFT data to perform phase shift compensation at the same time. In this way, the phase shift error coefficients of the transmission channels corresponding to each chirp sequence are respectively used for phase shift compensation for the data corresponding to each chirp sequence in the range dimension FFT data. Then, from the Doppler FFT data finally obtained, the data corresponding to the chirp sequences corresponding to the two transmission channels (TX0, TX1) are separated, and the range gate FFT data (range gate FFT compensation data) after phase shift compensation corresponding to each Doppler FFT data can be distinguished as the data after phase shift compensation processing of the range gate FFT data corresponding to the chirp sequence sent by which transmission channel using the phase shift error coefficient of which transmission channel.

[0077] Generally, the range-dimensional FFT data is composed of a plurality of range-gate FFT data. Accordingly, for the data corresponding to each chirp sequence in the range-dimensional FFT data, the phase shift error coefficients of the transmission channels corresponding to each chirp sequence are respectively used to perform phase shift compensation, which may include: for each range-gate FFT data, the phase shift error coefficients of the transmission channels corresponding to each chirp sequence are respectively used to perform phase shift compensation, and the range-gate FFT compensation data corresponding to each range-gate FFT data is obtained. Correspondingly, performing Doppler-dimensional Fourier transform processing on the range-dimensional FFT compensation data to obtain Doppler-dimensional FFT data may include: performing Doppler-dimensional Fourier transform processing on each range-gate FFT compensation data to obtain Doppler-dimensional FFT data corresponding to each range-gate FFT compensation data.

[0078] For example, Figure 3a FIG. 1 is a schematic diagram of the process of performing distance dimension FFT processing on a single receiving channel. In the figure, the echo signal received by the receiving channel, i.e., the chirp sequence (C 0 , C 1 , …, C N ) contains the chirp sequences of the two transmission channels corresponding to TX0 and TX1. By performing distance-dimensional FFT processing on the echo signal, the distance-dimensional FFT data (P 0 , P 1 ,…,P N-1 ), forming Figure 3b The figure shows a 1D FFT data plane diagram. In the 1D FFT data plane, there are two dimensions: the range dimension and the Doppler dimension. The range dimension includes M range gates, each of which is the range dimension FFT data of the same distance but different chirp numbers; the Doppler dimension includes N Doppler gates, each of which is the range dimension FFT data of a chirp pulse.

[0079] Combination Figure 3a and Figure 3b It can be seen that the distance dimension FFT data P corresponding to each receiving channel 0 , P 1 ,…,P N-1 With chirp sequence C 1 , C 2 , …, C N is corresponding, and the chirp sequence C 1 , C 2 , …, C N It contains the echo signals of the two transmission channels corresponding to TX0 and TX1, that is, the chirp sequences corresponding to the two transmission channels. The range dimension FFT data is composed of multiple range gate FFT data. Therefore, when performing phase shift compensation on the range dimension FFT data, Figure 3bPhase shift compensation is performed individually for each range gate.

[0080] For example Figure 4a In the example of the Mth range gate, the 1D FFT data of the Mth range gate (also called the "Mth range gate data slice") is phase-compensated using the phase error coefficients of the transmission channels of the transmitting antennas TX0 and TX1 along the Doppler dimension to obtain two corresponding range gate FFT compensated data slices. Since the phase compensation for each range gate FFT data slice is performed using the phase compensation coefficient of each transmission channel, multiple range gate FFT compensated data slices equal to the number of transmission channels can be obtained for each range gate FFT data slice.

[0081] In this way, after the range gate FFT compensation data slices are subsequently processed by 2D FFT, multiple 2D FFT data slices having the same number as the transmission channels can be obtained. For example, Figure 4a 2D FFT data slice 0 and 2D FFT data slice 1 in. Since 2D FFT data slice 0 and 2D FFT data slice 1 respectively contain the data corresponding to the chirp sequences of the two transmission channels (TX0, TX1), after obtaining the 2D FFT data slices, the data corresponding to each transmission channel can be separated, thereby separating the Doppler dimension FFT data obtained after performing phase shift compensation on the range dimension FFT data of a certain transmission channel based on the phase shift compensation coefficient of the transmission channel.

[0082] For example Figure 4b In the example, we still take the Mth range gate as an example, and slice P of the Mth range gate data 0 , P 1 ,…,P N-1 Using the phase error coefficient W 0 , W 1 , …, W N-1 Perform phase shift compensation. For example, the compensation algorithm is to multiply the two in phase, and the 1D FFT data slice Q after phase shift compensation can be obtained. 0 , Q 1 , …, Q N-1 ; Then, the compensated data is processed by 2D FFT to obtain a 2D FFT data slice, which contains two peaks corresponding to two transmission channels (the peaks of the 2D FFT corresponding to TX0 and TX1). By determining the transmission channel to which the phase error coefficient used in the phase compensation process belongs, it can be learned that the two 2D FFT peaks are the 2D FFT data corresponding to which transmission channel's range gate FFT data is subjected to phase compensation processing using the phase error coefficient of which transmission channel.

[0083] In a DDM radar sensor, the process of determining the speed information of a target object based on Doppler FFT data is different depending on the number of transmitting channels and receiving channels in the transceiver channel. This is because when there are two or more receiving channels, it is necessary to consider merging the 2D FFT data of different receiving channels.

[0084] Taking a receiving channel as an example, for example, when the transceiver channel is composed of at least two transmitting channels and one receiving channel, determining the speed information of the target object based on Doppler dimension FFT data may include: determining the speed information of the target object based on all Doppler gate FFT data.

[0085] In the aforementioned embodiment, the processing process of obtaining multiple Doppler gate FFT data through one range gate FFT data in the phase shift compensated 2D FFT data formed by two transmit channels, i.e., transmit channels corresponding to TX0 and TX1 and one receive channel (single receive channel) has been described. Since it is a single receive channel, there is no need to consider merging the 2D FFT data for a certain transmit channel between channels. Therefore, after obtaining multiple Doppler gate FFT data corresponding to each range gate FFT data, CFAR processing can be performed based on all Doppler gate FFT data to remove false targets from the energy dimension, thereby obtaining information on the distance and speed of the target object.

[0086] When the transceiver channel includes two receiving channels on two machines, it is necessary to consider merging the 2DFFT data of different receiving channels. For example, when the transceiver channel consists of at least two transmitting channels and at least two receiving channels, the velocity information of the target object can be determined based on the Doppler dimension FFT data, which can be achieved through the following steps (a to d).

[0087] Step a: extracting Doppler gate FFT data in each receiving channel respectively, wherein the Doppler gate FFT data in each receiving channel includes Doppler gate FFT data compensated by phase shift error coefficients of different transmitting channels.

[0088] Step b: combining the Doppler gate FFT data in all receiving channels that have been compensated by the phase shift error coefficient of the same transmitting channel to obtain multiple groups of Doppler gate FFT data corresponding to each transmitting channel.

[0089] Step c: for each set of Doppler gate FFT data, extract target Doppler gate FFT data therefrom, where the target Doppler gate FFT data is the echo signal of the transmission channel corresponding to the set of Doppler gate FFT data.

[0090] Step d: synthesizing the target Doppler gate FFT data extracted from each set of Doppler gate FFT data, and determining the velocity information of the target object based on the target Doppler gate FFT data extracted from each set of Doppler gate FFT data.

[0091] For example, in Figure 4, for Figure 5 As shown in the figure, in the 2D FFT data slices (Doppler gate FFT data) of multiple receiving channels, each slice covers the data of multiple receiving channels and multiple transmitting channels. Therefore, it is necessary to perform multi-channel merging detection on each 2D FFT data slice, determine the 2D FFT data of each transceiver channel (one receiving channel + one transmitting channel constitutes a transceiver channel), and extract the FFT vector of the target echo of the transmission signal of each transmitting channel, synthesize the FFT vectors of different transmitting channels into a MIMO virtual array, form the FFT vector of the receiving channel, and then determine the speed information of the target object after solving the angle of the target based on the synthesized FFT vector.

[0092] In step a, for example, the processing process shown in FIG. 4 may be used. For each range gate data slice of each receiving channel, for example, for the Mth range gate FFT data slice, two Doppler gate FFT data slices may be obtained, wherein one Doppler gate FFT data slice is obtained by performing phase shift compensation based on the phase shift error coefficient of TX0, recorded as 2D FFT data slice 0, and the other Doppler gate FFT data slice is obtained by performing phase shift compensation based on the phase shift error coefficient of TX1, recorded as 2D FFT data slice 1. In other words, for the Doppler gate FFT data of an entire receiving channel, two types of Doppler gate FFT data may be obtained by extracting the phase shift error coefficients of different transmitting channels used in the phase shift compensation process, one type of Doppler gate FFT data is obtained by performing phase shift compensation based on the phase shift error coefficient of TX0, and the other type of Doppler gate FFT data is obtained by performing phase shift compensation based on the phase shift error coefficient of TX1.

[0093] In step b, after obtaining the Doppler gate FFT data of each receiving channel and classifying and extracting the phase error coefficients of different transmitting channels used in the phase shift compensation process, the extracted Doppler gate FFT data compensated by the phase error coefficient of the same transmitting channel can be merged to obtain multiple groups of Doppler gate FFT data corresponding to the corresponding transmitting channels. Figure 4a The M-th range gate shown in , by merging all the 2D FFT data slices 0 obtained by phase shift compensation based on the TX0 phase shift error coefficient of each receiving channel, can obtain the 2D FFT data slices 0 of multiple receiving channels ( Figure 5Similarly, all 2D FFT data slices 1 obtained by performing phase shift compensation based on the TX1 phase shift error coefficient of each receiving channel are combined to obtain 2D FFT data slices 1 of multiple receiving channels ( Figure 5 ).

[0094] It should be noted that in the 2D FFT data slice 0 of the multiple receiving channels, although all Doppler gate FFT data are obtained by performing phase shift compensation based on the TX0 phase shift error coefficient, since there are 2D FFT data corresponding to the echo signals of the two transmitting channels, there are two phase shift compensation situations when performing phase shift compensation based on the TX0 phase shift error coefficient: one is the phase shift compensation performed on the echo signal of the transmitting channel of TX0 based on the TX0 phase shift error coefficient, and the other is the phase shift compensation performed on the echo signal of the transmitting channel of TX1 based on the TX0 phase shift error coefficient. Obviously, only the first situation, that is, the phase shift compensation performed on the echo signal of the transmitting channel of TX0 based on the TX0 phase shift error coefficient, is an effective phase shift compensation. The corresponding 2D FFT data slice 0 is the effective Doppler gate FFT data.

[0095] Similarly, in the 2D FFT data slice 1 of the multiple receiving channels, although all Doppler gate FFT data are obtained by phase shift compensation based on the TX1 phase shift error coefficient, since there are 2D FFT data corresponding to the echo signals of the two transmitting channels, there are two phase shift compensation situations when phase shift compensation is performed based on the TX1 phase shift error coefficient: one is the phase shift compensation performed for the echo signal of the transmitting channel of TX0 based on the TX1 phase shift error coefficient, and the other is the phase shift compensation performed for the echo signal of the transmitting channel of TX1 based on the TX1 phase shift error coefficient. Obviously, only the second situation, that is, the phase shift compensation performed for the echo signal of the transmitting channel of TX1 based on the TX1 phase shift error coefficient, is an effective phase shift compensation. The corresponding 2D FFT data slice 1 is the effective Doppler gate FFT data.

[0096] Therefore, in step c, it is necessary to extract valid Doppler gate FFT data from each set of Doppler gate FFT data corresponding to each transmission channel. That is, for each set of Doppler gate FFT data, it is necessary to determine which Doppler gate FFT data is the echo signal of the corresponding transmission channel (which can be determined by Figure 5 The multi-channel combined detection is realized in the method), and the Doppler gate FFT data of the echo signal of the corresponding transmission channel are used as the target Doppler gate FFT data and extracted from the Doppler gate FFT data of the group.

[0097] For example Figure 5In the above, for each group of Doppler gate FFT data corresponding to the transmit channel of TX0, it is necessary to extract the part of Doppler gate FFT data that is the echo signal of the transmit channel of TX0 as the target Doppler gate FFT data (the corresponding operation is to extract the FFT vector of the target echo of TX0), and for each group of Doppler gate FFT data corresponding to the transmit channel of TX1, it is necessary to extract the part of Doppler gate FFT data that is the echo signal of the transmit channel of TX1 as the target Doppler gate FFT data (the corresponding operation is to extract the FFT vector of the target echo of TX1).

[0098] In step d, all target Doppler gate FFT data are merged, that is, the Doppler gate FFT data obtained by the effective phase shift compensation processing received by all receiving channels are merged to obtain a MIMO virtual array as a 2D FFT data vector of multiple receiving channels (the corresponding operation is Figure 5 The MIMO virtual array is synthesized in the MIMO virtual array to obtain the receiving FFT vector. The CFAR processing is performed on the 2D FFT data vector obtained after the combination to remove the false targets from the energy dimension. At this time, the target information carries the information of the distance and speed dimensions.

[0099] The steps of obtaining the phase shift error coefficient will be described below through at least one embodiment.

[0100] In some embodiments, the phase shift error coefficient of the transmit channel may be obtained by the following steps:

[0101] The chirp sequence corresponding to the transmission channel is compared with the standard chirp sequence generated by the standard phase shifter to obtain the phase shift error coefficient of the transmission channel.

[0102] For example, the chirp sequence sent by each transmitting channel of the DDM radar sensor under the modulation of the built-in phase shifter can be detected in advance, and a standard chirp sequence can be generated by a standard phase shifter. Then, the chirp sequence corresponding to the transmitting channel is compared with the standard chirp sequence to determine the compensation calculation relationship between the two and obtain the phase shift error coefficient of the transmitting channel.

[0103] The compensation calculation relationship may be, but is not limited to, a multiplication or addition calculation relationship.

[0104] In some embodiments, using the phase shift error coefficient of the transmission channel to perform phase shift compensation on the range-dimensional FFT data to obtain the range-dimensional FFT compensation data may include: multiplying or adding the phase shift error coefficient of the transmission channel with the range-dimensional FFT data to obtain the range-dimensional FFT compensation data.

[0105] For example Figure 4bIn the example, for the Mth range gate slice in the 1D FFT data plane, it can be multiplied by the phase shift error coefficient to obtain the compensated 1D FFT data slice.

[0106] In some embodiments, the phase shift error coefficient may include a phase shift phase error and / or a phase shift amplitude error.

[0107] The above method will be supplemented by at least one embodiment below, adding a processing process for determining the angle information of the target object.

[0108] In some embodiments, the target detection method further includes: determining angle information of the target object based on Doppler dimension FFT data.

[0109] For example, when radar detects a target object, in addition to detecting the position and speed of the target object, another important detection content is to determine the angle information of the target object, that is, the angle information of the target object relative to the monitoring point.

[0110] In some embodiments, determining the angle information of the target object based on the Doppler dimension FFT data may include: extracting and compensating the complex data at the peak of the Doppler dimension FFT data generated by the corresponding transmission channel from the Doppler dimension FFT data, and synthesizing the receiving vector to determine the angle information of the target object.

[0111] For example Figure 5 In the example, after obtaining the target 2D FFT data slices corresponding to TX0 and TX1, the complex values ​​at the 2D FFT peaks corresponding to TX0 and TX1 are extracted from the two sets of 2D FFT data slices, and a receiving vector is synthesized, which will expand the receiving array and improve the angular resolution. Due to the corresponding compensation, these 2D FFT complex values ​​are accurate, and after angle calculation, the accurate angle information of the target object can be obtained.

[0112] In some embodiments, the transmission link corresponding to the transmission channel in the embodiments of the present application may include an analog signal source and a phase shifter. The analog signal source may be configured to provide an initial analog signal, and the phase shifter may be configured to generate a phase-shifted signal, and shift the phase of the initial analog signal based on the phase-shifted signal, so as to perform a preset phase shift operation on the initial analog signal, thereby realizing the DDM wave transmission method; wherein, the phase shifter may be a delay line phase shifter or a digital phase shifter.

[0113] In some embodiments, when the phase shifter is a digital phase shifter, the digital phase shifter includes a digital phase shift signal source, a digital-to-analog converter and a mixer. The digital phase shifter can be configured to generate a digital phase shift signal, the digital-to-analog converter can be configured to convert the received digital phase shift signal into an analog phase shift signal, and the mixer can be configured to use the received analog phase shift signal to perform a mixing operation on the received initial analog signal so as to perform a preset phase shift operation on the initial analog signal. Figure 6 This is a simplified schematic diagram of a signal transmission link of a digital phase shifter architecture in an embodiment of the present application. Figure 7 Based on Figure 6 A schematic diagram of a digital phase shifter architecture in a signal transmission link is shown; Figure 8 It is a schematic diagram of a transmission link including a specific compensation module in an embodiment of the present application.

[0114] An embodiment of the present application provides a signal transmission link, including a transmission link for transmitting an electromagnetic wave signal and a receiving link for receiving an echo signal formed based on the electromagnetic wave signal; the transmission link includes an analog signal source and a digital phase shifter, the analog signal source can be configured to provide an initial analog signal, the digital phase shifter can be configured to generate a digital phase shift signal, and based on the digital phase shift signal, the initial analog signal is phase shifted to perform a preset phase shift operation on the initial analog signal; the receiving link includes an analog-to-digital converter and a digital baseband processing module, the analog-to-digital converter can be configured to perform analog-to-digital conversion on the received echo signal to obtain a digital baseband signal, and the digital baseband signal processing module can be configured to perform distance-dimensional Fourier transform and speed-dimensional Fourier transform on the digital baseband signal in sequence; wherein the digital baseband processing module includes a phase shift compensation unit, the phase shift compensation unit can be configured to use the phase shift error coefficient of the digital phase shifter in the transmission link to obtain phase shift compensation for distance-dimensional FFT data by performing distance-dimensional Fourier transform; and the digital baseband processing module can be configured to perform speed-dimensional Fourier transform based on the compensated distance-dimensional FFT data.

[0115] like Figure 6As shown, in some optional embodiments, a signal transmission link of a digital phase shifter architecture may include an analog signal source 21, a digital phase shifter (Digital PS) 22 and a transmitting antenna 23, that is, the analog signal source 21 may be configured to provide an LO signal, and the digital phase shifter 22 may be configured to perform a preset phase shift operation on the received LO signal, so that the phase-shifted LO signal is radiated to a preset spatial area through the transmitting antenna 23. The analog signal source 21 may also be an architecture including a phase-locked loop PLL, which may provide an electromagnetic wave (such as a laser, microwave, etc.) signal. The analog signal source 21, the digital phase shifter 22 and the transmitting antenna 23 may be integrated into an integrated device, or may be separate components; for example, the analog signal source 21 and the digital phase shifter 22 may be integrated into a package to form a SoC chip, and the transmitting antenna 23 may be connected through the peripheral port of the chip, and formed on a carrier such as a PCB board. Meanwhile, in some optional embodiments, the transmitting antenna 23 may also be integrated on the chip package to form AiP or AoP, with a chip structure having a packaged antenna.

[0116] like Figure 6 As shown, the digital phase shifter 22 in the embodiment of the present application may include a mixer (Mixer) 221, a digital-to-analog converter (i.e., DAC) 222 and a phase shift signal source (such as a digital baseband signal source Baseband) 223, etc., that is, the phase shift signal source 223 can be configured to provide a digital phase shift signal; the digital-to-analog converter 222 can be configured to perform analog-to-digital conversion on the received digital phase shift signal to convert the digital phase shift signal into an analog phase shift signal; the mixer 221 can be configured to perform a mixing operation on the received analog phase shift signal with the received electromagnetic wave signal from the analog signal source 21, so as to achieve a phase shift operation of setting the above-mentioned electromagnetic wave signal using the digital phase shift signal. Optionally, when the above-mentioned signal transmission link provides a swept frequency signal, such as providing an FMCW laser signal or an FMCW microwave signal, a swept frequency electromagnetic wave signal can be provided based on the analog signal source 21, and / or a swept frequency digital phase shift signal can be provided based on the phase shift signal source 223, so that after mixing by the mixer 221, a swept frequency continuous wave signal is output.

[0117] In some optional embodiments, based on Figure 6In the structure shown, the analog signal source 21 can be configured to provide FMCW signals in the centimeter wave band or millimeter wave band (such as 3.1 GHz, 24 GHz, 60 GHz, 77 GHz, etc.) of microwaves, and the phase-shifted signal source 223 can be configured to provide a digital phase-shifted signal at the MHz level (for example, 3 MHz to 5 MHz, such as 3 MHz, 4 MHz, 5 MHz, etc.), that is, the digital-to-analog converter 222 performs digital-to-analog conversion on the digital phase-shifted signal at the MHz level to obtain an analog phase-shifted signal in the corresponding frequency range, and the mixer 221 can be configured to perform up-mixing or down-mixing operations on the received millimeter wave band FMCW signal based on the received analog phase-shifted signal of the fixed frequency band, so as to realize the operation of preset phase shifting of the FMCW signal.

[0118] In some optional embodiments, the centimeter wave signal in the 3.1 GHz frequency band may include 3.1 GHz to 10.6 GHz, for example, 3.1 GHz, 5 GHz, 5 GHz, 6 GHz, 8 GHz, 10.6 GHz, etc.; the millimeter wave signal in the 77 GHz frequency band may include 76 GHz to 81 GHz signals, for example, swept frequency signals such as 76 GHz to 77 GHz, 77 GHz to 79 GHz, 79 GHz to 81 GHz, or fixed frequency band signals such as 76 GHz, 77 GHz, 78 GHz, 79 GHz, 80 GHz, 81 GHz, etc.

[0119] based on Figure 6 In the structure shown, since the phase-shift signal source 223 provides a digital signal, in order to further adapt the signal characteristics, the mixer 221 can be set as an IQ Mixer, and the digital-to-analog converter 222 is an IQ DAC. At the same time, the phase-shift signal source 223 can be configured to provide a digital baseband signal source (DDFS) for phase shifting and / or provide a corresponding source signal as a waveform controller (WaveformControl).

[0120] For example, Figure 7As shown, the signal transmission link (TX digital phase shifter architecture) of the digital phase shifter architecture may include a digital baseband signal source (Baseband), a direct digital frequency synthesizer (Direct Digital Frequency Synthesizer, DDFS for short), an IQ digital to analog converter (Digital to Analog Convertor, DAC for short), a low-pass filter (Low-Pass Filter, LPF for short), an IQ modulator (IQ modulator / IQ Mixer), a power amplifier (Power Amplifier, PA), etc., that is, the baseband signal source is configured to provide a digital phase shift source signal, and the direct digital frequency synthesizer may be configured to implement CDM (Code-Division Multiplexing), DDM (Doppler Division Multiplexing), TDM (Time-Division Multiplexing), SDM (Space Division Multiplexing), CSD (Circuit Switch Data), Digital IF (Digital Intermediate Multiplexing) based on the received source signal. Frequency, digital intermediate frequency) and other signal waveforms and wave transmission methods to achieve flexible configuration of signal transmission form and transmission waveform. Among them, the signal amplified by the power amplifier can be radiated to a preset space area through a transmitting antenna packaged as an integral unit or externally.

[0121] For the signal transmission link of the digital phase shifter architecture in the embodiment of the present application, since the digital phase shifter architecture is configured to generate a baseband signal sequence in the digital domain, and can generate an analog baseband signal through a DAC, and then modulate the transmission signal to a high frequency through an orthogonal mixer, that is, because the baseband signal of the architecture is generated in the digital domain, it has better orthogonality and lower sidelobes, so its phase shift phase can be generated very accurately, resulting in higher phase modulation accuracy.

[0122] In some optional embodiments, for the signal transmission link of the digital phase shifter architecture, when the RF LO frequency sweep is used to realize that the transmitted signal is an FMCW signal, a compensation module can be added to the signal transmission link to solve the TX IQ imbalance (Imbalance), signal leakage (such as TX LO Leakage) and harmonic distortion (HD) caused by IQ mismatch. As shown in FIG4, a compensation module (TX compensation) can be set between the TX DDFS and the IQ DAC to perform calibration and compensation operations on the signal transmission link of the digital phase shifter architecture to achieve an operation to solve at least one of the above problems. Among them, the HD caused by the third-order nonlinearity of the baseband can be referred to as HD3.

[0123] In some optional embodiments, such as Figure 7-8 As shown, the compensation module (TX compensation) may include at least one of an LO compensation unit (TX LO leakage compensation), an IQ compensation unit (TX IQ Imbalance compensation) and an HD3 compensation unit (TX HD3 compensation), that is, the LO compensation unit may be configured to perform compensation operations for signal leakage, the IQ compensation unit may be configured to perform compensation operations for IQ imbalance, and the HD3 compensation unit may be configured to perform compensation operations for the above-mentioned HD3. The LO compensation unit is configured to be used to perform compensation operations for at least one of an IQ modulator imbalance (TX IQ Modulator imbalance) and an IQ channel imbalance (IQ channel imbalance). In addition, when the compensation module includes at least two of the LO compensation unit, the IQ compensation unit and the HD3 compensation unit, compensation may be performed synchronously (such as in parallel) or sequentially (such as in series) according to actual needs and signal characteristics. Figure 5 As shown, IQ compensation may be performed first, then LO compensation, and finally HD3 compensation.

[0124] In some optional embodiments, the signal transmission link of the digital phase shifter architecture may also include an error correction module (TX DAC Board Error Correction) for DAC and an AWGN (additive white gaussian noise) module for Gaussian white noise, etc., which are not shown in the figure and can be added or deleted according to actual needs. Among them, I in IQ mentioned in the embodiments of the present application can be expressed as the abbreviation of In-Phase (i.e., in-phase), Q can be expressed as the abbreviation of Quadrature (i.e., orthogonal), and RF can be expressed as the abbreviation of Radio Frequency (i.e., radio frequency).

[0125] like Fig. 9 As shown, a transceiver link may include a transmitting link and a receiving link, etc. The transmitting link may include a digital baseband signal source (Baseband), a direct digital frequency synthesizer (TX DDFS), an IQ digital-to-analog converter (IQDAC), a low-pass filter (LPF), an IQ modulator (IQ Modulator), a power amplifier (PA), etc. connected in sequence, and the signal amplified by the power amplifier is radiated to a preset spatial area through a transmitting antenna. The receiving link may include a low noise amplifier (LNA), a real mixer (Real Mixer), a trans-impedance amplifier (TIA), a low-pass filter (LPF), a high-pass filter (HPF), a real digital-to-analog converter (Real ADC), etc. connected in sequence. That is, the echo signal received by the receiving antenna is processed by the above-mentioned LNA, Real Mixer, TIA, LPF, HPF and Real ADC in sequence and converted into a real digital baseband signal. The subsequent digital signal processing module processes the real digital baseband signal to obtain parameter information of the target such as distance, speed, angle, height and micro-motion characteristics.

[0126] In the above transceiver link, the frequency of the TX-LO signal received by the IQ modulator in the transmitting link and the frequency of the RX-LO signal received by the Real Mixer in the receiving link may be the same. Figure 6 As shown, if the signal output by the Baseband is a sine wave of x MHz, then the TX-LO signal and the RX-LO signal can both be sine waves of z GHz.

[0127] exist Fig. 9In the embodiment shown, the transmitting link adopts a digital phase-shifting architecture, while the receiving link can adopt components of an analog architecture, that is, no IQ components are used, so it can effectively be compatible with sensors of the receiving link of the analog architecture, effectively reducing the development cost of the entire transceiver link system.

[0128] Optionally, in an embodiment of the present application, the receiving link may include a receiving antenna, that is, the receiving antenna may be connected through a peripheral port of the chip and formed on a carrier such as a PCB board. At the same time, in some optional embodiments, the receiving antenna may also be integrated on the chip package to form AiP or AoP, that is, a chip structure with a packaged antenna.

[0129] In some optional embodiments, in order to match the transmission link of the digital phase shifter architecture, relevant adjustments may be made to the receiving link, such as Figure 7 The transceiver link shown may include Figure 6 The similar transmission link architecture and receiving link in FIG. 1 (in order to avoid redundancy, the same parts will not be described here), that is, Figure 6 The RealMixer in the receiving link is adjusted to an IQ demodulator (IQ Demodulator), and the Real ADC is adjusted to an IQ ADC. That is, the receiving link may include a low noise amplifier (LNA), an IQ demodulator (IQ Demodulator), a transimpedance amplifier (TIA), a low pass filter (LPF), a high pass filter (HPF), an IQ digital-to-analog converter (IQ ADC), etc. connected in sequence. That is, the echo signal received by the receiving antenna is processed by the above-mentioned LNA, IQ Demodulator, TIA, LPF, HPF and IQ ADC in sequence and converted into an IQ digital baseband signal. The subsequent digital signal processing module processes the IQ digital baseband signal to obtain parameter information of the target such as distance, speed, angle, height and micro-motion characteristics (i.e., micro-Doppler).

[0130] In some optional embodiments, Fig. 9 In the signal transmission and reception link of the digital phase shifter architecture shown, the digital baseband processing module in the receiving link (not shown in the figure, and can be arranged after the IQADC) can also include a phase compensation unit for compensating for the phase shifter error. The phase compensation unit can obtain the distance dimension FFT data by Fourier transforming the distance dimension, and use the phase error coefficient of the transmitting channel (or transmitting link) to perform phase compensation on the distance dimension FFT data to obtain the distance dimension FFT compensation data; subsequently, based on the distance dimension FFT compensation data, such as speed dimension FFT and wave arrival estimation are performed to realize the detection of target data. Specifically, it can be implemented according to the method described in any embodiment of the present application.

[0131] It should be noted that the above examples in this embodiment are all illustrative descriptions for easy understanding and do not constitute a limitation on the technical solutions of the present invention.

[0132] The step division of the above methods is only for the purpose of clear description. When implemented, they can be combined into one step or some steps can be split and decomposed into multiple steps. As long as they include the same logical relationship, they are all within the scope of protection of this patent; adding insignificant modifications to the algorithm or process or introducing insignificant designs without changing the core design of the algorithm and process are all within the scope of protection of this patent.

[0133] Another embodiment of the present application relates to an integrated circuit. The implementation details of the integrated circuit of this embodiment are specifically described below. The following content is only for the convenience of understanding the implementation details provided, and is not necessary for implementing this solution. The schematic diagram of the integrated circuit of this embodiment can be as follows Fig.10 As shown, it includes: a signal transceiver channel 31, which is used to transmit radio signals and receive echo signals formed by the radio signals being reflected by the target object; a target detection module 32, which is used to detect the target object according to the echo signal based on the method described in any of the above embodiments.

[0134] Optionally, the integrated circuit may specifically be a chip structure, such as a millimeter wave radar chip, etc. Of course, the integrated circuit may also be implemented by other hardware.

[0135] In some embodiments, the target detection module 32 may include: a first processing unit, used to perform a range-dimensional Fourier transform on the echo signal received by the receiving channel to obtain a range-dimensional FFT data, and use the phase shift error coefficient of the transmitting channel to perform phase shift compensation on the range-dimensional FFT data to obtain a range-dimensional FFT compensated data; a second processing unit, used to perform a Doppler-dimensional Fourier transform on the range-dimensional FFT compensated data to obtain a Doppler-dimensional FFT data; a detection unit, used to determine the velocity information of the target object based on the Doppler-dimensional FFT data.

[0136] It is not difficult to find that this embodiment is an integrated circuit embodiment corresponding to the above method embodiment, and this embodiment can be implemented in conjunction with the above method embodiment. The relevant technical details and technical effects mentioned in the above embodiment are still valid in this embodiment, and in order to reduce repetition, they are not repeated here. Accordingly, the relevant technical details mentioned in this embodiment can also be applied in the above embodiment.

[0137] It is worth mentioning that all modules involved in this embodiment are logic modules. In practical applications, a logic unit can be a physical unit, a part of a physical unit, or a combination of multiple physical units. In addition, in order to highlight the innovative part of this application, this embodiment does not introduce units that are not closely related to solving the technical problems proposed by this application, but this does not mean that there are no other units in this embodiment.

[0138] Another embodiment of the present application relates to a radio device, including: a carrier; an integrated circuit as in the above embodiment, disposed on the carrier; and an antenna, disposed on the carrier and used for transmitting and receiving radio signals.

[0139] The integrated circuit is connected to the antenna via a first transmission line for transmitting and receiving radio signals. The carrier may be a printed circuit board PCB, and the first transmission line may be a PCB trace. In addition, the integrated circuit may be integrated with the antenna to form a device such as AiP, AoP or AoC.

[0140] Another embodiment of the present application relates to a terminal device, comprising: a device body; a radio device as described in any one of the above items arranged on the device body, the radio device being used for target detection and / or communication.

[0141] Specifically, based on the above embodiments, in one embodiment of the present application, the radio device can be arranged outside the device body, in another embodiment of the present application, the radio device can also be arranged inside the device body, and in other embodiments of the present application, the radio device can also be arranged partially inside the device body and partially outside the device body. The present application does not limit this, and it depends on the specific situation.

[0142] It should be noted that radio devices can achieve functions such as target detection and communication by transmitting and receiving signals.

[0143] In an optional embodiment, the above-mentioned device body can be a component and product used in fields such as smart housing, transportation, smart home, consumer electronics, monitoring, industrial automation, cabin detection and health care; for example, the device body can be intelligent transportation equipment (such as cars, bicycles, motorcycles, ships, subways, trains, etc.), security equipment (such as cameras), smart wearable devices (such as bracelets, glasses, etc.), smart home equipment (such as TVs, air conditioners, smart lights, etc.), various communication equipment (such as mobile phones, tablet computers, etc.), and gates, smart traffic lights, smart signs, traffic cameras and various industrial manipulators (or robots), etc., and can also be various instruments for detecting life characteristic parameters and various equipment equipped with the instrument. The radio device can be the radio device described in any embodiment of the present application. The structure and working principle of the radio device have been described in detail in the above embodiments, and will not be repeated here.

[0144] Another embodiment of the present application relates to a computer-readable storage medium storing a computer program, which implements the above method embodiment when executed by a processor.

[0145] That is, those skilled in the art can understand that all or part of the steps in the above-mentioned embodiment method can be completed by instructing the relevant hardware through a program, and the program is stored in a storage medium, including a number of instructions to enable a device (which can be a single-chip microcomputer, chip, etc.) or a processor to execute all or part of the steps of the method described in each embodiment of the present application. The aforementioned storage medium includes: U disk, mobile hard disk, read-only memory (Read-Only Memory, referred to as: ROM), random access memory (Random Access Memory, referred to as: RAM), disk or optical disk and other media that can store program codes.

[0146] Each embodiment in this specification is described in a progressive manner, and the same or similar parts between the embodiments can be referred to each other, and each embodiment focuses on the differences from other embodiments. In particular, for the integrated circuit embodiment, since it is basically similar to the method embodiment, the description is relatively simple, and the relevant parts can be referred to the partial description of the method embodiment. The device embodiment described above is only schematic, in which the modules described as separate components may or may not be physically separated, and the components displayed as modules may or may not be physical modules, that is, they may be located in one place, or they may be distributed on multiple network units. Some or all of the modules can be selected according to actual needs to achieve the purpose of the scheme of this embodiment. Ordinary technicians in this field can understand and implement it without paying creative work.

[0147] Those skilled in the art will appreciate that the above embodiments are specific embodiments for implementing the present application, and in actual applications, various changes may be made thereto in form and detail without departing from the spirit and scope of the present application.

Claims

1. A target detection method, characterized in that: Applied to a DDM radar sensor having at least one transceiver channel, the method comprises: Performing range-dimensional Fourier transform processing on the echo signal received by the receiving channel to obtain range-dimensional FFT data, and performing phase shift compensation on the range-dimensional FFT data using the phase shift error coefficient of the transmitting channel to obtain range-dimensional FFT compensation data; Performing Doppler dimension Fourier transform processing on the distance dimension FFT compensation data to obtain Doppler dimension FFT data; Based on the Doppler FFT data, velocity information of the target object is determined.

2. The target detection method according to claim 1, characterized in that: The adopting the phase shift error coefficient of the transmitting channel to perform phase shift compensation on the distance dimension FFT data comprises: For the data corresponding to each chirp sequence in the range-dimensional FFT data, phase shift compensation is performed using the phase shift error coefficients of the transmission channels corresponding to each chirp sequence.

3. The target detection method according to claim 1, characterized in that: The method further comprises: Based on the Doppler FFT data, angle information of the target object is determined.

4. The target detection method according to claim 3, characterized in that: The determining the angle information of the target object based on the Doppler FFT data includes: The complex data at the peak of the Doppler dimension FFT data generated by the corresponding transmission channel is extracted and compensated from the Doppler dimension FFT data, and a receiving vector is synthesized to determine the angle information of the target object.

5. The target detection method according to claim 2, characterized in that: The range dimension FFT data is composed of a plurality of range gate FFT data, and the phase shift compensation is performed using the phase shift error coefficients of the transmission channels corresponding to the chirp sequences in the range dimension FFT data, respectively, including: For each of the range gate FFT data, phase shift compensation is performed using the phase shift error coefficients of the transmission channels corresponding to each of the chirp sequences to obtain range gate FFT compensation data corresponding to each of the range gate FFT data.

6. The target detection method according to claim 5, characterized in that: The performing Doppler dimension Fourier transform processing on the distance dimension FFT compensation data to obtain Doppler dimension FFT data includes: The Doppler dimension Fourier transform processing is performed on each of the range gate FFT compensation data to obtain Doppler gate FFT data corresponding to each of the range gate FFT compensation data.

7. The target detection method according to claim 6, characterized in that: The transceiver channel is composed of at least two transmitting channels and one receiving channel, and the speed information of the target object is determined based on the Doppler dimension FFT data, including: Based on all the Doppler gate FFT data, the velocity information of the target object is determined.

8. The target detection method according to claim 6, characterized in that: The transceiver channel is composed of at least two transmitting channels and at least two receiving channels, and the speed information of the target object is determined based on the Doppler dimension FFT data, including: Extracting Doppler gate FFT data in each receiving channel respectively, wherein the Doppler gate FFT data in each receiving channel includes Doppler gate FFT data compensated by phase shift error coefficients of different transmitting channels; Combining the Doppler gate FFT data in all the receiving channels that have been compensated by the phase shift error coefficient of the same transmitting channel to obtain multiple groups of Doppler gate FFT data corresponding to each of the transmitting channels; For each group of the Doppler gate FFT data, extract target Doppler gate FFT data therefrom, wherein the target Doppler gate FFT data is the echo signal of the transmission channel corresponding to the group of the Doppler gate FFT data; The target Doppler gate FFT data extracted from each set of the Doppler gate FFT data are synthesized, and the velocity information of the target object is determined based on the target Doppler gate FFT data extracted from each set of the Doppler gate FFT data.

9. The target detection method according to any one of claims 2 to 8, characterized in that: The phase shift error coefficient of the transmitting channel is obtained by the following steps: The chirp sequence corresponding to the transmission channel is compared with a standard chirp sequence generated by a standard phase shifter to obtain a phase shift error coefficient of the transmission channel.

10. The target detection method according to claim 9, characterized in that: The adopting the phase shift error coefficient of the transmitting channel to perform phase shift compensation on the distance dimension FFT data to obtain the distance dimension FFT compensation data comprises: The phase shift error coefficient of the transmitting channel is multiplied or added to the range-dimensional FFT data to obtain range-dimensional FFT compensation data.

11. The target detection method according to claim 2, characterized in that: The phase shift error coefficient includes a phase shift error and / or a phase shift amplitude error.

12. The target detection method according to claim 4, characterized in that: The angle information of the target object includes the pitch angle and / or azimuth angle of the target object.

13. The target detection method according to any one of claims 1 to 12, characterized in that: The transmission link corresponding to the transmission channel includes an analog signal source and a phase shifter, the analog signal source can be configured to provide an initial analog signal, the phase shifter can be configured to generate a phase-shifted signal, and shift the phase of the initial analog signal based on the phase-shifted signal to perform a preset phase shift operation on the initial analog signal; wherein the phase shifter is a delay line phase shifter or a digital phase shifter.

14. The target detection method according to claim 13, characterized in that: When the phase shifter is a digital phase shifter, the digital phase shifter includes a digital phase shift signal source, a digital-to-analog converter and a mixer. The digital phase shifter can be configured to generate a digital phase shift signal. The digital-to-analog converter can be configured to convert the received digital phase shift signal into an analog phase shift signal. The mixer can be configured to perform a mixing operation on the received initial analog signal using the received analog phase shift signal so as to perform a preset phase shift operation on the initial analog signal.

15. A signal transceiver link, characterized in that: It includes a transmitting link for transmitting an electromagnetic wave signal and a receiving link for receiving an echo signal formed based on the electromagnetic wave signal; The transmission link includes an analog signal source and a digital phase shifter, wherein the analog signal source may be configured to provide an initial analog signal, and the digital phase shifter may be configured to generate a digital phase-shifted signal, and perform a phase shift on the initial analog signal based on the digital phase-shifted signal, so as to perform a preset phase shift operation on the initial analog signal; The receiving link includes an analog-to-digital converter and a digital baseband processing module, wherein the analog-to-digital converter may be configured to perform analog-to-digital conversion on the received echo signal to obtain a digital baseband signal, and the digital baseband signal processing module may be configured to perform distance-dimensional Fourier transform and velocity-dimensional Fourier transform on the digital baseband signal in sequence; The digital baseband processing module includes a phase compensation unit, and the phase compensation unit can be configured to adopt a phase error coefficient of a digital phase shifter in the transmission link to obtain a phase compensation for the distance-dimensional FFT data by Fourier transforming the distance-dimensional FFT data; and The digital baseband processing module may be configured to perform the speed dimension Fourier transform according to the compensated distance dimension FFT data.

16. A computer-readable storage medium storing a computer program, characterized in that: When the computer program is executed by a processor, the target detection method according to any one of claims 1 to 14 is implemented.

17. An integrated circuit, characterized in that: include: A signal transceiver channel, used for transmitting a radio signal and receiving an echo signal formed by the radio signal being reflected by a target object; A target detection module, configured to detect a target object according to the echo signal based on the method according to any one of claims 1 to 14.

18. The integrated circuit according to claim 17, characterized in that The target detection module comprises: The first processing unit is used to perform a range-dimensional Fourier transform process on the echo signal received by the receiving channel to obtain range-dimensional FFT data, and use the phase shift error coefficient of the transmitting channel to perform phase shift compensation on the range-dimensional FFT data to obtain range-dimensional FFT compensation data; A second processing unit is used to perform Doppler dimension Fourier transform processing on the distance dimension FFT compensation data to obtain Doppler dimension FFT data; The detection unit is used to determine the speed information of the target object based on the Doppler dimension FFT data.

19. A radio device, characterized in that: include: Carrier; The integrated circuit as claimed in any one of claims 17 to 18, arranged on the carrier; The antenna is arranged on the carrier and is used for sending and receiving radio signals.

20. A terminal device, characterized in that: include: Equipment body; The radio device according to claim 19 is arranged on the device body, and is used for target detection and / or communication.

Images