Phase-shift control method for DAB converter without DC bias

Abstract

The present invention discloses a phase-shift control method for a DAB converter without DC bias. This method introduces a new control variable D4 to control the drive signal, and is applicable to various phase-shift control strategies, including single-phase, extended-phase, dual-phase, and triple-phase. The proposed method only requires sampling the input and output voltage values, eliminating the need for additional hardware circuitry. The initial and final values ​​of the auxiliary inductor current within a single switching cycle are both zero, and the inductor current is completely symmetrical within each switching cycle. This solves the power shortage problem in the first half of the switching cycle, which is present in commonly used DC bias elimination methods, and ensures good dynamic response performance from a control perspective. The present invention is simple to implement while ensuring good dynamic response performance.

Description

Technical Field

[0001] The present invention relates to the field of power electronic control, in particular to a phase-shift control method for a DAB converter without direct current bias. Background Art

[0002] Due to their excellent soft-switching characteristics, wide voltage regulation range, electrical isolation, and high transmission power levels, DAB converters have been widely used in energy storage systems, electric vehicle charging, and DC transformers (L.Li, G.Xu, D.Sha, Y.Liu, Y.Sun, and M.Su, "Review of Dual-Active-Bridge Converters With Topological Modifications," in IEEE Transactions on Power Electronics, vol. 38, no. 7, pp. 9046-9076, July 2023, doi: 10.1109 / TPEL.2023.3258418.). DAB converters widely adopt phase-shift control strategies such as single phase shift (SPS), extended phase shift (EPS), double phase shift (DPS), and three phase shift (TPS). These strategies change the direction and magnitude of the converter's transmitted power by controlling the phase shift angle. However, during converter startup or load switching, the phase shift angle between adjacent switching cycles can change suddenly, causing a DC bias current to flow through the high-frequency transformer and auxiliary inductor (S.Mu, Z.Guo and Y.Luo, "Universal Modulation Scheme to Suppress TransientDC Bias Current in Dual Active Bridge Converters," in IEEE Transactions on Power Electronics, vol. 37, no. 2, pp. 1322-1333, Feb. 2022, doi: 10.1109 / TPEL.2021.3104628.). This DC bias increases the current stress on the switching devices, reducing converter efficiency and shortening their service life. Excessive DC bias current can even cause magnetic saturation in the transformer and auxiliary inductor, and in severe cases, can directly burn out the circuit.

[0003] Currently, there are two main control methods for suppressing and eliminating DC bias current: one is to correct the inductor current in the first half of the switching cycle (Chen Yandong, Wang Zili, Liang Yunfei, et al. DC Bias Elimination Method and Control System for Inductor Current in DAB Converters [P]. Zhejiang Province: CN115995981A, 2023-04-21). The other is to add an additional control time between two adjacent switching cycles to correct the inductor current (Yang Ping, Wu Wenrong, Wang Mixin, et al. A Method for Suppressing Transient Bias in DAB Four-Degree-of-Freedom Asymmetric Modulation [P]. Sichuan Province: CN117013793A, 2023-11-07). Method 1 results in an imbalance in the transmitted power between the first and second half of the switching cycle, resulting in a power shortage; method 2 has high control complexity and cannot guarantee the amount of transmitted power during transients. While both existing control methods suppress the current peak amplitude, they also reduce the system's dynamic response speed. Summary of the Invention

[0004] The technical problem to be solved by the present invention is to provide a phase shift control method for a DAB converter without DC bias in view of the shortcomings of the existing technology, so as to completely eliminate the generation of DC bias.

[0005] To solve the above technical problems, the technical solution adopted by the present invention is: a phase shift control method for a DAB converter without DC bias, comprising the following steps:

[0006] S1, sampling DAB converter input voltage V in With the output voltage V o ;

[0007] S2, output reference voltage V oref With the output voltage V o The voltage error is obtained by making a difference, and the voltage error is used as the input of the voltage controller to obtain the output power reference value P oref ;

[0008] S3. Define the current reference power P b =N*V in *V o / (2*f s *L r ), where N is the transformer ratio, f s is the switching frequency, L r The auxiliary inductance value is defined as P per unit power. n =P oref / P b , define the voltage conversion ratio d=N*V o / V in; Define D1 and D2 as the duty cycle of the primary and secondary full-bridge voltage waveforms respectively, D3 as the phase shift angle of the rising edge of the primary and secondary full-bridge voltage waveforms, and D4 as the phase shift angle of the forward shift of the drive waveform;

[0009] S4, based on per unit power P n The value of the voltage conversion ratio d determines the working mode of the DAB converter, and calculates the values ​​of D1, D2, and D3 corresponding to the current mode. in 、V o And the phase shift angle D4 of the driving waveform is obtained by calculating the D1, D2, and D3 modal parts;

[0010] S5. Generate a PWM signal according to the obtained values ​​of D1, D2, D3, and D4, and use the PWM signal as a driving signal for the DAB converter.

[0011] The present invention introduces a new control degree of freedom D4 to control the drive signal, so that the initial value and the final value of the inductor current in a single switching cycle are both 0, realizing the decoupling of the inductor current in adjacent switching cycles, and can completely eliminate the generation of DC bias in control, ensuring the smooth transition of the DAB converter in transient processes such as load jump; the initial value and the final value of the inductor current in each switching cycle can be made 0, realizing the decoupling of the inductor current in adjacent switching cycles, completely eliminating the DC bias current of the inductor from the control, and solving the problem of power shortage in the first half of the switching cycle, and having good dynamic response performance.

[0012] The modes of operation of the DAB converter include: when d < 1 and 0 ≤ P n ≤d*(1-d) / 2, the DAB converter operates in mode 1f; when d<1 and d*(1-d) / 2 <P n ≤0.25, the DAB converter operates in mode 2f; when d>1 and 0≤P n ≤(d-1) / 2 / d 2 When d>1 and (d-1) / 2 / d 2 <P n ≤0.25, the DAB converter operates in mode 4f; when d<1 and -d*(1-d) / 2≤P n <0, the DAB converter operates in mode 1r; when d<1 and -0.25≤P n <-d*(1-d) / 2, the DAB converter operates in mode 2r; when d>1 and -(d-1) / 2 / d 2 ≤P n <0, the DAB converter operates in mode 3r; when d>1 and -0.25≤P n <(d-1) / 2 / d 2When , the DAB converter works in mode 4r; among them, mode 1f, mode 2f, mode 3f, and mode 4f work in the forward power transmission mode; mode 1r, mode 2r, mode 3r, and mode 4r work in the reverse power transmission mode.

[0013] The expressions of D1, D2, and D3 are as follows:

[0014]

[0015]

[0016]

[0017]

[0018]

[0019]

[0020]

[0021]

[0022] The calculation formula of the phase shift angle D4 of the driving waveform is:

[0023]

[0024] The driving waveform generated by the duty cycles D1, D2, and D3 is corrected according to the phase shift angle D4 of the driving waveform.

[0025] Compared with the prior art, the beneficial effects of the present invention are as follows: in order to eliminate the DC bias current that occurs during transient processes such as startup and load switching of the DAB converter, the present invention proposes a phase-shift control method without DC bias. The proposed control method introduces a new control variable D4 to control the drive signal, and is applicable to various phase-shift control strategies such as single phase shift, extended phase shift, double phase shift, and triple phase shift. The proposed method only needs to sample the input voltage and output voltage values, without adding additional hardware circuits. The initial value and final value of the auxiliary inductor current are both 0 in a single switching cycle, and the inductor current is completely symmetrical in each switching cycle, which solves the problem of power shortage in the first half of the switching cycle that exists in the commonly used DC bias elimination method, and ensures good dynamic response performance from the control perspective. The implementation process of the present invention is simple, while ensuring good dynamic response performance. BRIEF DESCRIPTION OF THE DRAWINGS

[0026] Figure 1 This is the topological structure diagram of the DAB converter;

[0027] Figure 2Schematic diagram of the three-phase shift modulation strategy and inductor current waveform of the DAB converter;

[0028] Figure 3 The drive and current waveforms without and with DC bias optimization during forward power transmission.

[0029] Figure 4 The driving and current waveforms without and with DC bias optimization during reverse power transmission.

[0030] Figure 5 This is a control block diagram of a control method according to an embodiment of the present invention;

[0031] Figure 6 The inductor current waveform is not optimized for DC bias when the load changes.

[0032] Figure 7 Inductor current waveform optimized for adding DC bias when the load changes. DETAILED DESCRIPTION

[0033] To make the objectives, technical solutions, and advantages of the embodiments of the present invention more clear, the technical solutions in the embodiments of the present invention will be clearly and completely described below in conjunction with the accompanying drawings in the embodiments of the present invention. Obviously, the described embodiments are only part of the embodiments of the present invention, not all of the embodiments. Based on the embodiments of the present invention, all other embodiments obtained by ordinary technicians in this field without making creative efforts shall fall within the scope of protection of the present invention.

[0034] like Figure 1 The topology diagram of the DAB converter is shown in the figure. The converter consists of eight switching tubes and a high-frequency transformer. Among them, P1, P2, P3, and P4 are the primary switching devices of the converter, S1, S2, S3, and S4 are the secondary switching devices of the converter, and T r With L r They are high frequency transformer and auxiliary inductor respectively, and the transformer ratio is N:1. Its input end is connected to a DC power supply with an input voltage of V in , C i is the input filter capacitor; the output end is connected to the load R, and the output voltage is V o , C ois the output filter capacitor. The source of switch tube P1 is connected to the drain of P2 and to the same-name terminal of the transformer primary winding; the source of switch tube P3 is connected to the drain of P4 and to the non-same-name terminal of the transformer primary winding; the drains of switch tubes P1 and P3 are both connected to the positive pole of the voltage source, and the sources of switch tubes P2 and P4 are both connected to the negative pole of the voltage source. The source of switch tube S1 is connected to the drain of S2 and to the same-name terminal of the transformer secondary winding; the source of switch tube S3 is connected to the drain of S4 and to the non-same-name terminal of the transformer secondary winding; the drains of switch tubes S1 and S3 are both connected to the positive terminal of the load, and the sources of switch tubes S2 and S4 are both connected to the negative terminal of the load.

[0035] like Figure 2 The figure shows the schematic diagram of the three-phase shift modulation strategy and inductor current waveform of the DAB converter. D1 and D2 are the duty cycles of the primary and secondary full-bridge voltage waveforms, respectively, and D3 is the phase shift angle of the rising edge of the primary and secondary full-bridge voltage waveforms. D1′, D2′, and D3′ are the corresponding values ​​of the three phase shift angles in the second switching cycle. When the load suddenly changes and causes the phase shift angle of two adjacent switching cycles to change, the characteristic of the inductor current not to change suddenly will cause a DC bias to appear in the next switching cycle, such as Figure 2 The ideal inductor current in the second switching cycle does not have a DC bias, as shown in the dashed line. Figure 2 The gray solid line shows this. In a real circuit, the DC bias current of the inductor gradually approaches zero due to the damping in the circuit. However, the transient DC bias can produce large current peaks, seriously threatening the normal and stable operation of the DAB converter.

[0036] The basic principle of the phase shift control method for a DAB converter without DC bias proposed in the embodiment of the present invention is as follows: Figure 3 and Figure 4 shown. Figure 3 (a) shows the driving signal and inductor current waveforms during forward power transmission without DC bias optimization; Figure 3 (b) shows the driving signal and inductor current waveform after DC bias optimization during forward power transmission; Figure 4 (a) shows the driving signal and inductor current waveforms during reverse power transmission without DC bias optimization; Figure 4 (b) shows the driving signal and inductor current waveform after DC bias optimization during reverse power transmission.

[0037] For forward power transfer, as Figure 3 As shown, by sampling the input voltage V in , output voltage V o, and the calculated D1, D2, D3, the phase shift angle D4 at the zero-crossing point of the inductor current can be calculated. At this time, D4>0, and the generation of the drive signal can be adjusted by D4 to obtain the optimized drive signal as follows Figure 3 As shown in (b) in .

[0038] For reverse power transfer, such as Figure 4 As shown, a similar method can be used to calculate the phase shift angle D4 at the zero-crossing point of the inductor current. At this time, D4<0. By adjusting the generation of the drive signal through D4, the optimized drive signal can be obtained as shown in FIG. Figure 4 As shown in (b) in .

[0039] By introducing a new control variable, D4, to adjust the drive waveform, the initial and final values ​​of the inductor current within each switching cycle are both set to zero. This decouples the inductor currents between adjacent switching cycles, completely eliminating the inductor's DC bias current. This also addresses the power shortfall during the first half of the switching cycle, resulting in excellent dynamic response. This method is applicable to a variety of phase-shift control strategies, including single-phase, extended-phase, dual-phase, and triple-phase.

[0040] Figure 6 and Figure 7 The inductor current waveforms are shown before and after DC bias optimization when the load changes.

[0041] The specific implementation methods are as follows:

[0042] Figure 5 The figure shows the implementation process of the proposed control method in the DAB converter.

[0043] First, sample the input voltage V in With the output voltage V o , calculate the voltage conversion ratio of the converter d = N * V o / V in With reference power P b =N*V in *V o / (2*f s *L r ). Where N is the transformer ratio, f s is the switching frequency of the DAB converter, L r is the inductance of the auxiliary inductor. oref and the output voltage sampling value V o The voltage error obtained by the difference is output as the output power reference value P through the voltage controller. oref , calculate the per unit power as P n =P oref / P b .

[0044] The calculation of the primary-secondary duty cycle and phase shift angles D1, D2, and D3 is determined by the converter's primary and secondary voltages and the power they process. Numerous researchers have optimized the calculation methods for D1, D2, and D3 to optimize the reactive circulating current of DAB converters. The present invention focuses on eliminating the converter's DC bias by introducing a new control variable, D4, rather than the calculation method for D1, D2, and D3. Therefore, the following analysis uses the maximum current stress optimization modulation strategy proposed in the literature (S.Shao, M.Jiang, W.Ye, Y.Li, J.Zhang and K.Sheng, "Optimal Phase-Shift Control to Minimize Reactive Power for a Dual ActiveBridge DC-DC Converter," in IEEE Transactions on Power Electronics, vol. 34, no. 10, pp. 10193-10205, Oct. 2019, doi: 10.1109 / TPEL.2018.2890292.) as an example to calculate the primary-secondary duty cycle and phase-shift angles D1, D2, and D3. Based on the calculated values ​​of D1, D2, and D3, the calculation method for the phase-shift angle D4, which shifts the drive waveform forward, is analyzed.

[0045] According to d and P n The circuit can be divided into 8 working modes. When d<1 and 0≤P n ≤d*(1-d) / 2, the converter operates in mode 1f; when d<1 and d*(1-d) / 2 <P n ≤0.25, the converter operates in mode 2f; when d>1 and 0≤P n ≤(d-1) / 2 / d 2 When d>1 and (d-1) / 2 / d 2 <P n ≤0.25, the converter operates in mode 4f; when d<1 and -d*(1-d) / 2≤P n <0, the converter operates in mode 1r; when d<1 and -0.25≤P n <-d*(1-d) / 2, the converter operates in mode 2r; when d>1 and -(d-1) / 2 / d 2 ≤P n <0, the converter operates in mode 3r; when d>1 and -0.25≤P n <(d-1) / 2 / d 2When , the converter works in mode 4r. For the above eight working modes, the expressions for calculating D1, D2, and D3 in the sub-mode are as follows:

[0046]

[0047]

[0048]

[0049]

[0050]

[0051]

[0052]

[0053]

[0054] Based on D1, D2, D3, V in With V o The value of the new control variable D4 introduced in the embodiment of the present invention is calculated.

[0055] For mode 1f and mode 3f, the initial value and the final value within the switching cycle are both 0, and no optimization is required, that is, D4 = 0.

[0056] For mode 2f and mode 4f, the power transmission direction is forward. The drive control waveforms without DC bias optimization and with DC bias optimization are as follows: Figure 3 As shown. Based on the volt-ampere characteristics of the inductor, the expression of the inductor current changing with time can be obtained as:

[0057]

[0058] Among them, t1, t2, t3, t4 can be expressed as:

[0059]

[0060] From the volt-second balance characteristics of the inductor, we can get:

[0061] i L4 =-i L0

[0062] Combining the above three equations, we can get the inductor current values ​​at t1, t2, t3, and t4 respectively:

[0063]

[0064] To eliminate the DC bias, a new control variable D4 is introduced to adjust the drive signal so that the initial and final values ​​of the inductor current in a single switching cycle are both 0. This yields:

[0065]

[0066] The calculation expression of D4 under mode 2f and mode 4f is:

[0067]

[0068] For mode 1r and mode 3r, their driving signals and current waveforms are symmetrical with those of mode 1f and mode 3f. The rising edge of the secondary bridge driving signal leads the primary bridge driving signal by a phase shift angle of D3, so D4 = D3 in this mode.

[0069] For mode 2r and mode 4r, the power transmission direction is opposite. The drive control waveforms without DC bias optimization and with DC bias optimization are as follows: Figure 4 As shown. Based on the volt-second balance characteristic of the inductor, the current at each segment point is:

[0070]

[0071] To eliminate the DC bias, a new control variable D4 is introduced to adjust the drive signal so that the initial and final values ​​of the inductor current in a single switching cycle are both 0. This yields:

[0072]

[0073] The calculation expression of D4 under mode 2r and mode 4r is:

[0074]

[0075] Based on the calculated D4, the driving waveform generated by D1, D2, and D3 is modified. The optimized driving waveform is as follows: Figure 3 (b) and Figure 4 This control method can make the initial and final values ​​of the inductor current in a single switching cycle both zero, achieving the decoupling of the inductor currents in adjacent switching cycles, thereby eliminating the generation of inductor bias current.

[0076] The core idea of ​​the control method proposed in the embodiment of the present invention is to introduce a new control variable D4 to adjust the drive signal so that the initial and final values ​​of the inductor current in a single switching cycle are both 0. This method is applicable to various phase shift control strategies such as single phase shift, extended phase shift, double phase shift, and triple phase shift.

[0077] When generating a PWM signal based on the values ​​of D1, D2, D3, and D4, the embodiment of the present invention does not require changing the duty cycle of the primary and secondary full-bridge voltage waveforms or the phase shift angle of the rising edge of the primary and secondary full-bridge voltage waveforms. Instead, the drive signal is simply shifted based on the calculated forward phase shift angle D4. When this control method is used to drive a DAB converter, the auxiliary inductor current maintains a completely symmetrical current within a single switching cycle, eliminating the power shortfall during the first half of the transient switching cycle.

[0078] Although the preferred embodiments of the present application have been described, those skilled in the art may make additional changes and modifications to these embodiments once they have learned the basic creative concept. Therefore, the appended claims are intended to be interpreted as including the preferred embodiments and all changes and modifications that fall within the scope of the present application.

[0079] Obviously, those skilled in the art may make various changes and modifications to this application without departing from the spirit and scope of this application. Thus, if these modifications and variations of this application fall within the scope of the claims of this application and their equivalents, this application is intended to include these modifications and variations.

Claims

1. A phase shift control method for a DAB converter without DC bias, characterized in that: The following steps are involved: S1, sampling DAB converter input voltage V in With the output voltage V o ; S2, output reference voltage V oref With the output voltage V o The voltage error is obtained by making a difference, and the voltage error is used as the input of the voltage controller to obtain the output power reference value P oref ; S3. Define the current reference power P b =N*V in *V o / (2*f s *L r ), where N is the transformer ratio, f s is the switching frequency, L r The auxiliary inductance value is defined as P per unit power. n =P oref / P b , define the voltage conversion ratio d=N*V o / V in ; Define D1 and D2 as the duty cycle of the primary and secondary full-bridge voltage waveforms respectively, D3 as the phase shift angle of the rising edge of the primary and secondary full-bridge voltage waveforms, and D4 as the phase shift angle of the forward shift of the drive waveform; S4, based on per unit power P n The value of the voltage conversion ratio d determines the working mode of the DAB converter, and calculates the values ​​of D1, D2, and D3 corresponding to the current mode. in 、V o And the phase shift angle D4 of the driving waveform is obtained by calculating the D1, D2, and D3 modal parts; S5. Generate a PWM signal according to the obtained values ​​of D1, D2, D3, and D4, and use the PWM signal as a driving signal for the DAB converter.

2. The phase shift control method for a DAB converter without DC bias according to claim 1, wherein: The modes of operation of the DAB converter include: when d < 1 and 0 ≤ P n ≤d*(1-d) / 2, the DAB converter operates in mode 1f; when d<1 and d*(1-d) / 2 <P n ≤0.25, the DAB converter operates in mode 2f; when d>1 and 0≤P n ≤(d-1) / 2 / d 2 When d>1 and (d-1) / 2 / d 2 <P n ≤0.25, the DAB converter operates in mode 4f; when d<1 and -d*(1-d) / 2≤P n <0, the DAB converter operates in mode 1r; when d<1 and -0.25≤P n <-d*(1-d) / 2, the DAB converter operates in mode 2r; when d>1 and -(d-1) / 2 / d 2 ≤P n <0, the DAB converter operates in mode 3r; when d>1 and -0.25≤P n <(d-1) / 2 / d 2 When , the DAB converter works in mode 4r; among them, mode 1f, mode 2f, mode 3f, and mode 4f work in the forward power transmission mode; mode 1r, mode 2r, mode 3r, and mode 4r work in the reverse power transmission mode.

3. The phase shift control method for a DAB converter without DC bias according to claim 2, wherein: The expressions of D1, D2, and D3 are as follows: Mode 1f: Mode 2f: Mode 3f: Mode 4f: Mode 1r: Mode 2r: Mode 3r: Mode 4r:

4. The phase shift control method for a DAB converter without DC bias according to claim 2 or 3, characterized in that: The calculation formula of the phase shift angle D4 of the driving waveform is:

5. The phase shift control method for a DAB converter without DC bias according to claim 4, characterized in that: The driving waveform generated by the duty cycles D1, D2, and D3 is corrected according to the phase shift angle D4 of the driving waveform.

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