Wide-range differential pair frequency stabilization micro-vibration circuit device

Abstract

The invention discloses a wide-range differential pair frequency stabilization micro-vibration circuit device, relates to the technical field of radio and television electronic information and integrated circuits, and solves the problems that an existing oscillation circuit cannot meet existing applications in the aspects of frequency accuracy, amplitude stability, frequency adjustment convenience, parameter calculation difficulty, manufacturing simplicity and cost, integration feasibility and the like. The circuit device includes a differential pair oscillation circuit, a regulator circuit, and a current source. Under the condition of high power supply voltage, compared with a working environment with low power supply voltage, the working current is not increased much; not only can output sine wave signal frequency and amplitude stable in a wide power supply voltage range be maintained, but also normal operation can be realized under extremely low power supply voltage. The circuit has the characteristic of stable amplitude, the output amplitude of the circuit is only related to two factors of PN junction forward voltage drop and equivalent load resistance, and the circuit is ensured to always keep the amplitude of output sinusoidal signals unchanged when the frequency is subjected to large-range adjustment change application.

Description

Technical Field

[0001] The present invention relates to the field of broadcasting and television electronic information and integrated circuit technology, and in particular to a wide-range differential pair frequency-stabilizing micro-vibration circuit device, which can be applied to a sine wave oscillation circuit with a wider voltage range and is easy to integrate. Background Art

[0002] An electronic oscillator is a device that cleverly converts direct current into alternating current with precise frequency, amplitude and phase. This feature makes it an indispensable component of modern electronic information systems. Whether it is to provide stable and reliable clock signals for processors, timers and signal processing circuits in cutting-edge intelligent systems, or to generate carriers and modulated waves at the transmitting end of radio, television and communication systems, or even to generate local oscillator signals for down-conversion in receivers and generate carrier signals for synchronous demodulation, electronic oscillators are indispensable components. In addition, in power converters and mobile devices such as mobile phones, electronic oscillators must be used to generate high-frequency alternating current in order to produce high-efficiency switching power supplies and audio power amplifiers. In today's era of rapid development of intelligent devices, the close combination of electronic oscillators and signal processing technology has built the cornerstone of the functions of electronic information systems from demodulation, decoding, filtering to modulation. A series of complex and precise operations are derived from the AC oscillation signals generated by electronic oscillators. Therefore, in the field of electronic information technology, as a core component embedded in various types of mixed digital and analog integrated circuits, the stability of the output signal of the electronic oscillator is the primary criterion for measuring its performance, and the quality of its performance directly affects the performance indicators of the entire electronic system. Common circuit structures include quartz crystal oscillators, LC oscillators, ring oscillators, and RC oscillators. Quartz crystal oscillators are known for their highly stable frequencies, but their frequency adjustment range is very limited and difficult to change continuously and significantly. In contrast, although ring oscillators and RC oscillators are easy to integrate, they are significantly inferior to LC oscillators in terms of frequency stability, waveform fidelity, high frequency, and anti-interference. Therefore, in application scenarios such as broadcast and television channel switching reception, which require large-scale precise changes and stable frequencies, LC oscillators have become the preferred solution due to their excellent frequency stability performance.

[0003] Power supply voltage, temperature, and component parameter drift are the main factors that affect the stability of electronic oscillators. With the increasing popularity of mobile smart devices, these devices not only require electronic systems to have low voltage operating capabilities and low energy consumption, but also hope to be able to operate stably within a wide power supply voltage range. For example, a mobile phone charger needs to change the output voltage over a wide range according to the fast charging communication protocol. Therefore, if the oscillation circuit in the electronic system can operate within a wide power supply voltage range and can always maintain a high degree of stability in signal frequency and amplitude, it can better meet the dual needs of product performance and user experience.

[0004] There are many basic circuit structures to choose from when designing the LC oscillator circuit, including transformer feedback oscillation circuit, capacitor three-point oscillator, inductor three-point oscillation circuit and differential pair oscillation circuit. The optimal perspective of the circuit solution should be comprehensively considered from multiple dimensions such as accurate frequency, stable amplitude, convenient frequency adjustment, easy parameter calculation, simple and low cost production, and integration feasibility.

[0005] The convenience of frequency adjustment is of great significance for achieving circuit flexibility and adaptability. In order to flexibly adjust the oscillation frequency, a varactor diode is often used as the key capacitor element in the LC frequency selection circuit. If the varactor diode can be directly connected to the positive pole of the power supply or to the ground, it will be particularly easy to design a frequency conversion control circuit. For example, when building a phase-locked loop circuit, the control voltage can be directly applied to the varactor diode without any conversion, thereby achieving precise control of the oscillation frequency and phase.

[0006] The ease of production is a key indicator to measure the feasibility and economy of a design. If the design of tapped inductor components is avoided in circuit design, it can not only simplify the circuit structure and reduce production costs, but also significantly optimize the analysis and calculation process of the circuit, so that the circuit parameters can be analyzed and calculated intuitively with a simple model, further improving the design efficiency and accuracy.

[0007] Transistors are the most basic and easiest to integrate components in integrated circuits, especially bipolar transistors and field-effect transistors, which are the core of modern integrated circuits. Resistors and capacitors are also relatively easy to integrate, but their accuracy and temperature stability are not very good. For resistors, although they can also be integrated, considering the accuracy and temperature stability issues, the number of resistors is usually minimized in circuit design. As for capacitors, since they occupy a large area of ​​the chip, integrated capacitors are usually avoided in integrated circuit design, and external capacitors are used as much as possible to meet circuit requirements.

[0008] Taking the above factors into consideration, the present invention improves on the basis of the traditional differential pair LC oscillator circuit. The newly designed circuit is not only easy to integrate, but also can work stably under extremely low power supply voltage. It is worth mentioning that it can always maintain the stability of the frequency and amplitude of the output sine wave signal within a wide power supply voltage range. Even under higher power supply voltage conditions, the increase in its operating current is very small compared to the low power supply voltage working environment, thereby meeting the use requirements of various application scenarios. Summary of the invention

[0009] The present invention provides a wide-range differential pair frequency stabilization micro-oscillation circuit device to solve the problems that the existing oscillation circuit cannot meet the existing applications in terms of frequency accuracy, amplitude stability, frequency adjustment convenience, parameter calculation difficulty, manufacturing simplicity and cost, and integration feasibility.

[0010] A wide-range differential pair frequency-stabilized micro-vibration circuit device, the circuit device comprising a differential pair oscillation circuit, a voltage regulator circuit and a current source;

[0011] The differential pair oscillator circuit includes a transistor Q1, a transistor Q2, a parallel resonant circuit, and a load resistor RL.

[0012] The voltage stabilizer circuit includes transistor Q3, transistor Q4, transistor Q5, transistor Q6, transistor Q8, diode D1, diode D2, resistor R1, resistor R3 and resistor R4;

[0013] The base of the transistor Q1 is respectively connected to the collector of the transistor Q2, one end of the parallel resonant circuit, one end of the load resistor RL and one end of the output terminal J1;

[0014] The emitter of the transistor Q1 is connected to the emitter of the transistor Q2 and the current source respectively;

[0015] The collector of the transistor Q1 is respectively connected to the base of the transistor Q2, the emitter of the transistor Q3, the collector of the transistor Q5, the other end of the parallel resonant circuit, one end of the resistor R1, the other end of the load resistor RL, the positive end of the diode D1, the other end of the output terminal J1 and the power supply VCC;

[0016] The collector of the transistor Q3 is connected to the base of the transistor Q5 and the current source respectively;

[0017] The base of the transistor Q3 is connected to the collector of the transistor Q4;

[0018] The base of the transistor Q4 is connected to the collector of the transistor Q6 and the collector of the transistor Q8 respectively;

[0019] The emitter of the transistor Q4 is connected to one end of the resistor R4;

[0020] The emitter of the transistor Q5 is connected to the base of the transistor Q6;

[0021] The emitter of the transistor Q6 is respectively connected to the current source, one end of the resistor R3, the other end of the resistor R4 and the ground terminal GND;

[0022] The emitter of the transistor Q8 is connected to the other end of the resistor R1;

[0023] The base of the transistor Q8 is connected to the other end of the resistor R3 and the negative end of the diode D2 respectively;

[0024] The cathode terminal of the diode D1 is connected to the anode terminal of the diode D2.

[0025] Furthermore, the parallel resonant circuit is composed of an inductor L1 and a capacitor C1 connected in parallel.

[0026] Further, the current source is composed of a transistor Q7 and a resistor R2;

[0027] The base of the transistor Q7 is connected to the collector of the transistor Q3 and the base of the transistor Q5 respectively;

[0028] The collector of transistor Q7 is connected to the emitter of transistor Q1 and the emitter of transistor Q2 respectively;

[0029] The emitter of the transistor Q7 is connected to one end of the resistor R2 , and the other end of the resistor R2 is respectively connected to one end of the resistor R3 , the emitter of the transistor Q6 , the other end of the resistor R4 and the ground terminal GND.

[0030] Furthermore, assuming that the operating current of the circuit device is IX, the resistance value calculation formula of the resistor R2 is:

[0031] IX=0.7 / r2

[0032] Where r2 is the resistance value of resistor R2.

[0033] Furthermore, the amplification factor of the transistor Q3 is set to B3, the amplification factor of the transistor Q4 is set to B4, and the amplification factor of the transistor Q7 is set to B7, then the maximum resistance value r1 of the resistor R1 is required to be MAX for:

[0034] r1 MAX =B3*B4*B7*r2

[0035] Set the amplification factor of transistor Q8 to B8, then the maximum resistance value of resistor R3 is r3 MAX for:

[0036] r3 MAX =B8*r1

[0037] Where r1 is the resistance value of resistor R1.

[0038] Furthermore, the maximum resistance value r4 of the resistor R4 is set MAX for:

[0039] r4 MAX =B3*B7*r2.

[0040] Furthermore, the resistance value of the resistor R3 is set to be 10 times greater than the resistance value of the resistor R1 , and the resistance value of the resistor R4 is one tenth of the resistance value of the resistor R1 .

[0041] Furthermore, the transistor Q3 serves as a power adjustment tube of the voltage stabilizing circuit. When the power supply voltage exceeds 1.4V, it works in the amplification region to provide a stable power supply voltage for the current source. When the power supply voltage drops to a low voltage below 1.4V, the transistor Q3 works in the saturation region.

[0042] Beneficial effects of the present invention:

[0043] The circuit device described in the present invention has frequency stability in a wide power supply range. For example, under the power supply condition of 1.4V to 32V, the minimum frequency of the output signal of the experimental circuit is 36.40KHz and the maximum frequency is 36.21KHz. Although the voltage changes by more than 20 times, the relative error is only 0.52%.

[0044] The circuit device described in the present invention has amplitude stability in a wide power supply range. For example, under the power supply condition of 1.4V to 32V, the minimum amplitude of the output signal of the experimental circuit is 0.65V and the maximum amplitude is 0.69V. Although the voltage changes by more than 20 times, the relative error is only 6.1%.

[0045] The circuit device described in the present invention has a working current that does not increase much under higher power supply voltage conditions compared to a low power supply voltage working environment; it can not only keep the frequency and amplitude of the output sinusoidal wave signal stable within a wider power supply voltage range, but also can work normally under extremely low power supply voltage.

[0046] The circuit device of the present invention has the characteristic of stable amplitude, that is, the output amplitude is only related to two factors: the forward voltage drop of the PN junction and the equivalent load resistance. By utilizing this characteristic, it can be ensured that the amplitude of the output sinusoidal signal remains unchanged when the frequency of the circuit is adjusted and changed over a wide range. In contrast, in order to change the frequency, the traditional LC oscillator will change the inductance or capacitance, which will affect the amplitude of the output signal due to the change of the feedback amount of the circuit. BRIEF DESCRIPTION OF THE DRAWINGS

[0047] Figure 1 It is a schematic diagram of the wide-range differential pair frequency-stabilizing micro-vibration circuit device of the present invention;

[0048] Figure 2 The effect diagram of the 0.78V power supply test;

[0049] Figure 3 This is the effect diagram of 0.79V power supply test;

[0050] Figure 4 This is the effect diagram of 0.80V power supply test;

[0051] Figure 5 This is the effect diagram of 0.90V power supply test;

[0052] Figure 6 This is the effect diagram of 1.20V power supply test;

[0053] Figure 7 This is the effect diagram of 1.40V power supply test;

[0054] Figure 8 This is the test result under the condition of maximum 32V power supply;

[0055] Fig. 9 is a schematic diagram of a curve showing the relationship between the working current and the supply voltage;

[0056] Fig.10 It is a schematic diagram of the relationship between the output frequency and the supply voltage;

[0057] Fig.11 This is a schematic diagram of the relationship between the no-load output amplitude and the supply voltage. DETAILED DESCRIPTION

[0058] Specific implementation method 1. Combination Figure 1 The present embodiment is described, and a wide-range differential pair frequency-stabilized micro-vibration circuit device includes a power supply VCC, a ground terminal GND and an output terminal J1, a differential pair oscillation circuit composed of a transistor Q1, a transistor Q2, an inductor L1, a capacitor C1 and a load resistor RL; and a voltage stabilization circuit composed of a transistor Q3, a transistor Q4, a transistor Q5, a transistor Q6, a transistor Q8, a diode D1, a diode D2, a resistor R1, a resistor R3 and a resistor R4; and a current source composed of a resistor R2 and a transistor Q7;

[0059] The base of the transistor Q1 is electrically connected to the collector of the transistor Q2, the lower end of the inductor L1, the lower end of the capacitor C1, the lower end of the load resistor RL, and the lower end of the output terminal J1 respectively;

[0060] The emitter of transistor Q1 is electrically connected to the emitter of transistor Q2 and the collector of transistor Q7 respectively;

[0061] The collector of transistor Q3 is electrically connected to the base of transistor Q7 and the base of transistor Q5 respectively;

[0062] The base of transistor Q3 is electrically connected to the collector of transistor Q4;

[0063] The base of transistor Q4 is electrically connected to the collector of transistor Q6 and the collector of transistor Q8 respectively;

[0064] The emitter of transistor Q8 is electrically connected to the lower end of resistor R1;

[0065] The emitter of transistor Q5 is electrically connected to the base of transistor Q6;

[0066] The base of transistor Q8 is electrically connected to the upper end of resistor R3 and the negative end of diode D2 respectively;

[0067] The negative terminal of the diode D1 is electrically connected to the positive terminal of the diode D2;

[0068] The collector of transistor Q1 is electrically connected to the base of transistor Q2, the emitter of transistor Q3, the collector of transistor Q5, the upper end of inductor L1, the upper end of capacitor C1, the upper end of resistor R1, the upper end of load resistor RL, the positive end of diode D1, the upper end of output terminal J1, and power supply VCC respectively;

[0069] The emitter of transistor Q4 is electrically connected to the left end of resistor R4;

[0070] The emitter of the transistor Q6 is electrically connected to the lower end of the resistor R2 , the lower end of the resistor R3 , the right end of the resistor R4 , and the ground terminal GND respectively.

[0071] In application scenarios that require fast, accurate and wide-range adjustment of the oscillator output frequency, varactor diodes are often used as core capacitor components in LC frequency selection circuits. At this time, by adjusting the voltage applied to both ends of the varactor diode, the frequency of the output oscillation signal can be easily adjusted, that is, the voltage-controlled oscillator function is realized. This technology is widely used in phase-locked loop circuits of various radio, television and communication systems, making the switching of communication channels or the adjustment of radio and television channels easy and convenient, and is used to meet the needs of users to change channels or tune in.

[0072] In this embodiment, the capacitor C1 is directly connected to the power supply end, that is, the capacitor C1 realizes AC grounding; when the circuit device provided by the present invention is applied to a voltage-controlled oscillator, a varactor diode connected in series with a DC blocking capacitor can be used to form an equivalent capacitor to replace the capacitor C1; when the varactor diode is connected to the power supply VCC, the voltage applied across the varactor diode is equal to the voltage difference between the output end of the external control circuit and the power supply VCC; the characteristic of capacitor AC grounding can be further upgraded and deformed, that is, the end of the varactor diode originally connected to the power supply VCC is changed to ground, and the voltage applied across the varactor diode is directly equal to the voltage at the output end of the external control circuit; therefore, this capacitor AC grounding characteristic can simplify the relevant design of the control circuit required for this voltage-controlled oscillator.

[0073] In this embodiment, the inductor L1 does not need to be designed with a tap, and the oscillation frequency f of the circuit device is only determined by the inductance L of the inductor L1 and the capacitance C of the capacitor C1, and can be directly calculated using the formula;

[0074]

[0075] In this embodiment, ensuring that the oscillation circuit obtains a continuous and constant current supply is one of the core elements in the entire circuit design. For this purpose, it is crucial to design a high-performance voltage-stabilizing circuit that can provide a stable bias voltage for the current source. It is the basis for ensuring that the entire circuit device can operate stably within a wide range of power supply voltages. In this embodiment, a stable and reliable circuit structure (voltage-stabilizing circuit) is provided to ensure that the circuit can perform at its best under various voltage conditions, which incorporates two core characteristics: one is good adaptability in low-voltage environments, and the other is the ability to effectively reduce energy consumption under high-voltage conditions.

[0076] 1. Analyze the static operating point of the circuit device described in this embodiment:

[0077] 1. Designed in a wide operating voltage range:

[0078] Common analog oscillator circuits usually approximately follow Ohm's law, and their operating current is directly proportional to the supply voltage, that is, the power consumption or power consumption is proportional to the square of the supply voltage. This means that when the power supply voltage doubles, the operating current will also double accordingly, causing the power consumption to surge to four times the original. Therefore, in order to ensure normal operation under low voltage environments, traditional circuit designs often lead to a significant increase in operating current under high voltage power supply conditions. This phenomenon not only greatly increases energy consumption, but also requires that sufficient power dissipation capacity must be reserved when designing each component in the integrated circuit due to the sharp increase in power heat dissipation requirements to cope with potential overheating and damage risks.

[0079] The circuit device described in this embodiment can increase the operating voltage from 1.4V to 32V, and the operating current will not even double even though the voltage is increased by more than 20 times; what is particularly striking is that the circuit device shows excellent adaptability and can maintain a stable operating state even under extremely low voltage power supply conditions as low as 0.8V; this excellent performance is largely due to the key role played by the specially designed voltage and current stabilization circuit.

[0080] In this embodiment, the operating current of the differential pair oscillator circuit determines the load capacity of the circuit device. In subsequent analysis, it can be seen that the load capacity of the circuit device is proportional to the operating current. Therefore, when designing the circuit, the operating current is first determined according to the required load capacity. If the designed operating current is IX, only the resistance value of the resistor R2 needs to be designed. The specific formula is:

[0081] IX=0.7 / r2

[0082] If the amplification factor of transistor Q3 is B3, the amplification factor of transistor Q4 is B4, and the amplification factor of transistor Q7 is B7, then the maximum resistance of resistor R1 must not exceed the value calculated by the following formula;

[0083] r1 MAX =B3*B4*B7*r2

[0084] If the gain of transistor Q8 is B8, the maximum value of R3 cannot exceed the value calculated by the following formula;

[0085] r3 MAX =B8*r1

[0086] In order to reduce energy consumption under high voltage working conditions, the resistance value of resistor R3 can be designed to be as large as possible.

[0087] In this embodiment, when the power supply voltage just reaches the conduction threshold after the two PN junctions are connected in series, that is, about 1.4 volts, since the transistor Q3 enters the saturation state at this time, and the transistor Q4 begins to show the conduction characteristics, this change causes the current in the collector of the transistor Q4 to rise significantly, thereby causing unnecessary power loss. In order to optimize the energy efficiency of the circuit in this sensitive critical state, a current limiting resistor R4 is specially designed to be connected in series with the emitter of the transistor Q4. The resistance value of the resistor R4 plays a key role in the transition of the working state of the current limiting circuit from the saturated power supply as much as possible to the constant current control power supply state during the transition period when the power supply voltage transitions from the low voltage of 1.2 volts to the high voltage of 1.4 volts. If the resistance value of the resistor R4 is set to be relatively small, or even zero, its impact will only increase the energy consumption under the critical transition condition, but will not affect the output signal of the circuit. In other words, if the value of the resistor R4 is larger, it is beneficial to save power, but the upper limit of the resistance value of the resistor R4 should not exceed the value calculated by the following formula;

[0088] r4 MAX =B3*B7*r2

[0089] Under the ordinary silicon planar process conditions of integrated circuits, the NPN transistors produced are of a longitudinal structure, which can have a higher current magnification factor, and its value can exceed one hundred; while the PNP transistors Q3 and Q8 produced are of a transverse structure, and the current magnification factor is slightly lower, and its value is generally a dozen to several dozen; therefore, designing the resistor R1 as large as possible is conducive to reducing energy consumption, and designing the resistance value of resistor R3 to be 10 times greater than the resistance value of resistor R1, and the resistance value of resistor R4 is about one tenth of the resistance value of resistor R1, which is more reasonable. Because resistors exceeding 100 kilo-ohms are generally not integrated in the integrated circuit process, after considering various factors, the resistance value of resistor R3 is designed to be 100 kilo-ohms, the resistance value of R1 is 10 kilo-ohms, and the resistance value of R4 is 2 kilo-ohms. The design selection of these resistance values ​​does not require high precision, and the room for selection is very wide. If the resistance value is larger, it is conducive to reducing energy consumption in high voltage occasions, and if the resistance value is smaller, it is beneficial to resist external noise interference. Even if the specific resistance value differs by several times, it will not affect the normal operation of the circuit. Therefore, the design of this embodiment not only meets the circuit requirements, but also complies with the process limitations of integrated circuits.

[0090] In this embodiment, the transistor Q3 is used as a power supply adjustment tube of the voltage stabilization circuit. When the power supply voltage exceeds 1.4V, it works in the amplification region to provide a stable power supply voltage for the current source. When the power supply voltage drops below 1.4V, the transistor Q3 works in the saturation region to provide the current source with the highest possible power supply voltage to maintain the differential pair oscillation circuit to continue working. Next, the static operating points of the two situations of low voltage power supply below 1.4V and ordinary power supply above 1.4V are analyzed respectively.

[0091] Under low voltage power supply conditions, all node voltages in the voltage stabilizing circuit are lower than the power supply voltage. Because the power supply voltage is lower than the overall conduction voltage after the emitter of transistor Q5 and the emitter of transistor Q6 are connected in series, transistors Q5 and Q6 are both in the cut-off state. Because the power supply voltage is lower than the threshold conduction voltage after the diode D1 and the diode D2 are connected in series at this time, the current flowing through the resistor R3 can all flow through the emitter of the transistor Q8, and after being amplified, the current passing through the resistor R1 can all flow through the emitter of the transistor Q4, causing the transistor Q4 to be in the conduction state, and can drive the transistor Q3 to be saturated and turned on.

[0092] When the power supply voltage is higher than 1.4V, because the emitters of transistor Q5 and transistor Q6 can all be turned on, when the collector voltage of transistor Q3 increases with the power supply voltage, the current flowing through the base of transistor Q5 increases. This current is amplified by transistor Q5 and flows into the base of transistor Q6, and then amplified by transistor Q6 and outputted from the collector of transistor Q6 in an inverted manner. The effect is to pull down the base voltage of transistor Q4 while dividing most of the current at the circuit node where the base of transistor Q4 is located, so that the collector current of transistor Q4 is reduced, that is, the base current of transistor Q3 is reduced, and finally the collector voltage of transistor Q3 is reduced; it can be seen that this process is a voltage negative feedback, so that the output voltage of the voltage regulator circuit will always be equal to the sum of the forward voltage drops of the two emitters of transistor Q5 and transistor Q6 connected in series. Therefore, as long as the power supply voltage is higher than 1.4V, a stable double PN forward voltage drop can be maintained at the base of transistor Q7, so that transistor Q7 can always output a fixed current to the differential pair oscillation circuit.

[0093] 2. Analyze the static operating point of the differential pair oscillation circuit:

[0094] In the differential pair oscillation circuit with transistors Q1 and Q2 as the main components, because the DC resistance of inductor L1 is very small, it can be considered as a wire under DC conditions, so the base voltage and collector voltage of transistors Q1 and Q2 are the same, so that the working points of the two transistors are located at the critical intersection of the amplification region and the saturation region. This feature is different from the situation where the static working points of the two transistors are both in the amplification region in the ordinary differential pair amplifier circuit. In addition, in the ordinary differential pair oscillation circuit, in order to ensure that the upper half and lower half of the alternating current generated by the oscillation can be perfectly symmetrical, the two transistors on both sides and other connected components need to be designed in a balanced manner, that is, both sides need to be designed not only with the same DC working point, but also with the same DC load and the same AC load. More importantly, the performance of the transistors on both sides, especially the amplification factor, is also required to be as similar as possible. In the present embodiment, from the perspective of power supply, the current flowing through the base of transistor Q1 and the current flowing through the collector of transistor Q2 must pass through inductor L1, which causes an imbalance between transistor Q1 and transistor Q2 under AC working conditions; from the perspective of load conditions, only transistor Q2 has a load, while transistor Q1 does not; and at the same time, the performance of transistor Q1 and transistor Q2 is not required to be consistent.

[0095] In the DC static state, the sum of the current flowing through the emitter of transistor Q1 and the current flowing through the emitter of transistor Q2 is equal to the collector current of transistor Q7 in the current source. The current of the transistor with a higher amplification factor or a lower body resistance will definitely be higher than the current of the other transistor, but which current is slightly larger will not affect the normal operation of the circuit. In other words, the circuit device of this embodiment does not require the current balance of the two paths.

[0096] 2. Analysis of the oscillation starting process of the circuit device described in this embodiment:

[0097] 1. Critical saturation;

[0098] Because the operating current of ordinary low-power transistors is usually in the order of milliamperes, the voltage value at the critical intersection of the amplification region and the saturation region is equal to the forward voltage drop value of a PN junction. Therefore, for low-power transistors made of silicon materials, at the critical intersection of the amplification region and the saturation region, the voltage difference from the collector to the emitter of the transistor is usually about 0.7V; that is, if the voltage difference between the collector and the emitter of a transistor is about 0.7V, then the operating point of the transistor is located at the critical intersection of the amplification region and the saturation region. Once the operating point of the transistor approaches the saturation region, its amplification ability will weaken; when it fully enters the saturation region, the transistor no longer has the amplification ability.

[0099] 2. Initial power-on;

[0100] When the circuit is just powered on, although transistors Q1 and Q2 are both in the on state, the circuit does not start to oscillate at this time. Due to the presence of random noise in the circuit, this will cause the current flowing through Q1 and Q2 to fluctuate very slightly at a certain moment. Because the total current flowing through the two transistors is controlled by the current source and is always a fixed value, when the current of one transistor changes, the current of the other transistor will also change accordingly. If the current of one transistor increases, the current of the other transistor must decrease to keep the total current unchanged. The two transistors show a "one increases while the other decreases" relationship in current; because the amplification factors and other basic parameters of the two transistors are allowed to be different, and there must be performance differences in actual components, the transistor with a slightly larger amplification factor or a slightly larger conductivity will have a more obvious current change when subjected to a small current fluctuation, so it will be turned on smoothly first, while the other transistor may become poorly turned on or even cut off due to the decrease in current.

[0101] In this embodiment, the inductor L1 and the capacitor C1 form a parallel resonant circuit with a frequency selection function; among all the random noises of all frequencies, only the amplitude of the frequency signal with the same frequency as the parallel resonant frequency is the largest at both ends of the parallel resonant circuit, and the noise amplitudes of other frequencies are suppressed; therefore, at a certain moment after the circuit is powered on, when the voltage at the upper end of the inductor L1 is higher than the voltage at the lower end, this slight change immediately causes the emitter current of the transistor Q1 to decrease, and at the same time, the emitter current of the transistor Q2 increases accordingly, that is, the conduction of the transistor Q2 is smoother than that of the transistor Q1; because the transistor Q2 has the amplification capability at this time, its base current is significantly smaller than the collector current, and because the transistor emitter current is approximately equal to the collector current, the effect of the voltage at both ends of the inductor L1 being high at the top and low at the bottom is further enhanced, and the result of this positive feedback process is that the transistor Q2 can charge the parallel resonant circuit and increase the energy reserve in the resonant circuit.

[0102] After the parallel resonant circuit is charged for half a resonant cycle, the voltage across the inductor L1 changes to high at the bottom and low at the top. At this time, the emitter voltage of transistor Q1 is higher than the emitter voltage of transistor Q2. Therefore, according to the PN junction equation, the base current of transistor Q1 is higher than the base current of transistor Q2. After this current difference is amplified by transistor Q1 and transistor Q2, positive feedback acts on this differential pair circuit, resulting in the emitter current of transistor Q1 being greater than the emitter current of transistor Q2. Because the resonant circuit has just been charged in the previous cycle, the circuit state from this time to the next half resonant cycle is much smoother than the conduction of transistor Q1 than that of transistor Q2 compared with the situation in the previous half cycle just experienced. In addition, the resonant circuit only discharges but does not charge in this half resonant cycle, and the overall energy in the resonant circuit only decreases but does not increase. This is different from the characteristics of the ordinary differential pair tube oscillation circuit, which charges the resonant circuit regardless of the positive half cycle or the negative half cycle of the oscillation.

[0103] During a series of repeated cycles of charging energy storage and discharging energy release in the resonant circuit, if the total amount of accumulated energy storage continues to exceed the released energy, the voltage amplitude at both ends of the circuit will gradually increase; the gradual increase in this voltage amplitude, in turn, exacerbates the difference in conduction smoothness between transistors Q1 and Q2 in each cycle, forming a positive feedback mechanism, further promoting the charging energy storage injected into the resonant circuit in each oscillation cycle, thereby prompting the circuit to spontaneously enter an oscillation state, and applying the generated sinusoidal AC signal voltage to both ends of the load resistor RL, which can be output from the output terminal J1.

[0104] 3. Analysis of the amplitude control of the circuit device described in this embodiment:

[0105] When the circuit device described in this embodiment enters the oscillation state, as the amplitude of the oscillation voltage at both ends of the resonant circuit gradually increases, the working states of the transistors Q1 and Q2 in the circuit device will show significant dynamic changes, which are specifically reflected in the following two key features: first, the transistors Q1 and Q2 will alternately enter the saturation conduction and cut-off states. This means that at a certain moment, when one of the transistors is in the saturation conduction state, the other transistor must be in the cut-off state; secondly, the working points of the transistors Q1 and Q2 will be located at the critical intersection of the amplification region and the saturation region only in the short moment when the instantaneous voltage of the resonant circuit oscillation is very close to zero; in this short moment, both transistors have a certain current amplification capability; however, when the instantaneous voltage of the resonant circuit oscillation deviates from zero, one of the transistors will quickly enter the saturation region, while the other transistor will quickly enter the cut-off region, and at this time they both lose the function of current amplification.

[0106] In summary, after the circuit device enters the oscillation state, the working states of transistors Q1 and Q2 will be dynamically adjusted with the change of the oscillation voltage of the resonant circuit, showing the characteristics of alternate saturation conduction and cutoff and having amplification capability only when the instantaneous voltage is close to zero.

[0107] When the transistor Q2 is saturated and turned on to charge the resonant circuit, because the transistor Q7 in the current source circuit has a current limiting function, it is equivalent to using a constant current source to charge the LC parallel resonant circuit. In this case, the voltage across the LC parallel resonant circuit will change according to the sinusoidal law; that is, during the saturation conduction of the transistor Q2, the lower end of the output terminal J1 connected to the inductor in the resonant circuit generates a lower half cycle of a sinusoidal voltage. When the transistor Q2 is turned off, the resonant circuit discharges according to the zero input response law of the LC parallel resonant circuit, so that the lower end of the output terminal J1 generates a positive half cycle of a sinusoidal voltage.

[0108] In a conventional circuit, when two transistors work in a saturated conduction and cutoff mode alternately, this rapid switching action causes the circuit to output a square wave signal; in the circuit device described in this embodiment, although the transistors Q1 and Q2 also follow the law of saturated conduction and cutoff alternately, a key difference is that there is a current source in the circuit specifically used to charge the LC parallel resonant circuit; because according to circuit theory, when the LC parallel resonant circuit is excited by a constant current, the voltage across the circuit changes according to the sinusoidal law, so in this embodiment, the output waveform of the circuit is not a square wave but a sine wave, which is different from the output characteristics of the conventional circuit.

[0109] Any circuit input end that receives a signal has an electronic parameter called input impedance, which can be regarded as an equivalent load resistance RL in the circuit device of this embodiment, that is, the external circuit is connected in parallel with the load resistance to form a new equivalent load resistance RL for calculation. When this embodiment transmits a sinusoidal AC signal to the external circuit through the output terminal J1, the input impedance of the external circuit will affect the voltage amplitude of the output signal of this circuit device. In other words, the resistance value of the load resistor RL can affect the voltage amplitude of the output sinusoidal AC signal. Therefore, in practical applications, there are two different application scenarios for analysis and calculation. One is that when the load resistance is small and the output signal amplitude is required to be large, it is necessary to ensure that the circuit has sufficient load driving capability; the other is when the load resistance is large and the expected output signal amplitude is small, the method of adjusting the resistance value of the load resistor RL can be used to reduce the output signal amplitude of the circuit. Specifically, the influence of the load resistance RL on the output signal amplitude can be divided into two situations. One is that when the product of the constant current value of the current source and the resistance value of the load resistance RL is greater than the forward voltage drop of the PN junction, it can be considered that the resistance value of RL is in the "large" category, and the circuit has the effect of stable output. Otherwise, it is considered that the resistance value of RL is in the "small" category, which will reduce the output signal amplitude of the circuit.

[0110] When the load resistor RL is relatively large, even large enough to be in an open circuit state, the voltage amplitude at both ends of the LC parallel resonant circuit can reach a maximum value. At this time, since the collector and base of transistors Q1 and Q2 each contain a PN junction, when these two transistors are connected in parallel with the LC parallel resonant circuit in a manner in which the PN junctions between their collectors and bases are reversely connected to each other, a limiting circuit is formed, which limits the maximum amplitude of the voltage at both ends of the LC parallel resonant circuit to not exceed the forward conduction voltage of a PN junction, which can be specifically calculated using the PN junction equation; for a PN junction of ordinary silicon material, at the milliampere level commonly used in electronic signals, this voltage is approximately 0.7V. In other words, in this embodiment, when the load resistor RL is relatively large, the output signal voltage amplitude has a self-stabilizing characteristic and is always approximately stable at 0.7V.

[0111] When the load resistance RL is relatively small, the amplitude of the output oscillation signal will no longer remain constant, but will decrease as the load resistance RL decreases, which is specifically equal to the current value of the current source multiplied by the resistance value of the load resistance RL divided by 2; that is, there is a linear relationship between the output amplitude and the resistance value of the load resistance RL, and in this case, only the amplitude of the output oscillation signal will change significantly, while the frequency and waveform shape of the signal will not change significantly; using this feature, the load resistance RL can be reduced to generate a smaller amplitude signal, or the linear relationship between amplitude and load can be used for small signal amplitude modulation applications.

[0112] In practical application circuits, when the load of the oscillator is a fixed circuit, the resistance of its equivalent load resistance RL is usually fixed by the circuit parameters. When the equivalent resistance is relatively small, because the amplitude of the output oscillation signal is proportional to the current value of the current source, when performing analog amplitude modulation, the current source can be controlled to change the current. In other words, the resistance of resistor R2 can be dynamically changed, and the modulation signal can be superimposed on resistor R2. The modulation signal can also be superimposed on the emitter of transistor Q7. These flexible methods can realize analog amplitude modulation function and have good modulation linearity.

[0113] In summary, the circuit device described in this embodiment has the characteristic of stable amplitude, that is, the output amplitude is only related to two factors: the forward voltage drop of the PN junction and the equivalent load resistance. By utilizing this characteristic, it can be ensured that the amplitude of the output sinusoidal signal is always kept unchanged when the frequency of the circuit is adjusted and changed over a wide range. In contrast, in order to change the frequency, the traditional LC oscillator changes the inductance or capacitance, which will affect the amplitude of the output signal due to the change in the feedback amount of the circuit. In this embodiment, whether the inductor L1 or the capacitor C1 is short-circuited or open-circuited, it only causes the output signal of the circuit to be abnormal, but will not damage any other components in the circuit. In other words, the integrated circuit designed using this embodiment can ensure the safety of the chip itself when an external component fails unexpectedly.

[0114] Specific implementation method 2: Combination Figures 2 to 11 This embodiment is described as a specific example of the wide-range differential pair frequency stabilization micro-vibration circuit device described in the first embodiment.

[0115] The experiment was conducted by assembling the circuit with discrete components, testing the minimum operating voltage for the circuit to oscillate under open-load conditions, and measuring the total current, oscillation frequency, and signal amplitude under various supply voltages. An oscilloscope was used to observe whether the oscillation signal could always remain a sine wave and measure the signal amplitude. A frequency meter was used to measure the output frequency (although an oscilloscope can measure frequency, its accuracy is low), and a spectrum analyzer was used to observe whether there was parasitic oscillation.

[0116] Combination Figures 2 to 8As shown, in the 0.78V power supply test, the circuit can be seen from the oscilloscope to oscillate; in the 0.79V power supply test, the signal amplitude is slightly increased from the oscilloscope; in the 0.80V power supply test, the signal amplitude is slightly increased from the oscilloscope, but still small; in the 0.90V power supply test, the signal amplitude displayed on the oscilloscope increases significantly; in the 1.20V power supply test, the signal amplitude is already significantly larger; when the power supply is 1.40V, the amplitude is already very large, and it can be considered that if the voltage continues to rise, it will enter the stable output state; the highest test under the 32V power supply condition, compared with the 1.4V power supply condition, the signal frequency is slightly reduced, and the amplitude is almost unchanged. Table 1 is a list of data for experimental testing of the circuit device.

[0117] Table 1

[0118] Serial number Supply voltage (V) Supply current (mA) Output frequency(KHz) Output amplitude (V) 1 0.78 0.338 37.25 0.08 2 0.79 0.370 37.21 0.10 3 0.80 0.403 37.19 0.11 4 0.90 0.726 36.98 0.20 5 1.00 1.051 36.82 0.31 6 1.10 1.378 36.68 0.40 7 1.20 1.691 36.53 0.50 8 1.30 1.767 36.33 0.63 9 1.40 1.835 36.29 0.65 10 1.50 1.955 36.40 0.66 11 1.60 2.074 36.28 0.67 12 1.70 2.168 36.26 0.68 13 1.80 2.220 36.25 0.68 14 1.90 2.241 36.25 0.69 15 2.00 2.236 36.24 0.69 16 3.00 2.371 36.23 0.69 17 4.00 2.415 36.23 0.69 18 5.00 2.460 36.22 0.69 19 6.00 2.481 36.22 0.69 20 7.00 2.502 36.22 0.69 21 8.00 2.508 36.22 0.69 22 9.00 2.511 36.22 0.69 23 10.00 2.511 36.22 0.69 24 20.00 2.593 36.21 0.69 25 30.00 2.701 36.21 0.69 26 32.00 2.723 36.21 0.69

[0119] like Figures 9 to 11 As shown, the relationship between current consumption, frequency and no-load amplitude and supply voltage is plotted using experimental data.

[0120] A signal generator is used to output a sinusoidal AC signal with a frequency of 1KHz and superimposed on both ends of resistor R2. At this time, observing the display on the spectrum analyzer, it can be clearly found that a new signal with an intensity lower than the original oscillation signal appears on both sides of the originally stable output oscillation signal, and the frequency interval between the two new signals and the original oscillation signal is exactly 1KHz; this phenomenon is a typical feature of the amplitude modulation signal spectrum diagram, in which the original oscillation signal is called the carrier signal, the new signal with a higher frequency than the carrier signal is called the upper sideband signal, and the signal with a lower carrier signal is called the lower sideband signal. The signal output by the signal generator is called the modulated signal.

[0121] By using an oscilloscope to observe the waveform of the signal after modulation, we can see that the amplitude of the carrier signal has changed periodically, from a constant amplitude wave without modulation to an amplitude modulated wave; the envelope of the amplitude modulated wave is also a sine wave, and its frequency is the same as the modulation signal; if the frequency of the modulation signal is changed, the frequencies of the two sideband signals will also change accordingly; however, no matter how the modulation signal changes, the frequency difference between the two sideband signals and the carrier signal will always be equal to the frequency of the modulation signal; and in the entire experimental observation process, no other parasitic frequencies are shown on the spectrum diagram, indicating that this carrier oscillation circuit has good modulation performance.

[0122] The above experimental data analysis shows that the circuit device described in this embodiment has the following advantages:

[0123] 1. It has ultra-low voltage working capability and can start oscillating even at the lowest power supply of 0.78V. It can output sinusoidal AC signals and ensure normal operation at an ultra-low power supply voltage of 0.80V.

[0124] 2. It has high voltage working capability and can output sinusoidal AC power at a maximum power supply of 32V.

[0125] 3. It has frequency stability in a wide power supply range. When the power supply is from 1.4V to 32V, the minimum frequency of the output signal is 36.40KHz and the maximum frequency is 36.21KHz. Although the voltage changes more than 20 times, the relative error is only 0.52%.

[0126] 4. It has amplitude stability in a wide power supply range. When the power supply is from 1.4V to 32V, the minimum amplitude of the output signal is 0.65V and the maximum amplitude is 0.69V. Although the voltage changes more than 20 times, the relative error is only 6.1%.

[0127] 5. The circuit device has the ability to work at ultra-low voltage and is energy-saving at high operating voltage. Under the power supply condition of 1.4V to 32V, the minimum operating current is 1.835mA and the maximum operating current is 2.723mA, which is a 48% increase in current. That is to say, compared with the low voltage condition, although the voltage increases by more than 20 times, the current only increases by half.

[0128] 6. The power supply voltage is between 0.78 and 1.4V, which is a low voltage power supply situation. The experimental data is consistent with the theoretical value.

[0129] 7. Even at an extremely low power supply voltage of 0.78 volts, the output frequency changes from 37.25 kHz to 36.29 kHz, a frequency change of about 2.5%, compared to the 1.4 volt power supply case.

[0130] 8. Whether it is powered by an ultra-low voltage of 0.78V or a high voltage of 32V, within a wide range of power supply voltage, the quality of the sinusoidal signal output by the experimental circuit is relatively good. The waveform image seen on the oscilloscope not only maintains a sinusoidal curve at all times, but also has stable amplitude and frequency. At the same time, in this process, no other parasitic frequency signals are seen on the spectrum analyzer.

[0131] 9. In the simulated amplitude modulation experiment, the envelope of the amplitude modulated wave can be seen through the oscilloscope and is also a sine wave; through the spectrum analyzer, it can be seen that the center frequency of the amplitude modulated wave has not changed, and compared with the unmodulated state, there are only two additional upper and lower sideband components, and no other parasitic sideband components.

[0132] The technical features of the above-described embodiments may be arbitrarily combined. To make the description concise, not all possible combinations of the technical features in the above-described embodiments are described. However, as long as there is no contradiction in the combination of these technical features, they should be considered to be within the scope of this specification.

[0133] The above-mentioned embodiments only express several implementation methods of the present invention, and the descriptions thereof are relatively specific and detailed, but they cannot be understood as limiting the scope of the invention patent. It should be pointed out that, for ordinary technicians in this field, several variations and improvements can be made without departing from the concept of the present invention, and these all belong to the protection scope of the present invention. Therefore, the protection scope of the patent of the present invention shall be subject to the attached claims.

Claims

1. A wide-range differential pair frequency-stabilized micro-vibration circuit device, the circuit device comprising a differential pair oscillation circuit, a voltage regulator circuit and a current source; The differential pair oscillator circuit includes a transistor Q1, a transistor Q2, a parallel resonant circuit, and a load resistor RL. The voltage stabilizing circuit includes transistor Q3, transistor Q4, transistor Q5, transistor Q6, transistor Q8, diode D1, diode D2, resistor R1, resistor R3 and resistor R4; The base of the transistor Q1 is respectively connected to the collector of the transistor Q2, one end of the parallel resonant circuit, one end of the load resistor RL and one end of the output terminal J1; The emitter of the transistor Q1 is connected to the emitter of the transistor Q2 and the current source respectively; The collector of the transistor Q1 is respectively connected to the base of the transistor Q2, the emitter of the transistor Q3, the collector of the transistor Q5, the other end of the parallel resonant circuit, one end of the resistor R1, the other end of the load resistor RL, the positive end of the diode D1, the other end of the output terminal J1 and the power supply VCC; The collector of the transistor Q3 is connected to the base of the transistor Q5 and the current source respectively; The base of the transistor Q3 is connected to the collector of the transistor Q4; The base of the transistor Q4 is connected to the collector of the transistor Q6 and the collector of the transistor Q8 respectively; The emitter of the transistor Q4 is connected to one end of the resistor R4; The emitter of the transistor Q5 is connected to the base of the transistor Q6; The emitter of the transistor Q6 is respectively connected to the current source, one end of the resistor R3, the other end of the resistor R4 and the ground terminal GND; The emitter of the transistor Q8 is connected to the other end of the resistor R1; The base of the transistor Q8 is connected to the other end of the resistor R3 and the negative end of the diode D2 respectively; The cathode terminal of the diode D1 is connected to the anode terminal of the diode D2.

2. The wide-range differential pair frequency stabilization micro-vibration circuit device according to claim 1, characterized in that: The parallel resonant circuit is composed of an inductor L1 and a capacitor C1 connected in parallel.

3. The wide-range differential pair frequency stabilization micro-vibration circuit device according to claim 2, characterized in that: The oscillation frequency f of the circuit device is determined by the inductance L of the inductor L1 and the capacitance C of the capacitor C1 and is expressed by the formula: Wherein, L is the inductance of inductor L1, and C is the capacitance of capacitor C1.

4. The wide-range differential pair frequency stabilization micro-vibration circuit device according to claim 1, characterized in that: The current source is composed of a transistor Q7 and a resistor R2; The base of the transistor Q7 is connected to the collector of the transistor Q3 and the base of the transistor Q5 respectively; The collector of transistor Q7 is connected to the emitter of transistor Q1 and the emitter of transistor Q2 respectively; The emitter of the transistor Q7 is connected to one end of the resistor R2 , and the other end of the resistor R2 is respectively connected to one end of the resistor R3 , the emitter of the transistor Q6 , the other end of the resistor R4 and the ground terminal GND.

5. The wide-range differential pair frequency-stabilizing micro-vibration circuit device according to claim 4, characterized in that: Assuming the operating current of the circuit device is IX, the resistance value calculation formula of the resistor R2 is: IX=0.7 / r2 Where r2 is the resistance value of resistor R2.

6. The wide-range differential pair frequency-stabilized micro-vibration circuit device according to claim 5, characterized in that: The amplification factor of transistor Q3 is set to B3, the amplification factor of transistor Q4 is set to B4, and the amplification factor of transistor Q7 is set to B7. The maximum resistance value r1 of resistor R1 is required to be MAX for: r1 MAX =B3*B4*B7*r2 Set the amplification factor of transistor Q8 to B8, then the maximum resistance value of resistor R3 is r3 MAX for: r3 MAX ＝B8*r1 Where r1 is the resistance value of resistor R1.

7. The wide-range differential pair frequency-stabilizing micro-vibration circuit device according to claim 6, characterized in that: Set the maximum resistance value of resistor R4 to r4 MAX for: r4 MAX = B3*B7*r2.

8. The wide-range differential pair frequency-stabilizing micro-vibration circuit device according to claim 1, characterized in that: The resistance value of the resistor R3 is set to be 10 times greater than the resistance value of the resistor R1 , and the resistance value of the resistor R4 is one tenth of the resistance value of the resistor R1 .

9. The wide-range differential pair frequency stabilization micro-vibration circuit device according to claim 1, characterized in that: The transistor Q3 is used as a power supply adjustment tube of the voltage stabilizing circuit. When the power supply voltage exceeds 1.4V, it works in the amplification area to provide a stable power supply voltage for the current source; when the power supply voltage drops to a low voltage below 1.4V, the transistor Q3 works in the saturation area.

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