Controller for resonant converter and method of operating resonant converter

Abstract

A controller for a resonant converter is described. If the measured current signal is greater than the upper limit low load current threshold, the controller sets an upper limit voltage threshold based on the measured current signal. If the measured current signal is not greater than the upper limit low load current threshold, the controller sets the upper limit voltage threshold based on the power setting signal but independent of the measured current signal. If the measured current signal is less than the lower limit low load current threshold, the controller sets the lower limit voltage threshold based on the measured current signal. If the measured current signal is not less than the lower limit low load current threshold, the controller sets the lower limit voltage threshold based on the power setting signal but independent of the measured current signal. In response to the measured voltage signal exceeding an upper voltage threshold, the controller opens the first switch and closes the second switch. In response to the measured voltage signal falling below a lower limit voltage threshold, the controller opens the second switch and closes the first switch.

Description

Technical Field

[0001] The present disclosure relates to a controller for a resonant converter, and a method of operating a resonant converter. Background Art

[0002] The resonant power converter can use a half-cycle-by-half-cycle shutdown mechanism. Summary of the invention

[0003] According to a first aspect of the present disclosure, a controller for a resonant converter is provided, wherein the resonant converter is used to supply electric energy from a supply source to a load, and the resonant converter includes:

[0004] a first switch and a second switch, the first switch and the second switch being connected in series with each other between a supply source and a reference terminal, wherein when the first switch is closed and the second switch is open, the resonant converter has a high-side switching half-cycle, and wherein when the first switch is open and the second switch is closed, the resonant converter has a low-side switching half-cycle; and

[0005] a resonant tank electrically connected to the first switch and the second switch, wherein the resonant tank includes a resonant capacitor;

[0006] The controller is configured as:

[0007] receiving a measured current signal representative of a current flowing in the resonant tank;

[0008] receiving a measured voltage signal representative of a voltage at a predetermined point in the resonant tank;

[0009] receiving a power setting signal defining a requested power level for the load;

[0010] If the measured current signal is greater than the upper low load current threshold, setting an upper voltage threshold based on the measured current signal;

[0011] if the measured current signal is not greater than the upper low load current threshold, setting the upper voltage threshold based on the power setting signal but independent of the measured current signal;

[0012] If the measured current signal is less than a lower low load current threshold, setting a lower voltage threshold based on the measured current signal;

[0013] If the measured current signal is not less than a lower low load current threshold, setting a lower voltage threshold based on the power setting signal but independent of the measured current signal;

[0014] In response to the measured voltage signal exceeding an upper voltage threshold, opening the first switch and closing the second switch; and

[0015] In response to the measured voltage signal falling below a lower voltage threshold, the second switch is opened and the first switch is closed.

[0016] In one or more embodiments, the controller is additionally configured to:

[0017] setting a value of an upper low load current threshold based on the received power setting signal; and

[0018] A value of a lower low load current threshold is set based on the received power setting signal.

[0019] In one or more embodiments, the controller is further configured to, if the received power setting signal is greater than a high load threshold, then:

[0020] setting the value of the upper low load current threshold to a high load constant value; and

[0021] Set the value of the lower low load current threshold to the high load constant value.

[0022] In one or more embodiments, the controller is further configured to, if the received power setting signal is less than the high load threshold, then:

[0023] setting the value of the upper low load current threshold to a value less than the high load constant value; and

[0024] Set the value of the lower low load current threshold to a value smaller than the high load constant value.

[0025] In one or more embodiments, the controller is further configured to, if the received power setting signal is less than the high load threshold, then:

[0026] setting the value of the upper low load current threshold as a function of the received power setting signal; and

[0027] The value of the lower low load current threshold is set as a function of the received power setting signal.

[0028] In one or more embodiments, the controller is configured to:

[0029] If the received power setting signal is zero, setting the value of the upper low load current threshold to a zero load current value; and

[0030] If the received power setting signal is zero, the value of the lower low load current threshold is set to a zero load current value.

[0031] In one or more embodiments, the controller is further configured to, if the received power setting signal is less than the high load threshold, then:

[0032] Set the value of the upper low load current threshold to:

[0033] a first function of the received power setting signal, conditional upon the received power setting signal being greater than a medium load threshold; and

[0034] a second function of the received power setting signal, conditional upon the received power setting signal being less than a medium load threshold; and

[0035] Set the value of the lower low load current threshold to:

[0036] a first function of the received power setting signal, conditional upon the received power setting signal being greater than a medium load threshold; and

[0037] A second function of the received power setting signal is conditional upon the received power setting signal being less than a medium load threshold.

[0038] In one or more embodiments, the first function and the second function are linear functions. The first function may be steeper than the second function.

[0039] In one or more embodiments, the controller is further configured to, if the power setting signal is less than the offset current applied power threshold, then:

[0040] adjusting the measured current signal by adding an offset current signal to the measured current signal, and

[0041] The adjusted measured current signal is used instead of the measured current signal.

[0042] In one or more embodiments, the magnitude of the offset current signal increases as the magnitude of the power setting signal decreases.

[0043] In one or more embodiments, the magnitude of the offset current signal increases linearly as the magnitude of the power setting signal decreases.

[0044] In one or more embodiments, when the magnitude of the power setting signal is zero, the magnitude of the offset current signal has a maximum value.

[0045] Also disclosed is a method of operating a resonant converter, wherein the resonant converter is used to supply electrical energy from a supply source to a load, the resonant converter comprising:

[0046] a first switch and a second switch, the first switch and the second switch being connected in series with each other between a supply source and a reference terminal, wherein when the first switch is closed and the second switch is open, the resonant converter has a high-side switching half-cycle, and wherein when the first switch is open and the second switch is closed, the resonant converter has a low-side switching half-cycle; and

[0047] a resonant tank electrically connected to the first switch and the second switch, wherein the resonant tank includes a resonant capacitor;

[0048] The methods include:

[0049] receiving a measured current signal representative of a current flowing in the resonant tank;

[0050] receiving a measured voltage signal representative of a voltage at a predetermined point in the resonant tank;

[0051] receiving a power setting signal defining a requested power level for the load;

[0052] If the measured current signal is greater than the upper low load current threshold, setting an upper voltage threshold based on the measured current signal;

[0053] if the measured current signal is not greater than the upper low load current threshold, setting the upper voltage threshold based on the power setting signal but independent of the measured current signal;

[0054] If the measured current signal is less than a lower low load current threshold, setting a lower voltage threshold based on the measured current signal;

[0055] If the measured current signal is not less than a lower low load current threshold, setting a lower voltage threshold based on the power setting signal but independent of the measured current signal;

[0056] In response to the measured voltage signal exceeding an upper voltage threshold, opening the first switch and closing the second switch; and

[0057] In response to the measured voltage signal falling below a lower voltage threshold, the second switch is opened and the first switch is closed.

[0058] Although the present disclosure allows for various modifications and alternative forms, the details of the present disclosure have been shown in the drawings by way of example and will be described in detail. However, it should be understood that other embodiments than the specific embodiments described are also possible. All modifications, equivalents and alternative embodiments falling within the spirit and scope of the appended claims are also encompassed.

[0059] The above discussion is not intended to present every example embodiment or every implementation within the scope of the current or future claim sets. The figures and the following detailed description also illustrate various example embodiments. The various example embodiments may be more fully understood by considering the following detailed description in conjunction with the figures. BRIEF DESCRIPTION OF THE DRAWINGS

[0060] One or more embodiments will now be described, by way of example only, with reference to the accompanying drawings, in which:

[0061] Figure 1 A resonant converter is shown as part of the overall power supply;

[0062] Figure 2A and 2B The basic principle of capacitor voltage control for a resonant converter is shown, where the resonant capacitor is on the ground side (shunt);

[0063] Figure 3 shows a block diagram of a resonant converter with capacitor voltage control having a resonant capacitor on the ground side;

[0064] Figure 4 shows a capacitor voltage control (e.g., Figure 3 a signal of a resonant converter of the capacitor voltage control in the resonant converter, the resonant converter including a delay compensation mechanism;

[0065] Figure 5 The situation without load is shown;

[0066] Figure 6 A first step of adding the slope compensation signal to the Vcap level (i.e., upper and lower voltage thresholds) is shown;

[0067] Figure 7 Convergence to a single well-defined trajectory is shown;

[0068] Figure 8 A state plane representation showing operation according to an aspect of the present disclosure is shown;

[0069] Fig. 9 The case with zero load (Δv=0, resulting in Vcaph=Vcapl=0) and without adding Ipeak_ofs is shown;

[0070] Fig.10 The same situation with zero load and Ipeak_ofs added is shown;

[0071] Fig.11 shows a capacitor voltage control method based on the capacitor voltage control method explained above. Fig.10 how it is possible to achieve a clearly defined trajectory in a given situation;

[0072] Fig.12 Shown for Fig.10 How the trajectory (again shown as a dotted line) converges from a larger trajectory to a stable trajectory in the case of

[0073] Fig.13 Shown with Fig.12 Same setup, but now with additional delay;

[0074] Fig.14 The case from a smaller trajectory and convergence to a steady-state solution is shown;

[0075] Fig.15 shows a graph of vcaph level and vcapL level as a function of ctrl_p;

[0076] Fig.16 shows a graph showing how Ipeak (ie, upper and lower low-load current thresholds) can be defined as a function of ctrl_p (ie, a power setting signal);

[0077] Fig.17 The case of half load is shown (ctrl_p=1);

[0078] Fig.18 Shows Fig.16 The situation of the area where the functionality shown in is beginning to be enabled (Ctrl-p=Ctrl_p_border1);

[0079] Fig.19 shows a situation in the middle of the region where Ipeak decreases based on the value of ctrl_p, but the sloped region of the upper and lower current thresholds does not affect the control of the resonant converter;

[0080] Fig. 20 The situation at the edge of the upper current threshold and the lower current threshold where the slope area takes control is shown;

[0081] Fig.21 A further narrowing of the trajectory for smaller ctrl_p is shown;

[0082] Fig. 22 A zero load situation is shown;

[0083] Figure 23-25 shows how the power (Ctrl_p) decreases (and finally Fig.25 becomes 0 in the middle), how does ΔV become 0 under zero load?

[0084] Fig.26 The offset current applied power threshold is shown;

[0085] Fig. 27 Demonstrate soft start and soft stop;

[0086] Fig.28A and 28B An embodiment of a controller for a resonant converter according to the present invention is shown;

[0087] Fig.29 Shows Fig.28A and 28B Simulink implementation of the two blocks;

[0088] Fig.30Include more details of the first sub-block;

[0089] Fig.31 Includes more details of the second sub-block;

[0090] Fig.32 A lookup table structure for generating an Ipeakref signal is shown;

[0091] Fig.33 shows the lookup table structure used to generate the refscale function in Mathcad and the resulting graph;

[0092] Fig.34 shows the definition of VCRH and VCRL signals as a function of ctrl_p;

[0093] Fig.35 The calculation and graph results in Mathcad of the overall main function block are shown;

[0094] Fig.36A and 36B shows the signal during operation in low load;

[0095] Fig.37 shows a low power mode complete switching cycle in a state plane representation; and

[0096] Fig.38 An example embodiment of a method of operating a resonant converter according to the present disclosure is shown. DETAILED DESCRIPTION

[0097] The resonant power converter may use a half-cycle-by-half-cycle shutdown mechanism that involves sensing the resonant capacitor voltage (Vcap) and comparing Vcap to a power reference level for closed-loop operation. This may be referred to as capacitor voltage control. Instead of directly controlling the switch by forcing the switching frequency, the switching frequency is generated by the Vcap control mechanism.

[0098] Figure 1 A resonant converter is shown as part of the overall power supply, which in this example is used in an adapter. The adapter includes the following blocks:

[0099] ●Active bridge controller (power rectification);

[0100] PFC controller section (power factor correction using boost converter architecture);

[0101] The resonant LLC controller section (driving half-bridge stage plus LLC resonant tank including tapped winding transformer); and

[0102] ●SR controller (synchronous rectifier stage for generating the rectified output voltage Vout).

[0103] The resonant converter can be sized using the first harmonic approximation. In addition, controlling the output power using the capacitor voltage control method has the following advantages:

[0104] ●Possibility to use a dedicated low power mode.

[0105] • Better dynamics, since the delivered power follows the set point of the resonant tank variable almost immediately. This simplifies the control loop design.

[0106] The resonant converter may have a resonant capacitor in series with the half-bridge node, and the information for controlling the switches may be derived from the voltage across the resonant capacitor (the side not connected to the half-bridge node).

[0107] This method actually senses the voltage across the transformer, and therefore the DC component of the sensed signal is zero. In some applications, the high voltage signal at the resonant capacitor / transformer node should be attenuated to a low voltage signal that can be processed by an integrated circuit (IC). For this purpose, a capacitive voltage divider can be used, because a resistive voltage divider will produce undesirable low-pass filtering. Due to the fact that the DC component is zero, no information about the DC component is needed, so a capacitive voltage divider (which cannot transmit a DC component) is sufficient.

[0108] As an alternative, the resonant capacitor can also be placed in the ground branch in series with the primary winding of the transformer, e.g. Figure 1 As shown in .

[0109] Figure 2A and 2B The basic principle of capacitor voltage control for a resonant converter is shown, where the resonant capacitor is on the ground side (shunted). That is, the resonant capacitor is shunted between ground and the voltage source (Vbus) in order to reduce the ripple current in the ground path and the supply path.

[0110] Figure 2A and 2B The timing control principle of the switch using this embodiment is also shown.

[0111] Figure 2A and 2B The resonant converter includes a controller (not shown) that operates a first controllable switch and a second controllable switch arranged in series. These switches are connected between a supply source (Vbus) and a reference terminal (ground). The controller controls the first high-side switch using a signal marked as gh. The controller controls the second low-side switch using a signal marked as gl. The voltage at the half-bridge node (the node between the high-side switch and the low-side switch) is marked as Vhb.

[0112] The controller receives a measured voltage signal (Vcr) representing the voltage at a predetermined point in the resonant tank. In this example, the measured voltage signal (Vcr) represents the voltage across the resonant capacitor. Figure 2A and 2B As shown in , the controller compares the measured voltage signal (Vcr) with the upper voltage threshold (VCRH) and the lower threshold (VCRL). Based on the comparison, the switching half cycle ends. More specifically, in response to the measured voltage signal (Vcr) exceeding the upper voltage threshold (VCRH), the second low-side switch is closed and the first high-side switch is disconnected. In response to the measured voltage signal (Vcr) falling below the lower voltage threshold (VCRL), the second low-side switch is disconnected and the first high-side switch is closed. It should be understood that in an alternative embodiment, the polarity of the signal can be reversed so that: in response to the measured voltage signal (Vcr) falling below the lower voltage threshold, the second low-side switch is closed and the first high-side switch is disconnected; and in response to the measured voltage signal (Vcr) exceeding the upper voltage threshold, the second low-side switch is disconnected and the first high-side switch is closed. In this example, a non-overlapping time is implemented after one of the switches is opened and before the other switch is closed. In this way, there is a slight delay between the end of one half cycle and the beginning of the next half cycle. For simplicity, the attenuation of the Vcr voltage for the controller IC is not included Figure 2A and 2B middle.

[0113] for Figure 2A and 2B As shown in the graph, the resonant converter uses an input voltage Vbus of 400V, and therefore Vhb switches between 400V and 0V. For this example:

[0114] deltaV=360-40=320V;

[0115] VCRH = Vbus / 2 + deltaV / 2 = 200 + 160 = 360V; and

[0116] VCR1=Vbus / 2-deltaV / 2=200-160=40V.

[0117] gh and gl have an amplitude of 1V, which is a convenient representation of true = 1 and false = 0.

[0118] Figure 2A and 2B The setting of also takes into account the DC component of the resonant capacitor. The DC component is defined by the input voltage to the half-bridge (Vbus) multiplied by the duty cycle of the half-bridge voltage (typically Vbus / 2 at 50% duty cycle operation).

[0119] In practice, a capacitive voltage divider cannot represent the DC component of the resonant capacitor due to leakage, and a resistive voltage divider cannot reliably attenuate the high frequency portion of the Vcr signal due to parasitic capacitance. In the case of a resistive and capacitive voltage divider in parallel, it is difficult to maintain the same attenuation ratio due to component tolerances. The DC component across the capacitive voltage divider (and optionally the resistive voltage divider in parallel) can be restored in a practical circuit by adding a so-called symmetry loop.

[0120] The symmetry loop compares the measured duty cycle to the expected value of 50% and corrects the capacitive divider DC component so that a 50% duty cycle is achieved. The switch, together with the voltage divider, produces a DC component at the voltage divider. This DC component, together with the vcrh level and the vcrl level, defines the duty cycle and symmetry / asymmetry. The DC symmetry loop senses the duty cycle and adjusts the DC value so that a 50% duty cycle occurs with a symmetrical half cycle. The symmetry loop can rely on completing each half cycle by Vcr crossing one of the thresholds (VRCH or VCRL) so that a direct correlation is maintained with the divided resonant capacitor voltage. This includes the DC level and the difference between the two threshold levels (VRCH or VCRL).

[0121] The following includes calculations of Figure 2A and 2B The converted power of the resonant converter is ΔV, which is the difference between two threshold levels VCRH and VCRL (these thresholds may also be referred to as vcaph and vcapl).

[0122] ●ΔV across Cr between the start and end of each half cycle => charge difference

[0123] ○Power = average current × input voltage I x Vbus

[0124] ○I=Q x Fswitch

[0125] ○Q=Cr xΔV

[0126] ○Power = Cr x ΔV x Fswitch x Vbus can be set by switching cycle

[0127] It can be seen that the converted energy per switching cycle is proportional to ΔV. In theory, this means that when deltaV=0, the converted power can become zero. Values ​​less than 0 are not possible because it would cause an undefined frequency drift to a maximum value, and therefore there is an infinite number of possible solutions for the frequency at deltaV=0. Therefore, as the converter causes an abrupt jump to maximum frequency when approaching zero power, at least a maximum frequency limit is required.

[0128] In a practical power converter, the path between: i) the comparator comparing the measured voltage signal Vcr with the voltage threshold; and ii) the half-bridge node includes parasitic delays. This means that the effective level of the measured voltage signal Vcr at the moment of half-bridge switching is greater than the applied Vcap level that triggers the switching operation. The result is a shift to a converted power greater than that defined by the voltage threshold level. This becomes a problem especially for high frequency resonant converters, e.g., with nominal operating frequencies up to 500kHz and maximum operating frequencies up to 1Mhz when starting with low Vout.

[0129] This can also lead to a situation where the power cannot go to 0 while switching, while jumping to the maximum frequency at a small % power, and then the power suddenly goes to 0 and the frequency goes to Fmax. Therefore, when the load requires a power level below this small percentage power, there may not be a stable way to reach this power level. This creates erratic switching, which is unacceptable.

[0130] In some applications, it may be important to keep the converter switching when unloaded. One or more of the examples described herein may address this problem.

[0131] Figure 3 A block diagram of a resonant converter with capacitor voltage control having a resonant capacitor on the ground side is shown. The resonant converter also has a symmetrical loop, a cycle-by-cycle comparator and delay compensation based on adding a scaled version (k1) of the primary current.

[0132] exist Figure 3 In , delay compensation is performed by adding a signal proportional to the derivative of Vcr to the original Vcr signal. Figure 3 301. When Vcr is the integral of the resonant current, it means that the derivative of Vcr is related to the resonant current itself. Therefore, a scaled portion of the primary current is added to Vcr in order to obtain a slightly phase-shifted signal, as if there was a negative delay. This negative delay can then partially compensate for the effect of the positive delay. This functionality will not be described in detail as it is not the focus of the present disclosure.

[0133] This solution provides for a high frequency resonant converter to compensate for the effects of delays above about 5% of the maximum load, thereby resolving the inaccuracy in the relationship between ΔV and power. However, it may not be able to prevent complications when operating at no load.

[0134] Another operating mode is burst mode operation. In this case, power is concentrated into larger power bursts followed by intervals where no switching occurs. This works fine, but burst mode can generate audible noise. Therefore, the bursts can have soft edges (called soft start and soft stop). This means that the bursts start with zero power and even greater frequency in order to additionally reduce the magnetizing current, thereby also generating soft edges. (The magnetizing current is the main root cause of audible noise because it is responsible for the attraction between the transformer core halves that causes vibrations and thus generates audible noise.)

[0135] However, the effect of the delay mentioned above may prevent the possibility of going to zero power, and may also prevent the possibility of increasing the switching frequency required for the soft edge of the magnetizing current. The examples disclosed herein provide a solution to this problem.

[0136] One or more of the examples described herein solve the problems in the prior art, even in Figure 3 The delay compensation provided by the component identified by reference 301 in the figure can also prevent no-load and reduction of magnetizing current. In this way, the examples disclosed herein can provide a high-frequency resonant converter that can no-load when switching and also allow burst mode with soft start and soft stop, which can reduce magnetizing current during soft start and soft stop. This can bring the audible noise back to an acceptable level.

[0137] Figure 4 shows a capacitor voltage control (e.g., Figure 3 The converter is operated at a medium power level.

[0138] Figure 4 The left plot shows the signal in the time domain:

[0139] ●Vcap_recon: divided and DC biased resonant capacitor voltage.

[0140] ●Vcaprecon_corr: Vcap_recon delay compensated by the scaled resonant current.

[0141] - Vcaph, Vcapl: Control levels of the cycle-by-cycle comparator (which may also be referred to as upper and lower voltage thresholds).

[0142] ●Iprimsensed: The sensed resonant current.

[0143] The right graph shows the same signal graph in a state plane representation.

[0144] In the state plane, the voltage across the resonant capacitor (a capacitance-divided version thereof) is plotted as a horizontal axis relative to the resonant current plotted at the vertical axis. In the left graph, time flows from left to right. The resonant current between t1 and t2 represents the secondary side rectifier conduction interval, giving a high resonant frequency. In the state plane, time refers to points of the curve moving clockwise around in time. The interval t1-t2 is plotted as a dashed line 402.

[0145] Plotting such a state plane has advantages because information can be obtained from the plotted trajectory about the dynamic behavior of the converter. When the axes are scaled appropriately, the portion of the trajectory during the interval when the secondary diode is conducting is followed by the portion of the circle with the midpoint of the voltage across the resonant component (called the forcing voltage). During the interval t1-2, the forcing voltage is the half-bridge node voltage reflected output voltage = Vbus reflected output voltage. Vbus is the resonant converter supply voltage. In this example, since the capacitive divider is biased with a 2.5V on-chip reference voltage via a large series resistor, the divided resonant capacitor voltage (at Figure 3 and 4 The Vcap_recon (referred to as vcap_recon) has a DC level of 2.5V.

[0146] With fixed upper and lower voltage thresholds, a well-defined trajectory is produced for the region in which power is delivered to the load, such as Figure 4 as shown in .

[0147] Figure 5 The situation with no load (neglecting delays) is shown.

[0148] exist Figure 5 , ΔV (which is equal to Vcaph-Vcapl; i.e., upper voltage threshold minus lower voltage threshold) is set to 0 in order to produce zero power. Due to zero power, the resonant current (Iprim on the vertical axis) is then only the magnetizing current. There are now different trajectories as indicated by the larger outer trajectory 503 and the smaller inner trajectory 504 as 2 of many examples. This means that both the magnetizing current and the switching frequency are not defined. Smaller and larger refer to the smaller or larger peak value, which determines the size of the trajectory.

[0149] When delays are taken into account, it follows from the simulation that even at low load (i.e. not necessarily zero load), the power cannot be properly controlled and the frequency can drift to the maximum frequency while the power drifts to 0. With the overall feedback loop that regulates the output voltage closed, the result is unstable regulation.

[0150] Ways to address these issues will now be described.

[0151] Figure 6The first step of adding the slope compensation signal to the Vcap level (i.e., the upper and lower voltage thresholds) is shown. The slope is based on the resonant current (at Figure 6 , referred to as Iprim in the state plane). This makes the Vcap level Vcaph=Vcaph0+ax Iprim and Vcapl=Vcapl0+ax Iprim. Where: Vcaph is the upper voltage threshold; Vcaph0 is a constant that defines the position of the upper voltage threshold slope in the state plane; Vcapl is the lower voltage threshold; Vcapl0 is a constant that defines the position of the lower voltage threshold slope in the state plane; a is a constant that defines the gradient of the slope; and Iprim is a constant that defines the resonant current, which can also be referred to as the measured current signal. Vcaph0 is generated when ax Iprim=0, so it defines the intersection with the x-axis. Since 'a' is set to a specific value, Vcaph0 can be regarded as defining the vertical position of the inclined line in the curve graph. The same is true for Vcapl0.

[0152] Vcaph (upper voltage threshold) Figure 6 605. Vcapl (lower voltage threshold) is shown in Figure 6 606. During operation of the resonant converter, as shown by Figure 6 As indicated by the dashed lines in , the state of the switches of the resonant converter changes when the trajectory intersects the upper voltage threshold 605 and the lower voltage threshold 606. In this way, the system is forced to follow a trajectory where the effective Vcap level (ie, the upper voltage threshold and the lower voltage threshold) is zero.

[0153] Due to slope compensation, the effective Vcap level will become positive when the trajectory is too small and negative when the trajectory is too large. Therefore, the switching cycle will be forced to follow Figure 6 The desired convergence of the desired stable trajectory is shown in .

[0154] Figure 7 Convergence to a single well-defined trajectory is shown. Figure 7 It is shown how this convergence works using the choice of constructing a state plane trajectory defined by the excitation voltage and radius.

[0155] Starting at point A, the system will follow the dotted line trajectory defined by the current dependent vcap levels 705, 706, and finally converge to the dashed line trajectory 707. That is, starting from point A:

[0156] • Operation continues according to the trace labeled 708a until the measured voltage signal crosses the lower voltage threshold 706;

[0157] • Operation continues according to the trace labeled 708 b until the measured voltage signal crosses the upper voltage threshold 705 ;

[0158] • Operation continues according to the trace labeled 708c until the measured voltage signal crosses the lower voltage threshold 706. As can be seen, trace 708c has a larger range than trace 708a;

[0159] • Operation continues according to the trace labeled 708d until the measured voltage signal crosses the upper voltage threshold 705. As can be seen, trace 708d has a larger range than trace 708b;

[0160] ● The operation has now converged on a final well-defined trajectory, which is Figure 7 It is shown as a dotted line 707 in FIG.

[0161] The points that still need to be resolved are:

[0162] • How to set the appropriate Vcap level to obtain the desired accuracy of the primary current or output current as a function of ΔV; and

[0163] ●How to define it so that it can be used in a stable manner over the entire load range.

[0164] The problem with the first point arises from the fact that the Vcap level at the turn-off moment determines the energy to be converted to the load, while the main peak current at the turn-off moment does not directly clear to zero, but will affect the circulating energy. Therefore, it turns out that the converted energy depends on unknown variables.

[0165] According to a first aspect of the present disclosure, examples disclosed herein may apply the slope only during a well-defined portion of the Vcr window to prevent multiple trajectories.

[0166] Figure 8 805 and the lower voltage threshold 806 may be set to: depending on the value of the measured current signal (Iprim) - this is Figure 8 or independent of the value of the measured current signal (Iprim) - this is Figure 8 The vertical area shown in , and can be considered as a fixed Vcap (or threshold) level.

[0167] In this way, the slope is applied only during part of the trajectory, where multiple trajectories are possible. An upper voltage threshold 805 and a lower voltage threshold 806 are also set to define the transition from the region with the slope to the region with the fixed Vcap level in a specific way. The fixed Vcap level is used in the region where a precise resonant current amplitude is required.

[0168] Now refer to Figure 8 The state plane representation describes an example embodiment of a controller for a resonant converter. The resonant converter is used to supply electrical energy from a supply source to a load. As described above, the resonant converter includes a first switch and a second switch connected in series with each other between the supply source and a reference terminal (e.g., a ground terminal). When the first switch is closed and the second switch is open, the resonant converter has a high-side switching half-cycle. When the first switch is open and the second switch is closed, the resonant converter has a low-side switching half-cycle. The resonant converter also has a resonant tank electrically connected to the first switch and the second switch. The resonant tank includes at least a resonant capacitor. In some examples, the resonant tank also includes a resonant inductor and a transformer, and optionally includes a resonant capacitor at the other side of the transformer (for a CLLC converter). Most of these features of the resonant converter are included in various figures of the present application, including Figure 1 .

[0169] The controller receives a measured current signal (Iprim) representing the current flowing in the resonant tank. The controller also receives a measured voltage signal (Vcap) representing the voltage at a predetermined point in the resonant tank. In addition, the controller receives a power setting signal (ctrl_p, which will be discussed in more detail below) that defines a requested power level for the load.

[0170] If the measured current signal (Iprim) is greater than the upper low load current threshold (low) 809, the controller sets the upper voltage threshold (Vcaph) 805 based on the measured current signal (Iprim). Figure 8 The slope region of the upper voltage threshold (Vcaph) 805 is shown in FIG. Figure 7 As discussed, the ramp region enables a well-defined switching frequency to be achieved for low load operation. If the measured current signal (Iprim) is not greater than the upper low load current threshold (low) 809, the controller sets the upper voltage threshold (Vcaph) 805 based on the power setting signal (ctrl_p) but independently of the measured current signal (Iprim). Figure 8 805. This represents the conventional operation of the resonant converter such that it can be controlled according to the requested power level of the load, without the need for compensation provided by the sloped region of the upper voltage threshold (Vcaph) 805 under operating conditions. That is, the load is not so low that the resonant converter can reliably operate according to its conventional closed-loop control.

[0171] The controller operates in a similar manner for the lower voltage threshold (Vcapl) 806. That is, if the measured current signal (Iprim) is less than the lower low load current threshold (-low) 812, the controller sets the lower voltage threshold (Vcapl) 806 based on the measured current signal (Iprim). Figure 8 806. If the measured current signal (Iprim) is not less than the lower low load current threshold (-low) 810, the controller sets the lower voltage threshold (Vcapl) 806 based on the power setting signal (ctrl_p) but independently of the measured current signal (Iprim). Figure 8 The vertical region of the lower voltage threshold (Vcapl) 806 is shown in FIG.

[0172] Once the upper voltage threshold (Vcaph) 805 and the lower voltage threshold (Vcapl) 806 have been set in this manner, the controller may:

[0173] In response to the measured voltage signal exceeding an upper voltage threshold (Vcaph) 805, opening the first switch and closing the second switch; and

[0174] In response to the measured voltage signal falling below a lower voltage threshold (Vcapl) 806, the second switch is opened and the first switch is closed.

[0175] It should be understood that it is these switching operations that cause the trajectory in the state plane diagram to change direction.

[0176] In this way, the reference resonant current levels Ipeak=low (upper low load current threshold 809) and Ipeak=-low (lower low load current threshold 810):

[0177] The Vcap level becomes independent of the resonant current for Iresonant < low; and

[0178] ●Vcap level becomes independent of resonant current for Iresonant>- low.

[0179] This example produces a stable trajectory because the trajectory size increases when it is too small and decreases when it is too large. Therefore, it converges to a stable solution.

[0180] The ability to increase the track when the track is less than the desired result is such that for the region with a fixed Vcap level (i.e., the vertical portion of the upper voltage threshold (Vcaph) 805 and the lower voltage threshold (Vcapl) 806), ΔV = Vcaph - Vcap > 0. Therefore, the track area will increase due to the positive value of ΔV, thereby increasing the track with each complete track cycle. This will make the track less than Figure 8 Each trajectory shown as a dashed line in converges to the dashed line, which represents a stable trajectory. Similarly, Figure 8 The trajectory shown as a dashed line in FIG will also converge towards the dashed line trajectory because, due to the slope, the Vcap level (i.e., upper voltage threshold (Vcaph) 805 and lower voltage threshold (Vcapl) 806) becomes negative when the trajectory is greater than the dashed line. This is because a negative ΔV makes the trajectory smaller for each complete loop (= switching cycle), as already mentioned about Figure 7 Explained.

[0181] Ipeak_ofs

[0182] Figure 8 The ΔV value in is shown as the variable 'Ipeak_ofs' as the distance from Ipeak = low (upper low load current threshold 809) to a point on the upper voltage threshold line (Vcaph) 805, where Vcaph becomes 2.5V. The same is shown for the distance from Ipeak = -low (upper low load current threshold 809) to a point on the lower voltage threshold line (Vcapl) 806, where Vcapl. As discussed above, the slope at the Vcap level improves performance for no-load conditions in order to provide convergence to a single well-defined trajectory (see Figure 7 ).

[0183] The Ipeak_ofs value (which can be considered as an offset value of the sensed current signal) defines the slope towards the positive Vcap value ( Figures 9 to 14 curve) and negative Vcapl values ​​( Figures 9 to 14 The area where the dashed curve in ( ) extends.

[0184] Fig. 9 The case with zero load (Δv=0, resulting in Vcaph=Vcapl=0) and no Ipeak_ofs is shown. In this case, based on the Figure 7 With the capacitor voltage control method explained, no well-defined trajectory is possible, since an infinite number of switching frequencies are possible.

[0185] Fig.10 The same situation is shown with zero load and Ipeak_ofs added.

[0186] Fig.11 shows a capacitor voltage control method based on the capacitor voltage control method explained above. Fig.10 How is it possible to achieve a clearly defined trajectory in a given situation? Fig.11 It is shown how the trajectory (shown as dashed lines) converges from a smaller trajectory to a stable trajectory.

[0187] Fig.12Shown for Fig.10 In the case of , how the trajectory (again shown as a dotted line) converges from a larger trajectory to a stable trajectory.

[0188] In a practical case with additional delay, each half cycle is completed later than the actual time at which the measured voltage signal actually crosses the Vcaph, Vcapl values.

[0189] Fig.13 Shown with Fig.12 Same setup, but now with an additional delay between: i) the cycle-by-cycle comparator that determines the crossing of the measured voltage signal with one of the Vcaph, Vcapl values; and ii) the voltage at the half-bridge node that reacts.

[0190] The effect is that at each crossing of the Vcap level, a trajectory larger than the limit is followed. This produces a stable but larger than expected trajectory (see double dotted line 1320). The trajectory achieved in the absence of additional delay and the switching that occurs immediately when the vcap level crosses is shown as long dashed line 1321.

[0191] Fig.14 The case from a smaller trajectory and convergence to a steady-state solution is shown.

[0192] Furthermore, it is evident from the figure here that a trajectory larger than that desired will be followed. To maintain zero power, the effective vcap level should be zero. The effective Vcap level is the value of the Vcr coordinate of p(Vcr,Iresonant) at the moment the half-bridge node switches. In the trajectory, this can be considered as the moment when the trajectory changes direction. Fig.13 and 14 In , these levels are still zero, so they define zero load as required.

[0193] For larger delays, it may not be possible to reach zero power because the trace will become larger with larger delays. To address this issue, two things can be done:

[0194] 1. Slope compensation can be made steeper, forcing a larger trajectory to be completed at a smaller Vcaph value and a larger Vcapl value.

[0195] 2. ΔV can be made negative and the Ipeak_ofs value can be increased to allow for stable convergence to a steady state trajectory.

[0196] Definition of settings as a function of load

[0197] For the main feedback loop regulating the output voltage of the overall converter, it is important to keep the loop gain under control because the loop gain is related to the limits for the stability of the main regulation loop and the bandwidth (the ability to react to load transients at a certain stable speed).

[0198] Regarding the cycle-by-cycle part, the contribution to this loop gain is defined by the transfer of the average output current from the main control parameter (ΔV) to the current in the rectifier diode during the switching cycle. For the capacitor voltage control method, ΔV is proportional to the converted energy per switching cycle. Figure 2A and 2B Explain this situation.

[0199] As also discussed above, examples of the present disclosure may add slope compensation and shift the vcap level. An abrupt transition to this situation may result in an undesirable discontinuity in the transfer from ΔV to the converted energy in each switching cycle. Therefore, features of some aspects of the present disclosure will activate the functionality discussed above as a function of the control parameter ΔV.

[0200] In this relationship, it is important to understand that for medium to full power operation, the functionality described above with respect to defining upper and lower voltage thresholds as a function of the measured current signal may not be enabled, thereby enabling ΔV to be proportional to the primary control variable (ctrl_p) of the converter according to the following equation:

[0201] ●Vcaph=Ctrl_p*gain1

[0202] ●Vcapl=-Ctrl_p*gain1

[0203] Fig.15 A graph of vcaph and vcapL levels as a function of ctrl_p is shown, where gain1=1 for both equations above.

[0204] The gain from ctrl_p to ΔV is determined in a practical converter by some settings (e.g., the choice of resonant tank components and capacitive divider ratios), but for the purposes of this discussion, only the principle is important. Assuming no load involves ctrl_p=0 and maximum load involves ctrl_p=2, the result is that for a high frequency LLC converter with internal delays, the actual output power will be shifted by some offset due to the delay to reach a more positive value. Using the Figure 3 The delay compensation method represented by component 301 in , reasonably compensates for the shift. However, for small powers, such as less than about 5% of full load, the converter may not be able to operate in a stable manner.

[0205] To address this issue, as will now be discussed, examples of the present disclosure may activate its functionality in a specific way so that the transfer from ctrl_p to the output current remains as constant as possible, and preferably, it does not go to zero or even change sign.

[0206] Ipeak Definition

[0207] This section describes how to set the upper low load current threshold and the lower low load current threshold. Figure 8 These thresholds are described and will be collectively identified as Ipeak in this section.

[0208] The first part of this section deals with the peak current which defines the slope compensation for values ​​above the resonant current greater than Ipeak.

[0209] Fig.16 A graph showing how Ipeak (ie, the upper and lower low-load current thresholds) may be defined as a function of ctrl_p (ie, the power setting signal) is shown.

[0210] As will be discussed below, and as Fig.16 It indicates that the controller of the resonant converter sets the values ​​of the upper low-load current threshold and the lower low-load current threshold based on the received power setting signal (ctrl_p).

[0211] During medium to high power levels, no slope should be applied, and therefore the Ipeak level should always be higher than the peak value of the resonant current. Therefore, the high load threshold (Ctrl_p_border1) 1623 is used to define the boundary value of ctrl_p. Fig.16 The value of the high load threshold (Ctrl_p_border1) 1623 in Ctrl_p is 0.2. When ctrl_p is above the high load threshold (Ctrl_p_border1) 1623, Ipeak is set to a value that is not expected to affect the operation of the resonant converter at such relatively high loads (in Fig.16 ). Therefore, no slope will occur and the Vcap level (ie, the upper and lower voltage thresholds) is fixed and independent of the resonant current defined by ctrl_p.

[0212] In this way, if the received power setting signal (ctrl_p) is greater than the high load threshold (Ctrl_p_border1) 1623, the controller of the resonant converter:

[0213] Set the value of the upper low load current threshold (which can be considered as Ipeak) to the high load constant value ( Fig.16 4A in ). The high load constant value has a relatively high magnitude so that the slope does not affect the operation of the resonant converter for relatively high load operation; and

[0214] Set the value of the lower low load current threshold (which can be considered as -Ipeak) to the high load constant value ( Fig.16 -4A in).

[0215] like Fig.16 As shown in FIG. 1 , if the received power setting signal (ctrl_p) is less than the high load threshold (Ctrl_p_border1) 1623, the controller:

[0216] setting the value of the upper low load current threshold (Ipeak) to a value smaller than the high load constant value; and

[0217] The value of the lower low load current threshold (-Ipeak) is set to a value smaller than the high load constant value.

[0218] More precisely, for Fig.16 For example, if the received power setting signal (ctrl_p) is less than the high load threshold (Ctrl_p_border1) 1623, the controller:

[0219] setting the value of the upper low load current threshold (Ipeak) as a function of the received power setting signal (ctrl_p) such that the value of the upper low load current threshold (Ipeak) decreases as the value of the received power setting signal (ctrl_p) decreases; and

[0220] The value of the lower low load current threshold (-Ipeak) is set as a function of the received power setting signal (ctrl_p) so that the value of the lower low load current threshold (Ipeak) decreases as the value of the received power setting signal (ctrl_p) decreases.

[0221] As the value of the ctrl_p signal decreases, the slope defined by Vcaph and Vcapl (for example, Figure 8 805 and 806) are shifted towards the x-axis. There is a moment when the locus representing the operation of the resonant converter intersects the slope defined by Vcaph, Vcapl, and thus the slope of Vcaph, Vcapl begins to affect the turn-off moment of the resonant converter. This means that the gain from ctrl-p to the power at the output of the resonant converter is not only determined by the gain curve ( Fig.15 ) and is limited by the slope. Therefore, the effective gain becomes larger. Therefore, Fig.16 The transfer from ctrl_p to Ipeak (dIpeak / dCtrol_p) is limited to a maximum value in order to maintain stable operation. (Delays in the system are also taken into account to give additional dynamics.) This effect can lead to problematic regions when approaching zero power.

[0222] For example Fig.16This problematic region of near zero load for the example of has been identified as starting at ctrl_p values ​​less than 0.1. Therefore, an additional medium load threshold (Ctrl_p_border) 1624 is used. The medium load threshold (Ctrl_p_border) 1624 is set so that the point at which the slope begins to affect the transfer of power from ctrl_p to the resonant converter output is at Fig.16 The medium load threshold (Ctrl_p_border) 1624 is set to a level between medium and low of Ctrl_p in order to maintain the desired stable behavior, as discussed above. The level of the medium load threshold (Ctrl_p_border) 1624 is set according to the resonant peak current occurring when Ctrl_p = Ctrl_p_border. When Ctrl_p decreases below the medium load threshold (Ctrl_p_border) 1624, Ipeak gradually decreases to a zero load current value (e.g., Fig.16 This may be referred to as a 'low' level at zero load (ie when ctrl_p=0).

[0223] In this way, the controller:

[0224] If the received power setting signal (ctrl_p) is zero, the value of the upper low load current threshold (Ipeak) is set to a zero load current value (eg, Fig.16 0.9A in 10A); and

[0225] If the received power setting signal (ctrl_p) is zero, the value of the lower load current threshold (-Ipeak) is set to a zero load current value (eg, Fig.16 -0.9A in the range of 1.

[0226] like Fig.16 As shown in , the relationship between Ipeak (on the vertical axis) and ctrl_p (on the horizontal axis) is different for: i) values ​​of ctrl_p between the high load threshold (Ctrl_p_border1) 1623 and the medium load threshold (Ctrl_p_border) 1624; and ii) values ​​of crtl_p less than the medium load threshold (Ctrl_p_border) 1624.

[0227] More specifically, if the received power setting signal (ctrl_p) is less than the high load threshold (Ctrl_p_border1) 1623 (eg, Fig.16 0.2 in), then the controller:

[0228] Set the value of the upper low load current threshold (Ipeak) to:

[0229] A first function of the received power setting signal (ctrl_p) is provided that the received power setting signal (ctrl_p) is greater than a medium load threshold (Ctrl_p_border) 1624 (eg, Fig.16 0.1 in ); and

[0230] a second function of the received power setting signal (ctrl_p), conditional upon the received power setting signal (ctrl_p) being less than a medium load threshold (Ctrl_p_border) 1624; and

[0231] Set the value of the lower low load current threshold (-Ipeak) to:

[0232] A first function of the received power setting signal (ctrl_p) is provided that the received power setting signal (ctrl_p) is greater than a medium load threshold (Ctrl_p_border) 1624 (eg, Fig.16 0.1 in ); and

[0233] A second function of the received power setting signal (ctrl_p), conditional on the received power setting signal (ctrl_p) being less than a medium load threshold (Ctrl_p_border) 1624 .

[0234] like Fig.16 As shown in , the first function and the second function are linear functions. The first function is steeper than the second function because the value of Ipeak of the second first function decreases more rapidly relative to ctrl_p than the second function. It should be understood that in other examples, nonlinear functions may be used.

[0235] As about Fig. 9 As explained, the effect will be to force the trajectory to follow a stable path according to the value of ΔV and the resonant current defining the slope added to ΔV in this case. To illustrate this functionality, Figures 17 to 22 Given in Fig.16 In each of these figures, the upper voltage threshold is shown as a dotted line and the lower voltage threshold is shown as a dashed line.

[0236] The values ​​used in this example are:

[0237] Gain1 = 1,

[0238] slopegain = 0.5 (= dvcap / dIresonant), and

[0239] Ipeak_ofs = -0.1

[0240] Fig.17The case of half load (ctrl_p=1) is shown. In the context of evaluating the performance of a resonant converter at low load, Fig.17 Half the load can be considered as a relatively high load.

[0241] In this case, this can be considered as a normal situation where ΔV = Vcaph - Vcapl = 2. For the resonant current at values ​​high = 4, -4, the area where slope compensation starts is still outside the range. Slope compensation is such that at Ires = 'high' (points A and B), Ires has no effect on the vcap level, so the Vcaph level = Vcaph, as defined by Ctrl-p. Ipeak_ofs = -0.1 changes this value slightly, which will be discussed later. In any case, Fig.17 The operating trajectory in is expected to intersect the vertical regions of the upper and lower voltage thresholds.

[0242] Fig.18 Shows Fig.16 The load has been reduced, which has caused ΔV to decrease to 0.4, thus producing a ratio of Fig.17 But for the rest, the situation is the same Fig.17 The same situation in the slope compensation is when Ires = 'high' (in Fig.17 Starting at points A and B marked in FIG.

[0243] Fig.19 The figure shows that Ipeak decreases based on the value of ctrl_p, but the sloped region of the upper and lower current thresholds does not affect the situation in the middle of the region of control of the resonant converter. Since ctrl_p is between Ctrl_p_border1 and Ctrl_p_border, the peak current at Vcaph level = Vcaph is between high and medium. The peak current is between Fig.19 In the example, it is 3, which is the point A and B where the slope starts.

[0244] Fig. 20 The situation at the edge of the sloped region of the upper and lower current thresholds taking control is shown. At this point, the resonant peak current at the applied ctrl_p value begins to cross the region where the slope occurs. This means that the trajectory is now also forced to the desired stable path by the slope compensation and the fact that outside the region with slope compensation ΔV=Vcaph-Vcapl is still greater than 0. This ensures that for smaller trajectories, the situation converges to a single desired trajectory.

[0245] Fig.21It is shown that the trajectory is further narrowed by the applied slope at lower Ires values, and the trajectory is stabilized by the slope extending to positive and negative Vcaph levels.

[0246] at last, Fig. 22 shows the zero load condition, where the resonant peak current is reduced from the 'low' value (referenced above) Fig.16 The zero load current value described is limited.

[0247] The impact of Ipeak_ofs

[0248] Advantageously, slope compensation is not used for moderate and larger power levels because it would produce a Vcap level that is defined based on the resonant current, and therefore would inaccurately define the power. It is advantageous to have a smooth transition to low load situations where it is desirable to effectively use slope compensation in the region where the decision to complete a half cycle is taken. Therefore, in some examples, the controller may not immediately implement the slope for low loads. For this reason, the examples disclosed herein may keep the region where no slope occurs always in effect, but move points A and B (such as Fig.17 ) slowly shifts outside the active region as a function of Ctrl_p. From the above discussion, it should be understood that point A and point B represent upper and lower low-load current thresholds, which define the points at which the upper and lower voltage thresholds transition between: i) independent of the measured current signal; and ii) dependent on the measured current signal.

[0249] The effect of Ipeak_ofs is to ensure that ΔV remains positive for the region where slope compensation is not in effect and also for zero load, in order to ensure that the trajectory always converges to a stable trajectory.

[0250] However, without additional measures, this would mean that ΔV becomes 0 at zero load. This is demonstrated in Figure 23-25 In which the power (Ctrl_p) is small and finally Fig.25 In each of these figures, the upper voltage threshold is shown as a dotted line, and the lower voltage threshold is shown as a dashed line.

[0251] according to Fig.25 and Figures 9 to 11Related to the previous explanation, it is clear that in this case there is an undefined trajectory for zero load. However, the Ipeak_ofs value should not be activated immediately as a function of ctrl_p, because this would also give a sudden change in the transfer from ctrl_p to power, and would therefore have the risk of irregular and unstable behavior. On the other hand, it may not be desirable to have Ipeak_ofs act on normal operation (i.e., operation where the trajectory does not intersect the slope area defined by Vcaph and Vacpl). This is because Ipeak_ofs gives an additional offset with respect to the Vcap level as a function of ctrl_p, and therefore produces a shift in the output power of the resonant converter. Therefore, Ipeak_ofs is reduced in a smooth manner, so that the gain effect of reducing Ipeak_ofs to zero is spread out over a relatively large area, thereby accepting some errors in the definition of power in this area.

[0252] To make a smooth transition, the Ipeak_ofs value in this example gradually increases from 0 to the desired value defined by the variable 'refscale' from 0 to 1 over the ctrl_p region from Ctrl_p_border down to 0. Optionally, an additional factor α may be defined so that the increase starts at αx Ctrl_p_border. This is done in Fig.26 is shown in and can be referred to as the offset current applied power threshold.

[0253] like Figure 17-22 It has been shown in that ΔV for larger ctrl_p values ​​has much greater influence than Ipeak_ofs, so that the transfer from ctrl_p to power is almost unaffected by the Ipeak_ofs value, as it becomes dominant only at very small power levels close to zero.

[0254] Optionally, a ctrl_p dependent offset can be defined and added to ΔV to compensate for errors in the transfer from ctrl_p to power due to Ipeak_ofs at low loads. This can also allow the power to be tuned to 0 with an offset and Ipeak_ofs to be as large as needed for systems with larger delays. (See also Fig.14 's previous description. )

[0255] Definition of the magnetization reduction during the soft-start and soft-stop portions of the burst-on interval

[0256] During the soft start and soft stop intervals, it may be advantageous to further reduce the magnetizing current, since a sudden change in the magnetizing current causes a sudden change in the force with which the core halves attract each other. This may be the main cause of the audible noise. This explains the audible noise, since no magnetizing current flows during the end of the burst, while the normal resonant current changes suddenly during the on-burst.

[0257] The second effect is that the windings attract each other, which also causes audible noise. Therefore, during soft start and soft stop, the goal is to make both changes smooth in order to eliminate high frequency components.

[0258] To reduce audible noise due to these two effects, separate functionality is included as part of the present disclosure.

[0259] 1. Reduced audible noise from the attraction between windings due to modulation of the resonant portion of the current when delivering output power.

[0260] 2. By additionally reducing the magnetizing current, the audible noise from the attractive force between the core halves due to the magnetizing portion of the current is reduced.

[0261] Point 1 can be achieved through the functionality described so far to achieve zero power by reducing ctrl_p to 0 during the soft start and soft stop intervals. Fig. 27 This is shown in the right portion where ctrl_p is at a fixed level during the burst-on interval and decreases to zero in a smooth manner during the soft-start and soft-stop portions of the burst-on interval.

[0262] Point 2 can be achieved by subtracting an additional term 'Imagn_red' from the 'low' value to further reduce Ipeakref below the 'low' value. Fig. 22 As explained, the Ipeak value 'low' occurring at ctrl_p = 0 makes the output power zero, and the trajectory followed is defined by the Ipeak value 'low', which is the peak value of the magnetizing current. Therefore, the magnetizing current can be further reduced by subtracting the additional term 'Imagn_red'. The result will be a converter still producing zero load, but with a higher switching frequency.

[0263] For a resonant tank with a separate resonant inductor, the attraction of the core halves occurs due to the magnetizing current and the resonant current, since in this case the separate inductor also stores energy due to the output current slowdown, whereas the resonant transformer does not store energy in this case. Therefore, soft start and soft stop can also be applied by modulating ctrl_p for a circuit including a separate inductor.

[0264] Fig.28A and 28B An embodiment of a controller for a resonant converter according to the invention is shown.

[0265] An embodiment includes a resonant power converter using capacitor voltage control, the capacitor voltage control comprising:

[0266] ●Cycle-by-Cycle Comparator

[0267] ●Logic and drivers

[0268] ●Symmetrical loop

[0269] A capacitive voltage divider to divide the high voltage signal (vcr) across the resonant capacitor into a low voltage signal (Vcap_recon) as input for the symmetry regulator and the cycle-by-cycle comparator

[0270] An optional delay correction block that adds k1 times the sensed resonant current (Iprim) to the Vcap_recon signal to define a delay corrected version Vcap_recon_corr

[0271] A current sensing block that senses the current in the resonant capacitor via a capacitive shunt (C1, Rsense) and a differential amplifier that forms a noise-free version of the voltage across the sense resistor (Iprim)

[0272] The secondary rectifier stage, which includes the rectifier, output filter and output capacitor where the output voltage Vout appears

[0273] A secondary-side regulator that generates an error signal based on the output voltage and the desired output voltage

[0274] ●Power supply isolation between the secondary side of the error signal and the primary side of the error signal

[0275] ●Power definition block, which generates

[0276] ○Internal main control variable (Ctrl_p)

[0277] ○ 2 signals Vcrh_in and Vcrl_in, which are the unconditioned input signals of the cycle-by-cycle comparator

[0278] ●Main block representing the core of the present disclosure

[0279] exist Figures 28A to 30 The main blocks are described in more detail in , based on the implementation in the Simulink model. The same functionality is also modeled in Mathcad, as shown in Figures 32 to 35 Described in more detail in .

[0280] Fig.29 A Simulink implementation of two blocks for defining the Vcrh_out and Vcrl_out signals according to:

[0281] ●Sensed resonant tank current (Iprim_sensed2a)

[0282] ●Signals related to the required output power (Vcrh, Vcrl, Ctrl_p)

[0283] ● Settings for correct operation of the no-load method ((Ctrl_p_border, Ctrl_p_border1, low, medium, high, Imagn_red, α) (α is set inside the first sub-block, Iofs_max = scaling factor between refscale and Ipeak_ofs)

[0284] Fig.30 Includes more details of the first sub-block.

[0285] The output 'refscale' generates the signal at the upper diagram based on a lookup table with predefined points (Ctrl_p, refscale), where the points of the lookup table are defined using a mux structure (vertical thick line with arrows) and a table definition block with inputs:

[0286] ●u1 (continuous variable input Ctrl_p)

[0287] bp1 (horizontal axis coordinate of the lookup table point)

[0288] ●T((vertical axis coordinate of the lookup table point))

[0289] Regarding the refscale signal, for refscale=F(Ctrl_p), there are 3 points for the lookup table (see also Fig.26 )

[0290] 1.(Ctrl_p=0,refscale=1):

[0291] 2.(Ctrl_p=Ctrl_p_border,refscale=0)

[0292] 3.Ctrl_p=2,refscale=0)

[0293] 1. For the Ipeakref output, there is a similar way to use 4 input mux definitions (see also Fig.16 )’s 4 search points:

[0294] 2.(Ctrl_p=0,Ipeakref=low):

[0295] 3.(Ctrl_p=Ctrl_p_border,Ipeakref=med)

[0296] 4.(Ctrl_p=Ctrl_p_border1,Ipeakref=high)

[0297] 5.Ctrl_p=2,Ipeakref=high)

[0298] The signal 'Imagn_red' is a separate input that allows further reduction of the magnetizing current under zero load conditions, as described in relation to Fig. 27 Explained.

[0299] Fig.31 Includes more details of the second sub-block. The two output signals Vcaph_Icorr and Vcapl_Icorr are the result of 2 max functions, where the output is the maximum of both the input and the linear addition.

[0300] Implement the following functions in this block:

[0301] 1. Vcaph_Icorr=F(Ipeak_ofs,vcrh,iprimsensed2a,Ipeakref) is as follows:

[0302] a.Vcaph_Icorr=vcrh-max(Ipeak_ofs,(-iprimsensed2a-Ipeakref)*slopegain)

[0303] 2.Vcapl_Icorr=F(Ipeak_ofs,vcrl,iprimsensed2a,Ipeakref)

[0304] a.Vcapl_Icorr=vcrl-max(Ipeak_ofs,(iprimsensed2a-Ipeakref)*slopegain)

[0305] Figure 32-35 A Mathcad implementation of the two blocks is shown.

[0306] Fig.32 Shows the lookup table structure for generating the Ipeakref signal based on 2 4d vectors and the Iinterp command and the resulting plot.

[0307] Fig.33 The lookup table structure used to generate the refscale function in Mathcad and the resulting plot is shown.

[0308] Fig.34 The definition of the VCRH and VCRL signals as a function of ctrl_p is shown in FIG. Fig. 27 The Power Definition block passes the input signal for the Master block.

[0309] Fig.35The calculation and graph results in Mathcad of the overall main function block that generates the Vcrh_out and Vcrl_out signals (horizontal axis) from the sensed primary current (vertical axis) are shown. The reason the input signal is plotted on the vertical axis and the output signal is plotted on the horizontal axis is because it is part of the state plane representation where the resonant current is plotted vertically and the resonant capacitor voltage is plotted horizontally.

[0310] The function Vcaph_Icorr(Iprim) is a redefinition of the full function Vcaph_Icorr=f (all relevant variables and subfunctions). (See Fig.35 All dependent variables on the right side of . )

[0311] This is done for practical reasons when plotting in Mathcad, to prevent the text at the horizontal axis lines from not matching the required width of the plot, which would make the plot unnecessarily wide and less clear to the reader what is being shown.

[0312] In an embodiment, the features described in relation to the above 'definition of the magnetization reduction amount during the soft start and soft stop functions of the burst on interval' include soft start and soft stop functions. Fig. 27 .

[0313] The examples described herein may be applied in high power mode or low power mode to limit the power in the lower region to zero power.

[0314] For light loads and no loads, a dedicated low power mode is available.

[0315] In low power mode (LP mode), a complete switching cycle can be divided into parts called 'energy conversion cycles' (EC cycles), where power is delivered to the load and is identical to the normal high power mode (HP mode) switching cycle.

[0316] An interval of 2 EC cycles (or optionally more than 2) can be added to be able to put the converter in a quiescent state so that no losses occur. Fig.36A and 36B In the quiescent state, the quiescent state is called the 'park interval'. In the park interval, the resonant current is zero and the resonant tank is disconnected from the half-bridge node since both switches are off. The states 'dump ready' and 'dump' pause the resonant tank in a state such that when the 'resume' interval is activated after the park interval, the resonant tank returns to the same symmetrical trajectory as in high power mode.

[0317] Since the EC cycle is substantially the same as the EC cycle in the normal HP mode, the examples described herein are also applicable to the EC cycle in the LP mode. In the LP mode, the recovery state uses the same VCRH level as the VCRH level during the EC cycle. Fig.37 This is shown using a state plane representation.

[0318] Fig.37 A low power mode complete switching cycle in a state plane representation is shown.

[0319] As from Fig.37 It can be seen that the vcrh level after the recovery interval is the same as the vcrh level at the end of the HSS half cycle during the EC interval. Therefore, under the effect of the present invention, the vcrh level remains the same for both times.

[0320] In order to make the two EC half cycles of the LP mode symmetrical, there is only one value for the Vcrdump level that gives this symmetry. Therefore, an additional feature of the present disclosure is to define the function Vcrdump(vcrh, vcrl) to obtain the best symmetry (in fact, this is Fig.28A and 28B VCRH_out and VCRL_out in the output).

[0321] Compared to the known prior art, the examples disclosed herein can provide stable zero-load operation while switching a resonant converter with a high operating frequency controlled by a capacitor voltage control method. Such converters are thus forced to use burst mode or frequency control. For certain applications with two output voltages, burst mode is possible. By frequency control, it is directly possible to use a dedicated low-power mode that allows greater partial load efficiency. Therefore, the examples disclosed herein can use a dedicated low-power mode to achieve zero load and effective partial load operation.

[0322] The examples disclosed herein use slope compensation, but within a specific resonant current range. The slope of the Vcap level and the point at which the slope occurs can be defined in a specific way to force the converter into a zero load condition or a condition where the magnetizing current is reduced in a way that reduces the likelihood of instability, while still being compatible with a certain amount of delay in the capacitor voltage control portion.

[0323] The examples disclosed herein may replace the slope with a fixed Vcap level in a specific manner to force the resonant tank to follow

[0324] ●Stable low load or

[0325] Zero load track or

[0326] ● Trajectory with reduced magnetizing current required according to soft start and soft stop.

[0327] The examples disclosed herein are applicable to both high frequency resonant converters and low frequency resonant converters where better definition of soft start and soft stop during burst mode is needed in order to further reduce audible noise.

[0328] Examples disclosed herein relate to a method of enabling no-load operation of a high frequency resonant converter, wherein an internal delay may otherwise prevent the possibility of no-load occurring during switching.

[0329] Features of the present disclosure include:

[0330] 1. A resonant converter controller including a capacitor voltage control method for limiting power

[0331] 2. Feature 1 includes a method for defining the effective vcap level in a stable manner to perform the following

[0332] ○ No-load operation is possible during switching, and

[0333] ○ Further reduction of magnetizing current makes it possible to realize soft start and soft stop functions

[0334] 3. The method of step 2 has one or more of the following features:

[0335] ○ The slope compensation signal is added to the Vcap level derived from the primary current (producing a vcap level of Vcap0 + a xIresonant) to force the switching cycle to converge to the desired stable trajectory as desired ( Figure 6 and 7 )

[0336] ○ Apply slope only during a well-defined portion of the Vcr window to prevent multiple trajectories (limiting convergence to only 1 frequency) ( Figure 8 )

[0337] ○ Reduce Vcap0 to keep it positive outside the slope region to force the trajectory to the desired and stable magnetizing current ( Figure 9-14 and Figure 23-25 )

[0338] o The Ipeak current is defined by a predefined curve as a function of the main power control variable.

[0339] ( Figure 16-22 and 26)

[0340] 4. Method 3 is applied in the high power mode or the low power mode to limit the power in the lower region to zero power.

[0341] ○ In LP mode, the recovery state uses the same VCRH level as the VCRH level during the EC cycle ( Figures 36A-37 ).

[0342] ○ Define the function Vcrdump(vcrh,vcrl) to obtain the best symmetry ( Figures 36A-37 )

[0343] 5. Apply method 3 to define the magnetization reduction during the soft start and soft stop portions of the burst on interval ( Fig. 27 )

[0344] 6. Apply Method 3 to limit the converted power during the soft start and soft stop portions of the burst on interval ( Fig. 27 )

[0345] Fig.38 An example embodiment of a method of operating a resonant converter according to the present disclosure is shown.The resonant converter may be any of the resonant converters disclosed herein.

[0346] Methods include:

[0347] At step 3880, receiving a measured current signal representative of a current flowing in the resonant tank;

[0348] At step 3881, receiving a measured voltage signal representing a voltage at a predetermined point in the resonant tank; and

[0349] At step 3882, a power setting signal is received defining a requested power level for the load.

[0350] At step 3883, the method includes:

[0351] If the measured current signal is greater than the upper low load current threshold, setting an upper voltage threshold based on the measured current signal;

[0352] if the measured current signal is not greater than the upper low load current threshold, setting the upper voltage threshold based on the power setting signal but independent of the measured current signal;

[0353] If the measured current signal is less than a lower low load current threshold, setting a lower voltage threshold based on the measured current signal; and

[0354] If the measured current signal is not less than the lower low load current threshold, a lower voltage threshold is set based on the power setting signal but independent of the measured current signal.

[0355] At step 3884, in response to the measured voltage signal exceeding the upper voltage threshold, the method includes opening the first switch and closing the second switch.

[0356] At step 3885 , in response to the measured voltage signal falling below a lower voltage threshold, the method includes opening the second switch and closing the first switch.

[0357] Unless a specific order is explicitly stated, the instructions and / or flow chart steps in the above figures can be executed in any order. Moreover, those skilled in the art will recognize that although an example instruction set / method has been discussed, the materials in this specification can also be combined in various ways to produce other examples, and should be understood within the context provided in this specific embodiment.

[0358] In some example embodiments, the instruction set / method steps described above are implemented as functions and software instructions embodied as an executable instruction set, which is implemented on a computer or a machine programmed and controlled with the executable instructions. Such instructions are loaded for execution on a processor (e.g., one or more CPUs). The term processor includes a microprocessor, a microcontroller, a processor module or subsystem (including one or more microprocessors or microcontrollers), or other control or computing devices. A processor may refer to a single component or multiple components.

[0359] In other examples, the instruction sets / methods shown herein and the data and instructions associated therewith are stored in corresponding storage devices, which are implemented as one or more non-transitory machine or computer readable or computer usable storage media. Such computer readable or computer usable storage media are considered to be part of an article (or product). An article or product may refer to any manufactured single component or multiple components. Non-transitory machine or computer usable media as defined herein do not include signals, but such media may be able to receive and process information from signals and / or other transient media.

[0360] Example embodiments of the materials discussed in this specification may be implemented in whole or in part via a network, computer or data-based device and / or service. These may include a cloud, the Internet, an intranet, a mobile device, a desktop computer, a processor, a lookup table, a microcontroller, a consumer device, an infrastructure, or other enabling devices and services. As may be used herein and in the claims, the following non-exclusive definitions are provided.

[0361] In one example, one or more instructions or steps discussed herein are automated. The term automation or automatic (and similar variations thereof) means the use of computers and / or mechanical / electrical devices to control the operation of equipment, systems and / or processes without the need for human intervention, observation, effort and / or decision making.

[0362] It should be understood that any components that are said to be coupled may be coupled or connected directly or indirectly. In the case of indirect coupling, additional components may be placed between the two components that are said to be coupled.

[0363] In this specification, example embodiments have been presented based on a selected set of details. However, one of ordinary skill in the art will appreciate that many other example embodiments including different selected sets of these details may be practiced. It is intended that the appended claims cover all possible example embodiments.

Claims

1. A controller for a resonant converter, characterized in that: The resonant converter is used to supply electric energy from a supply source to a load, and the resonant converter includes: a first switch and a second switch, the first switch and the second switch being connected in series with each other between the supply source and a reference terminal, wherein when the first switch is closed and the second switch is open, the resonant converter has a high-side switching half-cycle, and wherein when the first switch is open and the second switch is closed, the resonant converter has a low-side switching half-cycle; and a resonant tank electrically connected to the first switch and the second switch, wherein the resonant tank includes a resonant capacitor; The controller is configured to: receiving a measured current signal representative of a current flowing in the resonant tank; receiving a measured voltage signal representative of a voltage at a predetermined point in the resonant tank; receiving a power setting signal defining a requested power level for the load; if the measured current signal is greater than an upper low load current threshold, setting an upper voltage threshold based on the measured current signal; if the measured current signal is not greater than the upper low-load current threshold, setting the upper voltage threshold based on the power setting signal but independent of the measured current signal; If the measured current signal is less than a lower low load current threshold, setting a lower voltage threshold based on the measured current signal; if the measured current signal is not less than the lower low load current threshold, setting the lower voltage threshold based on the power setting signal but independent of the measured current signal; In response to the measured voltage signal exceeding the upper voltage threshold, opening the first switch and closing the second switch; and In response to the measured voltage signal falling below the lower voltage threshold, the second switch is opened and the first switch is closed.

2. The controller according to claim 1, characterized in that: The controller is further configured to: setting a value of the upper low load current threshold based on a received power setting signal; and A value of the lower low load current threshold is set based on the received power setting signal.

3. The controller according to claim 2, characterized in that: The controller is further configured to, if the received power setting signal is greater than a high load threshold, then: setting the value of the upper low load current threshold to a high load constant value; and The value of the lower low load current threshold is set to a high load constant value.

4. The controller according to claim 3, characterized in that: The controller is further configured to, if the received power setting signal is less than the high load threshold, then: setting the value of the upper low load current threshold to a value less than the high load constant value; as well as The value of the lower low-load current threshold is set to a value smaller than the high-load constant value.

5. The controller according to claim 3 or claim 4, characterized in that: The controller is further configured to, if the received power setting signal is less than the high load threshold, then: setting the value of the upper low load current threshold as a function of the received power setting signal; and The value of the lower low load current threshold is set as a function of the received power setting signal.

6. The controller according to claim 5, characterized in that: The controller is configured to: If the received power setting signal is zero, setting the value of the upper low load current threshold to a zero load current value; and If the received power setting signal is zero, then the value of the lower low load current threshold is set to a zero load current value.

7. The controller according to claim 5 or claim 6, characterized in that: The controller is further configured to, if the received power setting signal is less than the high load threshold, then: The value of the upper low load current threshold is set to: a first function of the received power setting signal, conditional upon the received power setting signal being greater than a medium load threshold; and a second function of the received power setting signal, conditional upon the received power setting signal being less than the medium load threshold; as well as The value of the lower low load current threshold is set to: said first function of said received power setting signal, conditional upon said received power setting signal being greater than said medium load threshold; and The second function of the received power setting signal is conditioned on the received power setting signal being less than the medium load threshold.

8. The controller according to claim 7, characterized in that: The first function and the second function are linear functions, and wherein the first function is steeper than the second function.

9. A controller according to any preceding claim, characterised in that The controller is further configured to, if the power setting signal is less than an offset current applied power threshold, then: adjusting the measured current signal by adding an offset current signal to the measured current signal, and An adjusted measured current signal is used instead of the measured current signal.

10. A method of operating a resonant converter, characterized in that The resonant converter is used to supply electric energy from a supply source to a load, and the resonant converter includes: a first switch and a second switch, the first switch and the second switch being connected in series with each other between the supply source and a reference terminal, wherein when the first switch is closed and the second switch is open, the resonant converter has a high-side switching half-cycle, and wherein when the first switch is open and the second switch is closed, the resonant converter has a low-side switching half-cycle; and a resonant tank electrically connected to the first switch and the second switch, wherein the resonant tank includes a resonant capacitor; The method comprises: receiving a measured current signal representative of a current flowing in the resonant tank; receiving a measured voltage signal representative of a voltage at a predetermined point in the resonant tank; receiving a power setting signal defining a requested power level for the load; if the measured current signal is greater than an upper low load current threshold, setting an upper voltage threshold based on the measured current signal; if the measured current signal is not greater than the upper low-load current threshold, setting the upper voltage threshold based on the power setting signal but independent of the measured current signal; If the measured current signal is less than a lower low load current threshold, setting a lower voltage threshold based on the measured current signal; if the measured current signal is not less than the lower low load current threshold, setting the lower voltage threshold based on the power setting signal but independent of the measured current signal; In response to the measured voltage signal exceeding the upper voltage threshold, opening the first switch and closing the second switch; and In response to the measured voltage signal falling below the lower voltage threshold, the second switch is opened and the first switch is closed.

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