Control circuit and control method of vehicle charger

Abstract

According to the control circuit and the control method of the vehicle charger disclosed by the invention, the rectifier bridge of the bridge PFC is removed, so that the number of switching devices is reduced, the topological cost is reduced, and the loss is reduced; power factor correction is realized through average current control, and stable 400V direct current is output for post-stage power supply after power grid side 220V alternating current is rectified; the rear-stage resonant converter is highly symmetrical in structure and has the advantages of high power density, strong energy interactivity, input and output isolation and easy realization of soft switching; a hybrid modulation strategy is adopted for the resonant converter, frequency modulation and extended phase shift segmented modulation are adopted under the normal load condition, the problem that the frequency modulation range is narrow during voltage reduction is solved, intermittent Bang-Bang charge control is switched under the light load condition, and the problems that the output voltage drift is high and the efficiency is reduced during light load are solved; therefore, the charging efficiency of the vehicle-mounted charger is improved.

Description

Technical Field

[0001] The present invention belongs to the technical field of charging circuit control, and particularly relates to a control circuit of a vehicle-mounted charger, and also relates to a control method of a vehicle-mounted charger. Background Art

[0002] With the rapid development of China's automobile industry, the vehicle ownership has increased year by year. Electric vehicles, which use electric energy as fuel, have the advantages of energy conservation, environmental protection, and can help alleviate the shortage of energy resources and achieve sustainable development, and have become a research hotspot in recent years. For electric vehicles, the issues that users are most concerned about are the driving range, charging speed, and charging convenience. The driving range of electric vehicles is currently mainly limited by the development level of battery technology, while the charging speed and convenience depend on the improvement of charging technology.

[0003] The charging methods of electric vehicles are mainly divided into two types: on-board chargers and off-board chargers. The off-board charger can directly convert alternating current into direct current, and then charge the battery of the electric vehicle. It has a large output power and a fast charging speed, but the fixed location of the charging pile and sometimes mismatched power bring many inconveniences to users. The on-board charger is installed inside the electric vehicle. It is directly connected to the industrial frequency alternating current, converts the alternating current into direct current inside the vehicle to charge the battery, has a lower device cost, and also improves the flexibility of the charging method. Users can choose to directly use single-phase alternating current to charge the electric vehicle at a suitable location, and can also use the low electricity price during the night off-peak period to charge, reducing the charging cost. Summary of the Invention

[0004] The first object of the present invention is to provide a control circuit of a vehicle-mounted charger.

[0005] The second object of the present invention is to provide a control method of a vehicle-mounted charger, which is beneficial to improving the charging efficiency of the vehicle-mounted charger by using appropriate modulation strategies for the front and rear stages.

[0006] The first technical solution adopted by the present invention is a control circuit of a vehicle-mounted charger, including a bridge-less Boost PFC circuit, a capacitor C i , a resonant converter, and a capacitor C o ; The input of the bridge-less Boost PFC circuit is connected to the power grid, and the capacitor C o is connected to the load; The bridge-less Boost PFC circuit includes an inductor L. The inductor L is successively connected in series with a diode D1 and a diode D2. The power grid, the inductor L, the diode D1, and the diode D2 form a closed circuit a, and a switch circuit is connected in parallel to the closed circuit a; the switch circuit includes a switching transistor Q1 and a switching transistor Q2 connected in series; the capacitor Ci , capacitor C i One end is connected to the closed circuit a, and the capacitor C i The other end is connected to the switch circuit a; The resonant converter includes a primary full-bridge circuit, a high-frequency transformer T, and a secondary full-bridge circuit. The primary full-bridge circuit is connected to the secondary full-bridge circuit through the high-frequency transformer T; The primary full-bridge circuit and the capacitor C i are connected in parallel, and the secondary full-bridge circuit and the capacitor C o are connected in parallel.

[0007] The characteristics of the first technical solution of the present invention also lie in: The primary full-bridge circuit includes switch tubes S1, S2, S3, and S4. Switch tubes S1 and S3 are connected to one end of the capacitor C i , and switch tubes S2 and S4 are connected to the other end of the capacitor C i . Switch tubes S1 and S2 are connected in series, and switch tubes S3 and S4 are connected in series; The primary full-bridge circuit further includes a resonant inductor L connected to the primary side of the high-frequency transformer T r1 , an exciting inductor L m and a resonant capacitor C r1 . One end of the resonant inductor L r1 is connected to the connection point of switch tubes S1 and S2. The other end of the resonant inductor L r1 is connected to one end of the exciting inductor L m . One end of the resonant capacitor C r1 is connected to the connection point of switch tubes S3 and S4. The other end of the resonant capacitor C r1 is connected to the other end of the exciting inductor L m .

[0008] The secondary full-bridge circuit includes switch tubes S5, S6, S7, and S8. Switch tubes S5 and S7 are connected to one end of the capacitor C o , and switch tubes S6 and S8 are connected to the other end of the capacitor C o . Switch tubes S5 and S6 are connected in series, and switch tubes S7 and S8 are connected in series; The secondary full-bridge circuit further includes a resonant inductor L connected to the secondary side of the high-frequency transformer T r2 and a resonant capacitor C r2 . The resonant inductor L r2 is connected to the connection point of switch tubes S7 and S8. The resonant capacitor C r2 is connected to the connection point of switch tubes S5 and S6.

[0009] The second technical solution adopted by the present invention is a control method for a vehicle-mounted charger, which uses the control circuit of the vehicle-mounted charger described above and is specifically implemented according to the following steps: S1: The bridgeless Boost PFC circuit adopts average current control, takes the average value of the actual input inductor L current as the sampled current, and uses a control method with a current inner loop and a voltage outer loop to adjust the average value of the input current, so as to control the average value of the input current to follow the input grid voltage; S2: The resonant converter adopts hybrid modulation. Under normal load conditions, frequency modulation + extended phase-shifted segmented modulation is adopted; under light load conditions, intermittent Bang-Bang charge control is switched.

[0010] The characteristics of the second technical solution of the present invention also lie in: Specifically, S1 is as follows: Collect the input current, input voltage and output voltage of the bridgeless Boost PFC circuit; transmit the output voltage to the voltage loop compensator, and the feedback voltage signal output by the voltage loop compensator and the input voltage signal are operated by a multiplier to obtain a current reference signal. The current reference signal and the input current signal enter the current loop compensator, and the current loop compensator outputs a PWM signal to control the on and off times of the switching tube.

[0011] Specifically, S2 is as follows: Sample the output voltage of the resonant converter and the output current Calculate the output power through a multiplier , and compare the result with the reference value , where is the rated power; the normal load condition is: , ; the light load condition is: ; .

[0012] When under normal load and the resonant converter is boosting forward, pulse frequency modulation is adopted: The error obtained by subtracting the current output voltage from the reference voltage is fed into a proportional-integral regulator to obtain the required switching frequency , and then is converted into a modulation signal with a corresponding frequency through a voltage-controlled oscillator. Finally, the modulation signal is compared with a carrier signal through a comparator to obtain the driving signal of the primary switching tube, and the duty cycle remains unchanged at 50%. The secondary switching tubes are all turned off, and uncontrolled rectification is performed by the anti-parallel diodes; The switching frequency is as shown in Equation (1): (1) In formula (1), is the initial value of set to (2 - 3) . is the proportional coefficient of the PI regulator, is the integral coefficient of the PI regulator, is the and error when boosting the voltage.

[0013] Under normal load, when the resonant converter steps down in the forward direction, EPS modulation is adopted: The error obtained by subtracting the current from is sent to the PI regulator to obtain the phase shift angle , and then the phase shift angle is converted into the delay time t. Pulse signals with a fixed duty cycle are delayed by different degrees to obtain the drive signals of the primary and secondary switching tubes respectively, thereby controlling the turn-on and turn-off of the primary and secondary switching tubes of the resonant converter. The duty cycle is 50% and the switching frequency is always equal to the resonant frequency . The phase shift angle within the primary bridge and the phase shift angle between the primary and secondary bridges satisfy , ensuring the lowest switching loss of the resonant converter; The phase shift angle is represented by formulas (2) and (3): (2) (3) In formulas (2) and (3), is the phase shift angle, is the initial value of is the proportional coefficient of the PI regulator, is the integral coefficient of the PI regulator, is the and error when stepping down the voltage.

[0014] The delay time is represented by formula (4): (4) In formula (4), is the delay time, is the phase shift angle, is the resonant frequency.

[0015] Under light load, first stop the switching tubes of the resonant converter, and at the same time detect the output voltage . When When it drops to the minimum value, it enters the intermittent mode, and the switching transistor and conduct. The voltage of the primary-side resonant capacitor is compared with the reference value by using a hysteresis comparator. When , turn off and , and turn on and . When , turn off and , until drops to the minimum value again and cycle this process; The calculation method of the reference value is as follows: The net input charge of the resonant converter power supply side per half switching cycle can be calculated by Equation (5): (5) In Equation (5), is the primary-side resonant capacitor, , , are the voltages of the primary-side resonant capacitor , , at the moments of respectively; When operating stably, , substituting it into Equation (5) to get the input charge in one switching cycle as: (6) In Equation (6), is the primary-side resonant capacitor, is the voltage of the primary-side resonant capacitor at the moment of ; After compensating for the charge exchange occurring during the dead time, the input charge of the resonant converter per switching cycle is: (7) In Equation (7), is the voltage value of the resonant capacitor when the switch is turned off. When , and are known, the charge output by the power supply can be obtained from the voltage of the resonant capacitor . The input charge reflects the input power. When the resonant converter operates without load, the no-load input charge is obtained as: (8) In Equation (8), is the primary-side resonant capacitor, is the switch when it is turned off, the voltage value of the resonant capacitor ; When the input power under no-load condition is all output to the filter capacitor, the change in the output voltage is: (9) In Equation (9), is the primary-side resonant capacitor, is the switch when it is turned off, the voltage value of the resonant capacitor , is the output capacitor; When the resonant converter operates under no-load condition, the voltage fluctuation is the largest, denoted as , so that ΔV is less than , from which we can obtain: (10) In Equation (10), is the output capacitor, is the primary-side resonant capacitor, is the maximum voltage fluctuation when the resonant converter operates under no-load condition; Find the load value when the output efficiency is the highest, and obtain the capacitor voltage at this time as , then the reference value is: (11) In Equation (11), is the switch when it is turned off, the maximum voltage of the resonant capacitor , is the capacitor voltage value when the output efficiency is the highest.

[0016] The beneficial effects of the present invention are as follows: The bridgeless Boost PFC circuit of the present invention removes the rectifier bridge of the bridged PFC, reduces the number of switching devices, lowers the topology cost, and reduces losses; realizes power factor correction through average current control, rectifies the 220V AC power on the grid side and outputs stable 400V DC power to supply the subsequent stage; the subsequent resonant converter is highly symmetric in structure, has the advantages of high power density, strong energy interaction, input-output isolation, and easy implementation of soft switching; adopts a hybrid modulation strategy for the resonant converter, uses frequency modulation + extended phase-shift segmented modulation under normal load conditions to solve the problem of narrow frequency modulation range during step-down, and switches to intermittent Bang-Bang charge control under light load conditions to solve the problems of high output voltage drift and efficiency drop under light load; thus improving the charging efficiency of the on-vehicle charger. Brief Description of the Drawings

[0017] Figure 1 is a schematic diagram of the topological structure of the control circuit of the vehicle-mounted charger of the present invention; Figure 2 is a control block diagram of the bridge-less Boost PFC circuit in the control circuit of the vehicle-mounted charger of the present invention; Figure 3 is a waveform diagram of the input voltage and current of the bridge-less Boost PFC circuit in the control circuit of the vehicle-mounted charger of the present invention after average current control; Figure 4 is a waveform diagram of the total harmonic distortion rate of the input current in the control circuit of the vehicle-mounted charger of the present invention; Figure 5 is a complete block diagram of the resonant converter in the control circuit of the vehicle-mounted charger of the present invention adopting a hybrid modulation strategy. Detailed Embodiments

[0018] Next, the technical solutions in the embodiments of the present application will be clearly and completely described in conjunction with the accompanying drawings in the embodiments of the present application. Obviously, the described embodiments are only a part of the embodiments of the present application, rather than all the embodiments. Based on the embodiments in the present application, all other embodiments obtained by those of ordinary skill in the art without creative efforts shall fall within the protection scope of the present application.

[0019] Embodiment 1 As Figure 1 shown, the control circuit of the vehicle-mounted charger disclosed in this embodiment includes a bridge-less Boost PFC circuit, a capacitor C i , a resonant converter, and a capacitor C o connected in sequence; The input of the bridge-less Boost PFC circuit is connected to the power grid, and the capacitor C o is connected to the load; The bridge-less Boost PFC circuit includes an inductor L. The inductor L is sequentially connected in series with a diode D1 and a diode D2. The power grid, the inductor L, the diode D1, and the diode D2 form a closed circuit a, and a switching circuit is connected in parallel to the closed circuit a; the switching circuit includes a switching transistor Q1 and a switching transistor Q2 connected in series; the capacitor C i , the capacitor C i One end is connected to the closed circuit a, and the other end of the capacitor C i is connected to the switching circuit a; The resonant converter includes a primary full-bridge circuit, a high-frequency transformer T, and a secondary full-bridge circuit. The primary full-bridge circuit is connected to the secondary full-bridge circuit through the high-frequency transformer T; The primary full-bridge circuit is connected in parallel with the capacitor C i , and the secondary full-bridge circuit is connected in parallel with the capacitor C o .

[0020] In this embodiment, the 220V AC power grid can be rectified into 400V DC power by the front-stage bridge-less Boost PFC circuit, and then the resonant converter is used to complete step-up and step-down output to supply 300V - 500V to the power battery (load) of the electric vehicle. Among them, the inductor L in the input part is used for energy storage and current smoothing to reduce the input current ripple; the Boost conversion part mainly has two switching tubes , to control the energy transfer from the inductor to the output capacitor and regulate the output voltage; two diodes , to prevent current backflow; the output capacitor stores energy and stabilizes the output voltage to reduce ripple; the overall function of rectification and power factor correction is completed, and a stable 400V voltage is output to supply power to the subsequent stage. When the circuit operates in the positive half-cycle of the input voltage, is always on, controlling to turn on and off; when operating in the negative half-cycle, is always on, controlling to turn on and off.

[0021] Embodiment 2 On the basis of Embodiment 1, the primary full-bridge circuit includes switching tubes S1, S2, S3, and S4. Switching tubes S1 and S3 are connected to one end of capacitor C i , switching tubes S2 and S4 are connected to the other end of capacitor C i , switching tubes S1 and S2 are in series, and switching tubes S3 and S4 are in series; The primary full-bridge circuit also includes a resonant inductor L r1 , an exciting inductor L m and a resonant capacitor C r1 . One end of the resonant inductor L r1 is connected to the connection point of switching tubes S1 and S2, and the other end of the resonant inductor L r1 is connected to one end of the exciting inductor L m . One end of the resonant capacitor C r1 is connected to the connection point of switching tubes S3 and S4, and the other end of the resonant capacitor C r1 is connected to the other end of the exciting inductor L m .

[0022] Embodiment 3 On the basis of Embodiment 2, the secondary full-bridge circuit includes switching tubes S5, S6, S7, and S8. Switching tubes S5 and S7 are connected to one end of capacitor C o , switching tubes S6 and S8 are connected to the other end of capacitor C oThe other ends are connected. The switching tube S5 and the switching tube S6 are connected in series, and the switching tube S7 and the switching tube S8 are connected in series; The secondary full-bridge circuit further includes a resonant inductor L connected to the secondary side of the high-frequency transformer T r2 and a resonant capacitor C r2 , the resonant inductor L r2 is connected to the connection point of the switching tube S7 and the switching tube S8, and the resonant capacitor C r2 is connected to the connection point of the switching tube S5 and the switching tube S6.

[0023] In Embodiment 2 and Embodiment 3, the filter capacitor and , are used to filter high-frequency noise and stabilize the voltage; D1~D8 are the anti-parallel diodes of the SiC MOSFET switching tubes S1~S8 respectively; C S1 ~C S8 are the parasitic capacitors of the SiC MOSFET switching tubes S1~S8 respectively; T is a high-frequency transformer with a turns ratio of n:1; the resonant cavity includes two resonant inductors and two resonant capacitors as well as an exciting inductor ; overall, the input 400V DC is stepped up and down to 300~500V to supply power to the load, which can meet the charging requirements of most mainstream electric vehicles on the current market.

[0024] Embodiment 4 As Figure 2 shown, the control method of the vehicle charger disclosed in this embodiment uses the control circuit of the vehicle charger disclosed in any one of Embodiments 1-3, and is specifically implemented according to the following steps: S1: The bridgeless Boost PFC circuit adopts average current control, takes the average value of the actual input inductor L current as the sampling current, and uses the control method of current inner loop and voltage outer loop to adjust the average value of the input current, and controls the average value of the input current to follow the input grid voltage; S2: The resonant converter adopts hybrid modulation. Under normal load conditions, frequency modulation + extended phase-shifted segmented modulation is adopted; under light load conditions, intermittent Bang-Bang charge control is switched.

[0025] In this embodiment, after consulting the data and repeatedly simulating and verifying, using 20% full load as the sign of light load to switch the modulation strategy has the best effect. Therefore, when the load is higher than 20% full load, the strategy of frequency modulation + extended phase-shifted segmented modulation is adopted to solve the problem of narrow frequency modulation range when the converter steps down in the forward direction; when the load is lower than 20% full load, that is, under light load conditions, the intermittent Bang-Bang charge control strategy is switched to solve the problems of output voltage drift and efficiency decline under light load.

[0026] Embodiment 5 Based on Embodiment 4, S1 is specifically as follows: Collect the input current, input voltage, and output voltage of the bridgeless Boost PFC circuit; transmit the output voltage to the voltage loop compensator. The feedback voltage signal output by the voltage loop compensator and the input voltage signal are operated by a multiplier to obtain a current reference signal. The current reference signal and the input current signal enter the current loop compensator, and the current loop compensator outputs a PWM signal to control the on and off times of the switching transistor.

[0027] In this embodiment, the voltage loop regulates the output voltage to maintain the stability of the output voltage, while the current loop controls the average current of the input inductor to make it follow the current reference signal. The sampled input voltage signal and the output signal of the voltage loop are operated by a multiplier to obtain a current reference signal. By reasonably designing the voltage loop compensator Gvc(s) and the current loop compensator Gic(s), the PFC converter can achieve the goals of stable output voltage, power factor correction, and low harmonic distortion. Km is the multiplier gain, and the sampling of the input voltage, output voltage, and inductor current is detected by Kv(s), Ko(s), and Ki(s) respectively. As Figure 3 and Figure 4 shown, after the bridgeless Boost PFC circuit is controlled by average current, the input current waveform approaches a sine wave, is in phase with the input voltage, and the total harmonic distortion rate is lower than 5%. This control method can achieve power factor correction.

[0028] Embodiment 6 As Figure 5 shown, based on Embodiment 4, sample the output voltage and output current of the resonant converter, calculate the output power through a multiplier, and compare with the reference value . At this time, , is the rated power; the normal load condition is: ; the light load condition is: .

[0029] Under normal load, when the resonant converter steps up (gain M≥1) in the forward direction, pulse frequency modulation is adopted: Subtract the current output voltage from the reference voltage , send the obtained error into a proportional-integral regulator to obtain the required regulated switching frequency , then convert into a modulation signal of the corresponding frequency through a voltage-controlled oscillator, and finally obtain the drive signal of the primary switching transistor after comparing the modulation signal with the carrier signal through a comparator., the duty cycle remains unchanged at 50%, and the secondary side switching transistors are all turned off. The anti-parallel diodes are used for uncontrolled rectification; Switching frequency As shown in Equation (1): (1) In Equation (1), is the initial value of. Generally, in order to prevent excessive charging current of the output capacitor from damaging the circuit at the moment of startup, a soft start method with reduced frequency is used to start the converter. is set to (2~3) , is the proportional coefficient of the PI regulator, is the integral coefficient of the PI regulator, is when boosting the error between

[0030] Under normal load, when the resonant converter is stepping down in the forward direction (gain M<1), EPS modulation is used: The error obtained by subtracting the current from is sent to the PI regulator to obtain the phase shift angle . Then, the phase shift angle is converted into the delay time t. The pulse signal with a fixed duty cycle is delayed by different degrees to obtain the drive signals of the primary side and secondary side switching transistors respectively, and then the on and off of the primary side and secondary side switching transistors of the resonant converter are controlled. The duty cycle is 50% for both, and the switching frequency is always equal to the resonant frequency . The phase shift angle within the primary side bridge and the phase shift angle between the primary and secondary side bridges satisfy , ensuring the lowest switching loss of the resonant converter; The phase shift angle is represented by Equations (2) and (3): (2) (3) In Equations (2) and (3), is the phase shift angle, is the initial value of, is the proportional coefficient of the PI regulator, is the integral coefficient of the PI regulator, is when stepping down the error between

[0031] The delay time is represented by Equation (4): (4) In Equation (4), is the delay time, is the phase shift angle, is the resonant frequency.

[0032] Under light load, the switching intermittent Bang-Bang charge control is adopted. First, the switching device of the resonant converter stops working, and at the same time, the output voltage is detected , when decreases to the minimum value, it enters the intermittent mode, and the switching device , conducts, and the voltage of the primary-side resonant capacitor is compared with the reference value . When , , are turned off, , are turned on. When , , are turned off until drops to the minimum value again and this process is cycled; The calculation method of the reference value is as follows: The net input charge of the power supply side of the resonant converter in each half switching cycle can be calculated by Equation (5): (5) In Equation (5), is the primary-side resonant capacitor, , , are respectively , , the voltages of the primary-side resonant capacitor at moments; When operating stably, , substituting it into Equation (5), the input charge in one switching cycle is: (6) In Equation (6), is the primary-side resonant capacitor, is the voltage of the primary-side resonant capacitor at moment; Although the parasitic capacitance of the power device is very small, the charge and discharge during the dead time also generate charge exchange, which should not be ignored. The analysis shows that the output capacitances C S5 ~C S8 of the secondary-side switching devices do not participate in the energy exchange with the power supply, but are related to the output capacitances C S1 ~C S4Similarly, charge and discharge occur during the dead time. Therefore, after compensating for the power exchange that occurs during the dead time, the input power for each switching cycle of the resonant converter should be: (7) In Equation (7), is the voltage of the resonant capacitor when the switch is turned off. When , and are known, the power output by the power supply can be obtained from the voltage of the resonant capacitor. The input power reflects the input power. Therefore, by adjusting the capacitance voltage value , the input power can be adjusted.

[0033] When the resonant converter operates under no-load conditions, the latter term in Equation (7) is much smaller than the former term, which is used to offset the actual line losses. The no-load input power is obtained as: (8) In Equation (8), is the primary-side resonant capacitor, is the voltage of the resonant capacitor when the switch is turned off; The no-load input power is all output to the filter capacitor, and the change in the output voltage is: (9) In Equation (9), is the primary-side resonant capacitor, is the voltage of the resonant capacitor when the switch is turned off, is the output capacitor; When the resonant converter operates under no-load conditions, the voltage fluctuation is the largest, denoted as . To reliably control the output voltage, it is necessary to make ΔV less than . From this, we can obtain: (10) In Equation (10), is the output capacitor, is the primary-side resonant capacitor, is the maximum voltage fluctuation when the resonant converter operates under no-load conditions; Considering the output efficiency, after consulting the data and performing simulation analysis, the load value at which the output efficiency is the highest is found. After testing, the capacitance voltage at this time is , then the reference value is: (11) In formula (11), is the switch when turned off, the resonant capacitor the maximum value of the voltage, is the capacitor voltage value at the highest output efficiency.

[0034] Finally, it should also be noted that in this document, relational terms such as first and second are only used to distinguish one entity or operation from another entity or operation, and do not necessarily require or imply any actual relationship or order between these entities or operations. Moreover, the terms "comprising", "including" or any other variant thereof are intended to cover non-exclusive inclusion, such that a process, method, article or device comprising a series of elements includes not only those elements but also other elements not expressly listed, or elements inherent to such process, method, article or device. Without further limitation, an element defined by the statement "comprising an..." does not exclude the presence of additional identical elements in the process, method, article or device comprising the element.

[0035] The various embodiments in this specification are described in a progressive manner, with each embodiment focusing on the differences from other embodiments. The same or similar parts between the various embodiments can be referred to each other.

[0036] The above description of the disclosed embodiments enables those skilled in the art to implement or use the present application. Various modifications to these embodiments will be obvious to those skilled in the art, and the general principles defined herein can be implemented in other embodiments without departing from the spirit or scope of the present application. Therefore, the present application will not be limited to the embodiments shown herein, but rather to the broadest scope consistent with the principles and novel features disclosed herein.

Claims

1. The control circuit of the vehicle charger is characterized in that: It includes a bridgeless Boost PFC circuit, capacitor C i , resonant converter and capacitor C o ; The input of the bridgeless Boost PFC circuit is connected to the grid, and the capacitor C o Connect with load; The bridgeless Boost PFC circuit includes an inductor L, which is connected in series with a diode D1 and a diode D2. The power grid, the inductor L, the diode D1 and the diode D2 form a closed circuit a, and the closed circuit a is connected in parallel with a switch circuit; the switch circuit includes a switch tube Q1 and a switch tube Q2 connected in series; the capacitor C i , capacitor C i One end is connected to a closed circuit a, capacitor C i The other end is connected to the switch circuit a; The resonant converter includes a primary full-bridge circuit, a high-frequency transformer T and a secondary full-bridge circuit, and the primary full-bridge circuit is connected to the secondary full-bridge circuit through the high-frequency transformer T; Primary full bridge circuit and capacitor C i In parallel, the secondary full bridge circuit and capacitor C o in parallel.

2. The control circuit of the vehicle charger according to claim 1, characterized in that: The primary full-bridge circuit includes switch tube S1, switch tube S2, switch tube S3, and switch tube S4. Switch tube S1 and switch tube S3 are connected to capacitor C i One end is connected to the switch tube S2 and the switch tube S4 and the capacitor C i The other end is connected, the switch tube S1 is connected in series with the switch tube S2, and the switch tube S3 is connected in series with the switch tube S4; The primary full-bridge circuit also includes a resonant inductor L connected to the primary side of the high-frequency transformer T. r1 , Excitation inductance L m and resonant capacitor C r1 , resonant inductor L r1 One end is connected to the connection point of switch tube S1 and switch tube S2, and the resonant inductor L r1 The other end is connected to the excitation inductor L m One end of the resonant capacitor C r1 One end is connected to the connection point of switch tube S3 and switch tube S4, and the resonant capacitor C r1 The other end is connected to the excitation inductor L m the other end.

3. The control circuit of the vehicle charger according to claim 2, characterized in that: The secondary full-bridge circuit includes switch tube S5, switch tube S6, switch tube S7, and switch tube S8. Switch tube S5 and switch tube S7 are connected to capacitor C o One end is connected to the switch tube S6 and the switch tube S8 and the capacitor C o The other end is connected, the switch tube S5 is connected in series with the switch tube S6, and the switch tube S7 is connected in series with the switch tube S8; The secondary full-bridge circuit also includes a resonant inductor L connected to the secondary side of the high-frequency transformer T. r2 and resonant capacitor C r2 , resonant inductor L r2 The connection point between the switch tube S7 and the switch tube S8, the resonant capacitor C r2 A connection point connecting the switch tube S5 and the switch tube S6.

4. A method for controlling a vehicle charger, using the control circuit of the vehicle charger according to any one of claims 1 to 3, characterized in that: Follow the steps below to implement it: S1: The bridgeless Boost PFC circuit adopts average current control, takes the average value of the actual input inductor L current as the sampling current, and uses the current inner loop and voltage outer loop control method to adjust the average value of the input current, and controls the input current average value to follow the input grid voltage; S2: The resonant converter adopts hybrid modulation. Under normal load conditions, frequency modulation + extended phase shift segmented modulation is adopted; under light load conditions, intermittent Bang-Bang charge control is switched.

5. The control method of the vehicle charger according to claim 4, characterized in that: S1 is specifically: The input current, input voltage and output voltage of the bridgeless Boost PFC circuit are collected; the output voltage is transmitted to the voltage loop compensator, and the feedback voltage signal output by the voltage loop compensator and the input voltage signal are multiplied by a multiplier to obtain a current reference signal. The current reference signal and the input current signal enter the current loop compensator, and the current loop compensator outputs a PWM signal to control the on and off time of the switch tube.

6. The control method of the vehicle charger according to claim 4, characterized in that: S2 is specifically: sampling the output voltage of the resonant converter and output current The output power is calculated by the multiplier ,Will With reference value For comparison, , is the rated power; the normal load condition is: ; Light load conditions are: .

7. The control method of the vehicle charger according to claim 6, characterized in that: Under normal load, the resonant converter uses pulse frequency modulation when boosting forward: The current output voltage and reference voltage The error obtained by subtraction is sent to the proportional integral regulator to obtain the switching frequency required for adjustment. , and then through the voltage controlled oscillator The modulation signal is converted into a modulation signal of the corresponding frequency, and finally the modulation signal is compared with the carrier signal by a comparator to obtain the driving signal of the primary switch tube. , the duty cycle remains unchanged at 50%, the secondary switch tubes are all turned off, and the anti-parallel diode is used for uncontrolled rectification; The switching frequency As shown in formula (1): （1） In formula (1), for The initial value of Set to (2~3) , is the proportional coefficient of the PI regulator, is the integral coefficient of the PI regulator, For boost and of error.

8. The control method of the vehicle charger according to claim 6, characterized in that: Under normal load, when the resonant converter is in forward buck mode, EPS modulation is used: Set the current and The error obtained by subtraction is sent to the PI regulator to obtain the phase shift angle , and then the phase shift angle The pulse signal with a fixed duty cycle is converted into a delay time t, and the driving signals of the primary and secondary switch tubes are obtained after different degrees of delay, thereby controlling the on and off of the primary and secondary switch tubes of the resonant converter. The duty cycle is 50% and the switching frequency is Always equal to the resonant frequency , the phase shift angle within the primary bridge and the phase shift angle between the primary and secondary bridges meet , ensuring the lowest switching loss of the resonant converter; The phase shift angle is It is reflected by formula (2) and formula (3): （2） （3） In formula (2) and formula (3), is the phase shift angle, for The initial value of is the proportional coefficient of the PI regulator, is the integral coefficient of the PI regulator, When the pressure is reduced and The error of The delay time is expressed by formula (4): （4） In formula (4), is the delay time, Phase shift angle, is the resonant frequency.

9. The control method of the vehicle charger according to claim 6, characterized in that: When the load is light, the switch of the resonant converter is stopped first, and the output voltage is detected at the same time. ,when When it drops to the minimum value, it enters intermittent mode and the switch , The hysteresis comparator is used to convert the primary side resonant capacitor voltage With reference value In comparison, When , , open , , When , ,until Once again, it drops to the minimum value and repeats this process.

10. The control method of the vehicle charger according to claim 9, characterized in that: The calculation method of the reference value is: The net input power of the resonant converter power supply side in each half switching cycle is calculated by equation (5): （5） In formula (5), is the primary side resonant capacitor, , , They are , , The primary side resonant capacitor at the moment Voltage; When running stably, , substituting into formula (5), the input power in one switching cycle is obtained as: （6） In formula (6), is the primary side resonant capacitor, yes The primary side resonant capacitor Voltage; After compensating for the energy exchange that occurs during the dead time, the input energy of the resonant converter in each switching cycle is: （7） In formula (7), For switch Resonant capacitor at turn-off The voltage value is , and When is known, the amount of power output by the power supply can be calculated from the resonant capacitor voltage to obtain; When the resonant converter is running at no-load, the no-load input power is: （8） In formula (8), is the primary side resonant capacitor, For switch Resonant capacitor at turn-off Voltage value; The no-load input power is all output to the filter capacitor, and the output voltage change is: （9） In formula (9), is the primary side resonant capacitor, For switch Resonant capacitor at turn-off The voltage value, is the output capacitor; When the resonant converter operates under no-load conditions, the voltage fluctuation is the largest, which is recorded as , so that ΔV is less than , from which we can get: （10） In formula (10), is the output capacitor, is the primary side resonant capacitor, It is the maximum value of voltage fluctuation when the resonant converter works under no-load condition; Find the load value when the output efficiency is the highest, and get the capacitor voltage at this time: , then the reference value for: （11） In formula (11), For switch Resonant capacitor at turn-off The maximum voltage, It is the capacitor voltage value when the output efficiency is the highest.

Images