Radar device, radar signal processing method, and radar signal processing circuit

Abstract

The invention provides a radar apparatus, a radar signal processing method, and a radar signal processing circuit. A radar apparatus includes: a signal generation circuit that generates a baseband signal; a code generation circuit for generating a plurality of code sequences; a phase rotation circuit that adds a phase rotation based on a portion of the plurality of encoding sequences to the baseband signal and generates a plurality of encoded and multiplexed transmission signals; and a plurality of transmission antennas that respectively transmit the plurality of transmission signals. A code length of the plurality of coding sequences is greater than a coding multiplexing number for the plurality of transmission signals.

Description

[0001] This application is a divisional application of the application with the application date of "October 21, 2020", application number "2020800787190", and invention name "Radar Device". Technical Field

[0002] The present disclosure relates to a radar apparatus. Background Art

[0003] In recent years, radar systems that use short-wavelength radar transmission signals, including microwaves and millimeter waves, which achieve high resolution, have been gaining popularity. Furthermore, to improve safety outdoors, there is a demand for radar systems that can detect not only vehicles but also small objects such as pedestrians over a wide angle (e.g., so-called "wide-angle radar systems").

[0004] As a structure of a radar device with a wide-angle detection range, for example, there is a structure using the following method (Direction of Arrival (DOA) estimation method), which receives reflected waves from a target (or "object target") using an array antenna composed of multiple antennas (or, also called "antenna elements"), and estimates the direction of arrival (or, "angle of arrival") of the reflected wave based on the reception phase difference relative to the element spacing (antenna spacing).

[0005] For example, as a method for estimating the angle of arrival, the Fourier method (FFT (Fast Fourier Transform) method) can be listed, or, as a method that can obtain high resolution, the Capon method, MUSIC (Multiple Signal Classification) and ESPRIT (Estimation of Signal Parameters via Rotational Invariance Techniques, signal parameter estimation based on rotational invariance technology) can be listed.

[0006] In addition, as a radar device, for example, the following structure has been proposed (sometimes also called "MIMO (Multiple Input Multiple Output) radar"): in addition to the receiving side, the transmitting side also has multiple antennas (array antennas), and beam scanning is performed by signal processing using the transmitting and receiving array antennas (for example, refer to non-patent document 1).

[0007] Prior art literature

[0008] Patent Literature

[0009] Patent Document 1: U.S. Patent No. 9,541,638

[0010] Non-patent literature

[0011] Non-patent document 1: J. Li, and P. Stoica, "MIMO Radar with Colocated Antennas", Signal Processing Magazine, IEEE Vol. 24, Issue 5, pp. 106-114, 2007

[0012] Non-patent document 2: M. Kronauge, H. Rohling, "Fast two-dimensional CFAR procedure", IEEE Trans. Aerosp. Electron. Syst., 2013, 49, (3), pp. 1817-1823

[0013] Non-patent literature 3: Direction-of-arrival estimation using signal subspace modeling Cadzow, JA; Aerospace and Electronic Systems, IEEE Transactions on Volume: 28, Issue: 1 Publication Year: 1992, Page (s): 64-79

[0014] Non-Patent Document 4: V. Winkler, “Novel Waveform Generation Principle for Short-Range FMCW-Radars,” in Proc. German Microw. Conf., 2009, pp. 1-4.

[0015] Non-patent document 5: Y.Kozawa and H.Habuchi, "Theoretical Analysis of Atmospheric Optical DS / SS with On-Off Orthogonal M-sequence Pairs", Sixth International Conference on Information, Communications and Signal Processing (ICICS2007), P0686 (Dec.2007) Summary of the Invention

[0016] However, methods of detecting targets in radar devices (eg, MIMO radars) have not been fully studied.

[0017] The non-limiting embodiments of the present disclosure contribute to providing a radar device that improves target detection accuracy.

[0018] A radar device according to an embodiment of the present disclosure includes: a signal generation circuit for generating a baseband signal; a code generation circuit for generating a plurality of code sequences; a phase rotation circuit for adding a phase rotation based on a portion of the plurality of code sequences to the baseband signal to generate a plurality of code-multiplexed transmission signals; and a plurality of transmitting antennas for respectively transmitting the plurality of transmission signals; the plurality of code sequences having a code length greater than a code-multiplexing number for the plurality of transmission signals.

[0019] Furthermore, these broad or specific embodiments may be implemented by systems, apparatuses, methods, integrated circuits, computer programs, or recording media, or by any combination of systems, apparatuses, methods, integrated circuits, computer programs, and recording media.

[0020] According to one embodiment of the present disclosure, the target object detection accuracy of a radar device can be improved.

[0021] Further advantages and effects of an embodiment of the present disclosure will be clearly presented in the description and drawings. The above advantages and / or effects are provided by several embodiments and features described in the description and drawings, but it is not necessary to provide all of them in order to obtain one or more of the same features. BRIEF DESCRIPTION OF THE DRAWINGS

[0022] Figure 1 This is a block diagram showing a configuration example of a radar device according to the first embodiment.

[0023] Figure 2 This is a diagram showing an example of a transmission signal and a reflected wave signal when a chirp pulse is used.

[0024] Figure 3 This is a diagram showing an example of the Doppler shift amount in the first embodiment.

[0025] Figure 4 This is a block diagram showing a configuration example of a radar device according to the first embodiment.

[0026] Figure 5 This is a diagram showing an example of computer simulation results of aliasing determination processing.

[0027] Figure 6 This is a diagram showing an example of computer simulation results of aliasing determination processing.

[0028] Figure 7 This is a diagram showing an example of calculation results of the extended Doppler frequency index.

[0029] Figure 8 This is a block diagram showing a configuration example of a radar device according to a second embodiment.

[0030] Figure 9 This is a diagram showing an example of a transmission signal and a reflected wave signal when chirped pulses are used.

[0031] Figure 10 This is a block diagram showing a configuration example of a radar device according to variation 1 of embodiment 2.

[0032] Figure 11 This is a diagram showing an example of a transmission signal and a reflected wave signal when chirped pulses are used.

[0033] Figure 12 This is a block diagram showing a configuration example of a radar device according to variation 2 of embodiment 2.

[0034] Figure 13 This is a diagram showing an example of a transmission signal and a reflected wave signal when chirped pulses are used.

[0035] Figure 14 This is a diagram showing a configuration example of a radar transmitting unit according to another modification 1.

[0036] Figure 15 This is a diagram showing a configuration example of a radar transmitting unit according to another modification 1. DETAILED DESCRIPTION

[0037] MIMO radar, for example, transmits multiplexed signals (radar transmission waves) using time division, frequency division, or code division from multiple transmitting antennas (or "transmitting array antennas"). Next, the MIMO radar receives signals reflected by surrounding objects (radar reflection waves) using multiple receiving antennas (or "receiving array antennas"), demultiplexing and receiving the multiplexed transmission signal from each received signal. This process allows the MIMO radar to obtain a propagation path response represented by the product of the number of transmitting antennas and the number of receiving antennas, and perform array signal processing on these received signals as a virtual receiving array.

[0038] In addition, in MIMO radar, by appropriately configuring the element spacing in the transmitting and receiving array antenna, the antenna aperture can be virtually enlarged to achieve improved angular resolution.

[0039] As an example, the following focuses on a MIMO radar using coded multiplexing transmission, which is one of methods for simultaneously multiplexing transmission signals from a plurality of transmission antennas and transmitting them (for example, see Patent Document 1).

[0040] For example, a MIMO radar using code-multiplexed transmission repeatedly applies phase modulation based on a different code string (hereinafter also referred to as "code" or "code sequence") to each transmitting antenna for each repetition of a transmission signal (e.g., a chirp signal), and performs code-multiplexed transmission using multiple (e.g., M) transmitting antennas. Furthermore, the MIMO radar extracts distance information from the code-multiplexed received signal by performing detection processing on the signal received using multiple (e.g., N) receiving antennas.

[0041] Furthermore, a MIMO radar, for example, performs a Fourier transform on the distance information obtained with each repetition of the transmitted signal in M ​​velocity directions. The MIMO radar applies a phase correction based on the detected velocity component to the Fourier transform results in M ​​velocity directions, and multiplies this result by the inverse code string used to separate the code strings assigned to each transmitting antenna, thereby separating the code-multiplexed received signals. This MIMO radar structure allows the radar to suppress mutual interference between the code-multiplexed received signals and separate them, even when the relative velocity between the target and the MIMO radar is non-zero.

[0042] However, in the configuration of the MIMO radar described above, the velocity direction Fourier transform processing is performed in multiple (for example, M) steps. Therefore, in the MIMO radar, the velocity direction Fourier transform processing is performed at intervals of M transmission cycles. Therefore, the maximum Doppler frequency specified by the sampling theorem that does not produce Doppler aliasing is one-Mth (=1 / M) of the number of transmitting antennas used in code-multiplexed transmission. In the case of Doppler frequency components exceeding the maximum Doppler frequency specified by the sampling theorem that does not produce Doppler aliasing, the Doppler frequency cannot be determined, resulting in ambiguity. Thus, compared to transmission without code-multiplexing (single-antenna transmission, M=1), the Doppler range in which Doppler components can be unambiguously detected is reduced to 1 / M. In other words, the "Doppler range" is equivalent to the "relative velocity range of the target." In other words, the “Doppler range in which the Doppler component can be detected unambiguously” is a Doppler range in which the Doppler frequency can be determined unambiguously, and is hereinafter referred to as the “unambiguously detectable Doppler range”.

[0043] In this regard, one embodiment of the present disclosure describes a method for expanding the Doppler frequency range that generates no ambiguity during code-multiplexed transmission. A radar device according to one embodiment of the present disclosure suppresses mutual interference between code-multiplexed signals, even when Doppler fluctuations are caused by the movement of the target or the radar device, and expands the Doppler range that can be unambiguously detected to the same level as during single-antenna transmission. This improves target detection accuracy over a wider Doppler frequency range.

[0044] Hereinafter, an embodiment of the present disclosure will be described in detail with reference to the accompanying drawings. In the embodiments, the same components are denoted by the same reference numerals, and their descriptions are omitted due to duplication.

[0045] The following describes a configuration (in other words, a MIMO radar configuration) in which, in a radar apparatus, different transmission signals are simultaneously multiplexed and sent from multiple transmission antennas in a transmission branch, and each transmission signal is separated and received in a reception branch.

[0046] In addition, the following describes, as an example, a radar system using frequency-modulated pulse waves such as chirped pulses (also known as "fast chirp modulation") for transmission. However, modulation methods are not limited to frequency modulation. For example, one embodiment of the present disclosure can also be applied to a radar system using pulse compression radar, which transmits a pulse train using phase or amplitude modulation.

[0047] In addition, the radar device performs code-multiplexing and transmits the signal.

[0048] (Implementation 1)

[0049] [Structure of radar device]

[0050] Figure 1 It is a block diagram showing a configuration example of the radar device 10 according to this embodiment.

[0051] The radar device 10 includes a radar transmitting unit (transmitting branch) 100 and a radar receiving unit (receiving branch) 200 .

[0052] The radar transmitting unit 100 generates a radar signal (radar transmission signal), and transmits the radar transmission signal at a predetermined transmission cycle using a transmission array antenna composed of a plurality of transmission antennas 106 (eg, Nt).

[0053] The radar receiver 200 uses a receiving array antenna comprising multiple receiving antennas 202 (e.g., Na) to receive radar transmission signals, i.e., reflected wave signals, that have been reflected by a target (not shown). The radar receiver 200 processes the reflected wave signals received by each receiving antenna 202, for example, to detect the presence of a target object or estimate the range, Doppler frequency (in other words, relative velocity), and direction of arrival of the reflected wave signals, and outputs information related to the estimation results (in other words, positioning information).

[0054] Furthermore, the target is an object to be detected by the radar device 10 , and includes, for example, a vehicle (including four-wheeled and two-wheeled vehicles), a person, a rock, or a curb.

[0055] [Configuration of Radar Transmitter 100]

[0056] The radar transmitting unit 100 includes a radar transmission signal generating unit 101 , a code generating unit 104 , a phase rotating unit 105 , and a transmitting antenna 106 .

[0057] Radar transmission signal generator 101 generates a radar transmission signal (in other words, a baseband signal). Radar transmission signal generator 101 includes, for example, a modulation signal generator 102 and a VCO (Voltage Controlled Oscillator) 103. The following describes the components of radar transmission signal generator 101.

[0058] For example, Figure 2 As shown in the upper layer of FIG, the modulation signal generating unit 102 generates a sawtooth-shaped modulation signal (in other words, a modulation signal for VCO control) according to the radar transmission period Tr.

[0059] The VCO 103 outputs a frequency modulation signal (hereinafter, for example, referred to as a "frequency chirp signal" or "chirp signal") to the phase rotation unit 105 and the radar receiving unit 200 (a mixer unit described later) based on the radar transmission signal (modulation signal) output from the modulation signal generating unit 102.

[0060] The code generation unit 104 generates different codes for each transmitting antenna 106 performing code-multiplexed transmission. The code generation unit 104 outputs the phase rotation amount corresponding to the generated code to the phase rotation unit 105. Furthermore, the code generation unit 104 outputs information related to the generated code to the radar receiving unit 200 (output switching unit 209, described later).

[0061] Phase rotation unit 105 applies a phase rotation amount input from code generation unit 104 to the chirp signal input from VCO 103 and outputs the phase-rotated signal to transmit antenna 106. For example, phase rotation unit 105 includes a phase shifter and a phase modulator (not shown). The output signal from phase rotation unit 105 is amplified to a predetermined transmit power and radiated into space from each transmit antenna 106. In other words, radar transmit signals are code-multiplexed and transmitted from multiple transmit antennas 106 by applying a phase rotation amount corresponding to the code.

[0062] Next, an example of the coding (for example, orthogonal code) set in the radar device 10 will be described.

[0063] The code generation unit 104 generates a different code for each transmission antenna 106 that performs code-multiplexed transmission, for example.

[0064] For example, in the following, the number of transmitting antennas 106 for code multiplexing transmission is set to "Nt", and the code multiplexing number is set to "N CM ".exist Figure 1 In, N CM =Nt.

[0065] The code generation unit 104 generates N contained in a code sequence (for example, orthogonal code sequences (or simply referred to as "codes" or "orthogonal codes")) having a code length (in other words, the number of code elements) Loc. allcode (hereinafter sometimes also recorded as "N allcode N in (Loc) ") orthogonal codes CM Orthogonal codes are set as codes for code multiplexing transmission.

[0066] For example, the coding multiplexing number N CM Less than the orthogonal code number N allcode , N CM <N allcode In other words, the code length Loc of the orthogonal code is greater than the code multiplexing number N CM For example, the code length Loc is N CM Code ncm =[OC ncm (1), OC ncm (2),…,OC ncm (Loc)]. Here, “OC ncm (noc)" represents the ncmth orthogonal code Code ncm In addition, "ncm" represents the index of the orthogonal code used for code multiplexing, ncm = 1, ..., N CM In addition, “noc” is the index of the coding element, noc=1,…,Loc.

[0067] Here, the code length Loc is N allcode (N allcode -N CM ) orthogonal codes will not be used by the code generation unit 104 (in other words, will not be used for code multiplexing transmission). allcode -N CM ) orthogonal codes not used by the code generation unit 104 are referred to as "unused orthogonal codes." At least one of the unused orthogonal codes is used, for example, for aliasing determination of Doppler frequency in the aliasing determination unit 212 of the radar receiving unit 200 described later (an example will be described later).

[0068] By using unused orthogonal codes, the radar device 10 can, for example, separately receive individual signals transmitted through code-multiplexing from multiple transmitting antennas 106 while suppressing inter-code interference, and can expand the range of detectable Doppler frequencies (an example will be described later).

[0069] As described above, N generated in the code generation unit 104 CM Orthogonal codes are, for example, codes that are orthogonal to one another (in other words, uncorrelated codes). For example, Walsh-Hadamard codes can be used in orthogonal code sequences. The code length of a Walsh-Hadamard code is a power of 2, and each Walsh-Hadamard code of that length contains the same number of orthogonal codes as the code length. For example, Walsh-Hadamard codes of code lengths of 2, 4, 8, or 16 contain 2, 4, 8, or 16 orthogonal codes, respectively.

[0070] As an example, the number of codes is set to N in order to satisfy the following formula (1). CM The code length of the orthogonal code sequence is Loc.

[0071]

[0072] Here, ceil[x] is an operator (ceil function) that outputs the smallest integer greater than the real number x. In the case of a Walsh-Hadamard code with a code length of Loc, N allcode (Loc) = Loc. For example, a Walsh-Hadamard code with a code length of Loc = 2, 4, 8, or 16 contains 2, 4, 8, or 16 orthogonal codes, respectively. Therefore, N allcode (2) = 2, N allcode (4) = 4, N allcode (8) = 8 and N allcode (16)=16. The code generation unit 104 uses, for example, N contained in the Walsh-Hadamard code of code length Loc. allcode N of (Loc) codes CM orthogonal codes.

[0073] Here, we explain longer code lengths. For example, when the target or conventional radar device's moving speed includes acceleration, longer code lengths increase susceptibility to inter-symbol interference. Furthermore, longer code lengths increase the candidate Doppler aliasing range used in Doppler aliasing determination, described later. Therefore, if targets with multiple Doppler frequencies exist within different aliasing ranges at the same range index, the probability of duplicate Doppler frequency indices detected within these different aliasing ranges increases, increasing the probability that conventional radar devices will have difficulty properly determining aliasing.

[0074] Therefore, from the perspective of the amount of computation required for aliasing determination in the aliasing determination unit 212 of the radar receiver 200, which will be described later, the radar device 10 may use a code with a shorter code length. As an example, the radar device 10 may use an orthogonal code sequence with the shortest code length among the code lengths Loc that satisfy equation (1).

[0075] In addition, when the Walsh-Hadamard code of code length Loc includes, for example, the encoding of code length Loc [OC(1), OC(2), ..., OC(Loc-1), OC(Loc)], the Walsh-Hadamard code of code length Loc also includes the encoding [OC(1), -OC(2), ..., OC(Loc-1), -OC(Loc)], the odd-numbered encoding elements of the encoding are the same, while the signs of the even-numbered encoding elements are reversed.

[0076] In addition, even if it is another code different from the Walsh-Hadamard code of code length Loc, for example, in the case of the code [OC(1), OC(2), ..., OC(Loc-1), OC(Loc)] including the code length Loc, the code of code length Loc can be the code [OC(1), -OC(2), ..., OC(Loc-1), -OC(Loc)], in which the odd-numbered code elements are the same and the signs of the even-numbered code elements are reversed; or it can be the code [-OC(1), OC(2), ..., -OC(Loc-1), OC(Loc)], in which the even-numbered code elements are the same and the signs of the odd-numbered code elements are reversed.

[0077] The number of unused orthogonal codes (N allcode -N CM) is 2 or greater, the radar device 10 may select codes so that a group of orthogonal codes that does not include the codes with the aforementioned relationship is not used. For example, within the group of codes with the aforementioned relationship, one code may be used for code-multiplexed transmission while the other code is included in the unused orthogonal codes. The selection of these unused orthogonal codes can improve the accuracy of Doppler frequency aliasing determination in the aliasing determination unit 212 of the radar receiving unit 200, described later (an example will be described later).

[0078] The following describes the coding multiplexing number N. CM An example of orthogonal codes in .

[0079] <N CM =2 or 3>

[0080] In N CM = 2 or 3, for example, Walsh-Hadamard codes with code lengths Loc = 4, 8, 16, 32, ... can also be applied. In the case of these code lengths Loc, N CM <N allcode (Loc). In addition, when the coding multiplexing number is N CM =2 or 3, the Walsh-Hadamard code with the shortest code length among these code lengths Loc (for example, Loc=4) can be used.

[0081] For example, a Walsh-Hadamard code with a code length of Loc is expressed as “WH Loc (nwhc)". In addition, nwhc represents the coding index contained in the Walsh-Hadamard code with a code length of Loc, nwhc = 1, ..., Loc. For example, the Walsh-Hadamard code with a code length of Loc = 4 includes orthogonal codes WH4(1) = [1, 1, 1, 1], WH4(2) = [1, -1, 1, -1], WH4(3) = [1, 1, -1, -1] and WH4(4) = [1, -1, -1, 1].

[0082] Here, WH4(1) = [1, 1, 1, 1] and WH4(2) = [1, -1, 1, -1] in the Walsh-Hadamard code with code length Loc = 4 are a set of codes in which the odd-numbered code elements are the same, while the codes for the even-numbered code elements are reversed. Furthermore, WH4(3) = [1, 1, -1, -1] and WH4(4) = [1, -1, -1, 1] are also a set of codes that have the same relationship as the set of WH4(1) and WH4(2).

[0083] For example, when the number of unused orthogonal codes (N allcode -N CM ) is 2 or more, the radar device 10 may select codes so as not to include a group of codes having such a relationship among the unused orthogonal codes.

[0084] For example, in the coding multiplexing number N CM = 2, the code generation unit 104 determines two orthogonal codes in the Walsh-Hadamard code with code length Loc = 4 as the codes for code multiplexing transmission. In this case, the number of unused orthogonal codes (N allcode -N CM ) is 2.

[0085] For example, the code generation unit 104 may select the code for code-multiplexed transmission so that the unused orthogonal codes do not include the code group of WH4(1) and WH4(2) or the code group of WH4(3) and WH4(4). For example, the combination of codes (Code1 and Code2) for code-multiplexed transmission may be a combination of Code1=WH4(1) (=[1,1,1,1]) and Code2=WH4(3) (=[1,1,-1,-1]), a combination of Code1=WH4(1) and Code2=WH4(4), a combination of Code1=WH4(2) and Code2=WH4(3), or a combination of Code1=WH4(2) and Code2=WH4(4).

[0086] In addition, when the coding multiplexing number N CM =2, for example, the aliasing determination unit 212 in the radar receiving unit 200 may be N with a code length of Loc=4. allcode = 2 of the 4 Walsh-Hadamard codes not used by the code generation unit 104 (in other words, not used for code-multiplexed transmission) (= N allcode -N CM ) At least one of the unused orthogonal codes is used for aliasing determination (an example will be described later).

[0087] Next, the code length Loc N allcode The unused orthogonal codes among the orthogonal codes are recorded as "UnCode nuc =[UOC nuc (1),UOC nuc (2),…,UOC nuc (Loc)]". In addition, UnCode nuc Indicates the nucth unused orthogonal code. In addition, nuc represents the index of the unused orthogonal code, nuc = 1, ..., (N allcode -N CM ). In addition, UOC nuc (noc) represents the nucth unused orthogonal code UnCode nuc In addition, noc represents the index of the coding element, noc=1, ..., Loc.

[0088] For example, when the coding multiplexing number is N CM =2, and the codes for code-multiplexed transmission determined by the code generation unit 104 are Code1 = WH4(1) (= [1, 1, 1, 1]) and Code2 = WH4(3) (= [1, 1, -1, -1]), the unused orthogonal codes are UnCode1 = WH4(2) (= [1, -1, 1, -1]) and UnCode2 = WH4(4) (= [1, -1, -1, 1]). In addition, the combination of the unused orthogonal codes (UnCode1 and UnCode2) is not limited to the combination of WH4(2) and WH4(4), and may be a combination of other codes.

[0089] Similarly, in the coding multiplexing number N CM = 3, the code generation unit 104 determines, for example, three orthogonal codes in the Walsh-Hadamard code with a code length of Loc = 4 as the codes for code multiplexing transmission. In this case, the number of unused orthogonal codes (N allcode -N CM ) is 1.

[0090] For example, the code generation unit 104 may select Code1=WH4(3)=[1, 1, -1, -1], Code2=WH4(4)=[1, -1, -1, 1], and Code3=WH4(2)=[1, -1, 1, -1].

[0091] In addition, the aliasing determination unit 212 of the radar receiving unit 200 sets the N allcode = 1 of 4 Walsh-Hadamard codes (= N allcode -N CM ) Orthogonal codes are not used for aliasing determination (an example will be described later). For example, when the code multiplexing number is N CM =3, and the codes for code-multiplexed transmission determined by the code generation unit 104 are Code1 = WH4(3) = [1, 1, -1, -1], Code2 = WH4(4) = [1, -1, -1, 1], and Code3 = WH4(2) = [1, -1, 1, -1], the unused orthogonal code is UnCode1 = WH4(1) = [1, 1, 1, 1]. Furthermore, the combinations of the codes for code-multiplexed transmission (Code1, Code2, and Code3) and the unused orthogonal code (UnCode1) are not limited to these combinations, and other combinations of codes may also be used.

[0092] <N CM =4, 5, 6 or 7>

[0093] In N CM= 4, 5, 6 or 7, for example, Walsh-Hadamard codes with code lengths Loc = 8, 16, 32, ... can be applied. In the case of these code lengths Loc, N CM <N allcode (Loc). In addition, when the coding multiplexing number is N CM When Loc=4, 5, 6, or 7, the Walsh-Hadamard code with the shortest code length among these code lengths Loc (for example, Loc=8) can be used.

[0094] For example, a Walsh-Hadamard code with a code length of Loc=8 includes the following 8 orthogonal codes.

[0095] WH8(1)=[1 1 1 1 1 1 1 1],

[0096] WH8(2)=[1-1 1-1 1-1 1-1],

[0097] WH8(3)=[1 1-1-1 1 1-1-1],

[0098] WH8(4)=[1-1-1 1 1-1-1 1],

[0099] WH8(5)=[1 1 1 1-1-1-1-1],

[0100] WH8(6)=[1-1 1-1-1 1-1 1],

[0101] WH8(7)=[1 1-1-1-1-1 1 1],

[0102] WH8(8)=[1-1-1 1-1 1 1-1]

[0103] Here, WH8(1) and WH8(2) in the Walsh-Hadamard code with code length Loc = 8 are a set of codes in which the codes for the odd-numbered code elements are the same, while the codes for the even-numbered code elements are reversed. Similarly, the set of WH8(3) and WH8(4), the set of WH8(5) and WH8(6), and the set of WH8(7) and WH8(8) are also sets of codes with the same relationship as the set of WH8(1) and WH8(2).

[0104] For example, as the number of unused orthogonal codes (N allcode -N CM) is greater than 2, an example of selecting a code in a manner that does not include a group of codes having such a relationship in unused orthogonal codes, and a code for code multiplexing transmission can be selected in a manner that does not include a group of codes of WH8(1) and WH8(2), a group of codes of WH8(3) and WH8(4), a group of codes of WH8(5) and WH8(6), or a group of codes of WH8(7) and WH8(8) in unused orthogonal codes.

[0105] For example, in the coding multiplexing number N CM = 4, the code generation unit 104 determines the four orthogonal codes in the Walsh-Hadamard code with code length Loc = 8 as the codes for code multiplexing transmission. In this case, the number of unused orthogonal codes (N allcode -N CM ) is 4.

[0106] For example, in the code generation unit 104, the combination of codes (Code 1, Code 2, Code 3, and Code 4) for code-multiplexed transmission may be a combination of Code 1 = WH8(1), Code 2 = WH8(3), Code 3 = WH8(5), and Code 4 = WH8(7), or a combination of Code 1 = WH8(1), Code 2 = WH8(4), Code 3 = WH8(5), and Code 4 = WH8(8). Furthermore, the combination of codes (Code 1, Code 2, Code 3, and Code 4) for code-multiplexed transmission is not limited to these combinations.

[0107] In addition, when the coding multiplexing number N CM = 4, for example, the aliasing determination unit 212 in the radar receiving unit 200 sets the N allcode = 4 of the 8 Walsh-Hadamard codes not used by the code generation unit 104 (= N allcode -N CM ) Part or all of the unused orthogonal codes are used for aliasing determination (an example will be described later).

[0108] For example, in the coding multiplexing number N CM =4, and the codes for code multiplexing transmission determined by the code generation unit 104 are Code1=WH8(1), Code2=WH8(3), Code3=WH8(5), and Code4=WH8(7), the unused orthogonal codes are UnCode1=WH8(2), UnCode2=WH8(4), UnCode3=WH8(6), and UnCode4=WH8(8). In addition, for example, when the code multiplexing number is N CM=4, and the codes used for coded multiplexing transmission determined by the code generation unit 104 are Code1=WH8(1), Code2=WH8(4), Code3=WH8(5) and Code4=WH8(8), the unused orthogonal codes are UnCode1=WH8(2), UnCode2=WH8(3), UnCode3=WH8(6) and UnCode4=WH8(7).

[0109] Similarly, for example, in the case of code multiplexing number N CM = 5, the code generation unit 104 determines 5 orthogonal codes in the Walsh-Hadamard code with code length Loc = 8 as the codes for code multiplexing transmission. In this case, the number of unused orthogonal codes (N allcode -N CM ) is 3.

[0110] For example, in the code generation unit 104, the combination of codes (Code 1, Code 2, Code 3, Code 4, and Code 5) for code-multiplexed transmission may be Code 1 = WH8(1), Code 2 = WH8(3), Code 3 = WH8(5), Code 4 = WH8(7), and Code 5 = WH8(8), or Code 1 = WH8(1), Code 2 = WH8(4), Code 3 = WH8(5), Code 4 = WH8(7), and Code 5 = WH8(8). Furthermore, the combination of codes (Code 1, Code 2, Code 3, Code 4, and Code 5) for code-multiplexed transmission is not limited to these combinations.

[0111] In the coding multiplexing number N CM =5, for example, the aliasing determination unit 212 in the radar receiving unit 200 sets the N allcode = 3 of the 8 Walsh-Hadamard codes not used by the code generation unit 104 (= N allcode -N CM ) Part or all of the unused orthogonal code is used for aliasing determination (an example will be described later).

[0112] For example, when the coding multiplexing number is N CM =5, and the codes for code multiplexing transmission determined by the code generation unit 104 are Code1=WH8(1), Code2=WH8(3), Code3=WH8(5), Code4=WH8(7), and Code5=WH8(8), the unused orthogonal codes are UnCode1=WH8(2), UnCode2=WH8(4), and UnCode3=WH8(6). In addition, for example, when the code multiplexing number is N CM=5, and the codes used for coded multiplexing transmission determined by the code generation unit 104 are Code1=WH8(1), Code2=WH8(4), Code3=WH8(5), Code4=WH8(7) and Code5=WH8(8), the unused orthogonal codes are UnCode1=WH8(2), UnCode2=WH8(3) and UnCode3=WH8(6).

[0113] Similarly, for example, in the case of code multiplexing number N CM = 6, the code generation unit 104 determines 6 orthogonal codes in the Walsh-Hadamard code with code length Loc = 8 as the codes for code multiplexing transmission. In this case, the number of unused orthogonal codes (N allcode -N CM ) is 2.

[0114] For example, in the code generation unit 104, the combination of codes for code-multiplexed transmission (Code 1, Code 2, Code 3, Code 4, Code 5, and Code 6) may be Code 1 = WH8(1), Code 2 = WH8(2), Code 3 = WH8(3), Code 4 = WH8(4), Code 5 = WH8(5), and Code 6 = WH8(8). Furthermore, the combination of codes for code-multiplexed transmission (Code 1, Code 2, Code 3, Code 4, Code 5, and Code 6) is not limited to these combinations.

[0115] In addition, when the coding multiplexing number N CM = 6, for example, the aliasing determination unit 212 in the radar receiving unit 200 sets the N allcode = 2 of the 8 Walsh-Hadamard codes not used by the code generation unit 104 (= N allcode -N CM ) Part or all of the unused orthogonal codes are used for aliasing determination (an example will be described later).

[0116] For example, when the coding multiplexing number is N CM =6, and the codes used for coded multiplexing transmission determined by the code generation unit 104 are Code1=WH8(1), Code2=WH8(2), Code3=WH8(3), Code4=WH8(4), Code5=WH8(5) and Code6=WH8(8), the unused orthogonal codes are UnCode1=WH8(6) and UnCode2=WH8(7).

[0117] Similarly, for example, in the case of code multiplexing number N CM= 7, the code generation unit 104 determines 7 orthogonal codes in the Walsh-Hadamard code with code length Loc = 8 as the codes for code multiplexing transmission. In this case, the number of unused orthogonal codes (N allcode -N CM ) is 1.

[0118] For example, the code generation unit 104 may select Code 1 = WH8(1), Code 2 = WH8(2), Code 3 = WH8(3), Code 4 = WH8(4), Code 5 = WH8(5), Code 6 = WH8(6), and Code 7 = WH8(7) from the codes used for code-multiplexed transmission. The combinations of codes used for code-multiplexed transmission are not limited to these.

[0119] In addition, the aliasing determination unit 212 in the radar receiving unit 200 can be N allcode = 1 of the 8 Walsh-Hadamard codes not used by the code generation unit 104 (= N allcode -N CM ) Orthogonal codes are not used for aliasing determination (an example will be described later).

[0120] For example, in the coding multiplexing number N CM =7, and the codes used for coded multiplexing transmission determined by the code generation unit 104 are Code1=WH8(1), Code2=WH8(2), Code3=WH8(3), Code4=WH8(4), Code5=WH8(5), Code6=WH8(6) and Code7=WH8(7), the unused orthogonal code is UnCode1=WH(8).

[0121] The above describes the coding multiplexing number N. CM =4, 5, 6 or 7.

[0122] In addition, even if the code multiplexing number N CM =8 or more, the radar device 10 can also be used with the code multiplexing number N CM = 2 to 7, the codes for code-multiplexed transmission and the orthogonal codes are determined in the same manner.

[0123] For example, the code generation unit 104 may select N in the Walsh-Hadamard code with a code length Loc shown in equation (2). CM Orthogonal codes are used as codes for code multiplexing transmission. In this case, N CM <Loc=N allcode (Loc).

[0124]

[0125] In addition, the aliasing determination unit 212 in the radar receiving unit 200 can be N allcode = (N allcode -N CM ) unused orthogonal codes are used for aliasing determination (an example will be described later). In addition, the number of unused orthogonal codes (N allcode -N CM ) is two or more, the code generation unit 104 may select the codes for code-multiplexed transmission, for example, in the following manner, that is, the codes in which the odd-numbered code elements and the even-numbered code elements of the codes in the Walsh-Hadamard code of the code length Loc are the same, and the codes in which the signs of the other of the odd-numbered code elements and the even-numbered code elements are reversed are not included in the unused orthogonal codes.

[0126] In other words, it can also be that the codes in the Walsh-Hadamard code with code length Loc have the same coding elements for one of the odd-numbered coding elements and the even-numbered coding elements, and the signs of the coding elements of the other of the odd-numbered coding elements and the even-numbered coding elements are reversed, and either code of the group of codes is included in the unused orthogonal code, while the other code is not included in the unused orthogonal code.

[0127] Furthermore, the elements constituting the orthogonal code sequence are not limited to real numbers, but may also include complex values.

[0128] In addition, the code may be another orthogonal code different from the Walsh-Hadamard code, for example, an orthogonal M-sequence code or a pseudo-orthogonal code.

[0129] The above describes the coding multiplexing number N. CM An example of orthogonal codes in .

[0130] Next, an example of the phase rotation amount based on the code for code-multiplexed transmission generated in the code generation unit 104 will be described.

[0131] For example, the radar device 10 performs code multiplexing transmission using different orthogonal codes for the transmission antennas Tx#1 to Tx#Nt that perform code multiplexing transmission. Therefore, the code generation unit 104 sets the code based on the orthogonal code Code given to the ncm-th transmission antenna Tx#ncm in the m-th transmission cycle Tr. ncm The phase rotation ψ ncm (m), and output to the phase rotation unit 105. Here, ncm=1, ..., N CM .

[0132] For example, as the phase rotation amount ψ ncm(m), during the transmission period of code length Loc times, the orthogonal code Code is cyclically assigned as shown in the following formula (3): ncm Loc encoding elements OC ncm (1),…,OC ncm (Loc) corresponds to the phase amount respectively.

[0133] ψ ncm (m)=angle[OC ncm (OC_INDEX)] (3)

[0134] Here, angle(x) is an operator that outputs the radian phase of the real number x, angle(1) = 0, angle(-1) = π, angle(j) = π / 2, and angle(-j) = -π / 2. j is an imaginary unit. In addition, OC_INDEX indicates the orthogonal code sequence Code ncm The orthogonal code element index of the element can be cyclically changed in the range of 1 to Loc according to the transmission period (Tr) as shown in the following equation (4).

[0135] OC_INDEX=mod(m-1,Loc)+1 (4)

[0136] Here, mod(x, y) is a modulus operator and is a function that outputs the remainder after dividing x by y. Furthermore, m = 1, ..., Nc. Nc is the number of transmission cycles (hereinafter referred to as the "radar transmission signal transmission count") specified by the radar device 10 for radar positioning. Furthermore, the radar device 10 transmits radar transmission signals a number Nc that is an integer multiple of Loc (e.g., Ncode times). For example, Nc = Loc × Ncode.

[0137] Furthermore, the code generation unit 104 outputs the orthogonal code element index OC_INDEX to the output switching unit 209 of the radar receiving unit 200 at each transmission cycle (Tr).

[0138] The phase rotation unit 105 includes, for example, a phase shifter or a phase modulator corresponding to each of the Nt transmitting antennas 106. The phase rotation unit 105 applies a phase rotation amount ψ input from the code generation unit 104 to the chirp signal input from the radar transmission signal generation unit 101, for example, at each transmission period Tr. ncm (m).

[0139] For example, the phase rotation unit 105 assigns the orthogonal code Code assigned to the ncm-th transmission antenna Tx#ncm to the chirp signal input from the radar transmission signal generation unit 101 at the transmission cycle Tr. ncm The phase rotation ψ ncm(m). Here, ncm = 1, ..., N CM , m=1,…,Nc.

[0140] The outputs from the phase rotation unit 105 to the Nt transmitting antennas 106 are amplified to predetermined transmission power, for example, and then radiated into space from the Nt transmitting antennas 106 (for example, transmitting array antennas).

[0141] As an example, the number of transmitting antennas Nt=3 and the number of code multiplexing N CM = 3 when code multiplexing is performed. In addition, the number of transmitting antennas Nt and the number of code multiplexing N CM It is not limited to these values.

[0142] For example, in the mth transmission cycle Tr, the phase rotation amounts ψ1(m), ψ2(m), and ψ3(m) are output from the code generation unit 104 to the phase rotation unit 105 .

[0143] The first (ncm = 1) phase rotation unit 105 (in other words, the phase shifter corresponding to the first (e.g., Tx#1) transmitting antenna 106) applies a phase rotation amount as shown in equation (5) below to the chirp signal generated by the radar transmission signal generation unit 101 at each transmission period Tr. The output of the first phase rotation unit 105 is transmitted by transmitting antenna Tx#1. Here, cp(t) represents the chirp signal at every mth transmission period Tr.

[0144] exp[jψ1(1)]cp(t),exp[jψ1(2)]cp(t),exp[jψ1(3)]cp(t),...,exp[jψ1(Nc)]cp(t)(5)

[0145] Similarly, the second phase rotation unit 105 (ncm = 2) applies a phase rotation amount expressed in equation (6) below to the chirp signal generated by the radar transmission signal generation unit 101 at the transmission period Tr. The output of the second phase rotation unit 105 is transmitted from the transmission antenna Tx#2.

[0146] exp[jψ2(1)]cp(t),exp[jψ2(2)]cp(t),exp[jψ2(3)]cp(t),...,exp[jψ2(Nc)]cp(t)(6)

[0147] Similarly, the third (ncm=3) phase rotation unit 105 applies a phase rotation amount expressed by the following equation (7) to the chirp signal generated by the radar transmission signal generation unit 101 at the transmission period Tr. The output of the third phase rotation unit 105 is transmitted from the transmission antenna Tx#3.

[0148] exp[jψ3(1)]cp(t),exp[jψ3(2)]cp(t),exp[jψ3(3)]cp(t),...,exp[jψ3(Nc)]cp(t)(7)

[0149] Furthermore, when the radar device 10 continuously performs radar positioning, the orthogonal code Code may be variably set for each radar positioning (for example, for each Nc transmission period (Nc×Tr)). ncm The encoding.

[0150] Furthermore, the radar device 10 may variably set, for example, Nt transmit antennas 106 from which the output of the phase rotation unit 105 is transmitted (in other words, transmit antennas 106 corresponding to respective outputs of the phase rotation unit 105). For example, the correspondence between the plurality of transmit antennas 106 and the code sequences used for code-multiplexed transmission may differ for each radar positioning performed by the radar device 10. For example, when the radar device 10 receives a signal due to interference from other radars that varies for each transmit antenna 106, the code-multiplexed signal output from the transmit antenna 106 will vary for each radar positioning performed, thereby achieving an effect of randomizing the influence of the interference.

[0151] The configuration example of the radar transmitting unit 100 has been described above.

[0152] [Configuration of Radar Receiver 200]

[0153] exist Figure 1 In the figure, radar receiver 200 includes Na receive antennas 202 (e.g., also referred to as "Rx#1 to Rx#Na"), forming an array antenna. Furthermore, radar receiver 200 includes Na antenna system processing units 201-1 to 201-Na, a CFAR (Constant False Alarm Rate) unit 211, an aliasing determination unit 212, a code multiplexing separation unit 213, and a direction estimation unit 214.

[0154] Each receiving antenna 202 receives a reflected wave signal, which is a radar transmission signal reflected by a target, and outputs the received reflected wave signal to the corresponding antenna system processing unit 201 as a received signal.

[0155] Each antenna system processing unit 201 includes a wireless receiving unit 203 and a signal processing unit 206 .

[0156] The wireless receiving unit 203 includes a mixer unit 204 and an LPF (low pass filter) 205. The mixer unit 204 mixes the received reflected wave signal with the chirp signal input from the radar transmission signal generating unit 101. The LPF 205 performs LPF processing on the output signal of the mixer unit 204, thereby outputting a beat signal whose frequency corresponds to the delay time of the reflected wave signal. For example, Figure 2 As shown in the lower layer of , the difference frequency between the frequency of the transmitted chirp signal (transmitted frequency modulated wave) and the frequency of the received chirp signal (received frequency modulated wave) is obtained as the beat frequency.

[0157] The signal processing unit 206 of each antenna system processing unit 201 - z (where z=one of 1 to Na) includes an AD (Analog Digital) converter 207 , a beat frequency analyzer 208 , an output switch 209 , and a Doppler analyzer 210 .

[0158] The signal (for example, the beat signal) output from the LPF 205 is converted into discrete sampling data by the AD converter 207 in the signal processing unit 206 , which is discretely sampled.

[0159] The beat frequency analysis unit 208 analyzes the N signals obtained within a predetermined time range (range gate) according to the transmission cycle Tr. data The signal processing unit 206 then outputs a spectrum where the beat frequency corresponding to the delay time of the reflected wave signal (radar reflected wave) has a peak value. In addition, as an FFT process, the beat frequency analysis unit 208 may also multiply the beat frequency by a window function coefficient such as a Han window or a Hamming window. In addition, by using the window function coefficient, the radar device 10 can suppress the side lobes generated around the beat frequency peak value. In addition, in N data When the number of discrete sampling data is not a power of 2, the beat frequency analysis unit 208 may perform FFT processing using an FFT size that is a power of 2, for example, by including zero-padded data.

[0160] Here, the beat frequency response obtained by the m-th chirped pulse transmission and output from the beat frequency analysis unit 208 in the z-th signal processing unit 206 is represented by the RFT z (f b ,m). Here, f b represents the beat frequency index, which corresponds to the index of FFT (binary number). b =0,…,N data / 2, z=0,...,Na,m=1,...,N C . Beat frequency index f bThe smaller the beat frequency, the shorter the delay time of the reflected wave signal (in other words, the closer the distance to the target object).

[0161] In addition, the beat frequency index f b The following formula (8) can be used to convert the distance information R(f b ). Therefore, the beat frequency index f is also b It is called the “distance index f b ”.

[0162]

[0163] Here, B w represents the frequency modulation bandwidth of the chirp signal within the range gate, and C0 represents the speed of light.

[0164] The output switching unit 209 selectively switches the output of the beat frequency analysis unit 208 for each transmission cycle to the OC_INDEX-th Doppler analysis unit 210 among the Loc Doppler analysis units 210 based on the orthogonal code element index OC_INDEX output from the code generation unit 104. In other words, the output switching unit 209 selects the OC_INDEX-th Doppler analysis unit 210 in the m-th transmission cycle Tr.

[0165] The signal processing unit 206 includes Loc Doppler analysis units 210-1 to Doppler analysis units 210-Loc. For example, the output switching unit 209 inputs data to the noc-th Doppler analysis unit 210 at each Loc transmission cycle (Loc×Tr). Therefore, the noc-th Doppler analysis unit 210 uses data from the Ncode transmission cycle out of the Nc transmission cycles (e.g., the beat frequency response RFT output from the beat frequency analysis unit 208). z (f b ,m)), indexed by distance f b Perform Doppler analysis. Here, noc is the index of the coding element, noc=1, ..., Loc.

[0166] For example, when Ncode is a power of 2, FFT processing can also be applied to Doppler analysis. In this case, the FFT size is Ncode, and the maximum Doppler frequency that does not cause aliasing is ±1 / (2Loc×Tr) derived from the sampling theorem. In addition, the Doppler frequency index f s The Doppler frequency interval is 1 / (Ncode×Loc×Tr), and the Doppler frequency index f s The range of f s =-Ncode / 2,…,0,…,Ncode / 2-1.

[0167] For example, the output VFT of the Doppler analysis unit 210 of the z-th signal processing unit 206 is z noc (f b ,f s ) is represented by the following formula (9). In addition, j is an imaginary unit, and z = 1 to Na.

[0168]

[0169] In addition, when Ncode is not a power of 2, for example, by including zero-padded data, FFT processing can be performed as a power of 2 data size (FFT size). For example, when the FFT size in the Doppler analysis unit 210 is set to N when the zero-padded data is included, codewzero In the case of the z-th signal processing unit 206, the output VFT of the Doppler analysis unit 210 z noc (f b ,f s ) is represented by the following formula (10).

[0170]

[0171] Here, noc is the index of the coding element, noc = 1, ..., Loc. In addition, the FFT size is N codewzero According to the sampling theorem, the maximum Doppler frequency that does not cause aliasing is ±1 / (2Loc×Tr). In addition, the Doppler frequency index f s The Doppler frequency interval is 1 / (N codewzero ×Loc×Tr), Doppler frequency index f s The range of f s =-N codewzero / 2,…,0,…,N codewzero / 2-1.

[0172] Hereinafter, as an example, a case where Ncode is a power of 2 will be described. In addition, when zero padding is used in the Doppler analysis unit 210, in the following description, Ncode is replaced by N codewzero , can be applied in the same way and can achieve the same effect.

[0173] Furthermore, the Doppler analysis unit 210 may multiply the FFT processing by a window function coefficient such as a Hanning window or a Hamming window. By applying the window function, the radar device 10 can suppress side lobes generated around the beat frequency peak.

[0174] The above describes the processing in each component of the signal processing unit 206.

[0175] exist Figure 1 The CFAR unit 211 performs CFAR processing (in other words, performs adaptive threshold determination) using the outputs of the Loc Doppler analysis units 210 of the first to Nath signal processing units 206, and extracts the range index f that gives the peak signal. b_cfar and the Doppler frequency index f s_cfar .

[0176] The CFAR unit 211 converts the outputs of the Doppler analysis unit 210 of the first to Na-th signal processing units 206 into VFTs, for example, as shown in the following equation (11). z noc (f b ,f s ) are added to perform two-dimensional CFAR processing including the range axis and the Doppler frequency axis (equivalent to relative velocity), or CFAR processing combined with one-dimensional CFAR processing. For example, the processing disclosed in Non-Patent Document 2 can be applied to two-dimensional CFAR processing or CFAR processing combined with one-dimensional CFAR processing.

[0177]

[0178] The CFAR unit 211 adaptively sets a threshold and indexes the distance where the received power is greater than the threshold. b_cfar , Doppler frequency index f s_cfar and received power information PowerFT(f b_cfar ,f s_cfar ) is output to the aliasing determination unit 212.

[0179] Next, explain Figure 1 FIG. 2 shows an example of the operation of the aliasing determination unit 212 .

[0180] The aliasing determination unit 212 is based on, for example, the distance index f extracted in the CFAR unit 211. b_cfar and the Doppler frequency index f s_cfar The output of the Doppler analysis unit 210, that is, the Doppler component VFT z noc (f b_cfar ,f s_cfar ) to perform aliasing determination. Here, z = 1, ..., Na, noc = 1, ..., Loc.

[0181] The aliasing determination unit 212 performs Doppler aliasing determination processing, assuming that the target Doppler range is ±1 / (2×Tr), for example.

[0182] Here, for example, when Ncode is a power of 2, the Doppler analysis unit 210 applies FFT processing to each code element, and therefore performs FFT processing using the output from the beat frequency analysis unit 208 in a (Loc×Tr) cycle. Therefore, the Doppler range of the Doppler analysis unit 210 that does not cause aliasing according to the sampling theorem is ±1 / (2Loc×Tr).

[0183] Thus, the aliasing determination unit 212 assumes a target Doppler range that is larger than the Doppler range where aliasing does not occur in the Doppler analysis unit 210. For example, the aliasing determination unit 212 assumes a Doppler range of ±1 / (2×Tr) times the Doppler range where aliasing does not occur in the Doppler analysis unit 210, and performs aliasing determination processing.

[0184] An example of aliasing determination processing in the aliasing determination unit 212 will be described below.

[0185] Here, as an example, the following case is described, that is, the coding multiplexing number N CM =3, and the code generation unit 104 uses three orthogonal codes Code1=WH4(3)=[1,1,-1,-1], Code2=WH4(4)=[1,-1,-1,1] and Code3=WH4(2)=[1,-1,1,-1] in the Walsh-Hadamard code with code length Loc=4.

[0186] The aliasing determination unit 212 sets the N allcode = 1 of 4 Walsh-Hadamard codes (= N allcode -N CM ) Orthogonal codes are not used for aliasing determination. For example, when the number of coded multiplexing is N CM =3, and the codes used for coded multiplexing transmission determined by the code generation unit 104 are Code1 = WH4(3) = [1, 1, -1, -1], Code2 = WH4(4) = [1, -1, -1, 1], and Code3 = WH4(2) = [1, -1, 1, -1], the unused orthogonal code is UnCode1 = WH4(1) = [1, 1, 1, 1].

[0187] For example, when radar apparatus 10 performs code-multiplexed transmission using an orthogonal code with a code length of Loc = 4, as described above, Doppler analysis unit 210 applies FFT processing to each code element. Therefore, FFT processing is performed using the output from beat frequency analysis unit 208 with a period of (Loc × Tr) = (4 × Tr). Consequently, the Doppler range of Doppler analysis unit 210, which prevents aliasing according to the sampling theorem, is ±1 / (2Loc × Tr) = ±1 / (8 × Tr).

[0188] The aliasing determination unit 212 performs aliasing determination within a range of Loc times the code length of the orthogonal code sequence, compared to the range of Doppler analysis (Doppler range) in the Doppler analysis unit 210. For example, the aliasing determination unit 212 performs aliasing determination processing assuming a Doppler range of 4 times (= Loc) of the Doppler range of ±1 / (8×Tr) where aliasing does not occur (= ±1 / (2×Tr)).

[0189] Here, the distance index f extracted in the CFAR unit 211 is b_cfar and the Doppler frequency index f s_cfar The Doppler component VFT is the output of the corresponding Doppler analysis unit 210. z noc (f b_cfar , f s_cfar ) For example, it is possible to include the following within the Doppler range of ±1 / (2×Tr): Figure 3 (a) and Figure 3 (b) shows the Doppler component including aliasing.

[0190] For example, Figure 3 As shown in (a), at f s_cfar When <0, within the Doppler range of ±1 / (2×Tr), there may be f s_cfar -Ncode,f s_cfar 、f s_cfar +Ncode and f s_cfar +2Ncode, these 4 (=Loc) Doppler components.

[0191] In addition, for example, Figure 3 As shown in (b), at f s_cfar When > 0, within the Doppler range of ±1 / (2×Tr), there may be f s_cfar -2Ncode、f s_cfar -Ncode,f s_cfar and f s_cfar +Ncode these 4 (=Loc) Doppler components.

[0192] The aliasing determination unit 212 uses, for example, unused orthogonal codes. Figure 3 For example, the aliasing determination unit 212 may also perform code separation processing on the non-orthogonal code. Figure 3 The phase changes of the four (=Loc) Doppler components shown, including aliasing, are corrected.

[0193] Next, the aliasing determination unit 212 determines whether each Doppler component contains aliasing based on the received power of the Doppler components that have been code-separated without using orthogonal codes. For example, the aliasing determination unit 212 detects the Doppler component with the lowest received power among the Doppler components containing aliasing and determines the detected Doppler component as a true Doppler component. In other words, the aliasing determination unit 212 determines Doppler components with received powers other than the minimum received power among the Doppler components containing aliasing as pseudo-Doppler components.

[0194] This aliasing determination process reduces the ambiguity of the Doppler range containing aliasing. Furthermore, this aliasing determination process expands the range in which the Doppler frequency can be unambiguously detected to a range greater than -1 / (2Tr) and less than 1 / (2Tr), compared to the Doppler range of the Doppler analysis unit 210.

[0195] By performing code separation based on unused orthogonal codes, for example, the phase variation of the true Doppler component is accurately corrected, maintaining the orthogonality between the orthogonal codes used for code-multiplexed transmission and the unused orthogonal codes. Consequently, the unused orthogonal codes are uncorrelated with the code-multiplexed transmission signal, and the received power is approximately at noise level.

[0196] On the other hand, for example, the pseudo-Doppler component's phase change is incorrectly corrected, and the orthogonality between the orthogonal code used for code-multiplexed transmission and the unused orthogonal code is not maintained. This generates a correlation component (interference component) between the unused orthogonal code and the code-multiplexed transmission signal, enabling detection of received power greater than the noise level, for example.

[0197] As described above, the aliasing determination unit 212 can thereby determine the Doppler component with the minimum received power among the Doppler components code-separated without using orthogonal codes as a true Doppler component and determine other Doppler components with received powers different from the minimum received power as pseudo-Doppler components.

[0198] For example, the aliasing determination unit 212 corrects the phase change of the Doppler component including the aliasing based on the output of the Doppler analysis unit 210 in each antenna system processing unit 201, and calculates the phase change using the unused orthogonal code UnCode according to the following formula (12): nuc The received power after coding separation DeMulUnCode nuc (f b_cfar , f s_cfar , DR).

[0199]

[0200] In equation (12), the unused orthogonal code UnCode is calculated for the output of the Doppler analysis unit 210 in all antenna system processing units 201. nuc The sum of the received power after code separation using the orthogonal code is calculated. This improves the accuracy of aliasing determination even when the received signal level is low. However, instead of using equation (12), the received power after code separation using non-orthogonal codes may be calculated for the output of the Doppler analysis unit 210 in some antenna system processing units 201. Even in this case, for example, it is possible to maintain aliasing determination accuracy within a sufficiently high received signal level range and reduce the amount of computational processing.

[0201] In addition, in formula (12), nuc=1, ..., N allcode -N CM DR is an index indicating the Doppler aliasing range, and for example, takes an integer value in the range of DR=ceil[-Loc / 2], ceil[-Loc / 2]+1, ..., 0, ..., ceil[Loc / 2]-1.

[0202] In addition, in formula (12), the operator It represents the product of each element of vectors with the same number of elements. For example, for the n-th order vector A=[a1,..,a n ] and B=[b1,..,b n ], the product of each element is expressed by the following formula (13).

[0203]

[0204] In Formula (12), the operator "·" represents a vector inner product operator. In Formula (12), the superscript T represents a vector transposition, and the superscript * (asterisk) represents a complex conjugate operator.

[0205] In formula (12), α(f s_cfar ) represents the “Doppler phase correction vector”. s_cfar ), for example, when the output range of the Doppler analysis unit 210 (in other words, the Doppler range) is set to not include Doppler aliasing, the Doppler frequency index f extracted by the CFAR unit 211 is s_cfar The Doppler phase rotation amount caused by the time difference of the Doppler analysis between the Loc number of Doppler analysis units 210 is corrected.

[0206] For example, the Doppler phase correction vector α(f s_cfar ) is expressed as the following equation (14). The Doppler phase correction vector α(f s_cfar) is a vector with a Doppler phase correction coefficient as an element. The Doppler phase correction coefficient corrects the following phase rotation amount, which is the output VFT of the first Doppler analysis unit 210. z 1 (f b_cfar ,f s_cfar ) is used as the basis for the Doppler analysis time, and the output VFT of the second Doppler analysis unit 210 is used as the basis for the Doppler analysis time. z 2 (f b_cfar ,f s_cfar ) to the Locth Doppler analysis unit VFT z Loc (f b_cfar ,f s_cfar The Doppler frequency index f generated by the time delay of Tr, 2Tr, ..., (Loc-1)Tr in each output of s_cfar The amount of phase rotation in the Doppler component of .

[0207]

[0208] In equation (12), β(DR) represents an "aliasing phase correction vector." Taking into account the presence of Doppler aliasing, the aliasing phase correction vector β(DR) corrects the Doppler phase rotation by an integer multiple of 2π, for example, caused by the time difference in Doppler analysis between the Loc Doppler analysis units 210.

[0209] For example, the aliasing phase correction vector β(DR) is expressed as shown in the following equation (15).

[0210]

[0211] For example, when Loc=4, taking integer values ​​of DR=-2, -1, 0, 1, the aliasing phase correction vector β(DR) is expressed as shown in Equations (16), (17), (18), and (19).

[0212] β(-2)=[1,-1,1,-1] (16)

[0213]

[0214] β(0)=[1,1,1,1] (18)

[0215]

[0216] For example, in the case of Loc=4, Figure 3 (a) or Figure 3 The Doppler frequency index f is detected as the output of the Doppler analysis unit 210 in (b).s_cfar The Doppler range of the Doppler component of DR = 0 (eg, -1 / 8 Tr to +1 / 8 Tr) corresponds to DR = 0. In addition, according to the Doppler frequency index f for DR = 0 s_cfar The Doppler phase rotation of an integer multiple of 2π (e.g., β(1), β(-1), and β(-2)) is performed to calculate the Doppler component corresponding to the Doppler range of DR=1 (e.g., 1 / 8Tr to 3 / 8Tr), the Doppler component corresponding to the Doppler range of DR=-1 (e.g., -3 / 8Tr to -1 / 8Tr), and the Doppler component corresponding to the Doppler range of DR=-2 (e.g., -1 / 2Tr to -3 / 8Tr and 3 / 8Tr to 1 / 2Tr).

[0217] In addition, in formula (12), VFTALL z (f b_cfar ,f s_cfar ) For example, the output VFT of the Loc Doppler analysis unit 210 in the z-th antenna system processing unit 201 is expressed in vector form as follows (20): z noc (f b ,f s ) and the distance index f extracted in the CFAR unit 211 b_cfar and the Doppler frequency index f s_cfar Corresponding component VFT z noc (f b_cfar ,f s_cfar )(where noc=1,…,Loc).

[0218] VFTALL z (f b_cfar ,f s_cfar )=[VFT z 1 (f b_cfar ,f s_cfar ),VFT z 2 (f b_cfar ,f s_cfar )...,VFT z Loc (f b_cfar ,f s_cfar )] (20)

[0220] For example, the aliasing determination unit 212 calculates the unused orthogonal code UnCode using the phase change correction for the Doppler component including the aliasing in the range of DR = ceil[-Loc / 2], ceil[-Loc / 2]+1, ..., 0, ..., ceil[Loc / 2]-1 according to equation (12). nuc The received power after coding separation DeMulUnCode nuc (f b_cfar ,f s_cfar ,DR).

[0221] Next, the aliasing determination unit 212 detects the received power DeMulUnCode in the range of each DR. nuc (f b_cfar ,f s_cfar ,DR) with the smallest DR. Next, as shown in the following formula (21), the received power in the range of each DR is DeMulUnCoded nuc (f b_cfar ,f s_cfar ,DR) The minimum DR is recorded as "DR min ”.

[0222]

[0223] The following describes the reason why Doppler aliasing determination can be performed by the aliasing determination process described above.

[0224] For example, if the noise component is ignored, the VFTALL shown in equation (20) is z (f b_cfar ,f s_cfar ) is expressed as follows in the form of equation (22) and transmitted from the ncm-th transmitting antenna 106 (for example, Tx#ncm).

[0225]

[0226] Here, γz,ncm represents the complex reflection coefficient when the radar transmission signal transmitted from the ncm-th transmission antenna 106 is reflected by the target and received by the z-th antenna system processing unit 201. true Indicates the following index, which represents the true Doppler aliasing range. DR true The index value is set to be in the range of ceil[-Loc / 2], ceil[-Loc / 2]+1, ..., 0, ..., ceil[Loc / 2]-1. min =DR ture method to determine.

[0227] For the 1st to NthCM The radar transmission signal component sent from the transmitting antenna 106 uses the unused orthogonal code UnCode nuc The sum of the received power after coding separation PowDeMul(nuc,DR,DR true ) is represented by the following formula (23).

[0228]

[0229] In addition, PowDeMul(nuc,DR,DR) shown in formula (23) true ) is equivalent to the formula (12), The evaluation value of the item.

[0230] In equation (23), when DR=DRtrue, the orthogonal code UnCode is not used. nuc Orthogonal code Code used for coded multiplexing transmission ncm The correlation value between them is zero (for example, UnCode nuc * ·{Code ncm} T =0), therefore, PowDeMul(nuc,DR,DR true )=0.

[0231] On the other hand, in formula (23), when DR≠DRtrue, the output depends on Orthogonal code Code used for coded multiplexing transmission ncm The correlation value between PowDeMul(nuc,DR,DR true ). Here, in all UnCode nuc PowDeMul(nuc,DR,DR true ) is not zero, for example, as long as the following formula (24) is satisfied, when DR=DR true In the case of PowDeMul(nuc,DR true ,DR true ) is the smallest, the aliasing determination unit 212 can detect DR true (=DR min ). In other words, the aliasing determination unit 212 can perform Doppler aliasing determination according to equation (12).

[0232]

[0233] For example, in order to satisfy formula (24), as long as The term does not overlap with other unused orthogonal codes UnCode nuc2 Here, nuc2≠nuc.

[0234] Therefore, when there is one unused orthogonal code, equation (24) is satisfied. In addition, when there are multiple unused orthogonal codes, for example, the code generation unit 104 may also make The code for coded multiplexing transmission is selected in a manner that is not consistent with other unused orthogonal codes.

[0235] Here, when using codes such as Walsh-Hadamard codes or orthogonal M-sequence codes, the orthogonal code of code length Loc may include a code group in which the odd-numbered code elements are the same and the signs of the even-numbered code elements are reversed.

[0236] On the other hand, since β(0)=[1,1,…,1],β(-Loc / 2)=[1,-1,1,-1,…1,-1], The items are converted to UnCode nuc The odd-numbered code elements are the same, while the signs of the even-numbered code elements are reversed.

[0237] Therefore, when the number of unused orthogonal codes (N allcode -N CM ) is more than 2, for example, the code generation unit 104 may select codes for code-multiplexed transmission or unused orthogonal codes in the following manner, that is, a group of codes in which the odd-numbered code elements and one of the even-numbered code elements between codes in the orthogonal code of the code length Loc are the same, and the signs of the other of the odd-numbered code elements and the even-numbered code elements are reversed is not included in the unused orthogonal codes.

[0238] For example, the Walsh-Hadamard code with code length Loc=4 includes WH4(1)=[1, 1, 1, 1] and WH4(2)=[1, -1, 1, -1]. or, Therefore, for example, the code generation unit 104 may select a code for code-multiplexed transmission or an unused orthogonal code so that the group of WH4(1) and WH4(2) is not included in the plurality of unused orthogonal codes. Furthermore, WH4(3) = [1, 1, -1, -1] and WH4(4) = [1, -1, -1, 1] also have the same relationship. Therefore, for example, the code generation unit 104 may select a code for code-multiplexed transmission or an unused orthogonal code so that the group of WH4(3) and WH4(4) is not included in the plurality of unused orthogonal codes.

[0239] In addition, when there are multiple unused orthogonal codes UnCode nuc In the case of receiving power DeMulUnCode, nuc (fb_cfar ,f s_cfar ,DR), the following formula (25) is used to use the received power DeMulUnCodeAll (f b_cfar ,f s_cfar ,DR).

[0240]

[0241] By obtaining the received power after code separation using all the unused orthogonal codes, the aliasing determination unit 212 can improve the aliasing determination accuracy even when the received signal level is low.

[0242] For example, the aliasing determination unit 212 calculates DeMulUnCodeAll(f in each range of DR=ceil[-Loc / 2], ceil[-Loc / 2]+1, ..., 0, ..., ceil[Loc / 2]-1. b_cfar ,f s_cfar ,DR), detect the received power DeMulUnCodeAll(f b_cfar ,f s_cfar ,DR) the smallest DR (in other words, DR min ). When using equation (25), the DR that gives the minimum received power within the DR range is described as “DR min ”.

[0243]

[0244] In addition, the aliasing determination unit 212 may also perform the following processing, for example, that is, when comparing the unused orthogonal code UnCode nuc The minimum receiving power after coding separation DeMulUnCode nuc (f b_cfar ,f s_cfar ,DR min ) and the received power, thereby determining (in other words, measuring) the certainty of the aliasing determination. In this case, the aliasing determination unit 212 can also determine the certainty of the aliasing determination according to the following equations (27) and (28), for example.

[0245] DeMulUnCode nuc (f b_cfar ,f s_cfar ,DR min )<Threshold DR ×PowerFT(f b_cfar ,f s_cfar ) (27)

[0247] DeMulUnCode nuc (f b_cfar ,f s_cfar ,DR min )≥Threshod DR ×PowerFT(f b_cfar ,f s_cfar ) (28)

[0249] For example, when using the unused orthogonal code UnCode nuc The minimum receiving power after coding separation DeMulUnCode nuc (f b_cfar ,f s_cfar ,DR min ) is less than the distance index f extracted in the CFAR unit 211 b_cfar and the Doppler frequency index f s_cfar The received power value PowerFT(f b_cfar ,f s_cfar ) multiplied by the specified value Threshold DR When the obtained value is (for example, equation (27)), the aliasing determination unit 212 determines that the aliasing determination is sufficiently certain. In this case, the radar device 10 performs the subsequent processing (for example, code separation processing), for example.

[0250] On the other hand, for example, when using the unused orthogonal code UnCode nuc The minimum receiving power after coding separation DeMulUnCode nuc (f b_cfar ,f s_cfar ,DR min ) is equal to or greater than the received power value PowerFT(f b_cfar ,f s_cfar ) multiplied by Threshold DR If the obtained value is (e.g., equation (28)), the aliasing determination unit 212 determines that the accuracy of aliasing determination is insufficient (e.g., a noise component). In this case, the radar device 10 may not perform subsequent processing (e.g., code separation processing).

[0251] By this process, the aliasing determination error in the aliasing determination unit 212 can be reduced, and the noise component can be removed. DR For example, it can be set in the range of greater than 0 and less than 1. As an example, if noise components are included, the Threshold can also be set in the range of about 0.1 to 0.5. DR .

[0252] In addition, when there are multiple unused orthogonal codes UnCode nuc In the case of aliasing determination unit 212, the following processing can also be performed, that is, instead of receiving power DeMulUnCode nuc (f b_cfar ,f s_cfar ,DR), use DeMulUnCodeAll(f b_cfar ,f s_cfar ,DR) is compared with the received power to determine (in other words, measure) the certainty of the aliasing determination. In this case, the aliasing determination unit 212 may also use DeMulUnCodeAll (f b_cfar ,f s_cfar ,DR) replaces DeMulUnCode in formula (27) and formula (28) nuc (f b_cfar ,f s_cfar By determining the received power after code separation using all unused orthogonal codes, the aliasing determination unit 212 can improve the accuracy of the aliasing determination even when the received signal level is low.

[0253] In addition, the unused orthogonal code UnCode is used instead of equation (12). nuc The received power after coding separation DeMulUnCode nuc (f b_cfar ,f s_cfar ,DR) can also be calculated as follows (29).

[0254]

[0255] In formula (29), The term does not depend on the index of the Doppler component (Doppler frequency index) f s Therefore, for example, by tabulating it in advance, the amount of calculation in the aliasing determination unit 212 can be reduced.

[0256] The above describes an example of the operation of the aliasing determination unit 212 .

[0257] Next, an operation example of the code multiplexing separation unit 213 will be described.

[0258] The code multiplexing separation unit 213 performs separation processing of the code multiplexed signal based on the aliasing determination result in the aliasing determination unit 212 and the code for code multiplexed transmission.

[0259] For example, the code multiplexing separation unit 213 uses the aliasing determination result in the aliasing determination unit 212, ie, DR, in the following formula (30).min The aliasing phase correction vector β(DR min ), and the distance index f extracted in the CFAR unit 211 b_cfar and the Doppler frequency index f s_cfar The Doppler component VFTALL is the output of the corresponding Doppler analysis unit 210. z (f b_cfar ,f s_cfar The aliasing determination unit 212 can determine the index of the true Doppler aliasing range within the Doppler range greater than -1 / (2Tr) and less than 1 / (2Tr) (in other words, it can make the DR min =DR true Therefore, in the code multiplexing separation unit 213, the correlation value between the orthogonal codes used for code multiplexing can be set to zero within a Doppler range greater than -1 / (2Tr) and less than 1 / (2Tr), thereby enabling separation processing that suppresses interference between code multiplexing signals.

[0260]

[0261] Here, DeMul z ncm (f b_cfar ,f s_cfar ) is the distance index f used for the Doppler analysis unit 210 in the z-th antenna system processing unit 201. b_cfar and the Doppler frequency index f s_cfar The output orthogonal code Code ncm , the output obtained by performing code separation on the code multiplexed signal (for example, the code separation result). In addition, z = 1, ..., Na, ncm = 1, ..., N CM .

[0262] Furthermore, the code multiplexing separation unit 213 may use the following equation (31) instead of equation (30).

[0263]

[0264] In formula (31), Term (but in formula (31), DR = DR min ) does not depend on the index of the Doppler component (e.g., the Doppler frequency index) f s Therefore, for example, by tabulating it in advance, the amount of calculation in the coding multiplexing separation unit 213 can be reduced.

[0265] Through the code separation process described above, the radar device 10 can obtain the orthogonal code Code assigned to the ncm-th transmitting antenna Tx#ncm based on the result of the aliasing determination performed by the aliasing determination unit 212 based on the Doppler range ±1 / (2Loc×Tr) of the Loc times the Doppler range where aliasing does not occur in the Doppler analysis unit 210. ncm The signal obtained by separating the signal sent by code multiplexing.

[0266] Furthermore, the radar device 10 performs Doppler phase correction including Doppler aliasing (for example, based on the aliasing phase correction vector β (DR )) on the output of the Doppler analysis unit 210 for each code element during the code separation process. min ) processing). Therefore, mutual interference between code-multiplexed signals can be reduced to, for example, the noise level. In other words, in the radar device 10, inter-symbol interference can be reduced, thereby suppressing the impact of degradation on the detection performance of the radar device 10.

[0267] Figure 4 Another configuration example of the radar device 10 is shown. Figure 1 In the configuration of the radar device 10 shown in FIG. , as shown in equations (12), (29), (30), and (31), The items are used in common by the aliasing determination unit 212 and the coding multiplexing separation unit 213. For example, Figure 4 The radar device 10a shown in the figure includes a phase correction unit 215, which can convert the Doppler component VFTALL z (f b_cfar ,f s_cfar ) multiplied by the Doppler phase correction vector α(f s_cfar ) The aliasing determination unit 212a and the code multiplexing separation unit 213a may output the aliasing determination unit 212a and the code multiplexing separation unit 213a. By performing calculations on the terms, repeated calculations of the above terms can be reduced in the radar device 10a.

[0268] The above describes an example of the operation of the code multiplexing separation unit 213.

[0269] exist Figure 1 In the example, the direction estimation unit 214 calculates the distance index f based on the distance index f input from the code multiplexing separation unit 213. b_cfar , Doppler frequency index f s_cfar The code separation result DeMul outputted from the corresponding Doppler analysis unit 210 z ncm (f b_cfar ,f s_cfar), and perform target direction estimation processing.

[0270] For example, the direction estimation unit 214 generates the virtual receiving array correlation vector h(f b_cfar ,f s_cfar ), and perform direction estimation processing.

[0271] Virtual receiving array correlation vector h(f b_cfar ,f s_cfar ) contains the product of the number of transmitting antennas Nt and the number of receiving antennas Na, that is, Nt×Na elements. The virtual receiving array correlation vector h(f b_cfar ,f s_cfar ) is used to perform direction estimation processing on the reflected wave signal from the target based on the phase difference between each receiving antenna 202. Here, z = 1, ..., Na.

[0272]

[0273] The direction estimation unit 214 changes the direction estimation evaluation function value P within a predetermined angle range, for example. H (θ,f b_cfar ,f s_cfar The direction estimation unit 214 extracts a predetermined number of maximum peaks from the calculated spatial distribution in descending order and outputs the direction of the maximum peak as an estimated direction of arrival value (e.g., positioning output).

[0274] In addition, the direction estimation evaluation function value P H (θ,f b_cfar ,f s_cfar ) There are various methods depending on the arrival direction estimation algorithm. For example, the estimation method using an array antenna disclosed in Non-Patent Document 3 can also be used.

[0275] For example, in Nt×Na virtual receiving arrays with equal intervals d H When the arrangement is linear, the beamforming method can be expressed as in the following equations (33) and (34). In addition, methods such as Capon and MUSIC can also be applied in the same manner.

[0276] P H (θ u ,f b_cfar ,f s_cfar )=|a H (θ u )D cal h(f b_cfar ,f s_cfar )| 2 (33)

[0277]

[0278] Here, the superscript H is the Hermitian transpose operator. u ) represents the direction relative to θ u The direction vector of the arriving wave to the virtual receiving array.

[0279] In addition, the azimuth direction θ u is a vector that changes at azimuth intervals DStep within the azimuth range θmin to θmax for direction of arrival estimation. For example, θ can be set as follows u .

[0280] θ u =θmin+uDStep、u=0,…,NU

[0281] NU=floor[(θmax-θmin) / DStep]

[0282] Here, floor(x) is a function that returns the maximum integer value that does not exceed the real number x.

[0283] In addition, in formula (33), D cal It is a square matrix of (Nt×Na) order, including array correction coefficients for correcting phase and amplitude deviations between transmit antennas and receive array antennas, and coefficients for reducing the influence of inter-element coupling between antennas. When the coupling between antennas in the virtual receive array can be ignored, D cal It becomes a diagonal matrix, and the diagonal components include array correction coefficients for correcting phase deviations and amplitude deviations between transmitting array antennas and receiving array antennas.

[0284] The direction estimation unit 214 may also output the direction estimation result, and further output the direction estimation result based on the distance index f b_cfar The distance information, the Doppler frequency index f based on the target b_cfar and the determination result DR of the aliasing determination unit 212 min The Doppler velocity information of the target is used as the positioning result.

[0285] The direction estimation unit 214 may also be configured based on the Doppler frequency index f s_cfar and the determination result of the aliasing determination unit 212, ie, DR min , calculate the Doppler frequency index f according to formula (35) es_cfar . Doppler frequency index f es_cfar For example, it corresponds to the Doppler index when the FFT size of the Doppler analysis unit 210 is expanded to Loc×Ncode. es_cfar It is called the "Extended Doppler Frequency Index".

[0286] f es_cfar =f s_cfar +DR min ×Ncode (35)

[0287] Furthermore, assuming a Doppler range up to ±1 / (2×Tr), the extended Doppler frequency index f corresponding to this Doppler range is es_cfar The range is -Loc×Ncode / 2≦f es_cfar <Loc×Ncode / 2, therefore, the calculation result in formula (35) is f es_cfar <-Loc×Ncode / 2, f es_cfar +Loc×Ncode is set to f es_cfar In addition, in f es_cfar ≧Loc×Ncode / 2, f es_cfar -Loc×Ncode is set to f es_cfar .

[0288] In addition, the Doppler frequency information can also be converted into a relative velocity component and output. es_cfar Converted into relative velocity component v d (f es_cfar ), the following formula (36) can also be used for conversion. Here, λ is the wavelength of the carrier frequency of the RF (Radio Frequency) signal output from the wireless transmitter (not shown). In addition, Δ f is the Doppler frequency interval in the FFT processing of the Doppler analysis unit 210. For example, in this embodiment,

[0289] Δ f =1 / {Loc×N code ×T r}.

[0290]

[0291] Figure 5 (A) Figure 5 (B) Figure 5 (C) Figure 5 (D) and Figure 6 An example of computer simulation results of aliasing determination processing is shown.

[0292] Figure 5 (A) Figure 5 (B) Figure 5 (C) and Figure 5 As an example, the coding multiplexing number is N CM=3, the code length is Loc=4, and the codes used for coded multiplexing transmission determined by the code generation unit 104 are Code1=WH4(3)=[1,1,-1,-1], Code2=WH4(4)=[1,-1,-1,1] and Code3=WH4(2)=[1,-1,1,-1], and the unused orthogonal code is UnCode1=WH4(1)=[1,1,1,1].

[0293] Figure 5 (A) Figure 5 (B) Figure 5 (C) and Figure 5 (D) indicates that, for example, in the aliasing determination unit 212, for the distance index f extracted in the CFAR 211 unit b and the Doppler frequency index f s_cfar The goal is to calculate the received power DeMulUnCode1 (f b , f s_cfar , DR) results.

[0294] also, Figure 5 (A) shows the DeMulUnCode1 (f b , f s_cfar , DR) calculation results example, Figure 5 (B) shows the DeMulUnCode1 (f b , f s_cfar , DR) calculation results example, Figure 5 (C) shows the DeMulUnCode1 (f b , f s_cfar , DR) calculation results example, Figure 5 (D) is a plot of DeMulUnCode1 (f b , f s_cfar , DR) calculation results example.

[0295] In addition, Figure 5 (A) Figure 5 (B) Figure 5 (C) and Figure 5 In (D), the target Doppler frequency ftarget can be changed within the range of -fr / 2 or more and less than fr / 2, and each DeMulUnCode1(f b , f s , DR).

[0296] Figure 5 (A) Figure 5 (B) Figure 5 (C) and Figure 5 (D) shows the target Doppler frequency ftarget on the horizontal axis and DeMulUnCode1 (ftarget) on the vertical axis. b , f s_cfar , DR). In addition, for example, fr=1 / Tr is set.

[0297] Figure 6 For example, according to the following example, Figure 5 (A) Figure 5 (B) Figure 5 (C) and Figure 5 The detection result shown in (D) is that the received power DeMulUnCode1 (f b , f s_cfar ,DR) The smallest DR is DR min An example of plotting with the vertical axis.

[0298] according to Figure 5 (A) Figure 5 (B) Figure 5 (C) Figure 5 (D) and Figure 6 , when the Doppler frequency range of the target is -fr / 8≦ftarget<fr / 8, DR min = 0, when the Doppler frequency range of the target is fr / 8≦ftarget<3fr / 8, DR min =1, when the Doppler frequency range of the target is -3fr / 8≦ftarget<-fr / 8, DR min = -1, when the Doppler frequency range of the target is -fr / 2≦ftarget<-3fr / 8 and 3r / 8≦ftarget<fr / 2, DR min = -2. This is similar to the above Figure 3 (a) or Figure 3 (b) is consistent with the description of Figure 6 DR shown min As a result, it can be seen that the Doppler range corresponding to the target Doppler frequency range ftarget can be accurately detected.

[0299] in addition, Figure 7 is a diagram showing the following example, which is based on Figure 6 As a result, the direction estimation unit 214 plots the Doppler frequency index fs_char and the determination result in the aliasing determination unit 212, ie, DR min , calculate the extended Doppler frequency index f according to formula (35) es_cfar The result obtained.

[0300] Figure 7 The following example shows that when the Doppler frequency ftarget of the target shown on the horizontal axis is changed within the range of -fr / 2 or more and less than fr / 2, the calculated extended Doppler frequency index f es_cfar It is converted to Doppler frequency ftarget_est and plotted on the vertical axis.

[0301] In addition, the Doppler frequency index f is expanded es_cfar When converting to the Doppler frequency ftarget_est, the Doppler frequency interval Δ obtained by the FFT process in the Doppler analysis unit 210 can be used. f That is, you can use ftarget_est=f es_cfar ×Δ f .

[0302] according to Figure 7 The results can be confirmed by using the extended Doppler frequency index f es_cfar The Doppler frequency ftarget_est can be accurately detected in a range where the target Doppler frequency ftarget is greater than or equal to -fr / 2 and less than fr / 2, that is, within a range where the target Doppler frequency ftarget is greater than or equal to -1 / (2Tr) and less than 1 / (2Tr).

[0303] As described above, in this embodiment, radar device 10 receives, at receiving antenna 202, reflected wave signals resulting from a radar transmission signal, which has been code-multiplexed and transmitted using a portion of a plurality of orthogonal code sequences, and then, in aliasing determination unit 212, determines Doppler frequency domain aliasing in the reflected wave signal using other orthogonal code sequences from the plurality of orthogonal code sequences that are different from the portion used for code-multiplex transmission. For example, in a MIMO radar employing code-multiplex transmission, radar device 10 performs code-multiplex transmission using an orthogonal code sequence having a code length that allows the generation of a greater number of orthogonal codes than the number of code-multiplex transmissions.

[0304] With this configuration, radar device 10, for example, uses orthogonal codes not used for code-multiplexed transmission to determine Doppler aliasing for received signals (e.g., the output of Doppler analysis unit 210 for each code element of a code-multiplexed signal). For example, radar device 10 can determine aliasing within a Doppler range that is multiple times the code length of the orthogonal code sequence, compared to the Doppler analysis range of Doppler analysis unit 210. Thus, according to this embodiment, radar device 10 can expand the Doppler range that can be unambiguously detected to a level equivalent to that achieved with single-antenna transmission.

[0305] In addition, when the radar device 10 performs code separation based on the judgment result of Doppler aliasing, for example, it is possible to suppress the mutual interference between the code multiplexed signals to the noise level by performing Doppler phase correction including aliasing, thereby suppressing the degradation of radar detection performance and performing code multiplexing transmission of MIMO radar.

[0306] Therefore, according to this embodiment, the radar device 10 suppresses the generation of mutual interference between the code-multiplexed signals, even in the case of Doppler fluctuations caused by the movement of the target or the radar device 10, and sets the Doppler range that can be detected without ambiguity to the same Doppler range as when transmitting with a single antenna, thereby improving the target detection accuracy within a wider Doppler frequency range.

[0307] (Implementation Method 2)

[0308] In the first embodiment, the following case is described (for example, referring to Figure 2 ), in the radar device 10, when chirped pulses are repeatedly transmitted Nc times as radar transmission signals, the center frequency of the chirped signal is kept constant. However, the center frequency of the chirped signal is not limited to being constant.

[0309] In this embodiment, a case where the center frequency of the chirp signal is variably set will be described.

[0310] [Structure of radar device]

[0311] Figure 8 1 is a block diagram showing a configuration example of a radar device 10b according to this embodiment. Figure 8 In the embodiment 1 ( Figure 1 ) The same structure is assigned the same figure mark and its description is omitted.

[0312] Next, for example, the radar device 10b transmits a radar transmission signal in which the center frequency fc of the chirp signal is changed by Δf at each transmission period Tr (for example, increased when Δf>0 and decreased when Δf<).

[0313] The radar transmission signal generation unit 101 b in the radar transmission unit 100 b includes a modulation signal generation unit 102 , a VCO 103 b , and a transmission frequency control unit 107 .

[0314] For example, the modulation signal generator 102 periodically generates a sawtooth-shaped modulation signal for VCO control. Here, the transmission period is Tr.

[0315] The transmission frequency control unit 107 controls the center frequency fc of the frequency modulation signal (chirp signal) output from the VCO 103b at the transmission period Tr. For example, the transmission frequency control unit 107 may change the center frequency fc of the frequency modulation signal by Δf at the transmission period Tr.

[0316] The VCO 103 b outputs a frequency modulation signal to the phase rotation unit 105 and the radar receiving unit 200 (for example, the mixer unit 204 ) based on the outputs of the transmission frequency control unit 107 and the modulation signal generation unit 102 .

[0317] Figure 9 An example of a frequency modulated signal (hereinafter referred to as a chirp signal) after frequency modulation is shown.

[0318] exist Figure 9 For example, in the first transmission cycle Tr#1, VCO103b outputs a chirp signal with a center frequency fc(1) of f0. Figure 9 As shown, VCO103b outputs a chirp signal with a center frequency fc(2) of f0+Δf in the second transmission cycle Tr#2. Figure 9 In the mth transmission cycle Tr#m, the VCO 103b outputs a chirp signal having a center frequency fc(m) of f0+(m-1)Δf. In this way, the VCO 103b changes the center frequency of the chirp signal by Δf at each transmission cycle Tr.

[0319] That is, in Figure 9 In the Nth c Transmission cycle Tr#N c The center frequency of the chirp signal fc(N c ) is f0+Δf×(N c -1).

[0320] Furthermore, each chirp signal can use, for example, the time range T of the range gate A The chirp signal has the same frequency modulation bandwidth Bw as in Figure 9 The example shown shows the case where Δf>0 (in other words, the case where the center frequency fc increases), but the same applies to the case where Δf<0 (in other words, the case where the center frequency fc decreases).

[0321] Figure 8 Other operations in the radar transmitting unit 100 b shown may be the same as those in the first embodiment.

[0322] Next, an example of the operation of the radar receiving unit 200b of the radar device 10b will be described.

[0323] In the radar receiver 200b, the processing of signals received by the receive antenna 202 by each antenna system processing unit 201, and the subsequent operations in the CFAR unit 211, aliasing determination unit 212, and code multiplexing separation unit 213 are similar to those in Embodiment 1. Furthermore, in the radar receiver 200b, the direction estimation processing performed by the direction estimation unit 214b using the output of the code multiplexing separation unit 213 is also similar to that in Embodiment 1.

[0324] In the radar receiving unit 200 b , for example, the conversion process of the Doppler velocity information on the target in the direction estimating unit 214 b is different from that in the first embodiment.

[0325] In addition, the distance information R(f b ) As in embodiment 1, the radar receiving unit 200b can use the beat frequency index (or distance index) f based on equation (8), for example. b Output distance information R(f b ).

[0326] The direction estimation unit 214b may also be configured based on the Doppler frequency index f s_cfar and the determination result of the aliasing determination unit 212, ie, DR min , calculate the Doppler frequency index f according to formula (37) es_cfar . Doppler frequency index f es_cfar For example, it corresponds to the Doppler index when the FFT size of the Doppler analysis unit 210 is expanded to Loc×Ncode. es_cfar It is called the "Extended Doppler Frequency Index".

[0327] f es_cfar =f s_cfar +DR min ×Ncode (37)

[0328] Furthermore, assuming a Doppler range up to ±1 / (2×Tr), the extended Doppler frequency index f corresponding to this Doppler range is es_cfar The range is -Loc×Ncode / 2≦f es_cfar <Loc×Ncode / 2, therefore, the calculation result in formula (37) is f es_cfar <-Loc×Ncode / 2, f es_cfar +Loc×Ncode is set to f es_cfarIn addition, in f es_cfar ≧Loc×Ncode / 2, f es_cfar -Loc×Ncode is set to f es_cfar .

[0329] In addition, the direction estimation unit 214b can use, for example, the extended Doppler frequency index f es_cfar and distance index f b_cfar , output the Doppler velocity information v of the detected target in the following way d .

[0330] For example, when using a radar transmission signal in which the center frequency fc of a chirp signal varies by Δf with a transmission period Tr, the chirp signal's center frequency fc varies with the transmission period Tr even when the target's relative velocity is zero. Therefore, the received signal from radar device 10b includes phase rotation caused by the variation in the chirp signal's center frequency with each transmission period Tr.

[0331] For the target distance R target The center frequency fc in the mth transmission cycle Tr changes by (m-1)Δf based on the first center frequency, taking into account the distance R from the target. target The arrival time of the reflected wave (2R target / Co), and the phase rotation amount Δη(m,R target ) is expressed by Equation (38-1). Furthermore, the following Equation (38-1) represents the relative phase rotation amount with respect to the phase of the first transmission period Tr. C0 represents the speed of light. Therefore, the output of each of the Loc Doppler analysis units 210 of the radar device 10b includes the phase rotation caused by the change in the center frequency of the chirp signal for each transmission period Tr.

[0332]

[0333] Thus, as shown in equation (38-2), the direction estimation unit 214b calculates the Doppler velocity information v based on the conversion equation that takes into account the change in the center frequency fc of the chirp signal per transmission period Tr, that is, Δf. d (f es_cfar ,f b_cfar ).

[0334] The first term in equation (38-2) corresponds to equation (36) and is the extended Doppler frequency index f es_cfarThe second term in equation (38-2) is the Doppler velocity component generated by changing the center frequency fc of the chirp signal by Δf according to the transmission period Tr. The direction estimation unit 214b can calculate the original relative Doppler velocity v of the target by removing the Doppler component of the second term from the first term, as shown in equation (38-2). d (f es_cfar ,f b_cfar ). Here, R(f b_cfar ) is the beat frequency index f according to formula (8) b_cfar The distance information R(f b_cfar ).

[0335]

[0336] In addition, since the Doppler range is assumed to be up to ±1 / (2×Tr), d v d <-C0 / (4f0Tr), the direction estimation unit 214b can output the Doppler velocity information v of the detected target according to the following formula (39): d .

[0337]

[0338] Similarly, since the Doppler range is assumed to be up to ±1 / (2×Tr), d v d >C0 / (4f0Tr), the direction estimation unit 214b can output the Doppler velocity information v of the detected target according to the following formula (40): d .

[0339]

[0340] As described above, in this embodiment, radar device 10b varies the center frequency fc of the chirp signal based on the transmission period Tr of the radar transmit signal. For example, radar device 10b varies the center frequency fc of the chirp signal by Δf (e.g., increasing when Δf > 0, decreasing when Δf <) according to the transmission period Tr and transmits this chirp signal as the radar transmit signal. Even in this case, radar device 10b (e.g., a MIMO radar) can employ code-multiplexed transmission. Furthermore, similar to Embodiment 1, radar device 10b can determine Doppler aliasing using the output of Doppler analysis unit 210 (in other words, the received signal) for each code element of the code-multiplexed signal and the absence of orthogonal codes.

[0341] Furthermore, similar to Embodiment 1, radar device 10b performs Doppler phase correction that includes aliasing during code separation, thereby reducing the unambiguously detectable Doppler frequency range to ±1 / (Tr) and suppressing mutual interference between code-multiplexed signals to approximately the noise level. Thus, according to this embodiment, it is possible to suppress degradation in radar detection performance and implement code-multiplexed transmission for MIMO radar.

[0342] Furthermore, in this embodiment, the radar device 10b transmits a radar transmission signal by varying the center frequency fc of the chirp signal by Δf according to a transmission period Tr. This allows for improved range resolution based on the amplitude of the chirp signal's center frequency variation (see, for example, Non-Patent Document 4). This embodiment improves range resolution based on the amplitude of the chirp signal's center frequency variation, allowing the chirp sweep bandwidth (e.g., Bw) to be reduced compared to transmitting the chirp signal with a constant center frequency. By reducing the chirp sweep bandwidth, for example, range resolution can be improved while shortening the transmission period Tr, further expanding the Doppler range that can be unambiguously detected during code-multiplexed transmission.

[0343] (Variation 1 of Implementation Method 2)

[0344] The period for changing the center frequency of the chirp signal is not limited to the transmission period Tr. In Variation 1, a case is described in which the center frequency of the chirp signal is set to be variable every transmission period (Loc × Tr) of the code length Loc of one orthogonal code used for code-multiplexed transmission (hereinafter referred to as the "coded transmission period").

[0345] [Structure of radar device]

[0346] Figure 10 1 is a block diagram showing a configuration example of a radar device 10c according to a modification 1. Figure 10 In the embodiment 1 ( Figure 1 ) or implementation method 2 ( Figure 8 ) The same structure is assigned the same figure mark and its description is omitted.

[0347] In variation 1, for example, the radar device 10c transmits a radar transmission signal in which the center frequency fc of the chirp signal is varied by Δf (for example, increased when Δf>0 and decreased when Δf<) at the code transmission period (Loc×Tr).

[0348] The radar transmission signal generating unit 101 c in the radar transmitting unit 100 c includes a modulation signal generating unit 102 , a VCO 103 c , and a transmission frequency control unit 107 c .

[0349] For example, the modulation signal generator 102 periodically generates a sawtooth-shaped modulation signal for VCO control. Here, the transmission period is Tr.

[0350] The transmission frequency control unit 107c controls the center frequency fc of the frequency modulation signal (chirp signal) output from the VCO 103c at the code transmission period (Loc×Tr) based on the orthogonal code element index OC_INDEX output from the code generation unit 104c.

[0351] For example, during a transmission cycle Tr where OC_INDEX = 1, the transmission frequency control unit 107c can vary the center frequency fc of the frequency modulated signal output from the VCO 103c by Δf per transmission cycle Tr. In other words, during a transmission cycle Tr where OC_INDEX ≠ 1, the transmission frequency control unit 107c controls the center frequency fc of the frequency modulated signal output from the VCO 103c in the same manner as the center frequency fc during the previous transmission cycle Tr. This control allows the transmission frequency control unit 107c to control the center frequency fc so that it varies by Δf per the code transmission cycle (Loc × Tr).

[0352] The VCO 103 c outputs a frequency modulation signal to the phase rotation unit 105 and the radar reception unit 200 (for example, the mixer unit 204 ) based on the outputs of the transmission frequency control unit 107 c and the modulation signal generation unit 102 .

[0353] Figure 11 An example of a frequency modulated signal (hereinafter referred to as a chirp signal) after frequency modulation is shown.

[0354] exist Figure 11 For example, in the first transmission cycle Tr#1 (e.g., OC_INDEX=1), VCO103c outputs a chirp signal with a center frequency fc(1) of f0. Figure 11 As shown, VCO 103c outputs a chirp signal with a center frequency fc(2) of f0 in the second transmission cycle Tr#2 (e.g., OC_INDEX=2). Similarly, VCO 103c outputs chirp signals with center frequencies fc(3) to fc(Loc) of f0 in the third transmission cycle (e.g., OC_INDEX=3, not shown) through the Loc-th transmission cycle Tr#Loc (e.g., OC_INDEX=Loc), respectively.

[0355] In the (Loc+1)th transmission cycle Tr#(Loc+1), VCO 103c outputs a chirp signal having a center frequency fc(Loc+1) of f0+Δf. Furthermore, in the (Loc+2)th transmission cycle Tr#(Loc+2) to the (2Loc)th transmission cycle Tr#(2Loc), VCO 103c outputs chirp signals having center frequencies fc(Loc+2) to fc(2Loc) of f0+Δf, respectively.

[0356] Similarly, in the mth transmission cycle Tr#m, the VCO 103c outputs a chirp signal having a center frequency fc(m) of f0+floor[(m-1) / Loc]Δf.

[0357] That is, in Figure 11 In the Nth c Transmission cycle Tr#N c The center frequency of the chirp signal fc(N c ) is f0+(Ncode-1)Δf. Here, Ncode=N c / Loc.

[0358] Furthermore, each chirp signal can use the time range T of the range gate, for example A The chirp signal has the same frequency modulation bandwidth Bw as in Figure 11 The example shown shows the case where Δf>0 (in other words, the case where the center frequency fc increases), but the same applies to the case where Δf<0 (in other words, the case where the center frequency fc decreases).

[0359] Figure 10 Other operations in the radar transmitting unit 100 c shown may be the same as those in the first embodiment.

[0360] Next, an example of the operation of the radar receiving unit 200 c of the radar device 10 c will be described.

[0361] In the radar receiver 200c, the processing of signals received by the receive antenna 202 by each antenna system processing unit 201 and the subsequent operation of the CFAR unit 211 are the same as those in Embodiment 1. Furthermore, in the radar receiver 200c, the direction estimation process performed by the direction estimation unit 214c using the output of the code multiplexing separation unit 213 is also the same as that in Embodiment 1.

[0362] In the radar receiving unit 200 c , for example, the operation of the aliasing determination unit 212 c , the operation of the code multiplexing separation unit 213 c , and the conversion process of the Doppler velocity information on the target in the direction estimation unit 214 c are different from those in the first embodiment.

[0363] Hereinafter, an example of operation of the aliasing determination unit 212 c that is different from that of the first embodiment will be described.

[0364] For example, when using a radar transmission signal in which the chirp signal's center frequency fc varies by Δf per the code transmission period (Loc × Tr), the chirp signal's center frequency fc varies per the code transmission period (Loc × Tr) even when the target's relative velocity is zero. Therefore, the output of each of the Loc Doppler analysis units 210 of the radar device 10c includes phase rotation caused by the variation in the chirp signal's center frequency per code transmission period (Loc × Tr).

[0365] That is, for the target distance R target The center frequency fc in the mth transmission cycle Tr is based on the center frequency fc in the first transmission cycle Tr and changes by floor[(m-1) / Loc]Δf. Therefore, considering the distance R from the target target The arrival time of the reflected wave (2R target / Co), the phase rotation amount Δη(m, R target ) is expressed by equation (40-1). In addition, equation (40-1) expresses the relative phase rotation amount when the phase of the first transmission cycle Tr is used as a reference. C0 represents the speed of light.

[0366]

[0367] Since the switching period of the Doppler analysis unit 210 for each code element is consistent with the code transmission period (Loc×Tr) that causes the center frequency fc of the chirp signal to change by Δf, each of the Loc Doppler analysis units 210 performs Doppler analysis including phase rotation shown in equation (40-1).

[0368] Therefore, the difference is that the aliasing determination unit 212c corrects the Doppler phase rotation caused by the time difference of Doppler analysis between the Loc Doppler analysis units 210 by dividing the Doppler phase correction vector α (f s_cfar ), the center frequency change correction vector ξ(f b_cfar ) corrects the phase. That is, the aliasing determination unit 212c uses Instead of α(f s_cfar ). In addition, R(f b_cfar ) is the beat frequency index f according to formula (8) b_cfar Distance information R(f b_cfar ).

[0369]

[0370] In formula (40-2), according to theb_cfar ) of the reflected wave arrival time (2R(f b_cfar ) / Co), the phase rotation amount becomes 2πΔf×(2R(f b_cfar ) / Co), therefore, the phase rotation caused by the time difference of Doppler analysis between the Loc Doppler analysis units 210 is derived for each noc-th Doppler analysis unit 210 by (noc-1) / Loc times, with the first Doppler analysis unit 210 as a reference. Furthermore, noc=1, ..., Loc.

[0371] By setting the period for changing the center frequency fc of the chirp signal by Δf to the code transmission period (Loc×Tr), the switching period of the Doppler analysis unit 210 is made consistent for each code element. Therefore, the aliasing determination unit 212c can easily perform phase correction (excluding the Doppler phase correction vector α(f)) in the separation process of the code-multiplexed signal using the unused code. s_cfar ), the center frequency change correction vector of formula (40-2) is also used.

[0372] Based on the above reasons, the aliasing determination unit 212c can replace equation (12) with equation (41) to calculate the unused orthogonal code UnCode. nuc The received power after coding separation DeMulUnCode nuc (f b_cfar ,f s_cfar , DR). The difference between formula (41) and formula (12) is that formula (41) uses To replace α(f s_cfar ). Here, nuc=1,…,N allcode -N CM DR is an index indicating the Doppler aliasing range, and takes an integer value in the range of DR=ceil[-Loc / 2], ceil[-Loc / 2]+1, ..., 0, ..., ceil[Loc / 2]-1.

[0373]

[0374] In addition, the aliasing determination unit 212c may use equation (42) instead of equation (29).

[0375]

[0376] Next, an example of operation in the code multiplexing separation unit 213c that is different from that in Embodiment 1 is described. In the code multiplexing separation unit 213c, for the same reason as in the above-mentioned explanation of the example of operation of the aliasing determination unit 212c, equation (43) is used instead of equation (30), and the aliasing determination result in the aliasing determination unit 212c, i.e., DR min , and the distance index f extracted in the CFAR unit 211 b_cfar and the Doppler frequency index f s_cfar The Doppler component VFTALL is the output of the corresponding Doppler analysis unit 210. z (f b_cfar ,f s_cfar ) is used for coding separation. The difference between formula (43) and formula (30) is that To replace α(f s_cfar ).

[0377]

[0378] In addition, the coding multiplexing separation unit 213c can also use equation (44) instead of equation (31) and use the aliasing determination result DR of the aliasing determination unit 212c min , and the distance index f extracted in the CFAR unit 211 b_cfar and the Doppler frequency index f s_cfar The Doppler component VFTALL is the output of the corresponding Doppler analysis unit 210. z (f b_cfar ,f s_cfar ) to separate the coded multiplexed signal. In formula (44), The term does not depend on the index f of the Doppler component s , so by pre-tabulating, the amount of calculation can be reduced.

[0379]

[0380] By aligning the period of change of the chirp signal center frequency fc by Δf with the code transmission period (Loc×Tr), the switching period of the Doppler analysis unit 210 for each code element can be aligned, making it easier to perform phase correction in the code multiplexing separation process.

[0381] Next, an example of operation of the direction estimation unit 214 c that is different from that of the first embodiment will be described.

[0382] The direction estimation unit 214c may also be based on the Doppler frequency index f s_cfar and the determination result of the aliasing determination unit 212c, DR min , calculate the Doppler frequency index f according to formula (45) es_cfar. Doppler frequency index f es_cfar For example, it corresponds to the Doppler index when the FFT size of the Doppler analysis unit 210 is expanded to Loc×Ncode. es_cfar It is called the "Extended Doppler Frequency Index".

[0383] f es_cfar =f s_cfar +DR min ×Ncode (45)

[0384] Furthermore, assuming a Doppler range up to ±1 / (2×Tr), the extended Doppler frequency index f corresponding to this Doppler range is es_cfar The range is -Loc×Ncode / 2≦f es_cfar <Loc×Ncode / 2, therefore, the calculation result in formula (45) is f es_cfar <-Loc×Ncode / 2, f es_cfar +Loc×Ncode is set to f es_cfar In addition, in f es_cfar ≧Loc×Ncode / 2, f es_cfar -Loc×Ncode is set to f es_cfar .

[0385] A radar transmission signal in which the center frequency fc of the chirp signal changes by Δf according to the coded transmission period (Loc×Tr) includes a phase rotation caused by the change in the center frequency of the chirp signal even when the relative velocity of the target is zero, because the center frequency fc of the chirp signal changes according to the coded transmission period (Loc×Tr). Therefore, each coded transmission period (Loc×Tr) also includes a phase rotation caused by the change in the center frequency of the chirp signal.

[0386] Target distance R target The center frequency fc in the mth transmission cycle Tr changes by floor[(m-1) / Loc]Δf. Therefore, considering the distance R from the target target The arrival time of the reflected wave (2R target / Co), the phase rotation amount Δη(m, R target ) is expressed by equation (46). In addition, equation (46) expresses the relative phase rotation amount when the phase of the first transmission period Tr is used as a reference. C0 represents the speed of light.

[0387]

[0388] Therefore, the direction estimation unit 214c can use the extended Doppler frequency index f es_cfar and distance index f b_cfar, according to formula (47), the Doppler velocity information v of the detected target is output d (f es_cfar ,f b_cfar ). The first item in equation (47) is equivalent to equation (36), which is the Doppler frequency index f es_cfar The relative Doppler velocity component represented by . In addition, the second term in equation (47) is the Doppler velocity component generated by changing the center frequency fc of the chirp signal by Δf according to the code transmission period (Loc×Tr).

[0389] The direction estimation unit 214c can calculate the original relative Doppler velocity v of the target by removing the Doppler component of the second term from the first term in equation (47): d (f es_cfar ,f b_cfar ). Here, R(f b_cfar ) is the beat frequency index f according to formula (8) b_cfar Distance information R(f b_cfar ).

[0390]

[0391] As shown in equation (47), the direction estimation unit 214c calculates the Doppler velocity information v based on a conversion equation that takes into account the change in the center frequency fc of the chirp signal per coding transmission period (Loc×Tr), that is, Δf. d .

[0392] In addition, since the Doppler range is assumed to be up to ±1 / (2×Tr), d v d <-C0 / (4f0Tr), the direction estimation unit 214c can output the Doppler velocity information v of the detected target according to the following formula (48): d .

[0393]

[0394] Similarly, since the Doppler range is assumed to be up to ±1 / (2×Tr), d v d >C0 / (4f0Tr), the direction estimation unit 214c can output the Doppler velocity information v of the detected target according to the following equation (49): d .

[0395]

[0396] As described above, in variant 1, radar device 10c varies the center frequency fc of the chirp signal based on the transmission period (Loc × Tr) of a single orthogonal code sequence. For example, radar device 10c varies the center frequency fc of the chirp signal by Δf (e.g., increasing when Δf > 0, decreasing when Δf <) according to the code transmission period (Loc × Tr) and transmits this chirp signal as a radar transmission signal. Even in this case, radar device 10c (e.g., a MIMO radar) can employ code-multiplexed transmission. Furthermore, similar to embodiment 1, radar device 10c can detect Doppler aliasing using the output of Doppler analysis unit 210 (in other words, the received signal) for each code element of the code-multiplexed signal and the use of unused orthogonal codes.

[0397] Furthermore, according to Variation 1, radar device 10c, similar to Embodiment 1, performs Doppler phase correction including aliasing during code separation. This allows the unambiguously detectable Doppler frequency range to be set to ±1 / (Tr), while suppressing mutual interference between code-multiplexed signals to approximately the noise level. Thus, Variation 1 allows code-multiplexed transmission of MIMO radar to be performed while suppressing degradation of radar detection performance.

[0398] Furthermore, according to variation 1, when the period for causing the center frequency fc of the chirp signal to change by Δf is a plurality of transmission periods Tr, by making the coding transmission period (Loc×Tr) consistent, and also consistent with the switching period of the Doppler analysis unit 210 for each coding element, it is possible to easily perform separation processing of the coded multiplexed signal using unused coding in the aliasing determination unit 212c and phase correction in the coded multiplexing separation processing of the coded multiplexing separation unit 213c.

[0399] In addition, in variation 1, the radar device 10c, for example, causes the center frequency fc of the chirp signal in the radar transmission signal to change by Δf according to the coded transmission period (Loc×Tr) and then transmits it. Therefore, the center frequency change amplitude of the chirp signal is Δf×Ncode, and the distance resolution is 0.5C0 / (Δf×Ncode).

[0400] Thus, by increasing Δf×Ncode, according to variation 1, the range resolution can be improved based on the variation in the chirp signal's center frequency. This allows the chirp sweep bandwidth (e.g., Bw) to be reduced compared to transmitting the chirp signal with a constant center frequency. Reducing the chirp sweep bandwidth, for example, improves range resolution and shortens the transmission period Tr, further expanding the Doppler range that can be unambiguously detected during code-multiplexed transmission.

[0401] Furthermore, in Variation 1 of Embodiment 2, a radar transmission signal is described in which the center frequency fc of the chirp signal is varied by Δf per the code transmission period (Loc × Tr). However, a radar transmission signal in which the center frequency fc of the chirp signal is varied by Δf per (divisor of Loc × Tr) may also be used. Furthermore, when using 1 as a divisor of Loc, the center frequency fc is varied by Δf per Tr, similar to Embodiment 2.

[0402] When using a radar transmission signal in which the center frequency fc of the chirp signal changes by Δf with a transmission period ε, which is a multiple of Loc, that is, every ε transmission period (ε×Tr), the center frequency fc of the chirp signal changes by Δf×Loc / ε with each coded transmission period (Loc×Tr). Therefore, Δf in equations (47), (48), and (49) is replaced by Δf×Loc / ε. In addition, the center frequency change correction vector ξ(f b_cfar ) using formula (50). As described above, the same effect as in the above modification 1 can be obtained. Here, ε is a divisor of Loc.

[0403]

[0404] Furthermore, Variation 1 of Embodiment 2 may be implemented in combination with Embodiment 1, but the coding multiplexing method described in Embodiment 1 may not be applied.

[0405] For example, the code generation unit 104c sets N contained in the code sequence of code length Loc to allcode The number of code multiplexing in the orthogonal code N CM and orthogonal code number N allcode The phase rotation unit 105 uses the N contained in the code sequence of code length Loc. allcode The radar transmit signal is output by code-multiplexing all orthogonal codes and changing the center frequency fc of the chirp signal by Δf according to the code transmission period (Loc×Tr).

[0406] Thus, by increasing Δf×Ncode, the radar device 10c can improve range resolution by varying the chirp signal's center frequency. This allows the chirp sweep bandwidth (e.g., Bw) to be reduced compared to transmitting a chirp signal with a constant center frequency. Consequently, by reducing the chirp sweep bandwidth, the radar device 10c can, for example, improve range resolution and shorten the transmission period Tr.

[0407] Furthermore, the radar device c of variant 1 of embodiment 2, when not combined with embodiment 1, does not employ the aliasing determination unit 212 of the radar device 10, and thus has a Doppler frequency range of ±1 / (2Loc×Tr). However, since the chirp sweep frequency band is reduced to below 1 / Loc by increasing Δf×Ncode, the range resolution can be improved, and the transmission period can be shortened to below Tr / Loc.

[0408] Therefore, the radar device 10c of variant 1 of embodiment 2 uses a radar transmission signal in which the center frequency fc of the chirp signal varies by Δf according to the coded transmission period (Loc×Tr), thereby achieving the effect of expanding the Doppler range that can be detected without ambiguity even with conventional coded multiplexing transmission.

[0409] (Variation 2 of Implementation Method 2)

[0410] The method for controlling the center frequency fc of the frequency modulation signal (chirp signal) is not limited to that of the second embodiment ( Figure 9 ) and variant 1( Figure 11 In variation 2, another method for controlling the center frequency fc of the frequency modulation signal (chirp signal) is described.

[0411] [Structure of radar device]

[0412] Figure 12 1 is a block diagram showing a configuration example of a radar device 10d according to a second modification. Figure 12 In the embodiment 1 ( Figure 1 ) or implementation method 2 ( Figure 8 ) The same structure is assigned the same figure mark and its description is omitted.

[0413] In variation 2, for example, radar device 10d periodically changes the center frequency fc of the chirp signal over multiple transmission cycles within the code transmission cycle (Loc×Tr). In this case, by aligning the timing of the change in the chirp signal center frequency fc for one cycle with the code transmission cycle (Loc×Tr), radar device 10d can perform aliasing detection processing in aliasing detection unit 212d and code demultiplexing processing in code demultiplexing unit 213d (details will be described later).

[0414] The radar transmission signal generating unit 101d in the radar transmitting unit 100d includes a modulation signal generating unit 102, a VCO 103d, and a transmission frequency control unit 107d.

[0415] For example, the modulation signal generator 102 periodically generates a sawtooth-shaped modulation signal for VCO control. Here, the transmission period is Tr.

[0416] The transmission frequency control unit 107d controls the center frequency fc of the frequency modulation signal (chirp signal) output from the VCO 103d at the transmission period Tr based on the orthogonal code element index OC_INDEX output from the code generation unit 104d.

[0417] For example, during the transmission cycle Tr with OC_INDEX = 1, the transmission frequency control unit 107d sets the center frequency fc of the frequency modulated signal output from the VCO 103d to f0. Furthermore, during the transmission cycle Tr with OC_INDEX = 2, the transmission frequency control unit 107d sets the center frequency fc of the frequency modulated signal output from the VCO 103d to f0 + Δf. Similarly, during the transmission cycles Tr with OC_INDEX = 3 to Loc, the transmission frequency control unit 107d sets the center frequency fc of the frequency modulated signal output from the VCO 103d to (f0 + 2Δf) to (f0 + (Loc - 1)Δf), respectively.

[0418] By this control, the transmission frequency control unit 107d can perform control so that the center frequency fc of the chirp signal is periodically changed at each code transmission period (Loc×Tr), for example.

[0419] The VCO 103 d outputs a frequency modulation signal to the phase rotation unit 105 and the radar reception unit 200 (eg, the mixer unit 204 ) based on the outputs of the transmission frequency control unit 107 d and the modulation signal generation unit 102 .

[0420] Figure 13 An example of a frequency modulated signal (hereinafter referred to as a chirp signal) after frequency modulation is shown.

[0421] exist Figure 13 For example, in the first transmission cycle Tr#1, VCO 103d outputs a chirp signal having a center frequency fc(1) of f0. Furthermore, in the second transmission cycle Tr#2, VCO 103d outputs a chirp signal having a center frequency fc(2) of f0+Δf. Similarly, in the third transmission cycle Tr#3 through the Loc-th transmission cycle Tr#Loc, VCO 103d outputs chirp signals having center frequencies fc(3) to fc(Loc) of (f0+2Δf) to (f0+(Loc-1)Δf), respectively.

[0422] Furthermore, VCO 103d outputs a chirp signal having a center frequency fc(Loc+1) of f0 during the (Loc+1)th transmission cycle Tr#(Loc+1). Similarly, VCO 103d outputs chirp signals having center frequencies fc(Loc+2) to fc(2Loc) of (f0+Δf) to (f0+(Loc-1)Δf) during the (Loc+2)th transmission cycle Tr#(Loc+2) to the (2nd)th transmission cycle Tr#(2Loc), respectively.

[0423] Similarly, in the mth transmission cycle Tr#m, the VCO 103d outputs a chirp signal having a center frequency fc(m) of the chirp signal of f0+mod(m-1,Loc)Δf.

[0424] That is, in Figure 13 In the Nth c Transmission cycle Tr#N c The center frequency of the chirp signal fc(N c ) is f0+(Loc-1)Δf.

[0425] So, in Figure 13 In the embodiment, the change of the center frequency fc of the chirp signal is repeated once with a transmission period of the radar transmission signal that is approximately several times the code length of a plurality of coding sequences (for example, orthogonal codes).

[0426] Furthermore, each chirp signal can use, for example, the time range T of the range gate A The chirp signal has the same frequency modulation bandwidth Bw as in Figure 13 The example shown shows the case where Δf>0 (in other words, the case where the center frequency fc increases), but the same applies to the case where Δf<0 (in other words, the case where the center frequency fc decreases).

[0427] Figure 12 Other operations in the radar transmitting unit 100d shown may be the same as those in the first embodiment.

[0428] Next, an operation example of the radar receiving unit 200d of the radar device 10d will be described.

[0429] In the radar receiver 200d, the processing of the signals received by the receive antenna 202 by each antenna system processing unit 201 and the subsequent operation of the CFAR unit 211 are the same as those in Embodiment 1. Furthermore, in the radar receiver 200d, the direction estimation processing performed by the direction estimation unit 214d using the output of the code multiplexing separation unit 213d is also the same as that in Embodiment 1.

[0430] In the radar receiving unit 200d, for example, the operation of the aliasing determination unit 212d and the operation of the code multiplexing separation unit 213d are different from those in the first embodiment.

[0431] An example of operation of the aliasing determination unit 212 d that is different from that of the first embodiment will be described below.

[0432] For example, when using a radar transmission signal in which the center frequency fc of the chirp signal is changed to f0, f0+Δf, ..., f0+(Loc-1)Δf according to the transmission period Tr within each coded transmission period (Loc×Tr), the radar reflected wave when the chirp signal with the transmission center frequency fc being f0, f0+Δf, ..., f0+(Loc-1)Δf is input as the received signal to the 1st, 2nd, ..., Loc Doppler analysis units 210.

[0433] Therefore, each of the Loc Doppler analysis units 210 has the same center frequency as the input radar reflected wave.

[0434] On the other hand, since the center frequency of the chirp signal differs between the Loc Doppler analysis units 210, the radar receiving unit 200d uses the Doppler phase correction vector α (f s_cfar ), the center frequency change correction vector ζ(f b_cfar ) to correct the phase. That is, Equation (51) uses Instead of α(f s_cfar ).

[0435]

[0436] Here, R(f b_cfar ) is the beat frequency index f according to formula (8) b_cfar The distance estimate R(f b_cfar ). Formula (51) is derived based on the following situation. When the center frequency of the chirp signal transmitted by the first Doppler analysis unit 210 is used as a reference, the center frequencies of the chirp signals transmitted by the second to Loc-th Doppler analysis units 210 are different in the form of Δf, ..., (Loc-1)Δf. Therefore, the value from R(f b_cfar ) of the reflected wave arrival time (2R(f b_cfar ) / Co) have different phase rotation amounts.

[0437] That is, the phase rotation amount in the noc-th Doppler analysis unit 210 when the output of the first Doppler analysis unit 210 is used as the phase reference is 2π(noc-1)Δf×(2R(f b_cfar) / Co). In order to eliminate this phase rotation, the center frequency change correction vector shown in equation (51) is derived

[0438] ζ(f b_cfar ). Here, noc=1, ..., Loc.

[0439] In this way, in the following case, that is, when the center frequency fc of the chirp signal is periodically changed within a plurality of chirp transmission periods within the coding transmission period (Loc×Tr), since the switching period of the Doppler analysis unit 210 for each coding element coincides with the timing of one cycle of the change in the center frequency fc of the chirp signal, the aliasing determination unit 212d can easily perform phase correction in the separation process of the coded multiplexed signal using unused coding.

[0440] As described above, the aliasing determination unit 212d can replace equation (12) with equation (52) to calculate the unused orthogonal code UnCode. nuc The received power after coding separation DeMulUnCode nuc (f b_cfar ,f s_cfar , DR). Formula (52) uses To replace α(f s_cfar ). Here, nuc=1,…,N allcode -N CM DR is an index indicating the Doppler aliasing range, and takes an integer value in the range of DR=ceil[-Loc / 2], ceil[-Loc / 2]+1, ..., 0, ..., ceil[Loc / 2]-1.

[0441]

[0442] In addition, equation (53) can be used instead of equation (29).

[0443]

[0444] Next, an operation example of the code multiplexing separation unit 213d in the second variant of embodiment 2 is described. The code multiplexing separation unit 213d uses equation (54) instead of equation (30) for the same reason as the operation example of the aliasing determination unit 212d in the second variant of embodiment 2, and uses the aliasing determination result in the aliasing determination unit 212d, i.e., DR min , and the distance index f extracted in the CFAR unit 211 b_cfar and the Doppler frequency index f s_cfar The Doppler component VFTALL is the output of the corresponding Doppler analysis unit 210. z (f b_cfar ,fs_cfar ) is used for coding separation. To replace α(f s cfar ).

[0445]

[0446] In addition, the coding multiplexing separation unit 213d can also use equation (55) instead of equation (31) and use the aliasing determination result DR in the aliasing determination unit 212d min , and the distance index f extracted in the CFAR unit 211 b_cfar and the Doppler frequency index f s_cfar The Doppler component VFTALL is the output of the corresponding Doppler analysis unit 210. z (f b_cfar ,f s_cfar ) is used to separate the coded multiplexed signal. In addition, in formula (55), The term does not depend on the index f of the Doppler component s Therefore, by pre-tabulating, the amount of calculation in the coding multiplexing separation unit 213d can be reduced.

[0447]

[0448] In this way, the coding multiplexing separation unit 213d periodically changes the center frequency fc of the chirp signal within multiple chirp transmission periods within the coding transmission period (Loc×Tr), thereby aligning the switching period of the Doppler analysis unit 210 for each coding element with the timing of one cycle of change in the center frequency fc of the chirp signal, thereby making it easy to perform phase correction in the separation processing of the coding multiplexed signal.

[0449] Next, an operation example of the direction estimation unit 214d in the second modification of the second embodiment will be described.

[0450] For example, the direction estimation unit 214d may also be based on the Doppler frequency index f s_cfar and the determination result of the aliasing determination unit 212d, DR min , calculate the Doppler frequency index f according to formula (56) es_cfar .

[0451] f es_cfar =f s_cfar +DR min ×Ncode (56)

[0452] In addition, the Doppler frequency index f es_cfarFor example, it corresponds to the Doppler index when the FFT size of the Doppler analysis unit 210 is expanded to Loc×Ncode. es_cfar It is called "extended Doppler frequency index". In addition, it is assumed that the Doppler range is up to ±1 / (2×Tr), and the extended Doppler frequency index f corresponding to this Doppler range is es_cfar The range is -Loc×Ncode / 2≦f es_cfar <Loc×Ncode / 2, therefore, the calculation result of formula (56) is, at f es_cfar <-Loc×Ncode / 2, f es_cfar +Loc×Ncode is set to f es_cfar In addition, in f es_cfar ≧Loc×Ncode / 2, f es_cfar -Loc×Ncode is set to f es_cfar .

[0453] In addition, the direction estimation unit 214d can use, for example, the extended Doppler frequency index f es_cfar , use Equation (57) to output the Doppler velocity information of the detected target.

[0454]

[0455] As described above, in variant 2, radar device 10d periodically changes the center frequency fc of the chirp signal over multiple transmission cycles. In this case, the timing of the chirp signal center frequency fc changing over one cycle coincides with the code transmission period (Loc×Tr). Furthermore, radar device 10d (e.g., a MIMO radar) can also employ code-multiplexed transmission. Similarly to embodiment 1, radar device 10d can detect Doppler aliasing using the output of Doppler analysis unit 210 (in other words, the received signal) for each code element of the code-multiplexed signal and the absence of orthogonal codes.

[0456] Furthermore, according to Variation 2, radar device 10d, similar to Embodiment 1, performs Doppler phase correction including aliasing during code separation. This allows the unambiguously detectable Doppler frequency range to be set to ±1 / (Tr), while suppressing mutual interference between code-multiplexed signals to approximately the noise level. Thus, Variation 2 allows code-multiplexed transmission of MIMO radar to be performed while suppressing degradation of radar detection performance.

[0457] Furthermore, according to variant 2, by setting the timing of the change of the center frequency fc of the chirp signal through one cycle to the code transmission period (Loc×Tr), thereby coinciding with the switching period of the Doppler analysis unit 210 for each code element (i.e., the transmission period of the code used for code multiplexing), it is possible to easily perform the separation processing of the code multiplexing signal using unused codes in the aliasing determination unit 212d and the phase correction in the code multiplexing separation processing of the code multiplexing separation unit 213d.

[0458] Furthermore, according to variation 2, the timing of the change of the center frequency fc of the chirp signal for one cycle is set to the coding transmission period (Loc×Tr), but it can also be set to (divisor of Loc×Tr). When the timing of the change of the center frequency fc of the chirp signal for one cycle is set to the period of (divisor ε of Loc×Tr), that is, ε times the transmission period (ε×Tr), equation (58) can be used as the center frequency change correction vector ζ(f b_cfar ). According to the above, the same effect as the above-mentioned variation 1 can be obtained. Wherein, ε is a divisor of Loc, and ε>1.

[0459]

[0460] Furthermore, according to variation 2, a radar transmission signal is used in which the center frequency fc of the chirp signal is set to f0, f0+Δf, ..., f0+(Loc-1)Δf according to the transmission period Tr in the coded transmission period (Loc×Tr) and changes in integer multiples of Δf. However, this is not limited to this and the frequency can also be changed at any arbitrary frequency.

[0461] For example, the center frequency fc of the chirp signal may be changed to f0, f0+Δf1, f0+Δf2, ..., f0+Δf according to the transmission period Tr of the coded transmission period (Loc×Tr). Loc-1 Here, Δf1, Δf2, ..., Δf Loc-1 is the frequency variable value of the center frequency fc of the chirp signal at each transmission period Tr of the coded transmission period (Loc×Tr). In this case, equation (59) is used as the center frequency variation correction vector ζ(f b_cfar ). As described above, the same effect as the above-mentioned variation 1 can be obtained.

[0462]

[0463] In the above, one embodiment of the present disclosure has been described.

[0464] (Other variants 1)

[0465] The transmitting antenna of the radar device may have a subarray structure. For example, the radar device may perform Doppler multiplexing transmission using both subarray beamforming (subarray BF) and code multiplexing transmission.

[0466] By combining several transmit antennas into subarrays, the beamwidth of the transmit directional beam pattern can be narrowed, increasing the transmit directional gain. This narrows the detectable angular range but increases the detectable distance range. Furthermore, by making the beam weight coefficient used to generate the directional beam variable, the beam direction can be variably controlled.

[0467] Figure 14 1 is a block diagram showing a configuration example of the radar transmitter 100e of this modification. Figure 14 In the Figure 1 The same code is attached to the structure of the radar transmitter 100 shown in FIG. 1 and its description is omitted. In addition, the radar receiver of this variant has the same basic structure as the radar transmitter 100. Figure 1 The radar receiving unit 200 shown is common, so it is referenced Figure 1 Provide explanation.

[0468] Figure 14 The radar transmitter 100 e shown includes a plurality of transmission antennas 106 e capable of controlling the transmission phase on the output side of the phase rotation unit 105 .

[0469] For example, for the outputs of the Nt phase rotation units 105, N SA Subarrays SA#1 to SA#N of the transmitting antennas 106e SA (For example, Nt groups of subarrays). In addition, the subarray structure of the transmitting antenna 106e is not limited to Figure 14 For example, the number of transmitting antennas contained in the subarray of the output of each phase rotation unit 105 (in other words, N SA ) may be different in number or different between the phase rotation units 105. SA is an integer greater than 1. SA =1, and Figure 1 same.

[0470] exist Figure 14 In the example, the beam weight generation unit 108 generates a beam weight for directing the main beam direction of the transmission beam toward a predetermined direction using the subarray. SA The subarray of transmitting antennas is spaced d apart from each other. SA The transmission beam direction in the case of a straight line configuration is represented by θ TxBF In this case, the beam weight generation unit 108 generates the beam weight W as shown in the following equation (60), for example. Tx(Index_TxSubArray,θ TxBF ).

[0471]

[0472] Here, Index_TxSubArray represents the element index of the subarray, which is Index_TxSubArray=1, ..., N SA In addition, λ represents the wavelength of the radar signal, d SA Indicates the subarray antenna spacing.

[0473] Each beam weight multiplication unit 109 multiplies the output from the corresponding phase rotation unit 105 by the beam weight coefficient W input from the beam weight generation unit 108. Tx (Index_TxSubArray,θ TxBF ). Multiplied by the beam weight W Tx (Index_TxSubArray,θ TxBF ) is sent by N SA Sub-array antennas transmit. Here, Index_TxSubArray = 1, ..., N SA .

[0474] The subsequent operations are performed by replacing the transmitting antennas 106 (eg, Tx#1 to Nt) in the first embodiment with Figure 14 The transmitting antenna 106e (eg, transmitting sub-array antennas Tx#1 to Tx#N) is shown. ) , performing the same actions, the same effect as implementation mode 1 or implementation mode 2 can be obtained.

[0475] Through the above operation, the radar transmitter 100e can transmit code-multiplexed radar transmission signals using subarrays to direct the transmission directional beam in a predetermined direction. This improves the transmission directivity gain in the predetermined direction and expands the detectable range. Furthermore, by variably setting the beam weight coefficient used to generate the transmission directional beam, the radar transmitter 100e can variably control the beam direction.

[0476] In addition, Figure 14 , the phase rotation section 105 and the beam weight multiplication section 109 are shown as being performed separately. However, the present invention is not limited to this configuration. For example, the phase rotation section 105 may include a phase rotation corresponding to the beam weight multiplication section 109. Figure 15 3 is a block diagram showing a configuration example of a radar transmitter 100f having such a configuration.

[0477] exist Figure 15In the radar transmitter 100f shown in FIG. 1 , the phase adder 110 adds the phases of the sub-arrays SA#1 to SA#N output from the beam weight generator 108f. SA The phase rotation of is added to the phase rotation amount output from the code generation unit 104, and the result is output to the phase rotation unit 105f.

[0478] For example, the phase adding section 110 adds the phase rotation amount ψ outputted from the code generating section 104 at each transmission cycle Tr and given to the ncm-th transmission antenna Tx#ncm. ncm (m) and the beam weight generating unit 108f outputted from the beam weight generating unit 108f for each of the subarrays SA#1 to SA#N constituting the ncm-th transmitting antenna Tx#ncm. SA Phase rotation angle (W Tx (Index_TxSubArray,θ TxBF )) are added. Then, the phase adding unit 110 adds the added value (ψ ncm (m)+angle(W Tx (Index_TxSubArray,θ TxBF ))) output to N connected to the transmitting antenna Tx#ncm SA A phase rotation unit 105f (eg, a phase shifter) is provided.

[0479] Each phase rotation unit 105f applies phase rotation to the chirp signal output from the radar transmission signal generation unit 101 at the transmission period Tr based on the output from the phase addition unit 110. Here, ncm=1, ..., N CM , m=1,…,Nc.

[0480] In addition, the configuration for performing subarray transmission described in this variation is not limited to the configuration of Embodiment 1 (for example, Figure 1 ), and can also be applied to other variations or implementations (for example, Figure 4 、 Figure 8 、 Figure 10 or Figure 12 ).For example, Figure 14 and Figure 15 The structure of the radar transmitting unit 100e or 100f shown can also be applied to the radar devices 10a, 10b, 10c, and 10d.

[0481] (Other variants 2)

[0482] In the above embodiment, the case where Walsh-Hadamard codes are used as the codes generated by the code generation units 104, 104c, and 104d has been described. However, the codes are not limited to Walsh-Hadamard codes, and other codes may be used.

[0483] For example, orthogonal M-series coding or pseudo-orthogonal coding may be used as the orthogonal codes generated by the code generation units 104, 104c, and 104d. Hereinafter, an operation example in the case of using orthogonal M-series coding and pseudo-orthogonal coding will be described.

[0484] <Orthogonal M series encoding>

[0485] Orthogonal M-series coding, for example, is a code sequence in which code elements are appended to a sequence generated from an M-series so that the number of "1" and "-1" code elements is equal (in other words, balanced) (see, for example, Non-Patent Document 5). Furthermore, because codes containing these code elements are orthogonal to codes in which all code elements are "1," the following description will cover the case in which orthogonal M-series coding also includes codes in which all code elements are 1.

[0486] As an example, orthogonal M-series coding with a code length of 4 includes the following coding.

[0487] OMS4(1)=[1 1-1-1]

[0488] OMS4(2)=[-1 1 1-1]

[0489] OMS4(3)=[1-1 1-1]

[0490] OMS4(4)=[1 1 1 1]

[0491] Here, the orthogonal M-series code with code length Loc is represented as OMS Loc (nomc). Wherein, nomc represents the code index contained in the orthogonal M-series code of code length Loc. nomc = 1, ..., Loc.

[0492] For example, OMS4(1) and OMS4(2) are a group of codes in which the even-numbered code elements are the same and the signs of the odd-numbered code elements are reversed. allcode -N CM ) is two or more, the code generation unit 104, 104c, or 104d can select a code so as not to use an orthogonal code and not include a code group such as OMS4(1) and OMS4(2). This code selection enables, for example, Doppler frequency aliasing determination in the aliasing determination unit 212 of the radar receiving unit 200.

[0493] Similarly, OMS4(3) and OMS4(4) are groups of codes in which the odd-numbered code elements are the same and the signs of the even-numbered code elements are reversed. For example, when the number of orthogonal codes not used (N allcode -N CM) is two or more, the code generation unit 104, 104c, or 104d can select a code in such a manner that no orthogonal code is used and no code group including OMS4(3) and OMS4(4) is included. This code selection enables, for example, Doppler frequency aliasing determination in the aliasing determination unit 212 of the radar receiving unit 200.

[0494] Furthermore, the code length of the orthogonal M-series coding is not limited to Loc=4, and other code lengths may be used. For example, the orthogonal M-series coding with a code length of Loc=8 includes the following codes.

[0495] OMS4(1)=[-1 1-1-1 1 1 1-1]

[0496] OMS4(2)=[1-1 1-1-1 1 1-1]

[0497] OMS4(3)=[1 1-1 1-1-1 1-1]

[0498] OMS4(4)=[1 1 1-1 1-1-1-1]

[0499] OMS4(5)=[-1 1 1 1-1 1-1-1]

[0500] OMS4(6)=[-1-1 1 1 1-1 1-1]

[0501] OMS4(7)=[1-1-1 1 1 1-1-1]

[0502] OMS4(8)=[1 1 1 1 1 1 1 1]

[0503] For example, the code number N CM The code length Loc of an orthogonal M-series code can be expressed by the following equation (61).

[0504]

[0505] Here, ceil[x] is an operator (ceil function) that outputs the smallest integer greater than or equal to the real number x.

[0506] The code generation unit 104, 104c or 104d generates N contained in the orthogonal M series code of the code length Loc. allcode N of (Loc) codes CM For example, in the case of orthogonal M-series coding, N allcode For example, since the orthogonal M-series codes with code length Loc=4, 8 or 16 contain 4, 8 or 16 orthogonal codes respectively, Nallcode (4) = 4, N allcode (8) = 8 and N allcode (16)=16.

[0507] Pseudo-orthogonal codes

[0508] For example, in the first embodiment, in the code multiplexing separation unit 213 (for example, Figure 1 ), when using orthogonal codes, the orthogonal codes used in coded multiplexing transmission are used. ncm , the signal subjected to code multiplexing is separated. In contrast, when pseudo-orthogonal codes are used, the following aspects are different from orthogonal codes, that is, the code multiplexing separation unit 213 separates the pseudo-orthogonal codes Code used in code multiplexing transmission. ncm , using the inverse code "InvCode ncm ", separates the coded multiplexed signals.

[0509] In addition, for example, in the first embodiment, in the aliasing determination unit 212 (for example, Figure 1 ), in the case of using orthogonal code, use the unused orthogonal code UnCode nuc In contrast, when pseudo-orthogonal codes are used, the following aspects are different from orthogonal codes, that is, the aliasing determination unit 212 uses, for example, the pseudo-orthogonal code UnCode to separate the unused pseudo-orthogonal codes. nuc InvUnCode nuc " to make aliasing judgment.

[0510] In addition, the inverse code InvCode ncm It is a code derived from pseudo-orthogonal codes.

[0511] An example of operations in the aliasing determination unit 212 and the code demultiplexing unit 213 of this modification that are different from those in the first embodiment will be described below.

[0512] The code generation unit 104 generates N contained in a pseudo-orthogonal code sequence of code length Loc (in other words, a code sequence that is orthogonal to the corresponding inverse code). allcode N of the pseudo-orthogonal codes CM Pseudo-orthogonal codes are set as codes for code multiplexing transmission.

[0513] For example, the coding multiplexing number N CM Less than the number of pseudo-orthogonal codes N allcode , N CM <N allcode In other words, the code length Loc of the pseudo-orthogonal code is greater than the code multiplexing number N CM For example, the code length Loc is N CM Codencm =[OC ncm (1),OC ncm (2),…,OC ncm (Loc)].

[0514] In addition, the code length Loc of the pseudo-orthogonal code is not limited to a power of 2, and can be any code length. Thus, for example, the code length Loc of the pseudo-orthogonal code can be the code multiplexing number N CM The value obtained by adding 1 (Loc = N CM +1). In this case, for example, N allcode =N CM +1, and code length Loc=N allcode pseudo-orthogonal code sequence.

[0515] As an example, the following describes the case where the code length Loc is A portion of the Walsh-Hadamard code to generate the code number N CM However, the method of generating a pseudo-orthogonal code sequence is not limited to this.

[0516] As an example, the coding multiplexing number N is described. CM = 4 and uses a portion of a Walsh-Hadamard code of length 8 to generate N allcode =N CM +1=5 and the pseudo-orthogonal matrix with code length Loc=5.

[0517] For example, the code generation unit 104 can use 5 (=Loc) code elements in the Walsh-Hadamard code with a code length of 8 (for example, the first to fifth code elements: WH8(1) to WH8(5)) to generate the following pseudo-orthogonal codes PWH5(1) to PWH5(5).

[0518] PWH5(1)=[1 1 1 1 1]

[0519] PWH5(2)=[1-1 1-1 1]

[0520] PWH5(3)=[1 1-1-1 1]

[0521] PWH5(4)=[1-1-1 1 1]

[0522] PWH5(5)=[1 1 1 1-1]

[0523] For example, the inverse coding matrix "InvPOC" of the coding matrix "POC" having pseudo-orthogonal codes PWH5(1) to PWH5(5) as row elements can be derived according to the following equation (62): Here, the superscript H represents the complex conjugate transpose operator.

[0524] InvPOC=POC H (POC H Proof of Concept (PoC) -1 (62)

[0525] For example, the POC formed by the above-mentioned pseudo-orthogonal codes PWH5(1) to PWH5(5) is a coding matrix shown in the following equation (63).

[0526]

[0527] The inverse coding matrix InvPOC for the coding matrix POC shown in equation (63) is calculated, for example, by the following equation (64). In addition, each row of the inverse coding matrix InvPOC is expressed as inverse coding InvPWH5(1) to inverse coding InvPWH5(5).

[0528]

[0529] Here, the pseudo-orthogonal code PWH5 (npoc1) and the inverse code InvPWH5 (npoc2) have the relationship as shown in the following equation (65). Here, npoc represents the code index included in the pseudo-orthogonal code or the inverse code. For example, npoc1 = 1, ..., Loc, and npoc2 = 1, ..., Loc. Furthermore, npoc1 represents, for example, the code index npoc1 of the pseudo-orthogonal code PWH5, and npoc2 represents, for example, the code index npoc2 of the inverse code InvPWH5.

[0530]

[0531] For example, for the pseudo-orthogonal code PWH5(npoc) generated by the code generation unit 104, the code multiplexing number N CM = 4, the codes used for code-multiplexed transmission are Code1 = PWH5(1), Code2 = PWH5(2), Code3 = PWH5(3), and Code4 = PWH5(4). In this case, the unused pseudo-orthogonal code is UnCode1 = PWH5(5), and the inverse code for the unused pseudo-orthogonal code UnCode1 is InvUnCode1 = InvPWH5(5).

[0532] For example, the code generation unit 104 sets the number of pseudo-orthogonal codes not used (N allcode -N CM) is two or more, codes may be selected such that the inverse codes of the pseudo-orthogonal code with respect to the code length Loc have the same odd-numbered code elements, and the unused pseudo-orthogonal codes do not include any code groups in which the signs of the even-numbered code elements are reversed. This selection of unused pseudo-orthogonal codes can improve the accuracy of Doppler frequency aliasing determination in the aliasing determination unit 212 of the radar receiving unit 200.

[0533] Furthermore, in the inverse codes of the pseudo-orthogonal code with respect to the code length Loc, the odd-numbered code elements are the same, while the signs of the even-numbered code elements are reversed. For a group of codes, the odd-numbered code elements are the same, while the signs of the even-numbered code elements are reversed. Therefore, the code generation unit 104 does not use the number of pseudo-orthogonal codes (N allcode -N CM ) is two or more, codes may be selected such that the odd-numbered code elements of the codes in the pseudo-orthogonal code of code length Loc are identical, and the code groups in which the signs of the even-numbered code elements are reversed are not included in the unused pseudo-orthogonal codes. By selecting such unused pseudo-orthogonal codes, the accuracy of Doppler frequency aliasing determination in the aliasing determination unit 212 of the radar receiving unit 200 can be improved.

[0534] As another example, the coding multiplexing number N CM = 4 and uses a portion of a Walsh-Hadamard code of length 8 to generate N allcode =N CM +2=6 and the case of a pseudo-orthogonal matrix with code length Loc=6.

[0535] For example, the code generation unit 104 can use 6 (=Loc) code elements in the Walsh-Hadamard code with a code length of 8 (for example, the first to sixth code elements: WH8(1) to WH8(6)) to generate the following pseudo-orthogonal codes PWH6(1) to PWH6(6).

[0536] PWH6(1)=[1 1 1 1 1 1]

[0537] PWH6(2)=[1-1 1-1 1-1]

[0538] PWH6(3)=[1 1-1-1 1 1]

[0539] PWH6(4)=[1-1-1 1 1-1]

[0540] PWH6(5)=[1 1 1 1-1-1]

[0541] PWH6(6)=[1-1 1-1-1 1]

[0542] For example, the inverse codes InvPWH6(1) to InvPWH6(6) for the pseudo-orthogonal codes PWH6(1) to PWH6(6) can be calculated as follows. Here, the inverse code for the pseudo-orthogonal code PWH6(npoc) is InvPWH6(npoc). Where npoc = 1, ..., Loc.

[0543] InvPWH6(1)=0.25×[0 0 1 1 1 1]

[0544] InvPWH6(2)=0.25×[0 0 1-1 1-1]

[0545] InvPWH6(3)=0.25×[1 1-1-1 0 0]

[0546] InvPWH6(4)=0.25×[1-1-1 1 0 0]

[0547] InvPWH6(5)=0.25×[1 1 0 0-1-1]

[0548] InvPWH6(6)=0.25×[1-1 0 0-1 1]

[0549] Here, InvPWH6(1) and InvPWH6(2) in the inverse codes are a set of inverse codes in which the odd-numbered code elements are the same, and the signs of the even-numbered code elements are reversed (except for code elements that are 0). In addition, the codes PWH6(1) and PWH6(2) corresponding to InvPWH6(1) and InvPWH6(2) are also a set of codes in which the odd-numbered code elements are the same, and the signs of the even-numbered code elements are reversed.

[0550] Similarly, the group of InvPWH6(3) and InvPWH6(4), and the group of InvPWH6(5) and InvPWH6(6) are also inversely coded groups with the same relationship to each other as the group of InvPWH6(1) and InvPWH6(2). In addition, the group of InvPWH6(3) and InvPWH6(4), the group of InvPWH6(5) and InvPWH6(6), and the coded groups corresponding to the group of InvPWH6(1) and InvPWH6(2), namely, the group of PWH6(3) and PWH6(4), and the group of PWH6(5) and PWH6(6), are all coded groups with the same relationship to each other as the group of PWH6(1) and PWH6(2).

[0551] For example, when the number of pseudo-orthogonal codes not used (N allcode -N CM ) is two or more, the radar device 10 may select the code in such a manner that the pseudo-orthogonal code corresponding to the inverse code group of such a relationship is not included in the unused pseudo-orthogonal code.

[0552] For example, the combination of codes (Code 1, Code 2, Code 3, and Code 4) used for code-multiplexed transmission can be, for example, Code 1 = PWH6(1), Code 2 = PWH6(3), Code 3 = PWH6(5), and Code 4 = PWH6(6). In this case, the unused pseudo-orthogonal codes are, for example, UnCode 1 = PWH6(2) and UnCode 2 = PWH6(4). Furthermore, the combinations of codes used for code-multiplexed transmission and unused pseudo-orthogonal codes are not limited to these.

[0553] Next, an operation example of the aliasing determination unit 212 when using pseudo-orthogonal codes will be described.

[0554] The aliasing determination unit 212 uses, for example, a pseudo-orthogonal code of code length Loc and (N allcode -N CM The aliasing determination unit 212 performs aliasing determination based on the inverse codes corresponding to the non-pseudo-orthogonal codes. The operations in the aliasing determination unit 212 are the same as those in the first embodiment except that the pseudo-orthogonal codes are used instead of the orthogonal codes.

[0555] For example, the aliasing determination unit 212 corrects the phase change of the Doppler component including the aliasing based on the output of the Doppler analysis unit 210 in each antenna system processing unit 201, and calculates the phase change corresponding to the unused pseudo-orthogonal code UnCode according to the following equation (66) instead of using equation (12). nuc InvUnCode nuc The received power after coding separation DeMulUnCode nuc (f b_cfar , f s_cfar , DR).

[0556]

[0557] In equation (66), the outputs of the Doppler analysis unit 210 in all antenna system processing units 201 are used to calculate the pseudo-orthogonal code UnCode. nuc The corresponding inverse code InvUnCode nucThe sum of the received powers after code separation using the inverse code corresponding to the unused pseudo-orthogonal code may be used to calculate the received power after code separation. This improves the accuracy of aliasing determination even when the received signal level is low. However, instead of using equation (66), the received power after code separation using the inverse code corresponding to the unused pseudo-orthogonal code may be calculated for the output of the Doppler analysis unit 210 in some antenna system processing units 201. Even in this case, for example, it is possible to maintain aliasing determination accuracy within a range where the received signal level is sufficiently high and reduce the amount of computational processing.

[0558] In addition, in formula (53), nuc = 1, ..., N allcode -N CM DR is an index indicating the Doppler aliasing range, and for example, takes an integer value in the range of DR=ceil[-Loc / 2], ceil[-Loc / 2]+1, ..., 0, ..., ceil[Loc / 2]-1.

[0559] In addition, similarly, the unused pseudo-orthogonal code UnCode is used. nuc InvUnCode nuc The received power after coding separation DeMulUnCode nuc (f b_cfar , f s_cfar , the calculation formula of DR) can also be transformed into the following formula (67) to replace formula (29).

[0560]

[0561] Next, an operation example of the code multiplexing separation unit 213 when using pseudo-orthogonal codes will be described.

[0562] The code multiplexing separation unit 213 performs separation processing of the code multiplexing signal based on the aliasing determination result of the aliasing determination unit 212. For example, in the separation processing of the code multiplexing signal, the code multiplexing separation unit 213 separates the code multiplexing signal by using the pseudo-orthogonal code Code used for code multiplexing transmission. ncm The corresponding inverse code InvCode ncm Perform code separation on the code multiplexed signal.

[0563] For example, the code multiplexing separation unit 213 replaces the equation (30) with the following equation (68), based on the aliasing determination result in the aliasing determination unit 212, that is, DR min The aliasing phase correction vector β(DR min ), and the distance index f extracted in the CFAR unit 211 b_cfar and the Doppler frequency index f s_cfar The Doppler component VFTALL is the output of the corresponding Doppler analysis unit 210. z (fb_cfar ,f s_cfar ) to perform encoding separation processing.

[0564]

[0565] Here, DeMul z ncm (f b_cfar ,f s_cfar ) is the distance index f used for the Doppler analysis unit 210 in the z-th antenna system processing unit 201. b_cfar and the Doppler frequency index f s_cfar The output corresponds to the pseudo-orthogonal code Code ncm InvCode ncm , the code multiplexed signal is subjected to code separation (e.g., the code separation result). In addition, z = 1, ..., Na, ncm = 1, ..., N CM .

[0566] In addition, the code multiplexing separation unit 213 can use the following equation (69) instead of equation (68).

[0567]

[0568] In formula (69), (However, in formula (69), DR = DR min ) does not depend on the index of the Doppler component (e.g., the Doppler frequency index) f s Therefore, for example, by tabulating it in advance, the amount of calculation in the coding multiplexing separation unit 213 can be reduced.

[0569] By the above operations of the code generation unit 104 using the inverse code corresponding to the pseudo-orthogonal code, the aliasing determination unit 212 and the code multiplexing separation unit 213, the same effects as those of the first embodiment can be obtained even when the pseudo-orthogonal code is used.

[0570] In addition, the structure using coding described in this variation is not limited to the structure of the first embodiment (for example, Figure 1 ), and can also be applied to other variations or implementations (for example, Figure 4 、 Figure 8 、 Figure 10 or Figure 12 For example, the coding in this variation (eg, orthogonal M-series coding or pseudo-orthogonal coding) may be applied instead of the Walsh-Hadamard codes in the radar devices 10a, 10b, 10c, and 10d.

[0571] (Other variants 3)

[0572] For example, the operation of the aliasing determination unit 212 will be described in the following case: b_cfar In the image, there are multiple targets with approximately equal reception levels, and the intervals between the Doppler peaks of the multiple targets coincide with the intervals between Doppler aliasing.

[0573] Here, since the Doppler frequencies of multiple targets may differ, the relative speeds of movement between the targets and the radar device 10 may also differ. Therefore, by continuously performing radar observations in the radar device 10, even if aliasing is difficult to detect in the radar positioning output at a certain point in time, the radar positioning output at subsequent points in time can increase the likelihood that the distances between the multiple targets will be measured as different.

[0574] Furthermore, in order to more effectively separate the signals corresponding to multiple targets, for example, the radar device 10 can variably set at least one of the transmission period Tr and the code length of the orthogonal code (or pseudo-orthogonal code) generated in the code generation unit 104 according to each radar positioning performed (for example, according to Nc transmission periods (Nc×Tr)) when continuously performing radar positioning.

[0575] For example, the radar device 10 may switch the code length of the orthogonal code (or pseudo-orthogonal code) generated by the code generation unit 104 for each radar positioning, and perform code-multiplexed transmission.

[0576] For example, set Loc=N CM +δ. The radar device 10 can variably set δ for each radar positioning, making the Doppler aliasing interval variable. In this case, the radar device 10 can, for example, use orthogonal codes for transmission when Loc is a power of 2, or use pseudo-orthogonal codes for transmission when Loc is not a power of 2. For example, δ can be periodically and variably set to 1, 2, 1, 2, ... for each radar positioning.

[0577] Alternatively, the radar device 10 can switch between the following (a) and (b) for each radar positioning, thereby switching the code length and type of code used for code-multiplex transmission and performing code-multiplex transmission. This makes it possible to make the Doppler aliasing interval variable.

[0578] (a): The code generation unit 104 sets the code length to Walsh-Hadamard code or N in orthogonal M series code CM Orthogonal codes are used for coded multiplexing transmission.

[0579] (b): The code generation unit 104 sets the code length Loc=N CM +δ pseudo-orthogonal code N CM Pseudo-orthogonal codes are used for code multiplexing transmission. δ.

[0580] Therefore, for example, even if the reception levels of the Doppler peaks of multiple targets are at the same distance index f b_cfar If the intervals between Doppler peaks and Doppler aliasing are roughly equal, the probability that the intervals between Doppler aliasing will be different in other radar positioning (e.g., the next radar positioning) can be further increased. As a result, radar device 10 can more reliably separate signals corresponding to multiple targets.

[0581] Furthermore, the radar device 10 can obtain a similar effect by, for example, variably setting the transmission period Tr for each radar positioning operation to change the interval of Doppler aliasing.

[0582] In addition, the structure described in this variation is not limited to the structure of the first embodiment (for example, Figure 1 ), and can also be applied to other variations or implementations (for example, Figure 4 、 Figure 8 、 Figure 10 or Figure 12 For example, the setting of at least one of the transmission period Tr and the code length Loc in this variation can also be applied to the radar devices 10a, 10b, 10c, and 10d.

[0583] Other modifications 1 to 3 have been described above.

[0584] In a radar device according to an embodiment of the present disclosure, the radar transmitter and the radar receiver may be independently configured in physically separate locations. Furthermore, in a radar receiver according to an embodiment of the present disclosure, the direction estimation unit and other components may also be independently configured in physically separate locations.

[0585] Although not shown, a radar device according to one embodiment of the present disclosure includes, for example, a CPU (Central Processing Unit), a recording medium such as a ROM (Read Only Memory) storing a control program, and working memory such as a RAM (Random Access Memory). The functions of each of these components are implemented by the CPU executing the control program. However, the hardware structure of the radar device is not limited to this example. For example, the various functional components of the radar device may be implemented as integrated circuits (ICs). Each functional component may be independently implemented on a single chip, or some or all of them may be integrated on a single chip.

[0586] While various embodiments have been described above with reference to the accompanying drawings, the present disclosure is not limited to these examples. It is obvious that a person skilled in the art will be able to conceive of various variations or modifications within the scope of the claims, and it should be understood that these variations or modifications also fall within the technical scope of the present disclosure. Furthermore, the various components of the above-described embodiments may be arbitrarily combined without departing from the scope of the disclosure.

[0587] In addition, the expression "...part" in the above embodiments can also be replaced by other expressions such as "...circuitry", "...component", "device", "unit" or "module".

[0588] In the above-mentioned embodiments, an example in which the present disclosure is configured using hardware has been described. However, the present disclosure may also be implemented by software in cooperation with hardware.

[0589] In addition, the functional blocks used in the description of the above-mentioned embodiments are typically implemented as an integrated circuit, i.e., an LSI (Large Scale Integration). The integrated circuit can also control the functional blocks used in the description of the above-mentioned embodiments and include input terminals and output terminals. These functional blocks can be independently monolithic or monolithic in a manner that includes some or all of them. Although referred to as "LSI" here, it may also be referred to as "IC", "system LSI", "super LSI", or "extra LSI" depending on the degree of integration.

[0590] Furthermore, circuit integration is not limited to LSIs; dedicated circuits or general-purpose processors can also be used. Field Programmable Gate Arrays (FPGAs), which can be programmed after LSI fabrication, or reconfigurable processors, which allow reconfiguration of the connections and settings of circuit blocks within the LSI, can also be used.

[0591] Furthermore, if semiconductor technology or other derivative technologies develop and a technology for integrated circuits that replaces LSIs emerges, it would be possible to use this technology to integrate functional blocks. There is also the possibility of applying biotechnology, etc.

[0592] <Summary of the present disclosure>

[0593] A radar device according to an embodiment of the present disclosure includes: a signal generation circuit for generating a baseband signal; a code generation circuit for generating multiple code sequences; a phase rotation circuit for adding a phase rotation based on a portion of the multiple code sequences to the baseband signal to generate multiple code-multiplexed transmission signals; and multiple transmitting antennas for respectively transmitting the multiple transmission signals; wherein the code lengths of the multiple code sequences are greater than the code multiplexing number for the multiple transmission signals.

[0594] In one embodiment of the present disclosure, the multiple coding sequences are orthogonal code sequences, and the code length is a power of 2.

[0595] In one embodiment of the present disclosure, the multiple coding sequences are pseudo-orthogonal code sequences, and the code length is a value obtained by adding 1 to the coding multiplexing number.

[0596] In one embodiment of the present disclosure, in a first coding sequence and a second coding sequence included in the multiple coding sequences, the coding elements of one of the odd-numbered coding elements and the even-numbered coding elements are the same, while the signs of the coding elements of the other of the odd-numbered coding elements and the even-numbered coding elements are reversed, and one of the first coding sequence and the second coding sequence is included in other coding sequences that are different from a part of the multiple coding sequences.

[0597] In one embodiment of the present disclosure, each of the plurality of transmitting antennas transmits a transmission signal to which a different phase rotation is added for each positioning performed by the radar apparatus.

[0598] In one embodiment of the present disclosure, the code lengths of the multiple coding sequences are different according to each positioning performed by the radar apparatus.

[0599] In one embodiment of the present disclosure, a transmission period of the plurality of transmission signals is different for each positioning performed by the radar apparatus.

[0600] In one embodiment of the present disclosure, each of the multiple transmitting antennas is a sub-array structure.

[0601] A radar device according to one embodiment of the present disclosure includes: a signal generation circuit for generating a baseband signal; a code generation circuit for generating multiple code sequences; a phase rotation circuit for adding a phase rotation based on a portion of the multiple code sequences to the baseband signal to generate multiple code-multiplexed transmission signals; and multiple transmitting antennas for respectively transmitting the multiple transmission signals; the multiple transmission signals are chirp signals, and the center frequencies of the chirp signals vary according to the transmission period of the transmission signal or the transmission period of the code sequence.

[0602] A radar device according to one embodiment of the present disclosure includes: a signal generation circuit for generating a baseband signal; a code generation circuit for generating multiple code sequences; a phase rotation circuit for adding a phase rotation based on a portion of the multiple code sequences to the baseband signal to generate multiple code-multiplexed transmission signals; and multiple transmitting antennas for respectively transmitting the multiple transmission signals; the multiple transmission signals are chirp signals, and the center frequency of the chirp signals varies in a cycle corresponding to a transmission period of the transmission signal that is approximately several times the code length of the multiple code sequences.

[0603] A radar device according to one embodiment of the present disclosure includes: a receiving antenna for receiving a reflected wave signal, which is a signal reflected from a target by a transmission signal that has been code-multiplexed and transmitted based on a portion of a plurality of coding sequences; and a receiving circuit for determining Doppler frequency domain aliasing in the reflected wave signal based on another coding sequence that is different from the portion of the plurality of coding sequences.

[0604] In one embodiment of the present disclosure, the receiving circuit performs the aliasing determination within a range that is times the code length of the coding sequence compared to a Doppler analysis range for the reflected wave signal.

[0605] In one embodiment of the present disclosure, the receiving circuit performs code separation processing on the reflected wave signal based on the aliasing determination result and the portion of the code sequence.

[0606] The disclosure of Japanese Patent Application No. 2019-208153 filed on November 18, 2019, including the specification, drawings, and abstract, is incorporated herein by reference in its entirety.

[0607] Industrial Applicability

[0608] The present disclosure is suitable as a radar device capable of detecting a wide-angle range.

[0609] Description of Reference Numerals

[0610] 10, 10a, 10b, 10c, 10d Radar devices

[0611] 100, 100b, 100c, 100d, 100e, 100f radar transmitter

[0612] 101, 101b, 101c, 101d radar transmission signal generation unit

[0613] 102 Modulation signal generation unit

[0614] 103, 103b, 103c, 103d VCO

[0615] 104, 104c, 104d Code generation unit

[0616] 105, 105f phase rotation unit

[0617] 106, 106e, 106f Transmitting antennas

[0618] 107, 107c, 107d transmission frequency control unit

[0619] 108, 108f beam weight generation unit

[0620] 109 Beam weight multiplication unit

[0621] 110 Phase Adder

[0622] 200, 200a, 200b, 200c, 200d radar receiving unit

[0623] 201 Antenna System Processing Department

[0624] 202 Receiving Antenna

[0625] 203 Receiving Wireless Unit

[0626] 204 Mixer Section

[0627] 205 LPF

[0628] 206 Signal Processing Department

[0629] 207 AD conversion unit

[0630] 208 Beat Frequency Analysis Department

[0631] 209 Output switching unit

[0632] 210 Doppler Analysis Department

[0633] 211 CFAR Department

[0634] 212, 212a, 212c, 212d Aliasing determination unit

[0635] 213, 213a, 213c, 213d Coded multiplexing separation unit

[0636] 214, 214b, 214c, 214d Direction estimation unit

[0637] 215 Phase Correction Unit

Claims

1. A radar device, characterized in that: include: A signal generating circuit for generating a baseband signal; A code generation circuit generates a plurality of code sequences; a phase rotation circuit that adds a phase rotation based on the plurality of coding sequences to the baseband signal; as well as The baseband signal with the phase rotation added thereto is transmitted as a plurality of code-multiplexed transmission signals in each transmission cycle by a plurality of transmission antennas. The code lengths of the plurality of code sequences change every a predetermined number of the transmission cycles.

2. The radar device according to claim 1, wherein The code lengths of the plurality of code sequences are equal to or greater than a number obtained by adding 2 to the code multiplexing number for the plurality of transmission signals.

3. The radar device according to claim 1, wherein The multiple coding sequences are orthogonal code sequences, and the code length is a power of 2.

4. The radar device according to claim 1, wherein Each of the plurality of transmitting antennas transmits a transmission signal to which a different phase rotation is added for each positioning performed by the radar apparatus.

5. The radar apparatus according to claim 1, wherein The transmission cycles of the plurality of transmission signals are different for each of the predetermined number of transmission cycles.

6. The radar apparatus according to claim 1, wherein Each of the multiple transmitting antennas is a sub-array structure.

7. The radar apparatus according to claim 1, wherein Also includes: a receiving antenna for receiving a reflected wave signal, the reflected wave signal being a signal reflected at a target by a transmission signal that has been code-multiplexed and transmitted based on a portion of the plurality of code sequences; as well as The receiving circuit determines aliasing in the Doppler frequency domain in the reflected wave signal based on another code sequence that is different from the part of the plurality of code sequences.

8. A radar signal processing method, characterized in that: In the method, Generate baseband signal, Generate multiple coding sequences, adding a phase rotation based on the plurality of code sequences to the baseband signal, The baseband signal with the phase rotation added thereto is transmitted as a plurality of code-multiplexed transmission signals in each transmission cycle through a plurality of transmission antennas; The code lengths of the plurality of code sequences change every a predetermined number of the transmission cycles.

9. The radar signal processing method according to claim 8, wherein: The code lengths of the plurality of code sequences are equal to or greater than a number obtained by adding 2 to the code multiplexing number for the plurality of transmission signals.

10. The radar signal processing method according to claim 8, wherein: The multiple coding sequences are orthogonal code sequences, and the code length is a power of 2.

11. The radar signal processing method according to claim 8, wherein: Each of the plurality of transmitting antennas transmits a transmission signal to which a different phase rotation is added for each positioning performed by the radar apparatus.

12. The radar signal processing method according to claim 8, wherein: The transmission cycles of the plurality of transmission signals are different for each of the predetermined number of transmission cycles.

13. The radar signal processing method according to claim 8, wherein: Each of the multiple transmitting antennas is a sub-array structure.

14. The radar signal processing method according to claim 8, wherein: receiving, via a receiving antenna, a reflected wave signal, the reflected wave signal being a signal reflected at a target by a transmission signal that has been code-multiplexed and transmitted based on a portion of the plurality of code sequences; The determination of aliasing in the Doppler frequency domain in the reflected wave signal is performed based on another code sequence that is different from the part of the plurality of code sequences.

15. A radar signal processing circuit, characterized in that: include: A signal generating circuit for generating a baseband signal; A code generation circuit generates a plurality of code sequences; a phase rotation circuit that adds a phase rotation based on the plurality of coding sequences to the baseband signal; as well as A plurality of transmitting circuits transmit the baseband signal with the phase rotation added thereto as a plurality of code-multiplexed transmission signals in each transmission cycle through a plurality of transmitting antennas, The code lengths of the plurality of code sequences change every a predetermined number of the transmission cycles.

16. The radar signal processing circuit according to claim 15, wherein: The code lengths of the plurality of code sequences are equal to or greater than a number obtained by adding 2 to the number of code multiplexing operations for the plurality of transmission signals.

17. The radar signal processing circuit according to claim 15, wherein: The multiple coding sequences are orthogonal code sequences, and the code length is a power of 2.

18. The radar signal processing circuit according to claim 15, wherein: Each of the plurality of transmitting antennas transmits a transmission signal to which a different phase rotation is added for each positioning performed by the radar apparatus.

19. The radar signal processing circuit according to claim 15, wherein: The transmission cycles of the plurality of transmission signals are different for each of the predetermined number of transmission cycles.

20. The radar signal processing circuit according to claim 15, wherein: Each of the multiple transmitting antennas is a sub-array structure.

21. The radar signal processing circuit according to claim 15, wherein: The device further includes a receiving circuit, which receives a reflected wave signal through a receiving antenna. The reflected wave signal is a signal reflected at a target by a transmission signal that has been code-multiplexed and transmitted based on a portion of the multiple coding sequences. The receiving circuit determines Doppler frequency domain aliasing in the reflected wave signal based on another coding sequence that is different from the portion of the multiple coding sequences.

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