Multi-slot antenna and low-profile phased-array antenna using same

Abstract

The invention provides a multi-slot antenna. The multi-slot antenna comprises a grounding plate, a driving slot, a first parasitic slot, a second parasitic slot, a third parasitic slot and a fourth parasitic slot, the driving gap is located on the grounding plate and extends in the first direction. The first, second, third and fourth parasitic slots are all located on the ground plate, each parasitic slot extending in a first direction and a second direction different from the first direction. And the first parasitic slot and the third parasitic slot are symmetrical relative to the driving slot. The fourth parasitic slot is located on the grounding plate and located on the second side of the driving slot. And the second parasitic gap and the fourth parasitic gap are symmetrical relative to the driving gap.

Description

Technical Field

[0001] The present invention relates to communication technology, in particular to a multi-slot antenna for realizing wide-range beam control in communication and a low-profile phased array antenna using the same. Background Art

[0002] With the rapid development of intelligent transportation systems, vehicle-to-everything (V2X) communications have garnered significant attention in both industry and academia. V2X technology facilitates wireless communications between various devices, including vehicle-to-vehicle (V2V), vehicle-to-network (V2N), and vehicle-to-infrastructure (V2I). Antennas, as key components in wireless communication systems for transmitting and receiving electromagnetic waves (EM waves), significantly impact communication quality and stability.

[0003] Figure 1 This demonstration showcases the application of vehicle-mounted antennas in V2X communications. Because the relative positions of vehicles and devices change rapidly, flexible beam steering capabilities are crucial for providing reliable, real-time connectivity for V2X systems. Compared to traditional fixed-beam antennas, beam-steering antennas typically offer higher spatial resolution and longer communication ranges. Phased array antennas are a common method for achieving flexible beam steering.

[0004] Traditional phased array antennas typically experience significant gain loss at larger angles, resulting in a limited scanning range. This is primarily due to strong coupling between antenna elements and the narrow beam of their radiation patterns. Therefore, the current effective development strategy is to design wide-beam antenna elements that can extend the scanning range. Various techniques have been investigated to achieve wide-beam antenna elements. For example, wide beamwidth can be achieved by combining different resonant modes with complementary radiation patterns. Some literature has explored adding two metal strips to either side of a high-frequency patch antenna to enhance low-elevation-angle radiation by utilizing the vertical currents flowing through the strips. Other literature has also explored adding two suspended equivalent magnetic fluxes to achieve a radiation pattern complementary to the original forward radiation pattern. Recently, a wide-beam dielectric resonator antenna (DRA) has been designed by combining metal and dielectric loading. However, all of these wide-beam designs have a high profile.

[0005] In addition to static wide beams, dynamically reconfigurable wide beams also have practical value in phased array antenna design. Different radiation patterns of a reconfigurable design can together provide a wider scanning range. In some related literature, PIN diodes can be used to switch between two resonant modes with complementary radiation patterns. In another approach, a two-port DRA has been studied, which achieves a reconfigurable radiation pattern by adjusting the phase of the two ports. However, the use of reconfigurable radiation pattern technology will inevitably increase the design complexity and its associated costs. In addition, the overall profile of the reconfigurable antenna is usually higher.

[0006] In vehicular communications, low-profile antennas are often required to reduce wind resistance. By placing a high-impedance surface (HIS) or artificial magnetic conductor (AMC) beneath the horizontal dipole, a low-profile horizontal dipole antenna with a wide H-plane beamwidth can be achieved. When using a radiating slot or equivalent magnetic dipole as the primary radiation source, the low-profile characteristics can be maintained using a conventional metal reflector or a simple metasurface. However, such designs are typically implemented using multilayer printed circuit boards (PCBs), resulting in design costs and complexity that make them difficult to implement in practice. Furthermore, studies have explored designs that use single-layer substrate microstrip antennas to achieve wide-angle scanning. Related literature also investigates planar wide-beam multipole antennas based on single-layer substrates. These multipole antennas are further used as unit components to construct phased arrays with wide beam steering characteristics. Despite the simpler structures of some array designs, their performance has not met expectations.

[0007] Therefore, there is an urgent need to develop a phased array design for vehicular communications that meets the characteristics of low cost and low profile while maintaining competitive beam scanning performance. Summary of the Invention

[0008] The object of the present invention is to provide an apparatus and method to address the above-mentioned deficiencies and unmet needs in the prior art.

[0009] According to a first aspect of the present invention, a multi-slot antenna is provided, comprising a ground plane, a driven slot, a first parasitic slot, a second parasitic slot, a third parasitic slot, and a fourth parasitic slot. The driven slot is disposed on the ground plane and extends along a first direction. The first parasitic slot is disposed on the ground plane and located on a first side of the driven slot, wherein the first parasitic slot extends along the first direction and a second direction different from the first direction. The second parasitic slot is disposed on the ground plane and located on the first side of the driven slot, wherein the second parasitic slot extends along the first and second directions. The third parasitic slot is disposed on the ground plane and located on a second side of the driven slot, opposite the first side. A first end of the driven slot is located between the first and third parasitic slots, and the third parasitic slot extends along at least the first and second directions, with the first and third parasitic slots being symmetrical with respect to the driven slot. The fourth parasitic slot is disposed on the ground plane and located on the second side of the driven slot. The second end of the driving gap is opposite to the first end and is located between the second parasitic gap and the fourth parasitic gap. The fourth parasitic gap extends at least along the first direction and the second direction. The second parasitic gap and the fourth parasitic gap are symmetrical with respect to the driving gap.

[0010] According to a second aspect of the present invention, a multi-slot antenna array is provided. The multi-slot antenna array includes a plurality of multi-slot antennas, and the plurality of multi-slot antennas are arranged continuously in a horizontal direction.

[0011] Through this structure, a single-layer substrate-type multi-slot unit component with a wide H-plane beam width is provided. The wide-beam unit component is used to design an H-plane phased array and demonstrates a significant beam control range from -90° to +90°. In addition, a low-profile linear phased array antenna with a wide scanning range is provided. Each array unit component deploys multiple slots with a wide beam. These multiple slots can be divided into y-direction magnetic currents in the same direction and four equal-amplitude x-direction magnetic currents in different directions. The y-direction can provide a forward radiation pattern, while the x-direction enhances low-elevation-angle far-field radiation. Therefore, a flexible wide-beam H-plane radiation pattern can be obtained by combining these two types of magnetic currents. BRIEF DESCRIPTION OF THE DRAWINGS

[0012] The embodiments of the present invention are described in more detail below with reference to the accompanying drawings, in which:

[0013] Figure 1 A diagram of the application scenario of vehicle-mounted antennas for V2X communication is presented;

[0014] Figure 2A and Figure 2B The configuration diagrams of single-slot antenna and four-slot antenna are presented;

[0015] Figure 3A 、 Figure 3B 、 Figure 3C and Figure 3D Showed Figure 2A and Figure 2B The simulated current distribution diagram of each gap in the configuration on the ground and the equivalent magnetic flow diagram in the gap;

[0016] Figure 4A The simulation results of the YOZ plane normalized radiation pattern of the single slot antenna are presented;

[0017] Figure 4B The simulation results of the YOZ plane normalized radiation pattern of the four-slot antenna are presented;

[0018] Figure 5 The transverse electric field diagram in the slots of the four-slot antenna is presented to further illustrate the relationship between the electric field and the magnetic field.

[0019] Figure 6 The YOZ plane radiation patterns generated by the combination of the first magnetic current and the second magnetic current with different coefficients are shown;

[0020] Figure 7A A top view schematically shows a basic multi-slot antenna according to an embodiment of the present invention;

[0021] Figure 7B The simulated reflection coefficient diagram of the basic multi-slot antenna at different driving slot lengths according to an embodiment of the present invention is presented;

[0022] Figure 8A and Figure 8B A top view and a bottom view of an improved multi-slot antenna according to an embodiment of the present invention are respectively shown;

[0023] Figure 9 The simulated current distribution diagram and equivalent magnetic flux diagram at 8.6 GHz are presented, where (a) is the current distribution diagram of the basic multi-slot antenna; (b) is the equivalent magnetic flux diagram of the basic multi-slot antenna; (c) is the current distribution diagram of the improved multi-slot antenna; (d) is the equivalent magnetic flux diagram of the improved multi-slot antenna;

[0024] Figure 10 Shown for different L s3 Simulated normalized yoz plane radiation pattern of the proposed multi-slot antenna;

[0025] Figure 11 The basic multi-slot antenna is shown in L s1 =13.4 mm and the simulated reflection coefficient diagram of the improved multi-slot antenna under different L conditions;

[0026] Figure 12 Pictures of the proposed multi-slot component prototype were presented, where part (a) shows a perspective view and part (b) shows a top view;

[0027] Figure 13 Measurements and simulated reflection coefficient graphs of the component prototype were presented, showing good agreement between them;

[0028] Figure 14 Measurements and simulated radiation pattern graphs of the component prototype at 8.6 GHz were presented;

[0029] Figure 15A and Figure 15B Top and bottom view schematic diagrams of the antenna array according to an embodiment of the present invention were presented respectively;

[0030] Figure 16 Pictures of the multi-slot phased array prototype were presented, where part (a) is a perspective view, part (b) is a top view, and part (c) is a measurement environment diagram in anechoic chamber;

[0031] Figure 17 Measurements and simulated reflection coefficient graphs of components 1, 3 and 5 were presented, as well as an isolation diagram between components 4 and 5, that is, the measurement and simulation results of the multi-slot phased array antenna, such as |S 11 |, |S 33 |, |S 55 | and |S <� 45 |: <� <�

[0032] <� Figure 18 Measurements and simulated active VSWR graphs were presented, where part (a) is the simulation result of component 1, part (b) is the measurement result of component 1, part (c) is the simulation result of component 5, part (d) is the measurement result of component 5, part (e) is the simulation result of component 9, and part (f) is the measurement result of component 9; <� <��

[0033] <�� Figure 19 Measurement and simulation result graphs of the H-plane beam control performance of the multi-slot phased array under the condition of 8.6 GHz were presented; and <� <�

[0034] <� Figure 20 A table comparing the array design of the present invention with an existing wide-angle beam control linear phased array was presented. <� Detailed implementation manners <� <�

[0035] In the following description, multi-slot antennas and low-profile phased array antennas that utilize them to achieve a wide range of control in communications are described as preferred examples. Those skilled in the art will appreciate that modifications, including additions and / or substitutions, may be made without departing from the scope and spirit of the present invention. Specific details may be omitted to avoid obscuring the present invention; however, this disclosure is written to enable those skilled in the art to practice the teachings herein without undue experimentation.

[0036] Wide-angle scanning phased array antennas have been widely used in diverse fields, including radar, satellite communications, automotive sensing, and mobile communications. Wide-beam antenna unit assemblies can effectively expand the scanning range. However, existing research on wide-beam antenna unit assemblies for wide-angle scanning phased array antennas has at least one of the following drawbacks: prominent appearance, large footprint, complex structure, high cost, and limited scanning range. The present invention discloses a multi-slot antenna and a multi-slot antenna array using the same, designed to address these drawbacks.

[0037] First, the operating principle of hybrid magnetic flux technology is introduced. By combining two complementary radiation patterns, a wide-beam radiation pattern can be achieved. Slot antennas were chosen for their low profile and single-layer substrate structure. Traditional slot antennas typically have a forward radiation pattern. Therefore, achieving an end-on radiation pattern from a slot antenna was expected to be challenging.

[0038] Figure 2A and Figure 2B The configurations of a single slot antenna 10 and a quad-slot antenna 20 are shown. The single slot antenna 10 is fed via a microstrip line, while the quad-slot antenna 20 is directly excited by a lumped port using a high-frequency structure simulator (HFSS). Figure 3A 、 Figure 3B 、 Figure 3C and Figure 3D Showed Figure 2A and Figure 2B The simulated current distribution on the ground and the equivalent magnetic flux in the gap for each gap configuration in . Specifically, Figure 3A The current distribution diagram of the single slot antenna 10 is presented; Figure 3B The equivalent magnetic flux diagram of the single slot antenna 10 is presented; Figure 3C The current distribution diagram of the four-slot antenna 20 is presented; Figure 3D An equivalent magnetic flux diagram of the four-slot antenna 20 is shown.

[0039] like Figure 3A As shown in Figure 1, the current around the gap has components in both the x-direction and the y-direction. However, the y-direction currents in opposite directions cancel each other out, leaving only the x-direction current. Figure 3B The y-direction equivalent magnetic current in the y-direction is orthogonal. Therefore, the E field of the single slot antenna 10 in the yoz plane is φ-polarized. Figure 4A This can be verified by the simulation results of the YOZ plane normalized radiation pattern of the single slot antenna 10.

[0040] For the four-slot antenna 20, a more complicated situation occurs. Figure 3C , each slot of the four-slot antenna 20 can be modeled as a magnetic current in the x-direction. These four magnetic currents have the same amplitude, but the directions of the two adjacent magnetic currents are opposite, as shown in Figure 3D As shown in Figure 2 . For each magnetic current, its radiation field in the yoz plane is θ-polarized. However, because these magnetic currents are directed in opposite directions, their θ-polarized field components in the yoz plane cancel each other, leaving only their cross-polarized fields (φ-polarized components), which effectively become new co-polarized fields. In other words, the co-polarized and cross-polarized field components of the quad-slot antenna 20 are interchanged with the field components of a single x-slot antenna. Figure 4B The YOZ plane normalized radiation pattern of the four-slot antenna 20 is presented and the analysis is verified.

[0041] Figure 5 The transverse electric field in the slots of the four-slot antenna 20 is shown to further illustrate the relationship between the electric field and the magnetic field components. As shown in the figure, the electric field vectors are all along the y direction, and Figure 3D The x-direction equivalent magnetic flux shown in the figure is orthogonal. At the same time, it can be more clearly observed that the phases between any two adjacent gaps are opposite.

[0042] refer to Figure 4A and Figure 4B , it can be found that the maximum directivity of the slot antenna 10 and the four-slot antenna 20 appears in the forward direction (θ = 0°) and the end direction (θ = ±90°), respectively. In other words, the two radiation patterns are complementary within the wide beam angle range covered on the yoz plane. Here, the equivalent magnetic currents of the slot antenna 10 and the four-slot antenna 20 can be named the first magnetic current (MC1) and the second magnetic current (MC2), respectively. By superimposing the radiation patterns of these two magnetic currents with different weighting coefficients, a flexible beam shape can be obtained, as shown in the following formula: G(θ)=A·G1(θ)+(1-A)·G2(θ)…Formula (1) Wherein, G(θ) is the synthesized radiation pattern, G1(θ) and C2(θ) are the individual radiation patterns of the first magnetic current (MC1) and the second magnetic current (MC2), respectively, and their weighting coefficients are A and (1-A), respectively.

[0043] In order to better understand formula (1), Figure 6The synthetic YOZ plane radiation pattern generated by the first magnetic current and the second magnetic current at different coefficients A (i.e., the combination of the YOZ plane radiation pattern generated by MC1 and MC2 with different coefficients A) is shown. In order to directly use formula (1), the radiation pattern has been converted to a linear form. Figure 6 The resulting radiation patterns exhibit wide beams with varying center tilt angles. When the coefficient A = 0.5, the field strength in the central region can exceed 0.8, resulting in minimal center ripple and, consequently, gain fluctuations of less than 1 dB. Therefore, a flexible wide beam can be achieved by simultaneously exciting the first and second magnetic currents with controllable amplitudes.

[0044] Based on the mechanism discussed above, a design of a flexible wide-beam multi-slot antenna unit component is proposed. Figure 7A A top view of a multi-slot antenna 100A according to an embodiment of the present invention is shown. Figure 7A Also shown is a first direction D1 and a second direction D2. The first direction D1 is different from the second direction D2; in one embodiment, the first direction D1 and the second direction D2 are orthogonal to each other. For example, the first direction D1 can be a horizontal direction y, and the second direction D2 can be a vertical direction x.

[0045] The structure of the multi-slot antenna 100A can be composed of a single slot antenna 10 and a four-slot antenna 20. Specifically, the multi-slot antenna 100A includes a printed circuit board (PCB) 110, a driven slot 120, a first parasitic slot 130, a second parasitic slot 140, a third parasitic slot 150, and a fourth parasitic slot 160.

[0046] PCB 110 may include a ground plane 113 and a dielectric substrate (not shown), wherein ground plane 113 is disposed on the dielectric substrate. In one embodiment, ground plane 113 comprises a conductive material, such as a metal or an alloy. In one embodiment, PCB 110 is composed of ground plane 113 and a dielectric substrate, which is also referred to as a single-substrate structure.

[0047] Drive slot 120, first parasitic slot 130, second parasitic slot 140, third parasitic slot 150, and fourth parasitic slot 160 are located on ground plane 113. In one embodiment, parasitic slots can be formed by etching ground plane 113 so that the parasitic slots can be located on ground plane 113.

[0048] The driving slot 120 is located on the ground plate 113 and extends along the first direction D1. The driving slot 120 is located at the center of the ground plate 113. Here, "located at the center of the ground plate 113" means that the center of the driving slot 120 (e.g., the center of mass of the driving slot 120) coincides with the center of mass of the ground plate 113.

[0049] The first parasitic slot 130 , the second parasitic slot 140 , the third parasitic slot 150 and the fourth parasitic slot 160 can be excited by the driving slot 120 at the center in a desired phase relationship.

[0050] The first parasitic slot 130 and the second parasitic slot 140 are located on the ground plate and on one side 121 of the driving slot 120 (e.g., above the driving slot 120). The first parasitic slot 130 and the second parasitic slot 140 extend along the second direction D2, so the second parasitic slot 140 may be parallel to the first parasitic slot 130.

[0051] The third parasitic slot 150 and the fourth parasitic slot 160 are located on the ground plate and on the other side 123 of the driving slot 120, opposite to the first side 121 (e.g., below the driving slot 120). The third parasitic slot 150 and the fourth parasitic slot 160 extend along the second direction D2, so the fourth parasitic slot 160 can be parallel to the third parasitic slot 150.

[0052] Furthermore, the driving slot 120 has a first end 125 and a second end 127 that are opposite to each other. The first end 125 of the driving slot 120 is located between the first parasitic slot 130 and the third parasitic slot 150, wherein the first parasitic slot 130 and the third parasitic slot 150 are symmetrical with respect to the driving slot 120. The second end 127 of the driving slot 120 is located between the second parasitic slot 140 and the fourth parasitic slot 160, wherein the second parasitic slot 140 and the fourth parasitic slot 160 are symmetrical with respect to the driving slot 120.

[0053] With such a configuration, on the ground plate 113 , a combination of the driving slot 120 , the first parasitic slot 130 , the second parasitic slot 140 , the third parasitic slot 150 , and the fourth parasitic slot 160 presents an H-shape.

[0054] Figure 7B The simulated reflection coefficient diagram of the multi-slot antenna 100A of the embodiment of the present invention at different driving slot lengths is shown. Figure 7B As shown, by increasing the length L of the driving gap 120 along the direction D1 s1 , can make the driven gap mode closer to the parasitic gap mode. s1 = 15.4 mm, the driven slot mode and the parasitic slot mode merge, thus achieving a wider impedance bandwidth. However, the larger L s1 This means that the radiator size in the y direction is larger, which will limit the choice of unit component spacing in array design.

[0055] For phased array antennas with wide scanning range, the unit element spacing is usually required to be less than half a wavelength to avoid grating lobe problems. In this case, the electrical length of the driving slot 120 is about half a wavelength, which will result in a relatively large unit element spacing in the array design. In order to achieve a smaller unit element spacing, L can be selected. s1 = 13.4 mm, and finally the antenna is designed to operate only in parasitic slot mode.

[0056] about Figure 7B L in s1 = 13.4 mm, the parasitic slot mode operates at 8.6 GHz, with a size in the y-direction of 0.38λ0, where λ0 represents the wavelength in a vacuum at a given frequency. However, because the driven slot mode resonates at a higher frequency (9.75 GHz), the weighting coefficient of the y-direction magnetic flux (MC1) is smaller at 8.6 GHz, resulting in a large central dip in the radiation pattern in the yoz plane.

[0057] To further improve performance, an L-shaped parasitic slot can be used for beam adjustment. More specifically, to enhance the effect of the y-direction magnetic current, an L-shaped slot is used in the antenna design instead of a straight parasitic slot.

[0058] Figure 8A and Figure 8B A top view and a bottom view of a multi-slot antenna 100B according to an embodiment of the present invention are shown, respectively. Compared to the multi-slot antenna 100A, the multi-slot antenna 100B can be used as an improved model or a multi-slot antenna (that is, the multi-slot antenna 100A can be used as a basic model or a basic multi-slot antenna). It should be noted that although the improved multi-slot antenna 100B can be used as a next-generation model of the multi-slot antenna 100A, any other suitable modification or optimization of the multi-slot antenna 100B is also feasible and permissible. For ease of understanding, Figure 8A and Figure 8B A first direction D1 and a second direction D2 are also indicated. The first direction D1 is different from the second direction D2; in one embodiment, the first direction D1 and the second direction D2 are orthogonal to each other. For example, the first direction D1 can be the horizontal direction y, and the second direction D2 can be the vertical direction x.

[0059] The multi-slot antenna 100B has a similar configuration to the multi-slot antenna 100A, but differs in that the first parasitic slot 130 , the second parasitic slot 140 , the third parasitic slot 150 , and the fourth parasitic slot 160 of the multi-slot antenna 100B are all L-shaped.

[0060] Specifically, each of the first parasitic slot 130, the second parasitic slot 140, the third parasitic slot 150, and the fourth parasitic slot 160 extends along the first direction D1 and the second direction D2, thereby forming an L-shape. These slots (including the driving slot 120, the first parasitic slot 130, the second parasitic slot 140, the third parasitic slot 150, and the fourth parasitic slot 160) can be formed by etching on the ground plane 113 of the PCB 110, that is, they can be printed on the top surface of the PCB 110.

[0061] The driving slot 120 is located on the ground plate 113 and extends along the first direction D1. The driving slot 120 is located at the center of the ground plate. The driving slot 120 can be used as a driving slot for microstrip feeding and is etched at the center of the ground plate 113. The driving slot 120 has a length L along the first direction D1. s1 , which ranges from 10 mm to 15 mm, and has a width W along the second direction D2 s1 , which ranges from 0.15 mm to 0.25 mm. In one embodiment, the length L s1 About 13.4 mm, width W s1 About 0.2 mm.

[0062] In this embodiment, each of the first parasitic slot 130, the second parasitic slot 140, the third parasitic slot 150, and the fourth parasitic slot 160 includes a first extension portion 131, 141, 151, 161 and a second extension portion 133, 143, 153, 163. The following description takes the first extension portion 131 and the second extension portion 133 of the first parasitic slot 130 as an example, and the description is equally applicable to the other first extension portions 141, 151, 161 and second extension portions 143, 153, 163.

[0063] In the first parasitic slot 130, a first extension portion 131 extends along a first direction D1, and a second extension portion 133 is connected to the first extension portion 131 and extends along a second direction D2. The first extension portion 131 is shorter than the second extension portion 133 and is closer to the driving slot 120 than the distal end of the second extension portion 133, forming an L-shape. To further define the L-shape, the first extension portion 131 is closer to the center of the driving slot 120 than the second extension portion 133.

[0064] The first extending portion 131 has a length L along the first direction D1 s3 , which is between 1 mm and 1.5 mm. The second extension portion 133 has a length L along the second direction D2. s2 , which is between 5 mm and 15 mm, and has a width W along the first direction D1 s2, which is between 0.1 mm and 0.3 mm. In one embodiment, the length L s2 About 10 mm, width W s2 The L-shaped first parasitic gap 130 is about 0.2 mm, and the length L is about 1.2 mm. In addition, there is a gap between the L-shaped first parasitic gap 130 and the driving gap 120, which ranges from 0.5 mm to 1.5 mm. In one embodiment, the gap is about 1.05 mm.

[0065] The first extension portion 131 of the first parasitic slot 130 and the first extension portion 141 of the second parasitic slot 140 are located between the second extension portion 133 of the first parasitic slot 130 and the second extension portion 143 of the second parasitic slot 140. Therefore, the profiles / outlines of the L-shaped first parasitic slot 130 and the L-shaped second parasitic slot 140 are symmetrical with respect to the vertical axis of the driving slot 120, which means that the profiles / outlines of the L-shaped first parasitic slot 130 and the L-shaped second parasitic slot 140 are opposite (i.e., after horizontally mirroring the L-shaped first parasitic slot 130, its profile / outline is the same as the L-shaped second parasitic slot 140).

[0066] The first extension portion 131 of the first parasitic slot 130 and the first extension portion 151 of the third parasitic slot 150 are located between the second extension portion 133 of the first parasitic slot 130 and the second extension portion 153 of the third parasitic slot 150. Therefore, the profiles / outer shapes of the L-shaped first parasitic slot 130 and the L-shaped third parasitic slot 150 are symmetrical about the horizontal axis of the driving slot 120, which means that the profiles / outer shapes of the L-shaped first parasitic slot 130 and the L-shaped third parasitic slot 150 are opposite (i.e., after vertically mirroring the L-shaped first parasitic slot 130, its profile / outer shape is the same as the L-shaped third parasitic slot 150).

[0067] With this configuration, the first parasitic slot 130 and the third parasitic slot 150 are L-shaped and symmetrical with respect to the driving slot 120. The second parasitic slot 140 and the fourth parasitic slot 160 are also L-shaped and symmetrical with respect to the driving slot 120. In one embodiment, there are four identical L-shaped parasitic slots above and below the driving slot 120, and they are symmetrically etched on the ground plane 113.

[0068] Regarding the bottom view of the multi-slot antenna 100B, as shown in FIG. Figure 8BAs shown, the multi-slot antenna 100B further includes a microstrip line 170 for feeding the driven slot 120, located at the center of the dielectric substrate 115 of the PCB 110. The multi-slot antenna 100B also includes an SMP ground mounting area 172 for mounting a connector (not shown). The mounting area includes a plurality of ground metal vias 174 and solder pads 176. In one embodiment, the length L1 of the microstrip line 170 along the second direction D2 is approximately 8.4 mm, and the width W1 along the first direction D1 is approximately 1.82 mm.

[0069] In one embodiment, the PCB 110 has a square shape with a side length L g The thickness is about 30 mm, the thickness is about 0.813 mm, and the dielectric constant of the substrate is 3.38, thus forming a PCB with a single-layer substrate structure.

[0070] The parameters of the multi-slot antenna 100B are listed in Table I. Table I

[0071] The multi-slot antenna 100B can be designed for X-band and simulated using ANSYS HFSS. Figure 9 The simulated current distribution diagram and equivalent magnetic flux diagram at 8.6 GHz are presented, where part (a) is the current distribution diagram of the multi-slot antenna 100A; part (b) is the equivalent magnetic flux diagram of the multi-slot antenna 100A; part (c) is the current distribution diagram of the multi-slot antenna 100B; and part (d) is the equivalent magnetic flux diagram of the multi-slot antenna 100B.

[0072] like Figure 9 As shown in parts (a) and (c) of the diagram, the parasitic gap achieves effective coupling through the central driven gap. The corresponding equivalent configurations are shown in Figure 9 Parts (b) and (d) of Figure 9 From part (b), we can observe that Figure 9 The equivalent configuration of part (a) at 8.6 GHz consists of four magnetic currents (MC2) along the x-direction and a weaker magnetic current (horizontal dashed vector) along the y-direction. Figure 9 Part (d) shows Figure 9 The equivalent configuration of part (c) is similar to that of the L-shaped gap, except that a strong magnetic flux (MC1) along the y direction appears at the center due to the presence of the additional horizontal arm (L) of the L-shaped gap. It can be seen that the four L-shaped magnetic fluxes can be divided into MC1 and MC2. In addition, by adjusting L s3 , the weight coefficient A of MC1 can be flexibly controlled.

[0073] Figure 10 Shown for different L s3Simulated normalized yoz plane radiation patterns of the proposed multi-slot antenna 100A and multi-slot antenna 100B. For this purpose, the total length of the L-shaped slot is kept constant. The results show that there is a large central depression in the radiation pattern of the multi-slot antenna 100A. By increasing the horizontal arm (L) of the L-shaped slot in the multi-slot antenna 100B, the radiation pattern of the multi-slot antenna 100A is significantly improved. s3 ), this problem can be alleviated. The reason is that the y-direction magnetic current of the horizontal arm (L) enhances the forward radiation. It should be noted that as L s3 The resonant frequency and impedance matching are not significantly affected by the change of Figure 11 As shown, Figure 11 Shown in L s1 =13.4 mm, the multi-slot antenna 100A and different L s3 The simulated reflection coefficient diagram of the multi-slot antenna 100B under the condition of φ(A) is shown in FIG. It can be seen that this will greatly facilitate the design of the antenna.

[0074] The experimental results and discussions are described below.

[0075] Figure 12 Photos of the proposed multi-slit component prototype are shown, where part (a) shows a perspective view and part (b) shows a top view. Figure 12 The aforementioned parameter values ​​were applied. In addition, an RF choke was used to prevent backflow current to the SMP connector and feeder cable. The optimized outer diameter, height, and thickness used here were 12 mm, 7.5 mm, and 1 mm, respectively.

[0076] Figure 13 Measured and simulated reflection coefficient plots for the prototype component are presented, demonstrating good agreement. As can be seen from the plot, the measured and simulated 10-dB impedance bandwidths are 5.2% (8.44-8.89 GHz) and 3.1% (8.49-8.76 GHz), respectively. Referring again to the accompanying figure, the measured and simulated co-polarization gains are 2.7 dBi and 2.9 dBi, respectively, while the cross-polarization gains are both below -15 dBi (within the impedance bandwidth). It is noteworthy that the difference between the measured and simulated cross-polarization gain is greater than that of the co-polarization gain, as the cross-polarization component is generally more sensitive to manufacturing and experimental tolerances.

[0077] Figure 14 Measured and simulated radiation patterns of the prototype component at 8.6 GHz are shown. The co-polarized H-plane 3-dB beamwidth is very wide, with measured and simulated values ​​of 217° and 218°, respectively. As can be seen from the figure, the E-plane and H-plane cross-polarization levels are at least 15 dB lower than the co-polarization levels within their respective 3-dB beamwidths.

[0078] In various embodiments, a low-profile multi-slot linear phased array antenna is also provided. The multi-slot antenna 100B described above can be applied to the design of a linear H-plane phased array antenna using the same type of PCB (i.e., a single-layer dielectric structure). Notably, when the antenna array is vertically mounted on a large metal plate, such as a vehicle roof, its scanning range is largely unaffected.

[0079] Figure 15A and Figure 15B The top view and bottom view of the antenna array 200 according to an embodiment of the present invention are shown respectively. The configuration or structure of the antenna array 200 can adopt the structure of the multi-slot antenna 100B, further forming the configuration of the antenna array 200, wherein the unit component spacing is p=0.46λ0 and the length is L g , width W g The antenna array 200 can be designed to operate at the same frequency as the multi-slot antenna 100B. In some embodiments, the dimensional parameters of the multi-slot antenna unit components (e.g., the multi-slot antenna 100B) can be adjusted to parameters different from those previously described. For example, Table II lists the optimized design parameters for the antenna array 200, where the definitions of these parameters correspond to those in Table I above. Table II

[0080] In the antenna array 200, multi-slot antenna element components (e.g., multi-slot antenna 100B) can be arranged continuously in a horizontal direction and share the same substrate (e.g., a ground plane). In one embodiment, the parameters listed in Table II are applied to the antenna array 200 including nine multi-slot antenna element components. In other words, the parameters listed in Table II are set for the case where the antenna array 200 is composed of nine multi-slot antenna element components.

[0081] The experimental results and discussions of the antenna array are described as follows. Figure 16 Photos of the multi-slot phased array prototype are shown, where (a) is a perspective view, (b) is a top view, and (c) is a diagram of the measurement environment in an anechoic chamber. Figure 16 Photos of the array prototype and its testing environment are shown. RF chokes are used in the array to prevent unwanted current flow on the cable surface.

[0082] First, the reflection coefficient of the antenna array is studied. Since the array is symmetrical, only the leftmost unit cell (unit cell 1), the center unit cell (unit cell 5), and the unit cell between them (unit cell 3) are discussed here. Figure 17The measured and simulated reflection coefficient diagrams of components 1, 3, and 5, as well as the isolation diagram between components 4 and 5, that is, the measurement and simulation results of the multi-slot phased array antenna, such as |S 11 |, |S 33 |, |S 55 | and |S 45 |. The overlapping parts of the measured and simulated 10-dB bandwidths of the three unit components are 4.4% (8.46 - 8.84 GHz) and 2.6% (8.5 - 8.72 GHz), respectively. In addition, the measured and simulated |S 45 | is below -15 dB within the relevant bandwidth. Compared with the simulation results, the measured results show a 0.5% frequency upshift due to manufacturing and assembly errors.

[0083] In practical applications, the active reflection coefficient is an excellent indicator for evaluating the impedance matching degree in beam control. Here, unit component 1, unit component 5, and unit component 9 in the array are considered because the active reflection coefficient of the array is no longer symmetric; the reflection coefficients of the leftmost and rightmost unit components are usually not equal at different control angles. The active reflection coefficient of the m-th unit component can be calculated by the following formula: where S mn , k, d, and θ are the passive S parameter, wave number, element spacing, and scan angle, respectively. Equation (2) is used to calculate the active reflection coefficients of each element at different scan angles.

[0084] Figure 18 The effective VSWR diagrams of the measurement and simulation are presented, where part (a) is the simulation result of component 1, part (b) is the measurement result of component 1, part (c) is the simulation result of component 5, part (d) is the measurement result of component 5, part (e) is the simulation result of component 9, and part (f) is the measurement result of component 9. Figure 18 The active voltage standing wave ratio (VSWR) in the experiments of components 1, 5, and 9 at different scan angles is shown, and the differences are due to the imperfect conditions in the experiments. The figure shows that the active VSWR deteriorates at larger control angles due to the strong coupling between array components. At the center frequency (8.6 GHz), it can be observed that the active VSWR of three components is less than 3.

[0085] For a linear array with uniform amplitude and spacing, the progressive phase β required for a specific main beam direction θ is defined as: β = -kd sinθ… Equation (3)

[0086] Therefore, the theoretical progressive phase of the phased array at different scanning angles can be calculated using formula (3). Table III compares the theoretical progressive phase difference and the designed phase difference of the phased array at 8.6 GHz. The designed phase difference can be obtained by optimizing the simulation of the main beam arrival angle using ANSYS HFSS. It can be observed from Table III that at larger scanning angles, there is a small deviation between the theoretical phase difference and the designed phase difference. This is because the multi-slit array of the present invention is a finite small array. Generally, when the array size is small, the main beam direction of the phased array is also affected by the number of array components. Table III Comparison of theoretical phase difference and designed phase difference at different scanning angles

[0087] For the multi-slot array of the present invention, the measured beam steering performance can be measured using the active element pattern (AEP) method. In this method, the radiation pattern of a fully excited phased array can be expressed as the superposition of the radiation patterns of each element using the following formula: Among them, V q is the complex-valued feed voltage, The field generated by the qth unit excitation current distribution component can be called the unit excitation AEP (active unit direction pattern).

[0088] Equation (4) shows that, given a set of measured AEPs (including all mutual coupling effects), the array pattern can be synthesized for any set of complex feed signals. The measured AEPs are obtained by exciting one component and treating the remaining components with load terminations. By exciting the array element components of the present invention one by one with the same amplitude and phase, nine measured AEPs can be obtained. The measured AEPs are then processed by introducing the designed stepped phase and uniform amplitude listed in Table III in a post-processing process, and the beamsteering performance of the phased array of the present invention can be obtained using Equation (4).

[0089] Figure 19The measurement and simulation results of the H-plane beam steering performance of the multi-slot phased array at 8.6 GHz are shown. This confirms the beam steering capability of the H-plane multi-slot phased array of the present invention at 8.6 GHz. Due to the symmetry of the array structure, only the positive scanning range is given here for the sake of simplicity. As can be seen from the figure, the main beam of both measurement and simulation can be controlled from 0° to +90°. The peak achieved gains of measurement and simulation are 12.1 dBi and 12.9 dBi, respectively. Within the scanning range (0° to ±90°), the gain fluctuations of measurement and simulation are 2.1 dB and 1.8 dB, respectively, and their maximum sidelobe levels (SLLs) are less than -5 dB and -11.4 dB, respectively. The difference between the simulation and measurement results is mainly due to manufacturing and experimental errors, such as cable manufacturing errors, differences in the position of chokes on the cable, and deformation of the PCB during the measurement process.

[0090] In addition, it can be observed from the figure that the beam width of the main beam gradually widens with the increase of the scanning angle. This phenomenon can be explained from two aspects: on the one hand, for a uniformly excited linear phased array, its array factor beam width will increase as the main beam deviates from the normal direction, especially when it is close to the end-fire direction. On the other hand, the multi-slot antenna unit assembly of the present invention has a very wide 3dB measurement beam width (217°), which completely covers the upper half of the space. According to the pattern product principle, the influence of the unit radiation pattern of this wide beam on the main beam of the array radiation pattern is almost negligible. Therefore, the beam width characteristics of the controlled array pattern should be similar to the beam width characteristics of the array factor.

[0091] Figure 20 A table comparing the array design of the present invention with existing wide-angle beamsteering linear phased arrays is presented. As can be seen from the table, the multi-slot array antenna of the present invention exhibits moderate bandwidth characteristics. However, due to its ability to steer the main beam in the upper half-space with a low profile, the array design of the present invention remains competitive. Furthermore, the multi-slot array antenna of the present invention is manufactured using low-cost PCB technology.

[0092] As described above, the present invention provides a beam-broadening technology utilizing hybrid magnetic flux. The provided device can be used to construct a wide-beam, low-profile multi-slot antenna unit assembly. Research has shown that the equivalent magnetic flux of a multi-slot unit can be considered a combination of MC1 and MC2. By varying the short arm length of the L-shaped slot, the H-plane radiation pattern can be flexibly adjusted. The proposed wide-beam multi-slot antenna exhibits a very wide beamwidth of 217° in the H-plane.

[0093] Wide-beam antenna element assemblies are used to design a linear multi-slot phased array with a wide control range, boasting an H-plane scanning range of ±90°. Within this scanning range, the measured gain fluctuation is only 2.1 dB, and the maximum sidelobe level (SLL) is less than -5 dB. Compared to existing phased array antennas, the design provided by this invention is highly competitive in terms of scanning range and physical size. Therefore, it is well-suited for in-vehicle communications.

[0094] According to the embodiments of the present disclosure, the functional units and modules of the devices and methods can be implemented using computing devices, computing processors or electronic circuits, including but not limited to application specific integrated circuits (ASICs), field programmable gate arrays (FPGAs), microcontrollers and other programmable logic devices configured or programmed according to the present disclosure. Skilled software or electronics practitioners can easily write computer instructions or software codes that run on computing devices, computing processors or programmable logic devices based on the contents of the present disclosure.

[0095] All or part of the methods according to various embodiments may be executed on one or more computing devices, including server computers, personal computers, laptops, smart phones, tablet computers and other mobile computing devices.

[0096] Various embodiments may include computer storage media, transient and non-transitory storage devices storing computer instructions or software code that can be used to program or configure a computing device, computing processor, or electronic circuit to perform any of the processes of the present invention. Storage media, transient and non-transitory storage devices include, but are not limited to, floppy disks, optical disks, Blu-ray disks, DVDs, CD-ROMs, magneto-optical disks, ROMs, RAMs, flash memory devices, or any other medium or device suitable for storing instructions, code, and / or data.

[0097] Each functional unit and module according to various embodiments may also be implemented in a distributed computing environment and / or a cloud computing environment, where all or part of the machine instructions are executed by one or more processing devices in a distributed manner over a communication network, which may be an intranet, a wide area network (WAN), a local area network (LAN), the Internet, or other forms of data transmission media.

[0098] The foregoing description of the present invention has been presented for purposes of illustration and explanation. It is not intended to be exhaustive or to limit the invention to the precise forms disclosed. Numerous modifications and variations will occur to those skilled in the art.

[0099] The selection and description of the present embodiment are intended to best explain the principles of the present invention and its practical applications, so that others in the art can understand the various embodiments of the present invention and various modifications suitable for specific purposes.

Claims

1. A multi-slot antenna, characterized in that: include: ground plate; a driving gap, provided on the ground plate and extending along a first direction; a first parasitic gap disposed on the ground plate and located on a first side of the driving gap, wherein the first parasitic gap extends along the first direction and a second direction different from the first direction; a second parasitic gap disposed on the ground plate and located on the first side of the driving gap, wherein the second parasitic gap extends along the first direction and the second direction; a third parasitic slot disposed on the ground plate and located on a second side of the driving slot, opposite to the first side, wherein a first end of the driving slot is located between the first parasitic slot and the third parasitic slot, the third parasitic slot extends at least along the first direction and the second direction, and the first parasitic slot and the third parasitic slot are symmetrical with respect to the driving slot; as well as A fourth parasitic gap is arranged on the ground plate and is located on the second side of the driving gap, wherein the second end of the driving gap is opposite to the first end and is located between the second parasitic gap and the fourth parasitic gap, the fourth parasitic gap extends at least along the first direction and the second direction, and the second parasitic gap and the fourth parasitic gap are symmetrical with respect to the driving gap.

2. The multi-slot antenna according to claim 1, wherein The first parasitic slot, the second parasitic slot, the third parasitic slot, and the fourth parasitic slot are each formed into an L shape by extending along the first direction and the second direction.

3. The multi-slot antenna according to claim 2, wherein: The first parasitic gap, the second parasitic gap, the third parasitic gap, and the fourth parasitic gap each include: a first extending portion extending along the first direction; and The second extending portion is connected to the first extending portion and extends along the second direction, wherein the length of the first extending portion is smaller than the length of the second extending portion.

4. The multi-slot antenna according to claim 3, wherein: A length of the first extension portion in the first direction ranges from 1 mm to 1.5 mm.

5. The multi-slot antenna according to claim 3, wherein: A length of the second extending portion in the second direction ranges from 5 mm to 15 mm, and a width of the second extending portion in the first direction ranges from 0.1 mm to 0.3 mm.

6. The multi-slot antenna according to claim 2, wherein: The first parasitic gap and the third parasitic gap present a symmetrical L-shaped structure with respect to the driving gap.

7. The multi-slot antenna according to claim 2, wherein: The first extending portions of the first parasitic slot and the second parasitic slot are located between the second extending portions of the first parasitic slot and the second parasitic slot.

8. The multi-slot antenna according to claim 2, wherein: The first L-shaped parasitic gap is separated from the driving gap by a gap, and the gap ranges from 0.5 mm to 1.5 mm.

9. The multi-slot antenna according to claim 2, wherein: The first extending portion is closer to the driving gap than an end portion of the second extending portion.

10. The multi-slot antenna according to claim 1, wherein: The length of the driving gap in the first direction ranges from 10 mm to 15 mm, and the width of the driving gap in the second direction ranges from 0.15 mm to 0.25 mm.

11. A multi-slot antenna array, characterized in that: include: A plurality of multi-slot antennas according to any one of claims 1 to 10, wherein the plurality of multi-slot antennas are continuously arranged in a horizontal direction.

12. The multi-slot antenna array according to claim 11, wherein: The number of the multi-slot antennas is nine.

13. The multi-slot antenna array according to claim 12, wherein: The multi-slot antenna is symmetrically configured.

14. The multi-slot antenna array according to claim 13, wherein: The leftmost and rightmost components of the nine multi-slot antennas are symmetrical to the central component of the nine multi-slot antennas.

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