Wireless power transmission system based on anti-detuning rectifier and control method thereof

Abstract

The invention provides a wireless electric energy transmission system based on an anti-detuning rectifier and a control method thereof, and relates to the technical field of wireless electric energy transmission. According to the system, the input end of a first full bridge is connected to a direct-current power supply, and the output end is connected to a transmitting coil through a primary side compensation circuit; the receiving coil is connected to the input end of the second full bridge through a secondary side compensation circuit; the output end of the second full bridge is connected to a load; one end of the load is connected to one end of the second capacitor through the first capacitor; the other end of the load is connected to the other end of the second capacitor; the middle point of the first capacitor and the second capacitor is connected to the input end of the second full bridge through the two switching tubes in sequence; wherein the source electrodes of the two switch tubes are connected; the first full bridge is composed of four switch tubes, and the second full bridge is composed of rectifier diodes. The system is simple in structure, the cost can be effectively reduced, and high efficiency and a high power factor of the rectifier can be realized.

Description

Technical Field

[0001] The present invention relates to the technical field of wireless power transmission, and in particular to a wireless power transmission system based on an anti-detuning rectifier and a control method thereof. Background Art

[0002] In the field of power electronics, compared to traditional wired power transmission methods, charging through wireless power transfer (WPT) technology has the advantages of being convenient, safe, and free of sparks and electric shock hazards. It is not restricted by location and environment and has greater flexibility. With the rise of the electric vehicle industry, wireless power transmission technology has become a gradually popular technology in the field of electric vehicle charging technology. In actual application scenarios, the magnetic coupling mechanism has coil misalignment and changes in the battery equivalent resistance during the charging process. The WPT system (i.e., wireless power transmission system, hereinafter referred to as the system) needs to dynamically adjust the output power to meet the load characteristics. At this stage, in order to enable the system to have a wide range of output power adjustment capabilities, transmitter control and receiver control are mainly used.

[0003] Transmitter-side control primarily involves adjusting the inverter, and includes methods such as phase-shift control, frequency conversion control, phase-shift + frequency conversion control, subharmonic control, and pulse density control. These methods suffer from the following issues: the switch may not achieve soft switching; the system's operating frequency may deviate significantly from the optimal frequency, resulting in reduced power transmission capacity; and the bidirectional wireless communication module may cause slow dynamic response and even system instability.

[0004] The receiving end control can be implemented using a load-side cascade converter. This method is simple and easy to implement, but at the cost of generating large power conversion losses. As an alternative, an active rectifier (AR) can eliminate the load-side cascade converter, thereby achieving the required output regulation capability. However, AR will generate additional reactance in the resonant network, that is, the impedance mismatch problem, which seriously reduces the power transmission capability and efficiency of the system. To solve the impedance mismatch problem, a switch-controlled capacitor or a variable inductor is generally introduced to adjust the equivalent impedance of the compensation network. However, this method lacks gain adjustment capability and increases the system size and control complexity. Although the input voltage and input current of the AR can be modulated in phase, as a purely resistive equivalent load, soft switching is inevitably lost. In other words, the impedance matching and soft switching of the AR cannot be achieved at the same time. Summary of the Invention

[0005] In order to solve the above-mentioned technical problems existing in the prior art, the purpose of the present invention is to provide a rectifier that can adjust the output power without affecting the tuning conditions, does not require a two-way wireless communication module, and can also realize soft switching of the switch tube, thereby minimizing the switching loss and achieving high power factor and high efficiency of the system.

[0006] Specifically, in a first aspect, the present invention provides a wireless power transmission system based on an anti-detuning rectifier, the technical solution of which includes:

[0007] The input end of the first full bridge is connected to a DC power supply, and the output end is connected to the primary transmitting coil via a primary compensation circuit;

[0008] The secondary side receiving coil is connected to the input end of the second full bridge via the secondary side compensation circuit; the output end of the second full bridge is connected to the load;

[0009] One end of the load is connected to one end of the second capacitor via the first capacitor; the other end of the load is connected to the other end of the second capacitor;

[0010] The middle point between the first capacitor and the second capacitor is connected to the first input terminal of the second full bridge via the switch tube S5 and the switch tube S6 in sequence;

[0011] in,

[0012] The source of the switch tube S5 is connected to the source of the switch tube S6;

[0013] The first full bridge is composed of switch tubes S1 to S4, and the second full bridge is composed of rectifier diodes D1 to D4.

[0014] Optionally, both the primary-side compensation circuit and the secondary-side compensation circuit are SS compensation circuits, bilateral LCC compensation circuits, LCC-S compensation circuits, or S-LCC compensation circuits.

[0015] Preferably, both the primary side compensation circuit and the secondary side compensation circuit are SS compensation circuits.

[0016]

[0017] Where ω is the operating angular frequency of the wireless power transmission system, L p and L s are the self-inductance values ​​of the primary transmitting coil and the secondary receiving coil, respectively. p and C s They are the primary side compensation capacitor and the secondary side compensation capacitor respectively.

[0018] Preferably, both the primary side compensation circuit and the secondary side compensation circuit are bilateral LCC compensation circuits.

[0019]

[0020] Where, L a is the primary series compensation inductor, C a is the primary side parallel compensation capacitor, L b is the secondary side series compensation inductor, C bis the secondary side parallel compensation capacitor.

[0021] Preferably, both the primary side compensation circuit and the secondary side compensation circuit are LCC-S compensation circuits.

[0022]

[0023] Preferably, both the primary side compensation circuit and the secondary side compensation circuit are S-LCC compensation circuits.

[0024]

[0025] In a second aspect, the present invention further provides a control method for the anti-detuning rectifier of the above-mentioned wireless power transmission system, comprising:

[0026] The duty cycle of the control signal G5 of the switch tube S5 and the control signal G6 of the switch tube S6 are both set to D, and the period is both set to T.

[0027]

[0028] G6(t)=G5(tT / 2);

[0029] 0.5≤D≤1;

[0030] Where G5(t) is the value of the control signal G5 at time t, and G6(t) is the value of the control signal G6 at time t.

[0031] Compared to existing technologies, the technical solution provided by this invention, by introducing switches S5 and S6 connected in reverse series, can adjust the system's output power in response to magnetic coupling mechanism offsets and load changes. This eliminates the need for complex auxiliary circuits, complex magnetic coupling mechanisms, and bidirectional communication, effectively reducing system complexity and cost. Furthermore, the output voltage, output current, and output power can be adjusted by adjusting the duty cycle of switches S5 and S6. This simple implementation method achieves high efficiency and a high rectifier power factor. BRIEF DESCRIPTION OF THE DRAWINGS

[0032] Figure 1 This is a schematic diagram of the anti-detuning rectifier in the present invention.

[0033] Figure 2 Schematic diagram of key waveforms of the anti-detuning rectifier in the present invention.

[0034] Figure 3 Schematic diagram of analysis of mode 1 of the anti-detuning rectifier in the present invention.

[0035] Figure 4 Schematic diagram of analysis of mode 2 of the anti-detuning rectifier in the present invention.

[0036] Figure 5 Schematic diagram of analysis of mode 3 of the anti-detuning rectifier in the present invention.

[0037] Figure 6 Schematic diagram of analysis of mode 4 of the anti-detuning rectifier in the present invention.

[0038] Figure 7 This is a schematic diagram of the anti-detuning rectifier in the present invention applied to the current-type output SS compensation topology.

[0039] Figure 8 This is a schematic diagram of the anti-detuning rectifier in the present invention applied to the current-type output LCC-LCC compensation topology.

[0040] Figure 9 Schematic diagram of the system output current changing with duty cycle when the anti-detuning rectifier in the present invention is applied to the current-type output SS compensation topology.

[0041] Figure 10 This is a schematic diagram of the anti-detuning rectifier in the present invention applied to a voltage-type output LCC-S compensation circuit.

[0042] Figure 11 This is a schematic diagram of the anti-detuning rectifier in the present invention applied to the voltage-type output S-LCC compensation circuit.

[0043] Figure 12 Schematic diagram of the system output voltage changing with the duty cycle when the anti-detuning rectifier in the present invention is applied to the voltage-type output LCC-S compensation circuit.

[0044] Figure 13 The figures are circuit diagrams of two rectifiers; among them, (a) is the circuit diagram of the anti-detuning rectifier provided by the present invention, and (b) is the circuit diagram of a traditional active rectifier.

[0045] Figure 14 Schematic diagram of key waveforms of the anti-detuning rectifier in the present invention and the traditional half-bridge active rectifier.

[0046] Figure 15 Schematic diagram comparing the additional reactance introduced by the anti-detuning rectifier of the present invention and the traditional half-bridge active rectifier with the change of duty cycle.

[0047] Figure 16These are simulation waveforms of the anti-detuning rectifier and half-bridge active rectifier in the present invention using LCC-S and SS compensation topologies; among them, (a) is a simulation waveform of the half-bridge active rectifier regulation in the LCC-S compensated WPT system with an output of 300V / 1.8kW, (b) is a simulation waveform of the anti-detuning rectifier regulation in the LCC-S compensated WPT system with an output of 300V / 1.8kW, (c) is a simulation waveform of the half-bridge active rectifier regulation in the SS compensated WPT system with an output of 200V / 1.6kW, and (d) is a simulation waveform of the anti-detuning rectifier regulation in the SS compensated WPT system with an output of 200V / 1.6kW. DETAILED DESCRIPTION

[0048] Hereinafter, the technical solution provided by the present invention will be further elaborated in combination with embodiments and drawings.

[0049] Example 1

[0050] like Figure 1 As shown, an embodiment of the present invention provides a wireless power transmission system based on an anti-detuning rectifier, including a power transmitter and a power receiver. The power transmitter includes a DC power supply, a full-bridge inverter (first full-bridge), a primary compensation circuit, a primary transmitting coil L p The full-bridge inverter consists of four switches S1 to S4 with anti-parallel diodes, generating a square wave voltage v ab The voltage of the DC power supply is V in , the output voltage of the full-bridge inverter is the square wave voltage v ab , the output current is i ab ; The primary transmitting coil current is i p .

[0051] The power receiving end includes a secondary receiving coil L s , secondary side compensation circuit, diode full bridge rectifier (second full bridge) D R1 ~D R4 , output capacitor bridge arm C o1 ~C o2 , a pair of reverse series switches S5 and S6 and a load R o ; The secondary receiving coil current is i s ;Diode full bridge rectifier D R1 ~D R4 The input voltage is v cd , the input current is i cd ; The reverse series switch tube S5 and the switch tube S6 are connected to the diode branch D R3 ~D R4 (Since the full bridge is a symmetrical structure, it can also be a diode branch D R1 ~D R2, that is, any input terminal of the second full bridge) and the output capacitor bridge arm C o1 ~C o2 The midpoint of the diode full bridge rectifier forms a mixed mode rectifier (i.e., an anti-detuning rectifier); the output current of the diode full bridge rectifier is i rec ;The output voltage at the load end is V o , the output current is I o Optionally, the switch tube S5 and the switch tube S6 are MOSFET switch tubes.

[0052] Primary transmitting coil L p and the secondary receiving coil L s There is coupling between them, and the mutual inductance is M. There is no specific requirement for the compensation topology type, and the hybrid mode rectifier provided in this embodiment is applicable regardless of the compensation topology used.

[0053] In order to achieve the output power regulation of the system, the duty cycle D of the reverse series switch tube S5 and the switch tube S6 can be adjusted accordingly. The key control waveform is as follows: Figure 2 The formula is as follows:

[0054]

[0055] G6(t)=G5(tT / 2);

[0056] 0.5≤D≤1;

[0057] Where G5(t) is the value of the control signal G5 at time t, and G6(t) is the value of the control signal G6 at time t.

[0058] The working model of the hybrid mode rectifier is analyzed as follows. The equivalent circuits of each mode are as follows: Figures 3 to 6 As shown, v cd 、i cd They are diode full bridge rectifier D R1 ~D R4 The input voltage and input current.

[0059] The gate signals (control signals) of the switch tubes S5 and S6 are denoted as G5 and G6 respectively. The duty cycle of the switch tubes S5 and S6 is the same, denoted as D, and the range of D is [0.5, 1]. The control signals G5 and G6 are respectively connected to the diode full-bridge rectifier D at the moment of conduction. R1 ~D R4 The input current i cd When both the switch tube S5 and the switch tube S6 are in the on state, the diode full bridge rectifier D R1 ~D R4 The input voltage v cd The amplitude is V o / 2, otherwise, vcd The amplitude is V o .

[0060] Mode 1[t0≤t<t1] ： The equivalent circuit is Figure 3 As shown, in the time period t0≤t<t1, the diode full-bridge rectifier D R1 ~D R4 The input current i cd is negative, the reverse series connected switch tubes S5 and S6 are in the on state, and the diode D R1 , diode D R3 , diode D R4 In the cut-off state, the diode D R2 In the on state. At this time, it is the voltage doubler rectification mode, and the input current i cd The current flows through the drain-source of the switch tube S6 and the body diode of the switch tube S5. At this time, the drain-source voltage of the switch tube S5 is zero, realizing zero-voltage turn-on.

[0061] Mode 2[t1≤t <t2] ： The equivalent circuit is Figure 4 As shown, during the time period t1≤t<t2, the diode full-bridge rectifier D R1 ~D R4 The input current i cd Still negative, the switch tube S5 is in the on state, the switch tube S6 is in the off state, and the diode D R1 , diode D R4 In the cut-off state, the diode D R2 , diode D R3 In the on state. At this time, it is in full-bridge rectification mode, and the input current i cd Flowing through the rectifier diode D R2 、D R3 , no current flows through the branches of the reverse series connected switch tubes S5 and S6. At this time, the drain-source voltage of the switch tube S6 is V o / 2, the drain-source voltage of the switch tube S5 is 0.

[0062] Mode 3 [t2≤t<t3] ： The equivalent circuit is Figure 5 As shown, during the time period t2≤t<t3, the diode full-bridge rectifier D R1 ~D R4 The input current i cd The forward and reverse series connected switch tubes S5 and S6 are in the on state, and the diode D R2 , diode D R3 , diode D R4 In the cut-off state, the diode D R1In the on state. At this time, it is the voltage doubler rectification mode, and the input current i cd The current flows through the drain-source of the switch tube S5 and the body diode of the switch tube S6. At this time, the drain-source voltage of the switch tube S6 is zero, realizing zero-voltage turn-on.

[0063] Mode 4[t3≤t<t4[ ： The equivalent circuit is Figure 6 As shown, during the time period t3≤t<t4, the diode full-bridge rectifier D R1 ~D R4 The input current i cd Still positive, the switch tube S6 is in the on state, the switch tube S5 is in the off state, and the diode D R2 , diode D R3 In the cut-off state, the diode D R1 , diode D R4 In the on state. At this time, it is in full-bridge rectification mode, and the input current i cd Flowing through the rectifier diode D R , diode D R4 , no current flows through the branches of the reverse series connected switch tubes S5 and S6. At this time, the drain-source voltage of the switch tube S5 is V o / 2, the drain-source voltage of the switch tube S6 is 0.

[0064] Considering that in wireless power transmission systems, compensation circuits often need to be designed with two different output modes, namely current-type and voltage-type outputs. These configurations enhance the controllability of the output, thereby improving system performance. For example, the series-series (SS) compensation circuit and the bilateral LCC compensation circuit are classified as current-type output compensation circuits, while the LCC-S circuit and the S-LCC circuit are classified as voltage-type output compensation circuits. This embodiment has no specific requirements for the type of compensation circuit. Regardless of the compensation circuit used, the hybrid-mode rectifier proposed in this embodiment is applicable. Therefore, this embodiment will use the representative series-series (SS) compensation circuit and LCC-S compensation circuit as examples for case analysis. Before analysis, the following assumptions need to be made: the resonant tank is lossless, the switching devices are ideal, and the additional reactance caused by the hybrid rectifier can be ignored.

[0065] Current type output series-series (SS) compensation circuit such as Figure 7 In order to make the system work in a completely resonant state, the parameter design of the compensation network must meet the following requirements:

[0066]

[0067] Where ω is the operating angular frequency of the wireless power transmission system, L p and L sare the self-inductance values ​​of the primary transmitting coil and the secondary receiving coil, respectively. p and C s They are the primary side series compensation capacitor and the secondary side parallel compensation capacitor respectively;

[0068] Current-mode output bilateral inductor-capacitor-capacitor (bilateral LCC) compensation circuit Figure 8 In order to make the system work in a completely resonant state, the parameter design of the compensation network must meet the following requirements:

[0069]

[0070] Where, L a is the primary series compensation inductor, C a is the primary side parallel compensation capacitor, L b is the secondary side series compensation inductor, C b is the secondary side parallel compensation capacitor.

[0071] When the transmitter is in full resonance, the effective value of the secondary receiving coil current I cd The DC power supply voltage V in , the mutual inductance M is related to the system's operating angular frequency ω, and the formula is as follows:

[0072]

[0073] according to Figure 2 The output current of the full-bridge rectifier with diodes is i rec The waveform of the output current after output capacitor filtering is I o for:

[0074]

[0075] Rewrite the output current I o , the formula is as follows:

[0076]

[0077] Output current I of the current-mode output series-series (SS) compensation topology at different duty cycles o like Figure 9 As shown in Figure 2 , the hybrid-mode rectifier proposed in this embodiment can achieve a nearly doubled current regulation range. Circuit simulation of the WPT system verifies the analytical model expressed in the above equation. It can be seen that the output current decreases monotonically with increasing duty cycle D. Notably, the above equation remains valid regardless of whether the hybrid-mode rectifier introduces additional reactance at the receiving end.

[0078] Voltage type output LCC-S compensation circuit as follows Figure 10In order to make the system work in a completely resonant state, the parameter design of the compensation network must meet the following requirements:

[0079]

[0080] Voltage type output S-LCC compensation circuit as follows Figure 11 In order to make the system work in a completely resonant state, the parameter design of the compensation network must meet the following requirements:

[0081]

[0082] Calculate the induced voltage V of the receiving coil ind , the formula is as follows:

[0083]

[0084] from Figure 2 It can be seen that the input voltage It is a segmented periodic waveform. The input voltage v is decomposed by Fourier. cd Get its fundamental component v cd1 (t), the formula is as follows:

[0085]

[0086] Ignoring higher harmonics, the fundamental component v cd,1 The effective value of (t) is approximately equal to the induced voltage V of the receiving coil ind , then the average output voltage V o for:

[0087]

[0088] Output voltage V of voltage-type output LCC-S compensation topology at different duty cycles o like Figure 12 Obviously, the hybrid mode rectifier proposed in this embodiment can achieve almost double the voltage regulation range, and the output voltage increases monotonically with the increase of the duty cycle D. The simulation results are consistent with the above formula of the analytical model.

[0089] To regulate the output power at the receiving end, active rectifiers (ARs) are widely used in wireless power transmission systems, such as full-bridge and half-bridge active rectifiers, respectively. Figure 13 As shown in (a) and (b) in the figure. Figure 13 The half-bridge active rectifier shown in (b) is taken as an example to compare with the hybrid mode rectifier (HMR) proposed in this embodiment. Its working waveform is as follows Figure 14 As shown, G3 and G4 are the driving signals of switch tube S3 and switch tube S4 respectively, and the duty cycle is 0.5.cd,1 v cd The fundamental component of the input current is defined as dπ. The control angle of AR is defined as dπ, where d∈[0,1]. cd The negative zero crossing point of the switch is turned on after a delay of (1-d)π, and the switch tube S4 is turned on when the input current i cd It can be observed that the voltage v cd,1 Hysteresis current input current i cd , is the phase difference θ=(1-d)π / 2. Before the switch tube S4 (switch tube S3) is turned on, the input current i cd The positive (negative) current flows through its body diode. Therefore, both switch tubes S3 and S4 can achieve ZVS (Zero Voltage Switch).

[0090] Since the input current i cd and voltage v cd,1 The phase difference θ between the two, the half-bridge active rectifier and the load resistor together appear as a capacitive load, and the reactance X caused by the half-bridge active rectifier and the resistive load is e_AR It can be calculated using the following formula:

[0091]

[0092] Similarly, the reactance X caused by the hybrid mode rectifier in this embodiment is e_HMR The output current i can be calculated by rec and input current i cd The ratio is obtained as follows:

[0093]

[0094] ψ=(2D-1)π;

[0095] Where ψ is the intermediate calculation amount.

[0096] To investigate the reactance X e_AR , Reactance X e_HMR The change trend of reactance relative to d(D) is derived as follows:

[0097]

[0098]

[0099] By solving the above two equations, and And substituting the relevant parameters, we can get:

[0100]

[0101] Where, d extrema For X e_AR The duty cycle corresponding to the extreme value, D extrema The duty cycle corresponding to the extreme value of Xe_HMR.

[0102] It can be deduced that X e_AR and X e_HMR They all first decrease with the duty cycle, and then increase with the increase of the duty cycle. e_AR and X e_HMR Respectively in d extrema and d extrema Reached minimum value.

[0103] Assume that the load resistance R o If the reactance is 20Ω, the additional reactance caused by AR and HMR is Figure 15 Obviously, in the duty cycle range [0.5, 1], X e_HMR The magnitude is much smaller than X e_AR , can be ignored to a certain extent. In other words, the tuning condition of the WPT system is almost unaffected by HMR regulation. However, AR regulation may seriously detune the WPT system, especially when the duty cycle reaches extremes.

[0104] Figure 16 The simulation waveforms of the voltage-type and current-type output WPT systems using the half-bridge active rectifier (AR) and the hybrid mode rectifier (HMR) proposed in the present invention are shown. In (a) and (b), the LCC-S compensated WPT system with an output of 300V / 1.8kW was tested. It can be seen that under the same output power, the adjustment of the AR will lead to capacitive tuning of the system, while the adjustment of the HMR can maintain the unity power factor of the system. Due to the detuned working state, the inverter current i in (a) is ab and the rectified current i cd The RMS value of (b) is almost twice that of (b), resulting in high conduction loss.

[0105] exist Figure 16 In Figures (c) and (d), AR and HMR are used, respectively, to compensate for a 200V / 1.6kW SS-compensated WPT system. It can be seen that AR regulation results in inductive tuning of the system, while HMR regulation still maintains unity power factor. Similarly, detuned operation results in higher reactive current, which impairs overall system efficiency.

[0106] It can be seen that in this embodiment, the diode branch D R3 ~D R4 and output capacitor bridge arm C o1 ~C o2A pair of reverse-series switches, S5 and S6, are introduced at the midpoint of the WPT circuit. Controlling the on / off behavior of these switches switches S5 and S6 switches between two rectification modes: full-bridge rectification and voltage-doubler rectification. By controlling the duty cycle of switches S5 and S6, the WPT system's output power is adjusted without affecting tuning conditions and eliminating the need for bidirectional communication. This also achieves zero-voltage switching (ZVS) switching, minimizing switching losses and ensuring a high power factor and high efficiency.

[0107] As can be seen from the above embodiments and accompanying drawings, compared to the prior art, the technical solution provided by the present invention, by introducing switches S5 and S6 connected in reverse series, can adjust the system's output power in response to magnetic coupling mechanism offsets and load changes. This eliminates the need for complex auxiliary circuits, complex magnetic coupling mechanisms, and bidirectional communication, effectively reducing system complexity and cost. Furthermore, the output voltage, output current, and output power can be adjusted by adjusting the duty cycle of switches S5 and S6. This simple and easy implementation method achieves high efficiency and a high rectifier power factor.

Claims

1. A wireless power transmission system based on an anti-detuning rectifier, characterized in that: include: The input end of the first full bridge is connected to a DC power supply, and the output end is connected to the primary transmitting coil via a primary compensation circuit; The secondary side receiving coil is connected to the input end of the second full bridge via the secondary side compensation circuit; the output end of the second full bridge is connected to the load; One end of the load is connected to one end of the second capacitor via the first capacitor; the other end of the load is connected to the other end of the second capacitor; The middle point between the first capacitor and the second capacitor is connected to the input end of the second full bridge via the switch tube S5 and the switch tube S6 in sequence; in, The source of the switch tube S5 is connected to the source of the switch tube S6; The first full bridge is composed of switch tubes S1 to S4, and the second full bridge is composed of rectifier diodes D1 to D4.

2. A wireless power transmission system based on an anti-detuning rectifier according to claim 1, characterized in that: The primary side compensation circuit and the secondary side compensation circuit are both SS compensation circuits, bilateral LCC compensation circuits, LCC-S compensation circuits or S-LCC compensation circuits.

3. The wireless power transmission system based on the anti-detuning rectifier according to claim 1, characterized in that: Both the primary side compensation circuit and the secondary side compensation circuit are SS compensation circuits. Where ω is the operating angular frequency of the wireless power transmission system, L p and L s are the self-inductance values ​​of the primary transmitting coil and the secondary receiving coil, respectively. p and C s They are the primary side compensation capacitor and the secondary side compensation capacitor respectively.

4. The wireless power transmission system based on the anti-detuning rectifier according to claim 1, characterized in that: Both the primary side compensation circuit and the secondary side compensation circuit are bilateral LCC compensation circuits. Where ω is the operating angular frequency of the wireless power transmission system, L a is the primary series compensation inductor, C a is the primary side parallel compensation capacitor, L p and L s are the self-inductance values ​​of the primary transmitting coil and the secondary receiving coil, respectively. p is the primary compensation capacitor, L b is the secondary side series compensation inductor, C b is the secondary side parallel compensation capacitor.

5. The wireless power transmission system based on the anti-detuning rectifier according to claim 1, characterized in that: Both the primary side compensation circuit and the secondary side compensation circuit are LCC-S compensation circuits. Where ω is the operating angular frequency of the wireless power transmission system, L a is the primary series compensation inductor, C a is the primary side parallel compensation capacitor, L p and L s are the self-inductance values ​​of the primary transmitting coil and the secondary receiving coil, respectively. p and C s They are the primary side series compensation capacitor and the secondary side series compensation capacitor respectively.

6. The wireless power transmission system based on the anti-detuning rectifier according to claim 1, characterized in that: Both the primary side compensation circuit and the secondary side compensation circuit are S-LCC compensation circuits. Where ω is the operating angular frequency of the wireless power transmission system, L p and L s are the self-inductance values ​​of the primary transmitting coil and the secondary receiving coil, respectively. p is the primary compensation capacitor, L b is the secondary side series compensation inductor, C b is the secondary side parallel compensation capacitor, C s is the secondary side series compensation capacitor.

7. A control method for a wireless power transmission system based on an anti-detuning rectifier according to any one of claims 1 to 6, characterized in that: include: The duty cycle of the control signal G5 of the switch tube S5 and the control signal G6 of the switch tube S6 are both set to D, and the period is both set to T. G6(t)=G5(tT / 2); 0.5≤D≤1； Where G5(t) is the value of the control signal G5 at time t, and G6(t) is the value of the control signal G6 at time t.

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