Permanent magnet synchronous motor sensorless control method based on sliding mode observer

Abstract

The invention discloses a permanent magnet synchronous motor sensorless control method based on a sliding-mode observer, and the method comprises the steps: collecting the three-phase current and three-phase voltage of a permanent magnet synchronous motor, and converting the three-phase current and three-phase voltage into a two-phase current component and a two-phase voltage component under a static coordinate system; inputting two-phase current and voltage components into a novel sliding mode observer; according to the novel sliding-mode observer, a sinusoidal saturation switching function is adopted to replace a sign function, and a variable gain function based on cooperative control of a speed error and a state variable is adopted; taking the product of the sine type saturation switching function and the variable gain function as an equivalent control quantity to obtain a back electromotive force estimated value; inputting the back electromotive force estimated value into a phase-locked loop structure, and extracting a rotor estimated electrical angular velocity and a rotor estimated position angle through the phase-locked loop structure; and realizing sensorless vector control of the permanent magnet synchronous motor by using the estimated electrical angular velocity and the estimated position angle of the rotor. According to the method, the problem of phase lag is solved, and effective suppression of rotating speed buffeting is realized.

Description

Technical Field

[0001] This invention relates to the field of sensorless control technology for permanent magnet synchronous motors, and more specifically to a sensorless control method for permanent magnet synchronous motors based on a sliding mode observer. Background Technology

[0002] In recent years, permanent magnet synchronous motors (PMSMs) have been widely used in the robotics field due to their high power density, excellent dynamic response, and high efficiency. Especially in industrial robots, service robots, and automation systems, the performance advantages of PMSMs make them ideal drive sources. However, traditional control methods often rely on mechanical position sensors, such as photoelectric encoders and resolvers, which increases the system's size, weight, and cost. Furthermore, these sensors are susceptible to external environmental factors such as temperature, humidity, and vibration, reducing system reliability. Therefore, in recent years, research on sensorless control technology has gradually become a hot topic in the robotics field.

[0003] In sensorless control, common improvement strategies include extended sliding mode observers and higher-order sliding mode observers. Extended sliding mode observers can directly observe the back electromotive force (EMF) of the motor, thus avoiding system chattering problems caused by extracting the back EMF from the switching function. For example, in patent application CN118554813A, entitled "Anti-disturbance control method for permanent magnet direct drive motor based on extended sliding mode disturbance observer," an extended sliding mode disturbance observer is used to estimate the system disturbance term that cannot be accurately measured in real time, and the observed load torque value is fed forward to compensate the current regulator, effectively improving the dynamic response speed of the system. Higher-order sliding mode observers can achieve accurate estimation of motor rotor position and speed. For example, in patent application CN119341419A, entitled "Adaptive higher-order terminal sliding mode observer control method for permanent magnet synchronous motor," based on adaptive gain, extended equations, and a higher-order terminal sliding surface, the current signal control law and back EMF signal control law of the higher-order terminal sliding surface are designed, enabling the control system to maintain robustness and high-precision output simultaneously under high dynamic response requirements.

[0004] However, the aforementioned improvements make the observer structure extremely complex, significantly increasing the design workload and posing certain challenges in practical applications. Therefore, optimizing existing sliding mode observers to reduce chattering and improve estimation accuracy in order to meet the high control performance requirements of robotic applications is a problem that urgently needs to be solved by those skilled in the art. Summary of the Invention

[0005] In view of the above problems, the present invention provides a sensorless control method for permanent magnet synchronous motors based on sliding mode observers, so as to at least solve some of the technical problems mentioned in the background art.

[0006] To achieve the above objectives, the present invention adopts the following technical solution:

[0007] This invention provides a sensorless control method for permanent magnet synchronous motors based on sliding mode observers, applicable to a sensorless control system for permanent magnet synchronous motors based on sliding mode observers; the method includes:

[0008] S1. Collect the three-phase current and three-phase voltage of the permanent magnet synchronous motor;

[0009] S2. Through coordinate transformation, the three-phase current and three-phase voltage are converted into two-phase current components and two-phase voltage components in a stationary coordinate system;

[0010] S3. Input the two-phase current components and two-phase voltage components into the novel sliding mode observer; wherein, the novel sliding mode observer uses a sinusoidal saturation switching function to replace the sign function, and uses a variable gain function based on the coordinated control of speed error and state variables;

[0011] S4. The product of the sinusoidal saturation switching function and the variable gain function is used as the equivalent control quantity of the novel sliding mode observer, and the back electromotive force is estimated based on the equivalent control quantity.

[0012] S5. Input the back electromotive force estimate into a phase-locked loop structure, and extract the rotor estimated electric angular velocity and rotor estimated position angle through the phase-locked loop structure.

[0013] S6. Sensorless vector control of the permanent magnet synchronous motor is achieved by using the rotor to estimate the electric angular velocity and the rotor to estimate the position angle.

[0014] Furthermore, the novel sliding mode observer is represented as follows:

[0015]

[0016]

[0017]

[0018] Among them, i α Indicates the α-axis stator current component in the stationary coordinate system; i β This represents the β-axis stator current component in the stationary coordinate system. This represents the observed values ​​of the α-axis stator current component in the stationary coordinate system; These are the observed values ​​of the β-axis stator current components in the stationary coordinate system; p represents the differential operator; R s L represents the stator resistance. s Indicates stator inductance; u α Represents the α-axis stator voltage component in the stationary coordinate system; uβ The β-axis stator voltage component is represented in the stationary coordinate system; H represents the variable gain function based on the coordinated control of velocity error and state variables; f(s) α f(s) represents a sinusoidal saturation switching function acting on the α-axis; β ) represents a sinusoidal saturation switching function acting on the β axis.

[0019] Furthermore, the sinusoidal saturation switching function f(s) is expressed as:

[0020]

[0021] Where δ represents the frequency coefficient, and σ represents the boundary layer width; s represents the sliding surface.

[0022] Furthermore, the variable gain function H based on the coordinated control of velocity error and state variables is expressed as:

[0023]

[0024] Where, k b Represents the base gain, and k b >0; e x Indicates speed error, and μ represents the proportionality coefficient, and 0 < μ < 1; v represents the attenuation factor, and v > 0; s represents the sliding surface.

[0025] Furthermore, the base gain k b Conditions to be met:

[0026]

[0027] Among them, e α Represents the back electromotive force along the α axis; e β f(s) represents the back electromotive force along the β axis; α f(s) represents a sinusoidal saturation switching function acting on the α-axis; β ) represents a sinusoidal saturation switching function acting on the β axis.

[0028] Furthermore, the estimated back electromotive force is expressed as:

[0029]

[0030]

[0031] in, This represents the estimated value of the back electromotive force along the α axis; The value of the back electromotive force on the β-axis is represented by H; H represents the variable gain function based on the coordinated control of velocity error and state variables; f(s) αf(s) represents a sinusoidal saturation switching function acting on the α-axis; β ) represents a sinusoidal saturation switching function acting on the β-axis; k b Indicates the base gain; e x Indicates speed error; μ represents proportionality coefficient; v represents attenuation factor; s represents sliding surface; s α This represents the sliding surface acting on the α-axis; s β This represents the sliding surface acting on the β axis.

[0032] Further, in S5, the extraction of the rotor's estimated electrical angular velocity and estimated rotor position angle through the phase-locked loop structure specifically includes:

[0033] When the actual rotor position angle θ is different from the estimated rotor position angle Error satisfies At that time, the actual rotor position angle θ and the estimated rotor position angle The relationship is ;

[0034] Based on the actual rotor position angle θ and the estimated rotor position angle The relationship is used to calculate the back electromotive force difference Δe:

[0035]

[0036] Where k represents the magnitude of the estimated back electromotive force; This represents the estimated value of the back electromotive force along the α axis; The value represents the estimated back electromotive force of the β-axis; Δθ represents the difference between the actual rotor position angle θ and the estimated rotor position angle. angular error;

[0037] The back electromotive force difference Δe is passed through a gain of 1 / k. e The obtained angular error Δθ is then fed into the PI controller to obtain the estimated electric angular velocity of the rotor. ;

[0038] Estimate the electrical angular velocity of the rotor. Integrating the results yields the final estimated rotor position angle. .

[0039] Furthermore, the sensorless control system for the permanent magnet synchronous motor based on the sliding mode observer adopts a speed-current dual closed-loop structure.

[0040] Furthermore, the velocity-current dual closed-loop structure includes an outer velocity loop and an inner current loop;

[0041] The outer velocity ring is based on a preset target rotational speed. Estimating the electric angular velocity of the rotor The difference in mechanical speed, converted from 30 / πp, is used by a PI controller to generate two q-axis current command values ​​in the rotating coordinate system. ;

[0042] The inner current loop receives the q-axis current command value. And receive two d-axis current command values ​​in the rotating coordinate system. = 0 command, and respectively with q-axis feedback current i q and d-axis feedback current i d The comparison results are then processed by a PI controller to generate q-axis voltage commands u. q and d-axis voltage command u d ;

[0043] Based on the q-axis voltage command u q and d-axis voltage command u d It drives the permanent magnet synchronous motor by driving the inverter.

[0044] As can be seen from the above technical solution, compared with the prior art, the present invention discloses a sensorless control method for permanent magnet synchronous motors based on a sliding mode observer, which has the following beneficial effects:

[0045] This invention addresses the limitations of traditional sliding mode observers in suppressing rotational chattering by proposing a novel sliding mode observer structure. By replacing the sign function with a sinusoidal saturation function and eliminating the need for a low-pass filter, the phase lag problem is fundamentally solved.

[0046] This invention designs a dynamic gain function based on speed error and state variables, thereby effectively suppressing speed chattering and reducing errors.

[0047] The technical solution of the present invention will be further described in detail below with reference to the accompanying drawings and embodiments. Attached Figure Description

[0048] To more clearly illustrate the technical solutions in the embodiments of the present invention or the prior art, the drawings used in the description of the embodiments or the prior art will be briefly introduced below. Obviously, the drawings described below are only embodiments of the present invention. For those skilled in the art, other drawings can be obtained based on the provided drawings without creative effort.

[0049] Figure 1 A schematic flowchart of a sensorless control method for a permanent magnet synchronous motor based on a sliding mode observer, provided in an embodiment of the present invention.

[0050] Figure 2 This is a schematic diagram illustrating the principle of the rotational speed and position calculation process based on a phase-locked loop provided in an embodiment of the present invention.

[0051] Figure 3 A schematic diagram of the sensorless control system framework for a permanent magnet synchronous motor based on a sliding mode observer, provided in an embodiment of the present invention. Detailed Implementation

[0052] The technical solutions of the embodiments of the present invention will be clearly and completely described below with reference to the accompanying drawings. Obviously, the described embodiments are only some embodiments of the present invention, and not all embodiments. Based on the embodiments of the present invention, all other embodiments obtained by those skilled in the art without creative effort are within the scope of protection of the present invention.

[0053] This invention discloses a sensorless control method for permanent magnet synchronous motors based on a sliding mode observer, applicable to a sensorless control system for permanent magnet synchronous motors based on a sliding mode observer; wherein "sensorless" specifically refers to the absence of position sensors; such as Figure 1 As shown, the method specifically includes the following steps:

[0054] S1. Collect the three-phase current and three-phase voltage of the permanent magnet synchronous motor;

[0055] S2. Through coordinate transformation, the three-phase current and three-phase voltage are converted into two-phase current components and two-phase voltage components in a stationary coordinate system;

[0056] S3. Input the two-phase current components and the two-phase voltage components into the new sliding mode observer; wherein, the new sliding mode observer uses a sinusoidal saturation switching function to replace the sign function, and uses a variable gain function based on the coordinated control of speed error and state variables;

[0057] S4. The product of the sinusoidal saturation switching function and the variable gain function is used as the equivalent control quantity of the new sliding mode observer, and the back electromotive force is estimated based on the equivalent control quantity.

[0058] S5. Input the back EMF estimate into a phase-locked loop structure, and extract the rotor's estimated electric angular velocity and rotor's estimated position angle through the phase-locked loop structure.

[0059] S6. Sensorless vector control of permanent magnet synchronous motor is achieved by using rotor to estimate electric angular velocity and rotor to estimate position angle.

[0060] Next, each of the above steps will be explained in detail.

[0061] In step S1, the three-phase current (i) of the permanent magnet synchronous motor is collected. a i b i c ) and three-phase voltage (u a ub u c );

[0062] In step S2, the three-phase current and three-phase voltage are converted into two-phase current components and two-phase voltage components in a stationary coordinate system through coordinate transformation; expressed as:

[0063]

[0064]

[0065] In step S3, the two-phase current components and the two-phase voltage components are input to the novel sliding mode observer; wherein, the novel sliding mode observer uses a sinusoidal saturation switching function to replace the sign function, and employs a variable gain function based on the coordinated control of velocity error and state variables; specifically:

[0066] (1) The novel sliding mode observer is represented as:

[0067]

[0068]

[0069]

[0070] Among them, i α Indicates the α-axis stator current component in the stationary coordinate system; i β This represents the β-axis stator current component in the stationary coordinate system. This represents the observed values ​​of the α-axis stator current component in the stationary coordinate system; These are the observed values ​​of the β-axis stator current components in the stationary coordinate system; p represents the differential operator; R s L represents the stator resistance. s Indicates stator inductance; u α Represents the α-axis stator voltage component in the stationary coordinate system; u β The β-axis stator voltage component is represented in the stationary coordinate system; H represents the variable gain function based on the coordinated control of velocity error and state variables; f(s) α f(s) represents a sinusoidal saturation switching function acting on the α-axis; β ) represents a sinusoidal saturation switching function acting on the β axis.

[0071] (2) The sinusoidal saturation switching function f(s) is expressed as:

[0072]

[0073] Where δ represents the frequency coefficient, and σ represents the boundary layer width; s represents the sliding surface.

[0074] (3) The variable gain function H based on the coordinated control of velocity error and state variables is expressed as:

[0075]

[0076] Where, k b Represents the base gain, and k b >0; e x Indicates speed error, and μ represents the proportionality coefficient, and 0 < μ < 1; v represents the attenuation factor, and v > 0; s represents the sliding surface; e -v|s| This represents an exponential function with a sliding surface; in this embodiment of the invention, k can be set. b =181, μ=0.5, v=0.5;

[0077] The aforementioned basic gain k b Conditions to be met:

[0078]

[0079] Among them, e α Represents the back electromotive force along the α axis; e β This represents the back electromotive force along the β axis.

[0080] In step S4, the product of the sinusoidal saturation switching function and the variable gain function is used as the equivalent control quantity of the novel sliding mode observer, and the back electromotive force (EMF) estimate is obtained based on the equivalent control quantity; the back EMF estimate is expressed as:

[0081]

[0082]

[0083] in, This represents the estimated value of the back electromotive force along the α axis; The value of the back electromotive force on the β-axis is represented by H; H represents the variable gain function based on the coordinated control of velocity error and state variables; f(s) α f(s) represents a sinusoidal saturation switching function acting on the α-axis; β ) represents a sinusoidal saturation switching function acting on the β-axis; k b Indicates the base gain; e x Indicates speed error; μ represents proportionality coefficient; v represents attenuation factor; s represents sliding surface; s α This represents the sliding surface acting on the α-axis; s β This represents the sliding surface acting on the β axis.

[0084] In step S5, to reduce the observation errors of the observed rotational speed and position angle, a phase-locked loop (PLL) is combined with a sliding mode observer to construct a PLL-based calculation stage for rotational speed and position. Specifically, the estimated back electromotive force is input into a PLL structure, and the estimated rotor electrical angular velocity and estimated rotor position angle are extracted through the PLL structure. Figure 2 As shown, it specifically includes:

[0085] When the actual rotor position angle θ is different from the estimated rotor position angle Error satisfies At that time, the actual rotor position angle θ and the estimated rotor position angle The relationship is ;

[0086] Based on the actual rotor position angle θ and the estimated rotor position angle The relationship is used to calculate the back electromotive force difference Δe:

[0087]

[0088] Where k represents the magnitude of the estimated back electromotive force; This represents the estimated value of the back electromotive force along the α axis; The value represents the estimated back electromotive force of the β-axis; Δθ represents the difference between the actual rotor position angle θ and the estimated rotor position angle. angular error;

[0089] When the back electromotive force difference Δe approaches zero, the rotor estimated position angle Converging to the actual rotor position angle θ; the back electromotive force difference Δe is passed through a gain 1 / k e The obtained angular error Δθ is then fed into the PI controller to obtain the estimated electric angular velocity of the rotor. In a PI controller, K p K is the proportional term. i S is the integral term, and S is the Laplace operator;

[0090] Estimating the electrical angular velocity of the rotor Integrating the results yields the final estimated rotor position angle. .

[0091] In step S6, sensorless vector control of the permanent magnet synchronous motor is achieved by using the estimated electric angular velocity of the sub-engine and the estimated position angle of the rotor; this step can be implemented by a sensorless control system for permanent magnet synchronous motors based on a sliding mode observer.

[0092] In this embodiment of the invention, the system adopts a speed-current dual closed-loop structure, such as... Figure 3 As shown, it specifically includes an outer velocity loop and an inner current loop; wherein:

[0093] The outer speed ring is based on the preset target speed. Estimating the electric angular velocity of the rotor The difference in mechanical speed, converted from 30 / πp, is used by a PI controller to generate two q-axis current command values ​​in the rotating coordinate system. ;

[0094] The inner current loop then receives the q-axis current command value. And receive two d-axis current command values ​​in the rotating coordinate system. = 0 command, and respectively with q-axis feedback current i q and d-axis feedback current i d The comparison results are then processed by a PI controller to generate d-axis voltage commands u. d and q-axis voltage command u q Based on the q-axis voltage command u q and d-axis voltage command u d The permanent magnet synchronous motor (PMSM) is driven by an inverter via SVPWM.

[0095] The key to this system lies in its sensorless technology. It utilizes a sliding mode observer to monitor the input voltage command u in real time. α u β and the actual sampled current i α and i β This allows for the accurate deduction of the motor's real-time state. Through subsequent processing, the system ultimately obtains the estimated electrical angular velocity of the rotor. And rotor estimated position angle These two estimates are the core of the entire system: the estimated electrical angular velocity of the rotor. As feedback from the speed loop, the rotor estimates the position angle. This provides a precise angular reference for coordinate transformation in FOC control, thereby enabling complete closed-loop control.

[0096] In summary, this invention provides a sensorless control method for permanent magnet synchronous motors based on a sliding mode observer. Firstly, addressing the limitations of traditional sliding mode observers in suppressing speed chatter, a novel sliding mode observer structure is proposed. This structure replaces the sign function with a sinusoidal saturation function, eliminating the need for a low-pass filter and fundamentally solving the phase lag problem. Secondly, a dynamic gain function based on speed error and state variables is designed, thereby effectively suppressing speed chatter and reducing errors. Compared to traditional sliding mode observers, this invention improves system control accuracy and reduces system complexity, making it particularly suitable for applications such as robotics that require smooth control, providing a reliable method for reducing speed chatter. In this embodiment of the invention, the electrical parameters of the permanent magnet motor are as follows: inductance Ld=Lq=0.3mH, resistance Rs=0.5Ω, rotor flux linkage ψf=0.083Wb, number of permanent magnet pairs p=4, rated speed 1000rpm, control cycle Ts=0.0002s, motor moment of inertia J=0.0001, and friction coefficient B=0.0022.

[0097] The various embodiments in this specification are described in a progressive manner, with each embodiment focusing on the differences from other embodiments. The same or similar parts between the various embodiments can be referred to each other.

[0098] The above description of the disclosed embodiments enables those skilled in the art to make or use the invention. Various modifications to these embodiments will be readily apparent to those skilled in the art, and the general principles defined herein may be implemented in other embodiments without departing from the spirit or scope of the invention. Therefore, the invention is not to be limited to the embodiments shown herein, but is to be accorded the widest scope consistent with the principles and novel features disclosed herein.

Claims

1. A method for sensorless control of permanent magnet synchronous motor based on sliding mode observer, applied to a sensorless control system of permanent magnet synchronous motor based on sliding mode observer; characterized in that, The methods include: S1. Collect the three-phase current and three-phase voltage of the permanent magnet synchronous motor; S2. Through coordinate transformation, the three-phase current and three-phase voltage are converted into two-phase current components and two-phase voltage components in a stationary coordinate system; S3. Input the two-phase current components and two-phase voltage components to the sliding mode observer; wherein the sliding mode observer uses a sinusoidal saturation switching function to replace the sign function, and uses a variable gain function based on the coordinated control of speed error and state variables; S4. The product of the sinusoidal saturation switching function and the variable gain function is used as the equivalent control quantity of the sliding mode observer, and the back electromotive force is estimated based on the equivalent control quantity. S5. Input the back electromotive force estimate into a phase-locked loop structure, and extract the rotor estimated electric angular velocity and rotor estimated position angle through the phase-locked loop structure. S6. Sensorless vector control of the permanent magnet synchronous motor is achieved by using the rotor to estimate the electric angular velocity and the rotor to estimate the position angle.

2. The sensorless control method of permanent magnet synchronous motor based on sliding mode observer according to claim 1, characterized in that, The sliding mode observer is represented as follows: ； ； ； where i α represents the α-axis stator current component in the stationary coordinate system; i β represents the β-axis stator current component in the stationary coordinate system; represents the α-axis stator current component in the stationary coordinate system; i represents the β-axis stator current component in the stationary coordinate system; p represents the differential operator; R s represents the stator resistance; L s represents the stator inductance; u α represents the α-axis stator voltage component in the stationary coordinate system; u β represents the β-axis stator voltage component in the stationary coordinate system; H represents a variable gain function based on speed error and state quantity cooperation control; f(s α ) represents a sinusoidal saturation switching function acting on the α-axis; f(s β ) represents a sinusoidal saturation switching function acting on the β-axis.

3. The sensorless control method of permanent magnet synchronous motor based on sliding mode observer according to claim 1, characterized in that, The sinusoidal saturation switching function f(s) is expressed as: ； where δ denotes the frequency coefficient, and ; σ denotes the boundary layer width; s denotes the sliding surface.

4. The sensorless control method for a permanent magnet synchronous motor based on a sliding mode observer according to claim 1, characterized in that, The variable gain function H based on the coordinated control of velocity error and state variables is expressed as: ； Where, k b Represents the base gain, and k b >0; e x Indicates speed error, and μ represents the proportionality coefficient, and 0 < μ < 1; v represents the attenuation factor, and v > 0; s represents the sliding surface.

5. The sensorless control method for a permanent magnet synchronous motor based on a sliding mode observer according to claim 4, characterized in that, Base gain k b The condition must be met: ； where e α represents the α-axis back electromotive force; e β represents the β-axis back electromotive force; f(s α ) represents a sinusoidal saturation switching function acting on the α-axis; and f(s β ) represents a sinusoidal saturation switching function acting on the β-axis.

6. The sensorless control method for a permanent magnet synchronous motor based on a sliding mode observer according to claim 1, characterized in that, The estimated value of the back electromotive force is expressed as: ； ； in, This represents the estimated value of the back electromotive force along the α axis; The value of the back electromotive force on the β-axis is represented by H; H represents the variable gain function based on the coordinated control of velocity error and state variables; f(s) α f(s) represents a sinusoidal saturation switching function acting on the α-axis; β ) represents a sinusoidal saturation switching function acting on the β-axis; k b Indicates the base gain; e x Indicates speed error; μ represents proportionality coefficient; v represents attenuation factor; s represents sliding surface; s α This represents the sliding surface acting on the α-axis; s β This represents the sliding surface acting on the β axis.

7. The sensorless control method for a permanent magnet synchronous motor based on a sliding mode observer according to claim 1, characterized in that, In step S5, extracting the estimated rotor electrical angular velocity and estimated rotor position angle through the phase-locked loop structure specifically includes: When the actual rotor position angle θ is different from the estimated rotor position angle Error satisfies At that time, the actual rotor position angle θ and the estimated rotor position angle The relationship is ; Based on the actual rotor position angle θ and the estimated rotor position angle The relationship is used to calculate the back electromotive force difference Δe: ； Where k represents the magnitude of the estimated back electromotive force; This represents the estimated value of the back electromotive force along the α axis; The value represents the estimated back electromotive force of the β-axis; Δθ represents the difference between the actual rotor position angle θ and the estimated rotor position angle. angular error; The back electromotive force difference Δe is passed through a gain of 1 / k. e The obtained angular error Δθ is then fed into the PI controller to obtain the estimated electric angular velocity of the rotor. ; Estimate the electrical angular velocity of the rotor. Integrating the results yields the final estimated rotor position angle. .

8. A sensorless control method for a permanent magnet synchronous motor based on a sliding mode observer according to any one of claims 1-7, characterized in that, The sensorless control system for permanent magnet synchronous motors based on sliding mode observers adopts a speed-current dual closed-loop structure.

9. A sensorless control method for a permanent magnet synchronous motor based on a sliding mode observer according to claim 8, characterized in that, The velocity-current dual closed-loop structure includes an outer velocity loop and an inner current loop; The outer velocity ring is based on a preset target rotational speed. Estimating the electric angular velocity of the rotor The difference in mechanical speed, converted from 30 / πp, is used by a PI controller to generate two q-axis current command values ​​in the rotating coordinate system. ; The inner current loop receives the q-axis current command value. And receive two d-axis current command values ​​in the rotating coordinate system. = 0 command, and respectively with q-axis feedback current i q and d-axis feedback current i d The comparison results are then processed by a PI controller to generate q-axis voltage commands u. q and d-axis voltage command u d ; based on the q-axis voltage command u q and a d-axis voltage command u d to drive the permanent magnet synchronous motor in operation by driving the inverter.

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