Flexible interconnection equipment based on magnetic coupling transformer cascaded H bridge for medium-voltage 10kV power distribution network and low-frequency power elimination method thereof

Abstract

The invention provides a modularized multi-level flexible interconnection topology based on a magnetic coupling transformer and a low-frequency power elimination strategy of the modularized multi-level flexible interconnection topology, and aims to solve the problems that sub-modules in a flexible interconnection system of a medium-voltage power distribution network are large in capacitance and low-frequency power fluctuation is remarkable. In order to solve the problem that the requirement for a sub-module capacitor is large, a six-active-bridge magnetic coupling transformer structure is introduced, three-phase power is gathered in a magnetic coupling loop and counteracts the three-phase power, and therefore low-frequency voltage fluctuation of the sub-module capacitor is remarkably weakened. And a low-frequency power elimination control strategy is further adopted, so that the equivalent impedance at the position of the system is remarkably reduced, and the low-frequency power mainly flows through the MCTSAB instead of the capacitor. According to the topology, the output voltage can still be kept stable under the condition that the capacitance value is reduced by 90%, low-frequency ripples are effectively restrained, and a feasible scheme is provided for a flexible interconnection system of a medium-voltage power distribution network.

Description

Technical Field

[0001] This invention relates to the field of new energy power system technology, and in particular to a flexible interconnection device (Cascaded H-Bridge - magnetically coupled transformer Six-active-bridge, CHB-MCTSAB) for medium-voltage 10kV distribution networks and its low-frequency power elimination method. Background Technology

[0002] Due to the temporal and spatial randomness of distributed renewable energy and electric vehicle charging and discharging, distribution networks are prone to problems such as feeder overload and voltage exceeding limits, seriously affecting the safe operation of the distribution network. Flexible interconnection switches (FIS) based on power electronics technology can achieve rapid and accurate regulation of power flow, thereby optimizing the power flow distribution between distribution substations and providing a new opportunity to solve the above problems.

[0003] In existing demonstration projects of flexible interconnection in medium-voltage distribution networks, the solid-state transformer flexible interconnection switch (FIS-SST) is more suitable for medium-voltage urban distribution networks due to its small size and light weight. This type of FIS-SST often adopts a three-stage transformation structure consisting of an input stage, an isolation stage, and an output stage, such as... Figure 1 As shown in the diagram, the input and output stages employ a cascaded H-bridge structure, while the isolation stage uses a dual active bridge (DAB) structure. The three-stage converter is connected via capacitors in the input stage (Cascaded H-Bridge, CHB) and the output stage (CHB) submodule (SM). The power and voltage across this capacitor are affected by the power difference: the constant input power charging the capacitor minus the low-frequency fluctuation output power discharging from the CHB, and the low-frequency fluctuation power transmitted by the CHB through the AC feed lines at both ends is not in phase. Therefore, the capacitor voltages across the DAB both exhibit a second-harmonic power fluctuation component with different phases, which is a result of power balance.

[0004] To suppress power fluctuations at second harmonics and ensure that the voltage fluctuation of the SM capacitor remains within acceptable limits, large capacitors are essential. If electrolytic capacitors are used, considering their lifespan of approximately 3000 to 5000 hours, which is halved for every 10°C increase in temperature, this would drastically reduce the system's lifespan and reliability. Using large-capacity film capacitors would lead to a significant increase in system size. In CHB (Chain of Buses) systems, the SM capacitor accounts for more than 50% of the system's volume and weight. Therefore, reducing the capacitance value of CHB submodules is of great significance for practical engineering applications.

[0005] Therefore, how to eliminate low-frequency power in the flexible interconnection switch of the isolated solid-state transformer is a technical problem that urgently needs to be solved in the lightweight design and application of the flexible interconnection switch of the solid-state transformer. Summary of the Invention

[0006] To address the aforementioned problems, this invention provides a flexible interconnection device based on a cascaded H-bridge with magnetically coupled transformers. This invention also provides a low-frequency power elimination method suitable for this converter.

[0007] like Figure 2 As shown, in the flexible interconnection equipment based on a magnetically coupled transformer cascaded H-bridge, all SM input and output ports are cascaded at 10kV medium-voltage AC ports, belonging to a structure where input ports are in series and output ports are in series. Capacitor C i,1 and C i,2 Capacitors are provided to support the input and output ports of the submodule. The submodule uses a high-frequency transformer with a turns ratio of 1:1 to electrically isolate the two feeders, achieving fault isolation. The high-frequency isolation transformer adopts a six-port magnetic coupling structure to couple the three-phase submodules together, forming a second harmonic power cancellation channel.

[0008] The working objectives of the flexible interconnection device based on the cascaded H-bridge of magnetic coupling transformer are: (1) to realize the power flow transfer between medium-voltage 10kV distribution networks and improve the carrying capacity of the distribution network; (2) to eliminate the double frequency power on the capacitor voltage of the sub-module and realize the lightweight design of the capacitor of the sub-module; (3) the six-port magnetic coupling transformer couples the three-phase sub-modules together, which can realize the phase power self-balancing, so there is no need to use additional phase voltage balancing control, which reduces the control difficulty. The SM capacitor voltage on the feeder 1 side is controlled by global voltage equalization through the CHB on the feeder 1 side. Global voltage equalization control can ensure that the active power exchange between feeder 1 and CHB on the feeder 1 side is balanced. The average value u of the capacitor voltage of each phase H-bridge sub-module is calculated. ca1_1~n u cb1_1~n u cc1_1~n The values ​​are added together and averaged to obtain the average value u of the capacitor voltages of all submodules in CHB on feeder 1 side. dcavg Reference value u for capacitor voltage of all submodules in CHB on feeder 1 side. ref * with u dcavg The difference is input to the PI controller after subtraction to obtain the reference value i of the d-axis current in the CHB on feeder 1 side. d1 * Ultimately, this achieves a balance in the active power exchange between the CHB on feeder 1 and feeder 1; the CHB on feeder 2 adopts a power flow control strategy to control the three-phase voltage u of feeder 2. a2 u b2 u c2 Perform the Park transformation to obtain its d and q components u in the synchronous rotating coordinate system.d2 and u q2 By adjusting the reference value P2 of the active power transmitted by CHB on feeder 2. * The reference value for reactive power Q2 * This can change the active and reactive power of the interaction between feeder 1 and feeder 2. P2 * and Q2 * All currents are fed into the power control module, from which reference values ​​i for the d-axis and q-axis can be obtained. d2 * and i q2 * Ultimately, this enables the exchange of active power between feeder 1 and feeder 2.

[0009] During the transmission of active power from feeder 1 to feeder 2, low-frequency power needs to be provided for both feeders. This low-frequency power is supplied by the capacitors of the submodules, thus requiring large submodule capacitors, which severely impacts the lightweight design of solid-state transformer-type flexible interconnect equipment. Taking feeder 1 as an example, let's analyze the source of the low-frequency power. The voltage of the A-phase submodule is u. a1 =1.414U rms1 sin(w1t) / N, current is i a1 =1.414I rms1 sin(w1t- The instantaneous active power of the submodule is p. SMa1 =u SMa1 ×i SMa1 =P0×cos( )-p 2wt (0), where P0=U rms1 ×I rms1 / N、P 2wt (0)=P0cos(2w1t- It is evident that in addition to the active power component that varies with the phase angle, the instantaneous active power of the submodule also includes a second harmonic power component that does not vary with the phase angle. This invention proposes to isolate and connect the three phases using a magnetically coupled transformer structure, providing an elimination path for the second harmonic power. The active power of the submodule for phases A, B, and C are respectively p SMa1 =P0+p 2wt (0), p SMb1 =P0+p 2wt (-2pi / 3), p SMc1 =P0+p 2wt (2pi / 3), by collecting the instantaneous power of the three-phase SM to the magnetic coupling transformer of MCTSAB, the total power P of the three-phase SM can be obtained. SM-sum =p SMa1 +p SMb1 +p SMc1 =3P0, therefore, as Figure 3As shown, by combining the three-phase power from the CHB submodule to the MCTSAB converter, the second harmonic power components can be mutually canceled out, instead of being transmitted to the submodule capacitor, thus effectively reducing the capacitance value of the submodule capacitor. Attached Figure Description

[0010] Figure 1 CHB-DAB type solid-state transformer flexible interconnection equipment.

[0011] Figure 2 Flexible interconnection equipment based on magnetically coupled transformer cascaded H-bridge.

[0012] Figure 3 Second harmonic power flow diagram of flexible interconnection equipment based on magnetically coupled transformer cascaded H-bridge.

[0013] Figure 4 Control block diagram of CHB on AC feeder 1 side.

[0014] Figure 5 Control block diagram of CHB on side 2 of AC feeder.

[0015] Figure 6 This is the control block diagram of MCTSAB.

[0016] Figure 7 This is the LFPE control strategy diagram.

[0017] Figure 8 This is a simulation diagram of feeder power flow control.

[0018] Figure 9 This is a diagram of the DC-side input and output voltages when the SM capacitor value is 220uF.

[0019] Figure 10 This is a diagram of the DC-side input and output voltages when the SM capacitor value is 1mF.

[0020] Figure 11 A comparison chart of LFPE control and single-voltage control. Detailed Implementation

[0021] The technical solution of the present invention will be further described in detail below through specific embodiments.

[0022] Flexible interconnection equipment based on magnetically coupled transformer cascaded H-bridges, such as Figure 2 As shown, the three-phase submodules are connected together through a six-port magnetic coupling transformer to form a low-frequency power cancellation channel; the input and output ports of the submodules are all cascaded.

[0023] In a flexible interconnection device based on a magnetically coupled transformer cascaded H-bridge, all SM input and output ports are cascaded at 10kV medium-voltage AC ports, resulting in a structure where input ports are in series and output ports are in series. Capacitor C is... i,1 and C i,2 Capacitors are provided to support the input and output ports of the submodule. The submodule uses a high-frequency transformer with a turns ratio of 1:1 to electrically isolate the two feeders, achieving fault isolation. The high-frequency isolation transformer adopts a six-port magnetic coupling structure to couple the three-phase submodules together, forming a second harmonic power cancellation channel.

[0024] The control block diagram of the present invention is as follows: Figure 4 and Figure 5 The figures show the global voltage control block diagram for CHB on feeder 1 side and the power flow control block diagram for CHB on feeder 2 side, respectively. Both diagrams employ current decoupling control for the inner loop current. First, the current decoupling control is explained. The relationship between the three-phase currents of CHB and the three-phase currents of feeder 1 can be obtained from Kirchhoff's voltage law:

[0025] (1)

[0026] In the formula: u a1 u b1 u c1 The three-phase voltage output from CHB on feeder 1 side; R is the equivalent loss resistance of the system; u sa1 u sb1 u sc1 This represents the three-phase voltage of feeder 1.

[0027] Transforming equation (1) into a two-phase synchronous rotating coordinate system, we obtain the corresponding loop equation:

[0028] (2)

[0029] In the formula: i d1 i q1 These are the d-axis and q-axis components of the CHB current on feeder 1, respectively; u sd1 u sq1 These are the d-axis and q-axis components of the voltage of feeder 1, respectively; u d1 u q1 These are the d-axis and q-axis components of the CHB output voltage on feeder 1, respectively; u Cd1 u Cq1 These are the d-axis and q-axis components of the capacitor voltage of the CHB submodule on feeder 1, respectively. Due to the presence of the connecting reactor L1, i sd1 with i sq1 The controls are mutually influential and coupled, so current decoupling control is required.

[0030] The control block diagram of CHB on feeder 1 side is as follows: Figure 4As shown, the three-phase voltage u of feeder 1 sa1 u sb1 u sc1 As the input to the phase-locked loop, the corresponding angular frequency w1 is obtained as follows: Figure 4 As shown in the figure. In the figure: PLL, abc-dq represent synchronous rotating coordinate system phase-locked loops. The control block diagram for the CHB current decoupling control on feeder 1 side is as follows. Figure 4 As shown, the three-phase current i of CHB on feeder 1 side is... a1 i b1 i c1 Perform the Park transformation to obtain its d-axis and q-axis components i in the synchronous rotating coordinate system. d1 and i q1 Reference value i of the d-axis current in CHB on feeder 1 side. d1 * It can be obtained from the global equalization control section.

[0031] The CHB control block diagram on feeder 2 is as follows Figure 5 As shown, its current decoupling control section is the same as that of CHB on feeder 1 side, the difference being the reference value i of the d-axis current in CHB on feeder 2 side. d1 * This is obtained from power flow control. According to instantaneous reactive power theory, the power flow expression for feeder 2 is:

[0032] (3)

[0033] According to equation (3), it is not difficult to obtain the reference values ​​i of the d-axis and q-axis currents in CHB on feeder 2. d1 * and i q1 * .

[0034] The reference values ​​of the d-axis and q-axis currents of CHB on the feeder 1 (feeder 2) side are subtracted from the actual values, and the difference is input into the proportional-integral (PI) controller. At the same time, the d-axis and q-axis currents are decoupled and the d-axis and q-axis voltages of feeder 1 (feeder 2) are fed forward. Finally, after inverse Park transformation, the three-phase modulated voltage of CHB current decoupling control on the feeder 1 (feeder 2) side is obtained.

[0035] MCTSAB employs hybrid synchronous phase shift control (LFPE-HSPSC) with low-frequency power cancellation to control the low-frequency power cancellation of the output port voltage and the SM capacitor. The control logic of MCTSAB's HSPSC strategy is as follows: Figure 6 As shown, the primary side of MCTSAB has four switching transistors (M... p,1 -M p,4All full bridges on the primary side share the same set of complementary switching signals g. M_i,1 and g M_i,2 M p,1 and M p,4 (p=a,b,c) is determined by the switch signal g M_i,1 Drive, M p,2 and M p,3 By g M_i,2 Drive; the secondary side of the MCTSAB also has four switching transistors (S p,1 -S p,4 ), where S p,1 and S p,4 Having the same driving signal g Sp_i,1 S p,2 and S kp,3 They have the same driving signal g Sp_i,2 , where g Sp_i,1 and g Sp_i,2 They are a pair of complementary signals.

[0036] The purpose of employing the HSPSC strategy is to control the stability of the MCTSAB output voltage. For example... Figure 6 As shown, MCTSAB has four drive signals g. Sa_i g Sb_i g Sc_i g M_i , where g M_i It is g Sa_i g Sb_i g Sc_i The reference drive signal. Therefore, there are three independent phase shift angles between the drive signals: drive signal g Sa_i and g M_i The phase shift angle d between Sa_i Drive signal g Sb_i and g M_i The phase shift angle d between Sb_i Drive signal g Sc_i and g M_i The phase shift angle d between them is defined as follows: Sc_i By changing d Sa_i d Sb_i and d Sc_i It can achieve output voltage control of MCTSAB, and d Sa_i d Sb_i and d Sc_i It can be controlled independently.

[0037] To reduce the capacitance requirements of the submodules, MCTSAB needs to handle and transmit at a frequency of The fluctuating power component. Given that the quasi-proportional resonant (QPR) controller can achieve high gain and eliminate steady-state error at specific frequencies, this paper introduces a QPR control strategy into the control loop to enhance the system's performance. The gain at a given frequency has a transfer function form as shown in formula (4).

[0038] (4)

[0039] In the formula k p0 k r For coefficients, This is the cutoff frequency.

[0040] like Figure 7 The diagram shows the LFPE control strategy for the i-th CHB submodule, where p=(a,b,c) represents three phases. This control strategy first uses the difference u between the reference voltage value and the actual voltage value of the submodule capacitor. out * -u Cp2_i As the input signal, the control quantity D is obtained after passing through a PI circuit. Si At the same time, the output current i flowing to CHB CHB_i With the current i flowing out of MCTSAB M_i The difference is processed by a low-pass filter and then enters the discrete QPR controller. The output of the QPR controller then interacts with D... Si Superimposed, forming the phase shift angle control quantity d Sp_i .

[0041] Build such a system in Matlab / Simulink Figure 2 The simulation model of the 10kV flexible interconnection system is shown. To verify the power flow regulation capability of CHB-MCTSAB, three operating conditions were designed. Specifically: 1) 0~0.5s is condition 1, where feeder 2 supplies 0.2 pu of active power to feeder 1, and CHBs on both feeder 1 and feeder 2 sides supply 0.4 pu of reactive power to the feeders; 2) 0.5~1s is condition 2, where the reactive power of each feeder remains constant, and the active power of feeder 2 decreases and reverses; 3) 1.5~2s is condition 3, where the active power of feeder 2 remains constant, the reactive power of feeder 1 decreases and reverses, and the reactive power of feeder 2 increases. Figure 8 The simulation results for power flow control show that CHB-MTCSAB can achieve precise decoupling control of active and reactive power under all three operating conditions.

[0042] To verify the low-frequency power cancellation effect of MCTSAB, a comparative simulation of CHB-DAB and CHB-MCTSAB was conducted, such as... Figure 9 and Figure 10As shown. When using the CHB-DAB topology, the SM capacitor needs to smooth low-frequency power fluctuations, mainly around 100Hz. Therefore, when the SM capacitor is 220μF, the CHB-DAB topology cannot operate stably. The DC-side input and output waveforms of the CHB-DAB are as follows: Figure 9 As shown in (c) and (d). The CHB-MCTSAB provides a low-frequency power cancellation path; therefore, with an SM capacitor of 220μF, the average voltage can be stabilized at 3000V through output voltage control, as shown in (c) and (d). Figure 9 As shown in (a) and (b), the input voltage and output voltage ripple rates are only 2.7% and 3.3%, respectively.

[0043] When the SM capacitor is 1mF, such as Figure 10 As shown in (a) and (b), the CHB-MCTSAB still operates stably, and the output voltage ripple is significantly reduced. Figure 10 As shown in (c) and (d), the CHB-DAB can also operate stably, with DC-side input and output voltage ripples of 8V and 12V, respectively. However, since this paper does not employ phase-to-phase voltage control, the input voltage of the CHB-DAB is unbalanced. In contrast, the CHB-MCTSAB, using a six-port magnetically coupled transformer, can achieve phase-to-phase self-balancing, reducing control complexity.

[0044] Therefore, adopting the CHB-MCTSAB topology in flexible interconnection projects can reduce both the capacitance value of the SM capacitor and the control complexity.

[0045] To compare the performance of single-voltage control with the adopted LFPE control strategy, the control mode was switched from LFPE to single-voltage control at 0.1 s. Figure 11 As shown in the simulation results, the voltage fluctuation of the submodule capacitor increases from 57.7V to 147.8V after switching, and the voltage oscillation becomes more pronounced. This result indicates that LFPE control, by introducing low-frequency feedback of the submodule capacitor current, effectively enhances the MCTSAB's ability to absorb low-frequency power components, thereby significantly suppressing low-frequency voltage fluctuations. Therefore, even with a small capacitor, the system can maintain voltage stability and improve the quality of the output voltage waveform.

[0046] The above description is merely a preferred embodiment of the present invention and is not intended to limit the present invention. Any modifications, equivalent substitutions, improvements, etc., made within the spirit and principles of the present invention should be included within the protection scope of the present invention.

Claims

1. A flexible interconnection device based on a magnetically coupled transformer cascaded H-bridge for a medium-voltage 10kV distribution network, characterized in that, Includes a three-phase submodule, a six-port magnetic coupling transformer, and a submodule input support capacitor C. i,1 and output support capacitor C i,2 The input and output ports of each submodule (SM) are cascaded at 10kV medium-voltage AC ports, forming a topology of series input ports and series output ports. The six-port magnetic coupling transformer is a high-frequency isolation transformer with a turns ratio of 1:1, which couples the three-phase submodules to form a double-frequency power elimination channel, while realizing electrical isolation and fault isolation between the two feeders.

2. The flexible interconnection equipment based on a magnetically coupled transformer cascaded H-bridge for a medium-voltage 10kV distribution network according to claim 1, characterized in that, The six-port magnetic coupling transformer gathers the instantaneous power of the three-phase sub-module, causing the second harmonic power components of the three-phase sub-module to cancel each other out, thus preventing the second harmonic power from being transmitted to the sub-module capacitor; and achieves phase-to-phase power self-balancing through the magnetic coupling structure, eliminating the need for an additional phase-to-phase voltage balancing control module.

3. A low-frequency power elimination method applied to the flexible interconnection equipment according to claim 1 or 2, characterized in that, Includes the following steps: S1: Global voltage equalization control is applied to the cascaded H-bridge (CHB) on feeder 1 side, and the average value u of the capacitor voltage of all sub-modules on feeder 1 side is calculated. dcavg The voltage reference value u ref * with u dcavg The difference is input to the PI controller to obtain the d-axis current reference value i. d1 * This achieves active power balance between feeder 1 and CHB on feeder 1 side; S2: A power flow control strategy is adopted for CHB on feeder 2 side. Park transformation is performed on the three-phase voltage of feeder 2 to obtain the d and q components u in the synchronous rotating coordinate system. d2 and u q2 By adjusting the active power reference value P2 * and reactive power reference value Q2 * The power control module obtains the reference values ​​i for the d-axis and q-axis currents. d2 * and i q2 * This enables the exchange of active and reactive power between the two feeders; S3: The instantaneous power of the three-phase submodule is collected through a six-port magnetic coupling transformer. The active power of the three-phase submodule is p. SMa1 =P0+p 2wt (0), p SMb1 =P0+p 2wt (-2pi / 3), p SMc1 =P0+p 2wt (2pi / 3), total power P SM-sum =p SMa1 +p SMb1 +p SMc1 =3P0, cancels the second harmonic power component; S4: Hybrid synchronous phase-shift control with low-frequency power cancellation (LFPE-HSPSC) is employed to control the low-frequency power cancellation of the output port voltage and submodule capacitors of the flexible interconnect equipment. The LFPE-HSPSC strategy stabilizes the output voltage by controlling the phase shift angle of the drive signals of the primary and secondary side switches, and introduces quasi-proportional resonance (QPR) control to enhance the system's performance. Gain of frequency components; S5: The LFPE control strategy takes the difference between the reference value and the actual value of the submodule capacitor voltage as input, and obtains the control quantity D after passing through a PI circuit. Si The difference between the CHB output current and the MCTSAB output current is processed by a low-pass filter and then input into a discrete QPR controller, whose output is compared with D. Si The superposition forms the phase shift angle control quantity d Sp_i This suppresses low-frequency fluctuations in capacitor voltage.

4. The low-frequency power elimination method according to claim 3, characterized in that, In step S4, the driving signal for the hybrid synchronous phase-shift control includes the primary-side switch driving signal g. M_i,1 and g M_i,2 and secondary-side switch drive signal g Sp_i,1 g Sp_i,2 , and g M_i,1 With g M_i,2 Complementary, g Sp_i,1 With g Sp_i,2 Complementary; driving signal g Sa_i g Sb_i g Sc_i With g M_i As a reference signal, the three are adjusted independently with respect to g. M_i phase shift angle d Sa_i d Sb_i d Sc_i This enables output voltage control.

5. The low-frequency power elimination method according to claim 3, characterized in that, In step S4, the transfer function of the quasi-proportional resonance (QPR) controller is: In the formula k p0 k r For coefficients, This is the cutoff frequency.

6. The low-frequency power elimination method according to claim 3, characterized in that, In steps S1 and S2, the inner loop currents of CHB on both feeder 1 and feeder 2 sides are controlled by current decoupling. The three-phase currents are converted to a synchronous rotating coordinate system through Park transformation. Combined with voltage feedforward and PI control, the three-phase modulated voltage is obtained through inverse Park transformation.

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