Ultra wide band (UWB) residual sideband (RSB) calibration using dual phase shift keying (BPSK) signal sources

Abstract

A system for radio frequency (RF) residual sideband (RSB) calibration includes a complex (in-phase / quadrature (I / Q)) signal receiver; a signal generator configured to generate a transmit (Tx) signal; a first phase shifter operably coupled to the real signal transmitter; a first signal combiner configured to combine a receive (Rx) signal with the transmit (Tx) signal to generate a first combined signal; a second phase shifter configured to provide a selected phase shift to the first combined signal; and a complex down converter configured to alternately generate an in-phase Rx signal and a quadrature Rx signal using the in-phase LO signal and the quadrature LO signal to generate one or more signals indicative of relative Tx-Rx LO phase (theta), amplitude (A), Tx LO I / Q phase mismatch (epsilon), Rx I / Q amplitude mismatch (alpha), and Rx I / Q phase mismatch () at the output of the complex receiver.

Description

Technical Field

[0001] This disclosure relates generally to electronic devices, and more specifically to amplifiers in transceivers. Background Technology

[0002] Wireless communication devices and technologies are becoming increasingly prevalent, much like communication systems. Communication systems can be configured to use so-called ultra-wideband (UWB) communication technology, a radio technology that uses low-energy signals for short-range, high-bandwidth communication. One application of UWB radio includes location determination and ranging techniques, where radio frequency signals can be used to determine the location of user equipment (UE). For example, communication devices such as smartphones and receivers (called tags) can be used to determine the location of the tag. In some applications, UWB radio systems include receivers (Rx) capable of processing complex quadrature (in-phase (I) and quadrature (Q)) signals and transmitters that generate pulsed radio signals, such as dual phase-shift keying (BPSK) signals, which are real signals generated by modulating a constant-frequency carrier by changing the phase of the signal between 0 and 180 degrees.

[0003] Two indicators that can be used to determine the location of a UE using UWB technology include the time of arrival (ToA) and angle of arrival (AoA) of the radio signal. Therefore, it is desirable to have a way to accurately determine the ToA and AoA of the received signal. Summary of the Invention

[0004] The various embodiments of the systems, methods, and apparatuses within the scope of the appended claims each have several aspects, none of which individually fully encompasses the desired properties described herein. Certain prominent features are described herein without limiting the scope of the appended claims.

[0005] Details of one or more specific embodiments of the subject matter described in this specification are set forth in the accompanying drawings and the following description. Other features, aspects, and advantages will become apparent from the description, drawings, and claims. It should be noted that the relative dimensions in the following drawings may not be drawn to scale.

[0006] One aspect of this disclosure provides a radio frequency (RF) vestigial sideband (RSB) calibration circuit comprising: a complex (in-phase / quadrature (I / Q)) signal receiver; a signal generator configured to generate a transmitted (Tx) signal; a first phase shifter connected to the signal generator, the first phase shifter being configured to selectively process one of an in-phase local oscillator (LO) signal and a quadrature LO signal to create an initial phase between the received in-phase (I) signal and the quadrature (Q) signal, thereby generating a balanced I / Q signal; a first signal combiner configured to combine the received (Rx) signal with the transmitted (Tx) signal to generate a first combined signal; and a second phase shifter configured to provide a selected phase shift to the first combined signal. The circuit also includes: a downconverter configured to generate in-phase Rx signals and quadrature Rx signals; and an I / Q amplitude / phase error estimation element configured to determine the amplitude and phase errors of the in-phase and quadrature Rx signals, and configured to generate a phase compensation signal. The signal includes an amplitude compensation signal (α'); a first multiplier configured to generate a compensated in-phase Rx signal and a second multiplier configured to generate a compensated quadrature Rx signal; a second signal combiner configured to combine the compensated in-phase Rx signal and the compensated quadrature Rx signal to generate a frequency-independent compensation signal; and a filter (h(t)) configured to generate a frequency-dependent compensation signal.

[0007] Another aspect of this disclosure provides a method for radio frequency (RF) vestigial sideband (RSB) calibration, the method comprising: enabling a vestigial sideband (RSB) signal generation circuit; adjusting a first phase shifter in the signal generation circuit such that a received (Rx) baseband (BB) in-phase signal and an Rx BB quadrature signal have similar amplitudes; setting a second phase shifter to an initial code (dCT+ / -Δ); selecting a first (in-phase) local oscillator (LO) calibration signal (CAL_LO_I) at the RSB signal generation circuit; and measuring the Rx BB-I output (I) of an in-phase baseband filter (BBF I(t)) in response to the in-phase LO calibration signal (CAL_LO_I). TX I RX =Acos(θ)) and the Rx BB-Q output (I) of the quadrature baseband filter (BBF Q(t)). TX Q RX =αAcos(θ– +90°); select the second (quadrature) local oscillator (LO) calibration signal (CAL_LO_Q) at the RSB signal generation circuit; measure the Rx BB-I output (Q) of the in-phase baseband filter (BBF I(t)) in response to the quadrature LO calibration signal (CAL_LO_Q). TX I RX =Acos(θ+ε-90°)) and the Rx BB-Q output (Q) of the orthogonal baseband filter (BBF Q(t)). TX Q RX =αAcos(θ+ε- Derive the Rx I / Q phase error correction factor. c; Derive the Rx I / Q gain ratio αc; Calculate the Rx I / Q phase error correction factor c and the Rx I / Q gain ratio αc are applied to the Rx BB-I signal and the Rx BB_Q signal to obtain a frequency-independent compensation signal for input to the correction function (h(t)); the second phase shifter is adjusted to obtain an additional set of outputs from the in-phase baseband filter (BBF I(t)) and the quadrature baseband filter (BBF Q(t)); and the adjusted output of the in-phase baseband filter (BBF I(t)) after the second phase shifter is adjusted to obtain the additional set of outputs from the in-phase baseband filter (BBF I(t)) and the quadrature baseband filter (BBF Q(t)) is combined with the output of the correction function (h(t)) to generate a frequency-dependent compensation signal.

[0008] Another aspect of this disclosure provides an apparatus for performing radio frequency (RF) vestigial sideband (RSB) calibration, the apparatus comprising: components for enabling a vestigial sideband (RSB) signal generation circuit; components for adjusting a first phase shifter in the signal generation circuit such that the received (Rx) baseband (BB) in-phase signal and the Rx BB quadrature signal have similar amplitudes; components for setting a second phase shifter to an initial code (dCT+ / -Δ); components for selecting a first (in-phase) local oscillator (LO) calibration signal (CAL_LO_I) at the RSB signal generation circuit; and components for measuring the Rx BB-I output (I) of the in-phase baseband filter (BBF I(t)) in response to the in-phase LO calibration signal (CAL_LO_I). TX I RX =Acos(θ)) and the Rx BB-Q output (I) of the quadrature baseband filter (BBFQ(t)). TX Q RX =αAcos(θ– The components include: a +90° component; a component for selecting the second (quadrature) local oscillator (LO) calibration signal (CAL_LO_Q) at the RSB signal generation circuit; and a component for measuring the Rx BB-I output (Q) of the in-phase baseband filter (BBF I(t)) in response to the quadrature LO calibration signal (CAL_LO_Q). TX I RX =Acos(θ+ε-90°)) and the Rx BB-Q output (Q) of the orthogonal baseband filter (BBF Q(t)). TX Q RX =αAcos(θ+ε- The component used to derive the Rx I / Q phase error correction factor. The component c; the component used to derive the Rx I / Q gain ratio αc; the component used to correct the Rx I / Q phase error factor. The components include: c and the Rx I / Q gain ratio αc applied to the Rx BB-I signal and the Rx BB_Q signal to obtain a frequency-independent compensation signal for input to the correction function (h(t)); components for adjusting the second phase shifter to obtain an additional set of outputs from the in-phase baseband filter (BBF I(t)) and the quadrature baseband filter (BBF Q(t)); and components for combining the adjusted output of the in-phase baseband filter (BBF I(t)) after the second phase shifter is adjusted to obtain the additional set of outputs from the in-phase baseband filter (BBF I(t)) and the quadrature baseband filter (BBF Q(t)) with the output of the correction function (h(t)) to generate a frequency-dependent compensation signal.

[0009] Another aspect of this disclosure provides a system for radio frequency (RF) vestigial sideband (RSB) calibration, the system comprising: a complex (in-phase / quadrature (I / Q)) signal receiver; a signal generator configured to generate a transmit (Tx) signal; and a first phase shifter operatively coupled to a real signal transmitter, the first phase shifter being configured to selectively process one of an in-phase local oscillator (LO) signal and a quadrature LO signal to create a signal between the received in-phase (I) and quadrature (Q) signals. An initial phase is established to generate a balanced I / Q signal; a first signal combiner is configured to combine the received (Rx) signal with the transmitted (Tx) signal to generate a first combined signal; a second phase shifter is configured to provide a selected phase shift to the first combined signal; and a complex downconverter is configured to alternately use the in-phase LO signal and the quadrature LO signal to generate in-phase Rx signals and quadrature Rx signals to generate relative Tx-Rx LO phase (θ), amplitude (A), Tx LO I / Q phase mismatch (ε), Rx I / Q amplitude mismatch (α), and Rx I / Q phase mismatch (α) at ​​the output of the complex receiver. One or more signals. Attached Figure Description

[0010] In the accompanying drawings, unless otherwise indicated, similar reference numerals are used throughout the various views to refer to similar parts. For reference numerals with letter characters, such as "102a" or "102b", the letter characters distinguish two similar parts or elements in the same drawing. When the aim is to have the reference numerals cover all parts with the same reference numerals in all drawings, the letter characters of the reference numerals may be omitted.

[0011] Figure 1 This is a diagram illustrating communication between a wireless device and a wireless communication system.

[0012] Figure 2A This is a block diagram illustrating a wireless device in which exemplary technologies of this disclosure may be implemented.

[0013] Figure 2B This is a block diagram illustrating a wireless device in which exemplary technologies of this disclosure may be implemented.

[0014] Figure 3A A schematic diagram of a portion of a radio frequency integrated circuit (RFIC) with residual sideband (RSB) calibration circuitry is shown.

[0015] Figure 3B It shows having Figure 3A A schematic diagram showing a more detailed view of a portion of the radio frequency integrated circuit (RFIC) for residual sideband (RSB) calibration circuitry.

[0016] Figure 4 This is a graph showing the detected I and Q amplitudes (A) as a function of the digital capacitor tuning code.

[0017] Figure 5 This is a graph showing the relative Rx-Tx phase shift (θ) as a function of the digital capacitor tuning code.

[0018] Figure 6A and Figure 6B Together constitute Figure 3B A graphical example of the operation of the correction function h(t).

[0019] Figure 7A and Figure 7B Together, they show a flowchart illustrating an example of the operation of a method for residual sideband (RSB) calibration in an ultra-wideband (UWB) transmitter.

[0020] Figure 8A and Figure 8B Together, a device for residual sideband (RSB) calibration in an ultra-wideband (UWB) transmitter is shown. Detailed Implementation

[0021] The word “exemplary” is used in this document to mean “serving as an example, instance, or illustration.” Any aspect described as “exemplary” in this document is not necessarily to be construed as preferred or superior to other aspects.

[0022] According to an exemplary embodiment, a residual sideband (RSB) calibration (CAL) circuit includes a method for calibrating the RSB of a transceiver using a real (BPSK) transmit signal generated by an ultra-wideband (UWB) transmitter.

[0023] AoA and ToA can be determined by analyzing the residual sideband (RSB) of the received signal, where the phase difference between the Tx and Rx signals is used to determine AoA. AoA can be determined by analyzing the phase difference of the received signals between different antenna chains. To accurately determine the relative phase difference between two receiver chains, the RX I / Q mismatch, which can be represented as RSB or signal pattern, should be calibrated below the receiver's sensitivity level; therefore, RSB calibration is required. There are several methods for performing RSB calibration, including factory calibration and the more cost-effective on-chip calibration. Factory calibration is less than ideal due to its relatively high cost and inability to track voltage-temperature variations, making on-chip calibration the preferred solution. However, in UWB radio systems, only the receiver can process complex signals, while the transmitter can only generate real (BPSK) signals; therefore, traditional on-chip RSB calibration is not feasible because there is no complex or single-sideband (SSB) transmit signal available to inject into the receiver path. Furthermore, while ToA is relatively unaffected by I / Q imbalance, AoA is more severely affected by I / Q imbalance. Therefore, when only the real (BPSK) transmit signal is available, it is difficult to determine the AoA of the received signal.

[0024] Figure 1 This diagram illustrates communication between wireless device 110 and wireless communication system 120. Wireless communication system 120 can be a Long Term Evolution (LTE) system, a Code Division Multiple Access (CDMA) system, a Global System for Mobile Communications (GSM) system, a Wireless Local Area Network (WLAN) system, a 5G NR (New Radio) system, or some other wireless system. The CDMA system can implement Wideband CDMA (WCDMA), CDMA 1X, Evolved Data Optimized (EVDO), Time Division Synchronous CDMA (TD-SCDMA), or some other version of CDMA. For simplicity, Figure 1 A wireless communication system 120 is shown, comprising two base stations 130 and 132 and a system controller 140. Generally, a wireless communication system may include any number of base stations and any set of network entities.

[0025] Wireless device 110 may also be referred to as user equipment (UE), mobile station, terminal, access terminal, subscriber unit, station, etc. Wireless device 110 may be a cellular phone, smartphone, tablet device, wireless modem, personal digital assistant (PDA), handheld device, laptop computer, smartbook, netbook, tablet computer, cordless phone, medical device, automobile, device configured to connect to one or more other devices (e.g., via the Internet of Things), wireless local loop (WLL) station, Bluetooth device, etc. Wireless device 110 may communicate with wireless communication system 120. Wireless device 110 may also receive signals from broadcast stations (e.g., broadcast station 134) and / or may communicate with satellites (e.g., one or more satellites 150 in a Global Navigation Satellite System (GNSS) or satellites capable of receiving signals from wireless device 110, etc. Wireless device 110 may support one or more radio technologies for wireless communication, such as LTE, WCDMA, CDMA 1X, EVDO, TD-SCDMA, GSM, 802.11, 802.15, 5G, Sub6 5G, 6G, UWB, etc.

[0026] Wireless device 110 may support carrier aggregation, such as that described in one or more LTE or 5G standards. In some implementations, carrier aggregation is used to transmit a single data stream on multiple carriers, as opposed to separate carriers used for each data stream. Wireless device 110 is capable of operating in a variety of communication frequency bands, including those used by LTE, WiFi, 5G, or other communication frequency bands within a wide frequency range. Wireless device 110 is also capable of communicating directly with other wireless devices without communicating through a network.

[0027] In some embodiments, the wireless communication system 120 may include a location device 152 that communicates with the wireless device 110. In some embodiments, the location device 152 and the wireless device 110 may include UWB communication technology, and the wireless device 110 may use the technology described herein to determine the location of the location device 152.

[0028] Figure 2A This is a block diagram illustrating a wireless device 200 in which exemplary technologies of the present disclosure may be implemented. The wireless device 200 may be, for example, Figure 1 The illustrated implementation scheme of wireless device 110.

[0029] Figure 2A An example of a transceiver 220 with a transmitter 230 and a receiver 250 is shown. Generally, the conditioning of the signals in the transmitter 230 and receiver 250 can be performed by one or more stages such as amplifiers, filters, up-converters, down-converters, etc. These circuit blocks are based on... Figure 2AThe configurations shown are arranged differently. Furthermore, Figure 2A Other circuit blocks, not shown, can also be used to regulate the signals in transmitter 230 and receiver 250. Unless otherwise indicated, Figure 2A Any signal in any of the other diagrams in the attached figures may be single-ended or differential. Figure 2A Some circuit blocks can also be omitted.

[0030] exist Figure 2A In the example shown, wireless device 200 typically includes transceiver 220 and data processor 210. Data processor 210 may include processor 296 operatively coupled to memory 298. Memory 298 may be configured to store data and program code generally shown using reference numeral 299, and may typically include analog and / or digital processing components. Processor 296 and memory 298 may cooperate to control, configure, program, or otherwise fully or partially control some or all of the operation of embodiments of the residual sideband (RSB) calibration (CAL) circuitry described herein.

[0031] Transceiver 220 includes a transmitter 230 and a receiver 250 supporting bidirectional communication. Generally, wireless device 200 may include any number of transmitters and / or receivers for any number of communication systems and frequency bands. All or part of transceiver 220 may be implemented on one or more analog integrated circuits (ICs), RF ICs (RFICs), mixed-signal ICs, etc.

[0032] Transmitters or receivers can be implemented using either a superheterodyne architecture or a direct conversion architecture. In a superheterodyne architecture, the signal undergoes multiple stages of frequency conversion between radio frequency (RF) and baseband; for example, for a receiver, this might involve switching from RF to intermediate frequency (IF) in one stage and then from IF to baseband in another. In a direct conversion architecture, the signal is converted between RF and baseband in a single stage. Superheterodyne and direct conversion architectures can utilize different circuit blocks and / or have different requirements. Figure 2A In the example shown, transmitter 230 and receiver 250 are implemented using a direct conversion architecture.

[0033] In the transmission path, data processor 210 processes the data to be transmitted and provides in-phase (I) analog output signals and quadrature (Q) analog output signals to transmitter 230. In an exemplary embodiment, data processor 210 includes digital-to-analog converters (DACs) 214a and 214b for converting digital signals generated by data processor 210 into I analog output signals and Q analog output signals (e.g., I output current and Q output current) for further processing. In other embodiments, DACs 214a and 214b are included in transceiver 220, and data processor 210 provides data (e.g., for I and Q) digitally to transceiver 220.

[0034] Within transmitter 230, baseband (e.g., low-pass) filters 232a and 232b filter the I and Q analog transmit signals, respectively, to remove unwanted image frequencies caused by the preceding digital-to-analog conversion. Amplifiers (Amps) 234a and 234b amplify the signals from baseband filters 232a and 232b, respectively, and provide the I and Q baseband signals. Upconverter 240, with upconverters 241a and 241b, uses the I TX LO and Q TX LO signals from transmit (TX) local oscillator (LO) signal generator 290 to upconvert the I and Q baseband signals, and provide the upconverted signals. Filter 242 filters the upconverted signals to remove unwanted image frequencies caused by frequency upconversion and noise in the receive band. Power amplifier (PA) 244 amplifies the signal from filter 242 to obtain the desired output power level and provide the transmit RF signal. The transmitted RF signal can be routed through a duplexer or switch 246 and transmitted via antenna 248. Although the examples discussed herein utilize I and Q signals, those skilled in the art will understand that components of the transceiver can be configured to utilize polarity modulation.

[0035] In the receiving path, antenna 248 receives communication signals and provides the received RF signal, which can be routed through duplexer or switch 246 and provided to low-noise amplifier (LNA) 252. Duplexer 246 is designed to operate with specific RX and TX duplexer frequencies, thus isolating the RX and TX signals. The received RF signal is amplified by LNA 252 and filtered by filter 254 to obtain the desired RF input signal.

[0036] Downconverter mixers 261a and 261b in downconverter 260 mix the output of filter 254 with the I RX LO signal and Q RX LO signal (i.e., LO_I and LO_Q) from receive (RX) LO signal generator 280 to generate I baseband and Q baseband signals. The I baseband and Q baseband signals are amplified by amplifiers 262a and 262b and further filtered by baseband (e.g., low-pass) filters 264a and 264b to obtain I analog input signals and Q analog input signals, which are provided to data processor 210. In the illustrated exemplary embodiment, data processor 210 includes analog-to-digital converters (ADCs) 216a and 216b for converting the analog input signals into digital signals to be further processed by data processor 210. In some embodiments, ADCs 216a and 216b are included in transceiver 220 and provide data digitally to data processor 210.

[0037] exist Figure 2A In this configuration, TX LO signal generator 290 generates I TX LO and Q TX LO signals for up-conversion, while RX LO signal generator 280 generates I RX LO and Q RX LO signals for down-conversion. Each LO signal is a periodic signal with a specific base frequency. Phase-locked loop (PLL) 292 receives a reference clock from a processor or an external clock source and generates control signals for adjusting the frequency and / or phase of the TX LO signals from LO signal generator 290. Similarly, PLL 282 receives timing information from data processor 210 and generates control signals for adjusting the frequency and / or phase of the RX LO signals from LO signal generator 280.

[0038] The wireless device 200 may support carrier aggregation (CA) and may (i) receive multiple downlink signals transmitted by one or more cells on multiple downlink carriers at different frequencies, and / or (ii) transmit multiple uplink signals to one or more cells on multiple uplink carriers. However, those skilled in the art will understand that the aspects described herein may be implemented in systems, devices, and / or architectures that do not support carrier aggregation.

[0039] Figure 2AThe transceiver 220 is functionally illustrated in the text, and the illustrated configuration may or may not represent the physical device configuration in certain specific implementations. For example, as described above, the transceiver 220 may be implemented in various integrated circuits (ICs), RF ICs (RFICs), mixed-signal ICs, etc. In some embodiments, the transceiver 220 is implemented on a substrate or board (such as a printed circuit board (PCB)) having various modules, chips, and / or components. For example, the power amplifier 244, filter 242, and duplexer 246 may be implemented in separate modules or as discrete components, while the remaining components illustrated in the transceiver 220 may be implemented in a single transceiver chip.

[0040] Power amplifier 244 may include one or more stages, such as driver stages, power amplifier stages, or other components that may be configured to amplify communication signals at one or more frequencies, in one or more frequency bands, and at one or more power levels. Depending on various factors, power amplifier 244 may be configured to operate using one or more driver stages, one or more power amplifier stages, one or more impedance matching networks, and may be configured to provide good linearity, efficiency, or a combination of good linearity and efficiency.

[0041] In exemplary implementations of the superheterodyne architecture, PA 244 and LNA 252 (and in some examples, filters 242 and 254) may be implemented separately from other components in transmitter 230 and receiver 250 (e.g., on a millimeter-wave integrated circuit). Figure 2B The example superheterodyne architecture is illustrated in the figure.

[0042] Figure 2B This is a block diagram illustrating a wireless device in which exemplary technologies of this disclosure may be implemented.

[0043] Figure 2B An example of a transceiver 220a with a transmitter 230 and a receiver 250 is shown. Generally, the conditioning of the signals in the transmitter 230 and receiver 250 can be performed by one or more stages such as amplifiers, filters, up-converters, down-converters, etc. These circuit blocks are based on... Figure 2B The configurations shown are arranged differently. Furthermore, Figure 2B Other circuit blocks, not shown, can also be used to regulate the signals in transmitter 230 and receiver 250. Unless otherwise indicated, Figure 2B Any signal in any of the other diagrams in the attached figures may be single-ended or differential. Figure 2BSome circuit blocks may also be omitted. In an exemplary embodiment, transceiver 220a may include an ultra-wideband (UWB) transmitter, which may be implemented in, for example, a wireless communication device and in a positioning or location device that can accurately estimate the distance between two UWB-equipped devices. The determination of AoA may be used in the positioning or location device to accurately determine the location or position of the device.

[0044] exist Figure 2B The image shows a specific implementation with a UWB transmitter 230. Figure 2B In the example shown, wireless device 200a typically includes transceiver 220a and data processor 210. Data processor 210 may include processor 296 operatively coupled to memory 298. Memory 298 may be configured to store data and program code generally shown using reference numeral 299, and may typically include analog and / or digital processing components. Processor 296 and memory 298 may cooperate to control, configure, program, or otherwise fully or partially control some or all of the operation of the embodiments of the residual sideband (RSB) calibration (CAL) circuitry described herein.

[0045] Transceiver 220 includes a transmitter 230 and a receiver 250 supporting bidirectional communication. Generally, wireless device 200a may include any number of transmitters and / or receivers for any number of communication systems and frequency bands. All or part of transceiver 220a may be implemented on one or more analog integrated circuits (ICs), RF ICs (RFICs), mixed-signal ICs, etc.

[0046] In the transmission path, data processor 210 processes the data to be transmitted and provides in-phase (I) or quadrature (Q) analog output signals to transmitter 230. In an exemplary embodiment, data processor 210 includes a digital-to-analog converter (DAC) 214a for converting digital signals generated by data processor 210 into I analog output signals or Q analog output signals (e.g., I output current or Q output current) for further processing. In other embodiments, DAC 214a is included in transceiver 220, and data processor 210 provides data (e.g., for I or Q) digitally to transceiver 220.

[0047] Within transmitter 230, a baseband (e.g., low-pass) filter 232a filters either the I-analog transmit signal or the Q-analog transmit signal to remove unwanted mirroring caused by the preceding digital-to-analog conversion. An amplifier (Amp) 234a amplifies the signal from baseband filter 232a and provides either the I-baseband signal or the Q-baseband signal. An upconverter 240 with upconversion mixer 241a, an I / Q multiplexer 311, and a coarse-tuned phase shifter 320 utilize the I TX LO signal or the Q TX LO signal from transmit (TX) local oscillator (LO) signal generator 290 to upconvert the I-baseband signal or the Q-baseband signal and provide the upconverted signal.

[0048] In an exemplary embodiment, a single PLL 292 provides a frequency reference signal to the TX LO signal generator 290 and the RX LO signal generator 280. The TX LO signal generator 290 provides an I LO signal and a Q LO signal to an I / Q multiplexer 311. The I / Q multiplexer selects either the I LO signal or the Q LO signal based on a control signal supplied by a data processor 210 or another controller. The I LO signal or the Q LO signal is provided to a coarse phase shifter 320. The coarse phase shifter 320 applies a coarse phase shift to delay the phase of the selected signal from the I / Q multiplexer 311 and provides an output to a mixer 241a. The output of the mixer 241a is provided to an amplifier 321 and a power amplifier 244. The amplifier 321 provides a signal loopback function and provides the I or Q output of the mixer 241a as a current-mode TX-RX loopback TX-RX signal to a fine phase shifter 330 via connection 322. In an exemplary embodiment, the fine phase shifter 330 acts as the load of the LNA 252 in the receiver 250. The load provided by the fine phase shifter 330 may take the form of an inductor (L) capacitor (C) (LC) resonant circuit and thus acts as an impedance that converts the current signal provided by the amplifier 321 into a voltage signal at the input of the downconverter 260. The signal provided by the amplifier 321 is "on-chip," making it possible to perform accurate analysis of the signal provided by the mixer 241a, as described below.

[0049] Power amplifier (PA) 244 amplifies the signal from mixer 241a to obtain a desired output power level and provide a transmit RF signal. The transmit RF signal can be routed through TX / RX switch 247 and transmitted via antenna 248. Although the examples discussed herein utilize I and Q signals, those skilled in the art will understand that transceiver components can be configured to utilize polarity modulation. In an exemplary embodiment, amplifier 321 provides an amplified signal to receiver 250.

[0050] In the receiving path, antenna 248 receives communication signals and provides the received RF signal, which can be routed through TX / RX switch 247 and provided to low-noise amplifier (LNA) 252. TX / RX switch 247 is designed to operate such that the RX signal is isolated from the TX signal. The received RF signal is amplified by LNA 252 and phase-shifted / filtered by fine phase shifter 330. In an exemplary embodiment, fine phase shifter 330 may include an inductor (L) capacitor (C) circuit, sometimes referred to as an LC resonant circuit, which provides filtering and phase shifting, as will be described in more detail below. The output of fine phase shifter 330 is provided to downconverter 260.

[0051] Downconverter mixers 261a and 261b in downconverter 260 mix the output of filter 254 with the I RX LO signal and Q RX LO signal (i.e., LO_I and LO_Q) from receive (RX) LO signal generator 280 to generate I baseband and Q baseband signals. The I baseband and Q baseband signals are amplified by amplifiers 262a and 262b and further filtered by baseband (e.g., low-pass) filters 264a and 264b to obtain I analog input signals and Q analog input signals, which are provided to data processor 210. In the illustrated exemplary embodiment, data processor 210 includes analog-to-digital converters (ADCs) 216a and 216b for converting the analog input signals into digital signals to be further processed by data processor 210. In some embodiments, ADCs 216a and 216b are included in transceiver 220 and provide data digitally to data processor 210.

[0052] exist Figure 2B In this configuration, TX LO signal generator 290 generates I TX LO and Q TX LO signals for up-conversion, while RX LO signal generator 280 generates I RX LO and Q RX LO signals for down-conversion. Each LO signal is a periodic signal with a specific base frequency. Phase-locked loop (PLL) 292 receives timing information from data processor 210 and generates control signals for adjusting the frequency and / or phase of the TX LO signals from LO signal generator 290 and adjusting the frequency and / or phase of the RX LO signals from LO signal generator 290.

[0053] Figure 3A A schematic diagram of a portion of a radio frequency integrated circuit (RFIC) with residual sideband (RSB) calibration (CAL) circuitry is shown. In an exemplary embodiment, the RSB CAL circuitry 300 may be located in... Figure 2A or Figure 2BThe transceiver 220 is included in or is part of the transceiver 220. In an exemplary embodiment, the RSB CAL circuit 300 includes a low-noise amplifier (LNA) 303 configured to receive a received (Rx) signal via connection 302. The amplified Rx signal is provided by the LNA 303 to the signal combiner 306 via connection 304.

[0054] In an exemplary embodiment, the RSB CAL circuit 300 further includes an RSB transmit (Tx) circuit 310. The RSB Tx circuit 310 includes a multiplexer 311 configured to receive an in-phase calibration local oscillator (LO) signal (CAL_LO_I) via connection 308 and an orthogonal calibration local oscillator (LO) signal (CAL_LO_Q) via connection 309. The CAL_LO_I and CAL_LO_Q signals can be... Figure 2A or Figure 2B The Tx LO signal generator 290 is provided, or it is provided by another LO circuit. Multiplexer 311 selects whether to provide a CAL_LO_I signal or a CAL_LO_Q signal via connection 312 based on a control signal on connection 313 provided by data processor 210 or another controller. In an exemplary embodiment, RSB transmit (Tx) circuitry 310 may be referred to as a signal generator when performing RSB calibration as described herein.

[0055] The CAL_LO_I or CAL_LO_Q signal is provided to the coarse phase shifter 320. The coarse phase shifter 320 applies a phase shift to the signal on connection 312 and provides an output to the mixer 318 via connection 316. The phase shift applied by the coarse phase shifter 320 can be obtained from the data processor 210 via connection 315. Figure 2A or Figure 2BControlled by control signals from another controller application. In an exemplary embodiment, the coarse-adjustment phase shifter 320 may be implemented using an adjustable delay buffer, a low-pass resistance (R) capacitor (C) (RC) filter, or another circuit. When both the RX I signal and the RX Q signal are at similar amplitudes, the coarse-adjustment phase shifter 320 adjusts the TX-RX relative phase to obtain an optimized phase setting, where the similar amplitude indicates that the relative phase between TX and RX is close to 45 degrees*N, where N is any integer and is arbitrary as it will remain rotating from 0 to 2π and back to 0. The mixer 318 receives the baseband (BB) dual phase shift keying (BPSK) transmit signal via connection 317 and provides the up-converted BPSK signal via connection 319. The signal on connection 319 is the BBBPSK signal up-converted based on the CAL_LO_I signal or CAL_LO_Q signal from connection 317. The Tx signal on connection 319 is amplified by amplifier 321 and provided to signal combiner 306 via connection 322.

[0056] Signal combiner 306 combines the complex Rx signal on connection 304 with the real Tx signal on connection 322 and provides the combined signal on connection 307. The combined signal on connection 307 is provided to downconverter circuit 335. In an exemplary embodiment, fine phase shifter 330 is also connected to the output of signal combiner 306 via connection 324. In an exemplary embodiment, fine phase shifter 330 includes an inductor (L) capacitor (C) circuit, sometimes referred to as an LC resonant circuit. Fine phase shifter 330 provides a fine phase tuning stage by adjusting its L / C value. In an exemplary embodiment, fine phase shifter 330 can vary between inductor impedance, real impedance, and capacitor impedance to provide a slight phase shift relative to its initial value. In an exemplary embodiment, fine phase shifter 330 is a bandpass filter (BPF) and also an impedance and current-mode signal combiner. In an exemplary embodiment, fine phase shifter 330 includes an LC resonant circuit 339 with inductor 334 and adjustable capacitor 332. The adjustable capacitor 332 can be adjusted based on a control signal provided by the data processor 210 or another controller. The operation of the coarse phase shifter 320 and the fine phase shifter 330 will be described in more detail below.

[0057] The amplified combined signal on connection 307 is provided to downconverter circuit 335. Downconverter circuit 335 may include in-phase mixer 327 and quadrature mixer 331. In-phase mixer 327 receives in-phase Rx_LO_I signal via connection 328, and quadrature mixer 331 receives quadrature Rx_LO_Q signal via connection 333. The output of in-phase mixer 327 is provided to in-phase baseband filter (BBF I(t)) 337 via connection 334; and the output of quadrature mixer 331 is provided to quadrature baseband filter (BBF Q(t)) 338 via connection 336.

[0058] The output of (BBF I(t)) 337 is provided to analog-to-digital converter (ADC) 344 via connection 341, and the output of (BBF Q(t)) 338 is provided to ADC 346 via connection 342. The digital output of ADC 344 is provided via connection 347, and the digital output of ADC 346 is provided via connection 348. The signals on connections 347 and 348 are further processed by digital detection and compensation circuitry 349, as described below.

[0059] Figure 3B It shows having Figure 3A A schematic diagram of a portion of a radio frequency integrated circuit (RFIC) 301 for residual sideband (RSB) calibration (CAL) circuitry. In an exemplary embodiment, the RSB CAL circuitry 301 may be located in... Figure 2A or Figure 2B The transceiver 220 is included in or is part of the transceiver. In an exemplary embodiment, the RSB CAL circuit 300 includes a low-noise amplifier (LNA) 303 configured to receive a received (Rx) signal via connection 302. The amplified Rx signal is provided by the LNA 303 to the signal combiner 306 via connection 304.

[0060] In an exemplary embodiment, the RSB CAL circuit 300 further includes an RSB transmit (Tx) circuit 310. The RSB Tx circuit 310 includes a multiplexer 311 configured to receive an in-phase calibration local oscillator (LO) signal (CAL_LO_I) via connection 308 and an orthogonal calibration local oscillator (LO) signal (CAL_LO_Q) via connection 309. The CAL_LO_I and CAL_LO_Q signals can be... Figure 2A or Figure 2B The Tx LO signal generator 290 provides the signal, or it can be provided by another LO circuit. The multiplexer 311 selects whether to provide the CAL_LO_I signal or the CAL_LO_Q signal via connection 312 based on a control signal on connection 313 provided by data processor 210 or another controller.

[0061] The CAL_LO_I or CAL_LO_Q signal is provided to the coarse phase shifter 320. The coarse phase shifter 320 applies a phase shift to the signal on connection 312 and provides an output to the mixer 318 via connection 316. The phase shift applied by the coarse phase shifter 320 can be obtained from the data processor 210 via connection 315. Figure 2A or Figure 2B Controlled by control signals from another controller application. In an exemplary embodiment, the coarse-adjustment phase shifter 320 may be implemented using an adjustable delay buffer, a programmable low-pass resistor (R) capacitor (C) (RC) filter, or another circuit. When both the RX I and RX Q signals are at similar amplitudes, the coarse-adjustment phase shifter 320 adjusts the TX-RX relative phase to obtain an optimized setting where the similar amplitude indicates that the relative phase between TX and RX is close to 45 degrees*N, where N is 0 degrees, 90 degrees, 180 degrees, or 270 degrees. The mixer 318 receives the baseband (BB) dual phase shift keying (BPSK) transmit signal via connection 317 and provides the up-converted BPSK signal via connection 319. The signal on connection 319 is the BB BPSK signal on connection 317 up-converted based on the frequency of the CAL_LO_I or CAL_LO_Q signal. The Tx signal on connection 319 is amplified by amplifier 321 and provided to signal combiner 306 via connection 322.

[0062] Signal combiner 306 combines the complex Rx signal on connection 304 with the real Tx signal on connection 322 and provides the combined signal on connection 307. The combined signal on connection 307 is provided to Rx amplifier 325. In an exemplary embodiment, fine phase shifter 330 is also connected to the output of signal combiner 306 via connection 324. In an exemplary embodiment, fine phase shifter 330 includes an inductor (L) capacitor (C) circuit, sometimes referred to as an LC resonant circuit. In an exemplary embodiment, fine phase shifter 330 includes an LC resonant circuit 339 having an inductor 334 and an adjustable capacitor 332. The adjustable capacitor 332 can be adjusted based on a control signal provided by data processor 210 or another controller. Fine phase shifter 330 provides a fine phase tuning stage by adjusting the values ​​of capacitor 332 and inductor 334. In an exemplary embodiment, fine phase shifter 330 can vary between inductive impedance, real impedance, and capacitive impedance to provide a slight phase shift relative to its initial value. In an exemplary embodiment, the fine phase shifter 330 is a bandpass filter (BPF) and also an impedance and current-mode signal combiner. The operation of the coarse phase shifter 320 and the fine phase shifter 330 will be described in more detail below.

[0063] The amplified combined signal on connection 326 is provided to downconverter circuit 335. Downconverter circuit 335 may include in-phase mixer 327 and quadrature mixer 331. In-phase mixer 327 receives in-phase Rx_LO_I signal via connection 328, and quadrature mixer 331 receives quadrature Rx_LO_Q signal via connection 333. The output of in-phase mixer 327 is provided to in-phase baseband filter (BBF I(t)) 337 via connection 334; and the output of quadrature mixer 331 is provided to quadrature baseband filter (BBF Q(t)) 338 via connection 336.

[0064] The output of (BBF I(t)) 337 is provided to analog-to-digital converter (ADC) 344 via connection 341, and the output of (BBF Q(t)) 338 is provided to ADC 346 via connection 342. The digital output of ADC 344 is provided via connection 347, and the digital output of ADC 346 is provided via connection 348.

[0065] In an exemplary embodiment, the digital signals on connection 347 and connection 348 are provided to the I / Q amplitude / phase error estimation element 350 via corresponding connections 351 and 352. In an exemplary embodiment where the Q signal path is compensated, the output XI,comp(t) of ADC 346 on connection 347 can be used as a reference signal when compensating the output of ADC 344 on connection 348. The I / Q amplitude / phase error estimation element 350 provides amplitude and phase correction signals via connections 354, 356, and 357. (and α), and will be described in more detail below. In an exemplary embodiment, the processing performed by the I / Q amplitude / phase error estimation element 350 may be performed by processor 296 or another processor.

[0066] In an exemplary embodiment, the digital signal on connection 347 is provided to multiplier 358. The digital signal on connection 347 also represents the in-phase signal correction function XI,comp(t). In an exemplary embodiment, the digital signal on connection 348 is provided to multiplier 359. The output of multiplier 359 is provided to signal combiner 362 via connection 363. The output of multiplier 358 is also provided to signal combiner 362.

[0067] In an exemplary implementation, multiplier 358 combines the XI,comp(t) signal on connection 347 with the correction function tan(t) on connection 356. The output of multiplier 358 on connection 361 includes the corrected XI,comp(t) signal. The signal on connection 361 represents a digital signal, for example, an 8-bit signal at a clock rate of 998.4MHz, where the output of ADC 344 on connection 347 is corrected by the tan( ) function on connection 356. ') Correction.

[0068] In an exemplary implementation, multiplier 359 multiplies the Q output of ADC 346 connected to 348 with the correction function 1 / α'cos( ... The signal on connection 363 represents a digital signal, for example, an 8-bit signal at a clock rate of 998.4MHz, where the output of ADC 346 on connection 348 is multiplied by the correction function 1 / α'cos( ... ') Correction.

[0069] Signal combiner 362 combines the signal on connection 361 with the signal on connection 363. The output of signal combiner 362 is provided to I / Q amplitude / phase error estimation element 350 via connection 365 and to correction function (h(t)) 370 via connection 364. In an exemplary embodiment, the signals on connections 364 and 365 represent frequency-independent (FI) correction signals XQ,FI,comp(t). In an exemplary embodiment, I / Q amplitude / phase error estimation element 350 compares the output of signal combiner 362 on connection 365 again with the signal on connection 347. I / Q amplitude / phase error estimation element 350 then provides the frequency-dependent correction signal back to correction function (h(t)) 370 via connection 369. Correction function (h(t)) 370 equalizes the I and Q signals based on frequency and provides the frequency-dependent compensation signal XQ,FD,comp(t) via connection 372. The XQ,FD,comp(t) signal on connection 372 is also fed back to the I / Q amplitude / phase error estimation element 350. The I / Q amplitude / phase error estimation element 350 processes the XQ,FD,comp(t) signal on connection 372 by performing adaptive digital filter processing so that the filter coefficients XQ,FD,comp(t) of the correction function (h(t)) 370 are as close as possible to the reference signal XI,comp(t) on connection 347.

[0070] The output of the correction function (h(t)) 370 represents the quadrature correction signal XQ,FD,comp(t) and is provided via connection 372. The output at connection 347 is the signal XIcomp(t). The signals at connections 347 and 372 can be provided to the data processor 210 (or digital modem) for further processing to determine ToA and AoA.

[0071] Although shown as being in the “Q” path, in some embodiments the correction function (h(t)) 370 may also be located in the “I” path, and in other embodiments, portions of the correction function (h(t)) 370 may be located in both the “Q” path and the “I” path.

[0072] In an exemplary implementation, the RSB can be calibrated using a real BPSK Tx signal by solving for multiple variables, which can be generated by combining a BB BPSK signal, alternately up-converted by the CAL_LO_I or CAL_LO_Q signal on connection 317, with a received signal provided by LNA 303. By alternating the CAL_LO_I and CAL_LO_Q signals in multiple iterations, multiple equations with the variables listed below can be generated. The variables are as follows: θ is the phase relative to Tx-Rx LO.

[0073] A represents the amplitude.

[0074] ε is the Tx LO I / Q phase mismatch.

[0075] α is the amplitude mismatch of Rx I / Q.

[0076] It is an Rx I / Q phase mismatch.

[0077] The variable θ indicates the relative phase difference between the Tx and Rx signals. The term “θ” refers to the Tx-RxLO phase difference at the carrier frequency, and the phase difference between TX and RX can be determined based on θ, and this phase difference is approximated by c*θ / 2pi*Fc*N, where c is the speed of light, Fc is the carrier frequency, and N is an integer. The number N can be any number, as it will remain rotating from 0 to 2pi and back to 0. TX can be a transmitting source from a distant location, and RX can be a handheld device with two (2) or more receivers and associated antennas. If the two (2) receivers are close enough, for example less than 0.5 / 2*pi*Fc, then N will be the same, and the TX-RX distance difference between the two antenna paths can be determined, and since the distance between the two (2) antennas is known (the device manufacturer knows the distance between the antennas), the 2D angle of arrival (AoA) can be determined. Using a third antenna, the 3D angle (and AoA) can also be determined. I / Q errors occur at RX I and RX Q. UWB transmitters use only I or Q, so I / Q mismatch is not a problem in UWB transmitters because only real (biphase) signals are transmitted. Since the TX I and TX Q clocks are used as calibration sources, and because they are not perfectly separated by 90 degrees, TX I and TX Q errors should be determined not to be included in the RX I / Q error estimate, which could lead to overcompensation of I / Q at RX based on incorrect estimates. Returning to the reference RX I / Q error estimate, this I / Q error will limit the ability to determine the TX-RX phase difference, and knowing the relative TX-RX phase is crucial when determining AoA. Minimizing the I / Q phase error (or imbalance) improves the accuracy of AoA determination.

[0078] In an exemplary embodiment, for the transmitted signal, the coarse-adjustment phase shifter 320 is used to generate a phase difference (θ) of approximately 45° between the CAL_LO_I and CAL_LO_Q signals to maximize the signal swing (amplitude of the voltage swing) of the CAL_LO_I and CAL_LO_Q signals. In an exemplary embodiment, this is achieved by the data processor 210 ( Figure 2A or Figure 2B The control code provided to the coarse-tuning phase shifter 320 by the controller (or another controller) is swept until RXBB-I (at the in-phase baseband filter (BBF I(t)) 337) and RXBB_Q (at the quadrature baseband filter (BBF Q(t)) 338) have similar amplitudes, which means that θ (phase difference relative to Tx-RxLO) is close to 45° and both I and Q have the maximum signal-to-noise ratio (SNR).

[0079] In an exemplary embodiment, tuning the phase delay using coarse phase shifter 320 and fine phase shifter 330 will change the relative Tx-Rx LO phase difference θ, and will change the amplitude A. In an exemplary embodiment, tuning the phase delay using coarse phase shifter 320 and fine phase shifter 330 can change the Tx LO I / Q phase mismatch ε. In an exemplary embodiment, tuning the phase delay using coarse phase shifter 320 and fine phase shifter 330 will not change the Rx I / Q amplitude mismatch α, and will not change the Rx I / Q phase mismatch. .

[0080] In an exemplary embodiment, once the coarse-adjustment phase shifter 320 sets θ as close to 45° as possible, the data processor 210 ( Figure 2A or Figure 2B The control code provided to the fine phase shifter 330 by a controller (or another controller) is based on dCT+ / -Δ to set the fine phase shifter 330. The term "dCT" refers to the digital capacitor tuning code d_cap_tuning_code. For example, the LC resonant circuit 339 is set based on dCT+ / -Δ. For example, the impedance of the fine phase shifter 330 can be expressed as R+jX, where R is the real part of the impedance and X is the imaginary part of the impedance. The phase of this impedance can be expressed as arctan(X / R). When the value of capacitor 332 (e.g., by capacitor bank tuning) changes, both R and X will change slightly, so the phase change is represented by arctan(X+dX / R+dR)-arctan(X / R)=Δθfine. RX will have a default code to keep it resonant within the band (e.g., 8GHz). This code can be specified as d_cap_tuning_code. During this fine phase tuning stage, the code can be changed by a positive 1, a negative 1, or a small increment to provide a slight phase difference, thereby generating more equations than the number of unknowns.

[0081] In an exemplary embodiment, the CAL_LO_I signal on connection 308 is selected, and the BB BPSK transmit signal up-converted by the CAL_LO_I signal is provided to signal combiner 306. The Tx signal propagates through Rx amplifier 325, and the Rx signal is measured at in-phase baseband filter (BBF I(t)) 337 and at quadrature baseband filter (BBF I(t)) 338. In an exemplary embodiment, for the CAL_LO_I signal, in-phase baseband filter (BBF I(t)) 337 produces output I. TX I RX =Acos(θ), and the quadrature baseband filter (BBF Q(t)) 338 produces the output I. TX Q RX =αAcos(θ– +90°).

[0082] Next, the in-phase (I) LO input to mixer 318 is switched to the quadrature (Q) LO input, and the CAL_LO_Q signal on connection 309 is selected, while the BB BPSK transmit signal is provided to signal combiner 306. The Tx signal propagates through Rx amplifier 325, and the Rx signal is measured at in-phase baseband filter (BBF I(t)) 337 and at quadrature baseband filter (BBF I(t)) 338. In an exemplary embodiment, for the CAL_LO_Q signal, in-phase baseband filter (BBF I(t)) 337 produces output Q. TX I RX =Acos(θ+ε-90°), and the quadrature baseband filter (BBF Q(t)) 338 produces the output Q. TX Q RX =αAcos(θ+ε- ).

[0083] In this way, the following four equations are generated: I TX I RX =Acos(θ) I TX Q RX =αAcos(θ– +90°).

[0084] Q TX I RX =Acos(θ+ε-90°) Q TX Q RX =αAcos(θ+ε- ) Next, it is determined whether there are more resulting equations than unknowns. If there are more unknowns than resulting equations, the fine phase shifter 330 is adjusted based on dCT+ / -Δ (i.e., the adjustable components in the fine phase shifter 330 are set to different values ​​to select different phases), and the above steps are repeated to generate additional resulting equations after the fine phase shifter 330 is adjusted to different phases.

[0085] After the above two iterations, the following eight (8) equations can be used to solve for the unknowns θ0, θ1ε, and θ2ε. α, A0, A1: A 0 cos (θ 0 )=I TX I RX 0 A 1 cos (θ 1 )=I TX I RX 1 A 0 cos (90°+θ 0 +ε)=Q TX I RX 0 A 1 cos (90°+θ 1 +ε)=Q TX I RX 1 αA 0 cos (90°+ -θ 0 )=I TX Q RX 0 αA 1 cos (90°+ –θ 1 )=I TX Q RX 1 αA 0 cos (θ 0 +ε- )=Q TX Q RX 0 αA 1 cos (θ 1 +ε- )=Q TX QRX 1 Next, solve the above eight (8) equations to find the following unknowns θ0, θ1ε, The values ​​of θ0, θ1, and A1 can be obtained using an iterative method in an exemplary implementation. , α, A0, A1.

[0086] For example, in the first iteration step, θ can be estimated. 0,1 and A 0,1 Assuming the mismatch term (α, ε) is close to (1, 0), the first "seed" θ can be estimated. 0,1 and A 0,1 Given α, ε, θ, and approximate values ​​of A, we can start from... αA 0 cos (θ 0 +ε- )=Q TX Q RX 0-> arccos(Q TX Q RX 0 / α / A0)-ε-θo Solving in the middle .

[0087] This will produce: A0cos(θ0)=I TX I RX 0 A0cos(90°+θ0)=Q TX I RX 0 tan(θ0)=Q TX I RX 0 / I TX I RX 0 θ0=tan -1 (Q TX I RX 0 / I TX I RX 0) and A1cos(θ1)=I TX I RX 1 A1cos(90°+θ1)=Q TX I RX 1 tan(θ1)=Q TX IRX 1 / I TX I RX 1 θ1=tan -1 (Q TX I RX 1 / I TX I RX 1) If the initial absolute value of θ0 is greater than the initial absolute value of θ1: cosθ0 / cos(90°+θ0+ε)=I TX I RX 0 / Q TX I RX 0, and then first derive θ0 from this (initially assuming ε=0).

[0088] cosθ1 / cos(90°+θ1+ε)=I TX I RX 1 / Q TX I RX 1, and then θ1 is derived from this. cos(θ0+ε– ) / cos(90°+ –θ0)=Q TX Q RX 0 / I TX Q RX 0, and then derive from this cos(θ1+ε– ) / cos(90°+ –θ1)=Q TX Q RX 1 / I TX Q RX 1, and then ε is derived from this. Re-iterate the first equation (cosθ0 / cos(90°+θ0+ε)=I) TX I RX 0 / Q TX I RX 0, and then θ0 (initially assumed ε=0) is derived from it until it is solved.

[0089] If the initial absolute value of θ0 is less than the initial absolute value of θ1: cosθ1 / cos(90°+θ1+ε)=I TX I RX 1 / Q TX I RX 1, and from this θ1 is derived (initially assuming ε=0).

[0090] cosθ0 / cos(90°+θ0+ε)=ITX I RX 0 / Q TX I RX 0, and from this θ0 is derived cos(θ0+ε– ) / cos(90°+ –θ0)=Q TX Q RX 0 / I TX Q RX 0, and then derive from this cos(θ1+ε– ) / cos(90°+ –θ1)=Q TX Q RX 1 / I TX Q RX 1, and then ε is derived from this. Re-iterate the first equation (cosθ0 / cos(90°+θ0+ε)=I) TX I RX 0 / Q TX I RX 0, and then θ0 (initially assumed ε=0) is derived from it until it is solved.

[0091] Alternative methods for using iterative estimation algorithms For θ0=atan(-Q) TX I RX0 / I TX I RX0 The initial estimate begins. Estimate A0=I TX I RX0 / cos(θ0) Using A0 and θ0, estimate ε = acos(Q) TX I RX0 / A0)–π / 2–θ0 Using ε, estimate θ1=atan Using ε and θ1, estimate =θ1–acot Estimate A1+I using θ1 TX I RX1 / cos(θ1) Then, α=I TX Q RX1 / (A1*sin(θ1– )) The new θ0 for the next iteration can be calculated by averaging the two values: θ0=asin(ITX Q RX0 / α*A0)+ θ0=acos(Q TX Q RX0 / α*A0)+ -ε The iterations can be performed from a few to ten or more times. The simulation is based on floating-point numbers and does not use trigonometric approximations. Any DC offset will affect accuracy.

[0092] After solving for the variables θ0, A0, θ1, and A1, it is then determined whether optional additional fine-tuning of the phase is required as an additional check. For example, a third tuning code dCT+ / -Δ2 (dig_cap_tuning code + / -1 or + / -2 or some fine-tuning used to introduce Δθfine) can be applied to the fine phase shifter 330, and the above steps can be repeated to perform additional analysis on θ2 and A2.

[0093] If the third fine phase adjustment is not performed, the I / Q mismatch correction factor is generated by the I / Q amplitude / phase error estimation element 350. '(or c) and α' (or αc). For example, the correction factor can be expressed as a function tan( The correction factor α' is applied to mixer 358 via connection 356, and the correction factor α' can be used as a function 1 / α' cos( The output of multiplier 358 is provided to signal combiner 362 via connection 361, and the output of multiplier 359 is provided to signal combiner 362 via connection 363. In an exemplary embodiment, I / Q amplitude / phase error estimation element 350 uses the digital outputs of ADCs 344 and 346 and calculates variables θ, ε, α, ... The solution for A. The I / Q amplitude / phase error estimation element 350 can be a CPU, DSP, or dedicated HW to perform such calculations and return correction codes on connection 354 for those digital data on connections 347 and 348 applied by multipliers 358 and 359, respectively.

[0094] Other numerical methods for solving equations are possible, as shown in the examples above.

[0095] Start the frequency-dependent (FD) RSB calibration process.

[0096] After the frequency-independent (FI) part of the RSB calibration process is completed, the frequency-dependent (FD) part of the RSB calibration process then begins.

[0097] The baseband data input on connection 317 is switched to the RF calibration path via coarse adjustment phase shifter 320. For example, DAC214a provides a BB (DC) signal to mixer 318 via connection 317 (Figure 3), which becomes an RF carrier signal once up-converted (or the IF signal is provided to mixer 318).

[0098] The amplitudes and relative DC phases of XI and XQ should follow the previously derived Tx-Rx phase difference θn on the XI path and θn+π / 2 on the XQ path.

[0099] Next, the terms of the IQ BB correction function h(t)370 are derived. Since XI(t) approximates XQ,FD,comp(t) = XQ,FI,comp(t) * h(t), h(t) is determined such that XQ,FD,comp(t) approximates XI,comp(t) * tan(-θ0), and XI(t) and XQ(t) are captured at the ADC (344 / 346) output. The function h(t) can be estimated using an adaptive filter estimation algorithm. This operation can be referred to as the frequency correlation (FD) part of RSB calibration.

[0100] Figure 4 This is a graph 400 showing the detected I and Q amplitudes (A) as a function of the digital capacitor tuning code. The vertical axis 402 represents the amplitude (variable A) in millivolts (mV), and the horizontal axis 404 represents the digital tuning code (dCT) used to adjust the coarse tuning phase shifter 320 (Figure 3).

[0101] In an exemplary embodiment, as the control code applied to the coarse-tuning phase shifter 320 (FIG. 3) sweeps within a range, the amplitudes of the Rx in-phase and Rx quadrature signals received at the in-phase baseband filter (BBF I(t)) 337 and the quadrature baseband filter (BBF Q(t)) 338, respectively, change with the changing control code. In an exemplary embodiment, trace 412 shows the detected I amplitude, trace 414 shows the detected Q amplitude, trace 422 shows the detected I-swapped amplitude (when CAL_LO_I and CAL_LO_Q are swapped), and trace 424 shows the detected Q-swapped amplitude. The area in dashed box 415 typically shows the highest amplitudes of both the in-phase and quadrature signals and represents the point where the in-phase and quadrature signals are separated in phase as close as possible to 45°.

[0102] Figure 5This is a graph 500 showing the relative Rx-Tx phase shift (θ) as a function of the digital capacitor tuning code. The vertical axis 502 represents the relative phase difference (θ) between Tx and Rx in degrees, and the horizontal axis 504 represents the digital tuning code (dCT) used to adjust the coarse phase shifter 320 (Figure 3). Trace 512 represents the tuning code of the adjustable capacitor 332 in the fine phase shifter 330 (Figure 3), and point 515 represents the control code that provides a 45° phase shift between the RX BB I(t) signal on connection 341 and the RX BB Q(t) signal on connection 342 (Figure 3).

[0103] Figure 6A and Figure 6B The operations that together constitute the correction function h(t) in Figure 3 are illustrated graphically. Figure 6A In the figure, graph 600 shows the in-phase signal X. I The trace 612 of (t) (the output of the in-phase baseband filter (BBF I(t)) 337) and the quadrature signal X Q The trace 614 of (t) (output of the quadrature baseband filter (BBF Q(t)) 338).

[0104] exist Figure 6B In the figure, graph 620 shows the in-phase signal X. I The trace 612 represents the output of the in-phase baseband filter (BBF I(t)) 337 and the trace 624 represents the corrected quadrature signal XQ,FD,comp(t)=XQ,FI,comp(t)*h(t) (output of the quadrature baseband filter (BBF Q(t)) 338) after frequency-dependent RSB calibration (i.e., the operation of the correction function h(t) 370).

[0105] Figure 7A and Figure 7B Together, a flowchart illustrating an example of the operation of a method for calibrating residual sidebands (RSB) in an ultra-wideband (UWB) transmitter is shown. The blocks in method 700 may be performed in the order shown or not, and in some embodiments, may be performed at least partially in parallel.

[0106] In block 704, TX-RX phase calibration begins by sweeping control codes applied to coarse-tuned phase shifter 320 to generate a phase difference (θ) of approximately 45° between the CAL_LO_I and CAL_LO_Q signals to maximize the signal swing (amplitude of the voltage swing) of the CAL_LO_I and CAL_LO_Q signals. In an exemplary embodiment, this is achieved by data processor 210 ( Figure 2A or Figure 2BThe control code provided to the coarse-tuning phase shifter 320 by the controller (or another controller) is swept until RXBB-I (at the in-phase baseband filter (BBF I(t)) 337) and RXBB_Q (at the quadrature baseband filter (BBF Q(t)) 338) have similar amplitudes, which means that θ (phase difference relative to Tx-Rx LO) is close to 45° and both I and Q have the maximum signal-to-noise ratio (SNR).

[0107] In box 706, by data processor 210 ( Figure 2A or Figure 2B The control codes provided to the fine phase shifter 330 by the controller (or another controller) are based on dCT+ / -Δ to set the fine phase shifter 330, as described above.

[0108] In box 708, settings are configured for multiplexer 311 ( Figure 3B The LO input of the signal is selected to select the CAL_LO_I signal on connection 308, and the BB BPSK transmit signal, which is up-converted by the CAL_LO_I signal, is provided to the signal combiner 306. The output of the in-phase baseband filter (BBF I(t)) 337 on connection 341 is measured, and the output of the quadrature baseband filter (BBFI(t)) 338 on connection 342 is measured to obtain the output amplitude I, respectively. TX I RX =Acos(θ) and I TX Q RX =αAcos(θ– +90°).

[0109] In box 712, the switch is transferred to multiplexer 311 ( Figure 3B The LO input of the signal is selected to select the CAL_LO_Q signal on connection 309, and the BB BPSK transmit signal, which is up-converted from the CAL_LO_Q signal, is provided to the signal combiner 306. The output of the in-phase baseband filter (BBF I(t)) 337 on connection 341 and the output of the quadrature baseband filter (BBFI(t)) 338 on connection 342 are measured to obtain the output amplitude Q, respectively. TX I RX =Acos(θ+ε-90°) and Q TX Q RX =αAcos(θ+ε- ).

[0110] In this way, the following four equations are generated: I TX I RX =Acos(θ) I TX Q RX =αAcos(θ– +90°).

[0111] Q TX I RX =Acos(θ+ε-90°) Q TX Q RX =αAcos(θ+ε- ) In box 714, it is determined whether there are more resulting equations than unknowns. If there are more unknowns than resulting equations, the process returns to box 706, where the fine phase shifter 330 is adjusted based on dCT+ / -Δ (i.e., the adjustable components in the fine phase shifter 330 are set to different values ​​to select different phases), and the above steps are repeated to generate additional resulting equations after the fine phase shifter 330 has been adjusted to different phases. For example, data processor 210 ( Figure 2B (Or another controller) can determine whether there are more resulting equations than unknowns. If it is determined in block 714 that there are more resulting equations than unknowns, the process proceeds to block 716.

[0112] In box 716, the iterative method described above is used to solve for the unknowns θ0, θ1ε, and θ2ε. α, A0, A1, as described above. For example, data processor 210 ( Figure 2B (or another controller) to solve for the unknowns θ0, θ1ε, α, A0, A1, as described above.

[0113] In box 720, it is determined whether additional iterations of the fine phase shifter 330 need to be adjusted. If additional iterations are determined to be needed, the process returns to box 706. If additional iterations are determined not to be needed, the process proceeds to box 722.

[0114] In block 722, the I / Q mismatch correction factor (Rx I / Q phase error correction factor and Rx I / Q gain ratio correction factor α' on connection 354) is derived from the I / Q amplitude / phase error estimation element 350.

[0115] In box 724, frequency-dependent calibration begins. For example, the baseband data input on connection 317 is switched to the RF calibration path via coarse-tuning phase shifter 320. For example, DAC 214a provides a BB (DC) signal to mixer 318 via connection 317 (Figure 3), which becomes the RF carrier signal once up-converted (or the IF signal is provided to mixer 318). The Rx I / Q phase error correction factor and Rx I / Q gain ratio correction factor α' on connection 354 are applied to multipliers 358 and 359. The amplitudes and relative DC phases of XI and XQ follow the previously derived Tx-Rx phase difference θn on the XI path and θn+π / 2 on the XQ path.

[0116] In box 728, terms of the IQ BB correction function h(t)370 are derived. Since XI(t) approximates XQ,FD,comp(t) = XQ,FI,comp(t)*h(t), h(t) is determined such that XQ,FD,comp(t) approximates XI,comp(t).tan(-θ0), and XI(t) and XQ(t) are captured at the output of the ADC (344 / 346). The function h(t) can be estimated by the I / Q amplitude / phase error estimation element 350 using the adaptive filter estimation algorithm as described above.

[0117] Figure 8A and Figure 8B Together, an apparatus for residual sideband (RSB) calibration in an ultra-wideband (UWB) transmitter is illustrated. Apparatus 800 includes a component 804 for generating a phase difference (θ) of approximately 45° between the CAL_LO_I and CAL_LO_Q signals. In some embodiments, the component 804 for generating a phase difference (θ) of approximately 45° between the CAL_LO_I and CAL_LO_Q signals may be configured to perform method 700 (… Figure 7A One or more of the functions described in operation block 704. In an exemplary embodiment, component 804 for generating a phase difference (θ) of approximately 45° between the CAL_LO_I and CAL_LO_Q signals may include sweeping control codes applied to coarse-tuned phase shifter 320 to generate a phase difference (θ) of approximately 45° between the CAL_LO_I and CAL_LO_Q signals to maximize the signal swing (amplitude of voltage swing) of the CAL_LO_I and CAL_LO_Q signals.

[0118] The apparatus may also include a component 806 for setting a fine phase shifter based on dCT+ / -Δ. In some embodiments, the component 806 for setting a fine phase shifter based on dCT+ / -Δ may be configured to perform method 700. Figure 7AOne or more of the functions described in operation block 704 of the ) are included. In an exemplary embodiment, component 806 for setting the fine phase shifter based on dCT+ / -Δ may include functions that will be controlled by data processor 210 ( Figure 2A or Figure 2B The control codes provided by (or another controller) are applied to the fine phase shifter 330 to set the fine phase shifter 330 based on dCT+ / -Δ.

[0119] The apparatus may also include a component 808 for selecting the LO input of the multiplexer to select the CAL_LO_I signal. In some embodiments, the component 808 for selecting the LO input of the multiplexer to select the CAL_LO_I signal may be configured to perform method 700. Figure 7A One or more of the functions described in operation block 708. In an exemplary embodiment, component 808 for selecting the CAL_LO_I signal to the LO input of the multiplexer may include selection to multiplexer 311 ( Figure 3B The LO input is used to select the CAL_LO_I signal and to measure the output of the in-phase baseband filter (BBF I(t)) 337 and the output of the quadrature baseband filter (BBF I(t)) 338.

[0120] The apparatus may also include a component 812 for selecting the LO input of the multiplexer to select the CAL_LO_Q signal. In some embodiments, the component 812 for selecting the LO input of the multiplexer to select the CAL_LO_Q signal may be configured to perform method 700. Figure 7A One or more of the functions described in operation block 712 of the ) are included. In an exemplary embodiment, component 812 for selecting the CAL_LO_Q signal to the LO input of the multiplexer may include selection to multiplexer 311 ( Figure 3B The LO input is used to select the CAL_LO_Q signal and to measure the output of the in-phase baseband filter (BBF I(t)) 337 and the output of the quadrature baseband filter (BBF I(t)) 338.

[0121] The apparatus may also include a component 814 for determining whether there are more resulting equations than unknowns. In some embodiments, the component 814 for determining whether there are more resulting equations than unknowns may be configured to perform method 700. Figure 7A One or more of the functions described in operation block 714. In an exemplary embodiment, component 814 for determining whether there are more result equations than unknowns may include data processor 210. Figure 2B (or another controller), the data processor determines whether there are more resulting equations than unknowns.

[0122] The apparatus may also include a component 816 for solving for unknowns. In some embodiments, the component 816 for solving for unknowns may be configured to perform method 700. Figure 7A One or more of the functions described in operation block 716. In an exemplary embodiment, component 816 for solving unknowns may include data processor 210. Figure 2B (or another controller), this data processor is used to solve for the unknowns θ0, θ1ε, α, A0, A1, as described above.

[0123] The apparatus may also include a component 820 for determining whether additional iterations are needed. In some embodiments, the component 820 for determining whether additional iterations are needed may be configured to perform method 700. Figure 7A One or more of the functions described in operation block 720. In an exemplary embodiment, component 820 for determining whether additional iterations are needed may include data processor 210. Figure 2B (or another controller), the data processor is used to determine whether additional iterations are needed.

[0124] The apparatus may also include a component 822 for deriving an I / Q mismatch correction factor. In some embodiments, the component 822 for deriving the I / Q mismatch correction factor may be configured to perform method 700. Figure 7A One or more of the functions described in operation block 722. In an exemplary embodiment, component 822 for deriving the I / Q mismatch correction factor may include an I / Q amplitude / phase error estimation element 350 for deriving the Rx I / Q phase error correction factor. 'and Rx I / Q gain ratio correction factor α'.

[0125] The device may also include a component 824 for switching to the RF calibration path via a coarse phase shifter. In some embodiments, the component 824 for switching to the RF calibration path via a coarse phase shifter may be configured to perform method 700. Figure 7A One or more of the functions described in operation block 724. In an exemplary embodiment, component 824 for switching to the RF calibration path via a coarse phase shifter may include switching the baseband data input on connection 317 to the RF calibration path via coarse phase shifter 320. The amplitudes and relative DC phases of XI and XQ follow the previously derived Tx-Rx phase difference θn on the XI path and θn+π / 2 on the XQ path.

[0126] The apparatus may also include a component 828 for deriving terms of the IQ BB correction function h(t). In some embodiments, the component 828 for deriving terms of the IQ BB correction function h(t) may be configured to perform method 700. Figure 7A One or more of the functions described in operation block 728. In an exemplary embodiment, component 828 for deriving terms of the IQ BB correction function h(t) may include an I / Q amplitude / phase error estimation element 350, which is used to estimate the filter function of the correction function h(t) 370 using an adaptive filter estimation algorithm.

[0127] Specific implementation examples are described in the following numbered clauses: 1. A radio frequency (RF) vestigial sideband (RSB) calibration circuit, the RF (RF) vestigial sideband (RSB) calibration circuit comprising: a complex (in-phase / quadrature (I / Q)) signal receiver; a signal generator configured to generate a transmit (Tx) signal; and a first phase shifter connected to the signal generator, the first phase shifter being configured to selectively process one of an in-phase local oscillator (LO) signal and a quadrature LO signal to create an initial phase between the received in-phase (I) signal and the quadrature (Q) signal, thereby generating a balanced I / Q signal. The signal combiner is configured to combine the received (Rx) signal with the transmitted (Tx) signal to generate a first combined signal; a second phase shifter is configured to provide a selected phase shift to the first combined signal; a complex downconverter is configured to generate in-phase Rx signals and quadrature Rx signals; and an I / Q amplitude / phase error estimation element is configured to determine the amplitude and phase errors of the in-phase Rx signals and the quadrature Rx signals, and is configured to generate a phase compensation signal. The signal includes an amplitude compensation signal (α'); a first multiplier configured to generate a compensated in-phase Rx signal and a second multiplier configured to generate a compensated quadrature Rx signal; a second signal combiner configured to combine the compensated in-phase Rx signal and the compensated quadrature Rx signal to generate a frequency-independent compensation signal; and a filter (h(t)) configured to generate a frequency-dependent compensation signal.

[0128] 2. The RF RSB calibration circuit according to Clause 1, wherein the second phase shifter includes an inductor (L) capacitor (C) circuit, the inductor (L) capacitor (C) circuit including an inductor and an adjustable capacitor.

[0129] 3. The RF RSB calibration circuit according to any one of Clauses 1 to 2, wherein one of the in-phase LO signal and the quadrature LO signal is mixed with the baseband (BB) information signal.

[0130] 4. The RF RSB calibration circuit according to any one of Clauses 1 to 3, wherein the first phase shifter is a coarse adjustment phase shifter.

[0131] 5. The RF RSB calibration circuit according to Clause 4, wherein the phase shift generated by the second phase shifter is smaller than the phase shift provided by the first phase shifter.

[0132] 6. The RF RSB calibration circuit according to any one of Clauses 1 to 5, wherein the signal generator includes an ultra-wideband (UWB) transmitter configured to provide a real (bi-phase) transmission signal.

[0133] 7. The RF RSB calibration circuit according to any one of Clauses 1 to 6, wherein the filter (h(t)) is configured to adapt the frequency-dependent compensation signal to at least one of the compensated in-phase Rx signal and the compensated quadrature Rx signal.

[0134] 8. The RF RSB calibration circuit according to Clause 4, wherein the coarse adjustment phase shifter is configured to adjust the relative phase between the transmitted and received signals to obtain an optimized phase setting when both the in-phase Rx signal and the quadrature Rx signal are at similar amplitudes, the similar amplitude indicating that the relative phase between the transmitted and received signals is approximately 45 degrees.

[0135] 9. The RF RSB calibration circuit according to Clause 6, wherein the biphase transmit signal is alternately combined with the in-phase local oscillator (LO) signal and the quadrature LO signal.

[0136] 10. A method for radio frequency (RF) vestigial sideband (RSB) calibration, the method comprising: enabling a vestigial sideband (RSB) signal generation circuit; adjusting a first phase shifter in the signal generation circuit such that a received (Rx) baseband (BB) in-phase signal and an Rx BB quadrature signal have similar amplitudes; setting a second phase shifter to an initial code (dCT+ / -Δ); selecting a first (in-phase) local oscillator (LO) calibration signal (CAL_LO_I) at the RSB signal generation circuit; and measuring the Rx BB-I output (I) of an in-phase baseband filter (BBF I(t)) in response to the in-phase LO calibration signal (CAL_LO_I). TX I RX=Acos(θ)) and the Rx BB-Q output (I) of the quadrature baseband filter (BBF Q(t)). TX Q RX =αAcos(θ– +90°); a second (quadrature) local oscillator (LO) calibration signal (CAL_LO_Q) is selected at the RSB signal generation circuit; in response to the quadrature LO calibration signal (CAL_LO_Q), the Rx BB-I signal output (Q) of the in-phase baseband filter (BBF I(t)) is measured. TX I RX =Acos(θ+ε-90°)) and the Rx BB-Q signal output (Q) of the orthogonal baseband filter (BBF Q(t)). TX Q RX =αAcos(θ+ε- Derive the Rx I / Q phase error correction factor. c; Derive the Rx I / Q gain ratio αc; Use the Rx I / Q phase error correction factor c and the Rx I / Q gain ratio αc are applied to the Rx BB-I signal and the Rx BB-Q signal to obtain a frequency-independent compensation signal for input to the correction function (h(t)); the second phase shifter is adjusted to obtain an additional set of outputs from the in-phase baseband filter (BBF I(t)) and the quadrature baseband filter (BBF Q(t)); and the adjusted output of the in-phase baseband filter (BBF I(t)) after the second phase shifter is adjusted to obtain the additional set of outputs from the in-phase baseband filter (BBF I(t)) and the quadrature baseband filter (BBF Q(t)) is combined with the output of the correction function (h(t)) to generate a frequency-dependent compensation signal.

[0137] 11. The method according to Clause 10, wherein the Rx I / Q phase error correction factor is derived. c. Deriving the Rx I / Q gain ratio αc includes iteratively solving for the unknowns θ0, θ1ε. , α, A0, A1.

[0138] 12. The method according to any one of Clauses 10 to 11, the method further comprising: alternately mixing the in-phase LO signal and the quadrature LO signal with the baseband (BB) information signal.

[0139] 13. The method according to any one of Clauses 10 to 12, wherein the signal generation circuit provides a real (bi-phase) transmission signal.

[0140] 14. The method according to any one of Clauses 10 to 13, wherein the correction function (h(t)) adapts the frequency-dependent compensation signal to at least one of the compensated in-phase Rx signal and the compensated quadrature Rx signal.

[0141] 15. An apparatus for performing radio frequency (RF) vestigial sideband (RSB) calibration, the apparatus comprising: components for enabling a vestigial sideband (RSB) signal generation circuit; components for adjusting a first phase shifter in the signal generation circuit such that a received (Rx) baseband (BB) in-phase signal and an Rx BB quadrature signal have similar amplitudes; components for setting a second phase shifter to an initial code (dCT+ / -Δ); components for selecting a first (in-phase) local oscillator (LO) calibration signal (CAL_LO_I) at the RSB signal generation circuit; and components for measuring the Rx BB-I output (I) of an in-phase baseband filter (BBF I(t)) in response to the in-phase LO calibration signal (CAL_LO_I). TX I RX =Acos(θ)) and the RxBB-Q output (I) of the quadrature baseband filter (BBF Q(t)). TX Q RX =αAcos(θ– The components include: a component for selecting a second (quadrature) local oscillator (LO) calibration signal (CAL_LO_Q) at the RSB signal generation circuit; and a component for measuring the Rx BB-I output (Q) of the in-phase baseband filter (BBF I(t)) in response to the quadrature LO calibration signal (CAL_LO_Q). TX I RX =Acos(θ+ε-90°)) and the Rx BB-Q output (Q) of the orthogonal baseband filter (BBF Q(t)). TX Q RX =αAcos(θ+ε- The component used to derive the Rx I / Q phase error correction factor. The component c; the component for deriving the Rx I / Q gain ratio αc; the component for the Rx I / Q phase error correction factor The components include: c and the Rx I / Q gain ratio αc applied to the Rx BB-I signal and the Rx BB_Q signal to obtain a frequency-independent compensation signal for input to the correction function (h(t)); components for adjusting the second phase shifter to obtain an additional set of outputs from the in-phase baseband filter (BBF I(t)) and the quadrature baseband filter (BBF Q(t)); and components for combining the adjusted output of the in-phase baseband filter (BBF I(t)) after the second phase shifter is adjusted to obtain the additional set of outputs from the in-phase baseband filter (BBF I(t)) and the quadrature baseband filter (BBF Q(t)) with the output of the correction function (h(t)) to generate a frequency-dependent compensation signal.

[0142] 16. The apparatus according to Clause 15, wherein the Rx I / Q phase error correction factor is derived. c, and the component for deriving the Rx I / Q gain ratio αc includes: iteratively solving for the unknowns θ0, θ1ε, Components of α, A0, and A1.

[0143] 17. The device according to any one of Clauses 15 to 16, the device further comprising: a component for alternately mixing the in-phase LO signal and the quadrature LO signal with the baseband (BB) information signal.

[0144] 18. The device according to any one of Clauses 15 to 17, wherein the signal generation circuit provides a real (bi-phase) transmitted signal.

[0145] 19. The device according to any one of Clauses 15 to 18, wherein the correction function (h(t)) includes components for adapting the frequency-dependent compensation signal to at least one of the compensated in-phase Rx signal and the compensated quadrature Rx signal.

[0146] 20. A system for radio frequency (RF) vestigial sideband (RSB) calibration, the system comprising: a complex (in-phase / quadrature (I / Q)) signal receiver; a signal generator configured to generate a transmit (Tx) signal; and a first phase shifter operatively coupled to the signal generator, the first phase shifter being configured to selectively process one of an in-phase local oscillator (LO) signal and a quadrature LO signal to create an initial phase between the received in-phase (I) signal and the quadrature (Q) signal, from This generates a balanced I / Q signal; a first signal combiner configured to combine the received (Rx) signal with the transmitted (Tx) signal to generate a first combined signal; a second phase shifter configured to provide a selected phase shift to the first combined signal; and a complex downconverter configured to alternately use the in-phase LO signal and the quadrature LO signal to generate in-phase Rx signal and quadrature Rx signal to generate relative Tx-Rx LO phase (θ), amplitude (A), Tx LO I / Q phase mismatch (ε), Rx I / Q amplitude mismatch (α), and Rx I / Q phase mismatch (α) at ​​the output of the complex receiver. One or more signals.

[0147] 21. The system according to Clause 20, wherein the second phase shifter includes an inductor (L) capacitor (C) circuit, the inductor (L) capacitor (C) circuit including an inductor and an adjustable capacitor.

[0148] 22. The system according to any one of Clauses 20 to 21, wherein the in-phase LO signal and the quadrature LO signal are alternately combined with the baseband (BB) information signal.

[0149] 23. The system according to any one of Clauses 20 to 22, wherein the signal generator includes an ultra-wideband (UWB) transmitter configured to provide a real (bi-phase) transmission signal.

[0150] 24. The system according to any one of Clauses 20 to 23, wherein the first phase shifter is a coarse-adjustment phase shifter.

[0151] 25. The system according to Clause 24, wherein the phase shift generated by the second phase shifter is smaller than the phase shift provided by the first phase shifter.

[0152] The circuit architecture described in this article can be implemented on one or more ICs, analog ICs, RFICs, mixed-signal ICs, ASICs, printed circuit boards (PCBs), electronic devices, etc. The circuit architecture described in this article can also be manufactured using various IC process technologies, such as complementary metal-oxide-semiconductor (CMOS), N-channel MOS (NMOS), P-channel MOS (PMOS), bipolar junction transistors (BJTs), bipolar CMOS (BiCMOS), silicon-germanium (SiGe), gallium arsenide (GaAs), heterojunction bipolar transistors (HBTs), high electron mobility transistors (HEMTs), silicon-on-insulator (SOI), etc.

[0153] The apparatus for implementing the circuit described herein may be a standalone device or part of a larger device. The device may be (i) a standalone IC, (ii) a collection of one or more ICs that may include memory ICs for storing data and / or instructions, (iii) an RF IC such as an RF receiver (RFR) or an RF transmitter / receiver (RTR), (iv) an ASIC such as a mobile station modem (MSM), (v) a module that may be embedded in other devices, (vi) a receiver, a cellular phone, a wireless device, a mobile phone or mobile unit, (vii) and so on.

[0154] While selected aspects have been illustrated and described in detail, it should be understood that various substitutions and modifications may be made therein without departing from the spirit and scope of the invention, as defined in the appended claims.

Claims

1. A radio frequency (RF) vestigial sideband (RSB) calibration circuit, the radio frequency (RF) vestigial sideband (RSB) calibration circuit comprising: Complex (in-phase / quadrature (I / Q)) signal receiver; A signal generator configured to generate a transmit (Tx) signal; A first phase shifter, connected to the signal generator, is configured to selectively process one of an in-phase local oscillator (LO) signal and a quadrature LO signal to create an initial phase between the received in-phase (I) signal and the quadrature (Q) signal, thereby producing a balanced I / Q signal. A first signal combiner is configured to combine a received (Rx) signal with the transmitted (Tx) signal to generate a first combined signal; A second phase shifter is configured to provide a selected phase shift to the first combined signal; A downconverter configured to generate in-phase Rx signals and quadrature Rx signals; I / Q amplitude / phase error estimation element, the I / Q amplitude / phase error estimation element is configured to determine the amplitude and phase errors of the in-phase Rx signal and the quadrature Rx signal, and is configured to generate a phase compensation signal. ') and amplitude compensation signal (α'); A first multiplier configured to generate a compensated in-phase Rx signal and a second multiplier configured to generate a compensated quadrature Rx signal; A second signal combiner is configured to combine the compensated in-phase Rx signal and the compensated quadrature Rx signal to generate a frequency-independent compensated signal; and A filter (h(t)) is configured to generate a frequency-dependent compensation signal.

2. The RF RSB calibration circuit according to claim 1, wherein the second phase shifter includes an inductor (L) capacitor (C) circuit, the inductor (L) capacitor (C) circuit including an inductor and an adjustable capacitor.

3. The RF RSB calibration circuit according to claim 1, wherein one of the in-phase LO signal and the quadrature LO signal is mixed with the baseband (BB) information signal.

4. The RF RSB calibration circuit according to claim 1, wherein the first phase shifter is a coarse adjustment phase shifter.

5. The RF RSB calibration circuit of claim 4, wherein the phase shift generated by the second phase shifter is smaller than the phase shift provided by the first phase shifter.

6. The RF RSB calibration circuit of claim 1, wherein the signal generator comprises an ultra-wideband (UWB) transmitter configured to provide a real (bi-phase) transmission signal.

7. The RF RSB calibration circuit of claim 1, wherein the filter (h(t)) is configured to adapt the frequency-dependent compensation signal to at least one of the compensated in-phase Rx signal and the compensated quadrature Rx signal.

8. The RF RSB calibration circuit of claim 4, wherein the coarse adjustment phase shifter is configured to adjust the relative phase between the transmitted and received signals to obtain an optimized phase setting when both the in-phase Rx signal and the quadrature Rx signal are at similar amplitudes, the similar amplitude indicating that the relative phase between the transmitted and received signals is approximately 45 degrees.

9. The RF RSB calibration circuit of claim 6, wherein the biphase transmit signal is alternately combined with the in-phase local oscillator (LO) signal and the quadrature LO signal.

10. A method for radio frequency (RF) vestigial sideband (RSB) calibration, the method comprising: Enable the residual sideband (RSB) signal generation circuit; Adjust the first phase shifter in the signal generation circuit so that the received (Rx) baseband (BB) in-phase signal and the Rx BB quadrature signal have similar amplitudes; Set the second phase shifter to the initial code (dCT+ / -Δ); Select the first (in-phase) local oscillator (LO) calibration signal (CAL_LO_I) at the RSB signal generation circuit; In response to the in-phase LO calibration signal (CAL_LO_I), the Rx BB-I output (I) of the in-phase baseband filter (BBF I(t)) is measured. TX I RX =Acos(θ)) and the Rx BB-Q output (I) of the quadrature baseband filter (BBF Q(t)). TX Q RX =αAcos(θ– +90°)); Select a second (quadrature) local oscillator (LO) calibration signal (CAL_LO_Q) at the RSB signal generation circuit; In response to the quadrature LO calibration signal (CAL_LO_Q), the Rx BB-I signal output (Q) of the in-phase baseband filter (BBF I(t)) is measured. TX I RX =Acos(θ+ε-90°)) and the RxBB-Q signal output (Q) of the orthogonal baseband filter (BBF Q(t)). TX Q RX =αAcos(θ+ε- )); Derive the Rx I / Q phase error correction factor c); Derive the Rx I / Q gain ratio αc; The Rx I / Q phase error correction factor c and the Rx I / Q gain ratio αc are applied to the Rx BB-I signal and the Rx BB-Q signal to obtain a frequency-independent compensation signal for input to the correction function (h(t)). The second phase shifter is adjusted to obtain an additional set of outputs from the in-phase baseband filter (BBF I(t)) and the quadrature baseband filter (BBF Q(t)); as well as The adjusted output of the in-phase baseband filter (BBF I(t)) after the second phase shifter is adjusted to obtain an additional set of outputs from the in-phase baseband filter (BBF I(t)) and the quadrature baseband filter (BBF Q(t)) is combined with the output of the correction function (h(t)) to generate a frequency-dependent compensation signal.

11. The method of claim 10, wherein the Rx I / Q phase error correction factor is derived. c, and deriving the Rx I / Q gain ratio αc includes: Iteratively solve for the unknowns θ0, θ1, ε. , α, A0, A1.

12. The method according to claim 10, further comprising: The in-phase LO signal and the quadrature LO signal are alternately mixed with the baseband (BB) information signal.

13. The method of claim 10, wherein the signal generation circuit provides a real (bi-phase) transmitted signal.

14. The method of claim 10, wherein the correction function (h(t)) adapts the frequency-dependent compensation signal to at least one of the compensated in-phase Rx signal and the compensated quadrature Rx signal.

15. An apparatus for performing radio frequency (RF) vestigial sideband (RSB) calibration, the apparatus comprising: Components used to enable the residual sideband (RSB) signal generation circuit; The component used to adjust the first phase shifter in the signal generation circuit so that the received (Rx) baseband (BB) in-phase signal and the RxBB quadrature signal have similar amplitudes; Components used to set the second phase shifter to the initial code (dCT+ / -Δ); Components for selecting the first (in-phase) local oscillator (LO) calibration signal (CAL_LO_I) at the RSB signal generation circuit; Used to measure the RxBB-I output (I) of the in-phase baseband filter (BBF I(t)) in response to the in-phase LO calibration signal (CAL_LO_I). TX I RX =Acos(θ)) and the Rx BB-Q output (I) of the quadrature baseband filter (BBF Q(t)). TX Q RX =αAcos(θ– Components with a +90° angle; Components used to select the second (quadrature) local oscillator (LO) calibration signal (CAL_LO_Q) at the RSB signal generation circuit; The Rx BB-I output (Q) of the in-phase baseband filter (BBF I(t)) is used to measure the quadrature LO calibration signal (CAL_LO_Q). TX I RX =Acos(θ+ε-90°)) and the RxBB-Q output (Q) of the orthogonal baseband filter (BBF Q(t)). TX Q RX =αAcos(θ+ε- )) components; Used to derive the Rx I / Q phase error correction factor c's components; The component used to derive the Rx I / Q gain ratio αc; Used to adjust the Rx I / Q phase error correction factor c and the Rx I / Q gain ratio αc are applied to the Rx BB-I signal and the Rx BB_Q signal to obtain a frequency-independent compensation signal for use as input to the component of the correction function (h(t)); Components for adjusting the second phase shifter to obtain an additional set of outputs from the in-phase baseband filter (BBF I(t)) and the quadrature baseband filter (BBF Q(t)); and A component for combining the adjusted output of the in-phase baseband filter (BBF I(t)) after the second phase shifter is adjusted to obtain an additional set of outputs from the in-phase baseband filter (BBF I(t)) and the quadrature baseband filter (BBF Q(t)) with the output of the correction function (h(t)) to generate a frequency-dependent compensation signal.

16. The device of claim 15, wherein the Rx I / Q phase error correction factor is derived. c, and the components that derive the Rx I / Q gain ratio αc include: Used for iteratively solving for unknowns θ0, θ1, ε, Components of α, A0, and A1.

17. The apparatus of claim 15, further comprising: A component for alternately mixing the in-phase LO signal and the quadrature LO signal with the baseband (BB) information signal.

18. The apparatus of claim 15, wherein the signal generation circuit provides a real (bi-phase) transmitted signal.

19. The device of claim 15, wherein the correction function (h(t)) includes components for adapting the frequency-dependent compensation signal to at least one of the compensated in-phase Rx signal and the compensated quadrature Rx signal.

20. A system for radio frequency (RF) vestigial sideband (RSB) calibration, the system comprising: Complex (in-phase / quadrature (I / Q)) signal receiver; A signal generator configured to generate a transmit (Tx) signal; A first phase shifter, operatively coupled to the signal generator, is configured to selectively process one of an in-phase local oscillator (LO) signal and a quadrature LO signal to create an initial phase between the received in-phase (I) signal and the quadrature (Q) signal, thereby producing a balanced I / Q signal. A first signal combiner is configured to combine a received (Rx) signal with the transmitted (Tx) signal to generate a first combined signal; A second phase shifter is configured to provide a selected phase shift to the first combined signal; and A complex downconverter configured to alternately use the in-phase LO signal and the quadrature LO signal to generate in-phase Rx signals and quadrature Rx signals to generate relative Tx-Rx LO phase (θ), amplitude (A), Tx LO I / Q phase mismatch (ε), Rx I / Q amplitude mismatch (α), and Rx I / Q phase mismatch (α) at ​​the output of the complex receiver. One or more signals.

21. The system of claim 20, wherein the second phase shifter includes an inductor (L) capacitor (C) circuit, the inductor (L) capacitor (C) circuit including an inductor and an adjustable capacitor.

22. The system of claim 20, wherein the in-phase LO signal and the quadrature LO signal are alternately combined with the baseband (BB) information signal.

23. The system of claim 20, wherein the signal generator includes an ultra-wideband (UWB) transmitter configured to provide a real (bi-phase) transmission signal.

24. The system of claim 20, wherein the first phase shifter is a coarse-adjustment phase shifter.

25. The system of claim 24, wherein the phase shift generated by the second phase shifter is smaller than the phase shift provided by the first phase shifter.

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