Wireless power transmission system and method

Abstract

A wireless power transfer system, the wireless power transfer system comprising: a transmitter unit, the transmitter unit comprising a first transceiver, the first transceiver coupled to a first induction coil, and the first transceiver configured to drive the first induction coil; and a receiver unit, the receiver unit comprising a second transceiver coupled to a second induction coil, the second induction coil for inductively coupling with the first induction coil; wherein the receiver unit further comprises an injection locked oscillator coupled to the second transceiver for determining an oscillation frequency of the second transceiver, the injection locked oscillator being configured to be synchronized with a drive frequency of the transmitter unit.

Description

[0001] This work was supported by the Engineering and Physical Sciences Committee and ERPSRC approval numbers EP / N509486 / 1, EP / R513052 / 1 and EP / R029504 / 1. Technical Field

[0002] This disclosure relates to wireless power transmission systems and methods thereof. Background Technology

[0003] For example, wireless power transfer (WPT), particularly inductive power transfer (IPT), is becoming increasingly popular as a common technology for powering and charging electronic devices. Currently, wireless power transfer is most commonly used in relatively low-power applications. However, there is also considerable interest in applying inductive wireless power transfer to higher-power inductive power transfer (HP-IPT).

[0004] Active-passive (AP) systems are commonly used in IPT (Inductively Coupled Power Transmission). In an AP system, the receiver is a passive component and can only function as a receiver. Therefore, AP systems are only suitable for unidirectional power transmission. Another disadvantage of AP systems is that reflected reactance can cause system detuning, making it difficult to tune such systems for different operating conditions (e.g., different separation distances, different coil overlaps, different temperatures). For high-power systems, the losses caused by detuning are even more difficult to control.

[0005] Active-to-active (AP) systems are gaining increasing attention in the IPT (Internet Power Transmission) field. In AP systems, both the transmitter and receiver are active devices and can be reconfigured for bidirectional power transfer. AP systems can solve the tuning problems faced by active-passive systems and may be more efficient. However, AP systems are difficult to implement. In particular, obtaining a stable frequency and phase reference between the transmitter and receiver is challenging, which in turn leads to poor synchronization and thus lower power efficiency compared to many active-passive systems. In some systems, a separate communication link can be used to assist synchronization. However, such a separate communication link increases the complexity and cost of the system and is difficult to scale to high-frequency operation, such as megahertz. Summary of the Invention

[0006] The present invention aims to solve the above-mentioned problems.

[0007] In a first aspect, a wireless power transmission system is provided, comprising: a transmitter unit (first unit) including a first transceiver coupled to a first induction coil and configured to drive the first induction coil; and a receiver unit (second unit) including a second transceiver coupled to a second induction coil for inductive coupling with the first induction coil; wherein the receiver unit further includes an injection-locked oscillator coupled to the second transceiver for defining an oscillation frequency of the second transceiver, the injection-locked oscillator being configured to synchronize with a drive frequency of the transmitter unit (e.g., with a frequency at which the first coil is driven). For example, the injection-locked oscillator may be provided for synchronizing the oscillation frequency of the second transceiver with the drive frequency of the transmitter unit.

[0008] Injection-locked oscillators can also be referred to as injection-locked oscillators, for example, oscillators configured to be injection-locked to another frequency. Injection-locked oscillators are a term used in the field of radio frequency communication systems.

[0009] In this document, a transmitter unit is defined as a unit in a wireless power transmission system configured to inductively transmit energy. Equivalently, a receiver unit is defined as a supplementary unit in a wireless power transmission system configured to inductively couple with such a transmitter unit to receive and capture inductively transmitted energy.

[0010] As the reader will understand, since the transmitter unit of the first aspect includes a transceiver, it can alternatively be configured as a receiver (e.g., by switching the first transceiver from transmit mode to receive mode). Similarly, since the receiver unit of the first aspect includes a second transceiver, it can alternatively be configured as a transmitter (e.g., by switching the second transceiver to transmit mode). That is, the transmitter unit of the first aspect can be a transmit / receive unit, wherein the first transceiver is switched to operate as a transmitter; and the receiver unit of the first aspect can also be a transmit / receive unit, wherein the second transceiver is switched to operate as a receiver.

[0011] Bidirectional power transmission can be achieved by using a transceiver-based receiver (e.g., by switching the transmitter to operate as a receiver; and by switching the receiver to operate as a transmitter). Conversely, bidirectional power transmission cannot be achieved if a passive receiver (e.g., a receiver that does not contain a transceiver) is used. Since both the transmitter and receiver units in this disclosure are transceiver-based, the system constitutes an active-active system (rather than an active-passive system). More generally, by definition, an active component can be used as a transceiver (i.e., used as a transmitter or receiver).

[0012] The inventors discovered that, without being bound by theory, using an active-active system (compared to an active-passive system) can improve tuning performance. Similarly, without being bound by theory, this is believed to be because the transceiver uses a transistor-based rather than a diode-based system. In an active-passive system, the load reflected from the receiver to the transmitter is determined by the coupling coefficient, the rectifier circuit topology, and the load itself, and therefore cannot be independently controlled under given system operating conditions, and may contain a reactive component. However, considering that an active-active system includes transistor-controlled circuitry on both sides of the link, the reflected load can be controlled by changing the relative phase of the transistor's drive signals, thereby achieving tuning independent of system operating conditions by minimizing the reflected reactance.

[0013] In this paper, injection locking is defined as the process by which one side of a system locks to the frequency of another side, thereby synchronizing the two sides of the system. When the first component is injection-locked to the second component, it should be understood that the first component adopts the frequency of the second component. The inventors have discovered that an injection-locked oscillator can be used in the active-active system of this disclosure to obtain a stable frequency and phase reference at the receiver (particularly by locking to the drive frequency on the transmitter side). Therefore, synchronization performance is improved compared to prior art active-active systems, especially when operating at the megahertz level.

[0014] By coupling an injection-locked oscillator to a second transceiver, the drive frequency of the transmitter unit sets the oscillation frequency of the system, and the receiver unit locks to that drive frequency. This helps ensure that the drive frequency of the transmitter unit does not deviate from its defined ISM band. Typically, the transmitter will be configured to operate in the 13.56 MHz or 6.78 MHz band. However, those skilled in the art will understand that this disclosure is not limited to these bands, and other bands may also be used.

[0015] Synchronizing the receiver unit with the transmitter unit improves the efficiency of power transmission between them. Furthermore, it increases the system's tolerance to larger separation distances and / or smaller overlaps between the first and second coils. Therefore, both system efficiency and robustness are improved.

[0016] Synchronization between the first and second transceivers is achieved using an injection-locked oscillator on the receiver, eliminating the need for a separate communication link between the transmitter and receiver units. Frequency synchronization is guaranteed, and by extending the relative phase of the currents in the transceiver coils to maintain a fixed 90° angle, power transfer is maximized while reflected reactance is minimized or eliminated. This allows operation even under extremely low coupling conditions, such as large coil spacing or poor coil overlap. Consequently, system robustness is improved.

[0017] It is worth noting that if a receiver unit needs to be used with multiple different transmitter units at different times, and the drive frequency of each transmitter unit may be slightly different, injection locking is required to maximize efficiency when coupled to each transmitter unit. Similarly, the oscillation frequency of a crystal oscillator is also temperature-dependent. Therefore, injection locking is also required to maximize efficiency under temperature fluctuations.

[0018] The transmitter unit may also include an oscillator coupled to the first transceiver, which defines a stable oscillation frequency for the first transceiver. The oscillator of the transmitter unit may be a crystal oscillator with a fixed oscillation frequency. In some examples, the transmitter unit may (also) include an injection-locked oscillator (ILO). The ILO coupled to the second transceiver and the oscillator coupled to the first transceiver may be selected to have inherent oscillation frequencies similar to each other, for example, similar enough to achieve effective injection locking. For example, it may be selected such that the difference between the inherent oscillation frequency of the ILO and the oscillation frequency of the oscillator coupled to the first transceiver does not exceed 10% (e.g., not less than 90% and not more than 110% of the oscillator's oscillation frequency). For example, the difference between the inherent oscillation frequency of the ILO and the oscillation frequency of the oscillator may not exceed 5%, for example, the difference may not exceed 2%. In an exemplary implementation of transmission, the difference between the inherent oscillation frequency of the ILO and the oscillation frequency of the oscillator may be about 1%.

[0019] The injection-locked oscillator can be a voltage-controlled injection-locked oscillator with a voltage-tunable inherent oscillation frequency. Therefore, the injection-locked oscillator can be tuned to approximately match the driving frequency of the transmitting unit, thereby enhancing the injection-locking effect.

[0020] If the transmitter includes a crystal oscillator, the receiver's injection-locked oscillator can be configured to synchronize with the oscillation frequency of the crystal oscillator.

[0021] Injection-locked oscillators may have low Q factors, such as less than 200 or less than 100. In some examples, the Q factor may be below 90, for example, between 80 and 90. Injection-locked oscillators can also be nonlinear oscillators.

[0022] At least one of the first and second transceivers can be an EF-class transceiver. The inventors have found that EF-class transceivers are well-suited for applications with variable coupling coefficients, such as those caused by variations in coil spacing or coil overlap. These transceivers can be load-independent EF-class transceivers. In many cases, such EF-class transceivers have proven to further improve the robustness of the system.

[0023] The first and second transceivers can belong to the same class. In some examples, they can be substantially identical. The second transceiver can be tuned to implement a load independent of the first transceiver's load. In some examples, the two transceivers can be tuned to implement loads independent of each other. Therefore, they can be defined as load-independent transceivers.

[0024] At least one of the first and second coils can be an air coil. That is, at least one of the first and second coils can be free of ferromagnetic core. This helps to achieve unconstrained magnetic flux, thereby improving the efficiency of power transmission over a large range of relative coil positions.

[0025] The system may be a high-power system. For example, the system may be configured to operate at a power of at least 2kW, at least 4kW, and in some examples, at least 10kW or at least 15kW.

[0026] The receiver unit may also include a delay line configured to maintain a 90º phase offset relative to the transmitter unit. This delay line helps ensure maximum power delivery.

[0027] One problem with injection-locked oscillators is their sensitivity to temperature changes. Specifically, the inherent oscillation frequency of the injection-locked oscillator will change in response to temperature variations. To ensure effective injection locking in the system according to this disclosure, it may be necessary to ensure that the frequency difference between the injection-locked oscillator and the oscillator on the transmitting unit does not exceed a predetermined limit (if the frequency difference exceeds the predetermined limit, injection locking may not be possible). Therefore, it may be necessary to adjust the inherent oscillation frequency of the injection-locked oscillator according to the temperature.

[0028] Therefore, the injection-locked oscillator can be a voltage-controlled injection-locked oscillator with a voltage-tunable intrinsic oscillation frequency. This intrinsic oscillation frequency can be tuned by changing the voltage applied to the oscillator.

[0029] The receiver unit may also include a temperature sensor and a control unit, the control unit being configured to determine the temperature at the receiver unit based on the output of the temperature sensor.

[0030] The control unit can be configured to:

[0031] The temperature change at the receiver unit is determined based on the output from the temperature sensor.

[0032] Based on a lookup table, the variation of the intrinsic oscillation frequency of the injection-locked oscillator with temperature is determined;

[0033] Based on the determined change in the inherent oscillation frequency, determine the voltage change required to counteract that change;

[0034] This voltage change is applied to the injection-locked oscillator.

[0035] Therefore, even if the ambient temperature changes, the receiver unit can maintain a stable inherent oscillation frequency.

[0036] Over time, temperature changes can cause a phase shift between the transmitter and receiver units. Other environmental factors, such as electromagnetic interference, can also similarly introduce phase shift. Therefore, a delay line can be used to compensate for this phase shift between the receiver and transmitter units. Specifically, the control unit can be configured to:

[0037] This causes the delay line to introduce a relative phase disturbance into the oscillation signal sent from the injection-locked oscillator to the second transceiver;

[0038] For each phase disturbance, the power output or induced power of the receiver unit is determined based on the output of the second transceiver;

[0039] Based on the power output and the corresponding phase disturbance, determine the relative phase disturbance that maximizes the power output or induced power;

[0040] Control the delay line to keep the power output or induced power under maximum relative phase disturbance.

[0041] Therefore, the system can scan the phase changes on both sides of the current phase change and then select the phase change that maximizes the power output or induced power. Thus, the system can maintain a 90° phase relationship with the coil current to achieve maximum power output or induced power.

[0042] In a second aspect, a method is provided for controlling a receiver unit for an inductive power transmission system, the receiver unit comprising: a transceiver coupled to an induction coil; and a voltage-controlled injection-locked oscillator coupled to the transceiver for determining the oscillation frequency of the transceiver, the method comprising:

[0043] Determine the temperature change at the receiver unit;

[0044] Determine how the inherent oscillation frequency of an injection-locked oscillator changes with temperature;

[0045] Determine the voltage change required to counteract this change; and

[0046] A voltage change is applied to the injection-locked oscillator.

[0047] In a third aspect, a method is provided for controlling a receiver unit for an inductive power transmission system, the receiver unit comprising: a transceiver coupled to an inductive coil; a voltage-controlled injection-locked oscillator coupled to the transceiver for defining the oscillation frequency of the transceiver; and a delay line, the method comprising:

[0048] This causes the delay line to introduce a relative phase disturbance into the oscillation signal sent from the injection-locked oscillator to the transceiver;

[0049] For each phase disturbance, the induced power of the receiver unit is determined based on the output from the transceiver;

[0050] Based on the power output and the corresponding phase disturbance, determine the relative phase disturbance that maximizes the induced power;

[0051] Control the delay line to keep the power output or induced power under maximum relative phase disturbance.

[0052] This document also discloses a computer-readable medium (e.g., a non-transitory computer-readable medium) storing instructions that, when executed by a processor, cause the processor to perform steps according to the second or third aspect.

[0053] This document also discloses a system comprising a computer-readable medium (e.g., a non-transitory computer-readable medium) and a processor, wherein the computer-readable medium stores instructions that, when executed by the processor, cause the processor to perform steps according to the second or third aspect.

[0054] The appendix provides further information relating to the currently disclosed system and its operating principles. Therefore, the information in the appendix is ​​incorporated as part of this disclosure. Attached Figure Description

[0055] The following description of specific embodiments is provided by way of example only, with reference to the accompanying drawings, in which:

[0056] Figure 1 This is a block diagram of a wireless power transmission system according to this disclosure;

[0057] Figure 2 This explains Figure 1 The conceptual circuit diagram of the injection locking principle used in the system;

[0058] Figure 3a is Figure 1 A circuit diagram of an exemplary first transceiver and associated first coil in the system;

[0059] Figure 3b is Figure 1 A circuit diagram of an exemplary second transceiver and associated second coil in the system;

[0060] Figure 4 yes Figure 1 Circuit diagram of an injection-locked oscillator in a system;

[0061] Figure 5 Another inductive power transmission system according to this disclosure is shown;

[0062] Figure 6 It is shown in the example from Figure 1 or Figure 5 A flowchart of the method for forming the transmitter unit;

[0063] Figure 7 It is shown in the example from Figure 1 or Figure 5 A flowchart of the method for forming a receiver unit; and

[0064] Figure 8 It shows Figure 1 A block diagram of an example structure of the control unit used in the system. Detailed Implementation

[0065] This disclosure combines an active-active inductive power transmission (IPT) system with an injection-locked oscillator (ILO) located at an active receiver to achieve the advantages of an active-active system and to enable effective frequency and phase tracking at the receiver by providing the ILO.

[0066] Therefore, this disclosure includes two transceivers, each coupled to its respective coil. Inductive power transfer is possible between the transceivers when the two coils are inductively coupled to each other. The transceiver configured as a receiver is coupled to an injection-locked oscillator (ILO). The ILO sets the oscillation frequency of the transceiver configured as a receiver. Furthermore, the ILO synchronizes the frequency of the receiving transceiver with the frequency of the transmitting transceiver, thereby synchronizing the two transceivers. This synchronization process is called injection locking, and will be referred to below. Figure 2 To provide a more detailed description: Injecting the frequency of the receiver transceiver into the frequency of the transmitter transceiver can improve the efficiency of power transmission and also improve robustness against coil separation or misalignment.

[0067] Using ILO in this manner, the system disclosed herein achieves synchronization without the need for an additional out-of-band communication link. Therefore, complex signal processing tasks are unnecessary, and the device is relatively simple. Furthermore, since synchronization is achieved in an active-active configuration, the system exhibits high misalignment tolerance and can operate under low-coupling conditions and high-dynamic environments. The bidirectional transceiver can be used with wide-bandgap devices (i.e., devices operating at high frequencies) with extended operating frequencies. The bandgap of the wide-bandgap device can be higher than 2 eV. High frequencies can reach the MHz range.

[0068] Figure 1 A block diagram of a wireless power transmission system 100 is depicted, the system having two transceivers 140a and 140b. The wireless power transmission system includes a transmitter unit 110a and a receiver unit 110b. The transmitter unit 110a includes a first transceiver 140a coupled to a first induction coil 120a (also referred to as the primary coil 120a). In this example, the transmitter unit 110a also includes a first control unit 120a and an oscillator 120. The receiver unit 110b includes a second transceiver 140b coupled to a second induction coil 120b (also referred to as the secondary coil 120b) for inductive coupling with the first induction coil 120a. The receiver unit 110b also includes an injection-locked oscillator (ILO) 121. In this example, the receiver unit 110b also includes a second control unit 120b and a temperature sensor 131. The first induction coil 120a is inductively coupled to the second induction coil 120b. Oscillator 120 may be a crystal oscillator and is configured to define the oscillation frequency of the first transceiver 140a. ILO 121 defines the oscillation frequency of the second transceiver 140b and is configured to be injection-locked to the frequency of the first transceiver.

[0069] In the depicted example, the first transceiver 140a is part of the transmitter unit 110a and is therefore switched to a transmit state; while the second transceiver 140b is part of the receiver unit 110b and is therefore switched to a receive state. As the reader will understand, the first transceiver 140a (and therefore the transmitter unit 110a) can be switched to a receive state; and the second transceiver 140b (and therefore the receiver unit 110b) can also be switched to a transmit state. Therefore, the transmitter unit 110a can be referred to as the first unit, and the receiver unit 110b can be referred to as the second unit. For the purposes of this disclosure, a system is described herein in which the first transceiver 140a is configured to a transmit state and the second transceiver 140b is configured to a receive state.

[0070] Oscillator 120 may be a crystal oscillator with an oscillation frequency similar to the inherent oscillation frequency of injection-locked oscillator 121.

[0071] ILO 121 is configured to synchronize with the drive frequency of transmitter unit 110a. The drive frequency of transmitter unit 110a is the inherent oscillation frequency of oscillator 120. Synchronization is likely achieved due to a fixed phase shift in the second induction coil 120a during each oscillation cycle, resulting in a constant frequency offset over time corresponding to the drive frequency of transmitter unit 110a. ILO 121 is considered to be in an injection-locked state when the oscillation frequency of ILO 121 (and therefore the second transceiver 140b) matches the drive frequency of oscillator 120 (and therefore the first transceiver 140a).

[0072] When the second induction coil 120b is induced to couple with the first induction coil 120a, such that the coupling between the transmitter unit 110a and the receiver unit 110b is sufficient to overcome the difference between the inherent oscillation frequency of ILO 121 and the driving frequency of the receiver unit 110b, then ILO 121 will be pulled from its inherent oscillation frequency to the driving frequency of the receiver unit 110b.

[0073] To transmit high power between the first induction coil 120a and the second induction coil 120b, a fixed phase close to +90° is required, so that the induced voltage or induced power in the second induction coil 120b is in phase with the current (i.e., the reflected reactance is zero or minimal). A delay line (such as...) can be used. Figure 5 (As shown) to achieve a fixed phase close to +90°.

[0074] If the input voltage of the first induction coil 120a is forcibly short-circuited, the ILO 121 can still match the frequency of the circulating current in the first induction coil 120a, as long as the transistor in the first transceiver 140a still switches to the desired frequency. This is because there is a small excitation current on the first induction coil 120a, which is used to propagate the injected current at the desired frequency in the injection-locked oscillator 121, thereby stabilizing and synchronizing both sides of the system at the drive frequency. This drive frequency can be called the injection-locked frequency.

[0075] Figure 2 a is a conceptual circuit diagram illustrating the injection locking principle used in this disclosure.

[0076] like Figure 2 As shown, a simple injection-locked oscillator 200 includes an inductor L1 201, a capacitor C1 202, and a resistor R. P 203, transistor Q1, 204, and operational amplifier 205a form feedback loop 205b. A feedback current I exists in the feedback loop of the latch-up oscillator. OSC Furthermore, due to the inductive coupling between the injection-locked oscillator and the oscillator 120 operating at the drive frequency, the injection current I...inj It is pulled out from the feedback loop of the injection-locked oscillator. Injection current I inj The injection frequency is ω inj The injection-locked oscillator 200 pulls the frequency at which the receiver unit 110b operates to the frequency at which the transmitter unit 110a operates.

[0077] When the coupling between the inherent oscillation frequency of receiver unit 110b (i.e., the inherent oscillation frequency of ILO 200) and the inherent oscillation frequency (or drive frequency) of transmitter unit 110a is sufficient to overcome the difference between the first inherent oscillation frequency and the drive frequency, the first inherent oscillation frequency will be pulled to the second inherent oscillation frequency. This frequency change will produce a fixed phase shift between receiver unit 110b and transmitter unit 110a. This phase shift depends on the quality factor (Q factor) of ILO and the frequency difference between the drive frequency and the inherent oscillation frequency of ILO. For example, below Figure 4 The Q factor of the ILO shown is 86.

[0078] The output of the injection-locked oscillator 200 is the injection-locked voltage. Injecting lockout voltage The feedback input is fed to the second transceiver 140b to limit the oscillation frequency of the second transceiver 140b (and thus limit the oscillation frequency of the receiver 110b).

[0079] Figure 3a depicts the circuit diagram of the first transceiver A 300a, which can be used as... Figure 1 The first transceiver is 140a. Transceiver A 300a is an EF class inverter. Specifically, transceiver A 300a is a bidirectional EF class inverter. In Figure 3a, the transceiver load is modeled as a controlled voltage source V. dcA The transceiver 300a can receive wireless power (receive mode) and transmit wireless power (transmit mode). This transceiver can receive and store wireless power, and then transmit it to another receiving device. For example... Figure 3B As shown, the structure of transceiver 300b is the same as that of transceiver 300a. That is to say, it is also an EF class inverter and can be configured for transmit or receive mode.

[0080] Each transceiver 300a, 300b includes a first inductor L 1A L 1B Second inductor L 2A L 2B and the third inductor L 3A L 3B and the first capacitor C 1A C 1B The second capacitor C 2A C2B and the third capacitor C 3A C 3B Each transceiver 300a and 300b also includes a coil, which is represented by resistor R in the circuit diagram. coilA and R coilB This indicates that they respectively correspond to Figure 1 The primary coil 120a and secondary coil 120b are used. Each coil R... coilA and R coilB Each has a current i coilA and i coilB and voltage v PA and v PB Each transceiver 300a and 300b can sense a voltage V. MA and v MB It can be controlled by DC input current i dcA i dcB and input voltage V dcA V dcB Provide DC input power to the transceiver.

[0081] The drain voltage of transceiver A is represented by v. dsA This indicates that the drain voltage of transceiver B is represented by v. dsB This indicates that the source-drain voltage waveform can be measured at the drain of the transistor. For transceiver A, the transistor is represented by Q. 1A Indicated; for transceiver B, the transistor is represented by Q. 1B Indicated. Transistor Q 1A and Q 1B It can be a field-effect transistor. Transistor current is represented by i. dA and i dB This indicates that the gate-source voltage V GSA and V GSB This is the drive signal at the gate drive output. The voltage across the capacitor in the EF branch of transceiver A 300a is represented by V. c2A This indicates that the voltage of the capacitor in the EF branch of transceiver B 300b is represented by v. c2B The drain voltage waveform and the capacitor voltage waveform of the EF branch are examples of switching waveforms.

[0082] In use, transceiver A 300a can be used to wirelessly transmit power to transceiver B 300b. The oscillation frequency of transceiver A can be determined by... Figure 1 The oscillator 120 in the system is set. The oscillation frequency of transceiver B can be determined by... Figure 1 The ILO121 settings in the system. Each oscillator can be connected as shown in the figure, voltage v MA and v MB The results are shown as induction on each of transceivers 300a and 300b.

[0083] EF transceivers can be tuned to be load-independent of each other. US10170940B2 (incorporated hereby by reference only) describes how to achieve this load independence starting from column 7, line 63.

[0084] Figure 4 The circuit diagram of ILO 400 is depicted, which can be used as... Figure 1 The system uses ILO 121. The injection-locked oscillator 400 includes a first inductor L1, a first capacitor C1, a second capacitor C2, a third capacitor C3, a first resistor R1, a second resistor R2, a third resistor R3, a first transistor Q1, a second transistor Q2, a third transistor Q3, a fourth transistor Q4, and a variable capacitor 410a or 410b. The inductance of the first inductor L1 is 1µH, the capacitance of the first capacitor C1 is 1nF, the capacitance of the second capacitor C2 is 1nF, the capacitance of the third capacitor C3 is 42pF, the resistance of the first resistor R1 is 4.3kΩ, the resistance of the second resistor R2 is 100kΩ, and the resistance of the third resistor R3 is 100kΩ.

[0085] The variable capacitor circuit used can be either a varactor diode circuit 410a or a variable capacitor circuit 410b. The variable capacitor circuit can change the inherent oscillation frequency of the ILO 400.

[0086] The varactor diode circuit 410a includes a fourth capacitor C4, a fifth capacitor C5, a sixth capacitor C6, a fourth resistor R4, a first diode D1, and a second diode D2. The capacitance of the fourth capacitor C4 can be 0.1nF, the capacitance of the fifth capacitor C5 can be 0.8nF, the capacitance of the sixth capacitor C6 can be 0.8nF, and the resistance of the fourth resistor R4 can be 10kΩ.

[0087] The varactor diode circuit 410a and the variable capacitor 410b are optional and can be used to control the inherent oscillation frequency of the injection-locked oscillator 400. In practice, the varactor diode circuit 410a functions similarly to the variable capacitor 410b, to control the inherent oscillation frequency of the injection-locked oscillator 400 through the input voltage V. in This controls the resonant frequency of the resonant circuit comprising the first inductor L1 and the third capacitor C3. This makes it easier to inject and lock onto the first unit 110a by bringing the natural frequency of the injection-locked oscillator 400 closer to the natural frequency of the oscillator 120 in the first unit 110a or by generating a controlled VGS phase shift in the second unit 110b.

[0088] The injected signal originates from transmitter unit 110a and is the electromotive force of the voltage superimposed on the first inductor L1. The injected signal and the generated voltage are fed back to the first transistor Q1 through a high-pass filter comprising a second capacitor C2 and a third resistor R3. The first transistor Q1 is a nonlinear transistor, thus generating a sequence of pulsed currents at its collector. Since the sum of the collector currents of the first transistors Q1 and Q2 is constant, a sequence of pulsed currents with opposite signs is applied to the resonant circuit comprising the first inductor L1 and the third capacitor C3.

[0089] The injection-locked oscillator 400 can operate at a frequency of 13.56MHz with a fixed input voltage of 60V. The duty cycle can be fixed at 30%.

[0090] When ILO 400 is implemented as Figure 1 When ILO 130 is in the system, the output of the injection-locked oscillator 400 is the injection-locked voltage. The injected lockout voltage The feedback is sent to receiver unit 110b. This stabilizes the frequency and phase of the second transceiver 140b, which in turn stabilizes the frequency and phase of the second unit 100b, thus achieving a steady-state operating mode.

[0091] In some implementations, a delay module (such as...) is used at the output of the injection-locked oscillator 400. Figure 5 (As shown) to change phase V GS The delay module is designed to accommodate changes in the primary coil current without affecting the inherent frequency of the injection-locked oscillator 400. This delay module can be a DS1023-50 delay module. Optionally, a monostable circuit is placed before the gate drive circuit to ensure a fixed duty cycle of 30%.

[0092] Figure 5 Another example of an IPT system 500 according to this disclosure is shown, the system having a transmitter side 502, which includes a first transceiver 504, a first coil 506, a first monostable circuit 508, and a first crystal oscillator 510. The system 500 also has a receiver side 512, which includes a second transceiver 514, a second coil 516, a delay module 518, a second monostable circuit 520, an ILO 522, and a temperature sensor 524.

[0093] In the transmitting side 502, the oscillation voltage signal from the crystal oscillator 510 passes through the first monostable circuit 508 and is then transmitted to the first transceiver 504 to define the oscillation frequency of the first transceiver 504. This, in turn, drives the first coil 506, causing it to generate a varying current i. coilB .

[0094] A current I is induced in the second coil 516 on the receiving side 512. coilB The induced current is transmitted to the second transceiver 514. The oscillation frequency of the second transceiver 514 is set by ILO 522. The signal from ILO 522 is first transmitted to the delay module 518, then to the monostable circuit 520, and finally to the second transceiver 514 to limit the oscillation frequency of the second transceiver 514.

[0095] like Figure 5 As shown, temperature sensor 524 is used to feed temperature information back to the system, specifically ensuring that ILO 522 and delay unit 518 can adjust in response to any temperature-based changes on the receiving side 512 of system 500. For more details on temperature adjustment, please refer to [link to relevant documentation]. Figure 7 .

[0096] Finally, as well as Figure 5 As shown, a feedback loop exists between the second transceiver 514, the first transceiver 504, ILO 522, and the delay unit 518. It is this feedback loop that enables ILO 522 to be able to respond to the injected current I... inj Injecting current into the drive current I locked to the first coil 506 coilB middle.

[0097] Figure 6 It is shown in Figure 1 A flowchart of method 600 performed on transmitter unit 110a, for example, performed on control unit 130a of transmitter unit 110a. The method includes powering on first transceiver 140a at step 604. Optionally, before or after step 604, the method may include determining at step 602 whether receiver unit 110b is within a predetermined range of transmitter unit 110a, for example, within inductive coupling range.

[0098] Figure 7 It is shown in Figure 1A flowchart of method 700 performed at receiver unit 110b is provided, for example, performed on control unit 130b of receiver unit 110b. Method 700 controls receiver unit 110b for use in inductive power transmission system 100. Receiver unit 110b includes a transceiver 140b coupled to inductive coil 120b, and a voltage-controlled injection-locked oscillator 121 coupled to transceiver 140b for defining the oscillation frequency of transceiver 140a. The method includes energizing a second transceiver 140b at step 704. Optionally, before or after step 704, the method may include determining, at step 702, an induced current in the secondary coil 120b of receiver unit 110b resulting from inductive coupling with the primary coil 120a of transmitter unit 110a. The method includes, at step 706, determining a temperature change at receiver unit 110b based on a signal from temperature sensor 131 or 524. More specifically, the temperature change refers to the temperature change of the oscillator 121 of the receiver unit 110b.

[0099] Temperature is one of the major factors that varies over time, affecting the operation of the system's oscillator, the semiconductor devices used in the oscillator, and even causing changes in the inherent oscillation frequency. Due to the pre-characterized temperature-phase relationship, this can lead to unwanted phase injection and ultimately complete loss of synchronization. Phase injection or phase shift between the primary and secondary coil currents can be attributed to variations in the voltage across the first and second diodes of the varactor diode circuit 410a, as well as the influence of temperature near the injection-locked oscillator. Drastic temperature changes can prevent injection lock-in.

[0100] At step 708, the method includes obtaining an estimated phase shift using a pre-characterized temperature-phase relationship (e.g., from a predefined lookup table). In other words, determining how the inherent oscillation frequency of the injection-locked oscillator changes with temperature. At step 710, the method includes determining the input voltage required to counteract the estimated phase shift. This can also be described as determining the voltage change required to counteract the change. The voltage change required to counteract the estimated phase shift, or the input voltage, can then be applied to the injection-locked oscillator.

[0101] When receiver unit 110b also includes a delay line, at step 712, the method includes causing a phase change in the current in secondary coil 120b relative to the current in primary coil 120a via the delay line connected to receiver unit 110b. These phase changes, which may be referred to as phase disturbances, are introduced into the oscillation signal transmitted from injection-locked oscillator 121 to transceiver 140b.

[0102] For each phase disturbance, the method may include: determining the power output of the second transceiver 140a at the receiver unit 110b or the induced power at the receiver unit 110b based on the output of the transceiver 140b; determining the phase disturbance that maximizes the power output or induced power; and controlling the delay line to maintain the phase disturbance that maximizes the power output or induced power. In other words, at step 714, the method includes: for each introduced phase change, determining the power input at the receiver unit 110b and maintaining the phase change that maximizes the power input.

[0103] Final turn Figure 8 , Figure 8 This is a block diagram showing an example structure of control unit 130a or 130b.

[0104] Computer device 800 includes various data processing resources, such as processor 802 (specifically a hardware processor), which are connected to a central bus structure. Other data processing resources, such as memory 804, are also connected to the bus structure. Display adapter 806 connects display device 808 to the bus structure. One or more user input device adapters 810 connect user input devices 812 (e.g., keyboard and / or mouse) to the bus structure. One or more communication adapters 814 are also connected to the bus structure to provide connectivity with other computer systems 800 and other networks.

[0105] During operation, the processor 802 of the computer system 800 executes a computer program containing computer-executable instructions that can be stored in memory 804. When the computer-executable instructions are executed, they can cause the computer system 800 to perform one or more methods described herein, such as the methods of the second aspect, the methods of the third aspect, etc. Figure 6 Method or Figure 7 The method. The result of the processing performed can be displayed to the user through display adapter 806 and display device 808. User input for controlling the operation of computer system 800 can be received from user input device 812 through user input device adapter 810.

[0106] Obviously, Figure 6 Some functions of the computer system 800 shown may be absent in certain situations. For example, one or more of the computer devices 800 may not require a display adapter 806 or a display device 808. This is the case, for example, if some server-side computer devices 800 are only used for processing data and do not need to display information to the user. Similarly, user input device adapter 810 and user input device 812 may not be required. The simplest form of the computer device 800 includes a processor 802 and memory 804.

[0107] The above detailed description illustrates various exemplary arrangements and methods for controlling the IPT. However, the described arrangements and methods are merely examples, and those skilled in the art will understand that various modifications can be made thereto without departing from the scope of the appended claims.

[0108] More generally, it should be understood that the number of steps shown in the diagram is not intended to be limiting. Steps may be repeated multiple times as needed, and some steps may be omitted.

[0109] The aforementioned computer device can be a local computer or a server.

[0110] Although various specific combinations of components and method steps are described in the text, these are merely examples. Components and method steps can be combined in any suitable arrangement or combination. Certain components or method steps can also be omitted, resulting in any suitable combination of components or method steps.

[0111] The method can be implemented using computer-executable instructions. A computer program product or computer-readable medium may contain or store the computer-executable instructions. The computer program product or computer-readable medium may include hard disk drives, flash memory, read-only memory (ROM), CDs, DVDs, caches, random access memory (RAM), and / or any other storage medium, and the information may be stored for any duration (e.g., long-term, permanent, short-term, temporary buffered, and / or cached information). A computer program may contain the computer-executable instructions. The computer-readable medium may be a physical medium or a non-transitory computer-readable medium. The term "computer-readable" includes "machine-readable."

[0112] In the implementation, the modules, components and other functions described herein can be implemented as discrete components or integrated into the functions of hardware components of ASICs, FPGAs, DSPs or similar devices.

[0113] The singular forms “a” and “an” should not be interpreted as “the only one” but rather as “at least one” or “one or more”, unless otherwise stated. “Comprising” and its derivatives, including “comprises” and “comprise”, include each of the foregoing features, but do not exclude the inclusion of one or more other features.

[0114] The above implementations are merely examples and are illustrative in all respects, not restrictive. It should be understood that various modifications can be made to the described implementations without departing from the scope of this disclosure. Furthermore, many other modifications, though not described, are obviously within the scope of the appended claims.

[0115] The appendix provides further information about the system according to this disclosure and how it works. The information in the appendix relates to the system currently disclosed.

[0116] Synchronous operation of the high-frequency inductive power transmission system is achieved through injection locking.

[0117] Nunzio Pucci, Member, IEEE, Christos Papavassiliou, Senior Member, IEEE, and Paul D. Mitcheson, Senior Member, IEEE

[0118] Abstract—High-frequency inductive power transmission (IPT) systems can be designed to be highly tolerant of misalignment and atmospheric gaps, enabling operation in highly dynamic environments. Most examples in the literature employ a single active transmitter and a single passive receiver (active-passive scheme). Such systems are limited to unidirectional power flow, and the transmitter is prone to detuning due to variations in reflected reactance caused by diode nonlinearity. This also limits the coupling range for efficient system operation. Therefore, by adopting an active-active configuration, the application range of inductive power transmission systems can be greatly expanded. This enables bidirectional power transmission, power routing through multiple nodes, and dynamic retuning to eliminate reflected reactance. One of the biggest challenges in implementing an active secondary side in an IPT system is obtaining a stable frequency and phase reference relative to the transmitter coil current and thus the magnetic field for use by a synchronous rectifier / transceiver. Several synchronization methods have been proposed in the literature, but they either require a separate out-of-band communication link or are difficult to scale to megahertz levels. This paper proposes an alternative approach to existing schemes, namely, using an injection-locked oscillator to achieve optimal phase tracking. In addition, a series of candidate feedback configurations are proposed to improve system robustness. This paper elucidates the basic principle of injection locking and its application in synchronous IPT transceivers, and presents experimental results demonstrating its application in a bidirectional back-to-back Class EF transceiver configuration with an operating frequency of 13.56MHz, a coupling coefficient ranging from 1.9% to 8.4%, and a power consumption of up to 25W.

[0119] Keywords: EF class, resonant power converter, high frequency, wireless power transmission, synchronous rectification, injection lockout

[0120] I. Introduction

[0121] Inductive power transfer (IPT) has been a hot research topic over the past two decades. With the development of wide-bandgap devices, their operating frequencies have expanded to the megahertz range at medium power levels. To address challenges such as efficiency, coil spacing, and misalignment tolerance, increasingly sophisticated design schemes have been proposed, resulting in a growing number of applications that make IPT a viable solution in both the kilohertz and megahertz ranges.

[0122] A. Bidirectional power transmission and power routing

[0123] Although most of the current literature proposes designs that are limited to unidirectional power flow from a single transmitting inverter to a single passive receiving rectifier (active-passive system), the operation of bidirectional systems has become an increasingly important topic. The ability to reverse power flow through synchronous operation at both ends of a wireless link (active-active system) opens up new opportunities for IPT in applications such as vehicle-to-grid [2]-[4] and drone charging sensor networks [5].

[0124] Furthermore, controlling the phase of the current in each transceiver coil (with a fixed frequency reference) can be used to enable collaborative operation of multiple nodes in an IPT system, thereby achieving power routing and the formation of a magnetic field around the system. Such applications involving collaborative and reconfigurable transceiver networks can be beneficial in highly automated environments, such as factories, aerospace, or other applications requiring limited human intervention.

[0125] Furthermore, because active-active systems are self-regulating, they can operate at very low coupling coefficients, making them suitable for applications requiring large air gaps.

[0126] B. Advantages of synchronous rectification under low coupling

[0127] High-frequency inductive power transfer (HF-IPT) systems typically employ air-core coils to achieve unconstrained magnetic flux, making it easier to achieve long-distance, high-efficiency power transfer[6]-[9], and providing greater tolerance to coil misalignment

[10] ,

[11] . Work in

[12] -

[17] shows that such systems can even be designed with greater tolerance to load variations.

[0128] Despite the high efficiency achievable under a wide range of operating conditions, the development of HF-IPT systems still faces the following limitations: Passive rectifiers are a common choice due to their simple structure and considerable efficiency, but their impedance characteristics reflected back to the primary coil are significantly affected by the magnitude of the induced voltage in the secondary coil (as shown in

[18] ,

[19] , due to the nonlinear capacitance of the diode), making system tuning for such a wide range of operating conditions more challenging. Furthermore, low-coupling operation results in additional losses: the induced voltage in the secondary coil decreases as the coupling coefficient decreases. For a fixed power level, a larger current is required, which increases losses when using a passive rectifier due to the constant voltage drop across the diode.

[0129] Synchronous rectification solves these problems: the voltage drop across a transistor during conduction is typically much smaller than the voltage drop across a diode, resulting in lower losses. In synchronous rectifiers, the phase between the primary and secondary windings can also be controlled, allowing for tracking of the zero-reflection reactance state during coupling changes and achieving optimal operation.

[0130] In

[20] , a performance comparison between active-passive and active-active configurations for megahertz IPT systems showed that synchronous rectifiers may have an advantage in end-to-end efficiency, even at frequencies up to 27.12 MHz.

[0131] C. Challenges of Transceiver Synchronization at Megahertz

[0132] One of the main challenges in operating a synchronous or bidirectional HF-IPT system is clock synchronization between the two sides of the system: if there is a frequency mismatch between the primary and secondary sides, it is impossible to operate with a fixed relative phase. To transfer active power between the primary and secondary sides, a fixed phase of approximately ±90° is required to keep the induced voltage and current in the secondary resonator in phase (i.e., without reflected reactance). If a constant phase difference is introduced between the primary and secondary currents due to the frequency mismatch between the two sides, the average power transfer will be zero.

[0133] Synchronization solutions have been proposed for low-frequency [2], [4],

[21] and high-frequency

[12] ,

[22] ,

[23] active systems. However, some of these solutions are difficult to apply to EF-based HF-IPT systems due to the circuit configuration

[21] and / or the instrument bandwidth required for the high-frequency corresponding systems [2], [4]. While the work presented in

[22] requires an additional coil for measuring the current, which is feasible in that frequency range, it introduces additional components into the inverter, potentially altering the resonant link topology due to parasitic effects. In

[12] , the authors proposed a solution that is effective only under specific conditions: the system operates with a fixed on-time to achieve zero-voltage switching, but the off-time varies. This means that the system is effectively performing dynamic frequency tuning with a variable relative duty cycle. This can sometimes lead to instability. In

[23] , the authors used an auxiliary communication link.

[0134] To address some of the aforementioned challenges, this paper proposes a possible alternative to resolve some of the highlighted difficulties: this approach eliminates the need for a separate communication link, thus avoiding complex signal processing tasks, and allows operation under extremely low coupling conditions. This is achieved using a lock-in injection oscillator, enabling the secondary side of the system to naturally converge to the operating frequency of the primary side with a fixed relative phase shift.

[0135] The paper also demonstrates how this method can be combined with other techniques to track and correct for the optimal phase and achieve closed-loop control of the system. Experimental results show the system's startup behavior, the effect of temperature on oscillator behavior, and the stable operation of the system under different operating conditions (up to 25.7W).

[0136] This paper is summarized as follows: Part II provides an overview of Class EF transceivers and their operation in bidirectional power transmission. Part III describes the basic principles of injection locking and how it is applied to HF-IPT systems for synchronous rectification. Part IV presents experimental results obtained by operating a bidirectional 13.56MHz IPT system based on a back-to-back Class EF configuration. Part V explains how to implement a closed-loop configuration to improve system reliability under different operating conditions. Part VI summarizes the entire paper.

[0137] II. Implementing an EF transceiver for bidirectional wireless power transmission

[0138] like Figure 1As shown, the EF-class topology with a back-to-back bidirectional configuration is a type of coil driver commonly used in IPT systems operating in the megahertz range. This topology contains only one low-side switch, is easy to drive, and typically operates in open-loop mode at a fixed frequency and duty cycle. The EF-class topology has many similarities to the E-class coil driver, but it adds an additional LC branch, thus providing additional degrees of freedom in the design. This can be used to adjust the drain waveform, reduce its peak voltage (and thus reduce stress on the device), or to achieve desired system characteristics, such as load independence, by relaxing design constraints to use only zero-voltage switching (ZVS)

[12] .

[0139] In the specific context of this work, EF-class load-independent topologies are useful because they introduce additional system fault tolerance in terms of load variations, which is beneficial when coupling changes or phase search algorithms are executed, thereby ensuring the transceiver operates safely under various load scenarios.

[0140] As shown in

[12] ,

[24] ,

[25] , this topology can be used as either an inverter or a rectifier, the only difference being the relative phase between the currents in the transmitting and receiving coils: keeping this phase within ±90° ensures that each side of the system can operate as a transceiver, exchanging power without reflected reactance.

[0141] This work focuses on transceiver A and transceiver B ( Figure 1 The component values ​​used are reported in Table I. Although the specific component values ​​vary slightly due to differences in component length, both are tuned to achieve load independence. Detailed information on how to select components and system parameters (such as duty cycle and input voltage) for the EF class load-independent topology is reported in

[12] . This tuning scheme was specifically chosen because it can generate a nearly constant coil current over a wide load range.

[0142] Transceiver A

[0143]

[0144] Transceiver B

[0145]

[0146] Figure 1 A circuit diagram of two bidirectional Class EF transceivers with loads modeled as voltage sources.

[0147] Table I

[0148] Component values ​​for transceiver A and transceiver B.

[0149] EF class transceiver, Vdc=60V, δ=30%, with planar PCB coils on both sides from

[26] .

[0150]

[0151] In this study, the transceiver operated at a frequency of 13.56 MHz to utilize the improved quality factor of air-core coils in the megahertz band, thereby enabling unrestricted magnetic flux. This allows for efficient operation over greater distances and greater tolerance to misalignment. These coils can also be used in other megahertz ISM bands (e.g., 6.78 MHz).

[0152] III. Basic Principles of Injection Locking

[0153] Injection locking is a phenomenon that has been studied since 1946

[27] -

[31] . The core idea of ​​injection locking is that, under certain specific conditions, if there is coupling between two oscillators and the inherent oscillation frequencies of the oscillators are relatively close (i.e., operating within the locking range according to Equation 5), the frequencies of the two independent oscillators can be synchronized.

[0154] If the coupling between the oscillator with a natural oscillation frequency of ω0 (from the slave device side) and the oscillator with a natural oscillation frequency of ω1 (from the master device side) is sufficient to overcome the difference Then the frequency of the oscillator can be pulled from ω0 to ω1.

[0155] This introduces an inherent phase shift, which depends on the quality factor and frequency difference, as explained and outlined in

[29] . Figure 2 In the middle, of which I osc It is the current I in the feedback loop of the oscillator being injected into the lockout. inj It is the current pulled from the feedback loop of the oscillator due to coupling with the system operating at frequency ω1.

[0156]

[0157] (a) Phase shift of the oscillator being pulled

[0158]

[0159] (b) Conceptual circuit diagram of the oscillator during injection

[0160] Figure 2 Conceptual diagram of an injection-locked oscillator

[29] .

[0161] As shown in

[29] , the lock-in range ω of the oscillator can be obtained. L .like Figure 2When the second-order resonant circuit shown oscillates at frequency ω1, it will produce a phase shift α, with frequency ω1 near the resonant frequency ω0.

[0162]

[0163] It can be approximated as And rewritten as and :

[0164]

[0165] When the oscillator is injected with current I inj When the influence of I inj and I osc Presenting angle , among which is isI inj with I T The angle between them isI T with I osc The angle between them. When ω inj When (or ω1) deviates from ω0, the phase shift introduced by the resonant circuit will change with the angle. It increases with the increase of . This means that it will cause I to increase. osc Rotate counterclockwise. This can be written as:

[0166]

[0167] To find the locked range, this expression needs to be maximized, thus... At that time, This along a 90° angle at I inj and I T The distance is translated between them. Therefore, it can be written as... .

[0168] Use this information in conjunction with Equation 2 (set α= It can be written as:

[0169] .

[0170] When approaching the edge of the locked range (i.e., When ), it is approximately:

[0171]

[0172] More details about this derivation are provided in

[29] .

[0173] This characteristic is particularly useful in the context of synchronous rectification, where a switching signal orthogonal to the master signal needs to be generated. Therefore, changing the inherent oscillation frequency of such an oscillator can achieve two purposes: facilitating injection lock-in over a wider range, or providing a predetermined phase offset. As will be discussed in section V, the method of controlling the phase offset (by changing ω0 or introducing an external delay) depends on the specific circumstances.

[0174] From (5), we can see that in order to maximize ω L It must have a large value I inj This is especially true when attempting to pull high-quality factor oscillators. As reported in

[29] , the nonlinearity of the oscillator is another key factor, as ideal linear oscillators with high-quality factors cannot be pulled by injection lock.

[0175] Synchronization failure was monitored using an experimental setup when the distance between the primary and secondary coils exceeded 35 cm (approximately two coil diameters). The system entered a quasi-locked state and exhibited characteristic phase slippage with regular intervals of approximately half a second. For distances greater than 40 cm, frequency lock was completely lost, indicating that the system was operating far beyond the set locking range ω. L In

[29] Figure 7 The report presents an example of typical phase slip behavior caused by quasi-locking.

[0176] We observed that even when the input voltage of the two transceivers was reduced to around 10V, the corresponding coil current was about 500mA, and injection lock-in occurred, but there was no effective power exchange (i.e. the loss was higher than the transmitted power, but the coil current was synchronized).

[0177] The lock-in range depends on the magnitude of the injected current (see Equation 5), and therefore also on the distance between the two sides of the system. For a coil spacing of 25 cm, the lock-in range of the oscillator is estimated to be approximately 60 kHz. Using the attenuation method, the quality factor of the oscillator in Figure 3 is estimated to be approximately 86, corresponding to a current I... osc It is 1mA.

[0178]

[0179] (a) Complete schematic diagram of an injection-locked oscillator. Optional dashed elements are used to control the resonant frequency at a given input voltage.

[0180]

[0181] (b) is the resonant circuit of (a) that includes a varactor diode as an equivalent variable capacitor.

[0182] Figure 3. Circuit diagram of the oscillator.

[0183] One problem arising from injection-locked systems in HF-IPT is the possibility of simply synchronizing independent crystals using the same principle. While theoretically feasible, this task demonstrates significant practical challenges: the high quality factor of the crystals and their relatively small package size (minimizing coupling with the injection current) make it difficult to simply synchronize the two sides of the system without using an oscillator specifically designed for this task. In this study, we attempted to implement this task using experimental setups from a portion of IV. Even placing the master-side coil directly above the slave-side crystal, our attempts with a separate SG-210 STF CMOS oscillator were unsuccessful.

[0184] IV. System Design and Experimental Results

[0185] The system (such as) Figure 4 (As shown) consists of two back-to-back EF class transceivers, such as in

[25] . Figure 1 As shown: The input voltage on each side of this system is provided by a source-drain configuration in which a constant-voltage mode electronic load (drain) and a power supply (source) operate in parallel. This ensures that each side of the system receives the same input voltage and enables bidirectional operation.

[0186]

[0187] Figure 4 Experimental setup.

[0188] The link consists of two PCB planar coils (two turns, 20 cm outer diameter) with an inductance of 1.18 µH and a quality factor greater than 500 at 13.56 MHz. Further details regarding the design and characteristics of the coils have been reported in

[26] and

[32] . These coils were chosen to reproduce the experiment and precisely control the distance between them. In this work, the change in coupling coefficient was achieved solely by altering the distance (z-direction) between the two coils, but in principle, the same result could be reproduced by selecting appropriate offsets in the x and y directions.

[0189] Table I summarizes the components of the two transceivers. One of the two sides of the system (the master device) uses a crystal oscillator to generate V. GS The other side (the slave device) uses the oscillator shown in Figure 3, matched to the master device's frequency via injection lock-in (as explained in Section III). It operates similarly to a pMOS differential LC oscillator, the only difference being that one side of the resonant circuit is grounded. The optional circuitry presented in Figure 3 can be used to control the inherent oscillation frequency of the injection-locked oscillator. In practice, the varactor diode acts as a variable capacitor to control the input voltage V. in To control the resonant frequency of the resonant circuit.

[0190] This makes it easier to make the oscillator's inherent frequency closer to the frequency of the crystal on the host device, or at V GS A controlled phase offset is generated on the slave device to inject lock onto the master device.

[0191] If the latter fails to achieve injection locking due to the device being pushed into a quasi-locked state, or if the system operates without optional circuitry, a delay module can be used at the oscillator output to change V. GS The phase of the current relative to the transmitting coil current is maintained without affecting the oscillator's natural frequency. This has been experimentally verified using a DS1023-50 delay module. The stage preceding the gate drive circuit is always a monostable circuit to ensure a fixed duty cycle of 30%.

[0192]

[0193] Figure 5 Coil current at system startup: blue for the master device and orange for the slave device (operated by an injection-locked oscillator). Three figures show details at different points in time.

[0194] The system is designed to allow for the selection of appropriate parameters. and This is used to set the resonant frequency of the resonant circuit. This combination is set to ensure that the resonant frequency varies from 13.5MHz to 13.6MHz, while the varactor diode D... 1,2 V caused by the change in capacitance in The voltage changes from 0V to 5V accordingly.

[0195] The circuit in Figure 3 operates as follows: the injected signal is the additional electromotive force of the voltage applied to L1. These two signals are fed back to Q1 through a high-pass filter C2-R3. Q1 is nonlinear, thus generating a pulse current sequence at the collector of Q1. Since the sum of the collector currents of Q1 and Q2 is constant, a pulse sequence with opposite signs is applied to the resonant circuit L1-C3. A detailed description of the pull-in and lock-out mechanism is given in

[31] .

[0196] The system operates at a frequency of 13.56MHz, with a fixed input voltage of 60V on both sides and a fixed duty cycle of 30%.

[0197] When both sides of the system are turned on simultaneously, the master device begins to generate a coil current at the same frequency as its crystal oscillator. After several cycles, the magnetic field generated by this current will be strong enough to pull the oscillator circuit on the receiver side to the frequency of the master device, thereby stabilizing its frequency and phase to reach a steady state. For ease of understanding, it is assumed that both sides have enough energy to start the system. In reality, it is irrelevant which side of the system turns on first, or whether the master device is located at the transmitter or receiver.

[0198] Figure 5 This process is summarized in the diagram, which shows detailed information about the coil currents on each side of the system, with the master coil in blue and the slave coil in orange. In this specific case, the system is set to zero phase offset to facilitate observation of successful phase locking; however, in practice, it operates with a phase offset of ±90° between the primary and secondary coil currents. Frequency and phase locking are achieved after a transient period of less than 30µs.

[0199]

[0200] Figure 6 Oscilloscope waveforms of the system under injection lockout. The top shows the coil current waveform, and the bottom shows the drain voltage waveform.

[0201] Although no dedicated coil was used in this work to synchronize the oscillator on the device side, this component could theoretically be added to simplify the injection locking process. In this work, the unconstrained magnetic field generated by the link produces the required injection current on the oscillator board without the need for additional components.

[0202] Figure 6 The oscilloscope waveforms (scope capture) of the system operating at the desired phase offset of ±90° are shown to achieve optimal power transfer efficiency. The coil current (top waveform) indicates that the desired phase offset has been achieved. The reported drain voltage waveform shown at the bottom indicates that soft switching has been implemented on both the transmitter and receiver sides.

[0203] As reported in

[33] , even with a passive rectifier, the injection-locked-in phenomenon can be utilized to match the frequency of the primary coil with the resonant frequency of the secondary coil, thereby achieving optimal power transfer efficiency. Similarly, in the proposed system, when the input voltage on the master side is forcibly short-circuited or open-circuited, as long as the transistors on the master side continue to switch at the desired frequency, the injection-locked-in oscillator on the slave side can still match the frequency of the current in the primary coil: if there is a small excitation current on the master side, the injection current will propagate at the correct frequency in the oscillator on the slave side, thereby stabilizing both sides of the system at a matched frequency.

[0204] When the system is running in the absence of a master device nearby, the oscillator will oscillate at its natural frequency, and the transceiver will still function normally because the oscillator's natural frequency is designed to be relatively close to the operating system frequency.

[0205] Table II shows the input and output power under different operating conditions. Power can be reliably transmitted even at coupling as low as 1.9%; synchronization performance is only lost when the coupling is below 0.9% (35cm spacing). At a coupling of 4.5%, 13.7W of power can be transmitted, with an end-to-end efficiency of 53.7%. These values ​​tend to be higher for higher coupling and power levels: the standby power consumption achieved in this design is approximately 4W per side. Efficiency was measured by monitoring the power at both ends of the system using a Yologawa WT332E digital power meter.

[0206] Table II

[0207] System operation for different coil spacing

[0208]

[0209] To further verify the applicability of the proposed method under difficult conditions, we specifically studied low-efficiency / loose-coupling scenarios.

[0210] Figure 8 The loss breakdown is presented for a 4.5% coupling scenario.

[0211] In any given external environment, the key to ensuring the reliable operation of such a system lies in designing the feedback loop. Although the previous section demonstrated the system's operation in open-loop mode, many factors can affect the system's operation, leading to poor system performance or even system failure.

[0212] Temperature is one of the main factors that changes over time and affects system operation: such as Figure 7 As shown, the operating temperature of the oscillator module can vary by up to 10°C in a controlled environment (and even more in field deployments). This can affect the operation of the semiconductor devices used in the oscillator and even cause changes in the oscillation's natural frequency (and the lockout range calculated according to Equation 5). The consequences could include unexpected phase injection into the complete loss of synchronization range: as explained in Part III, the phase changes as the oscillator's natural frequency deviates from that of the master device. Furthermore, the required injection current amplitude also increases.

[0213]

[0214] (a) Temperature of the oscillator module before operation

[0215]

[0216] (b) Temperature of the oscillator module near steady state

[0217] Figure 7 The system thermal imager photographs focus on the injection-locked oscillator module.

[0218]

[0219] Figure 8 Loss distribution at a coupling coefficient of 4.5%.

[0220]

[0221] Figure 9. Block diagram of the system feedback loop.

[0222] As shown in Figure 10, the temperature rose by nearly 10°C over a 10-minute time span, while the phase dropped by 30°. This is something to be avoided in operating systems. A closed-loop feedback system can mitigate this problem.

[0223]

[0224] Figure 10. Temperature dependence of the relative phase of coil current over time.

[0225] V. Closed-loop operation

[0226] The problem can be addressed by implementing the method described in

[34] , which includes techniques for estimating the reflected impedance in a Class EF system: providing a reference voltage level to monitor the reflected reactance and then adjusting the relative phase offset accordingly. However, a practical problem needs to be considered: the method is based on information extraction at a single frequency, but the system presented herein contains an oscillator whose frequency may change during feedback loop adjustments or when synchronization is lost.

[0227] When designing the same system, using a wider bandwidth filter can solve some of the problems, but the required filter transfer function must be very flat across the entire operating frequency range. Experimental observations show that this method is effective when the system is already in a steady state (and therefore the frequency remains constant); however, if the system is unstable, the reflection impedance estimation technique cannot be used to lock the oscillator: because the filter response is not constant across the entire operating range, two slightly different operating frequencies will result in the same reference voltage value for the reflection impedance estimation.

[0228] A simpler alternative is to control the phase (or inherent oscillation frequency) of the slave oscillator by directly measuring factors that may affect injection lock-in. While temperature is perhaps one of the easiest factors to observe, others could include ambient EMI, humidity, light, and radiation. In this work, the feedback loop is based solely on temperature, but in principle, suitable sensors can be used to monitor other of the aforementioned factors. As shown in Figure 9, this correction can be performed using a lookup table to adjust the oscillation frequency or phase as needed.

[0229] As can be seen from Part III, changing the oscillator's inherent oscillation frequency not only affects the phase in which the injection lock occurs, but also alters the lock range (as reported in (5)). This means that, in some cases, if all corrections are made through the input voltage V of the VCO in Figure 3... in If this is done, the locking range may be reduced to a critical value, thereby preventing the injection of locks.

[0230] Therefore, the only thing that needs to be done through V in The adjustments made are to compensate for inherent frequency variations in the oscillator (e.g., temperature variations). Other corrections that only account for phase shifts rather than frequency variations (e.g., system adjustments or variations in reflected reactance from the coupled resonant circuit) should be made via the delay module so that the optimized locking range remains unaffected.

[0231] This allows the system to operate at relatively stable ω0 and ω L It operates at a lower value, thus providing better suppression of interference from external sources, thereby improving the system's resilience.

[0232] The feedback, designed to provide the necessary adjustment to ω0 in response to temperature changes, is characterized by: Figure 3 showing the temperature change near the custom oscillator using a PT 100 temperature sensor and the voltage across the varactor diode (V). in The phase shift between coil currents caused by changes in V. Combining the relationships derived from these two experiments (as shown in Figure 11), a lookup table can be established to adjust the phase shift by changing V. in The value is used to correct the phase shift caused by temperature changes, and thus correct the phase between ω0 and the coil current.

[0233] An additional step to improve the robustness of the system is an additional feedback loop for phase correction via a delay module in response to potential system detuning. This can be achieved using a maximum power point (MPP) tracking algorithm to maximize the power received by the system from the slave device. Since the input voltage of the transceiver is constant in this design, the received power is estimated by the received current, which is measured by a current sensor. When the reflected reactance on the receiving side is minimum and the received power is maximum, the phase between the coil currents will converge to a value close to -90°, as described in

[23] . Bidirectional power transfer can be achieved simply by introducing a phase shift of π using a delay module.

[0234]

[0235] Figure 11. Phase between coil currents and raw temperature readings from PT100 and raw input / output V of ADC in The relationship.

[0236] The synchronization process is summarized as follows:

[0237] 1) Read the temperature to obtain an estimated phase shift value caused solely by temperature change (i.e., the phase difference due to the temperature-phase relationship pre-characterized in Figure 13).

[0238] 2) The VCO input voltage is varied according to the pre-characterized DAC input phase relationship to counteract the inherent frequency shift of the oscillator that might be caused by the phase difference estimated based on the temperature reading. These two steps can be combined using a lookup table to directly convert the temperature reading into the microprocessor-to-DAC output: for example, if the temperature reading indicates that the estimated phase is 30° lower than expected, the DAC will generate a corresponding VCO voltage value to introduce a reverse 30° phase shift. Therefore, the DAC is initially set so that its values ​​have sufficient head and legroom to produce the desired correction. This is achieved using the data in Figure 13 and interpolating the intermediate points.

[0239] 3) Start by introducing a phase change through the delay line and find the point with the maximum received power.

[0240] 4) Achieve convergence.

[0241] 5) Continuously monitor the temperature and periodically correct the input voltage of the VCO to prevent phase injection caused by temperature changes.

[0242] 6) Keep both points adjacent to the current operating point to obtain maximum received power, and adjust the delay line accordingly.

[0243] 7) Repeat steps 5 and 6 to ensure proper phase is maintained.

[0244] 8) (Optional) If the direction of the system power flow needs to be reconfigured, an additional π phase shift can be added.

[0245]

[0246] Figure 12. Oscilloscope waveforms of the system operating under injection lockout and feedback loop conditions. The bottom shows the coil current waveform, and the top shows the drain voltage waveform.

[0247]

[0248] Figure 13. Phase and power of the receiver coil current in each iteration of the MPP tracking algorithm using an exhaustive search between -30° and 180° phase.

[0249]

[0250] Figure 14. Improved MPP tracking algorithm based on gradient change.

[0251] Even under these conditions, the transmitter may still operate in a suboptimal state due to potential tuning mismatches on both sides: as shown in Figure 12, the transmitter can still operate with non-zero reflected reactance. In this experiment, the system was intentionally pulled above its designed power level. Figure 12 actually shows that while the receiver's drain voltage waveform exhibits soft-switching characteristics as expected, the transmitter exhibits hard-switching characteristics, absorbing most of the end-to-end system losses, thus minimizing losses on the receiver side.

[0252] Losses can be balanced if the system is designed to optimize maximum efficiency rather than simply using information from the receiver for MPP tracking. However, this requires additional hardware and communication links to acquire information from the transmitter, and in this work, the feedback loop is implemented entirely based on measurements from the receiver.

[0253] This work is similar to the algorithm in

[23] , employing the MPP tracking algorithm. First, a quadrant check is performed, testing four points spaced 90° apart, and then the search range is narrowed down to a specific quadrant. After that, the algorithm performs a fine search to track the optimal phase, which corresponds to the maximum received power (negative power indicates received power). The operation of the algorithm is summarized in Figure 13.

[0254] This version of the algorithm has an average runtime of 5.6 seconds because each point is measured multiple times and the average is taken. However, execution time can be problematic, especially under high power levels and poor system conditions: the transmitter or receiver may overheat and be damaged.

[0255] To address this issue, the algorithm was modified to skip intermediate points during the fine-grained search and test the two adjacent points of the last found optimum. Another modification is that the fine-grained search is interrupted when a positive gradient above a certain threshold is detected during measurement. These modifications are illustrated in Figure 14. These changes, along with a reduction in the number of measurements per test point, reduced the execution time to 1.6 seconds (20 steps, averaging 80ms per step). The first step is a quadrant search, testing four distinct points spaced 90° apart. Then, two points near the current optimum are tested, followed by a coarse search with a step size of 4°. This step aims to find explicit changes in the gradient, corresponding to local minima. Two points near the candidate local minima are then tested to ensure convergence to the optimum.

[0256] The average phase error obtained using the first method is 2°, while the average error obtained using the second method is 5°. This can be attributed to the reduced measurement time per point and the fact that the second method skips intermediate points in the fine-grained search: if samples were not skipped, the values ​​of adjacent samples would likely have slightly similar values. This means that the impact of noise in a particular sample on the final convergence value is minimized because adjacent samples play an additional averaging role.

[0257] To address this issue, post-convergence corrections are performed at fixed intervals of 5 seconds after convergence. In addition to reducing the error to 2°, these post-convergence corrections help ensure that the system can still operate under MPP even when conditions change, such as coupling or the introduction of foreign objects.

[0258] Another advantage of using a faster algorithm is that slow temperature changes near the oscillator circuit have a negligible effect on convergence. If the algorithm is extremely slow, MPP tracking can be affected by changes in ambient temperature, which can cause variations in ω0, resulting in a phase shift.

[0259] This paper proposes a solution that requires no separate communication link, no complex signal processing, and allows operation under extremely low coupling conditions. This auxiliary circuit requires no high-performance instrumentation, thus it can be integrated into the system as a low-cost active-active operating solution.

[0260] The closed-loop system was tested under different coupling configurations, achieving a maximum power transfer of 25.7W at a coupling degree of 8.4% and an end-to-end efficiency of 60.9%.

[0261] VI. Conclusion

[0262] This work demonstrates a technique for achieving frequency and phase synchronization in an HF-IPT system, where both sides are active, thus enabling synchronization and bidirectional operation.

[0263] This paper discusses the basic principles of injection locking and demonstrates how it can be applied to a bidirectional HF-IPT system operating at 13.56 MHz with coupling between 1.2% and 8.4%. Experimental results show the power exchanged during each coupling.

[0264] This paper proposes the possibility of closed-loop operation and presents experimental results of the closed-loop design to account for temperature variations and the MPP tracking method, thereby fine-tuning the optimal phase.

Claims

1. A wireless power transmission system, the wireless power transmission system comprising: A transmitter unit, the transmitter unit including a first transceiver coupled to a first induction coil, and the first transceiver configured to drive the first induction coil; as well as The receiver unit includes a second transceiver coupled to a second induction coil, the second induction coil being used for inductive coupling with the first induction coil; The receiver unit further includes an injection-locked oscillator coupled to the second transceiver to define the oscillation frequency of the second transceiver, the injection-locked oscillator being configured to synchronize with the drive frequency of the transmitter unit.

2. The wireless power transfer system of claim 1, wherein, The transmitter unit further includes an oscillator coupled to the first transceiver, the oscillator defining a stable oscillation frequency for the first transceiver.

3. The wireless power transfer system of claim 1 or 2, wherein, The injection-locked oscillator has a low Q factor.

4. A wireless power transfer system according to any preceding claim, wherein, The injection-locked oscillator is a nonlinear oscillator.

5. A wireless power transfer system according to any preceding claim, wherein, At least one of the first transceiver and the second transceiver is an EF class transceiver.

6. A wireless power transfer system according to any preceding claim, wherein, At least one of the first induction coil and the second induction coil includes an air coil.

7. A wireless power transfer system according to any preceding claim, wherein, The first transceiver and the second transceiver belong to the same category.

8. A wireless power transfer system according to any preceding claim, wherein, The receiver unit also includes a delay line configured to maintain a 90º phase offset relative to the transmitter unit.

9. A wireless power transfer system according to any preceding claim, wherein, The second transceiver is tuned to implement a load independent of the first transceiver.

10. A wireless power transfer system according to any preceding claim, wherein, The injection-locked oscillator is a voltage-controlled injection-locked oscillator, which has a voltage-tunable inherent oscillation frequency.

11. A wireless power transfer system according to any preceding claim, wherein, The receiver unit also includes a temperature sensor and a control unit, the control unit being configured to determine the temperature at the receiver unit based on the output from the temperature sensor.

12. A wireless power transfer system as claimed in claim 11 when dependent on claim 10, wherein, The control unit is configured to: The temperature change at the receiver unit is determined based on the output from the temperature sensor. Based on a lookup table, the change in the inherent oscillation frequency of the injection-locked oscillator with temperature is determined; Based on the determined change in the inherent oscillation frequency, the voltage change required to counteract the change is determined; The voltage change is applied to the injection-locked oscillator.

13. A method of controlling a receiver unit for an inductive power transfer system, the receiver unit comprising: A transceiver coupled to an induction coil, and a voltage-controlled injection-locked oscillator coupled to the transceiver for defining the oscillation frequency of a second transceiver, the method comprising: Determine the temperature change at the receiver unit; Determine how the inherent oscillation frequency of the injection-locked oscillator changes with temperature; Determine the voltage change required to counteract the change; and The voltage change is applied to the injection-locked oscillator.

14. The wireless power transfer system of claim 11, wherein, The control unit is configured to: The delay line introduces a phase disturbance into the oscillation signal sent from the injection-locked oscillator to the second transceiver; For each phase disturbance, the induced power at the receiver unit is determined based on the output of the second transceiver; Determine the phase perturbation that maximizes the induced power; as well as The delay line is controlled to maintain the phase disturbance that maximizes the induced power.

15. A method of controlling a receiver unit for an inductive power transfer system, the receiver unit comprising: A transceiver coupled to an induction coil; A voltage-controlled injection-locked oscillator coupled to the transceiver is used to limit the oscillation frequency of the transceiver; and a delay line, the method comprising: The delay line introduces a phase disturbance into the oscillation signal sent from the injection-locked oscillator to the transceiver; For each phase disturbance, the induced power at the receiver unit is determined based on the output from the transceiver; Determine the phase perturbation that maximizes the induced power; The delay line is controlled to maintain the phase disturbance that maximizes the induced power.

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