Dual-mode millimeter wave radar channel amplitude-phase distortion compensation method and device and storage medium

Abstract

The invention discloses a dual-mode millimeter-wave radar channel amplitude-phase distortion compensation method and device and a storage medium, and the method comprises the steps: obtaining a transmitting-receiving control parameter, and carrying out the analysis of the transmitting-receiving control parameter, and obtaining a transmitting waveform parameter and a local oscillator waveform parameter; starting from the low-frequency end of the working bandwidth of the system, adjusting a transmitting waveform parameter and a local oscillator waveform parameter, and generating paired single-tone correction signals as a transmitting pre-driving signal and a local oscillator pre-driving signal respectively; generating an analog intermediate frequency signal according to the transmitting pre-driving signal and the local oscillator pre-driving signal, and extracting a distortion amplitude phase value corresponding to the current frequency point from the analog intermediate frequency signal; controlling the frequency of the paired single-tone correction signals to be gradually increased according to a preset step length, and obtaining distortion amplitude and phase values of a plurality of frequency points; performing interpolation fitting on the distortion amplitude-phase value, and calculating an equalizer coefficient; and step frequency is adjusted and calculation is repeated until all equalizer coefficients are obtained for performing dual-mode millimeter wave radar channel amplitude phase distortion compensation.

Description

Technical Field

[0001] This invention relates to the field of radar signal processing technology, specifically to a method, apparatus, and storage medium for compensating for amplitude and phase distortion in a dual-mode millimeter-wave radar channel. Background Technology

[0002] With the widespread application of millimeter-wave radar in automotive electronics, intelligent sensing, and other fields, the requirements for channel consistency and signal fidelity are increasing. Dual-mode radar, such as hybrid step-frequency and linear frequency modulation modes, is favored due to its combination of high resolution and flexible waveform configuration capabilities. However, in multi-channel receiving systems, factors such as device process variations, temperature drift, and frequency response nonlinearity can lead to amplitude and phase inconsistencies between channels, thereby affecting radar imaging quality and target detection accuracy.

[0003] Currently, the channel equalization techniques commonly used in broadband phased array radars mainly include two categories: analog equalization and digital equalization. Analog equalization methods are typically implemented using broadband analog filters, but this increases system size and complexity and makes it difficult to flexibly adapt to multi-channel calibration. Digital equalization methods, on the other hand, are based on digital filters or frequency domain compensation algorithms. While offering better flexibility, they require significant high-speed sampling and real-time processing resources, posing challenges related to cost and synchronization control, especially in large-scale arrays.

[0004] Since analog equalization and digital equalization, both based on fixed parameters, typically estimate and compensate for the channel amplitude and phase responses under a single operating mode and specific conditions, their compensation parameters remain constant throughout system operation. However, in dual-mode radar systems, the signal spectrum structure, excitation method, and effective channel model differ significantly between asynchronous frequency modulation modes and linear frequency modulation modes, causing the channel amplitude and phase distortion characteristics to dynamically change with the operating mode. Fixed-parameter compensation methods struggle to adapt to different modulation modes and their dynamic switching processes simultaneously, resulting in a significant decrease in compensation effectiveness under dual-mode dynamic switching scenarios, making it difficult to meet the requirements of high-precision imaging and target detection. Especially under broadband operating conditions, the amplitude and phase frequency response distortions of the channels further deteriorate, making traditional fixed-parameter compensation methods unsuitable for dual-mode dynamic switching scenarios.

[0005] In recent years, channel nonlinearity correction methods under deskew processing architectures have attracted attention. For example, Zhang Peng et al. proposed a "multi-channel time-varying correction factor" scheme in their paper "A Broadband Channel Digital Equalization Method Based on Deskew Processing." This method can alleviate channel nonlinearity inconsistency to some extent and compensates after deskewing based on linear frequency modulated (LFM) signals. However, this method is essentially only applicable to LFM waveforms because it uses LFM signals as excitation and relies on the stable single difference frequency structure and continuous linear phase evolution characteristics formed after deskew processing to achieve difference frequency extraction, reference signal construction, and time-varying correction factor modeling. However, stepped-frequency waveforms are essentially discrete frequency points and segmented constant-frequency signals, lacking continuous frequency sweep characteristics. In a single measurement, amplitude and phase distortion information at different frequency points cannot be extracted in the time domain. The key steps regarding difference frequency extraction, linear phase modeling, and time-varying nonlinear compensation do not hold true in stepped-frequency mode, making it difficult to directly apply this deskew-based channel nonlinearity correction method to the channel correction of stepped-frequency waveforms. Furthermore, this deskewing architecture requires different time-varying correction factors to be stored for different distorted waveforms, and cannot be directly extended to other waveform systems such as step frequency, nor does it address the coefficient adaptation problem under dual-mode dynamic switching conditions.

[0006] In related technologies, patent application CN118607448A proposes a channel equalizer design method with controllable frequency response error. By transforming the equalizer design problem into a second-order cone programming optimization problem, it can flexibly control the amplitude and phase frequency response errors, improving the robustness of engineering applications. However, this method relies on multiple frequency point measurements and linear interpolation, resulting in high computational complexity and sensitivity to measurement accuracy. It also involves a large number of matrix optimization solutions, placing high demands on processor computation and storage, potentially posing challenges in real-time or resource-constrained systems. Furthermore, it requires additional reference channels, making it difficult to adapt to the application requirements of low-cost, high-real-time dual-mode millimeter-wave radar systems, thus limiting its practical promotion and engineering application in such scenarios. Patent application CN119846631A proposes a phase synchronization and multi-channel equalization collaborative phase-preserving processing method. By collaboratively processing phase synchronization and channel equalization, it avoids the risk of destroying the synchronization phase in traditional methods. The reference channel conversion strategy effectively preserves the initial phase value, improving the accuracy of the SAR system and exhibiting good adaptability and robustness. However, this method relies on the accurate identification and demodulation of the synchronization signal channel, requiring high consistency of system data and channels. Furthermore, it may increase complexity when the number of channels is large or synchronization changes frequently. In addition, it does not address the nonlinearity, temperature drift, and amplitude and phase frequency distortion caused by multi-waveform switching in monostation radar systems, exhibiting limitations in waveform adaptability and multi-mode compensation. Patent application CN118707521A proposes a radar imaging method based on stepped-frequency deskewing and minimum entropy phase compensation. It estimates the phase difference between sub-pulses using conjugate multiplication of the overlapping spectrum of adjacent sub-pulses and achieves full-pulse phase compensation through recursive accumulation. While this method is simple to implement and computationally inexpensive, its estimation performance is highly dependent on the signal-to-noise ratio (SNR) and the degree of overlap between adjacent sub-bands. Its stability significantly decreases at low SNR or insufficient overlapping bandwidth. Simultaneously, the recursive accumulation structure causes errors to gradually accumulate and propagate backward, significantly amplifying the compensation error when the number of pulses is large. The patent application document with publication number CN110888131A proposes a multi-channel phase compensation method based on frequency-stepping SAR radar. It estimates the phase difference between frequency bands by traversing and searching the phase compensation values ​​and selecting the best value through pulse compression. Each candidate phase requires a full-frame pulse compression operation, which has high computational complexity and is difficult to meet the requirements of real-time or online correction. It is only suitable for offline processing. At the same time, it assumes that there is only a single fixed phase difference between frequency bands, the applicable model is relatively simple, it is sensitive to noise, the estimation accuracy is limited by the traversal resolution, and it lacks strict optimality and robustness guarantees.

[0007] In summary, current millimeter-wave radar channel distortion compensation techniques suffer from the following common shortcomings: First, most methods are only designed for a single waveform mode, lacking compatibility with dual-mode stepped-frequency and linear frequency modulation operating modes; second, existing equalization algorithms often neglect the real-time matching and smooth transition of equalization coefficients during mode switching; and third, the compensation process places strict requirements on the calibration signal and fails to fully consider time-varying drift caused by factors such as temperature and aging in actual systems. Therefore, there is an urgent need to develop a channel amplitude and phase distortion compensation method and device that can adapt to dual-mode operation, support dynamic coefficient switching, and possess good engineering feasibility, in order to improve the channel consistency, beam quality, and imaging performance of dual-mode millimeter-wave radar across the entire bandwidth. Summary of the Invention

[0008] This invention aims to propose a dual-mode channel equalization solution that takes into account both stepped frequency and linear frequency modulation dual-mode working scenarios. It is suitable for real-time correction of the amplitude and phase frequency response of the receiving channel in both stepped frequency and linear frequency modulation working modes.

[0009] The present invention solves the above-mentioned technical problems through the following technical means:

[0010] This invention proposes a method for compensating for amplitude and phase distortion in a dual-mode millimeter-wave radar channel, the method comprising: S1. Obtain the transmit and receive control parameters and parse the transmit and receive control parameters to obtain the transmit waveform parameters and the local oscillator waveform parameters; S2. Starting from the low-frequency end of the system's operating bandwidth, adjust the transmit waveform parameters and the local oscillator waveform parameters to generate paired single-tone correction signals, which are then used as the transmit pre-drive signal and the local oscillator pre-drive signal, respectively. S3. Generate an analog intermediate frequency signal based on the transmit pre-drive signal and the local oscillator pre-drive signal, and extract the distortion amplitude and phase values ​​corresponding to the current frequency point from the analog intermediate frequency signal; S4. Control the frequency of the paired single-tone correction signals to increase gradually according to the preset step frequency step size, and repeat steps S2 to S3 until the frequency reaches the high-frequency end of the system working bandwidth, and obtain the distortion amplitude and phase values ​​corresponding to multiple frequency points within the system working bandwidth. S5. For the distortion amplitude and phase values ​​of multiple frequency points, perform interpolation fitting within the sampling frequency range to calculate the equalizer coefficient of the current step frequency step size. S6. Adjust the preset step frequency step size and repeat steps S2 to S5 until the equalizer coefficients corresponding to all typical step frequency step sizes are obtained for use in dual-mode millimeter-wave radar channel amplitude and phase distortion compensation.

[0011] Furthermore, the dual-mode millimeter-wave radar channel amplitude and phase distortion compensation method is characterized in that, the step of acquiring and analyzing the transmit / receive control parameters to obtain the transmit waveform parameters and the local oscillator waveform parameters further includes: The transmit and receive control parameters are analyzed to obtain the serial frame in the transmit and receive control parameters, and the transmit waveform parameters, local oscillator waveform parameters and closed-loop correction selection signal are extracted in sequence. The closed-loop calibration selection signal is used to control the switching of the signal flow path between the calibration state and the normal operation state.

[0012] Further, the step of adjusting the transmit waveform parameters and local oscillator waveform parameters starting from the low-frequency end of the system's operating bandwidth to generate paired single-tone correction signals, which are then used as the transmit pre-drive signal and the local oscillator pre-drive signal, respectively, includes: The transmitted waveform parameters and the local oscillator waveform parameters are mapped to the transmitted waveform frequency parameters and the local oscillator waveform frequency parameters, respectively. Within the system's operating bandwidth, using the low-frequency end as the starting frequency of the operating bandwidth, the frequency parameters of the transmitted waveform and the local oscillator waveform are stepped according to the set step size until the total number of steps is reached. The transmitted waveform frequency and the local oscillator waveform frequency corresponding to each frequency step are calculated. The corresponding digital signal is generated based on the frequency of the transmitted waveform corresponding to each frequency step, and the corresponding digital local oscillator signal is generated based on the local oscillator waveform parameters corresponding to each frequency step. The transmitted digital signal corresponding to each frequency step is delayed to generate a transmitted digital delay signal. The transmitted digital delay signal is then mapped to a continuous time domain and mixed with a fixed frequency to obtain the transmitted pre-drive signal. The digital local oscillator signal corresponding to each frequency step is equalized to obtain the equalized local oscillator digital signal. The equalized local oscillator digital signal is then mapped to the continuous time domain and mixed with a fixed frequency to obtain the local oscillator pre-drive signal.

[0013] Further, the step of delaying the transmitted digital signal corresponding to each frequency step to generate a transmitted digital delay signal, and then mapping the transmitted digital delay signal to a continuous time domain and mixing it with a fixed frequency to obtain the transmitted pre-drive signal, includes: For each frequency step, the transmitted digital signal is subjected to channel digital path delay processing to generate a transmitted digital delay signal, expressed by the formula:

[0014] In the formula, To transmit digital delay signals, To transmit digital signals, is the base of the natural logarithm. The imaginary unit represents the channel digital path delay. , This represents the number of taps in the FIR equalizer filter. For the first The frequency of the transmitted waveform corresponding to the next frequency step. Sampling frequency, The number of sampling points for transmitting digital signals; The transmitted digital delay signal is converted from digital to analog to map it to the continuous time domain, and then mixed with a fixed frequency to obtain the transmitted pre-drive signal, as expressed by the formula:

[0015] In the formula, To transmit the pre-drive signal, To transmit the pre-drive signal amplitude, is the base of the natural logarithm. The imaginary unit represents the channel digital path delay. , This represents the number of taps in the FIR equalizer filter. For the first The frequency of the transmitted waveform corresponding to the next frequency step. For the fixed frequency of mixing, For time scale.

[0016] Further, the process of equalizing the digital local oscillator signal corresponding to each frequency step to obtain an equalized local oscillator digital signal, and then mapping the equalized local oscillator digital signal to a continuous time domain and mixing it with a fixed frequency to obtain the local oscillator pre-drive signal, includes: The digital local oscillator signal corresponding to each frequency step is equalized to obtain the equalized digital local oscillator signal, as expressed by the formula:

[0017] In the formula, The local oscillator digital signal after equalization. It is a digital local oscillator signal. The transfer function of the pre-calibrated unit equalizer. For discrete convolution symbols, This represents the number of taps in the FIR equalizer filter. These are the tap coefficients of the FIR equalizer filter. The original local oscillator digital signal, <1 hour Take the value of zero; At time 1, the aforementioned Normally, the value of the local oscillator digital signal is taken. ; The equalized local oscillator digital signal is converted from digital to analog and mapped to the continuous time domain. Then, it is mixed with a fixed frequency to obtain the local oscillator pre-drive signal, which is expressed by the following formula:

[0018] In the formula, This is the local oscillator pre-drive signal. The amplitude of the local oscillator pre-drive signal. is the base of the natural logarithm. The imaginary unit, For the first The local oscillator waveform frequency corresponding to the next frequency step. Sampling frequency, For the fixed frequency of mixing, For time scale.

[0019] Furthermore, the transfer function exist The expression for the domain is:

[0020] In the formula, It satisfies unit equilibrium.

[0021] Furthermore, the transmit pre-drive signal and the local oscillator pre-drive signal are a pair of single-tone signals with a fixed frequency difference, the fixed frequency difference being equal to a preset step frequency step size.

[0022] Further, the step of generating an analog intermediate frequency (IF) signal based on the transmit pre-drive signal and the local oscillator pre-drive signal, and extracting the distortion amplitude and phase values ​​corresponding to the current frequency point from the analog IF signal, includes: After filtering and power amplification of the pre-drive signal, a portion of the amplified radio frequency transmission signal is coupled as a coupling detection signal, and the other portion is sent to the transmission antenna as a transmission signal. After filtering and power amplification of the local oscillator pre-drive signal, a radio frequency receiving local oscillator signal is generated, wherein the digital transmission path delay corresponding to the transmit pre-drive signal and the local oscillator pre-drive signal is the same; Based on the closed-loop correction selection signal, the coupled detection signal is used as the receiving selection signal and is sequentially mixed, filtered and amplified with the radio frequency receiving local oscillator signal to generate an analog intermediate frequency signal. After performing analog-to-digital conversion on the analog intermediate frequency signal, the distortion amplitude and phase values ​​corresponding to the current frequency point are extracted.

[0023] Further, the step of extracting the distortion amplitude and phase values ​​corresponding to the current frequency point after performing analog-to-digital conversion on the analog intermediate frequency signal includes: The analog intermediate frequency signal is processed by analog-to-digital conversion to obtain compensated echo data, and multi-frequency correction data of the compensated echo signal is extracted within the working bandwidth. The formula for single-point frequency data is expressed as:

[0024] In the formula, It is a single-point frequency signal. The amplitude value of the mixed signal. The phase difference after mixing. For frequency difference, For phase, is the base of the natural logarithm. The imaginary unit, The time-domain sampling point number; Perform DFT transformation on the multi-frequency correction data, and extract the amplitude and phase values ​​corresponding to the mixing of the transmission frequency and the local oscillator frequency as the distortion amplitude and phase values ​​corresponding to the current frequency.

[0025] Furthermore, the equalizer coefficients for the current step frequency size are calculated by interpolating and fitting the distortion amplitude and phase values ​​at multiple frequency points within the sampling frequency range, including: For the distortion amplitude and phase values ​​at multiple frequency points, interpolation fitting is performed within the sampling frequency range, and inverse response calculation is performed on the amplitude and phase interpolation data. The frequency coordinates of the interpolation frequency points correspond one-to-one with the frequency points of the fast Fourier transform used to calculate the equalizer coefficients. Construct the DFT matrix of the inverse response of the amplitude-phase interpolation data based on the inverse response of the amplitude-phase interpolation data; Based on the system's operating bandwidth, sampling frequency, and the number of FFT points after frequency domain interpolation, a weighted vector of the inverse response of the amplitude-phase interpolation data is constructed. Based on the DFT matrix and weighting vector of the amplitude-phase interpolation inverse response, the equalizer coefficients for the current step frequency step size are solved using the weighted least squares method.

[0026] Furthermore, the interpolation fitting of the distortion amplitude and phase values ​​at multiple frequency points within the sampling frequency range, and the inverse response calculation of the amplitude and phase interpolation data, includes: For the distortion amplitude and phase values ​​at multiple frequency points, interpolation fitting is performed within the sampling frequency range, and the equalizer frequency response is designed to be the inverse response of the amplitude and phase interpolation data:

[0027] In the formula, This is the inverse response after amplitude and phase distortion interpolation. This represents the number of frequency points in the FFT after frequency domain interpolation. This represents the amplitude distortion value corresponding to the k-th FFT frequency point. This represents the phase distortion value corresponding to the k-th FFT frequency point; The frequency domain response vector is obtained by performing time delay compensation on the inverse response of the amplitude-phase interpolation data, as expressed by the formula:

[0028] In the formula, It is the frequency domain response vector. This is the inverse response after amplitude and phase distortion interpolation. This indicates element-wise multiplication. This is the time delay compensation phase vector. , This refers to the number of delay points introduced.

[0029] Furthermore, the construction of the DFT matrix of the amplitude-phase interpolation data inverse response based on the amplitude-phase interpolation data inverse response is expressed by the following formula:

[0030] In the formula, The DFT matrix, This represents the number of taps in the FIR equalizer filter. This represents the number of frequency points in the FFT after frequency domain interpolation. This refers to the sequence number of the frequency sampling point (FFT frequency point). This is the tap number (time-domain tap index) of the FIR equalizer filter.

[0031] Furthermore, the weighting vector is calculated using an amplitude-weighted method, where the in-band weighting amplitude of the equalizer's frequency response compensation is higher than that outside the band.

[0032] Furthermore, the step of constructing a weighted vector of the inverse response of the amplitude-phase interpolated data based on the system operating bandwidth, sampling frequency, and the number of FFT points after frequency domain interpolation includes: Based on the system's operating bandwidth, sampling frequency, and the number of FFT points after frequency domain interpolation, the frequency domain weighted vector is constructed as follows:

[0033] In the formula, For frequency domain weighted vectors, This represents the number of FFT points after frequency domain interpolation. For system operating bandwidth, The sampling frequency; The normalized weighting matrix is ​​calculated based on the frequency domain weighting vector:

[0034] In the formula, For normalized weighted matrices, This indicates that a diagonal matrix is ​​constructed using vector elements.

[0035] Furthermore, the equalizer coefficients for the current step frequency step size are solved using the weighted least squares method based on the DFT matrix and weighted vector of the amplitude-phase interpolation inverse response, including: The problem of constructing weighted least squares coefficients based on the DFT matrix and weighting vector of the amplitude-phase interpolation inverse response is as follows:

[0036] In the formula, This is the coefficient vector of the FIR equalizer filter. For normalized weighted matrices, The DFT matrix, It is the frequency domain response vector. To find the optimal The value is set to minimize the weighted sum of squared errors.

[0037] Solve the problem of weighted least squares coefficients, and calculate the weighted least squares solution. As the equalizer coefficient for the current step frequency step size This represents the number of taps in the FIR equalizer filter. These are the tap coefficient values ​​of the FIR equalizer filter.

[0038] Furthermore, the process of compensating for amplitude and phase distortion in dual-mode millimeter-wave radar channels includes: When switching to normal operation based on the closed-loop correction selection signal, the operating mode of the dual-mode millimeter-wave radar transceiver system is determined. When in step frequency working mode, select the equalizer coefficient corresponding to the current step frequency interval from the equalizer coefficient table to achieve amplitude and phase distortion compensation of the step frequency channel. When in linear frequency modulation (LFM) mode, the intermediate frequency (IF) is estimated based on prior information, and the equalizer coefficient with the step size closest to the estimated IF is selected from the equalizer coefficient table to achieve amplitude and phase distortion compensation for the LFM channel. The equalizer coefficient table is constructed based on the equalizer coefficients corresponding to all typical step frequency steps.

[0039] Furthermore, when in linear frequency modulation (LFM) mode, estimating the intermediate frequency based on prior information includes: Estimate the timing parameters of the received signal based on known distance or time interval information between transmitted signals; A mapping relationship is established using the linear correspondence between the time parameters and the frequency; Substitute the time parameter into the mapping relationship to calculate the corresponding intermediate frequency deviation or intermediate frequency value.

[0040] Furthermore, this invention also proposes a dual-mode millimeter-wave radar channel amplitude and phase distortion compensation device, the device comprising: The calibration processor is used to acquire and parse the transmit and receive control parameters to obtain the transmit waveform parameters and the local oscillator waveform parameters, and send them to the transmit waveform generator and the local oscillator waveform generator respectively. The transmit waveform generator is used to adjust the transmit waveform parameters starting from the low-frequency end of the system's operating bandwidth, generate a transmit pre-drive signal, and send it to the transmit component. The local oscillator waveform generator is used to adjust the local oscillator waveform parameters starting from the low-frequency end of the system's operating bandwidth, generate a local oscillator pre-drive signal, and send it to the receiving component. The transmitting component is used to convert the transmit pre-drive signal and send it to the receiving component, so that the receiving component can generate an analog intermediate frequency signal based on the converted transmit pre-drive signal and the received local oscillator pre-drive signal and send it to the correction processor; The calibration processor is also used to extract the distortion amplitude and phase values ​​corresponding to the current frequency point from the analog intermediate frequency signal, and control the frequencies of the transmit pre-drive signal generated by the transmit waveform generator and the local oscillator pre-drive signal generated by the local oscillator waveform generator to increase successively according to the preset step frequency step size until the frequency reaches the high-frequency end of the system operating bandwidth to obtain the distortion amplitude and phase values ​​corresponding to multiple frequency points within the system operating bandwidth; and to perform interpolation fitting on the distortion amplitude and phase values ​​of multiple frequency points within the sampling frequency range to calculate the equalizer coefficient of the current step frequency step size, readjust the preset step frequency step size and control the transmit waveform generator to regenerate the transmit pre-drive signal, control the local oscillator waveform generator to regenerate the local oscillator pre-drive signal until the equalizer coefficients corresponding to all typical step frequency step sizes are obtained for use in dual-mode millimeter-wave radar channel amplitude and phase distortion compensation.

[0041] Furthermore, the correction processor is connected to a signal processor, which is used to configure the transmit / receive control parameters; The transmit / receive control parameters include operating mode parameters, transmit link parameters, and receive link parameters.

[0042] Furthermore, the correction processor includes: The transmit / receive parameter parsing unit is used to parse the received transmit / receive control parameters to obtain the serial frame in the transmit / receive control parameters, extract the transmit waveform parameters, local oscillator waveform parameters and closed-loop correction selection signal in sequence, and send the closed-loop correction selection signal to the receiving component.

[0043] Furthermore, the transmitted waveform generator includes: The first direct frequency synthesizer is used to receive the transmit waveform parameters sent from the correction processor, map the transmit waveform parameters to transmit waveform frequency parameters, and within the system operating bandwidth, use the low frequency end as the starting frequency of the operating bandwidth to perform frequency stepping on the transmit waveform frequency parameters according to the set step size until the total number of steps is reached, calculate the transmit waveform frequency corresponding to each frequency step, generate the corresponding transmit digital signal from the transmit waveform frequency corresponding to each frequency step, and send it to the delay unit. The delay unit is used to perform delay processing on the transmitted digital signal to generate a transmitted digital delayed signal, and to send the transmitted digital delayed signal to the first digital-to-analog converter; The first digital-to-analog converter is used to convert the transmitted digital delay signal into a transmitted digital-to-analog converted signal and send it to the first mixer; The first mixer is used to mix the transmitted digital-to-analog conversion signal with a fixed frequency to obtain a transmitted pre-drive signal and send it to the transmitting component.

[0044] Furthermore, the local oscillator waveform generator includes: The second direct frequency synthesizer is used to receive the local oscillator waveform parameters sent from the correction processor and map the local oscillator waveform parameters to local oscillator waveform frequency parameters. Within the system operating bandwidth, the local oscillator waveform frequency parameters are stepped up according to a set step size, starting from the low-frequency end of the operating bandwidth, until the total number of steps is reached. The local oscillator waveform frequency corresponding to each frequency step is calculated so as to generate the corresponding digital local oscillator signal according to the local oscillator waveform parameters corresponding to each frequency step and send it to the equalizer. The equalizer is used to perform equalization processing on the digital local oscillator signal corresponding to each frequency step, obtain the equalized local oscillator digital signal, and send it to the second digital-to-analog converter. The second digital-to-analog converter is used to map the equalized local oscillator digital signal to the continuous time domain to obtain the local oscillator digital-to-analog converted signal and send it to the second mixer; The second mixer is used to mix the local oscillator digital-to-analog conversion signal with a fixed frequency to obtain the local oscillator pre-drive signal and send it to the receiving component.

[0045] Furthermore, the transmitting component includes: A bandpass filter is used to receive the transmit pre-drive signal from the transmit waveform generator module, filter the transmit pre-drive signal to obtain the transmit filtered signal, and send it to the power amplifier. The power amplifier is used to amplify the power of the transmitted filtered signal to obtain the amplified transmitted signal and send it to the coupler. The coupler is used to couple a portion of the amplified radio frequency transmitted signal as a coupling detection signal to the receiving component, and to send the other portion as a transmitted signal to the transmitting antenna.

[0046] Further, the receiving component includes: A low-noise amplifier is used to amplify the target echo signal received from the receiving antenna in a low-noise manner, obtain a low-noise amplified signal, and send it to the switch. The switch is used to select, based on the closed-loop correction selection signal received from the correction processor, the low-noise amplifier signal output by the low-noise amplifier or the partially coupled detection signal output by the transmitting component as the receiving selection signal and send it to the mixer amplifier. The mixer amplifier is used to mix and amplify the radio frequency received local oscillator signal sent by the filter amplifier with the receive selection signal to obtain a mixed signal and send it to the programmable filter. The filter amplifier is used to filter and amplify the local oscillator pre-drive signal sent by the local oscillator waveform generator to obtain the radio frequency received local oscillator signal and send it to the mixer amplifier. The programmable filter is used to sequentially mix, filter, and amplify the received selection signal and the radio frequency received local oscillator signal to generate an analog intermediate frequency signal.

[0047] Furthermore, the correction processor includes: An analog-to-digital converter is used to receive an analog intermediate frequency signal from a receiving component, perform analog-to-digital conversion on the analog intermediate frequency signal to obtain compensated echo data, and send it to a calibration data acquisition unit and a signal processor. The calibration data acquisition unit is used to convert the compensated echo data into multi-frequency calibration data and send it to the amplitude-phase conversion unit; The amplitude and phase extraction unit is used to perform DFT transformation on the multi-frequency correction data, extract the amplitude and phase values ​​corresponding to the transmission frequency and local oscillator frequency as the multi-frequency distortion amplitude and phase data corresponding to the current frequency point, and send them to the amplitude and phase interpolation unit. The amplitude-phase interpolation unit is used to perform interpolation fitting on multi-frequency distorted amplitude-phase data within the sampling frequency range to obtain amplitude-phase interpolation data and send it to the equalization coefficient calculation unit. The equalization coefficient calculation unit is used to calculate the equalizer coefficients based on amplitude and phase interpolation data and send them to the local oscillator waveform generator.

[0048] Furthermore, the equilibrium coefficient calculation unit includes: The amplitude-phase inverse response calculation subunit is used to perform inverse response calculation based on amplitude-phase interpolation data, obtain the amplitude-phase interpolation inverse response, and send it to the DFT matrix construction subunit. The DFT matrix construction subunit is used to construct the DFT matrix of the inverse response of the amplitude-phase interpolation data based on the inverse response of the amplitude-phase interpolation data and send it to the weighted vector construction subunit. The weighted vector construction subunit is used to construct a weighted vector of the inverse response of the amplitude-phase interpolation data based on the system operating bandwidth, sampling frequency and the number of FFT points after frequency domain interpolation, and send it to the weighted least squares coefficient solving subunit. The weighted least squares coefficient solving subunit is used to solve the equalizer coefficients of the current step frequency step size using the weighted least squares method based on the DFT matrix and weighting vector of the amplitude-phase interpolation inverse response, and then send them to the local oscillator waveform generator.

[0049] Furthermore, under normal operating conditions, the transmit / receive parameter parsing unit outputs a closed-loop correction selection signal to the receiving component module according to the currently selected operating mode, and calls the pre-stored equalizer coefficient table in the equalizer of the local oscillator waveform generator to obtain the equalizer coefficients corresponding to the operating mode, performs real-time equalization compensation processing on the received target echo signal, and outputs the compensated echo data.

[0050] Furthermore, the present invention also proposes a computer-readable storage medium storing a computer program thereon, which, when executed by a processor, implements the dual-mode millimeter-wave radar channel amplitude and phase distortion compensation method as described above.

[0051] Furthermore, the present invention also proposes a millimeter-wave radar transceiver system that uses a dual-mode millimeter-wave radar channel amplitude and phase distortion compensation device as described above.

[0052] The advantages of this invention are: (1) The unified distortion acquisition and coefficient generation mechanism adopted in this invention does not depend on the specific transmitted waveform form, but takes the equivalent frequency domain transfer function of the channel as the unified modeling object. This is because, regardless of the step frequency mode or the linear frequency modulation mode, their essence is to excite and observe the response of the same channel under different frequency sampling structures: the former acquires channel response samples at discrete frequency points, and the latter acquires channel response information within a continuous frequency band. Therefore, this invention starts from the low-frequency end of the system operating bandwidth, generates paired single-tone correction signals by adjusting the transmitted and local oscillator waveform parameters, and gradually changes their frequencies according to a preset step size to acquire the amplitude and phase distortion characteristics of multiple frequency points; interpolates and fits the amplitude and phase characteristics to calculate the equalizer coefficients, and updates them with the step frequency until the equalization parameters of the entire frequency band are obtained, generating an equalization coefficient table. By adjusting the transmit and local oscillator waveform parameters to generate paired single-tone correction signals, and gradually changing their frequencies according to a preset step size, different modes correspond to different waveform parameters / frequency scanning ranges and steps. This yields the channel amplitude and phase response for each mode, thereby enabling the unified mapping of the channel amplitude and phase responses obtained from different modes to common frequency coordinates and interpolation reconstruction. This forms a unified channel response model, and the equalizer coefficients generated based on this model serve as compensation coefficients. It can simultaneously support two mainstream radar operating modes: stepped frequency and linear frequency modulation (LFM) deskewing. It has optimal equalization compensation performance for received signals with corresponding frequency differences. Therefore, based on the generated compensation coefficients, amplitude and phase distortion compensation for both stepped frequency and LFM channels of dual-mode millimeter-wave radar can be achieved. This effectively solves the compensation mismatch problem caused by waveform mode switching in traditional methods, significantly enhances the multi-task adaptability of the radar system, and allows for flexible configuration of operating modes without changing the hardware architecture, improving the convenience of system deployment and upgrades.

[0053] (2) In the distortion parameter extraction stage, the present invention uses fixed-frequency discrete Fourier transform (DFT) instead of traditional fast Fourier transform (FFT) to extract amplitude and phase information for fixed frequency points in step scanning. This design avoids the repeated calculation of full bandwidth data by FFT, greatly reduces the computational complexity and resource consumption of digital signal processing, and is especially beneficial for achieving efficient and low-latency distortion correction in embedded or real-time platforms.

[0054] (3) The present invention can dynamically select the compensation coefficient that best matches the current working mode from a pre-generated equalizer coefficient table based on the specific parameters of the current working mode, such as the step frequency difference and the estimated intermediate frequency value of linear frequency modulation. This mechanism not only achieves accurate matching between the compensation parameters and the real-time waveform characteristics, but also further optimizes the coefficient selection by combining the target prior information, thereby maintaining high-precision channel consistency in complex working environments, improving the problem of amplitude and phase distortion caused by phase nonlinearity in existing systems that limits the signal-to-noise ratio, and improving the detection stability and imaging quality of the system in dynamic scenes.

[0055] Additional aspects and advantages of the invention will be set forth in part in the description which follows, and in part will be obvious from the description, or may be learned by practice of the invention. Attached Figure Description

[0056] The accompanying drawings, which form part of this invention, are used to provide a further understanding of the invention. The illustrative embodiments of the invention and their descriptions are used to explain the invention and do not constitute an improper limitation of the invention.

[0057] Figure 1 This is a flowchart illustrating a method for compensating for amplitude and phase distortion in a dual-mode millimeter-wave radar channel according to an embodiment of the present invention. Figure 2 This is a schematic diagram of the structure of a dual-mode millimeter-wave radar channel amplitude and phase distortion compensation device according to an embodiment of the present invention; Figure 3 This is a schematic diagram of the structure of a transmit waveform generator in one embodiment of the present invention; Figure 4 This is a schematic diagram of the structure of a local oscillator waveform generator in one embodiment of the present invention; Figure 5 This is a schematic diagram of the structure of the transmitting component in one embodiment of the present invention; Figure 6 This is a schematic diagram of the receiving component in one embodiment of the present invention; Figure 7 This is a schematic diagram of the structure of the calibration processor in one embodiment of the present invention; Figure 8 This is a schematic diagram of the structure of the equilibrium coefficient calculation unit in one embodiment of the present invention. Detailed Implementation

[0058] To make the objectives, technical solutions, and advantages of the embodiments of the present invention clearer, the technical solutions of the embodiments of the present invention will be clearly and completely described below in conjunction with the embodiments of the present invention. Obviously, the described embodiments are only some embodiments of the present invention, not all embodiments. Based on the embodiments of the present invention, all other embodiments obtained by those skilled in the art without creative effort are within the scope of protection of the present invention.

[0059] like Figure 1 As shown, the first embodiment of the present invention proposes a method for compensating for amplitude and phase distortion in a dual-mode millimeter-wave radar channel, the method comprising the following steps: S1. Obtain the transmit / receive control parameter a and parse the transmit / receive control parameter a to obtain the transmit waveform parameter b and the local oscillator waveform parameter e; S2. Starting from the low-frequency end of the system's operating bandwidth, adjust the transmit waveform parameter b and the local oscillator waveform parameter e to generate paired single-tone correction signals, which are then used as the transmit pre-drive signal c and the local oscillator pre-drive signal g, respectively. S3. Generate an analog intermediate frequency signal based on the transmit pre-drive signal and the local oscillator pre-drive signal, and extract the distortion amplitude and phase values ​​corresponding to the current frequency point from the analog intermediate frequency signal; S4. Control the frequency of the paired single-tone correction signals to increase gradually according to the preset step frequency step size, and repeat steps S2 to S3 until the frequency reaches the high-frequency end of the system working bandwidth, and obtain the distortion amplitude and phase values ​​corresponding to multiple frequency points within the system working bandwidth. S5. For the distortion amplitude and phase values ​​of multiple frequency points, perform interpolation fitting within the sampling frequency range to calculate the equalizer coefficient of the current step frequency step size. S6. Adjust the preset step frequency step size and repeat steps S2 to S5 until the equalizer coefficients corresponding to all typical step frequency step sizes are obtained, which can be used for amplitude and phase distortion compensation of dual-mode millimeter-wave radar channels.

[0060] It should be noted that the core challenge of channel amplitude and phase correction in dual-mode dynamic switching scenarios lies in the significant differences in the equivalent channel transfer function between different modes in terms of frequency domain sampling structure, frequency coverage, and statistical characteristics. This leads to abrupt changes in the channel amplitude and phase response with the switching of operating modes, making the compensation parameters estimated based on a single mode no longer applicable in another mode. The channel equivalent model changes dynamically with the modulation mode and operating state, while the correction parameters need to be accurately updated in a very short time. Furthermore, stable and reliable real-time estimation is difficult under conditions of strong noise, non-ideal hardware, and multi-channel coupling.

[0061] This embodiment switches the dual-mode millimeter-wave radar transceiver system from normal detection state to internal calibration state by configuring transmit and receive control parameters, and simultaneously triggers the calibration workflow. Starting from the low-frequency end of the system's operating bandwidth, it generates paired single-tone calibration signals by adjusting the transmit and local oscillator waveform parameters, and gradually changes their frequencies according to a preset step size to obtain the amplitude and phase distortion characteristics of multiple frequency points. The amplitude and phase characteristics are interpolated and fitted to calculate the equalizer coefficients, and updated with the step frequency until the equalizer parameters of the entire frequency band are obtained. The equalizer coefficients corresponding to all typical step frequency steps are generated, and an equalizer coefficient table is constructed accordingly. The unified distortion acquisition and coefficient generation mechanism adopted does not depend on the specific transmitted waveform form, but takes the channel's equivalent frequency domain transfer function as the unified modeling object. The equalizer coefficients generated accordingly serve as compensation coefficients to support both stepped frequency and linear frequency modulation modes. It has the ability to match and switch equalizer coefficients in real time, and the error is flexible and controllable. The equalizer coefficient table has the optimal equalization compensation performance for the received signal with the corresponding frequency difference, so that the equalizer coefficients can be flexibly adjusted according to the frequency deviation, and the equalizer coefficient table can achieve effective equalization compensation for the received signal under the corresponding frequency deviation conditions.

[0062] As a further preferred technical solution, step S1, which involves obtaining and parsing the transmit / receive control parameters to obtain the transmit waveform parameters and the local oscillator waveform parameters, further includes: The transmit / receive control parameter a is parsed to obtain the serial frame in the transmit / receive control parameter, and the transmit waveform parameter b, the local oscillator waveform parameter e, and the closed-loop correction selection signal h are extracted in sequence. The closed-loop correction selection signal h is used to control the switching of the signal flow path between the correction state and the normal operation state.

[0063] It should be noted that in this embodiment, the closed-loop correction selection signal h obtained by parsing the transmit and receive control parameters switches the radar system to the internal correction state. Starting from the low-frequency end of the system's operating bandwidth, paired single-tone correction signals are generated by adjusting the transmit and local oscillator waveform parameters, and their frequencies are gradually changed according to a preset step size to obtain the amplitude and phase distortion characteristics of multiple frequency points. The amplitude and phase characteristics are interpolated and fitted to calculate the equalizer coefficients, which are updated with the step frequency until the equalization parameters of the entire frequency band are obtained. The equalizer coefficients corresponding to all typical step frequency step sizes are generated, and an equalization coefficient table is constructed accordingly. Then, after switching to the normal operating state according to the closed-loop correction selection signal h, the target echo signal is received, and amplitude and phase distortion compensation of the dual-mode millimeter-wave radar channel is performed based on the pre-generated equalization coefficient table. This method supports both step frequency and linear frequency modulation modes, has real-time coefficient matching and switching capabilities, and significantly improves the amplitude and phase consistency and system imaging performance under multi-channel conditions.

[0064] It should be understood that the radar switches to calibration mode when it is initially shipped from the factory or when the system hardware status changes; after calibration is completed and an effective compensation coefficient is obtained, it switches to normal operating mode.

[0065] As a further preferred technical solution, step S2: starting from the low-frequency end of the system's operating bandwidth, adjusting the transmit waveform parameters and the local oscillator waveform parameters to generate paired single-tone correction signals, which are respectively used as the transmit pre-drive signal and the local oscillator pre-drive signal, specifically including the following steps: S21. Map the transmitted waveform parameter b and the local oscillator waveform parameter e to the transmitted waveform frequency parameter and the local oscillator waveform frequency parameter, respectively. Specifically, in this embodiment, the low-frequency end of the system's operating bandwidth is set to... Set the frequency step size to The frequency of the transmitted waveform is set based on the transmitted waveform parameter b. Based on the local oscillator waveform parameter e, the frequency of the local oscillator waveform is set as follows: .in, Indicates the starting frequency of the operating bandwidth. This indicates the interval between two consecutive frequency steps.

[0066] Within the operating bandwidth range, at the low frequency end Starting from the step size, the step size is as follows: Perform frequency stepping, the first The transmit waveform frequency corresponding to the next step With local oscillator waveform frequency They respectively satisfy the following relations:

[0067]

[0068] in, It represents the number of steps, starting from 1, indicating the first step within the bandwidth frequency range. One step, Indicates the first The frequency of the transmitted waveform corresponding to the next frequency step. Indicates the first The frequency of the local oscillator waveform corresponding to the next frequency step.

[0069] In practice, the low-frequency end of the system's operating bandwidth is determined. Choose a suitable frequency step size Set the number of steps and set a loop variable. Starting from 1, until a certain maximum value is reached. This value can be calculated based on the actual frequency range and step frequency to determine the total number of steps. in, It is the high-frequency end of the system's operating bandwidth. The total number of steps is a positive integer, and the method used is... Integer function.

[0070] S22. Within the system's operating bandwidth, using the low-frequency end as the starting frequency of the operating bandwidth, the frequency parameters of the transmitted waveform and the local oscillator waveform are stepped according to the set step size until the total number of steps is reached. The transmitted waveform frequency and the local oscillator waveform frequency corresponding to each frequency step are calculated. Specifically, in this embodiment, a loop variable is set according to the number of steps. In each frequency stepping cycle, based on the current step number According to the formula Calculate the corresponding first One transmit waveform frequency parameter In each frequency stepping cycle, based on the current step number... According to the formula Calculate the corresponding first Local oscillator waveform frequency parameters .

[0071] S23. Generate the corresponding digital signal based on the frequency of the transmitted waveform corresponding to each frequency step, and generate the corresponding digital local oscillator signal based on the local oscillator waveform parameters corresponding to each frequency step. Specifically, this embodiment is based on the first... One transmit waveform frequency parameter Generate the corresponding transmitted digital signal It can be expressed in complex exponential form:

[0072] in, , It is the number of sampling points of the signal. It is the sampling frequency. is the base of the natural logarithm. For the imaginary unit to satisfy .

[0073] In obtaining the first Local oscillator waveform frequency parameters Then, a corresponding digital local oscillator signal is generated based on the local oscillator waveform frequency parameters. Its expression in the discrete-time domain is:

[0074] in, Indicates the first The frequency parameters of the local oscillator waveform corresponding to the next frequency step. It is the number of sampling points of the signal. It is the sampling frequency. is the base of the natural logarithm. For the imaginary unit to satisfy .

[0075] It should be noted that after generating the transmit digital signal and the local oscillator digital signal corresponding to the current step, the step number... The process of generating the transmitted digital signal and digital local oscillator signal corresponding to the current step number is incremented and repeated until the step number reaches the preset maximum value. This completes the process of generating multi-frequency signals covering the entire operating bandwidth.

[0076] S24. Delay the transmitted digital signal corresponding to each frequency step to generate a transmitted digital delay signal, and then map the transmitted digital delay signal to the continuous time domain and mix it with a fixed frequency to obtain the transmitted pre-drive signal. Specifically, this embodiment transmits digital signals. To ensure causality in the system, a channel digital path delay is introduced. ,in It is the number of taps in the FIR equalizer filter, which generates the transmitted digital delay signal. :

[0077] Then transmit the digital delay signal Perform digital-to-analog conversion to map discrete signals to the continuous time domain. Then with a fixed frequency The pre-drive signal is obtained by mixing. :

[0078] simplify :

[0079] in, To transmit the pre-drive signal amplitude, For the fixed frequency of mixing, For time scale.

[0080] S25. Perform equalization processing on the digital local oscillator signal corresponding to each frequency step to obtain the equalized local oscillator digital signal, and then map the equalized local oscillator digital signal to the continuous time domain and mix it with a fixed frequency to obtain the local oscillator pre-drive signal.

[0081] Specifically, this embodiment uses the digital local oscillator signal. The signal is fed into a pre-corrected unit equalizer (a fixed delay circuit used before obtaining effective equalizer coefficients, where the coefficients are constants) to obtain the equalized local oscillator digital signal. This equalized local oscillator digital signal is then converted from digital to analog and mapped to the continuous time domain. Finally, it is mixed with a fixed frequency to obtain the local oscillator pre-drive signal. : 1) Transfer function of pre-corrected unit equalizer for First-order FIR equalization filter, its Domain expression:

[0082] in, The tap coefficients of the FIR equalizer filter satisfy "unity equalization" (amplitude response ≈ 1, phase response compensates for channel distortion).

[0083] The digital local oscillator signal is discretely convolved with the pre-corrected unit equalizer to obtain the equalized local oscillator signal e2:

[0084] in, Represents discrete convolution. The original local oscillator digital signal, i.e., the digital local oscillator signal. The nk-th sampling point represents the digital local oscillator signal. The k-beat-delayed sample values ​​on the discrete time axis are used to compare with the tap coefficients of the FIR equalization filter. Perform convolution operations. When <1, the Take the value of zero; At time 1, the aforementioned Normally, the value of the local oscillator digital signal is taken.

[0085] 2) The equalized local oscillator digital signal Mapping to the continuous time domain, we obtain the local oscillator digital-to-analog conversion signal e3. and with a fixed frequency The local oscillator pre-drive signal is obtained by mixing:

[0086] Among them, simplification For equalizer in The frequency response at that point can be further simplified to:

[0087] in, The amplitude of the local oscillator pre-drive signal. For the fixed frequency of mixing, For time scale, Let be the transfer function of the pre-calibrated unit equalizer, with the number of taps being . .

[0088] It should be noted that in practical applications, during step frequency correction, the sampling frequency... The system operating bandwidth is 512 MHz. 400MHz; pulse width for Number of sampling points 1024; Number of taps in the FIR equalizer filter The step size is 64; the step frequency is [missing information]. for The fixed frequency of mixing It is 75GHz.

[0089] As a further preferred technical solution, the transmit pre-drive signal and the local oscillator pre-drive signal are a pair of single-tone signals with a fixed frequency difference, wherein the fixed frequency difference is equal to a preset step size. .

[0090] As a further preferred technical solution, step S3: generating an analog intermediate frequency signal based on the transmit pre-drive signal and the local oscillator pre-drive signal, and extracting the distortion amplitude and phase values ​​corresponding to the current frequency point from the analog intermediate frequency signal, specifically includes the following steps: S31. After filtering and power amplifying the pre-drive signal, a portion of the amplified radio frequency transmission signal is coupled as a coupling detection signal, and the other portion is sent to the transmission antenna as a transmission signal. Specifically, this embodiment transmits a pre-drive signal. After filtering and power amplification, a portion of the signal is coupled as a coupling detection signal. Specifically, it includes: 1) Use a bandpass filter to process the transmit pre-drive signal, retaining only the system operating bandwidth. internal signal components Mathematical expression:

[0091] In the formula, This indicates that a bandpass filter is applied to the time-domain signal.

[0092] 2) Next, the signal amplitude is increased by a power amplifier, mathematically expressed as:

[0093] In the formula, This is the amplified radio frequency transmission signal. This is the transmit power gain.

[0094] 3) Coupling a small portion (proportion) from the amplified RF transmitted signal. Mathematical expression:

[0095] In the formula, For the coupled detection signal, the coupling coefficient It is usually much less than 1.

[0096] 4) The generated coupling detection signal is: .

[0097] S32. After filtering and power amplifying the local oscillator pre-drive signal, an RF receiving local oscillator signal is generated, wherein the digital transmission path delay corresponding to the transmitting pre-drive signal and the local oscillator pre-drive signal is the same. Specifically, the local oscillator pre-drive signal g is simultaneously filtered and amplified to generate the radio frequency receiving local oscillator signal g1. The specific process includes: 1) Similarly, out-of-band spurious signals are filtered out for the local oscillator pre-drive signal g, retaining only the components within the system bandwidth:

[0098] In the formula, This indicates that the local oscillator pre-drive signal g is located within the system bandwidth.

[0099] 2) Amplify the local oscillator signal to the amplitude required by the receiving link using an amplifier:

[0100] In the formula, For local oscillator power gain, The radio frequency receiver receives the local oscillator signal g1.

[0101] 3) Generate the RF receiving local oscillator signal g1:

[0102] in, This is the power gain of the local oscillator.

[0103] S33. Based on the closed-loop correction selection signal, the coupled detection signal j is used as the receiving selection signal and mixed, filtered and amplified sequentially with the radio frequency receiving local oscillator signal g1 to generate an analog intermediate frequency signal k. Specifically, the core of frequency mixing is to achieve frequency shifting by multiplying the radio frequency signal with the local oscillator signal. The mathematical essence is to use the product-difference property of trigonometric functions to retain the difference frequency component (i.e., the intermediate frequency signal).

[0104] Mixer output signal for:

[0105] in, Represents the complex conjugate of the local oscillator signal. To couple the detection signal amplitude, The amplitude of the mixed signal. The difference frequency after mixing. for , for , The link phase difference after mixing. Add phase to the link that couples the detection signal. Add phase to the link for receiving the local oscillator signal at radio frequency.

[0106] S34. After performing analog-to-digital conversion on the analog intermediate frequency signal k, extract the distortion amplitude and phase values ​​corresponding to the current frequency point.

[0107] As a further preferred technical solution, step S34: extracting the distortion amplitude and phase values ​​corresponding to the current frequency point after performing analog-to-digital conversion on the analog intermediate frequency signal k, specifically includes the following steps: S341. Perform analog-to-digital conversion on the analog intermediate frequency signal k to obtain compensated echo data, and extract multi-frequency correction data of the compensated echo signal within the working bandwidth range; Specifically, analog intermediate frequency signal The signal obtained through analog-to-digital conversion It is a complex signal, which can be represented as a single-point frequency signal:

[0108] in, It is the amplitude value of the mixed signal. It is the phase difference after mixing. It is a frequency difference .

[0109] For phase, This is the time-domain sampling point sequence number, indicating which sampling point the ADC outputs; S342. Perform DFT transformation on the multi-frequency correction data, and extract the amplitude and phase values ​​corresponding to the transmission frequency and local oscillator frequency as the distortion amplitude and phase values ​​corresponding to the current frequency.

[0110] Specifically, in practical applications, a DFT transform is performed directly on the multi-frequency correction data after mixing to extract the amplitude and phase values ​​corresponding to the transmit and local oscillator frequencies; the amplitude is extracted using a modulo operation. Phase extraction is achieved by using... and the angle(.) function: , The amplitude of the distorted signal. For the phase of the distorted signal, (.) represents the modulus function. (.) represents the phase unwinding function. (.) is used to find angle functions for complex numbers. For multi-frequency point serial numbers, (.) represents multi-frequency correction data. The frequency of the signal after mixing. (.) represents the DFT transformation operation.

[0111] It should be noted that in the distortion parameter extraction stage, this embodiment uses fixed-frequency discrete Fourier transform (DFT) instead of traditional fast Fourier transform (FFT) to extract amplitude and phase information for fixed frequency points in step scanning. This design avoids the repeated calculation of full-bandwidth data by FFT, greatly reducing the computational complexity and resource consumption of digital signal processing. It is especially beneficial for achieving efficient and low-latency distortion correction in embedded or real-time platforms.

[0112] As a further preferred technical solution, step S4: controlling the frequency of the paired single-tone correction signals to increase sequentially according to a preset step frequency size, and repeating steps S2 to S3 until the frequency reaches the high-frequency end of the system operating bandwidth, and obtaining the distortion amplitude and phase values ​​corresponding to multiple frequency points within the system operating bandwidth, specifically includes the following steps: 1) Step-by-step traversal

[0113] Generate the first The transmitted digital signal / digital local oscillator signal at each frequency point is processed through steps S2 to S3 to obtain the analog intermediate frequency signal. Then, analog-to-digital conversion is performed to obtain a digital signal. : 2) Constructing multi-frequency distortion amplitude and phase vectors The distortion amplitude and phase values ​​corresponding to multiple frequency points within the bandwidth are extracted. The functional expression for the distortion amplitude and phase values ​​is: ,in It represents the maximum number of steps.

[0114] The multi-frequency distortion amplitude and phase vector is constructed as follows:

[0115] In the formula, For multi-frequency distortion amplitude and phase vectors, For the first The amplitude distortion value corresponding to each frequency point For the first The phase distortion value corresponding to each frequency point.

[0116] As a further preferred technical solution, step S5: interpolating and fitting the distortion amplitude and phase values ​​at multiple frequency points within the sampling frequency range to calculate the equalizer coefficients for the current step frequency size, specifically including the following steps: S51. For the distortion amplitude and phase values ​​at multiple frequency points, interpolation fitting is performed within the sampling frequency range, and inverse response calculation is performed on the amplitude and phase interpolation data. The frequency coordinates of the interpolation frequency points correspond one-to-one with the frequency points of the fast Fourier transform used to calculate the equalizer coefficients. Specifically, the frequency response of the equalizer is designed to be the inverse response after amplitude-phase distortion interpolation:

[0117] in, This is the inverse response after amplitude and phase distortion interpolation. This represents the number of FFT points after frequency domain interpolation. This is the amplitude distortion value corresponding to the k-th frequency point after interpolation (the amplitude value of the signal is extracted by frequency step sampling, and the amplitude distortion value of the k-th FFT frequency point is calculated by frequency domain interpolation). This is the phase distortion value corresponding to the k-th frequency point after interpolation (the phase value extracted by frequency step sampling of the signal, and the phase distortion value of the k-th FFT frequency point is calculated by frequency domain interpolation).

[0118] Construct the delay compensation phase vector:

[0119] in, The delay compensation phase vector is given by the FIR equalizer filter tap number. The number of delay points introduced is ; Time delay compensation for the inverse response:

[0120] in, It is the frequency domain response vector. This is the inverse response after amplitude and phase distortion interpolation. This indicates element-wise multiplication. This is the delay compensation phase vector.

[0121] S52. Construct the DFT matrix of the inverse response of the amplitude-phase interpolation data based on the inverse response of the amplitude-phase interpolation data; Specifically, construct DFT matrix for:

[0122] Its matrix form is:

[0123] in, K is the number of taps in the FIR equalization filter, and K is the number of FFT points after frequency domain interpolation.

[0124] S53. Based on the system operating bandwidth, sampling frequency, and the number of FFT points after frequency domain interpolation, construct a weighted vector of the inverse response of the amplitude-phase interpolation data; Specifically, in this embodiment, the frequency domain weighting vector is constructed as follows:

[0125] In the formula, For frequency domain weighted vectors, This represents the number of FFT points after frequency domain interpolation. For system operating bandwidth, The sampling frequency; The normalized weighting matrix is ​​calculated based on the frequency domain weighting vector:

[0126] In the formula, For normalized weighted matrices, This indicates that a diagonal matrix is ​​constructed using vector elements.

[0127] S54. Based on the DFT matrix and weighting vector of the amplitude-phase interpolation inverse response, use the weighted least squares method to solve the equalizer coefficients for the current step frequency step size.

[0128] Specifically, the problem of constructing weighted least squares coefficients based on the DFT matrix and weighting vector of the amplitude-phase interpolation inverse response is as follows:

[0129] In the formula, This is the coefficient vector of the time-domain FIR equalization filter. For normalized weighted matrices, The DFT matrix, It is the frequency domain response vector. In order to find the optimal The value should be set to minimize the weighted sum of squared errors; The weighted least squares coefficient problem is solved, and the weighted least squares solution is obtained as follows:

[0130] in, It is the frequency domain response vector; It is the coefficient vector of the time-domain FIR equalization filter; This indicates the conjugate transpose. This represents the number of taps in the FIR equalizer filter. These are the tap coefficient values ​​of the FIR equalizer filter. Therefore, the equalizer coefficients... The calculated weighted least squares solution .

[0131] As a further preferred technical solution, pseudo-inverse is used in actual calculations to ensure numerical stability:

[0132] Store the final equalizer coefficients: To save as a file.

[0133] As a further preferred technical solution, step S6, the process of performing amplitude and phase distortion compensation for dual-mode millimeter-wave radar channels, includes the following steps: When switching to normal operation based on the closed-loop correction selection signal, the operating mode of the dual-mode millimeter-wave radar transceiver system is determined. When in step frequency working mode, select the equalizer coefficient corresponding to the current step frequency interval from the equalizer coefficient table to achieve amplitude and phase distortion compensation of the step frequency channel. When in linear frequency modulation (LFM) mode, the intermediate frequency (IF) is estimated based on prior information, and the equalizer coefficient with the step size closest to the estimated IF is selected from the equalizer coefficient table to achieve amplitude and phase distortion compensation for the LFM channel. The equalizer coefficient table is constructed based on the equalizer coefficients corresponding to all typical step frequency steps.

[0134] It should be noted that this embodiment first configures the transmit and receive parameters and switches to the calibration state; starting from the low-frequency end of the bandwidth, the transmit and local oscillator waveform parameters are adjusted to generate a pair of single-tone calibration signals, which serve as the transmit pre-drive signal and the local oscillator pre-drive signal; after filtering and power amplifying the transmit pre-drive signal, the coupling detection signal is obtained, and the local oscillator pre-drive signal is simultaneously filtered and amplified to generate the RF receive local oscillator signal; the coupling detection signal and the RF receive signal are mixed, filtered, and amplified to generate an analog intermediate frequency (IF) signal; the analog IF signal is converted from analog to digital and the distortion amplitude and phase values ​​are extracted; the frequency of the pair of single-tone calibration signals is controlled to gradually increase according to a preset step size to obtain the distortion amplitude and phase values ​​at multiple frequency points; the distortion amplitude and phase values ​​are interpolated and fitted to calculate the equalizer coefficients; the step frequency is adjusted and the calculation is repeated until all equalizer coefficients are obtained; after switching to normal operating state, the target echo signal is received. This method supports both step frequency and linear frequency modulation modes, has real-time coefficient matching and switching capabilities, and significantly improves the amplitude and phase consistency and system imaging performance under multi-channel conditions.

[0135] As a further preferred technical solution, estimating the intermediate frequency based on prior information when in linear frequency modulation mode specifically includes the following steps: Estimate the timing parameters of the received signal based on known distance or time interval information between transmitted signals; A mapping relationship is established using the linear correspondence between the time parameters and the frequency; Substitute the time parameter into the mapping relationship to calculate the corresponding intermediate frequency deviation or intermediate frequency value.

[0136] It should be noted that when the system switches to normal operating state, the low-noise amplifier signal i1 output from the low-noise amplifier unit is selected as the receiving selection signal i2 and mixed, filtered and amplified with the RF receiving local oscillator signal g1 to generate an analog intermediate frequency signal.

[0137] This embodiment can dynamically select the best-matching compensation coefficient from a pre-generated equalizer coefficient table based on the specific parameters of the current operating mode, such as the step frequency difference and the estimated intermediate frequency value of linear frequency modulation. This mechanism not only achieves precise matching between the compensation parameters and real-time waveform characteristics, but also further optimizes the coefficient selection by combining prior target information. This maintains high-precision channel consistency even in complex operating environments, improves the signal-to-noise ratio limitation caused by amplitude and phase distortion due to phase nonlinearity in existing systems, and enhances the detection stability and imaging quality of the system in dynamic scenes.

[0138] In addition, such as Figure 2As shown, the second embodiment of the present invention also proposes a dual-mode millimeter-wave radar channel amplitude and phase distortion compensation device, the device comprising: a correction processor 1, a transmit waveform generator 2, a transmit assembly 3, a transmit antenna 4, a local oscillator waveform generator 5, a receive assembly 6, a receive antenna 7, and a signal processor 8; The signal processor 8 is connected to the correction processor 1 and is used to output transmit / receive control parameters a to the correction processor 1. The transmit / receive control parameters a include working mode parameters (selection of normal detection mode / internal correction mode, selection of step frequency or linear frequency modulation waveform), transmit link parameters (carrier frequency or start frequency, step frequency step size or frequency modulation frequency) and receive link parameters (receive gain, bandwidth configuration). The correction processor 1 is used to acquire the transmit / receive control parameter a and parse the transmit / receive control parameter a to obtain the transmit waveform parameter b, the local oscillator waveform parameter e and the closed-loop correction selection signal h, and send them to the transmit waveform generator 2, the local oscillator waveform generator 5 and the receiving component 6 respectively. The correction processor 1 is also used to output the compensation echo data l to the signal processor 8. Transmit waveform generator 2 is used to adjust the transmit waveform parameter b starting from the low-frequency end of the system operating bandwidth, generate a transmit pre-drive signal c, and send it to the transmit component 3; Local oscillator waveform generator 5 is used to adjust the local oscillator waveform parameter e starting from the low-frequency end of the system operating bandwidth, generate a local oscillator pre-drive signal g, and send it to the receiving component 6; Transmitting component 3 is used to convert the transmit pre-drive signal c and send it to receiving component 6, so that receiving component 6 can generate an analog intermediate frequency signal k based on the converted transmit pre-drive signal and the received local oscillator pre-drive signal g and send it to correction processor 1; The calibration processor 1 is also used to extract the distortion amplitude and phase values ​​corresponding to the current frequency point from the analog intermediate frequency signal k, and control the frequencies of the transmit pre-drive signal c generated by the transmit waveform generator 2 and the local oscillator pre-drive signal g generated by the local oscillator waveform generator 5 to increase successively according to the preset step frequency step size until the frequency reaches the high-frequency end of the system operating bandwidth to obtain the distortion amplitude and phase values ​​corresponding to multiple frequency points within the system operating bandwidth; and to perform interpolation fitting on the distortion amplitude and phase values ​​of multiple frequency points within the sampling frequency range to calculate the equalizer coefficient of the current step frequency step size, adjust the preset step frequency step size and control the transmit waveform generator 2 to regenerate the transmit pre-drive signal c, and control the local oscillator waveform generator 5 to regenerate the local oscillator pre-drive signal g until the equalizer coefficients corresponding to all typical step frequency step sizes are obtained for use in dual-mode millimeter-wave radar channel amplitude and phase distortion compensation.

[0139] Furthermore, an equalizer coefficient table is constructed based on the equalizer coefficients corresponding to all typical step frequency steps and pre-stored in the equalizer 52 in the local oscillator waveform generator 5. This table is used to receive the target echo signal after switching to normal operation and to perform amplitude and phase distortion compensation for the dual-mode millimeter-wave radar channel based on the pre-generated equalizer coefficient table. As a further preferred technical solution, the correction processor 1 includes: The transmit / receive parameter parsing unit 16 is used to parse the received transmit / receive control parameter a to obtain the serial frame in the transmit / receive control parameter a, extract the transmit waveform parameter b, the local oscillator waveform parameter e, and the closed-loop correction selection signal h in sequence, and send the closed-loop correction selection signal h to the receiving component 6.

[0140] As a further preferred technical solution, such as Figure 3 As shown, the transmitted waveform generator 2 includes: a first direct frequency synthesizer 21, a delay unit 22, a first digital-to-analog converter 23, and a first mixer 24. The first direct frequency synthesizer 21 is connected to the correction processor 1 and the delay unit 22 respectively. It is used to receive the transmission waveform parameter b from the correction processor 1 and output the transmission digital signal b1 to the delay unit 22. Specifically, it maps the transmission waveform parameter to the transmission waveform frequency parameter, and within the range of the system's working bandwidth, it steps the transmission waveform frequency parameter according to the set step size with the low frequency end as the starting frequency of the working bandwidth until the total number of steps is reached. It calculates the transmission waveform frequency corresponding to each frequency step, generates the corresponding transmission digital signal b1 corresponding to each frequency step, and sends it to the delay unit 22. The delay unit 22 is used to delay the transmitted digital signal b1 to generate a transmitted digital delayed signal b2, and send the transmitted digital delayed signal b2 to the first digital-to-analog converter 23; The first digital-to-analog converter 23 is used to convert the transmitted digital delay signal b2 into a transmitted digital-to-analog converter signal b3 and send it to the first mixer 24; The first mixer 24 is used to mix the transmitted digital-to-analog conversion signal b3 with a fixed frequency to obtain a transmitted pre-drive signal c and send it to the transmitting component 3.

[0141] As a further preferred technical solution, such as Figure 4 As shown, the local oscillator waveform generator 5 includes a second direct frequency synthesizer 51, an equalizer 52, a second digital-to-analog converter 53, and a second mixer 54 connected in sequence, wherein: The second direct frequency synthesizer 51 is connected to the correction processor 1 and the equalizer 52. It is used to receive the local oscillator waveform parameter e from the correction processor 1 and output the digital local oscillator signal e1 to the equalizer 52. The specific process is as follows: the local oscillator waveform parameter e is mapped to the local oscillator waveform frequency parameter. Within the range of the system's working bandwidth, the local oscillator waveform frequency parameter is stepped up according to the set step size, starting from the low frequency end as the starting frequency of the working bandwidth, until the total number of steps is reached. The local oscillator waveform frequency corresponding to each frequency step is calculated so as to generate the corresponding digital local oscillator signal e1 according to the local oscillator waveform parameter corresponding to each frequency step and send it to the equalizer 52. The equalizer 52 is used to perform equalization processing on the digital local oscillator signal e1 corresponding to each frequency step, to obtain the equalized local oscillator digital signal e2 and send it to the second digital-to-analog converter 53; the equalizer 52 is also used to receive the equalizer coefficient f sent from the correction processor 1. The second digital-to-analog converter 53 is used to map the equalized local oscillator digital signal e2 to the continuous time domain to obtain the local oscillator digital-to-analog converted signal e3 and send it to the second mixer 54; The second mixer 54 is used to mix the local oscillator digital-to-analog conversion signal e3 with a fixed frequency to obtain the local oscillator pre-drive signal g and send it to the receiving component 6.

[0142] As a further preferred technical solution, such as Figure 5 As shown, the transmitting assembly 3 includes a bandpass filter 31, a power amplifier 32, and a coupler 33, wherein: The bandpass filter 31 is used to receive the transmit pre-drive signal c from the transmit waveform generator 2, and to filter the transmit pre-drive signal c to obtain the transmit filtered signal c1 and send it to the power amplifier 32. The power amplifier 32 is used to amplify the power of the transmitted filtered signal c1 to obtain the amplified transmitted signal c1 and send it to the coupler 33. The coupler 33 is used to couple a portion of the amplified radio frequency transmission signal c1 as a coupling detection signal j to the receiving component 6, and to send the other portion as a transmission signal d to the transmitting antenna 4.

[0143] As a further preferred technical solution, such as Figure 6 As shown, the receiving component 6 includes a low-noise amplifier 61, a switch 62, a mixer amplifier 63, a programmable filter 64, and a filter amplifier 65, wherein: The low-noise amplifier 61 is used to perform low-noise amplification processing on the target echo signal i received from the receiving antenna 7 to obtain a low-noise amplified signal i1 and send it to the switch 62. The switch 62 is used to select, according to the closed-loop correction selection signal h received from the correction processor 1, the low noise amplifier signal i1 output by the low noise amplifier or the partial coupling detection signal j output by the transmitter component 3 as the reception selection signal i2 and send it to the mixer amplifier 63. The mixer amplifier 63 is used to mix and amplify the radio frequency received local oscillator signal g1 sent by the filter amplifier 65 with the receive selection signal i2 to obtain the mixed signal i3 and send it to the programmable filter 64. The filter amplifier 65 is used to filter and amplify the local oscillator pre-drive signal g sent by the local oscillator waveform generator 5 to obtain the radio frequency received local oscillator signal g1 and send it to the mixer amplifier 63. The programmable filter 64 is used to sequentially mix, filter, and amplify the receive selection signal i3 and the radio frequency receive local oscillator signal g1 to generate an analog intermediate frequency signal k.

[0144] It should be noted that switch 62 is located in the receiving component 6 and is used to switch the signal flow path under the control of the correction processor 1: In the calibration state, the switch 62 receives the closed-loop calibration selection signal h from the calibration processor 1 and switches to the calibration working state to select the partial coupling detection signal j output from the transmitting component 3 as the receiving selection signal i2; In normal operation, the switch 62 switches to normal operation mode to select the low noise amplifier signal i1 output from the low noise amplifier 61 as the receive selection signal i2.

[0145] As a further preferred technical solution, such as Figure 7 As shown, the correction processor 1 includes an analog-to-digital converter 11, a correction data acquisition unit 12, an amplitude-phase extraction unit 13, an amplitude-phase interpolation unit 14, an equalization coefficient calculation unit 15, and a transmit / receive parameter parsing unit 16, wherein: The analog-to-digital converter 11 is used to receive the analog intermediate frequency signal k from the receiving component 6, perform analog-to-digital conversion processing on the analog intermediate frequency signal k to obtain the compensated echo data l, and send it to the calibration data acquisition unit 12 and the signal processor 8. The calibration data acquisition unit 12 is used to convert the compensation echo data l into multi-frequency point calibration data k1 and send it to the amplitude-phase conversion unit 13. The amplitude and phase extraction unit 13 is used to perform DFT transformation on the multi-frequency point correction data k1, extract the amplitude and phase values ​​corresponding to the transmission frequency and local oscillator frequency as the multi-frequency point distortion amplitude and phase data k2 corresponding to the current frequency point, and send it to the amplitude and phase interpolation unit 14. The amplitude-phase interpolation unit 14 is used to interpolate and fit the multi-frequency point distorted amplitude-phase data k2 within the sampling frequency range to obtain amplitude-phase interpolation data k3 and send it to the equalization coefficient calculation unit 15. The equalization coefficient calculation unit 15 is used to calculate the equalizer coefficient f based on the amplitude-phase interpolation data k3 and send it to the local oscillator waveform generator 5.

[0146] As a further preferred technical solution, such as Figure 8 As shown, the equalization coefficient calculation unit 15 includes an amplitude-phase inverse response calculation subunit 151, a DFT matrix construction subunit 152, a weighted vector construction subunit 153, and a weighted least squares coefficient solution subunit 154. Amplitude-phase inverse response calculation subunit 151 is used to perform inverse response calculation based on amplitude-phase interpolation data k3, obtain amplitude-phase interpolation inverse response k31 and send it to DFT matrix construction subunit; The DFT matrix construction subunit 152 is used to construct the DFT matrix k32 of the amplitude-phase interpolation data inverse response based on the amplitude-phase interpolation data inverse response k31 and send it to the weighted vector construction subunit 153. The weighted vector construction subunit 153 is used to construct a weighted vector k33 of the inverse response of the amplitude-phase interpolation data based on the system operating bandwidth, sampling frequency and the number of FFT points after frequency domain interpolation, and send it to the weighted least squares coefficient solving subunit 154. The weighted least squares coefficient solving subunit 154 is used to solve the equalizer coefficient f of the current step frequency step size using the weighted least squares method based on the DFT matrix k32 and weighting vector k33 of the amplitude-phase interpolation inverse response and send it to the local oscillator waveform generator 5.

[0147] It should be noted that the correction processor 1 includes two working states: a correction state and a normal working state. In the calibration state, the system constructs a transmit-receive closed-loop calibration architecture, and the calibration processor 1 is configured with two parallel working branches: In the first branch, the transmit / receive parameter parsing unit 16 of the correction processor 1 receives transmit / receive control parameter a from the signal processor 8, and outputs transmit waveform parameter b and local oscillator waveform parameter e after parsing and processing, which are respectively used as input control signals for the transmit waveform generator 2 and the local oscillator waveform generator 5. In the second branch, the analog-to-digital converter of the correction processor 1 receives the analog intermediate frequency signal k from the receiving component 6, and after analog-to-digital conversion, it is transmitted to the correction data acquisition unit. The distortion amplitude and phase values ​​are extracted by the amplitude and phase extraction unit, the interpolation fitting is performed by the amplitude and phase interpolation unit, and the equalization coefficient calculation unit is performed to calculate the equalizer coefficient. Under normal operating conditions, the system adapts to at least two preset operating modes: the transmit / receive parameter parsing unit 16 of the correction processor 1 outputs a closed-loop correction selection signal to the receiving component 6 according to the currently selected operating mode, and calls the equalizer coefficient corresponding to the operating mode to perform real-time equalization compensation processing on the received signal, and finally outputs the compensated echo data.

[0148] As a further preferred technical solution, under normal operating conditions, the transmit / receive parameter parsing unit 16 outputs a closed-loop correction selection signal to the receiving component 6 according to the currently selected operating mode, and calls the equalizer coefficient corresponding to the operating mode to perform real-time equalization compensation processing on the received target echo signal, and outputs the compensated echo data.

[0149] It should be noted that other embodiments or specific implementation methods of the dual-mode millimeter-wave radar channel amplitude and phase distortion compensation device of the present invention can refer to the above-described method embodiments, and will not be repeated here.

[0150] Furthermore, the third embodiment of the present invention also proposes a computer-readable storage medium storing a computer program thereon, which, when executed by a processor, implements the dual-mode millimeter-wave radar channel amplitude and phase distortion compensation method as described in the first embodiment above.

[0151] Furthermore, the fourth embodiment of the present invention also proposes a millimeter-wave radar transceiver system, which uses the dual-mode millimeter-wave radar channel amplitude and phase distortion compensation method as described in the first embodiment above, or is equipped with the dual-mode millimeter-wave radar channel amplitude and phase distortion compensation device as described in the second embodiment.

[0152] It should be noted that the dual-mode millimeter-wave radar channel amplitude and phase distortion compensation device proposed in this embodiment can be applied to a dual-mode millimeter-wave radar transceiver system. It only requires that the hardware of the dual-mode millimeter-wave radar transceiver system supports the compensation algorithm proposed in this invention, that is, there must be a coupling branch, and the computing resources of the system must support the calculation of the compensation algorithm. In this way, this invention can be ported to the software platform of the dual-mode millimeter-wave radar transceiver system.

[0153] In addition, the dual-mode millimeter-wave radar channel amplitude and phase distortion compensation device proposed in this embodiment can also be applied to stepped-frequency radar and conventional radar based on deskewing system.

[0154] It should be noted that the computer-readable medium disclosed in this embodiment may be a computer-readable signal medium or a computer-readable storage medium, or any combination thereof. A computer-readable storage medium may be, for example,—but not limited to—an electrical, magnetic, optical, electromagnetic, infrared, or semiconductor system, apparatus, or device, or any combination thereof. More specific examples of a computer-readable storage medium may include, but are not limited to: an electrical connection having one or more wires, a portable computer disk, a hard disk, random access memory (RAM), read-only memory (ROM), erasable programmable read-only memory (EPROM or flash memory), optical fiber, and portable compact disk read-only memory (CD-ROM). ROM, optical storage devices, magnetic storage devices, or any suitable combination thereof. In this disclosure, a computer-readable storage medium can be any tangible medium containing or storing a program that can be used by or in connection with an instruction execution system, apparatus, or device. In this disclosure, a computer-readable signal medium can include a data signal propagated in baseband or as part of a carrier wave, carrying computer-readable program code. Such propagated data signals can take various forms, including but not limited to electromagnetic signals, optical signals, or any suitable combination thereof. A computer-readable signal medium can also be any computer-readable medium other than a computer-readable storage medium, which can send, propagate, or transmit a program for use by or in connection with an instruction execution system, apparatus, or device. The program code contained on the computer-readable medium can be transmitted using any suitable medium, including but not limited to: wires, optical fibers, RF (radio frequency), etc., or any suitable combination thereof.

[0155] The aforementioned computer-readable medium may be included in the aforementioned electronic device; or it may exist independently and not assembled into the electronic device. The aforementioned computer-readable medium carries one or more programs, which, when executed by the electronic device, cause the electronic device to perform a zero-sample image anomaly detection method according to the above embodiments.

[0156] Computer program code for performing the operations of this disclosure can be written in one or more programming languages ​​or a combination thereof, including object-oriented programming languages ​​such as Java, Smalltalk, and C++, and conventional procedural programming languages ​​such as the "C" language or similar programming languages. The program code can be executed entirely on the user's computer, partially on the user's computer, as a standalone software package, partially on the user's computer and partially on a remote computer, or entirely on a remote computer or server.

[0157] In cases involving remote computers, the remote computer can be connected to the user's computer via any type of network—including a local area network (LAN) or a wide area network (WAN), or it can be connected to an external computer (e.g., via the Internet using an Internet service provider).

[0158] It should be understood that various parts of the present invention can be implemented in hardware, software, firmware, or a combination thereof. In the above embodiments, multiple steps or methods can be implemented in software or firmware stored in memory and executed by a suitable instruction execution system. For example, if implemented in hardware, as in another embodiment, it can be implemented using any one or a combination of the following techniques known in the art: discrete logic circuits having logic gates for implementing logical functions on data signals, application-specific integrated circuits (ASICs) having suitable combinational logic gates, programmable gate arrays (FPGAs), field-programmable gate arrays (FPGAs), etc.

[0159] In the description of this specification, references to terms such as "one embodiment," "some embodiments," "example," "specific example," or "some examples," etc., indicate that a specific feature, structure, material, or characteristic described in connection with that embodiment or example is included in at least one embodiment or example of the invention. In this specification, the illustrative expressions of the above terms do not necessarily refer to the same embodiment or example. Furthermore, the specific features, structures, materials, or characteristics described may be combined in any suitable manner in one or more embodiments or examples.

[0160] Furthermore, the terms "first" and "second" are used for descriptive purposes only and should not be construed as indicating or implying relative importance or implicitly specifying the number of indicated technical features. Thus, a feature defined as "first" or "second" may explicitly or implicitly include at least one of that feature. In the description of this invention, "a plurality of" or "several" means at least two, such as two, three, etc., unless otherwise explicitly specified.

[0161] Although embodiments of the present invention have been shown and described above, it is understood that the above embodiments are exemplary and should not be construed as limiting the present invention. Those skilled in the art can make changes, modifications, substitutions and variations to the above embodiments within the scope of the present invention.

Claims

1. A method for compensating for amplitude and phase distortion in a dual-mode millimeter-wave radar channel, characterized in that, include: S1. Obtain the transmit and receive control parameters and parse the transmit and receive control parameters to obtain the transmit waveform parameters and the local oscillator waveform parameters; S2. Starting from the low-frequency end of the system's operating bandwidth, adjust the transmit waveform parameters and the local oscillator waveform parameters to generate paired single-tone correction signals, which are then used as the transmit pre-drive signal and the local oscillator pre-drive signal, respectively. S3. Generate an analog intermediate frequency signal based on the transmit pre-drive signal and the local oscillator pre-drive signal, and extract the distortion amplitude and phase values ​​corresponding to the current frequency point from the analog intermediate frequency signal; S4. Control the frequency of the paired single-tone correction signals to increase gradually according to the preset step frequency step size, and repeat steps S2 to S3 until the frequency reaches the high-frequency end of the system working bandwidth, and obtain the distortion amplitude and phase values ​​corresponding to multiple frequency points within the system working bandwidth. S5. For the distortion amplitude and phase values ​​of multiple frequency points, perform interpolation fitting within the sampling frequency range to calculate the equalizer coefficient of the current step frequency step size. S6. Adjust the preset step frequency step size and repeat steps S2 to S5 until the equalizer coefficients corresponding to all typical step frequency step sizes are obtained, which can be used for amplitude and phase distortion compensation of dual-mode millimeter-wave radar channels.

2. The dual-mode millimeter-wave radar channel amplitude and phase distortion compensation method as described in claim 1, characterized in that, The step of acquiring and parsing the transmit / receive control parameters to obtain the transmit waveform parameters and the local oscillator waveform parameters further includes: The transmit and receive control parameters are analyzed to obtain the serial frame in the transmit and receive control parameters, and the transmit waveform parameters, local oscillator waveform parameters and closed-loop correction selection signal are extracted in sequence. The closed-loop calibration selection signal is used to control the switching of the signal flow path between the calibration state and the normal operation state.

3. The dual-mode millimeter-wave radar channel amplitude and phase distortion compensation method as described in claim 1, characterized in that, Starting from the low-frequency end of the system's operating bandwidth, adjusting the transmit waveform parameters and local oscillator waveform parameters to generate paired single-tone correction signals, which are then used as the transmit pre-drive signal and the local oscillator pre-drive signal, respectively, includes: The transmitted waveform parameters and the local oscillator waveform parameters are mapped to the transmitted waveform frequency parameters and the local oscillator waveform frequency parameters, respectively. Within the system's operating bandwidth, using the low-frequency end as the starting frequency of the operating bandwidth, the frequency parameters of the transmitted waveform and the local oscillator waveform are stepped according to the set step size until the total number of steps is reached. The transmitted waveform frequency and the local oscillator waveform frequency corresponding to each frequency step are calculated. The corresponding digital signal is generated based on the frequency of the transmitted waveform corresponding to each frequency step, and the corresponding digital local oscillator signal is generated based on the local oscillator waveform parameters corresponding to each frequency step. The transmitted digital signal corresponding to each frequency step is delayed to generate a transmitted digital delay signal. The transmitted digital delay signal is then mapped to a continuous time domain and mixed with a fixed frequency to obtain the transmitted pre-drive signal. The digital local oscillator signal corresponding to each frequency step is equalized to obtain the equalized local oscillator digital signal. The equalized local oscillator digital signal is then mapped to the continuous time domain and mixed with a fixed frequency to obtain the local oscillator pre-drive signal.

4. The dual-mode millimeter-wave radar channel amplitude and phase distortion compensation method as described in claim 3, characterized in that, The step involves delaying the transmitted digital signal corresponding to each frequency step to generate a transmitted digital delayed signal. This transmitted digital delayed signal is then mapped to a continuous time domain and mixed with a fixed frequency to obtain the transmitted pre-drive signal, including: For each frequency step, the transmitted digital signal is subjected to channel digital path delay processing to generate a transmitted digital delay signal, expressed by the formula: In the formula, To transmit digital delay signals, To transmit digital signals, is the base of the natural logarithm. The imaginary unit represents the channel digital path delay. , This represents the number of taps in the FIR equalizer filter. For the first The frequency of the transmitted waveform corresponding to the next frequency step. Sampling frequency, The number of sampling points for transmitting digital signals; The transmitted digital delay signal is converted from digital to analog to map it to the continuous time domain, and then mixed with a fixed frequency to obtain the transmitted pre-drive signal, as expressed by the formula: In the formula, To transmit the pre-drive signal, To transmit the pre-drive signal amplitude, is the base of the natural logarithm. The imaginary unit represents the channel digital path delay. , This represents the number of taps in the FIR equalizer filter. For the first The frequency of the transmitted waveform corresponding to the next frequency step. For the fixed frequency of mixing, For time scale.

5. The dual-mode millimeter-wave radar channel amplitude and phase distortion compensation method as described in claim 3, characterized in that, The process of equalizing the digital local oscillator signal corresponding to each frequency step to obtain an equalized local oscillator digital signal, and then mapping the equalized local oscillator digital signal to a continuous time domain and mixing it with a fixed frequency to obtain the local oscillator pre-drive signal, includes: The digital local oscillator signal corresponding to each frequency step is equalized to obtain the equalized digital local oscillator signal, as expressed by the formula: In the formula, The local oscillator digital signal after equalization. It is a digital local oscillator signal. The transfer function of the pre-calibrated unit equalizer. For discrete convolution symbols, This represents the number of taps in the FIR equalizer filter. These are the tap coefficients of the FIR equalizer filter. The original local oscillator digital signal, <1 hour Take the value of zero; 1 o'clock Normally, the value of the local oscillator digital signal is taken. ; The equalized local oscillator digital signal is converted from digital to analog and mapped to the continuous time domain. Then, it is mixed with a fixed frequency to obtain the local oscillator pre-drive signal, which is expressed by the following formula: In the formula, This is the local oscillator pre-drive signal. The amplitude of the local oscillator pre-drive signal. is the base of the natural logarithm. The imaginary unit, For the first The local oscillator waveform frequency corresponding to the next frequency step. Sampling frequency, For the fixed frequency of mixing, For time scale.

6. The dual-mode millimeter-wave radar channel amplitude and phase distortion compensation method as described in claim 5, characterized in that, The transfer function exist The expression for the domain is: In the formula, It satisfies unit equilibrium.

7. The dual-mode millimeter-wave radar channel amplitude and phase distortion compensation method as described in claim 1 or 3, characterized in that, The transmit pre-drive signal and the local oscillator pre-drive signal are a pair of single-tone signals with a fixed frequency difference, which is equal to a preset step frequency step size.

8. The dual-mode millimeter-wave radar channel amplitude and phase distortion compensation method as described in claim 2, characterized in that, The step of generating an analog intermediate frequency (IF) signal based on the transmit pre-drive signal and the local oscillator pre-drive signal, and extracting the distortion amplitude and phase values ​​corresponding to the current frequency point from the analog IF signal, includes: After filtering and power amplification of the pre-drive signal, a portion of the amplified radio frequency transmission signal is coupled as a coupling detection signal, and the other portion is sent to the transmission antenna as a transmission signal. After filtering and power amplification of the local oscillator pre-drive signal, a radio frequency receiving local oscillator signal is generated, wherein the digital transmission path delay corresponding to the transmit pre-drive signal and the local oscillator pre-drive signal is the same; Based on the closed-loop correction selection signal, the coupled detection signal is used as the receiving selection signal and is sequentially mixed, filtered and amplified with the radio frequency receiving local oscillator signal to generate an analog intermediate frequency signal. After performing analog-to-digital conversion on the analog intermediate frequency signal, the distortion amplitude and phase values ​​corresponding to the current frequency point are extracted.

9. The dual-mode millimeter-wave radar channel amplitude and phase distortion compensation method as described in claim 8, characterized in that, The step of extracting the distortion amplitude and phase values ​​corresponding to the current frequency point after performing analog-to-digital conversion on the analog intermediate frequency signal includes: The analog intermediate frequency signal is processed by analog-to-digital conversion to obtain compensated echo data, and multi-frequency correction data of the compensated echo signal is extracted within the working bandwidth. The formula for single-point frequency data is expressed as: In the formula, It is a single-point frequency signal. The amplitude value of the mixed signal. The phase difference after mixing. For frequency difference, For phase, is the base of the natural logarithm. The imaginary unit, The time-domain sampling point number; Perform DFT transformation on the multi-frequency correction data, and extract the amplitude and phase values ​​corresponding to the transmission frequency and local oscillator frequency as the distortion amplitude and phase values ​​corresponding to the current frequency.

10. The dual-mode millimeter-wave radar channel amplitude and phase distortion compensation method as described in claim 1, characterized in that, The equalizer coefficients for the current step frequency size are calculated by interpolating and fitting the distortion amplitude and phase values ​​at multiple frequency points within the sampling frequency range, including: For the distortion amplitude and phase values ​​at multiple frequency points, interpolation fitting is performed within the sampling frequency range, and inverse response calculation is performed on the amplitude and phase interpolation data. The frequency coordinates of the interpolation frequency points correspond one-to-one with the frequency points of the fast Fourier transform used to calculate the equalizer coefficients. Construct the DFT matrix of the inverse response of the amplitude-phase interpolation data based on the inverse response of the amplitude-phase interpolation data; Based on the system's operating bandwidth, sampling frequency, and the number of FFT points after frequency domain interpolation, a weighted vector of the inverse response of the amplitude-phase interpolation data is constructed. Based on the DFT matrix and weighting vector of the amplitude-phase interpolation inverse response, the equalizer coefficients for the current step frequency step size are solved using the weighted least squares method.

11. The dual-mode millimeter-wave radar channel amplitude and phase distortion compensation method as described in claim 10, characterized in that, The process of interpolating and fitting the distortion amplitude and phase values ​​at multiple frequency points within the sampling frequency range, and then performing inverse response calculations on the interpolated amplitude and phase data, includes: For the distortion amplitude and phase values ​​at multiple frequency points, interpolation fitting is performed within the sampling frequency range, and the equalizer frequency response is designed to be the inverse response of the amplitude and phase interpolation data: In the formula, This is the inverse response after amplitude and phase distortion interpolation. This represents the number of frequency points in the FFT after frequency domain interpolation. This represents the amplitude distortion value corresponding to the k-th FFT frequency point. This represents the phase distortion value corresponding to the k-th FFT frequency point; The frequency domain response vector is obtained by performing time delay compensation on the inverse response of the amplitude-phase interpolation data, as expressed by the formula: In the formula, It is the frequency domain response vector. This is the inverse response after amplitude and phase distortion interpolation. This indicates element-wise multiplication. This is the time delay compensation phase vector. , This refers to the number of delay points introduced.

12. The dual-mode millimeter-wave radar channel amplitude and phase distortion compensation method as described in claim 10, characterized in that, The DFT matrix of the amplitude-phase interpolation data inverse response, constructed based on the amplitude-phase interpolation data inverse response, is expressed by the following formula: In the formula, The DFT matrix, This represents the number of taps in the FIR equalizer filter. This represents the number of frequency points in the FFT after frequency domain interpolation. For FFT frequency point number, This refers to the tap number of the FIR equalizer filter.

13. The dual-mode millimeter-wave radar channel amplitude and phase distortion compensation method as described in claim 10, characterized in that, The weighted vector is calculated using an amplitude-weighted method, where the in-band weighted amplitude of the equalizer's frequency response compensation is higher than that outside the band.

14. The dual-mode millimeter-wave radar channel amplitude and phase distortion compensation method as described in claim 10, characterized in that, The step of constructing a weighted vector for the inverse response of the amplitude-phase interpolated data based on the system operating bandwidth, sampling frequency, and the number of FFT points after frequency domain interpolation includes: Based on the system's operating bandwidth, sampling frequency, and the number of FFT points after frequency domain interpolation, the frequency domain weighted vector is constructed as follows: In the formula, For frequency domain weighted vectors, This represents the number of FFT points after frequency domain interpolation. For system operating bandwidth, The sampling frequency; The normalized weighting matrix is ​​calculated based on the frequency domain weighting vector: In the formula, For normalized weighted matrices, This indicates that a diagonal matrix is ​​constructed using vector elements.

15. The dual-mode millimeter-wave radar channel amplitude and phase distortion compensation method as described in claim 10, characterized in that, The equalizer coefficients for the current step frequency step size are solved using the weighted least squares method based on the DFT matrix and weighted vector of the amplitude-phase interpolation inverse response, including: The problem of constructing weighted least squares coefficients based on the DFT matrix and weighting vector of the amplitude-phase interpolation inverse response is as follows: In the formula, This is the coefficient vector of the time-domain FIR equalization filter. For normalized weighted matrices, The DFT matrix, It is the frequency domain response vector. To find the optimal The value should be set to minimize the weighted sum of squared errors; Solve the problem of weighted least squares coefficients, and calculate the weighted least squares solution. As the equalizer coefficient for the current step frequency step size This represents the number of taps in the FIR equalizer filter. These are the tap coefficient values ​​of the FIR equalizer filter.

16. The dual-mode millimeter-wave radar channel amplitude and phase distortion compensation method as described in claim 2, characterized in that, The process of performing amplitude and phase distortion compensation for dual-mode millimeter-wave radar channels includes: When switching to normal operation based on the closed-loop correction selection signal, the operating mode of the dual-mode millimeter-wave radar transceiver system is determined. When in step frequency working mode, select the equalizer coefficient corresponding to the current step frequency interval from the equalizer coefficient table to achieve amplitude and phase distortion compensation of the step frequency channel. When in linear frequency modulation (LFM) mode, the intermediate frequency (IF) is estimated based on prior information, and the equalizer coefficient with the step size closest to the estimated IF is selected from the equalizer coefficient table to achieve amplitude and phase distortion compensation for the LFM channel. The equalizer coefficient table is constructed based on the equalizer coefficients corresponding to all typical step frequency steps.

17. The dual-mode millimeter-wave radar channel amplitude and phase distortion compensation method as described in claim 16, characterized in that, When in linear frequency modulation (LFM) mode, estimating the intermediate frequency (IF) based on prior information includes: Estimate the timing parameters of the received signal based on known distance or time interval information between transmitted signals; A mapping relationship is established using the linear correspondence between the time parameters and the frequency; Substitute the time parameter into the mapping relationship to calculate the corresponding intermediate frequency deviation or intermediate frequency value.

18. A dual-mode millimeter-wave radar channel amplitude and phase distortion compensation device, characterized in that, include: The calibration processor is used to acquire and parse the transmit and receive control parameters to obtain the transmit waveform parameters and the local oscillator waveform parameters, and send them to the transmit waveform generator and the local oscillator waveform generator respectively. The transmit waveform generator is used to adjust the transmit waveform parameters starting from the low-frequency end of the system's operating bandwidth, generate a transmit pre-drive signal, and send it to the transmit component. The local oscillator waveform generator is used to adjust the local oscillator waveform parameters starting from the low-frequency end of the system's operating bandwidth, generate a local oscillator pre-drive signal, and send it to the receiving component. The transmitting component is used to convert the transmit pre-drive signal and send it to the receiving component, so that the receiving component can generate an analog intermediate frequency signal based on the converted transmit pre-drive signal and the received local oscillator pre-drive signal and send it to the correction processor; The calibration processor is also used to extract the distortion amplitude and phase values ​​corresponding to the current frequency point from the analog intermediate frequency signal, and control the frequencies of the transmit pre-drive signal generated by the transmit waveform generator and the local oscillator pre-drive signal generated by the local oscillator waveform generator to increase successively according to the preset step frequency step size until the frequency reaches the high-frequency end of the system operating bandwidth to obtain the distortion amplitude and phase values ​​corresponding to multiple frequency points within the system operating bandwidth; and to perform interpolation fitting on the distortion amplitude and phase values ​​of multiple frequency points within the sampling frequency range to calculate the equalizer coefficient of the current step frequency step size, adjust the preset step frequency step size and control the transmit waveform generator to regenerate the transmit pre-drive signal, control the local oscillator waveform generator to regenerate the local oscillator pre-drive signal until the equalizer coefficients corresponding to all typical step frequency step sizes are obtained for use in dual-mode millimeter-wave radar channel amplitude and phase distortion compensation.

19. The dual-mode millimeter-wave radar channel amplitude and phase distortion compensation device as described in claim 18, characterized in that, The correction processor is connected to the signal processor, which is used to configure the transmit / receive control parameters. The transmit / receive control parameters include operating mode parameters, transmit link parameters, and receive link parameters.

20. The dual-mode millimeter-wave radar channel amplitude and phase distortion compensation device as described in claim 18, characterized in that, The correction processor includes: The transmit / receive parameter parsing unit is used to parse the received transmit / receive control parameters to obtain the serial frame in the transmit / receive control parameters, extract the transmit waveform parameters, local oscillator waveform parameters and closed-loop correction selection signal in sequence, and send the closed-loop correction selection signal to the receiving component.

21. The dual-mode millimeter-wave radar channel amplitude and phase distortion compensation device as described in claim 18, characterized in that, The transmitted waveform generator includes: The first direct frequency synthesizer is used to receive the transmit waveform parameters sent from the correction processor, map the transmit waveform parameters to transmit waveform frequency parameters, and within the system operating bandwidth, use the low frequency end as the starting frequency of the operating bandwidth to perform frequency stepping on the transmit waveform frequency parameters according to the set step size until the total number of steps is reached, calculate the transmit waveform frequency corresponding to each frequency step, generate the corresponding transmit digital signal from the transmit waveform frequency corresponding to each frequency step, and send it to the delay unit. The delay unit is used to perform delay processing on the transmitted digital signal to generate a transmitted digital delayed signal, and to send the transmitted digital delayed signal to the first digital-to-analog converter; The first digital-to-analog converter is used to convert the transmitted digital delay signal into a transmitted digital-to-analog converted signal and send it to the first mixer; The first mixer is used to mix the transmitted digital-to-analog conversion signal with a fixed frequency to obtain a transmitted pre-drive signal and send it to the transmitting component.

22. The dual-mode millimeter-wave radar channel amplitude and phase distortion compensation device as described in claim 18, characterized in that, The local oscillator waveform generator includes: The second direct frequency synthesizer is used to receive the local oscillator waveform parameters sent from the correction processor and map the local oscillator waveform parameters to local oscillator waveform frequency parameters. Within the system operating bandwidth, the local oscillator waveform frequency parameters are stepped up according to a set step size, starting from the low-frequency end of the operating bandwidth, until the total number of steps is reached. The local oscillator waveform frequency corresponding to each frequency step is calculated so as to generate the corresponding digital local oscillator signal according to the local oscillator waveform parameters corresponding to each frequency step and send it to the equalizer. The equalizer is used to perform equalization processing on the digital local oscillator signal corresponding to each frequency step, obtain the equalized local oscillator digital signal, and send it to the second digital-to-analog converter. The second digital-to-analog converter is used to map the equalized local oscillator digital signal to the continuous time domain to obtain the local oscillator digital-to-analog converted signal and send it to the second mixer; The second mixer is used to mix the local oscillator digital-to-analog conversion signal with a fixed frequency to obtain the local oscillator pre-drive signal and send it to the receiving component.

23. The dual-mode millimeter-wave radar channel amplitude and phase distortion compensation device as described in claim 21, characterized in that, The transmitting component includes: A bandpass filter is used to receive the transmit pre-drive signal from the transmit waveform generator, filter the transmit pre-drive signal to obtain the transmit filtered signal, and send it to the power amplifier. The power amplifier is used to amplify the power of the transmitted filtered signal to obtain the amplified transmitted signal and send it to the coupler. The coupler is used to couple a portion of the amplified radio frequency transmitted signal as a coupling detection signal to the receiving component, and to send the other portion as a transmitted signal to the transmitting antenna.

24. The dual-mode millimeter-wave radar channel amplitude and phase distortion compensation device as described in claim 23, characterized in that, The receiving component includes: A low-noise amplifier is used to amplify the target echo signal received from the receiving antenna in a low-noise manner, obtain a low-noise amplified signal, and send it to the switch. The switch is used to select, based on the closed-loop correction selection signal received from the correction processor, the low-noise amplifier signal output by the low-noise amplifier or the partially coupled detection signal output by the transmitting component as the receiving selection signal and send it to the mixer amplifier. The mixer amplifier is used to mix and amplify the radio frequency received local oscillator signal sent by the filter amplifier with the receive selection signal to obtain a mixed signal and send it to the programmable filter. The filter amplifier is used to filter and amplify the local oscillator pre-drive signal sent by the local oscillator waveform generator to obtain the radio frequency received local oscillator signal and send it to the mixer amplifier. The programmable filter is used to sequentially mix, filter, and amplify the received selection signal and the radio frequency received local oscillator signal to generate an analog intermediate frequency signal.

25. The dual-mode millimeter-wave radar channel amplitude and phase distortion compensation device as described in claim 18, characterized in that, The correction processor includes: An analog-to-digital converter is used to receive an analog intermediate frequency signal from a receiving component, perform analog-to-digital conversion on the analog intermediate frequency signal to obtain compensated echo data, and send it to a calibration data acquisition unit and a signal processor. The calibration data acquisition unit is used to convert the compensated echo data into multi-frequency calibration data and send it to the amplitude-phase conversion unit; The amplitude and phase extraction unit is used to perform DFT transformation on the multi-frequency correction data, extract the amplitude and phase values ​​corresponding to the transmission frequency and local oscillator frequency as the multi-frequency distortion amplitude and phase data corresponding to the current frequency point, and send them to the amplitude and phase interpolation unit. The amplitude-phase interpolation unit is used to perform interpolation fitting on multi-frequency distorted amplitude-phase data within the sampling frequency range to obtain amplitude-phase interpolation data and send it to the equalization coefficient calculation unit. The equalization coefficient calculation unit is used to calculate the equalizer coefficients based on amplitude-phase interpolation data and send them to the local oscillator waveform generator.

26. The dual-mode millimeter-wave radar channel amplitude and phase distortion compensation device as described in claim 25, characterized in that, The equilibrium coefficient calculation unit includes: The amplitude-phase inverse response calculation subunit is used to perform inverse response calculation based on amplitude-phase interpolation data, obtain the amplitude-phase interpolation inverse response, and send it to the DFT matrix construction subunit. The DFT matrix construction subunit is used to construct the DFT matrix of the inverse response of the amplitude-phase interpolation data based on the inverse response of the amplitude-phase interpolation data and send it to the weighted vector construction subunit. The weighted vector construction subunit is used to construct a weighted vector of the inverse response of the amplitude-phase interpolation data based on the system operating bandwidth, sampling frequency and the number of FFT points after frequency domain interpolation, and send it to the weighted least squares coefficient solving subunit. The weighted least squares coefficient solving subunit is used to solve the equalizer coefficients of the current step frequency step size using the weighted least squares method based on the DFT matrix and weighting vector of the amplitude-phase interpolation inverse response, and then send them to the local oscillator waveform generator.

27. The dual-mode millimeter-wave radar channel amplitude and phase distortion compensation device as described in claim 20, characterized in that, Under normal operating conditions, the transmit / receive parameter parsing unit outputs a closed-loop correction selection signal to the receiving component according to the currently selected operating mode, and calls the pre-stored equalizer coefficient table in the equalizer of the local oscillator waveform generator to obtain the equalizer coefficients corresponding to the operating mode, performs real-time equalization compensation processing on the received target echo signal, and outputs the compensated echo data.

28. A computer-readable storage medium having a computer program stored thereon, characterized in that, When the computer program is executed by the processor, it implements the dual-mode millimeter-wave radar channel amplitude and phase distortion compensation method as described in any one of claims 1 to 17.

29. A millimeter-wave radar transceiver system, characterized in that, The dual-mode millimeter-wave radar channel amplitude and phase distortion compensation method as described in any one of claims 1 to 17 is used, or a dual-mode millimeter-wave radar channel amplitude and phase distortion compensation device as described in any one of claims 18 to 27 is arranged.

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