A Ku-band low-profile flat panel slot antenna based on tight coupling aperture radiation

Abstract

This invention discloses a Ku-band low-profile planar slotted antenna array based on tightly coupled aperture radiation, belonging to the field of antenna array technology. It includes: an antenna body with an internal cavity structure, comprising at least one tightly coupled radiating element; an aperture radiating array, formed by the internal cavity structure of the antenna body, comprising aperture radiating elements and a power divider, the aperture radiating elements and the power divider being respectively located at the top and bottom of the antenna body; and the power divider. This invention employs an architecture combining tightly coupled aperture radiating elements with an H-plane waveguide feed network. Through tight coupling, capacitive-inductive resonance cancellation is achieved, enabling lossless transmission of electromagnetic energy between the cavity and the radiating aperture, thus improving aperture radiation efficiency. Simultaneously, the H-plane T-shaped branch structure compresses the longitudinal dimension, combined with plastic electroplating technology, reducing the overall thickness and weight compared to traditional waveguide horn arrays.

Description

Technical Field

[0001] This invention relates to the field of antenna array technology, and in particular to a Ku-band low-profile planar slot antenna array based on tightly coupled aperture radiation. Background Technology

[0002] With the rapid development of modern radar and communication systems, the requirements for antennas in communication systems are gradually moving towards miniaturization and high efficiency. Flat panel antennas, due to their advantages of small size, high efficiency, low profile, and light weight, perfectly meet the general needs of vehicle-mounted or airborne communication systems. Therefore, the development of flat panel antennas has become a focus in the antenna field. Flat panel antennas employ array antenna technology, integrating dozens, hundreds, or even thousands of antenna elements onto a single flat panel to achieve high gain. In satellite communication, to avoid adjacent satellite interference, certain requirements are usually placed on the sidelobe levels of the antenna pattern, especially the first sidelobe. Simultaneously, to meet certain gain requirements and improve the efficiency of electromagnetic energy radiation, the electromagnetic field at the aperture of the flat panel antenna needs to satisfy certain amplitude and phase distributions. Array antennas can achieve these required amplitude and phase distributions by designing their feed network, and then radiating the electromagnetic field with the required distribution outwards through the radiating array, thereby realizing the desired radiation pattern.

[0003] An array antenna consists of two parts: the design of the array elements and the design of the feeding network. The array elements must be selected according to specific needs. For example, traveling-wave antennas such as helical antennas and tapered slot antennas have wide frequency bands and high gain, but they are large in electrical size, have high profiles, and low aperture efficiency. Microstrip antennas are also a commonly used array antenna element. Compared with other types of antennas, microstrip antennas have certain structural advantages: low profile, small size, light weight, easy integration, easy conformal design, low cost, and can be integrated into the system design. Furthermore, microstrip antennas also have many advantages in physical performance, such as ease of implementing multiple polarization modes and the ability to operate at dual or multiple frequencies. Of course, microstrip antennas also have some disadvantages, the most important being their narrow operating bandwidth; typically, the impedance bandwidth of a microstrip antenna in resonant state is only a few percent. The surface waves generated on the surface of the microstrip antenna, along with conductor and dielectric losses, result in low radiation efficiency. In addition, due to the limitation of the dielectric substrate thickness, the voltage applied to the antenna cannot be too high, resulting in low power capacity for microstrip antennas. For microstrip transmission lines and antennas, dispersion loss becomes increasingly severe with increasing frequency, making them unsuitable for millimeter-wave operation. Waveguide slot antennas utilize slots on the waveguide surface, radiating by perturbing the surface current. The waveguide itself can synthesize the energy of the slot elements, and with low transmission loss, waveguide slot antennas offer high radiation efficiency. Based on the principles of array synthesis, low sidelobes can be achieved by weighting the amplitude and phase of the slot excitation signal, making them widely used in radar systems. However, in higher millimeter-wave bands, mutual coupling exists between slots in the waveguide slot array, significantly impacting the radiation pattern. While algorithms can be designed to reduce this mutual coupling, these algorithms are complex and require sophisticated waveguide fabrication.

[0004] Aperture-type antennas generally have relatively high radiation efficiency. When antenna height is limited, open waveguides and horn antennas can be selected. An open waveguide radiates electromagnetic waves through an opening at its end. It comes in various shapes, including rectangular, circular, and elliptical, with circular and rectangular apertures being the most widely used. However, the opening at the waveguide end causes the generation of higher-order modes, resulting in significant reflections and poor matching performance. Furthermore, the small size of the waveguide end opening leads to lower radiation efficiency. In practical applications, open waveguides are often modified to gradually increase the size of the waveguide end opening, thus forming horn antennas. These antennas have wide operating bandwidth, good matching performance, and strong directivity, meeting the design requirements for high gain.

[0005] Horn arrays offer advantages such as wide bandwidth, high efficiency, and low loss; however, the horn's profile is relatively high and thick, resulting in significant weight. Currently, to reduce the horn array's profile height, a hybrid feed network combining printed circuitry and waveguides is typically used. However, this complex structure cannot avoid transmission losses from the waveguide to the printed circuitry, leading to a decrease in the array antenna's radiation efficiency. E-plane waveguide power dividers have a smaller dimension in the array direction, allowing direct feeding of tightly spaced elements. However, their larger longitudinal dimension results in a thicker array. While non-standard waveguides can be used in the feed network design to reduce the longitudinal dimension to some extent, the effect remains limited and is not conducive to installation and use on vehicle-mounted or mobile platforms. Summary of the Invention

[0006] The purpose of this invention is to provide a Ku-band low-profile planar slot antenna array based on tightly coupled aperture radiation to solve the problems mentioned in the background art.

[0007] To achieve the above objectives, the present invention provides the following technical solution: a Ku-band low-profile planar slot antenna array based on tightly coupled aperture radiation, comprising:

[0008] The antenna body has an internal cavity structure, and the antenna body includes at least one tightly coupled radiating element;

[0009] The aperture radiation array is composed of the internal cavity structure of the antenna body. The aperture radiation array includes aperture radiation elements and power dividers. The aperture radiation elements and power dividers are respectively opened at the top and bottom of the antenna body, and multiple aperture radiation elements are connected to the top of the power divider.

[0010] The power divider adopts an H-plane T-shaped branch structure, which achieves longitudinal profile compression through narrow side branching of waveguides and high-density integration of planar layout through non-standard waveguides.

[0011] The composite network frame is used to array multiple antenna bodies, with non-equiphase output to shorten the transmission path and reduce loss, and to achieve equal amplitude and in-phase distribution of the array aperture field through the geometric flip arrangement of the antenna bodies.

[0012] Preferably, the aperture radiation unit includes a cavity, a radiation aperture, and a coupling port. The coupling port is used to connect the cavity to the power divider. The cavity and the radiation aperture are tightly coupled together. The coupling between the two is enhanced by adjusting the size of the cavity and the radiation aperture, so that capacitive-inductive resonance cancellation is achieved to realize lossless transmission of electromagnetic energy and broadband impedance matching. The tightly coupled connection specifically includes:

[0013] The equivalent inductance of the cavity and the equivalent capacitance of the radiation aperture form an LC resonant circuit. By adjusting the coupling coefficient, the capacitive reactance cancels the inductive reactance, thereby achieving a purely resistive input impedance, reducing reflection loss and improving radiation efficiency.

[0014] Preferably, the cavity supports TE02 mode, in which the radiation aperture is located at the position of strongest electromagnetic energy in the cavity. By adjusting the relative position and size between the radiation aperture and the cavity, the coupling is optimized, and the mode conversion from TE02 mode to TE10 mode is completed and radiated into free space.

[0015] Preferably, in TE02 mode, the tightly coupled radiation unit is equivalent to one divided into four, and the energy fed into the coupling port is evenly distributed on the four radiation aperture surfaces.

[0016] Preferably, the radiating aperture is rectangular and supports the TE10 mode, with the TE10 mode being the main mode. The long side of the rectangle is greater than or equal to half the cutoff wavelength of the main mode transmission to ensure the transmission of the TE10 main mode. The short side of the rectangle is matched with the free space impedance. By selecting an appropriate short side size, the coupling of energy to free space is enhanced, thereby optimizing the efficiency of radiation.

[0017] Preferably, the H-plane T-shaped branch structure makes the branch waveguide parallel to the magnetic field in the main waveguide, so as to achieve compression of the longitudinal dimension and suppress the increase of the overall thickness when the back-end synthesized network is a multi-layer structure, while maintaining the low loss characteristics of waveguide transmission.

[0018] Preferably, the non-standard waveguide is a flat waveguide, with its wide side dimension being greater than or equal to half of the cutoff wavelength of the main mode transmission to ensure the main mode transmission, and its narrow side dimension meeting the structural strength requirements from the perspective of engineering feasibility. This ensures electrical performance while avoiding excessively thin cavity walls, thereby achieving high-density integration of planar layout and reducing the difficulty of processing.

[0019] Preferably, the power divider includes a first matching step and a second matching step. The first matching step extends horizontally, and the second matching step extends vertically. The first matching step is used to adjust the power distribution ratio to achieve equal power distribution, so that electromagnetic energy is evenly distributed to the left and right arms. The second matching step is used for total port impedance matching to reduce reflected waves and decouple the power distribution and impedance matching functions.

[0020] Preferably, the power divider further includes an input port and a coupling output port. The input port is configured to receive external electromagnetic energy and feed it into the power divider. The coupling output port is used to output the energy distributed by the power divider to the coupling port to transmit electromagnetic energy into the cavity.

[0021] Preferably, the synthesized network frame has an output port with a preset phase difference, which is achieved by optimizing the transmission line path layout. The antenna body is arranged alternately in the forward and reverse directions, and the preset phase difference is automatically compensated by geometric flipping, so that the phase of the array aperture field is in phase and the maximum value points to zero degrees. The synthesized network frame achieves equal phase and unequal amplitude power distribution through cascaded power dividers. The unequal amplitude is weighted according to Taylor distribution to reduce the sidelobe level and meet the network access requirements.

[0022] The technical effects and advantages of this invention are as follows:

[0023] (1) The present invention adopts an architecture that combines a tightly coupled aperture radiation unit with an H-plane waveguide feed network. The capacitor-inductor resonance cancellation is achieved through tight coupling connection, so that electromagnetic energy is transmitted without loss between the cavity and the radiation aperture, and the aperture radiation efficiency is improved. At the same time, the longitudinal dimension is compressed by using the H-plane T-shaped branch structure and combined with the plastic electroplating process, the relative density of metal materials is reduced and the weight is reduced. This is superior to the narrow-band low efficiency characteristics of microstrip arrays, and meets the stringent requirements of vehicle and airborne platforms for low profile, lightweight and high efficiency.

[0024] (2) The present invention adopts the non-phase output of the synthetic network frame (preset 0° / 180° phase difference) combined with the geometric flip arrangement of the antenna body (alternating installation in the forward / reverse direction), and uses the structural mirror to automatically compensate for the phase difference. Without the need for additional electrically adjustable phase shifters or complex mutual coupling calculation algorithms, the equal amplitude and in-phase distribution of the array aperture field is realized. At the same time, through the cascade of multi-stage H-plane power dividers, the power distribution ratio is independently adjusted by the first matching step to realize Taylor distribution amplitude weighting, so that the sidelobe level decreases, which meets the network access requirements of satellite communication system for suppressing adjacent satellite interference, and reduces system complexity and manufacturing cost.

[0025] (3) The present invention adopts a design with a radiating aperture size much smaller than that of the traditional horn, which makes the space between the radiating aperture units larger, eliminates the thin-walled structure, provides enough space for the mechanical structure to arrange large-diameter fastening screws, and significantly increases the structural connection strength; at the same time, a non-standard flat waveguide is adopted, and by reasonably setting the narrow side size (to meet the structural strength requirements from the perspective of engineering feasibility), the cavity wall is avoided to be too thin, which reduces the risk of processing deformation and manufacturing difficulty, improves the product yield, and realizes the high-density integration and mass production feasibility of planar layout. Attached Figure Description

[0026] The accompanying drawings are provided to further illustrate the invention and form part of the specification. They are used together with the embodiments of the invention to explain the invention, but do not constitute a limitation thereof. In the drawings:

[0027] Figure 1 This is a schematic diagram of the overall structure of the antenna array of the present invention;

[0028] Figure 2 This is a schematic diagram of the antenna structure of the present invention;

[0029] Figure 3 This is one of the schematic diagrams of the aperture radiation array structure of the present invention;

[0030] Figure 4 This is a second schematic diagram of the porous radiation array structure of the present invention;

[0031] Figure 5 This is a schematic diagram of the power divider structure of the present invention;

[0032] Figure 6 This is a three-dimensional structural diagram of the radiation aperture of the present invention;

[0033] Figure 7 This is a top view of the structure at the radial aperture of the present invention;

[0034] Figure 8 This is a schematic diagram of the front structure at the coupling port of the present invention;

[0035] Figure 9 This is a partial structural diagram of the power divider of the present invention;

[0036] Figure 10 This is a schematic diagram of the structure of the radiation aperture of the present invention on the tightly coupled radiation unit;

[0037] Figure 11 This is a schematic diagram of the coupling port of the present invention on the tightly coupled radiation unit;

[0038] Figure 12 This is a schematic diagram of the structure of the synthetic network frame of the present invention;

[0039] Figure 13 The figure shows the simulation results of the standing wave ratio of this invention;

[0040] Figure 14 The image shows the simulation results of the radiation pattern at 14.5 GHz according to the present invention.

[0041] Figure 15 The image shows the simulation results of the radiation pattern at 14.925 GHz according to the present invention.

[0042] Figure 16 The above is a simulation result of the radiation pattern at 15.35 GHz according to the present invention.

[0043] Figure 17 This is a graph showing the VSWR test results of the present invention;

[0044] Figure 18 This is the 14.5GHz radiation pattern of the present invention;

[0045] Figure 19This is the 14.925GHz radiation pattern of the present invention;

[0046] Figure 20 This is the 15.35GHz radiation pattern of the present invention.

[0047] In the attached diagram: 100, antenna body; 101, tightly coupled radiating element; 200, aperture radiating array; 201, aperture radiating element; 211, cavity; 212, radiating aperture; 213, coupling port; 202, power divider; 221, first matching step; 222, second matching step; 223, input port; 224, coupled output port; 300, synthesizing network frame. Detailed Implementation

[0048] The technical solutions of the embodiments of the present invention will be clearly and completely described below with reference to the accompanying drawings. Obviously, the described embodiments are only some embodiments of the present invention, and not all embodiments. Based on the embodiments of the present invention, all other embodiments obtained by those skilled in the art without creative effort are within the scope of protection of the present invention.

[0049] Example 1: Ku-band low-profile planar slot antenna array based on tightly coupled aperture radiation

[0050] Reference Figures 1 to 12 This embodiment discloses a Ku-band low-profile planar slotted antenna array based on tightly coupled aperture radiation, suitable for vehicle-mounted or airborne satellite communication systems. The antenna array operates in the Ku-band (14.5-15.35 GHz) and has a thickness of only 22.5 mm, which is significantly thinner and lighter than traditional waveguide horn arrays. While achieving a low profile and lightweight design, it also meets the stringent electrical performance requirements of high gain and low sidelobes.

[0051] Overall structural configuration and assembly relationship:

[0052] like Figure 1 and Figure 2 As shown, the antenna array in this embodiment adopts a hierarchical cascaded architecture, including an antenna body 100, an aperture radiation array 200, and a composite network frame 300.

[0053] The antenna body 100 serves as the basic supporting structure and includes several tightly coupled radiating elements 101. The antenna body 100 is a solid shell structure composed of a plastic substrate and a surface electroplated layer, with an internal cavity structure forming an aperture radiating array 200. The aperture radiating array 200 is essentially an internal cavity enclosed by the solid walls of the antenna body 100, including the aperture radiating elements 201 and a power divider 202. The aperture radiating elements 201 are located at the top of the antenna body 100, and the power divider 202 is located at the bottom of the antenna body 100, forming a vertical energy transmission channel from bottom to top. Four aperture radiating elements 201 are connected to the top of the power divider 202.

[0054] The synthesizing network frame 300 is located on the back of the antenna body 100 and is used to array multiple antenna bodies 100 to achieve array-level power combining and distribution.

[0055] Lossless transmission and broadband matching mechanism of tightly coupled radiating element 101:

[0056] like Figure 3 , Figure 6 , Figure 10 and Figure 11 As shown, the aperture radiating unit 201 includes a cavity 211, several radiating apertures 212, and a coupling port 213. The coupling port 213 is used to connect the cavity 211 and the power divider 202 to achieve energy communication between the cavity 211 and the power divider 202. The size of the coupling port 213 is much smaller than that of a conventional Ku-band waveguide port to achieve tight coupling between the cavity 211 and the power divider 202. Unlike traditional horn radiating units, the size of the radiating apertures 212 in this embodiment is much smaller than that of traditional horn apertures, resulting in a larger space between the aperture radiating units 201. This eliminates the thin-walled structure between traditional horn units, reducing manufacturing difficulty and providing sufficient mechanical space to arrange larger diameter fastening screws (such as M2.5 or M3 specifications), significantly increasing structural connection strength and ensuring the integrity of the electrical structure.

[0057] The cavity 211 and the radiating aperture 212 are connected by tight coupling. The core of this tight coupling connection lies in constructing a capacitor-inductor resonance cancellation mechanism to achieve lossless transmission of electromagnetic energy and broadband impedance matching.

[0058] Specifically, energy is transferred from the power divider 202 to the cavity 211 via coupling port 213. The cavity 211 supports TE02 mode, and its electric field distribution is characterized by the weakest energy along the long side at the center of the cavity 211, with the electric field symmetrically distributed around the short side's central axis, forming two symmetrical energy concentration regions. The radiation aperture 212 is precisely positioned at the non-centrosymmetric location of the cavity 211 where the electromagnetic energy is strongest, i.e., at the energy peaks on both sides deviating from the short side's central axis. The strongest coupling is achieved by fine-tuning the relative position between the radiation aperture 212 and the cavity 211 (e.g., within ±0.3mm of offset along the long side) and the size of the radiation aperture 212.

[0059] From the perspective of equivalent circuit analysis, cavity 211 and radiating aperture 212 can be equivalent to two coupled LC resonant circuits. Cavity 211 is equivalent to having an equivalent inductance. The resonant circuit, with radiating aperture 212 equivalent to a resonant circuit with equivalent capacitance C, is tightly coupled through mutual inductance M, and the coupling coefficient is... When the value of K is large (usually K≥0.5), a tight coupling state is formed. At this time, the capacitive reactance of the strong coupling capacitance C forced by the radial aperture 212... The inductive reactance of the equivalent inductance L of cavity 211 is exactly offset. ( ), making the input impedance It exhibits pure resistive characteristics, with an extremely low standing wave ratio (VSWR) (measured <1.2), almost 100% energy transmission, and near-zero reflection loss, thus achieving lossless transmission of electromagnetic energy.

[0060] Furthermore, by adjusting the dimensions of cavity 211 and radiation aperture 212 and their relative positions, the coupling coefficient K is adjusted to achieve the dual resonant frequency of the coupled system. and When they move closer together, they form a W-shaped resonance characteristic, which reduces the natural frequency of each individual resonator. The narrowband characteristics are extended to a wideband response, making the antenna impedance bandwidth greater than 14% in the Ku band (14.5-15.35GHz), thus achieving wideband impedance matching.

[0061] Mode conversion and equivalent power distribution mechanism:

[0062] like Figure 6As shown, the tight coupling configuration enables mode conversion from TE02 mode to TE10 mode. Specifically, cavity 211 supports TE02 mode, with its electric field symmetrically distributed around the short side central axis, forming electromagnetic energy concentration regions on both sides off-center. Radiation aperture 212 is located at the position of strongest electromagnetic energy in cavity 211, supporting TE10 mode. Through tight coupling, the electromagnetic energy of TE02 mode is coupled to radiation aperture 212 and radiated into free space in TE10 mode, achieving uniform energy distribution and efficient radiation.

[0063] Meanwhile, the tightly coupled radiation unit 101 is functionally equivalent to a one-to-four power divider 202. Specifically, the cavity 211 itself can be equivalent to a one-to-two power divider 202, which divides the energy into two paths; and the tightly coupled connection further divides each path of energy into two radiation apertures 212, thereby evenly distributing the total energy fed into the coupling port 213 to the apertures of the four radiation apertures 212, achieving a uniform distribution of the aperture field and maximizing radiation efficiency.

[0064] Dual-function dimensional design of radial aperture 212:

[0065] like Figure 2 , Figure 6 and Figure 7 As shown, the radial aperture 212 has a rectangular structure, and its long and short sides have different functional design objectives.

[0066] The long side dimension of the radiation aperture 212 is set to be greater than or equal to half of the cutoff wavelength of the dominant mode transmission (≥ ),in The wavelength (approximately 20.7 mm) corresponding to the lowest frequency (14.5 GHz) at which the waveguide can transmit in the dominant TE10 mode is set to ≥10.35 mm to ensure the transmission of the dominant TE10 mode and suppress the generation of higher-order modes.

[0067] The short side dimension of the radiating aperture 212 is related to the free space impedance matching. By selecting an appropriate short side dimension (usually set to 1 / 3 to 1 / 2 of the long side dimension), the equivalent impedance of the radiating aperture 212 is adjusted to be close to the free space wave impedance (377Ω), thereby enhancing the impedance matching between the antenna and free space, allowing more energy to be coupled into free space, and thus achieving the highest efficiency radiation.

[0068] Profile compression and functional decoupling of H-plane waveguide feed networks:

[0069] like Figure 4 , Figure 5 , Figure 8 and Figure 9As shown, the power divider 202 adopts an H-plane T-type branch structure. This structure is configured so that the branch waveguide is parallel to the magnetic field in the main waveguide, thereby achieving longitudinal dimension compression.

[0070] Specifically, such as Figure 4 As shown, the branching structure of the E-plane T-type splitter (ET) is on the wide side of the waveguide, parallel to the main waveguide magnetic field, and its longitudinal thickness is approximately 1.5-2 times the width of the wide side of the waveguide. In contrast, the branching structure of the H-plane T-type splitter (HT) is on the narrow side of the waveguide, parallel to the main waveguide magnetic field, and its longitudinal thickness is only the width of the narrow side of the waveguide, approximately half the thickness of the ET splitter. Therefore, choosing the H-plane T-type branching structure can reduce the longitudinal profile height, making it particularly suitable for multi-layered back-end synthesizing networks, effectively suppressing the increase in overall thickness while maintaining the low-loss characteristics of waveguide transmission.

[0071] like Figure 5 and Figure 9 As shown, the power divider 202 adopts a non-standard flat waveguide design. Compared to the standard waveguide BJ140 (wide side dimension 15.799mm × narrow side dimension 7.899mm), this embodiment uses a non-standard flat waveguide, whose wide side dimension is set to ≥ (≥10.35mm) to ensure the transmission of the main mode TE10, while being much smaller than the standard waveguide width, thereby reducing the planar layout size; its narrow side size is set to ≥1.0mm from the perspective of engineering feasibility to ensure the adhesion strength of the electroplated layer and the structural rigidity, avoiding the processing deformation and yield reduction caused by excessively thin cavity walls, thus achieving high-density integration of the planar layout and reducing the processing difficulty.

[0072] Functional decoupling design of dual-matching steps:

[0073] like Figure 5 and Figure 9 As shown, the power divider 202 includes a first matching step 221 and a second matching step 222, which realizes the functional decoupling of power distribution and impedance matching.

[0074] The first matching step 221 extends horizontally (along the wide side of the waveguide), and its step height and width are configured to adjust the power distribution ratio. By precisely designing the geometric parameters of the first matching step 221, such as setting the step height to 1 / 4 to 1 / 3 of the narrow side dimension, the equivalent impedance of the branch waveguide is adjusted to achieve equal power distribution (1:1) or unequal power distribution, such as the power ratio required by the Taylor distribution, so that electromagnetic energy is evenly distributed to the left and right arms according to a preset ratio.

[0075] The second matching step 222 extends vertically (along the waveguide height direction) and is located approximately [distance from the total port]. At a point one-quarter of the waveguide wavelength, the step height is configured to achieve overall port impedance matching. Through the impedance transformation effect of the second matching step 222, the input impedance of the overall port is matched to the standard waveguide impedance, reducing reflected waves at the overall port, lowering reflection loss, and improving transmission efficiency.

[0076] Through the synergistic effect of the first matching step 221 and the second matching step 222, the power distribution function and the impedance matching function are independent of each other and can be optimized separately, avoiding the functional coupling and performance trade-offs in traditional single-step designs.

[0077] Input / output port configuration:

[0078] like Figure 9 As shown, the power divider 202 also includes an input port 223 and a coupled output port 224. The input port 223 is located at one end of the power divider 202 and is configured to receive external electromagnetic energy, such as from a power combining network or the pre-stage power divider 202, and feed it into the main waveguide cavity of the power divider 202. The coupled output port 224 is located at the top of the power divider 202 and is configured to output the energy distributed by the power divider 202 to the coupling port 213, thereby transmitting electromagnetic energy into the cavity 211 to form a complete feed link.

[0079] Array-level power combining and phase self-compensation mechanism:

[0080] like Figure 12 As shown, the composite network frame 300 is used to array multiple antenna bodies 100. Considering the long transmission line path of the composite network frame 300, a non-equiphase output design is adopted to reduce losses.

[0081] Specifically, the composite network frame 300 has output ports with a preset phase difference, and the phase distribution from top to bottom is 0°, 180°, 0°, 180°, 180°. This preset phase difference is achieved by optimizing the transmission line path layout: the port with a phase of 180° is placed in the shorter transmission path, and the port with a phase of 0° is placed in the longer transmission path, so that the actual total length of the transmission line for each port is slightly reduced compared to the total length of the transmission line when outputting with equal phase, thereby reducing path loss.

[0082] To achieve equal-amplitude and in-phase distribution of the array aperture field, the antenna body 100 is arranged alternately in forward and reverse orientations, utilizing geometric flipping to automatically compensate for the preset phase difference. The specific arrangement rules are as follows: from top to bottom, forward installation, reverse installation, forward installation, reverse installation, and reverse installation. In forward installation, the long side of the radiating aperture 212 is along the X-axis, and the feed port is located on the left. In reverse installation, the long side of the radiating aperture 212 is along the X-axis, but the feed port is located on the right (or rotated 180° around the Z-axis). This geometric flipping is equivalent to introducing a 180° phase inversion, which precisely cancels the 180° phase difference output by the synthesizing network frame 300, ensuring that the array aperture is in phase and that the maximum radiation pattern value points to zero degrees. Phase self-compensation can be achieved without additional electrically adjustable phase shifters or phase-shifting circuits, simplifying the system structure and reducing cost and power consumption.

[0083] Furthermore, the composite network frame 300 is configured to achieve equal-phase but unequal-amplitude power distribution through cascaded multi-stage H-plane power dividers 202. Each stage of the power divider 202 achieves different power division ratios by adjusting the step parameters of the first matching step 221, and performs amplitude weighting according to a Taylor distribution, thereby effectively reducing sidelobe levels and meeting the network access requirements of satellite communication systems for suppressing adjacent satellite interference.

[0084] Lightweight plastic electroplating process:

[0085] like Figure 6 and Figure 7 As shown, to achieve lightweight design, the cavity 211 of the antenna body 100 is made using a plastic electroplating process. The plastic substrate, as the structural support, has a lower manufacturing density compared to metal, resulting in a lighter overall weight. This avoids the bulkiness and heaviness issues of traditional all-metal waveguides.

[0086] Example 2: Multi-stage Cascaded Array

[0087] Based on the single antenna body 100 of Embodiment 1, larger-scale arrays can be further constructed. For example... Figure 12 As shown, 20 antenna bodies 100 are arranged in a 4×5 array and cascaded through a power combining network frame 300 to form the complete unit. The thickness of the entire unit is only 22.5mm (including the power combining network), and the weight is about 13.7kg. In the 14.5-15.35GHz frequency band, the gain is ≥42.58dBi and the sidelobe level is ≤-17dB, achieving synergistic optimization of high gain, low sidelobe, low profile, and lightweight design.

[0088] To verify the design effectiveness of the tightly coupled radiating element 101 and the H-plane waveguide feed network, electromagnetic simulation software was used to simulate and analyze the antenna performance.

[0089] Reference Figure 13The standing wave ratio (SWR) of the aperture radiating array 200 was simulated. The simulation results show that in the Ku band, the impedance bandwidth with an antenna SWR of less than 1.2 is greater than 14%, which verifies that the dual resonant frequency (W-shaped resonant characteristic) formed by the tight coupling connection between the cavity 211 and the radiating aperture 212 effectively expands the operating bandwidth. At the same time, the functional decoupling design of the first matching step 221 and the second matching step 222 achieves good impedance matching.

[0090] Reference Figures 14 to 16 Far-field radiation patterns were simulated at three frequency points: 14.5 GHz, 14.925 GHz, and 15.35 GHz. Simulation results show that the antenna pattern has a symmetrical main lobe, sidelobe levels below -13 dB, and uniform gain distribution. This verifies the uniform aperture field distribution characteristics achieved by tight coupling and the effectiveness of the short-side dimension matching design of the radiating aperture 212 with free-space impedance. Furthermore, the simulated aperture radiation efficiency exceeds 85%, demonstrating the lossless electromagnetic energy transmission effect achieved by the LC resonant cancellation mechanism.

[0091] Table 1 Summary of Test Results

[0092]

[0093] To further verify the overall performance, a physical prototype was fabricated based on the structural parameters of Example 1 and tested in a microwave anechoic chamber. The prototype measures 259mm × 246mm × 12mm and consists of 20 antenna bodies 100 arranged in a 4×5 array, cascaded together by a composite network frame 300 to form the complete unit. The overall thickness is 22.5mm, and the array weight is 13.7kg.

[0094] Reference Figure 17 A vector network analyzer was used to test the standing wave ratio (VSWR) of the entire system's input ports. The test results showed that within the operating frequency band of 14.5-15.35 GHz, the overall VSWR was less than or equal to 1.4, meeting the performance requirements. This result verifies the overall port impedance matching effect achieved by the combined network frame 300 through the cascaded power divider 202, and the good matching performance between the tightly coupled radiating unit 101 and the combined network frame 300.

[0095] Reference Figures 18 to 20 As shown in Table 1, the far-field radiation pattern of the entire device was tested in a microwave anechoic chamber. The test conditions were as follows: a standard gain horn was used as the reference antenna, the antenna under test was placed at the center of the turntable, and the rotation angle was ±90°. The gain and sidelobe level at each frequency were recorded.

[0096] Test results show that:

[0097] At the 14.5GHz frequency point (refer to) Figure 18(and Table 1): The antenna gain is 42.58 dBi, the sidelobe level is -17.06 dB, the main lobe width is narrow, and the directivity is strong;

[0098] At the 14.925GHz frequency point (refer to) Figure 19 (and Table 1): The antenna gain is 42.65 dBi, the sidelobe level is -16.9 dB, and the gain flatness is good;

[0099] At the 15.35GHz frequency point (refer to) Figure 20 (and Table 1): The antenna gain is 42.7 dBi, the sidelobe level is -17.5 dB, and the high-frequency band performance is stable.

[0100] The above test data verifies the following technical effects of the present invention:

[0101] High radiation efficiency verification: The measured gain is ≥42.58dBi. Combined with the array size and theoretical calculation, the aperture radiation efficiency reaches more than 85%, which is consistent with the simulation results. This verifies the effectiveness of the tight coupling connection in achieving lossless transmission of electromagnetic energy through LC resonance cancellation, as well as the correctness of the design of the short side dimension of the 212 radiation aperture and the free space impedance matching.

[0102] Low sidelobe characteristics verification: The measured sidelobe level is ≤-17.06dB, which is significantly better than the theoretical limit of -13dB (uniformly distributed array). This verifies the effectiveness of the synthetic network frame 300 in achieving Taylor distribution weighting through the cascade of multi-stage H-plane power dividers 202, as well as the guarantee of array pattern quality by the phase self-compensation mechanism (non-equiphase output and geometric flipping coordination).

[0103] Broadband matching characteristic verification: In the 14.5-15.35GHz frequency band, the VSWR is ≤1.4, which verifies the design of broadband impedance matching (W-shaped resonance) achieved by forming dual resonant frequencies through tight coupling connection, as well as the total port impedance matching function of the second matching step 222.

[0104] Low profile and lightweight verification: The overall thickness is 22.5mm, which is about 64% lower than that of traditional waveguide horn arrays and about 50% lighter, while maintaining the above electrical performance indicators. This verifies the synergistic effect of the H-plane T-shaped branch structure profile compression function and the lightweight design of plastic electroplating process.

[0105] In summary, the simulation and test data of this embodiment fully demonstrate that the Ku-band low-profile planar slot antenna array based on tightly coupled aperture radiation achieves excellent electrical performance with high gain (≥42.58dBi), low sidelobes (≤-17dB), and wideband matching (VSWR≤1.4, bandwidth>14%) while achieving low profile (thickness 22.5mm) and lightweight (weight reduction of 50%). It is suitable for the lightweight and low-profile deployment requirements of vehicle-mounted and airborne satellite communication systems.

[0106] Although embodiments of the invention have been shown and described, it will be understood by those skilled in the art that various changes, modifications, substitutions and alterations can be made to these embodiments without departing from the principles and spirit of the invention, the scope of which is defined by the appended claims and their equivalents.

Claims

1. A Ku-band low profile panel slot antenna array based on tight coupling aperture radiation, characterized in that, include: The antenna body (100) has an internal cavity structure, and the antenna body (100) includes at least one tightly coupled radiating element (101). The aperture radiation array (200) is formed by the internal cavity structure of the antenna body (100). The aperture radiation array (200) includes aperture radiation elements (201) and power dividers (202). The aperture radiation elements (201) and power dividers (202) are respectively opened at the top and bottom of the antenna body (100). Multiple aperture radiation elements (201) are connected to the top of the power dividers (202). The power divider (202) adopts an H-plane T-shaped branch structure, which achieves compression of the longitudinal profile through narrow side branching of the waveguide, and achieves high-density integration of planar layout through non-standard waveguides; The composite network frame (300) is used to array multiple antenna bodies (100), with non-equiphase output to shorten the transmission path and reduce loss, and to achieve equal amplitude and in-phase distribution of the array aperture field through the geometric flip arrangement of the antenna bodies (100).

2. The Ku-band low-profile planar slot antenna array based on tight coupling aperture radiation according to claim 1, characterized in that, The aperture radiation unit (201) includes a cavity (211), a radiation aperture (212), and a coupling port (213). The coupling port (213) is used to connect the cavity (211) and the power divider (202). The cavity (211) and the radiation aperture (212) are tightly coupled. The coupling between the two is enhanced by adjusting the size of the cavity (211) and the radiation aperture (212), so that the capacitor-inductor resonance cancels out, realizing lossless transmission of electromagnetic energy and broadband impedance matching. The tightly coupled connection specifically includes: The equivalent inductance of the cavity (211) and the equivalent capacitance of the radiation aperture (212) form an LC resonant circuit. By adjusting the coupling coefficient, the capacitive reactance cancels the inductive reactance, thereby achieving pure resistance of the input impedance, reducing reflection loss and improving radiation efficiency.

3. The Ku-band low-profile planar slot antenna array based on tight coupling aperture radiation according to claim 2, characterized in that, The cavity (211) supports the TE02 mode, in which the radiation aperture (212) is located at the position of the strongest electromagnetic energy in the cavity (211). By adjusting the relative position and size between the radiation aperture (212) and the cavity (211), the coupling is optimized, and the mode conversion from the TE02 mode to the TE10 mode is completed and radiated into free space.

4. The Ku-band low-profile planar slot antenna array based on tightly coupled aperture radiation according to claim 3, characterized in that, In TE02 mode, the tightly coupled radiation unit (101) is equivalent to being divided into four parts, and the energy fed into the coupling port (213) is evenly distributed on the four radiation apertures (212).

5. The Ku-band low-profile planar slot antenna array based on tightly coupled aperture radiation according to claim 3, characterized in that, The radiation aperture (212) is rectangular and supports the TE10 mode, which is the main mode. The long side of the rectangle is greater than or equal to half of the cutoff wavelength of the main mode transmission to ensure the transmission of the TE10 main mode. The short side of the rectangle is matched with the free space impedance. By selecting an appropriate short side size, the coupling of energy to free space is enhanced, thereby optimizing the efficiency of radiation.

6. The Ku-band low-profile planar slot antenna array based on tightly coupled aperture radiation according to claim 1, characterized in that, The H-plane T-shaped branch structure makes the branch waveguide parallel to the magnetic field in the main waveguide, thereby compressing the longitudinal dimension and suppressing the increase in overall thickness when the back-end synthesized network is a multi-layer structure, while maintaining the low loss characteristics of waveguide transmission.

7. The Ku-band low-profile planar slot antenna array based on tightly coupled aperture radiation according to claim 2, characterized in that, The non-standard waveguide is a flat waveguide, with its wide side dimension being greater than or equal to half of the cutoff wavelength of the main mode transmission to ensure the main mode transmission. Its narrow side dimension meets the structural strength requirements from the perspective of engineering feasibility. While ensuring electrical performance, it avoids excessively thin cavity walls, thereby achieving high-density integration of planar layout and reducing the difficulty of processing.

8. The Ku-band low-profile planar slot antenna array based on tightly coupled aperture radiation according to claim 7, characterized in that, The power divider (202) includes a first matching step (221) and a second matching step (222). The first matching step (221) extends horizontally, and the second matching step (222) extends vertically. The first matching step (221) is used to adjust the power distribution ratio to achieve equal power distribution, so that electromagnetic energy is evenly distributed to the left and right arms. The second matching step (222) is used for total port impedance matching to reduce reflected waves and decouple the power distribution and impedance matching functions.

9. A Ku-band low-profile planar slot antenna array based on tightly coupled aperture radiation according to claim 8, characterized in that, The power divider (202) also includes an input port (223) and a coupling output port (224). The input port (223) is configured to receive external electromagnetic energy and feed it into the power divider (202). The coupling output port (224) is used to output the energy distributed by the power divider (202) to the coupling port (213) to transmit electromagnetic energy into the cavity (211).

10. The Ku-band low-profile planar slot antenna array based on tightly coupled aperture radiation according to claim 1, characterized in that, The synthesized network frame (300) has an output port with a preset phase difference. The preset phase difference is achieved by optimizing the transmission line path layout. The antenna body (100) is arranged alternately in the forward and reverse directions. The preset phase difference is automatically compensated by geometric flipping, so that the phase of the array aperture field is in phase and the maximum value points to zero degrees. The synthesized network frame (300) is cascaded with power dividers (202) to achieve equal phase and unequal amplitude power distribution. The unequal amplitude is weighted according to Taylor distribution to reduce the sidelobe level to meet the network access requirements.

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