Transformers, switching power supplies, and air conditioning equipment

Abstract

This application discloses a transformer, a switching power supply, and an air conditioning device. By setting a first shielding layer between the magnetic core and the primary winding, and electrically connecting the first shielding layer to the primary ground, the magnetic core potential is clamped to the primary ground potential, thereby eliminating the displacement current of the magnetic core to ground. An auxiliary winding is set between the primary winding and the secondary winding. The corresponding terminals of the auxiliary winding are opposite to those of the primary winding. The auxiliary winding is connected to the primary ground. The auxiliary winding includes Nb turns of coil, the primary winding includes Np turns of coil, and the secondary winding includes Ns turns of coil, where 2Nb≤3Ns-Np. In this way, a reverse displacement current can be actively created inside the transformer to offset the original interference current, thereby significantly reducing the net current flowing to ground.

Description

Technical Field

[0001] This application relates to the field of power supply technology, and in particular to a transformer, a switching power supply, and an air conditioning device. Background Technology

[0002] In the design of switching power supplies for household appliances such as air conditioners, to meet electromagnetic compatibility requirements, related technologies typically connect a Y capacitor in parallel between the primary and secondary windings of the transformer to provide a low-impedance loop for common-mode interference currents. However, connecting a Y capacitor in parallel has some drawbacks: First, it has poor surge protection capability; high voltage generated by lightning strikes or other events can easily couple from the primary winding to the secondary winding through the Y capacitor, damaging the main control chip. Second, it reduces electrical isolation, potentially generating a high induced voltage between the secondary winding and ground, posing a safety hazard.

[0003] However, if the Y capacitor is removed, in order to compensate for the deterioration in EMC performance caused by the removal of the Y capacitor, it is often necessary to add a bulky and expensive filter device to make the power supply EMC qualified, which greatly increases the size and cost of the transformer and power supply. Summary of the Invention

[0004] This application aims to at least address one of the technical problems existing in the prior art. To this end, some embodiments of this application propose a transformer, a switching power supply, and an air conditioning device, designed to improve the EMC performance of the transformer.

[0005] To address the aforementioned technical problems, some embodiments of this application provide a transformer, the transformer comprising: A magnetic core, a primary winding, and a secondary winding, wherein the primary winding and the secondary winding are disposed on the magnetic core; A first shielding layer is disposed between the magnetic core and the primary winding and electrically connected to the primary ground, for clamping the potential of the magnetic core to the potential of the primary ground; An auxiliary winding is disposed between the primary winding and the secondary winding. The corresponding terminal of the auxiliary winding is opposite to the corresponding terminal of the primary winding. The first terminal of the auxiliary winding is suspended, and the second terminal of the auxiliary winding is electrically connected to the primary ground. Wherein, the number of turns in the auxiliary winding is Nb, the number of turns in the primary winding is Np, the number of turns in the secondary winding is Ns, and 2Nb≤|3Ns-Np|.

[0006] In power supply solutions of related technologies, to meet electromagnetic compatibility (EMC) requirements, a Y-capacitor is typically connected in parallel between the primary and secondary windings of the transformer to provide a low-impedance loop for common-mode interference currents. However, the introduction of the Y-capacitor leads to a decrease in the power supply's anti-interference capability and reduced electrical isolation. Removing the Y-capacitor results in common-mode interference caused by the core-to-ground displacement current Icg and the primary winding-to-secondary winding-to-ground current Isg. Furthermore, the current Isg can be decomposed into the primary winding-to-secondary winding displacement current Ips and the secondary winding-to-ground current, thus causing EMC problems.

[0007] To compensate for the deteriorated EMC performance caused by removing the Y capacitor, some embodiments of this application provide a transformer scheme for removing the Y capacitor. On one hand, by setting a first shielding layer between the magnetic core and the primary winding and electrically connecting the first shielding layer to the primary ground, the magnetic core potential is clamped to the primary ground potential to eliminate the displacement current Icg of the magnetic core to ground. On the other hand, an auxiliary winding is set between the primary winding and the secondary winding, and the corresponding terminal of the auxiliary winding is set to be opposite to the corresponding terminal of the primary winding. The auxiliary winding is connected to the primary ground. By setting the number of turns of the auxiliary winding to Nb, the induced voltage Vb on the auxiliary winding is adjusted so that the compensation current Ibs flowing from the secondary winding to the auxiliary winding is opposite to the direction of the original displacement current Ips flowing from the primary winding to the secondary winding. By setting 2Nb≤|3Ns-Np|, when the number of turns of the auxiliary winding is slightly less than the set number of turns (|(3Ns-Np) / 2|), it is in an undercompensated state. At this point, the compensation current Ibs is slightly smaller than the native current Ips, which can effectively suppress conducted interference (CE) while having a positive impact on passing radiated emission (RE) tests.

[0008] In some embodiments, 3Nb≥|(10Ns-3Np) / 7|.

[0009] In this embodiment, by setting 3Nb≥|(10Ns-3Np) / 7|, the number of turns of the auxiliary winding can be roughly determined based on the number of turns of the primary winding, the number of turns of the secondary winding, and the coil size parameters. This can minimize the original displacement current Ips and ensure a better cancellation effect in the actual product.

[0010] In some embodiments, this application provides a transformer, the transformer comprising: A magnetic core, a primary winding, and a secondary winding, wherein the primary winding and the secondary winding are disposed on the magnetic core; The first shielding layer is disposed between the magnetic core and the primary winding and is electrically connected to the primary ground to clamp the magnetic core potential to the primary ground potential in order to eliminate the displacement current of the magnetic core to ground. An auxiliary winding is disposed between the primary winding and the secondary winding. The corresponding terminal of the auxiliary winding is opposite to the corresponding terminal of the primary winding. The first terminal of the auxiliary winding is suspended, and the second terminal of the auxiliary winding is electrically connected to the primary ground. Wherein, the number of turns of the auxiliary winding is Nb, and the induced voltage Vb of the auxiliary winding when it is powered on causes the compensation current Ibs flowing from the secondary winding to the auxiliary winding to be in the opposite direction to the original displacement current Ips flowing from the primary winding to the secondary winding, and the absolute value of the difference between the compensation current Ibs and the original displacement current Ips is less than one-tenth of the original displacement current Ips.

[0011] To compensate for the deteriorated EMC performance caused by removing the Y capacitor, some embodiments of this application provide a transformer scheme for removing the Y capacitor. On the one hand, by setting a first shielding layer between the magnetic core and the primary winding and electrically connecting the first shielding layer to the primary ground, the displacement current Icg of the magnetic core to ground is eliminated by clamping the magnetic core potential to the primary ground potential. On the other hand, an auxiliary winding is set between the primary and secondary windings, and the corresponding terminals of the auxiliary winding are set to be opposite to those of the primary winding. The auxiliary winding is connected to the primary ground. By setting the number of turns of the auxiliary winding to Nb and adjusting the induced voltage Vb on the auxiliary winding, the compensation current Ibs flowing from the secondary winding to the auxiliary winding is made to flow in the opposite direction to the original displacement current Ips flowing from the primary winding to the secondary winding. Moreover, the absolute value of the difference between the compensation current Ibs and the original displacement current Ips is less than one-tenth of the original displacement current Ips. At this time, the compensation current Ibs can at least offset nine-tenths of the original displacement current Ips. In this way, a reverse displacement current can be actively created inside the transformer to offset the original interference current, thereby significantly reducing the net current flowing to the ground. Thus, superior EMC performance can be achieved without a Y capacitor, and interference problems under high-power applications can be solved.

[0012] In some embodiments, the current value of the compensation current Ibs is equal to the current value of the original displacement current Ips, and the total displacement current Isg of the secondary winding to ground is zero.

[0013] In some embodiments of this application, the displacement current Isg of the secondary winding to ground is one of the sources of common-mode interference voltage. When the switching power supply is working, the induced voltage Vb of the auxiliary winding when powered on causes the compensation current Ibs generated by the auxiliary winding and flowing from the secondary winding to the auxiliary winding to be equal in magnitude and opposite in direction to the original displacement current Ips flowing from the primary winding to the secondary winding, thereby making the total displacement current Isg of the secondary winding to ground approach zero. When the total displacement current Isg approaches zero, the potential difference between the secondary circuit and the ground will be greatly reduced or even eliminated, thereby directly and significantly reducing the common-mode noise in conducted EMI testing, enabling the entire device to pass EMC testing without using Y capacitors.

[0014] In some embodiments, the current value of the compensation current Ibs is less than the current value of the original displacement current Ips.

[0015] In some embodiments of this application, the amplitude of the compensation current Ibs is smaller than that of the original interference current Ips, and the direction of the compensation current Ibs is opposite to that of the original displacement current Ips. The transformer is in a slightly undercompensated state. In the slightly undercompensated state, the transformer can take into account both conducted and radiated EMI engineering conditions.

[0016] In some embodiments, the distance between the auxiliary winding and the secondary winding is d1, and the distance between the primary winding and the secondary winding is d2, where 5 / 3d1≤d2≤7 / 3d1.

[0017] In some embodiments of this application, the winding sequence of the transformer in this embodiment is magnetic core, first shielding layer, primary winding, auxiliary winding, and secondary winding. The auxiliary winding is located between the primary winding and the secondary winding. The position between the auxiliary winding and its adjacent winding coils can be roughly determined. Then, based on the size parameters of the coil winding, the number of turns of the auxiliary winding can be roughly determined. This can minimize the original displacement current Ips, ensure a better cancellation effect in the actual product, and help reduce the size of the transformer.

[0018] In some embodiments, the induced voltage Vb of the auxiliary winding, the voltage Vp across the primary winding, and the voltage Vs across the secondary winding satisfy the following relationship: Vb = (3Vs - Vp) / 2.

[0019] In some embodiments of this application, the voltage Vp across the primary winding and the voltage Vs across the secondary winding determine the original displacement current Ips. The induced voltage Vb of the auxiliary winding and the voltage Vs across the secondary winding determine the compensation current Ibs flowing from the secondary winding to the auxiliary winding. Since the auxiliary winding is located between the primary winding and the secondary winding, under a specific structure (the distance between the secondary winding and the auxiliary winding is half the distance between the primary and secondary windings, i.e., d2=2d1), Vb = (3Vs - Vp) / 2, which can make the compensation current Ibs offset the original displacement current Ips as much as possible, thereby ensuring that the offsetting effect is achieved in the actual product.

[0020] In some embodiments, the distributed capacitance Cps of the primary winding to the secondary winding is inversely proportional to the distance d2 between the primary winding and the secondary winding; and / or The distributed capacitance Cps of the primary winding to the secondary winding is directly proportional to the relative area Sps between the primary winding and the secondary winding.

[0021] In some embodiments of this application, the magnitude of the displacement current is determined by the driving voltage and the capacitive reactance of the parasitic capacitance. The native displacement current from the primary winding to the secondary winding is: Ips = (Vp - Vs) * 2πf * Cps, where f is the noise frequency and Cps is the distributed capacitance from the primary winding to the secondary winding (i.e., the primary-secondary parasitic capacitance). Since the distributed capacitance Cps from the primary winding to the secondary winding is inversely proportional to the distance d2 between the primary and secondary windings, and directly proportional to the relative area Sps between the primary and secondary windings, the magnitude of the native displacement current Ips from the primary winding to the secondary winding can be adjusted by adjusting the distance d2 between the primary and secondary windings, or by adjusting the relative area Sps between the primary and secondary windings. This ensures that the compensation current Ibs can offset the native displacement current Ips as much as possible, thus achieving a cancellation effect in the actual product.

[0022] In some embodiments, the distributed capacitance Cbs of the auxiliary winding to the secondary winding is inversely proportional to the distance d1 between the auxiliary winding and the secondary winding; and / or The distributed capacitance Cbs of the auxiliary winding to the secondary winding is directly proportional to the relative area Sbs between the auxiliary winding and the secondary winding.

[0023] In some embodiments of this application, the magnitude of the displacement current is determined by the driving voltage and the capacitive reactance of the parasitic capacitance. The displacement current of the secondary winding to the auxiliary winding is: Ibs = (Vs - Vb) * 2πf * Cbs, where f is the noise frequency and Cbs is the distributed capacitance of the auxiliary winding to the secondary winding (i.e., the parasitic capacitance between the secondary and auxiliary windings). Since the distributed capacitance Cbs of the auxiliary winding to the secondary winding is inversely proportional to the distance d1 between the auxiliary and secondary windings, and the distributed capacitance Cbs of the auxiliary winding to the secondary winding is directly proportional to the relative area Sbs between the auxiliary and secondary windings, the magnitude of the displacement current Ibs of the secondary winding to the auxiliary winding can be adjusted by adjusting the distance d1 between the auxiliary and secondary windings, or by adjusting the relative area Sbs between the auxiliary and secondary windings. This ensures that the compensation current Ibs offsets the original displacement current Ips as much as possible, thus achieving a cancellation effect in the actual product.

[0024] In some embodiments, the induced voltage Vb of the auxiliary winding is directly proportional to the number of turns Nb of the auxiliary winding.

[0025] In some embodiments of this application, when the switching power supply is operating, the changing voltage Vp on the primary winding Np generates an alternating magnetic field in the magnetic core. This alternating magnetic field induces voltages in all windings. The secondary winding voltage Vs and the auxiliary winding voltage Vb are both induced by the same main magnetic flux. Therefore, the voltage of each winding is strictly proportional to its number of turns. Specifically, the number of turns Nb in the auxiliary winding is related to the induced voltage Vb generated by the auxiliary winding when powered on. Reducing the number of turns Nb in the auxiliary winding will directly reduce |Vb|, thereby reducing |Ibs|. Setting the correct number of turns Nb to obtain the correct induced voltage Vb can ensure that the compensation current Ibs generated by the auxiliary winding is exactly equal to (or close to) the original displacement current Ips that needs to be canceled in amplitude, ultimately achieving the effect of eliminating common-mode interference and realizing a high-performance EMC design without Y capacitors.

[0026] A second aspect of this application also provides a switching power supply, including a transformer as described in any of the above embodiments, and a magnetic ring filter disposed between a power input terminal and the transformer, the magnetic ring filter being configured to filter the power input terminal.

[0027] In some embodiments of this application, in low-power power supply applications (power output less than 75 watts), the transformer in any of the above embodiments can eliminate most interference, ensuring the power supply meets EMC standards. However, in higher-power switching power supplies (e.g., power output greater than 75 watts), power lines introduce common-mode interference. Therefore, power lines are both a pathway for external grid interference to enter the device and a channel for internal interference to be emitted outwards. To eliminate power line interference, a magnetic ring filter can be installed between the power input terminal and the transformer. This magnetic ring filter filters external or external interference channels, complementing the transformer's internal handling of the switching power supply's own interference, thus forming a complete two-stage filtering system to cope with more complex interference environments.

[0028] In some embodiments, the magnetic ring filter includes a ferrite magnetic ring, and the power input terminal includes a live wire, a neutral wire, and a ground wire, wherein the live wire and the neutral wire of the power input terminal are wound around the ferrite magnetic ring, and the ground wire of the power input terminal is not wound around the ferrite magnetic ring.

[0029] In some embodiments of this application, the ferrite core exhibits high impedance characteristics at high frequencies, acting as a lossy inductor. The live wire (L) and neutral wire (N) of the power input are simultaneously wound in parallel on the same ferrite core. The ground wire (PE), serving as a discharge path for common-mode interference current, is directly grounded and not wound on the core. When the live and neutral wires pass through the core in the same direction, the magnetic fields generated by their differential-mode currents (normal operating currents) are equal in magnitude and opposite in direction within the core, canceling each other out. Therefore, the core has very low impedance to differential-mode currents and does not affect normal operation. For common-mode interference currents (equal in magnitude and direction on the live and neutral wires), their magnetic fields are superimposed in the same direction within the core, making the ferrite core behave as a high-impedance element, thereby significantly attenuating the transmission of common-mode interference.

[0030] A third aspect of some embodiments of this application also provides an air conditioning device, including a switching power supply as described in any of the above embodiments, the switching power supply being used to power the control board of the air conditioning device.

[0031] Additional aspects and advantages of this application will be set forth in part in the description which follows, and in part will be obvious from the description, or may be learned by practice of this application. Attached Figure Description

[0032] The above and / or additional aspects and advantages of this application will become apparent and readily understood from the description of the embodiments taken in conjunction with the following drawings, in which: Figure 1 This is a first schematic diagram of a transformer provided in some embodiments of this application; Figure 2 This is a schematic diagram of the transformer principle provided in some embodiments of this application. Figure 1 ; Figure 3 This is a schematic diagram of a power supply circuit provided in some embodiments of this application. Figure 1 ; Figure 4 This is a schematic diagram of a power supply circuit provided in some embodiments of this application. Figure 3 ; Figure 5 This is a schematic diagram of a power supply circuit provided in some embodiments of this application. Figure 4 ; Figure 6 This is a schematic diagram of a power supply circuit provided in some embodiments of this application. Figure 5 ; Figure 7 This is a schematic diagram of a power supply circuit provided in some embodiments of this application. Figure 6 ; Figure 8 This is a schematic diagram of a power supply circuit provided in some embodiments of this application. Figure 7 ; Figure 9 This is a schematic diagram of the test circuit provided in some embodiments of this application. Figure 1 ; Figure 10 This is a schematic diagram of the test circuit provided in some embodiments of this application. Figure 2 ; Figure 11 This is a schematic diagram of the transformer principle provided in some embodiments of this application. Figure 2 . Detailed Implementation

[0033] The embodiments of this application are described in detail below. The embodiments described with reference to the accompanying drawings are exemplary. The embodiments of this application are described in detail below.

[0034] In related technical solutions, to meet electromagnetic compatibility requirements, a Y capacitor is usually connected in parallel between the primary and secondary windings of the transformer to provide a low-impedance loop for common-mode interference current. However, after introducing the Y capacitor, the power supply suffers from reduced anti-interference capability and reduced electrical isolation.

[0035] To address the aforementioned technical problems, some embodiments of this application provide a transformer, see [link to relevant documentation]. Figure 1 As shown, the transformer includes: a magnetic core 100, a primary winding 210, and a secondary winding 220, with the primary winding 210 and the secondary winding 220 disposed on the magnetic core 100.

[0036] In this embodiment, combined with Figure 1As shown, the magnetic core 100 is provided with a primary winding 210 and a secondary winding 220. The primary winding 210 is located between the magnetic core 100 and the secondary winding 220. The same-name terminals of the primary winding 210 and the secondary winding 220 are opposite. When the switching transistor on the primary side is turned off, the induced voltage of the secondary winding 220 can make the rectifier diode conduct in the forward direction and release energy to the load.

[0037] In some embodiments, an air gap 101 is provided in the magnetic core 100. The air gap 101 can increase the magnetic circuit reluctance and prevent the magnetic core 100 from becoming magnetically saturated under large DC bias or instantaneous large current. This avoids the inductance from dropping sharply due to magnetic saturation. Under magnetic saturation, the transformer will not work properly and may cause device damage.

[0038] In some embodiments, the transformer in this embodiment further includes a first shielding layer, which is disposed between the magnetic core 100 and the primary winding 210 and electrically connected to the primary ground. The first shielding layer is used to clamp the potential of the magnetic core 100 to the primary ground potential to eliminate the displacement current of the magnetic core 100 to ground. In this embodiment, in the switching power supply scheme, after removing the Y capacitor, the displacement current Icg of the magnetic core 100 to ground is one of the main causes of common-mode interference. In order to compensate for the deteriorated EMC performance caused by removing the Y capacitor, a first shielding layer is provided between the magnetic core 100 and the primary winding 210, and the first shielding layer is electrically connected to the primary ground. This clamps the potential of the magnetic core 100 to the primary ground potential, making the displacement current Icg of the magnetic core 100 to ground close to zero.

[0039] In some embodiments, the first shielding layer is a shielding copper foil.

[0040] In some embodiments, the transformer in this embodiment further includes an auxiliary winding 230, which is disposed between the primary winding 210 and the secondary winding 220. The corresponding terminal of the auxiliary winding 230 is opposite to that of the primary winding 210. The first terminal of the auxiliary winding 230 is suspended, and the second terminal of the auxiliary winding 230 is connected to the primary ground. The number of turns of the auxiliary winding 230 is Nb, and the induced voltage Vb of the auxiliary winding 230 when energized causes the compensation current Ibs flowing from the secondary winding 220 to the auxiliary winding 230 to be in the opposite direction to the original displacement current Ips flowing from the primary winding 210 to the secondary winding 220. The absolute value of the difference between the compensation current Ibs and the original displacement current Ips is less than one-tenth of the original displacement current Ips.

[0041] In this embodiment, common-mode interference is also caused by the current Isg from the primary winding 210 to the secondary winding 220 and back to ground. This current Isg can be decomposed into the displacement current Ips from the primary winding 210 to the secondary winding 220 and the current from the secondary winding 220 to ground. To counteract the current Isg from the primary winding 210 to the secondary winding 220 and back to ground, an auxiliary winding 230 is provided between the primary winding 210 and the secondary winding 220. The corresponding terminals of the auxiliary winding 230 are opposite to those of the primary winding 210. The auxiliary winding 230 is connected to the primary ground. By setting the number of turns of the auxiliary winding 230 to Nb, the induced voltage Vb when the auxiliary winding 230 is energized is adjusted, so that the compensation current Ibs flowing from the secondary winding 220 to the auxiliary winding 230 is equal to the current flowing from the primary winding 210 to the secondary winding 220. The original displacement current Ips of winding 220 is in the opposite direction, and the absolute value of the difference between the compensation current Ibs and the original displacement current Ips is less than one-tenth of the original displacement current Ips. At this time, the compensation current Ibs can at least offset nine-tenths of the original displacement current Ips. In this way, a reverse displacement current can be actively created inside the transformer to offset the original interference current, thereby significantly reducing the net current flowing to the ground. Thus, a relatively excellent EMC performance can be achieved without a Y capacitor, and the interference problem under high power applications can be solved.

[0042] In some embodiments, the primary winding 210, the secondary winding 220, and the auxiliary winding 230 are enameled wires.

[0043] In some embodiments, combined with Figure 1 As shown, a retaining tape 310 is also provided between the primary winding 210, the secondary winding 220, the auxiliary winding 230 and the magnet 100.

[0044] In some embodiments, combined with Figure 1 As shown, insulating tape 320 is also provided between the primary winding 210 and the auxiliary winding 230, and insulating tape 320 is also provided between the secondary winding 220 and the auxiliary winding 230.

[0045] In some embodiments, combined with Figure 1 As shown, the output side of the primary winding 210, secondary winding 220, and auxiliary winding 230 is also provided with a frame 400.

[0046] In some embodiments, the current value of the compensation current Ibs is equal to the current value of the original displacement current Ips, the current direction of the compensation current Ibs is opposite to the current direction of the original displacement current Ips, and the total displacement current Isg of the secondary winding 220 to ground is zero.

[0047] In this embodiment, the displacement current Isg of the secondary winding 220 to ground is one of the main sources of common-mode interference voltage. When the switching power supply is working, the induced voltage Vb of the auxiliary winding 230 when powered on causes the compensation current Ibs generated by the auxiliary winding 230 and flowing from the secondary winding 220 to the auxiliary winding 230 to be equal in magnitude and opposite in direction to the original displacement current Ips flowing from the primary winding 210 to the secondary winding 220. This makes the total displacement current Isg of the secondary winding 220 to ground approach zero. When Isg approaches zero, the potential difference between the secondary circuit and ground will be greatly reduced or even eliminated, thereby directly and significantly reducing common-mode noise in conducted EMI testing, enabling the entire device to pass EMC testing without using Y capacitors.

[0048] In some embodiments, the distance between the auxiliary winding 230 and the secondary winding 220 is d1, and the distance between the primary winding 210 and the secondary winding 220 is d2, where 5 / 3d1≤d2≤7 / 3d1.

[0049] In this embodiment, based on the distance d1 between the auxiliary winding 230 and the secondary winding 220, and the distance d2 between the primary winding 210 and the secondary winding 220, the position between the auxiliary winding 230 and its adjacent winding coils can be roughly determined. Then, based on the size parameters of the coil windings, the number of turns of the auxiliary winding can be roughly determined, which can minimize the original displacement current Ips and ensure a better cancellation effect in the actual product.

[0050] In some embodiments, the distance between the auxiliary winding 230 and the secondary winding 220 is d1, and the distance between the primary winding 210 and the secondary winding 220 is d2, where d2 = 2d1.

[0051] In this embodiment, the winding sequence of the transformer is: magnetic core 100, first shielding layer, primary winding 210, auxiliary winding 230, and secondary winding 220. The auxiliary winding 230 is located between the primary winding 210 and the secondary winding 220. The auxiliary winding 230 allows the primary winding 210 and the secondary winding 220 to be approximately equally spaced. Thus, the primary winding 210, auxiliary winding 230, and secondary winding 220 are arranged as closely as possible, which helps to reduce the size of the transformer.

[0052] In some embodiments, the induced voltage Vb of the auxiliary winding 230, the voltage Vp across the primary winding 210, and the voltage Vs across the secondary winding 220 satisfy the following relationship: Vb = (3Vs - Vp) / 2.

[0053] In this embodiment, the voltage Vp across the primary winding 210 and the voltage Vs across the secondary winding 220 determine the original displacement current Ips. The induced voltage Vb of the auxiliary winding 230 and the voltage Vs across the secondary winding 220 determine the compensation current Ibs flowing from the secondary winding 220 to the auxiliary winding 230. Since the auxiliary winding 230 is located between the primary winding 210 and the secondary winding 220, under a specific structure (the distance between the secondary winding 220 and the auxiliary winding 230 is half the distance between the primary and secondary windings, i.e., d2=2d1), Vb = (3Vs-Vp) / 2, which can make the compensation current Ibs offset the original displacement current Ips as much as possible, thereby ensuring a better offsetting effect in the actual product.

[0054] In some embodiments, the distributed capacitance Cps of the primary winding 210 to the secondary winding 220 is inversely proportional to the distance d2 between the primary winding 210 and the secondary winding 220.

[0055] In this embodiment, combined with Figure 2 As shown, Vp is the voltage across the primary winding 210, Vs is the voltage across the secondary winding 220, the corresponding terminals of the auxiliary winding 230 are opposite to those of the primary winding 210, and the corresponding terminals of the auxiliary winding 230 are opposite to those of the secondary winding 220. The magnitude of the displacement current of the primary winding 210 is determined by the driving voltage and the capacitive reactance of the parasitic capacitance. The primary displacement current of the primary winding 210 to the secondary winding 220 is: Ips = (Vp - Vs) * 2πf * Cps, where f is the noise frequency and Cps is the distributed capacitance of the primary winding 210 to the secondary winding 220 (i.e., the primary-secondary parasitic capacitance). Since the distributed capacitance Cps of the primary winding 210 to the secondary winding 220 is inversely proportional to the distance d2 between the primary winding 210 and the secondary winding 220, and the distributed capacitance Cps of the primary winding 210 to the secondary winding 220 is directly proportional to the relative area Sps between the primary winding 210 and the secondary winding 220, the magnitude of the original displacement current Ips of the primary winding 210 to the secondary winding 220 can be adjusted by adjusting the distance d2 between the primary winding 210 and the secondary winding 220. This allows the compensation current Ibs to offset the original displacement current Ips as much as possible, ensuring a better offsetting effect in actual products.

[0056] In some embodiments, Formula 1 is obtained according to the distributed capacitance calculation formula: Cbs = (ε1 × s1) / d1; d1 is the distance between the secondary winding 220 and the auxiliary winding 230, s1 is the relative area between the secondary winding 220 and the auxiliary winding 230, and ε1 is the dielectric constant between the secondary winding 220 and the auxiliary winding 230.

[0057] Formula 2: Cps = (ε2 × s2) / d2, where d2 is the distance between the primary winding 210 and the secondary winding 220, s2 is the relative area between the primary winding 210 and the secondary winding 220, and ε2 is the dielectric constant between the primary winding 210 and the secondary winding 220.

[0058] In some embodiments, the distributed capacitance Cps of the primary winding 210 to the secondary winding 220 is directly proportional to the relative area Sps between the primary winding 210 and the secondary winding 220.

[0059] In this embodiment, the magnitude of the displacement current is determined by the driving voltage and the capacitive reactance of the parasitic capacitance. The original displacement current of the primary winding 210 to the secondary winding 220 can be obtained according to Formula 3: Ips = (Vp - Vs) ×2πf ×Cps, where f is the noise frequency and Cps is the distributed capacitance of the primary winding 210 to the secondary winding 220 (i.e., the primary-secondary parasitic capacitance). Since the distributed capacitance Cps of the primary winding 210 to the secondary winding 220 is inversely proportional to the distance d2 between the primary winding 210 and the secondary winding 220, and the distributed capacitance Cps of the primary winding 210 to the secondary winding 220 is directly proportional to the relative area Sps between the primary winding 210 and the secondary winding 220, the magnitude of the original displacement current Ips of the primary winding 210 to the secondary winding 220 can be adjusted by adjusting the relative area Sps between the primary winding 210 and the secondary winding 220. This allows the compensation current Ibs to offset the original displacement current Ips as much as possible, ensuring a better offsetting effect in actual products.

[0060] In some embodiments, the distributed capacitance Cbs of the auxiliary winding 230 to the secondary winding 220 is inversely proportional to the distance d1 between the auxiliary winding 230 and the secondary winding 220.

[0061] In this embodiment, the magnitude of the displacement current Ibs is determined by the driving voltage and the capacitive reactance of the parasitic capacitance. The displacement current of the secondary winding 220 to the auxiliary winding 230 can be obtained according to Formula 4: Ibs = (Vs - Vb) ×2πf ×Cbs, where f is the noise frequency and Cbs is the distributed capacitance of the auxiliary winding 230 to the secondary winding 220 (i.e., the parasitic capacitance of the secondary-auxiliary winding 230). Since the distributed capacitance Cbs of the auxiliary winding 230 to the secondary winding 220 is inversely proportional to the distance d1 between the auxiliary winding 230 and the secondary winding 220, and the distributed capacitance Cbs of the auxiliary winding 230 to the secondary winding 220 is directly proportional to the relative area Sbs between the auxiliary winding 230 and the secondary winding 220, the magnitude of the displacement current Ibs of the secondary winding 220 to the auxiliary winding 230 can be adjusted by adjusting the distance d1 between the auxiliary winding 230 and the secondary winding 220. This allows the compensation current Ibs to offset the original displacement current Ips as much as possible, ensuring a good offsetting effect in actual products.

[0062] In some embodiments, under the ideal condition that the current Isg of the primary winding 210 to the secondary winding 220 to ground is 0A, the compensation current Ibs = Ips. Within the same transformer, ε and s are essentially constant, only d is variable. If the thickness of the insulating tape is ignored, d2 can be considered as 2d1.

[0063] Substituting d2=2d1 into Equations 1 and 2 for distributed capacitance, we obtain Equation 5: Cbs / Cps=d2 / d1=2.

[0064] Substituting the condition: compensation current Ibs = Ips, formula 5 combined with formulas 3 and 4 yields formula 6: Vb = (3Vs - Vp) / 2.

[0065] Since the voltage on the winding is directly proportional to the number of turns, formula 7 can be derived from formula 6 regarding the relationship between the winding coils: Nb = (3Ns - Np) / 2, where Nb is an integer.

[0066] In this embodiment, assuming Ns=20 and Np=100, Nb=-20 can be calculated according to Formula 7. -20 indicates that the corresponding terminal of the auxiliary winding is opposite to the corresponding terminal of the primary winding 210, and the corresponding terminal of the auxiliary winding is opposite to the corresponding terminal of the secondary winding 220. Thus, assuming the compensation current Ibs=Ips and d2=2d1, if the number of turns of the secondary winding 220 is 20 and the number of turns of the primary winding 210 is 100, then setting the number of turns of the auxiliary winding 230 to 20 can minimize the elimination of the original displacement current Ips, ensuring a better compensation effect in the actual product.

[0067] In some embodiments, the auxiliary winding 230 has Nb turns, the primary winding 210 has Np turns, and the secondary winding 220 has Ns turns, where 2Nb ≤ |3Ns - Np|. When the number of turns in the auxiliary winding 230 is slightly less than the set number (|(3Ns - Np) / 2|), it is in an undercompensated state. In this case, the compensation current Ibs is slightly less than the original current Ips, which can effectively suppress conducted interference (CE) while positively impacting the pass of radiated emission (RE) tests.

[0068] In some embodiments, substituting d2 = (7 / 3)d1 into Equations 1 and 2 for the distributed capacitance yields Equation 8: Cbs / Cps=d2 / d1=7 / 3.

[0069] Substituting the condition: compensation current Ibs = Ips, formula 8 combined with formulas 3 and 4 yields formula 9: 7Vb = (10Vs - 3Vp).

[0070] Since the voltage on the winding is directly proportional to the number of turns, formula 9 can be used to derive formula 10 for the relationship between the winding coils: Nb = |(10Ns-3Np) / 7|, where Nb is an integer.

[0071] In this embodiment, assuming Ns=20 and Np=100, Nb=100 / 7 can be calculated according to formula 10. Nb can be 14, which means that the compensation current Ibs=Ips. In the case of (7 / 3)d1, if the number of turns of the secondary winding 220 is 20 and the number of turns of the primary winding 210 is 100, then the number of turns of the auxiliary winding 230 is set to 14, so as to eliminate the original displacement current Ips as much as possible and ensure a better compensation effect in the actual product.

[0072] In some embodiments, 3Nb≥|(10Ns-3Np) / 7|, so that the number of turns of the auxiliary winding can be roughly determined based on the number of turns of the primary winding 210, the number of turns of the secondary winding 220, and the coil size parameters, which can minimize the original displacement current Ips and ensure a better cancellation effect in the actual product.

[0073] In some embodiments, the distributed capacitance Cbs of the auxiliary winding 230 to the secondary winding 220 is directly proportional to the relative area Sbs between the auxiliary winding 230 and the secondary winding 220.

[0074] In this embodiment, the magnitude of the displacement current is determined by the driving voltage and the capacitive reactance of the parasitic capacitance. The displacement current of the secondary winding 220 to the auxiliary winding 230 is: Ibs = (Vs - Vb) ×2πf ×Cbs, where f is the noise frequency and Cbs is the distributed capacitance of the auxiliary winding 230 to the secondary winding 220 (i.e., the parasitic capacitance of the secondary-auxiliary winding 230). Since the distributed capacitance Cbs of the auxiliary winding 230 to the secondary winding 220 is inversely proportional to the distance d1 between the auxiliary winding 230 and the secondary winding 220, and the distributed capacitance Cbs of the auxiliary winding 230 to the secondary winding 220 is directly proportional to the relative area Sbs between the auxiliary winding 230 and the secondary winding 220, the magnitude of the displacement current Ibs of the secondary winding 220 to the auxiliary winding 230 can be adjusted by adjusting the relative area Sbs between the auxiliary winding 230 and the secondary winding 220. This allows the compensation current Ibs to offset the original displacement current Ips as much as possible, ensuring a better offsetting effect in actual products.

[0075] In some embodiments, the induced voltage Vb of the auxiliary winding 230 is directly proportional to the number of turns Nb of the auxiliary winding 230.

[0076] In this embodiment, when the switching power supply is working, the changing voltage Vp on the primary winding 210Np will generate an alternating magnetic field in the magnetic core 100. This alternating magnetic field will induce voltages in all windings. The voltage Vs of the secondary winding 220 and the voltage Vb of the auxiliary winding 230 are both generated by the same main magnetic flux. Therefore, the voltage of each winding is strictly proportional to its number of turns. Specifically, the number of turns Nb of the auxiliary winding 230 is related to the induced voltage Vb generated by the auxiliary winding 230 when powered on. Reducing the number of turns Nb of the auxiliary winding 230 will directly reduce |Vb|, thereby reducing |Ibs|. Setting the correct number of turns Nb to obtain the correct induced voltage Vb can ensure that the compensation current Ibs generated by the auxiliary winding 230 is exactly equal to (or close to) the original displacement current Ips that needs to be canceled in amplitude, ultimately achieving the effect of eliminating common-mode interference and realizing a high-performance EMC design without Y capacitors.

[0077] In some embodiments, combined with Figure 2 As shown, Cbs is the parasitic capacitance between the auxiliary winding 230 and the secondary winding 220, and Csg is the parasitic capacitance between the secondary winding 220 and ground. The first terminals of the auxiliary winding 230 and the primary winding 210 are connected to ground. Taking Np=100, Nb=20, and Ns=10 as an example, Np is the number of turns of the primary winding 210, NB is the number of turns of the auxiliary winding 230, and Ns is the number of turns of the secondary winding 220. The induced voltage of the auxiliary winding 230 is Vb=Vp×Nb / Np=0.2Vp, and the voltage across the secondary winding 220 is Vs=Vp×Ns / Np=0.1Vp.

[0078] Because the high voltage Vp across the primary winding 210 will generate a displacement current Ips to the secondary side through the parasitic capacitance Cps. At the same time, the voltage Vs across the secondary winding 220 will also generate current through Cbs. Thus, an induced voltage Vb is generated across the auxiliary winding 230. Since the first end of the auxiliary winding 230 is grounded, the voltage of the other end of the auxiliary winding 230 relative to ground is Vb. This voltage will generate a compensation current Ibs on Cbs. Since the corresponding terminals of the auxiliary winding 230 are opposite to those of the primary winding 210, the compensation current Ibs is less than 0A. Thus, the compensation current Ibs is designed to be opposite in direction to the interference current conducted from the primary winding. By adjusting the number of turns of the auxiliary winding 230 so that Vb=2Vs, the amplitude of the compensation current Ibs can be made exactly equal to the equivalent interference component of Ips on the secondary side, thereby canceling each other out. The total current Isg flowing from the transformer secondary winding 220 to the ground is 0A, constructing a current path that cancels out the interference current, achieving the effect of eliminating the Y capacitance to ground while meeting EMC standards.

[0079] In some embodiments, see Figure 3 As shown, this application embodiment provides a power supply circuit in which the Y capacitor is removed. See [link to relevant documentation]. Figure 3 As shown, the power supply circuit in this embodiment includes a transformer 200, which includes a magnetic core 100, a primary winding 210, a secondary winding 220, a first shielding layer 240, and an auxiliary winding 230. The primary winding 210 and the secondary winding 220 are disposed on the magnetic core 100. The first shielding layer 240 is disposed between the primary winding 210 and the magnetic core 100. The auxiliary winding 230 is disposed between the primary winding 210 and the secondary winding 220. The first shielding layer 240 and the auxiliary winding 230 are electrically connected to the primary ground GND1.

[0080] In this embodiment, to overcome the drawbacks of removing the Y capacitor, a power supply scheme that removes the Y capacitor is provided. After removing the Y capacitor, common-mode interference is mainly caused by the displacement current Icg of the magnetic core 100 to ground and the current Isg generated by the primary winding 210 to the secondary winding 220 and then to ground. The current Isg can be further decomposed into the displacement current Ips of the primary winding 210 to the secondary winding 220 and the current of the secondary winding 220 to ground.

[0081] Combination Figure 1 As shown, in this embodiment, the first shielding layer 240 is disposed between the magnetic core 100 and the primary winding 210 and is electrically connected to the primary ground GND1. The first shielding layer 240 is a conductive shielding layer, and the connection between the first shielding layer 240 and the magnetic core 100 is an electrical connection.

[0082] In this embodiment, the power supply circuit can be a switching power supply circuit. The first shielding layer 240 is used to clamp the potential of the magnetic core 100 to the primary ground GND1 potential, which can eliminate the displacement current of the magnetic core 100 to ground. In the switching power supply scheme, after removing the Y capacitor, the displacement current Icg of the magnetic core 100 to ground is one of the main causes of common-mode interference. In order to compensate for the deterioration of EMC performance due to the removal of the Y capacitor, the first shielding layer 240 is provided between the magnetic core 100 and the primary winding 210, and the first shielding layer 240 is electrically connected to the primary ground GND1. This clamps the potential of the magnetic core 100 to the primary ground GND1 potential, making the displacement current Icg of the magnetic core 100 to ground close to zero.

[0083] On the other hand, an auxiliary winding 230 is provided between the primary winding 210 and the secondary winding 220, and the corresponding terminal of the auxiliary winding 230 is set to be opposite to the corresponding terminal of the primary winding 210. The auxiliary winding 230 is connected to the primary ground GND1. By setting the number of turns of the auxiliary winding 230 to Nb, the induced voltage Vb when the auxiliary winding 230 is powered on is adjusted, so that the compensation current Ibs flowing from the secondary winding 220 to the auxiliary winding 230 is opposite to the direction of the original displacement current Ips flowing from the primary winding 210 to the secondary winding 220. At this time, a reverse displacement current can be actively created inside the transformer 200 to cancel the original interference current, thereby significantly reducing the net current flowing to the ground. Thus, a better EMC performance can be achieved without a Y capacitor, and the interference problem under high power application can be solved.

[0084] In some embodiments, combined with Figure 4 As shown, the power supply circuit in this embodiment includes an input buffer module 410. The first end of the input buffer module 410 and the first end of the primary winding 210 are both connected to the power input terminal 600. The second end of the input buffer module 410 is connected to the primary ground GND1.

[0085] In this embodiment, the input buffer module 410 is connected in parallel with the primary winding 210 and can be used to absorb the peak energy of the input power supply. The input buffer module 410 can also form a resonant network with the leakage inductance of the transformer 200. In the flyback topology power supply structure, it can also absorb the peak voltage generated by the leakage inductance of the transformer 200 when the switching transistor is turned off, protecting the switching transistor from being broken down.

[0086] In some embodiments, combined with Figure 4 As shown, the power supply circuit in this embodiment includes a first switch module 430. The first end of the first switch module 430 is connected to the second end of the primary winding 210, and the second end of the first switch module 430 is connected to the primary ground GND1.

[0087] In this embodiment, the first switching module 430 is connected in series with the primary winding 210. The capacitor in the input buffer module 410 and the leakage inductance of the transformer 200 form a resonant network. By controlling the switching frequency of the first switching module 430, the first switching module 430 can switch under zero voltage or zero current conditions, thereby significantly reducing switching losses and electromagnetic interference.

[0088] In some embodiments, combined with Figure 4 As shown, the power supply circuit in this embodiment includes an output filter module 420. The first end of the output filter module 420 is connected to the first end of the secondary winding 220, and the second end of the output filter module 420 and the second end of the secondary winding 220 are connected to the secondary ground GND2.

[0089] In this embodiment, the output filter module 420 utilizes its charging and discharging characteristics to store energy when the secondary winding 220 outputs current, and provides current to the downstream load when the output of the secondary winding 220 stops, thereby filling voltage troughs and achieving the effect of stabilizing the output voltage. Furthermore, when the downstream load requires a large current for a certain period, the output filter module 420 can quickly discharge to replenish the current energy gap and maintain voltage stability.

[0090] In some embodiments, see Figure 5 As shown, the power supply circuit also includes: a unidirectional conduction module 440, the input terminal of which is connected to the first terminal of the secondary winding 220, and the output terminal of which is connected to the first terminal of the output filter module 420.

[0091] In this embodiment, the secondary winding 220 of the transformer outputs high-frequency alternating current. When the polarity of the induced voltage in the secondary winding 220 is positive (positive at the top, negative at the bottom), the unidirectional conduction module 440 conducts in the forward direction, allowing current to flow. When the voltage polarity reverses (negative at the top, positive at the bottom), the unidirectional conduction module 440 cuts off in the reverse direction, preventing reverse current flow. Through this periodic conduction and cutoff, the high-frequency pulsed alternating current output by the transformer is converted into unidirectional pulsating direct current, providing energy to the downstream load. Furthermore, the unidirectional conduction module 440 on the secondary side also serves as a freewheeling protection mechanism. When the first switch module 430 on the primary side is turned off, the magnetic field of the secondary winding 220 collapses and releases energy. Without the unidirectional conduction module 440, the polarity of the induced voltage in the secondary winding 220 would reverse, and the resulting reverse electromotive force might break down the first switch module 430 on the primary side, or prevent energy from being effectively transferred to the load. The unidirectional conduction module 440 cuts off during this period, providing isolation and protection.

[0092] In some embodiments, combined with Figure 6As shown, the unidirectional conduction module 440 includes a first diode D1, the anode of the first diode D1 is connected to the first end of the secondary winding 220, and the cathode of the first diode D1 is connected to the output filter module 420.

[0093] In this embodiment, the unidirectional conduction module 440 has the ability to conduct current unidirectionally. The first diode D1 rectifies the output current of the secondary winding 220. Utilizing the charging and discharging characteristics of the first diode D1, energy is stored when the first diode D1 outputs current. When the first diode D1 is off, the output filter module 420 provides current to the downstream load RO, thereby filling voltage troughs and achieving a stable output voltage. When the downstream load RO requires a large current for a certain period, the output filter module 420 can quickly discharge to replenish the current energy gap and maintain voltage stability.

[0094] In some embodiments, combined with Figure 6 As shown, the input buffer module 410 includes an input capacitor Cin1, a primary winding 210 connected in series with the first switching module 430, a first end of the input capacitor Cin1 connected to the first end of the primary winding 210, and a second end of the input capacitor Cin1 connected to the first switching module 430.

[0095] In some embodiments, combined with Figure 6 As shown, the first switching module 430 includes a first switching transistor Q1, the drain of the first switching transistor Q1 is connected to the primary winding 210, and the source of the first switching transistor Q1 is connected to the primary ground GND1.

[0096] In some embodiments, combined with Figure 6 As shown, the output filter module 420 includes an output capacitor Co1. The anode of the output capacitor Co1 is connected to the first end of the secondary winding 220 via a first diode D1. The second end of the output capacitor Co1 and the second end of the secondary winding 220 are connected to the secondary ground GND2.

[0097] In some embodiments, combined with Figure 7 As shown, when the output power of the power supply circuit is large, there are a first interference source 610 and a second interference source 620 in the power supply circuit. The first interference source 610 exists in the input power line, while the second interference source 620 exists in the transformer 200. The interference in the transformer 200 can be reduced by generating a reverse current through the auxiliary winding 230. Therefore, suppressing the common-mode interference introduced by the power line has become an urgent technical problem to be solved.

[0098] To address interference issues on the power input side, see [link / reference]. Figure 8As shown, the power supply circuit also includes a magnetic ring filter 450 disposed between the power input terminal 600 and the transformer 200, the magnetic ring filter 450 being configured to filter the power input terminal 600.

[0099] In this embodiment, in low-power power supply applications (power output less than 75 watts), the transformer 200 in any of the above embodiments can eliminate most of the interference, ensuring the power supply meets EMC standards. However, in higher-power switching power supplies (e.g., power output greater than 75 watts), the power lines introduce common-mode interference. Therefore, the power lines are both a pathway for external grid interference to enter the device and a channel for internal interference to be emitted outwards. To eliminate power line interference, a magnetic ring filter 450 can be installed between the power input terminal 600 and the transformer 200. The magnetic ring filter 450 filters external or external interference channels, complementing the transformer's internal handling of the switching power supply's own interference, thus forming a complete two-stage filtering system to cope with more complex interference environments.

[0100] In some embodiments, the magnetic ring filter 450 includes a ferrite magnetic ring, and the power input terminal 600 includes a live wire, a neutral wire, and a ground wire, wherein the live wire and the neutral wire of the power input terminal 600 are wound around the ferrite magnetic ring, and the ground wire of the power input terminal 600 is not wound around the ferrite magnetic ring.

[0101] In this embodiment, the ferrite core exhibits high impedance characteristics at high frequencies, acting as a lossy inductor. The live wire (L) and neutral wire (N) of the power input terminal 600 are simultaneously wound in parallel on the same ferrite core. The ground wire (PE), serving as a discharge path for common-mode interference current, is directly grounded and not allowed to be wound on the core. When the live and neutral wires pass through the core together in the same direction, the magnetic fields generated by the differential-mode currents (normal operating currents) they carry are equal in magnitude and opposite in direction within the core, canceling each other out. Therefore, the core has very low impedance to differential-mode currents and does not affect normal operation. For common-mode interference currents (equal in magnitude and same in direction on the live and neutral wires), their magnetic fields are superimposed in the same direction within the core, making the ferrite core behave as a high-impedance element, thereby significantly attenuating the transmission of common-mode interference.

[0102] In some embodiments, where the number of turns of the auxiliary winding 230 is roughly determined in the above embodiments, this application also proposes a test circuit. This test circuit is applied to the power supply circuit of any of the above embodiments. Through this test circuit, the specific number of turns of the auxiliary winding 230 can be further determined, and it can be judged whether the power supply circuit conforms to EMC standards under the current setting of the number of turns of the auxiliary winding 230 of the transformer. See also Figure 9As shown, the test circuit includes: a first test terminal 710, which is configured to acquire the voltage of the common node of the first switching module 430 and the primary winding 210 to obtain a reference voltage signal.

[0103] In this embodiment, the voltage of the common node of the first switching module 430 and the primary winding 210 is used to obtain a reference voltage signal, which can be considered as the main source of native common-mode noise. Its rapidly changing dv / dt is coupled through the transformer parasitic capacitance to generate a native displacement current Ips. This native displacement current Ips is detected by the first test terminal 710 and used as the reference voltage signal.

[0104] In some embodiments, see Figure 9 As shown, the test circuit also includes a second test terminal 720, a third test terminal 730, a test module 741, and a differential sampling module 742. The second test terminal 720 and the third test terminal 730 are respectively configured to connect to the primary ground GND1 and the secondary ground GND2. The two ends of the test module 741 are respectively connected to the second test terminal 720 and the third test terminal 730, and are configured to establish an impedance loop between the primary ground GND1 and the secondary ground GND2. The differential sampling module 742 is connected to the second test terminal 720 and the third test terminal 730, and is configured to perform differential sampling on the second test terminal 720 and the third test terminal 730 to obtain a differential sampling voltage signal.

[0105] In this embodiment, the test module 741 is connected to the second test terminal 720 and the third test terminal 730 at both ends, thereby connecting the test module 741 between the primary ground GND1 and the secondary ground GND2. The acquired differential sampling signal is positively correlated with the net common-mode current flowing through the circuit. The net common-mode current is the sum of the original displacement current Ips and the compensation current Ibs generated by the auxiliary winding 230.

[0106] In some embodiments, see Figure 9 As shown, the test circuit also includes a main control module 750, which is connected to the first test terminal 710 and the differential sampling module 742. The main control module 750 is configured to compare the differential sampling voltage signal with the reference voltage signal, and if the differential sampling voltage signal and the reference voltage signal are in phase, it is determined that the number of turns of the auxiliary winding 230 meets the preset electromagnetic compatibility conditions.

[0107] In this embodiment, the main control module 750 compares the differential sampling voltage signal with the reference voltage signal. If the differential sampling voltage signal and the reference voltage signal are in phase, it means that the phase of the net common mode current is the same as the phase of the noise source signal. The compensation current Ibs cancels out the original displacement current Ips to a certain extent. At this time, the main control module 750 can determine that the number of turns of the auxiliary winding 230 meets the preset electromagnetic compatibility conditions.

[0108] In some embodiments, the main control module 750 is further configured to detect the peak-to-peak value of the differential sampling voltage signal, and if the peak-to-peak value is less than 3000mV, determine that the number of turns of the auxiliary winding 230 meets the preset electromagnetic compatibility conditions.

[0109] In this embodiment, when the common-mode noise is suppressed to below this level, the conducted emissions of the switching power supply have a high probability of meeting the requirements of common EMC standards such as CISPR 32 Class B. This voltage value (3000mV) can be converted into the magnitude of the common-mode current to ground by the test module 741 (e.g., 100kΩ). Therefore, when the peak-to-peak value is less than 3000mV, the degree of "undercompensation" is moderate, that is, the residual common-mode noise has been suppressed to a safe level sufficient to pass the conducted emissions (CE) test.

[0110] In some embodiments, the magnetic core 100 of the transformer 200 is connected to the primary ground GND1 via a first shielding layer 240 (copper foil). In the power supply circuit, the primary ground GND1 is not a "cold ground" connected to earth, but a "hot ground" directly connected to the high-voltage switching network, and its potential fluctuates drastically at a high frequency. The magnetic core 100 of the transformer 200 itself carries a high-voltage potential. If absolutely reliable insulation is not provided, contact with the magnetic core 100 by an operator or any potentially conductive component will result in an electric shock hazard, which is completely non-compliant with safety standards.

[0111] In this embodiment, the transformer 200 is wrapped in multiple layers with polyester film tape with a withstand voltage of up to 3000V. The effect is to establish a strong and reliable insulation barrier between the transformer's live parts (including the magnetic core and internal windings) and the external environment / personnel, preventing electric shock and meeting the requirements of safety standards such as IEC / EN 61558 and IEC / EN 60950 for reinforced or double insulation.

[0112] In some embodiments, the auxiliary winding 230 and the first shielding layer 240 are both connected to the primary ground GND1. In this case, the magnetic core 100 can be considered as a device on the primary side. Furthermore, the secondary winding 220 to the primary winding 210 and the secondary winding 220 to the magnetic core 100 need to meet the AC withstand voltage of 3000VAC and the leakage current of less than 3mA, so as to ensure the safety requirements (AC3.0KV / 3mA / 1min) under this application condition.

[0113] In some embodiments, the transformer 200 is wrapped with safety tape to prevent it from coming into contact with surrounding components.

[0114] In this embodiment, the insulating safety tape used around the transformer 200 suppresses the high-frequency electric field radiation of the winding to the outside to a certain extent, which indirectly helps to optimize radiated emissions (RE).

[0115] In some embodiments, the transformer 200 is surrounded by at least two layers of insulating safety tape, which can be made of polyester film, with a long-term temperature resistance of 130°C and an electrical strength (withstand voltage) of 3000V.

[0116] In some embodiments, the type of insulating safety tape used around the transformer 200 can be CT(b)(g)*.

[0117] In some embodiments, the main control module 750 is further configured to detect the compensation current Ibs flowing from the secondary winding 220 to the auxiliary winding 230 and the original displacement current Ips flowing from the primary winding 210 to the secondary winding 220 when the auxiliary winding 230 is powered on, and to determine that the number of turns of the auxiliary winding 230 meets the preset electromagnetic compatibility conditions if the compensation current Ibs is less than the original displacement current Ips and the absolute value of the difference between the compensation current Ibs and the original displacement current Ips is less than one-tenth of the original displacement current Ips.

[0118] In this embodiment, when the differential sampling voltage signal and the reference voltage signal are in phase, it indicates that the phase of the net common-mode current is the same as that of the noise source. This means that the amplitude of the compensation current Ibs is smaller than the original interference current Ips, but the directions are roughly the same, and the system is in an "undercompensated" state. If the two are out of phase, it indicates that the compensation current Ibs not only tries to cancel Ips, but also "overcompensates," producing reverse net interference, and the system is in an "overcompensated" state. "Slight undercompensation" is a better engineering state that balances conducted and radiated EMI. Therefore, the main control module 750 uses "phase consistency" as a key and necessary criterion for judging whether the compensation effect of the auxiliary winding 230 meets the standard.

[0119] In some embodiments, the resistance of the test module 741 is 80-120k ohms.

[0120] In some embodiments of this application, the test module 741 is connected between the primary ground GND1 and the secondary ground GND2, providing a signal path for measurement. By setting the test module 741 to have a certain resistance value, the current signal that cannot be directly measured can be converted into a measurable voltage signal. When the switching power supply is working, the displacement current Ips of the primary winding 210 to the secondary winding and the compensation current Ibs of the secondary winding 220 to the auxiliary winding 230 will form a loop. According to Ohm's law, the net common-mode current flowing through this resistor will generate a voltage drop across it. The magnitude and waveform of the differential sampling voltage signal directly reflect the characteristics of the net common-mode current. By measuring this voltage through the differential sampling module 742 (e.g., a differential probe), the cancellation effect of the displacement current can be indirectly and quantitatively evaluated.

[0121] In some embodiments, see Figure 10 As shown, Figure 10 This is a schematic diagram of a testing device used in a transformer 200. In the switching power supply, the primary winding 210 is connected in series with the first switching module 430. The output terminal of the first switching module 430 and the first terminal of the input capacitor Cin1 are connected to the primary ground GND1. The other terminal of the input capacitor Cin1 is connected to the power input terminal 600. The first terminal of the secondary winding 220 is connected to the first terminal of the output capacitor Co1 and the first terminal of the load Ro via an output diode. The second terminal of the secondary winding 220, the second terminal of the output capacitor Co1, and the second terminal of the load Ro are connected to the secondary ground GND2.

[0122] In this embodiment, a test resistor Rtest with a preset resistance value is connected in parallel between the primary ground GND1 and the secondary ground GND2. The voltage waveform between the primary ground GND1 and the secondary ground GND2 is measured by a differential probe 700. The differential probe 700 uploads the measured voltage waveform to the main control module 750. The main control module 750 defines the voltage waveform between the primary ground GND1 and the secondary ground GND2 as the measured waveform. The main control module 750 also acquires the voltage waveform of the common node between the first switching module 430 and the primary winding 210, and uses this voltage waveform as the reference voltage waveform. The main control module 750 determines the phase relationship between the measured voltage waveform and the reference voltage waveform. If the two waveforms are similar, it is determined to be an undercompensated state, and the number of turns of the auxiliary winding 230 should be reduced. If the two waveforms are opposite, it is determined to be an overcompensated state, and the number of turns of the auxiliary winding 230 should be increased.

[0123] After adjusting the number of turns of the auxiliary winding 230, the voltage waveform between the primary ground GND1 and the secondary ground GND2 is measured by the differential probe 700 to obtain the measured voltage waveform. The main control module 750 then determines the phase relationship between the measured voltage waveform and the reference voltage waveform until the measured voltage waveform is similar to the reference voltage waveform and the voltage amplitude of the measured voltage waveform and the reference voltage waveform is less than the preset threshold. In this case, the corresponding EMC standard is met.

[0124] In some embodiments, the number of turns of the auxiliary winding 230 should be set to an undercompensated state, that is, the measured voltage waveform is similar to the reference voltage waveform, but has a certain voltage amplitude difference. A slight undercompensated state is beneficial to radiated EMI.

[0125] In this embodiment, overcompensation means that the compensation current Ibs is greater than the original current Ips. This actually creates a new, reverse net interference source, which may cause EMC tests to exceed the limits at other frequencies, or even bring unpredictable noise problems. In this embodiment, by selecting a controllable and slight undercompensation state, the uncertainty risk brought about by overcompensation is actively avoided.

[0126] In some embodiments, the preset threshold can be 2000mV-3000mV.

[0127] In some embodiments, the preset threshold can be 3000mV or 2500mV.

[0128] In some embodiments, combined with Figure 11 As shown, a first shielding layer 240 is provided on the surface of the magnetic core 100, which is directly connected to the primary ground GND1. The first shielding layer 240 blocks the direct interference path from the primary winding 210 to the magnetic core 100, so that the displacement current Icg of the magnetic core 100 to ground is 0.

[0129] In some embodiments, combined with Figure 10 As shown, the auxiliary winding 230 is wrapped around the secondary winding 220 and is also connected to the primary ground GND1. It not only blocks the interference from the secondary to the magnetic core 100, but more importantly, it forms a low-impedance "bridge" with the primary ground GND1.

[0130] In this embodiment, the interference generated on the primary side is iph1. Due to the common-mode interference generated when the primary winding 210 is working, part of it flows to the magnetic core 100 through the parasitic capacitance. However, the first shielding layer 240 clamps the potential of the magnetic core 100 to the primary ground GND1, so the interference cannot escape through the magnetic core 100. The interference generated on the secondary side is lsh1. The interference current generated by the secondary winding 220 would normally flow into the ground through the parasitic capacitance Csg. However, due to the presence of the auxiliary winding 230, which acts as a secondary shielding layer, the interference current is forced to change course after reaching the secondary shielding layer and flows back to the primary ground GND1 along the red dotted line. At this time, the interference current forms an internal loop between the auxiliary winding 230 → primary ground GND1 → first shielding layer 240 → primary winding 210. By utilizing its special winding structure (primary-auxiliary-secondary) and specific turn ratio, an active compensation network is constructed inside the transformer. The voltage Vb generated by the auxiliary winding 230 induces a reverse current Ibs on the parasitic capacitance Cbs, thereby canceling the interference current Isg that would normally flow to the ground. This achieves the effect of eliminating the Y capacitance to ground while still meeting EMC standards.

[0131] Some embodiments of this application also provide a power supply circuit, which includes the test circuit in any of the above embodiments. In this test circuit, the internal test module 741 includes a test switch. By controlling the switching state of the test switch, the test switch can be turned on in the test mode, thereby enabling the detection of the transformer 200.

[0132] In some embodiments, the test module 741 includes a test switch and a test resistor Rtest.

[0133] In this embodiment, a test resistor Rtest with a preset resistance value is connected in parallel between the primary ground and the secondary ground, and the voltage waveform between the primary ground and the secondary ground is measured by a differential probe 700. The differential probe 700 uploads the measured voltage waveform to the main control module. The main control module defines the voltage waveform between the primary ground and the secondary ground as the measured waveform. The main control module also acquires the voltage waveform of the common node between the first switch and the primary winding 210, and uses this voltage waveform as the reference voltage waveform. The main control module determines the phase relationship between the measured voltage waveform and the reference voltage waveform. If the two waveforms are similar, it is determined to be an undercompensated state, and the number of turns of the auxiliary winding 230 should be reduced. If the two waveforms are opposite, it is determined to be an overcompensated state, and the number of turns of the auxiliary winding 230 should be increased.

[0134] After adjusting the number of turns of the auxiliary winding 230, the voltage waveform between the primary ground and the secondary ground is measured through the differential probe 700 to obtain the measured voltage waveform. The main control module then determines the phase relationship between the measured voltage waveform and the reference voltage waveform until the measured voltage waveform is similar to the reference voltage waveform and the voltage amplitude of the measured voltage waveform and the reference voltage waveform is less than the preset threshold. In this case, the corresponding EMC standard is met.

[0135] In some embodiments, the number of turns of the auxiliary winding 230 should be set to an undercompensated state, that is, the measured voltage waveform is similar to the reference voltage waveform, but has a certain voltage amplitude difference. A slight undercompensated state is beneficial to radiated EMI.

[0136] In this embodiment, overcompensation means that the compensation current Ibs is greater than the original current Ips. This actually creates a new, reverse net interference source, which may cause EMC tests to exceed the limits at other frequencies, or even bring unpredictable noise problems. In this embodiment, by selecting a controllable and slight undercompensation state, the uncertainty risk brought about by overcompensation is actively avoided.

[0137] In some embodiments, the preset threshold can be 2000mV-3000mV.

[0138] In some embodiments, the preset threshold can be 3000mV or 2500mV.

[0139] In some embodiments, combined with Figure 4 As shown, a first shielding layer 240 is provided on the surface of the magnetic core 100 and is directly connected to the primary ground. The first shielding layer 240 blocks the direct interference path from the primary winding to the magnetic core, so that the displacement current Icg of the magnetic core 100 to ground is 0.

[0140] In some embodiments, combined with Figure 4 As shown, the auxiliary winding 230 is wrapped around the outer layer of the secondary winding and is also connected to the primary ground. It not only blocks interference from the secondary to the magnetic core, but more importantly, it forms a low-impedance "bridge" with the primary ground.

[0141] In this embodiment, the interference generated on the primary side is iph1. Due to the common-mode interference generated when the primary winding is working, part of it flows to the magnetic core through the parasitic capacitance. However, the magnetic core potential is clamped to the primary ground by the first shielding layer 240, so the interference cannot escape through the magnetic core 100. The interference generated on the secondary side is lsh1. The interference current generated by the secondary winding 220 would normally flow into the ground through the parasitic capacitance Csg. However, due to the presence of the auxiliary winding 230, which acts as a secondary shielding layer, the interference current is forced to change course after reaching the secondary shielding layer and flows back to the primary ground along the red dotted line. At this time, the interference current forms an internal loop between the auxiliary winding 230 → primary ground GND1 → first shielding layer 240 → primary winding 210. By utilizing its special winding structure (primary-auxiliary-secondary) and specific turn ratio, an active compensation network is constructed inside the transformer. The voltage Vb generated by the auxiliary winding induces a reverse current Ibs on the parasitic capacitance Cbs, thereby canceling the interference current Isg that would normally flow to the ground. This achieves the effect of eliminating the Y capacitance to ground while still meeting EMC standards.

[0142] In some embodiments, the differential sampling module 742 includes a differential sampling switch. In test mode, the differential sampling switch is turned on to perform differential sampling on the second test terminal 720 and the third test terminal 730 to obtain a differential sampling voltage signal. The main control module 750 compares the differential sampling voltage signal with the reference voltage signal. If the differential sampling voltage signal and the reference voltage signal are in phase, it is determined that the number of turns of the auxiliary winding 230 meets the preset electromagnetic compatibility conditions. In this way, after the power supply circuit has been working for a period of time, the transformer 200 can be tested by the test circuit to determine whether it meets the preset electromagnetic compatibility conditions.

[0143] Some embodiments of this application also provide an air conditioning device, including a switching power supply as described in any of the above embodiments, the switching power supply being used to power the control board of the air conditioning device.

[0144] Other configurations and operations of the switching power supply according to some embodiments of this application are known to those skilled in the art and will not be described in detail here.

[0145] In the description of this specification, the references to terms such as "one embodiment," "some embodiments," "illustrative embodiment," "example," "specific example," or "some examples," etc., refer to specific features, structures, materials, or characteristics described in connection with that embodiment or example, which are included in at least one embodiment or example of this application. In this specification, the illustrative expressions of the above terms do not necessarily refer to the same embodiment or example.

[0146] Although embodiments of this application have been shown and described, those skilled in the art will understand that various changes, modifications, substitutions and alterations can be made to these embodiments without departing from the principles and spirit of this application, the scope of which is defined by the claims and their equivalents.

Claims

1. A transformer, characterized in that, The transformer includes: A magnetic core, a primary winding, and a secondary winding, wherein the primary winding and the secondary winding are disposed on the magnetic core; A first shielding layer is disposed between the magnetic core and the primary winding and electrically connected to the primary ground, for clamping the potential of the magnetic core to the potential of the primary ground; An auxiliary winding is disposed between the primary winding and the secondary winding. The corresponding terminal of the auxiliary winding is opposite to the corresponding terminal of the primary winding. The first terminal of the auxiliary winding is suspended, and the second terminal of the auxiliary winding is electrically connected to the primary ground. Wherein, the number of turns in the auxiliary winding is Nb, the number of turns in the primary winding is Np, the number of turns in the secondary winding is Ns, and 2Nb≤|3Ns-Np|.

2. The transformer according to claim 1, characterized in that, 3Nb≥|(10Ns-3Np) / 7|.

3. A transformer, characterized in that, The transformer includes: A magnetic core, a primary winding, and a secondary winding, wherein the primary winding and the secondary winding are disposed on the magnetic core; A first shielding layer is disposed between the magnetic core and the primary winding and electrically connected to the primary ground to clamp the magnetic core potential to the primary ground potential, thereby eliminating the displacement current of the magnetic core to ground. An auxiliary winding is disposed between the primary winding and the secondary winding. The corresponding terminal of the auxiliary winding is opposite to the corresponding terminal of the primary winding. The first terminal of the auxiliary winding is suspended, and the second terminal of the auxiliary winding is electrically connected to the primary ground. The compensation current Ibs flowing from the secondary winding to the auxiliary winding is in the opposite direction to the original displacement current Ips flowing from the primary winding to the secondary winding, and the absolute value of the difference between the compensation current Ibs and the original displacement current Ips is less than one-tenth of the original displacement current Ips.

4. The transformer according to claim 3, characterized in that, The compensation current Ibs is equal to the original displacement current Ips, and the total displacement current Isg of the secondary winding to ground is zero. The compensation current Ibs is less than the original displacement current Ips.

5. The transformer according to claim 3, characterized in that, The distance between the auxiliary winding and the secondary winding is d1, and the distance between the primary winding and the secondary winding is d2, where 5 / 3d1≤d2≤7 / 3d1.

6. The transformer according to any one of claims 1-5, characterized in that, The distributed capacitance Cps of the primary winding to the secondary winding is inversely proportional to the distance d2 between the primary and secondary windings; and / or The distributed capacitance Cps of the primary winding to the secondary winding is directly proportional to the relative area Sps between the primary winding and the secondary winding.

7. The transformer according to any one of claims 1-5, characterized in that, The distributed capacitance Cbs of the auxiliary winding to the secondary winding is inversely proportional to the distance d1 between the auxiliary winding and the secondary winding; and / or The distributed capacitance Cbs of the auxiliary winding to the secondary winding is directly proportional to the relative area Sbs between the auxiliary winding and the secondary winding.

8. A switching power supply, characterized in that, The transformer includes any one of claims 1-7, and a magnetic ring filter disposed between the power input terminal and the transformer, the magnetic ring filter being configured to filter the power input terminal.

9. The switching power supply according to claim 8, characterized in that, The magnetic ring filter includes a ferrite magnetic ring, and the power input terminal includes a live wire, a neutral wire, and a ground wire. The live wire and neutral wire of the power input terminal are wound around the ferrite magnetic ring, while the ground wire of the power input terminal is not wound around the ferrite magnetic ring.

10. An air conditioning device, characterized in that, Includes the switching power supply as described in claim 8 or 9, the switching power supply being used to power the control board of the air conditioning equipment.

Images