A control method for permanent magnet synchronous motor

Abstract

This invention belongs to the field of permanent magnet synchronous motor speed control technology, and provides a disturbance rejection control method for permanent magnet synchronous motors. The method includes: establishing a mathematical model and mechanical motion equations of the permanent magnet synchronous motor in a d-q synchronous rotating coordinate system; simultaneously inputting the error e between the electrical angle setpoint and the observed electrical angle to a linear PI control channel and a nonlinear variable damping channel, outputting a q-axis current reference component; constructing a manifold characteristic operator, and adjusting the gain of the extended state observer in real time according to the manifold characteristic operator, outputting a real-time observed value of the total system disturbance; converting the real-time observed value into a feedforward current compensation component, and subtracting it from the q-axis component of the actual current to obtain the final q-axis control quantity; inputting the final q-axis control quantity to the current loop to achieve disturbance rejection control of the permanent magnet synchronous motor. This invention achieves smooth start-up without sacrificing response speed and possesses operating condition awareness capabilities.

Description

Technical Field

[0001] This invention relates to the field of speed control technology for permanent magnet synchronous motors, and specifically to a disturbance rejection control method for permanent magnet synchronous motors. Background Technology

[0002] Permanent magnet synchronous motors (PMSMs) are widely used in high-end manufacturing fields such as industrial robots and electric vehicles due to their high power density and high efficiency. Traditional speed loop control typically employs PI controllers or linear extended state observers (LESOs), but in high-performance applications, existing technologies suffer from the following drawbacks: 1. The contradiction between overshoot and speed: In order to pursue fast start-up, traditional PI controllers often require a large proportional gain, which inevitably leads to speed overshoot, causing mechanical shock and wear.

[0003] 2. The contradiction between bandwidth and noise: The observation bandwidth of traditional LESO is usually fixed. If the bandwidth is set too large, it will introduce a lot of high-frequency measurement noise, resulting in electromagnetic torque pulsation; if the bandwidth is too small, there will be a lag in the observation of load changes, resulting in a large drop in speed.

[0004] 3. Limitations of algorithm complexity: Although there are solutions in the existing technology that use intelligent algorithms such as neural networks to optimize parameters, their computational load is huge and it is difficult to run in real time in low-cost embedded chips.

[0005] Therefore, there is an urgent need for a control strategy that is computationally simple, has clear physical meaning, and can simultaneously resolve the aforementioned contradictions. Summary of the Invention

[0006] To address the shortcomings of existing technologies, this invention provides a disturbance rejection control method for permanent magnet synchronous motors, which enables smooth start-up without sacrificing response speed and provides operational condition sensing capabilities.

[0007] This invention provides a disturbance rejection control method for a permanent magnet synchronous motor, comprising: Establish the mathematical model and mechanical motion equations of the permanent magnet synchronous motor in the dq synchronous rotating coordinate system; Constructing a dual-mode variable damping speed controller: The electrical angle setpoint... With the observed electrical angle The error e is simultaneously input to both the linear PI control channel and the nonlinear variable damping channel, outputting the q-axis current reference component. ; Constructing an extended state observer based on an error manifold: This involves constructing a manifold characteristic operator and adjusting the gain of the extended state observer in real time based on the manifold characteristic operator. Output the real-time observation of the total system disturbance. ; Real-time observations Converted into feedforward current compensation component and the q-axis component of the actual current The final q-axis control value is obtained by subtraction. ; final q-axis control quantity The input is fed into the current loop to achieve disturbance rejection control of the permanent magnet synchronous motor.

[0008] As can be seen from the above technical solution, the disturbance rejection control method for permanent magnet synchronous motors provided by the present invention constructs a dual-mode variable damping controller, and realizes energy dissipation and overshoot suppression in the start-up stage by introducing a nonlinear damping term with respect to the error; constructs an ESO based on the error manifold characteristics, and uses the manifold operator to sense the operating conditions in real time and adaptively adjust the observation gain.

[0009] Optionally, the mathematical model of the permanent magnet synchronous motor in the dq synchronous rotating coordinate system is as follows: ;in, , Here are the d-axis and q-axis components of the actual current, and is the phase resistance of the motor. , For d-axis and q-axis inductance, Electric angular velocity, It is a permanent magnet flux chain. , These are the d-axis and q-axis voltages; The electromagnetic torque equation is: ;in, For electromagnetic torque, It is the extreme logarithm; when using Under the vector control strategy, the electromagnetic torque equation simplifies to: ; The equation of motion for the machine is: ;in, The moment of inertia of the motor. For load torque, The coefficient of viscous friction, This represents the mechanical angular velocity of the motor.

[0010] Optionally, the control law of the dual-mode variable damping speed controller is: , in, , The proportional integral coefficient, The variable damping gain coefficient is... It is a nonlinear function. The core design parameter for the nonlinear intensity of the fal function is... This is the filter factor.

[0011] Optionally, the manifold feature operator , Output rotational speed for the observer Compared with actual speed The residual, express The first derivative, This indicates taking the absolute value, where c is the weighting coefficient; Observer Gain Follow Dynamic adjustment, specifically: , denoted by , where k represents the base gain and k represents the gain adjustment coefficient.

[0012] Optionally, the state equation of the extended state observer is: , In the formula, b is the gain parameter. , For the observer's base gain, , For the output of the extended state observer, , They are respectively , With respect to the first derivative over time .

[0013] Optionally, feedforward current compensation component ; Final q-axis control quantity .

[0014] By adopting the above technical solution, this application has the following beneficial effects: 1. Completely eliminate overshoot: Through a variable damping mechanism, the speed overshoot is suppressed to zero without sacrificing response speed, thus achieving smooth start-up; 2. Intelligent disturbance rejection: By utilizing the error manifold characteristics, the observer is equipped with "operating condition perception" capability, achieving an adaptive effect of "noise filtering in static conditions and disturbance rejection in dynamic conditions"; 3. Improved power quality: Compared with traditional methods, the present invention significantly reduces electromagnetic torque pulsation and stator current harmonics, which helps to extend the life of the motor. Attached Figure Description

[0015] To more clearly illustrate the specific embodiments of the present invention or the technical solutions in the prior art, the accompanying drawings used in the description of the specific embodiments or the prior art will be briefly introduced below. In all the drawings, similar elements or parts are generally identified by similar reference numerals. In the drawings, the elements or parts are not necessarily drawn to scale.

[0016] Figure 1 A flowchart of a disturbance rejection control method for a permanent magnet synchronous motor provided by an embodiment of the present invention is shown; Figure 2 A schematic diagram of a permanent magnet synchronous motor control system provided in an embodiment of the present invention is shown; Figure 3 The waveform diagram shows the dynamic response of the rotational speed under the traditional PI control strategy. Figure 4 The image shows the dynamic response waveform of the rotational speed under the dual-modal control strategy provided in this embodiment of the invention. Figure 5 The waveform diagram of electromagnetic torque pulsation under traditional PI control; Figure 6 The electromagnetic torque optimization waveform diagram is provided under the strategy in the embodiment of the present invention; Figure 7 The waveform diagram of the three-phase stator current under traditional PI control; Figure 8 The three-phase stator current waveform diagram under the strategy provided in the embodiment of the present invention. Detailed Implementation

[0017] The embodiments of the technical solution of the present invention will now be described in detail with reference to the accompanying drawings. These embodiments are only used to more clearly illustrate the technical solution of the present invention and are therefore merely examples, and should not be construed as limiting the scope of protection of the present invention. It should be noted that, unless otherwise stated, the technical or scientific terms used in this application should have the ordinary meaning as understood by one of ordinary skill in the art to which this invention pertains.

[0018] In one embodiment, such as Figure 1 As shown, a disturbance rejection control method for a permanent magnet motor is provided, including: S1. Establish the mathematical model and mechanical motion equations of the permanent magnet synchronous motor in the dq synchronous rotating coordinate system.

[0019] (1) The mathematical model of the permanent magnet synchronous motor in the dq synchronous rotating coordinate system is: ; in, , Here are the d-axis and q-axis components of the actual current, and is the phase resistance of the motor. , For d-axis and q-axis inductance, Electric angular velocity, It is a permanent magnet flux chain. , These are the d-axis and q-axis voltages.

[0020] (2) The electromagnetic torque equation is: ; in, For electromagnetic torque, It is the extreme logarithm; when using Under the vector control strategy, the electromagnetic torque equation simplifies to: ; (3) The equation of motion for the machine is: ; in, The moment of inertia of the motor. For load torque, The coefficient of viscous friction, This represents the mechanical angular velocity of the motor.

[0021] S2. Construct a dual-mode variable damping speed controller: Set the electrical angle setpoint... With the observed electrical angle The error e is simultaneously input to both the linear PI control channel and the nonlinear variable damping channel, outputting the q-axis current reference component. ; The control law of the dual-mode variable damping speed controller is: , in, , The proportional integral coefficient, The variable damping gain coefficient is... It is a nonlinear function. The core design parameter for the nonlinear intensity of the fal function is... , where is the filtering factor. When the error is large, the nonlinear term dominates, and the system exhibits a high-damping mode to suppress overshoot; when the error is driven to zero, the linear term dominates, and the system exhibits a zero-steady-state-error mode.

[0022] As the basic steady-state control channel of the system, the linear PI control channel is responsible for the steady-state speed regulation function without steady-state error, ensuring the speed control accuracy of the motor under steady-state conditions, maintaining the steady-state energy balance of the system, and fully retaining the advantages of the excellent steady-state performance of traditional PI control.

[0023] As the core channel for transient regulation of the system, the nonlinear variable damping channel dynamically generates a damping injection term based on the amplitude of the real-time speed error through a nonlinear feedback mechanism, adaptively adjusting the damping characteristics of the system and solving the overshoot problem in the transient process.

[0024] The system monitors the speed error e in real time. When the error is large (during startup or transients), the controller automatically enters the "high-damping mode" and injects a nonlinear damping current component through the NLSEF module. , In a physical sense, this is equivalent to increasing the system's viscous friction coefficient, thereby absorbing excess energy during motor acceleration and eliminating overshoot from a physical mechanism perspective.

[0025] S3. Constructing an extended state observer based on the error manifold: Constructing manifold characteristic operators The gain of the extended state observer is adjusted in real time based on the manifold characteristic operator. Output the real-time observation of the total system disturbance. ; manifold characteristic operators , Output rotational speed for the observer Compared with actual speed The residual, express The first derivative of , where c is the weighting coefficient.

[0026] By defining the residual between the observed speed output by the observer and the actual feedback speed of the motor as the core observed variable, an error manifold characteristic operator is constructed based on this residual and its dynamic characteristics. This operator can quantify the current operating conditions of the system in real time, accurately identify different operating conditions such as steady-state operation, load abrupt changes, and parameter perturbations, and provide a basis for adaptive adjustment of the observation bandwidth. When the load torque TL experiences a step change, When a sudden change occurs, the observer gain automatically slides along the manifold to achieve time-delay-free estimation and compensation of the total disturbance.

[0027] Observer Gain Follow Dynamic adjustment, specifically: , Let represent the base gain, and k represent the gain adjustment coefficient. This ensures that under sudden load changes (… The observation bandwidth is automatically widened under steady-state conditions. (Approaching zero) The bandwidth automatically shrinks to filter out noise. For the speed control system of permanent magnet synchronous motor, all internal and external disturbances affecting the speed control performance, such as sudden changes in load torque, motor parameter perturbations, and unmodeled dynamics of the system, are combined and defined as the total disturbance of the system. This total disturbance is set as a new extended state of the system, and a mathematical model of the extended state observer containing this extended state is constructed to provide a model basis for the real-time observation of the total disturbance.

[0028] The state equation of the extended state observer is: , In the formula, b is the gain parameter. , For the observer's base gain, , For the output of the extended state observer, , They are respectively , The first derivative over time (i.e., the real-time observation of the electrical angle and the total disturbance of the system in the next stage). .

[0029] S4. Real-time observation values Converted into feedforward current compensation component and the q-axis current component The final q-axis control value is obtained by subtraction. ; final q-axis control quantity The input is fed into the current loop to achieve disturbance rejection control of the permanent magnet synchronous motor.

[0030] Feedforward current compensation component Ultimately, the q-axis control quantity .

[0031] To verify the convergence of the dual-modal variable damping control strategy proposed in this embodiment, the following derivation is based on Lyapunov stability theory.

[0032] Define the speed tracking error as According to the PMSM equations of motion, the system's error dynamic equation can be described as: , In the dual-modal control law design of this invention, the feedforward compensation term has already offset the observed disturbance. Therefore, the closed-loop error equation of the system can be reconstructed as: , (1) Constructing Lyapunov candidate functions Choose a positive definite Lyapunov function for: , Obviously, for any All have ,and .

[0033] (2) Deriving the derivative of the function right Find the time derivative : , Expanding, we get: , (3) Convergence test Examine the sign characteristics of the two terms on the right side of the above equation: First item because The first term is always negative (a dissipation term). For the second term, according to the definition of the fal function: , e represents the input error, which is moderate. The nonlinear factor is a core design parameter for the nonlinear strength of the fal function, and its value range is strictly limited to [value range missing]. . This is the filter factor.

[0034] It can be known that e and Same number, that is Therefore, as long as the damping coefficient The second item It is also always wrong.

[0035] In conclusion, when hour, .

[0036] According to the Lyapunov stability criterion, the system is asymptotically stable over a wide range at the equilibrium point e=0. This means that no matter how large the initial error is, the controller provided in this embodiment can ensure that the speed error converges to zero, theoretically verifying the inevitability of the "no overshoot" characteristic.

[0037] To verify the effectiveness of the embodiments of the present invention, a PMSM vector control system was built in the MATLAB / Simulink environment, and a comparative experiment was conducted with the traditional PI control strategy.

[0038] Figure 3 The speed response under the traditional PI control strategy is shown, and compared. Figure 4 The speed response of the embodiment of the present invention shown in the figure exhibits that traditional PI control suffers from overshoot during the startup phase, and the speed drops significantly during a sudden load change of 0.3s. However, after applying the strategy of the present invention, the speed curve shows a perfectly smooth upward trend, completely eliminating overshoot, and the recovery time under load disturbance is shortened by about 40%, proving that the dual-modal mechanism effectively improves the dynamic stiffness of the system.

[0039] Figure 5 The electromagnetic torque waveform under the traditional PI control strategy is shown, and compared. Figure 6 As can be seen from the electromagnetic torque waveform of the embodiment of the present invention, under conventional control, the electromagnetic torque exhibits severe high-frequency oscillations, which exacerbates the mechanical wear and noise of the motor. However, in contrast,… Figure 6 In this embodiment of the invention, the error manifold (ESO) enables smooth observation of disturbances, significantly reducing high-frequency noise and making the torque waveform more stable, which is beneficial to extending the service life of the motor.

[0040] The quality of the current waveform directly reflects the steady-state accuracy of the control algorithm. Figure 7 The diagram shows the three-phase stator current under a traditional PI control strategy. The current contains numerous glitches and exhibits a high harmonic distortion (THD). (Comparison) Figure 8 The three-phase stator currents shown in this invention exhibit better sinusoidal characteristics and purer waveforms. This indicates that the invention effectively suppresses current harmonics, reduces iron losses and temperature rise in the motor, and improves the system's energy efficiency ratio.

[0041] The above embodiments are only used to provide a detailed description of the technical solutions of this application. However, the descriptions of the above embodiments are only for the purpose of helping to understand the methods of the embodiments of the present invention and should not be construed as limiting the embodiments of the present invention. Any variations or substitutions that can be easily conceived by those skilled in the art should be covered within the protection scope of the embodiments of the present invention.

Claims

1. A disturbance rejection control method for a permanent magnet synchronous motor, characterized in that, include: Establish the mathematical model and mechanical motion equations of the permanent magnet synchronous motor in the dq synchronous rotating coordinate system; Constructing a dual-mode variable damping speed controller: The electrical angle setpoint... With the observed electrical angle The error e is simultaneously input to both the linear PI control channel and the nonlinear variable damping channel, outputting the q-axis current reference component. ; Constructing an extended state observer based on an error manifold: This involves constructing a manifold characteristic operator and adjusting the gain of the extended state observer in real time based on the manifold characteristic operator. Output the real-time observation of the total system disturbance. ; Real-time observations Converted into feedforward current compensation component and the q-axis component of the actual current The final q-axis control value is obtained by subtraction. ; final q-axis control quantity The input is fed into the current loop to achieve disturbance rejection control of the permanent magnet synchronous motor.

2. The method according to claim 1, characterized in that, The mathematical model of the permanent magnet synchronous motor in the dq synchronous rotating coordinate system is as follows: ;in, , Here are the d-axis and q-axis components of the actual current, and is the phase resistance of the motor. , For d-axis and q-axis inductance, Electric angular velocity, It is a permanent magnet flux linkage. , These are the d-axis and q-axis voltages; The electromagnetic torque equation is: ;in, For electromagnetic torque, It is the extreme logarithm; when using Under the vector control strategy, the electromagnetic torque equation simplifies to: ; The equation of motion for the machine is: ;in, For the moment of inertia of the motor, For load torque, The coefficient of viscous friction, This represents the mechanical angular velocity of the motor.

3. The method according to claim 2, characterized in that, The control law of the dual-mode variable damping speed controller is: ， in, , The proportional integral coefficient, The variable damping gain coefficient is... It is a nonlinear function. The core design parameter for the nonlinear intensity of the fal function is... This is the filter factor.

4. The method according to claim 3, characterized in that, The manifold feature operator , Output rotational speed for the observer Compared with actual speed The residual, express The first derivative, This indicates taking the absolute value, where c is the weighting coefficient; Observer Gain Follow Dynamic adjustment, specifically: ， denoted by , where k represents the base gain and k represents the gain adjustment coefficient.

5. The method according to claim 4, characterized in that, The state equation of the extended state observer is: ， Where b is the gain parameter, , For the observer's base gain, , For the output of the extended state observer, .

6. The method according to claim 5, characterized in that, Feedforward current compensation component ; Final q-axis control quantity .

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