Printed circuit board, method for determining a current space vector, converter, printed circuit board and series of converters

Abstract

Printed circuit board for an inverter, comprising control electronics, wherein the printed circuit board includes at least means for current sensing, characterized by the fact that The means each comprise a first and a second measuring amplifier circuit arranged in parallel to this first, wherein the first measuring amplifier circuit has a different gain than the second measuring amplifier circuit. wherein only the output of the first measuring amplifier circuit or only the output of the second measuring amplifier circuit is supplied as a measured current value to the control electronics of the inverter, where the first measuring amplifier circuit and the second measuring amplifier circuit are implemented in hardware.

Description

[0001] The invention relates to a printed circuit board, a method for determining a current space vector, a converter, a printed circuit board and a series of converters.

[0002] In a digitally controlled, pulse-width modulated (PWM) output voltage converter, achieving sufficient control accuracy requires determining a current space vector for each PWM period that corresponds to the average value of the actual current space vector calculated over that PWM period. With an inductive load, such as an electric motor, the PWM output voltage results in a current ripple. The current space vector must be determined without being affected by this current ripple. In some state-of-the-art converters, the current sensors are located in the motor leads, resulting in continuous current signals. These signals are filtered analogously to remove the current ripple and can then be used for control.

[0003] From the book “Practice of field-oriented three-phase drive control” by Nguyen Phung Quang and Jörg-Andreas Dittrich, second edition, from 1999, pulse-width modulated converters are known from pages 110-113, in which current sensing is carried out using three ( Fig. .1) or even just two ( Fig. .3, 5.5, 5.6, 5.9, 5.11) are provided with potential-isolating current sensors in the output branches. A current space vector with two degrees of freedom is determined from the current measurements ( Fig. .3, 5.5, 5.6, 5.9, 5.11). Furthermore, pages 76 to 77 explain that to detect the fundamental frequency of the motor currents without harmonics, i.e., without current ripple, the current measurements must be taken at suitable times. The deviation from the fundamental frequency is referred to as the differential current space vector. The current measurements are then taken at the zero crossing of the differential current space vector. A disadvantage of these known converters is that at least two expensive, galvanically isolated current measuring devices are required.

[0004] From DE 196 81 189 T1 is a pulse-width modulated converter with intermediate circuit capacitor ( Fig. 1, reference numeral 3) and a connected output stage are known, which includes only a current sensor for detecting the intermediate circuit current. Depending on the angle of the output voltage phasor, the current detection thus provides information about a single phase current or sums of phase currents. A disadvantage is that, at certain angles of the output voltage phasor, determining a current space vector is not satisfactorily feasible.

[0005] From the publication by Francesco Parasiliti, "Low cost current sensing in DSP Based Drives," Industrial Electronics, 1999, ISIE'99, Proceedings of the IEEE, Volume 3, 1999, International Symposium, Volume 3, 1999, a converter is known in which shunt resistors are arranged in all lower branches of the half-bridges as a means of sensing the respective currents. All three current measurements are taken, with the measurement impulses occurring in the middle of the time interval within a pulse-width modulation period in which the discrete switching state (000), i.e., the zero vector v0, is present (page 1287, left column). Fig. 7) In the actual implementation, there is a slight time offset from this center point due to the consideration of dead times, signal propagation delays, and the like. The current space vector is then determined from the corresponding measured values. A disadvantage of this, however, is that in the shaded areas of the hexagon (page 1287, left column, last section and Fig. 8) No or only an inaccurate determination of the current space vector is possible, since either no zero vector is used in these shaded areas or only for a very short time. If the zero vector is not present or is only present for a very short time, then automatically no or only a very short current measurement signal is present in a phase, which is therefore not available for evaluation or only available inaccurately. This results in the following: in the case of an output voltage space vector averaged over a pulse width modulation period from the shaded area (page 1287, Fig. 8) at least one current measurement is falsified.

[0006] From US 5 815 391 A, a converter is known in which means for detecting the respective currents are arranged in all lower branches of the half-bridges ( Fig. 2A). In the corresponding table ( Fig. 2B) explains that the current in phase A is measurable depending on the switching states (for example, line 1: measurable;) or can be calculated from the two other current measurements (line 5: calculable;). However, if the lower switch of phase A and another lower switch are open, the current measurement of phase A cannot be determined. Such switching states are marked as "unknown" for phase A in this document.

[0007] The aforementioned Table 2B and the described problem refer to the switching states, i.e., the respective instantaneous state of the inverter.

[0008] The aforementioned Table 2B refers only to phase A. However, corresponding tables for phases B and C can be compiled. It is readily apparent that for some switching states, namely (111), (110), (101), (011), no current measurement or only one current measurement can be determined, and therefore a current space vector cannot be calculated for these instantaneous states. US 5,815,391 A does not explain how to determine a current space vector that corresponds to the average value of the actual current space vector calculated over one pulse-width modulation period. In particular, it does not describe how to proceed so that the current space vector is not distorted by current ripple.

[0009] The duration of a "known" switching state within a pulse width modulation period depends on the following factors: Carrier frequency of pulse width modulation, The type of pulse width modulation method. If the duration of a "known" switching state is too short, the current space vector cannot be determined.

[0010] How the pulse-width modulation method is implemented is not explained in detail in this document; however, it is clear to those skilled in the art that the document teaches that these "unknown" states pose a problem. Column 1, lines 40-44, mentions a "backward calculation," but this is not described in a feasible manner. The document teaches those skilled in the art that the problem of the "unknown" states can be circumvented by following the procedure described below. Fig. 5 and Fig. 6. Measuring instruments, in particular potential-free measuring instruments such as Rogowski coils or the like, are used in both the upper and lower branches of the half-bridges. By cleverly combining the measurement signals, especially adding the measurement signals from the lower and upper measuring instruments, a continuous current measurement signal is achieved and the "unknown" states no longer pose a problem.

[0011] This principle from US 5,815,391 A, to use measuring instruments in both the upper and lower branches of half-bridges, would also be applicable to the aforementioned IEEE document. In this way, current measurement in the shaded areas would be easily and accurately feasible. However, this would require complex and therefore expensive measuring instruments, particularly those for overcoming the potential barrier. The solutions would thus be very complex and costly.

[0012] From DE 102 48 375 A1, a method for determining a current space vector is known as the closest prior art.

[0013] A motor control system is known from US 5 406 150 A.

[0014] The invention is therefore based on the objective of further developing a converter in which as many costly parts as possible can be dispensed with and, in particular, cost-effective manufacturing can be achieved in mass production, and yet a method for determining a current space vector can be carried out.

[0015] According to the invention, the problem is solved in the printed circuit board according to the features specified in claim 1, in the method for determining a current space vector according to claims 3, 12 or 13, in the converter according to the features specified in claim 9 and in the series according to the features specified in claim 14.

[0016] Important features of the invention in the method are that it is designed to determine a current space indicator, especially for use in a tax and / or regulatory procedure, for a pulse-width modulated inverter, comprising signal electronics, a power output stage with power switches arranged in three half-bridges, each having a lower and an upper branch, and in Means for scanning the respective currents arranged on the three upper or lower branches of the half-bridge, the scanning means can be operated with selectable different gains.

[0017] A key advantage is that the gain can be adjusted during operation as needed, thus adapting the current resolution to different operating conditions. Alternatively, a first gain can be activated and used with a first type of circuit breaker and / or shunt resistor, and a second gain with a second type of circuit breaker and / or shunt resistor. This means that the same control electronics with current sensing amplifier circuits can always be used within a product series – even if different circuit breakers are employed.

[0018] In a preferred embodiment, the means each comprise an amplifier circuit whose gain can be selectively adjusted to different values. The advantage here is that the gain can be adapted to the specific requirements.

[0019] In particular, the means each comprise two optionally operable amplifier circuits, each with a different gain.

[0020] In an advantageous design, the gain is switched during the ongoing control process, particularly to achieve better resolution in current measurement. A further advantage is that a suitable resolution can be selected depending on the operating state.

[0021] In an advantageous embodiment, the same sensing means are used for different circuit breakers, where the breakers can be used for different maximum permissible currents or power ratings. The advantage here is that a high variance of converters can be produced using a small number of parts and thus low storage costs.

[0022] In an advantageous embodiment, after modifying the amplification of the current sensing signals assigned to a respective phase, the offset and / or the amplification of the current sensing signal assigned to that phase is first determined and then taken into account. The advantage here is that the current sensing can be implemented more reliably, thus enabling trouble-free switching. Furthermore, the current sensing of the phase currents can be implemented more uniformly, resulting in improved control behavior.

[0023] In an advantageous embodiment, a current is impressed and measured in each bridge branch to determine the gain. The advantage here is that a defined current is provided, thus enabling a uniform determination of the gain in the various bridge branches assigned to the phases.

[0024] Important features of the inverter, especially the pulse-width modulated inverter, are that the inverter includes signal electronics, a power output stage with power switches arranged in three half-bridges, each having a lower and an upper branch, and means for sensing the respective currents arranged in the three upper or lower branches of the half-bridge. where the means for sampling can be operated with different gains depending on the state of a selection instrument.

[0025] In particular, each device comprises at least two amplifier circuits, with only one being operated at any given time. A key advantage is that hardware components, such as the circuit board with signal electronics and the current sensing amplifier, are reusable and can therefore be used with different power classes of semiconductor switches. This allows for a high degree of variability with a small number of parts.

[0026] In a preferred embodiment, the selection device is implemented as a selector switch or as another hardware-implemented device. The advantage here is that the switch is easy to implement.

[0027] In an advantageous embodiment, the selection method is implemented as a parameter value or other software-based method. The advantage here is that no additional hardware is required.

[0028] Key features of the printed circuit board are that it is designed for an inverter, wherein the inverter includes control electronics, wherein the printed circuit board includes at least means for current sensing, wherein the means each have at least two measuring amplifier circuits, wherein only the output of one of the measuring amplifier circuits is supplied as the sensed current value to the control electronics of the inverter.

[0029] An advantage is that the circuit board can be used for different types of power electronics, especially different power classes of power electronics.

[0030] Key features of the circuit board are that it is designed for an inverter, the inverter including control electronics, where the circuit board includes at least means for current sensing, wherein the means each comprise a measuring amplifier circuit switchable to at least two measuring gains, wherein a control line from the control electronics to the measuring amplifier circuit is provided for controlling the switching.

[0031] A key advantage is that the amplification can be switched during operation, i.e., while the control process is running, thus allowing the resolution of the current measurement to be changed and adapted to the specific requirements. This improves the effectiveness of control within a specifically selected operating range.

[0032] Key features of the inverter series are that the series includes at least a first and second variant of inverters, with the variants differing at least in their maximum permissible rated power. all variants feature the same printed circuit board, which includes at least the current sensing amplifier circuits, wherein each current detection is assigned two measuring amplifier circuits, whereby only the output of one of the measuring amplifier circuits is supplied as the detected current value to the control electronics of the inverter.

[0033] The advantage here is that a high variance in the product series can be achieved with a small number of parts.

[0034] Key features of the inverter series are that the series comprises at least a first and second variant of inverters, the variants differing at least in their maximum permissible rated power, and that all variants use the same printed circuit board which includes at least the current sensing amplifier circuits, with each current sensing circuit being assigned a current sensing amplifier circuit switchable to at least two sensing gains, and with a control line from the control electronics to the current sensing amplifier circuit being provided to control the switching.

[0035] An advantage of this is that switching is possible during operation, i.e., during the ongoing control process.

[0036] In an advantageous embodiment, an equal current is impressed into each phase to determine the measurement gain, and the corresponding measurement gain for each phase is then determined. The advantage here is that a defined current can be specified, and the gain can be determined from it, particularly uniformly for all current phases.

[0037] In an advantageous embodiment, the current in a switched-off bridge branch is measured to determine the offset of the current measurement assigned to each phase, particularly before determining the respective measurement gain. The advantage here is that the determination of the gain is improved and can be carried out with higher accuracy.

[0038] In an advantageous embodiment, an equal current is impressed in each phase and then determined, and the measurement gain associated with the respective phase is determined from this.

[0039] A key advantage is that the recorded current values ​​are adjustable, and in particular, adaptable. This allows for consistent current measurement across all three phases. The recorded current is therefore undistorted. For example, if a constant current vector occurs, the reconstruction of the current space vector after recording the three phase currents is undistorted, meaning it remains consistent regardless of the angular position.

[0040] In an advantageous embodiment, the current in a switched-off bridge branch is measured to determine the offset of the current measurement assigned to each phase, particularly before determining the respective measurement gain. An advantage here is that the offset can be subtracted from the measurement result, thus enabling further improved current measurement. In combination, this allows for a very accurate determination of the current space vector, the magnitude of which depends on the absolute value of the measurement gains, which remain unknown. However, the invention enables an angularly undistorted determination of the current space vector, which can even be determined without offset. In this way, the torque modeling of the converter's control is very good, and the control characteristics, such as control accuracy, are significantly improved.

[0041] In an advantageous embodiment, the converter is equipped with a final stage comprising bridge branches such that - in a first step at least one upper and the associated lower bridge branch, which are assigned to a first phase, are switched off and the upper bridge branch, which is assigned to a second phase, and the lower bridge branch, which is assigned to a third phase, are switched on, in particular whereby a unipolar current is impressed, in particular for such a long period of time that a constant current is impressed, - in at least the switched-on bridge branch, at least one current sample value is determined using the means for sampling the current, and the current measurement gain assigned to the switched-on bridge branch for current detection is determined and taken into account, - in a second and third step the first step is repeated, whereby the bridge branches assigned to the three phases are cyclically exchanged.

[0042] The advantage here is that, without requiring any additional equipment beyond that necessary for current measurement, the measurement gains in the individual phases can be adjusted easily and quickly, particularly in a very short time. By applying a voltage, a current is only impressed for such a short time as to reach a steady state. Therefore, only additional software is required to implement the invention.

[0043] In an advantageous embodiment, in each intermediate step arranged after the aforementioned first, second, and third steps, the corresponding lower bridge branch of the same phase is switched on instead of the upper bridge branch, and the upper bridge branch is switched on instead of the lower bridge branch, thereby reversing the impressed current direction. In particular, each intermediate step is arranged immediately after the first, second, and third steps; that is, the first step is followed by an intermediate step, the second by a corresponding intermediate step, and the third by a corresponding intermediate step. An advantage of this is that an impressed current is reversed, thus enabling a torque-free determination of the measurement gains.

[0044] In an advantageous embodiment, the current measurements for the control procedure are divided by the measurement gain associated with each phase, i.e., normalized current measurements are used. An advantage of this is that a simple comparison of the measurement gains of the phases with each other is made possible.

[0045] In an advantageous embodiment, the current in a switched-off bridge branch is measured to determine the offset of the current measurement assigned to each phase, particularly before determining the respective measurement gain. The advantage here is that the offset can be determined in a particularly simple manner.

[0046] In an advantageous design, - to control the half-bridges, a pulse pattern is generated within a time interval equal to one pulse width modulation period, such that at a first sampling time lying within the pulse width modulation period, at least one upper or one lower bridge branch, equipped with means for sampling the current, is switched off. - in the switched-off bridge branch, at least one current sample value is determined using the means for sampling the current, and from this a current offset value for current detection is determined and taken into account. - the switched-off bridge branch is chosen differently depending on the mean output voltage space vector and The current space vector is determined from the current sampling values, with the mean output voltage space vector being determined by the differences of the output potentials of the three output phases averaged over a pulse width modulation period.

[0047] An advantage of this is that the offset determination can be easily carried out using a converter.

[0048] In an advantageous design, - to control the half-bridges, a symmetrical pulse pattern is generated within a time interval equal to one pulse width modulation period, such that at a further sampling point located in the middle of the pulse width modulation period, one pair of the three upper or one pair of the three lower bridge branches, equipped with means for sampling the current, are switched on. - in each of the two switched-on bridge branches, at least one current sample value is determined using the means for sampling the current, - the pair of switched-on bridge branches is chosen differently depending on the mean output voltage space vector and - determined from the current sampling values ​​of the current space vector, whereby the mean output voltage space vector is determined by the differences of the output potentials of the three output phases averaged over one pulse width modulation period.

[0049] In particular, - to control the half-bridges, a pulse pattern is generated within a time interval equal to one pulse width modulation period, such that at a first sampling time lying within the pulse width modulation period, at least one upper or one lower bridge branch, equipped with means for sampling the current, is switched off. - in the switched-off bridge branch, at least one current sample value is determined using the means for current sampling, and a current offset value for current detection is determined and taken into account from this. - the switched-off bridge branch is chosen differently depending on the mean output voltage space vector and - determined from the current sampling values ​​of the current space vector, whereby the mean output voltage space vector is determined by the differences of the output potentials of the three output phases averaged over one pulse width modulation period.

[0050] An advantage of this approach is that a current offset value is determined, thus enabling offset-free current measurement and resulting in improved control behavior of the inverter. Even in the event of temperature drift in the measuring amplifier's offset value, a consistently valid value is always determined. Since the drift in typical industrial operating conditions exhibits a characteristic time constant that is larger than the pulse width modulation period corresponding to a pulse width modulation frequency in the range of 1 kHz to 100 kHz, it is not necessary to determine an offset value in every pulse width modulation period; rather, it suffices to perform this at longer intervals. Furthermore, the invention eliminates the need for a sample-hold circuit, even though current measurements for multiple phases can be acquired and only one analog-to-digital converter is required.

[0051] Since, in the variant with current detection devices arranged in all three lower branches of the half-bridges, the phase in which the corresponding pulse-width modulation signal is shorter than a minimum duration in the LOW state is always excluded, reliable determination of the current space vector is ensured using a suitable pulse-width modulation method, in particular a symmetrical pulse-width modulation method. In the respective LOW state, the lower power switch is closed and the upper one is open. The opposite is true in a HIGH state.

[0052] Furthermore, as many costly components as possible can be eliminated, especially potential-isolating current sensors, since the currents are measured not in the output branches, but in the lower or upper branches of the half-bridges. Mass production allows for cost-effective manufacturing.

[0053] In particular, with two of the three current measurement methods, the corresponding current samples are not acquired simultaneously within the pulse width modulation period and / or the corresponding measurement impulses are not executed simultaneously. An advantage of this is that the inverter can use a microcontroller with only a single analog-to-digital converter and no additional sample-hold circuit, i.e., no external sample-hold circuit.

[0054] In particular, at least one of the two means is used to record a current sample value more than once per pulse width modulation period and / or to provide more than one measurement impulse within a pulse width modulation period.

[0055] In particular, a current sample is acquired exactly twice per pulse-width modulation period using the first of the two means, and the measurement trigger for the second of the two means is located in the middle of the pulse-width modulation period. Advantageously, the two measurement triggers of the first of the two means are equidistant from the middle of the pulse-width modulation period, meaning that the time interval between the first measurement trigger of the first means and the first measurement trigger of the second means is equal to the time interval between the measurement trigger of the second means and the second measurement trigger of the first means, with the latter measurement trigger of the second means located in the middle of the pulse-width modulation period.

[0056] In the variant with current sensing devices arranged in all three lower branches of the half-bridges, the midpoint of the pulse-width modulation period corresponds to the midpoint of the respective LOW states. Current measurement signals are present at the current sensing devices during these states.

[0057] To determine the current space vector, the current measurements from the phases belonging to the two aforementioned devices are sufficient. All three current measurements from the two devices are always available because an advantageous pulse-width modulation (PWM) method is used, such as a symmetrical PWM method, and the output voltage space vector is limited to the inscribed circle of the hexagon. This also means, for example, that no corner vectors of the hexagon occur continuously over an entire PWM period.

[0058] To ensure the current space vector remains unaffected by the current ripple caused by pulse-width modulation, the measurement impulses must be triggered at appropriate times. Ideally, the measurement impulses for both instruments would be triggered simultaneously and in the middle of the pulse-width modulation period. However, this would require either two analog-to-digital converters or external sample-hold circuits, which would be costly. Instead, two measurement impulses are performed with the first instrument, offset in time. The measurement impulse for the second instrument is placed in the middle of the pulse-width modulation period. The two measurement impulses for the first instrument are each equidistant from the middle of the pulse-width modulation period.Thus, as a result of the averaging and because of the symmetry of the current ripple, which is symmetrical to the middle of the pulse width modulation period due to the symmetrical pulse width modulation method, a current measurement value is determined using the first mean, which is equal to the current measurement value that could be measured with the first mean at the time of the middle of the pulse width modulation period.

[0059] In particular, the current space vector is determined for each pulse width modulation period. An advantage of this is that even at low pulse width modulation frequencies, the current space vector can be determined as frequently as possible to improve the control performance of the inverter's control method.

[0060] In an advantageous embodiment, the first sampling point lies in the middle of the pulse-width modulation period or in the middle of a time interval during which the respective bridge branch is switched off. Advantageously, any current occurring in the preceding time interval has thus decayed and no residual currents remain.

[0061] In an advantageous embodiment, several initial sampling points occur within a single time interval, and the average of the acquired current measurements is used to determine the offset value. A key advantage is that a multiplexer allows the measured values ​​from only one analog-to-digital converter to be acquired, which sequentially acquires the current measurements from different bridge branches.

[0062] In an advantageous embodiment, current sampling values ​​are acquired in a further switched-off bridge branch in a time-symmetrical manner, both before and after the first sampling time, to determine the offset value. The advantage here is that the acquired values ​​for different switched-off bridge branches essentially correspond to a time-synchronous acquisition.

[0063] In preferred embodiments, the current measurement signals are generated at the lower DC link potential, with current values ​​assigned to the respective half-bridges being determined from current sampling values ​​derived from the current measurement signals. In particular, the reference potential of the signal electronics, which includes a control unit for the inverter, corresponds to the reference potential at which the current measurement signals are generated. An advantage of this is that optocouplers for galvanic isolation can be omitted. The reference potential of the signal electronics also corresponds to the lower DC link potential. A significant advantage is that the control signals for the lower power switches in the half-bridges can be generated by the signal electronics without a large voltage difference that would necessitate complex galvanic isolation.Only the control signals of the upper circuit breakers need to be controlled via optocouplers or other potential-isolating devices. Particularly with a converter that has no connections for an encoder, such as a speed or position encoder, or for other potential-isolating devices, this represents a significant step towards cost savings and a reduction in the number of parts.

[0064] Advantageously, power semiconductor switches, such as npn type IGBTs, can be used as lower circuit breakers in half-bridges, which can be controlled with control voltages that have the lower DC link potential as their reference potential.

[0065] When using complementary power semiconductor switches, such as IGBT type pnp, the upper DC link potential must be selected as the reference potential for current measurement and for the signal electronics, and the inverter must be designed accordingly.

[0066] In a preferred implementation, the pair is selected differently depending on the angle of the output voltage space vector and not on the pulse-width modulation pattern. An advantage of this is that the programmatic implementation is particularly simple.

[0067] In another implementation, a time interval of two or more pulse-width modulation periods is used instead of a single pulse-width modulation period. The advantage here is that the current space vector can be determined even at high switching frequencies. While this does introduce a slight distortion, it is minimal at high switching frequencies.

[0068] The acquisition of the current sample value, acquired by the first current measuring device, is performed midway through the pulse-width modulation (PWM) period. Specifically, a second current measuring device captures a first and second corresponding current sample value symmetrically before and after the current sample value acquired by the first current measuring device. The advantage of this approach is that current measurements can be determined with only one analog-to-digital converter, without the need for external sample-hold circuits, and these measurements are not distorted by current ripple caused by PWM. Therefore, determining the current measurements is particularly simple and requires minimal computational effort.

[0069] By using additional external sample-hold circuits, the corresponding current samples could be acquired simultaneously using two of the three current acquisition methods, and / or the corresponding measurement impulses could be executed simultaneously. In particular, the measurement impulses would be executed in the middle of the pulse-width modulation period. The advantage of this approach is that determining the current measurements is particularly simple, yet there are no distortions caused by current ripple.

[0070] In another preferred embodiment, the first of the two current sensing means acquires a corresponding current sample value at least one pulse-width modulation period after acquiring a current sample value associated with the second current sensing means. In particular, the measurement impulses associated with the two current sensing means occur at different pulse-width modulation periods. An advantage of this is that even at high switching frequencies, a current space vector can be determined using only a single analog-to-digital converter and without additional sample-hold circuits, especially without distortions caused by current ripple.

[0071] In a preferred embodiment, the first current sample of the pair is acquired more than once per pulse-width modulation period, and an interpolated value and / or average is calculated from the acquired values ​​according to the times of the respective acquisitions. Specifically, a first current sample of the pair, assigned to a first half-bridge, is acquired before and after the second current sample of the pair, assigned to a second half-bridge. An advantage of this approach is that, through interpolation or averaging, a fictitious current measurement can be determined with good accuracy, which is available simultaneously with the other acquired current sample.

[0072] In a preferred embodiment, the measured current space vector corresponds to the mean value of the current space vector over one pulse-width modulation period. An advantage of this is that the current ripple resulting from the pulse-width modulation does not distort the result and therefore also the control procedures.

[0073] Important features of the invention of the inverter, comprising signal electronics and a power output stage, which includes power switches arranged in three half-bridges, each comprising a lower and an upper branch, wherein the inverter can be operated with pulse width modulation, are that - means for recording the respective currents are arranged either in all three lower branches or in all three upper branches of the half-bridges, - current measurement signals acquired by the three means of current acquisition are fed and / or can be fed to only a single analog-to-digital converter via a multiplexer.

[0074] The advantage here is that costly potential-separating means can be saved and the size of the inverter can be reduced.

[0075] In a preferred embodiment, the current sensing means comprise resistors, in particular shunt resistors. An advantage of this is that the current sensing is extremely cost-effective.

[0076] In a preferred embodiment, the current sensing means are arranged in the half-bridges such that they are connected to either the upper or the lower DC link potential. An advantage of this arrangement is that galvanic isolation means can be eliminated. In a further preferred embodiment, the signal electronics have a reference potential that also serves as the reference potential for the current sensing means. If this reference potential is Uz-, then galvanic isolation means for the control signals of the lower circuit breakers of the half-bridges can be eliminated. If this reference potential is Uz+, then galvanic isolation means for the control signals of the upper circuit breakers of the half-bridges can be eliminated.

[0077] In a preferred embodiment, the signal electronics have a reference potential that also serves as the reference potential for the current sensing devices. This is advantageous because costly galvanic isolators can be omitted. If the current sensing devices were located in the motor's supply lines, costly galvanic isolators would be necessary.

[0078] In a preferred embodiment, any current sensing means can be assigned to a single analog-to-digital converter by means of a multiplexer and / or switch. An advantage of this is that the converter can be implemented cost-effectively, particularly by eliminating the need for additional analog-to-digital converters.

[0079] In principle, it would also be possible to assign more than one analog-to-digital converter (ADC) to the current sensing means. The advantage here would be that, with the use of appropriately arranged multiplexers, the selected pair of current sensing means could be assigned to, for example, two ADCs, thus eliminating the need for additional sample-hold circuits. Specifically, each current sensing means could be assigned its own ADC. The advantage here is that no additional sample-hold circuits are necessary and synchronous acquisition of the current samples can be ensured. Preferably, a sample-hold circuit could also be placed between each current sensing means and its assigned ADC. The advantage here would be that the current space vector could be determined without distortion from current ripple, and a single ADC would suffice.Preferably, the signal electronics could include a microcontroller with a single analog-to-digital converter and a sample-hold circuit not integrated into the microcontroller, i.e., an additional external component. The advantage here would be the use of a cost-effective microcontroller that only includes a single analog-to-digital converter.

[0080] In a preferred embodiment, the signal electronics comprise means for generating pulse-width modulated control signals for the circuit breakers, and the signal electronics have a reference potential that also serves as the reference potential for the current sensing means. An advantage of this is that means for potential isolation can be eliminated.

[0081] In an advantageous design, a single analog-to-digital converter can be assigned a specific current sensing device by means of a multiplexer and / or switch. This eliminates the need for additional, costly analog-to-digital converters.

[0082] In another advantageous embodiment, the essential features are that - means for recording the respective currents are arranged either in all three lower branches or in all three upper branches of the half-bridges, - the pulse width modulation frequency is greater than a minimum frequency, - within a time interval of two pulse width modulation periods, at least one corresponding current sample value is determined using any two of the three means of current detection, and the same mean output voltage space vector is output in both pulse width modulation periods, - with a first mean of the pair, a first current sample value is captured in the middle of the first of the two pulse width modulation periods and a corresponding current sample value is captured with a second instrument of the pair in the middle of the second, i.e. immediately following, pulse width modulation period, - a current space vector is formed from the current sampling values ​​determined with this pair of means, or the current values ​​in the output branches are formed, in particular for use in a control and / or regulation procedure, and - this pair of means is selected differently depending on the mean output voltage space vector, - and where the mean output voltage space vector is determined by the differences of the output potentials of the three output phases averaged over one pulse width modulation period.

[0083] A key advantage is that even at high pulse-width modulation frequencies, a single analog-to-digital converter is sufficient, eliminating the need for external sample-hold circuits. If the frequency falls below a minimum threshold, such as 10 kHz, the system can switch to the previously described method. Therefore, these advantages are present at all pulse-width modulation frequencies.

[0084] Further advantageous embodiments result from the dependent claims.

[0085] The invention will now be explained in more detail with the help of illustrations: In the Fig. Figure 1 shows a schematic diagram of a power output stage for a three-phase inverter. The power switches with associated freewheeling diodes are represented by six switch symbols S. Ro' S So' , S To , S Ru , S Su and S Tudesignated. In the lower branches of the half-bridges, shunt resistors R are used as a means of current detection. R , R S and R T These are thus arranged with the lower reference potential U. Z_ connected, which is also simultaneously the reference potential of the signal electronics of the inverter comprising a control and regulation device 1. The motor-side output potentials of the inverter are U R , U S and U T ; the motor currents are denoted by I R , I S and I T designated.

[0086] The resistors are connected to amplifier circuits V R , V S and V T connected, each outputting a current measurement signal I RM , I SM and I TM generate. Each phase in the Fig. 1 and Fig. 2. Each amplifier circuit is assigned and illustrated in the drawing. According to the invention, however, each of these amplifier circuits is implemented in hardware from two optionally usable amplifier circuits arranged in parallel and having different gains. Alternatively, each amplifier circuit can also be implemented with switchable gain.

[0087] According to the invention, the gains of the amplifier circuits are selectable. This makes it possible to use the entire control electronics of the various embodiments for different power classes of inverters, which, according to their power class, have different inverter stages with switches S. The switches S are designed for different maximum permissible currents and / or power levels.

[0088] Therefore, if different switches are used for different power classes, resulting in different current values, the measurement gains can be selected accordingly, allowing, for example, the same shunt resistor or at least the same shunt resistor value to be used. In this way, only the power section, comprising the inverter's switches, needs to be adapted to the power class.

[0089] The inverter can therefore be implemented with a single circuit board that includes the current sensing amplifier circuits. This circuit board can thus be used in multiple inverters with different power ratings and therefore different power levels with corresponding power switches. A wide range of models within a single series can thus be manufactured with a small number of parts.

[0090] Using the offset determination described below and the gain determination also described below, switching during operation is quick and easy. For example, if the expected current measurement is small, but a higher resolution of the current measurement is required for better control accuracy, switching is possible even while the motor is operating and being controlled. This is achieved without any abrupt changes in the measured current curve, such as those caused by different offsets, or angle-dependent variations in the measured current value, such as those caused by different gains in the individual current sensors.

[0091] From the three current measurement signals according to Fig. 1. Current samples are derived by analog-to-digital conversion, from which current measurements are generated. The current space vector is determined from these current measurements, whereby the half-bridges corresponding to the current measurements are selected depending on the angle of the voltage space vector. Since the motor currents are determined according to Kirchhoff's law I R + I S + I T If the current space vector is zero, it has two degrees of freedom. It therefore has two independent parameters. In principle, it is sufficient to use only the current measurements assigned to two branches to determine the current space vector. However, it must be noted that pulse-width modulation signals from the inverter do not have to be constant within a pulse-width modulation period; the switching states can change in such a way that switching states may occur which, on their own, would not allow for the determination of the current space vector!

[0092] For the control procedure of the inverter, it is sufficient to achieve adequate control quality by determining a current space vector as a calculated variable once per pulse width modulation period, which represents the mean value of the physical current space vector formed over one pulse width modulation period.

[0093] The pulse width modulation signals (PWM) generated by the control device 1 R (t), PWM S (t) and PWM T (t) determine the state of the circuit breakers S Ro , S So , S To , S Ru , S Su and S Tu The pulse width modulation signals are provided as follows: If the respective pulse width modulation signal, for example PWM, R (t), 1, the associated upper circuit breaker, i.e., for example S Ro , closed and the associated lower circuit breaker, for example S Ru, opened. In this case, the corresponding output voltage potential is U. Z+ If the pulse width modulation signal is 0, i.e., in the LOW state, the corresponding power switches are in the opposite state and the corresponding output voltage potential is U. Z- The so-called dead time included in the practical implementation, during which the upper and lower switches are open, is not relevant to the basic function of the invention.

[0094] When a pulse-width modulation signal belonging to a half-bridge is in the LOW state, the corresponding motor current flows in the lower branch of the associated half-bridge and thus through the respective shunt resistor. For accurate current measurements, the corresponding motor current must flow through the shunt resistor for a minimum duration. This minimum duration depends on the filtering effect of the measuring amplifier circuits, comprising amplifier circuits V. R , V S and V T A filtering effect is achieved through appropriate circuitry to suppress noise or interference components in the current measurements. For example, with a pulse width modulation frequency of 16 kHz, a filter time constant in the range of 0.5 µs to 2 µs is advantageous.

[0095] Advantageously, the exemplary embodiment uses a so-called symmetrical pulse-width modulation (PWM) method. In such symmetrical PWM methods, the time-averaged value calculated from the time of the first switching state change of the PWM signal in a first phase and the time of the corresponding subsequent switching state change in the same phase has the same value as the corresponding average values ​​of the other two phases. Therefore, the switching state changes from HIGH to LOW and back are symmetrical about the center of the PWM period in all three phases.

[0096] Advantageously, in one embodiment, the amplitude of the averaged output voltage space vector is set to UZ / 3 The output is limited so that a sinusoidal rotating voltage system can always be generated. This limitation defines the maximum output level. Overdriving, i.e., leaving this operating range defined by the limitation, is always avoided.

[0097] Advantageously, this embodiment only applies to inverters that are operated with pulse-width modulation (PWM) in such a way that no pulse-width modulation period results in such a high output level that only a single active switching state occurs for a full pulse-width modulation period. In such an active switching state, the pulse-width modulation signals (PWM) R (t), PWM S (t) and PWM T(t) the values ​​(110), (101), (100), (001), (010) or (011), which are also referred to as discrete active output voltage space vectors. Active switching states therefore do not include switching states (111) and (000). The latter two switching states are also referred to as discrete zero voltage space vectors or zero vectors.

[0098] So-called boundary vectors are the output voltage space vectors averaged over a pulse-width modulation period, which would theoretically result if a single active switching state were applied over an entire pulse-width modulation period. Advantageously, in one embodiment, the output voltage space vector averaged over a pulse-width modulation period never assumes the value of such a boundary vector due to the aforementioned limitation of the operating range.

[0099] The image shows the aforementioned IEEE document by Francesco Parasiliti, according to the one there. Fig. As discussed in section 8 on page 1287, the limitation of the output voltage space vector averaged over a pulse-width modulation period, as described above, means that it lies within the maximum incircle of the hexagon. The averaged output voltage space vector never assumes the value of a boundary vector, and therefore, in this embodiment, never lies at the corners of the hexagon.

[0100] Since the output voltage space vector is limited, various discrete switching states alternate. By carefully selecting the timing of the measurement impulses within a pulse-width modulation period, a current space vector can be determined for each pulse-width modulation period. The alternating states include, among others, states for which determining a current space vector would not be possible if they were continuously present over a pulse-width modulation period, such as state (101) according to the last row of Table 2B from US 5,815,391 A. Specifically, for example, a measurement impulse is executed at a first time point in a first phase, a second measurement impulse at a second time point in a second phase, and a third measurement impulse at a third time point in the first phase.By averaging the first and third measured values, the determinable current space vector is not distorted by current ripple in the pulse-width modulation method permissible for the embodiment, particularly symmetrical pulse-width modulation methods where the output voltage space vector is limited to the inner circle of the hexagon. Thus, only a single analog-to-digital converter is necessary, especially without the need for external sample-hold circuits.

[0101] In the Fig. 2 is a feasible variant for part of the control and regulating device 1 from Fig. Figure 1 shows a sample-hold circuit 21, which receives the three current measurement signals, and a microcontroller 22 with an integrated analog-to-digital converter. The microcontroller 22 controls the sampling times, also called measurement pulses, via the signals S / H1, S / H2, and S / H3. At these times, the sample-hold circuit 21 holds current measurement signals until the analog-to-digital conversion. The microcontroller 22 then only needs to include a single analog-to-digital converter along with an analog multiplexer or switch 23. The advantage of this is that simultaneous acquisition of current samples is achieved with only a single analog-to-digital converter, even though a sequential analog-to-digital conversion takes place. The microcontroller 22 then determines current measurements or a current space vector from the current samples.The microcontroller 22 controls the sample hold circuit via the signals S / H1, S / H2 and S / H3, and thus the measurement impulses according to the procedures described below.

[0102] In another embodiment, according to Fig. 3 in contrast to the variant after Fig. 2. The sample-hold circuit 21 can be omitted. However, it is necessary to acquire the current sample values ​​sequentially according to the procedures described below.

[0103] In the Fig. Figure 4 shows exemplary time courses of the motor-side output potentials U, averaged over one pulse width modulation period. R , U S and U T of the inverter over one output voltage period, where the motor-side output potentials U R , U S and U TThe values ​​are shown in a normalized form, and the output voltage angle α extends over the range from 0 to 2π. According to the normalization, the potential value U corresponds to... Z- the value -1 and U Z+ the value +1.

[0104] In the Fig. Figure 4 shows that the mean values ​​of the output potentials contain a third harmonic.

[0105] Further are in Fig. Four angular ranges 1 to 6 of the output voltage phasor are shown, in which different means of current sensing are used.

[0106] For an output voltage angle of α = π / 6, the mean value of the output potential U is S Zero. The mean value of the initial potential U. R lies close to the positive maximum value of 1. The mean value of the initial potential U Tlies close to the minimum value of -1. The mean values ​​of the output potentials therefore almost reach the maximum output level of the inverter at this output voltage angle and with the exemplary chosen amplitude.

[0107] In the Fig. 5 are for the initial potentials from Fig. 4. Again at the output voltage angle α = π / 6, the corresponding time profiles of the pulse width modulation signals (PWM) are shown. R (t), PWM S (t) and PWM T (t) sketched. From these, the control signals for the circuit breakers S are derived. Ro , S So , S To , S Ru , S Su and S Tu derived. If the respective pulse width modulation signal, for example PKM R (t), 1, the associated upper circuit breaker, i.e., for example S Ro , closed and the associated lower circuit breaker, for example S Ru, opened. In this case, the corresponding output voltage potential is U. R = U Z+ If the pulse width modulation signal is 0, the corresponding power switches go to the other state and the corresponding output voltage potential is U. Z- The so-called dead time incorporated into the practical implementation, during which the upper and lower switches are open, is fundamentally irrelevant to the function of the exemplary embodiment. In the Fig. Figure 5 shows two pulse width modulation periods in the abscissa direction.

[0108] The output voltage potential U averaged over one pulse width modulation period S = 0 is according to Fig. 5 through a PWM S (t) - Signal reached that is 1 for 50% of the pulse width modulation period and 0 for 50% of the pulse width modulation period. The pulse width modulation ratios of the other pulse width modulation signals (PWM) R (t) and PWMT (t) are distributed according to their output voltage potential averaged over a pulse width modulation period.

[0109] In the Fig. 6 are the exemplary motor currents I corresponding to the two mentioned pulse width modulation periods. R (t), I S (t) and I T (t) is shown in the output branches. The current ripple resulting from the pulse-width modulated output potentials is also indicated.

[0110] In the Fig. Figure 7 is the idealized time course of the current measurement signals I corresponding to the two pulse width modulation periods mentioned. RM (t), I SM (t) and I TM (t) shown without the filtering effect of the amplifier circuits V R , V S and V TA non-zero current reading only occurs during those time intervals when the pulse-width modulation (PWM) signal of the corresponding half-bridge is zero; i.e., when the lower switch is closed. When the upper switch is closed, the motor current does not flow through the shunt resistor of the half-bridge, so during this time the current measurement signal is zero, regardless of the actual current. Therefore, the current measurement signals do not continuously represent the motor currents, but only during those time intervals when the PWM signals are zero. R (t), PWM S (t) and PWM T (t) are zero. In the shown curve, the current measurement signal I RM (t) is only different from zero for a very short period of time. At full output voltage, this period would disappear entirely. Fig. 7 It is therefore evident that determining a current space vector fails under such operating conditions if the current space vector is formed using current measurement signals that are only different from zero for such a short time as in Fig. 7 examples I RM (t) that they are significantly distorted by the filter effect at the time of acquisition, i.e., at the measurement trigger. However, a minimum filter effect is necessary to suppress measurement noise and interference signals.

[0111] Therefore, in one embodiment, for each pulse-width modulation period, the "optimal" pair exhibiting the wider pulse-width modulation signals is selected from the three half-bridges; that is, only current measurement signals from these two specific half-bridges are used. The selection is made such that the current measurement signal whose lower switch remains closed for a shorter time than the other two lower switches within a given pulse-width modulation period is not used.

[0112] In the Fig. Figure 7 shows an operating state for two pulse width modulation periods, in which the selected pair I SM (t) and I TM (t). The times at which the current sampling values ​​are recorded, also referred to as measurement impulses, are indicated by jagged arrows.

[0113] Two basic types of procedures can be used.

[0114] For a first method, the measurement points are indicated by solid, not dashed, jagged arrows. The current sampling values ​​for the selected pair I are then recorded. SM (t) and I TM (t) simultaneously acquired. Several implementations are possible. In one implementation, each current sample value is generated by a separate analog-to-digital converter. In a second implementation, the three current measurement signals are processed according to Fig. Two signals are each fed to a sample-hold amplifier of the sample-hold circuit 21, with the sample-hold amplifiers switching to the hold state at the time of sampling. Thus, the analog-to-digital converter can sequentially convert the held current measurement signals using the multiplexer or switch. In a third embodiment, two analog-to-digital converters and corresponding switches are used. Further embodiments also allow for hybrid configurations.

[0115] The first method advantageously uses a symmetrical pulse-width modulation (PWM) technique, where the switching frequency is not higher than 20 kHz and the maximum modulation level described above is not exceeded. Thus, within each PWM period, there is always a pulse pattern in which a PWM signal with a LOW state shorter than the minimum duration occurs in at most one output branch. This means that both PWM signals are never simultaneously in the LOW state for less than the minimum duration in two phases. In the event that the LOW state of one of the PWM signals, for example, PWM, is R Even if (t) falls below the minimum duration, it is still ensured that the current space vector is determined correctly. This is because the two other pulse-width modulation signals, in this example PWM, S (t) and PWM T(t) do not fall below the minimum duration. Thus, in one embodiment, a correct determination of the current space vector is always guaranteed, provided that, for the respective type of pulse width modulation method, it is ensured that at most only one pulse width modulation signal remains in the LOW state for a shorter duration than the minimum duration.

[0116] In a second method, the measurement impulse for I lies in the middle of the pulse width modulation period. TM The first method is replaced by two measurement impulses, each with a time offset Δt. The first measurement impulse for I is performed sequentially. TM , then the one for I SM and then the second measurement impulse for I TM It is advantageous to use the measurement impulse for I SMto place in the middle of the pulse-width modulation period. The time offset Δt should be as small as possible, but larger than the analog-to-digital converter time. From the first and second current samples for I TM The average value is calculated and used to determine the current space vector. This average value corresponds to the fictitious current sampling value for I. TM , which simultaneously serves to record the current sampling value I SM would be detectable. Thus, the current space vector determined in this way is free from the influence of the current ripple. This is because, when using symmetrical pulse-width modulation methods, the current ripple also exhibits a point-symmetrical course relative to the respective current function value at t=0.5, around the middle of the pulse-width modulation period, as is also the case in Fig. 6 is recognizable as an example. Fig. 6. The midpoint of the first pulse width modulation period lies at t=0.5 and the midpoint of the second pulse width modulation period at t=1.5. Therefore, since the two current sampling values ​​I TM (t = 0.5 - Δt) and I TM (t = 0.5 + Δt) point-symmetric to the current function value I TM (t = 0.5) is the average of the two current sampling values ​​I. TM (t = 0.5- Δt) and I TM (t = 0.5 + Δt) the current function value I TM (t = 0.5).

[0117] This method makes it possible to determine the current space vector without distortion from current ripple using only a single analog-to-digital converter. Since all current samples can be performed sequentially, additional sample-hold circuits are unnecessary. These additional sample-hold circuits are not the same as those that are standard features in conventional analog-to-digital converters.

[0118] The second method also advantageously uses a symmetrical pulse-width modulation (PWM) technique, where the switching frequency is preferably not higher than 20 kHz and the maximum modulation level described above is not exceeded. Thus, within each PWM period, a pulse pattern is always present in which a PWM signal with a LOW state shorter than the minimum duration occurs in at most one output branch. In the PWM technique used in the second method, at least one PWM signal is always present whose LOW state lasts longer than the sum of the minimum duration and twice the time offset, i.e., 2*Δt.Thus, in one embodiment, correct determination of the current space vector is always guaranteed, provided that the pulse-width modulation (PWM) signals have a sufficient pulse width for the respective type of PWM method. While flat-top methods are generally symmetrical PWM methods, they are not advantageous for use with low output voltages.

[0119] In the exemplary embodiment according to Fig. 7. Furthermore, one or more current samples are recorded during those time periods in which a vanishing current sample value is expected. For example, in Fig. 7 showed that for phase R the measured value I RMoff is recorded at two time points, using the mean of the two recorded measurements, and for phase S the measured value I SMoff. In phase T, the time range for an interference-free acquisition of a current measurement is too short, and therefore no current is recorded in the time periods in which a vanishing current sample value is expected.

[0120] If the voltage values ​​to be generated in the individual phases change in subsequent pulse-width modulation periods, and thus a longer period of expected zero current occurs in phase T, a current sample is also acquired in phase T. Particularly when generating a rotating magnetic field, the current measurement is therefore distributed evenly across the three phases.

[0121] In this way, the offset of the individual measuring amplifiers in the sample-hold circuit can be determined. This offset is taken into account during current measurement by subtracting the current offset value from the measured value. This results in an improved measurement and thus a correspondingly improved motor control. Advantageously, this offset determination is possible online, i.e., during the ongoing operation of the control system. Only additional current measurements are performed, which do not cause any significant additional load on the control unit or slow down the control process. A further advantage is that temperature-induced drift in the offset can be determined and counteracted accordingly.

[0122] The current sampling point is at least 6 µs, preferably at least 8 µs, apart from the preceding time period in which the corresponding bridge branch was switched on. This ensures that oscillations have ceased and the current has reliably decayed.

[0123] In a further embodiment according to the invention, instead of the two measurements in phase R, only a single measured value is recorded, which is not recorded simultaneously with the measured value of phase S.

[0124] In continuing education, a moving average of some recently recorded measurements is used as the currently recorded offset value.

[0125] Preferably, the measured values ​​are recorded symmetrically around the first sampling time.

[0126] In this embodiment, only pulse width modulation methods are used that have a switching state in the middle of the pulse width modulation period such that at least two lower switches are closed.

[0127] In another embodiment, the current samples are not determined in the same pulse-width modulation period. Instead, the first current sample is acquired in the first pulse-width modulation period, and the second in one of the subsequent pulse-width modulation periods. Again, only a single analog-to-digital converter is required, without an additional sample-hold circuit. This is particularly advantageous at high switching frequencies and thus short pulse-width modulation periods. No distortion due to current ripple occurs. Only the change in the angle of the current space vector during this time can cause distortion. The selection of the pair is carried out according to... Fig. 8.

[0128] In Fig. Figure 4 shows angle ranges 1 to 6, corresponding to the different angle ranges of the output voltage phasor. Fig. In section 8, the pair to be used for each of these angle ranges is indicated by jagged arrows. For example, in angle range 1, only S and T are used, i.e., I. SM and I TM , to use. In angle range 2, only R and T are used, i.e., I. RM and I TM , to be used. In those angular ranges where no current measurement is performed in a given phase, the measurement of the offset value is provided. This is in Fig. 8 with entry I RMoff , I SMoff and I TMoff designated.

[0129] Each bridge branch assigned to a respective phase R, S,T is according to Fig. 1 and Fig. 2 is assigned a respective current sensing means. This comprises, as indicated in the figures, a respective measuring amplifier. According to the invention, the measuring gain of the three means assigned to the three phases is determined at the beginning or at another time interval and / or at least adjusted to each other.

[0130] In a first step, one of the three bridge branches is switched off, and the other two are energized such that the current flowing in the bridge branch assigned to the second phase is equal in magnitude to that flowing in the bridge branch assigned to the third phase. A current sample value is then recorded in both phases.

[0131] Preferably, in a second step, the applied voltage is reversed, i.e., the current direction is inverted, whereby the same duration of current application as in the first step is provided, such that the corresponding current-time area is essentially the same for both steps. In this way, any torque-generating effect is prevented, at least on average. After the current has settled in the second time step, corresponding current sampling values ​​are again recorded for both phases.

[0132] The first and second steps are repeated, with the resources assigned to the phases being exchanged cyclically.

[0133] In this way, the three gains of the current sensing devices assigned to the three phases, comprising their respective measuring amplifiers, are determined. Preferably, this determination is preceded by determining the offset value by acquiring a current sample value with each bridge branch switched off.

[0134] After determining the three measurement gains, current measurement in all three phases can be standardized; that is, the current sample value divided by the measurement gain can be used for the inverter's control procedure. A current space vector that is constant in magnitude but changes in angle is thus also measured as a constant current, and the control system achieves improved control characteristics, in particular higher control accuracy. Specifically, the torque determined from these current measurements is more accurate and realistic.

[0135] Furthermore, if the measurement gains are determined repeatedly at time intervals, a drift in the measurement gains can be compensated for and taken into account.

[0136] As described above, in the second method, the measurement impulse for a current sample value of the first method, which lies symmetrically in the middle of the pulse width modulation period, is replaced by two measurement impulses, each with a time offset Δt. Fig. Figure 9 shows how the measurement pulses are to be performed depending on the angular ranges of the output voltage space vector. It specifically shows in which half-bridge two measurement pulses are to be performed and in which half-bridge the single, centrally located measurement pulse is to be performed. Thus, in angular range 1, a first measurement pulse for I is performed. TM , then a single, centrally located measuring impulse for I SM and finally a second measurement impulse for I TMThe measurement impulses have a time interval of Δt.

[0137] In angle range 3, a first measurement impulse for I therefore takes place. RM , then a single, centrally located measuring impulse for I TM and finally a second measurement impulse for I RM .

[0138] For each angular range, there is a phase with a solid flash, indicating a current sample taken in the center of that phase, and another phase with two dashed flashes, indicating a double current sample taken in that phase by the time offset Δt before and after the center point. In one embodiment, only pulse-width modulation methods are used whose pulse pattern is always such that the lower switches in the respective angular range of the phases marked with flashes are closed for a period of time around the respective current samples. In the phase with two dashed flashes, the corresponding lower switch is closed from before the first measurement pulse until after the second measurement pulse.

[0139] A characteristic of the usable pulse width modulation methods is that at least at the time of the middle of the pulse width modulation period, two lower switches are always closed, with the one of these two lower switches that is assigned to the phase with the double current sampling remaining closed for at least a time interval 2Δt.

[0140] Again, in those angular ranges where no current measurement is performed in a given phase, the measurement of the offset value is provided. This is in Fig. 9 with entry I RMoff , I SMoff and I TMoff marked. Thus, the otherwise unused areas are utilized for determining the offset value of the current measurement, i.e., all the means for this purpose, such as measuring amplifiers, multiplexers and analog / digital converters.

[0141] In other embodiments, according to Fig. 7 the two current sampling values ​​of the current measurement signal ITM not symmetrical around the measurement impulse for capturing the current sample value of the current measurement signal I SM around, i.e. not with a time interval Δt before and after the detection of I SM , but with different time intervals. Then, instead of the average value, an interpolated value is calculated that takes the corresponding time intervals into account, whereby motor-characteristic parameters as well as the type and duration of the switching states must be considered during interpolation.

[0142] In other embodiments, more current samples are acquired instead of the two mentioned. Furthermore, each current sample can, in principle, be replaced by multiple current samples. Thus, a further reduction in measurement noise is foreseeable.

[0143] In further embodiments, the time response of the amplifier circuits V R , V S and V TThis is taken into account by delaying all measurement impulses by the filter time constant of the amplifier circuits. The filter time constant is less than half the minimum duration.

[0144] The method also works at pulse width modulation frequencies higher than 20 kHz. Reference symbol list 1 Control and regulating device 21 Sample Hold Circuit 22 microcontrollers 23 multiplexers

Claims

[1] Printed circuit board for an inverter, comprising control electronics, wherein the printed circuit board includes at least means for current sensing, characterized by , that The means each comprise a first and a second measuring amplifier circuit arranged in parallel to this first, wherein the first measuring amplifier circuit has a different gain than the second measuring amplifier circuit. wherein only the output of the first measuring amplifier circuit or only the output of the second measuring amplifier circuit is supplied as a measured current value to the control electronics of the inverter, where the first measuring amplifier circuit and the second measuring amplifier circuit are implemented in hardware. [2] Method for determining a current space vector, especially for use in a tax and / or regulatory procedure, for a pulse-width modulated inverter comprising signal electronics, a power output stage with power switches arranged in three half-bridges, each having a lower and an upper branch, and means for sampling the respective currents arranged in the three upper or lower branches of the half-bridge, wherein the means for sampling each comprise a first and a second measuring amplifier circuit arranged in parallel to this first, wherein the first measuring amplifier circuit has a different gain than the second measuring amplifier circuit, wherein only the output of the first measuring amplifier circuit or only the output of the second measuring amplifier circuit is supplied as a measured current value to the control electronics of the inverter, where the first measuring amplifier circuit and the second measuring amplifier circuit are implemented in hardware. [3] Method according to claim 2, characterized by, that the amplification is switched during the ongoing control and regulation process, particularly to achieve better resolution in current measurement. [4] Method according to any one of claims 2 to 3, characterized by , that the same means of sensing are used for different circuit breakers, one of which is suitable for a larger maximum permissible switching current than another. [5] Method according to any one of claims 2 to 4, characterized by , that After changing the gain of the current measurement assigned to a respective phase, the offset and / or the gain of the current measurement assigned to the respective phase is first determined and then taken into account. in particular, where a current is impressed and determined to determine the amplification in the respective bridge branch. [6] Inverters, in particular pulse-width modulated inverters, characterized by , that the converter comprises signal electronics, a power output stage with power switches arranged in three half-bridges, each having a lower and an upper branch, and means for sampling the respective currents arranged in the three upper or lower branches of the half-bridge, wherein the means for sampling each comprise a first and a second measuring amplifier circuit arranged in parallel to this first, wherein the first measuring amplifier circuit has a different gain than the second measuring amplifier circuit, wherein only the output of the first measuring amplifier circuit or only the output of the second measuring amplifier circuit is supplied as a measured current value to the control electronics of the inverter, where the first measuring amplifier circuit and the second measuring amplifier circuit are implemented in hardware. [7] Inverter according to claim 6, characterized by , that -a selector switch or other hardware-implemented means is used as the selection device or - a parameter value or other means implemented in software is used as a means of selection. [8] Inverter according to one of claims 6 to 7, characterized by , that to determine the measurement gains in each phase, an equal current is impressed and then measured, and from this the measurement gain corresponding to the respective phase is determined. [9] Inverter according to any one of claims 6 to 8, characterized by , that for the offset determination of the current measurement assigned to a respective phase, the current in a switched-off bridge branch is measured, in particular before the determination of the respective measurement gain. [10] Series of converters, the series includes at least a first and second variant of inverters, the variants differ at least in their maximum permissible rated power, characterized by , that All variants use the same circuit board, which includes at least the current sensing amplifier circuits, wherein each current detection is assigned two measuring amplifier circuits, each with a different gain, wherein only the output of one of the measuring amplifier circuits of each current detection is electrically connected to the control electronics of the inverter for transmitting the detected current value.

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