Method for operating an electric motor

Abstract

Method for operating an electric motor in which a flux sensor of a control electronics determines a stator flux vector depending on a predetermined torque requirement, a voltage sensor of the control electronics determines a scaled stator voltage vector depending on the determined stator flux vector, an inverter of the control electronics switches an electrical DC voltage provided by an intermediate circuit depending on the determined scaled stator voltage vector and generates a multi-phase AC voltage by means of the switching, and the control electronics operate an electric motor by applying the generated multi-phase AC voltage; as well as control electronics for an electric motor.

Description

[0001] The invention relates to a method for operating an electric motor, in which a flux sensor of a control electronics unit determines a stator flux vector depending on a predetermined torque requirement, a voltage sensor of the control electronics unit determines a scaled stator voltage vector depending on the determined stator flux vector, an inverter of the control electronics unit switches a DC voltage supplied by an intermediate circuit depending on the determined scaled stator voltage vector, and generates a multiphase AC voltage by means of the switching, and the control electronics unit operates an electric motor by applying the generated multiphase AC voltage. The invention further relates to control electronics for an electric motor.

[0002] Methods of the type mentioned above, in various forms, represent the state of the art and serve to operate an electric motor using a direct current voltage supplied by an intermediate circuit. The direct current voltage of the intermediate circuit is usually supplied by a battery and is assumed to be at least substantially constant over time.

[0003] A so-called inverter of a control electronics unit, which is also referred to as a voltage converter, an inverter or a power electronics unit, generates a multi-phase alternating voltage by switching the supplied DC voltage, with which the control electronics supply the electric motor.

[0004] The electric motor typically comprises a stator with multiple stator windings, each with two terminals. Often, the stator windings are connected in a star configuration, meaning that the first terminals of each stator winding are unconnected and electrically linked to the control electronics, while the second terminals are electrically connected to each other, forming a neutral point for the stator windings.

[0005] Furthermore, the electric motor usually comprises a rotor rotatably mounted in the stator, containing a plurality of permanent magnets. The magnetization of the permanent magnets need not be constant. For example, CN 1 12 234 894 A discloses a method for operating an electric motor in which the magnetization of a permanent magnet of an electric motor rotor is selectively varied.

[0006] A multiphase alternating voltage comprises a plurality of alternating voltages corresponding to the plurality of stator windings, all of which have identical angular frequencies but different phase angles. Each alternating voltage of the multiphase alternating voltage is also referred to as a phase.

[0007] The inverter supplies each stator winding of the electric motor with exactly one phase of the multi-phase alternating voltage. If the electric motor has three stator windings, for example – as is usually the case, especially if it is designed as a drive motor for an electric vehicle – the phases of the three-phase alternating voltage are usually designated by the letters U, V, and W.

[0008] Each switching operation of the inverter occurs at a specific switching point and involves connecting a free terminal of a stator winding to a pole of the DC link or disconnecting the free terminal of the stator winding from a pole of the DC link. A recurring temporal sequence of switching points, i.e., a switching rhythm of the inverter, is referred to as an inverter clocking pattern. Each clocking pattern implies a form of the multiphase AC voltage, i.e., a time waveform of the multiphase AC voltage.

[0009] To generate the multi-phase alternating voltage, the control electronics determine a stator space voltage vector depending on a specified torque requirement and an operating point of the electric motor.

[0010] A stator space vector, for example the stator space voltage vector, is a vector quantity given with respect to a two-dimensional coordinate system fixed relative to the stator of the electric motor, i.e., a stator-fixed coordinate system. The two coordinate axes of the stator-fixed two-dimensional coordinate system are usually denoted by α and iβ.

[0011] A modulator in the inverter determines each switching point of the inverter based on the specified stator voltage vector. Generally, four switching modes of the multiphase AC voltage are distinguished: asynchronous pulse width modulation (PWM), synchronous pulse width modulation, overmodulation (OVM), and block switching (6-step).

[0012] If the specified torque requirement varies, for example due to a varying acceleration request from the driver of the electric vehicle, and / or the operating point of the electric motor varies, for example due to a varying load on the electric motor, a deadbeat element of the control electronics can determine the stator space voltage phasor in such a way that, ideally, within the shortest possible time period, i.e., within the fewest possible periods of the multiphase alternating voltage, steady-state operation of the electric motor at a new operating point is achieved again.

[0013] DE 10 2006 052 042 A1 discloses a method for operating an electric motor in which a deadbeat element of the electric motor's control electronics completely or at least largely compensates for a discontinuity of an operating parameter of the electric motor or the control electronics, for example a sudden voltage drop of an intermediate circuit, within a switching cycle.

[0014] EP 2 469 692 A1 also describes a method for operating an electric motor in which a control electronics minimizes a difference between an estimated stator space flux vector of an electric motor, i.e. a stator space flux vector of a magnetic flux of the stator, and a stator space flux vector determined by a control electronics by varying at least one switching point of a predetermined sequence of time switching points which depends on a modulation level and is provided by an assignment table.

[0015] Common control electronics, such as those mentioned above, do not provide all four of the aforementioned clock cycles, or at least not one of them, with a practically acceptable computational effort. If a control electronics unit provides at least two different clock cycles, artifacts can occur in the multiphase AC voltage when switching between them. These artifacts are accompanied by a brief vibration of the electric motor or a brief drop in power or torque, which is undesirable.

[0016] It is therefore an object of the invention to propose a method for operating an electric motor which provides each of the aforementioned four clock cycles with a practically acceptable computational effort and avoids artifacts when switching between the clock cycles. Furthermore, it is an object of the invention to provide control electronics for an electric motor.

[0017] One aspect of the invention is a method for operating an electric motor, in which a flux sensor of a control electronics is controlled depending on a predetermined torque requirement T*. em a stator space flux pointer ψ* αβ determined, a voltage generator of the control electronics depending on the determined stator flux vector ψ* αβ a scaled stator space voltage vector V*' αβ determined, an inverter of the control electronics depending on the determined scaled stator space voltage vector V*' αβ a DC electrical voltage V supplied by an intermediate circuit dc switches and, by means of switching, a multi-phase alternating voltage V* UVW generated and the control electronics drive an electric motor by applying the generated multi-phase alternating voltage V* UVW operates. The stator flux indicator ψ* αßis a setpoint or manipulated variable. Quantities marked with an asterisk (*) are to be understood here as setpoints or manipulated variables. Such operating methods are implemented particularly in electrically powered vehicles. Accordingly, numerous and diverse application possibilities for the invention arise.

[0018] According to the invention, a flow angle actuator of the flow sensor determines the torque angle depending on the specified torque requirement T*. em a stator flux angle δ* αβ , represents an MTPA (Maximum Torque per Ampere) actuator of the flux encoder depending on the specified torque requirement T* em a first stator space flux amplitude ψ MTPA Provides an operating point actuator of the flux encoder depending on the electrical DC voltage V dc, an angular frequency ω e the multiphase alternating voltage V* UVW , a certain estimated stator space current pointer Î αβof the electric motor and an ohmic stator resistance R S of the electric motor a second stator flux amplitude ψ R A flux calculator of the flux sensor is ready and determined depending on the determined stator flux angle δ*. αβ and a ratio of the provided first stator space flux amplitude ψ MTPA to the provided second stator flux amplitude ψ R a trajectory of the stator space flux pointer ψ* aβ and depending on the specific trajectory and the specific stator flux angle δ* αβ the stator flux pointer ψ* αβ Quantities marked with ^ are to be understood here as estimates. The trajectory of the stator space flux vector ψ* aβ is also referred to as stator space flux trajectory. The direct current voltage V dc, the angular frequency ω e , and the estimated stator space current pointer Î αβ define an operating point of the electric motor.

[0019] The flux sensor determines the stator flux angle δ* αβ and the stator flux amplitudes ψ MTPA , ψ R first separately from each other and then combined to determine the specific stator flux angle δ* αβ and the stator flux amplitudes ψ MTPA , ψ R to the stator flux pointer ψ* αβ The flux calculator determines the stator space flux vector ψ* αß or the trajectory of the stator space flux vector ψ* αß in the two-dimensional stator-fixed coordinate system in a polar representation, i.e., each stator space flux vector ψ* aβ or each point of the trajectory is defined as a 2-tuple consisting of the stator flux angle δ* αβ and one of the stator flux angle δ* αβ dependent stator flux amplitude |ψ* αβ | specified. In other words, the specific trajectory is defined together with the stator flux angle δ*. αβa length of the stator space flux pointer ψ* αβ .

[0020] Determining the stator flux amplitudes ψ separately MTPA , ψ R and the stator flux angle δ* αβ This enables the flux encoder to have great flexibility and high speed when determining the trajectory of the stator space flux vector ψ*. αβ .

[0021] The trajectory can firstly be defined as a circle with the provided first stator space flux amplitude ψ MTPA as a radius if the ratio is less than or equal to 0.5*√3. If the provided second stator flux amplitude ψ R Interpreted as a side length of a regular hexagon concentric with the circle, the specified range of the ratio includes every regular hexagon completely enclosed by the circle and maximally inscribed in the circle.

[0022] Advantageously, a flow calculator's amplitude conversion table (LUT) overstates the ratio nonlinearly when the ratio is greater than 0.5√3. The LUT contains multiple discrete pairs of values ​​for a function that is linear between 0 and 0.5√3 and nonlinear above 0.5√3. Thanks to these pairs of values, the nonlinear function does not need to be calculated. A required pair of values ​​is simply approximated by a pair of values ​​read from the LUT, resulting in a reduction in computation time.

[0023] The ratio is exaggerated non-linearly in such a way that a circle initially inscribed within the regular hexagon is continuously transformed into a circle circumscribed by the regular hexagon.

[0024] Secondly, the trajectory can thus be defined as a largest circle with a product of the provided second stator flux amplitude ψ Rand the non-linearly exaggerated ratio as a radius and a regular hexagon concentric with the circle with the provided second stator flux amplitude ψ R The closed curve inscribed on a side length can be determined if the non-linearly exaggerated ratio is less than 0.5*√3.

[0025] The flow calculator determines the trajectory as a combination of respective, internally arranged and interconnected segments of the circle and the regular hexagon at their intersections. Each segment connects two adjacent points of intersection between the circle and the regular hexagon. The largest inscribed closed curve is a hybrid circle-hexagon trajectory, which allows for a continuous deformation of the trajectory from the circle to the regular hexagon. Thanks to this continuous deformation of the trajectory, abrupt transitions between the different clock cycles of the multiphase alternating voltage V* are avoided. UVW and consequently, artifacts when switching between the different timings were avoided.

[0026] Apart from that, calculating the largest inscribed closed curve requires little computational effort. Consequently, the flow computer can determine the trajectory practically in real time if the specified torque requirement T* is met. em or the operating point of the electric motor varies constantly.

[0027] Thirdly, the trajectory can be represented as a regular hexagon with the provided second stator flux amplitude ψ R as a side length if the non-linearly exaggerated ratio is equal to 0.5*√3.

[0028] At the specified value, the regular hexagon is inscribed in the circle enlarged using the amplitude conversion table.

[0029] Fourthly, the trajectory can be determined as an eighteen-sided polygon inscribed in the regular hexagon if the non-linearly exaggerated ratio is equal to 0.5*√3 and the flow calculator is given a reduction factor k. fThe specified reduction factor is less than one. Reducing the flux amplitude allows the corners of the regular hexagon to "fold in," resulting in an eighteen-sided trajectory. The smaller the specified reduction factor k, the more... f The larger the amplitude, the larger the "folded-in" corner of the regular hexagon, and vice versa. The reduced second stator flux amplitude ψ Rf = k f · ψ R defines a distance of the folded corner from the center of the regular hexagon. If the given reduction factor k f Starting from zero and continuously varied, the regular hexagon is accordingly continuously transformed into the eighteen-sided polygon.

[0030] The circle, the regular hexagon, and the eighteen-sided polygon are special forms of the trajectory. Stator space flux vector ψ* αβOn the circular trajectory, synchronous pulse-width modulation (PWM) or asynchronous PWM is produced. Stator flux vectors on the hybrid circular-hexagonal trajectory produce overmodulation (OVM). Accordingly, the product of the provided second stator flux amplitude and the non-linearly increased ratio can be expressed as a stator flux overmodulation amplitude ψ. R-OVM are designated as stator flux indicators ψ* αβ The block timing (6-step) is achieved on the regularly hexagonal trajectory. Stator flux indicator ψ* αβ on the eighteen-sided trajectory, a synchronous triple switching (3-pulse switching) is achieved.

[0031] It is noted that the control electronics do not calculate the aforementioned trajectories in parallel, but rather alternatively depending on the case, i.e., exactly one of the aforementioned trajectories is calculated at any given time, which results in a short computing time.

[0032] Preferably, a PI torque actuator of the flux angle actuator adjusts the torque according to the specified torque requirement T*. em a first rotor spatial angle δ* PI Provides an angle conversion table (look-up table, LUT) of the flux angle actuator depending on the specified torque requirement T* em a second rotor spatial angle δ* LUT The flux angle actuator is ready and determines the stator flux angle δ*. αβ depending on a rotor angle θ r of the electric motor, the provided first rotor solid angle δ* PI and the provided second rotor spatial angle δ* LUTThe angle conversion table (look-up table, LUT) contains a plurality of discrete pairs of values ​​for a nonlinear function. Consequently, the nonlinear function does not need to be calculated. A required pair of values ​​is simply approximated by a pair of values ​​read from the angle conversion table, which reduces computation time. The angle conversion table allows for an increased dynamic range of the PI torque actuator or the flux angle actuator.

[0033] The first rotor spatial angle δ PI and the second rotor solid angle δ LUT are given in a two-dimensional rotor-fixed coordinate system. The two coordinate axes of the two-dimensional rotor-fixed coordinate system are usually denoted by d and iq.

[0034] Furthermore, preferably a deadbeat element of the voltage generator is determined by means of the determined stator space flux vector ψ* αβ a stator space voltage pointer V*αβ and determines a voltage pointer scaler of the voltage generator depending on the determined stator space voltage pointer V*. αβ the scaled stator space voltage pointer V*' αβ .

[0035] The deadbeat element can determine the stator space voltage pointer V*. αβ according to Vαβ*=Ψαβ*−Ψ^αβ*Te+RSl^αβ determine, whereby Ψ^αβ* an estimated stator space flux indicator, T e one period of the multiphase alternating voltage V* UVW , R S an ohmic stator resistance of the electric motor and Î αβ The estimated stator space current phasor is used. The voltage phasor scaler can determine the amplitude of the scaled voltage phasor.

[0036] An inverter modulator can determine a switching point dependent on a specific scaled stator space voltage vector, independently of a clock cycle of the control electronics. For example, the clock cycle, i.e., a calculation period T, can t The control electronics' interval can be 100 µs, which corresponds to a control electronics processing frequency of 10 kHz. In contrast, a switching clock of the control electronics, i.e., one period T, can be... e the multiphase alternating voltage V* UVW , 1.176 ms, which corresponds to a switching frequency of 850 Hz. In particular, a ratio of the processing clock T requires t and the switching cycle T e or that the ratio of the computing frequency to the switching frequency is not an integer.

[0037] Preferably, the modulator orders within one processing cycle T. t no switching point, one switching point, or two switching points. In other words, within one processing cycle T,t Not switched, switched exactly once, or switched exactly twice. If within the calculation cycle T t If the clock is switched exactly once, this can, for example, occur in an earlier (left) half of the processing cycle T. t or in a later (right) half of the calculation cycle T t This occurs within the calculation cycle T. t If the switching occurs exactly twice, for example, the first switching can take place in an earlier (left) half of the switching cycle T. t and a second switch T t in a later (right) half of the switching cycle. By means of the switching points within the calculation cycle T t A duty cycle of the processing clock T is determined. t defined.

[0038] Ideally, the flow computer continuously deforms the trajectory for each modulation level m in a range from 0 to 23 / π continuous. The modulation level m is defined as a ratio of the amplitude of a fundamental oscillation of the polyphase alternating voltage V*. UVW to an amplitude of a periodic modulation alternating voltage provided by the modulator, which is also called a carrier. If the amplitude of the modulation alternating voltage is denoted as V dc / √3 and the maximum amplitude of the fundamental oscillation as 2V dc The maximum modulation level is determined by selecting / π. m=23 / π=1,1027.

[0039] The modulation range from 0 to 1 is called a pulse-width modulation (PWM) range. At a modulation level m in the pulse-width modulation range, each phase of the multiphase AC voltage V applied to the electric motor is modulated. UVWEssentially sinusoidal. If the ratio of a frequency of the polyphase AC voltage to a frequency of the carrier is an integer, the resulting pulse-width modulation is called synchronous. If the ratio of the two frequencies is not an integer, the resulting pulse-width modulation is called asynchronous.

[0040] The modulation range between 1 and 23 / π This is referred to as an overmodulation range. At a modulation level m within the overmodulation range, the phases of the multiphase AC voltage applied to the electric motor are not sinusoidal. At the maximum modulation level m = 1.1027, each phase of the multiphase AC voltage applied to the electric motor essentially has a rectangular shape (block switching, 6-step).

[0041] The overmodulation region can have a first overmodulation subregion (OVM I) and a second overmodulation subregion (OVM II) that differs from the first. For the first overmodulation subregion, 1 < m ≤ 1.05. For the second overmodulation subregion, 1.05 < m < 1.1027.

[0042] Another aspect of the invention is a control electronics unit for an electric motor, comprising an inverter, a modulator, and a deadbeat circuit. Such control electronics units are widely used, thus offering numerous application possibilities for the invention, particularly in the field of e-mobility, i.e., electrically powered vehicles.

[0043] According to the invention, the control electronics further comprise an MTPA actuator, an operating point actuator, a flux angle actuator, and a flux computer, and are configured to operate an electric motor together with an intermediate circuit and the electric motor using a method according to the invention. The control electronics allow the electric motor to be operated practically without latency and without artifacts when switching between significantly different clock speeds of the multiphase alternating voltage.

[0044] A significant advantage of the method according to the invention is that any known clock frequency of a multiphase alternating voltage is provided for operating an electric motor with a practically acceptable computational effort, and artifacts during transitions between different clock frequencies are avoided. In this way, the electric motor is operated across the entire modulation range without momentary vibration or momentary drops in power or torque.

[0045] The invention is schematically illustrated in the drawings with reference to one embodiment and is further described with reference to the drawings. It shows: Fig. 1 a block diagram of a control electronics according to an embodiment of the invention; Fig. 2 a section of one of the in Fig. 1 control electronics shown, defined hexagonal trajectory and a voltage hexagon corresponding to a stator space flux pointer; Fig. 3 two of the in Fig. 1. Control electronics shown 1. Specific trajectories; Fig. 4 one to the in Fig. The 3 eighteen-sided trajectories shown correspond to a multiphase alternating voltage; Fig. 5 one to the in Fig. The 3 hexagonal trajectories shown correspond to a multiphase alternating voltage; Fig. 6 four of the in Fig. Trajectories generated by the control electronics shown in section 1.

[0046] Fig. Figure 1 shows a block diagram of a control electronics unit 1 according to an embodiment of the invention. The control electronics unit 1 comprises a flux sensor 12, a voltage sensor 11, and an inverter 10.

[0047] The flux encoder 12 includes a flux calculator 13, which may comprise an amplitude conversion table 130. Furthermore, the flux encoder 12 may include a flux angle actuator 14 with a flux angle conversion table 140 and a PI torque actuator 141, an MTPA actuator 15, and an operating point actuator 16.

[0048] The voltage generator 11 can include a deadbeat circuit 112 and a voltage pointer scaler 111. Furthermore, the control electronics 1 can include an averaging circuit 17, an estimating circuit 18, and a current limiter 19.

[0049] The control electronics 1 are configured to operate an electric motor 2 together with an intermediate circuit 3 and the electric motor 2 using a method according to an embodiment of the invention as follows. The electric motor 2 comprises, by way of example and without limitation, three star-connected stator windings. The method can readily be adapted to electric motors with five, seven, or nine star-connected stator windings.

[0050] Depending on a given torque requirement 4, the flux sensor 12 of the control electronics 1 determines a stator space flux indicator 132.

[0051] The estimating element 18 of the control electronics 1 determines, depending on a specific scaled stator space voltage phasor 110, a rotor angle 20 of the electric motor 2 and a measured stator current 21 of the electric motor 2, an estimated torque 180, an estimated stator space flux amplitude 181, an estimated stator space flux phasor 182 and an estimated stator space current phasor 183 of the electric motor 2.

[0052] The averaging element 17 determines a mean estimated torque 170 and a mean estimated stator flux amplitude 171 depending on the estimated torque 180 and the estimated stator flux amplitude 181.

[0053] Depending on a given maximum current 6 and a stator space current 184, the current-limited maximum torque 190 determines a current-limited maximum torque 190, which limits the given torque requirement 4.

[0054] The flux angle actuator 14 of the flux encoder 12 determines a stator flux angle 142 depending on the specified torque requirement 4. For this purpose, the PI torque actuator 141 of the flux angle actuator 14 can provide a first rotor flux angle 1410 depending on the specified torque requirement 4 and, in particular, depending on the mean estimated torque 180. An angle conversion table 140 of the flux angle actuator 14 can provide a second rotor flux angle 1400 depending on the specified torque requirement 4 and, in particular, depending on the mean estimated stator flux amplitude 181. The flux angle actuator 14 can then determine the stator flux angle 142 depending on a rotor angle 20 of the electric motor 2, the provided first rotor flux angle 1410, and the provided second rotor flux angle 1400.

[0055] Depending on the specified torque requirement 4, the MTPA actuator 15 of the flux encoder 12 provides a first stator flux amplitude 150. Depending on the DC voltage, an angular frequency of the multiphase AC voltage 100, a specific estimated stator current phasor 183 of the electric motor 2, and an ohmic stator resistance 22 of the electric motor 2, the operating point actuator 16 of the flux encoder 12 provides a second stator flux amplitude 160.

[0056] Depending on the determined stator flux angle 142 and a ratio of the provided first stator flux amplitude 150 to the provided second stator flux amplitude 160, the flux calculator 133 of the flux pointer 132 determines a trajectory 131.

[0057] The trajectory 131 can be determined as a circle with the provided first stator space flux amplitude 150 as a radius if the ratio is less than or equal to 0.5*√3.

[0058] The amplitude conversion table 130 of the flow calculator 13 can non-linearly overstate the ratio if the ratio is greater than 0.5*√3.

[0059] The trajectory 1311 is determined as a largest closed curve 1311 inscribed in a circle 1310 with a product of the provided second stator flux amplitude 160 and the non-linearly superimposed ratio as a radius and a regular hexagon 1312 concentric with the circle 1310 with the provided second stator flux amplitude 160 as a side length, if the non-linearly superimposed ratio is less than 0.5*√3.

[0060] The trajectory 131 can be determined as a regular hexagon 1312 with the provided second stator space flux amplitude 160 as one side length, if the non-linearly exaggerated ratio is equal to 0.5*√3.

[0061] The trajectory 131 can be determined as a largest closed curve 1311 inscribed in a circle 1310 with a product of the provided second stator flux amplitude 160 and the non-linearly superimposed ratio as a radius and a regular hexagon 1312 concentric with the circle 1310 with the provided second stator flux amplitude 160 as a side length, if the non-linearly superimposed ratio is greater than 0.5*√3.

[0062] The trajectory 131 can be determined as an eighteen-sided polygon 1313 inscribed in the regular hexagon 1312 if the non-linearly exaggerated ratio is equal to 0.5*√3 and the flow calculator 13 is given a reduction factor 5 less than one.

[0063] Depending on the determined trajectory 131 and the determined stator flux angle 142, the flux calculator 132 determines the stator flux pointer 132.

[0064] Depending on the determined stator flux pointer 132, the voltage generator 11 of the control electronics 1 determines a scaled stator voltage pointer 110.

[0065] In particular, the deadbeat element 112 of the voltage generator 11 determines a stator voltage phasor 1120 using the determined stator flux phasor 132, and the voltage phasor scaler 111 of the voltage generator 11 determines the scaled stator voltage phasor 110 depending on the determined stator voltage phasor 1120. The deadbeat element 112 determines the stator voltage phasor 1120 depending on the determined estimated stator flux phasor 182 and the determined estimated stator current phasor 183.

[0066] Depending on the specific scaled stator space voltage pointer 110, the inverter 10 of the control electronics 1 switches a DC voltage provided by the intermediate circuit 3 and generates a multi-phase AC voltage 100 by means of switching.

[0067] The control electronics 1 operate the electric motor 2 by applying the generated multi-phase alternating voltage 100.

[0068] A (not shown) modulator of the inverter 10 can determine a switching time of the inverter 10 that depends on the scaled stator space voltage phasor 110 independently of a calculation clock 7 (see Fig. 2 to 5) of the control electronics 1.

[0069] The modulator can arrange no switching point, one switching point, or two switching points within a calculation cycle 7.

[0070] The flow computer 13 can continuously calculate the trajectory 131 for every modulation level in a range from 0 to 23 / π deform continuously.

[0071] Fig. 2 a) shows a section of one of the in Fig. The control electronics shown in Figure 1 have a defined hexagonal trajectory 1312 and a voltage hexagon 101 of the modulator corresponding to a stator flux vector 132. The section comprises four calculation cycles 7. Four stator flux vectors are plotted on the hexagonal trajectory 1312, corresponding to calculation times k-1, k, k+1, k+2. Each calculation cycle 7 between two adjacent calculation times k-1, k, k+1, k+2 is, for example, 100 µs. One corner of the hexagonal trajectory 1312 is located within the calculation cycle k. Accordingly, the calculation cycle k comprises two clock segments 7a, 7b in a defined duty cycle.

[0072] Fig. 2 b) shows three phases of the multiphase alternating voltage 100, each as phase-to-zero alternating voltages V U , V V , V WDuring the four calculation cycles, the modulator assigns a switching point of phase V within an earlier (left) half of the calculation cycle k, according to the defined duty cycle. V on. The two phases V U and V W are not switched within the calculation clock k.

[0073] Fig. 3 shows two of the in Fig. The control electronics shown in Figure 1 exhibit specific trajectories 1312 and 1313. The hexagonal trajectory 1312 and the eighteen-sided trajectory 1313 are represented in a stator-fixed coordinate system. Furthermore, Figure 1 shows Fig. 3 a stator flux pointer 132 belonging to the eighteen-sided trajectory 1313 with a stator flux angle 142. In Fig. Figure 3 shows the second stator flux amplitude 160 and the second stator flux amplitude 161 reduced by means of the reduction factor 5.

[0074] Fig. 4 shows one of the ones in Fig. 3 eighteen-sided trajectory shown 1313 corresponding multiphase alternating voltage 100. In Fig. 4 a) is a period 1001 of two phases U, V of the multiphase alternating voltage 100 shown as a phase-to-phase alternating voltage, i.e. V UV . In Fig. 4b) is the period 1001 of a phase of the multiphase alternating voltage 100 as a phase-to-zero alternating voltage, for example V U , shown. In Fig. 4c) are three phases of the multiphase alternating voltage 100 each as phase-to-zero alternating voltages V U , V V , V W The graph shows the periods, each phase-shifted by 120°. For example, a period of 1001 corresponds to 1.167 ms, or a frequency of 1000 of 850 Hz. In contrast, a calculation period is 100 µs, which corresponds to a calculation frequency of 10 kHz. The ratio between the frequency of 1000 and the calculation frequency is not an integer. The switching points each lie within the calculation periods.

[0075] Fig. 5 shows one of the ones in Fig. 3 hexagonal trajectories shown 1312 corresponding multiphase alternating voltage 100. In Fig. 5 a) is a period 1001 of two phases U, V of the multiphase alternating voltage 100 shown as a phase-to-phase alternating voltage, i.e. V UV , shown. In Fig. 5 b) is the period 1001 of a phase of the multiphase alternating voltage 100 as a phase-to-zero alternating voltage V U shown. In Fig. 5c) are three phases of the multiphase alternating voltage 100 as phase-to-zero alternating voltages V U , V V , V W The diagram shows the signals, each phase-shifted by 120°. The ratio of the frequency (1000) to the calculation frequency is not an integer. The switching points each lie within calculation periods.

[0076] Fig. 6 shows four of the in Fig. The control electronics shown in Figure 1 generated trajectories, which are labeled (a), (b), (c), and (d). Trajectories (a) and (b) are each circles 1310. Trajectory (c) is a maximum closed curve 1311 inscribed in a circle and a regular hexagon intersecting the circle and concentric with the circle. Trajectory (d) is a regular hexagon 1312.

[0077] Furthermore, it shows Fig. 6. For each trajectory (a), (b), (c), and (d), a plurality of stator space voltage phasors 1120 are defined, each belonging to a different calculation cycle. The plurality of stator space voltage phasors 1120 for trajectory (a) corresponds to a modulation level m = 1. A modulation level m = 1 marks an upper limit of the pulse width modulation range. The stator space voltage phasor 1120 follows a circle in a sequence of calculation cycles.

[0078] The majority of stator space voltage vectors 1120 for the trajectory (b) corresponds to the modulation level m = 1.1027. The modulation level m = 1.1027 marks the block clocking. The stator space voltage vector 1120 points exclusively to the vertices of a regular hexagon in a sequence of calculation cycles.

[0079] The majority of stator space voltage phasors 1120 to the trajectories (b), (c) correspond to modulation intensities 1 < m < 1.1027. Modulation intensities 1 < m < 1.1027 lie within the overmodulation subregion (OM). The overmodulation subregion comprises a first overmodulation subregion (OM I) and a second overmodulation subregion (OM II) distinct from the first.

[0080] The majority of stator space voltage phasors 1120 for the trajectory (b) correspond to a modulation level m = 1.05. The modulation level m = 1.05 marks an upper limit of the first overmodulation sub-region. The stator space voltage phasor 1120 follows a regular hexagon in a sequence of calculation cycles.

[0081] The majority of stator space voltage phasors 1120 for trajectory (c) correspond to a modulation level of 1.05 < m < 1.1027. The modulation level of 1.05 < m < 1.1027 lies within the second overmodulation sub-region. In a sequence of calculation cycles, the stator space voltage phasor 1120 points exclusively to corner regions of a regular hexagon. REFERENCE MARK LIST: 1 Control electronics 10 inverters 100 multi-phase alternating voltage V* UVW 1000 angular frequency ω e 1001 Period T e = 2π / ω e 101 Voltage hexagon 11 voltage generators 110 scaled stator space voltage pointer V*' αβ 111 Voltage pointer scale 112 Deadbeat Link 1120 Stator space voltage pointer V* αβ 12 flow meters 13 flow calculators 130 Amplitude Conversion Table 131 Trajectory 1310 district 1311 largest inscribed closed curve 1312 regular hexagon 1313 Octadecagon 132 Stator space flux indicator ψ* αβ 14 Flux angle actuator 140 Angle Conversion Table 1400 second rotor spatial angle δ* LUT 141 PI torque actuator 1410 first rotor spatial angle δ PI 142 Stator flux angle δ* αβ 15 MTPA actuator 150 first stator flux amplitude ψ MTPA 16 Operating point actuator 160 second stator flux amplitude ψ R 161 reduced second stator flux amplitude ψ Rf 17th mediating element 170 mean estimated torque T̂ em-avg 171 mean estimated stator space flux amplitude 18 estimator 180 estimated torque T̂ em 181 estimated stator space flux amplitude 182 estimated stator space flux indicator Ψ^αβ* 183 estimated stator space current indicator Î αβ 184 estimated stator space current 19 Current limiting element 190 current-limited maximum torque T* em max -cl 2 electric motors 20 Rotor angle θ r 21 measured stator current i measured 22 ohmic stator resistance R S 3 Intermediate circle 30 DC voltage V dc 4 specified torque requirement T* em 5 Reduction factor k f 6 Maximum current i max 7 Calculation clock T t

Claims

[1] Method for operating an electric motor (2) wherein - a flux sensor (12) of a control electronics (1) determines a stator space flux indicator (132) depending on a specified torque requirement (4); - a voltage generator (11) of the control electronics (1) determines a scaled stator voltage vector (110) depending on the determined stator flux vector (132); - an inverter (10) of the control electronics (1) switches a DC voltage provided by an intermediate circuit (3) depending on the specific scaled stator space voltage phasor (110) and generates a multi-phase AC voltage (100) by means of switching; - the control electronics (1) operates an electric motor (2) by applying the generated multi-phase alternating voltage (100); - a flux angle actuator (14) of the flux sensor (12) determines a stator space flux angle (142) depending on the specified torque requirement (4); - an MTPA actuator (15) of the flux sensor (1) provides a first stator space flux amplitude (150) depending on the specified torque requirement (4); - an operating point actuator (16) of the flux encoder (12) provides a second stator flux amplitude (160) depending on the electrical DC voltage, an angular frequency of the multiphase AC voltage (100), a certain estimated stator space current phasor (183) of the electric motor (2) and an ohmic stator resistance (22) of the electric motor (2); - a flux calculator (13) of the flux sensor (12) determines a trajectory (131) of the stator flux pointer (132) depending on the determined stator flux angle (142) and a ratio of the provided first stator flux amplitude (150) to the provided second stator flux amplitude (160) and determines the stator flux pointer (132) depending on the determined trajectory (131) and the determined stator flux angle (142). [2] Method according to claim 1, wherein the trajectory (131) is determined as a circle with the provided first stator space flux amplitude (150) as a radius when the ratio is less than or equal to 0.5*√3. [3] Method according to claim 1 or 2, wherein an amplitude conversion table (130) of the flow computer (13) non-linearly overscores the ratio when the ratio is greater than 0.5*√3. [4] Method according to claim 3, wherein the trajectory (131) is determined as a largest closed curve (1311) inscribed in a circle (1310) with a product of the provided second stator flux amplitude (160) and the nonlinearly superimposed ratio as a radius and a regular hexagon (1312) concentric with the circle (1310) with the provided second stator flux amplitude (160) as a side length, if the nonlinearly superimposed ratio is less than 0.5*√3, and / or wherein the trajectory (131) is determined as a regular hexagon (1312) with the provided second stator flux amplitude (160) as a side length, if the nonlinearly superimposed ratio is equal to 0.5*√3. [5] Method according to claim 4, wherein the trajectory (131) is determined as an eighteen-sided polygon (1313) inscribed in the regular hexagon (1312) when the non-linearly superimposed ratio is equal to 0.5*√3 and the flow computer (13) is given a reduction factor (5) less than one. [6] A method according to any one of claims 1 to 5, wherein a PI torque actuator (141) of the flux angle actuator (14) provides a first rotor solid angle (1410) depending on the specified torque requirement (4), an angle conversion table (140) of the flux angle actuator (14) provides a second rotor solid angle (1400) depending on the specified torque requirement (4), and the flux angle actuator (14) determines the stator solid angle (142) depending on a rotor angle (20) of the electric motor (2), the provided first rotor solid angle (1410), and the provided second rotor solid angle (1400), and / or wherein a deadbeat element (112) of the voltage generator (11) determines a stator solid voltage vector (1120) by means of the determined stator solid flux vector (132), and a voltage vector scaler (111) of the voltage generator (11) depending on the determined stator space voltage phasor (1120) the scaled stator space voltage phasor (110) is determined. [7] Method according to any one of claims 1 to 6, wherein a modulator of the inverter (10) determines a switching time of the inverter (10) dependent on the scaled stator space voltage vector (110) independently of a calculation clock (7) of the control electronics (1). [8] Method according to claim 7, wherein the modulator arranges no switching time, one switching time or two switching times within a calculation cycle. [9] Method according to any one of claims 1 to 8, wherein the flow computer (13) continuously calculates the trajectory (131) for each modulation level applied by the modulator (11) in a range from 0 to 23 / π constantly deformed. [10] Control electronics (1) for an electric motor (2), comprising an inverter (10), a voltage sensor (11), a flux sensor (12) with a flux computer (13) and a flux angle actuator (14), an MTPA actuator (15) and an operating point actuator (16), and configured to operate an electric motor (2) together with an intermediate circuit (3) and the electric motor (2) according to one of claims 1 to 9.

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