Sensor component with improved overload and interference resistance

Abstract

A sensor component is disclosed comprising a transducer and an electrical circuit arranged in a single package. The electrical circuit includes a bias voltage coupled to a first electrode of the transducer. A non-inverting amplifier stage is coupled to a second electrode of the transducer. An inverting amplifier stage is coupled to an output of the non-inverting amplifier stage. A negative feedback path is located between an output of the inverting amplifier stage and the first electrode of the transducer. An electrical output signal of the transducer can be attenuated by applying an attenuation signal, based on a filtered electrical signal, to the first electrode via the negative feedback path.

Description

AREA OF REVELATION

[0001] The present disclosure relates generally to sensor components and in particular to sensor components that have improved overload performance, as well as electrical circuits for such components. BACKGROUND

[0002] Microelectromechanical systems (MEMS) microphone components, comprising a transducer and an ASIC housed in a package with a device interface, are well-known and used in in-the-ear hearing aids, consumer electronics, appliances, devices, and other host systems. These and other microphones incorporate a capacitive MEMS motor or other transducer that generates an electrical output signal in response to detected acoustic signals. The electrical output signal is buffered or amplified at a front end of the ASIC before further processing and output at the device interface. However, large-amplitude electrical signals generated by the transducer can overload the ASIC's front end and impair the microphone's acoustic overload point (AOP) performance.AOP (Auditory Operational Performance) in the hearing aid industry is typically defined as the input sound pressure level (dB SPL) at which the total harmonic distortion (THD) of the microphone output signal reaches 10% at 1 kHz. Furthermore, some amplifiers commonly used to reduce intermodulation distortion (IMD) in MEMS microphones can negatively impact AOP performance. Therefore, there is a continuous need for microphones and other sensor components with improved overload performance, as well as electrical circuitry for such sensors. BRIEF DESCRIPTION OF THE DRAWINGS

[0003] The objectives, features, and advantages of the present disclosure are explained in more detail below with reference to the detailed description and the accompanying claims in conjunction with the accompanying drawings. The drawings show only representative embodiments and implementations and do not limit the teaching of the disclosure, the scope of which is defined by the accompanying claims. Fig. Figure 1 is an exploded view of a representative sensor component. Fig. 2 is a cross-sectional view of the sensor. Fig. 2. Fig. Figure 3 is a block diagram of a representative electrical circuit implementation for a sensor component. Fig. Figure 4 is a representative electrical circuit implementation of the block diagram from Fig. 3. Fig. 5 is a representative alternative inverting amplifier stage for the electrical circuit consisting of Fig. 4. Fig. Figure 6 is a block diagram of an alternative electrical circuit for a sensor component. Fig. Figure 7 is a representative circuit implementation of the block diagram from Fig. 6. Fig. Figure 8 shows diagrams of the microphone output signals for different attenuation levels. Fig. Figure 9 shows diagrams of the sensor output signal THD compared to the input signal sound pressure level (SPL) for different attenuation levels.

[0004] Those skilled in the art will understand that the drawings are presented for the sake of simplicity and clarity and therefore may not be to scale and may not include known features, that the sequence of actions or steps may differ from the sequence described, that some or all of these actions or steps may be carried out simultaneously unless otherwise stated, and that the terms and expressions used herein will have meanings known to those skilled in the art unless expressly assigned a different meaning here. DETAILED DESCRIPTION

[0005] The disclosure relates generally to sensor components and, in particular, to sensor components with improved acoustic overload point (AOP) performance and electrical circuits for such sensor components. Acoustic sensor components, which include microphones and vibration sensors, are commonly integrated into in-the-ear hearing devices, mobile phones, PCs, televisions, and smart speakers, as well as many other consumer devices. These and other sensor components are also integrated into vehicles and appliances, durable goods, machinery and equipment, and other hosts.

[0006] The sensor component generally comprises a transducer and an electrical circuit, arranged in a housing with an external electrical interface. Fig. Figure 1 comprises a representative sensor 100, a housing with a cover 102 mounted on a base 104. A protective ring 106 is optionally located between the cover and the substrate. The cover, protective ring, and base may include conductive materials to electrically shield the transducer and the electrical components within the housing. The cover may consist of a metal housing or metallized PCB materials. The base may also be a PCB or another substrate with one or more layers.

[0007] The transducer is electrically connected to the electrical circuit, and the electrical circuit is electrically connected to the external electrical interface of the enclosure. The transducer generally comprises one or more electrodes (e.g., a membrane or diaphragm) that are movable relative to a fixed electrode. The displacement of the movable electrode relative to the fixed electrode in response to a detected state forms the basis for generating an electrical signal that represents the detected state. Representative transducers include capacitive microelectromechanical systems (MEMS), also known as MEMS dies or MEMS motors, electrocapacitive motors (ECMs), piezoelectric devices and photocells, and other devices that incorporate both fixed and movable electrodes. The electrical circuit can be implemented as one or more integrated circuits (ICs) or application-specific integrated circuits (ASICs).

[0008] In the Fig. 1 and Fig. 2 The representative sensor component 100 is implemented as a microphone comprising an acoustic transducer 110, which is arranged above an acoustic port 114 in the base 104. In Fig. 2. The transducer 110 comprises a diaphragm 111 that is movable relative to a perforated backplate 113 to respond to changes in acoustic pressure entering the interior of the housing through the sound opening 114. Alternatively, the acoustic transducer can be arranged above an acoustic port in the cover. In other implementations, the sensor is a vibration sensor or accelerometer without an acoustic port in the housing. The transducer of a vibration sensor or accelerometer can be configured as a test mass. Other implementations include sensor components for detecting gas under other conditions.

[0009] In Fig. 1 is an IC 112 arranged in the package and electrically connected to contacts 108 on the base. In Fig. 2. The MEMS converter 110 is wire-bonded to the IC 112, and the IC is wire-bonded to contacts on the base. Alternatively, the IC can be surface-mounted on the contacts. The contacts on the base can be electrically connected to the electrical interface 120 on the outside of the package via vias extending through the base. Alternatively, the IC can be mounted on another surface inside the package, such as the cover, and connected to the electrical interface via conductors extending through the side walls or another structure of the package.

[0010] The electrical circuit generally comprises a bias voltage coupled to a first electrode of the converter, a non-inverting amplifier stage with an input coupled to a second electrode of the converter, an inverting amplifier stage coupled to an output of the non-inverting amplifier stage, a negative feedback path between an output of the inverting amplifier stage and the first electrode of the converter, and a filter configured to filter an electrical signal from the circuit. An electrical output signal of the converter can be attenuated by a negative feedback signal (attenuation signal) applied to the first electrode via the negative feedback path, with the attenuation signal superimposed on the filtered electrical signal.In some implementations, the attenuation signal is selectively applied to the transducer, based on a signal detected at the input of the inverting amplifier stage. Configured this way, the sensor can detect signals of interest without interference from other signals. Representative electrical circuits are described here.

[0011] In the Fig. 3 and Fig. 6 comprises an electrical circuit 200, a voltage source 202, which is coupled to a converter 110. In the Fig. 4 and Fig. Figure 7 comprises a representative voltage source 202 and a multi-stage charge pump 204, which has a low-pass filter (LPF) at one output coupled to the first electrode of the transducer 110. The representative LPF of the charge pump offers a high output impedance and includes an equivalent resistance of 10 TΩ, coupled to a capacitor C. QRESIt is coupled by a 15 pF capacitor. Alternatively, other resistor and capacitor values ​​can be used.

[0012] In the Fig. 3 and Fig. 6 is a non-inverting amplifier stage 206 coupled with the converter 110. In the Fig. 4 and Fig. 7 is an input V IN The non-inverting amplifier 206 is coupled to the second electrode of the converter 110. The representative converter 110 is a capacitive motor C. MOTOR A high-impedance interface of the non-inverting amplifier stage can prevent charge discharge from the capacitive motor C. MOTOR limit the one connected to entrance V IN is coupled. A pair of diodes connected in parallel with reversed polarity between the input V IN and a reference voltage V REFX This ensures high impedance at input V INThe input impedance is on the order of 100 TΩ, for example 300 TΩ. The non-inverting amplifier stage 206 in Fig. 4 is implemented as a PMOS device. Alternatively, the non-inverting amplifier stage can be implemented as an NMOS device or another low-power device.

[0013] In the Fig. 3 and Fig. In section 4, the non-inverting amplifier stage 206 is coupled to an inverting amplifier stage 208 via a filter network described below. Fig. Section 5 comprises an alternative inverting amplifier stage, an inverting amplifier 216 coupled to an output driver 218. The gain of the inverting amplifier stages 208 in the Fig. 4 and Fig. 5 is based on a ratio of an input capacitor C IN and a feedback capacitor C FB A feedback resistor RFB It provides a DC bias voltage for the inverting amplifier stage 208. The feedback resistor R FB It can be chosen to provide a flat frequency response. For example, a flat audio frequency response can be achieved if R FB 800 MΩ and the capacitor C IN The capacitance is approximately 10 pF. Other resistor and capacitor values ​​can also provide a flat frequency response in this and other sensor implementations, depending on the desired gain. The inverting amplifier is implemented as an NMOS device. Alternatively, the inverting amplifier can be implemented as a PMOS device or another low-power component. In another alternative implementation, the inverting amplifier could be replaced by a non-inverting amplifier, and the output driver could provide the signal inversion.

[0014] In the Fig. 6 and Fig. 7 is the non-inverting amplifier stage 206 coupled with the inverting amplifier stage 208. In Fig. Figure 7 of the inverting amplifier stage comprises an inverting amplifier 216 coupled to a driver 218. The gain of the inverting amplifier stage is based on the ratio between an input capacitor C and the input capacitor C. IN and a feedback capacitor C FBThe DC bias resistors R and 2R can be chosen to provide an internal loop gain (1 + 2R / R) for the driver 218. This internal loop gain can reduce the signal swing of the inverting amplifier 216 required to deliver a desired output. The inverting amplifier is implemented as an NMOS device. Alternatively, the inverting amplifier can be implemented as a PMOS device or another low-power device, among others. Alternatively, the inverting amplifier stage 208 can be made of Fig. 7 in the electrical circuit Fig. 4 and the inverting amplifier stage 208 from Fig. 4 in the electrical circuit Fig. 7 must be implemented.

[0015] In general, the output signal of the sensor component can be based on the output of the non-inverting amplifier stage or the inverting amplifier stage. In the Fig. 4 and Fig. 5 can be the output V OUT1 the non-inverting amplifier stage 206 or the output V OUT The inverting amplifier stage 208 is directly coupled to the external electrical interface of the sensor housing. Fig. 7 can similarly be the output V OUT1 the non-inverting amplifier stage 206 or the output V OUTThe driver 218 is coupled to the electrical interface on the outside of the sensor housing. However, the output of the inverting amplifier stage is phase-shifted relative to the signal at the input of the non-inverting stage. In some implementations, a switch such as a multiplexer may be provided to select the output signal source depending on the sensor's application. Furthermore, one or more additional signal processing circuits may be coupled to the selected output to further process the output signal before it reaches the external electrical interface.

[0016] In Fig. 3 The filter network between the non-inverting amplifier stage 206 and the inverting amplifier stage 208 is a bandpass filter (BPF) 210, which includes an intermediate low-pass filter (LPF) 212 coupled to an intermediate high-pass filter (HPF) 214. Fig. 4 The bandpass filter is located between the output V OUT1 the non-inverting amplifier stage 206 and the input of the inverting amplifier stage 208. The LPF section of the BPF includes a resistor R LPF between the non-inverting output V OUT1 and a node V FLT as well as a capacitor C LPF , which is between node V FLT and is connected to ground or another voltage reference. The HPF section of the BPF includes a capacitor C. HPF between node VFLT and a node V G2 and a resistor R HPF , which is between node V G2 and is connected to ground or another voltage reference. The resistors shown are R LPF and R HPFThe resistors are variable to allow for adjustment of the filter cutoff frequencies. Alternatively, the resistors can have fixed values ​​while the capacitors are adjustable. In microphone sensors, the bandpass filter can have a low cutoff frequency between 20 Hz and 200 Hz and a high cutoff frequency between 4 kHz and 20 kHz. In another implementation, the bandpass filter has a low cutoff frequency of no more than 300 Hz and a high cutoff frequency of no less than 4 kHz. In acoustic vibration sensors, the low and high frequency cutoffs can be selected within a device-specific linear operating range of the sensor. In one implementation, the low frequency cutoff is no more than 300 Hz and the high frequency cutoff is no less than 3 kHz. More generally, the bandpass filter cutoff frequencies can be selected to optimize the acquisition of other signals depending on the sensor application.

[0017] In some embodiments, the electrical circuit includes a feedback control circuit 222 that selectively applies the negative feedback signal to the converter, as in the Fig. 3 and Fig. Figure 6 shows the feedback control circuit. The circuit controls the application of the damping signal to the converter based on the detection of one or more signals downstream of the filter between the non-inverting and inverting amplifier stages, for example at node V. G2 . In Fig. 4. The feedback control circuit is implemented by a switching network 224, which selectively activates / deactivates the damping signal. The switching network 224 can be controlled by a control unit (shown in Fig. 3) can be actuated based on one or more signal levels detected (e.g., by a threshold detector) at the input of the inverting amplifier stage 208. The switch can be controlled based on a time average or other algorithmic analysis of one or more signal levels detected at the output of the filter network.

[0018] In the Fig. 6 and Fig. 7 A negative feedback path 220 is located between an output of the inverting amplifier stage 208 and the converter 110. The output V OUT The driver 218 is connected to the capacitive motor C via a bandstop filter, which is described further herein. MOTOR coupled. In Fig. 7 The bandstop filter 226 comprises a low-pass filter (LPF) and a high-pass filter (HPF), which are located between the output of the driver 218 and the base plate of the C QRES-Filter capacitor of the bias source 202. The LPF includes a capacitor C LPF and a resistor R LPF between the output of driver 218 and ground or another voltage reference. The HPF includes a resistor R. HPF and a capacitor C HPF parallel to the LPF capacitor C LPF A junction between the resistor R HPF and a capacitor C HPF is via capacitor C QRES the preload with the capacitive motor C MOTOR coupled. The resistors R LPF and R HPF The resistors are variable to allow for adjustment of the cutoff frequencies. Alternatively, the resistors can have fixed values ​​and the capacitors can be variable.

[0019] In the electrical circuits of Fig. 6 and Fig. 7. The negative feedback signal (attenuation signal) attenuates predominantly signals from the transducer at frequencies outside the suppressed band of the band-stop filter. Thus, the low and high cutoff frequencies of the band-stop filter can be selected to optimize the transmission of the signals of interest with minimal interference from other unwanted signals. In microphone sensors, the band-stop filter can have a low cutoff frequency between 20 Hz and 200 Hz and a high cutoff frequency between 4 kHz and 20 kHz. In another implementation, the band-stop filter has a low cutoff frequency of no more than 300 Hz and a high cutoff frequency of no less than 4 kHz. In acoustic vibration sensors, the low and high frequency cutoff frequencies can be selected within a device-specific linear operating range of the sensor.In one implementation, the low-frequency cutoff frequency is no more than 300 Hz and the high-frequency cutoff frequency is no less than 3 kHz. More generally, the cutoff frequencies of the bandstop filter can be selected to optimize the acquisition of other signals depending on the sensor application.

[0020] Fig. Figure 8 shows graphically simulated output signals from sensor components for different levels of signal attenuation applied to an input signal of 600 mVpk (~122 dB SPL). The DC average is 450 mV, and the attenuation threshold is ±200 mV. At 0 dB attenuation, the output signal is subject to significant limiting, resulting in a total harmonic distortion (THD) of 11.33%, which is above the 10% guideline for medical hearing devices. At 3 dB SPL attenuation, the limiting of the output signal is reduced, resulting in a THD of 6.484%, which is below the 10% guideline. At 6 dB SPL attenuation, the clipping is eliminated, but the THD increases to 9.202%. For reference, see Fig. 8 an undistorted output signal generated by a supply voltage of 0.9 V without overdriving.

[0021] Fig.Figure 9 graphically shows the THD of a 1 kHz output signal compared to the input sound pressure level (dB SPL) for various signal attenuation levels. At 0 dB attenuation, the THD begins to exceed 10% at approximately 122 dB SPL. At 3 dB attenuation, the THD begins to exceed 10% at approximately 123 dB SPL. At 4.5 dB attenuation, the THD begins to exceed 10% at slightly more than 123 dB SPL. At 6 dB attenuation, the THD begins to exceed 10% at approximately 124 dB SPL.

[0022] While the disclosure and the currently considered best embodiment have been described in a manner that establishes ownership and enables a person skilled in the art to manufacture and use them, it is, however, self-evident and understandable that there are many equivalents to the representative embodiments described herein and that countless modifications and variations can be made to them without departing from the scope and spirit of the invention, which is limited not by the described embodiments but by the attached claims and their equivalents.

[0023] Claimed is:

Claims

[1] Sensor component, comprising: a converter arranged in a housing; an electrical circuit arranged within the housing and electrically connected to the transducer and an electrical interface on an outside of the housing, the electrical circuit comprising: a voltage source connected to a first electrode of the converter; a non-inverting amplifier stage that has a second electrode of the converter; an inverting amplifier stage coupled to an output of the non-inverting amplifier stage; a negative feedback path between an output of the inverting amplifier stage and the first electrode of the converter; a filter configured to filter an electrical signal from an electrical circuit, wherein an electrical output signal of the converter is attenuated by an attenuation signal applied to the first electrode via the negative feedback path, the attenuation signal being based on the filtered electrical signal. [2] Sensor component according to claim 1, wherein the filter is a bandpass filter located between the non-inverting amplifier stage and the inverting amplifier stage. [3] Sensor component according to claim 2, which further comprises a signal detector between the bandpass filter and the inverting amplifier stage and an attenuation signal actuator which applies the attenuation signal to the converter on the basis of a signal detected by the signal detector. [4] Sensor component according to claim 2, wherein the filter is a bandpass filter having a lower cutoff frequency of not more than 300 Hz and an upper cutoff frequency of not less than 4 kHz. [5] Sensor component according to claim 2, wherein the filter has a bandpass filter having a lower cutoff frequency between 20 Hz and 200 Hz and an upper cutoff frequency between 4 kHz and 20 kHz. [6] Sensor component according to claim 2, wherein the inverting amplifier stage comprises an inverting amplifier coupled to a driver and the negative feedback path is located between an output of the driver and the converter. [7] Sensor component according to claim 6, which further comprises a signal detector at an input of the inverting amplifier stage and an attenuation signal actuator which applies the attenuation signal based on a signal detected by the signal detector. [8] Sensor component according to claim 7, wherein the damping signal actuator applies the damping signal to the transducer when a signal level detected by the signal detector reaches a threshold value. [9] Sensor component according to claim 1, wherein the filter is a bandstop filter between the output of the inverting amplifier stage and the converter, wherein the attenuation signal predominantly attenuates signals from the converter at frequencies outside a rejected band of the bandstop filter. [10] Sensor component according to claim 9, wherein the bandstop filter has a low cutoff frequency of not more than 300 Hz and a high cutoff frequency of not less than 4 kHz. [11] Sensor component according to one of claims 4, 5 and 9, wherein the sensor component is an acoustic sensor. [12] Sensor component according to claim 1, wherein the transducer comprises a MEMS transducer (microelectromechanical system) arranged above a sound opening in the housing. [13] Sensor component according to claim 12, wherein the input of the non-inverting amplifier stage has an impedance of more than 100 TΩ. [14] Sensor component according to claim 13, wherein the inverting amplifier stage has an input coupled to an output of the non-inverting amplifier stage.

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