Current source type motor drive control for ac motor

Abstract

To provide a controller for a multi-phase AC motor having multiple coils.SOLUTION: A controller has: multiple AC current sources each of which is connected to one of multiple coils to supply sine-wave current to one of the multiple coils; and a control logic which is connected to the multiple AC current sources and forms a sine-wave voltage reference to control the action of an AC induction motor in input to each of the AC current sources driving the coils connected to each of the AC current sources by using the sine-wave AC current of a specified phase, a variable frequency and a variable amplitude.SELECTED DRAWING: Figure 2

Description

[Technical field]

[0001] This application claims priority to Provisional Application No. 62 / 774,932, filed December 4, 2018, the entire contents of which are incorporated herein by reference.

[0002] The present disclosure relates to a novel control concept for controlling AC induction motors that relies on current source type control rather than traditional voltage source type control. [Background technology]

[0003] For decades, AC motors have served every industry, and AC motors continue to be the workhorse of many industrial applications around the world. In many cases, such as direct AC voltage applications, these motors are directly connected to a three-phase AC voltage source that meets the specific motor rating. For both variable speed applications and certain control applications, a three-phase Voltage Source Inverter (VSI) with a DC link voltage capable of resolving the line voltage is used to control the AC induction motor.

[0004] VSI controls AC induction motors by regulating both voltage and frequency, a standard prior art approach using Volts per Hertz (V / F) and Field Oriented Control techniques. Motor control using VSI is achieved by sinusoidal modulation of the pole voltages per phase based on incremental integral volt-seconds with the line or motor phase inductance, resulting in quasi-sinusoidal motor currents.

[0005] This approach has a very long history and is well established. However, the complexities of providing a high DC-link voltage and of performing inverter phase-leg modulation result in a motor control solution that has many drawbacks. These drawbacks include the need for large and costly DC bus capacitors, a large form factor with excessive weight, power losses and high frequency losses, high electromagnetic interference (EMI) losses and electromagnetic compatibility (EMC) issues, high stress on the motor insulation, and bearing currents.

[0006] Although the present disclosure describes a specific embodiment for controlling an asynchronous AC motor (i.e., an induction motor), those skilled in the art will recognize that the solution provided herein is equally applicable to the control of both synchronous and asynchronous AC motors in that it eliminates the above-mentioned drawbacks of VSI control and power stage modulation by using a directly coupled current source to control the AC motor without the need to rely on a rated AC voltage. Thus, the teachings disclosed herein may be applied to many AC motor applications, and are particularly well suited to cooling fan motors that are required to operate in environments requiring extremely low EMI.

[0007] By way of further background, in the prior art, AC motors, both synchronous and asynchronous, have commonly been controlled by applying pulse-width modulation (PWM) techniques to the output of a voltage source inverter (VSI). This prior art approach often requires a high DC link voltage between the rectifier and inverter sections of the VSI.

[0008] As an example, in many aerospace and other industrial applications, AC motors may be coupled to both single-phase and three-phase sinusoidal voltages, such as an AC generator feed, or directly to the output of a carrier rejection filter when using PWM to generate sinusoidal output currents from a VSI. In all such cases, the AC motor typically receives a high DC link voltage to generate the required sinusoidal currents for each phase.

[0009] During operation of a three-phase or polyphase asynchronous motor (e.g., induction motor) or synchronous motor (e.g., permanent magnet AC (PMAC) motor), the sinusoidal currents for each phase flowing in the stator coils are separated from each other in phase (e.g., by 120° for a three-phase motor) and generate a rotating magnetic field. This rotating magnetic field, or AC flux swing, then passes through the rotor windings and induces voltages in them. These induced voltages generate their own magnetic field due to the current flowing in the shorted rotor windings. In response, the rotor field attempts to align with the rotating stator field. Similarly, in the case of a PM motor, the rotor magnets attempt to align with the rotating stator field.

[0010] In both cases, these interactions produce a torque on the rotor that causes it to begin to rotate. For the rotor of an asynchronous induction motor to remain energized (i.e., producing torque) in the motoring mode, the rotational speed of the motor is always less than the synchronous speed of the stator excitation frequency. The difference between these speeds is known as the slip frequency in the motoring mode of operation.

[0011] As mentioned above, pulse-width modulated (PWM) drives or power controllers are commonly used to drive and control asynchronous and synchronous AC motors over a wide range of speed and load operating conditions. For traditional power controllers employing VSI, the DC supply voltage must be sized such that the inverter AC output line voltage is less than the DC supply voltage. The modulation strategy, whether it is space vector, third harmonic, or sinusoidal, determines the maximum DC voltage utilization.

[0012] With a PWM controlled drive, the motor current 12 (see Figure 1) per phase depends on the motor and line inductances and the "on-time" of the sinusoidal modulation cycle. This current is also called the "integral volt-second". For sinusoidal modulation, the "on-time" is a function of the following transfer function and the comparative relationship between a sine function and a trapezoidal ramp carrier signal: In particular, the high and low pulse on-times or pole voltage durations can be defined as:

number

number

number

[0013] For such a control scheme to be effective, the AC motor is rated at a unique voltage such that the time integral of the voltage (volt-seconds) (see reference number 12 in FIG. 1) guarantees a full cycle sinusoidal current swing that meets the AC motor requirements. In this regard, it should be noted that all AC motors are current-controlled devices.

[0014] Generally, prior art voltage source inverters work well but require high DC link supply voltages. Prior art voltage source inverters also have the drawbacks, design challenges, and long term reliability issues mentioned above. These issues include power stage switching losses and high stress on the motor insulation, high frequency losses, bearing currents, possible shoot-through causing reliability issues, EMI noise, and the need to use output sine wave filters to remove pole voltage swings when working with long line motor and controller systems.

[0015] Furthermore, during motor or generator mode operation of an asynchronous or synchronous motor, rotation of the rotor and currents injected into the polyphase windings of the armature induce rotating back EMF voltages in the stator and / or rotor armature.

[0016] At zero speed and during the initial starting condition, the AC motor draws much higher phase current due to the absence of this back EMF, and the AC motor momentarily sees the full supply voltage at starting. However, once the AC motor begins to rotate, the increased back EMF has the effect of reducing the voltage seen by the motor (due to the effect of the simultaneous operation of the motor and generator), thereby reducing the motor phase current to its normal rated value.

[0017] To continue normal operation and establish electrical balance where the back-EMF has a constant average value, the control system must pull down the line-potential to overcome the effects of the back-EMF. This process of power supply potential degradation due to back-EMF generation requires a significant input of energy, in the form of voltage, from the power supply to maintain normal operation.

[0018] The biggest obstacle of prior art motor drives, especially those involving three-phase voltage source inverter (VSI) topologies, is the need for large DC link voltages. Typically, AC induction motors are energized using either sinusoidal voltages or modulated switch node pole voltages, so the current is a factor of integral volt-seconds over the sinusoidal modulation window. Generating these large DC link voltages is a challenge in itself, especially when many motors require hundreds of volts and higher DC link voltages to generate the currents required for their unique torque requirements.

[0019] In this regard, the aerospace industry is currently considering using higher voltages in the range of 540VDC to 805VDC to drive lower phase currents because using higher voltages would result in lighter transmission cables. This trend toward higher DC voltages makes the use of correspondingly higher DC link voltages for motor control much more difficult.

[0020] A second challenge with prior art VSI systems is the use of switch-node pole voltage modulation, where the inverter pole node swings from ground to the full DC-link voltage, as in the case of a two-level modulation scheme, or at a reduced voltage level, as in the case of a multi-level configuration. Either way, this complicates the design from a modulation control perspective, where EMI and electromagnetic compatibility (EMC) are major considerations. Extremely large voltage swings with rise times of 100-300ns will almost certainly lead to common-mode noise emissions, along with the possibility of radiated emissions due to the switch-node modulation, inherent inductances in the VSI, parasitic capacitive elements, and cabling. Summary of the Invention [Means for solving the problem]

[0021] To overcome and mitigate the above-mentioned shortcomings of prior art motor control using voltage source inverters, the present disclosure presents a solution for the control of both synchronous and asynchronous AC motors that eliminates the VSI and PWM modulation techniques and uses directly coupled current sources to provide current control to the AC motor.

[0022] The features and advantages of the present disclosure will be more fully understood from the following more detailed description taken in conjunction with the accompanying drawings, in which: [Brief description of the drawings]

[0023] [Figure 1] An illustrative example square wave pole voltage modulation waveform 10 is shown in the top chart, as commonly used in the prior art, with the resulting integrated volt-seconds 12 (current) shown in the bottom chart. [Diagram 2] FIG. 1 illustrates an illustrative high level block diagram of a current source motor control technique in accordance with the present disclosure. [Diagram 3] FIG. 2 illustrates a circuit for an exemplary current amplifier that can be used as a linear current source in accordance with the present disclosure. [Figure 4]FIG. 2 illustrates a more detailed architecture diagram of one embodiment according to the present disclosure. [Diagram 5] FIG. 1 illustrates an example 120 degree phase shift clock logic for use in accordance with one embodiment of the present disclosure. [Figure 6] FIG. 1 illustrates an example voltage-to-frequency converter for use in accordance with an embodiment of the present disclosure. [Figure 7] FIG. 1 illustrates an example circuit embodiment for a variable frequency sine wave voltage generating circuit using a switched capacitor filter for use in accordance with an embodiment of the present disclosure. [Figure 8] FIG. 2 illustrates an example circuit for generating a variable amplitude sine wave voltage using an analog multiplier for use in accordance with an embodiment of the present disclosure. DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0024] As mentioned above, one of the challenges in designing aerospace equipment is to minimize electromagnetic interference (EMI). Prior art motor controllers based on voltage source inverter (VSI) technology use pulse width modulation (PWM) techniques to switch the DC link voltage using a sinusoidal modulation scheme. As a result, the integral volt-seconds of the motor inductance (or an in-line sine wave filter) generates a quasi-sinusoidal current with a ripple term, as shown in FIG. 1 at reference number 12.

[0025] This is the standard by which virtually all prior art VSI motor controllers operate, but among other drawbacks, PWM creates technical challenges for reducing EMI. This is especially true when highly sensitive equipment, such as RF systems, demands the highest immunity to EMI. Under these circumstances, for example, cooling fan motors can become a major source of EMI.

[0026] In all cases, a polyphase motor requires an AC current to be injected into its windings to generate a rotating magnetic field. From a conventional operating point of view, the motor current is the difference between the AC supply voltage and the back EMF voltage divided by the motor impedance. As explained above, according to the usual prior art methods, the AC current is provided by using a fixed AC voltage or by generating the AC voltage from a voltage source inverter with a high DC link voltage and using PWM techniques to provide a sinusoidal current. In the motor operating mode, the supply voltage must always be greater than the back EMF (BEMF) voltage.

[0027] According to this disclosure, a circuit topology is described in which the required sinusoidal currents are directly injected into each phase of a multi-phase stator winding, without PWM switching of a high DC link voltage.

[0028] This can be accomplished using a current source with sufficient voltage required to source the current, especially in motor starting where the current needs to be sized to overcome the impedance of the windings. Such a current source topology can operate without the need for high DC voltages, so that, for example, sinusoidal PWM generates sinusoidal AC current in integral volt-seconds per phase.

[0029] According to the present disclosure, the current source is not limited to a particular type, for example, in one embodiment, the current source according to the present disclosure may be a current source type synchronous down converter, also known in the art as a buck converter.

[0030] Although a buck converter provides a highly efficient approach, many other current source topologies can be used, including, as another example, a linear current amplifier as shown in Figure 3. The selection of a particular current source can depend on the type of application and the motor current requirements.

[0031] In accordance with the present disclosure, each phase of a multi-phase current source is provided with a variable amplitude and variable frequency sinusoidal voltage reference that represents both the amplitude and frequency of the current that is to be supplied to each phase of a multi-phase motor. For control purposes, one of these phases is designated as a master phase reference, as described further herein.

[0032] FIG. 2 shows a high level block diagram of the current source control topology disclosed herein for driving a three-phase AC motor. With reference thereto, both the amplitude and frequency of the current reference per phase are controlled by a voltage ramp generator 20, whose voltage starts at zero and rises with a controlled slope until it reaches a maximum value. The controlled slope may start at a low level and then increase as the three-phase AC motor starts and approaches its rated speed. As is known in the art, the voltage ramp may be generated by an integrator circuit, by a digital-to-analog converter, or by other means. The maximum voltage value that the ramp reaches is designed to correspond to the maximum motor current (per phase) and the maximum or nominal fundamental frequency. By way of example, this may be 10A at 400Hz.

[0033] 2, the voltage output of the ramp generator 20 is input to a voltage-to-frequency converter (V2F) 22, which is also synchronized to the output of a master clock 24 to reduce jitter in its frequency output. The output of the master clock 24 is also input to a phase-locked loop (PLL) 26, which receives the output of the voltage-to-frequency converter 22 via a buffer amplifier 23.

[0034] 2, a measure of the zero crossings of the back EMF (BEMF) 28 manifested at the output of a master current source (arbitrarily selected from one of three AC current sources A, B, and C that drive the AC motor, referenced by the numerals 32, 34, and 36, respectively) is provided by a comparator 50 and also input to the PLL 26. The PLL 26 thus provides a frequency output to the control logic 30 whose phase is locked to the zero crossings of the BEMF. The purpose of ultimately locking the phases of the currents output from the three AC current sources 32, 34, 36 to the zero crossings of the motor's BEMF is to increase the output power factor.

[0035] For clarity, phase A may be arbitrarily selected as the master phase. With a single BEMF master phase, all three current source references are phase shifted by 120 degrees, so specifying the corresponding master phase current sources ensures that the currents per phase are correctly phase locked to the BEMF.

[0036] From an incremental control standpoint, the motor BEMF is phase shifted from the applied voltage by the phase impedance. When the master phase current is progressively phase adjusted relative to the master BEMF using PLL control, this implies that the transfer of real power to the motor, i.e., the power factor, is maximized. (Reactive power is due to the non-zero power factor angle resulting from the phase impedance.)

[0037] Optionally, to reduce the magnitude of the BEMF applied to the PLL 26, a back EMF suppression network 52 may be added across each phase of the multi-phase winding to absorb most of the motor's BEMF.

[0038] 2, the control logic 30, in response to receiving inputs from the phase-locked loop 26 and the V2F 22, generates three sinusoidal voltage references (each phase-shifted 120° relative to one another) that drive respective AC current sources 32, 34, and 36 according to the frequency output from the V2F 22 to energize respective stator windings 38 of the three-phase AC motor. As shown, a common mode filter 401 is provided between the output of the AC current sources and the stator windings 38 to provide high impedance rejection of external common mode conducted emissions, thereby protecting the controller. As further shown, respective current loops 40, 42, 43 at the output of the AC current sources 32, 34, 36 provide feedback signals to the control logic 30 via respective buffers 44, 45, and 46. These feedback signals are used by the control logic 30 to verify that the frequencies of the three sinusoidal voltage references at the inputs to the AC current sources 32, 34, 36 are correct and that the sum of the three phase currents is zero, thereby providing a balanced drive to the three-phase motor and ensuring that all three phases are driven equally.

[0039] As will be explained further below with respect to FIG. 4 and for an exemplary embodiment in which the motor is an AC induction motor (ACIM), the current in each phase of the polyphase windings may also be monitored by such feedback signals and used for slip control purposes, as in the case of a conventional scalar control structure under V / F control. Thus, if the measured RMS motor current begins to increase above the maximum nominal current due to an increase in load torque and the motor RPM subsequently drops below the nominal fundamental frequency, the control logic 30 performs slip compensation, as is known in the art of AC induction motors, to correct the imbalance by adjusting the slip frequency. As is also understood in the art of AC motors, slip control is not required or important when controlling a synchronous AC motor.

[0040] Specifically, under these conditions, slip compensation adjusts the operating trajectory such that the magnitude of the current and frequency supplied to the AC motor are both increased, thereby gradually increasing the motor RPM back up to the maximum nominal range. At this point, the motor current for each phase of the polyphase winding is finally within the normal operating range, and the motor RPM reflects the relationship between the current excitation frequency and the number of motor poles. This slip compensation mechanism is always in place to maintain motor slip under conditions of load uncertainty.

[0041] As is evident from the above description of FIG. 2, one purpose of the control logic 30 is to maintain phase alignment between the sinusoidal current applied to the master phase winding of the polyphase winding and the master back EMF. This is accomplished by using the PLL 26 to synchronize the phase of the sinusoidal voltage reference used to control the sinusoidal current to the zero crossings of the back EMF. The phase alignment of the current to the BEMF voltage drives a higher power factor. In this regard, it should be stated that this motor controller operates, in conventional terms, as a scalar controller, where the ratio of both the applied voltage V and the frequency F is maintained.

[0042] This V / F ratio (e.g., 115VAC / 400Hz=0.287) maintains constant torque as the motor is soft started and voltage ramped up to its base operating frequency. Because the process of current injection is not based on integral volt-seconds applied to the motor inductance by PWM, the current can be adjusted in magnitude and frequency in a much shorter time than a typical VSI drive, so the controller disclosed herein exhibits a much faster dynamic response than conventional controllers. This drives increased bandwidth when handling fast load changes.

[0043] Since the motor currents are directly injected as sinusoidal currents per phase, the V / F relationship needs to be maintained to maintain constant torque over a full frequency range and up to the fundamental frequency. This implies that the currents must be controlled in both amplitude and frequency at each of the current sources per phase. The controller disclosed herein is also capable of performing field weakening by increasing the frequency of the motor currents above the fundamental frequency in a constant power operating mode.

[0044] A more detailed architecture overview showing further details of the control logic 30 is provided in FIG. 4 and is further described below, although it will be understood that in various embodiments the current source architecture disclosed herein may be implemented by discrete electronic circuit blocks or components (i.e., without software or firmware) or, alternatively, by including software or firmware to reduce hardware complexity using a microcontroller, microprocessor, or the like.

[0045] Referring to Figure 4, the same reference numbers used in Figure 2 have been reused in Figure 4 to designate similar components. Furthermore, for ease of explanation, the V2F converter 22 and PLL 26 are shown generally within a block labeled 30 in Figure 4, which corresponds to the control logic 30 in Figure 2. The important components and functions of this control logic block 30 are further detailed in Figure 4 and described below.

[0046] A. Synchronous Phase-Shifted Sine Wave Voltage Generation From a current source control point of view, the current controller requires three sinusoidal voltage references that can provide both variable amplitude and variable frequency. For a three-phase AC motor, the phase displacement of these voltage references relative to each other must be 120 degrees. In the embodiment of FIG. 4, these three phase-shifted voltage references are generated by a phase-delay clock generator in a three-phase clock (CLK) logic block 60, which in an exemplary embodiment may use the circuit shown in FIG. 5.

[0047] Referring to FIG. 5, this circuit receives the frequency output from the voltage to frequency converter 22 (see FIG. 4) which can be divided by a ratio (n) to give a frequency at the output of the binary counter 54 which is six times the maximum excitation frequency of the AC motor. As an example, for a 400 Hz motor this would be 2400 Hz. By incorporating three D flip-flops 55, 56, 57 using synchronous clocks and cascaded QD ripple-through inputs, the circuit of FIG. 5 is able to further divide this frequency by six, thereby giving three 120 degree phase shifted clocks at a maximum frequency of 400 Hz at the respective outputs of buffers 58, 59, and 61. As shown in FIG. 4, these outputs from the 3-phase CLK logic block 60 are applied to respective switched capacitor filters 62, 64, and 66 which generate the required sinusoidal waveforms.

[0048] In this regard, the voltage frequency converter V2F22, whose variable frequency output is applied to the input of the three-phase CLK logic block 60, may incorporate a synchronous clock to minimize jitter. A circuit that may be used for an exemplary voltage frequency converter is shown in FIG. 6. In brief, as shown in FIG. 6, the voltage ramp input 601 used for frequency conversion is constrained using an amplitude limit 600. This ensures that the output frequency of V2F is limited to the upper frequency of the motor (e.g., 400 Hz). The ramp is then band-limited using a two-pole active low-pass filter 602, e.g., a Bessel filter with a Q value of 0.577. The input filtering, together with the synchronous clock 604, gives V2F22 a jitter-free output frequency.

[0049] Referring again to the block diagram of Figure 4, application of an 8-bit binary count (from 0 (0x00) to 8-bit (0xFF)) to a Digital-to-Analog Converter (DAC) can be used to generate a voltage ramp output from block 20 to drive V2F 22. The ramp voltage output from block 20 is used by downstream circuitry in Figure 4 to provide two different ramp rates.

[0050] In particular, as shown in FIG. 4, a first ramp is applied directly to control the voltage-to-frequency converter 22, and the resulting frequency is used to control the output frequency of the current source voltage reference generated by the three-phase CLK logic 60 and ultimately the frequency at the output of each of the SC filters 62, 64, and 66.

[0051] A second ramp having a different ramp rate is then derived from the first ramp and used to control the amplitude of the current source voltage reference output from the SC filters 62, 64, 66. Such amplitude control may be achieved by using a voltage multiplier circuit as shown in FIG. 8 which receives the output from the SC filters 62, 64, 66 and provides amplitude control of the three-phase sinusoidal voltage reference. This is achieved by multiplying the output of the SC filters 62, 64, 66 by the second ramp using respective voltage multipliers 800, 802, 804 to linearly increase the gain of the sinusoidal voltage reference. As will be apparent, the two ramps are scaled accordingly and have different slopes to perform their particular circuit functions.

[0052] 4, generation of the sinusoidal current source voltage references applied to the inputs of each of the AC current sources 32, 34, and 36 that power each of the multi-phase windings can be accomplished as follows: A voltage to frequency (V2F) converter 22 provides variable frequency, phase shifted, square wave clock signals per phase that are synchronized to a higher frequency clock and output respectively from 3-phase CLK Logic 60.

[0053] As shown in FIG. 4 and in more detail in FIG. 7, in one embodiment, these three square wave signals for each phase are then passed through respective 10th order elliptic switched capacitor filters (10 thA corner frequency of 100 Hz is added to the square wave SC filter (SC filter 62, 64, 66). (The ratio of the corner frequency to the fundamental sine wave frequency output from the 3-phase CLK logic 60 may be set to 96.) To eliminate clock frequency feedthrough, the outputs of the SC filters 62, 64, 66 are passed to low pass analog SINC correction filters 702, 704, 706. These SINC correction low pass filters attenuate the harmonics in the square wave SC filter clock to a level where they can be ignored, thereby producing a phase shifted sinusoidal waveform at the respective outputs of the SINC correction low pass filters. This attenuation is typically -60 dB at the clock frequency. For a 400 Hz motor frequency, this would be a clock frequency of 38.4 kHz for clarity. Since both the fundamental frequency clock and the corner frequency clock are synchronous and vary with the ramp reference, the SINC correction LP filters 702, 704, and 706 generate a tracking sine wave output that is variable in frequency (e.g., from 0 Hz to 400 Hz fundamental) to drive an AC motor. This is because the SC filter tracks a range of frequencies, so the input frequency to the SC filter is also the same as the corner frequency of the filter. The square wave from the 3-phase CLK logic 60 is presented to the SC filters 62, 64, and 66, and then to the SINC correction LP filters 702, 704, and 706, so that the higher harmonics are attenuated, leaving only the properly phase-shifted sine wave fundamental to be output from the respective filters 702, 704, and 706.

[0054] B. Input Voltage Monitoring and Verification According to the present disclosure, another function of the control logic 30 is to monitor and verify the DC voltages powering each of the AC current sources per phase. If these voltages fall outside predetermined limits, the monitor will inhibit motor operation or initiate a power down soft stop process if the motor is running. This function may be implemented by a comparator that monitors the DC input voltage used to power the AC current sources with an input deglitch filter.

[0055] C. Phase-by-phase synchronous current source converter As previously explained, each of the current sources per phase may be configured, for example, as a linear regulator or as a synchronous current source down converter (i.e., buck converter) or equivalent AC current source. For AC induction motor applications, such as three-phase cooling fans, linear current sources may be used due to the relatively low phase currents and simplicity of circuit function. For motors requiring higher current operation, current source down converters will generally be more efficient than linear regulators, since they use pulse width modulation for control purposes.

[0056] According to this disclosure, the control objective in all cases is to provide sinusoidal currents per phase to each of the stator windings. The current converters must also be capable of sourcing full-rate sinusoidal phase currents, which is based on the control objectives of torque (current) and frequency. Although this disclosure describes exemplary embodiments using linear current sources as well as current source type down converters (buck converters), AC current sources with suitable characteristics based on other circuit topologies may alternatively be used as the synchronous current sources disclosed herein and are encompassed by the scope of this disclosure.

[0057] D. Phase current feedback As shown in Figure 4, current transformers (e.g., current loops 40, 42, and 43) are connected to the output of each of the synchronous current source regulators for each phase. These are used to generate voltage analogs of the phase currents for current feedback control purposes. Being sinusoidal current source type drives, the line frequency of the current is typically a few hundred Hertz. Therefore, with scalar V / F control, instantaneous sensing and correction of the current is not necessary and an RMS sensing method is sufficient.

[0058] E. Phase current RMS verification and slip compensation Each of the sinusoidal currents per phase is used in a closed loop control structure to ensure that the stator current per phase meets the commanded current. The highest of the three phase currents, monitored by the current loops 40, 42, 43, is converted to RMS terms in the SUM / High Wins block 68. The slip control block 70 uses the highest of the three RMS currents to detect motor drag or other conditions where the load torque is higher than normal (resulting in a lower motor RPM than normal operation). The motor RPM can be detected by using an inductive pickup in the motor or by measuring the back EMF voltage. In this situation, the slip control block 70 provides a signal that is fed back to the input of the AC current sources via multipliers 74, 76 and 78 and to the V2F 22. This feedback causes the current sources per phase to make a small increase in excitation frequency and output current to compensate for the increased load torque and pull the motor back to normal operating conditions.

[0059] In particular, as the excitation frequency increases, the motor RPM also increases progressively, and the slip term between electrical frequency and rotor RPM is reduced to a level where a nominal slip of 2-3% is appropriate and the motor current is once again within normal limits. Once the effects of the high load on the motor are removed, the slip compensator progressively reduces the electrical frequency applied to each of the stator windings per phase so that the motor operates at the correct fundamental frequency and within normal RPM limits. This mode of slip control is a low bandwidth compensator and operates in a manner that does not introduce abrupt transient changes in RPM. Slip control regulation is performed with maximum limits imposed on both frequency and current imposed by limit block 72.

[0060] The control logic 30 also uses the measured AC current of each phase to compensate for phase current imbalances and ensure that each phase is driven equally. This is accomplished by taking the sum of all three phases in the SUM / High Wins block 68. A sum close to zero indicates balanced current drive, which can be detected by a window comparator. This implies that the sum of all three phases should equal zero.

[0061] F. Motor Tachometer Feedback The motor RPM can be determined by tracking the back EMF voltage or by using a Hall sensor as a tachometer. This information is used to inform the centralized controller so the slip compensator can scale the correct trajectory of compensation in both current and frequency based on the motor applied electrical frequency and the actual motor RPM.

[0062] G. Motor Position Synchronization Using PLL for Optimal Power Factor It is important to ensure that the motor BEMF is phase matched (i.e., in phase) with the sourced currents so that the power factor can be optimized by the control logic 30. This is achieved by synchronous control of the current sources for the BEMF per phase and using a phase-locked loop 26 to progressively phase match the motor BEMF to the phase currents.

[0063] H. Variable frequency, variable voltage sinusoidal current command The control logic 30 generates a voltage analog of the current command, which is variable amplitude and frequency. The current command to each of the AC current sources is the output of the V2F22 and a sinusoidal voltage regulation scheme. This is also augmented by a slip controller which adjusts the current to the motors as needed. All three commands per phase are phase shifted by 120 degrees. This is important when considering starting high inertia loads and reacting to fast changes in load torque.

[0064] I. Phase Locked Loop (PLL) As described herein, the phase-locked loop 26 is used to phase-lock the BEMF with the phase of the master current control reference. This ensures that the motor current frequency per phase is aligned to the BEMF. Since each of the current references (sinusoidal voltages) per phase are phase-shifted simultaneously, using a dedicated phase to lock the PLL ensures that all phases are aligned with respect to their own BEMF for the applied phase current.

[0065] As disclosed herein, a current source controller topology performs voltage frequency control, slip control, and torque control of an AC motor. The specific implementation of such a controller may be accomplished by incorporating a digital signal processor (DSP) or by implementation using separate analog and / or digital solutions.

[0066] The following significant advantages can be obtained by using the novel current source controller architecture disclosed herein compared to older prior art VSI topologies:

[0067] Importantly, the current source approach disclosed herein results in a reduction in the VSI and the high voltage DC bus required to provide the high link voltage to the VSI. Thus, the DC link voltage per current source only needs to be sized to withstand (i.e., source) the maximum current required for sufficient torque versus frequency.

[0068] The reduction in VSI eliminates the need to employ pulse width modulation of the pole voltage, which reduces the complex PWM control logic required for the sinusoidal output filter.

[0069] Because the current source type control disclosed herein focuses on controlling current, it is capable of driving fast changes in current and frequency, which can be used to manage load uncertainties and increase torque.

[0070] Because the topology disclosed herein is based on a current source type motor drive, it provides an excellent design for so-called "long line" motor and controller applications where there are significant wire lengths between the motor and the controller. In particular, current source controllers are long line capable and can be used without the EMI issues that arise with standard VSI topologies that use switch node modulation of the voltage.

[0071] Another advantage is that this type of controller can also be used to control single-phase AC motors by using a single-phase section and a controller with reduced complexity.

[0072] There are numerous applications for this current source control scheme, including, but not limited to, control of both synchronous and asynchronous three-phase motors, control of single-phase AC motors, pump controllers using AC induction motors, long line AC motor applications, control of PMAC motors, and three-phase AC fans and blowers, to name just a few applications.

[0073] Furthermore, the efficiency gains that can be achieved using the disclosed current source topology can be significant, especially when optimally designed synchronously controlled down converters incorporating SiC technology are used.

[0074] In addition to the advantages mentioned above, strategic gains for this topology include faster time to market design and implementation, reduced form factor and weight, reduced system complexity, and reduced non-recurring costs.

[0075] Briefly, the above disclosure describes an embodiment of control for a three-phase motor using a current source type control topology. Such a current source type topology relies on pole voltage modulation, is EMC quiet, is long line capable, provides control with improved efficiency over the prior art, and eliminates problems caused by stand-alone prior art VSI circuit topologies.

[0076] While illustrative embodiments of the present disclosure have been shown and described in detail, various modifications and improvements thereon will become readily apparent to those skilled in the art, all of which are intended to be encompassed by the following claims.

Claims

1. A controller for a polyphase AC motor having several windings, comprising: a plurality of AC current sources, the number of AC current sources equal to the number of windings, each AC current source being coupled to a different winding to provide a sinusoidal current to said different winding without pulse width modulation switching of the DC link voltage; control logic coupled to the plurality of AC current sources, the control logic generating a common variable frequency sinusoidal voltage reference at a respective input of each of the AC current sources, each of the AC current sources driving a connected winding of each of the AC current sources with a sinusoidal AC current of a specified phase and variable amplitude of the common variable frequency; a voltage-to-frequency converter coupled to the control logic and receiving as an input a variable lamp voltage; A controller comprising:

2. The controller of claim 1 , wherein the plurality of AC current sources comprises linear current amplifiers.

3. 2. The controller of claim 1, wherein the control logic comprises: clock logic and a switched capacitor filter for generating the sinusoidal voltage reference.

4. 4. The controller of claim 3, wherein a voltage-to-frequency converter is connected to an input of the clock logic to generate a plurality of clock signals having different phases.

5. The controller of claim 4 , wherein the clock signal is input to the switched capacitor filter to generate the sinusoidal voltage reference.

6. 2. The controller of claim 1, further comprising a phase-locked loop for locking a phase of an induced voltage generated by the AC motor to a phase of the sinusoidal AC current output from one of the plurality of AC current sources.

7. 1. A controller for a three-phase AC induction motor having three stator windings, comprising: three AC current sources, each coupled to a different one of the stator windings to supply a sinusoidal current to the different one of the stator windings without pulse width modulation switching of a DC link voltage; control logic coupled to the three AC current sources, the control logic generating a common variable frequency sinusoidal voltage reference coupled to a respective input of each AC current source, thereby controlling each AC current source to drive its associated winding with a sinusoidal AC current of specified phase and variable amplitude at the common variable frequency; a voltage-to-frequency converter coupled to the control logic and receiving as an input a variable lamp voltage; A controller comprising:

8. The controller of claim 7 , wherein the AC current source comprises a linear current amplifier.

9. 8. The controller of claim 7, wherein the control logic comprises: clock logic and a switched capacitor filter for generating the sinusoidal voltage reference.

10. 10. The controller of claim 9, wherein a voltage-to-frequency converter is coupled to an input of the clock logic to generate a plurality of clock signals having phase shifted 120 degrees relative to one another.

11. The controller of claim 10 , wherein the phase shifted clock signal is input to the switched capacitor filter to generate the sinusoidal voltage reference.

12. 11. The controller of claim 10, further comprising a phase-locked loop for locking a phase of an induced voltage of the motor to a phase of the sinusoidal current output from one of the plurality of AC current sources.

13. 1. A method for controlling a three-phase AC motor, the three-phase AC motor having three windings, coupling an AC current source to each winding to supply a sinusoidal current to the winding, the amplitude and frequency of each current source being controlled by a voltage ramp generator; generating sinusoidal voltage references of different voltage levels and phases; coupling the sinusoidal voltage reference to each of the AC current sources to drive each of the current sources to output a variable current at a specific phase and at a variable frequency to control operation of the three-phase motor; The method includes:

14. The method of claim 13 , wherein the step of coupling an AC current source further comprises the step of coupling a linear current amplifier.

15. 14. The method of claim 13, wherein the generating step further comprises using clock logic and a switched capacitor filter to generate the sinusoidal voltage reference.

16. 16. The method of claim 15, wherein the generating step further comprises using a voltage-to-frequency converter to generate a clock having a different phase than the clock logic.

17. 14. The method of claim 13, wherein an induced voltage generated by the three-phase motor is phase locked to one of the currents output from the current source.

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