Controllable transfer networks for high frequency power conversion.

Abstract

In one aspect, a system for diverting a radio frequency (RF) signal between a source and a load may include a first port and a second port connected to the source and the load, respectively; a first filter connected to the first port; a second filter connected to the second port, where at least one of the first filter and the second filter is operable to provide a variable reactance using dynamic frequency tuning (DFT); a two-port switching network connected between the first filter and the second filter, the switching network comprising a plurality of switches; and a controller coupled to the switching network and the source, the controller configured to operate the plurality of switches according to a switching pattern to dynamically adjust a frequency of the RF signal to provide a controllable impedance matching between the first port and the second port.

Description

[Technical field]

[0001] CROSS-REFERENCE TO RELATED APPLICATIONS

[0001] This application claims the benefit under 35 U.S.C. § 119 of U.S. Provisional Patent Application No. 63 / 289,809, filed December 15, 2021, and is incorporated by reference in its entirety into this specification. [Background technology]

[0002]

[0002] Many semiconductor processing applications, such as etching, require the delivery of radio frequency (RF) power to a plasma load. To efficiently and effectively deliver the required power from the RF amplifier, the variable impedance of the plasma load must be dynamically matched to the impedance required by the RF amplifier or RF generator. Furthermore, to achieve the desired functionality, this power delivery must be precisely controllable over a wide power range with a wide bandwidth (e.g., controllable for large, rapid steps in commanded power), even when the plasma's load impedance varies rapidly. Finally, the need for precise plasma control over large spatial regions calls for an RF power delivery system that can separately control RF power delivery to multiple outputs (e.g., of multiple spatially distributed applicators).

[0003] <Variable impedance matching>

[0003] To facilitate effective use of RF power amplifiers / generators / inverters and to facilitate power control to the load, it is often necessary to provide a dynamically adjustable impedance match between the RF source and the plasma load. Maintaining a well-controlled impedance match in such systems can be important to maintain high efficiency of the RF power amplifier, enable desired power delivery to the load, and ensure precision in controlling the RF power. This is typically accomplished using a Tunable Matching Network (TMN), sometimes referred to as a "Tunable Impedance Matching Network," "Automatic Matching Unit" (AMU), or "Antenna Tuning Unit" (ATU), which can sense and dynamically adjust the impedance match between the RF source and the load over a specified operating range.

[0004] FIG. 1 shows an impedance Z S and a source 102 having an impedance Z L 1 illustrates an RF system 100 including a TMN 106 coupled between a source 102 and a load 104 having a impedance of 100 MHz. A controller 108, which may be implemented using digital and / or analog circuitry, is coupled to provide control signals to the TMN 106 for controlling operation of the TMN 106. The TMN 106 has a desired impedance transformation characteristic in response to such control signals. That is, the TMN 106 may be controlled to adjust, tune, change, or otherwise manipulate the impedance presented by the TMN 106 to the source 102 and / or load 104. The TMN 106 may, for example, transform a desired impedance Z from the source 102 to the TMN 106. S,IN , and the desired impedance Z from the load 104 to the TMN 106 L,IN can be controlled to show

[0005]

[0005] The source 102, the TMN 106, the controller 108, and other elements of the RF system 100 may be coupled to a supply voltage (e.g., VDC) and ground (not shown). The controller 108 may control the operation of the TMN 106 based at least in part on information received from an optional feedforward circuit (not shown) coupled to the source 102 and / or an optional feedback circuit (not shown) coupled to the load 104. The feedforward information may include information regarding the effective input impedance of the TMN 106, the timing of the RF waveform, specified signal levels and / or impedance levels, etc. The feedback information may include measurement information regarding the effective load impedance and / or power reflected from the load, the timing of the RF waveform, etc.

[0006]

[0006] One method of achieving dynamic impedance matching includes matching networks (such as L-networks, T-networks, or π-networks) in which the passive component values ​​can be dynamically adjusted. FIG. 1A shows an exemplary embodiment of an ideally lossless lumped-element TMN 120 based on a step-up L-matching topology that matches the impedance of a load 124 to the impedance of a source (input) 122. The embodiment of the TMN 120 includes tunable elements 126a, 126b that can perform a "step-up" impedance matching. The range of impedances that can be matched with such techniques is shown in FIG. 1B using a Smith chart 140. The tunable elements 126a, 126b can be provided as a variety of different tuning mechanisms, such as mechanically driven tunable capacitors, switched capacitor and / or inductor banks, varactors (where the bias voltage of the varactor is adjusted to adjust the capacitance), and phase-switched impedance modulation (PSIM), in which a passive energy storage element is switched at RF frequencies to adjust the effective impedance. Various such techniques may be characterized based on the range of load impedances that can be matched, the resolution with which the match can be performed, and the speed at which the desired impedance match can be achieved (e.g., limited by the tuning speed of the components).

[0007] There is a need for improved impedance control techniques and structures for use in applications requiring fast response over a wide range of power and load impedances, such as plasma applications.

[0008] <High-speed power control> Another challenge is to accurately control RF power during large power steps, such as may occur in plasma "pulsing" used in modern RF plasma processing. During large power steps, significant short-term (e.g., tens of microseconds) RF power overshoots may occur, which may have detrimental effects in semiconductor processing applications. This may be addressed by adapting the impedance match fast enough, but short-term control and limiting of RF power delivery on very fast (e.g., 10 microsecond) time scales may be considered a phenomenon worth addressing in itself. One possibility to address this is the development of RF power amplifiers or RF generators with a wide enough bandwidth of power control capability in the face of large command steps and large load impedance changes.

[0009] <Multiple output power distribution control>

[0009] A further area of ​​interest is the development of RF power delivery systems that can separately control RF power for multiple outputs (possibly from a single RF power source). Multiple outputs can allow for spatial control of the plasma and tuning of uniformity. While multiple "parallel" delivery systems that can address mutual load pull effects are one possibility, it would also be desirable to develop systems that are scalable to multiple outputs (e.g., two, three, or even more outputs) from a single RF generator / power amplifier and / or a single RF source and matching system. Summary of the Invention

[0010] <Controllable conversion network>

[0010] Improved RF power conversion structures and techniques are described herein to address the problems discussed above. The key approach introduced here is the use of a two-port switching network that can achieve a controllable voltage transfer from an RF input to an RF output. The switching network operates synchronously with the RF signal to be transferred and can act approximately as a controllable transformer. The switching network can act to have an effective turns ratio that is dynamically variable, i.e., depending on the timing of the switching relative to the RF signal.

[0011]

[0011] According to one aspect of the present disclosure, a system for diverting radio frequency (RF) signals between a source and a load may include a first port and a second port connected to the source and the load, respectively; a first filter connected to the first port; a second filter connected to the second port, where at least one of the first filter and the second filter is operable to provide a variable reactance using dynamic frequency tuning (DFT); a two-port switching network connected between the first filter and the second filter, the switching network comprising a plurality of switches; and a controller coupled to the switching network and the source, the controller configured to operate the plurality of switches according to a switching pattern to dynamically adjust a frequency of the RF signal to provide a controllable impedance matching between the first port and the second port.

[0012]

[0012] In some embodiments, the first filter may be configured to extract a fundamental voltage of the RF signal. In some embodiments, the first filter may comprise a capacitor and an inductor connected in parallel across the terminals of the first port. In some embodiments, the second filter may be configured to extract a fundamental current of the RF signal. In some embodiments, the second filter may comprise a capacitor and an inductor connected in series to one terminal of the second port. In some embodiments, the controller may operate a plurality of switches in synchronism with the RF signal and at a controllable phase shift relative to the RF signal. In some embodiments, the controller may operate two or more switches according to a switching pattern defined by an angle β relative to the RF signal. In some embodiments, the controller may operate a load resistor R L and reactance X L In some embodiments, the controller may be configured to calculate β by L and reactance X L The switching frequency may be selected by:

[0013] In some embodiments, the load may include a plasma load. In some embodiments, the controller may be configured to provide a controllable impedance match between the first port and the second port during a power step of at least 2x that occurs in 10 microseconds or less.

[0014] In some embodiments, the plurality of switches may include four switches arranged in a bridge circuit. In some embodiments, the four switches may include unidirectional blocking, bidirectional pass switches.

[0015]

[0015] In some embodiments, the four switches may include a first switch having a first terminal connected to the first terminal of the first filter and a second terminal connected to the first terminal of the second filter; a second switch having a first terminal connected to the second terminal of the first filter and a second terminal connected to the second terminal of the second filter; a third switch having a first terminal connected to the second terminal of the first switch and to the first terminal of the second filter; and a fourth switch having a first terminal connected to the second terminal of the second switch and to the second terminal of the second filter and having a second terminal connected to the second terminal of the third switch.

[0016]

[0016] In some embodiments, the four switches may include a first switch having a first terminal connected to the first terminal of the first filter and a second terminal connected to the first terminal of the second filter; a second switch having a first terminal connected to the first terminal of the second filter and to the second terminal of the first switch and having a second terminal connected to the second terminal of the first filter; a third switch having a first terminal connected to the second terminal of the first filter and to the second terminal of the second switch and having a second terminal connected to the second terminal of the second filter; and a fourth switch having a first terminal connected to the second terminal of the second filter and to the second terminal of the third switch and having a second terminal connected to the third terminal of the first filter.

[0017]

[0017] According to another aspect of the present disclosure, a controllable diversion network for diverts a radio frequency (RF) signal between a source and a load may include a first port and a second port connected to the source and the load, respectively, a first filter connected to the first port and configured to extract a fundamental voltage of the RF signal, a second filter connected to the second port and configured to extract a fundamental current of the RF signal, where at least one of the first filter and the second filter is operable to provide a variable reactance using dynamic frequency tuning (DFT), and a switching network connected between the first filter and the second filter, the switching network comprising a bridge circuit having four switches.

[0018] In some embodiments, the first filter may comprise a capacitor and an inductor connected in parallel across the terminals of the first port. In some embodiments, the second filter may comprise a capacitor and an inductor connected in series to one terminal of the second port. In some embodiments, the four switches may be connected in the manner described above.

[0019]

[0019] It is to be understood that individual elements of the various embodiments described herein may be combined to form other embodiments not specifically described above. Various elements that are described in the context of a single embodiment may be provided separately or in any suitable subcombination. It is also to be understood that other embodiments not specifically described herein are within the scope of the appended claims.

[0020]

[0020] The manner of making and using the disclosed subject matter may be understood by reference to the detailed description in conjunction with the drawings, in which like reference numerals identify like elements. [Brief description of the drawings]

[0021] [Figure 1] FIG. 1 is a block diagram of a prior art RF system with a variable impedance matching network (TMN). [Figure 1A]

[0022] FIG. 1 is a circuit diagram showing an exemplary implementation of an ideally lossless lumped element TMN based on a step-up L-matching topology. [Figure 1B]

[0023] 1B is a Smith chart showing the range of load impedances that can be matched to a desired 50 ohm input impedance using a TMN such as that shown in FIG. 1A. [Diagram 2]

[0024] FIG. 1 is a block diagram of an RF system including a controllable transformation network (CTN) according to an embodiment of the present disclosure. [Figure 2A]

[0025] FIG. 1 is a circuit diagram illustrating how a CTN can be implemented using a two-port switching network and effective variable reactance acting as a controllable transformer, according to some embodiments. [Figure 2B] FIG. 1 is a circuit diagram illustrating how a CTN can be implemented using a two-port switching network and effective variable reactance acting as a controllable transformer, according to some embodiments. [Diagram 3]

[0026] A block diagram showing a CTN architecture including an input filter, a switching network, and an output filter according to some embodiments. [Figure 3A]

[0027] FIG. 4 is a circuit diagram illustrating an implementation of the CTN architecture of FIG. 3 according to some embodiments. [Figure 3B] FIG. 4 is a circuit diagram illustrating an implementation of the CTN architecture of FIG. 3 according to some embodiments. [Figure 4]

[0028] FIG. 3B is a circuit diagram showing the circuit of FIG. 3A connected to an RF source and a load. [Figure 4A]

[0029] FIG. 5 illustrates switching patterns and associated waveforms that may be used to control a CTN circuit such as that of FIG. 4 in accordance with some embodiments. [Figure 5A]

[0030] FIG. 4 is a circuit diagram illustrating another implementation of the CTN architecture of FIG. 3 in accordance with some embodiments. [Figure 5B] FIG. 4 is a circuit diagram illustrating another implementation of the CTN architecture of FIG. 3 in accordance with some embodiments. [Figure 6]

[0031] 5A-5C are diagrams illustrating representative waveforms that may result from switching in a CTN circuit such as FIG. 4, FIG. 5A, or FIG. 5B in accordance with the present disclosure. [Figure 7]

[0032] FIG. 5B illustrates an ideal control angle curve that may be generated by a CTN circuit such as that of FIG. 4, FIG. 5A, or FIG. 5B in accordance with some embodiments. [Figure 8]

[0033] 8A and 8B are diagrams illustrating other switching patterns and associated waveforms that may be used in a CTN circuit such as FIG. 4, FIG. 5A, or FIG. 5B, according to some embodiments. [Figure 9]

[0034] FIG. 2 is a circuit diagram illustrating a multiple-output CTN according to some embodiments. [Figure 10]

[0035] FIG. 1 is a block diagram of a multiple output RF power transfer system with combined matching and RF power distribution in accordance with some embodiments. [Figure 11]

[0036] FIG. 1 is a circuit diagram of a controllable transformation power amplifier (CTPA) according to some embodiments. [Figure 11A]

[0037] 12A-12C are diagrams illustrating exemplary switching patterns that may be used in the CTPA of FIG. 11 according to some embodiments. [Figure 12]

[0038] 1 is a circuit diagram of another CTPA, according to some embodiments. [Figure 12A]

[0039] 13A-13C are diagrams illustrating example switching patterns that may be used in the CTPA of FIG. 12 in accordance with some embodiments. DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0022]

[0040] The drawings are not necessarily to scale, and do not include all elements of the system, with emphasis generally being placed upon illustrating the concepts, structures, and techniques sought to be protected herein.

[0023] <Controllable conversion network>

[0041] Figure 2 shows the impedance Z S and an RF source 202 having an impedance Z L 2 illustrates an RF system 200 with a controllable transformation network (CTN) 206 that may be used to provide a variable impedance match between a load 204 having an impedance Z 1 , and a source 202 having a load 204 having a impedance Z 2 . A controller 208, which may be implemented using digital and / or analog circuitry, is coupled to provide control signals 210 to the CTN 206 to control the operation of the CTN 206. In response to such control signals, the CTN 206 has a desired impedance transformation characteristic. That is, the CTN 206 may be controlled to adjust, tune, change, or otherwise manipulate the impedance presented by the CTN 206 to the source 202 and / or load 204. The CTN 206 may be, for example, a tunable impedance matching circuit that may be used to match the impedance Z 1 , and / or the load 204 to a desired impedance. L The source side port of the CTN 206 presents the desired impedance Z to the source 202 even in the face of S,IN 206 to the load 204. SThe controller 208 may also be coupled to provide a control signal 212 to the source 202, for example to dynamically adjust the frequency of the source's RF signal. The controller 208 may, in some embodiments, receive a signal from the load 204, such as an impedance measurement looking into the load, as represented by line 211.

[0024]

[0042] The source 202, the CTN 206, the controller 208, and other elements of the RF system 200 may be coupled to a power supply voltage (eg, VDC) and ground (not shown).

[0025]

[0043] Turning to FIG. 2A, an exemplary CTN 220 can include a two-port switching network 222 configured to provide controllable voltage conversion (e.g., variable matching). The two-port switching network 222 acts as the primary means of conversion from an RF impedance level Z1 on one side of the CTN 220 to an RF impedance level Z2 on the other side of the CTN 220. In this case, the RF input can correspond to either a source or a load, and the RF output can correspond to the other. The switching network 222 can operate in synchronism with the RF signal to be converted and act approximately as a controllable transformer, as shown in FIG. 2A (specific implementations are described below in connection with subsequent figures). That is, the switching network can act to have a dynamically variable effective turns ratio (N1:N2) depending on the timing of switching relative to the RF signal.

[0026]

[0044] A secondary tuning means 224 may be coupled to the switching network 222 to implement a CTN that may be used to implement a variable impedance match between the RF source and the load. The secondary tuning means 224 of FIG. 2A may implement tuning of a series reactance. The secondary tuning means 224 may be a variable reactance X having a reactive impedance jX, implemented by a filter coupled to the switching network 222, as described further below. The variable reactance may be implemented in some embodiments by a high quality factor "Q" resonant tank and dynamic frequency tuning (DFT). DFT is a technique in which the frequency of an RF power source or source is adjusted such that the impedance or admittance provided by a frequency sensitive reactive or susceptible network is modified in a desired manner, and is used for impedance tuning in plasma and other applications.

[0027]

[0045] Secondary tuning means 224 is coupled to one output terminal of switching network 222 in the example of FIG. 2A.

[0028]

[0046] 2B shows another example of a CTN 240 in which a secondary tuning means 244, such as a susceptance admittance Y=jB, is coupled across both input terminals of a two-port switching network 242. The secondary tuning means 244 of FIG. 2B may provide tuning of the shunt susceptance.

[0029]

[0047] 3 illustrates an architecture that may be used to implement a CTN according to some embodiments. An example CTN architecture 300 may include a first (or “input”) RF port 302, a first (or “input side”) filter 304, a switching network 306, a second (or “output side”) filter 308, and a second (or “output”) RF port 310 coupled as shown. The first RF port 302 may be coupled to either an RF source or a load, and the second RF port 310 may be coupled to the other.

[0030]

[0048] The first filter 304 may include a fundamental voltage selective filter and the second filter 308 may include a fundamental current selective filter as shown in FIG. 3. One or both of the filters 304, 308 may be used with DFT and have variable reactance. The switching network 306 may be switched synchronously with the RF signal provided at the first port 302 to provide variable commutation. The switching network 306 and the filters 304, 308 together enable a controllable impedance match over a wide bandwidth between the first RF port 302 and the second RF port 310.

[0031]

[0049] Figures 3A and 3B show an example implementation of a CTN circuit utilizing the architecture of Figure 3. It will be appreciated that many implementations of this general approach are possible.

[0032]

[0050] Turning to FIG. 3A, an exemplary circuit 320 comprises a first RF port 322 coupled to a fundamental voltage-selective filter 324 and a second RF port 330 coupled to a fundamental current-selective filter 328. A switching network comprising switches 326a-326d may be coupled between the filters 324 and 328 as shown. The switches 326a-326d may be comprised as field-effect transistors (FETs), high-electron-mobility transistors (HEMTs), bipolar junction transistors (BJTs), or heterojunction bipolar transistors (HBTs) in some examples, although other active or passive components may be used in some cases. In general, any type of unidirectional blocking, bidirectional passing switching element may be used. This is generally true for any of the CTN embodiments shown or described herein.

[0033]

[0051] The fundamental voltage selection filter 324 may comprise a first capacitor 324a and a first inductor 324b coupled in parallel across the terminals of the first RF port 322, as shown. The fundamental current selection filter 328 may comprise a second capacitor 328a and a second inductor 328b coupled in series between one terminal of the second RF port 330 and the midpoint of the switches 326c and 326d.

[0034]

[0052] The filters 324 and 328 may be designed to extract the fundamental voltage and current, respectively, of the RF signal being converted. Thus, the values ​​of the capacitors 324a, 328a and the inductors 324b, 328b may be selected based on the nominal operating frequency of a given RF application in which the circuit 320 is placed. They may also be selected to provide high frequency selectivity or net quality factor Q, taking into account the respective impedance levels they interface with. This is generally true for any of the CTN implementations shown or described herein.

[0035]

[0053] 3B, another exemplary circuit 340 comprises a first RF port 342 coupled to a fundamental voltage selection filter 344 and a second RF port 350 coupled to a fundamental current selection filter 348. A switching network comprising switches 346a, 346b may be coupled between the filters 344 and 348 as shown. The fundamental voltage selection filter 344 may comprise a first capacitor 344a and a first inductor 344b coupled in parallel across terminals of the first RF port 342. The fundamental current selection filter 348 may comprise a second capacitor 348a and a second inductor 348b coupled in series between one terminal of the second RF port 350 and a midpoint of the switches 346a and 346b. Although only two switches 346a, 346b are shown in FIG. 3B, because the switches are bidirectional blocking and bidirectional passing, each of the illustrated switches may be implemented as two unidirectional blocking bidirectional passing switches (e.g., in “reverse” series) or two bidirectional blocking unidirectional passing switches (e.g., in “reverse” parallel), as the case may be.

[0036]

[0054] FIG. 4, together with FIG. 4A, illustrates the switching operation of an implementation of a CTN circuit based on the CTN circuit of FIG. 3A according to some embodiments.

[0037]

[0055] The exemplary circuit 400 includes a first RF port 402 coupled to a fundamental voltage selective filter 404, a second RF port 410 coupled to a fundamental current selective filter 408, and a switching network 406 coupled between the filters 404 and 408, as shown. In the example of Figure 4, a source 412 (e.g., a voltage source) is coupled to the first port 402 and a load 413 (e.g., a resistive load) is coupled to the second port 410. In other examples, a load may be coupled to the first port 402 and a source may be coupled to the second port 410.

[0038]

[0056] The source 412 has an input impedance Z S Knowing the input, voltage V S and current I S The source 412 may provide an RF signal having an impedance Z S The load 413 may have an equivalent output impedance (or Thévenin output impedance) that may be L and the voltage V L and current I LThe switching network 406 may receive a transformed / converted RF signal having a voltage V1 and a current I1. The fundamental voltage selection filter 404 may filter the input RF signal and provide the filtered RF signal to a first port 414 of the switching network 406, the filtered RF signal having a voltage V1 and a current I1, where Z1 indicates the impedance at the port 414. The impedance Z1 refers to the ratio of the fundamental complex voltage component to the fundamental complex current component at the first port. The switching network 406 includes four switches 406w-406z arranged as shown and operable according to one or more switching patterns (i.e., "switching waveforms"). A controller (e.g., the controller 208 of FIG. 2) may provide control signals to the individual switches 406w-406z to turn the switches on and off from time to time according to the switching patterns, in more detail. The switching network 406 can then output a shunted RF signal at a second port 416, the shunted RF signal having a voltage V2 and a current I2, where Z2 represents the ratio of the fundamental complex voltage component V2 to the fundamental complex current component I2, and thus indicates the impedance at the port 416. The fundamental current selection filter 408 filters the shunted RF signal to produce an output signal with an impedance Z L and the output signal can be supplied to a load 413 having a voltage V L and current I L has.

[0039]

[0057] Turning to FIG. 4A, in the circuit 400 of FIG. 4, the step-down of the fundamental voltage from the first port 414 to the second port 416 (i.e., from V1 to V2) and the step-up of the fundamental current between these ports (i.e., from I1 to I2) can be controlled according to a switching angle β, where β is between 0 and π.

[0040]

[0058] A first plot 420 has a horizontal axis 420x representing angle (ωt) and a vertical axis 420y representing voltage. A second plot 430 has a horizontal axis 430x representing angle (ωt) and a vertical axis 430y representing current. Each of the four switches 406w-406z may be switched on and off once per RF cycle according to β, with only one cycle shown in FIG. 4A. For example, as shown in legend 440, between angles 0 and β, switches 406w, 406x, and 406y may be on and switch 406z may be off; between angles β and π, switches 406x, 406y, and 406z may be on and switch 406w may be off; between angles π and π+β, switches 406y, 406z, and 406w may be on and switch 406x may be off; and between angles π and π+β, switches 406z, 406w, and 406x may be on and switch 406y may be off, as shown in legend 440. In practice, it should be recognized that there may be some additional dead time (when both switches of a complementary pair are held off) between complementary switch pairs, such as switches W and Z, and switches X and Y, to allow for zero-voltage switching transitions, as is known in the art of switching converters, that is not shown in the timing diagram. x represents a phase shift between the fundamental component of voltage V2 and the fundamental component of current I1 that may occur in the desired operation of the system.

[0041]

[0059] Referring to plot 420, waveform 422 represents V1 and waveform 424 represents V2. As can be seen, V2 is zero between β and π when switch 406x is off, and is zero between π+β and 2π when switch 406y is off. Elsewhere, V2 coincides with V1 (i.e., waveforms 422 and 424 overlap).

[0042]

[0060] Referring to plot 430, waveform 432 represents I1 and waveform 434 represents I2. I1 is zero between β and π when switch 406x is off, and zero between π+β and 2π when switch 406y is off, as can be seen. Elsewhere, I1 coincides with I2 (i.e., waveforms 432 and 434 overlap). This idealization ignores any transitions in voltages V2 and I1 that may occur in practice, along with the dead times discussed above.

[0043]

[0061] The switching network 406 effectively acts like a non-ideal transformer for the fundamental components of the voltage and current waveforms between the ports 414 and 416. The switching network 406 is not ideal in that there may be a phase shift between the fundamental components of the voltage and / or current at the ports 414, 416 depending on the angle β. The switching network nevertheless effectively acts to provide a timing-dependent step-down of the fundamental voltage and a step-up of the fundamental current from the first port 414 to the second port 416. In some embodiments, the effect of any phase shift through the switching network is compensated for using secondary control handles (e.g., DFTs related to admittance jB and / or impedance jY, such as using input and / or output filter networks), thereby achieving an overall impedance match between the ports 414 and 416.

[0044]

[0062] The advantage of using the circuit topology of Figure 4 in combination with the switching pattern of Figure 4A is that the switching pattern requires only one switching cycle per device per RF cycle, and all switching transitions can be done with zero-voltage switching (ZVS) to keep losses low when the switching pattern includes adequate dead time. Of course, many other topologies and switching patterns may be realized using two-port switching structures and associated elements to provide impedance matching.

[0045]

[0063] 5A and 5B show two different views of another CTN circuit implementation according to an embodiment of the present disclosure. The exemplary circuit 500 is similar to the circuit 400 of FIG. 4, and both are based on the CTN architecture of FIG. 3. Here, the basic voltage selection filter includes an inductor 502 (L P ) and two capacitors 504a, 504b (C P ), the switching network includes four switches 506w to 506z, and the fundamental current selection filter includes an inductor 508 and a capacitor 510.

[0046]

[0064] Circuit 500 therefore differs from the embodiment of FIG. 4 by including a replicated capacitor C between the left and right half bridges of the switching network as shown. P Each of the capacitors 504a, 504b (C P ) may be used half of the RF cycle during operation, i.e., in each half cycle of operation, one bridge leg may be shorted and the other may be switched for controllable commutation so that one of the capacitors 504a, 504b is used in each half cycle.

[0047]

[0065] The exemplary circuit 500 is shown as stepping up the impedance from a load at the second port 514 to a source at the first port 512. The circuit 500 can also operate in the reverse direction by stepping down the load impedance at the first port 512 to the source impedance at the second port 514.

[0048]

[0066] The switching patterns / waveforms used in the circuit 500 of FIG. 5 may be the same or similar to those shown in FIG. 4A, and the switching timing may be similar. Each device (i.e., switch) therefore switches once per cycle, and all devices switch in ZVS. In some embodiments, a basic current selection filter (i.e., inductor 508 and capacitor 510, or L S -C S ) is realized with a high quality factor, and changing the RF frequency can be used to vary the effective impedance seen thereby (i.e., by the DFT). That is, a controller (e.g., controller 208 of FIG. 2) can send a control signal to adjust the frequency of the RF power source (e.g., source 202 of FIG. 2). Thus, according to some embodiments, the DFT can be used to perform reactance compensation for the effects of the load impedance, transformation network, and / or input filter network. Using an appropriate DFT and switching control angle β, the load impedance R L The desired resistive input impedance Z S can be realized.

[0049]

[0067] In comparison to many matching systems in which a variable effective reactance (i.e., a single-port element) is used, the embodiments of Figures 4, 5A, and 5B utilize a two-port switching network that acts as a variable transformer (i.e., a variable two-port device).

[0050]

[0068] 6 is a plot 600 having a horizontal axis 600x representing the signal angle (ωt) and a vertical axis 600y representing the value of the normalized signal f(t). The voltage, current and impedance transformations performed by the controllable transformation networks of FIGS. 4, 5A and 5B can be determined using information about the amplitude and phase of the fundamental component of the waveform f(t) 602 defined below.

[0051]

number

[0052]

[0069] The fundamental component of this waveform can be expressed as:

[0053]

number

[0054] In this formula,

number

[0055]

[0070] Analyzing the waveform in Figure 4A, the input impedance Z S to the desired value Z0, the secondary reactance X comp and the ideal values ​​of the switching angle β can be identified. The particular ideal control variables can be found by solving the following equations:

[0056]

number

[0057]

[0071] Using the above equations, a series of curves can be generated, as shown in Figure 7. A first plot 700 shows R plotted against switching angle β (horizontal axis 700x, in degrees). L A second plot 720 includes a curve 702 of X / Z0 (vertical axis 700y) plotted against switching angle β (horizontal axis 720x, in degrees). net / RL (vertical axis 720y) includes curve 722.

[0058]

[0072] Curves similar to 702, 704 may be generated using the general concepts and techniques disclosed herein for various CTN circuit implementations and switching patterns disclosed herein, including but not limited to those of Figures 4, 4A, 5A, and 5B.

[0059]

[0073] Curves 702 and 722 are plotted against a load resistance R L and reactance X L Therefore, the secondary reactance X comp and the switching angle β (and possibly X comp and β) (where X comp =X net -X L ). X comp To select β, specifically, the following procedure may be followed.

[0060] 1. Known load resistance R L , load reactance X L , and the desired impedance value Z0. 2. Use curve 702 to find β. 3. Use curve 722 to create X net / R L and thus X net Request. 4.X net / R L From X comp Determine where X comp isX net and X L This is the difference between...

[0061]

[0074] The above procedure may be implemented and / or applied in various disclosed CTN systems and circuits to provide impedance matching between an RF source and a load while minimizing phase shift. For example, the controller 208 of FIG. 2 may include hardware and / or software configured to generate a curve similar to that shown in FIG. 7 and to rapidly select and adjust an RF frequency and switching angle β using the above procedure, where the compensation reactance X comp , L S and C S As another example, the selected switching angle β may be applied to any of the circuits of FIGS.

[0062]

[0075] In the above model, it may be assumed that the input filter network is at resonance (i.e., exhibits infinite shunt impedance) and that the desired value of the output reactance is achievable by DFT or some other means. Furthermore, it may be assumed that the input and output filters address harmonics generated by the switching network and possibly include additional filtering not shown.

[0063]

[0076] In some embodiments, the controller in a wide bandwidth system does not just follow the ideal curve of FIG. 7, but also uses some combination of feedback and feedforward to adjust X net to the control variable X comp may also be selected. This may include the use of digital pre-distortion, adaptive feedforward, etc., to dynamically determine the control points as operating conditions change.

[0064]

[0077] While the curves in FIG. 7 assume that an output filter network is used to generate a variable reactance for matching, in some embodiments, a high Q input tank may be used as well, and the susceptance B (imaginary part of admittance Y) of the input shunt filter may be tuned by DFT in conjunction with the operation of the switching network to achieve impedance matching. In some cases, both input and output susceptances / reactances may be used together to achieve matching. It may be advantageous to use a higher Q filter network at the output (to reduce reactive power handled by the switching network) or at the input (to facilitate matching in the case of multiple outputs described below). According to embodiments of the present disclosure, either or both may be utilized through DFT or some other means (e.g., switched reactance bank, PSIM network, varactors, mechanically variable reactance, etc.) to generate a variable reactance.

[0065]

[0078] Figures 8A and 8B show other switching patterns that may be used within CTN-based RF matching systems, architectures, and circuits, such as the RF system of Figure 2, the CTN architecture of Figure 3, and / or the CTN circuit implementations of Figures 4, 5A, and 5B. These patterns include the ability to perform matching in fixed frequency operation, potentially providing improved control capabilities.

[0066]

[0079] FIG. 8A shows a plot 800 with a horizontal axis 800x representing angle (ωt), a vertical axis 800y representing voltage, a first waveform 802 representing V1, and a second waveform 804 representing V2 (where V1 and V2 correspond to voltages at different ports of the switching network as previously discussed). In this example, different switching angles β1 and β2 are used in each half cycle. The switching transitions may be more specifically performed at angles 0, β1, π, and β2+π. Again, there may be a possible dead time between complementary switch pairs. This may allow both the basic voltage commutation and some degree of phase shift through the switching network to be controllable. This may be useful for further tuning or to allow fixed frequency operation over a range (e.g., without DFT).

[0067]

[0080] FIG. 8B shows a plot 820 with a horizontal axis 820x representing angle (ωt), a vertical axis 820y representing voltage, a first waveform 822 representing V1, and a second waveform 824 representing V2 (where V1 and V2 correspond to voltages at different ports of the switching network as previously discussed). In this example, two switching transitions per half cycle are utilized, defined by switching angles α and β. The switching transitions may occur at angles α, β, α+π, and β+π in more detail. Again, there may be additional dead times between complementary switch pairs not shown, as is known in the art. Although the additional switching transitions provide additional degrees of freedom in controlling the phase shift through the switching network, it should be understood herein that the added transitions may be less suitable for ZVS switching. Thus, in some embodiments, additional circuitry may be added to facilitate ZVS switching while still using the switching pattern of FIG. 8B. It is worth noting that if the switching angle α=π-β is selected, the commutation by the switching network exhibits no phase shift and acts like an "ideal transformer" of the fundamental component. Alternatively, the switching angles α and β can be selected together to control both the commutation ratio and the phase shift, allowing impedance matching at a fixed frequency.

[0068]

[0081] 9, the structures and techniques disclosed above can be extended to realize a CTN-based impedance matching system that handles multiple outputs (e.g., N=2, 3, 5 outputs, etc.). A multiple-output CTN circuit 900 can match a source 902 to N loads 904a-n as shown. The exemplary circuit 900 includes an input-side filter 906 (e.g., a fundamental voltage-selective filter), N switching networks 908a-n, and N output-side filters 910a-n (e.g., fundamental current-selective filters).

[0069]

[0082] DFT can be used to control the system of Figure 9 to provide a single tuning handle (e.g., using a high Q input filter network and a low Q output filter close to resonance). Each switching network can have a separate control angle β (or a set of control angles as shown in Figure 8). In combination, a set of load impedances presents the source with the desired input port impedance Z S On the other hand, multiple load impedances Z L1 From Z LN It may be possible to control how power is distributed between the . This therefore represents a combination of a wide bandwidth tunable matching network and a controllable RF power splitter, as shown in RF power transfer system 1000 of FIG.

[0070] <Controllable switching power amplifier>

[0083] Turning to FIG. 11, as described above, there is a need for an RF inverter (or power amplifier or RF generator) that can provide generation and controlled delivery of RF power to a highly variable load impedance over a wide bandwidth. According to some embodiments, the switching networks, CTNs, and related techniques described above can be utilized to provide a power amplifier system with exactly these characteristics. One approach is to cascade a conventional (e.g., switched mode) RF amplifier with a CTN. Another approach is to provide a controllable conversion power amplifier (CTPA).

[0071]

[0084] 11 shows an example of a CTPA according to some embodiments. The exemplary CTPA 1100 includes a voltage source 1102, an input port 1104, an input side filter 1106, a switching network 1108 with four switches 1108w-1108z, an output port 1110, an output side filter 1112, and another port 1114 that can be connected to a load 1116, as shown. On the input side, the CTPA 1100 can operate in a manner similar to an “inverse class D” or “current mode class D” switching power amplifier, but the switching angle of the embedded switching network 1108 is controlled to achieve controllable voltage transfer (and therefore power control).

[0072]

[0085] FIG. 11A shows examples of switching patterns and associated waveforms 1120 and 1140 that may be used in the CTPA of FIG.

[0073]

[0086] According to the switching pattern and associated waveform 1120, between angles 0 and β, switches 1108w, 1108x, 1108y are on and switch 1108z is off; between angles β and π, switches 1108x, 1108y, 1108z are on and switch 1108w is off; between angles π and π+β, switches 1108y, 1108z, 1108w are on and switch 1108x is off; and between angles π+β and 2π, switches 1108z, 1108w, 1108x are on and switch 1108y is off.

[0074]

[0087] According to the switching pattern and associated waveforms 1140, between angles 0 (approximately) and β, all switches are on except switch 1108z, between angles β and π (approximately) all switches are on except switch 1108w, between angles π (approximately) and π+β all switches are on except switch 1108x, and between angles π+β and 2π (approximately) all switches are on except switch 1108y.

[0075]

[0088] The overlap of switching waveforms shown in Figure 11A may be used to allow for changes in switching frequency while maintaining ZVS switching for all devices with high Q input resonances, such as enabling DFT while still allowing devices to switch at ZVS. (The control range may be optimal for switching frequencies below the input resonance frequency.) DFT of the switching frequency with a high Q output network allows for management of variations in load reactance.

[0076]

[0089] The circuit of FIG. 11 and the corresponding switching pattern of FIG. 11A can provide a power amplifier that can handle fast power control into a wide range of rapidly changing load impedances. sw Control of provides the necessary control handle for a wide range of load reactance and resistance values. It should also be appreciated that a multiple output version of the proposed power amplifier system can be realized using, for example, an approach similar to that of FIG.

[0077]

[0090] It will be further understood that other power amplifier topologies may be integrated with such switching / controllable conversion networks and associated controls. For example, Figure 12 shows a circuit 1200 implemented based on a voltage mode class D configuration, including the option to provide multiple outputs. Figure 12A shows a switching pattern 1220 that may be used in the circuit of Figure 12.

[0078]

[0091] As used herein, the terms "processor" and "controller" are used to describe electronic circuitry that performs a function, operation, or sequence of operations. The function, operation, or sequence of operations may be hard-coded into the electronic circuitry or soft-coded by instructions held in a memory device. The function, operation, or sequence of operations may be performed using digital values ​​or using analog signals. The processor or controller may be embodied in an application specific integrated circuit (ASIC), which may be an analog or digital ASIC, a microprocessor with associated program memory, and / or a separate electronic circuit, which may be analog or digital. The processor or controller may incorporate an internal processor or module that performs a function, operation, or a portion of a sequence of operations. A module may similarly incorporate an internal processor or module that performs a function, operation, or a portion of a sequence of operations of the module.

[0079]

[0092] As used herein, the term "preset," when referring to a value or signal, is used to refer to a value or signal that is set or fixed at the factory during manufacture or thereafter by external means, e.g., programming. As used herein, the term "predetermined," when referring to a value or signal, is used to refer to a value or signal that is specified by the circuit after manufacture and during operation.

[0080]

[0093] The electronic circuits shown in the figures herein may be shown in the form of analog or digital blocks, but it will be understood that the analog blocks may be replaced with digital blocks performing the same or similar functions, and the digital blocks may be replaced with analog blocks performing the same or similar functions, and analog-to-digital or digital-to-analog conversions may not be explicitly shown in the figures, but will be understood.

[0081]

[0094] In the foregoing detailed description, various features have been grouped together in one or more separate embodiments for the purpose of streamlining the disclosure. This method of disclosure is not to be interpreted as reflecting an intention that the claims require more features than are expressly recited therein. Rather, aspects of the invention may exist in fewer than all features of each disclosed embodiment.

[0082]

[0095] References in this disclosure to "one embodiment," "embodiments," "some embodiments," or variations of such phrases indicate that the embodiment being described may have a particular feature, structure, or characteristic, and that all embodiments may have the particular feature, structure, or characteristic. Moreover, such phrases do not necessarily refer to the same embodiment. Furthermore, when a particular feature, structure, or characteristic is described in accordance with the knowledge of one skilled in the art to which the invention pertains, such feature, structure, or characteristic may be affected in relation to other embodiments, whether or not it is explicitly described.

[0083]

[0096] The disclosed subject matter is not limited in its application to the details of construction and the arrangement of components set forth in the following description or illustrated in the drawings. The disclosed subject matter is capable of other embodiments and can be practiced and carried out in various ways. Thus, those skilled in the art to which the invention pertains will appreciate that the conception on which this disclosure is based may be readily utilized as a basis for the design of other structures, methods and systems for carrying out the several purposes of the disclosed subject matter. The claims should therefore be regarded as including such equivalent constructions insofar as they do not depart from the spirit and scope of the disclosed subject matter.

[0084]

[0097] While the disclosed subject matter has been described and illustrated in the foregoing exemplary embodiments, it will be understood that the disclosure is presented by way of example only and that many changes may be made in the details of the embodiments of the disclosed subject matter without departing from the spirit and scope of the disclosed subject matter.

[0085]

[0098] All publications and references cited herein are expressly incorporated by reference in their entirety.

Claims

1. 1. A system for transferring radio frequency (RF) signals between a source and a load, comprising: a first port and a second port connected to the source and the load, respectively; a first filter connected to the first port; a second filter connected to the second port, at least one of the first filter and the second filter being operable to provide a variable reactance using dynamic frequency tuning (DFT); a two-port switching network connected between the first filter and the second filter, the two-port switching network comprising a plurality of switches; a controller coupled to the switching network and the source, the controller configured to operate the plurality of switches according to a switching pattern to dynamically adjust a frequency of the RF signal to provide a controllable impedance match between the first port and the second port; A system comprising:

2. The system of claim 1 , wherein the first filter is configured to extract a fundamental voltage of the RF signal.

3. 3. The system of claim 2, wherein the first filter comprises a capacitor and an inductor connected in parallel across terminals of the first port.

4. The system of claim 1 , wherein the second filter is configured to extract a fundamental current of the RF signal.

5. The system of claim 4 , wherein the second filter comprises a capacitor and an inductor connected in series to one terminal of the second port.

6. The system of claim 1 , wherein the controller operates the plurality of switches in synchronization with and with a controllable phase shift relative to the RF signal.

7. The system of claim 1 , wherein the controller operates the two or more switches according to a switching pattern defined by an angle β relative to the RF signal.

8. The controller has a load resistance R Ｌ and reactance X Ｌ The system of claim 7 , configured to calculate β by:

9. The controller has a load resistance R Ｌ and reactance X Ｌ The system of claim 7 configured to select a switching frequency by:

10. The system of claim 1 , wherein the load comprises a plasma load.

11. 2. The system of claim 1, wherein the controller is configured to provide controllable impedance matching between the first port and the second port during at least a twofold power step that occurs in 10 microseconds or less.

12. The system of claim 1 , wherein the plurality of switches of the switching network includes four switches arranged in a bridge.

13. The four switches are a first switch having a first terminal connected to the first terminal of the first filter and having a second terminal connected to the first terminal of the second filter; a second switch having a first terminal connected to the second terminal of the first filter and having a second terminal connected to the second terminal of the second filter; a third switch having a first terminal connected to the second terminal of the first switch and to the first terminal of the second filter; a fourth switch having a first terminal connected to the second terminal of the second switch and to the second terminal of the second filter, and having a second terminal connected to the second terminal of the third switch; The system of claim 12 , comprising:

14. The four switches are a first switch having a first terminal connected to the first terminal of the first filter and having a second terminal connected to the first terminal of the second filter; a second switch having a first terminal connected to the first terminal of the second filter and to the second terminal of the first switch, and having a second terminal connected to the second terminal of the first filter; a third switch having a first terminal connected to the second terminal of the first filter and to the second terminal of the second switch, and having a second terminal connected to the second terminal of the second filter; a fourth switch having a first terminal connected to the second terminal of the second filter and to the second terminal of the third switch, and having a second terminal connected to the third terminal of the first filter; The system of claim 12 , comprising:

15. The system of claim 12 , wherein the four switches include unidirectional blocking, bidirectional pass switches.

16. 1. A controllable diversion network for diverts radio frequency (RF) signals between a source and a load, comprising: a first port and a second port connected to the source and the load, respectively; a first filter connected to the first port and configured to extract a fundamental voltage of the RF signal; a second filter connected to the second port and configured to extract a fundamental current of the RF signal, at least one of the first filter and the second filter being operable to provide a variable reactance using dynamic frequency tuning (DFT); and a switching network connected between the first filter and the second filter, the switching network comprising a bridge circuit having four switches; A controllable transfer network comprising:

17. 17. The controllable commutation network of claim 16, wherein the first filter comprises a capacitor and an inductor connected in parallel across the terminals of the first port.

18. 17. The controllable commutation network of claim 16, wherein the second filter comprises a capacitor and an inductor connected in series to one terminal of the second port.

19. The four switches are a first switch having a first terminal connected to the first terminal of the first filter and having a second terminal connected to the first terminal of the second filter; a second switch having a first terminal connected to the second terminal of the first filter and having a second terminal connected to the second terminal of the second filter; a third switch having a first terminal connected to the second terminal of the first switch and to the first terminal of the second filter; a fourth switch having a first terminal connected to the second terminal of the second switch and to the second terminal of the second filter, and having a second terminal connected to the second terminal of the third switch; 17. The controllable transformation network of claim 16, comprising:

20. The four switches are a first switch having a first terminal connected to the first terminal of the first filter and having a second terminal connected to the first terminal of the second filter; a second switch having a first terminal connected to the first terminal of the second filter and to the second terminal of the first switch, and having a second terminal connected to the second terminal of the first filter; a third switch having a first terminal connected to the second terminal of the first filter and to the second terminal of the second switch, and having a second terminal connected to the second terminal of the second filter; a fourth switch having a first terminal connected to the second terminal of the second filter and to the second terminal of the third switch, and having a second terminal connected to the third terminal of the first filter; 17. The controllable transformation network of claim 16, comprising:

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