Radar signal processing circuit and radar signal processing method

Abstract

To suppress deterioration of radar detection performance caused by circuit errors.SOLUTION: A radar signal processing circuit includes: a transmission phase rotation unit that imparts phase rotation based on a plurality of different phase rotation amounts controlled by a series of a plurality of mutually different phase variable patterns to a transmission signal for each transmission period; a radar transmission radio unit that outputs a plurality of different radar transmission signals to which multiplexing processing has been applied to a plurality of transmission antennas; and a radar reception radio unit that inputs one or more reflected wave signals resulting from at least one of the plurality of different radar transmission signals being reflected by a target via a plurality of reception antennas. The plurality of different phase rotation amounts are a series in which a shift amount that is an integer multiple of two or greater of 2π is sequentially added for each of a plurality of transmission periods for each transmission period, and the plurality of mutually different phase variable patterns are patterns that have elements of each of the plurality of transmission periods and vary the amount of phase rotation for each of the plurality of transmission periods.SELECTED DRAWING: Figure 1

Description

Technical Field

[0001] The present disclosure relates to a radar signal processing circuit and a radar signal processing method.

Background Art

[0002] In recent years, studies have been underway on radar devices using radar transmission signals with short wavelengths including microwaves or millimeter waves capable of obtaining high resolution. In addition, in order to improve outdoor safety, there is a demand for the development of a radar device (wide-angle radar device) that detects objects (targets) including pedestrians over a wide angle range, in addition to vehicles.

[0003] For example, as a radar device, a pulse compression radar device is known. When detecting a vehicle / pedestrian with a pulse compression radar device, the reflected wave from a pedestrian is weak compared to the reflected wave from a vehicle. Therefore, in the radar transmission unit, a transmission configuration that transmits a pulse compression wave with a low range side lobe is required, and in the radar reception unit, a reception configuration with a wide reception dynamic range is required.

[0004] As a pulse compression code for obtaining low range side lobe characteristics, for example, it is known to use a Barker code, a PN sequence code, a complementary code, or the like. Hereinafter, as an example, the case of using a complementary code will be described. The complementary code consists of codes that form a pair (hereinafter, complementary codes a n , b n . Here, n = 1,..., L. L is the code length). The autocorrelation operation of each of the two codes is represented by the following equations (1) and (2).

Equation

Equation

[0005] In equations (1) and (2), when n > L and n < 1, a n = 0, b nIt is equal to 0. The complementary code has the property that, by adding the autocorrelation operation results of each of the two codes with the shift time τ made the same, the correlation values other than τ = 0 become zero and the range side lobe becomes zero as shown in the following formula (3).

Number

[0006] Such complementary codes a n , b n are known to be time-division transmitted in each predetermined radar transmission period in a pulse compression radar.

[0007] Regarding the method for generating complementary codes, it is disclosed in Non-Patent Document 1. According to Non-Patent Document 1, for example, based on the complementary A = [a 1, a2] =

[0011] , B = [b 1, b2] = [1 -1] composed of elements '1' or '-1', complementary codes with code lengths L = 4, 8, 16, 32, …, 2 P can be sequentially generated. The pulse compression radar can increase the received signal level of the radar reflected wave by repeatedly transmitting the above-described pulse compression code a predetermined number of times in each radar transmission period.

Prior Art Documents

Patent Documents

[0008]

Patent Document 1

Patent Document 2

Non-Patent Documents

[0009]

Non-Patent Document 1

Non-Patent Document 2

Non-Patent Document 3

Summary of the Invention

Problems to be Solved by the Invention

[0010] When attempting to implement the above-described pulse compression radar using a radio frequency band (RF: Radio Frequency) such as millimeter waves, circuit errors are introduced into the RF circuit or analog baseband circuit in the radar transmitter or radar receiver of the radar device, and the radar detection performance (or radar ranging performance) deteriorates compared to the ideal characteristics without circuit errors.

[0011] One aspect of the present disclosure provides a radar signal processing circuit and a radar signal processing method capable of suppressing deterioration of radar detection performance due to circuit errors.

Means for Solving the Problems

[0012] A radar signal processing circuit according to an aspect of the present disclosure includes a signal generation unit that generates a transmission signal, a transmission phase rotation unit that applies a phase rotation based on a plurality of different phase rotation amounts controlled by a series of a plurality of mutually different phase variable patterns to the transmission signal for each transmission period, a multiplexing unit that applies multiplexing processing to a plurality of different radar transmission signals that are the transmission signals to which the phase rotation is applied, a radar transmission radio unit that outputs the plurality of different radar transmission signals to which the multiplexing processing is applied to a plurality of transmission antennas, and a radar reception radio unit that inputs, via a plurality of reception antennas, one or more reflected wave signals in which at least one of the plurality of different radar transmission signals is reflected by a target. The plurality of different phase rotation amounts is a series obtained by sequentially adding, for each of a plurality of transmission periods, a shift amount that is an integer multiple of 2 or more of 2π, and the plurality of mutually different phase variable patterns has elements for each of the plurality of transmission periods and is a pattern that varies the phase rotation amount for each of the plurality of transmission periods.

[0013] A radar signal processing method according to an aspect of the present disclosure includes generating a transmission signal, applying a phase rotation based on a plurality of different phase rotation amounts controlled by a series of a plurality of mutually different phase variable patterns to the transmission signal for each transmission period, applying multiplexing processing to a plurality of different radar transmission signals that are the transmission signals to which the phase rotation is applied, outputting the plurality of different radar transmission signals to which the multiplexing processing is applied to a plurality of transmission antennas, inputting, via a plurality of reception antennas, one or more reflected wave signals in which at least one of the plurality of different radar transmission signals is reflected by a target, the plurality of different phase rotation amounts being a series obtained by sequentially adding, for each of a plurality of transmission periods, a shift amount that is an integer multiple of 2 or more of 2π, the plurality of mutually different phase variable patterns having elements for each of the plurality of transmission periods and being a pattern that varies the phase rotation amount for each of the plurality of transmission periods.

[0014] Note that these general or specific aspects may be implemented in a system, method, integrated circuit, computer program, or recording medium, or may be implemented in any combination of a system, device, method, integrated circuit, computer program, and recording medium.

Advantages of the Invention

[0015] According to one aspect of the present disclosure, it is possible to suppress the deterioration of radar detection performance due to circuit errors.

[0016] Further advantages and effects in one aspect of the present disclosure will be apparent from the specification and drawings. Such advantages and / or effects are provided by some embodiments and the features described in the specification and drawings respectively, but not all necessarily need to be provided in order to obtain one or more identical features.

Brief Description of the Drawings

[0017]

Figure 1

Figure 2

Figure 3

Figure 4A

Figure 4B

Figure 5

Figure 6

Figure 7

Figure 8

Embodiments for Carrying Out the Invention

[0018] In a radar device, as circuit errors, IQ mismatch, DC offset, etc. occur in the quadrature modulation circuit / quadrature demodulation circuit, phase noise occurs in the frequency conversion section, and quantization noise occurs in the AD converter and DA converter.

[0019] When a DC offset is included as a circuit error, a phenomenon occurs in which the noise level increases compared to the case where there is no error. Therefore, in a radar device, when the reflected wave from a target is lower than the noise level, the target is not detected, leading to a decrease in the detection rate and deterioration of the radar detection performance.

[0020] In response to this problem, conventionally, a method of removing the DC offset by the circuit configuration of the radar transmission section or the radar reception section has been proposed. Specifically, Patent Document 1 discloses a configuration in which a high-pass filter is arranged for discrete data output from an AD converter to remove the DC offset. Further, Patent Document 2 discloses a configuration in which a band-pass filter is arranged at the input stage of the AD converter to remove the DC offset. Also, Non-Patent Document 2 discloses a circuit configuration for correcting IQ imbalance circuit errors.

[0021] However, in the prior art, since it is necessary to provide a DC offset removal circuit or a correction circuit for IQ imbalance circuit errors, the circuit configuration becomes complicated. Further, when a high-pass filter or a band-pass filter is provided for DC offset removal, a desired radar reflection wave component other than the DC offset component may be attenuated, or amplitude distortion or phase distortion may occur due to the filter response, resulting in deterioration of the radar detection performance.

[0022] Also, in a configuration with a DC offset removal circuit, if the DC offset component or the IQ imbalance component cannot be completely removed and a circuit error component remains, the circuit error component will also be integrated by the coherent integration process in the radar reception process, resulting in deterioration of the radar detection performance. For example, even if the residual component of the circuit error is minute, the residual component will increase by about 30 to 40 dB due to the integration effect, so a high-precision error detection mechanism is required, and the hardware configuration of the radar device will become complicated.

[0023] On the other hand, Non-Patent Document 3 discloses a radar device that transmits a code combining phase modulation during a plurality of pulse transmission periods and cancels the received DC offset component by performing coherent integration processing on the reflected wave.

[0024] As an example, a code A = [a1, a2, …, a L used for pulse compression on the transmission side of the radar device, and a code -A = [-a1, -a2, …, -a L that is 180 degrees out of phase with respect to the code A are used. The principle that the received DC offset component is canceled by transmitting the codes A and -A in two transmission periods, performing correlation processing on the reception side, and performing coherent integration is shown below.

[0025] Note that hereinafter, the case where there is no noise component and the sum of each element of the code A is not zero as shown in the following equation (4) will be described.

Equation

[0026] <Example 1: When the received signal of the radar reflected wave includes a received DC offset component> (1-1) When transmitting the code A, the received signal of the code A including the received DC offset component α Rx (γA + α Rx ) and the autocorrelation value obtained by the autocorrelation operation with the code A is represented by the following equation (5).

Equation

[0027] In Equation (5), γ represents the complex received response of the radar reflected wave, and the asterisk (*) is the complex conjugate operator.

[0028] When transmitting the (1-2) code -A, the received DC offset component α Rx is included in the received signal of code -A (-γA + α Rx ), and the autocorrelation value obtained by the autocorrelation operation between the code -A and itself is expressed by the following Equation (6).

Number

[0029] On the receiving side, when performing coherent integration processing on the autocorrelation value (Equation (5)) obtained only by the above (1-1), even if γ is zero (that is, even if there is no radar reflected wave), the two items including the received DC offset component α Rx are integrated. For this reason, the noise level (floor level) increases over all distance ranges, and the radar detection performance deteriorates.

[0030] On the other hand, on the receiving side, when performing coherent integration processing on the autocorrelation values (Equation (5) and Equation (6)) obtained by the above (1-1) and (1-2), as shown in the following Equation (7), the received DC offset component α Rx can be canceled.

Number

[0031] Thereby, an increase in the noise level can be prevented, and the deterioration of the radar detection performance of the radar device can be suppressed. Also, for the received signal of the radar reflected wave including Doppler fluctuations, similarly, the received DC offset component α Rx can be canceled, so an increase in the noise level (floor level) can be prevented, and the deterioration of the radar detection performance can be suppressed.

[0032] <Example 2: When the transmitted DC offset component α is included in the received signal of the radar reflected wave without Doppler fluctuation> Tx <When it is included>

[0033] (2-1) When transmitting symbol A, the received signal (γA + α) including the transmitted DC offset component α Tx and the autocorrelation value obtained by the autocorrelation operation between the received signal and symbol A is expressed by the following equation (8). Tx

Equation

[0034] (2-2) When transmitting symbol -A, the received signal (-γA + α) of symbol -A including the transmitted DC offset component α Tx and the autocorrelation value obtained by the autocorrelation operation between the received signal and symbol -A is expressed by the following equation (9). Tx

Equation

[0035] On the receiving side, when the autocorrelation values (Equations (8) and (9)) obtained in (2-1) and (2-2) above are subjected to coherent integration processing, similar to the case where the received DC offset component α shown in <Example 1> is included, the transmitted DC offset component α Rx is canceled, preventing an increase in the noise level (floor level) and suppressing deterioration of the radar detection performance. Tx

[0036] As described in <Example 1> and <Example 2>, by canceling the transmitted DC offset component or the received DC offset component, deterioration of the radar detection performance can be suppressed.

[0037] ​​​However, even with the above method, if there is a transmitted DC offset component (including the carrier leakage component) in the radar transmitter and there is Doppler fluctuation in the radar reflected wave, the transmitted DC offset component will remain. This is because the transmitted DC offset component is subject to Doppler fluctuation, resulting in a cancellation error in the above method. Therefore, there is a problem that the noise level of the Doppler component included in the remaining transmitted DC offset component increases and the radar detection performance deteriorates.

[0038] Hereinafter, the case where the radar reflected wave includes Doppler fluctuation will be described.

[0039] <Example 3: When the received signal of the radar reflected wave including Doppler fluctuation contains the transmitted DC offset component α Tx is included>

[0040] Here, the Doppler fluctuation included in the radar reflected wave is regarded as "exp(j2πf d ×T r ) = exp(jΨ d )" (f d : Doppler frequency, T r : Radar transmission period. However, the condition is that the Doppler fluctuation within the symbol can be regarded as constant).

[0041] (3-1) When transmitting symbol A, the autocorrelation value obtained by the autocorrelation operation between the received signal γ(A + α Tx ) containing the transmitted DC offset component α and symbol A is expressed by the following formula (10). Tx

Equation

[0042] (3-2) When transmitting symbol -A, the autocorrelation value obtained by the autocorrelation operation between the received signal γ(-A + α Tx ) of symbol -A containing the transmitted DC offset component α and symbol -A is expressed by the following formula (11). Tx ) exp(jΨ d )

Equation

[0043] On the receiving side, when the autocorrelation values (Equations (10) and (11)) obtained in the above (3-1) and (3-2) are subjected to coherent integration processing, depending on γ, the transmission DC offset component α as shown in the following Equation (12) Tx Two terms including are integrated.

Equation

[0044] Generally, for the reception processing of radar reflected waves including Doppler fluctuations, coherent integration processing using Doppler frequency analysis is applied. Therefore, the noise level (floor level) of the Doppler frequency components corresponding to the two terms in Equation (12) increases, and the radar detection performance deteriorates. Also, the received power of the transmission DC offset component α Tx is proportional to |γα Tx | 2 Therefore, the higher the received power of the radar reflected wave, the greater the increase in the noise level (floor level), and the greater the deterioration of the radar detection performance.

[0045] Next, in a pulse compression radar, even when a transmission DC offset (carrier leak) and a reception DC offset are included, a method for preventing an increase in the noise level (floor level) and suppressing deterioration of the radar detection performance without adding a high-precision correction circuit for circuit error correction will be described.

[0046] In the method of transmitting a pulse compression code with phase modulation added in the above-described plurality of pulse transmission periods, when a transmission DC offset component (including a carrier leak component) exists in the radar transmission unit, if there is a Doppler fluctuation in the radar reflected wave, the transmission DC offset component remains, and the noise level of the specific Doppler component increases. To address this issue, the transmission DC offset component can be reduced by the following method.

[0047] Specifically, the radar device uses a code A = [a1, a2, …, a L for pulse compression on the transmission side, and a code -A = [-a1, -a2, …, -a L obtained by inverting the phase of code A. In four radar transmission cycles, codes A, -A, -A, and A are transmitted respectively, and correlation processing is performed on the receiving side, followed by coherent integration processing.

[0048] The following specifically describes this method. Hereinafter, similar to the above, the case where there is no noise component and the sum of the elements of code A is not zero (refer to Equation (4)) will be described.

[0049] <Example 4: When the received signal of the radar reflected wave containing Doppler fluctuations includes a transmission DC offset component α Tx >

[0050] Here, the Doppler fluctuations included in the radar reflected wave are regarded as "exp(j2πf d ×T r ) = exp(jΨ d )" (f d : Doppler frequency, T r : Radar transmission period. However, this is under the condition that the Doppler fluctuations within the code can be considered constant).

[0051] (4 - 1) When transmitting code A, the autocorrelation value obtained by autocorrelation operation between the received signal γ(A + α Tx ) including the transmission DC offset component α and code A is expressed by the following Equation (13). Tx ) and code A is represented by the following Equation (13). [Equation]

[0052] (4 - 2) When transmitting code -A, the autocorrelation value obtained by autocorrelation operation between the received signal γ(-A + α Tx ) including the transmission DC offset component α of code -A and code -A is expressed by the following Equation (14). Tx ) exp(jΨ d ) and code -A is represented by the following Equation (14). [Number]

[0053] When transmitting symbol - A, the received signal γ (-A + α Tx ) containing the transmission DC offset component α Tx ) exp(j2Ψ d ) and the autocorrelation value obtained by the autocorrelation operation between symbol - A and itself is expressed by the following equation (15). [Number]

[0054] When transmitting symbol A, the received signal γ (A + α Tx ) containing the transmission DC offset component α Tx ) exp(j3Ψ d ) and the autocorrelation value obtained by the autocorrelation operation between symbol A and itself is expressed by the following equation (16). [Number]

[0055] On the receiving side, when performing coherent integration processing on the autocorrelation values (Equations (13) to (16)) obtained in the above (4 - 1) to (4 - 4), two terms containing the transmission DC offset component α depending on γ are integrated as shown in the following equation (17). Tx [Number]

[0056] In Equation (17), if the Doppler phase fluctuation Ψ d is in the range smaller than π / 6, then "|1 - exp(j2Ψ d )| < 1", so the residual component of the transmission DC offset can be reduced more than in the case shown in <Example 3> (see Equation (12)). However, in <Example 4> as well, the residual component of the transmission DC offset cannot be completely canceled.

[0057] As described above, for the reception processing of radar reflected waves including Doppler fluctuations, coherent integration processing using Doppler frequency analysis is applied. For this reason, the noise level (floor level) of the Doppler frequency component including the remaining transmission DC offset component (the second term in Equation (17)) increases, and the radar detection performance deteriorates. For example, as described in <Example 4>, on the transmission side, codes A, -A, -A, A are transmitted every four radar transmission cycles, and on the reception side, correlation reception processing using the transmission code is performed on the radar reflected wave, and the output obtained by performing coherent integration processing every two radar transmission cycles with the transmission codes (A, -A) for canceling the reception DC offset as one unit is Doppler frequency-analyzed. In this case, the noise level (floor level) of the Doppler frequency component included in the remaining transmission DC offset component increases, and the radar detection performance deteriorates.

[0058] Here, in the Doppler frequency analysis, the reason why the noise level (floor level) of a specific frequency component increases due to the remaining transmission DC offset component is that the output of the coherent integration processing has a fixed phase fluctuation.

[0059] Therefore, in one aspect of the present disclosure, a random phase fluctuation is given to the output of the coherent integration processing with the transmission codes (A, -A) for canceling the reception DC offset as one unit so that the output of the coherent integration processing on the reception side does not have a fixed phase fluctuation.

[0060] Specifically, when performing coherent integration processing using Doppler frequency analysis on the reception side, a radar device according to one aspect of the present disclosure randomly switches whether to use a code with a phase inversion (a code with a phase change of π) added to each unit of transmission codes (corresponding to A, -A transmitted in two radar transmission cycles in the case of <Example 1> to <Example 4> above) for canceling the reception DC offset so that the Doppler phase fluctuation does not become a steady phase shift amount (phase change amount).

[0061] As a result, variations occur in the phase fluctuations of the output obtained by performing coherent integration processing on the radar reflected waves for each of the above-mentioned single-unit transmission codes, and the remaining transmitted DC offset component can be whitened in the Doppler frequency domain. Thereby, an increase in the noise level (floor level) of a specific Doppler frequency component can be prevented, and deterioration of the radar detection performance can be suppressed.

[0062] Hereinafter, embodiments according to one aspect of the present disclosure will be described in detail with reference to the drawings. In the embodiments, the same reference numerals are assigned to the same components, and the description thereof will be omitted since it is redundant.

[0063] [Embodiment 1] [Configuration of Radar Device] FIG. 1 is a block diagram showing the configuration of a radar device 10 according to the present embodiment.

[0064] The radar device 10 includes a radar transmission unit 100 and a radar reception unit 200.

[0065] The radar transmission unit 100 generates a high-frequency (radio frequency: RF) radar signal (radar transmission signal). Then, the radar transmission unit 100 transmits the radar transmission signal at a predetermined transmission period.

[0066] The radar reception unit 200 receives a reflected wave signal that is a radar transmission signal reflected by a measurement target. The radar reception unit 200 performs processing synchronized with the radar transmission unit using, for example, a reference signal (not shown). Further, the radar reception unit 200 may perform signal processing on the received reflected wave signal, for example, processing such as detection of the presence or absence of a target and direction estimation. The measurement target is an object to be detected by the radar device 10 and includes, for example, a vehicle (including four-wheel and two-wheel vehicles) or a person.

[0067] [Configuration of Radar Transmission Unit 100] The radar transmission unit 100 includes a radar transmission signal generation unit 101, a transmission radio unit 106, and a transmission antenna 107.

[0068] The radar transmission signal generation unit 101 generates a baseband radar transmission signal (pulse compression signal) obtained by modulating a code with a code length L every radar transmission period (T r ).

[0069] The radar transmission signal generation unit 101 operates based on a transmission reference clock obtained by multiplying a reference signal (not shown) by a predetermined multiple. Hereinafter, the transmission reference clock frequency is denoted as f TxBB . Here, the radar transmission period (T r ) is an integer N TxBB times (N TxBB × (1 / f r )) of the discrete time interval (1 / f r ) determined by the transmission reference clock frequency (f TxBB ) obtained by multiplying the reference signal by a predetermined multiple.

[0070] The radar transmission signal generation unit 101 includes a code generation unit 102, a phase rotation control unit 103, a transmission phase rotation unit 104, and a modulation unit 105.

[0071] Specifically, the code generation unit 102 generates a transmission code with a code length L every radar transmission period (T r ). Specifically, the code generation unit 102 generates a transmission code Code(m) with a code length L in the m-th radar transmission period.

[0072] Hereinafter, each element of the transmission code Code(m) is denoted as C n (m). That is, the transmission code Code(m) consists of L elements {C1(m), C2(m),..., C L (m)}. Also, the element C n (m) of the transmission code consists of two values such as {-1, 1} or four values such as {1, -1, -j, j}. Here, j is the imaginary unit. Also, n = 1, 2,..., L and m = 1, 2,..., Q. Here, Q represents the number of radar transmission periods used when the radar device 10 performs measurements such as distance, Doppler, and arrival direction.

[0073] As the transmission code, for example, it is preferable to apply a Barker code, a complementary code, an M-sequence code, a Gold code, etc. that can obtain low-range side lobe characteristics. Also, the transmission codes in each radar transmission cycle may be the same code or different codes. Alternatively, a plurality of transmission codes may be switched in each radar transmission cycle.

[0074] The phase rotation control unit 103 has a plurality of N e radar transmission cycles (= N e × T r ) as a unit, and within a plurality of (N e ) radar transmission cycles, it outputs a phase rotation amount signal for applying a phase rotation that is an integer multiple of 2π (2πN s ) to the transmission code to the transmission phase rotation unit 104 and the radar reception unit 200 (reception phase rotation unit 206). Here, N e is an integer greater than 1, and N s is an integer of 1 or more.

[0075] The phase rotation control unit 103 outputs a phase rotation amount signal that has a constant phase shift amount (phase change amount) across each radar transmission cycle within a period of a plurality of (N e ) radar transmission cycles (= N e × T r ). For example, the phase rotation control unit 103 outputs a phase rotation amount signal that periodically applies phase rotation amounts of φ×0, φ×1, φ×2, …, φ(N r -1) for each radar transmission cycle (T e ). Here, φ = 2πN s / N e . For example, when N e = 4 and N s = 1, the phase shift amount φ for each radar transmission cycle (T r ) is π / 2. In this case, the phase rotation amounts output in each of the 4 (= N e ) radar transmission cycles (T r ) are, for example, 0, π / 2, π, and 3π / 2.

[0076] When using complementary codes as transmission codes (including cases where multiple complementary codes are combined for transmission, such as in Spano codes), the phase shift amount for the code pairs that make up the complementary codes is set to zero. That is, the same phase rotation is applied to the code pairs that make up the complementary codes. This results in the effect of maintaining the high side lobe suppression characteristics of the complementary codes.

[0077] For example, when using complementary codes, the phase rotation control unit 103, for an even N e number of radar transmission periods (T r ), outputs a phase rotation amount signal that periodically applies phase rotation amounts of φ×0, φ×0, φ×1, φ×1, φ×2, φ×2, …, φ(N e -1), φ(N e -1). Here, φ = 2πN s / (N e / 2) = 4πN s / N e . For example, when N e = 8 and N s = 1, the phase shift amount φ is π / 2. In this case, the phase rotation amounts output in each of the 8 (= N e ) radar transmission periods (T r ) are, for example, 0, 0, π / 2, π / 2, π, π, 3π / 2, 3π / 2.

[0078] Also, when the phase shift amount φ is π / 2, the phase rotation can be realized by swapping the I signal component and the Q signal component (accompanied by sign conversion of positive and negative), so a multiplier can be made unnecessary in the transmission phase rotation unit 104.

[0079] Also, by setting N e / N s > 2 (in the case of complementary codes, N e / N s > 4), the effect of being able to cancel the DC offset component even when it has I signal and Q components can be obtained.

[0080] Furthermore, the phase rotation control unit 103, for a plurality of N e number of radar transmission periods (= Ne ×T r ) as a unit, a phase rotation amount variable signal that controls the phase rotation amount according to a predetermined phase variable pattern is output to the transmission phase rotation unit 104 and the radar reception unit 200 (reception phase rotation unit 206). That is, the phase rotation control unit 103 has a plurality of N e The period corresponding to the radar transmission cycle of times (= N e ×T r ) The pattern of the phase rotation amount with respect to the radar transmission signal within is changed every period (N e ×T r ).

[0081] Here, as the phase variable pattern, a random pattern that varies the phase rotation may be used. For example, the phase rotation control unit 103 uses a pseudo-random code (PN code), an M-sequence code, or a Gold code as the phase variable pattern, and controls the phase rotation amount according to the code polarity of each code of the phase variable pattern.

[0082] As an example, the phase variable pattern consists of N PP elements, and each element consists of two values of {-1, 1}. Hereinafter, each element of the phase variable pattern is denoted as PP(q). Here, q = 1, 2,..., N PP . The phase rotation control unit 103 sequentially reads out each element PP(q) of the phase variable pattern, and repeats the output of the value of the same element over N e radar transmission cycles (= N e ×T r ). That is, the phase rotation amount variable signal PC(m) output from the phase rotation control unit 103 in the m-th radar transmission cycle is represented by the following equation (18).

Equation

[0083] Here, m = 1,..., N e ×N d . Note that N d is a parameter defined in the Doppler analysis unit 208 described later. Note that the number of elements N PPis less than the number Q (= N e × N d ) of the radar transmission periods, the phase rotation control unit 103 cyclically reads out the phase variable pattern in Q radar transmission periods.

[0084] That is, the phase variable pattern is set such that each element PP(q) randomly changes every N e times of the radar transmission periods (T r ).

[0085] The transmission phase rotation unit 104 applies a phase rotation to the transmission code output from the code generation unit 102 based on the phase rotation amount signal and the phase rotation amount variable signal instructed by the phase rotation control unit 103. The transmission phase rotation unit 104 outputs the transmission code with the phase rotation applied to the modulation unit 105. For example, as shown in the following formula (19), the transmission phase rotation unit 104 outputs a signal GP(m) obtained by applying a phase rotation to the transmission code Code(m) output from the code generation unit 102 in the m-th radar transmission period.

Equation

[0086] In this way, the transmission phase rotation unit 104 applies a phase rotation to the transmission code (radar transmission signal) according to the phase variable pattern (PC(m)).

[0087] As described above, when the transmission code is a complementary code, by not performing a phase shift between the codes constituting the pair of complementary codes (applying the same phase rotation), an effect of canceling the distance side lobe between the codes constituting the pair of complementary codes (high side lobe suppression characteristic) can be obtained. That is, as shown in the following formula (20), when the transmission code is a complementary code, the transmission phase rotation unit 104 outputs a signal GP(m) obtained by applying a phase rotation to the transmission code Code(m) so that the phase shift within the two transmission periods in which the codes constituting the pair of complementary codes are transmitted becomes zero.

Equation

[0088] The modulation unit 105 performs pulse modulation (amplitude shift keying (ASK) or phase shift keying (PSK)) on the transmission code output from the transmission phase rotation unit 104, and outputs a modulated signal (radar transmission signal) to the transmission radio unit 106.

[0089] For example, when the modulation unit 105 uses phase shift keying (PSK), the phase modulation where the transmission code is binary such as {-1, 1} is BPSK, and the phase modulation where the transmission code is quaternary such as {1, -1, -j, j} is QPSK or four-phase PSK, and a predetermined modulation symbol on the IQ phase plane is assigned.

[0090] Further, the modulation unit 105 outputs a modulated signal restricted within a predetermined band by passing the modulated signal obtained by modulating the transmission code through a band-limiting filter (not shown).

[0091] Here, if the in-phase component of the modulated signal is represented as I(n s ) and the quadrature component is represented as Q(n s ), the modulated signal G(n s ) can be expressed as in the following equation (21).

Equation

[0092] Here, n s is a natural number representing discrete time. Also, the discrete time interval is (1 / f TxBB ), and f TxBB is the transmission reference clock frequency obtained by multiplying the reference signal by a predetermined multiple.

[0093] Further, the modulation unit 105 performs modulation on each transmitted code output from the transmission phase rotation unit 104 using No samples of the transmission reference clock per code. As a result, for the transmitted code with a code length L, in the radar signal section Tw, Nw = No × L samples are included. Also, in the no-signal section (T r ) in the radar transmission period (T r - Tw), Nu samples (= N r - Nw) of the transmission reference clock are included (for example, refer to FIG. 2). Therefore, the modulation signal in the m-th radar transmission period can be expressed as in the following equation (22).

Equation

[0094] The transmission radio unit 106 quadrature-modulates the signal output from the modulation unit 105, performs frequency conversion to generate a radar transmission signal in the carrier frequency (Radio Frequency: RF) band, amplifies it to a predetermined transmission power by a transmission amplifier, and outputs it to the transmission antenna 107. The transmission antenna 107 radiates the radar transmission signal output from the transmission radio unit 106 into space.

[0095] Note that a common reference signal is applied to the local oscillators of the transmission radio unit 106 and the reception radio unit 202 described later. Thereby, synchronization can be achieved between the local oscillators of the transmission radio unit 106 and the reception radio unit 202.

[0096] Also, the radar transmission unit 100 may include a radar transmission signal generation unit 101a shown in FIG. 3 instead of the radar transmission signal generation unit 101. The radar transmission signal generation unit 101a does not have the code generation unit 102, the phase rotation control unit 103, the transmission phase rotation unit 104, and the modulation unit 105 shown in FIG. 1, but instead includes a code storage unit 111 and a DA conversion unit 112. The code storage unit 111 stores in advance the code sequence generated in the code generation unit 102 (FIG. 1) and sequentially reads out the stored code sequence cyclically. The DA conversion unit 112 converts the code sequence (digital signal) output from the code storage unit 111 into an analog signal.

[0097] [Configuration of Radar Receiver Unit 200] In FIG. 1, the radar receiver unit 200 includes a receiving antenna 201, a receiving radio unit 202, and a signal processing unit 203.

[0098] The receiving antenna 201 receives a signal (reflected wave signal) in which the radar transmission signal in the RF band transmitted from the radar transmitter unit 100 is reflected by a reflecting object including a measurement target, and outputs the received reflected wave signal as a received signal to the receiving radio unit 202.

[0099] The receiving radio unit 202 amplifies the received signal output from the receiving antenna 201 to a predetermined level, converts the received signal in the high-frequency band to the baseband band in terms of frequency, and converts the received signal in the baseband band to a received signal in the baseband band including an I signal (In-Phase signal component) and a Q signal (Quadrature signal component).

[0100] The signal processing unit 203 includes an AD conversion unit 204, a correlation calculation unit 205, a received phase rotation unit 206, a coherent integration unit 207, and a Doppler analysis unit 208.

[0101] Note that each unit in the signal processing unit 203 operates based on a received reference clock obtained by multiplying a reference signal (not shown) by a predetermined factor. Hereinafter, the received reference clock frequency is denoted as f RxBB Let's assume. Here, the radar transmission period (T r ) is an integer N RxBB times (N RxBB ) of the discrete time interval (1 / f v ) determined by the received reference clock frequency (f v ) obtained by multiplying the reference signal by a predetermined factor. Also, hereinafter, the transmission reference clock frequency f RxBB ) is assumed to have a relationship f TxBB = f RxBB ×N TR with the received reference clock frequency f TxBB = f RxBB ×N TR ).

[0102] The AD conversion unit 204 samples the baseband signal including the I signal and the Q signal output from the receiving radio unit 202 at discrete time (1 / f RxBB ) based on the reception reference clock frequency f RxBB to convert the I signal and the Q signal into digital data.

[0103] In the following description, the baseband reception signal including the I signal and the Q signal at discrete time k is represented as a complex number signal x(k) = I r (k) + jQ r (k). Also, hereinafter, the discrete time k is based on the timing at which the m-th radar transmission cycle (T r ) starts (k = 1), and the signal processing unit 203 periodically performs measurements until the radar transmission cycle T r ends. That is, k = 1, …, N v . Here, j is the imaginary unit.

[0104] Therefore, the output signal of the AD conversion unit 204 in the m-th radar transmission cycle can be expressed as in the following equation (23). Hereinafter, X(k) is referred to as a complex baseband signal.

Equation

[0105] The correlation operation unit 205 performs a correlation operation between the complex baseband signal X(N r (m - 1) + k) output from the AD conversion unit 204 and the transmission code C v (m) transmitted in the radar transmission unit 100 for each radar transmission cycle T n . Here, n = 1, …, L. For example, the correlation operation value AC(k, m) of the sliding correlation operation at discrete time k in the m-th radar transmission cycle is calculated based on the following equation (24).

Equation

[0106] In Equation (24), the asterisk (*) represents the complex conjugate operator. Also, k = 1, …, N v is true.

[0107] Note that the correlation calculation unit 205 is not limited to performing correlation calculations for k = 1, …, N v and may limit the measurement range (i.e., the range of k) according to the existence range of the target to be measured by the radar device 10. Thereby, in the radar device 10, it is possible to reduce the amount of arithmetic processing of the correlation calculation unit 205. For example, the correlation calculation unit 205 calculates k = Nw / N TR + 1, …, (Nu - Nw) / N TR may limit the measurement range. In this case, the radar device 10 does not perform measurements in the time interval corresponding to the code transmission interval Tw.

[0108] Thereby, even when the radar transmission signal directly loops into the radar reception unit 200, the radar device 10 does not perform processing by the correlation calculation unit 205 during the period when the radar transmission signal loops, so that it is possible to perform measurements that exclude the influence of the loop. Also, when limiting the measurement range (the range of k), the same processing with the measurement range (the range of k) limited may be applied to the processing of the reception phase rotation unit 206, the coherent integration unit 207, and the Doppler analysis unit 208 described below. Thereby, the processing amount in each component can be reduced, and the power consumption in the radar reception unit 200 can be reduced.

[0109] The reception phase rotation unit 206 applies a phase rotation in the opposite direction (canceling direction) of the phase rotation applied by the transmission phase rotation unit 104 of the radar transmission unit 100, PC(m)exp[-j{(m - 1)modN r}φ], to the signal (correlation calculation value) output from the correlation calculation unit 205 every radar transmission period (T e ). That is, in the m-th radar transmission period (T r ), the reception phase rotation unit 206 outputs a signal ACP(k, m) with a phase rotation applied to the output AC(k, m) of the correlation calculation unit 205 as shown in the following Equation (25). [Number]

[0110] That is, the reception phase rotation unit 206 applies a phase rotation in the direction opposite to the phase rotation of the transmission phase rotation unit 104 to the output AC(k, m) (reflected wave signal) of the correlation operation unit 205 according to the phase variable pattern (PC(m)).

[0111] For example, N e = 4, N s = 1, when PC(m) = 1, the phase shift amount -φ per radar transmission period (T r ) is -π / 2. In this case, the phase rotation amounts output in each of the 4 (= N e ) radar transmission periods (T r ) are, for example, 0, -π / 2, -π, -3π / 2. Also, when N e = 4, N s = 1, PC(m) = -1, the phase shift amount -φ per radar transmission period (T r ) is -π / 2. However, when PC(m) = -1, due to the relationship exp(jπ) = -1, the phase rotation amounts output in each of the 4 (= N e ) radar transmission periods (T r ) are, for example, π, π / 2, 0, -π / 2. That is, the difference in the corresponding phase rotation amounts -φ (here {0, -π / 2, -π, -3π / 2} and {π, π / 2, 0, -π / 2}) between different phase variable patterns (PC(m) = 1, -1) is π each.

[0112] Note that when complementary codes are used as transmission codes, in order to make the phases of the code pairs constituting the complementary codes the same, the reception phase rotation unit 206 outputs the signal ACP(k, m) obtained by applying a phase rotation to the output AC(k, m) of the correlation operation unit 205 so that the phase shift within the transmission periods of the two codes constituting the complementary code becomes zero, as shown in the following equation (26).

Equation

[0113] For example, N e = 8, N s = 1, when PC(m) = 1, the phase shift amount -φ becomes -π / 2. For 8 (= N e ) radar transmission cycles (T r ), the phase rotation amounts output in each of them are, for example, 0, 0, -π / 2, -π / 2, -π, -π, -3π / 2, -3π / 2. Also, N e = 8, N s = 1, when PC(m) = -1, the phase shift amount -φ becomes -π / 2. However, when PC(m) = -1, due to the relationship exp(jπ) = -1, for 8 (= N e ) radar transmission cycles (T r ), the phase rotation amounts output in each of them are, for example, π, π, π / 2, π / 2, 0, 0, -π / 2, -π / 2.

[0114] The coherent integrator 207 uses the correlation operation value ACP(k, m) output from the reception phase rotation unit 206 for each discrete time k of the m-th radar transmission cycle, and adds (coherent integration) the correlation operation value ACP(k, m) for each discrete time k over a period of a predetermined number N e of radar transmission cycles, and calculates the coherent integration value ACC(k, v) for each discrete time k. Here, k = 1, …, N v is satisfied.

[0115] Specifically, the v-th coherent integration value ACC(k, v) is calculated as shown in the following equation (27).

Equation

[0116] Here, the period of the radar transmission cycle (that is, the integration section of the coherent integrator 207) in which the correlation operation value ACP(k, v) is added for each discrete time k in the coherent integrator 207 is a predetermined number N eBy setting it as such, even when the received DC offset is included in the correlation operation value ACP(k, v), based on the following principle, the received DC offset component can be canceled, and together with the suppression of the noise component, the deterioration of the radar detection performance due to the received DC offset can be prevented.

[0117] That is, for the received DC offset component α Rx , and the transmitted DC offset component α dTx including the Doppler frequency fluctuation f Tx exp(j2πf dTx ×T r ), when it is included, the output AC(k, m) of the correlation operation unit 205 includes the component shown in the following equation (28) regardless of k and m.

Equation

[0118] Also, the output ACP(k, m) of the reception phase rotation unit 206 includes the component shown in the following equation (29) regardless of k.

Equation

[0119] Therefore, the output ACC(k, v) of the coherent integration unit 207 includes the component shown in the following equation (30) regardless of k.

Equation

[0120] As shown in equation (30), in the integration interval of the coherent integration unit 207,

Equation

Equation

[0121] On the other hand, when the Doppler frequency fluctuation f dTx is not zero, that is, when the reflected wave signal includes a Doppler frequency fluctuation, the output ACC(k, v) of the coherent integration unit 207 includes a transmitted DC offset component shown in the following Equation (31) regardless of k.

Equation

[0122] The phase rotation control unit 103 controls to randomly change the phase variable pattern PC(N e (that is, the integration interval of the coherent integration unit 207) to 1 or -1 for each predetermined number N e (v - 1)+m). As a result, in the output of the Doppler analysis unit 208, the remaining transmitted DC offset component included in the output ACC(k, v) of the coherent integration unit 207 shown in Equation (31) is whitened in the Doppler frequency domain. Therefore, an increase in the noise level (floor level) of a specific Doppler frequency component can be prevented, and deterioration of the radar detection performance can be suppressed.

[0123] The Doppler analysis unit 208 performs Doppler frequency analysis on the output signal of the coherent integration unit 207. Specifically, the Doppler analysis unit 208 performs Doppler frequency analysis with the timing of the discrete time k aligned, using the N d outputs ACC(k, 1) to ACC(k, N d ) of the coherent integration unit 207 obtained for each discrete time k as one unit. As described above, the N d outputs ACC(k, 1) to ACC(k, N dSince the remaining transmitted DC offset components included in each of are whitened in the Doppler frequency domain, in the Doppler frequency analysis by the Doppler analysis unit 208, it is possible to suppress an increase in the noise level (floor level) of a specific frequency component due to the transmitted DC offset component.

[0124] Specifically, as shown in the following equation (32), the Doppler analysis unit 208 performs coherent integration after correcting the phase fluctuations Ψ(f f ΔΨ) corresponding to 2N s different Doppler frequencies f s ) = 2πf s (T r ×N e ).

Equation

[0125] Here, FT_CI(k, f s ) represents the coherent integration result of the Doppler frequency f s ΔΨ at the discrete time k of the Doppler analysis unit 208. Note that f s = -N f +1,..., 0,..., N f and k = 1,..., (N r +N u )N s / N o , and ΔΨ is the phase rotation unit.

[0126] As a result, the signal processing unit 203 obtains, for each discrete time k, the coherent integration results FT_CI(k, -N f +1),..., FT_CI(k, N f -1) corresponding to 2N f Doppler frequency components during each period of T r in the radar transmission cycle, which is a multiple of N e ×N d (T r ×N e ×N d = T r ×Q). Note that j is the imaginary unit.

[0127] ΔΨ=1 / N d In this case, the processing of the Doppler analysis unit 208 is performed at a sampling interval (T r ×N e ), sampling frequency 1 / (T r ×N e ) is equivalent to subjecting the output of coherent integrator 207 to discrete Fourier transform (DFT).

[0128] Also, N f By setting N to a power of 2, the Doppler analysis unit 208 can apply Fast Fourier Transform (FFT) processing, thereby reducing the amount of calculation processing. f >N d So, q>N d In the domain where ACC(k,N d By performing zero padding such that (w-1)+q+1)=0, FFT processing can be similarly applied, thereby reducing the amount of calculation processing.

[0129] The radar device 10 receives the output FT_CI(k, f s ) to estimate the range / Doppler frequency (relative velocity) of the radar measured target.

[0130] That is, the output FT_CI(k, f s ) squared absolute value |FT_CI(k, f s )| 2 is the Doppler frequency f s This corresponds to the reflected wave reception level from the target for each time. Based on the discrete time k at which the noise level reaches a peak power value equal to or greater than a predetermined value and the Doppler f, the radar device 10 calculates the distance R(k) to the target and the relative velocity v based on the Doppler frequency. d (f s ) can be estimated as follows: s =-N f +1,..,0,...,N f and k=1,…, (N r +N u )Ns / N o is as follows.

[0131] When converting the time information k into the distance information R(k), the following equation (33) is used. Here, Tw represents the symbol transmission interval, L represents the pulse code length, and C0 represents the speed of light.

Equation

[0132] Also, for converting the Doppler frequency information f s into the relative velocity component v d (f s ), the following equation (34) is used. Here, λ is the wavelength of the carrier frequency of the radar transmission signal in the RF band output from the transmission radio unit 106.

Equation

[0133] The results of confirming the effects of the operations of the phase rotation control unit 103, the transmission phase rotation unit 104, and the reception phase rotation unit 206 described above using computer simulation are shown below.

[0134] Figures 4A and 4B show the output (received level) of the Doppler analysis unit 208 under the condition that one measurement target moves away from the radar device 10 at a speed of 20 km / h in the direction away from the 5 m point.

[0135] In addition, in Figures 4A and 4B, it is assumed that there is a transmission DC offset (carrier leakage) in the transmission radio unit 106. Also, Figures 4A and 4B show, as the radar transmission code, a complementary code (code length L = 64), the number of additions N e = 32 of the coherent integration unit 207, and the number of samples N d = 512 in the Doppler analysis unit 208, showing the results of computer simulation.

[0136] Further, FIG. 4A shows the result when the phase inversion (PC(m)) is not randomly switched as in the conventional case, unlike the operation of the radar device 10 according to the present embodiment. On the other hand, FIG. 4B shows the result when the phase rotation control unit 103 randomly switches the phase inversion by the phase variable pattern PC(m) as described above.

[0137] In both FIGS. 4A and 4B, sharp peaks appear at the coordinates where the speed is 20 km / h at a distance of 5 m, and it can be seen that the desired measurement target is detected.

[0138] However, in FIG. 4A, in addition to the peak of the desired measurement target at a distance of 5 m and a speed of 20 km / h, peaks with weak reception levels appear uniformly in all distance ranges (0 to 200 m) at the Doppler frequency component of 20 km / h. These peaks with weak reception levels are frequency components that do not originally exist and are factors causing false detection.

[0139] On the other hand, in FIG. 4B showing the result based on the operation according to the present embodiment, in addition to the peak of the desired measurement target at a distance of 5 m and a speed of 20 km / h, there is no phenomenon in which peaks with weak reception levels appear uniformly in all distance ranges at a specific Doppler frequency component as detected in FIG. 4A. That is, according to the present embodiment, it can be confirmed that there is no deterioration in radar performance that causes false detection even under the condition that there is a transmission DC offset (carrier leak) in the transmission radio unit 106.

[0140] As described above, the radar device 10 has a plurality of N e cycles of radar transmission period (T r ) corresponding period (N e × T r ) for the phase rotation amount of the radar transmission signal (transmission code) within the phase variable pattern PC(m), and within the period (N e × T r)A phase rotation control unit 103 that changes each time, a transmission phase rotation unit 104 that applies phase rotation (first phase rotation) to the radar transmission signal according to a phase variable pattern, and a reflected wave signal (correlation operation value ACC(k,m) in FIG. 1), and a reception phase rotation unit 206 that applies a phase rotation in the opposite direction to the first phase rotation according to a phase variable pattern.

[0141] That is, by the operations of the phase rotation control unit 103, the transmission phase rotation unit 104, and the reception phase rotation unit 206, the radar device 10 uses a plurality of transmission codes that cancel the received DC offset within a period (N e ×T r ) as one unit, and randomly switches whether to make a code with a phase inversion added to each unit of the transmission code (a code with a π phase change).

[0142] As a result, in the coherent integration unit 207 of the radar device 10, the output of the coherent integration process does not become a fixed phase fluctuation, and the remaining transmitted DC offset component is whitened in the Doppler frequency domain. Therefore, it is possible to prevent the noise level (floor level) of a specific Doppler frequency component from rising in the output of the Doppler analysis unit 208, and it is possible to suppress the deterioration of the radar detection performance.

[0143] Therefore, according to the present embodiment, it is possible to suppress the deterioration of the radar detection performance due to circuit errors.

[0144] Further, according to the present embodiment, the radar device 10 can prevent the deterioration of the radar detection performance caused by the transmitted DC offset without having a circuit configuration for correcting the transmitted DC offset (carrier leak) with high accuracy, so the configuration of the radar device 10 can be simplified.

[0145] In FIG. 1, the case where the reception phase rotation unit 206 of the radar device 10 is arranged at the subsequent stage of the correlation operation unit 205 is shown. However, as shown in FIG. 5, even if the reception phase rotation unit 206 is arranged at the preceding stage of the correlation operation unit 205, the same results as in the above embodiment can be obtained.

[0146] [Embodiment 2] Since the radar device according to this embodiment has the same basic configuration as the radar device 10 according to Embodiment 1, it will be described with reference to FIG. 1.

[0147] In this embodiment, as shown in FIG. 6, a case where a plurality of radar devices 10 according to Embodiment 1 (FIG. 1) are provided will be described. Hereinafter, as an example, a case where two radar devices A and B shown in FIG. 6 are provided will be described.

[0148] In this embodiment, in a plurality of radar devices 10, the phase-variable patterns controlled by the phase rotation control unit 103 of each radar device 10 are made different from each other. For example, in the radar device A shown in FIG. 6, the phase-variable pattern PP (1) (q) (where q = 1, 2,..., N pp ) is set, and in the radar device B, a phase-variable pattern PP (1) different from PP (2) (q) (where q = 1, 2,..., N pp ) is set.

[0149] As an example, the phase rotation control unit 103 of each radar device 10 may set different phase-variable patterns by shifting the transmission timing of the same M-sequence code. For example, the phase-variable pattern PP (1) (q) of the radar device A is a pattern using an M-sequence code with a code length of 511 (where q = 1,..., 511). In this case, the phase-variable pattern PP (2) (q) of the radar device B may be a pattern obtained by shifting the transmission timing of the M-sequence code used in the radar device A by N shift only. That is, PP (2) (q) = PP (1) (q + N shift ).

[0150] For example, in FIG. 6, a radar receiving unit 200 of a radar device A receives a received signal (desired signal), which is a reflected wave of a radar transmission signal transmitted by the radar device A, and a reflected wave (interference signal) of a radar transmission signal transmitted by a radar device B.

[0151] Even in this case, as described above, by varying the phase-variable pattern between the radar devices A and B, the transmitted DC offset components remaining in the received signals from the respective radar devices A and B are whitened in the Doppler frequency domain, similar to the first embodiment. Therefore, it is possible to prevent the noise level (floor level) of a specific Doppler frequency component from increasing, and it is possible to suppress the deterioration of the radar detection performance of the radar device A. That is, an effect of reducing the mutual interference between a plurality of radar devices 10 in which the same frequency band or a part of the frequency bands overlap can be obtained.

[0152] FIG. 7 shows the results of computer simulation evaluation regarding the amount of mutual interference when the radar device A and the radar device B transmit radar transmission signals using different phase-variable patterns.

[0153] In FIG. 7, the radar device A and the radar device B use complementary codes as transmission codes, and the phase shift in the phase rotation control unit 103 is N e = 16, N s = 2. Also, N d = 512 is used in the Doppler analysis units 208 of the radar device A and the radar device B.

[0154] The horizontal axis of FIG. 7 indicates the time shift amount of the transmission timing of the phase-variable pattern in units of the radar transmission period (T r ), and the vertical axis indicates the received power of the radar device A (Desired Power [plotted with black circles]), the received power of the radar device B (Undesired Power [plotted with x marks]), and the signal-to-interference power ratio (SIR [plotted with white circles]) of the radar device A.

[0155] From the simulation results shown in FIG. 7, as the phase variable patterns of radar device A and radar device B, at the transmission timing of the M-sequence code, if there is a time shift in the transmission timing of 16 (= N e ) or more radar transmission cycles, it can be confirmed that the SIR of radar device A is improved by about 20 dB.

[0156] The phase rotation control unit 103 sequentially reads out each element PP(q) of the phase variable pattern, and for N e radar transmission cycles (= N e ×T r ), the values of the same element are repeatedly output. From this, if N shift ≧1, there is a time shift in the transmission timing of 16 (= N e ) or more radar transmission cycles between different phase variable patterns, and it can be seen that the interference suppression effect can be enhanced. Note that the improvement amount of the SIR depends on the code length N pp of the code used in the phase variable pattern and the parameter N d used in the Doppler analysis unit 208 (for example, N d = 512). Therefore, from the viewpoint of SIR improvement, it is more preferable to use as long N p and N d as possible within the allowable measurement time.

[0157] As described above, in the present embodiment, the phase variable patterns are different from each other among the plurality of radar devices 10. For example, the transmission timings of the same M-sequence code used in the phase variable patterns of each of the plurality of radar devices 10 are shifted. By doing so, the mutual interference between the plurality of radar devices 10 can be reduced.

[0158] Note that in the above case, since the transmission timings of the plurality of radar devices 10 may accidentally coincide, there is a possibility that a case where the mutual interference cannot be suppressed probabilistically may occur. On the other hand, the mutual interference can be reduced probabilistically by the following method.

[0159] Specifically, in the plurality of radar devices 10, the code length N ppAmong the M-sequence codes, Preferred pairs with low cross-correlation are used for each phase-variable pattern.

[0160] Or, for the code length N pp Among the M-sequence codes, it is known that N pp different Gold codes can be generated from Preferred pairs with low cross-correlation. Therefore, in a plurality of radar devices 10, Gold codes generated from Preferred pairs with low cross-correlation may be used for each phase-variable pattern.

[0161] Or, in a plurality of radar devices 10, mutual interference may be probabilistically reduced by randomly varying each phase-variable pattern for each radar measurement (or for every predetermined number of measurements).

[0162] [Embodiment 3] In Embodiment 2, a method of reducing mutual interference between a plurality of radar devices 10 by making the phase-variable patterns different between the plurality of radar devices 10 was described. In contrast, in the present embodiment, a configuration of a MIMO (Multiple Input Multiple Output) radar using a plurality of transmit-receive antennas instead of the plurality of radar devices 10 will be described.

[0163] That is, in the present embodiment, the phase-variable patterns are different from each other between the plurality of transmit antennas (that is, between the MIMO streams).

[0164] FIG. 8 is a block diagram showing a configuration example of the radar device 20 according to the present embodiment. In FIG. 8, the same components as those in Embodiment 1 (FIG. 1) are denoted by the same reference numerals, and the description thereof is omitted.

[0165] The radar device 20 shown in FIG. 8 shows the configuration of a time-division MIMO radar that switches a plurality of transmission antennas in time division to transmit different radar transmission signals that are time-division multiplexed, and separates each radar transmission signal to perform reception processing. Note that the configuration of the radar device is not limited to this, and the radar device 20 may be configured to send out different transmission signals that are frequency-division multiplexed or code-division multiplexed from a plurality of transmission antennas, separate each transmission signal, and perform reception processing.

[0166] [Configuration of Radar Transmission Unit 300] The radar transmission unit 300 of the radar device 20 includes a radar transmission signal generation units 101-1 to 101-Nt, a switching control unit 301, a transmission switching unit 302, and a transmission array antenna unit 303.

[0167] The transmission array antenna unit 303 is composed of Nt transmission antennas (Tx#1 to Tx#Nt).

[0168] The radar transmission signal generation units 101-1 to 101-Nt are provided corresponding to the Nt transmission antennas (Tx#1 to Tx#Nt), respectively. Each radar transmission signal generation unit 101 performs the same operation as in Embodiment 1 (FIG. 1). However, the phase variable patterns set for each radar transmission signal generation unit 101 are different from each other.

[0169] For example, the phase variable pattern in each of the Nt radar transmission signal generation units 101 is set by shifting the transmission timing of the same M-sequence code by one or more code elements.

[0170] Specifically, let the phase variable pattern in the radar transmission signal generation unit 101-1 be PP (1) (q) (where q = 1,..., N pp ). In this case, let the phase variable pattern in the radar transmission signal generation unit 101-2 be PP (2) (q + N shift ), and let the phase variable pattern in the radar transmission signal generation unit 101-3 be PP (3) (q + 2N shift)(q+(Nt-1)N), and similarly hereinafter, the phase variable pattern in the radar transmission signal generation unit 101-Nt is set to PP (Nt) (q+(Nt-1)N shift ) may be used. However, N shift ≧1.

[0171] Note that, as the phase variable pattern for each of the Nt radar transmission signal generation units 101, a Preferred pair with low cross-correlation among the M-sequence codes of code length N pp may be used. Alternatively, among the M-sequence codes of code length N pp , it is known that N pp different Gold codes can be generated from the Preferred pair with low cross-correlation. Therefore, the Gold codes generated from the Preferred pair with low cross-correlation may be used for the phase variable pattern of each of the Nt radar transmission signal generation units 101. Alternatively, the phase variable pattern of each of the Nt radar transmission signal generation units 101 may be randomly varied for each radar measurement (or for each predetermined number of measurements) to probabilistically reduce mutual interference.

[0172] The switching control unit 301 outputs a control signal (hereinafter referred to as a switching control signal) for instructing the switching timing of the transmission antennas (Tx#1 to Tx#Nt) of the transmission array antenna unit 303 (that is, the output switching of the radar transmission signal) to the transmission switching unit 302 and the radar reception unit 400 (correlation calculation unit 205 and output switching unit 403).

[0173] Based on the switching control signal from the switching control unit 301, the transmission switching unit 302 selects one of the Nt transmission antennas of the transmission array antenna unit 303, and among the outputs of the Nt radar transmission signal generation units 101, the output signal of the radar transmission signal generation unit 101 corresponding to the selected transmission antenna is used as the input to the selected transmission antenna. The transmission switching unit 302 frequency-converts the output signal (baseband radar transmission signal) of the selected radar transmission signal generation unit 101 to a predetermined radio frequency band and outputs it to the selected (connected) transmission antenna.

[0174] The transmission array antenna unit 303 radiates the radar transmission signal output from the transmission switching unit 302 into space from the transmission antenna selected (connected) to the transmission switching unit 302.

[0175] Hereinafter, the control operation of the transmission switching unit 302 by the switching control unit 301 will be described. The control operation of the radar receiving unit 400 by the switching control unit 301 will be described later in the operation description of the radar receiving unit 400.

[0176] The switching control unit 301 e outputs a switching control signal to the transmission switching unit 302 to sequentially switch the radar transmission signal generation unit 101 and the transmission antenna every N radar transmission cycles.

[0177] For example, in the first N e radar transmission cycles (N e ×T r ), the output signal of the radar transmission signal generation unit 101-1 is used as the input of the transmission switching unit 302, and the signal converted into a high-frequency signal is output to the transmission antenna (Tx#1) of the transmission array antenna unit 303.

[0178] The switching control unit 301 e in the next N e radar transmission cycles (N r ×T), uses the output signal of the radar transmission signal generation unit 101-2 as the input of the transmission switching unit 302, and outputs the signal converted into a high-frequency signal to the transmission antenna (Tx#2) of the transmission array antenna unit 303.

[0179] The switching control unit 301 repeats the same operation, uses the output signal of the radar transmission signal generation unit 101-Nt as the input of the transmission switching unit 302, and outputs the signal converted into a high-frequency signal to the transmission antenna (Tx#Nt) of the transmission array antenna unit 303.

[0180] Also, in the next N e radar transmission cycles (N e ×T rIn [0], again, the output signal of the radar transmission signal generation unit 101-1 is used as the input to the transmission switching unit 302, and the signal converted into a high-frequency signal is output to the transmission antenna (Tx#1) of the transmission array antenna unit 303.

[0181] The switching control unit 301 repeats the above operations a predetermined number of times (N d ×N t times).

[0182] In the above operations, the radar transmission unit 300 operates so as to sequentially read out GP(m) (for example, refer to Equation (19) or Equation (20)) described in Embodiment 1 as the signal generated by each radar transmission signal generation unit 101. Here, m = 1, …, N e ×N d is satisfied.

[0183] [Configuration of Radar Reception Unit 400] The radar reception unit 400 of the radar device 20 includes a reception array antenna unit 401, an antenna system processing unit 402, and a direction estimation unit 404.

[0184] The reception array antenna unit 401 is composed of Na reception antennas (Rx#1 to Rx#Na). The Na reception antennas respectively receive the signal (reflected wave signal) obtained by reflecting the radar transmission signal transmitted from the radar transmission unit 300 by a reflecting object including the radar measurement target. Each signal received by the Na reception antennas is input as a reception signal to the antenna system processing unit 402 corresponding to each reception antenna (Rx#1 to Rx#Na).

[0185] Each antenna system processing unit 402 includes a reception radio unit 202 and a signal processing unit 203.

[0186] The reception radio unit 202 of the z-th antenna system processing unit 402-z amplifies the reception signal from the z-th reception antenna (Rx#z) to a predetermined level, converts the reception signal in the high-frequency band into a reception signal in the baseband band, and converts the reception signal in the baseband band into a reception signal in the baseband band including an I signal and a Q signal. Here, z = 1, …, Na.

[0187] The signal processing unit 203 of the z-th antenna system processing unit 402-z is composed of an A / D conversion unit 204, a correlation operation unit 205, a reception phase rotation unit 206, an output switching unit 403, a coherent integration unit 207, and a Doppler analysis unit 208. Note that the signal processing unit 203 includes Nt coherent integration units 207 and Doppler analysis units 208 respectively corresponding to the transmission antennas (Tx#1 to Tx#Nt).

[0188] Hereinafter, for each component of the signal processing unit 203 of the z-th antenna system processing unit 402-z, mainly the operations different from those in Embodiment 1 will be described.

[0189] The correlation operation unit 205 performs a correlation operation between the complex baseband signal X(N v (m - 1)+k) (for example, refer to Equation (23)) output from the AD conversion unit 204 and the transmission code generated in the radar transmission signal generation unit 101 selected every N e times of the radar transmission period (N e ×T r ) by the switching control unit 301.

[0190] The reception phase rotation unit 206 applies a phase rotation in the opposite direction (canceling direction) to the phase rotation applied by the transmission phase rotation unit 103 of the radar transmission signal generation unit 101 selected by the switching control unit 301 every N e times of the radar transmission period (N e ×T r ) to the signal (correlation operation value) output from the correlation operation unit 205.

[0191] The output switching unit 403 switches the output to the coherent integration units 207-1 to 207-Nt respectively corresponding to the transmission antenna numbers (#1 to #Nt) selected by the switching control unit 301 every N e times of the radar transmission period (N e ×T r ).

[0192] For example, when the transmission antenna (Tx#1) is selected by the switching control unit 301, the output switching unit 403 switches the signal from the reception phase rotation unit 206 to the coherent integration unit 207-1 corresponding to the transmission antenna (Tx#1) and outputs it.

[0193] Also, when the transmission antenna (Tx#2) is selected by the switching control unit 301, the output switching unit 403 switches the signal from the reception phase rotation unit 206 to the coherent integration unit 207-2 corresponding to the transmission antenna (Tx#2) and outputs it.

[0194] The output switching unit 403 repeats the same operation. When the transmission antenna (Tx#Nt) is selected by the switching control unit 301, the output switching unit 403 switches the signal from the reception phase rotation unit 206 to the coherent integration unit 207-Nt corresponding to the transmission antenna (Tx#Nt) and outputs it.

[0195] The N D th coherent integration unit 207-N D performs coherent integration processing in units of a plurality of N e periods (N e ×T r ) for the output of the reception phase rotation unit 206 selected every N e radar transmission cycles (N r ×N e ). Here, N D = 1,…, Nt.

[0196] The Doppler analysis unit 208 performs Doppler frequency analysis on the output signal of the coherent integration unit 207. That is, the Doppler analysis unit 208 performs Doppler frequency analysis by aligning the timing at discrete time k using the N d outputs from the coherent integration unit 207 obtained at each discrete time k.

[0197] As described above, in the radar device 20, for a predetermined number of times N e(That is, for each integration period of the coherent integration unit 207), the transmission antenna is switched by the switching control unit 301, and different phase variable patterns PC(m) are set between the transmission antennas. As a result, in the output of the Doppler analysis unit 208 corresponding to each transmission antenna, similar to the first embodiment, the remaining transmitted DC offset component included in the output ACC(k,v) (see Equation (31)) of the coherent integration unit 207 for the radar transmission signal transmitted from each transmission antenna is whitened in the Doppler frequency domain. Therefore, it is possible to prevent an increase in the noise level (floor level) of a specific Doppler frequency component and suppress the deterioration of the radar detection performance. Furthermore, by setting different phase variable patterns between the transmission antennas, it is possible to reduce the mutual interference between the radar transmission signals transmitted by each transmission antenna.

[0198] In the following description, the w-th output FT_CI (z) (1) (k, fs, w) , …, FT_CI (z) (Na) (k, fs, w) obtained by performing the same processing in each of the antenna system processing units 402-1 to 402-Na from the antenna system processing unit 402-1 are collectively expressed as a virtual reception array correlation vector h(k, fs, w) as shown in the following equations (35) and (36). The virtual reception array correlation vector h(k, fs, w) includes Nt×Na elements, which is the product of the number of transmission antennas Nt and the number of reception antennas Na. The virtual reception array correlation vector h(k, fs, w) is used in the description of the process of performing direction estimation based on the phase difference between the reception antennas for the reflected wave signal from the target, which will be described later. Here, z = 1, …, Nt, N D = 1, …, Na. Note that f s = -N f + 1, …, 0, …, N f .

Equation

Equation

[0199] The above describes the processing in each component of the signal processing unit 203 of the antenna system processing unit 402-z.

[0200] For the virtual array correlation vector h(k, fs, w) from the w-th Doppler analysis unit 208 at each discrete time k output from the antenna system processing units 402-1 to 402-Na, the direction estimation unit 404 corrects the phase shift deviation and amplitude deviation between the transmission antennas of the transmission array antenna unit 303 and between the reception antennas of the reception array antenna unit 401 with the array correction value h_cal [b] By multiplying, the virtual received array correlation vector h _after_cal (k, fs, w) that corrects the antenna deviation is calculated. The virtual received array correlation vector h _after_cal (k, fs, w) is represented by the following equation (37). Here, b = 1,.., (Nt × Na).

Equation

[0201] The virtual received array correlation vector h _after_cal (k, fs, w) that corrects the antenna deviation is a column vector consisting of Na × N r elements. Hereinafter, each element of the virtual received array correlation vector h _after_cal (k, fs, w) is denoted as h1(k, fs, w), …, h Na×Nr (k, fs, w) and is used in the description of the direction estimation process.

[0202] Then, the direction estimation unit 404 performs a direction estimation process based on the phase difference of the reflected wave signals between the reception antennas using the virtual received array correlation vector h _after_cal (k, fs, w).

[0203] The direction estimation unit 404 is the direction estimation evaluation function value P HThe spatial profile is calculated by making the azimuth direction θ in (θ, k, fs, w) variable within a predetermined angular range, and a predetermined number of maximum peaks of the calculated spatial profile are extracted in descending order, and the azimuth direction of the maximum peak is used as the estimated arrival direction value.

[0204] Note that the evaluation function value P H (θ, k, fs, w) has various types depending on the arrival direction estimation algorithm. For example, the estimation method using an array antenna disclosed in Reference Non-Patent Document 1 may be used.

[0205] (Reference Non-Patent Document 1) Direction-of-arrival estimation using signal subspace modeling Cadzow, J.A.; Aerospace and Electronic Systems, IEEE Transactions on Volume: 28, Issue: 1 Publication Year: 1992, Page(s): 64 - 79

[0206] For example, the beamformer method can be expressed as the following equations (38) and (39). Other methods such as Capon and MUSIC can also be applied in the same way.

Number

Number

[0207] Here, the superscript H is the Hermitian transpose operator. Also, a H (θ u ) represents the direction vector of the virtual receiving array for the incoming wave in the azimuth direction θ u . Also, θ u is changed at a predetermined azimuth interval β1 within the azimuth range for performing arrival direction estimation. For example, θ u is set as follows. θ u = θmin + uβ1, u = 0, …, NU NU = floor[(θmax - θmin) / β1] + 1

[0208] Here, floor(x) is a function that returns the largest integer value not exceeding the real number x.

[0209] The configuration of the radar receiving unit 400 has been described above.

[0210] As described above, in the present embodiment, in the radar apparatus 20 which is a MIMO radar, by varying the phase variable pattern for the radar transmission signals transmitted by each transmission antenna, it is possible to reduce the mutual interference between the radar transmission signals transmitted by switching a plurality of transmission antennas.

[0211] Also, in the present embodiment, since the mutual interference between the radar transmission signals transmitted by switching a plurality of transmission antennas can be reduced, the time interval for switching the plurality of transmission antennas can be narrowed, and the detection time can also be shortened.

[0212] The embodiments according to one aspect of the present disclosure have been described above.

[0213] Note that the above-described embodiments and the operations according to the respective variations may be appropriately combined and implemented.

[0214] [Other Embodiments] (1) In the above embodiment, the radar apparatuses 10 and 20 may be provided with a circuit configuration for simply correcting the transmission DC offset. Thereby, the radar apparatuses 10 and 20 can further suppress an increase in the noise level caused by the transmission DC offset by using in combination the configuration for controlling the phase variable pattern described above and the configuration for correcting the transmission DC offset component.

[0215] (2) In the above embodiment, the case of using a coded pulse radar has been described, but the present disclosure is also applicable to a radar system using a frequency-modulated pulse wave such as a chirp pulse radar.

[0216] (3) In the radar devices 10 and 20 shown in FIGS. 1, 5, 6, and 8, the radar transmission units 100 and 300 and the radar reception units 200 and 400 may be individually arranged at physically separated locations.

[0217] (4) Although not shown in the drawings, the radar devices 10 and 20 have, for example, a CPU (Central Processing Unit), a storage medium such as a ROM (Read Only Memory) storing a control program, and a working memory such as a RAM (Random Access Memory). In this case, the functions of the above-described respective units are realized by the CPU executing the control program.

[0218] As described above, various embodiments have been described with reference to the drawings. Needless to say, the present disclosure is not limited to such examples. It is obvious that those skilled in the art can conceive of various modification examples or correction examples within the scope described in the claims, and it is naturally understood that those also belong to the technical scope of the present disclosure. Also, within the scope not departing from the gist of the disclosure, the respective components in the above embodiments may be arbitrarily combined.

[0219] In the above embodiments, the present disclosure has been described as an example configured using hardware. However, the present disclosure can also be realized by software in cooperation with hardware.

[0220] Also, each functional block used in the description of the above embodiments is typically realized as an LSI which is an integrated circuit. The integrated circuit may control each functional block used in the description of the above embodiment and may include an input terminal and an output terminal. These may be individually formed into one chip, or may be formed into one chip so as to include a part or all of them. Here, it is described as an LSI, but depending on the degree of integration, it may also be referred to as an IC, a system LSI, a super LSI, or an ultra LSI.

[0221] In addition, the method of integrating circuits is not limited to LSI, and it may also be realized using an application-specific circuit or a general-purpose processor. After manufacturing the LSI, an FPGA (Field Programmable Gate Array) that can be programmed, or a reconfigurable processor that can reconfigure the connection or setting of circuit cells inside the LSI may be used.

[0222] Furthermore, if a technology for integrating circuits that replaces LSI appears due to the progress of semiconductor technology or another derived technology, of course, functional blocks may be integrated using that technology. The application of biotechnology or the like is a possible example.

[0223] <Summary of the present disclosure> The radar device of the present disclosure is a radar device including a radar transmitter that transmits a radar signal at a predetermined transmission period, and a radar receiver that receives a reflected wave signal obtained by reflecting the radar signal by a target, wherein the radar transmitter changes, for each period, a pattern of a phase rotation amount to be applied to the radar signal within a period corresponding to a plurality of times Ne (Ne is an integer greater than 1) of the transmission period, the phase rotation amount is obtained by dividing an integer multiple of 2π by the plurality of times Ne, the pattern of the phase rotation amount is obtained by multiplying the phase rotation amount by a code polarity of a predetermined code sequence, and includes a phase rotation control unit, and a transmission phase rotation unit that applies a first phase rotation to the radar signal according to the pattern of the phase rotation amount, and the radar receiver includes a reception phase rotation unit that applies a second phase rotation in a direction opposite to the first phase rotation to the reflected wave signal according to the pattern of the phase rotation amount.

[0224] In the radar device of the present disclosure, when the code polarity of the predetermined code sequence is +1, the phase change of the phase rotation amount is 0, and when the code polarity of the predetermined code sequence is -1, the phase change of the phase rotation amount is π.

[0225] In the radar device of the present disclosure, the predetermined code sequence is based on a pseudo-random code, an M-sequence code, or a Gold code.

[0226] In the radar device of the present disclosure, when generating the radar signal using complementary codes, the transmission phase rotation unit applies the same phase rotation amount within two transmission periods for transmitting the pair of codes constituting the complementary codes, and applies the phase rotation amount changed every two transmission periods.

[0227] In the radar device of the present disclosure, a predetermined code sequence is different among a plurality of radar devices.

[0228] In the radar device of the present disclosure, the radar transmission unit includes a plurality of transmission antennas, and the predetermined code sequence is different among the transmission antennas.

[0229] The radar method of the present disclosure changes a pattern of a phase rotation amount applied to a radar signal within a period corresponding to a plurality of transmission periods Ne (Ne is an integer greater than 1) for each period, applies a first phase rotation to the radar signal according to the pattern of the phase rotation amount, transmits the radar signal to which the first phase rotation has been applied at the transmission period, receives a reflected wave signal obtained by reflecting the radar signal to which the first phase rotation has been applied by a target, and applies a second phase rotation in a direction opposite to the first phase rotation to the reflected wave signal according to the pattern of the phase rotation amount. The phase rotation amount is obtained by dividing an integer multiple of 2π by a plurality of Ne times, and the pattern of the phase rotation amount is obtained by multiplying the phase rotation amount by the code polarity of a predetermined code sequence.

Industrial Applicability

[0230] The present disclosure is suitable as a radar device for detecting a wide-angle range.

Explanation of Signs

[0231] 10, 20 Radar device 100, 300 Radar transmission unit 101 Radar transmission signal generation unit 102 Code generation unit 103 Phase rotation control unit 104 Transmission Phase Rotation Unit 105 Modulation Unit 106 Transmission Radio Unit 107 Transmission Antenna 111 Code Memory Unit 112 DA Conversion Unit 200, 400 Radar Receiver 201 Receiver Antenna 202 Receiver Radio Unit 203 Signal Processing Unit 204 AD Conversion Unit 205 Correlation Calculation Unit 206 Receiver Phase Rotation Unit 207 Coherent Integration Unit 208 Doppler Analysis Unit 301 Switching Control Unit 302 Transmission Switching Unit 303 Transmission Array Antenna Unit 401 Receiver Array Antenna Unit 402 Antenna System Processing Unit 403 Output Switching Unit 404 Direction Estimation Unit

Claims

1. A signal generation unit that generates a transmission signal; A transmission phase rotation unit that applies phase rotation based on a plurality of different phase rotation amounts controlled by a series of a plurality of mutually different phase variable patterns to the transmission signal for each transmission period; A multiplexing unit that applies multiplexing processing to a plurality of different radar transmission signals that are the transmission signals to which the phase rotation is applied; A radar transmission radio unit that outputs the plurality of different radar transmission signals to which the multiplexing processing is applied to a plurality of transmission antennas; A radar reception radio unit that inputs, via a plurality of reception antennas, one or more reflected wave signals obtained by reflecting at least one of the plurality of different radar transmission signals from a target; Comprising: The plurality of different phase rotation amounts are a series obtained by sequentially adding, for each of a plurality of transmission periods, a shift amount that is an integer multiple of 2 or more of 2π; The plurality of mutually different phase variable patterns have elements for each of the plurality of transmission periods, and are patterns that vary the phase rotation amount for each of the plurality of transmission periods; A radar signal processing circuit.

2. The signal generation unit outputs a signal series that constitutes the transmission signal; The plurality of different phase rotation amounts include a phase rotation amount with a shift amount of zero with respect to one signal constituted by the signal series; The radar signal processing circuit according to Claim 1.

3. The multiplexing unit outputs, for each of the plurality of transmission periods, one of the plurality of different radar transmission signals to one antenna selected from the plurality of transmission antennas; The radar signal processing circuit according to Claim 1.

4. The multiplexing unit simultaneously outputs the plurality of different radar transmission signals to the plurality of transmission antennas for each of the plurality of transmission periods; The radar signal processing circuit according to Claim 1.

5. Generate a transmission signal, Apply phase rotation based on a plurality of different phase rotation amounts controlled by a series of a plurality of mutually different phase variable patterns to the transmission signal for each transmission period, Apply multiplexing processing to a plurality of different radar transmission signals that are the transmission signals to which the phase rotation is applied, Output the plurality of different radar transmission signals to which the multiplexing processing is applied to a plurality of transmission antennas, Input, via a plurality of reception antennas, one or more reflected wave signals obtained by reflecting at least one of the plurality of different radar transmission signals from a target The plurality of different phase rotation amounts is a series obtained by sequentially adding a shift amount that is an integer multiple of 2π or more for each of a plurality of transmission periods, The plurality of mutually different phase variable patterns have elements for each of the plurality of transmission periods, and are patterns for varying the phase rotation amount for each of the plurality of transmission periods. Radar signal processing method.

6. The transmission signal has a signal series, The plurality of different phase rotation amounts include a phase rotation amount with a shift amount of zero for one signal constituted by the signal series. The radar signal processing method according to claim 5.

7. The multiplexing process outputs one radar transmission signal selected from the plurality of different radar transmission signals to one antenna selected from the plurality of transmission antennas for each of the plurality of transmission periods. The radar signal processing method according to claim 5.

8. The multiplexing process simultaneously outputs the plurality of different radar transmission signals to the plurality of transmission antennas respectively for each of the plurality of transmission periods. The radar signal processing method according to claim 5.

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