Radar apparatus and radar signal processing method

Abstract

To improve detection accuracy of a target in a radar device.SOLUTION: The radar device includes a transmission antenna unit including a plurality of transmission antenna elements arranged at different positions, and a transmission circuit that multiplexes and transmits a transmission signal of a first linearly polarized wave and a transmission signal of a second linearly polarized wave different from the first linearly polarized wave in each transmission cycle from the transmission antenna unit, the transmission circuit applies a phase rotation amount in which a phase between the transmission signal of the first linearly polarized wave and the transmission signal of the second linearly polarized wave is ξ or -ξ, the transmission antenna unit radiates the transmission signal of the first linearly polarized wave and the transmission signal of the second linearly polarized wave from different positions, A transmission signal of a first linearly polarized wave is radiated from a first position of the different positions, and a transmission signal of a second linearly polarized wave is radiated from a second position and a third position of the different positions, the first position is a position sandwiched between the second position and the third position, and the first linearly polarized wave and the second linearly polarized wave have a same center frequency.SELECTED DRAWING: Figure 1

Description

[Technical Field]

[0001] The present disclosure relates to a radar device. [Background technology]

[0002] In recent years, radar devices using short-wavelength radar transmission signals, including microwaves and millimeter waves, which can provide high resolution, have been studied. Furthermore, to improve outdoor safety, there is a demand for the development of radar devices that can detect small objects, such as pedestrians, over a wide angle range (e.g., called wide-angle radar devices).

[0003] A radar device having a wide detection range may be configured to receive reflected waves from a target using an array antenna composed of multiple antennas (also called antenna elements), and estimate the direction of arrival (also called the angle of arrival) of the reflected waves using a signal processing algorithm based on the received phase difference relative to the element spacing (antenna spacing) (Direction of Arrival (DOA) estimation). Examples of DOA estimation methods include the Fourier method, or methods that can achieve high resolution, such as the Capon method, MUSIC (Multiple Signal Classification), and ESPRIT (Estimation of Signal Parameters via Rotational Invariance Techniques).

[0004] Furthermore, a radar device has been proposed that includes, for example, a receiver and a transmitter that are equipped with multiple antennas (array antennas), and that performs beam scanning by signal processing using the transmit and receive array antennas (sometimes referred to as MIMO (Multiple Input Multiple Output) radar) (see, for example, Non-Patent Document 1). [Prior art documents] [Patent documents]

[0005] [Patent Document 1] Special Publication No. 2011-526371 [Patent Document 2] U.S. Patent No. 9,541,638 [Patent Document 3] U.S. Patent Publication No. 2019 / 0064337 [Patent Document 4] U.S. Patent Publication No. 2020 / 0363497 [Patent Document 5] Japanese Patent Publication No. 2020-148754 [Non-patent literature]

[0006] [Non-Patent Document 1] J. Li, and P. Stoica, "MIMO Radar with Colocated Antennas", Signal Processing Magazine, IEEE Vol. 24, Issue: 5, pp. 106-114, 2007 [Non-patent document 2] M. Kronauge, H. Rohling, "Fast two-dimensional CFAR procedure", IEEE Trans. Aerosp. Electron. Syst., 2013, 49, (3), pp. 1817-1823 [Non-patent document 3] Direction-of-arrival estimation using signal subspace modeling Cadzow, JA; Aerospace and Electronic Systems, IEEE Transactions on Volume: 28 , Issue: 1 Publication Year: 1992 , Page(s): 64 - 79 Summary of the Invention [Problem to be solved by the invention]

[0007] However, methods for detecting targets in radar devices (for example, MIMO radars) have not been fully studied.

[0008] Non-limiting examples of the present disclosure contribute to providing a radar device that improves target detection accuracy. [Means for solving the problem]

[0009] A radar device according to one embodiment of the present disclosure includes a plurality of transmitting antennas including a first transmitting antenna that radiates a first linearly polarized wave and a second transmitting antenna that is adjacent to the first transmitting antenna and radiates a second linearly polarized wave different from the first linearly polarized wave, and a transmitting circuit that multiplexes and transmits, from the plurality of transmitting antennas, transmission signals that have been given a phase rotation amount that causes a phase difference of φ or −φ between the first transmitting antenna and the second transmitting antenna in each transmission period.

[0010] These comprehensive or specific embodiments may be realized as a system, an apparatus, a method, an integrated circuit, a computer program, or a recording medium, or may be realized as any combination of a system, an apparatus, a method, an integrated circuit, a computer program, and a recording medium. [Effects of the Invention]

[0011] According to an embodiment of the present disclosure, it is possible to improve the detection accuracy of a target in a radar device.

[0012] Further advantages and benefits of an embodiment of the present disclosure will become apparent from the specification and drawings. Such advantages and / or benefits may be provided by some of the embodiments and features described in the specification and drawings, respectively, but not necessarily all of them may be provided to obtain one or more identical features. [Brief explanation of the drawings]

[0013] [Figure 1] Block diagram showing an example of the configuration of a radar device [Figure 2]FIG. 1 is a diagram showing an example of a transmission signal when a chirp pulse is used. [Figure 3] FIG. 10 is a diagram showing an example of allocation of Doppler shift amounts and orthogonal codes. [Figure 4] FIG. 10 is a diagram showing an example of allocation of Doppler shift amounts and orthogonal codes. [Figure 5] FIG. 10 is a diagram showing an example of allocation of Doppler shift amounts and orthogonal codes. [Figure 6] FIG. 1 shows an example of a transmitting antenna. [Figure 7] FIG. 1 shows an example of a transmitting antenna. [Figure 8] FIG. 1 shows an example of a transmitting antenna. [Figure 9] A diagram showing an example of the directivity of left-handed circularly polarized waves [Figure 10] A diagram showing an example of the directivity of right-handed circularly polarized waves [Figure 11] FIG. 1 shows an example of a transmission signal and a reception signal when a chirp pulse is used. [Figure 12] FIG. 10 is a diagram showing an example of Doppler region compression processing. [Figure 13] FIG. 10 is a diagram showing an example of allocation of Doppler shift amounts and orthogonal codes. [Figure 14] FIG. 10 is a diagram showing an example of Doppler aliasing determination; [Figure 15] Diagram showing an example of antenna placement [Figure 16] Diagram showing an example of antenna placement [Figure 17] Diagram showing an example of antenna placement [Figure 18] Diagram showing an example of antenna placement [Figure 19] Block diagram showing an example of the configuration of a radar device [Figure 20] Diagram showing an example of antenna placement [Figure 21] Block diagram showing an example of the configuration of a radar device [Figure 22] FIG. 10 is a diagram showing an example of switching control of a transmission signal; [Figure 23] FIG. 10 is a diagram showing an example of switching control of a transmission signal; [Figure 24] FIG. 10 is a diagram showing an example of a transmission switching control table. [Figure 25]Block diagram showing an example of the configuration of a radar device [Figure 26] FIG. 10 is a diagram showing an example of a Doppler multiplexing allocation table. [Figure 27] FIG. 10 is a diagram showing an example of a Doppler multiplexing allocation table. [Figure 28] FIG. 10 is a diagram showing an example of a Doppler multiplexing allocation table. DETAILED DESCRIPTION OF THE INVENTION

[0014] A MIMO radar transmits signals (radar transmission waves) multiplexed using, for example, time division, frequency division, or code division from multiple transmission antennas (also called transmission array antennas), receives signals (radar reflected waves) reflected by surrounding objects using multiple reception antennas (also called reception array antennas), and separates and receives the multiplexed transmission signals from each received signal. Through this processing, the MIMO radar performs array signal processing on these received signals as a virtual reception array.

[0015] In addition, in MIMO radar, by appropriately arranging the element spacing in the transmitting and receiving array antennas, it is possible to enlarge the antenna aperture of the virtual receiving array and improve the angular resolution.Alternatively, in MIMO radar, by arranging the antenna spacing of the virtual receiving array more closely, it is possible to suppress side lobes or grating lobes.

[0016] For example, Patent Document 1 discloses a MIMO radar (hereinafter referred to as "time division multiplexing MIMO radar") that uses time division multiplexing transmission, which transmits signals by shifting the transmission time for each transmitting antenna, as a multiple transmission method for MIMO radar. The time division multiplexing MIMO radar outputs transmission pulses, which are an example of transmission signals, while sequentially switching the transmitting antennas at a specified cycle. The time division multiplexing MIMO radar receives signals that are the transmission pulses reflected by an object using multiple receiving antennas, and after correlation processing between the received signals and the transmission pulses, performs, for example, spatial FFT (Fast Fourier Transform) processing (processing to estimate the arrival direction of the reflected wave).

[0017] A time-division multiplexing MIMO radar sequentially switches between transmitting antennas that transmit transmission signals (e.g., transmission pulses or radar transmission waves) at a specified period. This allows the time-division multiplexing MIMO radar to extract a channel response represented by the product (=Nt×Na) of the number of transmitting antennas Nt and the number of receiving antennas Na, and performs array signal processing on these (Nt×Na) received signals as a virtual receiving array. For example, it is difficult to use transmitting antennas that exceed the number of transmitting antennas (e.g., the time-division multiplexing number) that multiplex the transmission signals by switching them in a time-division manner. For example, when a radar device transmits transmission signals at a time-division multiplexing number Nt using Nt transmitting antennas, it is difficult to extract more than (Nt×Na) channel responses. Therefore, when the number of antennas is limited due to constraints such as the cost or installation location of the radar device, the angular resolution or sidelobe suppression effect may be limited, and angle measurement performance may not be improved.

[0018] Next, as an example, attention will be focused on a method of simultaneously multiplexing and transmitting transmission signals from a plurality of transmission antennas.

[0019] As a method for simultaneously multiplexing and transmitting transmission signals from multiple transmitting antennas, there is, for example, a method of transmitting signals so that the multiple transmission signals can be separated in the Doppler frequency domain at the receiving unit (hereinafter referred to as "Doppler multiplexing transmission") (for example, see Non-Patent Document 2).

[0020] In Doppler multiplexing, for example, a transmitter applies different Doppler shifts to transmission signals transmitted from a reference transmitting antenna and from a transmitting antenna other than the reference transmitting antenna, and the transmission signals are simultaneously transmitted from multiple (e.g., Nt) transmitting antennas. In Doppler multiplexing, signals received using multiple (e.g., Na) receiving antennas are filtered in the Doppler frequency domain, so that the transmission signals transmitted from each transmitting antenna are separated and received. As a result, a MIMO radar using Doppler multiplexing (hereinafter referred to as a "Doppler multiplexing MIMO radar") can extract a propagation path response expressed as the product (=Nt×Na) of the number of transmitting antennas Nt and the number of receiving antennas Na, and performs array signal processing on these (Nt×Na) received signals as a virtual receiving array. For example, it is difficult to use transmitting antennas that exceed the number of transmitting antennas used for Doppler multiplexing (e.g., the Doppler multiplexing number). For example, when a radar device transmits a transmission signal using Nt transmission antennas with a Doppler multiplexing number of Nt, it is difficult to extract more than (Nt×Na) propagation path responses.

[0021] Another method for simultaneously multiplexing and transmitting transmission signals from multiple transmitting antennas is code multiplexing (see, for example, Patent Document 2). For example, a MIMO radar using code multiplexing (hereinafter referred to as "code multiplexing MIMO radar") repeatedly applies phase modulation based on a code sequence (hereinafter also referred to as a code or a code sequence) that differs for each transmitting antenna to each repeated transmission of a transmission signal (e.g., a chirp signal), and performs code multiplexing transmission from multiple (e.g., Nt) transmitting antennas. Furthermore, the code multiplexing MIMO radar extracts distance information from the code-multiplexed received signal, for example, by detecting signals received using multiple (e.g., Na) receiving antennas. Furthermore, the code multiplexing MIMO radar divides the distance information obtained for each repeated transmission of the transmission signal into M parts and performs Fourier transform processing in the velocity direction (M, for example, uses the code length of the code sequence). Code-multiplexed MIMO radar separates the code-multiplexed received signals by applying phase correction based on the detected velocity components to the results of Fourier transform processing in the M velocity directions and multiplying them by an inverse code sequence that separates the code sequence assigned to each transmitting antenna.

[0022] With this configuration of the code-multiplexing MIMO radar, for example, even when the relative velocity between the target and the code-multiplexing MIMO radar is not zero, the code-multiplexing MIMO radar can suppress mutual interference between code-multiplexed received signals and separate the code-multiplexed received signals. As a result, the code-multiplexing MIMO radar can extract a propagation path response represented by the product (=Nt×Na) of the number of transmitting antennas Nt and the number of receiving antennas Na, and performs array signal processing on these (Nt×Na) received signals as a virtual receiving array. For example, it is difficult to use transmitting antennas that exceed the number of transmitting antennas used for code-multiplexing transmission (e.g., the code multiplexing number). For example, when a radar device transmits transmission signals using Nt transmitting antennas with a code multiplexing number Nt, it is difficult to extract more than (Nt×Na) propagation path responses.

[0023] Furthermore, there is a technology for improving radar detection or discrimination performance by using an antenna that emits radio waves of different polarizations or an antenna that receives radio waves of different polarizations (see, for example, Patent Documents 3 or 4). A radar device that uses multiple polarized waves is also called, for example, a "polarimetric radar."

[0024] For example, Patent Document 3 and Patent Document 4 disclose a method for detecting and identifying an object by transmitting a transmission signal from an antenna using vertical polarization or horizontal polarization and using the signal received by the antenna using vertical polarization or horizontal polarization. Patent Document 4 also discloses a method for detecting and identifying an object by transmitting a transmission signal from an antenna using left-handed circular polarization or right-handed circular polarization and using the signal received by an antenna using left-handed circular polarization or right-handed circular polarization. Antennas that use linear polarization such as vertical polarization or horizontal polarization, or circular polarization such as left-handed circular polarization or right-handed circular polarization are also called "polarized antennas."

[0025] Such polarized radar improves the radar's detection or identification performance while using multiple polarized antennas of different types. For example, to transmit four types of polarized waves, including vertical and horizontal polarizations as well as left-handed and right-handed circular polarizations, four transmitting antennas are used. Furthermore, if a MIMO radar is configured for each polarization, more transmitting antennas are used. For example, to perform MIMO multiplexing using Nt transmitting antennas for each of the four polarizations, 4×Nt transmitting antennas are used. For example, in one embodiment of the present disclosure, a radar device (also referred to as a polarized radar device or a polarized MIMO radar) generates new polarized waves (e.g., circularly polarized waves) by combining polarized antennas, thereby performing multiplexing using more polarized waves with fewer transmitting antennas. As a result, the radar device of one embodiment of the present disclosure can suppress an increase in the number of transmitting antennas and utilize more virtual receiving antennas, thereby improving the angle measurement performance of the radar device and improving target detection accuracy.

[0026] Hereinafter, an embodiment according to an example of the present disclosure will be described in detail with reference to the drawings. In the embodiment, the same components are denoted by the same reference numerals, and redundant descriptions thereof will be omitted.

[0027] The following describes a configuration (e.g., MIMO radar configuration) in which a radar device transmits different multiplexed transmission signals simultaneously from multiple transmission antennas in a transmission branch, and a reception branch separates the transmission signals and performs reception processing.

[0028] In the following, as an example, a configuration of a radar system using a frequency-modulated pulse wave such as a chirp pulse (also called fast chirp modulation) will be described. However, the modulation system is not limited to frequency modulation. For example, an embodiment of the present disclosure can also be applied to a radar system using a pulse compression radar that transmits a pulse train after phase modulation or amplitude modulation.

[0029] The radar device also performs, for example, Doppler multiplexing. Furthermore, the radar device encodes (for example, CDM (Code Division Multiplexing)) signals to which different phase rotations (for example, phase shifts) corresponding to the number of Doppler multiplexes are applied (hereinafter referred to as "Doppler multiplexed transmission signals") and multiplexes the signals (hereinafter referred to as "Coded Doppler Multiplexing").

[0030] [Radar device configuration] The radar device 10 in FIG. 1 includes a radar transmitter (transmitting branch) 100, a radar receiver (receiving branch) 200, and a positioning output unit 300.

[0031] The radar transmitter 100 generates a radar signal (radar transmission signal) and transmits the radar transmission signal at a specified transmission period (hereinafter referred to as the "radar transmission period") using a transmission array antenna configured with multiple transmission antennas 109 (e.g., Nt antennas).

[0032] The radar receiver 200 receives reflected wave signals, which are radar transmission signals reflected by targets (not shown), using a receiving array antenna including multiple receiving antennas 202-1 to 202-Na. The radar receiver 200 performs signal processing on the reflected wave signals received by each receiving antenna 202, for example, to detect the presence or absence of a target or estimate the arrival distance, Doppler frequency (e.g., relative velocity), and arrival direction of the reflected wave signals, and outputs information related to the estimation results (e.g., positioning information).

[0033] The positioning output unit 300 performs positioning output processing based on information relating to the estimated result of the arrival direction input from the radar receiving unit 200.

[0034] The radar device 10 may be mounted on a moving body such as a vehicle, and the positioning output from the positioning output unit 300 (e.g., information on the estimation result) may be connected to a control device ECU (Electronic Control Unit) (not shown) such as an Advanced Driver Assistance System (ADAS) that improves collision safety or an autonomous driving system, and used for vehicle drive control or alarm call control.

[0035] The radar device 10 may be attached to a relatively high structure (not shown), such as a roadside utility pole or a traffic light. The radar device 10 may be used, for example, as a sensor in an assistance system for improving the safety of passing vehicles or pedestrians or in a system for preventing the intrusion of suspicious individuals (not shown). The positioning output of the radar receiving unit 200 may be connected to a control device (not shown) in the assistance system for improving safety or in the system for preventing the intrusion of suspicious individuals, and used for alarm generation control or abnormality detection control. The uses of the radar device 10 are not limited to these, and the radar device 10 may be used for other purposes.

[0036] Furthermore, a target is an object to be detected by the radar device 10, and includes, for example, a vehicle (including two-wheeled and four-wheeled vehicles), a person, a block, or a curb.

[0037] [Configuration of radar transmitter 100] The radar transmitter 100 includes a radar transmission signal generator 101 , a phase rotation amount setting unit 105 , a phase rotator 108 , and a transmission antenna 109 .

[0038] The radar transmission signal generation unit 101 generates a radar transmission signal. The radar transmission signal generation unit 101 includes, for example, a transmission signal generation control unit 102, a modulation signal generation unit 103, and a VCO (Voltage Controlled Oscillator) 104. Each component of the radar transmission signal generation unit 101 will be described below.

[0039] The transmission signal generation control unit 102 sets, for example, the timing of generating a transmission signal for each radar transmission period, and outputs information about the set timing of generating a transmission signal to the modulation signal generating unit 103 and the phase rotation amount setting unit 105 (for example, the Doppler shift setting unit 106). Here, the radar transmission period is defined as "Tr."

[0040] The modulation signal generating unit 103 periodically generates, for example, a sawtooth modulation signal based on information relating to the timing of transmission signal generation for each radar transmission period Tr input from the transmission signal generation control unit 102 .

[0041] Based on the modulation signal input from the modulation signal generating unit 103, the VCO 104 outputs a frequency modulation signal (hereinafter, referred to as a frequency chirp signal or chirp signal, for example) to the phase rotation unit 108 and the radar receiving unit 200 (mixer unit 204, described later) as a radar transmission signal (radar transmission wave) as shown in FIG. 2, for example.

[0042] The phase rotation amount setting unit 105 sets the amount of phase rotation (e.g., the amount of phase rotation corresponding to coded Doppler multiplexing) to be applied to the radar signal for each radar transmission period Tr in the phase rotation unit 108, based on information relating to the timing of transmission signal generation for each radar transmission period Tr input from the transmission signal generation control unit 102. The phase rotation amount setting unit 105 includes, for example, a Doppler shift setting unit 106 and an encoding unit 107.

[0043] The Doppler shift setting unit 106 sets the amount of phase rotation corresponding to the amount of Doppler shift to be applied to the radar transmission signal (for example, a chirp signal) based on, for example, information relating to the timing of transmission signal generation for each radar transmission cycle Tr.

[0044] The encoding unit 107 sets the amount of phase rotation corresponding to the encoding, for example, based on information regarding the timing of transmission signal generation for each radar transmission cycle Tr. The encoding unit 107 calculates the amount of phase rotation for the phase rotation unit 108, for example, based on the amount of phase rotation input from the Doppler shift setting unit 106 and the amount of phase rotation corresponding to the encoding, and outputs the calculated amount of phase rotation to the phase rotation unit 108. The encoding unit 107 also outputs information regarding the code sequence (for example, each element of the orthogonal code sequence) used for the encoding to the radar receiving unit 200 (for example, the output switching unit 209).

[0045] The coded Doppler multiplexing number for the Doppler multiplexed signal set by coding section 107 does not have to depend on the amount of phase rotation (amount of Doppler shift) of each transmitting antenna 109 set by phase rotation section 108. For example, even if phase rotation section 108 sets the same amount of phase rotation (amount of Doppler shift) for a pair of adjacent transmitting antennas 109, coding section 107 may set the coded Doppler multiplexing number to the same value or to different values.

[0046] Phase rotation section 108 applies the amount of phase rotation input from encoding section 107 to the chirp signal input from VCO 104, and outputs the phase-rotated signal to transmitting antenna 109. For example, phase rotation section 108 includes a phase shifter, a phase modulator, etc. (not shown).

[0047] The output signal of the phase rotation unit 108 is amplified to a specified transmission power and radiated into space from each transmitting antenna 109. For example, a radar transmission signal is multiplexed and transmitted from the multiple transmitting antennas 109 by adding a phase rotation amount corresponding to the Doppler shift amount and the orthogonal code sequence.

[0048] Next, an example of a method for setting the amount of phase rotation in phase rotation setting section 105 will be described.

[0049] The Doppler shift setting unit 106 determines the amount of Doppler shift DOP ndm The phase rotation amount φ for adding ndm and output to the encoding unit 107. Here, ndm=1, ∼, N DM N DM is the number of different Doppler shift amounts that can be set, and is hereinafter referred to as the "Doppler multiplex number."

[0050] In the radar device 10, since encoding by the encoding unit 107 is also performed, the Doppler multiplexing number N DM may be set to be smaller than the number Nt of transmitting antennas 109 used for multiplex transmission. DM must be 2 or more.

[0051] Doppler shift amount DOP1, DOP2, ~, DOP N_DM ("N_DM" is "N DM For example, the Doppler shift amounts DOP1, DOP2, . . . , DOP may be set to be equal intervals, or may be set to be uneven intervals. N_DM In order to use the encoding by the encoding unit 107 described later, for example, 0≦DOP1, DOP2, . . . , DOP N_DM <1 / (TrL oc ) may be set to satisfy the Doppler shift amounts DOP1, DOP2, ~, DOP N_DM may be set to satisfy, for example, equation (1).

number

[0052] Also, for example, the Doppler shift amounts DOP1, DOP2, ~, DOP N_DM The minimum Doppler shift interval Δf MinInterval The Doppler shift intervals may satisfy the following equation (2). N_DM Here, Loc represents the number of code elements. For example, Loc represents the code length of the code used in the encoding unit 107. In the following, Loc=2 is used as an example (an example will be described later).

number

[0053] In addition, each Doppler shift amount DOP1, DOP2, ~, DOP N_DM The phase rotation amount φ for adding ndm may be assigned as shown in the following equation (3), for example.

number

[0054] In addition, the intervals are equal to Δf MinInterval When the Doppler shift amount is set to be equal to the Doppler shift amount DOP ndm The phase rotation amount φ for adding ndm is assigned, for example, as shown in the following equation (4).

number

[0055] The minimum Doppler shift interval Δf MinIntervalThe narrower the interval between the Doppler multiplexed signals, the more likely it is that interference between the Doppler multiplexed signals will occur, and the more likely it is that the target detection accuracy will be reduced (for example, deteriorated). Therefore, it is preferable to widen the interval between the Doppler shift amounts within the range that satisfies the constraints of equation (2). For example, when the equality sign is established in equation (2) (for example, Δf MinInterval =1 / (T r N DM L OC )) can maximize the interval between Doppler multiplexed signals in the Doppler region (hereinafter referred to as "maximum equal interval Doppler shift amount setting"). In this case, the Doppler shift amounts DOP1, DOP2, ..., DOP N_DM The phase rotation range is from 0 to less than 2π. DM Each signal is equally divided into 100 and a different phase rotation amount is assigned to each of them. For example, the Doppler shift amount DOP ndm The phase rotation amount φ for adding ndm is assigned as shown in the following equation (5): Note that angles are expressed in radians hereinafter.

number

[0056] In equation (5), for example, the Doppler multiplexing number N DM = 2, the phase rotation amount φ1 for imparting the Doppler shift amount DOP1 is 0, and the phase rotation amount φ2 for imparting the Doppler shift amount DOP2 is π. Similarly, in equation (5), for example, when the Doppler multiplexing number N DM In the case of φ=4, the phase rotation amount φ1 to impart the Doppler shift amount DOP1 is 0, the phase rotation amount φ2 to impart the Doppler shift amount DOP2 is π / 2, the phase rotation amount φ3 to impart the Doppler shift amount DOP3 is π, and the phase rotation amount φ4 to impart the Doppler shift amount DOP4 is 3π / 2. For example, ndm The phase rotation amount φ ndm are equally spaced.

[0057] The Doppler shift amounts DOP1, DOP2, ~, DOP N_DMThe allocation of the amount of phase rotation to be applied is not limited to this allocation method. For example, the allocation of the amount of phase rotation shown in equation (5) may be shifted. For example, φ ndm =2π(ndm) / N DM Alternatively, the phase rotation amounts may be assigned as follows: Alternatively, a phase rotation amount assignment table may be used to assign the Doppler shift amounts DOP1, DOP2, ..., DOP NDM The phase rotation amounts φ1, φ2,~, φ N_DM (However, "N_DM" is N DM (corresponding to

[0058] In addition, when setting the amount of Doppler shift at equal intervals, the amount of phase rotation φ ndm If the denominator of is set to an integer and the phase rotation amount is set to an integer value in degree units, the phase rotation amount can be easily set. For example, Δf MinInterval =1 / (T r (N DM +N int )L OC ), the phase rotation amount φ shown in equation (4) is ndm The denominator of the equation (6) is set to an integer value. DM +N int ) is a factor of 360 int When this is set, the amount of phase rotation is set to an integer value, making it easy to set the amount of phase rotation.

number

[0059] where N int takes an integer value greater than or equal to 0. For example, N DM If = 7, N int When =1 is set, φ ndm =2π(ndm-1) / (N DM +N int )=π(ndm-1) / 4, φ1, φ2,.., φ N_DMare integer values ​​in degree units such as 0°, 45°, 90°, 135°, ..., 270°, making it easy to set the amount of phase rotation.

[0060] In addition, in equation (6), N int If = 0, the maximum equidistant Doppler shift amount is set.

[0061] The encoding unit 107 receives the N DM The phase rotation amount φ1,~,φ that gives the Doppler shift amount N_DM For each of the above, one or N CM The encoder 107 sets a phase rotation amount based on a plurality of orthogonal code sequences (not more than 100). The encoder 107 also sets a phase rotation amount based on both the Doppler shift amount and the orthogonal code sequences, for example, an "encoded Doppler phase rotation amount" for generating an encoded Doppler multiplexed signal, and outputs the result to the phase rotation unit 108.

[0062] An example of the operation of the encoding unit 107 will now be described.

[0063] For example, the encoding unit 107 may select the number of codes (for example, the number of code multiplexing) N CM orthogonal code sequences are used.

[0064] Below, we use N CM The orthogonal code sequences are ncm ={OC ncm (1), OC ncm (2), ~, OC ncm (Loc)}. OC ncm (noc) is the ncmth orthogonal code sequence Code ncm represents the noc-th code element in,where noc is the index of the code element, and noc=1,∼,Loc.

[0065] In this embodiment, for example, the number of codes (for example, the number of code multiplexing) N CM For example, an orthogonal code sequence with code length Loc=2 may be used.CM As orthogonal code sequences for noc=2, Code1={OC1(1), OC1(2)} and Code2={OC2(1), OC2(2)} may be used. Each code element takes a real or complex value. For example, when the noc-th code element OC1(noc) in the first code is used as a reference, the noc-th code element OC2(noc) in the second code may use a code element with a phase difference of +90° when noc=1 and -90° when noc=2. Alternatively, a code element with a phase difference of -90° when noc=1 and +90° when noc=2 may be used.

[0066] For example, N CM As orthogonal code sequences for .times. ... 2 =-1).

[0067] In general terms, the orthogonal code sequence used in this embodiment may be expressed as Code2={j×A, −j×B} or Code2={−j×A, j×B} for the first code Code1={A, B}. These codes are orthogonal to each other, and when the noc-th code element OC1(noc) in the first code is used as a reference, the noc-th code element OC2(noc) in the second code is a code element that differs in phase by +90° when noc=1 and −90° when noc=2, or by −90° when noc=1 and +90° when noc=2. Here, A and B are real or complex numbers, and the absolute values ​​of A and B are equal, for example, |A|=|B|.

[0068] Here, when the noc-th code element OC1(noc) in the first code is used as a reference, the noc-th code element OC2(noc) in the second code is a code element that has a phase difference of +90° when noc=1 and -90° when noc=2, or a phase difference of -90° when noc=1 and +90° when noc=2, but the phase difference is not limited to a phase difference of ±90°.

[0069] For example, a phase difference of ±ξ may be applied to the noc-th code element in the first code and the second code. ξ may be in the range of π / 6 to 5π / 6 radians (=30° to 150°). For example, when the phase deviation between the transmitting antennas 109 is corrected in advance, a circularly polarized wave whose main beam direction of the transmitted beam changes depending on ξ is generated by beam transmission, as described below. For example, when the spacing between the transmitting antennas transmitting the beam is λ / 2 and ξ = 90°, a circularly polarized wave whose main beam direction is 0° in the front direction is generated. Also, when ξ = 30°, for example, a circularly polarized wave is generated whose main beam direction is shifted approximately -15° from the front direction. Also, when ξ = 150°, for example, a circularly polarized wave is generated whose main beam direction is shifted approximately +15° from the front direction. Here, λ is the wavelength of the high-frequency signal output from the transmitting antenna 109 (when a chirp signal is used, the wavelength at the center frequency of the chirp signal).

[0070] For example, when the main beam direction can be different from the front direction (or when the angle with a good axial ratio for circular polarization is set as the front direction), when the noc-th code element OC1(noc) in the first code is used as a reference, the noc-th code element OC2(noc) in the second code may be a code element with a phase difference of +ξ when noc=1 and -ξ when noc=2, or a code element with a phase difference of -ξ when noc=1 and +ξ when noc=2. Here, for example, ξ may be in the range of π / 6 to 5π / 6 radians (=30° to 150°).

[0071] Such an orthogonal code sequence can be generally expressed as Code2={exp(jξ)×A, -exp(jξ)×B} or Code2={-exp(jξ)×A, exp(jξ)×B} for the first code Code1={A, B}. These codes are orthogonal to each other, and when the noc-th code element OC1(noc) in the first code is used as a reference, the noc-th code element OC2(noc) in the second code is a code element with a phase difference of +ξ when noc=1 and -ξ when noc=2, or a radian phase difference of -ξ when noc=1 and +ξ when noc=2. Here, A and B are real or complex numbers, and the absolute values ​​of A and B are equal, for example, |A|=|B|.

[0072] In the following explanation, an example will be shown in which Code1 = {1, 1} and Code2 = {j, -j} are mainly used, but this is not limited to this, and codes of Code1 = {A, B}, Code2 = {j×A, -j×B} or Code2 = {-j×A, j×B}, or codes of Code1 = {A, B}, Code2 = {exp(jξ)×A, - exp(jξ)×B} or Code2 = {- exp(jξ)×A, exp(jξ)×B} may also be used, and similar effects will be obtained.

[0073] In addition to this embodiment, in Modification 1 and Modification 2 of Embodiment 1 described below, examples are shown in which Code1={1, 1} and Code2={j, -j} are mainly used, but the present invention is not limited to this. Code1={A, B}, Code2={j×A, -j×B}, or Code2={-j×A, j×B}, or Code1={A, B}, Code2={exp(jξ)×A, -exp(jξ)×B}, or Code2={-exp(jξ)×A, exp(jξ)×B} may also be used, with similar effects. Here, for example, ξ may be in the range of π / 6 to 5π / 6 radians (=30° to 150°).

[0074] In the encoding unit 107, the ndm-th Doppler shift amount DOP input from the Doppler shift setting unit 106 is ndm The number of code multiplexes when encoding a Doppler multiplexed signal using DOP_CODE (ndm)" where ndm=1,~, N DM is.

[0075] The encoding unit 107 may, for example, select the number of encoded Doppler multiplexes N when encoding the Doppler multiplexed signal. DOP_CODE (1), N DOP_CODE (2), ~, and N DOP_CODE (N DM ) is set to the number of coded Doppler multiplexing Nt so that the sum of the DOP_CODE For example, the encoding unit 107 sets the number of coded Doppler multiplexes N (ndm) so as to satisfy the following equation (7): DOP_CODE (ndm) is set. This enables the radar device 10 to perform multiplex transmission in the Doppler domain and the code domain (hereinafter referred to as coded Doppler multiplex transmission) using the Nt transmitting antennas 109.

number

[0076] Furthermore, the encoding unit 107 uses, for example, the uniform Doppler shift amount setting including the maximum uniform Doppler shift amount setting to determine the number of coded Doppler multiplexes N DOP_CODE (1), N DOP_CODE (2),~, N DOP_CODE (N DM ) for 1 or more N CM For example, the encoding unit 107 may set the code number N for all of the coded Doppler multiplex numbers. CM At least one Doppler shift DOP ndm The number of coded Doppler multiplexes N corresponding to DOP_CODE (ndm) to N CM Therefore, the Doppler shift amount DOP ndmand orthogonal code sequences, at least one Doppler shift amount DOP ndm The number of multiplexings (coded Doppler multiplexing number) N DOP_CODE (ndm) may be different from the coded Doppler multiplexing numbers associated with other Doppler shift amounts. For example, the encoding unit 107 sets the coded Doppler multiplexing numbers for Doppler-multiplexed signals non-uniformly. With this setting, the radar device 10 can individually separate and receive the coded Doppler multiplexed signals transmitted from the multiple transmitting antennas 109 over a Doppler range of ±½Tr, for example, by aliasing determination processing in the reception processing described later.

[0077] Alternatively, the encoding unit 107 may set the number of coded Doppler multiplexes N to N by using a uniform Doppler shift amount setting that is narrower than the maximum uniform Doppler shift amount setting. DOP_CODE (1), N DOP_CODE (2),~, N DOP_CODE (N DM ) for 1 or more N CM For example, the encoding unit 107 may set the number of codes N for all the coded Doppler multiplex numbers. CM Therefore, the Doppler shift amount DOP ndm In multiple combinations of the orthogonal code sequence and the Doppler shift amount DOP ndm The number of multiplexings (coded Doppler multiplexing number) N by the orthogonal code sequence associated with each DOP_CODE The (ndm) may be the same. For example, the encoding unit 107 uniformly sets the coded Doppler multiplexing number for the Doppler multiplexed signals. This setting enables the radar device 10 to individually separate and receive the coded Doppler multiplexed signals transmitted from the multiple transmitting antennas 109 over a Doppler range of ±1 / (2×Loc×Tr) by, for example, performing aliasing detection processing in the reception processing described later.

[0078] Alternatively, the encoding unit 107 may set the number of coded Doppler multiplexes N DOP_CODE (1), N DOP_CODE(2),~, N DOP_CODE (N DM ) for 1 or more N CM For example, the encoding unit 107 may set the number of codes N for all the coded Doppler multiplex numbers. CM For example, the encoding unit 107 uniformly sets the number of encoded Doppler multiplexing signals. In this setting, for example, aliasing determination processing in the reception processing described later is not applied. In addition, the radar device 10 may set, for example, ±1 / (2Loc×N DM ×Tr), signals that have been coded Doppler multiplexed from the multiple transmitting antennas 109 can be individually separated and received.

[0079] The encoding unit 107 encodes the ndm-th Doppler shift amount DOP in the m-th transmission period Tr. ndm The phase rotation amount φ ndm , the coded Doppler phase rotation amount ψ shown in the following equation (8) ndop_code(ndm), ndm (m) and outputs it to the phase rotation section 108.

number

[0080] Here, the subscript "ndop_code(ndm)" represents the Doppler shift DOP ndm The phase rotation amount φ ndm Number of coded Doppler multiplexes N for DOP_CODE (ndm) represents the index below. For example, ndop_code(ndm)=1,…, N DOP_CODE (ndm). Also, angle[x] is an operator that outputs the radian phase of real number x, for example, angle[1]=0, angle[-1]=π, angle[j]=π / 2, angle[-j]=-π / 2. Also, floor[x] is an operator that outputs the largest integer not exceeding real number x, where j is the imaginary unit.

[0081] For example, as shown in equation (8), the coded Doppler phase rotation amount ψndop_code(ndm), ndm (m) is the Doppler shift amount DOP during the period of the transmission cycle of the code length Loc used for encoding. ndm The amount of phase rotation to be applied is kept constant (for example, the first term of equation (8)), and the code Code ndop_code(ndm) Each of the Loc code elements OC ndop_code(ndm) (1),…,OC ndop_code(ndm) (Loc) is assigned a corresponding phase rotation amount (the second term of equation (8)).

[0082] The encoding unit 107 also outputs the orthogonal code element index OC_INDEX to the radar receiver 200 (the output switching unit 209, which will be described later) for each transmission period (Tr). ndop_code(ndm) and varies cyclically within the range of 1 to Loc for each transmission period (Tr), as shown in the following equation (9):

number

[0083] Here, mod(x, y) is the modulo operator, which is a function that outputs the remainder after dividing x by y. Also, m=1, ∼,Nc. Nc is the number of transmission periods used for radar positioning (hereinafter referred to as the "radar transmission signal transmission count"). Also, the radar transmission signal transmission count Nc is set to be an integer multiple (Ncode multiple) of Loc. For example, Nc=Loc×Ncode.

[0084] Next, in the encoding unit 107, the number of coded Doppler multiplexing signals N DOP_CODE An example of a method for setting (ndm) non-uniformly will be described.

[0085] For example, the encoding unit 107 determines the number of orthogonal code sequences (for example, the number of code multiplexes or the number of codes) N CM For example, the number of orthogonal code sequences N CM and the number of Doppler multiplexes N DM satisfies the following relationship with respect to the number Nt of transmitting antennas 109 used for multiplex transmission: (Number of orthogonal code sequences N CM ) × (Doppler multiplex number N DM )>Number of transmit antennas used for multiplex transmission Nt

[0086] For example, the number of orthogonal code sequences that satisfy the above conditions is N CM and the number of Doppler multiplexes N DM Of these, the product (N CM ×N DM ) is more suitable in terms of characteristics and circuit complexity. However, the number of orthogonal code sequences N that satisfies the above condition is CM and the number of Doppler multiplexes N DM Of these, the product (N CM ×N DM ) is not limited to a combination with a smaller value, and other combinations are also applicable.

[0087] In this embodiment, as an example, the number of codes (for example, the number of code multiplexing) N CM =2 orthogonal code sequence is used.

[0088] For example, if Nt=5, then N DM =3 and N CM The combination of =2 is preferable.

[0089] As an example, Figure 3 shows the case where Nt=5, N DM =3, N CM For example, the allocation of Doppler shift amounts DOP1, DOP2, and DOP3 and orthogonal codes Code1 and Code2 is as shown in FIG. DOP_CODE (1), N DOP_CODE (2) and N DOP_CODE Determined according to the setting in (3).

[0090] For example, in Figure 3(a), N DOP_CODE (1)=2, N DOP_CODE (2)=2, N DOP_CODE (3)=1, and (b) in Figure 3 shows an example where N DOP_CODE (1)=1, N DOP_CODE (2)=2, N DOP_CODE(3)=2, and (c) in Figure 3 shows an example where N DOP_CODE (1)=2, N DOP_CODE (2)=1, N DOP_CODE Here is an example where (3)=2.

[0091] In addition, in (a) and (b) of FIG. 3, the coded Doppler multiplexing number N DOP_CODE Code 1 is used for the Doppler shift amount corresponding to (ndm)=1, but is not limited to this. For example, when the number of coded Doppler multiplexing is N CM For smaller settings, Code2 may be used instead of Code1, as shown in FIG. 3(c).

[0092] As shown in Figure 3(a), N DOP_CODE (1)=N DOP_CODE (2)=2, N DOP_CODE (3)=1, and N DOP_CODE (1)=N DOP_CODE (2)≠N DOP_CODE As shown in (2), the coded Doppler multiplexing number N DOP_CODE are set non-uniformly. In such a setting, the Doppler frequency range can be set to be equivalent to the maximum Doppler velocity when transmitting from one antenna, for example (details will be described later).

[0093] Also, for example, when Nt=6 or 7, N DM =4 and N CM The combination of =2 is preferable.

[0094] As an example, Figure 4 shows the case where Nt=6, N DM =4, N CM For example, the allocation of Doppler shift amounts DOP1, DOP2, DOP3, and DOP4 and orthogonal codes Code1 and Code2 is as shown in FIG. DOP_CODE (1), N DOP_CODE (2), N DOP_CODE (3) and N DOP_CODE Determined according to the setting of (4).

[0095] For example, in Figure 4(a), NDOP_CODE (1)=N DOP_CODE (2)=2, N DOP_CODE (3)=N DOP_CODE (4)=1, and (b) in Fig. 4 shows an example where N DOP_CODE (1)=N DOP_CODE (3)=2, N DOP_CODE (2)=N DOP_CODE Here is an example where (4)=1.

[0096] In FIG. 4, the number of coded Doppler multiplexes N DOP_CODE Code 1 is used for the Doppler shift amount corresponding to (ndm)=1, but is not limited to this. For example, when the number of coded Doppler multiplexing is N CM For smaller settings, Code2 may be used instead of Code1, as shown in FIG. 4(a), or Code1 and Code2 may be mixed, as shown in FIG. 4(b).

[0097] Also, for example, as shown in FIG. DM =4, N CM If N = 2, there are two Doppler shifts that do not use all the codes. DM Of the 4 combinations of Doppler shift amounts that do not use any of the codes, there are 6 combinations (=4C2) of selecting two Doppler shift amounts from the four Doppler shift amounts, and in each combination, there are 4 combinations (=N CM ×N CM ) Therefore, Nt=6, N DM =4, N CM When DOP=2, there are a total of 24 combinations of the Doppler shift amount DOP and the orthogonal code Code assignment.

[0098] Similarly, for example, if Nt=8, N DM =5 and N CM For example, when Nt=9, the combination of N DM =5 and N CM For example, when Nt=10, the combination of N DM =6 and N CMA combination of Nt=11 or more is preferable. Note that the number Nt of transmitting antennas 109 is not limited to the above example, and an embodiment of the present disclosure can also be applied to cases where Nt=11 or more.

[0099] Next, in the encoding unit 107, the number of coded Doppler multiplexing signals N DOP_CODE An example of a method for uniformly setting (ndm) will be described.

[0100] In the encoding unit 107, the number of coded Doppler multiplexing signals N DOP_CODE The method for uniformly setting (ndm) is to use the number of orthogonal code sequences N that satisfy the following conditions: CM and the number of Doppler multiplexes N DM Of these, the product (N CM ×N DM ) is more suitable in terms of characteristics and circuit complexity. CM ×N DM ) is not limited to a combination with a smaller value, and other combinations are also applicable.

[0101] For example, the encoding unit 107 determines the number of orthogonal code sequences (for example, the number of code multiplexes or the number of codes) N CM For example, the number of orthogonal code sequences N CM and the number of Doppler multiplexes N DM satisfies the following relationship with respect to the number Nt of transmitting antennas 109 used for multiplex transmission: (Number of orthogonal code sequences N CM ) × (Doppler multiplex number N DM ) = Number of transmitting antennas used for multiplex transmission Nt

[0102] For example, if Nt=4, then N DM = 2 and N CM For example, when Nt=6, the combination of N DM =3 and N CM For example, when Nt=8, the combination of N DM = 4 and N CMFor example, when Nt=10, the combination of N DM =5 and N CM For example, when Nt=12, the combination of N DM = 6 and N CM The combination of =2 is preferable.

[0103] The number Nt of transmitting antennas 109 is not limited to the above example, and an embodiment of the present disclosure can be applied. In this case, the number Nt of orthogonal code sequences CM >1, Doppler multiplex number N DM > 1, and (the number of orthogonal code sequences N CM ) × (Doppler multiplex number N DM )=the number of transmitting antennas Nt used for multiplex transmission, the number of transmitting antennas Nt used for multiplex transmission may be set to an Nt that is equal to or greater than 4 and satisfies the above condition.

[0104] Next, the coded Doppler phase rotation amount ψ ndop_code(ndm), ndm An example of setting (m) is explained below.

[0105] For example, in the encoding unit 107, the number of transmitting antennas used for multiplexing is Nt=4, and the number of Doppler multiplexing is N DM =2, code multiplex number N CM = 2, and an orthogonal code sequence Code1={1,1} and Code2={j,-j} with a code length Loc=2 is used. In this case, for example, if the number of coded Doppler multiplexes is N DOP_CODE (1)=2, N DOP_CODE If (2)=2, encoding section 107 calculates the amount of encoded Doppler phase rotation ψ as shown in the following equations (10) to (13). 1, 1 (m), ψ 2, 1 (m), ψ 1, 2 (m), ψ 2, 2 (m) and outputs it to the phase rotation section 108.

number

number

number

number

[0106] Here, as an example, the Doppler shift amount DOP ndm The amount of phase rotation to be given is φ ndm =2π(ndm-1) / N DM When the phase rotation amount φ1=0 for imparting the Doppler shift amount DOP1 and the phase rotation amount φ2=π for imparting the Doppler shift amount DOP2 are used, the encoding unit 107 calculates the encoded Doppler phase rotation amount ψ as shown in the following equations (14) to (17): 1, 1 (m), ψ 2, 1 (m),ψ 1, 2 (m), ψ 2, 2 (m) is set and output to phase rotation section 108. Here, m=1, ∼, Nc. Note that here, a modulo operation by 2π is performed, and the range of radians is described as being equal to or greater than 0 and less than 2π (the same applies to the following explanations).

number

number

number

number

[0107] As shown in equations (14) to (17), the amount of phase rotation is φ, which is an equal division of 2π. ndm =2π(ndm-1) / N DM When the coded Doppler phase rotation amount ψ is set to 1, 1 (m), ψ 2, 1 (m), ψ 1, 2 (m), ψ 2, 2 (m) is N DM ×N CM = 2 × 2 = 4 transmission cycles.

[0108] Or, as another example, the Doppler shift amount DOP ndm The amount of phase rotation to be applied is φ ndm =2π(ndm) / N DM and the phase rotation amount φ1 to impart the Doppler shift amount DOP1 may be set to φ1=π, and the phase rotation amount φ2 to impart the Doppler shift amount DOP2 may be set to φ2=0. In this case, encoding section 107 calculates the encoded Doppler phase rotation amount ψ as shown in the following equations (18) to (21). 1, 1 (m), ψ 2, 1 (m), ψ 1, 2 (m), ψ 2, 2 (m) is set and output to the phase rotation unit 108, where m=1, ∼, Nc.

number

number

number

number

[0109] Furthermore, as shown in equations (14) to (17) or equations (18) to (21), the number of phases (for example, two, 0 and π) used for the amount of phase rotation (for example, the amount of phase rotation that imparts a Doppler shift) is smaller than the number of transmitting antennas 109 used for multiplex transmission, Nt=4. For example, as shown in equations (14) to (17) or equations (18) to (21), the number of phases (for example, two, 0 and π) used for the amount of phase rotation that imparts a Doppler shift is smaller than the number of Doppler shifts used for multiplex transmission (for example, the number of Doppler multiplexes) Nt DM =2.

[0110] In addition, for example, in the encoding unit 107, the number of transmitting antennas used for multiplex transmission is Nt=6, and the number of Doppler multiplexing N DM =4, code multiplex number N CM= 2, and an orthogonal code sequence Code1={1,1} and Code2={j,-j} with a code length Loc=2 is used. In this case, for example, if the number of coded Doppler multiplexes is N DOP_CODE (1)=1, N DOP_CODE (2)=1, N DOP_CODE (3)=2, N DOP_CODE If (4)=2, encoding section 107 calculates the amount of encoded Doppler phase rotation ψ as shown in the following equations (22) to (27). 1, 1 (m), ψ 1, 2 (m), ψ 1, 3 (m) , ψ 2, 3 (m) , ψ 1, 4 (m) , ψ 2, 4 (m) is set and output to the phase rotation unit 108, where m=1, ∼, Nc.

number

number

number

number

number

number

[0111] Here, as an example, the Doppler shift amount DOP ndm The amount of phase rotation to be applied is φ ndm =2π(ndm-1) / N DM and the phase rotation amount φ1=0 to impart Doppler shift amount DOP1, the phase rotation amount φ2=π / 2 to impart Doppler shift amount DOP2, the phase rotation amount φ3=π to impart Doppler shift amount DOP3, and the phase rotation amount φ4=3π / 2 to impart Doppler shift amount DOP4 are used, the encoding unit 107 calculates the encoded Doppler phase rotation amount ψ as shown in the following equations (28) to (33). 1, 1 (m), ψ1, 2 (m), ψ 1, 3 (m) , ψ 2, 3 (m) , ψ 1, 4 (m) , ψ 2, 4 (m) is set and output to the phase rotation unit 108, where m=1, ∼, Nc.

number

number

number

number

number

number

[0112] As shown in equations (28) to (33), the amount of phase rotation is φ, which is an equal division of 2π. ndm =2π(ndm-1) / N DM When the coded Doppler phase rotation amount ψ is set to 1, 1 (m), ψ 1, 2 (m), ψ 1, 3 (m) , ψ 2, 3 (m) , ψ 1, 4 (m) , ψ 2, 4 (m) is N DM ×N CM = 4 × 2 = 8 transmission cycles.

[0113] Furthermore, as shown in equations (28) to (33), the number of phases (for example, four, 0, π / 2, π, and 3π / 2) used for the amount of phase rotation (for example, the amount of phase rotation that imparts a Doppler shift) is smaller than the number of transmitting antennas 109 used for multiplex transmission, Nt=6. For example, as shown in equations (28) to (33), the number of phases (for example, four, 0, π / 2, π, and 3π / 2) used for the amount of phase rotation that imparts a Doppler shift is smaller than the number of Doppler shifts used for multiplex transmission (for example, the number of Doppler multiplexes) Nt DM =4.

[0114] Here, as an example, the number of transmitting antennas 109 is Nt=4, and the number of Doppler multiplexing N DM = 2, and the number of transmitting antennas 109 is Nt = 6, and the number of Doppler multiplexing N DM The setting of the phase rotation amount in the case of Nt = 4 has been explained, but the number of transmitting antennas 109 and the number of Doppler multiplexing N DM is not limited to these values. For example, whatever the value of the number Nt of transmitting antennas 109, the number of phases used for the amount of phase rotation may be set to be less than the number Nt of transmitting antennas 109 used for multiplex transmission. Also, the number of phases used for the amount of phase rotation to impart a Doppler shift is set to be less than the number Nt of Doppler shifts used for multiplex transmission. DM may be set equal to

[0115] Furthermore, as in the above example, the phase rotation amount may be set using the phase rotation amount setting shown in the maximum equal-interval Doppler shift amount setting, or the phase rotation amount setting shown in the equal-interval Doppler shift amount setting, for example, equation (6), may be used.

[0116] The method for setting the amount of phase rotation in phase rotation amount setting section 105 has been described above.

[0117] In FIG. 1, the phase rotation unit 108 rotates the coded Doppler phase rotation amount ψ set in the phase rotation amount setting unit 105. ndop_code(ndm), ndm Based on (m), a phase rotation amount is given to the chirp signal input from the radar transmission signal generator 101 for each transmission period Tr. Here, ndm=1, ∼, NDM and ndop_code(ndm)=1,~, N DOP_CODE (ndm).

[0118] Number of coded Doppler multiplexes N DOP_CODE (1), N DOP_CODE (2),~,N DOP_CODE (N DM ) is set equal to the number Nt of transmitting antennas 109, and the Nt coded Doppler phase rotation amounts are input to Nt phase rotation units 108, respectively.

[0119] The Nt phase rotation units 108 rotate the chirp signal input from the radar transmission signal generation unit 101 by the input coded Doppler phase rotation amount ψ ndop_code(ndm), ndm The outputs from the Nt phase rotation units 108 (for example, called coded Doppler multiplexed signals) are amplified to a specified transmission power and then radiated into space from the Nt transmission antennas 109 of the transmitting array antenna unit.

[0120] In the following, the coded Doppler phase rotation amount ψ ndop_code(ndm), ndm The phase rotation unit 108 that adds (m) is expressed as "phase rotation unit PROT#[ndop_code(ndm), ndm]". Similarly, the transmitting antenna 109 that radiates the output of the phase rotation unit PROT#[ndop_code(ndm), ndm] into space is expressed as "transmitting antenna Tx#[ndop_code(ndm), ndm]". Here, ndm=1, ∼, N DM and ndop_code(ndm)=1,~, N DOP_CODE (ndm).

[0121] In this embodiment, the number of codes (for example, the number of code multiplexing) N CM = 2 orthogonal code sequences are used. DOP_CODE (ndm)=1 or 2 and ndop_code(ndm)≦2.

[0122] For example, when the number of transmitting antennas used for multiplex transmission is Nt=4, the number of Doppler multiplexing NDM =2, code multiplex number N CM = 2, the orthogonal code sequence Code1 = {1, 1} and Code2 = {j, -j} with code length Loc = 2, and the number of coded Doppler multiplexes is N DOP_CODE (1)=2, N DOP_CODE The case where (2)=2 will be described. In this case, the encoding unit 107 transmits to the phase rotation unit 108 the amount of encoded Doppler phase rotation ψ 1, 1 (m), ψ 2, 1 (m), ψ 1, 2 (m), ψ 2, 2 (m) is input for each transmission period.

[0123] For example, the phase rotation unit PROT#[1, 1] rotates the chirp signal generated by the radar transmission signal generation unit 101 for each transmission period by a phase rotation amount ψ as shown in the following equation (34): 1, 1 (m) is added. The output of the phase rotation unit PROT#[1, 1] is output from the transmitting antenna Tx#[1, 1]. Here, cp(t) represents the chirp signal for each transmission period.

number

[0124] Similarly, the phase rotation unit PROT#[2, 1] rotates the chirp signal generated by the radar transmission signal generation unit 101 for each transmission period by a phase rotation amount ψ as shown in the following equation (35): 2, 1 The output of the phase rotation unit PROT#[2, 1] is output from the transmitting antenna Tx#[2, 1].

number

[0125] Similarly, the phase rotation unit PROT#[1, 2] rotates the chirp signal generated by the radar transmission signal generation unit 101 for each transmission period by a phase rotation amount ψ as shown in the following equation (36): 1, 2 The output of the phase rotation unit PROT#[1, 2] is output from the transmitting antenna Tx#[1, 2].

number

[0126] Similarly, the phase rotation unit PROT#[2, 2] rotates the chirp signal generated by the radar transmission signal generation unit 101 for each transmission period by a phase rotation amount ψ as shown in the following equation (37): 2, 2 The output of the phase rotation unit PROT#[2, 2] is output from the transmitting antenna Tx#[2, 2].

number

[0127] The coded Doppler phase rotation amount ψ ndop_code(ndm), ndm An example of setting (m) was explained.

[0128] In this embodiment, for example, the polarization of the transmitting antenna 109, the arrangement of the transmitting antenna 109, and the allocation of the coded Doppler phase rotation amount are associated as follows: This association enables the radar device 10 to use, in radar processing, not only the transmitting antenna 109 that performs multiplexing but also a transmitting antenna with a polarization (e.g., circular polarization) different from the polarization (e.g., horizontal polarization and vertical polarization) of the transmitting antenna that performs multiplexing (an example will be described later).

[0129] For example, at least one pair of adjacent transmitting antennas 109 are antennas that emit (or radiate) mutually different orthogonal polarizations (e.g., horizontal polarization and vertical polarization), and transmit radar transmission signals using the same Doppler multiplexing (e.g., Doppler shift amount).

[0130] In this embodiment, the number of codes (for example, the number of code multiplexing) N CM Since an orthogonal code sequence with N = 2 is used, DOP_CODE (ndm _BF )=2.

[0131] For example, adjacent N DOP_CODE(ndm _BF )=2 transmit antennas 109 are provided with phase rotation units PROT#[1, ndm _BF ], phase rotation part PROT#[2, ndm _BF ] is assigned to the transmit antenna Tx#[1, ndm _BF ], transmitting antenna Tx#[2, ndm _BF ], where ndm _BF is 1,~, N DM For example, the Doppler shift amount DOP ndm Among the multiple combinations of the orthogonal code sequence and the multiple transmitting antennas 109, adjacent transmitting antennas 109 are antennas that emit different orthogonal polarized waves, and in the combinations associated with each antenna, the Doppler shift amount is the same (for example, ndm=ndm _BF )

[0132] For example, one or more combinations (for example, pairs) that satisfy the association between the allocation of the coded Doppler phase rotation amount and the polarization and arrangement of the transmitting antenna 109 may be included.

[0133] As an example, when the number of transmitting antennas used for multiplex transmission is Nt=4, the number of Doppler multiplexing N DM =2, code multiplex number N CM = 2, the orthogonal code sequence Code1 = {1, 1} and Code2 = {j, -j} with code length Loc = 2, and the number of coded Doppler multiplexes is N DOP_CODE (1)=2, N DOP_CODE The case where (2)=2 is explained. Note that the number of beam transmitting antennas N BF = 2, and the index of the Doppler multiplexed signal used for the beam transmitting antenna is ndm _BF =1 and 2 are used.

[0134] In Figure 6, for example, the Nt=4 horizontally adjacent transmitting antennas 109 are, from the left antenna, transmitting antenna Tx#[1,1], transmitting antenna Tx#[2,1], transmitting antenna Tx#[1,2], and transmitting antenna Tx#[2,2].

[0135] In Figure 6, from the left, 2(=N DOP_CODE (1)) Adjacent transmitting antennas Tx#[1,1] and Tx#[2,1] transmit radar transmission signals using the same Doppler multiplexing (Doppler shift amount = DOP1). Furthermore, transmitting antennas Tx#[1,1] and Tx#[2,1] transmitting radar transmission signals using the same Doppler multiplexing (Doppler shift amount = DOP1) each use antennas that emit linearly polarized radio waves that are orthogonal to each other. For example, in the example shown in FIG. 6, transmitting antenna Tx#[1,1] may be an antenna that emits horizontally polarized radio waves (e.g., a "horizontally polarized antenna"), and transmitting antenna Tx#[2,1] may be an antenna that emits vertically polarized radio waves (e.g., a "vertically polarized antenna"). However, without being limited thereto, a vertically polarized antenna may be used for transmitting antenna Tx#[1,1], and a horizontally polarized antenna may be used for transmitting antenna Tx#[2,1]. In addition, the black circles (●) in FIG. 6 indicate the phase centers of the respective transmitting antennas.

[0136] Also, in Figure 6, from the right, 2(=N DOP_CODE (2)) Adjacent transmitting antennas Tx#[1,2] and Tx#[2,2] transmit radar transmission signals using the same Doppler multiplexing (Doppler shift amount = DOP2). Furthermore, transmitting antennas Tx#[1,2] and Tx#[2,2] transmitting radar transmission signals using the same Doppler multiplexing (Doppler shift amount = DOP2) each use antennas that emit linearly polarized radio waves that are orthogonal to each other. For example, in the example shown in FIG. 6, a horizontally polarized antenna may be used for transmitting antenna Tx#[1,2], and a vertically polarized antenna may be used for transmitting antenna Tx#[2,2]. However, without being limited to this, a vertically polarized antenna may be used for transmitting antenna Tx#[1,2], and a horizontally polarized antenna may be used for transmitting antenna Tx#[2,2].

[0137] As another example, when the number of transmitting antennas used for multiplex transmission is Nt=5, the Doppler multiplexing number N DM =3, code multiplex number N CM= 2, the orthogonal code sequence Code1 = {1, 1} and Code2 = {j, -j} with code length Loc = 2, and the number of coded Doppler multiplexes is N DOP_CODE (1)=2, N DOP_CODE The case where (2)=2 is explained. Note that the number of beam transmitting antennas N BF = 2, and the index of the Doppler multiplexed signal used for the beam transmitting antenna is ndm _BF1 =1, and ndm _BF2 Use =2.

[0138] In Fig. 7, for example, the Nt=5 horizontally adjacent transmitting antennas 109 are, from the left antenna, transmitting antenna Tx#[1,1], transmitting antenna Tx#[2,1], transmitting antenna Tx#[1,2], transmitting antenna Tx#[2,2], and transmitting antenna Tx#[1,3]. Note that the black circles (●) in Fig. 7 indicate the phase centers of each transmitting antenna.

[0139] In Figure 7, from the left, 2(=N DOP_CODE (1)) Adjacent transmitting antennas Tx#[1,1] and Tx#[2,1] transmit radar transmission signals using the same Doppler multiplexing (Doppler shift amount = DOP1). Furthermore, transmitting antennas Tx#[1,1] and Tx#[2,1] transmitting radar transmission signals using the same Doppler multiplexing (Doppler shift amount = DOP1) each use antennas that emit linearly polarized radio waves that are orthogonal to each other. For example, in the example shown in FIG. 7, a horizontally polarized antenna may be used for transmitting antenna Tx#[1,1], and a vertically polarized antenna may be used for transmitting antenna Tx#[2,1]. However, this is not limiting, and a vertically polarized antenna may be used for transmitting antenna Tx#[1,1], and a horizontally polarized antenna may be used for transmitting antenna Tx#[2,1].

[0140] Also, the 2(=N) on the right side of the transmitting antennas Tx#[1, 1] and Tx#[2, 1] DOP_CODE(2)) Adjacent transmitting antennas Tx#[1,2] and Tx#[2,2] transmit radar transmission signals using the same Doppler multiplexing (Doppler shift amount = DOP2). Furthermore, transmitting antennas Tx#[1,2] and Tx#[2,2] transmitting radar transmission signals using the same Doppler multiplexing (Doppler shift amount = DOP2) each use antennas that emit linearly polarized radio waves that are orthogonal to each other. For example, in the example shown in FIG. 7, a horizontally polarized antenna may be used for transmitting antenna Tx#[1,2], and a vertically polarized antenna may be used for transmitting antenna Tx#[2,2]. However, this is not a limitation, and a vertically polarized antenna may be used for transmitting antenna Tx#[1,2], and a horizontally polarized antenna may be used for transmitting antenna Tx#[2,2]. Furthermore, transmitting antenna Tx#[1,3] may use an arbitrary polarized antenna.

[0141] Furthermore, for example, when the radar device 10 includes a plurality of sets (or pairs) of adjacent horizontally polarized antennas and vertically polarized antennas, the radar device 10 may set a different Doppler shift amount for each of the plurality of sets. For example, in Fig. 6 and Fig. 7, the set Doppler shift amount may be different for the set of transmitting antennas Tx#[1,1] and Tx#[2,1] and the set of transmitting antennas Tx#[1,2] and Tx#[2,2].

[0142] As described above, at least one pair of adjacent transmitting antennas 109 (e.g., corresponding to a first transmitting antenna) includes a transmitting antenna 109 that radiates horizontally polarized waves and a transmitting antenna 109 that radiates vertically polarized waves (e.g., corresponding to a second transmitting antenna). The same Doppler shift amount is set for at least one pair of adjacent transmitting antennas 109, and radar transmission signals are transmitted using the same Doppler multiplexing. For example, at least one pair of adjacent transmitting antennas 109 perform code-multiplexed transmission using the same Doppler multiplexing, and radiate radar transmission signals into space using different polarization antennas.

[0143] Here, received signals for each transmission period corresponding to radar transmission signals code-multiplexed using the same Doppler multiplexing can be considered as received signals corresponding to beam transmission by multiple transmitting antennas 109. For example, the transmission from at least one pair of adjacent transmitting antennas 109 described above is equivalent to beam transmission by a subarray formed by the adjacent transmitting antennas 109. For example, when the radar device 10 transmits radar transmission signals with equal power from at least one pair of adjacent transmitting antennas 109 described above, the radar device 10 can treat the transmission as transmission by a new transmitting antenna (hereinafter referred to as a "beam transmitting antenna") with the midpoint position of the adjacent transmitting antennas 109 as the phase center of the subarray (details will be described later in the section on reception processing). Note that the open circles (◯) in FIGS. 6 and 7 indicate the phase centers of the respective beam transmitting antennas.

[0144] Furthermore, polarized antennas that emit orthogonal linearly polarized radio waves, such as a horizontally polarized antenna and a vertically polarized antenna, may be used for at least one pair of adjacent transmitting antennas 109. As a result, the received signals for each transmission cycle corresponding to radar transmission signals code-multiplexed using the same Doppler multiplexing can be regarded as received signals corresponding to transmissions from transmitting antennas that emit circularly polarized radio waves from at least one pair of adjacent transmitting antennas 109. This principle will be explained below using computer simulations.

[0145] This paper describes a computer simulation in which two types of orthogonal linearly polarized antennas (for example, a horizontally polarized antenna and a vertically polarized antenna) are combined to generate different polarized waves (right-handed circularly polarized waves or left-handed circularly polarized waves).

[0146] For example, as shown in Fig. 8, the results of a computer simulation will be described for the case where the feed phases to the horizontally polarized antenna (ANT#1) and the vertically polarized antenna (ANT#2) are set to 90 degrees or -90 degrees with ANT#1 as the reference (e.g., 0 degrees) and signals are transmitted simultaneously. Here, as an example, the element spacing of the horizontally polarized antenna (ANT#1) and the vertically polarized antenna (ANT#2) is set to 0.5 wavelengths.

[0147] The element spacing is not limited to 0.5 wavelengths and may be wider. Widening the element spacing narrows the beamwidth of the main beam formed by the two antennas, narrowing the field of view within which the axial ratio can be maintained. Furthermore, if the element spacing is wider than one wavelength, grating lobes will occur in a direction different from the direction of the main beam formed by the two antennas, resulting in the emission of radio waves that are circularly polarized in both the main beam direction and the grating lobe direction.

[0148] 8 shows an example of a linearly polarized antenna using a planar patch antenna with a single feed point, but the present invention is not limited to this, and an antenna element with a two-point feed type planar patch antenna may also be used. Also, an antenna element in which a plurality of antenna elements are used to form an array may also be used.

[0149] For example, when ANT#1 is used as the reference and the phase of ANT#2 is shifted by 90 degrees, a left-handed circularly polarized wave with the directivity shown in Fig. 9 is emitted. Fig. 9(a) shows the directivity in the horizontal plane, and Fig. 9(b) shows the directivity in the vertical plane. Furthermore, when ANT#1 is used as the reference and the phase of ANT#2 is shifted by -90 degrees, a right-handed circularly polarized wave with the directivity shown in Fig. 10 is emitted. Fig. 10(a) shows the directivity in the horizontal plane, and Fig. 10(b) shows the directivity in the vertical plane.

[0150] Here, the two phases [0°, 90°] and [0°, −90°] for generating left-handed circularly polarized waves or right-handed circularly polarized waves using ANT#1 and ANT#2 are coded by the coding unit 107 as N CMThis corresponds to the case where Code1={1, 1} and Code2={j, -j} are used as orthogonal code sequences for 0, 1, 2, 3, 4, 5, 6, 7, 8, 9, 10, 11, 12, 13, 14, 15, 16, 17, 18, 19, 20, 21, 22, 23, 24, 25, 26, 27, 28, 29, 30, 31, 32, 33, 34, 35, 36, 37, 38, 39, 40, 41, 42, 43, 44, 45, 46, 47, 48, 49, 50, 51, 52, 53, 54, 55, 56, ncm When (noc) is used as the base, the noc-th code element OC of the second code ncm (noc) is a code element with a phase difference of +90° or −90°, where noc=1 or 2. For example, among multiple orthogonal codes, the phase of the code element OC corresponding to each transmission period differs by 90° between the first code Code1={1, 1} for the radar transmission signal transmitted from ANT#1 and the second code Code2={j, −j} for the radar transmission signal transmitted from ANT#2.

[0151] The radar transmission signals, which are code-multiplexed using the same Doppler multiplexing to encode signals encoded by the encoding unit 107, are transmitted from the multiple transmitting antennas 109 with phase differences of +90° or −90°. Therefore, the received signals corresponding to these radar transmission signals can be considered to be received signals corresponding to transmissions from transmitting antennas that emit circularly polarized radio waves.

[0152] For example, when Code1={1, 1} and Code2={j, -j} are used, in a first transmission period, a phase rotation amount corresponding to the first code element "1" of Code1 may be imparted to the radar transmission signal transmitted from ANT#1, and a phase rotation amount corresponding to the first code element "j" of Code2 may be imparted to the radar transmission signal transmitted from ANT#2. As a result, left-handed circularly polarized radio waves are emitted by ANT#1 and ANT#2 in the first transmission period. Furthermore, in a second transmission period following the first transmission period, a phase rotation amount corresponding to the second code element "1" of Code1 may be imparted to the radar transmission signal transmitted from ANT#1, and a phase rotation amount corresponding to the second code element "-j" of Code2 may be imparted to the radar transmission signal transmitted from ANT#2. As a result, right-handed circularly polarized radio waves are emitted by ANT#1 and ANT#2 in the second transmission period.

[0153] As described above, for example, the radar device 10 (radar transmitter 100) multiplexes and transmits radar transmission signals from ANT#1 and ANT#2 by imparting a phase rotation amount to the transmission signals such that the phases of ANT#1 and ANT#2 differ by 90° in each transmission period. This transmission method enables the radar device 10 to use transmission antennas that exceed the number Nt of transmission antennas for multiplexing, and to use transmission antennas with polarizations (e.g., circular polarizations) different from the polarizations (e.g., horizontal polarization and vertical polarizations) of the transmission antenna 109.

[0154] 6 and 7 have been described taking the example of the transmitting antenna 109 arranged in the horizontal direction, but the arrangement of the transmitting antenna 109 is not limited to this. For example, the transmitting antenna 109 may be arranged in the vertical direction, or arranged in a planar manner in the horizontal and vertical directions. Furthermore, the antenna constituting the transmitting antenna 109 may be composed of multiple subarray elements arranged in the horizontal direction, multiple subarray elements arranged in the vertical direction, or multiple subarray elements arranged in a planar manner in the horizontal and vertical directions. Furthermore, the antennas shown in FIGS. 6 and 7 may be some of the multiple antennas included in the radar device 10.

[0155] In this way, in this embodiment, the Doppler shift amount DOP ndm and orthogonal code sequence Code ncm At least one of the Doppler shift amounts DOP is different. ndm and orthogonal code sequence Code ncm and a combination (for example, assignment) of each of them.

[0156] In this embodiment, the number of coded Doppler multiplexing signals N DOP_CODE When (ndm) is set to non-uniform, the Doppler shift amount DOP ndm and orthogonal code sequence Code ncm In combination with each Doppler shift amount DOP ndm The orthogonal code sequence Code corresponding to ncm The number of multiplexing (e.g., the number of coded Doppler multiplexing N DOP_CODE 3, the Nt transmitting antennas 109 may include at least a plurality of (for example, two) transmitting antennas 109 that transmit transmission signals that are code-multiplexed with different orthogonal code sequences, and at least one transmitting antenna 109 that transmits a transmission signal that is not code-multiplexed. For example, the radar transmitting signal transmitted from the radar transmitting unit 100 may include at least the coded Doppler multiplexing number N DOP_CODE (ndm) to the code number N CM The coded Doppler multiplex signal set to and the coded Doppler multiplex number N DOP_CODE (ndm) to the code number N CM and a coded Doppler multiplexed signal set to a smaller value.

[0157] In this embodiment, the number of coded Doppler multiplexing signals N DOP_CODE When (ndm) is set uniformly, the Doppler shift amount DOP ndm and orthogonal code sequence Code ncm In combination with the Doppler shift DOP ndm The corresponding orthogonal code sequence Codencm The number of multiplexing (e.g., the number of coded Doppler multiplexing N DOP_CODE (ndm)) can be the same.

[0158] [Configuration of radar receiver 200] 1, the radar receiving unit 200 includes Na receiving antennas 202 forming an array antenna. The radar receiving unit 200 also includes Na antenna system processing units 201-1 to 201-Na, a CFAR (Constant False Alarm Rate) unit 211, a coded Doppler demultiplexing unit 212, a Doppler demultiplexing unit 213, and a direction estimating unit 214.

[0159] Each receiving antenna 202 receives a reflected wave signal, which is a radar transmission signal reflected by a target, and outputs the received reflected wave signal to the corresponding antenna system processing unit 201 as a received signal.

[0160] Each antenna system processing unit 201 includes a receiving radio unit 203 and a signal processing unit 206 .

[0161] The radio reception unit 203 includes a mixer unit 204 and an LPF (low pass filter) 205. The mixer unit 204 of the radio reception unit 203 mixes the received reflected wave signal with a chirp signal, which is a transmission signal input from the radar transmission signal generation unit 101, and passes the resulting signal through the LPF 205. This extracts a beat signal whose frequency corresponds to the delay time of the reflected wave signal. For example, as shown in FIG. 11, the difference frequency between the frequency of the transmitted chirp signal (transmitted frequency modulated wave) and the frequency of the received chirp signal (received frequency modulated wave) can be obtained as the beat frequency.

[0162] The signal processing unit 206 of each antenna system processing unit 201-z (where z=1 to Na) has an AD conversion unit 207, a beat frequency analysis unit 208, an output switching unit 209, and a Doppler analysis unit 210.

[0163] The signal (for example, a beat signal) output from the LPF 205 is converted into discrete sample data by the AD conversion unit 207 in the signal processing unit 206, which is discretely sampled.

[0164] The beat frequency analysis unit 208 analyzes N data The discrete sample data is subjected to FFT processing. As a result, the signal processing unit 206 outputs a frequency spectrum in which a peak appears at a beat frequency corresponding to the delay time of the reflected wave signal (radar reflected wave). During the FFT processing, the beat frequency analysis unit 208 may multiply the signal by a window function coefficient such as a Han window or a Hamming window. By using the window function coefficient, it is possible to suppress side lobes that occur around the beat frequency peak.

[0165] Here, the beat frequency response output from the beat frequency analysis unit 208 in the z-th signal processing unit 206 obtained by transmitting the m-th chirp pulse is called RFT z (f b , m), where f b represents the beat frequency index, which corresponds to the FFT index (bin number). For example, f b =0,~,N data / 2-1, z=1,~,Na, m=1,~,N C The beat frequency index f b The smaller the beat frequency, the shorter the delay time of the reflected wave signal (for example, the closer the distance to the target).

[0166] Also, the beat frequency index f b is calculated using the following equation (38): b ) can be transformed into the beat frequency index f b Let "distance index f b " is called.

number

[0167] where B w represents the frequency modulation bandwidth within the range gate of the chirp signal, and C0 represents the speed of light.

[0168] The output switching unit 209 selectively switches and outputs the output of the beat frequency analysis unit 208 for each transmission period to the OC_INDEX-th Doppler analysis unit 210 out of the Loc Doppler analysis units 210 based on the orthogonal code element index OC_INDEX input from the encoding unit 107 of the phase rotation amount setting unit 105. For example, in the m-th transmission period Tr, the output switching unit 209 selects the OC_INDEX-th Doppler analysis unit 210 obtained by equation (9).

[0169] The signal processing unit 206 has Loc (here, as an example, Loc=2) Doppler analysis units 210-1 to 210-Loc. For example, data is input to the noc-th Doppler analysis unit 210 every Loc transmission periods (Loc×Tr) by the output switching unit 209. Therefore, the noc-th Doppler analysis unit 210 receives data (for example, beat frequency response RFT input from the beat frequency analysis unit 208) for Ncode transmission periods out of the Nc transmission periods. z (f b , m)) to obtain the distance index f b Doppler analysis is performed for each, where noc is the index of the code element, noc=1, ~, Loc.

[0170] For example, if Ncode is a power of 2, FFT processing can be applied in Doppler analysis. In this case, the FFT size is Ncode, and the maximum Doppler frequency at which aliasing does not occur, as derived from the sampling theorem, is ±1 / (2Loc×Tr). Also, the Doppler frequency index f s The Doppler frequency interval is 1 / (Ncode×Loc×Tr), and the Doppler frequency index f s The range of f s = -Ncode / 2, ~, 0, ~, Ncode / 2-1.

[0171] In the following, as an example, a case where Ncode is a power of 2 will be described. If Ncode is not a power of 2, for example, by including zero-padded data, FFT processing can be performed with a data size (FFT size) that is a power of 2. Furthermore, the Doppler analysis unit 210 may multiply by a window function coefficient such as a Han window or a Hamming window during FFT processing. Applying a window function can suppress side lobes that occur around the Doppler frequency peak.

[0172] For example, the output VFT of the Doppler analysis unit 210 of the z-th signal processing unit 206 z noc (f b , f s ) is expressed by the following equation (39): where j is the imaginary unit and z=1 to Na.

number

[0173] The processing in each component of the signal processing unit 206 has been described above.

[0174] In FIG. 1, the CFAR unit 211 performs CFAR processing (for example, adaptive threshold determination) using the outputs of the Loc Doppler analyzers 210 of the first to Na-th signal processors 206, and calculates the distance index f that gives the peak signal. b_cfar and the Doppler frequency index f s_cfar Extract.

[0175] The CFAR unit 211 calculates the output VFT of the Doppler analysis unit 210 of the first to Na-th signal processing units 206, as shown in the following equation (40), for example. z noc (f b , f s) are power-added, and two-dimensional CFAR processing consisting of a distance axis and a Doppler frequency axis (corresponding to relative velocity) or CFAR processing combining one-dimensional CFAR processing is performed. For the two-dimensional CFAR processing or the CFAR processing combining one-dimensional CFAR processing, the processing disclosed in Non-Patent Document 2 may be applied, for example.

number

[0176] The CFAR unit 211 adaptively sets a threshold value and calculates a distance index f that has a received power greater than the threshold value. b_cfar , the Doppler frequency index f s_cfar , and received power information PowerFT(f b_cfar , f s_cfar ) to the coded Doppler demultiplexing unit 212.

[0177] The Doppler shift amount DOP ndm The phase rotation amount φ for adding ndm For example, when equation (5) is used, the intervals of the Doppler shift amounts in the Doppler frequency domain in the output of the Doppler analysis unit 210 are equal, and when the intervals of the Doppler shift amounts ΔFD are expressed in terms of the intervals of the Doppler frequency indexes, ΔFD=Ncode / N DM Therefore, in the output of the Doppler analysis unit 210, peaks are detected at intervals of ΔFD for each signal that is Doppler shift multiplexed in the Doppler frequency domain. ndm When equation (5) is used, Ncode and N DM Depending on the situation, ΔFD may not be an integer. In such cases, ΔFD can be made an integer value by using equation (59) described later. In the following, the reception processing operation will be described assuming that ΔFD is an integer value.

[0178] Figure 12(a) shows the N DM12(a), when reflected waves from three targets are observed at Doppler frequency indexes f1, f2, and f3, the reflected waves are also observed at Doppler frequency indexes at intervals of ΔFD for each of f1, f2, and f3 (e.g., f1-ΔFD, f2-ΔFD, f3-ΔFD+Ncode).

[0179] Therefore, the CFAR unit 211 may divide each output of the Doppler analysis unit 210 into ranges of intervals ΔFD of the Doppler shift amount, and for each divided range, align the peak positions of each signal multiplexed by Doppler shift and perform power addition (for example, called "Doppler domain compression") as shown in the following equation (41), and then perform CFAR processing (for example, called "Doppler domain compression CFAR processing"). s_comp =-ΔFD / 2,~,-ΔFD / 2-1. For example, ΔFD=Ncode / N DM If f s_comp =Ncode / (2N DM ),~,Ncode / (2N DM )-1.

number

[0180] However, in equation (41),

number

[0181] Similarly, in equation (41),

number

[0182] This reduces the Doppler frequency range of the CFAR processing to 1 / N.DM This reduces the amount of CFAR processing and simplifies the circuit configuration. DM Since the power of each of the Doppler shift multiplexed signals can be added, the SNR (Signal to Noise Ratio) can be increased to (N DM ) 1 / 2 This can improve the radar detection performance of the radar device 10.

[0183] 12(b) shows an example of an output after applying the Doppler domain compression process shown in equation (41) to the output of the Doppler analysis unit 210 shown in FIG. 12(a). As shown in FIG. 12(b), N DM If Ncode / 2 is 2, the CFAR unit 211 adds the power component of Doppler frequency index f1 and the power component of f1-ΔFD by Doppler domain compression processing and outputs the result. Similarly, as shown in FIG. 12(b), the CFAR unit 211 adds the power component of Doppler frequency index f2 and the power component of f2-ΔFD and outputs the result. Furthermore, for the power component of Doppler frequency index f3, since f3-ΔFD is smaller than -Ncode / 2, the CFAR unit 211 adds the power component of Doppler frequency index f3 and f3-ΔFD+Ncode (for example, Ncode / 2). DM If = 2, the power component of f3 + ΔFD is added and output.

[0184] As a result of Doppler domain compression, the range of the Doppler frequency index fs_comp in the Doppler frequency domain is -ΔFD / 2 or more, ~, ΔFD / 2-1 or less (ΔFD=Ncode / N DM In the case of -Ncode / (2N DM ) or more,~,Ncode / (2N DM )-1 or less), and the range of CFAR processing is compressed, so the amount of calculation in CFAR processing can be reduced. Also, in FIG. 12, for example, the reflected waves from three targets are power-added, so the SNR of the signal component is improved. Note that, since the noise component is also power-combined, the improvement effect of SNR is, for example, (N DM ) 1 / 2 This is an improvement to some extent.

[0185] The CFAR unit 211 using the Doppler domain compression CFAR process adaptively sets a threshold value, for example, and calculates the distance index f b_cfar , the Doppler frequency index f s_comp_cfar , and N DM The Doppler frequency index (f s_comp_cfar +(nfd-ceil(N DM / 2)-1)×ΔFD) b_cfar , f s_comp_cfar +(nfd-ceil(N DM / 2)-1)×ΔFD), nfd=1,…,N DM to the coded Doppler demultiplexing unit 212.

[0186] The CFAR unit 211 also calculates, for example, a distance index f b_cfar , the Doppler frequency index f s_comp_cfar is output to the Doppler demultiplexing unit 213.

[0187] The Doppler shift amount DOP ndm The phase rotation amount φ for adding ndm is not limited to Equation (5). For example, the phase rotation amount φ at which peaks are detected at regular intervals in the Doppler frequency domain output from the Doppler analysis unit 210 for each of the Doppler shift multiplexed signals is ndm If so, the CFAR unit 211 can apply Doppler domain compression CFAR processing.

[0188] For example, by setting the Doppler shift amount at equal intervals, Δf MinInterval =1 / (Tr(N DM +N int )L OC ) is set, the phase rotation amount φ ndm is set according to equation (6), and each Doppler-shift-multiplexed signal is output from the Doppler analysis unit 210 as ΔFD=Ncode / (N DM +N int) are detected as peaks at intervals of . In this case, too, the CFAR unit 211 can apply the Doppler domain compression CFAR processing.

[0189] Next, a description will be given of an example of the operation of the coded Doppler demultiplexing unit 212 shown in Fig. 1. The coded Doppler demultiplexing unit 212 separates linearly polarized signals (for example, horizontally polarized signals and vertically polarized signals) using, for example, both the output of the first Doppler analysis unit 210 and the output of the second Doppler analysis unit 210.

[0190] In the following, an example of processing by coded Doppler demultiplexing section 212 when Doppler domain compression CFAR processing is used in CFAR section 211 will be described.

[0191] The coded Doppler demultiplexing unit 212 receives the distance index f b_cfar , the Doppler frequency index f s_comp_cfar , and N DM The Doppler frequency index (f s_comp_cfar +(nfd-ceil(N DM / 2)-1)×ΔFD) b_cfar , f s_comp_cfar +(nfd-ceil(N DM / 2)-1)×ΔFD), nfd=1,~,N DM Based on this, the output of the Doppler analysis unit 210 is used to separate the coded Doppler multiplexed signals, and the transmitting antenna 109 is identified (also referred to as judgment or identification) and the Doppler frequency (e.g., Doppler velocity or relative velocity) is determined.

[0192] As described above, when the equal-interval Doppler shift amount setting including the maximum equal-interval Doppler shift amount setting is used, the encoding unit 107 of the phase rotation amount setting unit 105 can set, for example, N DM Number of coded Doppler multiplexes N DOP_CODE (1), N DOP_CODE (2),…, N DOP_CODE (N DM ) to N CMDo not set the number of coded Doppler multiplexes to N, and set at least one coded Doppler multiplex number to N CM For example, the coded Doppler demultiplexing unit 212 (1) performs code demultiplexing processing and sets the number of coded Doppler multiplexes to N. CM The coded Doppler multiplexing unit 212 detects coded Doppler multiplexed signals set to less than 1 (for example, detects unused coded Doppler multiplexed signals not used for multiplex transmission) and performs aliasing determination. Thereafter, the coded Doppler demultiplexing unit 212 (2) performs Doppler code separation processing of the coded Doppler multiplexed signals used for multiplex transmission based on the aliasing determination result.

[0193] The processes (1) and (2) in the coded Doppler demultiplexing unit 212 will be described below.

[0194] <(1) Alias ​​detection process (detection process for unused coded Doppler multiplexed signals)> The coded Doppler demultiplexing unit 212 performs the Doppler aliasing determination process, for example, with the Doppler range of the assumed target set to ±1 / (2Tr).

[0195] Here, for example, if Ncode is a power of 2, the Doppler analysis unit 210 applies FFT processing to each code element, and therefore performs FFT processing using the output from the beat frequency analysis unit 208 at a period of (Loc×Tr). Therefore, the Doppler range in which aliasing does not occur in the Doppler analysis unit 210 according to the sampling theorem is ±1 / (2Loc×Tr). This Doppler range ±1 / (2Loc×Tr) is further multiplied by the Doppler multiplexing number N DM Therefore, the coded Doppler demultiplexing unit 212 performs Doppler multiplexing using a Doppler range of ±1 / (2Loc×N DM ×Tr) for Loc×N DM The aliasing determination process is performed assuming a Doppler range of up to ±1 / (2Tr).

[0196] Here, as an example, Nt=3, and the Doppler multiplexing number N DM =2, code multiplex number N CMHere, the case where the Doppler shift amount DOP ndm The phase rotation amount φ for adding ndm are assigned as shown in equation (5) based on the maximum equally spaced Doppler shift amount setting, for example. In this case, the phase rotation amount φ1 to impart the Doppler shift amount DOP1 is φ1=0, and the phase rotation amount φ2 to impart the Doppler shift amount DOP2 is φ2=π. The encoding unit 107 uses two orthogonal codes Code1={1,1} and Code2={j,-j} with a code length Loc=2. As shown in FIG. 13(a), N DOP_CODE (1)=2, N DOP_CODE Use (2)=1.

[0197] In this case, the coded Doppler demultiplexing unit 212 is configured to operate within a Doppler range of ±1 / (2Loc×N DM ×Tr)=±1 / (8Tr), 4 times (=Loc×N DM The aliasing determination process is performed assuming a Doppler range of ±1 / (2Tr).

[0198] Here, the distance index f extracted in the CFAR unit 211 is b_cfar and the Doppler frequency index f s_comp_cfar The Doppler component VFT, which is the output of the Doppler analysis unit 210 corresponding to z noc (f b_cfar ,f s_comp_cfar ) may contain Doppler components including aliasing as shown in (a) and (b) in FIG. 14, for example, in the Doppler range of ±1 / (2Tr).

[0199] For example, as shown in FIG. 14(a), s_comp_cfar <0, in the Doppler range of ±1 / (2Tr), f s_comp_cfar -Ncode / N DM , f s_comp_cfar , f s_comp_cfar +Ncode / N DM , and f s_comp_cfar +2Ncode / N DM 4(=Loc×N DM) possible Doppler components (ΔFD=Ncode / N DM Using f s_comp_cfar -ΔFD, f s_comp_cfar , f s_comp_cfar +ΔFD, and f s_comp_cfar It can also be expressed as +2ΔFD.

[0200] Also, for example, as shown in (b) of FIG. 14, f s_comp_cfar >0, in the Doppler range of ±1 / (2Tr), f s_comp_cfar -2Ncode / N DM , f s_comp_cfar -Ncode / N DM , f s_comp_cfar , and f s_comp_cfar +Ncode / N DM , 4(=Loc×N DM ) possible Doppler components (ΔFD=Ncode / N DM Using f s_comp_cfar -2ΔFD, f s_comp_cfar -ΔFD, f s_comp_cfar , and f s_comp_cfar It can also be expressed as +ΔFD. s_comp_cfar For these possible Doppler components (4(=Loc × N DM ) street) f s_comp_cfar We call these 4(=Loc×N DM ) Doppler component candidates exist in each Doppler region, the index "D r " is used to express this. D r is an index indicating the Doppler fold range, e.g., D r ∈{-ceil(Loc×N DM / 2),…, ceil(Loc×N DM In Figure 14, D r =-2,~,1. Note that D r The region where D = 0 is the region where there is no Doppler folding, r The area where D ≠ 0 indicates that Doppler aliasing occurs. rThe larger the absolute value of D r This indicates that the Doppler range is far from the Doppler range indicated by =0.

[0201] The coded Doppler demultiplexing unit 212 detects 4 (=Loc×N) signals including aliasing in the Doppler range of ±1 / (2Tr) as shown in FIG. DM ) phase changes corresponding to the Doppler components are corrected, and the number of coded Doppler multiplexes is set to N CM The coded Doppler multiplexing signal set to less than 1000 (for example, unused coded Doppler multiplexing signals) is subjected to coded Doppler multiplexing demultiplexing processing.

[0202] Then, the coded Doppler demultiplexing unit 212 determines whether each Doppler component candidate is a true Doppler component based on the received power of the component obtained by performing coded Doppler demultiplexing processing on the unused coded Doppler multiplexed signal.

[0203] For example, the coded Doppler demultiplexing unit 212 s_comp_cfar Among the Doppler component candidates for f, the Doppler component with the smallest reception power obtained by performing coded Doppler demultiplexing processing based on unused coded Doppler multiplexed signals may be detected, and the detected Doppler component may be determined to be the true Doppler component. s_comp_cfar Among the Doppler component candidates for the signal, the Doppler components having reception powers other than the minimum reception power may be determined to be false Doppler components.

[0204] This aliasing detection process can resolve the ambiguity in the Doppler range of ±1 / (2Tr). Also, this aliasing detection process resolves the ambiguity in the Doppler range of ±1 / (2Loc×N) where aliasing due to Doppler multiplexing does not occur. DM Compared with the conventional method where the Doppler frequency is detected without ambiguity, the range in which the Doppler frequency can be detected without ambiguity can be expanded to a range of -1 / (2Tr) or more and less than 1 / (2Tr).

[0205] This is because, by performing coded Doppler demultiplexing based on unused coded Doppler multiplexed signals, for example, the phase change of the true Doppler component is correctly corrected, and orthogonality between the coded Doppler multiplexed signals used for multiplexing and the unused coded Doppler multiplexed signals is maintained. Therefore, there is no correlation between the coded Doppler multiplexed signal code used for multiplexing and the unused coded Doppler multiplexed signals, and the received power is approximately at the noise level.

[0206] On the other hand, for example, with respect to a false Doppler component, the phase change of the Doppler component is erroneously corrected, and the orthogonality between the coded Doppler multiplexed signal used for multiplex transmission and the unused coded Doppler multiplexed signal is not maintained. Therefore, a correlation component (interference component) occurs between the coded Doppler multiplexed signal code used for multiplex transmission and the unused coded Doppler multiplexed signal, and for example, a received power greater than the noise level may be detected. Therefore, as described above, the coded Doppler demultiplexing unit 212 performs coded Doppler demultiplexing based on the unused coded Doppler multiplexed signal. s_comp_cfar Among the Doppler component candidates for the Doppler component, the Doppler component with the smallest received power may be determined to be the true Doppler component, and other Doppler components with received power different from the smallest received power may be determined to be false Doppler components.

[0207] For example, the coded Doppler demultiplexing unit 212 calculates f based on the output of the Doppler analysis unit 210 in each antenna system processing unit 201. s_comp_cfar The phase change corresponding to each Doppler component of the Doppler component candidate for DAR (f b_cfar ,f s_comp_cfar ,D r , nuc,nud) are calculated according to equation (42).

[0208] Here, nuc and nud represent the index of the orthogonal code that becomes the unused coded Doppler multiplexed signal and the index of the Doppler multiplexed signal. For example, in the case of (b) of Figure 13, the unused coded Doppler multiplexed signal is indicated by an x ​​in the figure, and Code2 is assigned as the code and a Doppler shift amount of DOP1 is assigned. Therefore, the indexes of the orthogonal code to which the unused coded Doppler multiplexed signal is assigned are nuc=2 and nud=1.

[0209] Hereinafter, a set of an index of an orthogonal code used for a coded Doppler multiplexed signal and an index of a Doppler multiplexed signal will be referred to as "DCI (index of orthogonal code, index of Doppler multiplexed signal)." DCI(nuc,nud) represents, for example, an index of an orthogonal code to which an unused coded Doppler multiplexed signal is assigned and an index of the Doppler multiplexed signal. For example, in the case of (b) of FIG. 13, an unused coded Doppler multiplexed signal is assigned to DCI(2,1). Similarly, for example, in the case of (a) of FIG. 4, unused coded Doppler multiplexed signals are assigned to DCI(2,3) and DCI(2,4).

number

[0210] where Y z (f b_cfar ,f s_comp_cfar ,D r , nuc, nud) is calculated based on the output of the Doppler analysis unit 210 in the z-th antenna system processing unit 201 as shown in the following equation (43): s_comp_cfar is the received signal after correcting the phase change according to each Doppler component of the Doppler component candidate for , and separating the unused coded Doppler multiplexed signal to which DCI (nuc, nud) is assigned.

number

[0211] In equations (42) and (43), in order to separate the unused coded Doppler multiplexed signals to which DCI (nuc, nud) is assigned, the output VFTALL of the Doppler analyzer 210 in the z-th antenna system processor 201 is z (f b_cfar ,f s_comp_cfar ,D r , nud), the unused orthogonal code Code nuc is calculated, and the sum of the received powers is calculated for all antenna system processors 201. This makes it possible to improve the accuracy of aliasing detection even when the received signal level is low. However, instead of using equation (42), the received power after separation of unused coded Doppler multiplexed signals may be calculated for the output of the Doppler analyzer 210 in some of the antenna system processors 201. Even in this case, for example, in a range where the received signal level is sufficiently high, it is possible to reduce the amount of calculation processing while maintaining the accuracy of aliasing detection.

[0212] In addition, in equations (42) and (43), D r is an index indicating the Doppler fold range, e.g., D r ∈{-ceil(Loc×N DM / 2),~, ceil(Loc×N DM It takes an integer value in the range { / 2)-1}.

[0213] Also, in equation (43),

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[0214] In addition, in equation (43), the superscript T represents a vector transpose, and the superscript * (asterisk) represents a complex conjugate operator.

[0215] In equation (43), α(f s_comp_cfar ,D r ) represents the "Doppler phase correction vector." The Doppler phase correction vector α(f s_comp_cfar ,D r ) is, for example, the Doppler frequency index f extracted in the CFAR unit 211. s_comp_cfar is the output range (for example, Doppler range) of the Doppler analysis unit 210 that does not include Doppler aliasing, the Doppler aliasing range D r The Doppler phase rotation caused by the time difference between the Doppler analyses of the Loc Doppler analyzers 210 at the time of the Doppler analysis is corrected.

[0216] For example, the Doppler phase correction vector α(f s_comp_cfar ,D r ) is expressed as the following equation (45). The Doppler phase correction vector α(f s_comp_cfar ,D r ) is, for example, the output VFT of the first Doppler analysis unit 210. z 1 (f b_cfar , f s_comp_cfar ) as a reference for the Doppler analysis time, the output VFT of the second Doppler analysis unit 210 z 2 (f b_cfar , f s_comp_cfar ) to the Doppler analysis unit VFT of the Loc number z Loc (f b_cfar , f s_comp_cfar The Doppler frequency index f caused by the time delays of Tr, 2Tr, ~, (Loc-1)Tr at each s_comp_cfar Doppler fold range D r is a vector whose elements are Doppler phase correction coefficients that correct the phase rotation in the Doppler component in r N code / N DM The term is ΔFD=Ncode / NDM Using D r It can also be written as ΔFD. Therefore, ΔFD=Ncode / N DM This can be applied not only to:

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[0217] Such a Doppler phase correction vector α(f s_comp_cfar , D r ) phase correction by f s_comp_cfar This corresponds to correcting the phase change according to each Doppler component in the Doppler component candidates for the Doppler component.

[0218] Also, in equation (43), VFTALL z (f b_cfar , f s_comp_cfar , D r , nud) is, for example, the output VFT of the Loc Doppler analyzers 210 in the z-th antenna system processor 201, as shown in the following equation (46): z noc (f b , f s ), the distance index f extracted by the CFAR unit 211 b_cfar and the Doppler frequency index f s_comp_cfar Corresponding to the Doppler fold range D r In this case, the DCI (nuc,ndu) is assigned to the unused coded Doppler multiplexed signal, and the extracted components are expressed in vector form. r ={-ceil(Loc×N DM / 2),~, ceil(Loc×N DM It takes an integer value in the range { / 2)-1}.

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[0219] In equation (46), N code F R (D r , nud) / NDM is the Doppler fold range D r In the nudth Doppler multiplex signal, f s_comp_cfar represents the offset value of the Doppler index for N code F R (D r , nud) / N DM The term is ΔFD=Ncode / N DM Using F R (D r , nud) ΔFD. Therefore, ΔFD=Ncode / N DM This can be applied not only to DM is.

[0220] F R (D r , nud) is the Doppler fold range D r and the Doppler shift amounts DOP1, DOP2, ~, DOP N_DM The phase rotation amounts φ1, φ2, ~, φ N_DM Therefore, for example, the coded Doppler demultiplexing unit 212 can set the Doppler aliasing range D r and the amount of phase rotation, F R (D r , nud) and the Doppler aliasing range D r and based on the amount of phase rotation, F R (D r , nud) may be read out. Also, for example, the Doppler shift amounts DOP1, DOP2, . . . DOP N_DM The phase rotation amounts φ1, φ2, ~, φ N_DM -π≦φ1<φ2<…<φ N_DM <π, F R (D r , nud) can be expressed as the following equation (47).

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[0221] For example, the coded Doppler demultiplexing unit 212 calculates the received power P after code separation using the unused coded Doppler multiplexed signal to which DCI (nuc, nud) is assigned according to equations (42) and (43). DAR (f b_cfar ,f s_comp_cfar ,D r , nuc, nud) for each D r ∈{-ceil(Loc×N DM / 2),…, ceil(Loc×N DM / 2)-1}.

[0222] Then, the coded Doppler demultiplexing unit 212 demultiplexes each D r Within the range of DAR (f b_cfar ,f s_comp_cfar ,D r , nuc,nud) is minimized. r In the following, as shown in the following equation (48), r Within the range of DAR (f b_cfar ,f s_comp_cfar ,D r , nuc,nud) is minimized. r "D rmin " is expressed as ".

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[0223] If there are a plurality of unused coded Doppler multiplexed signals, the coded Doppler demultiplexing unit 212 demultiplexes the received power P DAR (f b_cfar ,f s_comp_cfar ,D r , nuc, nud), the received power Pall after code separation using all unused orthogonal codes is calculated as follows: DAR (f b_cfar ,f s_comp_cfar ,D r ) may also be used.

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[0224] By calculating the received power after code separation using all unused orthogonal codes, the accuracy of the folding back process can be improved even when the received signal level is low.

[0225] For example, the coded Doppler demultiplexing unit 212 r ∈{-ceil(Loc×N DM / 2),~, ceil(Loc×N DM / 2)-1} DAR (f b_cfar ,f s_comp_cfar ,D r ) and Pall DAR (f b_cfar ,f s_comp_cfar ,D r ) is the smallest D r (For example, D rmin For example, when equation (49) is used, each D r D, which gives the minimum received power in the range r "D rmin " is expressed as ".

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[0226] Furthermore, the coded Doppler demultiplexing unit 212 calculates, for example, the minimum received power Pall after code separation using an unused coded Doppler multiplexed signal to which DCI (nuc, nud) is assigned. DAR (f b_cfar ,f s_comp_cfar ,D rmin ) and the received power PowerFT_comp(f b_cfar ,f s_comp_cfar ) and perform a process of determining (for example, measuring) the likelihood of aliasing detection. In this case, the coded Doppler demultiplexing unit 212 may determine the likelihood of aliasing detection according to, for example, the following equations (51) and (52).

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[0227] For example, the coded Doppler demultiplexing unit 212 demultiplexes the distance index f extracted in the CFAR unit 211. b_cfar and the Doppler frequency index f s_comp_cfar PowerFT_comp(f b ,f s_comp_cfar ) to the specified value Threshold DR The minimum received power Pall after code separation using unused coded Doppler multiplexed signals to which DCI (nuc, nud) is assigned is greater than the value multiplied by DAR (f b_cfar ,f s_comp_cfar ,D rmin ) is small (for example, equation (51)), it is determined that the aliasing determination is sufficiently accurate. In this case, the radar device 10 may perform, for example, subsequent processing (for example, code separation processing).

[0228] On the other hand, for example, the coded Doppler demultiplexing unit 212 calculates PowerFT_comp(f b ,f s_comp_cfar ) and Threshold DR The minimum received power Pall after code separation using unused coded Doppler multiplexed signals to which DCI (nuc, nud) is assigned is greater than the value multiplied by DAR (f b_cfar ,f s_comp_cfar ,D rmin ) is equal to or larger than (for example, equation (52)), the accuracy of the aliasing determination is insufficient, and the reliability of the aliasing determination is determined to be low (for example, a noise component). In this case, the radar device 10 may not need to perform subsequent processing (for example, code separation processing), for example.

[0229] By this processing, it is possible to reduce the judgment error in the aliasing judgment and also to remove noise components. DRmay be set in the range of 0 to less than 1. For example, considering that a noise component is included, Threshold DR may be set in the range of about 0.1 to 0.5.

[0230] An example of the operation of the return process has been described above.

[0231] <(2) Doppler code separation processing of coded Doppler multiplexed signals used in multiplex transmission> The coded Doppler demultiplexing unit 212 performs coded Doppler demultiplexing processing on the coded Doppler multiplexed signal used for multiplex transmission, based on the result of the aliasing determination.

[0232] For example, the coded Doppler demultiplexing unit 212 calculates D rmin By applying equation (43) based on this, the coded Doppler multiplexing demultiplexer 212 demultiplexes and receives the coded Doppler multiplexed signals to which the DCI (ncm, ndm) used for multiplexing is assigned. For example, the coded Doppler demultiplexer 212 performs demultiplexing processing using the following equation (53) to demultiplex and receive the coded Doppler multiplexed signals to which the DCI (ncm, ndm) used for multiplexing is assigned. In the aliasing determination processing, the index (D rtrue ) can be determined (for example, D rmin =D rtrue Therefore, in the coded Doppler demultiplexing unit 212, the correlation value between the orthogonal codes used for code multiplexing can be set to zero in the Doppler range of -1 / (2Tr) or more and less than 1 / (2Tr), and demultiplexing processing can be performed with interference between code-multiplexed signals suppressed.

number

[0233] where Y z (f b_cfar ,f s_comp_cfar ,D rmin, ncm, ndm) is the distance index f of the Doppler analysis unit 210 in the z-th antenna system processing unit 201. b_cfar and the Doppler frequency index f s_comp_cfar At the output of rmin The ndmth coded Doppler multiplex signal VFTALL z (f b_cfar ,f s_comp_cfar ,D rmin , ndm), the orthogonal code Code ncm This is the output (for example, the coded Doppler demultiplexing result) of the code-multiplexed signal obtained by code separation using the above, and it is possible to separate the coded Doppler multiplexed signal to which DCI (ncm, ndm) used for multiplex transmission is assigned. Note that z=1,~,Na and ncm=1,~,N CM is.

[0234] By the code separation process described above, the radar device 10 can separate and receive the coded Doppler multiplexed signal to which the DCI (ncm, ndm) used for multiplex transmission is assigned, based on the aliasing determination result of the Doppler analysis unit 210, which assumes a Doppler range of up to ±1 / (2Tr) that is Loc times the Doppler range ±1 / (2Loc×Tr) where aliasing does not occur.

[0235] Furthermore, since the coded Doppler multiplexed signal to which DCI (ncm, ndm) is assigned is transmitted from the transmitting antenna Tx#[ncm, ndm], it is also possible to determine the transmitting antenna 109. For example, the radar device 10 can separate and receive the coded Doppler multiplexed signal to which DCI (ncm, ndm) is assigned, which is transmitted from the transmitting antenna Tx#[ncm, ndm].

[0236] Furthermore, the radar device 10 performs, for example, a Doppler phase correction including Doppler aliasing (for example, a Doppler phase correction vector α(f s_comp_cfar , D r ) are used for phase correction. These phase corrections are s_comp_cfarThis corresponds to correcting the phase change according to each Doppler component in the Doppler component candidate for the code-multiplexed signals. Therefore, the mutual interference between the code-multiplexed signals can be reduced to, for example, the noise level. For example, in the radar device 10, the inter-symbol interference can be reduced, and the influence of the inter-symbol interference on the deterioration of the detection performance of the radar device 10 can be suppressed.

[0237] An example of the operation of the coded Doppler demultiplexing unit 212 has been described above.

[0238] 1, the Doppler demultiplexing unit 213 includes, for example, Doppler demultiplexing units 213-1 to 213-Loc corresponding to the outputs of the Loc number of Doppler analysis units 210. In the example shown in FIG. 1, Loc=2, and the Doppler demultiplexing unit 213 may include a first Doppler demultiplexing unit 213 (or referred to as the Doppler demultiplexing unit 213-1) and a second Doppler demultiplexing unit 213 (or referred to as the Doppler demultiplexing unit 213-2).

[0239] The first Doppler demultiplexing unit 213 receives the distance index f b_cfar and the Doppler frequency index f s_comp_cfar The first Doppler analysis unit 210 (also referred to as Doppler analysis unit 210-1) outputs the output of the first Doppler analysis unit 210 to the direction estimation unit 214. At this time, the first Doppler demultiplexing unit 213 outputs, for example, the Doppler aliasing determination result D rmin may be used.

[0240] For example, in the example shown in FIG. 1, the first Doppler demultiplexing unit 213 receives the output VFT of the first Doppler analysis unit 210 (Doppler analysis unit 210-1). z 1 (f b_cfar , f s_comp_cfar +(N code F R (D rmin , ndm_BF) / N DM )) to the direction estimation unit 214. Here, ndm_BF is 1, ∼, N DMThe plurality of transmitting antennas 109 to which the ndm_BF-th Doppler multiplexed signal is assigned is, for example, a combination of transmitting antennas 109 that satisfy the above-mentioned condition of adjacent arrangement.

[0241] The first Doppler demultiplexing unit 213 separates the circularly polarized signal using, for example, the output of the first Doppler analysis unit 210. As a result, the first Doppler demultiplexing unit 213 obtains a received signal of a reflected wave due to the transmitted signal, which is a circularly polarized wave. Note that z=1 to Na.

[0242] In addition, in FIG. 1, the second Doppler demultiplexing unit 213 receives the distance index f b_cfar and the Doppler frequency index f s_comp_cfar The second Doppler analysis unit 210 (or the Doppler analysis unit 210-2) outputs the output of the second Doppler analysis unit 210 to the direction estimation unit 214. At this time, the second Doppler demultiplexing unit 213 outputs the output of the Doppler aliasing determination result D rmin may be used.

[0243] For example, in the example shown in FIG. 1, the second Doppler demultiplexing unit 213 receives the output VFT of the second Doppler analysis unit 210 (Doppler analysis unit 210-2). z 2 (f b_cfar , f s_comp_cfar +(N code F R (D rmin , ndm_BF) / N DM )) to the direction estimation unit 214. Here, ndm_BF is 1, ..., N DM The plurality of transmitting antennas 109 to which the ndm_BF-th Doppler multiplexed signal is assigned is, for example, a combination of transmitting antennas 109 that satisfy the above-mentioned condition of adjacent arrangement.

[0244] The second Doppler demultiplexing unit 213 separates the circularly polarized signal using, for example, the output of the second Doppler analysis unit 210. As a result, the second Doppler demultiplexing unit 213 obtains a received signal of a reflected wave from the transmitted signal, which is a circularly polarized wave. Note that z=1 to Na.

[0245] Furthermore, with respect to the received signal of a reflected wave from a circularly polarized transmission signal, which is the output of the first Doppler demultiplexing unit 213, the second Doppler demultiplexing unit 213 receives a received signal of a reflected wave corresponding to the transmission signal, which is a circularly polarized wave with a rotation direction different from the rotation direction of the circularly polarized wave corresponding to the output of the first Doppler demultiplexing unit 213. For example, if the circularly polarized wave corresponding to the output of the first Doppler demultiplexing unit 213 is a left-handed circularly polarized wave, the circularly polarized wave corresponding to the output of the second Doppler demultiplexing unit 213 will be a right-handed circularly polarized wave. Also, for example, if the circularly polarized wave corresponding to the output of the first Doppler demultiplexing unit 213 is a right-handed circularly polarized wave, the circularly polarized wave corresponding to the output of the second Doppler demultiplexing unit 213 will be a left-handed circularly polarized wave.

[0246] Hereinafter, the circularly polarized wave corresponding to the output of the first Doppler demultiplexing unit 213 will be referred to as a "positive circularly polarized wave" based on the rotation of the circularly polarized wave. The circularly polarized wave corresponding to the output of the second Doppler demultiplexing unit 213 will be referred to as a "negative circularly polarized wave" because it is the negative rotation of the positive circularly polarized wave.

[0247] In FIG. 1, the direction estimation unit 214 receives the distance index f b_cfar , the Doppler frequency index f s_comp_cfar Doppler aliasing judgment result D rmin Based on this, the separated received signal Y z (f b_cfar ,f s_comp_cfar ,D rmin , ncm,ndm), target direction estimation processing (hereinafter referred to as "direction estimation processing for linearly polarized waves") is performed.

[0248] In addition, the direction estimation unit 214 performs target direction estimation processing (hereinafter referred to as "direction estimation processing for positive circular polarization") based on the output from the first Doppler analysis unit 210 (Doppler analysis unit 210-1 in Figure 1) input from the first Doppler multiplex separation unit 213.

[0249] In addition, the direction estimation unit 214 performs target direction estimation processing (hereinafter referred to as "direction estimation processing for counter-circularly polarized waves") based on the output from the second Doppler analysis unit 210 (Doppler analysis unit 210-2 in Figure 1) input from the second Doppler multiplexing separation unit 213.

[0250] The direction estimation process in the direction estimation unit 214 may include, for example, a direction estimation process for linearly polarized waves, a direction estimation process for normal circularly polarized waves, and a direction estimation process for counter-circularly polarized waves. Each of these direction estimation processes will be described below.

[0251] <Direction estimation processing for linearly polarized waves> For example, the direction estimator 214 calculates a virtual receiving array correlation vector h(f b_cfar , f s_comp_cfar ) is generated and direction estimation processing is performed.

[0252] Virtual receiving array correlation vector h(f b_cfar , f s_comp_cfar ) includes Nt×Na elements, which is the product of the number of transmitting antennas Nt and the number of receiving antennas Na.

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[0253] The direction estimation unit 214 calculates the virtual receiving array correlation vector h(f b_cfar , f s_comp_cfar ) is used to perform processing to estimate the direction of the reflected wave signal from the target based on the phase difference between the receiving antennas 202.

[0254] Here, the virtual receiving array correlation vector h(fb_cfar , f s_comp_cfar ) includes reflected wave reception signals of signals transmitted from different linearly polarized antennas (for example, vertically polarized antennas and horizontally polarized antennas). Therefore, the direction estimation unit 214 calculates the virtual receiving array correlation vector h(f b_cfar , f s_comp_cfar ), elements that are combinations of the polarization of a given transmitting antenna 109 and the polarization of a receiving antenna 202 may be extracted, and direction estimation processing may be performed using a virtual receiving array correlation vector consisting of the extracted elements. This allows direction estimation results to be obtained for each polarization of a given transmitting antenna 109 and each polarization of a receiving antenna 202.

[0255] <Direction estimation process for positive circularly polarized waves> The direction estimation unit 214 calculates a positive circular polarization virtual receiving array correlation vector h as shown in the following equation (55) based on the output of the first Doppler analysis unit 210 input from the first Doppler demultiplexing unit 213. c1 (f b_cfar , f s_comp_cfar ) is generated and the direction of the target is estimated relative to the normal circular polarization.

[0256] Here, the correlation vector h c1 (f b_cfar , f s_comp_cfar ) is the output of the first Doppler analysis unit 210 (for example, VFT z 1 (f b_cfar , f s_comp_cfar +(N code F R (D rmin , ndm_BF) / N DM The received signal includes a reflected wave signal of a signal that is code-multiplexed using the same Doppler multiplexing based on the above-mentioned 104), beam-transmitted by adjacent transmitting antennas 109 of different linear polarization, and transmitted as a circularly polarized wave. For example, when the beam-transmitting antennas are N BF (e.g., ndm_BF=1,~, N BF ), the positive circularly polarized virtual receiving array correlation vector h c1 (f b_cfar , f s_comp_cfar ) is NBF ×Na elements. The direction estimation unit 214 calculates the positive circular polarization virtual receiving array correlation vector h c1 (f b_cfar , f s_comp_cfar ) may be used to perform processing to estimate the direction of the reflected wave signal from the target based on the phase difference between the receiving antennas 202.

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[0257] <Direction estimation process for counter-circularly polarized waves> The direction estimation unit 214 performs direction estimation processing for the counter-circularly polarized waves of the target based on the output of the second Doppler analysis unit 210 input from the second Doppler demultiplexing unit 213. The direction estimation unit 214 estimates the direction of the counter-circularly polarized waves virtual receiving array correlation vector h c2 (f b_cfar , f s_comp_cfar ) and performs direction estimation processing for the counter-circularly polarized wave of the target.

[0258] Here, the counter-circularly polarized virtual receiving array correlation vector h c2 (f b_cfar , f s_comp_cfar ) is the output of the second Doppler analyzer 210 (for example, VFT z 2 (f b_cfar , f s_comp_cfar +(N code F R (D rmin , ndm_BF) / N DM The received signal includes a reflected wave signal of a signal that is code-multiplexed using the same Doppler multiplexing based on the above-mentioned 104), beam-transmitted by adjacent transmitting antennas 109 of different linear polarization, and transmitted as a circularly polarized wave. For example, when the beam-transmitting antennas are N BF (e.g., ndm_BF=1,~, N BF ), the counter-rotating virtual receiving array correlation vector h c2 (f b_cfar , f s_comp_cfar ) is N BF ×Na elements.

[0259] The direction estimation unit 214 calculates the counter-circularly polarized virtual receiving array correlation vector h c2 (f b_cfar , f s_comp_cfar ) is used to perform processing to estimate the direction of the reflected wave signal from the target based on the phase difference between the receiving antennas 202.

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[0260] [Antenna placement example 1] Hereinafter, examples of direction estimation processing for linearly polarized waves, direction estimation processing for normal circularly polarized waves, and direction estimation processing for counter-circularly polarized waves in the direction estimation unit 214 will be described using examples of antenna arrangements.

[0261] For example, when the number of transmitting antennas used for multiplex transmission is Nt=4, the number of Doppler multiplexing N DM =2, code multiplex number N CM = 2, the orthogonal code sequence Code1 = {1, 1} and Code2 = {j, -j} with code length Loc = 2, and the number of coded Doppler multiplexes is N DOP_CODE (1)=2, N DOP_CODE The case where (2)=2 is explained. Note that the number of beam transmitting antennas N BF = 2, and the index of the Doppler multiplexed signal used for the beam transmitting antenna is ndm _BF =1 and ndm _BF Use =2.

[0262] In FIG. 15, for example, in the radar device 10, the four transmitting antennas 109 (Tx#1, Tx#2, Tx#3, and Tx#4) arranged horizontally are, from the left antenna, a horizontally polarized (represented as "H") transmitting antenna Tx#[1,1], a vertically polarized (represented as "V") transmitting antenna Tx#[2,1], a horizontally polarized transmitting antenna Tx#[1,2], and a vertically polarized transmitting antenna Tx#[2,2].

[0263] In FIG. 15, the two adjacent transmitting antennas Tx#1 (Tx#[1,1]) and Tx#2 (Tx#[2,1]) from the left side code-multiplex and transmit radar transmission signals using the same Doppler multiplexing (Doppler shift amount = DOP1). Therefore, in FIG. 15, a beam transmitting antenna is formed by Tx#1 and Tx#2, and circularly polarized waves are transmitted. In addition, the transmitting antennas Tx#1 (Tx#[1,1]) and Tx#2 (Tx#[2,1]) may switch between transmitting normal circularly polarized waves and transmitting reverse circularly polarized waves in each transmission cycle depending on the phase difference between the codes during code-multiplexing transmission.

[0264] Similarly, in FIG. 15, the two adjacent transmitting antennas Tx#3 (Tx#[1,2]) and Tx#4 (Tx#[2,2]) from the right side code-multiplex and transmit radar transmission signals using the same Doppler multiplexing (Doppler shift amount = DOP2). Therefore, in FIG. 15, Tx#3 and Tx#4 form a beam transmitting antenna and transmit circularly polarized waves. Furthermore, transmitting antennas Tx#1 (Tx#[1,2]) and Tx#2 (Tx#[2,2]) may switch between transmitting normal circularly polarized waves and transmitting reverse circularly polarized waves in each transmission cycle depending on the phase difference between the codes during code-multiplexing transmission.

[0265] In Fig. 15, the number of beam transmitting antennas is N BF = 2. Hereinafter, the beam transmitting antennas by Tx#1 and Tx#2 in Fig. 15 may be referred to as "Tx#5." Also, the beam transmitting antennas by Tx#3 and Tx#4 in Fig. 15 may be referred to as "Tx#6." For example, Tx#5 and Tx#6 can be treated equivalently as antennas that switch between forward and reverse circularly polarized waves for each transmission cycle.

[0266] Here, we will explain the case where Tx#5 and Tx#6 use codes that transmit circularly polarized waves with the same rotation direction, either forward or reverse, for each transmission period, but this is not limited to this, and Tx#5 and Tx#6 may use circularly polarized waves with different rotation directions.

[0267] 15, the number of receiving antennas Na is eight (for example, Rx#1 to Rx#8), including antennas of four different types of polarization. In the example shown in Fig. 15, Rx#1 and Rx#2 are horizontally polarized (H) receiving antennas, Rx#3 and Rx#4 are vertically polarized (V) receiving antennas, Rx#5 and Rx#6 are forward circularly polarized (C) receiving antennas, and Rx#7 and Rx#8 are reverse circularly polarized (R) receiving antennas.

[0268] The number of receiving antennas Na is not limited to eight and may be, for example, another number. The types of polarization used by the receiving antenna 202 are also not limited to four and may be three, two, or one. Furthermore, instead of being limited to horizontally polarized waves and vertically polarized waves, linearly polarized waves tilted in an oblique direction, for example, polarized waves tilted in a ±45° direction, may also be used.

[0269] For example, when radar transmission signals are transmitted at equal power from adjacent Tx#1 (Tx#[1,1]) and Tx#2 (Tx#[2,1]), the midpoint between Tx#1 and Tx#2 becomes the phase center of beam transmitting antenna Tx#5 (indicated by the x symbol in FIG. 15(a)). Similarly, when radar transmission signals are transmitted at equal power from adjacent Tx#3 (Tx#[1,2]) and Tx#4 (Tx#[2,2]), the midpoint between Tx#3 and Tx#4 becomes the phase center of beam transmitting antenna Tx#6 (indicated by the x symbol in FIG. 15(a)).

[0270] In addition, when the radar transmission signals are not transmitted with equal power from the transmitting antennas 109 constituting the beam transmitting antenna, the position corresponding to the ratio of the transmission power of each transmitting antenna 109 constituting the beam transmitting antenna (the position of the center of gravity of the transmission power from each transmitting antenna) can be treated as the transmission from the beam transmitting antenna with the phase center of the sub-array.

[0271] The arrangement of transmitting antennas Tx#1 to Tx#4 (e.g., Nt transmitting antennas 109) and receiving antennas Rx#1 to Rx#8 (e.g., Na receiving antennas 202) as shown in (a) of Figure 15 constitutes the arrangement VA#1 to VA#32 of virtual receiving antennas (or MIMO virtual antennas) as shown in (b) of Figure 15.

[0272] In addition, in FIG. 15(a), the positive circularly polarized beam transmitting antennas Tx#5 and Tx#6 (for example, N BF The arrangements of the transmitting antennas Rx#1 to Rx#8 and the receiving antennas Rx#1 to Rx#8 form arrangements CA#1 to CA#16 of positive circular polarized virtual receiving antennas as shown in (c) of Fig. 15. The arrangements CA#1 to CA#16 of positive circular polarized virtual receiving antennas may also be represented as VA#33 to VA#48, for example.

[0273] In addition, in FIG. 15(a), the reversed circularly polarized beam transmitting antennas Tx#5 and Tx#6 (for example, N BF The arrangements RA#1 to RA#16 of counter-circular polarized virtual receiving antennas are configured as shown in (d) of Figure 15 from the arrangements of the transmitting antennas Rx#1 to Rx#8 and the receiving antennas Rx#1 to Rx#8. The arrangements RA#1 to RA#16 of counter-circular polarized virtual receiving antennas are also represented as VA#33 to VA#48, for example. The arrangements RA#1 to RA#16 of counter-circular polarized virtual receiving antennas are the same as the arrangements CA#1 to CA#16 of forward circular polarized virtual receiving antennas, respectively.

[0274] Here, the arrangement of the virtual receiving antenna (virtual receiving array) may be expressed as in the following equation (57), based on the position (e.g., the position of the feed point) of the transmitting antenna 109 that constitutes the transmitting array antenna and the position (e.g., the position of the feed point) of the receiving antenna 202 that constitutes the receiving array antenna.

number

[0275] Here, the position coordinates of the transmitting antenna 109 (for example, Tx#n) constituting the transmitting array antenna are expressed as (XT_#n ,Y T_#n ) (e.g., n=1,~, Nt+N BF ), and the position coordinates of the receiving antenna 202 (for example, Rx#m) constituting the receiving array antenna are expressed as (X R_#m ,Y R_#m ) (for example, m=1,~,Na), and the position coordinates of the virtual antenna VA#k that constitutes the virtual receiving array antenna are expressed as (X V_#k ,Y V_#k ) (e.g., k=1,~, (Nt+N BF ) × Na).

[0276] In equation (57), for example, VA#1 is expressed as the position reference (0,0) of the virtual receiving array.

[0277] Figure 15(b) shows an example of a virtual receiving antenna arrangement when using the arrangement of Tx#1 to #4 and Rx#1 to #8 shown in Figure 15(a). The virtual receiving antenna includes 32 antenna elements, and their respective arrangements are shown as VA#1 to VA#32.

[0278] Note that the combination of transmitting antenna polarization (e.g., H or V) and receiving antenna polarization (e.g., H, V, C, or R) for each virtual receiving antenna is different for each antenna, and is therefore written as "(transmitting antenna polarization / receiving antenna polarization)." For example, a virtual receiving antenna in the case where horizontally polarized waves are transmitted and horizontally polarized waves are received is written as "H / H." (c) and (d) of Figure 15 are written in the same way. The following notations also follow this format.

[0279] The arrangement of Tx#1 to #4 and Rx#1 to #8 in (a) of Figure 15 is a one-dimensional arrangement on the X axis (horizontal direction in Figure 15), so the virtual receiving antennas are also arranged on the X axis. Note that since some arrangements overlap on the X axis, they are shown shifted vertically in (b) of Figure 15, but are actually arranged one-dimensionally at positions on the X axis. (c) and (d) of Figure 15 are written in the same way. The following notations also follow this format.

[0280] Figure 15(c) shows a virtual receiving antenna arrangement when using antennas Tx#5 and #6 transmitting beams with positive circular polarization and Rx#1 to #8. The virtual receiving antenna is an array arrangement of 16 antenna elements (e.g., CA#1 to CA#16). CA#1 to CA#16 correspond to VA#33 to VA#48, respectively.

[0281] 15(d) shows a virtual receiving antenna arrangement when using antennas Tx#5 and #6 transmitting beams with counter-circular polarization and Rx#1 to #8. The virtual receiving antenna has an antenna arrangement of 16 antenna elements (e.g., RA#1 to RA#16) similar to that shown in FIG. 15(c). RA#1 to RA#16 correspond to VA#33 to VA#48, respectively.

[0282] In this way, the virtual receiving antenna arrangement using the beam transmitting antenna enables the radar device 10 to transmit using more polarized waves. Furthermore, by combining the polarized waves of the transmitting antenna 109 and the polarized waves of the receiving antenna 202, it is possible to obtain received signals with a combination of orthogonal polarized waves (also called "cross-polarized waves") (for example, a combination of horizontally polarized waves and vertically polarized waves, or a combination of left-handed circularly polarized waves and right-handed circularly polarized waves) in addition to a combination of transmitting and receiving antennas with the same polarized waves. The radar device 10 can improve its detection or identification performance by taking advantage of the fact that the reception characteristics of the target's reflected waves change depending on the combination of transmitting and receiving polarized waves.

[0283] <Direction estimation processing for linearly polarized waves> For example, when performing direction estimation processing using a horizontally polarized antenna (H) for both transmission and reception, the direction estimation unit 214 extracts four virtual receiving antennas consisting of VA#1, VA#2, VA#17, and VA#18, and performs direction estimation processing.

[0284] Virtual receiving array correlation vector h(f b_cfar , f s_comp_cfar) is a column vector, and the first element to the Nt×Na-th element contained therein represent the received signals of virtual antennas VA#1 to VA# (Nt×Na). For example, the direction estimation unit 214 calculates a virtual receiving array correlation vector h(f b_cfar , f s_comp_cfar The direction of arrival is estimated using a partial array correlation vector obtained by extracting the 1st, 2nd, 17th, and 18th elements of h sub (f b_cfar , f s_comp_cfar ) and h sub (f b_cfar , f s_comp_cfar ) the number of dimensions is N sub Let's say.

[0285] The direction estimation unit 214 calculates, for example, a direction estimation evaluation function value P H (θ, f b_cfar , f s_comp_cfar ) is varied within a specified angle range to calculate a spatial profile. The direction estimation unit 214 extracts a predetermined number of maximum peaks from the calculated spatial profile in descending order, and outputs the azimuth direction of the maximum peak as an arrival direction estimate (e.g., positioning output).

[0286] The direction estimation evaluation function value P H (θ, f b_cfar , f s_comp_cfar There are various methods for estimating the direction of arrival (DOA) depending on the algorithm. For example, the estimation method using an array antenna disclosed in Non-Patent Document 3 may be used.

[0287] For example, the beamformer method can be expressed as in the following equation (58): Other methods such as Capon and MUSIC can also be applied in the same way.

number

[0288] Here, in equation (58), the superscript H is the Hermitian transpose operator.sub (θ u ) is the azimuth direction θ u For example, if the virtual receiving antennas consisting of VA#1, #2, #17, and #18 are arranged at equal intervals D H When the virtual receiving arrays VA#1, VA#2, VA#17, and VA#18 are linearly arranged, the direction vectors of the virtual receiving arrays VA#1, VA#2, VA#17, and VA#18 can be expressed as in the following equation (59).

number

[0289] Also, the azimuth direction θ u is a vector obtained by varying the azimuth interval β1 within the azimuth range for estimating the direction of arrival. For example, θ u is set as follows: θ u =θmin + uβ1, u=0,…, NU NU=floor[(θmax-θmin) / β1]+1 Here, floor(x) is a function that returns the largest integer value that does not exceed the real number x.

[0290] Also, in equation (58), D cal is an array correction coefficient for correcting the phase deviation and amplitude deviation between the antennas of the partial virtual receiving arrays VA#1, #2, #17, and #18, and N including a coefficient for reducing the influence of inter-element coupling between the antennas. sub If the coupling between the antennas of the virtual receiving array can be ignored, then D cal is a diagonal matrix, and the diagonal elements include array correction coefficients that correct the phase deviation and amplitude deviation between the transmitting array antennas and the receiving array antennas.

[0291] Furthermore, for example, when performing direction estimation processing when a vertically polarized antenna is used for both transmission and reception, the direction estimation unit 214 extracts four virtual receiving antennas consisting of VA#11, VA#12, VA#27, and VA#28, and performs direction estimation processing similar to the processing described above.

[0292] Furthermore, for example, the direction estimation unit 214 may perform direction estimation processing using cross-polarized waves. For example, when performing direction estimation processing using a vertically polarized antenna for transmission and a horizontally polarized antenna for reception, the direction estimation unit 214 extracts four virtual receiving antennas consisting of VA#9, VA#10, VA#25, and VA#26, and performs direction estimation processing similar to the processing described above.

[0293] The combination of the polarized antenna for transmission and the polarized antenna for reception is not limited to the above, and different combinations of polarized antennas for transmission and reception may be used.

[0294] For example, the direction estimation unit 214 is not limited to direction estimation using a virtual receiving array with the same combination of transmitting and receiving polarized antennas, but may also perform direction estimation using a virtual receiving array with a different combination of transmitting and receiving polarized antennas. For example, the direction estimation unit 214 may extract all virtual receiving antennas obtained from the output of the coded Doppler demultiplexing unit 212, consisting of VA#1 to VA#32, and perform direction estimation similar to the above-described process. In this case, the direction estimation unit 214 performs direction estimation using received signals containing different types of polarization combinations (e.g., H / V, H / C, etc.), thereby obtaining direction estimation results that are less dependent on polarization. Furthermore, since the direction estimation unit 214 performs direction estimation using all virtual receiving antennas, the received SNR is improved, thereby improving the detection performance of the radar device 10. Furthermore, since the direction estimation process is performed using the maximum aperture length of the available virtual receiving antennas, angular resolution is also improved.

[0295] The direction estimation unit 214 may perform, for example, a direction estimation process using the same type of polarized antenna for both transmission and reception as described above, and a direction estimation process combining different types of polarization, and use both of these direction estimation results as the direction estimation process result. This provides a direction estimation process result that is highly dependent on polarization and a direction estimation process result that is less dependent on polarization. The results of such direction estimation processes may be input to a target identification processing unit (not shown), where target identification processing is performed.

[0296] <Direction estimation process for positive circularly polarized waves> For example, when performing direction estimation processing using a circularly polarized antenna (C) for both transmission and reception, the direction estimation unit 214 extracts four virtual receiving antennas consisting of CA#5, CA#6, CA#13, and CA#14, and performs direction estimation processing.

[0297] Circularly polarized virtual receiving array correlation vector h c1 (f b_cfar , f s_comp_cfar ) is a column vector, containing elements 1 to N BF ×Na-th element is the virtual antenna VA#(Nt×Na+1) to VA#(Nt+N BF ) × Na received signal, or CA#1 to CA#(N BF ×Na) is a received signal. For example, the direction estimation unit 214 calculates a positive circular polarization virtual receiving array correlation vector h corresponding to the received signals of CA#5, CA#6, CA#13, and CA#14. c1 (f b_cfar , f s_comp_cfar The direction of arrival is estimated using the partial array correlation vector obtained by extracting the 5th, 6th, 13th, and 14th elements of h c1sub (f b_cfar , f s_comp_cfar ) and h c1sub (f b_cfar , f s_comp_cfar ) the number of dimensions is N c1sub Let's say.

[0298] The direction estimation unit 214 calculates, for example, a direction estimation evaluation function value P Hc1 (θ, f b_cfar , f s_comp_cfar ) is varied within a specified angle range to calculate a spatial profile. The direction estimation unit 214 extracts a predetermined number of maximum peaks from the calculated spatial profile in descending order, and outputs the azimuth direction of the maximum peak as an arrival direction estimate (e.g., positioning output).

[0299] The direction estimation evaluation function value P Hc1 (θ, fb_cfar , f s_comp_cfar There are various methods for estimating the direction of arrival (DOA) depending on the algorithm. For example, the estimation method using an array antenna disclosed in Non-Patent Document 3 may be used.

[0300] For example, the beamformer method can be expressed as in the following equation (60): Other methods such as Capon and MUSIC can also be applied in the same way.

number

[0301] Also, a c1sub (θ u ) is the azimuth direction θ u 10 shows the direction vectors of the extracted partial virtual receiving arrays CA#5, CA#6, CA#13, and CA#14 with respect to the incoming wave.

[0302] Also, D c1cal is an N-th element including an array correction coefficient for correcting the phase deviation and amplitude deviation between the antennas of the partial virtual receiving arrays CA#5, CA#6, CA#13, and CA#14 and a coefficient for reducing the influence of inter-element coupling between the antennas. c1sub If the coupling between the antennas of the virtual receiving array can be ignored, then D c1cal is a diagonal matrix, and the diagonal elements include array correction coefficients that correct the phase deviation and amplitude deviation between the transmitting array antennas and the receiving array antennas.

[0303] Furthermore, for example, the direction estimation unit 214 may perform direction estimation processing using cross-polarized waves. For example, when performing direction estimation processing using a forward circularly polarized antenna for transmission and a reverse circularly polarized antenna for reception, the direction estimation unit 214 extracts four virtual receiving antennas consisting of CA#7, CA#8, CA#15, and CA#16, and performs direction estimation processing similar to the processing described above.

[0304] For example, the direction estimation unit 214 is not limited to direction estimation using a virtual receiving array with the same combination of transmitting and receiving polarized antennas, but may also perform direction estimation using a virtual receiving array with a different combination of transmitting and receiving polarized antennas. For example, the direction estimation unit 214 may extract all virtual receiving antennas obtained from the output of the first Doppler demultiplexing unit 213, which includes CA#1 to CA#16, and perform direction estimation similar to the above-described process. In this case, the direction estimation unit 214 performs direction estimation using received signals containing different types of polarization combinations (e.g., C / H, C / V, etc.), thereby obtaining direction estimation results with low polarization dependency. Furthermore, since the direction estimation unit 214 performs direction estimation using all virtual receiving antennas, the reception SNR is improved, thereby improving the detection performance of the radar device 10. Furthermore, since the direction estimation process is performed using the maximum aperture length of the available virtual receiving antennas, angular resolution is also improved.

[0305] The direction estimation unit 214 may perform, for example, a direction estimation process using the same type of polarized antenna for both transmission and reception as described above, and a direction estimation process combining different types of polarization, and use both of these direction estimation results as the direction estimation process result. This allows for a direction estimation process result that is highly dependent on polarization and a direction estimation process result that is less dependent on polarization. The results of such direction estimation processes may be input to a target identification processing unit (not shown), where target identification processing is performed.

[0306] <Direction estimation process for counter-circularly polarized waves> For example, when performing direction estimation processing using a counter-circularly polarized antenna for both transmission and reception, the direction estimation unit 214 extracts four virtual receiving antennas consisting of RA#7, RA#8, RA#15, and RA#16, and performs direction estimation processing.

[0307] Counter-circularly polarized virtual receiving array correlation vector h c2 (f b_cfar , f s_comp_cfar ) is a column vector, containing elements 1 to N BF×Na-th element is the virtual antenna VA#(Nt×Na+1) to VA#(Nt+N BF ) × Na received signals, or RA#1 to RA#(N BF ×Na) are received signals. For example, the direction estimation unit 214 calculates the counter-circularly polarized virtual receiving array correlation vectors h corresponding to the received signals RA#7, RA#8, RA#15, and RA#16. c2 (f b_cfar , f s_comp_cfar The direction of arrival is estimated using the partial array correlation vector obtained by extracting the 7th, 8th, 15th, and 16th elements of h c2sub (f b_cfar , f s_comp_cfar ) and h c2sub (f b_cfar , f s_comp_cfar ) the number of dimensions is N c2sub Let's say.

[0308] The direction estimation unit 214 calculates, for example, a direction estimation evaluation function value P Hc2 (θ, f b_cfar , f s_comp_cfar ) is varied within a specified angle range to calculate a spatial profile. The direction estimation unit 214 extracts a predetermined number of maximum peaks from the calculated spatial profile in descending order, and outputs the azimuth direction of the maximum peak as an arrival direction estimate (e.g., positioning output).

[0309] The direction estimation evaluation function value P Hc2 (θ, f b_cfar , f s_comp_cfar There are various methods for estimating the direction of arrival (DOA) depending on the algorithm. For example, the estimation method using an array antenna disclosed in Non-Patent Document 3 may be used.

[0310] For example, the beamformer method can be expressed as in the following equation (61): Other methods such as Capon and MUSIC can also be applied in the same way.

number

[0311] Also, a c2sub (θ u ) is the azimuth direction θ u 10 shows the direction vectors of the extracted partial virtual receiving arrays RA#7, RA#8, RA#15, and RA#16 for the arriving waves of .

[0312] Also, D c2cal is an N-ary array including an array correction coefficient for correcting the phase deviation and amplitude deviation between the antennas of the partial virtual receiving arrays RA#7, RA#8, RA#15, and RA#16 and a coefficient for reducing the influence of inter-element coupling between the antennas. c2sub If the coupling between the antennas of the virtual receiving array can be ignored, then D c2cal is a diagonal matrix, and the diagonal elements include array correction coefficients that correct the phase deviation and amplitude deviation between the transmitting array antennas and the receiving array antennas.

[0313] Furthermore, for example, the direction estimation unit 214 may perform direction estimation processing using cross-polarized waves. For example, when performing direction estimation processing using a reverse circularly polarized antenna for transmission and a forward circularly polarized antenna for reception, the direction estimation unit 214 extracts four virtual receiving antennas consisting of RA#5, RA#6, RA#13, and RA#14, and performs direction estimation processing similar to the processing described above.

[0314] For example, the direction estimation unit 214 is not limited to direction estimation using a virtual receiving array with the same combination of transmitting and receiving polarized antennas, but may also perform direction estimation using a virtual receiving array with a different combination of transmitting and receiving polarized antennas. For example, the direction estimation unit 214 may extract all virtual receiving antennas obtained from the output of the second Doppler demultiplexing unit 213, consisting of RA#1 to RA#16, and perform direction estimation similar to the above-described process. In this case, the direction estimation unit 214 performs direction estimation using received signals containing different types of polarization combinations (e.g., R / H, R / V, etc.), thereby obtaining direction estimation results with low polarization dependency. Furthermore, since the direction estimation unit 214 performs direction estimation using all virtual receiving antennas, the reception SNR is improved, thereby improving the detection performance of the radar device 10. Furthermore, since the direction estimation process is performed using the maximum aperture length of the available virtual receiving antennas, angular resolution is also improved.

[0315] The direction estimation unit 214 may perform, for example, a direction estimation process using the same type of polarized antenna for both transmission and reception as described above, and a direction estimation process combining different types of polarization, and use both of these direction estimation results as the direction estimation process result. This allows for a direction estimation process result that is highly dependent on polarization and a direction estimation process result that is less dependent on polarization. The results of such direction estimation processes may be input to a target identification processing unit (not shown), where target identification processing is performed.

[0316] An example of the direction estimation process for counter-circularly polarized waves has been described above.

[0317] Furthermore, the direction estimation unit 214 may perform direction estimation processing using the outputs of the first and second Doppler demultiplexing units 213, for example.

[0318] For example, when a circularly polarized antenna (e.g., C / C) is used for both transmission and reception, and a counter-circularly polarized antenna (e.g., R / R) is used for both transmission and reception, the direction estimation unit 214 extracts four virtual receiving antennas consisting of CA#5, CA#6, CA#13, and CA#14 from the output of the first Doppler demultiplexing unit 213, extracts four virtual receiving antennas consisting of RA#7, RA#8, RA#15, and RA#16 from the output of the second Doppler demultiplexing unit 213, and calculates the extracted partial array correlation vectors as h c1sub (f b_cfar , f s_comp_cfar ) and h c2sub (f b_cfar , f s_comp_cfar ) is used to perform direction estimation processing.

[0319] The direction estimation unit 214 calculates, for example, a direction estimation evaluation function value P Hc12 (θ, f b_cfar , f s_comp_cfar ) is varied within a specified angle range to calculate a spatial profile. The direction estimation unit 214 extracts a predetermined number of maximum peaks from the calculated spatial profile in descending order, and outputs the azimuth direction of the maximum peak as an arrival direction estimate (e.g., positioning output).

[0320] The direction estimation evaluation function value P Hc12 (θ, f b_cfar , f s_comp_cfar There are various methods for estimating the direction of arrival (DOA) depending on the algorithm. For example, the estimation method using an array antenna disclosed in Non-Patent Document 3 may be used.

[0321] For example, the beamformer method can be expressed as in the following equation (62): Other methods such as Capon and MUSIC can also be applied in the same way.

number

[0322] Furthermore, for example, when a combination of circularly polarized antennas (e.g., C / R and R / C) in which transmission and reception are cross-polarized is used, the direction estimation unit 214 extracts four virtual receiving antennas consisting of CA#7, CA#8, CA#15, and CA#16 from the output of the first Doppler demultiplexing unit 213, extracts four virtual receiving antennas consisting of RA#5, RA#6, RA#13, and RA#14 from the output of the second Doppler demultiplexing unit 213, and calculates the respective extracted partial array correlation vectors as h c1sub (f b_cfar , f s_comp_cfar ) reaches h c2sub (f b_cfar , f s_comp_cfar ) is used to perform a similar direction estimation process.

[0323] In this way, when the direction estimation unit 214 performs direction estimation processing using the outputs of the first and second Doppler demultiplexing units 213, it can use more virtual antennas, improving the reception SNR and the target detection performance of the radar device 10.

[0324] Furthermore, the direction estimating unit 214 may perform direction estimation processing using the output of the coded Doppler demultiplexing unit 212 and either or both of the outputs of the first and second Doppler demultiplexing units 213 .

[0325] For example, the direction estimation unit 214 extracts all virtual receiving antennas obtained from the output of the coded Doppler demultiplexing unit 212 consisting of VA#1 to #32, extracts all virtual receiving antennas obtained from the output of the first Doppler demultiplexing unit consisting of CA#1 to CA#16, extracts all virtual receiving antennas obtained from the output of the second Doppler demultiplexing unit 213 consisting of RA#1 to RA#16, and calculates each of the extracted partial array correlation vectors as h sub (f b_cfar , f s_comp_cfar ), h c1sub (f b_cfar , f s_comp_cfar ) reaches h c2sub (f b_cfar , f s_comp_cfar ) is used to perform direction estimation processing.

[0326] The direction estimation unit 214 calculates, for example, a direction estimation evaluation function value P Hall (θ, f b_cfar , f s_comp_cfar ) is varied within a specified angle range to calculate a spatial profile. The direction estimation unit 214 extracts a predetermined number of maximum peaks from the calculated spatial profile in descending order, and outputs the azimuth direction of the maximum peak as an arrival direction estimate (e.g., positioning output).

[0327] The direction estimation evaluation function value P Hall (θ, f b_cfar , f s_comp_cfar There are various methods for estimating the direction of arrival (DOA) depending on the algorithm. For example, the estimation method using an array antenna disclosed in Non-Patent Document 3 may be used.

[0328] For example, the beamformer method can be expressed as in the following equation (63): Other methods such as Capon and MUSIC can also be applied in the same way.

number

[0329] In the above case, the direction estimation unit 214 performs direction estimation processing using received signals including a combination of different types of polarization, thereby obtaining direction estimation results that are less dependent on polarization. Furthermore, since the direction estimation unit 214 performs direction estimation processing using all virtual receiving antennas, the receiving SNR is improved, and the detection performance of the radar device 10 can be improved. Furthermore, since the direction estimation processing is performed using the maximum aperture length of the available virtual receiving antennas, the angular resolution is also improved. Since more virtual antennas can be used, the receiving SNR is improved, and target detection performance can be improved.

[0330] Here, for example, to configure two-transmission MIMO for each of four polarizations, i.e., horizontal polarization, vertical polarization, forward circular polarization, and reverse circular polarization, eight transmission antennas are used in existing methods. In contrast, in this embodiment, to configure two-transmission MIMO for each of the four polarizations, four transmission antennas 109 can be used, resulting in an effect of reducing the number of transmission antennas. Furthermore, in this embodiment, code-multiplexed transmission is performed on Doppler-multiplexed signals, thereby shortening the transmission time. For example, compared to switching four transmission antennas in a time-division manner, an effect of halving the transmission time can be obtained.

[0331] The direction estimation unit 214 may also perform direction estimation processing using received signals that include a combination of different types of polarization. In this case, direction estimation results that are less dependent on polarization can be obtained, and performing direction estimation processing using more virtual antennas improves the reception SNR, thereby improving the detection performance of the radar device 10. In addition, since direction estimation processing is performed using the maximum aperture length of the available virtual receiving antennas, angular resolution is also improved. For example, when direction estimation processing is performed using all virtual receiving antennas, a maximum of (Nt×Na+2×N BF ×Na) virtual receive antennas can be used, and more than Nt ×Na virtual receive antennas can be used.

[0332] The direction estimation unit 214 may also perform, for example, direction estimation processing using the same type of polarized antenna for both transmission and reception as described above, and direction estimation processing combining different types of polarization, and use both of these direction estimation results as the direction estimation processing result. This provides a direction estimation processing result that is highly dependent on polarization and a direction estimation processing result that is less dependent on polarization. The results of such direction estimation processing may be input to a target identification processing unit (not shown), where target identification processing is performed.

[0333] The direction estimation unit 214 outputs, for example, a distance index f b_cfarand Doppler velocity information of the target based on the Doppler frequency determination result of the target (the Doppler aliasing determination process result in the coded Doppler demultiplexing unit 212).

[0334] When, for example, equation (5) is used as the amount of phase rotation, the Doppler frequency information is D rmin Using this, it can be calculated in an expanded range as shown in the following equation (64).

number

[0335] Furthermore, when the amount of phase rotation is expressed by, for example, Equation (6), the Doppler frequency information is expressed by D rmin Using this, it can be calculated in an expanded range as in the following equation (65).

number

[0336] The Doppler frequency information may be converted into a relative velocity component and output. rmin Using the Doppler frequency index f out The relative velocity component v d (f out ), the following equation (66) may be used. Here, λ is the wavelength of the carrier frequency of the RF signal output from the transmitting radio unit (not shown) (when a chirp signal is used, the wavelength at the center frequency of the chirp signal is used). Also, Δ f is the Doppler frequency interval in the FFT processing in the Doppler analysis unit 210. For example, in this embodiment, Δ f =1 / {N code ×Loc×T r}.

number

[0337] In the above antenna arrangement example, four types of polarized wave receiving antennas are used, but the present invention is not limited to this, and two types of polarized wave receiving antennas may be used, for example. In this case, two types of polarized wave receiving antennas are included, for example, circularly polarized waves (right-handed polarized waves, left-handed polarized waves) or different linearly polarized waves (for example, horizontally or vertically polarized waves).

[0338] Alternatively, a single type of polarized receiving antenna may be used, such as a circularly polarized receiving antenna (right-handed or left-handed) or a different linearly polarized receiving antenna (e.g., horizontal or vertical).

[0339] Below, examples of arrangements in which the receiving antenna 202 has two types of polarized waves and one type will be described.

[0340] [Antenna placement example 2] FIG. 16 shows an example of antenna arrangement when the receiving antenna 202 has two types of polarized waves (for example, C and R).

[0341] As shown in (a) of Figure 16, the arrangement of transmitting antennas Tx#1 to Tx#4 (e.g., Nt transmitting antennas 109) and receiving antennas Rx#1 to Rx#8 (e.g., Na receiving antennas) forms arrangements VA#1 to VA#32 of virtual receiving antennas (or MIMO virtual antennas) as shown in (b) of Figure 16.

[0342] In addition, in FIG. 16(a), the positive circularly polarized beam transmitting antennas Tx#5 and Tx#6 (for example, N BF The arrangements of the transmitting antennas Rx#1 to Rx#8 and the receiving antennas Rx#1 to Rx#8 form arrangements CA#1 to CA#16 of positive circular polarized virtual receiving antennas as shown in (c) of Fig. 16. The arrangements CA#1 to CA#16 of positive circular polarized virtual receiving antennas may also be represented as VA#33 to VA#48, for example.

[0343] In addition, in FIG. 16(a), the reversed circularly polarized beam transmitting antennas Tx#5 and Tx#6 (for example, N BF The arrangements of the transmitting antennas Rx#1 to Rx#8 and the receiving antennas Rx#1 to Rx#8 form arrangements RA#1 to RA#16 of counter-circularly polarized virtual receiving antennas as shown in (d) of Figure 16. The arrangements RA#1 to RA#16 of counter-circularly polarized virtual receiving antennas are also represented as VA#33 to VA#48, for example. The arrangements RA#1 to RA#16 of counter-circularly polarized virtual receiving antennas are the same as the arrangements CA#1 to CA#16 of forward-circularly polarized virtual receiving antennas, respectively.

[0344] Note that (b) to (d) of Figure 16 include overlapping arrangements on the X axis (horizontal direction in Figure 16), so in (b) to (d) of Figure 16, the arrangements are shown shifted vertically, but are each arranged one-dimensionally at a position on the X axis. Also, since the combination of transmitting antenna polarization and receiving antenna polarization for each virtual receiving antenna differs for each antenna, it is written as (transmitting antenna polarization / receiving antenna polarization). For example, a virtual receiving antenna that transmits horizontally polarized waves and receives normal circularly polarized waves is written as "H / C."

[0345] In addition, in (a) of Figure 16, as an example, a case has been described in which the receiving antenna 202 includes a forward circularly polarized antenna and a reverse circularly polarized antenna, but the types and combinations of polarized antennas included in the receiving antenna 202 are not limited to this, and other types and combinations of polarized antennas may also be applied.

[0346] [Antenna placement example 3] FIG. 17 shows an example of antenna arrangement when the receiving antenna 202 has one type of polarization (for example, C).

[0347] As shown in (a) of Figure 17, the arrangement of transmitting antennas Tx#1 to Tx#4 (e.g., Nt transmitting antennas 109) and receiving antennas Rx#1 to Rx#8 (e.g., Na receiving antennas) forms arrangements VA#1 to VA#32 of virtual receiving antennas (or MIMO virtual antennas) as shown in (b) of Figure 17.

[0348] In addition, in FIG. 17(a), the positive circularly polarized beam transmitting antennas Tx#5 and Tx#6 (for example, N BF The arrangements of the transmitting antennas Rx#1 to Rx#8 and the receiving antennas Rx#1 to Rx#8 form arrangements CA#1 to CA#16 of positive circular polarized virtual receiving antennas as shown in (c) of Fig. 17. The arrangements CA#1 to CA#16 of positive circular polarized virtual receiving antennas may also be represented as VA#33 to VA#48, for example.

[0349] In addition, in FIG. 17(a), the reversed circularly polarized beam transmitting antennas Tx#5 and Tx#6 (for example, N BF The arrangements of the transmitting antennas Rx#1 to Rx#8 and the receiving antennas Rx#1 to Rx#8 form arrangements RA#1 to RA#16 of counter-circularly polarized virtual receiving antennas as shown in (d) of Figure 17. The arrangements RA#1 to RA#16 of counter-circularly polarized virtual receiving antennas are also represented as VA#33 to VA#48, for example. The arrangements RA#1 to RA#16 of counter-circularly polarized virtual receiving antennas are the same as the arrangements CA#1 to CA#16 of forward-circularly polarized virtual receiving antennas, respectively.

[0350] Note that Figure 17(b) includes overlapping arrangements on the X-axis (horizontal direction in Figure 17), so they are shown shifted vertically in Figure 17(b), but are each arranged one-dimensionally at positions on the X-axis. Also, the combination of transmitting antenna polarization and receiving antenna polarization for each virtual receiving antenna is different for each antenna, so it is described as (transmitting antenna polarization / receiving antenna polarization). For example, a virtual receiving antenna that transmits horizontally polarized waves and receives vertically polarized waves is described as "H / V."

[0351] In (a) of Figure 17, as an example, a case has been described in which the receiving antenna 202 includes a normal circularly polarized antenna, but the type of polarized antenna included in the receiving antenna 202 is not limited to this, and other types of polarized antennas may also be applied.

[0352] In the above-described antenna arrangement examples 1 to 3, virtual receiving antennas with the same polarization as the transmitting and receiving antennas are arranged at uniform intervals DH (For example, D H The above example has been explained in which the antennas are arranged at intervals of 0.5 wavelengths. However, the arrangement is not limited to this. The antennas may be arranged at intervals of D H (For example, D H An example of such an antenna arrangement will be described below.

[0353] [Antenna placement example 4] Figure 18 shows a virtual receiving antenna arrangement with a spacing D H (For example, D H This shows an arrangement including a 0.5 wavelength spacing.

[0354] As shown in (a) of Figure 18, the arrangement of transmitting antennas Tx#1 to Tx#4 (e.g., Nt transmitting antennas 109) and receiving antennas Rx#1 to Rx#4 (e.g., Na receiving antennas) forms arrangements VA#1 to VA#16 of virtual receiving antennas (or MIMO virtual antennas) as shown in (b) of Figure 18.

[0355] In addition, in FIG. 18(a), the positive circularly polarized beam transmitting antennas Tx#5 and Tx#6 (for example, N BF The arrangements of the transmitting antennas CA#1 to CA#8 and the receiving antennas Rx#1 to Rx#8 form positive circular polarized virtual receiving antenna arrangements CA#1 to CA#8 as shown in (c) of Fig. 18. The positive circular polarized virtual receiving antenna arrangements CA#1 to CA8# are also represented as VA#17 to VA#24, for example.

[0356] In addition, in FIG. 18(a), the reversed circularly polarized beam transmitting antennas Tx#5 and Tx#6 (for example, N BFThe arrangements of the transmitting antennas Rx#1 to Rx#8 and the receiving antennas Rx#1 to Rx#8 form the arrangements RA#1 to RA#8 of counter-circularly polarized virtual receiving antennas as shown in (d) of Figure 18. The arrangements RA#1 to RA#8 of counter-circularly polarized virtual receiving antennas are also represented as VA#17 to VA#24, for example. The arrangements RA#1 to RA#8 of counter-circularly polarized virtual receiving antennas are the same as the arrangements CA#1 to CA#8 of forward-circularly polarized virtual receiving antennas, respectively.

[0357] Note that (b) of Figure 18 includes overlapping arrangements on the X-axis (horizontal direction in Figure 18), so they are shown shifted vertically in (b) of Figure 18, but are each arranged one-dimensionally at positions on the X-axis. Also, the combination of transmitting antenna polarization and receiving antenna polarization for each virtual receiving antenna is different for each antenna, so it is described as (transmitting antenna polarization / receiving antenna polarization). For example, a virtual receiving antenna that transmits horizontally polarized waves and receives vertically polarized waves is described as "H / V."

[0358] As shown in FIG. 18(a), the distance between the beam transmitting antennas Tx#5 and Tx#6 is 2D. H (For example, D H = 0.5 wavelength intervals, 1 wavelength), and the intervals between the receiving antennas Rx#1 to Rx#4 are set to 3D H (For example, D H 18(c) and the reversed circularly polarized virtual receiving antenna arrangement shown in FIG. 18(d) have the spacing D H and interval 2D H As a result, the direction estimation for the normal and reverse circularly polarized waves becomes a virtual receiving antenna arrangement including a spacing of 2D, and while expanding the aperture length, the spacing D H This arrangement allows for enhanced angular resolution while also suppressing grating lobes.

[0359] In antenna arrangement example 4, the beam transmitting antennas Tx#5 and Tx#6 transmit beams using a plurality of transmitting antennas 109, so that the spacing between the beam transmitting antennas is equal to or greater than one wavelength. Therefore, the difference between the spacing between the beam transmitting antennas and the spacing between the receiving antennas 202 is the spacing D H The antenna arrangement is such that the spacing between the forward and reverse circularly polarized virtual receiving antennas is D H Including interval D H This results in a virtual receiving antenna arrangement that includes a wider spacing than D. This allows for direction estimation for normal and reverse circular polarization by enlarging the aperture length and increasing the spacing D. H Since the optical fiber includes the optical fiber, the angular resolution can be improved and grating lobes can be suppressed.

[0360] In the above-described example of arrangement, the virtual receiving antennas are arranged one-dimensionally in the horizontal direction, for example, but the arrangement is not limited to this, and transmitting and receiving antennas arranged on a two-dimensional plane including the horizontal and vertical directions may be used. By using such an arrangement, it is possible to obtain a two-dimensional direction estimation result in the horizontal and vertical directions, for example, as the target direction estimation process.

[0361] Also, D H is not limited to 0.5 wavelength, and may be set to, for example, about 0.45 wavelength to 0.8 wavelength (for example, any value in the range of 0.45λ to 0.8λ).

[0362] The above describes examples of antenna arrangements.

[0363] As described above, in this embodiment, the radar device 10 multiplexes and transmits, from the multiple transmitting antennas 109 in each transmission period, radar transmission signals to which a phase rotation amount (for example, a phase rotation amount corresponding to an orthogonal code sequence) has been imparted such that the phase between the horizontally polarized antenna and the vertically polarized antenna, which are at least one pair of adjacent transmitting antennas 109, differs by 90° or −90°.

[0364] As a result, the received signals for each transmission cycle corresponding to signals transmitted from at least one pair of adjacent transmitting antennas 109 can be regarded as received signals corresponding to radar transmission signals transmitted with circular polarization (for example, normal circular polarization or reverse circular polarization) different from the linear polarization (for example, horizontal polarization or vertical polarization) corresponding to each transmitting antenna 109. Therefore, the radar device 10 can perform multiplexed transmission using circular polarization in addition to linear polarization by using, for example, multiple transmitting antennas 109 that are linearly polarized antennas.

[0365] Therefore, according to this embodiment, even when transmitting four types of polarized waves, for example, left-handed circularly polarized waves and right-handed circularly polarized waves in addition to vertically polarized waves and horizontally polarized waves, it is possible to improve the target detection accuracy of the radar device 10 while suppressing an increase in the number of transmitting antennas 109 used.

[0366] The codes Code1 and Code2 used in this embodiment represent codes when the phase deviation between the transmitting antennas 109 is corrected in advance. Therefore, the phase difference at the feed point of each transmitting antenna 109 is the phase difference between the code elements of the codes Code1 and Code2 assigned to each transmitting antenna 109. Here, when the radar device 10 transmits using Code1={OC1(1), OC1(2)} and Code2={OC2(1), OC2(2)}, the phase difference between each code element is given by angle[OC2(1)]-angle[OC1(1)] and angle[OC2(2)]-angle[OC1(2)]. Therefore, in this embodiment, the radar device 10 transmits N signals with a code length Loc=2 using the same Doppler multiplexed signal from the two transmitting antennas 109. CM For example, when transmitting using Code1={1, 1} and Code2={j, -j} as orthogonal code sequences for OC2(1), OC1(2), OC2(3), OC1(4), OC2(5), OC3(6), OC4(7), OC5(8), OC6(9), OC7(10), OC8(11), OC9(12), OC101, OC112, OC113, OC114, OC115, OC116, OC117, OC118, OC119, OC220, OC221, OC222, OC223, OC224, OC225, OC226, OC227, OC228, OC229, OC230, OC231, OC232, OC233, OC234, OC

[0367] Similarly, in this embodiment, the radar device 10 transmits N signals having a code length Loc=2 using the same Doppler multiplexed signals from the two transmitting antennas 109. CM For example, when transmitting using Code1={A, B} and Code2={-j×A, j×B} as orthogonal code sequences of OC2(1), OC1(2), OC2(3), OC1(4), OC2(5), OC1(6), OC2(7), OC2(8), OC2(9), OC3(10), OC4(11), OC5(12), OC6(13), OC7(14), OC8(15), OC9(16), OC10(17), OC11(18), OC12(19), OC13(20), OC14(21), OC15(22), OC16(23), OC17(24), OC18(25), OC19(26), OC20, OC21, OC3(10), OC4(11), OC5(12), OC6(

[0368] Similarly, in this embodiment, the radar device 10 transmits N signals having a code length Loc=2 using the same Doppler multiplexed signals from the two transmitting antennas 109. CM For example, when transmitting using Code1={A, B} and Code2={exp(jξ)×A, -exp(jξ)×B} as orthogonal code sequences with θ=2, the phase difference between the feed points of the two transmitting antennas 109 is angle[OC2(1)]-angle[OC1(1)]=ξ and angle[OC2(2)]-angle[OC1(2)]=-ξ, and the phase difference between the feed points of the two transmitting antennas for each transmission period is ξ or -ξ. Here, ξ may be in the range of π / 6 to 5π / 6 radians (=30° to 150°), for example. The same applies to the following Modification 1 and Modification 2.

[0369] (First Modification of First Embodiment) In the first embodiment, the Doppler multiplexing number N DM = 1, and MIMO multiplexing transmission may be performed using code multiplexing without using Doppler multiplexing.

[0370] 19 shows a configuration example of a radar device 10a according to Modification 1 of Embodiment 1. Since Modification 1 of Embodiment 1 does not use Doppler multiplexing, the configuration related to the transmission operation and reception operation of Doppler multiplexing (for example, the Doppler shift setting unit 106 and the coded Doppler demultiplexing unit 212) is removed from the configuration of the radar device 10 in FIG. 1, and a code demultiplexing unit 215 is provided instead of the coded Doppler demultiplexing unit 212.

[0371] In the example of FIG. 19, the radar device 10a has a Doppler multiplexing number N DM =1, and code multiplexing is performed using a code with a code length of 2. Therefore, in FIG. 19, the configuration has the number of transmitting antennas Nt=2. The transmitting antenna 109 may perform code-multiplexing transmission as in the first embodiment by using, for example, different linearly polarized antennas (for example, horizontally polarized and vertically polarized antennas). This allows the radar device 10a to transmit signals as positive circularly polarized waves and negative circularly polarized waves in beam transmission during code multiplexing, in addition to signals received by different linearly polarized antennas (for example, horizontally polarized and vertically polarized antennas), and thereby obtain positive circularly polarized and negative circularly polarized received signals.

[0372] In this case, the radar device 10a is configured with different linearly polarized antennas (for example, horizontally polarized and vertically polarized antennas), beam transmitting antennas with forward and reverse circular polarization, one transmitting antenna for each polarization, and when multiple Na receiving antennas 202 are provided, a SIMO (Single in Multiple Output) configuration is achieved in which 1 × Na virtual receiving antennas are obtained for each polarization.

[0373] An example of the operation of each component of the radar device 10a shown in FIG. 19 will be described below.

[0374] In the radar device 10a shown in FIG. 19, the radar transmitter 100a operates in the following manner: DM =1, the phase rotation amount φ1 that gives the first Doppler shift amount DOP1 = 0, N DOP_CODE (1)=N CM= 2 and the number of transmitting antennas Nt = 2, the operation is the same as that of the first embodiment, and therefore a description of the operation will be omitted.

[0375] Next, an example of the operation of the radar receiver 200a of the radar device 10a shown in FIG. 19 will be described.

[0376] The operations of the components from the receiving antenna 202 to the antenna system processing unit 201 are the same as those in the first embodiment, and therefore a description of the operations will be omitted.

[0377] Since the CFAR unit 211 does not perform Doppler multiplexing (for example, when the Doppler multiplexing number N DM =1), the operation is performed without using Doppler domain compression CFAR processing.

[0378] The code demultiplexer 215 receives the distance index f b_cfar , the Doppler frequency index f s_cfar , and the output of the Doppler analysis unit 210. The code demultiplexing unit 215 outputs the demultiplexed received signal of the code-multiplexed signal to the direction estimation unit 214.

[0379] For example, the code demultiplexing unit 215 performs demultiplexing and reception of the code-multiplexed signal by multiplying the complex conjugate of the code used for multiplex transmission as shown in the following equation (67): z (f b_cfar ,f s_cfar , ncm) is output to the direction estimation unit 214.

number

[0380] where Y z (f b_cfar ,f s_cfar , ncm) is the distance index f of the Doppler analysis unit 210 in the z-th antenna system processing unit 201. b_cfar and the Doppler frequency index fs_cfar The output of VFTALL z (f b_cfar ,f s_cfar ) is transmitted using an orthogonal code Code ncm This is the output of the code-multiplexed signal separated using z=1,~,Na and ncm=1,~,N CM is.

[0381] Here, the Doppler phase correction vector α(f s_cfar ) is expressed as in the following equation (68). The Doppler phase correction vector α(f s_cfar ) is, for example, the output VFT of the first Doppler analysis unit 210. z 1 (f b_cfar , f s_comp_cfar ) as a reference for the Doppler analysis time, the output VFT of the second Doppler analysis unit 210 z 2 (f b_cfar , f s_comp_cfar ) the Doppler frequency index f caused by the time delay of Tr s_cfar is a vector whose elements are Doppler phase correction coefficients that correct the phase rotation in the Doppler component of

number

[0382] In addition, VFTALL shown in equation (67) z (f b_cfar , f s_cfar ) is, for example, the output VFT of the two Doppler analyzers 210 in the z-th antenna system processor 201, as shown in the following equation (69): z noc (f b , f s ), the distance index f extracted by the CFAR unit 211 b_cfar and the Doppler frequency index f s_cfar The components extracted based on the above are expressed in vector format, where noc=1,2.

number

[0383] In FIG. 19, the direction estimation unit 214 receives a distance index f b_cfar , the Doppler frequency index f s_cfar Demultiplexed received signal Y z (f b_cfar ,f s_cfar , ncm), target direction estimation processing (hereinafter referred to as direction estimation processing for linearly polarized waves) is performed.

[0384] Furthermore, the direction estimation unit 214 performs target direction estimation processing (hereinafter referred to as direction estimation processing for positive circularly polarized waves) based on the output from the first Doppler analysis unit 210 (Doppler analysis unit 210-1 in FIG. 19).

[0385] Furthermore, the direction estimation unit 214 performs target direction estimation processing (hereinafter referred to as direction estimation processing for counter-circularly polarized waves) based on the output from the second Doppler analysis unit 210 (Doppler analysis unit 210-2 in FIG. 19).

[0386] The direction estimation process in the direction estimation unit 214 may include, for example, a direction estimation process for linearly polarized waves, a direction estimation process for normal circularly polarized waves, and a direction estimation process for counter-circularly polarized waves. Each of these direction estimation processes will be described below.

[0387] <Direction estimation processing for linearly polarized waves> For example, the direction estimator 214 calculates a virtual receiving array correlation vector h(f b_cfar , f s_cfar ) is generated and direction estimation processing is performed.

[0388] Virtual receiving array correlation vector h(f b_cfar , f s_cfar ) includes Nt×Na elements, which is the product of the number of transmitting antennas Nt and the number of receiving antennas Na.

number

[0389] The direction estimation unit 214 calculates the virtual receiving array correlation vector h(f b_cfar , f s_cfar ) is used to perform direction estimation processing based on the phase difference between the receiving antennas 202 for the reflected wave signal from the target.

[0390] Here, the virtual receiving array correlation vector h(f b_cfar , f s_cfar ) includes reflected wave reception signals of signals transmitted from different linearly polarized antennas (for example, vertically polarized antennas and horizontally polarized antennas). Therefore, the direction estimation unit 214 calculates the virtual receiving array correlation vector h(f b_cfar , f s_cfar ), elements that are combinations of the polarization of a given transmitting antenna 109 and the polarization of a receiving antenna 202 may be extracted, and direction estimation processing may be performed using a virtual receiving array correlation vector consisting of the extracted elements. This allows direction estimation results to be obtained for each polarization of a given transmitting antenna 109 and each polarization of a receiving antenna 202.

[0391] <Direction estimation process for positive circularly polarized waves> Based on the output of the first Doppler analysis unit 210, the direction estimation unit 214 calculates a positive circularly polarized virtual receiving array correlation vector h c1 (f b_cfar , f s_cfar ) is generated and the direction of the target is estimated relative to the normal circular polarization.

[0392] Here, the correlation vector h c1 (f b_cfar , f s_cfar ) is the output of the first Doppler analysis unit 210 (for example, VFT z 1 (f b_cfar , f s_cfar), and includes the reflected wave reception signal of the signal transmitted as a circularly polarized wave by adjacent transmitting antennas 109 with different linear polarizations. For example, when there is one beam transmitting antenna, the positive circularly polarized virtual receiving array correlation vector h c1 (f b_cfar , f s_cfar ) includes 1×Na elements. The direction estimation unit 214 calculates the positive circular polarization virtual receiving array correlation vector h c1 (f b_cfar , f s_cfar ) is used to perform processing to estimate the direction of the reflected wave signal from the target based on the phase difference between the receiving antennas 202.

number

[0393] <Direction estimation process for counter-circularly polarized waves> The direction estimation unit 214 performs a process of estimating the direction of the counter-circularly polarized wave of the target based on the output of the second Doppler analysis unit 210. The direction estimation unit 214 estimates the direction of the counter-circularly polarized wave virtual receiving array correlation vector h c2 (f b_cfar , f s_cfar ) and performs direction estimation processing for the counter-circularly polarized wave of the target.

[0394] Here, the counter-circularly polarized virtual receiving array correlation vector h c2 (f b_cfar , f s_cfar ) is the output of the second Doppler analyzer 210 (for example, VFT z 2 (f b_cfar , f s_ cfar ), and includes the reflected wave reception signal of the signal transmitted as a circularly polarized wave by adjacent transmitting antennas 109 with different linear polarizations. For example, when there is one beam transmitting antenna, the counter-rotation virtual receiving array correlation vector h c2 (f b_cfar , f s_cfar ) contains 1 × Na elements.

[0395] The direction estimation unit 214 calculates the counter-circularly polarized virtual receiving array correlation vector h c2 (f b_cfar , f s_cfar ) is used to estimate the direction of the reflected wave signal from the target based on the phase difference between the receiving antennas 202.

number

[0396] Hereinafter, the virtual receiving array correlation vector h(f b_cfar , f s_cfar ) for linearly polarized waves, and the correlation vector h for the positive circularly polarized wave virtual receiving array. c1 (f b_cfar , f s_cfar ) and the direction estimation process for the normal circular polarization using the anti-circular polarization virtual receiving array correlation vector h c2 (f b_cfar , f s_cfar The operation of the direction estimation process for the counter-circularly polarized wave using the azimuth angle θ is the same as that in the first embodiment, and therefore the description of the operation will be omitted.

[0397] The direction estimation unit 214 may perform direction estimation processing using a received signal containing a combination of different types of polarization. In this case, a direction estimation result that is less dependent on polarization can be obtained. Furthermore, by performing direction estimation processing using more virtual antennas, the reception SNR can be improved, and the detection performance of the radar device 10a can be improved. Furthermore, since direction estimation processing is performed using the maximum aperture length of the available virtual receiving antennas, the angular resolution can also be improved. For example, when direction estimation processing is performed using all virtual receiving antennas, Nt=2,N BF = 1, so maximum (Nt × Na + 2 × N BF ×Na)=4Na virtual receiving antennas can be used, which is more than 2Na (=Nt ×Na).

[0398] The direction estimation unit 214 may also perform direction estimation processing using the same type of polarized antenna for both transmission and reception as described above, and direction estimation processing combining different types of polarization for transmission and reception, and use both of these direction estimation results as the direction estimation processing result. This provides a direction estimation processing result that is highly dependent on polarization and a direction estimation processing result that is less dependent on polarization. The results of such direction estimation processing may be input to a target identification processing unit (not shown) to perform target identification processing.

[0399] Here, in order to configure one transmission SIMO for each of four types of polarization (different linearly polarized antennas (e.g., vertically polarized antennas and horizontally polarized antennas) and forward / reverse circular polarization), four transmission antennas are used in the existing method. In contrast, in the configuration of FIG. 19, two transmission antennas 109 are used to configure one transmission SIMO for each of the four polarizations, which has the effect of reducing the number of transmission antennas. Furthermore, the radar device 10a performs code multiplexing transmission, which reduces the transmission time. For example, compared to switching between four transmission antennas 109 in a time-division manner, the transmission time can be halved.

[0400] Furthermore, compared to the first embodiment, the number of transmitting antennas is smaller, but the same effects as those of the first embodiment can be obtained. Specific examples of antenna arrangements are shown below.

[0401] [Antenna placement example 5] FIG. 20 shows an example of antenna arrangement when the polarization of the receiving antenna 202 is one type (for example, V). As shown in FIG. 20(a), the arrangement of transmitting antennas Tx#1 to Tx#2 (for example, Nt transmitting antennas 109) and receiving antennas Rx#1 to Rx#3 (for example, Na receiving antennas) constitutes arrangements VA#1 to VA#6 of virtual receiving antennas (or MIMO virtual antennas) as shown in FIG. 20(b). Also, in FIG. 20(a), the positive circularly polarized beam transmitting antenna Tx#3 (for example, N BFThe arrangements of the transmitting antennas CA#1 to CA#3 and the receiving antennas Rx#1 to Rx#3 form positive circular polarized virtual receiving antenna arrangements CA#1 to CA#3 as shown in (c) of Fig. 20. The positive circular polarized virtual receiving antenna arrangements CA#1 to CA#3 may also be represented as VA#7 to VA#9, for example.

[0402] In addition, in FIG. 20(a), the reversed circularly polarized beam transmitting antenna Tx#3 (for example, N BF The arrangements of the transmitting antennas (Rx#1 to Rx#3) and the receiving antennas Rx#1 to Rx#3 form the arrangements RA#1 to RA#3 of the counter-circularly polarized virtual receiving antennas as shown in (d) of Figure 20. The arrangements RA#1 to RA#3 of the counter-circularly polarized virtual receiving antennas are also represented as VA#7 to VA#9. The arrangements RA#1 to RA#3 of the counter-circularly polarized virtual receiving antennas are the same as the arrangements CA#1 to CA#3 of the forward-circularly polarized virtual receiving antennas, respectively.

[0403] In addition, the combination of transmitting antenna polarization and receiving antenna polarization for each virtual receiving antenna is different for each antenna, so it is written as (transmitting antenna polarization / receiving antenna polarization). For example, a virtual receiving antenna that transmits horizontally polarized waves and receives vertically polarized waves is written as "H / V."

[0404] 20(a) has been described as an example in which the receiving antenna 202 includes a vertically polarized antenna, but the type of polarized antenna included in the receiving antenna 202 is not limited to this, and other types of polarized antennas may be applied. For example, a horizontally polarized antenna or a circularly polarized antenna may be applied. Alternatively, multiple types of polarized antennas may be applied to any of Rx#1 to Rx#3.

[0405] Here, the arrangement shown in FIG. 20(a) is such that the antenna spacing D between the transmitting antennas Tx#1 and Tx#2 of different linear polarizations that constitute the beam transmitting antenna Tx#3 is H On the other hand, the receiving antennas Rx#1, Rx#3, and Rx#2 (adjacent receiving antennas 202) are spaced apart by an antenna spacing of 2D in the X-axis direction (for example, horizontal direction). HSome of the receiving antennas Rx#1, Rx#3, and Rx#2 are arranged at an antenna interval D in the Y-axis direction (for example, vertically) relative to the other antennas. v By arranging them in this way, the following effects can be obtained.

[0406] For example, as shown in FIG. 20(a), the arrangement of the receiving antenna 202 is as follows: receiving antennas Rx#1 and Rx#2 are arranged in a line in the X-axis direction (for example, the horizontal direction), and receiving antennas Rx#2 and Rx#3 are arranged in the orthogonal Y-axis direction (for example, the vertical direction) at an interval D V Since the Rx#3 is included and arranged offset by , two-dimensional direction estimation in the X-axis direction and the Y-axis direction (for example, the vertical direction and the horizontal direction) is possible. For example, V By setting the spacing between the beams at half wavelength intervals, it is possible to suppress the occurrence of grating lobes in angle measurement processing in the vertical direction within a range of ±90°.

[0407] In addition, D V The intervals are not limited to half wavelength intervals, but may be set at intervals shorter than the wavelength (λ) of the radar transmission signal. For example, D V and may be set to approximately 0.45λ to 0.8λ (for example, any value in the range of 0.45λ to 0.8λ). Here, λ represents the wavelength of the carrier frequency of the radar transmission signal. For example, if a chirp signal is used as the radar transmission signal, λ is the wavelength of the center frequency in the frequency sweep band of the chirp signal.

[0408] As shown in FIG. 20(a), the transmitting antennas Tx#1 and Tx#2 are arranged linearly in the X-axis direction with an antenna spacing D H As shown in the arrangement of the virtual receiving antennas in FIG. 20(b), the antenna spacing between VA#1 and VA#4, VA#2 and VA#5, and VA#3 and VA#6 is D H This becomes:

[0409] As shown in FIG. 20(a), the receiving antennas Rx#1, Rx#3, and Rx#2 are arranged in the X-axis direction (for example, the horizontal direction) with an antenna spacing of 2D. HTherefore, the distance between the transmitting antennas Tx#1 and Tx#2 (for example, D H ), the difference in the spacing between the receiving antennas 202 is the spacing D H The virtual receiving antenna arrangement has a spacing D H For example, as shown in the arrangement of virtual receiving antennas in FIG. 20(b), VA#1, VA#4, VA#3, VA#6, VA#2, and VA#5 are spaced apart by D H (However, VA#3 and VA#6 are spaced apart by D in the Y-axis direction (for example, vertical direction). V offset placement).

[0410] For example, D H By setting the antenna spacing to half wavelength, it is possible to suppress the occurrence of grating lobes in angle measurement processing in the horizontal range of ±90°. H Since the antennas are arranged in this manner, it is possible to suppress side lobes in the angle measurement process, thereby improving the detection performance of multiple targets.

[0411] In addition, D H The intervals are not limited to half wavelength intervals, but may be set at intervals shorter than the wavelength (λ) of the radar transmission signal. For example, D H and may be set to approximately 0.45λ to 0.8λ (for example, any value in the range of 0.45λ to 0.8λ). Here, λ represents the wavelength of the carrier frequency of the radar transmission signal. For example, if a chirp signal is used as the radar transmission signal, λ is the wavelength of the center frequency in the frequency sweep band of the chirp signal. The same applies to the following arrangement examples.

[0412] Furthermore, in the direction estimation process for linearly polarized waves performed by the direction estimation unit 214, when the radar device 10a extracts elements that are combinations of the polarization of a predetermined transmitting antenna 109 and the polarization of a receiving antenna 202 and performs direction estimation process using a virtual receiving array correlation vector consisting of the extracted elements, for example, when the cross-polarized wave combination is a combination of a horizontally polarized transmitting antenna and a vertically polarized antenna, the direction estimation process is performed using virtual receiving antennas VA#1, VA#2, and VA3, and a direction estimation result that depends on the transmission and reception characteristics between the cross-polarized antennas is obtained.

[0413] Furthermore, for example, when a combination of a vertically polarized transmitting antenna and a vertically polarized antenna is used as a combination of homogeneous polarizations, the radar device 10a performs direction estimation processing using virtual receiving antennas VA#4, VA#5, and VA#6, and can obtain direction estimation results that depend on the transmission and reception characteristics of homogeneous polarizations. By using the direction estimation results of the former combination of transmitting antenna and receiving antenna and the latter combination, it is possible to improve detection and identification performance.

[0414] In addition, when the radar device 10a extracts elements that are combinations of the polarization of the predetermined transmitting antenna 109 and the polarization of the receiving antenna 202 and performs direction estimation processing using a virtual receiving array correlation vector consisting of the extracted elements, the antenna interval D H is half the wavelength, the element spacing in the X-axis direction (for example, horizontal direction) is the antenna spacing 2D H This results in one wavelength, so grating lobes occur in angle measurement processing in the horizontal range of ±90°.

[0415] Therefore, if the detection angle range assumed by the radar device 10a is wider than the angle at which grating lobes occur, the radar device may be more likely to erroneously detect false peaks caused by grating lobes as targets within the detection angle range, which may result in a deterioration in the detection performance of the radar device 10a.

[0416] However, the direction estimation unit 214 can suppress grating lobes by performing direction estimation processing using received signals containing a combination of different types of polarization. In this case, the direction estimation unit 214 performs direction estimation processing using, for example, virtual receiving antennas VA#1 to VA#6 as received signals containing a combination of different types of polarization. As a result, the element spacing in the X-axis direction (for example, the horizontal direction) is reduced to the antenna spacing D H This arrangement allows for the suppression of grating lobes in angle measurement processing within a horizontal range of ±90°.

[0417] Furthermore, the direction estimation unit 214 performs direction estimation processing using the maximum aperture length of the available virtual receiving antennas, thereby improving angular resolution. Furthermore, the direction estimation unit 214 performs direction estimation processing using all available virtual receiving antennas, thereby improving reception SNR.

[0418] Furthermore, the direction estimation unit 214 performs direction estimation processing for forward circularly polarized waves using CA#1, CA#2, and CA#3. Similarly, the direction estimation unit 214 performs direction estimation processing for reverse circularly polarized waves using RA#1, RA#2, and RA#3. This uses a combination of a forward or reverse circularly polarized wave transmitting antenna and a vertically polarized antenna, making it possible to obtain direction estimation results that depend on the transmission and reception characteristics of the polarization, thereby improving detection and identification performance.

[0419] In the direction estimation process using CA#1, CA#2, and CA#3 or the direction estimation process using RA#1, RA#2, and RA#3, the antenna interval D H is half the wavelength, the element spacing in the X-axis direction (for example, horizontal direction) is the antenna spacing 2D H As a result, a grating lobe occurs in the angle measurement process in the horizontal ±90° range. However, the direction estimation unit 214 can suppress the grating lobe by performing the direction estimation process using a received signal that includes a combination of different types of polarization.

[0420] In this case, the direction estimation unit 214 performs direction estimation processing using, for example, virtual receiving antennas VA#1 to VA#6, CA#1, CA#2, CA#3, RA#1, RA#2, and RA#3 as a received signal containing a combination of different types of polarization. As a result, the element spacing in the X-axis direction (for example, the horizontal direction) is equal to the antenna spacing D H This arrangement includes the antennas 211 and 212, making it possible to suppress grating lobes in angle measurement processing in the horizontal ±90° range. Furthermore, the direction estimation unit 214 performs direction estimation processing using the maximum aperture length of the available virtual receiving antennas, thereby improving angular resolution. Furthermore, since direction estimation processing is performed using all available virtual receiving antennas, the reception SNR is also improved.

[0421] (Modification 2 of Embodiment 1) The configuration of Figure 19 in variant 1 of embodiment 1 shows a configuration in which the number of transmitting antennas Nt = 2, and one transmitting SIMO for each of four types of polarization (different linearly polarized antennas (e.g., vertically polarized antenna and horizontally polarized antenna) and forward / reverse circular polarization), but it is also possible to combine time division multiplexing and switch the code-multiplexed transmitting antenna 109 in a time division manner to form a MIMO configuration for each of the four types of polarization.

[0422] 21 shows a configuration example of a radar device 10b according to Modification 2 of Embodiment 1. The radar device 10b has, for example, four transmit antennas Nt=4, a configuration in which pairs of two transmit antennas 109 that are code-multiplexed are switched in a time-division manner, and two-transmission MIMO is performed for each of four types of polarized waves.

[0423] In the radar device 10b shown in FIG. 21, a transmission switching control unit 110 and a transmission switching unit 111 are added to the configuration of the radar device 10a shown in FIG.

[0424] The following mainly describes the operation of the radar device 10b shown in FIG. 21 that differs from the configuration of the radar device 10a shown in FIG.

[0425] The operations of the radar transmission signal generator 101, the phase rotation amount setting unit 105, and the phase rotator 108 in the radar transmitter 100b of the radar device 10b shown in FIG. 21 are performed in accordance with the Doppler multiplexing number N DM =1, the phase rotation amount φ1 that gives the first Doppler shift amount DOP1 = 0, N DOP_CODE (1)=N CM = 2 and the number of transmitting antennas Nt = 2, the operation is the same as that of the first embodiment, and therefore a description of the operation will be omitted.

[0426] The transmission switching control unit 110 controls switching between the two transmission antennas 109 output from the transmission switching unit 111 every predetermined transmission period. For example, as shown in Fig. 22, the transmission switching control unit 110 controls switching between the two transmission antennas 109 output from the transmission switching unit 111 every two transmission periods, which is the transmission period for a code length Loc=2 assigned by the encoding unit 107. Note that Fig. 22 shows an example in which the codes Code1=

[0011] and Code2=[j -j] are assigned by the encoding unit 107.

[0427] Here, the two transmitting antennas 109 that form the switching unit are a pair of adjacent transmitting antennas 109 that generate different linearly polarized waves (for example, horizontally polarized wave and vertically polarized wave). The transmission switching control unit 110 performs an operation of switching these pairs every two transmission cycles.

[0428] Hereinafter, the number of pairs of adjacently arranged transmitting antennas 109 that transmit different linearly polarized waves will be denoted as "Nsw." Furthermore, among the Nt transmitting antennas 109, the pair of transmitting antennas 109 that transmits first will be referred to as the "first transmitting pair," and the pair of transmitting antennas 109 that is switched in the following two transmission cycles will be referred to as the "second transmitting pair." Hereinafter, when Nt / 2 transmission pairs are included in the Nt transmitting antennas 109, each transmission pair will be represented by a transmission pair index "nsw," where nsw = 1, .., Nt / 2. Furthermore, the transmission switching control unit 110 outputs the transmission pair index nsw to the output switching unit 209 for each transmission cycle.

[0429] 21 and 22, the transmitting antenna 109 of the nsw-th transmitting pair that radiates the output of the phase rotation unit PROT#[1,1] into space is represented as "transmitting antenna Tx#[1,nsw]". The transmitting antenna 109 that radiates the output of the phase rotation unit PROT#[ndop_code(1),1]=PROT#[2,1] into space is represented as "transmitting antenna Tx#[ndop_code(ndm),nsw]=Tx#[2,nsw]". In FIG. 21 and FIG. 22, Nt=4, and Tx#[1,1], Tx#[ndop_code(1),1] (=Tx#[2,1]), Tx#[1,2], and Tx#[ndop_code(1),2] (=Tx#[2,2]) are assigned to the four transmitting antennas 109. Also, ndop_code(1)=2.

[0430] The transmission switching unit 111 performs an operation of pairing adjacently arranged transmission antennas that emit different linearly polarized waves every two transmission cycles, based on the control of the above-mentioned transmission switching control unit 110. For example, as shown in Fig. 22, the radar device 10b transmits radar transmission signals from each of a plurality of pairs of transmission antennas 109 in a time-division multiplexed manner.

[0431] Next, an example of the operation of the radar receiver 200b of the radar device 10b shown in FIG. 21 will be described.

[0432] The operations of the components from the receiving antenna 202 to the antenna system processing unit 201 are the same as those in the first embodiment, and therefore a description of the operations will be omitted.

[0433] The output switching unit 209 selectively switches and outputs the output of the beat frequency analysis unit 208 for each transmission period to the (nsw-1)×Loc+OC_INDEX-th Doppler analysis unit 210 out of the Loc×Nsw (in FIG. 21, Loc=2) Doppler analysis units 210 based on the orthogonal code element index OC_INDEX input from the encoding unit 107 of the phase rotation amount setting unit 105 and the transmission pair index nsw input from the transmission switching control unit 110. For example, in the m-th transmission period Tr, the output switching unit 209 selects the Doppler analysis unit 210 obtained by the following equation (73), where Loc=2.

number

[0434] The signal processing unit 206 has Loc×Nsw Doppler analysis units 210-1 to 210-Loc×Nsw. For example, data is input to the ncsw-th Doppler analysis unit 210 by the output switching unit 209 for every Loc×Nsw transmission periods (Loc×Nsw×Tr). Here, ncsw=1, to, Loc×Nsw. Therefore, the ncsw-th Doppler analysis unit 210 analyzes data (for example, beat frequency response RFT input from the beat frequency analysis unit 208) for Ncsub=Nc / (Loc×Nsw) transmission periods out of the Nc transmission periods. z (f b , m)) to obtain the distance index f b Doppler analysis is performed every 10 seconds. Note that Nc is set to an integer multiple of (Loc × Nsw).

[0435] For example, if Ncsub is a power of 2, FFT processing can be applied to Doppler analysis. In this case, the FFT size is Ncsub, and the maximum Doppler frequency at which aliasing does not occur, as derived from the sampling theorem, is ±1 / (2 Loc × Nsw × Tr). Also, the Doppler frequency index f s The Doppler frequency interval is 1 / (Loc×Nsw×Ncsub×Tr), and the Doppler frequency index f s The range of f s= -Ncsub / 2, ~, 0, ~, Ncsub / 2-1. ncsw=1,~, (Loc×Nsw)

[0436] For example, the output VFT of the Doppler analysis unit 210 of the z-th signal processing unit 206 z ncsw (f b , f s ) is expressed by replacing Ncode with Ncsub, Loc with (Loc×Nsw), and noc with ncsw in equation (39), and the other operations are the same as those in the first embodiment.

[0437] In FIG. 21, the CFAR unit 211 performs CFAR processing (for example, adaptive threshold determination) using the outputs of the (Loc×Nsw) Doppler analyzers 210 of the first to Na-th signal processors 206, and calculates the distance index f b_cfar and the Doppler frequency index f s_cfar Extract.

[0438] The CFAR unit 211 calculates the output VFT of the first to Nsw×Loc-th Doppler analyzers 210 in the first to Na-th signal processors 206 as shown in the following equation (74), for example. z 1 (f b , f s )~VFT z Nsw×Loc (f b , f s ) is added to the power b , f s ) is used to perform two-dimensional CFAR processing consisting of a distance axis and a Doppler frequency axis (corresponding to relative velocity), or CFAR processing that combines one-dimensional CFAR processing.

[0439] The CFAR unit 211 adaptively sets a threshold value and calculates a distance index f that has a received power greater than the threshold value. b_cfar , the Doppler frequency index f s_cfar , and received power information PowerFT(f b_cfar , f s_cfar) to the code demultiplexing unit 215. Note that the CFAR unit 211 does not perform Doppler multiplexing (Doppler multiplexing number N DM =1), the operation is performed without using Doppler domain compression CFAR processing.

number

[0440] The code demultiplexer 215 receives the distance index f b_cfar , the Doppler frequency index f s_cfar , and the output of the Doppler analysis unit 210. The code demultiplexing unit 215 outputs the demultiplexed received signal of the code-multiplexed signal to the direction estimation unit 214.

[0441] For example, the code demultiplexing unit 215 performs demultiplexing and reception of the code-multiplexed signal by multiplying the complex conjugate of the code used for multiplexing, as shown in the following equation (75). z (f b_cfar ,f s_cfar , ncm, nsw) are output to the direction estimation unit 214.

number

[0442] where Y z (f b_cfar ,f s_cfar , ncm, nsw) is the distance index f of the Doppler analyzer 210 in the z-th antenna system processor 201 when transmitted from the transmitting antenna 109 of the nsw-th transmitting pair. b_cfar and the Doppler frequency index f s_cfar The output of VFTALL z (f b_cfar ,f s_cfar , nsw) are transmitted using an orthogonal code Code ncm The output is a demultiplexed code-multiplexed signal using z=1,~,Na and ncm=1,~,NCM ,nsw=1,~,N SW is.

[0443] Here, the Doppler phase correction vector α(f s_cfar ) is expressed as in the following equation (76). The Doppler phase correction vector α(f s_cfar ) is, for example, the output VFT of the first Doppler analysis unit 210. z 1 (f b_cfar , f s_cfar ) as a reference for the Doppler analysis time, the output VFT of the ncsw-th Doppler analysis unit 210 z ncsw (f b_cfar , f s_cfar ) transmission time delay (ncsw-1)×Tr / (Loc×N SW ) resulting in the Doppler frequency index f s_cfar is a vector whose elements are Doppler phase correction coefficients that correct the phase rotation in the Doppler component of

number

[0444] Also, in equation (75), VFTALL z (f b_cfar , f s_cfar , nsw) is, for example, the output VFT of the two Doppler analyzers 210 in the z-th antenna system processor 201, as shown in the following equation (77): z ncsw (f b , f s ), the distance index f extracted by the CFAR unit 211 b_cfar and the Doppler frequency index f s_cfar The components extracted based on the above are expressed in vector format, where noc=1,2.

number

[0445] In FIG. 21, the direction estimation unit 214 receives a distance index f b_cfar , the Doppler frequency index f s_cfar Demultiplexed received signal Y z (f b_cfar ,f s_cfar , ncm, nsw), target direction estimation processing (hereinafter referred to as direction estimation processing for linearly polarized waves) is performed.

[0446] In addition, the direction estimation unit 214 performs target direction estimation processing (hereinafter referred to as direction estimation processing for positive circular polarization) using the output of the first to second Nsw Doppler analysis units 210 (in Figure 21, Doppler analysis units 210-1 to 2Nsw) that becomes the received signal of the transmission that becomes positive circular polarization.

[0447] In addition, the direction estimation unit 214 performs target direction estimation processing (hereinafter referred to as direction estimation processing for the counter-circularly polarized wave) using the output of the first to second Nsw Doppler analysis units 210 (in FIG. 21, Doppler analysis units 210-1 to 2Nsw) that becomes the received signal of the transmitted counter-circularly polarized wave.

[0448] The direction estimation process in the direction estimation unit 214 may include a direction estimation process for linearly polarized waves, a direction estimation process for normal circularly polarized waves, and a direction estimation process for counter-circularly polarized waves. The operation of each direction estimation process will be described below.

[0449] <Direction estimation processing for linearly polarized waves> For example, the direction estimator 214 calculates a virtual receiving array correlation vector h(f b_cfar , f s_cfar ) is generated and direction estimation processing is performed.

[0450] Virtual receiving array correlation vector h(f b_cfar , f s_cfar ) includes Nt×Na elements, which is the product of the number of transmitting antennas Nt and the number of receiving antennas Na.

number

[0451] where hsw nsw (f b_cfar , f s_cfar ) is a reception vector of the reception signal that is code-separated when transmitted from the transmitting antenna 109 of the nsw-th transmission pair, and includes 2Na elements, which is the product of the number of transmitting antennas 2 and the number of receiving antennas Na, as shown in the following equation (79).

number

[0452] The direction estimation unit 214 calculates the virtual receiving array correlation vector h(f b_cfar , f s_cfar ) is used to perform direction estimation processing based on the phase difference between the receiving antennas 202 for the reflected wave signal from the target.

[0453] Here, the virtual receiving array correlation vector h(f b_cfar , f s_cfar ) includes reflected wave reception signals of signals transmitted from different linearly polarized antennas (for example, vertically polarized antennas and horizontally polarized antennas). Therefore, the direction estimation unit 214 calculates the virtual receiving array correlation vector h(f b_cfar , f s_cfar ), elements that are combinations of the polarization of a given transmitting antenna 109 and the polarization of a receiving antenna 202 may be extracted, and direction estimation processing may be performed using a virtual receiving array correlation vector consisting of the extracted elements. This allows direction estimation results to be obtained for each polarization of a given transmitting antenna 109 and each polarization of a receiving antenna 202.

[0454] <Direction estimation process for positive circularly polarized waves> The direction estimation unit 214 uses the output of the first to second Nsw Doppler analysis units 210 that is a reception signal of a transmission that is a positive circularly polarized wave to calculate a positive circularly polarized virtual reception array correlation vector h c1 (f b_cfar , f s_cfar) is generated and the direction of the target is estimated relative to the normal circular polarization.

[0455] Here, the correlation vector h c1 (f b_cfar , f s_cfar ) includes Nsw×Na elements. The direction estimation unit 214 calculates the positive circular polarization virtual receiving array correlation vector h c1 (f b_cfar , f s_cfar ) is used to perform direction estimation processing based on the phase difference between the receiving antennas 202 for the reflected wave signal from the target.

[0456] For example, as shown in Figure 22, when Tx#[1,1] is a horizontally polarized antenna, Tx#[2,1] is a vertically polarized antenna, Tx#[1,2] is a horizontally polarized antenna, and Tx#[2,2] is a vertically polarized antenna, and Code1=[1,1] and Code2=[j -j] are used in code multiplexing, the positive circularly polarized virtual receiving array correlation vector h c1 (f b_cfar , f s_cfar ) can be expressed as the following equation (80):

number

[0457] <Direction estimation process for counter-circularly polarized waves> The direction estimation unit 214 calculates the counter-circularly polarized virtual receiving array correlation vector h using the output of the first to second Nsw Doppler analysis units 210 that is the received signal of the transmitted counter-circularly polarized wave. c2 (f b_cfar , f s_cfar ) and performs direction estimation processing for the counter-circularly polarized wave of the target.

[0458] Here, the counter-circularly polarized virtual receiving array correlation vector h c2 (f b_cfar , f s_cfar ) includes Nsw×Na elements. The direction estimation unit 214 calculates the counter-circularly polarized virtual receiving array correlation vector h c2 (f b_cfar, f s_cfar ) is used to perform direction estimation processing based on the phase difference between the receiving antennas 202 for the reflected wave signal from the target.

[0459] For example, as shown in Figure 22, if Tx#[1,1] is a horizontally polarized antenna, Tx#[2,1] is a vertically polarized antenna, Tx#[1,2] is a horizontally polarized antenna, and Tx#[2,2] is a vertically polarized antenna, and Code1=[1,1] and Code2=[j -j] are used in code multiplexing, then the counter-circularly polarized virtual receiving array correlation vector h c2 (f b_cfar , f s_cfar ) can be expressed as the following equation (81).

number

[0460] Hereinafter, the virtual receiving array correlation vector h(f b_cfar , f s_cfar ) for linearly polarized waves, and the correlation vector h for the positive circularly polarized wave virtual receiving array. c1 (f b_cfar , f s_cfar ) and the direction estimation process for the normal circular polarization using the anti-circular polarization virtual receiving array correlation vector h c2 (f b_cfar , f s_cfar The operation of the direction estimation process for the counter-circularly polarized wave using the azimuth angle θ is the same as that in the first embodiment, and therefore the description of the operation will be omitted.

[0461] The direction estimation unit 214 may perform direction estimation processing using received signals containing a combination of different types of polarization. In this case, direction estimation results that are less dependent on polarization can be obtained. Furthermore, by performing direction estimation processing using more virtual antennas, the reception SNR can be improved, and the detection performance of the radar device 10b can be improved. Furthermore, since direction estimation processing is performed using the maximum aperture length of the available virtual receiving antennas, angular resolution can also be improved. For example, when direction estimation processing is performed using all virtual receiving antennas, a maximum of (Nt×Na+2×N BF×Na) virtual receive antennas can be used, which is greater than Nt ×Na virtual receive antennas.

[0462] The direction estimation unit 214 may also perform direction estimation processing using the same type of polarized antenna for both transmission and reception as described above, and direction estimation processing combining different types of polarization, and use both of these direction estimation results as the direction estimation processing result. This allows for direction estimation processing results that are highly dependent on polarization and direction estimation processing results that are less dependent on polarization. The results of such direction estimation processing may be input to a target identification processing unit (not shown) to perform target identification processing.

[0463] As described above, in the second modification of the first embodiment, adjacent pairs of different linearly polarized antennas (for example, horizontally (H) and vertically (V) polarized antennas) are code-multiplexed using orthogonal codes that are 90° out of phase with each other and orthogonal to each other (for example, orthogonal codes [0°, 90°] and [0°, −90°]). Furthermore, when there are multiple such pairs of transmitting antennas 109, the radar device 10b switches the transmission time for each pair and performs time-division multiplexing transmission.

[0464] This operation allows for the reception of reflected waves from transmissions using two types of circular polarization (e.g., right-handed and left-handed circular polarizations with opposite rotation directions) in addition to transmissions from different linearly polarized antennas, thereby improving target detection or identification performance.

[0465] Note that some of the transmitting antennas 109 may overlap between transmission pairs. For example, as shown in Fig. 23, the transmission switching control unit 110 controls switching between two transmitting antennas output from the transmission switching unit 111 every two transmission periods, which is the transmission period of the code length Loc=2 assigned by the encoding unit 107. Also, an example is shown in which the encoding unit 107 assigns codes Code1=

[0011] and Code2=[j -j].

[0466] Here, the transmission switching control unit 110 may hold information related to transmission switching control shown in FIG. 24 (for example, a "transmission switching control table"). The transmission switching control unit 110 may assign codes to the transmission antennas #1 to #4 and perform switching control based on the transmission switching control table. For example, for transmission pair index nsw=1, the transmission switching control unit 110 uses transmission antennas Tx#1 and Tx#2 and assigns codes Code1=

[0011] and Code2=[j -j], respectively. For transmission pair index nsw=2, the transmission switching control unit 110 uses transmission antennas Tx#2 and Tx#3 and assigns codes Code1=

[0011] and Code2=[j -j], respectively. Furthermore, for transmission pair index nsw=3, the transmission switching control unit 110 uses transmission antennas Tx#3 and Tx#4 and assigns codes Code1=

[0011] and Code2=[j -j], respectively. In this example, some antennas (Tx#2) overlap between transmission pairs with transmission pair indexes nsw=1 and 2. Also, some antennas (Tx#3) overlap between transmission pairs with transmission pair indexes nsw=3 and 4.

[0467] Based on such control by the transmission switching control unit 110, the encoding unit 107, the phase rotation unit 108, and the transmission switching unit 111 may be controlled in the same manner.

[0468] The radar receiver 200b also performs the same operations as those described with reference to FIG.

[0469] As described above, by overlapping some antennas between transmitting pairs, the direction estimator 214 calculates the positive circularly polarized virtual receiving array correlation vector h c1 (f b_cfar , f s_cfar ) and the counter-circularly polarized virtual receiving array correlation vector h c2 (f b_cfar , f s_cfar) can be increased, and the reception SNR of direction estimation processing for forward circularly polarized waves and direction estimation processing for reverse circularly polarized waves can be improved. Also, for example, by devising the antenna arrangement, it is possible to reduce grating lobes or side lobes during direction estimation. Also, by devising the antenna arrangement, it is possible to increase the aperture length and improve the angular resolution.

[0470] (Embodiment 2) In the first embodiment, an operation using MIMO multiplexing transmission that combines Doppler multiplexing and code multiplexing has been described. In the first embodiment, in a pair of adjacent antenna arrangements of different linearly polarized antennas (for example, a pair of a horizontally polarized antenna and a vertically polarized antenna), code-multiplexing transmission is performed using a common Doppler-multiplexed signal (for example, the same Doppler shift amount). Furthermore, the codes used for code multiplexing are orthogonal codes (for example, orthogonal codes

[0011] , [j -j]) in which the code elements differ in phase by 90° and are orthogonal to each other. Furthermore, when there are multiple pairs of adjacent antenna arrangements of different linearly polarized antennas, code-multiplexing transmission is performed using different Doppler-multiplexed signals (for example, different Doppler shift amounts) in those pairs.

[0471] These operations allow for transmission from different linearly polarized antennas as well as transmission using two types of circularly polarized waves (for example, right-handed and left-handed circularly polarized waves with mutually opposite rotation directions) to be received, enabling transmission using more types of polarized waves than the types of polarized waves of the transmitting antenna 109, and enabling transmission using many polarized waves with a small number of transmitting antennas. Furthermore, by combining the polarized waves of the transmitting antenna 109 with the polarized waves of the receiving antenna 202, more combinations of transmitting and receiving polarized waves can be obtained. In addition, by performing direction estimation processing in the direction estimator 214 for each combination of polarized waves of the transmitting and receiving antennas, it is possible to improve target detection or identification performance, for example, when a reflected wave with a polarized wave characteristic of each target is obtained. In addition, by using MIMO multiplexing transmission, which combines Doppler multiplexing and code multiplexing, it is possible to obtain an effect of shortening the transmission time compared to transmission by sequentially switching antennas in a time-division manner.

[0472] The present invention is not limited to operations using MIMO multiplexing as described in the first embodiment, and similar effects can be obtained in operations using other MIMO multiplexing.

[0473] For example, in this embodiment, in MIMO multiplexing transmission using Doppler multiplexing, an operation is described in which a transmission process that uses multiple linearly polarized antennas to perform Doppler multiplexing (for example, a process that applies different Doppler shift rates to perform multiplexing) and a transmission process that uses adjacent linearly polarized antennas with different antenna arrangements to perform Doppler multiplexing of beam transmission are switched over in time.

[0474] Even with this type of operation, in addition to transmission from different linearly polarized antennas, a received signal of a reflected wave can be obtained by transmission using at least one type of circular polarization (for example, one of right-handed and left-handed circular polarizations that have opposite rotation directions). For example, transmission using more types of polarization than the types of polarization of the transmitting antennas is possible, and transmission using many polarized waves is possible with fewer transmitting antennas, thereby achieving the same effects as those shown in embodiment 1.

[0475] FIG. 25 shows an example of the configuration of a radar device 10c according to this embodiment.

[0476] In comparison with the configuration of embodiment 1 (FIG. 1), in FIG. 25, a transmission switching control unit 112 is added that switches over time between transmission that performs Doppler multiplexing using a plurality of linearly polarized antennas and transmission that performs Doppler multiplexing of beam transmission using adjacent linearly polarized antennas with different antenna arrangements. Also, in FIG. 25, instead of encoding unit 107, a transmission weight generation unit 113 and a transmission weight multiplication unit 114 are provided that perform multiplication of adjacent linearly polarized antennas with different antenna arrangements by weight coefficients that result in circular polarization, or set and multiply zero amplitude weights (operation equivalent to transmission off control) to turn off transmission for antennas that are not transmitting.

[0477] The following mainly describes the parts that perform operations different from those in the first embodiment.

[0478] [Configuration of radar transmitter 100c] The operation of the radar transmission signal generating unit 101 is the same as in the first embodiment, and therefore a description thereof will be omitted.

[0479] The transmission switching control unit 112 controls the transmission antennas Tx#1 to #Nt and the beam transmission antennas Num_BF=1 to N BF The transmission switching control unit 112 holds information on the amount of Doppler shift to be periodically assigned at the Doppler shift variable setting cycle Nsw (hereinafter referred to as a "Doppler multiplexing assignment table" as an example). The transmission switching control unit 112 controls each component of the phase rotation amount setting unit 105, for example, based on the Doppler multiplexing assignment table. Furthermore, the transmission switching control unit 112 outputs information on the Doppler shift variable setting cycle (for example, a transmission switching index nsw, which will be described later) to the output switching unit 209.

[0480] 26 shows an example of a Doppler multiplexing allocation table. The table shown in FIG. 26 is for Tx#1 to #4 (number of transmitting antennas Nt=4) in the antenna arrangement shown in FIG. 25, and for the number of beam transmitting antennas N BF = 2 and the Doppler shift variable setting period Nsw = 2.

[0481] In the example shown in FIG. 26, in a transmission cycle of nsw=1, two beam transmitting antennas Tx#5 (transmitting a beam with positive circular polarization using Tx#1 and Tx#2) and Tx#6 (transmitting a beam with positive circular polarization using Tx#3 and Tx#4) transmit a Doppler multiplexing number N DM (1)=2 gives the Doppler shift amounts (DOP1 and DOP2).

[0482] In the example shown in FIG. 26, in a transmission cycle of nsw=2, the Doppler multiplexing number N DM (2)=4 gives a Doppler shift.

[0483] Here, the transmission cycle when nsw=1 is a transmission cycle that satisfies mod(m-1, Nsw)+1=1. Also, the transmission cycle when nsw=2 is a transmission cycle that satisfies mod(m-1, Nsw)+1=2. Here, m is an index indicating the number of transmissions in the transmission cycle, and m=1, to, Nc. Hereinafter, nsw will be referred to as the "transmission switching index."

[0484] In the following, the number of Doppler multiplexes in the transmission period of nsw is N DM (nsw). Here, N DM (1)=2, N DM (2)=4, but is not limited to this and may be set as long as it satisfies the following conditions.

[0485] <Setting conditions for the number of Doppler multiplexes in the Doppler shift variable setting period Nsw> The sum of the Doppler multiplexing numbers in the Doppler shift variable setting period Nsw is calculated by the following equation (82) when the number of transmitting antennas Nt and the number of transmitting beams N BF The sum of (Nt+N BF ) or more.

number

[0486] In addition, the number of transmitting antennas Nt and the number of transmitting beams N BF The setting may be such that transmission is performed at least once from each of the transmitting antennas. In addition, the setting may be such that the transmitting antenna 109 and the transmitting antenna used for beam transmission do not overlap in each Doppler shift variable setting period.

[0487] In addition, the Doppler multiplexing number N in the transmission period of each nsw DM (nsw) is Nt≧N DM It may be set such that (nsw) ≥ 1, where nsw = 1, ~,Nsw.

[0488] For example, a Doppler shift variable setting cycle of 3 transmission cycles may be set as Nsw = 3. For example, the transmission switching index nsw = 1 may be set to transmit a forward circularly polarized wave by the beam transmitting antenna, nsw = 2 may be set to transmit a reverse circularly polarized wave by the beam transmitting antenna, and nsw = 3 may be set to transmit from each transmitting antenna 109 (horizontally polarized wave or vertically polarized wave).

[0489] 27 shows an example of a Doppler multiplexing allocation table. The table shown in FIG. 27 is for Tx#1 to #4 (number of transmitting antennas Nt=4) in the antenna arrangement shown in FIG. 25, and for the number of beam transmitting antennas N BF 27, in the transmission period of nsw=1, the Doppler multiplexing number N is transmitted from two beam transmitting antennas Tx#5 (transmitting a beam with positive circular polarization using Tx#1 and Tx#2) and Tx#6 (transmitting a beam with positive circular polarization using Tx#3 and Tx#4). DM (1)=2 gives the Doppler shift amounts (DOP1 and DOP2).

[0490] In the example shown in FIG. 27, in a transmission cycle of nsw=2, two beam transmitting antennas Tx#5 (transmitting a beam with counter-circular polarization using Tx#1 and Tx#2) and Tx#6 (transmitting a beam with counter-circular polarization using Tx#3 and Tx#4) transmit a Doppler multiplex number N DM In the example shown in FIG. 27, in a transmission cycle of nsw=3, the Doppler multiplexing number N DM (3)=4 gives a Doppler shift.

[0491] Furthermore, for example, in each Doppler shift variable setting period, the radar transmitter 100c may be configured to transmit a mixture of circularly polarized and linearly polarized waves, such as transmitting forward or reverse circularly polarized waves from a beam transmitting antenna and transmitting from a transmitting antenna (horizontally polarized or vertically polarized) different from the beam transmitting antenna. For example, when transmitting from at least one pair of adjacent transmitting antennas 109, the radar transmitter 100c may perform, in the same transmission period, a transmission process of multiplexing radar transmission signals by applying the same Doppler shift amount and transmission weights with a phase difference of 90°, and a transmission process of multiplexing radar transmission signals by applying different Doppler shift amounts.

[0492] 28 shows an example of a Doppler multiplexing allocation table. The table shown in FIG. 28 is for Tx#1 to #4 (number of transmitting antennas Nt=4) in the antenna arrangement shown in FIG. 25, and for the number of beam transmitting antennas N BF 28, in the transmission period of nsw=1, the Doppler multiplexing number N is set from one beam transmitting antenna Tx#5 (transmitting a beam with normal circular polarization using Doppler shift amount DOP1 from Tx#1 and Tx#2) and two transmitting antennas Tx#3 and Tx#4 (transmitting using Doppler shift amounts DOP2 and DOP3, respectively). DM (2)=3 gives a Doppler shift.

[0493] In the example shown in FIG. 28, in the transmission period of nsw=2, the other beam transmitting antenna Tx#6 (transmitting a beam with positive circular polarization using Doppler shift amount DOP3 from Tx#3 and Tx#4) and the two transmitting antennas Tx#1 and Tx#2 (transmitting using Doppler shift amounts DOP1 and DOP2, respectively) transmit a Doppler multiplexing number N DM (2)=3 gives a Doppler shift.

[0494] The transmission switching control unit 112 performs the following control on the Doppler shift setting unit 106 based on, for example, the above-mentioned Doppler multiplexing allocation table.

[0495] The Doppler shift setting unit 106 sets the Doppler multiplex number N in the transmission cycle of nsw under the control of the transmission switching control unit 112. DM Using (nsw), the Doppler shift amount DOP ndm The phase rotation amount φ for adding (nsw) ndm (nsw) is set and output to the transmission weight generating unit 113. Note that antennas that do not transmit in the transmission period of nsw (antennas with transmission turned off) may be included. Here, ndm=1, ∼, N DM (nsw), where nsw=1,~,Nsw.

[0496] The number of Doppler multiplexes N in the phase rotation amount setting unit 105 DM Doppler shift DOP using (nsw) ndm The phase rotation amount φ for adding (nsw) ndm The setting is the same as in the first embodiment, and the amount of Doppler shift may be set at equal intervals, or may be set at unequal intervals.

[0497] For example, the following equation (83) may be used as the amount of Doppler shift at equal intervals.

number

[0498] Furthermore, for example, the following equation (84) may be used as the amount of Doppler shift at irregular intervals.

number

[0499] where N int (nsw) takes an integer value of 0 or greater.

[0500] The transmission weight generation unit 113 generates transmission weights W consisting of amplitudes and phases for the transmission antennas Tx#1 to #Nt (the number of transmission antennas Nt) under the control of the transmission switching control unit 112. ntx(nsw) is set as follows, where ntx=1 to Nt.

[0501] (1) When the beam transmitting antenna is on, the transmission weight generating unit 113 sets a weight coefficient for the transmitting antenna 109 that constitutes the beam transmitting antenna that is on to be circularly polarized using a different linearly polarized antenna.

[0502] For example, when a transmission weight coefficient of Amp×exp[jη] is set for one transmitting antenna 109, the transmission weight coefficient for the other transmitting antenna 109 is Amp×exp[j(η+π / 2)] or Amp×exp[j(η-π / 2)]. For example, the phase difference of transmission weights for circular polarization using different linearly polarized antennas is +90° or -90°. By multiplying such transmission weights, a transmission signal that is a right-handed or left-handed circularly polarized wave is transmitted. Hereinafter, when one of the circular polarization directions is referred to as a normal rotation, the other circular polarization direction may be referred to as a reverse rotation. Here, ξ is an arbitrary phase.

[0503] (2) A transmitting antenna that is not included in the beam transmitting antennas that are to be turned on for transmission and that is to be turned on for transmission is given a predetermined amplitude value Amp that is not zero.

[0504] For the transmitting antennas that are not included in (1) or (2) and whose transmission is to be turned off, a weight of zero amplitude (an operation equivalent to transmission-off control) is set to turn off transmission.

[0505] For example, when the transmission switching control unit 112 performs control based on the Doppler multiplexing allocation table shown in FIG. 26, the transmission weight generation unit 113 may set the following transmission weights.

[0506] If nsw=1, W1(1)=Amp×exp[jη], W2(1)=Amp×exp[j(η+π / 2)], W3(1)=Amp×exp[jη], W4(1)= Amp×exp[j(η+π / 2)] or W1(1)=Amp×exp[jη], W2(1)=Amp×exp[j(η-π / 2)], W3(1)=Amp×exp[jη], W4(1)= Amp×exp[j(η-π / 2)] When nsw=2, W1(2)=W2(2)=W3(2)=W4(2)= Amp

[0507] When a beam transmitting antenna is on, the transmission weight generation unit 113 may set a transmission weight coefficient that results in circular polarization using a different linearly polarized antenna for the transmitting antennas 109 that constitute the beam transmitting antenna that is on for transmission. For example, when the transmission weight generation unit 113 sets a transmission weight of Amp×exp[jη] for one transmitting antenna 109, it may set a transmission weight of Amp×exp[j(η+ξ)] or Amp×exp[j(η-ξ)] for the other transmitting antenna 109.

[0508] Here, ξ may be in the range of π / 6 to 5π / 6 radians (=30° to 150°). For example, if the phase deviation between the transmitting antennas 109 is corrected in advance, a circularly polarized wave is generated by beam transmission, with the main beam direction of the transmitted beam varying depending on ξ. For example, if the spacing between transmitting antennas that transmit beams is λ / 2 and ξ=90°, a circularly polarized wave is generated with the main beam direction facing 0°.

[0509] Furthermore, for example, when ξ = 30°, the main beam direction is shifted by approximately -15° from the front direction, resulting in a circularly polarized wave. Furthermore, for example, when ξ = 150°, the main beam direction is shifted by approximately +15° from the front direction, resulting in a circularly polarized wave. Here, λ is the wavelength of the high-frequency signal output from the transmitting antenna (when a chirp signal is used, the wavelength at the center frequency of the chirp signal is used).

[0510] The transmission weight generation unit 113 generates a transmission weight (hereinafter referred to as a "Doppler multiplex transmission weight") WD that includes the amount of Doppler phase rotation in the set transmission weight. ntx and outputs it to the transmission weight multiplication unit 114.

[0511] The following equation (85) is the Doppler multiplex transmission weight WD for the ntx-th transmitting antenna in the m-th transmission period Tr. ntx where ntx=1,~,Nt.

number

[0512] Note that floor[x] is an operator that outputs the largest integer that does not exceed the real number x. j is the imaginary unit.

[0513] The Nt transmission weight multiplication units 114 multiply the chirp signal cp(t) input from the radar transmission signal generation unit 101 by the Doppler multiplex transmission weights WD for each of the Nt transmission antennas 109 for each transmission period Tr. ntx (m), where ntx=1, ..,Nt. The outputs from the Nt transmission weight multiplication units 114 are amplified to a specified transmission power, and then radiated into space from the Nt transmission antennas 109 of the transmitting array antenna unit.

[0514] For example, the ntx-th transmission weight multiplication unit 114 multiplies the m-th chirp signal cp(t) generated by the radar transmission signal generation unit 101 by a Doppler multiplex transmission weight WD ntx (m) and outputs the result to the ntx-th transmitting antenna 109. For example, the ntx-th transmission weight multiplier 114 multiplies the Nc chirp signals cp(t) generated by the radar transmission signal generator 101 for each transmission period by a Doppler multiplex transmission weight WD ntx (nsw) is multiplied as in the following equation (86) and output to the ntx-th transmitting antenna.

number

[0515] In this way, when transmitting from at least one pair of adjacent transmitting antennas 109 (for example, a pair of a horizontally polarized antenna and a vertically polarized antenna), the radar transmitter 100c switches over time between a transmission process in which radar transmit signals are multiplexed by applying the same Doppler shift amount and transmission weights with a phase difference of 90°, and a transmission process in which radar transmit signals are multiplexed by applying different Doppler shift amounts. For example, by using transmission weights with a phase difference of 90°, as in the first embodiment, the radar device 10c can use transmitting antennas that exceed the number Nt of transmitting antennas for multiplexing, and can use transmitting antennas with polarizations (for example, circular polarizations) different from the polarizations (for example, horizontally polarized and vertically polarized) of the transmitting antennas 109.

[0516] [Configuration of the radar receiver 200c] In FIG. 25, the operation of the radar receiver 200c from the mixer unit 204 to the beat frequency analyzer 208 is the same as that in the first embodiment.

[0517] The output switching unit 209 selectively switches the output of the beat frequency analysis unit 208 for each transmission period to the nsw-th Doppler analysis unit 210 out of the Nsw Doppler analysis units 210 based on the transmission switching index nsw input from the transmission switching control unit 112. For example, the output switching unit 209 selects the mod(m-1, Nsw)+1-th Doppler analysis unit 210 in the m-th transmission period Tr, where nsw=1, .., Nsw.

[0518] The signal processing unit 206 has Nsw Doppler analysis units 210-1 to 210-Nsw (or referred to as the first to Nsw-th Doppler analysis units 210). For example, data is input to the nsw-th Doppler analysis unit 210 every Nsw transmission periods (Nsw×Tr) by the output switching unit 209. The nsw-th Doppler analysis unit 210 analyzes data for Ncsub=Nc / Nsw transmission periods out of the Nc transmission periods (for example, beat frequency response RFT input from the beat frequency analysis unit 208). z (fb , m)) to obtain the distance index f b Doppler analysis is performed every 10 seconds. Note that Nc is set to an integer multiple of Nsw.

[0519] For example, if Ncode is a power of 2, FFT processing can be applied in Doppler analysis. In this case, the FFT size is Ncsub, and the maximum Doppler frequency at which aliasing does not occur, as derived from the sampling theorem, is ±1 / (2Nsw × Tr). Also, the Doppler frequency index f s The Doppler frequency interval is 1 / (Nsw×Ncsub×Tr), and the Doppler frequency index f s The range of f s = -Ncsub / 2,~, 0,~, Ncsub / 2-1.

[0520] For example, the output VFT of the Doppler analysis unit 210 of the z-th signal processing unit 206 z nsw (f b , f s ) is expressed by replacing Ncode with Ncsub, Loc with Nsw, and noc with nsw in equation (39), and other than this, the operation is the same as in the first embodiment.

[0521] In FIG. 25, the CFAR unit 211 performs CFAR processing (for example, adaptive threshold determination) using the outputs of the Nsw Doppler analyzers 210 of the first to Na-th signal processors 206, and determines the distance index f that gives the peak signal. b_cfar and the Doppler frequency index f s_cfar Extract.

[0522] The CFAR unit 211 calculates the output VFT of the nsw-th Doppler analyzer 210 in the first to Na-th signal processors 206, for example, as shown in the following equation (87). z nsw (f b , f s ) PowerFT with power added nsw (f b , f s), and performs two-dimensional CFAR processing consisting of a distance axis and a Doppler frequency axis (corresponding to relative velocity), or CFAR processing that combines one-dimensional CFAR processing. The CFAR unit 211 adaptively sets a threshold and selects a distance index f that has a received power greater than the threshold. b_cfar (nsw), Doppler frequency index f s_cfar (nsw), and received power information PowerFT nsw (f b_cfar (nsw), f s_cfar (nsw)) is output to the Doppler demultiplexing unit 216.

number

[0523] Or, the number of Doppler multiplexes in each nsw is the same (N DM (1)=…=N DM (Nsw)), and the amount of Doppler shift in each nsw is also the same, the CFAR unit 211 calculates PowerFT(f b , f s ) and perform CFAR processing in common. The CFAR unit 211 adaptively sets a threshold value and outputs a distance index f b_cfar , the Doppler frequency index f s_cfar , and received power information PowerFT (f b_cfar , f s_cfar ) is output to the Doppler demultiplexer.

number

[0524] The Doppler shift amount DOP ndm The phase rotation amount φ for adding ndmFor example, when equation (83) is used, the intervals of the Doppler shift amounts in the Doppler frequency domain in the output of the Doppler analysis unit 210 are equal, and when the intervals of the Doppler shift amounts ΔFD(nsw) are expressed in terms of the intervals of the Doppler frequency indexes, ΔFD(nsw)=Ncsub / N DM (nsw). Therefore, in the output of the Doppler analysis unit 210, peaks are detected at intervals of ΔFD for each signal that is Doppler shift multiplexed in the Doppler frequency domain, and the Doppler domain compression CFAR processing described as the processing of the CFAR unit 211 in the first embodiment can be applied. In this case, f s_comp =-ΔFD(nsw) / 2, ∼,-ΔFD(nsw) / 2−1. In addition, the CFAR unit 211 calculates, for example, the distance index f b_cfar , the Doppler frequency index f s_comp_cfar (nsw) is output to the Doppler demultiplexing unit 216.

[0525] In addition, the Doppler shift amount DOP ndm The phase rotation amount φ for adding ndm For example, when equation (84) is used, the intervals of the Doppler shift amounts in the Doppler frequency domain in the output of the Doppler analysis unit 210 are irregular, and the intervals of the Doppler frequency indexes are integer multiples of the interval ΔFD(nsw) of the Doppler shift amounts. Here, ΔFD(nsw)=Ncsub / (N DM (nsw)+N int Therefore, in the output of the Doppler analysis unit 210, peaks are detected at intervals of ΔFD(nsw) or intervals of an integer multiple of ΔFD(nsw) for each signal that is Doppler shift multiplexed in the Doppler frequency domain, and the Doppler domain compression CFAR processing described as the processing of the CFAR unit 211 in the first embodiment can be applied. In this case, f s_comp =-ΔFD(nsw) / 2, ∼,-ΔFD(nsw) / 2−1. In addition, the CFAR unit 211 calculates, for example, the distance index f b_cfar , the Doppler frequency index f s_comp_cfar (nsw) is output to the Doppler demultiplexing unit 216.

[0526] The operation when Doppler domain compression CFAR processing is applied will be described below.

[0527] In FIG. 25, the nsw-th Doppler demultiplexing unit 216 receives the distance index f b_cfar (nsw) and the Doppler frequency index f s_comp_cfar (nsw) and the output of the nsw-th Doppler analyzer 210, the Doppler multiplexed signal is separated.

[0528] For example, the separation of the non-uniformly spaced Doppler multiplexed signals using equation (84) is described in Patent Document 5, and a detailed description thereof will be omitted. b_cfar (nsw) and the Doppler frequency index f s_comp_cfar Output VFT of nsw-th Doppler analyzer 210 in (nsw) z nsw (f b_cfar , f s_comp_cfar (nsw)) and VFT z nsw (f b_cfar , f s_comp_cfar By using the power information of the Doppler frequency index, which is (nsw) + ΔFD(nsw) × (integer multiple)), the Doppler multiplexed signal DOP ndm The Doppler demultiplexing unit 216 can detect the transmitting antenna 109 assigned to the Doppler multiplexed signal based on the Doppler multiplexing assignment table. The Doppler demultiplexing unit 216 can detect the Doppler frequency of the target within the range of ±1 / (2Nsw×Tr).

[0529] Furthermore, for example, in separating the equally spaced Doppler multiplexed signals using equation (83), the Doppler multiplexing separation unit 216 separates the Doppler frequency of the target into ±1 / (2Nsw×N DM (nsw) × Tr), and the distance index f b_cfar (nsw) and the Doppler frequency index f s_comp_cfar Output VFT of nsw-th Doppler analyzer 210 in (nsw) znsw (f b_cfar , f s_comp_cfar (nsw)) and VFT z nsw (f b_cfar , f s_comp_cfar Doppler multiplex signal DOP is calculated using the Doppler frequency index (nsw) + ΔFD(nsw) × (integer multiple)). ndm Furthermore, the Doppler multiplexing separation unit 216 can detect the transmitting antenna 109 assigned to the Doppler multiplexed signal based on the Doppler multiplexing assignment table.

[0530] In the following, the distance index f input from the CFAR unit 211 is b_cfar (nsw) and the Doppler frequency index f s_comp_cfar For (nsw), the Doppler frequency of the ndmth Doppler multiplex signal of the detected target is expressed as FDP(ndm, f s_comp_cfar It is written as (nsw)).

[0531] By the above operation, the nsw-th Doppler demultiplexing unit 216 receives the distance index f b_cfar and the Doppler frequency index f s_comp_cfar and the output of the nsw-th Doppler analysis unit 210. The nsw-th Doppler demultiplexing unit 216 then performs Doppler demultiplexing based on the distance index f b_cfar and N DM The output VFT of the nsw-th Doppler analysis unit 210 of the Doppler frequency components of the (nsw) Doppler multiplexed signals z nsw (f b_cfar , FDP(1,f s_comp_cfar )), VFT z nsw (f b_cfar , FDP(2,f s_comp_cfar )),~, VFT z nsw (f b_cfar , FDP(N DM (nsw),f s_comp_cfar) is output to the direction estimation unit 214 as a separated reception signal of the Doppler multiplexed signal, where z=1, ∼, Na and nsw=1, ∼, Nsw.

[0532] In FIG. 25, the direction estimation unit 214 receives a distance index f b_cfar , the Doppler frequency index f s_comp_cfar The direction of the target is estimated based on the separated received signals of the Doppler multiplexed signals.

[0533] The direction estimation unit 214 receives the output VFT of each nsw-th Doppler analysis unit 210, which is a separated reception signal of the Doppler multiplexed signal. z nsw (f b_cfar , FDP(1,f s_comp_cfar )), VFT z nsw (f b_cfar , FDP(2,f s_comp_cfar )),~, VFT z nsw (f b_cfar , FDP(N DM (nsw),f s_comp_cfar ) for each virtual receiving array correlation vector hs nsw ( b_cfar , f s_comp_cfar ) is generated and direction estimation processing is performed. As shown in equation (89), each virtual receiving array correlation vector hs nsw ( b_cfar , f s_comp_cfar ) is N DM The direction estimator 214 calculates Nsw virtual receiving array correlation vectors hs1(f b_cfar , f s_comp_cfar )~hs Nsw (f b_cfar , f s_comp_cfar ) is used to perform direction estimation processing based on the phase difference between the receiving antennas 202 for the reflected wave signal from the target.

number

[0534] The direction estimation unit 214 calculates Nsw virtual receiving array correlation vectors hs1(f b_cfar , f s_comp_cfar )~hs Nsw (f b_cfar , f s_comp_cfar ), elements that are combinations of the polarization of a given transmitting antenna and the polarization of a receiving antenna are extracted, and direction estimation processing may be performed using a virtual receiving array correlation vector consisting of the extracted elements. This allows direction estimation results to be obtained for each polarization of a given transmitting antenna and each polarization of a receiving antenna.

[0535] For example, when the transmission switching control unit 112 performs control based on the Doppler multiplexing allocation table of FIG. 26, Nsw=2 virtual receiving array correlation vectors hs1(f b_cfar , f s_comp_cfar ) and hs2(f b_cfar , f s_comp_cfar ) is obtained.

[0536] Here, the virtual receiving array correlation vector hs1(f b_cfar , f s_comp_cfar ) includes elements such as those in the following equation (90), and the Doppler multiplexing number N DM (1)=2. In addition, since the signals Doppler multiplexed using DOP1 and DOP2 are reflected reception signals for the transmitted signal of positive circular polarization, the direction estimator 214 estimates the direction of the positive circular polarization by using the virtual receiving array correlation vector hs1(f b_cfar , f s_comp_cfar ) is used to perform direction estimation processing.

number

[0537] In addition, the virtual receiving array correlation vector hs2(f b_cfar , f s_comp_cfar ) includes elements such as those in the following equation (91), and the Doppler multiplexing number N DM(1)=4. Furthermore, since the signals Doppler multiplexed using DOP1, DOP2, DOP3, and DOP4 are reflected reception signals of horizontally polarized or vertically polarized transmission signals, the direction estimator 214 estimates the direction of linear polarization by using the virtual receiving array correlation vector hs2(f b_cfar , f s_comp_cfar ) is used to perform direction estimation processing.

number

[0538] Hereinafter, the virtual receiving array correlation vector h s2 (f b_cfar , f s_cfar ) for linearly polarized waves, and the correlation vector h for the positive circularly polarized wave virtual receiving array. s2 (f b_cfar , f s_cfar The direction estimation process for the normal circularly polarized wave using the azimuth angle θ is the same as that in the first embodiment, and the description of the operation will be omitted.

[0539] The direction estimation unit 214 may perform direction estimation processing using received signals containing a combination of different types of polarization. In this case, direction estimation results that are less dependent on polarization are obtained, and performing direction estimation processing using more virtual antennas improves the reception SNR, thereby improving the detection performance of the radar device 10c. In addition, performing direction estimation processing using the maximum aperture length of the available virtual receiving antennas also improves angular resolution.

[0540] The direction estimation unit 214 may also perform the direction estimation process using the same type of polarized antenna for both transmission and reception, as well as a direction estimation process combining different types of polarization, and use both of these direction estimation results as the direction estimation process result. This provides a direction estimation process result that is highly dependent on polarization and a direction estimation process result that is less dependent on polarization. The results of such direction estimation processes may be input to a target identification processing unit (not shown) to perform target identification processing.

[0541] The transmission weight W used in this second embodiment is ntx (nsw) represents the sign when the phase deviation between the transmitting antennas 109 is corrected in advance. Therefore, when the beam transmitting antenna is on, the transmission weight generation unit 113 calculates that the phase difference at each feed point of the two transmitting antennas that make up the beam transmitting antenna is the phase difference between the transmission weights assigned to each transmitting antenna.

[0542] Here, when a transmission weight of Amp×exp[jη] is set for one transmitting antenna 109 as the transmission weight to be assigned to the two transmitting antennas that make up the beam transmitting antenna, and a transmission weight of Amp×exp[j(η+π / 2)] or Amp×exp[j(η-π / 2)] is assigned to the other transmitting antenna 109, the phase difference between the feeding points of the two transmitting antennas for each transmission period will be 90° or -90°.

[0543] Similarly, when a transmission weight of Amp×exp[jη] is set for one transmitting antenna 109 as the transmission weight to be assigned to the two transmitting antennas that make up the beam transmitting antenna, and a transmission weight of Amp×exp[j(η+ξ)] or Amp×exp[j(η-ξ)] is assigned to the other transmitting antenna 109, the phase difference between the feed points of the two transmitting antennas in each transmission cycle will be ξ or -ξ. Here, for example, ξ may be in the range of π / 6 to 5π / 6 radians (= 30° to 150°).

[0544] The embodiments of the present disclosure have been described above.

[0545] [Other embodiments] (1) In the antenna arrangement examples shown in Figures 15 to 18, the polarization types of the transmitting antennas 109 have been described as being arranged as "H", "V", "H", and "V" from left to right as an example, but the arrangement of the transmitting antennas 109 is not limited to this, and it is sufficient that adjacent transmitting antennas 109 are arranged as a pair of a horizontally polarized antenna (H) and a vertically polarized antenna (V).

[0546] For example, in the antenna arrangements shown in Figures 15 to 18, the polarization types of the transmitting antenna 109 may be arranged from left to right as "H", "V", "V", and "H". In this arrangement, for example, isolation may be increased. Also, for example, in the antenna arrangements shown in Figures 15 to 18, the polarization types of the transmitting antenna 109 may be arranged from left to right as "V", "H", "H", and "V".

[0547] (2) In a radar device according to an embodiment of the present disclosure, the radar transmitter and the radar receiver may be located separately in physically separated locations. Also, in a radar receiver according to an embodiment of the present disclosure, the direction estimator and other components may be located separately in physically separated locations.

[0548] (3) The number of transmitting antennas Nt, the number of receiving antennas Na, and the number of Doppler multiplexing N used in one embodiment of the present disclosure DM , number of codes N CM , the number of beam transmitting antennas N BF The numerical values ​​of the parameters such as the number of pairs of transmitting antennas and the Doppler shift variable setting period are merely examples, and the present invention is not limited to these values.

[0549] Although not shown, a radar device according to an embodiment of the present disclosure includes, for example, a central processing unit (CPU), a storage medium such as a read-only memory (ROM) storing a control program, and a working memory such as a random access memory (RAM). In this case, the functions of the above-described units are realized by the CPU executing the control program. However, the hardware configuration of the radar device is not limited to this example. For example, each functional unit of the radar device may be realized as an integrated circuit (IC). Each functional unit may be implemented individually on a single chip, or a single chip may include some or all of the functional units.

[0550] Although various embodiments have been described above with reference to the drawings, it goes without saying that the present disclosure is not limited to such examples. It is clear that a person skilled in the art can conceive of various modifications or alterations within the scope of the claims, and it is understood that these also naturally fall within the technical scope of the present disclosure. Furthermore, the components of the above-described embodiments may be combined in any manner without departing from the spirit of the disclosure.

[0551] Furthermore, the notation "... section" in the above-described embodiments may be replaced with other notations such as "... circuitry," "... assembly," "... device," "... unit," or "... module."

[0552] In each of the above embodiments, the present disclosure has been described as an example configured using hardware, but the present disclosure can also be realized by software in cooperation with hardware.

[0553] Furthermore, each functional block used in the description of each of the above embodiments is typically realized as an LSI, which is an integrated circuit. The integrated circuit controls each functional block used in the description of the above embodiments and may have input and output terminals. These may be individually integrated into single chips, or some or all of them may be integrated into a single chip. While the term LSI is used here, it may also be called an IC, system LSI, super LSI, or ultra LSI depending on the level of integration.

[0554] Furthermore, the method of integration is not limited to LSI, but may be realized using a dedicated circuit or a general-purpose processor. It is also possible to use a field programmable gate array (FPGA) that can be programmed after LSI manufacturing, or a reconfigurable processor that can reconfigure the connections or settings of circuit cells inside the LSI.

[0555] Furthermore, if an integrated circuit technology that can replace LSI emerges due to advances in semiconductor technology or other derivative technologies, it is natural that such technology may be used to integrate functional blocks. The application of biotechnology, etc. is also a possibility.

[0556] Summary of this disclosure A radar device according to one embodiment of the present disclosure includes a plurality of transmitting antennas including a first transmitting antenna that radiates a first linearly polarized wave and a second transmitting antenna that is adjacent to the first transmitting antenna and radiates a second linearly polarized wave different from the first linearly polarized wave, and a transmitting circuit that multiplexes and transmits, from the plurality of transmitting antennas, transmission signals that have been given a phase rotation amount that causes a phase difference of ξ or −ξ between the first transmitting antenna and the second transmitting antenna in each transmission period.

[0557] In one embodiment of the present disclosure, the transmitting circuit code-multiplexes and transmits the transmission signal using a plurality of orthogonal codes, and the phases of the code elements corresponding to each transmission period differ by ξ or -ξ between a first orthogonal code for the transmission signal transmitted from the first transmitting antenna and a second orthogonal code for the transmission signal transmitted from the second transmitting antenna among the plurality of orthogonal codes.

[0558] In one embodiment of the present disclosure, ξ is a value ranging from 30° to 150°.

[0559] In one embodiment of the present disclosure, the transmission circuit sets the same Doppler shift amount for the first transmission antenna and the second transmission antenna.

[0560] In one embodiment of the present disclosure, when the plurality of transmitting antennas includes a plurality of pairs of the first transmitting antenna and the second transmitting antenna, the transmitting circuit sets a different Doppler shift amount for each of the plurality of pairs.

[0561] In one embodiment of the present disclosure, the antenna further includes a first Doppler analysis circuit that performs Doppler analysis using reflected wave signals of the transmission signals transmitted from the first transmitting antenna and the second transmitting antenna, reflected by a target, in either an odd-numbered or even-numbered transmission cycle; a second Doppler analysis circuit that performs Doppler analysis using the reflected wave signals in the other of the odd-numbered or even-numbered transmission cycles; a first separation circuit that separates a circularly polarized signal using either the output of the first Doppler analysis circuit or the output of the second Doppler analysis circuit; and a second separation circuit that separates at least one of the first linearly polarized signal and the second linearly polarized signal using both the output of the first Doppler analysis circuit and the output of the second Doppler analysis circuit.

[0562] In one embodiment of the present disclosure, the antenna further comprises a plurality of receiving antennas, the plurality of receiving antennas including antennas for receiving at least one of the first linearly polarized wave, the second linearly polarized wave, a circularly polarized wave in a first direction, and a circularly polarized wave in a second direction opposite to the first direction.

[0563] In one embodiment of the present disclosure, when the plurality of transmitting antennas includes a plurality of pairs of the first transmitting antenna and the second transmitting antenna, the transmitting circuit time-division multiplexes the transmitting signal from each of the plurality of pairs.

[0564] In one embodiment of the present disclosure, the transmission circuit switches over time between a first transmission process in which the transmission signals are multiplexed by applying the same Doppler shift amount and transmission weights with different phases of ξ or -ξ when transmitting from each of the first transmission antenna and the second transmission antenna, and a second transmission process in which the transmission signals are multiplexed by applying different Doppler shift amounts.

[0565] In one embodiment of the present disclosure, the transmission circuit performs, in the same transmission period, a first transmission process in which the transmission signals are multiplexed by applying the same Doppler shift amount and transmission weights with different phases of ξ or -ξ when transmitting from each of the first transmission antenna and the second transmission antenna, and a second transmission process in which the transmission signals are multiplexed by applying different Doppler shift amounts.

[0566] In one embodiment of the present disclosure, when the plurality of transmitting antennas includes a plurality of pairs of the first transmitting antenna and the second transmitting antenna, the transmitting circuit sets a different Doppler shift amount for each of the plurality of pairs in the first transmitting process.

[0567] In one embodiment of the present disclosure, the system further comprises a plurality of receiving antennas, wherein the plurality of transmitting antennas include a plurality of pairs of the first transmitting antenna and the second transmitting antenna, and the difference between the spacing between the phase centers of each of the plurality of pairs and the spacing between adjacent receiving antennas among the plurality of receiving antennas is a specified value based on the wavelength of the transmitting signal.

[0568] In one embodiment of the present disclosure, the specified value is any value in the range of 0.45 to 0.8 times the wavelength.

[0569] In one embodiment of the present disclosure, the system further comprises a plurality of receiving antennas, wherein in a first direction, the distance between the first transmitting antenna and the second transmitting antenna and the distance between adjacent receiving antennas among the plurality of receiving antennas are a first specified value based on the wavelength of the transmission signal, and in a second direction perpendicular to the first direction, the distance between some receiving antennas and other receiving antennas among the plurality of receiving antennas is a second specified value based on the wavelength of the transmission signal.

[0570] In one embodiment of the present disclosure, each of the first specified value and the second specified value is any value in a range of 0.45 to 0.8 times the wavelength. [Industrial Applicability]

[0571] The present disclosure is suitable for a radar device that detects a wide angle range. [Explanation of symbols]

[0572] 10, 10a, 10b, 10c radar equipment 100, 100a, 100b, 100c Radar transmitter 101 Radar transmission signal generator 102 Transmission signal generation control section 103 Modulation signal generator 104 VCO 105 Phase rotation amount setting unit 106 Doppler shift setting section 107 Encoding section 108 Phase Rotation Unit 109 Transmitting Antenna 110, 112 Transmission switching control unit 111 Transmission switching unit 113 Transmission weight generation unit 114 Transmission weight multiplication unit 200 Radar receiver 201 Antenna system processing unit 202 Receiving antenna 203 Receiving Radio Unit 204 Mixer section 205 LPF 206 Signal Processing Unit 207 AD conversion section 208 Beat Frequency Analysis Unit 209 Output switching unit 210 Doppler analysis unit 211 CFAR Department 212 Coded Doppler demultiplexer 213,216 Doppler demultiplexing unit 214 Direction estimation part 215 Code demultiplexer 300 Positioning output unit

Claims

1. a transmitting antenna unit including a plurality of transmitting antenna elements arranged at different positions; a transmission circuit that multiplexes and transmits from the transmission antenna a transmission signal of a first linearly polarized wave and a transmission signal of a second linearly polarized wave different from the first linearly polarized wave in each transmission period; Equipped with the transmission circuit applies a phase rotation amount that causes a phase difference of ξ or −ξ between the first linearly polarized transmission signal and the second linearly polarized transmission signal; The transmitting antenna unit radiating the first linearly polarized transmit signal and the second linearly polarized transmit signal from different positions; radiating the first linearly polarized transmit signal from a first one of the different positions; radiating the second linearly polarized transmit signal from a second position and a third position among the different positions; the first position is a position sandwiched between the second position and the third position, the first linearly polarized wave and the second linearly polarized wave have the same center frequency; Radar equipment.

2. the transmission circuit code-multiplexes and transmits the first linearly polarized transmission signal and the second linearly polarized transmission signal using a plurality of orthogonal codes; Among the plurality of orthogonal codes, a phase difference between a first orthogonal code for the first linearly polarized transmission signal and a second orthogonal code for the second linearly polarized transmission signal, the phases of code elements corresponding to each transmission period being ξ or −ξ; The radar device according to claim 1 .

3. The ξ is a value in the range of 30° to 150°. The radar device according to claim 1 .

4. the transmission circuit sets the same amount of Doppler shift for the first linearly polarized transmission signal and the second linearly polarized transmission signal. The radar device according to claim 2 .

5. The transmitting antenna unit radiating the first linearly polarized transmit signal from a fourth position among the different positions; the third position is a position sandwiched between the first position and the fourth position, The transmission circuit a Doppler shift amount set for the first linearly polarized transmission signal radiated from the first position and the second linearly polarized transmission signal radiated from the second position is different from a Doppler shift amount set for the first linearly polarized transmission signal radiated from the fourth position and the second linearly polarized transmission signal radiated from the third position; The radar device according to claim 2 .

6. a first Doppler analysis circuit that performs Doppler analysis using reflected wave signals of the first linearly polarized transmission signal and the second linearly polarized transmission signal reflected by a target in either an odd-numbered or even-numbered transmission period; a second Doppler analysis circuit that performs Doppler analysis using the reflected wave signal in either the odd-numbered or even-numbered transmission period; a first separation circuit that separates a circularly polarized signal using either an output of the first Doppler analysis circuit or an output of the second Doppler analysis circuit; a second separation circuit that separates at least one of the first linearly polarized transmit signal and the second linearly polarized transmit signal using both the output of the first Doppler analysis circuit and the output of the second Doppler analysis circuit. The radar device according to claim 1 .

7. a receiving antenna section including a plurality of transmitting antenna elements arranged at different positions; the receiving antenna unit receives the first linearly polarized wave, the second linearly polarized wave, and a circularly polarized wave; the circular polarization includes at least one of a circular polarization in a first direction and a circular polarization in a second direction opposite to the first direction; The radar device according to claim 1 .

8. The transmitting antenna unit radiating the first linearly polarized transmit signal from a fourth position among the different positions; the third position is a position sandwiched between the first position and the fourth position, The transmission circuit time-division multiplexing and transmitting the first linearly polarized transmission signal radiated from the first position and the second linearly polarized transmission signal radiated from the second position; time-division multiplexing and transmitting the first linearly polarized transmission signal radiated from the fourth position and the second linearly polarized transmission signal radiated from the third position; The radar device according to claim 1 .

9. a receiving antenna section including a plurality of receiving antenna elements arranged at different positions; The transmitting antenna unit a difference between a distance between phase centers of adjacent transmitting antenna element pairs among the plurality of transmitting antenna elements and a distance between adjacent receiving antenna elements among the plurality of receiving antenna elements is a specified value based on wavelengths of the first linearly polarized transmission signal and the second linearly polarized transmission signal; The radar device according to claim 1 .

10. The specified value is any value in the range of 0.45 to 0.8 times the wavelength. The radar device according to claim 9.

11. a receiving antenna including a plurality of receiving antenna elements arranged at different positions; In a first direction, a distance between the first position and the second position and a distance between adjacent receiving antennas among the plurality of receiving antenna elements are first specified values ​​based on wavelengths of the first linearly polarized transmission signal and the second linearly polarized transmission signal; In a second direction orthogonal to the first direction, a distance between some of the plurality of receiving antenna elements and other receiving antenna elements is a second specified value based on wavelengths of the first linearly polarized transmission signal and the second linearly polarized transmission signal. The radar device according to claim 1 .

12. Each of the first specified value and the second specified value is a value in a range of 0.45 to 0.8 times the wavelength. The radar device according to claim 11.

13. applying a phase rotation amount that causes a phase difference of ξ or −ξ between the first linearly polarized transmit signal and the second linearly polarized transmit signal in each transmission period; the first linearly polarized transmission signal and the second linearly polarized transmission signal are multiplexed and transmitted from a transmission antenna unit including a plurality of transmission antenna elements arranged at different positions; A radar signal processing method, comprising: The transmitting antenna unit radiating the first linearly polarized transmit signal and the second linearly polarized transmit signal from different positions; radiating the first linearly polarized transmit signal from a first one of the different positions; radiating the second linearly polarized transmit signal from a second position and a third position among the different positions; the first position is a position sandwiched between the second position and the third position, the first linearly polarized wave and the second linearly polarized wave have the same center frequency; Radar signal processing method.

14. the first linearly polarized transmission signal and the second linearly polarized transmission signal are code-multiplexed and transmitted using a plurality of orthogonal codes; Among the plurality of orthogonal codes, a phase of a code element corresponding to each transmission period differs by ξ or −ξ between a first orthogonal code for the first linearly polarized transmission signal and a second orthogonal code for the second linearly polarized transmission signal; 14. The radar signal processing method of claim 13.

15. The ξ is a value in the range of 30° to 150°.

14. The radar signal processing method of claim 13.

16. the same Doppler shift amount is set for the first linearly polarized transmission signal and the second linearly polarized transmission signal; 15. The radar signal processing method of claim 14.

17. radiating the first linearly polarized transmit signal from a fourth position among the different positions; the third position is a position sandwiched between the first position and the fourth position, a Doppler shift amount set for the first linearly polarized transmission signal radiated from the first position and the second linearly polarized transmission signal radiated from the second position is different from a Doppler shift amount set for the first linearly polarized transmission signal radiated from the fourth position and the second linearly polarized transmission signal radiated from the third position; 15. The radar signal processing method of claim 14.

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