Reactance compensation circuit

Abstract

To provide a reactance compensation circuit capable of eliminating matching deviation based on the influence of the output capacity of a semiconductor switch as much as possible.SOLUTION: The phase shift amount control circuit 65 outputs a control signal for controlling the phase shift amount based on the capacitor voltage VCB. The variable phase shift circuit 62 shifts the phase based on the control signal. The bridge circuit 63c is driven by the drive signal shifted by the variable phase shift circuit 62, and the voltage outputted from the bridge circuit 63c is controlled to have a predetermined difference in phase from the voltage on the side of the power transmission source with respect to the bridge circuit 63c.SELECTED DRAWING: Figure 2

Description

[Technical Field]

[0001] The present invention relates to a reactance compensation circuit. [Background technology]

[0002] In a power supply system that transmits power via a coupler, a matching circuit composed of a capacitor and a coil is used to match the coupler's impedance with the impedance of the power source and the load, thereby transmitting power efficiently. However, under conditions in which the relative positions of the power transmitting and receiving sides change, the reactance component of the coupler's impedance changes, causing a mismatch with the matching circuit composed of a capacitor and a coil alone, making it impossible to transmit power efficiently. As a solution to this problem, a method has been proposed in which an automatic compensation circuit is connected between the power source and the load to compensate for the changed reactance component (see, for example, Patent Document 1).

[0003] This automatic compensation circuit is a bridge circuit consisting of a capacitor and a switch, which turns on and off at a phase difference of 90 degrees from the transmission voltage of the power source. When there is a phase difference between the transmission voltage and the transmission current, the charge and discharge of the capacitor become unbalanced according to the on / off operation of this switch, causing the capacitor voltage to rise. In this way, the automatic compensation circuit generates a voltage with an amplitude corresponding to the capacitor voltage and a phase difference of 90 degrees from the transmission voltage. This voltage generates a current to align the phase difference between the transmission voltage and the load current, which occurs due to a mismatch caused by a change in the reactance component. The capacitor voltage increases until the on / off timing and current phase are 90 degrees out of phase with the transmission voltage. In other words, the capacitor voltage automatically increases until the phase difference between the transmission voltage and the load current is 90 degrees. This circuit does not detect the voltage, current, or power applied to the power source or load and perform feedback control based on the detected information, but rather enables automatic reactance compensation with a simple configuration. [Prior art documents] [Patent documents]

[0004] [Patent Document 1] International Publication No. 12 / 164845 Summary of the Invention [Problem to be solved by the invention]

[0005] However, the phase of the bridge circuit's output voltage lags behind the on / off timing of the semiconductor switch, which is 90 degrees out of phase with the transmission voltage, by the charge / discharge time of the output capacitance. This phase lag causes a phase shift between the transmission voltage and current after automatic compensation for reactance. The amount of this shift is determined by the charge / discharge capacitance of the semiconductor switch, and becomes larger as the capacitor voltage increases. Because the capacitor voltage fluctuates depending on the magnitude of the matching mismatch, it is not possible to take measures such as shifting the gate drive phase by a fixed amount in advance.

[0006] As described above, the configuration of Patent Document 1 shown in the Background Art section can automatically compensate for reactance and achieve matching with a simple configuration, but mismatch due to the influence of output capacitance remains. An object of the present disclosure is to provide a reactance compensation circuit that can minimize mismatch caused by the influence of output capacitance. [Means for solving the problem]

[0007] The reactance compensation circuit of claim 1 comprises a bridge circuit, a voltage detection circuit, a phase shift amount control circuit, and a variable phase shift unit. The bridge circuit is connected in series between a high-frequency power supply and a load and comprises a capacitor and a switch. The voltage detection circuit detects the voltage of the capacitor in the bridge circuit.

[0008] The phase shift amount control circuit outputs a control signal that controls the amount of phase shift based on the detected voltage. The variable phase shift unit shifts the phase based on the control signal. The bridge circuit is driven by a drive signal with a phase shifted by the variable phase shift circuit, and the output voltage of the bridge circuit is controlled to have a predetermined phase difference with the voltage on the high-frequency power supply side of the bridge circuit, thereby minimizing mismatch due to the influence of output capacitance. [Brief explanation of the drawings]

[0009] [Figure 1] FIG. 1 is an electrical configuration diagram illustrating the overall configuration of a power supply device according to a first embodiment. [Figure 2] FIG. 1 is an electrical configuration diagram illustrating a schematic configuration of a reactance compensation circuit according to a first embodiment. [Figure 3] FIG. 10 is a diagram showing a change in the effective capacitance of a semiconductor switch with respect to the amplitude of a capacitor voltage in the first embodiment. [Figure 4] Configuration examples of a voltage detection circuit, a phase shift amount control circuit, and a variable phase shift circuit in the first embodiment [Figure 5] FIG. 1 is a diagram illustrating a relationship between voltages of various parts in the first embodiment. [Figure 6] Configuration examples of a voltage detection circuit, a phase shift amount control circuit, and a variable phase shift circuit in the second embodiment [Figure 7] FIG. 10 is a diagram schematically illustrating the relationship between voltages of various parts in the second embodiment. [Figure 8] Configuration examples of a voltage detection circuit, a phase shift amount control circuit, and a variable phase shift circuit in the third embodiment [Figure 9] Configuration examples of a voltage detection circuit, a phase shift amount control circuit, and a variable phase shift circuit in the fourth embodiment [Figure 10] Configuration examples of a voltage detection circuit, a phase shift amount control circuit, and a variable phase shift circuit in the fifth embodiment [Figure 11] FIG. 13 is a diagram schematically illustrating the relationship between voltages of various parts in the fifth embodiment. [Figure 12]FIG. 13 is a diagram showing a comparison of the relationship between the voltages of the various parts depending on the amount of phase change in the fifth embodiment. [Figure 13] Configuration examples of a voltage detection circuit, a phase shift amount control circuit, and a variable phase shift circuit in the sixth embodiment [Figure 14] 13A and 13B are circuit configuration examples of a voltage detection circuit, a phase shift amount control circuit, and a variable phase shift circuit according to the sixth embodiment. DETAILED DESCRIPTION OF THE INVENTION

[0010] Hereinafter, several embodiments of the reactance compensation circuit will be described. Components that perform the same functions in the various embodiments will be assigned the same reference numerals, and descriptions thereof will be omitted.

[0011] (First embodiment) The first embodiment will be described with reference to FIGS. 1 is configured by cascading a matching circuit 30, a coupler 40, a compensation circuit 50, and a reactance compensation circuit 60, and is provided to efficiently supply power from a power transmission power source 20 to a load 70. The power transmission power source 20 is a high-frequency power source that generates an AC voltage of, for example, 6.78 MHz, and outputs the AC voltage to the matching circuit 30.

[0012] The matching circuit 30 has a circuit configuration in which, for example, coils La1 and La2, and coils La3 and La4 are connected in series to the differential paths, respectively, and a capacitor Ca is connected in parallel between the common connection points of coils La1 and La2, and coils La3 and La4. The matching circuit 30 is provided on the power transmitting side to perform impedance matching between the power transmitting source 20 and the coupler 40. In an electric field coupling type wireless power transfer system, for example, the coupler 40 has a power transmitting electrode and a power receiving electrode facing each other, and can be described by an equivalent circuit using capacitors Cb1 and Cb2 as shown in FIG. 1. The compensation circuit 50 is provided on the power receiving side to compensate for the reactance of capacitors Cb1 and Cb2 in the equivalent circuit of the coupler 40, and is composed of coils Lc1 and Lc2.

[0013] Under operating conditions in which the relative positions of the power transmitting electrodes on the power transmitting side and the power receiving electrodes on the power receiving side change, the capacitances of capacitors Cb1 and Cb2 on the equivalent circuit of coupler 40 change. Therefore, a compensation circuit 50 determined by fixed circuit constants alone will cause a mismatch due to the change in capacitance of capacitors Cb1 and Cb2 on the equivalent circuit, making it impossible to transmit power efficiently. For this reason, a reactance compensation circuit 60 is provided between power transmitting power source 20 and load 70 in addition to matching circuit 30 and compensation circuit 50.

[0014] The reactance compensation circuit 60 of this embodiment is configured to compensate for reactance. As shown in Fig. 2, the reactance compensation circuit 60 includes the following components: a phase shift circuit 61, a variable phase shift circuit 62, a compensation voltage generation circuit 63, a voltage detection circuit 64, and a phase shift amount control circuit 65.

[0015] Although not shown in detail, the phase shift circuit 61 receives a control signal having the same phase as the power transmission voltage of the power transmission power source 20. The phase shift circuit 61 shifts the phase of the input control signal by a fixed phase difference (for example, 90°) and outputs the shifted signal to the variable phase shift circuit 62. The variable phase shift circuit 62 detects the capacitor voltage V of the capacitors C1 and C2 detected by the voltage detection circuit 64. CB The phase is further shifted by an amount of phase shift specified by a phase shift amount control circuit 65 based on the output of the variable phase shift circuit 62, and the phase is output to a compensation voltage generation circuit 63. The compensation voltage generation circuit 63 is driven by the output of the variable phase shift circuit 62. The compensation voltage generation circuit 63 outputs power from the power transmission power source 20 to a load 70 while automatically compensating for changes in reactance that could not be fully compensated for by the compensation circuit 50 with fixed circuit constants formed by the coils Lc1 and Lc2 alone.

[0016] The compensation voltage generating circuit 63 is composed of a first gate drive circuit 63a, a second gate drive circuit 63b, and a bridge circuit 63c. The bridge circuit 63c includes semiconductor switches M1 to M4 and capacitors C1 to C4 as shown in the figure. The semiconductor switches M1 to M4 in this embodiment are composed of N-channel MOSFETs.

[0017] The first gate drive circuit 63a and the second gate drive circuit 63b receive the signal shifted by the variable phase shift circuit 62 as a drive signal and drive the gates of the semiconductor switches M1 to M4.

[0018] The bridge circuit 63c includes a circuit in which a series connection circuit of capacitors C1 and C2 is connected to a circuit in which the drain sources of semiconductor switches M1 and M2 are connected in series, and a circuit in which a series connection circuit of capacitors C3 and C4 is connected to a circuit in which the drain sources of semiconductor switches M3 and M4 are connected in series. The compensation circuit 50 described above outputs transmission power between a common connection point Ni1 of the semiconductor switches M1 and M2 and a common connection point Ni2 of the semiconductor switches M3 and M4.

[0019] The reactance compensation circuit 60 is provided to compensate for reactance and outputs a voltage with a phase difference of 90° with respect to the phase of the transmission voltage. The reactance compensation circuit 60 outputs a differential voltage to the load 70 from between the common connection point No. 1 of the capacitors C1 and C2 and the common connection point No. 2 of the capacitors C3 and C4.

[0020] The voltage detection circuit 64 detects the voltage V of the series circuit of the capacitors C1 and C2. CB This capacitor voltage V CB is the voltage applied to the semiconductor switches M1 to M4 when the semiconductor switches M1 to M4 are in the OFF state, and has a positive correlation with the charge / discharge capacitance of the semiconductor switches M1 to M4. Hereinafter, the voltage V of the series connection circuit of the capacitors C1 and C2 will be referred to as CB The capacitor voltage V CB The voltage detection circuit 64 detects the capacitor voltage V, which appears at several hundred volts. CB is converted to a level of about several volts that can be used to control the variable phase shift circuit 62.

[0021] The phase shift amount control circuit 65 detects the capacitor voltage V CB The phase shift amount is determined based on the above and a control signal for the phase shift amount is output to the variable phase shift circuit 62.

[0022] The charge / discharge time of the output capacitance of the semiconductor switches M1 to M4 is proportional to the time-converted effective capacitance value of the semiconductor switches M1 to M4. As shown in Figure 3, as the drain-source voltage of the semiconductor switches M1 to M4 increases, that is, as the amplitude of the output voltage of the bridge circuit 63c increases, the effective capacitance value also increases, and the phase delay due to the charge / discharge time also increases. The variable phase shift circuit 62 compensates for the phase delay that occurs in accordance with the magnitude of the amplitude of the output voltage by advancing the phase by the time equivalent to the charge / discharge time of the output capacitance of the semiconductor switches M1 to M4.

[0023] By adopting such a configuration, it is possible to compensate for the phase delay caused by the charge / discharge time of the output capacitance of the semiconductor switches M1 to M4. Regardless of the degree of mismatch, the phase shift between the voltage and current can be suppressed, and power can be transmitted efficiently.

[0024] <Specific Examples of the Voltage Detection Circuit 64, the Phase Shift Amount Control Circuit 65, and the Variable Phase Shift Circuit 62> Specific examples of the voltage detection circuit 64, the phase shift amount control circuit 65, and the variable phase shift circuit 62 shown in FIG. 2 will be described below with reference to FIG.

[0025] 4, the voltage detection circuit 64 includes an operational amplifier OP1 and resistors Ra1 to Ra4. The resistors Ra1 to Ra4 are connected to the input terminal of the operational amplifier OP1, and the input capacitor voltage V CB is compressed to about 1 / 100 of the voltage V A It outputs the voltage V A is input to the phase shift amount control circuit 65.

[0026] The phase shift amount control circuit 65 includes an inverting amplifier circuit in the front stage that combines an operational amplifier OP2 and resistors Rb1 to Rb3, and a characteristic inverting circuit in the rear stage that includes an operational amplifier OP3, resistors Rb4 to Rb6, and a capacitor C1, and the capacitor voltage V CB It outputs a voltage that changes linearly with changes in the detected voltage.

[0027] The phase shift amount control circuit 65 generates a voltage VA The inverted and amplified voltage V B The phase shift amount control circuit 65 also outputs the voltage V B The voltage V is inverted upside down around 2.5V by the characteristic inversion circuit in the subsequent stage. C The voltage V B -V A characteristics, voltage V C -V A This shows the characteristic of the capacitor voltage V CB The direction of the phase shift can be adjusted by adjusting the voltage V C is input to the variable phase shift circuit 62.

[0028] The variable phase shift circuit 62 is configured by a digital circuit including an A / D converter 62a and a programmable timing IC 62b. C is input to the A / D converter 62a, the A / D converter 62a outputs a voltage V C The programmable timing IC 62b converts the transmitted voltage into digital form and outputs it to the programmable timing IC 62b. A drive signal synchronized with the transmitted voltage is input to the programmable timing IC 62b. This drive signal has a fixed phase difference (for example, a phase difference of 90°) with respect to the phase of the transmitted voltage.

[0029] The programmable timing IC 62b shifts the drive signal by an amount of phase shift corresponding to the output digital value of the A / D converter 62a and outputs the shifted drive signal. CB The drive signal synchronized with the transmission voltage is shifted by an amount of phase shift corresponding to the value of .

[0030] At this time, the phase shift amount control circuit 65 controls the capacitor voltage V CB Since the variable phase shift circuit 62 outputs a linear value in response to the detected value of the capacitor voltage V CB The higher the detected value, the smaller the phase shift amount. CBThe amount of phase shift can be changed in accordance with the change in the output capacitance, and the phase delay due to the charging and discharging time of the output capacitance can be compensated for.

[0031] As described above, according to this embodiment, the phase shift amount control circuit 65 controls the capacitor voltage V CB The variable phase shift circuit 62 determines the amount of phase shift based on this and outputs a control signal for the amount of phase shift. Based on this control signal, the variable phase shift circuit 62 shifts the drive signal synchronized with the transmission voltage by the amount of phase shift and outputs the resulting drive signal to drive the compensation voltage generation circuit 63. As a result, it is possible to minimize mismatching due to the influence of the output capacitance of the semiconductor switches M1 to M4. Furthermore, control using the programmable timing IC 62b makes circuit implementation and adjustment easy.

[0032] (Second embodiment) The second embodiment will be described with reference to Figures 6 and 7. The second embodiment differs from the first embodiment in that a phase shift amount control circuit 265 is used instead of the phase shift amount control circuit 65. The same parts as in the first embodiment are given the same reference numerals and their explanations will be omitted, and only the different parts will be described.

[0033] In the configuration of this embodiment, attention is paid to the nonlinearity of the relationship between the time-converted effective capacitance of the semiconductor switches M1 to M4 and the source-drain voltage of the semiconductor switches M1 to M4, and the phase shift amount control circuit 265 controls the voltage V A The detected value is converted into a nonlinear value and output.

[0034] 6, the phase shift amount control circuit 265 employs a configuration in which the inverting amplifier circuit in the upstream stage of the phase shift amount control circuit 65 is modified. The phase shift amount control circuit 265 includes a nonlinear amplifier circuit in the upstream stage that combines an operational amplifier OP2, resistors Rb2, Rb3, Rb31 to Rb34, and Zener diodes ZD1 to ZD3.

[0035] A plurality of resistors Rb31-Rb33 and a plurality of Zener diodes ZD1-ZD3 are connected in series, and this series connection circuit is connected in parallel between the inverting input terminal and the output terminal of the operational amplifier OP2. In addition, a resistor Rb34 is connected between the inverting input terminal and the output terminal of the operational amplifier OP2.

[0036] As shown in the upper diagram of Figure 7, the voltage V B and voltage V A The nonlinear amplifier circuit in the preceding stage of the phase shift amount control circuit 265 is connected to a relatively low voltage V A When the voltage V B A relatively large change in voltage V A When the voltage V B The nonlinear amplifier circuit in the previous stage of the phase shift amount control circuit 265 reduces the change in the input voltage V A When inverting and amplifying, a nonlinear conversion is performed to obtain the voltage V B It can be output as

[0037] Furthermore, the phase shift amount control circuit 265 is connected to the characteristic inversion circuit explained in the first embodiment in the subsequent stage of the nonlinear amplifier circuit in the previous stage, and outputs a voltage V C As a result, the voltage V A and voltage V C The relationship is also nonlinear.

[0038] The phase shift amount control circuit 265 outputs the voltage V C is input as a control signal to the variable phase shift circuit 62. At this time, the phase shift amount control circuit 265 controls the capacitor voltage V CB The variable phase shift circuit 62 outputs a nonlinear value in response to the detected value of the capacitor voltage V CB The higher the detected value, the smaller the rate of change in the amount of phase shift. CBThe amount of phase shift can be changed in accordance with the change in the reactance component, and the matching error can be minimized.

[0039] As a result, the phase delay due to charging and discharging of capacitors C1 to C4 can be compensated for more accurately, and power can be transmitted efficiently. Also, the phase shift amount control circuit 265 may be configured using a microcomputer, and in this case as well, the nonlinearity of the output capacitance of semiconductor switches M1 to M4 can be easily reproduced.

[0040] According to the configuration of this embodiment, the capacitor voltage V CB The nonlinearity of the detected value is realized by a combination of the Zener voltages of the Zener diodes ZD1 to ZD3 in the feedback loop of the operational amplifier OP2 and the feedback resistors Rb31 to Rb34. The configuration according to this embodiment makes it possible to compensate for the phase shift without performing complex feedback control. Moreover, the inherent advantage of the simple control system of the reactance compensation device 10 is not lost.

[0041] (Third embodiment) The third embodiment will be described with reference to Fig. 8. The third embodiment differs from the first embodiment in that a variable phase shift circuit 362 is used instead of the variable phase shift circuit 62. The same parts as those in the first embodiment are given the same reference numerals and their explanations will be omitted, and only the different parts will be described.

[0042] The variable phase shift circuit 362 includes a front-stage fixed amount shift unit 62c and a rear-stage variable phase shift circuit 62d. The fixed amount shift unit 62c includes an operational amplifier OP4, resistors Rd1 and Rd2, and a capacitor C2 in the illustrated form. The resistor Rd1 is connected between the inverting input terminal of the operational amplifier OP4 and the input node, and the capacitor C2 is connected between the non-inverting input terminal of the operational amplifier OP4 and the input node. The resistor Rd2 is connected between the non-inverting input terminal of the operational amplifier OP4 and ground.

[0043] The fixed amount shifter 62c receives a sine wave signal synchronized with the transmission voltage of the power transmission source 20, shifts the phase of this sine wave signal by a pre-adjusted fixed amount, and outputs the shifted signal to the downstream variable phase shift circuit 62d. This fixed amount of shift is pre-adjusted to be, for example, 90° relative to the phase of the transmission voltage.

[0044] The variable phase shift circuit 62d includes an operational amplifier OP5, a capacitor C3, resistors Rc1 to Rc5, and a MOSFET_Ma as shown in the figure. The MOSFET_Ma is a junction-type FET. A resistor Rc1 is connected between the inverting input terminal of the operational amplifier OP5 and the input node, and a capacitor C3 is connected between the non-inverting input terminal of the operational amplifier OP5 and the input node.

[0045] An externally controlled variable resistor Rr is connected between the non-inverting input terminal of the operational amplifier OP5 and ground. The variable resistor Rr is composed of resistors Rc4 and Rc5 and a MOSFET_Ma. The output voltage V of the phase shift amount control circuit 65 C is applied to the gate of MOSFET_Ma via a voltage divider circuit made up of resistors Rc2 and Rc3.

[0046] The output voltage V of the phase shift amount control circuit 65 C is lower than a predetermined value, the MOSFET_Ma is turned off, and the resistance value of the variable resistor Rr becomes equal to the resistance value of the resistor Rc5. As a result, the resistance value of the variable resistor Rr becomes relatively large. C is higher than a predetermined value, the MOSFET_Ma is turned on, and the combined resistance of the variable resistor Rr becomes the parallel resistance value of the resistors Rc4 and Rc5, which becomes relatively small.

[0047] Capacitor voltage V CB When is zero or very low, the mismatch is small. C becomes lower than a predetermined value, the MOSFET_Ma is turned off and the variable resistor Rr becomes large. Therefore, the negative amount of the phase shift can be made relatively large.

[0048] In this state, the variable phase shift circuit 62d adjusts the shift amount of the fixed amount shift unit 62c at the previous stage so that the phase difference between the transmission voltage and the output voltage of the bridge circuit 63c becomes 90 degrees. CB As the voltage Vcc rises, the channel of the MOSFET_Ma begins to open, decreasing the resistance value of the variable resistor Rr and reducing the negative amount of the phase shift.

[0049] That is, when the mismatch of the compensation circuit 50 becomes large, the capacitor voltage V CB The output of the variable phase shift circuit 62d is input to the comparator CP, which shapes the phase-adjusted voltage into a square wave and outputs it as a gate drive signal.

[0050] As a result, the phase delay due to the charging and discharging of the output capacitance of the semiconductor switches M1 to M4 can be compensated for, and the output voltage of the bridge circuit 63c can be maintained at a 90-degree phase with respect to the phase of the transmission voltage.

[0051] In the configuration of this embodiment, the resistance value of the variable resistor Rr connected to the input of the operational amplifier OP5 is adjusted by turning on and off the MOSFET_Ma to control the amount of phase shift. In this case, the amount of phase shift = -2 × tan -1 The resistance values ​​of the resistors Rc1, Rc4, and Rc5 and the capacitance value of the capacitor C3 can be determined using the relationship (RC / 2πf).This embodiment also provides the same effects as the previous embodiment.

[0052] (Fourth embodiment) The fourth embodiment will be described with reference to Fig. 9. The fourth embodiment differs from the third embodiment in that the phase shift amount control circuit 265, which was also described in the second embodiment, is used instead of the phase shift amount control circuit 65. The same parts as in the third embodiment are given the same reference numerals and their description will be omitted, and only the different parts will be described.

[0053] In the configuration of this embodiment, attention is paid to the nonlinearity of the relationship between the time-converted effective capacitance of the semiconductor switches M1 to M4 and the source-drain voltage of the semiconductor switches M1 to M4, and the phase shift amount control circuit 265 controls the voltage V A The detected value is subjected to nonlinear conversion and output.

[0054] Specifically, as shown in Fig. 9, the phase shift amount control circuit 265 has a configuration in which the inverting amplifier circuit in the front stage of the phase shift amount control circuit 65 is modified. This configuration has also been explained in the second embodiment, so the explanation will be omitted. Then, as shown in the lower diagram of Fig. 7, the voltage V A and voltage V C The relationship is nonlinear.

[0055] The phase shift amount control circuit 265 outputs the voltage V C is input as a control signal to variable phase shift circuit 62d. Variable phase shift circuit 62d shifts the phase based on this control signal, so that it can more accurately compensate for the phase delay caused by charging and discharging capacitors C1 to C4, thereby transmitting power efficiently. Furthermore, phase shift amount control circuit 265 may be configured using a microcomputer, and in this case as well, the nonlinearity of the output capacitance of semiconductor switches M1 to M4 can be easily reproduced.

[0056] According to the configuration of this embodiment, the capacitor voltage V CB By using a circuit that outputs a nonlinear value in response to the detected value, it is possible to more accurately compensate for the phase delay due to charging and discharging. According to the configuration of this embodiment, the phase shift can be compensated for without performing complex feedback control, and the inherent advantage of the simple control system of this circuit is not lost.

[0057] (Fifth embodiment) The fifth embodiment will be described with reference to Figures 10 to 12. The fifth embodiment differs from the second embodiment in that a variable phase shift circuit 462 is used instead of the variable phase shift circuit 62. The same parts as those in the second embodiment are given the same reference numerals and their explanations will be omitted, and only the different parts will be described.

[0058] The variable phase shift circuit 462 includes a differential square wave generating circuit 62e in the preceding stage and a variable phase shift circuit 62f in the subsequent stage. The differential square wave generating circuit 62e includes operational amplifiers OP6 to OP8, resistors Re1 to Re8, and capacitors C3 and C4 in the illustrated configuration.

[0059] The inverting input terminal of the operational amplifier OP6 receives a divided voltage obtained by dividing the power supply voltage Vcc using resistors Re1 and Re2, while the non-inverting input terminal receives the input voltage IN. The input voltage IN is a square wave signal synchronized with the transmission voltage and is set to a phase that is 90° out of phase with the transmission voltage of the transmission power source 20. The operational amplifier OP6 compares the input voltage IN with the divided voltage and outputs a voltage. The output of the operational amplifier OP6 is input to a circuit using the operational amplifier OP7.

[0060] Resistor Re3 is connected between the output terminal of operational amplifier OP6 and the inverting input terminal of operational amplifier OP7. Resistor Re4 and capacitor C3 are connected in parallel between the inverting input terminal and output terminal of operational amplifier OP7. Resistor Re5 is connected between the non-inverting input terminal of operational amplifier OP7 and ground. This causes operational amplifier OP7 to invert and amplify the output of operational amplifier OP6.

[0061] A filter consisting of a capacitor C4 and a resistor Re6 is configured at the output of the operational amplifier OP7, and outputs a triangular wave voltage to the non-inverting input terminal of the comparator CP1 that configures the variable phase shift circuit 62f (see FIG. 12). The triangular wave voltage is also input to the inverting input terminal of the operational amplifier OP8 via the resistor Re7.

[0062] A resistor Re8 is connected between the inverting input terminal and output terminal of the operational amplifier OP8. A capacitor C4 is connected between the non-inverting input terminal of the operational amplifier OP8 and ground. The operational amplifier OP8 inverts the triangular wave voltage output from the filter and outputs it to the non-inverting input terminal of the comparator CP2 that constitutes the variable phase shift circuit 62f.

[0063] On the other hand, in this embodiment, the biases of the resistors Ra1 to Ra4, Rb2 to Rb6, and Rb31 to Rb34 that constitute the voltage detection circuit 64 and the phase shift amount control circuit 265 are appropriately set, so that the voltage V B -Voltage V A , voltage V C -Voltage V A The input / output characteristic is nonlinear. The phase shift amount control circuit 265 converts the input voltage V A is level-shifted to a voltage V centered around zero volts. C Convert it to and output it.

[0064] The phase shift amount control circuit 265 controls the voltage V C to the inverting input terminals of comparators CP1 and CP2 of the variable phase shift circuit 62f. The variable phase shift circuit 62f includes comparators CP1 and CP2 and NOR gates D1 and D2 in the illustrated form. The NOR gates D1 and D2 are configured as so-called RS flip-flops. The outputs of the comparators CP1 and CP2 are input as the R input and S input of the RS flip-flop, respectively. The NOR gates D1 and D2 then output Q outputs or / Q outputs as the output signal OUT.

[0065] When the variable phase shift circuit 462 of this embodiment is used, as shown in FIG. C As the value of changes, the outputs of the comparators CP1 and CP2 change and the phase of the output signal OUT changes.

[0066] For example, voltage V C When the voltage V is relatively high, the output of the comparator CP2 rises relatively slowly, and the rise of the output signal OUT also slows. C When the voltage V is relatively low, the output of the comparator CP2 rises relatively quickly, and the output signal OUT also rises quickly. C It changes depending on the size of

[0067] Similarly, the voltage V C When the voltage V is relatively high, the output of the comparator CP1 rises relatively slowly, and the rise of the output signal OUT also slows. C When V is a relatively low voltage, the output of the comparator CP1 rises relatively quickly, and the output signal OUT also rises quickly. C The rising timing of comparators CP1 and CP2 changes depending on the magnitude of voltage V C Since the period and duty of the output signal OUT do not change in principle, the period and duty of the output signal OUT change in synchronization with the change in the magnitude of the output signal OUT.

[0068] The variable phase shift circuit 462 is connected to a voltage V C Since the output signal OUT, which detects the phase lead of the gate drive signal, is used as the gate drive signal, the phase delay can be compensated for. Moreover, even if the amount of phase shift changes, the duty of the gate drive signal can be kept constant. With the configuration of this embodiment, the duty of the gate drive signal can be kept constant while eliminating distortion.

[0069] In this embodiment, the input voltage IN is a square wave signal synchronized with the transmission voltage. When the input voltage IN is a square wave signal, the output of the variable phase shift circuit 362 described in the previous embodiment will be distorted if the variable phase shift circuit 362 is used. In this case, if the phase shift amount changes, the duty of the gate drive signal will change, and therefore the waveform must be shaped.

[0070] For this reason, in the above-described embodiment, a sine wave signal is input to the variable phase shift circuit 362. When the input voltage IN is a square wave voltage as in this embodiment, the variable phase shift circuit 462 can be used to digitally process the voltage, eliminating the need for waveform shaping.

[0071] (Sixth embodiment) The sixth embodiment will be described with reference to FIGS. Even if the imaginary / real part ratio of the load impedance remains the same and the mismatch continues, the larger the real part of the load impedance, the smaller the load current Is. As a result, it takes time to charge and discharge the effective capacitance of the bridge circuit 63c, and the phase lag increases.

[0072] In this case, the capacitor voltage V that determines the amplitude of the output voltage of the bridge circuit 63c CB It is desirable to control the variable phase shift circuit 62 taking into consideration the detection result of the load current Is flowing through the load 70 in addition to the detection result of the load current Is.

[0073] An example configuration is shown in Fig. 13. In the configuration shown in Fig. 13, a rectifier 71 and a main load 72 are connected as a load 70 downstream of a compensation voltage generating circuit 63, and a phase shift amount control circuit 665 is provided instead of the phase shift amount control circuit 65 in order to take into account the detection result of the load current Is. A shunt resistor Rs is connected between the rectifier 71 and the main load 72. A load current detection circuit 66 is connected to the shunt resistor Rs.

[0074] The load current detection circuit 66 converts the load current Is into a voltage using a shunt resistor Rs and detects it. The load current detection circuit 66 is a current-voltage conversion circuit configured as shown in the figure, including an operational amplifier OP9, resistors Rg1 to Rg5, and a MOSFET_M5. A phase shift amount control circuit 665 converts the capacitor voltage V detected by the voltage detection circuit 64 into a voltage. CB and the load current Is detected by the load current detection circuit 66, the phase shift amount control circuit 665 determines the phase shift amount and outputs it as a control signal to the variable phase shift circuit 62. The phase shift amount control circuit 665 causes the variable phase shift circuit 62 to change the phase using the determined phase shift amount.

[0075] In this case, the phase shift amount control circuit 665 may reduce the phase shift amount as the load current Is detected as smaller by the load current detection circuit 66. Conversely, the phase shift amount control circuit 665 may increase the phase shift amount as the load current Is detected as larger by the load current detection circuit 66. With this configuration, the phase shift can be appropriately compensated for even under conditions where the value of the load current Is changes significantly.

[0076] 14 shows a specific example of the phase shift amount control circuit 665. In the phase shift amount control circuit 665, the circuit of the operational amplifier OP3 at the output stage is configured as an adder. The output of the load current detection circuit 66 is given to the inverting input terminal of the operational amplifier OP3 through a resistor Rb4b. With this configuration, if the load current Is becomes smaller, the current detection result of the load current detection circuit 66 becomes smaller, and the output voltage of the load current detection circuit 66 becomes lower, and the phase shift amount control circuit 665 increases the output voltage V C can be lowered.

[0077] This results in a voltage V A Even under the same conditions, the smaller the load current Is, the greater the voltage V C This reduces the phase shift amount of the variable phase shift circuit 62. This increases the amount of lead in phase of the gate drive signal relative to the phase of the transmission voltage.

[0078] For example, voltage V A is a predetermined value, the phase of the output signal OUT should be adjusted to satisfy the condition that the phase of the output voltage of the bridge circuit 63c lags behind the phase of the transmission voltage by 270 degrees. This is because the ideal state is achieved where the phase of the output voltage of the bridge circuit 63c leads the phase of the transmission voltage by 90 degrees.

[0079] To satisfy this condition, the resistance values ​​of the resistors Rb4, Rb4b, Rb5, and Rb6 of the phase shift amount control circuit 665 are set to control the voltage V C Under this adjustment condition, when the load current Is decreases, the voltage V Cdecreases, and the amount of phase shift becomes smaller than 270 degrees. The phase of the output signal OUT advances by this amount. When the load current Is decreases, the charge / discharge time of the output capacitance of the semiconductor switches M1 to M4 increases, but the increase in charge / discharge time is compensated for by the phase advance of the output signal OUT, and the phase of the output voltage of the bridge circuit 63c can be maintained at a 90-degree advance with respect to the phase of the transmission voltage. According to this embodiment, the reactance can be compensated for taking the load current Is into consideration, and good follow-up to changes in the load current Is can be achieved.

[0080] (Other embodiments) The present disclosure is not limited to the above-described embodiment, and for example, the following modifications or extensions are possible. 1, the compensation circuit 50 has been described as having coils Lc1 and Lc2 connected in series between the input and output, but the present invention is not limited to this. For example, the compensation circuit 50 may be configured in a T-shape with a series coil, a parallel capacitor, and a series coil in this order, or in a π-shape with a parallel capacitor, a series coil, and a parallel capacitor in this order.

[0081] Although the reactance compensation circuit 60 has been described as being provided between the compensation circuit 50 and the load 70, it may also be provided between the power transmission power source 20 and the matching circuit 30. The matching circuit 30 may also be configured as a π type, or may be configured by swapping the coil and capacitor shown in FIG.

[0082] Although the variable phase shift unit is configured by combining the phase shift circuit 61 that shifts the phase by a fixed phase difference and the variable phase shift circuit 62, 362, 462 that shifts the phase by a variable shift amount in the above embodiment, the present invention is not limited to this. For example, if the variable phase shift circuit 62, 362, 462 has a function to offset the phase, the phase shift circuit 61 may be omitted.

[0083] Although the present disclosure has been described based on the above-described embodiment, it is understood that the present disclosure is not limited to the embodiment or the structure described in the embodiment. The present disclosure also encompasses various modifications and modifications within the equivalent range. In addition, various combinations and forms, as well as other combinations and forms including only one element, more than one element, or less than one element, are also within the scope and spirit of the present disclosure. [Explanation of symbols]

[0084] In the drawings, 20 is a power transmission power supply (high frequency power supply), 60 is a reactance compensation circuit, 61 is a phase shift circuit (variable phase shift section), 63c is a bridge circuit, 64 is a voltage detection circuit, 65, 265, 665 are phase shift amount control circuits, 62, 362, 462 are variable phase shift circuits (variable phase shift sections), 66 is a load current detection circuit, 70 is a load, M1 to M4 are semiconductor switches (switches), and C1 to C4 are capacitors.

Claims

1. a bridge circuit (63c) connected between the high frequency power supply (20) and the load (70) and consisting of a capacitor and a switch; a voltage detection circuit (64) for detecting the voltage of the capacitor of the bridge circuit; a phase shift amount control circuit (65; 265; 665) that outputs a control signal for controlling the phase shift amount based on the detected voltage; a variable phase shift unit (61, 62; 61, 362; 61, 462) that shifts the phase of the voltage on the side of the high frequency power supply based on the control signal, a reactance compensation circuit that drives the bridge circuit with a drive signal having the phase shifted by the variable phase shift unit, and controls the output voltage of the bridge circuit to have a predetermined phase difference with a voltage on the high-frequency power supply side of the bridge circuit.

2. 2. The reactance compensation circuit according to claim 1, wherein the variable phase shift unit reduces the amount of phase shift as the voltage of the capacitor detected by the voltage detection circuit increases.

3. 3. The reactance compensation circuit according to claim 2, wherein the variable phase shift unit reduces the rate of change of the amount of phase shift as the voltage of the capacitor detected by the voltage detection circuit increases.

4. a load current detection circuit (66) for detecting a load current; 2. The reactance compensation circuit according to claim 1, wherein the phase shift amount control circuit outputs the control signal that controls the phase shift amount based on the detected voltage of the voltage detection circuit and the load current of the load current detection circuit, and the variable phase shift unit shifts the phase based on the control signal.

5. 5. The reactance compensation circuit according to claim 4, wherein the phase shift amount control circuit (665) reduces the amount of phase shift as the detected load current becomes smaller.

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