Power Conversion Equipment

Abstract

To prevent a gain of a frequency transfer function from an output current command value to an actual output current of an inverter from deviating significantly from 1 or prevent an inverter output current phase from deviating from a phase of the output current command value at a command angular frequency due to fluctuations in motor equipment constants, resulting in a poor performance of an AC induction motor.SOLUTION: When a frequency of a fundamental component of an output current command value (Iαref, Iβref) is set as a command angular frequency (ωre), a gain of a closed-loop frequency transfer function from the output current command value (Iαref, Iβref) to a current detection value (iα, iβ), is set to almost 1 at a command angular frequency (ω re) and a phase of the closed-loop frequency transfer function from the output current command value (Iαref, Iβref) to the current detection value (iα, iβ) is set to almost 0° at the command angular frequency (ωre).SELECTED DRAWING: Figure 2

Description

[Technical field]

[0001] The present disclosure relates to a power conversion device that drives an AC induction motor. [Background technology]

[0002] Patent Document 1 discloses a power conversion device including an inverter that drives an AC induction motor by converting DC to three-phase AC through switching operation and outputting the converted AC, a current control unit that calculates a voltage command value, and a switching operation control unit that controls the switching operation of the inverter based on the voltage command value calculated by the current control unit. In this power conversion device, the current control unit determines a d-axis current and a q-axis current from the rotation angle and phase current of the AC induction motor, and generates a d-axis voltage command value so that the deviation between the d-axis current command value and the d-axis current is small, and generates a q-axis voltage command value so that the deviation between the q-axis current command value and the q-axis current is small. [Prior art documents] [Non-patent literature]

[0003] [Non-Patent Document 1] Kondo, Yuki, "Study on Application of Induction Motor Speed ​​Sensorless Vector Control to Railway Vehicle Drive", Transactions of the Institute of Electrical Engineers of Japan, Vol. 125, No. 1, pp. 1-8, January 1, 2005 Summary of the Invention [Problem to be solved by the invention]

[0004] However, in a power conversion device that drives an AC induction motor, there is a demand for making the output current of the inverter follow the AC output current command value without phase shift. In the non-patent document 1, current control is performed on the AC induction motor by speed sensorless vector control. In the speed sensorless vector control, the rotation coordinate transformation of the detected current is performed based on the motor equipment constant, so if the motor equipment constant fluctuates due to current change, temperature change, or individual variation, an error is likely to occur in the detected current after the rotation coordinate transformation. Therefore, at the command angular frequency that is the frequency of the fundamental wave component of the output current command value, the gain of the frequency transfer function from the output current command value to the actual output current of the inverter may significantly deviate from 1, or the phase of the inverter's output current may significantly deviate from the phase of the output current command value, which may result in failure to obtain good performance of the AC induction motor.

[0005] An object of the present disclosure is to prevent a situation in which good performance of an AC induction motor cannot be obtained due to fluctuations in the motor device constants, which causes the gain of the frequency transfer function from the output current command value to the actual output current of the inverter to deviate significantly from 1 at the command angular frequency, or the phase of the inverter's output current to deviate significantly from the phase of the output current command value. [Means for solving the problem]

[0006] The first aspect of the present disclosure is a power supply for a power converter including an inverter (12) that drives an AC induction motor (3) by converting DC to three-phase AC through a switching operation and outputting the converted AC. α *, V β A current control unit (20) that calculates a voltage command value (V α *, V β and a switching operation control unit (16) for controlling the switching operation of the inverter (12) based on a reference value (Iα*), the power conversion device further comprising a current detection unit (15) for detecting an output current of the inverter (12), and the current control unit (20) detects an output current command value (Iα*) of the inverter (12). ref , Iβ ref ) and the detection value (iu , i v , i w ) based on the current detection value (i α , i β ) based on the voltage command value (V α *, V β a current control operation for calculating an output current command value (Iα*) is performed for a U-phase current flowing through a U-phase coil of the AC induction motor (3), a V-phase current flowing through a V-phase coil of the AC induction motor (3), and a W-phase current flowing through a W-phase coil of the AC induction motor (3), respectively, or the current control operation is performed for an α-axis current and a β-axis current corresponding to the U-phase current, the V-phase current, and the W-phase current, respectively, ref , Iβ ref ) to the command angular frequency (ω re ), the output current command value (Iα ref , Iβ ref ) to the current detection value (i α , i β The gain of the closed loop frequency transfer function up to the command angular frequency (ω re ) is approximately 1, and the output current command value (Iα ref , Iβ ref ) to the current detection value (i α , i β ) is the phase of the closed loop frequency transfer function up to the command angular frequency (ω re ) is approximately 0°.

[0007] In the first mode, no rotational coordinate transformation is performed, and the output current of the inverter (12) is set to the output current command value (I α ref , I β ref ), the effect of fluctuations in the motor constants on the current control is small. This makes it possible to prevent the AC induction motor (3) from losing its good performance due to fluctuations in the motor constants.

[0008] In a second aspect of the present disclosure, in the first aspect, the current control operation is performed by controlling the output current command value (Iα ref , Iβref ) and current detection value (i α , i β ) based on the difference between the voltage command value (V α *, V β *) is calculated from the difference, and the voltage command value (V α *, V β *) is the gain of the frequency transfer function up to the command angular frequency (ω re ) is infinite, and the voltage command value (V α *, V β *) is the phase of the frequency transfer function up to the command angular frequency (ω re ) is approximately 0°.

[0009] In the second mode, the output current command value (Iα ref , Iβ ref ) and current detection value (i α , i β ) to obtain the voltage command value (V α *, V β *) The gain of the frequency transfer function is re ), the output current command value (Iα ref , Iβ ref ) and the detected current value (i α , i β ) command angular frequency (ω re ) component can be made almost zero.

[0010] A third aspect of the present disclosure is the second aspect, wherein the current control operation calculates the feedback value (FB) based on a difference between the difference and a feedback value (FB), and calculates the voltage command value (V α *, V β *), and the feedback value (FB) is the command angular frequency (ω re ) is characterized in that it includes an amplified value of the component.

[0011] In the third aspect, the feedback value (FB) is set to the command angular frequency (ω re) component, the voltage command value (V α *, V β *) is the gain of the frequency transfer function up to the command angular frequency (ω re ) can be made infinite.

[0012] A fourth aspect of the present disclosure is directed to the third aspect, wherein the current control operation is a first integral value (Z 1α ) is multiplied by a first gain (f1) to calculate a first proportional value, and the first integral value (Z 1α ) is integrated to obtain the second integral (Z 2α ) by a second gain (f2) to calculate a second proportional value, and the voltage command value (V α *, V β *) and the command angular frequency (ω re The amount of change in the first and second gains (f1, f2) when the command angular frequency (ω re ) is proportional to the square of the change in

[0013] In the fourth aspect, the first and second gains (f1, f2) are set to a command angular frequency (ω re ), the current control operation can be performed over a wide range of command angular frequencies (ω re ) can be adapted.

[0014] A fifth aspect of the present disclosure is a method for manufacturing a drive circuit according to the fourth aspect, wherein the command angular frequency (ω re ) squared change amount is D, and the output current command value (Iα ref , Iβ ref ) to the current detection value (i α , i β ) is the crossover angular frequency of the gain of the open loop frequency transfer function up to ωc, the inductance of the primary winding of the AC induction motor (3) is L1, the inductance of the secondary winding of the AC induction motor (3) is L2, and the mutual inductance between the primary winding and the secondary winding is M, re The amount of change in the first gain (f1) when L1 changes is D·(L1·L2-M 2) / L2, and the command angular frequency (ω re The amount of change in the second gain (f2) when ) changes is 3D ωc (L1 L2-M 2 ) / L2.

[0015] In the fifth embodiment, the current control unit (20) controls a wide range of command angular frequencies (ω re ) can perform control suitable for the AC induction motor (3).

[0016] A sixth aspect of the present disclosure is the fourth or fifth aspect, wherein the current control operation is performed by controlling the current detection value (i α , i β ) by a third gain (f3) to calculate a third proportional value, and the first proportional value, the second proportional value, and the third proportional value are subtracted from 0 to calculate the voltage command value (V α *, V β *), and the output current command value (Iα ref , Iβ ref ) to the current detection value (i α , i β ) is a crossover angular frequency of a gain of an open loop frequency transfer function up to ωc, an inductance of a primary winding of the AC induction motor (3) is L1, an inductance of a secondary winding of the AC induction motor (3) is L2, a mutual inductance between the primary winding and the secondary winding is M, a resistance of the primary winding is R1, a resistance of the secondary winding is R2, and the third gain (f3) is f3, the following formula (A) is established.

[0017]

number

[0018] In the sixth aspect, the current control unit (20) can perform control suitable for the AC induction motor (3) over a wide range of command angular frequencies.

[0019] A seventh aspect of the present disclosure is the first or second aspect, wherein the current control operation is performed by controlling the output current command value (Iα ref , Iβ ref) and current detection value (i α , i β ) and the voltage command value (V α *, V β *) is multiplied by a predetermined current control gain to obtain the voltage command value (V α *, V β *) is calculated.

[0020] An eighth aspect of the present disclosure is characterized in that, in any one of the first to seventh aspects, the inverter further includes a rectifier unit (11) that rectifies an AC voltage of an AC power source (2) to the DC voltage, and a capacitance element (14) connected between input terminals (12a, 12b) of the inverter (12), and the capacitance element (14) constitutes an LC filter (LC) together with an inductance component between the AC power source (2) and the capacitance element (14).

[0021] A ninth aspect of the present disclosure relates to the eighth aspect, and relates to a voltage command value (V ) for detecting an electrical physical quantity including a component of a resonant frequency of the LC filter (LC), and a voltage command value (V ) for detecting an electrical physical quantity including a component of a resonant frequency of the LC filter (LC) based on a value detected by the physical quantity detection unit (18). α *, V β The output voltage correction unit (50) is further provided for correcting the output voltage difference (*).

[0022] In the ninth aspect, the output voltage correction unit (50) α *, V β By correcting the above-mentioned resonance frequency component contained in the physical quantity, the voltage fluctuation of the capacitance element (14) caused by the resonance of the LC circuit (LC) can be suppressed.

[0023] A tenth aspect of the present disclosure is the ninth aspect, further comprising a reactor (13) connected between the AC power supply (2) and the capacitance element (14), and the physical quantity is a voltage (V L ) is characterized in that

[0024] In the tenth aspect, the output voltage correction unit (50) calculates a voltage command value (V α *, V β *) correction, the voltage (V L ) can be attenuated.

[0025] An eleventh aspect of the present disclosure is a method for manufacturing a capacitance element (14) according to the ninth aspect, wherein the physical quantity is a voltage (V dc ) is characterized in that

[0026] In an eleventh aspect, the output voltage correction unit (50) calculates a voltage command value (V α *, V β *) to obtain the voltage (V dc ) can be attenuated.

[0027] A twelfth aspect of the present disclosure is characterized in that, in the ninth aspect, the power supply further includes a reactor (13) connected between the AC power source (2) and the capacitance element (14), and the physical quantity is a current flowing through the reactor (13).

[0028] In the twelfth aspect, the output voltage correction unit (50) calculates a voltage command value (V α *, V β By correcting *, the component of the resonant frequency contained in the current flowing through the reactor (13) can be attenuated.

[0029] A thirteenth aspect of the present disclosure, in the ninth aspect, is characterized in that the physical quantity is a current flowing through the capacitance element (14).

[0030] In the thirteenth aspect, the output voltage correction unit (50) calculates a voltage command value (V α *, V β *) makes it possible to attenuate the component of the resonant frequency contained in the current flowing through the capacitance element (14).

[0031] The fourteenth aspect of the present invention is the ninth aspect, characterized in that the physical quantity is electrical energy stored in the capacitance element (14).

[0032] In the fourteenth aspect, the output voltage correction unit (50) calculates a voltage command value (V α *, V β *) makes it possible to attenuate the resonant frequency component contained in the electrical energy stored in the capacitance element (14).

[0033] The fifteenth aspect, in the ninth aspect, is characterized in that the physical quantity is an input current from the AC power supply (2).

[0034] In the fifteenth aspect, the output voltage correction unit (50) calculates a voltage command value (V α *, V β *) makes it possible to attenuate the resonant frequency component contained in the input current from the AC power supply (2).

[0035] In a sixteenth aspect, in any one of the first to fifteenth aspects, the current control unit (20) calculates a slip frequency (ω s ) and estimate the slip frequency (ω s ), the output current command value (Iα ref , Iβ ref ) is calculated.

[0036] In the sixteenth aspect, the current control unit (20) calculates the estimated slip frequency (ω s ) can be reflected in the output current of the inverter (12). [Brief description of the drawings]

[0037] [Figure 1] FIG. 1 is a block diagram showing a configuration of a power conversion system according to the first embodiment. [Diagram 2] FIG. 2 is a block diagram of a sine wave tracking controller. [Diagram 3]FIG. 3 is a block diagram of a power conversion system. [Figure 4] FIG. 4(a) is a gain diagram showing the gain of the closed-loop frequency transfer function from the output current command value to the current detection value, and FIG. 4(b) is a phase diagram showing the phase of the closed-loop frequency transfer function from the output current command value to the current detection value. [Diagram 5] FIG. 5(a) is a gain diagram showing the gain of the frequency transfer function from the output current command value to the voltage command value, and FIG. 5(b) is a phase diagram showing the phase of the frequency transfer function from the output current command value to the voltage command value. [Figure 6] FIG. 6 is a view corresponding to FIG. 1 of the second embodiment. [Figure 7] FIG. 7 is a block diagram showing the configuration of a repetitive controller. [Figure 8] FIG. 8 is a view corresponding to FIG. 1 of the third embodiment. [Figure 9] FIG. 9 is a view corresponding to FIG. 1 of the fourth embodiment. DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0038] Hereinafter, embodiments of the present disclosure will be described with reference to the drawings. Note that the following embodiments are essentially preferred examples and are not intended to limit the scope of the present invention, its applications, or its uses.

[0039] First Embodiment 1 shows a power conversion system (1). The power conversion system (1) includes a power conversion device (10) according to the first embodiment of the present disclosure, and an AC induction motor (3). The power conversion device (10) converts three-phase AC power source power input from a three-phase AC power source (2) into AC power having a desired frequency and a desired voltage, and supplies the AC induction motor (3).

[0040] The power conversion device (10) includes a converter (11) as a rectifier, an inverter (12), a reactor (13), a capacitance element (14), a current detection unit (15), a current control unit (20), and a PWM modulation unit (16) as a switching operation control unit.

[0041] The converter (11) rectifies the AC voltage of the three-phase AC power supply (2) to DC and outputs the DC to first and second output nodes (11a, 11b). More specifically, the converter (11) is a full-wave rectifier circuit. The converter (11) has six diodes (11D) connected in a bridge configuration. The cathodes of the diodes (11D) face the first output node (11a) and the anodes face the second output node (11b).

[0042] The inverter (12) converts the DC output from the converter (11) into a three-phase AC by switching and outputs the three-phase AC to drive the AC induction motor (3). More specifically, the inverter (12) has six switching elements (12S) and six freewheeling diodes (12D). The six switching elements (12S) are bridge-connected. More specifically, the inverter (12) has three switching legs connected between its first and second input terminals (12a, 12b). Each switching leg is formed by connecting two switching elements (12S) in series with each other.

[0043] In each of the three switching legs, the midpoint between the upper arm switching element (12S) and the lower arm switching element (12S) is connected to a coil of each phase (u-phase, v-phase, w-phase coil) of the AC induction motor (3). A free wheel diode (12D) is connected in inverse parallel to each switching element (12S).

[0044] One end of the reactor (13) is connected to a first output node (11a) of the converter (11), and the other end of the reactor (13) is connected to a first input terminal (12a) of the inverter (12).

[0045] The capacitance element (14) is connected between the first and second input terminals (12a, 12b) of the inverter (12). Thus, the reactor (13) is connected between the three-phase AC power supply (2) and the capacitance element (14).

[0046] The capacitance value of the capacitance element (14) is set so that it can hardly smooth the output voltage of the converter (11), but can suppress the ripple voltage caused by the switching operation of the inverter (12). The ripple voltage is a voltage fluctuation of the DC link voltage (Vdc) corresponding to the switching frequency of the switching element (12S). Therefore, the DC link voltage (Vdc), which is the voltage of the capacitance element (14), contains a pulsating component corresponding to the frequency of the AC voltage of the three-phase AC power source (2). Since the three-phase AC power source (2) is a three-phase power source, the pulsating component corresponding to the frequency of the three-phase AC power source (2) is six times the frequency of the three-phase AC power source (2).

[0047] The capacitance element (14) constitutes an LC filter (LC) together with an inductance component between the three-phase AC power supply (2) and the capacitance element (14). The inductance component includes a reactor (13).

[0048] The current detection unit (15) detects the output current of the inverter (12) and outputs a detected value (i u , i v , i w In detail, the current detection unit (15) detects a U-phase current flowing through a U-phase coil of the AC induction motor (3) and outputs a detection value (i u ), a U-phase current detection unit (15a) detects a V-phase current flowing through a V-phase coil of the AC induction motor (3) and outputs a detection value (i v ), and a V-phase current detection unit (15b) that detects a W-phase current flowing through a W-phase coil of the AC induction motor (3) and outputs a detection value (i wIn addition, although not shown in the embodiment, instead of the current detection unit (15) that directly detects the current of each phase, a shunt resistor or the like is used to detect the input current of the inverter (12), and the detection value (i u , i v , i w ) may be provided.

[0049] The current control unit (20) controls an α-axis output current command value (I α ref ), and β-axis output current command value (I β ref Then, the current control unit (20) calculates an output current command value (I α ref , I β ref ) and the current detection value (i α , i β ) and the voltage command value (V α *, V β *) is calculated for each of the α-axis current and the β-axis current.

[0050] Specifically, the current control unit (20) includes a uvw-αβ conversion unit (21), a primary magnetic flux estimator (22), a magnetic flux subtractor (23), a magnetic flux controller (24), a torque current command value generating unit (25), an MT-αβ conversion unit (26), and a sine wave tracking controller (27).

[0051] The UVW-αβ conversion unit (21) converts the detection values ​​(i u , i v , i w ) to the α-axis current detection value (i α ) and β-axis current detection value (i β ) and output.

[0052] The primary magnetic flux estimator (22) converts the α-axis current detection value (i α ) and β-axis current detection value (i β ) based on the primary magnetic flux (Ψ^) and phase (θ ψ The primary magnetic flux (Ψ^) and phase (θ ψ ^) can be estimated by a known method disclosed in "Position sensorless vector control of IPMSM based on primary flux linkage estimation on stationary coordinates and sinusoidal current tracking control system (Joint Study Group on Semiconductor Power Conversion and Motor Drives, General Semiconductor Power Conversion and AC Drives)" by Ryota Takahashi and 3 others, IEEJ Study Group Materials. SPC, IEEJ, August 28, 2015, pp. 51-56. Specifically, the voltage command value (V α *, V β The stator flux (Ψ 1α , Ψ 1β ) and estimate the stator flux (Ψ 1α , Ψ 1β ) based on the primary magnetic flux (Ψ^) and phase (θ ψ The α-axis component of the voltage drop is calculated by the resistance of the primary winding and the α-axis current detection value (i α ), and the β-axis component of the voltage drop is calculated by multiplying the resistance of the primary winding and the β-axis current detection value (i β ) can be calculated by multiplying it by . As the integration means, a circuit combining a phase compensator and an imperfect integration can be used.

[0053] The magnetic flux subtractor (23) subtracts the primary magnetic flux (Ψ^) estimated by the primary magnetic flux estimator (22) from the magnetic flux command value (Ψ*) and outputs the subtraction result.

[0054] The magnetic flux controller (24) calculates a magnetic flux current command value (i M )

[0055] The torque current command value generating unit (25) divides the torque command value (T*) by the number of pole pairs (Pn) and the armature flux linkage (ψ^) of the AC induction motor (3) to obtain a torque current command value (iT )

[0056] The MT-αβ converter (26) converts the phase (θ ψ ^), the flux current command value (i M ), and the torque current command value (i T ) to the α-axis output current command value (I α ref ) and β-axis output current command value (I β ref ) and output.

[0057] The sine wave tracking controller (27) determines the output current command value (I α ref , I β ref ) and the detection value (i u , i v , i w ) based on the current detection value (i α , i β ) and the voltage command value (V α *, V β A current control operation for calculating the α-axis current and the β-axis current corresponding to a U-phase current flowing through a U-phase coil of the AC induction motor (3), a V-phase current flowing through a V-phase coil of the AC induction motor (3), and a W-phase current flowing through a W-phase coil of the AC induction motor (3) is performed.

[0058] Specifically, as shown in FIG. 2, the sine wave tracking controller (27) has an α-axis voltage command value calculation section (27a) and a β-axis voltage command value calculation section (27b).

[0059] The α-axis voltage command value calculation unit (27a) and the β-axis voltage command value calculation unit (27b) respectively calculate the corresponding output current command value (I α ref , I β ref ) and the corresponding current detection value (iα , i β ) and the voltage command value (V α *, V β That is, the α-axis voltage command value calculation unit (27a) calculates the α-axis output current command value (I α ref ) and the α-axis current detection value (i α ) and the α-axis voltage command value (V α In addition, the β-axis voltage command value calculation unit (27b) calculates the β-axis output current command value (I β ref ) and the β-axis current detection value (i β ) and the β-axis voltage command value (V β *) is calculated.

[0060] Specifically, the α-axis voltage command value calculation unit (27a) and the β-axis voltage command value calculation unit (27b) each include a first subtractor (31), a second subtractor (32), a first integrator (33), a second integrator (34), a feedback value generation unit (35), a first proportional unit (36), a second proportional unit (37), a third proportional unit (38), and a third subtractor (39).

[0061] The first subtractor (31) calculates the output current command value (I α ref , I β ref ) to the current detection value (i α , i β ) is subtracted from the output current command value (I α ref , I β ref ) and current detection value (i α , i β ) to output the difference.

[0062] The second subtractor (32) subtracts the output of the first subtractor (31), i.e., the output current command value (I α ref , I β ref) and current detection value (i α , i β ) and outputs the difference between the difference and the feedback value (FB).

[0063] The first integrator (33) receives the output of the second subtractor (32), i.e., the output current command value (I α ref , I β ref ) and current detection value (i α , i β ) and the feedback value (FB) to obtain the first integral value (Z 1α ) to output.

[0064] The second integrator (34) calculates the first integral value (Z 1α ) and integrate the second integral (Z 2α ) to output.

[0065] The feedback value generating unit (35) calculates the second integral value (Z 2α ) to a specified command angular frequency (ω re ) to generate the feedback value (FB). re ) is the output current command value (I α ref , I β ref ) is the frequency of the fundamental component. The feedback value (FB) is the output current command value (I α ref , I β ref ) and current detection value (i α , i β ) command angular frequency (ω re ) includes the amplification value of the component.

[0066] The first proportional controller (36) calculates the first integral value (Z 1α ) is multiplied by a first gain (f1) to calculate a first proportional value.

[0067] The second proportional controller (37) (f1) is a second integral value (Z 2α ) is multiplied by a second gain (f2) to calculate a second proportional value.

[0068] The third proportional controller (38) converts the detected current value (i α , i β ) is multiplied by a third gain (f3) to calculate a third proportional value.

[0069] The third subtractor (39) subtracts the first proportional value calculated by the first proportional unit (36), the second proportional value calculated by the second proportional unit (37), and the third proportional value output by the third proportional unit (38) from 0 to obtain a voltage command value (V α *, V β *) is calculated.

[0070] In this way, the current control operation by the current control unit (20) is performed based on the output current command value (I α ref , I β ref ) and current detection value (i α , i β ) based on the difference between the voltage command value (V α *, V β The current control operation by the current control unit (20) calculates an output current command value (I α ref , I β ref ) and current detection value (i α , i β ) and the feedback value (FB), and calculates a voltage command value (V α *, V β The current control operation by the current control unit (20) calculates a voltage command value (V*) based on the first proportional value calculated by the first proportional unit (36) and the second proportional value calculated by the second proportional unit (37). α *, V β *) is calculated.

[0071] The PWM modulation unit (16) converts the voltage command value (V α *, V β *) based on the switching signal (S u , S v , S w ), thereby PWM-controlling the switching operation of the switching element (12S) of the inverter (12).

[0072] Here, the state equation of the AC induction motor (3) is as shown in the following equation (1).

[0073] In the following, R1 is the resistance of the primary winding, R2 is the resistance of the secondary winding, L1 is the inductance of the primary winding of the AC induction motor (3), L2 is the inductance of the secondary winding of the AC induction motor (3), M is the mutual inductance between the primary winding and the secondary winding of the AC induction motor (3), and ω re is the command angular frequency, i α is the α-axis current, i β is the β-axis current, ψ 1α is the α-axis component of the stator flux, ψ 1β is the β-axis component of the stator flux, v α is the α-axis voltage command value, v β indicates the β-axis voltage command value.

[0074]

number

[0075] Therefore, the block diagram of the power conversion system (1) is represented as shown in FIG.

[0076] If the first gain (f1) is f1, the second gain (f2) is f2, and the third gain (f3) is f3, the first to third gains (f1, f2, f3) are set as shown in the following formulas (2) to (4). c is the output current command value (I α ref , I β ref ) to the current detection value (i α , i β) shows the crossover angle frequency of the gain of the open loop frequency transfer function up to

[0077]

number

[0078]

number

[0079]

number

[0080] The first to third gains (f1, f2, f3) are calculated based on the block diagram of FIG. 3 as shown in the above formulas (2) to (4). In this way, the first to third gains (f1, f2, f3) determined based on the state equation of the AC induction motor (3) are used, so that the current control unit (20) can control a wide range of command angular frequencies (ω re ) can perform control suitable for the AC induction motor (3).

[0081] Therefore, the command angular frequency (ω re The amount of change in the first and second gains (f1, f2) when the command angular frequency (ω re Specifically, it is proportional to the change in the command angular frequency (ω re When the square of the change in the command angular frequency (ω re ) changes, the change in the first gain (f1) is D·(L1·L2-M 2 ) / L2, and the command angular frequency (ω re The amount of change in the second gain (f2) when ) changes is 3D ωc (L1 L2-M 2 ) / L2.

[0082] Figure 4(a) shows the command angular frequency (ω re When the command frequency is set to 120Hz, i.e., the command frequency is set to 754rad / s, the output current command value (I αref , I β ref ) to each current detection value (i α , i β ) is a gain diagram showing the gain of the closed loop frequency transfer function up to each output current command value (I α ref , I β ref ) to each current detection value (i α , i β The gain of the closed loop frequency transfer function up to the command angular frequency (ω re ), it is approximately 0 dB, that is, approximately 1. Here, "gain is approximately 1" means that the gain is 0.95 or more and 1.05 or less.

[0083] Figure 4(b) shows the command angular frequency (ω re When the output current command value (I α ref , I β ref ) to each current detection value (i α , i β ) is a phase diagram showing the phase of the closed loop frequency transfer function from α ref , I β ref ) to each current detection value (i α , i β The phase of the closed loop frequency transfer function up to the command angular frequency (ω re ) is approximately 0°. Here, "the phase is approximately 0°" means that the phase is between -5° and 5°.

[0084] Figure 5(a) shows the command angular frequency (ω re When the output current command value (I α ref , I β ref ) to each voltage command value (V α *, V β The gain diagram shows the gain of the frequency transfer function up to each output current command value (Iα ref , I β ref ) to each voltage command value (V α *, V β *) The gain of the frequency transfer function is re ), it is infinite. Therefore, the output current command value (I α ref , I β ref ) and current detection value (i α , i β ) to obtain the voltage command value (V α *, V β *) and the gain of the frequency transfer function up to the command angular frequency (ω re ) becomes infinity.

[0085] Figure 5(b) shows the command angular frequency (ω re When the output current command value (I α ref , I β ref ) to each voltage command value (V α *, V β *) is a phase diagram showing the phase of the frequency transfer function from α ref , I β ref ) to each voltage command value (V α *, V β *) The phase of the frequency transfer function is re ), it is approximately 0°. Therefore, the output current command value (I α ref , I β ref ) and current detection value (i α , i β ) to obtain the voltage command value (V α *, V β *) and the phase of the frequency transfer function up to the command angular frequency (ω re ) is approximately 0°.

[0086] Therefore, according to the first embodiment, the output current of the inverter (12) is calculated based on the output current command value (I α ref , I β ref ), the effect of fluctuations in the motor constants on the current control is small. This makes it possible to prevent the AC induction motor (3) from losing its good performance due to fluctuations in the motor constants.

[0087] In addition, the command angular frequency (ω re ), the output current command value (Iα ref , Iβ ref ) to the current detection value (i α , i β ) is approximately 1, it is possible to prevent the gain from deviating significantly from 1, which would prevent the AC induction motor (3) from achieving good performance.

[0088] In addition, the command angular frequency (ω re ), the output current command value (Iα ref , Iβ ref ) to the current detection value (i α , i β ) is approximately 0°, the phase of the output current of the inverter (12) is approximately 0°. ref , Iβ ref ) which would result in a significant phase shift from the normal phase, preventing the AC induction motor (3) from achieving good performance.

[0089] In addition, since the primary magnetic flux estimator (22) uses only the resistance of the primary winding of the AC induction motor (3) as the motor constant, the influence of the fluctuation of the motor constant on the primary magnetic flux estimation is small. Therefore, the calculated output current command value (Iα ref , Iβ ref ) is also less affected by fluctuations in the motor constants.

[0090] Furthermore, in order to calculate the first to third gains (f1, f2, f3) of the sine wave tracking controller (27), the inductance of the primary winding of the AC induction motor (3), the inductance of the secondary winding of the AC induction motor (3), the mutual inductance between the primary winding and the secondary winding, the resistance of the primary winding, and the resistance of the secondary winding are used as motor mechanical constants. Therefore, there is a possibility that a fluctuation in the motor mechanical constants will affect the gains (f1, f2, f3) and thus affect the responsiveness. However, the fluctuation in the motor mechanical constants will affect the output current of the inverter (12) by the output current command value (I α ref , I β ref ) is approximately equal to

[0091] In addition, the first and second gains (f1, f2) are set to the command angular frequency (ω re ), the current control operation can be performed over a wide range of command angular frequencies (ω re ) can be adapted.

[0092] Second Embodiment Fig. 6 is a diagram corresponding to Fig. 1 of the second embodiment. In the second embodiment, the current control unit (20) of the power conversion device (10) has a repetitive controller (40) instead of the sine wave tracking controller (27). As shown in Fig. 7, the repetitive controller (40) has an α-axis voltage command value calculation unit (40a) and a β-axis voltage command value calculation unit (40b).

[0093] The α-axis voltage command value calculation unit (40a) and the β-axis voltage command value calculation unit (40b) respectively calculate the corresponding output current command value (I α ref , I β ref ) and the corresponding current detection value (i α , i β ) and the corresponding voltage command value (V α *, V βThat is, the α-axis voltage command value calculation unit (40a) calculates the α-axis output current command value (I α ref ) and the α-axis current detection value (i α ) and the α-axis voltage command value (V α The β-axis voltage command value calculation unit (40b) calculates the β-axis output current command value (I β ref ) and the β-axis current detection value (i β ) and the β-axis voltage command value (V β *) is calculated.

[0094] Specifically, the α-axis voltage command value calculation section (40a) and the β-axis voltage command value calculation section (40b) each include a subtractor (41), an adder (42), an amplifier (43), and a delay element (44).

[0095] The subtractor (41) calculates the output current command value (I α ref , I β ref ) to the current detection value (i α , i β ) is subtracted from the output current command value (I α ref , I β ref ) and current detection value (i α , i β ) to output the difference.

[0096] The adder (42) calculates the output of the subtractor (41), i.e., the output current command value (I α ref , I β ref ) and current detection value (i α , i β ) and the voltage command value of the fundamental component one cycle before (V α *, V β *) and output the result.

[0097] The amplifier (43) receives the output of the adder (42), i.e., the output current command value (I α ref , I β ref ) and current detection value (i α , i β ) and the voltage command value of the fundamental component one cycle before (V α *, V β *) is multiplied by a specified current control gain to obtain the voltage command value (V α *, V β *) is calculated.

[0098] The delay element (44) delays the voltage command value (V α *, V β *) to the output current command value (I α ref , I β ref ) and output it with a delay of one period of the fundamental wave component. One period of the fundamental wave component is 2π times the command angular frequency (ω re ) can be calculated by dividing it by

[0099] The other configurations are the same as those in the first embodiment, so the same components are given the same reference numerals and detailed descriptions thereof are omitted.

[0100] In the second embodiment, each output current command value (I α ref , I β ref ) to each voltage command value (V α *, V β *) The gain of the frequency transfer function is re ), it becomes infinite, and each output current command value (I α ref , I β ref ) and current detection value (i α , i β ) to calculate the voltage command value (V α *, V β *) and the gain of the frequency transfer function up to the command angular frequency (ω re ) becomes infinite. Therefore, each output current command value (Iα ref , I β ref ) to each current detection value (i α , i β The gain of the closed loop frequency transfer function up to the command angular frequency (ω re ), it is approximately 0 dB, that is, approximately 1.

[0101] Third Embodiment Fig. 8 is a diagram corresponding to Fig. 1 of embodiment 3. In the present embodiment 3, the power conversion device (10) further includes a DC link voltage detection unit (17), a physical quantity detection unit (18), and an output voltage correction unit (50) in addition to the configuration of embodiment 2.

[0102] The DC link voltage detector (17) detects the DC link voltage (V dc In the first and second embodiments, the power conversion device (10) also detects the DC link voltage (V dc Although a DC link voltage detector (17) is provided to detect the DC link voltage, the illustration and description thereof are omitted in the first and second embodiments.

[0103] The physical quantity detection unit (18) detects a reactor voltage (V L ) is detected as a physical quantity. L ) is an electrical physical quantity that contains a component at the resonant frequency of an LC filter (LC).

[0104] In addition, the output voltage correction unit (50) corrects the reactor voltage (V L ) based on the reactor voltage (V L The voltage command value (V α *, V β The output voltage correction unit (50) has a polar coordinate conversion unit (51), a modulation factor calculation unit (52), a correction amount calculation unit (53), and a correction unit subtractor (54).

[0105] The polar coordinate conversion unit (51) converts the voltage command value (V α *, V β By performing polar coordinate conversion on the voltage command value (V*) indicating the voltage amplitude and the voltage phase (θv), a voltage command value (V*) indicating the voltage amplitude and a voltage phase (θv) are calculated.

[0106] The modulation factor calculation unit (52) calculates the voltage command value (V*) calculated by the polar coordinate conversion unit (51) and the DC link voltage (V dc ) and calculate the modulation factor (D**). The modulation factor (D**) is D**, the voltage command value (V*) is V*, and the DC link voltage (V dc ) is Vdc, the following equation (5) holds.

[0107] D**=V* * √2 / Vdc...Equation (5) The correction amount calculation unit (53) calculates the reactor voltage (V L ) based on the reactor voltage (V L ) is detected by the physical quantity detection unit (18). L ) is multiplied by a predetermined correction gain to calculate the correction amount (Dk).

[0108] The subtractor for correction (54) subtracts the correction amount (Dk) calculated by the correction amount calculation unit (53) from the modulation factor (D**) calculated by the modulation factor calculation unit (52), and outputs the subtraction result as the corrected modulation factor (D*).

[0109] The PWM modulation unit (16) generates a switching signal (S u , S v , S w ), thereby PWM-controlling the switching operation of the switching element (12S) of the inverter (12).

[0110] Therefore, according to the third embodiment, the output voltage correction unit (50) corrects the voltage command value (V α *, V β *) is corrected to obtain the reactor voltage (V L ) can be attenuated, and therefore the voltage fluctuation of the capacitance element (14) due to the resonance of the LC circuit (LC) can be suppressed.

[0111] First Modification of the Third Embodiment In the first modification of the third embodiment, the DC link voltage detector (17) constitutes a physical quantity detector (18). That is, the physical quantity detector (18) detects the DC link voltage (V dc ) is detected as a physical quantity.

[0112] In addition, the correction amount calculation unit (53) calculates the DC link voltage (V dc ) based on the DC link voltage (V dc In detail, the correction amount calculation unit (53) calculates a correction amount (Dk) so as to attenuate a component of a resonance frequency of the LC filter (LC) included in the physical quantity detection unit (18). dc ) to the DC link voltage (V dc ) and the ripple component in the DC link voltage (V dc ) to obtain the DC link voltage (V dc ) to extract a resonance frequency component of the LC filter (LC). Then, based on the extracted resonance frequency component, the correction amount calculation unit (53) calculates a correction amount (Dk) so as to suppress the resonance frequency component. Specifically, the correction amount calculation unit (53) calculates the correction amount (Dk) by multiplying the resonance frequency component by a predetermined correction gain.

[0113] In addition, the DC link voltage (V dcTo identify the ripple component contained in the three-phase AC power supply (2), first, a triangular wave is generated by multiplying the frequency of the zero-cross signal of the line voltage of the two-phase AC voltage among the AC voltages output by the three-phase AC power supply (2) by six. Then, by referring to a predetermined lookup table showing multiple phases and the ripple waveforms corresponding to each phase, the ripple waveform corresponding to the phase of the triangular wave is identified as the ripple component.

[0114] In addition, the DC link voltage (V dc The DC component of the three-phase AC power supply (2) may be determined by a low-pass filter, or the average value of one period of the AC voltage output by the three-phase AC power supply (2) may be determined as the DC component.

[0115] The other configurations are the same as those in the third embodiment, so detailed description thereof will be omitted.

[0116] Therefore, according to the first modification of the third embodiment, the output voltage correction unit (50) calculates the voltage command value (V α *, V β *) correction, the DC link voltage (V dc ) can be attenuated.

[0117] Second Modification of the Third Embodiment In the second modification of the third embodiment, the physical quantity detection unit (18) detects, as the physical quantity, a reactor current that is a current flowing through the reactor (13).

[0118] Further, the correction amount calculation unit (53) calculates a correction amount (Dk) based on the reactor current detected by the physical quantity detection unit (18) so as to attenuate a component of the resonant frequency of the LC filter (LC) contained in the reactor current. More specifically, the correction amount calculation unit (53) extracts a component of the resonant frequency of the LC filter (LC) contained in the reactor current by subtracting a ripple component and a DC component of the reactor current from the reactor current detected by the physical quantity detection unit (18). Then, based on the extracted component of the resonant frequency, the correction amount calculation unit (53) calculates a correction amount (Dk) so as to suppress the component of the resonant frequency. More specifically, the correction amount calculation unit (53) calculates the correction amount (Dk) by multiplying the component of the resonant frequency by a predetermined correction gain.

[0119] The ripple component included in the reactor current and the DC component of the reactor current are calculated based on the DC link voltage (V dc ) can be determined by a method similar to that used to determine the ripple and DC components.

[0120] The other configurations are the same as those in the third embodiment, so detailed description thereof will be omitted.

[0121] Therefore, according to the second modification of the third embodiment, the output voltage correction unit (50) α *, V β *) The compensation can attenuate the resonant frequency component contained in the reactor current.

[0122] Third Modification of Third Embodiment In the third modification of the third embodiment, the physical quantity detection section (18) detects, as the physical quantity, a capacitance current that is a current flowing through the capacitance element (14).

[0123] Further, the correction amount calculation unit (53) calculates a correction amount (Dk) based on the capacitance current detected by the physical quantity detection unit (18) so as to attenuate the component of the resonance frequency of the LC filter (LC) contained in the capacitance current. More specifically, the correction amount calculation unit (53) extracts the component of the resonance frequency of the LC filter (LC) contained in the capacitance current by subtracting the ripple component contained in the capacitance current from the capacitance current detected by the physical quantity detection unit (18). Then, the correction amount calculation unit (53) calculates a correction amount (Dk) based on the extracted component of the resonance frequency so as to suppress the component of the resonance frequency. More specifically, the correction amount calculation unit (53) calculates the correction amount (Dk) by multiplying the component of the resonance frequency by a predetermined correction gain.

[0124] In the first modification of the third embodiment, the ripple component included in the capacitance current is dc ) can be identified by a method similar to that used to identify the ripple component of

[0125] The other configurations are the same as those in the third embodiment, so detailed description thereof will be omitted.

[0126] Therefore, according to the third modification of the third embodiment, the output voltage correction unit (50) α *, V β *) correction makes it possible to attenuate the resonant frequency component contained in the capacitance current.

[0127] Fourth Modification of the Third Embodiment In the third modification of the third embodiment, the physical quantity detection unit (18) detects the electrical energy stored in the capacitance element (14) as a physical quantity. The physical quantity detection unit (18) detects the DC link voltage (V dc ) and a first multiplier (not shown) that calculates the square of the DC link voltage (V dc and a second multiplier (not shown) that calculates a value obtained by multiplying the square of (a) by 1 / 2.

[0128] Further, the correction amount calculation unit (53) calculates a correction amount (Dk) based on the electrical energy detected by the physical quantity detection unit (18) so as to attenuate a component of the resonant frequency of the LC filter (LC) contained in the electrical energy. More specifically, the correction amount calculation unit (53) extracts a component of the resonant frequency of the LC filter (LC) contained in the electrical energy by subtracting a ripple component and a DC component of the electrical energy from the electrical energy detected by the physical quantity detection unit (18). Then, based on the extracted component of the resonant frequency, the correction amount calculation unit (53) calculates a correction amount (Dk) so as to suppress the component of the resonant frequency. More specifically, the correction amount calculation unit (53) calculates the correction amount (Dk) by multiplying the component of the resonant frequency by a predetermined correction gain.

[0129] The ripple component contained in the electrical energy and the DC component of the electrical energy are calculated as the DC link voltage (V dc ) can be determined by a method similar to that used to determine the ripple and DC components.

[0130] The other configurations are the same as those in the third embodiment, so detailed description thereof will be omitted.

[0131] Therefore, according to the fourth modification of the third embodiment, the output voltage correction unit (50) α *, V β *) makes it possible to attenuate the resonant frequency component contained in the electrical energy stored in the capacitance element (14).

[0132] Fifth Modification of the Third Embodiment In the fifth modification of the third embodiment, the physical quantity detection unit (18) detects, as a physical quantity, an input current from the three-phase AC power supply (2), i.e., an input current to the converter (11). The physical quantity detection unit (18) detects the input current of any one phase of the three-phase input current from the three-phase AC power supply (2).

[0133] Further, the correction amount calculation unit (53) calculates a correction amount (Dk) based on the input current detected by the physical quantity detection unit (18) so as to attenuate a component of the resonance frequency of the LC filter (LC) contained in the input current. More specifically, the correction amount calculation unit (53) extracts a component of the resonance frequency of the LC filter (LC) contained in the input current by subtracting a ripple component and a DC component of the input current from the input current detected by the physical quantity detection unit (18). Then, based on the extracted component of the resonance frequency, the correction amount calculation unit (53) calculates a correction amount (Dk) so as to suppress the component of the resonance frequency. More specifically, the correction amount calculation unit (53) calculates the correction amount (Dk) by multiplying the component of the resonance frequency by a predetermined correction gain.

[0134] The ripple component included in the input current from the three-phase AC power source (2) and the DC component of the input current are calculated based on the DC link voltage (V dc ) can be determined by a method similar to that used to determine the ripple and DC components.

[0135] The other configurations are the same as those in the third embodiment, so detailed description thereof will be omitted.

[0136] Therefore, according to the fifth modification of the third embodiment, the output voltage correction unit (50) α *, V β *) makes it possible to attenuate the resonant frequency component contained in the input current from the three-phase AC power supply (2).

[0137] Fourth Embodiment Fig. 9 is a diagram corresponding to Fig. 1 of embodiment 4. In embodiment 4, a current control unit (20) of a power conversion device (10) further includes an αβ-MT conversion unit (61), a slip frequency estimating unit (62), a speed estimating unit (63), an adder (64), an estimated speed calculating unit (65), a frequency subtractor (66), and a speed controller (67) in addition to the configuration of embodiment 1.

[0138] The αβ-MT conversion unit (61) converts the α-axis current detection value (i α ) and β-axis current detection value (i β ) to the magnetic flux current detection value (i 1M ) and torque current detection value (i 1T ) and output.

[0139] The slip frequency estimation unit (62) estimates the magnetic flux current detection value (i 1M ) and torque current detection value (i 1T ) and the primary flux (Ψ^) estimated by the primary flux estimator (22), the slip frequency (ω s ) is estimated. s ) is estimated by the following equation (6). In equation (6), ωs is the slip frequency (ω s ), i 1T is the torque current detection value (i 1T ), Ψ1^ is the primary flux (Ψ^) estimated by the primary flux estimator (22), σ is the degree of coupling of the AC induction motor (3), i 1M is the magnetic flux current detection value (i 1M ) is shown.

[0140] ωs=R2 L1 i 1T / (L2·|Ψ1^|-σ·L1·L2·i 1M ) ···(6) The speed estimation unit (63) estimates the torque current detection value (i 1T ) and the primary magnetic flux (Ψ^) estimated by the primary magnetic flux estimator (22), the primary frequency (ω1) is estimated by the following equation (7). In equation (7), ω1 is the primary frequency (ω1), v 1T is the T-axis voltage command value, i 1T is the torque current detection value (i 1T ), and Ψ1^ denotes the primary flux (Ψ^) estimated by the primary flux estimator (22).

[0141] ω1=(v 1T -R1·i 1T) / |Ψ1^| (7) The adder (64) calculates the slip frequency (ω s ) and the primary frequency (ω1) estimated by the speed estimation unit (63).

[0142] The estimated speed calculation unit (65) calculates an estimated speed (ωm^) by dividing the addition result output by the adder (64) by the number of pole pairs (Pn) of the AC induction motor (3).

[0143] A frequency subtractor (66) subtracts the estimated speed (ωm^) from the command speed (ωm*) and outputs the subtraction result.

[0144] The speed controller (67) generates and outputs a torque command value (T*) based on the subtraction result output by the frequency subtractor (66) by, for example, proportional-integral (PI) control.

[0145] The other configurations are the same as those in the first embodiment, so the same components are given the same reference numerals and detailed descriptions thereof are omitted.

[0146] In this manner, the current control unit (20) controls the detection value (i u , i v , i w ) based on the slip frequency (ω s Then, the current control unit (20) estimates the estimated slip frequency (ω s ) based on the output current command value (I α ref , I β ref ) is calculated.

[0147] Therefore, according to the fourth embodiment, the current control unit (20) calculates the estimated slip frequency (ω s ) can be reflected in the output current of the inverter (12).

[0148] Other Embodiments In the first to fourth embodiments and the modified examples, the current control unit (20) controls the output current command value (I α ref , I β ref ) and the detection value (i u , i v , i w ) based on the current detection value (i α , i β ) and the voltage command value (V α *, V β The current control operation to calculate the α-axis current and the β-axis current corresponding to the U-phase current, the V-phase current, and the W-phase current was performed for each of them. However, the current control unit (20) may perform the current control operation for each of the U-phase current, the V-phase current, and the W-phase current. In other words, the current control unit (20) calculates the U-phase current command value, the V-phase current command value, and the W-phase current detection value (i u , i v , i w ) based on which the U-phase voltage command value, the V-phase voltage command value, and the W-phase voltage command value may be calculated. [Industrial Applicability]

[0149] INDUSTRIAL APPLICABILITY As described above, the present disclosure is useful for a power conversion device that drives an AC induction motor. [Explanation of symbols]

[0150] 2 Three-phase AC power supply 3 AC induction motor 10 Power conversion device 11 Converter (rectifier) 12 Inverter 12a First input terminal 12b Second input terminal 13 Reactor 14 Capacitance elements 15 Current detection section 16 PWM modulation section (switching operation control section) 18 Physical quantity detection section 20 Current control section 50 Output voltage correction section LC LC Filter V α * α-axis voltage command value V β * β-axis voltage command value i u , i v , i w Detection Value I α ref α-axis output current command value I β ref β-axis output current command value i α α-axis current detection value i β β-axis current detection value ω re Command Angular Frequency ω s Slip Frequency FB Feedback value Z 1α First integral value Z 2α Second integral value f1 first gain f2 Second gain f3 3rd gain

Claims

1. an inverter (12) that drives an AC induction motor (3) by converting DC to three-phase AC through a switching operation and outputting the converted AC; Voltage command value (V α *, V β *) calculating a current control unit (20); The voltage command value (V α *, V β and a switching operation control unit (16) that controls a switching operation of the inverter (12) based on a switching frequency of the inverter (12) and a switching operation control unit (16) that controls a switching operation of the inverter (12) based on a switching frequency of the inverter (12), a current detection unit (15) that detects an output current of the inverter (12), The current control unit (20) controls an output current command value (I α ref , I β ref ) and the detection value (i ｕ , i ｖ , i ｗ ) based on the current detection value (i α , i β ) based on the voltage command value (V α *, V β a current control operation for calculating a U-phase current flowing through a U-phase coil of the AC induction motor (3), a V-phase current flowing through a V-phase coil of the AC induction motor (3), and a W-phase current flowing through a W-phase coil of the AC induction motor (3), The output current command value (I α ref , I β ref ) is set to the command angular frequency (ω ｒｅ ), the output current command value (I α ref , I β ref ) to the current detection value (i α , i β The gain of the closed loop frequency transfer function up to the command angular frequency (ω ｒｅ) is approximately 1, and the output current command value (I α ref , I β ref ) to the current detection value (i α , i β ) is the phase of the closed loop frequency transfer function up to the command angular frequency (ω ｒｅ ) is approximately 0°.

2. 2. The power conversion device according to claim 1, The current control operation is performed by adjusting the output current command value (I α ref , I β ref ) and the current detection value (i α , i β ) based on the difference between the voltage command value (V α *, V β *) is calculated. The voltage command value (V α *, V β The gain of the frequency transfer function up to the command angular frequency (ω ｒｅ ) is infinite, and the voltage command value (V α *, V β *) is the phase of the frequency transfer function up to the command angular frequency (ω ｒｅ ) is approximately 0°.

3. 3. The power conversion device according to claim 2, The current control operation calculates the feedback value (FB) based on a difference between the difference and a feedback value (FB), and calculates the voltage command value (V α *, V β *) is calculated. The feedback value (FB) is the command angular frequency (ω ｒｅ ) an amplified value of the component.

4. The power conversion device according to claim 3, The current control operation is performed by integrating the difference to obtain a first integral value (Z 1α ) to the first gain (f 1 ) to calculate a first proportional value, and the first integral value (Z 1α ) is integrated to obtain the second integral value (Z ２α ) to the second gain (f 2 ) to calculate a second proportional value, and the voltage command value (V α *, V β *) is calculated. The command angular frequency (ω ｒｅ The first and second gains (f 1 , f 2 The change in the command angular frequency (ω ｒｅ ) is proportional to the square of the change in current.

5. The power conversion device according to claim 4, The command angular frequency (ω ｒｅ ) squared change amount is D, the output current command value (I α ref , I β ref ) to the current detection value (i α , i β ) is the crossover angular frequency of the gain of the open loop frequency transfer function up to the command angular frequency (ω ｒｅ ) when the first gain (f 1 The change in L1 is D (L1 L2 - M ２ ) / L2, and the command angular frequency (ω ｒｅ ) when the second gain (f 2 The change in the ２ ) / L2.

6. The power conversion device according to claim 4 or 5, The current control operation is performed by detecting the current detection value (i α , i β ) to the third gain (f ３ ) to calculate a third proportional value, and subtracting the first proportional value, the second proportional value, and the third proportional value from 0 to calculate the voltage command value (V α *, V β *) is calculated. The output current command value (I α ref , I β ref ) to the current detection value (i α , i β The crossover angular frequency of the gain of the open loop frequency transfer function up to １ , the inductance of the secondary winding of the AC induction motor (3) is L ２ , the mutual inductance between the primary winding and the secondary winding is M, and the resistance of the primary winding is R １ , the resistance of the secondary winding is R ２ and the third gain (f 3 ) to f ３ When the above equation (A) is satisfied, the power conversion device is characterized in that: [006]

7. 3. The power conversion device according to claim 1, The current control operation is performed by adjusting the output current command value (I α ref , I β ref ) and the current detection value (i α , i β ) and the voltage command value (V α *, V β *) is multiplied by a predetermined current control gain to obtain the voltage command value (V α *, V β A power conversion device characterized in that it calculates the

8. In the power conversion device according to any one of claims 1 to 7, a rectification unit (11) that rectifies an AC voltage of an AC power source (2) into the DC voltage; a capacitance element (14) connected between the input ends (12a, 12b) of the inverter (12), The power conversion device, wherein the capacitance element (14) constitutes an LC filter (LC) together with an inductance component between the AC power supply (2) and the capacitance element (14).

9. 9. The power conversion device according to claim 8, a physical quantity detection unit (18) that detects an electrical physical quantity including a component at a resonant frequency of the LC filter (LC); The voltage command value (V α *, V β and an output voltage correction unit (50) that corrects the output voltage difference (Vp)*.

10. 10. The power converter according to claim 9, a reactor (13) connected between the AC power supply (2) and the capacitance element (14); The power conversion device, wherein the physical quantity is a voltage of the reactor (13).

11. 10. The power converter according to claim 9, The power conversion device, wherein the physical quantity is a voltage of the capacitance element (14).

12. 10. The power converter according to claim 9, a reactor (13) connected between the AC power supply (2) and the capacitance element (14); The power conversion device, wherein the physical quantity is a current flowing through the reactor (13).

13. 10. The power converter according to claim 9, The power conversion device, wherein the physical quantity is a current flowing through the capacitance element (14).

14. 10. The power converter according to claim 9, The power conversion device, wherein the physical quantity is electrical energy stored in the capacitance element (14).

15. 10. The power converter according to claim 9, The power conversion device, wherein the physical quantity is an input current from the AC power source (2).

16. In the power conversion device according to any one of claims 1 to 15, The current control section (20) calculates a slip frequency (ω ｓ ) and estimate the slip frequency (ω ｓ ) based on the output current command value (I α ref , I β ref ) is calculated.

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