Motor Drive Unit

Abstract

To provide a motor drive device capable of forming a desired current waveform by a simple control system while suppressing current distortion and suppressing motor noise as much as possible.SOLUTION: A motor drive device 3 has one or more n converters 5 and drives a motor 4 with the outputs of the converters 5. A control unit 11 outputs control information according to an input command value. A pulse generation unit 212 generates pulses to be output to the converters 5 so as to operate the n converters 5 on the basis of the control information output from the controller 11. A back electromotive voltage correction unit 12c corrects the pulse width according to the back electromotive voltage generated in the motor 4 when the motor 4 is driven.SELECTED DRAWING: Figure 33

Description

[Technical field]

[0001] The present invention relates to a motor drive device. [Background technology]

[0002] Power generating devices for hybrid vehicles have been proposed in the past (see, for example, Patent Documents 1 and 2). The power generating device described in Patent Document 1 has a multi-phase converter connected to each phase of a permanent magnet synchronous motor, and the multi-phase converters connected to each phase are connected in parallel. The multi-phase converter, which has a multiplexed converter configuration, outputs a sinusoidal voltage by driving each phase while changing the phase, thereby reducing the output current ripple.

[0003] According to the power generating device described in Patent Document 2, the feedback control unit calculates a basic value of the output current to the capacitor and the motor side based on the difference between an offset command voltage for the motor and the actual voltage of the capacitor. The feedforward correction unit calculates the output command value of the current to the capacitor and the motor side by performing feedforward correction with the phase current of the motor. The pulse width calculation unit calculates the ON operation time of the chopper control based on this. This makes it possible to convert the capacitor voltage to the AC voltage required for the motor when desired.

[0004] According to the technology described in Patent Document 2, the control system becomes complicated because it requires voltage / current feedback and feedforward control. In addition, a large number of sensors are required, which leads to an increase in the size of the system and an increase in costs. Furthermore, the impact becomes more pronounced when the number of multiplexing is increased. When the control system becomes complicated, the responsiveness deteriorates, making it difficult to increase the operating frequency. Furthermore, if the back electromotive force is not taken into consideration and the pulse width is not corrected, distortion occurs in the drive current, which causes motor noise. [Prior art documents] [Patent documents]

[0005] [Patent Document 1] JP 2009-219299 A [Patent Document 2] JP 2009-194964 A Summary of the Invention [Problem to be solved by the invention]

[0006] An object of the present invention is to provide a motor drive device that is capable of shaping a current waveform as desired using a simple control system while suppressing current distortion and minimizing motor noise. [Means for solving the problem]

[0007] The invention of claim 1 is directed to a motor drive device that includes n converters (one or more) and drives a motor using the outputs of the converters. A control unit outputs control information according to an input command value. A pulse generation unit generates pulses to be output to the n converters to operate them based on the control information output by the control unit. A back electromotive force correction unit corrects the pulse width according to the back electromotive force generated in the motor when the motor is driven. The back electromotive force correction unit corrects the pulse width based on the amplitude and phase information of the back electromotive force.

[0008] When the back electromotive force correction unit corrects the back electromotive force by taking the back electromotive force into account, the motor's back electromotive force can be incorporated in the calculation of the pulse width in a feedforward manner, and the current waveform can be shaped into an ideal sine wave. This makes it possible to drive a motor with high efficiency, low noise, and low EMC using a simple control system with only current feedback control. Note that the "back electromotive force" referred to in this specification is sometimes called back electromotive force. [Brief description of the drawings]

[0009] [Figure 1] Overall electrical configuration diagram in the first embodiment [Diagram 2] Electrical configuration diagram for one phase in the first embodiment [Diagram 3]Control configuration diagram of a converter in the first embodiment [Figure 4] Driving voltage and driving current waveforms of the converter in the case of four-multiplexing in the first embodiment [Diagram 5] 1 is a flowchart outlining a multiple pulse generation process according to the first embodiment; [Figure 6] FIG. 1 is an explanatory diagram of a method for generating multiple pulses in the first embodiment; [Figure 7] Part 2 of a flowchart outlining a multiple pulse generation process in the first embodiment. [Figure 8] FIG. 1 is a diagram illustrating a relationship between an inductor current and an output current in the first embodiment. [Figure 9] FIG. 2 is a second explanatory diagram showing the relationship between the inductor current and the output current in the first embodiment. [Figure 10] FIG. 1 is an explanatory diagram showing an example of setting the number of multiple operations for a current command value in the first embodiment. [Figure 11] FIG. 2 is an explanatory diagram showing an example of setting the number of multiple operations for a current command value in the first embodiment. [Figure 12] FIG. 11 is an explanatory diagram showing the relationship between the rotation speed command value, the output current, and the number of multiplexed operations in the first embodiment. [Figure 13] 11 is a flowchart for explaining an outline of a pulse generation process in the second embodiment. [Figure 14] FIG. 13 is a diagram for explaining a problem that occurs when switching the number of multiplexed operations in the third embodiment. [Figure 15] FIG. 13 is an explanatory diagram illustrating an outline of a method for generating an interpolation pulse in a third embodiment. [Figure 16] Overall electrical configuration diagram of a modification related to the first to third embodiments [Figure 17] FIG. 13 is a diagram for explaining a schematic switching logic of the number of multiplexed operations with respect to a current command value in a modification related to the first to third embodiments; [Figure 18] FIG. 11 is a diagram showing a comparison result before and after application of a modification example related to the first to third embodiments. [Figure 19]FIG. 13 is an explanatory diagram of a defect of a comparative example technique with respect to a modification example related to the first to third embodiments. [Figure 20] FIG. 10 is a diagram for explaining the effects of modifications related to the first to third embodiments. [Figure 21] Overall electrical configuration diagram in the fourth embodiment [Figure 22] Example of generating multiple interpolated pulses in the fourth embodiment [Diagram 23] FIG. 13 is a diagram for explaining a method for generating an interpolated multiple pulse in the fourth embodiment. [Figure 24] 13 is a flowchart for explaining an outline of a method for generating an interpolated multiple pulse in the fourth embodiment. [Diagram 25] 13 is a timing chart for explaining a method for generating a multiple pulse in the fifth embodiment; [Figure 26] Generation example of interpolated multiple pulses in the fifth embodiment [Figure 27] Overall electrical configuration diagram of a modification related to the fourth and fifth embodiments [Figure 28] FIG. 13 is a diagram showing a comparison result between the fourth and fifth embodiments and related modified examples. [Figure 29] FIG. 13 is a diagram showing a comparison result before and after application of a modification example related to the fourth and fifth embodiments. [Diagram 30] FIG. 13 is an explanatory diagram of a defect of a comparative example technique with respect to a modification example related to the fourth and fifth embodiments. [Diagram 31] FIG. 10 is a diagram for explaining the effects of modifications related to the fourth and fifth embodiments. [Diagram 32] Example of current waveform affected by harmonic current distortion in conventional technology [Diagram 33] Overall electrical configuration diagram in the sixth embodiment [Diagram 34] FIG. 1 is a diagram for explaining an electrical configuration of a pulse generating unit according to a sixth embodiment. [Diagram 35] FIG. 2 is a second diagram for explaining an electrical configuration of a pulse generating unit in the sixth embodiment. [Diagram 36] Example of current waveform when the pulse width command value is corrected according to the back electromotive force in the sixth embodiment [Figure 37] Overall electrical configuration diagram of a modification related to the sixth and seventh embodiments. [Figure 38] FIG. 13 is a diagram illustrating the process of a back electromotive force correction unit in a modification related to the sixth and seventh embodiments. [Figure 39] FIG. 13 is a diagram showing a comparison result before and after application of a modification related to the sixth and seventh embodiments. [Diagram 40] Examples of changes in current waveforms before and after application of modifications related to the sixth and seventh embodiments [Diagram 41] FIG. 13 is an explanatory diagram of a voltage utilization rate improvement control unit in a modification related to the sixth and seventh embodiments. [Diagram 42] FIG. 13 is an explanatory diagram of a comparative example technique and upper and lower limit clamping action in a modification related to the sixth and seventh embodiments. [Diagram 43] Overall electrical configuration diagram in the eighth embodiment [Diagram 44] Overall electrical configuration diagram in the ninth embodiment DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0010] Hereinafter, several embodiments will be described with reference to the drawings. In the embodiments, the same or similar components are denoted by the same or similar reference numerals, and the description thereof may be omitted.

[0011] (First embodiment) The first embodiment will be described with reference to Fig. 1 to Fig. 12. As shown in Fig. 1, a three-phase inverter 3 serving as a power converter is connected to a battery 2. The battery 2 is a storage battery such as a nickel-metal hydride storage battery or a lithium storage battery. A motor 4 is connected to the three-phase inverter 3. The motor 4 is a power generating device that drives the wheels of a hybrid vehicle, for example, and is exemplified by a permanent magnet synchronous motor (PMSM).

[0012] The three-phase inverter 3 is provided with multi-phase converters 6u, 6v, 6w for multiple phases, namely U phase, V phase, and W phase. The three-phase inverter 3 is composed of multi-phase converters 6u, 6v, 6w, each of which is made up of n (2 or more) multiple parallel-connected converters 5u, 5v, 5w, which serve as basic units.

[0013] In FIG. 1, the converters 5u, 5v, and 5w, which are basic units, are indicated by the reference numerals suffixed with "1 to n." In the following description, each of the converters 5u1, 5u2, 5u3...5un, 5v1, 5v2, 5v3...5vn, 5w1, 5w2, 5w3...5wn may be abbreviated to simply "converter 5." In addition, the current flowing through the inductor (hereinafter referred to as the inductor current I L The converters 5u1, 5v1, and 5w1 that are the detection targets of the converters 5u1, 5v1, and 5w1 may be referred to as “master phase converters 5.” An example of the configuration of the converter 5 is shown in FIG.

[0014] For example, the U-phase multiphase converter 6u shares the total current flowing through the n converters 5 at 1 / n, and drives the converters 5u1...5un with a phase difference of T / n between them to equalize the drive current of each converter. The output currents of the converters 5u1...5un are combined to cancel out current ripples, thereby outputting a desired current waveform, a sine wave current in this case. The same applies to the V-phase and W-phase multiphase converters 6v and 6w.

[0015] As shown in FIG. 3, each of the converters 5 includes an upper arm switch SW1, a lower arm switch SW2, an inductor L, and a capacitor C in the illustrated form, and is configured as a step-down non-inverting buck converter that converts the voltage of the battery 2 as desired.

[0016] The upper arm switch SW1 and the lower arm switch SW2 are configured as power switches such as N-channel power MOSFETs, and freewheeling diodes D1 and D2 are connected between the drain and source of each switch to commutate the load current. In the following description, both or either of the upper arm switch SW1 and the lower arm switch SW2 may be referred to as a "power switch."

[0017] In the control system 1 according to this embodiment, a control device 10 is connected to a three-phase inverter 3, and control is mainly performed by the control device 10 shown in Fig. 1. The control device 10 is configured by a computer equipped with multiple cores, volatile and non-volatile memory 37, and the like, and functionally includes a control unit 11 and a pulse generating unit 12. The memory 37 is used as a storage unit that stores various data as a non-transient substantial storage medium. A voltage sensor 13 is installed in the battery 2, and the detected voltage of the voltage sensor 13 is input to the control unit 11.

[0018] A current sensor 14 is provided in the current path of the inductor L that constitutes the converters 5u1, 5v1, and 5w1 that are the master phases of the UVW phases. The control unit 11 detects the inductor current I L A voltage sensor 15 is provided to detect the output voltage Vout of the capacitor C, and the voltage detected by the voltage sensor 15 is input to the control unit 11.

[0019] The control unit 11 receives the current command value Io calculated according to the required torque as a command value, and outputs control information according to the current command value Io to the pulse generating unit 12. Note that the update cycle of the current command value Io is set to a time that is sufficiently shorter than the cycle of the AC frequency, and the current command value Io, which changes with the AC frequency, can be set finely.

[0020] A current sensor 16 is provided to detect phase currents Iu, Iv, Iw to be input to the motor 4, and the current detected by the current sensor 16 is input to the control unit 11. The control unit 11 also receives the rotor angle θ from a rotational position sensor 4a such as a resolver installed in the motor 4 and calculates information on the angular velocity ω. The control unit 11 outputs this information to the pulse generating unit 12 as feedback control information.

[0021] The pulse generating unit 12 generates multiple pulses (multiphase pulses) based on control information input from the control unit 11 so as to cause the 3×n converters 5 of the three-phase inverter 3 to perform multiple operations.

[0022] 2, the pulse generating unit 12 has functional configurations including a gate driving unit 21, a zero current detecting unit 22, and a pulse calculating unit 23. As shown in Fig. 3, the pulse calculating unit 23 has a counter 24 including a pulse width counter 24a, a start phase counter 24b, and an end phase counter 24c.

[0023] The pulse calculation unit 23 calculates each parameter of the on time Ton and off time Toff of the upper arm switch SW1 and the lower arm switch SW2 when each converter 5 of the multi-phase converters 6u, 6v, 6w of each phase operates in the current boundary mode, the period T, and the phase difference Td between multiple pulses to be input to multiple converters 5 of the same phase (e.g., U phase).

[0024] The current boundary mode is a mode in which the upper arm switch SW1 and the lower arm switch SW2 are switched on and off when the inductor current IL is detected to be zero. The parameters of the on time Ton, the off time Toff, the period T, and the phase difference Td are set to be the same between multiple pulses input to each converter 5 of the same-phase multiphase converter (e.g. 6u).

[0025] The gate driver 21 drives the three-phase inverter 3 based on the calculation result of the pulse calculator 23. The gate driver 21 drives the upper arm switch SW1 or the lower arm switch SW2 to turn on and off based on the on-time Ton, the period T, and the phase difference Td calculated by the pulse calculator 23.

[0026] In this embodiment, for example, when the current command value Io of a certain phase, for example the U phase, exceeds 0, the gate driver 21 drives the upper arm switch SW1 on and off while keeping the lower arm switch SW2 of each converter 5 constituting the multi-phase converter 6u off. When the upper arm switch SW1 is turned on, an inductor current I L The capacitor C is charged while gradually increasing.

[0027] After that, when the on-time Ton has elapsed, the gate driver 21 drives the upper arm switch SW1 to the off-state. Even when the upper arm switch SW1 is turned off, the inductor current I L continues to flow.

[0028] Inductor current I L gradually decreases, but the zero current detection unit 22 detects the inductor current I L The gate driver 21 detects the timing when the inductor current I L When the current command value Io becomes zero, the upper arm switch SW1 is turned on again. This operation is repeated as long as the current command value Io exceeds 0. L When zero is detected, the upper arm switch SW1 is turned on and the circuit operates in a current boundary mode.

[0029] Conversely, when the current command value Io is less than 0, the gate driver 21 drives the lower arm switch SW2 on and off while keeping the upper arm switch SW1 off. When the lower arm switch SW2 is turned on, the inductor current I LThe inductor current I is gradually decreased while discharging from the capacitor C. After that, when the on-time Ton has elapsed, the gate driver 21 drives the lower arm switch SW2 to the off-state. Even if the lower arm switch SW2 is turned off, the inductor current I L continues to flow.

[0030] Inductor current I L gradually increases, but the zero current detection unit 22 detects the inductor current I L The gate driver 21 detects the timing when the inductor current I L When the current command value Io becomes zero, the lower arm switch SW2 is turned on again. This operation is repeated as long as the current command value Io is below 0. Each converter 5 detects the inductor current I L When zero is detected, the lower arm switch SW2 is turned on in a current boundary mode.

[0031] FIG. 4 shows an example of the waveform when the number of multiplexed operations is n=4, the period is T, and the phase difference Td is T / n. L When the inductor current I L is energized in a triangular waveform.

[0032] The control device 10 gradually changes the current command value Io of each of the UVW phases in a sinusoidal shape based on the above-mentioned feedback information, and outputs control information to the pulse generating unit 12. The pulse generating unit 12 changes each parameter of the multiple pulse (on time Ton, off time Toff, cycle T, phase difference Td) as shown in FIG.

[0033] Each converter 5 supplies phase currents Iu, Iv, Iw to the motor 4 with the same phase difference Td=T / n, so that the output currents of each converter 5 can be superimposed by the multiplexing number n and supplied to each of the UVW phases of the motor 4. This makes it possible to control each of the phase currents Iu, Iv, Iw to the desired current command value Io, which here changes sinusoidally.

[0034] The maximum output of each phase current Iu, Iv, Iw is determined so as to be smaller than the inductor saturation current and to satisfy the heat generation requirement of the three-phase inverter 3, and the number n of multiplexed operations is preferably determined based on the upper limit that satisfies the maximum output. In addition, the frequency corresponding to the period T of the multiplexed pulses is preferably set higher than the audible frequency.

[0035] It is desirable to make the number of multiple operations n as large as possible because the larger it is, the greater the current ripple cancellation effect becomes. Note that if the number of multiple operations n is made too large, it will cause the control by the control device 10 to become complicated, so it is advisable to determine the number of multiple operations n according to the processing capacity based on the resources of the control device 10.

[0036] The processing operations of the multi-phase converters 6u, 6v, and 6w for each phase will be described in detail below using flowcharts. Since the operations of the UVW phases are generally the same, the processing operations of the multi-phase converter 6u for the U phase will be described, and the processing operations of the multi-phase converters 6v and 6w for the V and W phases will be omitted.

[0037] As shown in FIG. 5, the processing steps of the first multiplex converter 5u1 are shown. In FIG. 5, the zero current detection unit 22 detects the inductor current I L becomes zero, the gate driver 21 turns on the upper arm switch SW1 or the lower arm switch SW2 in S3 or S4. At this time, the pulse calculator 23 determines whether to turn on the upper arm switch SW1 or the lower arm switch SW2 in S2 based on the current command value Io.

[0038] Current command value I o If exceeds zero, the gate driver 21 turns on the upper arm switch SW1 in S3 and the current command value I o If is less than zero, the gate driver 21 turns on the lower switch SW2 at S4.

[0039] When driving the converter 5u1, the pulse calculation unit 23 calculates the on-time Ton for which the power switch SW1 or SW2 of each converter 5 is kept on, and the phase difference Td between the multiple pulses applied to each converter 5.

[0040] If the current command value of the phase current Iu is Io and the number of multiple converters 5 is n, the average current I of the converter 5 is Io / n. In this control system 1, the inductor current I L Since the inductor current I L Peak current I Lp is twice the average current I = 2I.

[0041] When the current command value Io>0, the pulse calculation unit 23 calculates the on time Ton and off time Toff of the upper arm switch SW1, as well as the period T and phase difference Td, based on the following equations (1-1) to (1-4), respectively.

number

[0042] When the current command value Io<0, the pulse calculation unit 23 calculates the on time Ton and off time Toff, as well as the period T and phase difference Td of the lower arm switch SW2 based on the following equations (2-1) to (2-4), respectively.

number

[0043] The pulse generating unit 12 generates pulses for driving the upper arm switch SW1 and the lower arm switch SW2, and the gate driving unit 21 drives the upper arm switch SW1 or the lower arm switch SW2 to ON in S3 and S4.

[0044] 6 shows an example of pulse generation for starting the first to n-th multiplexed converters 5. The pulse width counter 24a detects the inductor current I L On the condition that zero is detected, the counter value is set to a predetermined value and counted down to measure the on-time Ton.

[0045] 5, the pulse calculation unit 23 determines YES in S5 of Fig. 5 if the on-time Ton has elapsed, and turns off the on-power switch SW1 or SW2 in S6. This allows the power switch SW1 or SW2 to continue being on for the on-time Ton.

[0046] Furthermore, the pulse calculation unit 23 of the pulse generation unit 12 operates the start phase counter 24b simultaneously with the pulse width counter 24a. The start phase counter 24b is a counter for measuring the start timing of the pulses to be input to the second to nth multiplex converters 5u2...5un. The start phase counter 24b measures the time corresponding to the phase difference Td calculated by the pulse calculation unit 23 by setting the counter value to a predetermined value and counting down. The end phase counter 24c is a counter for measuring the end timing of the pulses to be input to the first to nth multiplex converters 5u1...5un. The end phase counter 24c is a counter for measuring the time corresponding to the phase difference Td calculated by the pulse calculation unit 23 by setting the counter value to a predetermined value and counting down.

[0047] 7 shows the processing steps of the m-th converter 5 (where m≧2). The pulse generating unit 12 performs the processing from S12 onwards on the condition that the power switch SWm-1 of the m-1th converter 5 is turned on in S11. The power switch SWm-1 shown here refers to either the upper arm switch SW1 or the lower arm switch SW2 of the m-1th converter 5, and the current command value I o Varies based on:

[0048] After the power switch SWm-1 of the m-1th converter 5 is turned on, in S12, the start phase counter 24b measures whether or not the phase difference Td=Ton / n, which is the start timing of the mth converter, has elapsed. When the condition of S12 is satisfied, in S13, the power switch SWm of the mth converter 5 is turned on. The power switch SWm refers to either the upper arm switch SW1 or the lower arm switch SW2 of the mth converter 5, and the current command value I o Varies based on:

[0049] Thereafter, each time the start phase counter 24b finishes counting, the pulse generating unit 12 generates a pulse to sequentially turn on the power switch SWm of the m-th multiplexed converter 5. This makes it possible to start outputting driving pulses with a phase difference Td between them.

[0050] On the other hand, as shown in FIG. 6, when the count by the pulse width counter 24a of the first multiplex converter 5u1 ends, it is determined that the on-time Ton has elapsed, and the pulse generating unit 12 stops the pulse output to the power switch SW1 or SW2 of the first multiplex converter 5u1. The pulse generating unit 12 starts counting by the end phase counter 24c at the timing when the count by the pulse width counter 24a ends. After that, the pulse generating unit 12 ends the count by the end phase counter 24c, and in S14, sequentially stops the pulse output to the power switch SWm of the m-th converter 5 each time the on-time Ton has elapsed. This sequentially turns off the m-th power switch SWm.

[0051] When the power switch SW1 or SW2 is turned off, hard switching occurs. However, if the charging time of the capacitor C provided at the output of the power switches SW1 and SW2 is made sufficiently longer than the switching time, the period during which the voltage rise is limited can be extended, enabling ZVS (Zero Voltage Switching). This makes it possible to reduce switching loss to almost zero. If the capacity of the output capacitor is insufficient, it is a good idea to add a capacitor C in parallel.

[0052] As a result, the pulse calculation unit 23 can generate multiple pulses to be input to the 1st...nth multiplex converters 5u1...5un by outputting pulses sequentially after the phase difference Td calculated as described above from the timing at which the pulse was output to the first multiplex converter 5u1.

[0053] The resonance time of each converter 5 is determined by the output capacitance of the power switches SW1 and SW2, the junction capacitance of the freewheeling diodes D1 and D2, and the inductance of the inductor L. It is preferable to set the on-timing of the power switches SW1 and SW2 to the lowest voltage point after a half resonance cycle determined by these capacitances and inductances. The lowest voltage point after a half resonance cycle is the point at which the inductor current I L This is the timing when the switching loss at turn-on is reduced to zero after overshooting zero. This enables ZCS (Zero-Current-Switching) and pseudo-ZVS (Zero-Voltage-Switching), and reduces switching loss at turn-on to virtually zero.

[0054] <Explanation of the multiplex number switching unit 25> The configuration and technical significance of the multiplex number switching unit 25, which is a feature of this embodiment, will be described below. As shown in FIG. 1, the pulse generating unit 12 switches the number of converters 5 to be simultaneously operated in multiplex operation, n a The multiplex number switching unit 25 corresponds to a multiplex operation number changing unit.

[0055] The multiplex number switching unit 25 switches the multiplex operation number n of the converter 5 according to the current command value Io or the maximum value of the current command value Io which changes depending on the input command rotation speed. a This is a block for changing the number of multiplex operations n by the multiplex number switching unit 25 according to the present application. a The switching method and its significance will be explained.

[0056] The multiplex number switching unit 25 changes the multiplex operation number n according to the current command value Io or the rotation speed command value. a and outputs it to the pulse calculation unit 23. Alternatively, the control device 10 calculates the number of multiple operations n a and inputs the multiplexing operation number n a Based on this data, the multiplexing number switching unit 25 outputs the multiplexing operation number n a Alternatively, the above may be switched.

[0057] Current command value I o When n is greater than 0, the multiplexing number switching unit 25 selects the multiplexing operation number n a Using these, the on-time Ton1, off-time Toff1, and period T of the upper arm switch SW1, and the phase difference Td between the converters are calculated based on equations (3-1) to (3-4), respectively.

number

[0058] The multiplex number switching unit 25 also switches the multiplex operation number n a In other words, the power switches SW1 and SW2 are kept off. As shown in FIG. 8 and FIG. 9, when the current command value Io is the same, the number of multiple operations n a The smaller the value, the longer the period and the wider the pulse width.

[0059] The multiplex number switching unit 25 changes the multiplex operation number n according to the current command value Io. a As shown in FIG. 10, the larger the current command value Io, the smaller the number of multiple operations n a The smaller the current command value Io, the smaller the number of multiple operations n a In addition, in FIG. 10, the switching timing of the current command value Io is reduced by the number of multiple operations n a However, the present invention is not limited to this.

[0060] As shown in FIG. 11, the multiplexing number switching unit 25 selects the multiplexing operation number na The timing of switching does not have to be the same as the timing of switching the current command value Io, and the number of multiple operations n is changed at a timing delayed from the current switching timing upon receiving a command to switch the current command value Io. a You can switch between multiple operations. a By reducing the pulse width, it is possible to suppress the shortening of the pulse width, and the converter 5 can output a pulse normally even when the current command value Io is relatively small.

[0061] As shown in FIG. 12, when the command speed is used as the command value, the number of multiple operations n a When a command rotation speed is input, the control device 10 changes the output current into a sinusoidal wave shape based on a current command value Io.

[0062] In this case, the number of multiple operations switching unit 25 increases the number of multiple operations n as the command rotation speed increases. a The smaller the command speed, the greater the number of multiple operations n a In addition, the multiplex number switching unit 25 changes the number of multiplex operations n according to the maximum value of the current command value Io, which varies depending on the input command rotation speed. a You can also switch between multiple operations n depending on the command speed. a You can also switch between the above.

[0063] According to this embodiment, the multiplexing number switching unit 25 switches the multiplexing operation number n a The number of multiple operations n can be changed. a By reducing the pulse width, the shortening of the pulse width can be suppressed. Therefore, even if the command rotation speed is relatively small, the converter 5 can be driven normally.

[0064] Second embodiment The second embodiment will be described with reference to Fig. 13. Regarding the second embodiment, differences from the first embodiment will be described, and a description of the same parts will be omitted.

[0065] Fig. 13 shows a flowchart replacing Fig. 7. As shown in Fig. 13, from the timing when the power switch SW1 or SW2 of the converter 5 of the first multiplex is turned on in S11a, the phase difference T of the m-th multiplex pulse with respect to the pulse of the first multiplex as a reference is calculated in S12a. dm After the time period Tm=Ton×(m−1) / n has elapsed, the power switch SWm of the m-th converter 5 may be turned on in S13.

[0066] The phase difference Tdm of the start phase of the m-th multiplex with respect to the start phase of the first multiplex can be calculated based on the relationship in the following formula (4).

number

[0067] For example, in the case of the U phase, the first multiple converter 5u1 may be set as a master phase, and the power switch SW1 or SW2 of the master phase converter 5u1 may be turned on, and then the power switches SW1 or SW2 of the second or more m-th multiple converters 5u2...5un may be set as slave phases and turned on in sequence. This embodiment also provides the same effects as the first embodiment.

[0068] Third embodiment The third embodiment will be described with reference to Figs. 14 and 15. The differences from the first and second embodiments will be described. a When switching between the current boundary modes, it may not be possible to continue the rules in the current boundary mode.

[0069] As shown in FIG. 14, the number of multiple operations n a Consider switching from 2 to 4 and switching the period T of the multiple pulses to period T2 = T / 2.

[0070] As shown in FIG. 14, before the switching timing ta, the first and third multiplex converters 5u1, 5u3 are operated to continue the current boundary mode and control the phase current Iu to the desired current command value Io.

[0071] However, after the switching timing ta, when the operations of the first to fourth multiplex converters 5u1 to 5u4 are switched in sequence at a period Td2=T2 / 4, the inductor current I L can be made zero, but the inductor current I L cannot be reduced to zero and operation in current boundary mode cannot continue.

[0072] In this way, the current number of multiple operations n is multiplied by the next number of multiple operations n a When switching to the current boundary mode, the pulse generating section 12 may continue the current boundary mode by receiving a switching output from the multiplex number switching section 25 and generating an interpolation multiple pulse Pu.

[0073] Number of multiple operations n a An example in which the pulse generating section 12 generates an interpolation multiple pulse Pu to continue the operation in the current boundary mode when switching between the current boundary modes will be described with reference to FIG.

[0074] Cycle before switching T, cycle after switching T a , the number of multiple operations before switching is n, and the number of multiple operations after switching is n a , where n <n a In this case, the phase difference before switching T d =T / n a , phase difference T after switching da =T a / n a It becomes.

[0075] Here, the phase difference before switching T d is the number of multiple operations n a 15, since the number of multiplexed operations before switching is n, there is no change in the phase current Iu before and after the switching timing tb.

[0076] In this case, if the period of the m-th interpolation multiple pulse in the interpolation multiple pulse Pu is Tm, the periods T1 to Tm can be calculated based on FIG. 15 and the following formula (5).

number

[0077] By generating an interpolated multiple pulse Pu having a period Tm of the m-th multiple pulse from the switching timing tb shown in FIG. 15, it is possible to switch the number n of multiple operations while continuing the current boundary mode.

[0078] The above method is merely an example, and the method of generating the interpolated multiple pulses Pu is not limited to the above method. For example, the phase difference T o The phase difference before switching, T / n a and phase difference after switching T a / n a It may be set to any value between

[0079] As described above, according to the third embodiment, when the multiple operation number n is switched, the interpolation multiple pulse Pu is generated in the middle of the switching. This makes it possible to stably continue the current boundary mode even if the multiple operation number n of the converter 5 is switched.

[0080] (Modifications of the feedback control configuration related to the first to third embodiments) Modified examples of the feedback control configuration related to the first to third embodiments will be described with reference to Fig. 16 to Fig. 20. A control device 10A may be configured as shown in Fig. 16. The control device 10A incorporates a control unit 11A, a pulse generating unit 12A, and a memory 37. The control unit 11A is configured by connecting a subtractor 30, a current controller 31, a two-phase to three-phase converter 32 as a phase converter, and a three-phase to two-phase converter 34 in the illustrated form.

[0081] The control unit 11A receives a d-axis current command value Id* and a q-axis current command value Iq*. A three-phase to two-phase converter 34 of the control unit 11A receives the phase currents Iu, Iv, and Iw, and also receives the rotor angle θ from a rotational position sensor 4a such as a resolver installed on the motor 4 to calculate information on the angular velocity ω. The three-phase to two-phase converter 34 converts the three-phase currents Iu, Iv, and Iw of the motor 4 into a d-axis current Id and a q-axis current Iq and outputs them to the subtractor 30.

[0082] The subtractor 30 subtracts the d-axis current Id and the q-axis current Iq from the d-axis current command value Id* and the q-axis current command value Iq*, respectively, and outputs the results to the current controller 31. The current controller 31 outputs the dq-axis current command values ​​Id_cmd and Iq_cmd to the two-phase to three-phase converter 32, for example, by proportional-integral control. The two-phase to three-phase converter 32 converts the dq-axis current command values ​​input from the current controller 31 into current command values ​​Iu_cmd, Iv_cmd, Iw_cmd for each of the three phases of the motor 4, and outputs them to the pulse generating unit 12A.

[0083] The pulse generating unit 12A includes pulse generating blocks 12u, 12v, and 12w for each of the three phases, and also includes one multiplex number switching unit 25 for changing the operation multiplex number. Each of the pulse generating blocks 12u, 12v, and 12w has functional configurations as a zero current detection unit 22, a pulse calculation unit 23, and a gate driving unit 21. The pulse calculation unit 23 of each of the pulse generating blocks 12u, 12v, and 12w receives current command values ​​Iu_cmd, Iv_cmd, and Iw_cmd for each of the three phases of the motor 4. The zero current detection unit 22 detects zero current for each phase and outputs it to the pulse calculation unit 23.

[0084] The pulse calculation unit 23 calculates each parameter of the on time Ton and off time Toff of the upper arm switch SW1 and the lower arm switch SW2 when each converter 5 of the multi-phase converters 6u, 6v, 6w of each phase operates in the current boundary mode, the period T, and the phase difference Td between multiple pulses to be input to multiple converters 5 of the same phase (e.g., U phase).

[0085] The gate driver 21 drives the three-phase inverter 3 based on the calculation result of the pulse calculator 23. The gate driver 21 drives the upper arm switch SW1 or the lower arm switch SW2 to turn on and off based on the on-time Ton, the period T, and the phase difference Td calculated by the pulse calculator 23.

[0086] The multiplex number switching unit 25A receives the current command value Iq_cmd of the q-axis current from the current controller 31 and switches the multiplex operation number n of the converter 5. a The multiplex number switching unit 25A changes the multiplex operation number n based on the current command value Iq_cmd as shown in FIG. a For example, if the current command value Iq_cmd exceeds 10, the number of multiple operations n a is set to 8, and if the current command value Iq_cmd is greater than 4 and less than 10, the number of multiple operations n a is set to 4, and if the current command value Iq_cmd is greater than 1 and less than 4, the number of multiple operations n a is set to 2, and if the current command value Iq_cmd is 1 or less, the number of multiple operations n a may be set to 1.

[0087] In this way, by changing the number of multiplex operations n every 2^n in accordance with the current command value Iq_cmd, it becomes possible to omit the phase difference calculation in the pulse calculation unit 23. a An example of the operation when changing is omitted here because it is similar to the flow of Figures 10 and 11. As shown in Figure 18, which shows examples of waveforms of three-phase currents Iu, Iv, and Iw before and after application, the effects of harmonic current distortion can be improved.

[0088] For example, as shown in Fig. 19, when the converter 5 is operating in 4-multiplex mode, if the multiplex number switching unit 25 switches to 3-multiplex mode, it becomes necessary to recalculate the signal related to the phase difference T / 3 with respect to the period T. As described above, the converter 5 to be operated can be changed by switching between operating and stopping the converter 5 based on predetermined conditions. As shown in Fig. 20 as an example of switching to 2-multiplex mode, it is only necessary to stop the second and fourth multiplexed signals, so there is no need to bother to calculate the signal related to the phase difference T / 2, and the amount of control processing can be reduced.

[0089] (Fourth embodiment) The fourth embodiment will be described with reference to Fig. 21 to Fig. 24. In this embodiment, an embodiment will be described in which the phase currents Iu, Iv, and Iw based on multiple pulses are smoothly changed in response to a change in the current command value Io while the number n of multiplex operations of the converter 5 is kept the same. That is, in this embodiment, the configuration of the multiplex number switching unit 25 described in the first to third embodiments may be omitted.

[0090] In order to continue operation in the current boundary mode even in a transient state in which the pulse width is changed in response to a change in the current command value Io, the pulse generating unit 112 may generate, by the function of the interpolating unit 12b, an interpolated multiple pulse Pu that interpolates between a plurality of multiple pulses sequentially output to each converter 5. As shown in Fig. 21, the pulse generating unit 112 has the function of the interpolating unit 12b.

[0091] For example, the interpolation unit 12b may set the value of the phase difference between the turn-off timings between the interpolated multiple pulses Pu between the value of the phase difference between the turn-off timings between the current first multiple pulses to be input to each converter 5 in each of the UVW phases and the value of the phase difference between the turn-off timings between the next second multiple pulses.

[0092] Specifically, it is desirable to generate the interpolated multiple pulse Pu by the function of the interpolation unit 12b as follows. As shown in FIG. 22, the phase difference between the turn-off timings of the first multiple pulses this time is P kThe phase difference between the turn-off timings of the next multiple pulses is P k+1 Then, the phase difference P between the turn-off timings of the interpolated multiple pulses Pu ka is calculated based on the following formula (6).

number

[0093] Here, the linear interpolation method is described using the formula (6) as an example, but is not limited to this. It is sufficient that the phase difference between the turn-off timings or the turn-on timings of the interpolated multiple pulses Pu is set to a value between the phase differences between the current and next multiple pulses. Then, the output current between the current and next multiple pulses can be smoothly interpolated, and the drive current output of the converter 5 can be leveled.

[0094] FIG. 23 shows an example of changes in the counter values ​​of the pulse width counter 24a, the start phase counter 24b, and the end phase counter 24c when an interpolation multiple pulse Pu is generated using the counter 24 described in the above embodiment, and an example of an output result of the interpolation multiple pulse Pu by the pulse calculation unit 23.

[0095] Hereinafter, a method for generating the interpolation multiple pulse Pu will be described with reference to the flowchart of Fig. 24. As shown in Fig. 24, the pulse generating unit 12 generates the inductor current I L If zero is detected, the counter value is set in S12.

[0096] The pulse calculation unit 23 sets the counter values ​​of the pulse width counter 24a, the start phase counter 24b, and the end phase counter 24c using a method similar to that shown in the first embodiment, thereby setting the on-time Ton, off-time Toff, and period T of the power switch SW1 or SW2, and the phase difference Td of the on / off timing between the master phase and slave phase converters 5u1...5un.

[0097] When the current command value Io does not change and the next pulse width of the multiple pulse is not changed from the current pulse width, the pulse generating unit 12 judges NO in S13 and outputs the next multiple pulse based on the set counter value in S18.

[0098] However, when the current command value Io changes and the next pulse width of the multiple pulse is to be changed from the current pulse width, the answer is YES in S13, the counter value is calculated in S14 so as to satisfy the condition for the phase difference between the interpolated multiple pulses Pu described above, and the interpolated multiple pulse Pu is output in S15.

[0099] On the other hand, when the pulse generating unit 12 detects zero current of the master phase of the interpolation multiple pulse Pu in S16, it sets the counter value again based on the pulse width and phase difference of the next multiple pulse in S17, and outputs the next multiple pulse in S18.

[0100] Although the above is illustrated by a flowchart to make the flow easier to understand, the processing steps may be processed in parallel as necessary.

[0101] When conventional technology is applied, it is necessary to calculate the width of the pulses to be input to the converter 5 of the master phase and each slave phase as the interpolation values ​​between the multiple pulses, and it has been confirmed that the output current temporarily increases while these interpolation pulses are being output.

[0102] According to this embodiment, the phase difference between the current first multiple pulse and the next second multiple pulse is considered, and the interpolation multiple pulse Pu is generated so as to linearly interpolate the phase difference by the function of the interpolation unit 12b. In the above-mentioned method, the phase difference between the first multiple pulses before the interpolation is generated as P k , the phase between the second multiple pulses after interpolation is P k+1 Then, the phase difference between the interpolated multiple pulses Pu is the phase difference P k and phase difference P k+1This allows smooth interpolation between the current multiple pulse and the next multiple pulse, and the drive current output of each converter 5 can be leveled.

[0103] In addition, according to this embodiment, the phase differences between the interpolated multiple pulses Pu can all be set to the same phase difference, so that the calculation process can be completed in one go. When the phase differences between the interpolated multiple pulses Pu are linearly interpolated, for example, the phase difference P k and the phase difference between the next multiple pulses P k+1 The width of the interpolated multiple pulse Pu and the phase difference between the interpolated multiple pulses Pu corresponding to the multiplexing number of the converter 5 may be mapped and stored in the memory 37, and the calculation may be performed using the map stored in the memory 37.

[0104] Fifth embodiment The fifth embodiment will be described with reference to Figures 25 and 26. Since this embodiment is a modification of the fourth embodiment, only the parts that differ from the fourth embodiment will be described.

[0105] 25, the pulse calculation unit 23 includes a start phase counter 24b that measures the start phase of each pulse of the multiple pulse, and a multiple pulse width counter 24d that controls the pulse width of each pulse of the multiple pulse. The pulse generation unit 112 inputs a pulse to the next multiple converter 5 every time the start phase counter 24b expires.

[0106] The start phase counter 24b measures the start phase at which each of the multiplex pulses starts to be output. The multiplex pulse width counter 24d measures the time of each pulse width input to the first to fourth multiplex converters 5u1...5u4, and measures the start phase measured by the start phase counter 24b as a reference and measures the pulse width of each of the multiplex pulses.

[0107] The pulse width of the group of multiple pulses input to the converter 5 of the master phase to the slave phase this time is T k The phase difference between these multiple pulses is P k In addition, the pulse width of a group of multiple pulses to be input to the converters 5 of the master phase to the slave phase next time is set as T k+1 , the phase difference between these multiple pulses is P k+1 Then, the pulse width T of the interpolated multiple pulse Pu shown in FIG. ka can be calculated based on the following formula (7).

number

[0108] The pulse calculation unit 23 calculates the phase difference P k , P k+1 are successively measured, and the pulse width T k , T ka , T k+1 By measuring the current multiple pulse, the interpolated multiple pulse Pu, and the next multiple pulse can be calculated in sequence. Since the method of this embodiment can also generate a multiple pulse, the same effects as those of the fourth embodiment can be achieved.

[0109] (Modifications of the feedback control configuration related to the third to fifth embodiments) Modified examples of the feedback control configuration related to the third to fifth embodiments will be described with reference to Fig. 27 to Fig. 31. A control device 10B may be configured as shown in Fig. 27. The control device 10B incorporates a control unit 11B, a pulse generating unit 12B, and a memory 37. The control unit 11B is configured by connecting a subtractor 30, a current controller 31, a two-phase to three-phase converter 32 as a phase converter, and a three-phase to two-phase converter 34 in the illustrated form.

[0110] The control unit 11B receives the d-axis current command value Id* and the q-axis current command value Iq*. The three-phase to two-phase converter 34 of the control unit 11 receives the phase currents Iu, Iv, and Iw, and also receives the rotor angle θ from a rotational position sensor 4a such as a resolver installed in the motor 4 to calculate information on the angular velocity ω. The three-phase to two-phase converter 34 converts the three-phase currents Iu, Iv, and Iw of the motor 4 into a d-axis current Id and a q-axis current Iq and outputs them to the subtractor 30.

[0111] The subtractor 30 subtracts the d-axis current Id and the q-axis current Iq from the d-axis current command value Id* and the q-axis current command value Iq*, respectively, and outputs the results to the current controller 31. The current controller 31 outputs the dq-axis current command values ​​Id_cmd and Iq_cmd to the two-phase to three-phase converter 32, for example, by proportional-integral control. The two-phase to three-phase converter 32 converts the dq-axis operation amounts input from the current controller 31 into current command values ​​Iu_cmd, Iv_cmd, and Iw_cmd for each of the three phases of the motor 4, and outputs them to the pulse generating unit 12B.

[0112] The pulse generating unit 12B includes pulse generating blocks 12u, 12v, and 12w for each of the three phases. Each of the pulse generating blocks 12u, 12v, and 12w includes functional configurations as a zero current detection unit 22, a pulse calculation unit 23Z, and a gate driving unit 21. The pulse calculation unit 23Z for each of the pulse generating blocks 12u, 12v, and 12w receives the current command values ​​Iu_cmd, Iv_cmd, and Iw_cmd for each of the three phases of the motor 4. The zero current detection unit 22 detects zero current for each phase and outputs the detected current to the pulse calculation unit 23Z.

[0113] The pulse calculation unit 23Z calculates each parameter of the on time Ton and off time Toff of the upper arm switch SW1 and the lower arm switch SW2 when each converter 5 of the multi-phase converters 6u, 6v, 6w of each phase operates in the current boundary mode, the period T, and the phase difference Td between the multiple pulses to be input to multiple converters 5 of the same phase.

[0114] The gate driver 21 drives the three-phase inverter 3 based on the calculation result of the pulse calculator 23. The gate driver 21 drives the upper arm switch SW1 or the lower arm switch SW2 to turn on and off based on the on-time Ton, the period T, and the phase difference Td calculated by the pulse calculator 23.

[0115] The pulse calculation unit 23Z of each phase calculates the on-time Ton and off-time Toff of the upper arm switch SW1, as well as the period T and phase difference Td, based on the values ​​of the current command values ​​Iu_cmd, Iv_cmd, and Iw_cmd. For example, in the U-phase pulse generating block 12u, when the current command value Iu_cmd>0, the pulse calculation unit 23Z calculates the on-time Ton and off-time Toff of the upper arm switch SW1, as well as the period T and phase difference Td, based on equations (1-1) to (1-4), respectively.

[0116] In the V-phase pulse generating block 12v, the pulse calculation unit 23Z calculates the on-time Ton and off-time Toff, as well as the period T and phase difference Td of the upper arm switch SW1 based on the equations (1-1) to (1-4) when the current command value Iv_cmd>0. In the W-phase pulse generating block 12w, the pulse calculation unit 23Z calculates the on-time Ton and off-time Toff, as well as the period T and phase difference Td of the upper arm switch SW1 based on the equations (1-1) to (1-4) when the current command value Iw_cmd>0.

[0117] The pulse calculation unit 23Z calculates the on-time Ton and off-time Toff, as well as the period T and phase difference Td of the upper arm switch SW1 based on the values ​​of the current command values ​​Iu_cmd, Iv_cmd, and Iw_cmd. For example, in the U-phase pulse generating block 12u, when the current command value Iu_cmd<0, the pulse calculation unit 23Z calculates the on-time Ton and off-time Toff, as well as the period T and phase difference Td of the lower arm switch SW2 based on the equations (2-1) to (2-4), respectively.

[0118] In the V-phase pulse generating block 12v, when the current command value Iv_cmd<0, the pulse calculation unit 23Z calculates the on-time Ton and off-time Toff of the lower arm switch SW2, as well as the period T and phase difference Td, based on the equations (2-1) to (2-4), respectively. In the W-phase pulse generating block 12w, when the current command value Iw_cmd<0, the pulse calculation unit 23Z calculates the on-time Ton and off-time Toff of the lower arm switch SW2, as well as the period T and phase difference Td, based on the equations (2-1) to (2-4), respectively.

[0119] 23 shows changes in the counter values ​​of the pulse width counter 24a, the start phase counter 24b, and the end phase counter 24c when generating an interpolated multiple pulse Pu using the counter 24. The pulse calculation unit 23 can also realize generation of an interpolated multiple pulse Pu with the hardware configuration shown in the upper part of FIG.

[0120] The pulse calculation unit 23Z executes the processes of the formulas (1-1) to (2-4) to input the on-time Ton and the period T of the upper arm switch SW1 and the lower arm switch SW2 to the pulse width counter 24a as parameters of the counter 24. Then, the pulse calculation unit 23 calculates the phase difference P k+1 is input to the start phase counter 24b and the end phase counter 24c.

[0121] When the phase difference between the interpolated multiple pulses Pu is, for example, linearly interpolated, the pulse calculation unit 23Z calculates the phase difference P k and the phase difference between the multiple pulses this time, P k+1 The average value of P k +P k+1 / 2 is calculated, and the average value is output to the end phase counter 24c, which corrects the end phase. In this way, the 1st to nth multiple interpolated multiple pulses Pu can be generated using the counter 24 as shown in Fig. 23. By generating the interpolated multiple pulse Pu using this method, the current boundary mode can be maintained when the pulse width is changed, but in cases where there is a large change in the phase voltages Vu, Vv, and Vw, for example, the current boundary mode may not be maintained.

[0122] In the region of maximum and minimum values ​​where the phase voltages Vu, Vv, and Vw are almost constant, no problem occurs with the interpolation pulse output of the pulse calculation unit 23. See the center diagram of FIG. 29. However, in the time region where the phase voltages Vu, Vv, and Vw change sharply, the voltage changes between the time when the first multiplex pulse that serves as the master is output and the time when the second to fourth multiplex pulses that serve as the slave are output. For this reason, the current for the on-time Ton of the pulse calculation unit 23 decreases or increases, and the period of the slave pulse changes, resulting in an unintended output. See the right diagram of FIG. 29. In the embodiment described later, correction is performed based on the back electromotive force, but there are cases where the current boundary mode cannot be maintained even if correction is performed based on the back electromotive force. An experimental example of the first and second multiplex pulses at this time is shown in the upper diagram of FIG. 31, but the current boundary mode cannot be maintained in region B.

[0123] In order to eliminate this problem, it is preferable to configure pulse calculation unit 23Z as the hardware shown in the lower part of Fig. 28. In this hardware configuration, pulse calculation unit 23Z corrects the end phase every time measurement is completed using start phase counter 24b and end phase counter 24c.

[0124] At this time, the pulse calculation unit 23Z executes the calculation process of the formulas (1-1) to (2-4) to input the on-time Ton and the period T of the upper arm switch SW1 and the lower arm switch SW2 to the pulse width counter 24a as parameters of the counter 24. Furthermore, the pulse calculation unit 23Z calculates the phase difference P k+1to the start phase counter 24b and the end phase counter 24c. In the same manner as described above, when the phase difference between the interpolated multiple pulses Pu is, for example, linearly interpolated, the pulse calculation unit 23Z inputs the phase difference P k and the phase difference between the multiple pulses this time, P k+1 The average value of P k +P k+1 The pulse calculation unit 23Z calculates / 2, outputs the average value to the end phase counter 24c, and corrects the end phase by the end phase counter 24c. Furthermore, every time the start phase counter 24b and the end phase counter 24c finish measurement, the pulse calculation unit 23Z outputs a correction value A to the end phase counter 24c based on the counter correction instruction, thereby updating the counter value.

[0125] As shown in the timing chart on the right side of Fig. 30, the value of the end phase counter 24c is gradually changed based on the value of the correction value A. Then, the phase currents Iu, Iv, and Iw can be maintained at the intended amplitudes. In this way, the pulse current boundary mode can be maintained as shown in the lower diagram of Fig. 31.

[0126] Sixth embodiment The sixth embodiment will be described with reference to FIGS.

[0127] The motor 4 generates a back electromotive force due to a change in magnetic flux according to the rotation speed, and the output voltage fluctuates. If the control response is insufficient, the back electromotive force affects the phase currents Iu, Iv, and Iw, causing harmonic current distortion as shown in FIG. 32. This high-frequency current component causes the motor 4 to generate noise. In this case, unless the control response is improved, the effect of the harmonic current distortion cannot be improved, and there is a risk that the control device 10, which has limited physical resources, cannot improve the effect. In particular, if the motor 4 is rotated at a speed higher than a predetermined speed, the output voltage Vout is significantly affected by the back electromotive force, and the current distortion may become even larger.

[0128] Therefore, in this embodiment, a method for correcting the pulse width of the multiple pulses based on the back electromotive force generated in the motor 4 will be described. As shown in Fig. 33, the control device 10 includes a pulse generating unit 212. The control device 10 includes a function as a back electromotive force correcting unit 12c.

[0129] At this time, the amplitude at which the back electromotive force is at its maximum is Vm, and the phase information of the back electromotive force is (ω e t+φ)=(2πf e t+φ), where t is time and ω e is the electrical angular velocity, φ is the phase difference with respect to the current, f e It is preferable that the output voltage Vout is operated at half the input voltage Vin, i.e., Vin / 2, and it is preferable that the output voltage Vout be controlled so as to satisfy the relationship shown in the following equation (8) taking into account the back electromotive force.

number

[0130] Furthermore, when the pulse generating unit 212 calculates the amplitude Vm of the back electromotive voltage, it is preferable to perform the calculation based on the rotation speed of the motor 4. Then, it is preferable that the pulse generating unit 212 calculates the output voltage Vout based on the rotation position of the motor 4, the amplitude Vm of the back electromotive voltage, and phase information, and corrects the pulse width of the multiple pulses based on this output voltage Vout. Specifically, for example, in the formulas (1-1) to (1-4), (2-1) to (2-4), and (3-1) to (3-4), Vout may be calculated as formula (8).

[0131] The back electromotive force is predictable because it is an inherent value based on the individual motor 4. For example, the back electromotive force can be calculated and predicted by multiplying the rotation speed of the motor 4 by a back electromotive force constant determined inherently for the motor 4. Furthermore, in order to correct the pulse width with high accuracy, it is sufficient to take into account the voltage generated in the winding resistance and inductance of the motor 4 in addition to the back electromotive force of the output voltage Vout.

[0132] The control device 10 determines the rotation speed of the motor 4 using the sensing results of the rotational position sensor 4a of the motor 4, and performs vector control by converting the three-phase currents Iu, Iv, Iw into currents in the magnetic force direction related to torque based on the rotor position of the motor 4.

[0133] An example of the feedback control configuration at this time will be described with reference to Fig. 34. The control unit 11 is configured by connecting a subtractor 30, a current controller 31, a two-phase to three-phase converter 32 as a phase converter, and a three-phase to two-phase converter 34 in the illustrated configuration.

[0134] The control unit 11 receives a current command value Io and performs current feedback control based on the difference current between the currents Iu, Iv, and Iw of the phases of the motor 4. The three-phase to two-phase converter 34 converts the three-phase currents into dq-axis currents, and the current controller 31 outputs the manipulated variables of the dq axes, for example, by proportional-integral control. The two-phase to three-phase converter 32 converts the manipulated variables of the dq axes input from the current controller 31 into manipulated variables for each of the three phases of the motor 4.

[0135] In this embodiment, the pulse generating unit 212 has the functions of the back electromotive force correcting unit 12c and the pulse calculating unit 33. The pulse calculating unit 33 calculates each parameter including the on-time Ton1, off-time Toff1, cycle T, and phase difference Td of the multiple pulses to be input to the converter 5 for each phase of the motor 4. When the current command value Io is positive, the on-time Ton1, off-time Toff1, and cycle T can be calculated using the on-time Ton1, off-time Toff1, cycle T, and phase difference Td of the above-mentioned equations (1-1) to (1-4). When the current command value Io is negative, the on-time Ton2, off-time Toff2, cycle T, and phase difference Td of the above-mentioned equations (2-1) to (2-4) can be calculated.

[0136] The function of the back electromotive force correction unit 12c here includes, for example, a process of calculating the amplitude Vm of the back electromotive force from the rotation speed of the motor 4, and a process of calculating the output voltage Vout based on the amplitude Vm of the back electromotive force and phase information. When the pulse calculation unit 33 calculates the above parameters for each phase of the motor 4, the pulse calculation unit 33 corrects the pulse width based on the output voltage taking the back electromotive force into consideration by the function of the back electromotive force correction unit 12c. Since the amplitude Vm of the back electromotive force is proportional to the rotation speed of the motor 4, it is desirable to calculate the amplitude Vm of the back electromotive force so that it is proportional to the rotation speed of the motor 4.

[0137] As the control method, the current control method has been exemplified above, but a control unit 211 may be configured instead of the control unit 11 as shown in FIG. 35. The control unit 211 includes a speed controller 35 and a subtractor 36 in addition to a subtractor 30, a current controller 31, a two-phase to three-phase converter 32, and a three-phase to two-phase converter 34. As shown in this control unit 211, the angular velocity ω of the motor 4 may be obtained using a rotational position sensor 4a, the difference between the angular velocity ω and the command angular velocity ω* may be detected by a subtractor 36, and the speed may be controlled by a speed controller 35. In this case, a non-interference control method that eliminates mutual interference between the d-axis current and the q-axis current may be used. This can eliminate the cause of instability that occurs when the motor 4 is driven. When the back electromotive force becomes large relative to the input voltage Vin, it is desirable to perform field weakening control.

[0138] Each converter 5 is configured by connecting step-down converters in parallel as shown in FIG. 33, and this configuration is suitable for the case where the output voltage Vout fluctuates within the range of the power supply voltage of the battery 2.

[0139] In the above, the back electromotive force is calculated from the rotation speed of the motor 4, but a map linking the rotation speed of the motor 4 with the amplitude Vm and phase information of the back electromotive force may be prepared in advance in the memory 37. The pulse generating unit 212 may correct the width of the multiple pulses by referring to the amplitude Vm and phase information of the back electromotive force stored in the memory 37 corresponding to the rotation speed of the motor 4. In addition, the output voltage Vout may be detected by the voltage sensor 15 and used in the calculation after taking into account a delay time based on the detection of the voltage sensor 15.

[0140] The pulse generating unit 212 may store the result of periodically correcting the width of the multiple pulse in accordance with the back electromotive force by the back electromotive force correcting unit 12c in the memory 37 in correspondence with the current command value Io or the command rotation speed. Also, while the pulse calculating unit 33 is periodically calculating the correction contents based on the next current command value Io, the width of the multiple pulse may be corrected by the function of the back electromotive force correcting unit 12c using the current or past calculation results related to the multiple pulse stored in the memory 37.

[0141] As shown in this embodiment, the back electromotive force correction unit 12c corrects the pulse width based on the back electromotive force, so that the back electromotive force of the motor 4 can be incorporated into the calculation of the pulse width in a feedforward manner. This makes it possible to more closely approximate the current waveform to the desired ideal sine wave, as shown in Fig. 36, thereby achieving high efficiency, low noise, and low EMC. In addition, this can be achieved by a simple control system using only current feedback control.

[0142] Seventh embodiment 3 and 33, when the upper arm switch SW1 as the first arm switch is turned on and off while the lower arm switch SW2 is turned off, the current is commutated through the inductor L and the freewheeling diode D2.

[0143] At this time, instead of the freewheeling action of the freewheeling diode D2, the lower arm switch SW2 opposite the upper arm switch SW1 may be turned on as a second arm switch to perform synchronous rectification.

[0144] Synchronous rectification can reduce losses even further. It also eliminates the need for an external freewheel diode D2, making it possible to reduce the size of the device. To achieve synchronous rectification, it is sufficient to turn on the opposing upper arm switch SW1 for a period of time that is set based on the above-mentioned formula for calculating the off-time Toff of the pulse (for example, the off-time Toff1 in formula (1-2), the off-time Toff2 in formula (2-2), or the off-time Toff1 in formula (3-2)). However, to prevent a short circuit between the power switches SW1 and SW2, it is desirable to provide a dead time between the on-time Ton of the upper arm switch SW1 and the lower arm switch SW2.

[0145] According to the above-mentioned synchronous rectification method, the synchronous rectification time can be predicted by calculation. In this case, it is not necessary to provide a separate external sensor to detect the timing of synchronous rectification, which contributes to the miniaturization and cost reduction of the system.

[0146] As explained above, when the current command value Io>0, the on-time Ton1, period T, and phase difference Td of the upper arm switch SW1 are set in the same manner as in equations (1-1), (1-3), and (1-4), respectively. However, the synchronous rectification time should be set to a time equal to the off-time Toff1 shown in equation (1-2) in accordance with the on-time Ton2 of the lower arm switch SW2 during synchronous rectification.

[0147] In addition, when the current command value Io<0, the on-time Ton2, period T, and phase difference Td of the lower arm switch SW2 are set in the same manner as in equations (2-1), (2-3), and (2-4), respectively, and a time equal to the off-time Toff2 shown in equation (2-2) is set in accordance with the on-time Ton1 of the upper arm switch SW1 during synchronous rectification.

[0148] (Modifications of the feedback control configuration related to the sixth and seventh embodiments) Modified examples of the feedback control configuration related to the sixth and seventh embodiments will be described with reference to Fig. 37 to Fig. 42. A control device 10C may be configured as shown in Fig. 37. The control device 10C incorporates a control unit 11C, a pulse generating unit 12C, and a memory 37. The control unit 11C is configured by connecting a subtractor 30, a current controller 31, a two-phase to three-phase converter 32 as a phase converter, and a three-phase to two-phase converter 34 in the illustrated form.

[0149] The control unit 11C receives the d-axis current command value Id* and the q-axis current command value Iq*. The three-phase to two-phase converter 34 of the control unit 11C receives the phase currents Iu, Iv, and Iw, and also receives the rotor angle θ from the rotational position sensor 4a to calculate information on the angular velocity ω. The three-phase to two-phase converter 34 converts the three-phase currents Iu, Iv, and Iw of the motor 4 into the d-axis current Id and the q-axis current Iq and outputs them to the subtractor 30.

[0150] The subtractor 30 subtracts the d-axis current Id and the q-axis current Iq from the d-axis current command value Id* and the q-axis current command value Iq*, respectively, and outputs the results to the current controller 31. The current controller 31 outputs the dq-axis current command values ​​Id_cmd and Iq_cmd to the two-phase to three-phase converter 32, for example, by proportional-integral control. The two-phase to three-phase converter 32 converts the dq-axis operation amounts input from the current controller 31 into current command values ​​Iu_cmd, Iv_cmd, and Iw_cmd for each of the three phases of the motor 4, and outputs them to the pulse generating unit 12A.

[0151] The pulse generating unit 12C includes pulse generating blocks 12u, 12v, and 12w for each of the three phases. Each of the pulse generating blocks 12u, 12v, and 12w includes a zero current detection unit 22, a pulse calculation unit 23, a gate driving unit 21, and a functional configuration of a back electromotive force correction unit 12c for correcting a pulse based on a back electromotive force. The pulse calculation unit 23 of each of the pulse generating blocks 12u, 12v, and 12w receives the current command values ​​Iu_cmd, Iv_cmd, and Iw_cmd for each of the three phases of the motor 4. The zero current detection unit 22 detects the zero current for each phase and outputs it to the pulse calculation unit 23.

[0152] The pulse calculation unit 23 calculates each parameter of the on time Ton and off time Toff of the upper arm switch SW1 and the lower arm switch SW2 when each converter 5 of the multi-phase converters 6u, 6v, 6w of each phase operates in the current boundary mode, the period T, and the phase difference Td between multiple pulses to be input to multiple converters 5 of the same phase (e.g., U phase).

[0153] The gate driver 21 drives the three-phase inverter 3 based on the calculation result of the pulse calculator 23. The gate driver 21 drives the upper arm switch SW1 or the lower arm switch SW2 to turn on and off based on the on-time Ton, the period T, and the phase difference Td calculated by the pulse calculator 23.

[0154] A map linking the rotation speed of the motor 4 with the amplitude Vm and phase difference φ of the back electromotive force is stored in advance in the memory 37. The back electromotive force correction unit 12c for each phase refers to the motor rotation speed dependent on the angle θ of the motor 4 by checking it against the stored contents of the memory 37, and inputs the amplitudes Vm_u, Vm_v, and Vm_w of the back electromotive force and the phase differences φ_u, φ_v, and φ_w for each phase.

[0155] The back electromotive voltage correction unit 12c of each phase calculates the output voltages Vout_u, Vout_v, and Vout_w based on the amplitudes Vm_u, Vm_v, and Vm_w of the back electromotive voltages and the phase differences φ_u, φ_v, and φ_w of the back electromotive voltages together with the power supply voltage Vin. At this time, as shown in the above-mentioned formula (8), as shown in FIG. 38, the back electromotive voltage correction unit 12c calculates the output voltage Vout. This makes it possible to correct the influence of the back electromotive voltage as in the above-mentioned embodiment. As shown in FIG. 39, the output voltage Vout can be corrected based on the back electromotive voltage, and as shown in FIG. 40, the influence of harmonic current distortion can be eliminated as much as possible to improve the output voltage Vout.

[0156] Furthermore, when the rotation speed of the motor 4 increases, the amplitude Vm of the back electromotive voltage increases. As the amplitude Vm of the output voltage Vout approaches the power supply voltage Vin, the voltage generated in the inductor L decreases, and no current can flow through the inductor L. In this case, there is a risk of the device becoming uncontrollable and inoperable. Therefore, as shown in FIG. 41, it is desirable to provide a power supply voltage utilization rate improvement control unit 21d that improves the utilization rate of the power supply voltage Vin. When the amplitude Vm of the back electromotive voltage exceeds a predetermined threshold, this voltage utilization rate improvement control unit 12d sets upper and lower limits on the output voltage Vout after the back electromotive voltage is corrected and outputs the output voltage Vout.

[0157] For example, the upper limit is set to a value slightly lower than the power supply voltage Vin, and the lower limit is set to a value slightly higher than the reference value 0 of the power supply voltage Vin. Here, the voltage utilization rate improvement control unit 12d clamps the output voltage Vout after the back electromotive force correction to the upper limit value if it exceeds the upper limit value, and clamps it to the lower limit value if it falls below the lower limit value.

[0158] This allows the pulse calculation unit 23 to perform pulse calculations while limiting the sine wave peak of the output voltage Vout for pulse width correction calculation by upper and lower limits. Since upper and lower limits are set for the output voltage Vout, it can operate without problems even when a relatively large back electromotive force is generated, and the operable range can be expanded. In addition, the voltage utilization rate can be increased.

[0159] Eighth embodiment The eighth embodiment will be described with reference to Fig. 43. In the sixth and seventh embodiments described above, the number of multiple operations n is not limited to 2 or more, and can also be applied to the case where n=1.

[0160] That is, as shown in FIG. 43, in each of the UVW phases, converters 5u1, 5v1, 5w1 having a multiplex operation number n=1 corresponding to the master phase may be used to generate a single pulse for each phase to drive the motor 4.

[0161] Ninth embodiment The ninth embodiment will be described with reference to FIG. 44. A control device 10D may be configured as shown in FIG. 44. The control device 10D includes the multiplex number switching unit 25A in the control device 10A, the pulse calculation unit 23Z in the control device 10B, and the back electromotive force correction unit 12c in the control device 10C, all of which have been described in the above-described embodiments. In other words, the control device 10D can be applied to a configuration including all of the features of the above-described embodiments. Here, the multiplex number switching unit may be 25, the pulse calculation unit may be 23, and the back electromotive force correction unit may be "12" or "12d".

[0162] (Other embodiments) The present invention is not limited to the above-described embodiment, and can be implemented in various modifications and variations without departing from the spirit and scope of the present invention. For example, the following modifications or extensions are possible.

[0163] In the above embodiment, the power converter uses a step-down converter 5, but the present invention is not limited to this, and for example, a step-up or step-up / step-down converter 5 can be used. The converter 5 may be a non-insulated type or an insulated type. The voltage sensor 15 may be provided as necessary.

[0164] In the above-described embodiment, the number of multiplexed operations n is set to 4 or 2, but the number of multiplexed operations n may be 3 or 5 or more. In the above-described embodiment, the power switches SW1 and SW2 are exemplified as N-channel power MOSFETs, but they may be configured with other types of power switching elements.

[0165] In the above embodiment, the free wheel diodes D1 and D2 are provided in parallel to the power MOSFETs constituting the power switches SW1 and SW2 to divert the load current, but this is not limited to the above. Body diodes added to the power MOSFETs may be used instead of the free wheel diodes D1 and D2. Also, a power switch having reverse conductivity (for example, a reverse conducting IGBT (RC-IGBT)) may be used.

[0166] Similarly to the first multiplex converters 5u1, 5v1, and 5w1, the second or more m-th multiplex converters 5u2...5un, 5v2...5vn, 5w2...5wn are each provided with a zero current detection unit 22, and the inductor current I L After detecting zero, the corresponding power switch SW1 or SW2 may be turned on. It is preferable to use the on-time Ton of the power switches SW1 and SW2 calculated as in the above formula (1-1), (2-1), or (3-1).

[0167] The components of the control device 10, namely the "control unit 11" and the "pulse generating units 12, 112, 212", may be realized using hardware that combines logic circuits, or may be realized by hardware such as a microcomputer executing a program.

[0168] The control devices 10, 10A, 10B, 10C, and 10D described herein may be implemented by a special-purpose computer provided by configuring a processor and memory programmed to perform one or more functions embodied in a computer program. Alternatively, the control devices 10, 10A, 10B, 10C, and 10D described herein may be implemented by a special-purpose computer provided by configuring a processor with one or more dedicated hardware logic circuits. Alternatively, the control devices 10, 10A, 10B, 10C, and 10D described herein may be implemented by one or more special-purpose computers configured by a combination of a processor and memory programmed to perform one or more functions and a processor configured with one or more hardware logic circuits. The computer program may also be stored in a computer-readable non-transitory tangible recording medium as instructions executed by the computer.

[0169] Although the present invention has been described based on the above-mentioned embodiment, it is understood that the present invention is not limited to the embodiment or structure. The present invention also includes various modifications and modifications within the equivalent range. In addition, various combinations and forms, and other combinations and forms including one, more, or less than one element, are also within the scope and concept of the present invention. [Explanation of symbols]

[0170] In the drawings, 3 indicates a three-phase inverter (motor driving device), 4 indicates a motor, 5, 5u1, 5v1, 5w1, 5u2, 5v2, 5w2, 5u3, 5v3, 5w3, 5un, 5vn, 5wn indicate converters, 6u, 6v, 6w indicate multi-phase converters, 10 and 10C indicate control devices, 11 and 11C indicate control units, 212 indicates a pulse generating unit, 12c indicates a back EMF correction unit, 22 indicates a zero current detection unit, 37 indicates a holding unit, SW1 indicates an upper arm switch (first arm switch, second arm switch), and SW2 indicates a lower arm switch (second arm switch, first arm switch).

Claims

1. A motor drive device (3) comprising one or more n converters and driving a motor using outputs of the converters, A control unit (11, 11C) that outputs control information according to an input command value; A pulse generating unit (212) that generates pulses to be output to the n converters so as to operate the converters based on the control information output by the control unit; a back electromotive voltage correction unit (12c) that corrects the width of the pulse in accordance with a back electromotive voltage generated in the motor when the motor is driven, The back electromotive force correction unit corrects the pulse width based on amplitude and phase information of the back electromotive force.

2. When the back electromotive voltage correction unit calculates the amplitude of the back electromotive voltage, 2. The motor drive device according to claim 1, wherein the amplitude of the back electromotive voltage is calculated based on the number of revolutions of the motor.

3. 3. The motor drive device according to claim 2, wherein the back electromotive force correction unit calculates the amplitude of the back electromotive force so as to be proportional to the number of revolutions of the motor.

4. When the back electromotive force correction unit obtains amplitude and phase information of the back electromotive force, 2. The motor drive device according to claim 1, wherein the amplitude and phase information of the back electromotive voltage are determined by referring to a map stored in correspondence with the number of revolutions of the motor.

5. 5. The motor drive device according to claim 1, wherein the back electromotive force correction unit corrects the pulse width by calculating an on-time, an off-time, and a period of the pulse based on the back electromotive force.

6. The converter is configured with an upper arm switch (SW1) and a lower arm switch (SW2) to which a power supply voltage is input, and a step-down converter using an inductor (L) and a capacitor (C), When applied to a multiplexed configuration where n is 2 or more, the pulse is input to the n converters as multiple pulses; When the current command value as the command value is positive, an upper arm switch of the converter is driven, and an on time Ton1, an off time Toff1, a period T, and a phase difference Td between the multiple pulses of the upper arm switch are set based on equations (1-1) to (1-4), respectively; 6. The motor drive device according to claim 5, wherein when a current command value as the command value is negative, a lower arm switch of the converter is driven, and an on time Ton2, an off time Toff2 of the lower arm switch, as well as a period T and a phase difference Td between the multiple pulses are set based on equations (2-1) to (2-4), respectively. Here, Vin is the input voltage, Vout is the output voltage, L is the inductance of the inductor, I is the average current of each converter, and I ＬＰ denotes the peak current of the inductor current, and n: the number of multiplexed operations of the converter. [0010] [0025]

7. The motor drive device of claim 6, further comprising a synchronous rectification control unit (212) that performs synchronous rectification by turning on a second arm switch that faces a first arm switch that is in the off time during a period corresponding to the off time of the upper arm switch or the off time of the lower arm switch.

8. A motor drive device (3) having one or more n converters and driving a motor using outputs of the converters, A control unit (11, 11C) that outputs control information according to an input command value; A pulse generating unit (212) that generates pulses to be output to the n converters so as to operate the converters based on the control information output by the control unit; a back electromotive voltage correction unit (12c) that corrects the width of the pulse in accordance with a back electromotive voltage generated in the motor when the motor is driven, A power supply voltage utilization rate improvement control unit (12d) for improving a utilization rate of a power supply voltage is provided, The power supply voltage utilization rate improvement control unit is a motor drive device that, when the amplitude of the back electromotive force exceeds a predetermined threshold, sets upper and lower limits on the output voltage after correcting the back electromotive force that exceeds the threshold.

9. 6. The motor drive device according to claim 1, wherein the converter is of a step-down type, a step-up type, or a step-up / step-down type.

10. A motor drive device (3) having one or more n converters and driving a motor using outputs of the converters, A control unit (11, 11C) that outputs control information according to an input command value; A pulse generating unit (212) that generates pulses to be output to the n converters so as to operate the converters based on the control information output by the control unit; a back electromotive voltage correction unit (12c) that corrects the width of the pulse in accordance with a back electromotive voltage generated in the motor when the motor is driven, a current controller (31) which receives a current command value, performs current feedback control based on a difference current between the current command value and a phase current to be input to the motor, and outputs a manipulated variable; a phase converter (32) for converting the operation amount input from the current controller into an operation amount for each phase of the motor; a pulse calculation unit (33) that calculates parameters including an on-time, an off-time, and a period of the pulse for each phase of the motor; When the back electromotive voltage correction unit corrects the pulse width in accordance with the back electromotive voltage, A motor drive device that calculates the pulse width when the pulse calculation unit calculates the parameters for each phase of the motor.

11. A motor drive device (3) having one or more n converters and driving a motor using outputs of the converters, A control unit (11, 11C) that outputs control information according to an input command value; A pulse generating unit (212) that generates pulses to be output to the n converters so as to operate the converters based on the control information output by the control unit; a back electromotive voltage correction unit (12c) that corrects the width of the pulse in accordance with a back electromotive voltage generated in the motor when the motor is driven, When applied to a multiplexed configuration where n is 2 or more, the pulse is input to the n converters as multiple pulses; a holding unit (37) for holding a result of the back electromotive force correction unit periodically correcting the width of the multiple pulses by changing the width of the multiple pulses, A motor drive device in which, while the periodic correction is being performed, the back electromotive force correction unit corrects the width of the multiple pulses using past calculation results regarding the multiple pulses stored in the holding unit.

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