Rotating electric machine control device and rotating electric machine control method

Abstract

To provide a control device for a rotary electric machine capable of effectively suppressing a current ripple, and a control method for the rotary electric machine.SOLUTION: A control device 100 is provided for controlling switching operations of a plurality of switching elements 42UA-44WA and 42UB-44WB in inverters 40A and 40B in which DC power of a power source 30 is converted into AC power and outputted by the switching operations of the plurality of switching elements 42UA-44WA and 42UB-44WB, in such a manner that electrification sections for electrification to coils 14A and 14B of a rotary electric machine 10 to which AC power outputted from the inverters 40A and 40B is supplied, become complementary.SELECTED DRAWING: Figure 1

Description

[Technical field]

[0001] The present invention relates to a control device for a rotating electric machine and a control method for a rotating electric machine. [Background technology]

[0002] For example, a large three-phase synchronous motor may have two windings for each of the U, V, and W phases, and two inverters for generating voltages to be applied to the windings. Each of the two windings is arranged with a predetermined electrical angle in the same phase. Of the two inverters, one inverter generates a voltage whose phase differs by a predetermined electrical angle from the voltage generated by the other inverter and applies it to the windings.

[0003] Fig. 18 is an explanatory diagram showing an example of a current change during power supply in a rotating electric machine (motor) having two systems of inverters and windings. In Fig. 18, space vector modulation, which will be described later, is used to control the voltage applied to the windings of the rotating electric machine. As shown in Fig. 18, a voltage vector (system A) is input as a voltage command to the inverter of system A, resulting in the generation of an inverter current (system A) indicated as "effective" in Fig. 18. Then, a voltage vector (system B) is input as a voltage command to the inverter of system B, which is the other system of the two inverter systems, resulting in the generation of an inverter current (system B) indicated as "effective" in Fig. 18.

[0004] The inverter current (system A) and inverter current (system B) generate inverter currents (A+B) between the power supplies that supply power to each inverter and each inverter. Since the inverter current (A+B) affects the smoothing of the power supply voltage as a current ripple, a capacitor is provided in parallel with the power supply as a filter for smoothing the voltage. The current of the capacitor becomes like the capacitor current shown in FIG. 18. The capacitor current is generated in a state in which the phase is inverted from that of the inverter current (A+B).

[0005] When the amplitude of the capacitor current is large, the capacitor implemented as a filter is required to have a large capacity, but implementing a capacitor with a large capacity poses problems in terms of cost and conflicts with the need to miniaturize the circuit.

[0006] One way to reduce the capacitance of the filter capacitor is to suppress the amplitude of the capacitor current. In Fig. 18, so-called non-energized sections 120 and 122 are present, in which neither the inverter of system A nor the inverter of system B generates a voltage to be applied to the windings of the rotating electric machine. If the non-energized sections are prominent, the amplitude of the inverter current (A+B) becomes large, and as a result, the amplitude of the capacitor current also becomes large.

[0007] Patent Document 1 proposes a motor drive device that reduces current ripples by shifting the PWM phase between two systems to cancel each other out.

[0008] Furthermore, Patent Document 2 proposes a power conversion device that offsets the phase of PWM control between two systems and offsets the phase difference between the systems of voltage commands by 30 degrees to cancel out the current ripples and reduce the current ripples.

[0009] Moreover, Patent Document 3 proposes a power conversion device that reduces current ripples by shifting the charge / discharge timing of two systems through duty shift in PWM control. [Prior art documents] [Patent documents]

[0010] [Patent Document 1] JP 2001-197779 A [Patent Document 2] JP 2012-50252 A [Patent Document 3] Patent No. 4941686 Summary of the Invention [Problem to be solved by the invention]

[0011] However, in each of the inventions disclosed in Patent Documents 1 to 3, the degree of freedom in the operation of the switching elements that constitute the inverter is low, and there is a concern that it may become difficult to effectively suppress current ripples.

[0012] The present invention has been made in view of the above-mentioned problems, and an object of the present invention is to provide a rotating electric machine control device and a rotating electric machine control method that can effectively suppress current ripples. [Means for solving the problem]

[0013] In order to achieve the above-mentioned object, the control device for a rotating electric machine according to the present invention includes a plurality of power conversion circuits (40A, 40B), each of which is provided with a plurality of switching elements (42UA to 44WA, 42UB to 44WB), which convert DC power of a common DC power source into AC power and output it through the switching operation of each of the plurality of switching elements (42UA to 44WA, 42UB to 44WB) provided in each of the plurality of power conversion circuits (40A, 40B), a rotating electric machine (10) to which the AC power output from each of the plurality of power conversion circuits (40A, 40B) is supplied, and a current supply control unit (100) that controls the switching operation of each of the plurality of switching elements (42UA to 44WA, 42UB to 44WB) provided in each of the plurality of power conversion circuits (40A, 40B) so that each of the current supply sections during which the plurality of power conversion circuits (40A, 40B) supply current to windings (14A, 14B) of the rotating electric machine (10) is complementary.

[0014] With this configuration, it is possible to obtain a control device for a rotating electric machine and a control method for a rotating electric machine that can effectively suppress current ripples. [Brief description of the drawings]

[0015] [Figure 1] FIG. 2 is an electric circuit diagram of a control system for a rotating electric machine. [Diagram 2] 4 is an explanatory diagram showing an example of a current change during power supply by an inverter in the control device for a rotating electric machine according to the present embodiment. FIG. [Diagram 3]FIG. 2 is a schematic diagram showing an example of energized sections and non-energized sections in a three-system inverter. [Figure 4] FIG. 4 is an explanatory diagram showing a case where a voltage command vector is expressed as a combination of effective voltage vectors; [Diagram 5] 1 is a table showing switching modes corresponding to voltage vectors of switching elements constituting an inverter, and U-phase voltages, V-phase voltages, and W-phase voltages corresponding to the voltage vectors. [Figure 6] 1A is a schematic diagram showing an example of the transition of a voltage vector and a switching pattern in one switching period of 60-degree modulation, and FIG. 1B is a schematic diagram showing an example of the transition of a voltage vector and a switching pattern in one switching period of 120-degree modulation. [Figure 7] 1 is a table showing the ratios of instantaneous space voltage vectors in 120 degree modulation. [Figure 8] (A) is a schematic diagram showing a section for 60 degree modulation, and (B) is a table showing the ratios of instantaneous spatial voltage vectors for 60 degree modulation. [Figure 9] FIG. 11 is an explanatory diagram of a case where a current vector is used for section discrimination. [Figure 10] FIG. 4 is a schematic diagram showing a manner in which a voltage command vector is expressed by a plurality of effective voltage vectors. [Figure 11] 1A is a schematic diagram showing the case where a voltage command vector is expressed by four effective voltage vectors, and FIG. 1B is a table showing the ratios of instantaneous spatial voltage vectors when a voltage command vector is expressed by four effective voltage vectors. [Figure 12] 1A is a schematic diagram showing the case where a voltage command vector is expressed by six effective voltage vectors, and FIG. 1B is a table showing the ratios of instantaneous spatial voltage vectors when a voltage command vector is expressed by six effective voltage vectors. [Figure 13] 1A is a schematic diagram showing the case where a voltage command vector is expressed by eight effective voltage vectors, and FIG. 1B is a table showing the ratios of instantaneous spatial voltage vectors when a voltage command vector is expressed by eight effective voltage vectors. [Figure 14] 4 is a flowchart showing an example of processing by the control device for a rotating electric machine according to the present embodiment. [Figure 15] 1A is a block diagram showing a case where a rotating electric machine has multiple winding systems corresponding to multiple inverters, and FIG. 1B is a block diagram showing a case where a rotating electric machine has multiple rotating electric machines corresponding to multiple inverters, respectively. [Figure 16] FIG. 1A is an explanatory diagram showing a case where the command values ​​of the U, V, and W phases are modulated to generate command values ​​for system A and system B, and PWM control is performed in comparison with a carrier wave, and FIG. 1B is an explanatory diagram showing a case where the command values ​​of two of the U, V, and W phases are offset so that they rise or fall at the timing of the peaks or valleys of the carrier wave, and PWM control is performed in comparison with the carrier wave. [Figure 17] 1 is an explanatory diagram showing an example of a change in voltage vector at low modulation and high modulation in three-phase modulation by PWM and modulation in which a command voltage is offset by a peak or valley of a carrier wave. FIG. [Figure 18] 1 is an explanatory diagram showing an example of a current change during power supply in a rotating electric machine having two systems of inverters and windings. FIG. DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0016] [First embodiment] A first embodiment will be described below with reference to the drawings. The rotating electric machine in this embodiment is used, for example, as a vehicle power source. However, the rotating electric machine can be widely used for industrial, vehicular, home appliance, office automation equipment, amusement machines, etc. In the following embodiments, the same reference numerals are used in the drawings to designate the same or equivalent parts, and the explanations of the parts with the same reference numerals are incorporated herein.

[0017] FIG. 1 is an electric circuit diagram of a control system for a rotating electric machine 10. The rotating electric machine 10 according to this embodiment is a three-phase synchronous motor.

[0018] 1 shows two sets of windings 14A, 14B, with the winding 14A consisting of a U-phase winding 14UA, a V-phase winding 14VA, and a W-phase winding 14WA, and the winding 14B consisting of a U-phase winding 14UB, a V-phase winding 14VB, and a W-phase winding 14WB. An A-system inverter 40A and a B-system inverter 40B are provided for each of the windings 14A and 14B. The A-system inverter 40A and the B-system inverter 40B are each configured as a full-bridge circuit having the same number of upper arm switches and lower arm switches as the number of phases of the phase windings. The A-system inverter 40A generates a voltage to be applied to the windings 14A, 14B from the power supplied from the power source 30 by turning on and off switching elements 42UA, 42VA, 42WA which are upper arm switches and switching elements 44UA, 44VA, 44WA which are lower arm switches, while the B-system inverter 40B generates a voltage to be applied to the windings 14A, 14B from the power supplied from the power source 30 by turning on and off switching elements 42UB, 42VB, 42WB which are upper arm switches and switching elements 44UB, 44VB, 44WB which are lower arm switches. Each of the switching elements 42UA, 42VA, 42WA, 44UA, 44VA, 44WA, 42UB, 42VB, 42WB, 44UB, 44VB, 44WB is, for example, a FET (field effect transistor) or an IGBT (insulated gate bipolar transistor).

[0019] The A-system inverter 40A and the B-system inverter 40B are controlled by a control device 100. Based on a command signal input from the outside and a current supply state, an output voltage calculation unit 110A calculates a voltage command vector V A The output voltage calculation unit 110B calculates the voltage command vector V B Then, the modulation unit 112 generates the input voltage command vector V A , V B From this, it generates gate signals to be applied to the gates of the switching elements constituting each of the A-system inverter 40A and the B-system inverter 40B.

[0020] As the A-system inverter 40A and the B-system inverter 40B operate, a power supply current 50, an inverter current 54, an A-system inverter current 54A, and a B-system inverter current 54B are generated in the circuit shown in FIG. 1, and a capacitor current 52 flows in the capacitor 20 that smoothes the power supply voltage.

[0021] Fig. 2 is an explanatory diagram showing an example of a current change when power is supplied by inverters 40A, 40B in a control device 100 for a rotating electric machine according to this embodiment, which includes two systems of inverters and windings. In Fig. 2, each of the inverters 40A, 40B is controlled by space vector modulation. In space vector modulation, the switching state of each of the switching elements 42UA-44WA, 42UB-44WB constituting each of the inverters 40A, 40B is expressed by eight types of vectors, and a voltage command vector, which is a command value, is expressed by a composition (sum) of the eight types of vectors.

[0022] For example, the switching modes of the switching elements 42UA to 44WA constituting the inverter 40A, which is a three-phase inverter, are 3 = 8 possible switching modes. Among them, in the switching mode in which each of the switching elements 42UA, 42VA, and 42WA, which are the upper arm switches, is turned on, no voltage is applied to the winding 14A of the rotating electric machine 10. Therefore, the vector expressing this switching mode is defined as a reactive voltage vector V7.

[0023] Similarly, in the switching state in which each of the switching elements 44UA, 44VA, and 44WA, which are the lower arm switches, is turned on, no voltage is applied to the winding 14A of the rotating motor 10, so the vector representing such a switching state is regarded as a reactive voltage vector V0.

[0024] Other vectors expressing switching modes in which voltages are applied to winding 14A of rotating electric machine 10 are effective voltage vectors V1, V2, V3, V4, V5, and V6. As shown in Fig. 5, which will be described later, the switching modes indicated by each of effective voltage vectors V1 to V6 do not include a case in which an upper arm switch and a lower arm switch of the same phase are simultaneously turned on. Therefore, so-called through current is not generated in switching elements 42UA to 44WA constituting inverter 40A by space vector modulation.

[0025] The active voltage vectors V1 to V6 and the reactive voltage vectors V0, V7 are similarly applied to the inverter 40B, and no shoot-through current occurs in the switching elements 42UB to 44WB that configure the inverter 40B, which is also the case with the inverter 40B.

[0026] In FIG. 2, the inverter current -I is the inverter current (A system) corresponding to the effective voltage vectors V2 and V6 enclosed by the dashed lines in the voltage vector (A system). W , -I V The voltage vector (system A) becomes a reactive voltage vector V0 following the effective voltage vectors V2 and V6, and a non-current-carrying section is created in which the inverter current (system A) is ineffective.

[0027] In the voltage vector (B system), when the voltage vector (A system) becomes the effective voltage vectors V2 and V6 as described above, it becomes the reactive voltage vector V7, and as a result, a non-energized section occurs in which the inverter current (B system) is invalid. The voltage vector (B system) becomes the effective voltage vectors V2 and V6 shown in the dashed line box following the reactive voltage vector V7, and the inverter current (B system) corresponds to the effective voltage vectors V2 and V6 and becomes the inverter current -I W , -I V In this way, a current-carrying section is formed in which each of the current-carrying sections occurs.

[0028] 2, the conduction section of the inverter current (system A) and the conduction section of the inverter current (system B) appear complementary to each other. Also, the non-conducting section of the inverter current (system A) and the non-conducting section of the inverter current (system B) appear complementary to each other. As a result, the amplitude (fluctuation) of the inverter current (A+B) is suppressed, and the amplitude (fluctuation) of the capacitor current 52 of the capacitor 20 that occurs in response to the inverter current (A+B) is also suppressed.

[0029] When the current-carrying and non-current-carrying sections appear complementarily, the current-carrying sections caused by each of inverters 40A, 40B and the non-current-carrying sections caused by each of inverters 40A, 40B being of approximately the same length can also be effective in suppressing the amplitude of capacitor current 52.

[0030] 1 and 2 show an example in which inverters 40A, 40B and windings 14A, 14B are provided, but as shown in FIG. 3, a configuration including inverters and windings of three or more systems, for example, inverters and windings of systems A, B, and C, may be provided. In such a configuration, the energized section ("active" in FIG. 3) and the non-energized section ("inactive" in FIG. 3) are made to appear complementarily, as in the case of the two systems A and B. In addition, the relationship between the energized section and the non-energized section in a configuration including three systems of inverters and windings as shown in FIG. 3 is length of the energized section: length of the non-energized section = 1 / 3: (1-1 / 3) = 1:2. In a configuration having inverters and windings of N systems (N is a natural number of 2 or more), the relationship between the energized section and the non-energized section is generalized as follows. The following relationship holds without contradiction even when inverters 40A, 40B and windings 14A, 14B are provided, namely, length of current-carrying section: length of non-current-carrying section = 1 / 2: (1-1 / 2) = 1:1. Length of energized section: Length of non-energized section = 1 / N:(1-1 / N)

[0031] FIG. 4 is an explanatory diagram showing a case where the voltage command vector 60 is expressed by a synthesis of effective voltage vectors. In space vector modulation, there are 120-degree modulation in which the voltage command vector 60 is expressed by a synthesis of effective voltage vectors whose phase difference is a natural number multiple of 120 degrees, and 60-degree modulation in which the voltage command vector 60 is expressed by a synthesis of effective voltage vectors whose phase difference is a natural number multiple of 60 degrees. FIG. 4 mainly describes the 120-degree modulation. In the 120-degree modulation, the quadrants of a virtual space in which the voltage vector exists are called sections A, B, C, D, E, and F. When the voltage command vector 60 is expressed as a synthesis of two effective voltage vectors, it is determined by which quadrant (section) the voltage command vector 60 exists in that the voltage command vector 60 is expressed by a synthesis of which effective voltage vectors. For example, as shown in FIG. 4, when the voltage command vector 60 exists in section A, the voltage command vector 60 is expressed by a synthesis of an effective voltage vector V2 and an effective voltage vector V6. In addition, in the 120-degree modulation, the reactive voltage vector becomes a reactive voltage vector V7. The voltage command vector 60 may be expressed as a combination of the effective voltage vector V2 and the effective voltage vector V6.

[0032] When the relationship between the voltage command vector 60 and the effective voltage vector in the 120-degree modulation is generalized, the voltage command vector 60 is expressed using the effective voltage vectors present in the sections adjacent to the section in which the voltage command vector 60 exists. In other words, the phase difference D 120 But 30 degrees <D 120 The voltage command vector 60 is expressed by the composition of two effective voltage vectors with angles <90 degrees.

[0033] For reference, Fig. 4 shows the composition of the effective voltage vector of the voltage command vector 60 in 60-degree modulation. In 60-degree modulation, the voltage command vector 60 is expressed as a composition of an effective voltage vector V1 and an effective voltage vector V2. Note that α and β in Fig. 4 are α-axis and β-axis that define the two-phase fixed coordinate system of the rotating electric machine 10 when three-phase is converted to two-phase.

[0034] 4, for the same voltage command vector 60, in the 120-degree modulation, the magnitude of the effective voltage vector that combines the voltage command vector 60 is larger than in the 60-degree modulation, so the magnitude of the expressible voltage command vector 60 is more limited than in the 60-degree modulation. In this embodiment, as will be described later, the modulation rate that is proportional to the magnitude of the voltage command vector is set to a predetermined threshold value M a In principle, 120 degree modulation is used when the a In the above cases, 60 degree modulation is used as a general rule.

[0035] In addition, since the phase difference of the effective voltage vector used in 120-degree modulation is larger than that in 60-degree modulation, 120-degree modulation is slightly more advantageous than 60-degree modulation when expanding the width of the energized section.

[0036] FIG. 5 shows switching behaviors of switching elements constituting inverters 40A and 40B corresponding to voltage vectors V0 to V7, and U-phase voltages v u , V phase voltage v v , W phase voltage v w 5. In Fig. 5, items indicated with "ON / off" etc. indicate the operation state of the upper arm switch on the left and the lower arm switch on the right. Also, Ed / 2 corresponds to the power supply voltage VDC which is the voltage on the positive side of the power supply 30, and -Ed / 2 corresponds to 0 which is the voltage on the negative side of the power supply 30.

[0037] The v shown in the bottom row of Figure 5 αβ are the voltage values ​​of each voltage vector in the two-phase fixed coordinate system αβ of the rotating electric machine 10. αβ is defined by the following equation (1), and as a result, v αβ The values ​​of are as shown in Fig. 5. As will be described later, formula (1) is also used when describing a space vector such as a voltage command vector.

[0038] JPEG0007608897000001.jpg30161

[0039] Fig. 6(A) is a schematic diagram showing an example of the transition of the voltage vector and the switching pattern in one switching period of 60-degree modulation, and Fig. 6(B) is a schematic diagram showing an example of the transition of the voltage vector and the switching pattern in one switching period of 120-degree modulation. As shown in Fig. 6, the switching pattern may be simpler in the case of 60-degree modulation.

[0040] Figure 7 is a table showing the ratio of instantaneous space voltage vectors in 120-degree modulation. As shown in Figure 7, the voltage command vector V ref is expressed as a sum with the ratio shown in Fig. 7 as a coefficient. As an example, when the voltage command vector exists in section A as shown in Fig. 5, it is expressed by the following formula (2).

[0041] JPEG0007608897000002.jpg19127

[0042] Each ratio shown in FIG. 7 is calculated as follows. In the 120-degree modulation, as shown in FIG. 4, six sections A to F are partitioned for each of six types of effective voltage vectors, each of which has a phase difference of 60 degrees. As an example, the voltage command vector V ref In this case, the U-phase voltage v u , V phase voltage v v , W phase voltage v w Using the voltage command vector V ref is expressed as the following equation (3).

[0043] JPEG0007608897000003.jpg45164

[0044] On the other hand, the voltage command vector V ref is expressed as the following formula (4): In formula (4), s and t are coefficients.

[0045] JPEG0007608897000004.jpg59154

[0046] By comparing the right-hand sides of the above equations (3) and (4), the coefficients s and t are calculated, and the voltage command vector V ref can be expressed as the above formula (2). Similarly, the voltage command vectors V ref The coefficients s and t are also calculated for the reactive voltage vector V0. For example, the coefficient of the reactive voltage vector V0 is 1+(3v i / E d ) (i=u, v, w), the coefficient of the reactive voltage vector V7 is, for example, 1-(3v i / E d ) (i=u, v, w), respectively.

[0047] Fig. 8(A) is a schematic diagram showing a section in 60-degree modulation, and Fig. 8(B) is a table showing the ratio of instantaneous spatial voltage vectors in 60-degree modulation. As shown in Fig. 8(A), in 60-degree modulation, the area between each of the effective voltage vectors V1 to V6 is defined as a corner section.

[0048] In the case of 60-degree modulation, as in the case of 120-degree modulation, the voltage command vector V ref The coefficients of the effective voltage vectors constituting the above vectors are calculated as shown in Fig. 8. Note that M in (1-M) / 2, which is the coefficient of the reactive voltage vectors V0 and V7, is a modulation factor command, and M≦1.0. A value obtained by subtracting the modulation factor command M related to the voltage application to the windings 14A and 14B of the rotating electric machine 10 from 1 is an invalid value that is not related to the voltage application to the windings 14A and 14B of the rotating electric machine 10. This invalid value is halved and distributed equally to each of the reactive voltage vectors V0 and V7, so that the coefficients (ratios) of the reactive voltage vectors V0 and V7 are made equal.

[0049] In Fig. 8(B), the reactive voltage vectors V0 and V7 are set to an equal ratio, but they do not have to be set to an equal ratio. The fact that the reactive voltage vectors V0 and V7 may or may not be set to an equal ratio also applies to Figs. 11(B), 12(B), and 13(B) described later.

[0050] Fig. 9 is an explanatory diagram of a case where a current vector is used for section discrimination. Fig. 9 shows sections 1 to 6 (hereinafter abbreviated as "S1 to S6" in the drawings) assuming 60-degree modulation, and in section 1, a voltage command vector 60 and a current vector 66 having a phase difference of Δθ with respect to the voltage command vector 60 are respectively shown. The current vector 66 is calculated based on the currents of the windings 14A, 14B detected by the current sensors 46, 48, the induced voltages generated in the windings 14A, 14B, and, if the rotating electric machine 10 is a brushless motor or the like with a rotor made of a permanent magnet, the position of the rotor detected by a Hall element or the like.

[0051] In section discrimination of the current vector 66, the switching state in the section to which the current vector 66 belongs is adopted, and the length of the adopted switching state is determined by the voltage command vector 60. In other words, this means that the effective voltage vector to be used is determined from the current vector 66, and each output time of the effective voltage vector to be used is determined from the voltage command vector 60.

[0052] In FIG. 9, both the voltage command vector 60 and the current vector 66 exist in section 1, but in section 1, a switching state is adopted based on the coefficients enclosed in a thick frame on the second line of FIG. 8(B) and the effective voltage vectors V1 and V2 corresponding to these constants, and the length of the adopted switching state is determined by the voltage command vector 60.

[0053] The matter of identifying the section by the current vector 66, determining the length of the switching state by the voltage command vector 60, and controlling the inverters 40A, 40B corresponds to expressing a vector having the same phase as the current vector 66 and the same amplitude as the voltage command vector 60 as the sum of multiple effective voltage vectors V1 to V6, and controlling the switching operation of each of the multiple switching elements 42UA to 44WB provided in each of the inverters 40A, 40B based on the multiple effective voltage vectors V1 to V6 and the coefficients of the multiple effective voltage vectors V1 to V6 that constitute the sum.

[0054] By determining the section using the current vector 66, an effect similar to retardation control, which delays the timing of applying voltage to the windings of a rotating electric machine, is produced in PWM control. As a result, the amplitude of the capacitor current can be suppressed, thereby reducing the so-called current ripple.

[0055] 9, when the section is determined by the current vector 66, the section may be determined by using either the current vector 66 or the voltage command vector 60 depending on the magnitude of the phase difference Δθ with the voltage command vector 60. As an example, when the phase difference Δθ is less than a predetermined angle (e.g., 5 degrees), the section is determined by the voltage command vector 60 rather than the current vector 66.

[0056] Furthermore, when section discrimination is performed using the current vector 66, if Δθ is, for example, 30 degrees or more, section discrimination may be performed using a phase shifted by 30 degrees from the voltage command vector 60.

[0057] 10 is a schematic diagram showing a mode in which a voltage command vector 60 is expressed by a plurality of effective voltage vectors V1 to V6. In space vector modulation, it is common to express the voltage command vector 60 using two effective voltage vectors, and in particular, in the conventional technology, 60-degree modulation using two effective voltage vectors is adopted. However, in the 60-degree modulation, the phase difference of the effective voltage vector used is relatively narrow at 60 degrees, and it is not suitable for expanding the energization section. However, in the case of 120-degree modulation, in which the phase difference of the effective voltage vector used is large, it is possible to increase the energization width compared to the 60-degree modulation, even when two effective voltage vectors are used.

[0058] In addition, the current supply section can be extended by using three or more effective voltage vectors. In Fig. 10, the voltage command vector 60 is expressed by using three effective voltage vectors, four effective voltage vectors, six effective voltage vectors, and eight effective voltage vectors, respectively. Details of expressing the voltage command vector 60 by using four effective voltage vectors, six effective voltage vectors, and eight effective voltage vectors, respectively, will be described later with reference to Figs. 11, 12, and 13.

[0059] FIG. 11(A) is a schematic diagram showing the case where a voltage command vector 60 is expressed by four effective voltage vectors, and FIG. 11(B) is a table showing the ratios of instantaneous spatial voltage vectors when the voltage command vector 60 is expressed by four effective voltage vectors.

[0060] As shown in FIG. 11A, when the voltage command vector 60 exists in section 6, in addition to expressing the voltage command vector 60 using effective voltage vectors V1 and V6 that sandwich section 6, the voltage command vector 60 is also expressed using effective voltage vectors V2 and V5.

[0061] The result obtained by subtracting a value unrelated to the voltage application to the windings 14A, 14B of the rotating electric machine 10 from the modulation factor command M, which is a parameter for adjusting how much of the current conduction section is output to the windings 14A, 14B of the rotating electric machine 10, can be used as a guide for expanding the current conduction section by the difference when the current conduction section determined from the voltage command vector is shorter than the expected current conduction section, and becomes the ratio (coefficient) of the effective voltage vector when expanding the current conduction section. The value unrelated to the voltage application to the windings 14A, 14B of the rotating electric machine 10 is expressed as follows when the voltage command vector 60 is expressed using the effective voltage vectors V2, V5 in addition to expressing the voltage command vector 60 using the effective voltage vectors V1, V6 as shown in FIG. 11(A): u -v v ) / E d It becomes.

[0062] This coefficient is halved and distributed equally to both ends of the area enclosed by the thick line in FIG. 11(B), thereby obtaining the coefficient of the effective voltage vector that expands the current-carrying section.

[0063] FIG. 12(A) is a schematic diagram showing the case where the voltage command vector 60 is expressed by six effective voltage vectors, and FIG. 12(B) is a table showing the ratios of instantaneous spatial voltage vectors when the voltage command vector 60 is expressed by six effective voltage vectors.

[0064] As shown in FIG. 12A, when the voltage command vector 60 exists in section 6, in addition to expressing the voltage command vector 60 using effective voltage vectors V1 and V6 that sandwich section 6, the voltage command vector 60 is expressed using effective voltage vectors V3 and V4, and further effective voltage vectors V1 and V6 in a superimposed manner.

[0065] In FIG. 12(B), the modulation rate command M and (v u -v v ) / E d The differences between these and the effective voltage vectors V1, V6, etc. are expressed as four effective voltage vectors, and are then divided into four and added to the coefficients shown in FIG. 8(B).

[0066] FIG. 13(A) is a schematic diagram showing the case where the voltage command vector 60 is expressed by eight effective voltage vectors, and FIG. 13(B) is a table showing the ratios of instantaneous spatial voltage vectors when the voltage command vector 60 is expressed by eight effective voltage vectors.

[0067] As shown in FIG. 13A, when the voltage command vector 60 exists in section 6, in addition to expressing the voltage command vector 60 using effective voltage vectors V1 and V6 that sandwich section 6, the voltage command vector 60 is expressed using effective voltage vectors V2, V3, V4, and V5, and further effective voltage vectors V1 and V6 in a superimposed manner.

[0068] In FIG. 13(B), the modulation rate command M and (v u -v v ) / Ed The differences between these are expressed by six effective voltage vectors in addition to the effective voltage vectors V1 and V6, and are then divided into six equal parts and added to the coefficients shown in FIG. 8(B).

[0069] 14 is a flowchart showing an example of processing by the control device 100 for a rotating electric machine according to this embodiment. In step 130, the current waveforms of the U, V, and W phases that drive the rotating electric machine 10 are monitored, and sensor information is detected to obtain rotation speed information, which is the actual rotation speed of the rotating electric machine 10.

[0070] In step 132, a voltage command vector 60 is generated. As an example, in step 132, the analog current values ​​of the U, V, and W phases are converted into digital current values ​​I u , I v , I w After converting to, the two-phase current value I α , I β Then, convert it to the current value I α , I β The current value I is calculated by converting the coordinate system from fixed coordinates to rotating coordinates. d , I q The voltage value V is calculated from the current value by correcting the deviation from the command value using PI control. d , V q Furthermore, the voltage value V d , V q The coordinate system of V is obtained by inverse coordinate transformation from rotating coordinates to fixed coordinates. α , V β is subjected to space vector transformation to generate a voltage command vector 60.

[0071] Steps 134, 136, and 140 are steps for calculating the time ratio (coefficient) of the voltage vector. In step 134, the modulation factor calculated based on the magnitude of the voltage command vector 60 generated in step 132 is compared with a predetermined threshold M a It is determined whether the modulation factor is less than the predetermined threshold value M a The specifics will be determined through experiments using actual equipment, etc.

[0072] In step 134, the modulation factor is set to a predetermined threshold M a If it is determined that the modulation rate is less than the predetermined threshold value M a If so, the procedure proceeds to step 140.

[0073] In step 136, 120-degree modulation is performed. In the 120-degree modulation, the voltage command vector 60 is expressed as an effective voltage vector based on the ratio of the instantaneous space voltage vectors shown in FIG.

[0074] In step 138, based on the result obtained in step 136, a gate signal to be applied to the gate of the switching element constituting each of inverters 40A, 40B is output, and the process ends.

[0075] In step 140, 60-degree modulation is performed. In the 60-degree modulation, the voltage command vector 60 is expressed as an effective voltage vector based on the ratio of the instantaneous space voltage vectors shown in, for example, Figures 8(B), 10(B), 11(B), and 12(B). In step 140, in addition to the 60-degree modulation, a PWM control based on triangular wave comparison, which will be described later, may be performed.

[0076] In step 142, based on the result obtained in step 140, a gate signal to be applied to the gate of the switching element constituting each of inverters 40A, 40B is output, and the process ends.

[0077] Figure 15(A) is a block diagram showing a case where a rotating electric machine 10 has multiple winding systems corresponding to multiple inverters 40A, 40B, respectively, and Figure 15(B) is a block diagram showing a case where a rotating electric machine 10 has multiple rotating electric machines 10A, 10B corresponding to multiple inverters 40A, 40B, respectively.

[0078] In the example shown in FIG. 15(A), the voltages generated by the multiple inverters 40A, 40B are applied to the windings of one rotating electric machine 10 corresponding to each of the multiple inverters 40A, 40B, so that a large output can be expected and smooth rotation can be obtained. The rotating electric machine 10 shown in FIG. 15(A) is suitable for applications requiring redundant design and large output, such as electric power steering and the main engine of an electric vehicle. In addition, the switching patterns of each system do not necessarily have to follow the same effective voltage vector. For example, a method of complementarily driving the A system using 60-degree modulation and the B system using 120-degree modulation may be adopted.

[0079] In the example shown in FIG. 15(B), the voltages generated by the multiple inverters 40A, 40B are applied to the windings of the rotating electric machines 10A, 10B corresponding to the multiple inverters 40A, 40B, respectively, so that the multiple rotating electric machines 10A, 10B can be controlled simultaneously. The rotating electric machines 10A, 10B shown in FIG. 15(B) are suitable for use in railroad cars equipped with multiple rotating electric machines, and industrial equipment such as fan and pump drive systems. The lengths of the energized sections of the windings of the rotating electric machines 10A, 10B do not necessarily have to be the same. For example, the energized section of the windings of the rotating electric machine 10A may be 80% of the entire energized section, whereas the energized section of the windings of the rotating electric machine 10B may be 20% of the entire energized section. Or, the energized section of the windings of the rotating electric machine 10A may be 20% of the entire energized section, whereas the energized section of the windings of the rotating electric machine 10B may be 80% of the entire energized section.

[0080] As described above, the control device 100 for the rotating electric machine according to this embodiment is configured to adjust the modulation factor to a predetermined threshold value M a If the modulation factor is less than the threshold value M a In the above cases, a modulation means other than the 120-degree modulation is used to expand the energization period by inverter 40A and the energization period by inverter 40B, and control is performed so that each energization period appears complementarily. One example of modulation other than the 120-degree modulation is 60-degree modulation, and more specifically, 60-degree modulation using two or more effective voltage vectors.

[0081] By making the current conduction section by inverter 40A and the current conduction section by inverter 40B appear complementary to each other, the amplitude of the inverter current is suppressed, and further the amplitude of the capacitor current flowing through capacitor 20, whose amplitude changes in response to the inverter current, is suppressed, thereby suppressing current ripple in the circuit including power supply 30 and inverters 40A and 40B. As a result, the capacitance of capacitor 20, which smoothes the voltage of the circuit, can be suppressed.

[0082] [Second embodiment] The second embodiment will be described below with reference to the drawings. In this embodiment, the capacitor current 52 is suppressed by a method other than space vector modulation. In this embodiment, the same components as those in the first embodiment are denoted by the same reference numerals, and detailed description of the components will be omitted.

[0083] FIG. 16(A) is an explanatory diagram showing a case where the command values ​​of the U, V, and W phases are each modulated (offset) to generate a command value for system A including inverter 40A and a command value for system B including inverter 40B, and PWM control is performed in comparison with carrier wave 82. FIG. 16(B) is an explanatory diagram showing a case where the command values ​​of two of the U, V, and W phases are offset so that they rise or fall at the timing of the peaks or valleys of carrier waves 82A and 82B, and PWM control is performed in comparison with carrier waves 82A and 82B.

[0084] In the example shown in FIG. 16(A), the U-phase command value is set to command voltages 80UA and 80UB, the V-phase command value is set to command voltages 80VA and 80VB, and the W-phase command value is set to command voltages 80WA and 80WB. Each command value is compared with a carrier wave 82 to generate a PWM control signal such as voltage vectors V0 to V7. In the example shown in FIG. 16(A), for example, the command values ​​that were originally one for each of the U, V, and W phases may be offset in the positive or negative direction according to the voltage command vector 60 or the current vector 66 to command voltages 80UA and 80UB, command voltages 80VA and 80VB, and command voltages 80WA and 80WB. In FIG. 16(A), the command value of the A system is offset in the negative direction and the command value of the B system is offset in the positive direction with respect to the original command value, but the command value of the B system may be offset in the negative direction and the command value of the A system may be offset in the positive direction.

[0085] By offsetting the command values ​​in this way, it becomes possible to shift the timing of generation of reactive voltages such as those shown as V0 and V7 between system A and system B while maintaining the line potential difference of the rotary electric machine 10.

[0086] In the example shown in Fig. 16(B), a disturbance is applied to the command voltage at the peaks or valleys of carrier waves 82A and 82B to offset it, and a desired voltage command vector 60 is realized in one electrical angle period. Carrier wave 82A is for system A including inverter 40A, and carrier wave 82B is for system B including inverter 40B. Carrier waves 82A and 82B have a predetermined phase difference so that a current conduction section by system A and a current conduction section by system B appear complementarily. Furthermore, applying a disturbance to the command voltage at the peaks or valleys of carrier waves 82A and 82B to offset it enables a kind of voltage compensation, and the current conduction section can be extended.

[0087] In Fig. 16(B), when carrier wave 82A becomes a valley, command voltage 84V is offset in the negative direction to become command voltage 86V, and command voltage 84W is offset in the positive direction to become command voltage 86W. Then, at time t1 when carrier wave 82A becomes a crest, command voltage 84V is offset in the positive direction to become command voltage 86V, and command voltage 84W is offset in the negative direction to become command voltage 86W. In the example shown in Fig. 16(B), since two-phase modulation of the V phase and W phase is performed, command voltage 86U, which is the command value for the U phase, does not change.

[0088] The voltage vector of system A in FIG. 16(B) switches from V5 to V3 as command voltage 86W falls and command voltage 86V rises at time t1 when carrier wave 82A reaches a peak.

[0089] In addition, in Fig. 16(B), if command voltage 86W falls and command voltage 86V rises at time t2 when carrier wave 82B peaks, the voltage vector of system B changes as shown by system B in Fig. 16(B). As a result, the current conduction section of system A and the current conduction section of system B appear complementary to each other.

[0090] Fig. 17 is an explanatory diagram showing an example of changes in voltage vectors at low and high modulation in three-phase modulation by PWM and modulation in which the command voltage is offset by the crest or trough of the carrier wave shown in Fig. 16(B). In three-phase modulation by PWM, the voltage amplitude indicated by the voltage vector becomes large both in the case of low modulation where the difference between the command voltages 88U, 88V, and 88W, which are the modulation rate commands, is relatively small, and in the case of high modulation where the difference between the command voltages 92U, 92V, and 92W, which are the modulation rate commands, is large, so that the effective voltage vector section is larger in the high modulation region than in the low modulation region.

[0091] In modulation that offsets the command voltage, which is a modulation rate command, at the peak or valley of the carrier wave 90, the effective voltage can be expanded more than with three-phase modulation by PWM while maintaining the amplitude of the voltage vector on average. For example, in the case of low modulation, in three-phase modulation by PWM, the non-conducting section by reactive voltage vectors V0 and V7 is longer than the conducting section by effective voltage vectors V2 and V3, but in modulation that offsets the command voltage, the conducting section by effective voltage vectors V2 and V4 is longer than the non-conducting section by reactive voltage vector V7.

[0092] Furthermore, in the case of high modulation, non-conducting intervals due to reactive voltage vectors V0 and V7 exist in three-phase modulation using PWM, but in modulation that offsets the command voltage, non-conducting intervals due to reactive voltage vectors V0 and V7 do not exist, and only conducting intervals due to effective voltage vectors V2, V3, and V4 exist. Therefore, in modulation that offsets the command voltage, even if triangular wave comparison modulation is used, it is possible to expect a voltage vector output similar to that of 120-degree modulation in space vector modulation or modulation using three or more effective voltage vectors.

[0093] As described above, according to this embodiment, the amplitude of the capacitor current 52 flowing through the capacitor 20 is suppressed by modulation that offsets the command voltage, which is a modulation rate command, at the time of the peak or valley of the carrier wave, thereby suppressing the current ripple in the circuit including the power supply 30 and the inverters 40A and 40B. As a result, the capacitance of the capacitor 20 that smoothes the voltage of the circuit can be suppressed. [Explanation of symbols]

[0094] 10, 10A, 10B rotating electric machine, 14A, 14B winding, 14UA, 14UB U-phase winding, 14VA, 14VB V-phase winding, 14WA, 14WB W-phase winding, 20 capacitor, 30 power supply, 40A, 40B inverter, 42UA, 42VA, 42WA, 44UA, 44VA, 44WA, 42UB, 42VB, 42WB, 44UB, 44VB, 44WB switching element, 46, 48 current sensor, 50 power supply current, 52 capacitor current, 54 inverter current, 54A A-system inverter current, 54B B-system inverter current, 60 voltage command vector, 66 current vector, 80UA command voltage, 80VA command voltage, 80WA command voltage, 100 control device, 110A, 110B output voltage calculation unit, 112 Modulation section, V0, V7 reactive voltage vectors, V1, V2, V3, V4, V5, V6 effective voltage vectors

Claims

1. a plurality of power conversion circuits (40A, 40B), each of which is provided with a plurality of switching elements (42UA to 44WA, 42UB to 44WB), which convert DC power of a common DC power source into AC power and output the AC power by switching operations of the respective switching elements (42UA to 44WA, 42UB to 44WB); a rotating electric machine (10) to which AC power output from each of the plurality of power conversion circuits (40A, 40B) is supplied; an energization control unit (100) that controls switching operations of a plurality of switching elements (42UA to 44WA, 42UB to 44WB) provided in each of the plurality of power conversion circuits (40A, 40B) so that each of energization sections in which the plurality of power conversion circuits (40A, 40B) energize the windings (14A, 14B) of the rotating electric machine (10) is complementary; Including, A relationship between a current-carrying section of each of the plurality of power conversion circuits (40A, 40B) and a current-deactivating section of each of the plurality of power conversion circuits (40A, 40B) to the windings (14A, 14B) of the rotating electric machine (10) is such that, when there are N systems of the plurality of power conversion circuits (40A, 40B), the length of the current-carrying section: the length of the current-deactivating section = 1 / N: (1 - 1 / N). A control device for a rotating electric machine.

2. The control device for a rotating electric machine according to claim 1, wherein the current control unit (100) controls the switching operation of each of a plurality of switching elements (42UA to 44WA, 42UB to 44WB) provided in each of the plurality of power conversion circuits (40A, 40B) based on the coefficients of each of the plurality of effective voltage vectors (V1 to V6) and the plurality of effective voltage vectors (V1 to V6) in a voltage command vector (60) expressed as the sum of a plurality of effective voltage vectors (V1 to V6) indicating each of the operation patterns of the plurality of switching elements (42UA to 44WA, 42UB to 44WB).

3. The rotating electric machine further includes a current detection unit (46, 48) for detecting a current in a winding (14A, 14B) of the rotating electric machine (10), The control device for a rotating electric machine according to claim 2, wherein the current control unit (100) expresses a vector having the same phase as a current vector 66 of the windings (14A, 14B) calculated based on the current detected by the current detection unit (46, 48) and having the same amplitude as a voltage command vector 60 as a sum of the effective voltage vectors (V1 to V6), and controls the switching operation of each of a plurality of switching elements (42UA to 44WA, 42UB to 44WB) provided in each of the plurality of power conversion circuits (40A, 40B) based on the plurality of effective voltage vectors (V1 to V6) in the sum and the coefficients of each of the plurality of effective voltage vectors (V1 to V6).

4. 4. The control device for a rotating electric machine according to claim 2, wherein the sum of the plurality of effective voltage vectors (V1 to V6) is a sum of three or more effective voltage vectors (V1 to V6).

5. 4. The control device for a rotating electric machine according to claim 2, wherein the sum of the plurality of effective voltage vectors (V1 to V6) is a sum of a plurality of effective voltage vectors (V1 to V6) whose phase difference is a natural number multiple of 120 degrees.

6. A control device for a rotating electric machine according to any one of claims 2 to 5, wherein, when a modulation rate indicated by the voltage command vector (60) is equal to or greater than a predetermined threshold, the sum of the multiple effective voltage vectors (V1 to V6) is a sum of multiple effective voltage vectors (V1 to V6) whose phase difference is a natural number multiple of 60 degrees, and, when the modulation rate is less than the predetermined threshold, the sum of the multiple effective voltage vectors (V1 to V6) is a sum of multiple effective voltage vectors (V1 to V6) whose phase difference is a natural number multiple of 120 degrees.

7. The rotating electric machine (10) includes a plurality of windings (14A, 14B) corresponding to the plurality of power conversion circuits (40A, 40B), 7. The control device for a rotating electric machine according to claim 1, wherein the plurality of power conversion circuits (40A, 40B) energize the corresponding windings (14A, 14B), respectively.

8. a plurality of rotating electric machines (10A, 10B) corresponding to the plurality of power conversion circuits (40A, 40B), The control device for a rotating electric machine according to any one of claims 1 to 6, wherein the plurality of power conversion circuits (40A, 40B) energize windings of the corresponding rotating electric machines (10A, 10B).

9. A method for controlling a rotating electric machine (10) to which AC power is supplied from a plurality of power conversion circuits (40A, 40B), each of which is provided with a plurality of switching elements (42UA to 44WA, 42UB to 44WB) and which converts DC power of a common DC power source into AC power and outputs the AC power by switching operations of the respective switching elements (42UA to 44WA, 42UB to 44WB), comprising: controlling switching operations of a plurality of switching elements (42UA to 44WA, 42UB to 44WB) provided in each of the plurality of power conversion circuits (40A, 40B) so that current supply sections in which the plurality of power conversion circuits (40A, 40B) supply current to the windings (14A, 14B) of the rotating electric machine (10) are complementary to each other; A relationship between a current-carrying section of each of the plurality of power conversion circuits (40A, 40B) and a current-deactivating section of each of the plurality of power conversion circuits (40A, 40B) to the windings (14A, 14B) of the rotating electric machine (10) is such that, when there are N systems of the plurality of power conversion circuits (40A, 40B), the length of the current-carrying section: the length of the current-deactivating section = 1 / N: (1 - 1 / N). A method for controlling a rotating electric machine.

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