MIMO Radar System

Abstract

To provide a MIMO radar system having an antenna array easily configured and capable of determining relative speed and a positioning angle of a radar object with high accuracy and a large range of uniqueness in short measurement time.SOLUTION: A MIMO radar system includes a control evaluation device, a transmission array having a plurality of transmission antennas spaced apart from one another in an angular resolution direction (y, z), and a reception array having a plurality of reception antennas spaced apart from one another in the angular resolution direction, where an antenna interval in each of the transmission array and the reception array is below a Nyquist limit for unambiguous angle measurement. A combined antenna interval of the transmission array and the reception array is configured to exceed the Nyquist limit.SELECTED DRAWING: Figure 5

Description

[Technical field]

[0001] The present invention relates to a MIMO radar system, in particular to a MIMO radar system for vehicles. [Background technology]

[0002] Radar systems are being used in vehicles to detect the traffic environment over a wider range and provide information on the distance, relative speed and direction angle of the vehicle or of located objects such as obstacles to one or more safety or comfort functions that relieve the driver's burden in steering the vehicle or completely or partially replace the human driver. In this context, multiple input-multiple output (MIMO) systems are increasingly being used, in which multiple transmit and receive antennas are used to achieve high angular resolution.

[0003] WO 2018 / 076005 mentions different types of MIMO radar systems. The transmitters and / or receivers can be located at different positions. Virtual channels can be created using mutually orthogonal codes. For channel separation, Time Division Multiple Access (TDMA) and / or Frequency Division Multiple Access (FDMA) can be used.

[0004] From DE 10 2014 212 284 A1 a MIMO radar measurement method is known in which a transmission signal is frequency modulated in a ramp manner with a modulation model in which ramp trains are assigned to different transmission switching states, which differ in terms of the selection of antenna elements used for transmission, and which overlap each other in time. A transmission switching state is in turn assigned to a number of trains which overlap each other in time. Based on the peak positions of the two-dimensional spectrum of the signal obtained for the train, a value of the relative velocity of a radar target, which is periodic with a predefined velocity period, is determined. The phase relationship of the spectral values ​​of the spectrum of the train of transmission switching states is compared with the phase relationship expected for the respective periodic value of the relative velocity, and based on the result of the comparison an estimate of the relative velocity is selected.

[0005] US Patent Application Publication No. 2017 / 0160380 describes a MIMO radar system in which multiple transmit antennas transmit simultaneously, using Pseudo-Random Phase Modulation (PRPM) to randomly vary the phase of the signals routed to each transmit antenna to obtain orthogonality between simultaneously emitted and received signals.

[0006] The digital modulation method using multiple carrier frequencies is known as orthogonal frequency division multiplexing (OFDM). The use of OFDM in radar systems is being increasingly considered. In OFDM, the frequency band is divided into multiple orthogonal sub-bands or sub-carriers (FDM, frequency division multiplexing), and OFDM symbols are transmitted sequentially. The transmitted signal for an OFDM symbol is composed of mutually orthogonal sub-carrier signals modulated according to the symbol modulation pattern, and transmitted simultaneously within the OFDM symbol period.

[0007] The applicant has already proposed a MIMO radar system in which the repetition rate of the multiplexed array is below the Nyquist limit for an unambiguous Doppler measurement, so that the measurement of the relative velocity is ambiguous, albeit with high resolution. To resolve this ambiguity, the proposed method exploits the fact that demultiplexing methods, in which signals that can be uniquely assigned to the individual transmitting antennas of the transmitting array are reconstructed from the received signals, only give high quality results if the phase offset due to the Doppler effect is appropriately corrected. However, this requires knowledge of the relative velocity of the object in question. The ambiguity is resolved by testing various ambiguity hypotheses and selecting the one that provides the highest quality standard signal during demultiplexing. [Prior art documents] [Patent documents]

[0008] [Patent Document 1] International Publication No. 2018 / 076005 [Patent Document 2] DE 102014212284 [Patent Document 3] US Patent Application Publication No. 2017 / 0160380 Summary of the Invention [Problem to be solved by the invention]

[0009] An object of the present invention is to provide a MIMO radar system having an easily configured antenna array that can determine the relative velocity and positioning angle of a radar target with high accuracy and a wide unambiguity range in a short measurement time. [Means for solving the problem]

[0010] This problem is solved according to the present invention by: a transmitting array having a plurality of transmitting antennas spaced apart from one another in an angularly resolved direction; a receiving array having a plurality of receiving antennas spaced apart from one another in an angularly resolved direction; - Control evaluation device and This problem is solved by a MIMO radar system equipped with - the antenna spacing of the transmit and receive arrays is above the Nyquist limit for unambiguous angle measurement (spatial frequencies below the Nyquist limit), but the antenna spacing of the combined transmit and receive arrays is below this Nyquist limit (spatial frequencies above the Nyquist limit); A control evaluation device - transmitting, in each of a plurality of repeated measurement cycles, a series of transmit signals divided into a plurality of measurement blocks via a transmit array, a uniform multiplexing pattern being applied within each measurement block, the multiplexing pattern being different for each measurement block; - for each measurement block received signal, performing a Doppler estimation and an angle estimation based on the receive array; - performing a Doppler correction of the received signal based on the Doppler estimation; - Demodulate the Doppler-corrected received signal, - Refine the Doppler and angle estimates based on the signals received in the various measurement blocks of the measurement cycle, taking into account the phases of the signals transmitted from the various transmit antennas. It is configured for:

[0011] The present invention modifies previously proposed methods in that the angle measurements are undersampled, i.e. sampled below the Nyquist limit, but not the Doppler measurements. To this end, either the transmit array aperture or the receive array aperture are selected to provide high angular separation, which results in ambiguous results, or are selected to fill in the gaps between individual antennas in virtual arrays representing various combinations of transmit and receive antennas, resulting in unambiguous yet high resolution results.

[0012] However, angle estimation based on a virtual array requires demultiplexing of the received signals, i.e., reconstructing signals that can be uniquely associated with the individual transmit antennas, which requires correction for the phase advance due to the Doppler effect.

[0013] A feature of the proposed system is that the period duration during which the multiplexing pattern is repeated extends over the entire duration of the measurement cycle and thus includes a large number of measurement blocks. For the purpose of a high-resolution and unambiguous Doppler measurement, the signals obtained in the various measurement blocks may be combined in such a way that, for a given total duration of the measurement cycle, the measurement time available for an accurate Doppler measurement is maximized. Thus, the relative velocity of the located object can be determined with greater accuracy. These values ​​form the basis for the Doppler correction, so that a higher signal quality is obtained during the reconstruction (demultiplexing) of the signals transmitted from the individual transmitting antennas, which increases the overall accuracy of the measurement results for a given complexity of the antenna array and a given amount of calculations during signal evaluation.

[0014] Advantageous embodiments and further variants of the invention emerge from the dependent claims.

[0015] Taking into account the extended multiplexing period, it is advantageous to correct migration effects in the signal evaluation due to the change in the distance of the localized object during the extended measurement period. For this purpose, the coordinates representing the relative speed and distance can be transformed in the multidimensional detection space in such a way that for the signals obtained in the different measurement blocks, the signal peaks representing the estimates of the distance and relative speed of the localized object occur at the same positions in all measurement blocks. This ensures that migration effects do not have a detrimental effect when the signals obtained in the different measurement blocks are evaluated together.

[0016] In one embodiment, the radar system is an FMCW or chirp sequence radar, in which the frequency of the transmitted signal is modulated according to a train of steep frequency ramps, so-called chirps, whose slope is so large that the Doppler effect on the ramps is negligible, and thus a pure propagation time measurement, i.e. a distance measurement, is performed. The relative velocity measurement is then performed by evaluating the ramp-to-ramp phase shift due to the Doppler effect.

[0017] According to the MIMO principle, appropriate multiplexing and demultiplexing methods are required to separate the signals transmitted from different transmit antennas from one another, in particular code division multiplexing and time division multiplexing methods.

[0018] In code division multiplexing, the signals transmitted simultaneously from the transmit antennas are coded with orthogonal or quasi-orthogonal codes set in a code matrix. The code distribution to the different transmit antennas remains unchanged within each measurement block, but varies from measurement block to measurement block.

[0019] If a complete measurement cycle includes, for example, nM measurement blocks each setting a different code distribution for a transmit antenna, the phases of the signals received at the different measurement blocks form a vector with nM components. For a square code matrix, nM is equal to the number of simultaneous transmit antennas of the transmit array. The received signal can then be decoded by multiplying the vector by the inverse of the code matrix. However, a prerequisite is that the orthogonality of the coded signals is maintained or at least can be restored when reflected by a radar target. If the relative velocity of the radar target is different from zero, the Doppler effect will result in some disturbance of the orthogonality, with the result that the decoded signal for a given transmit antenna will also contain signal components originating from other transmit antennas.

[0020] Therefore, to obtain measurement results of a high standard of quality, the received signal must be corrected according to the Doppler effect.

[0021] In other embodiments, a time division multiplexing scheme can be used instead of a code division multiplexing scheme. In that case, in each measurement block, only a single transmit antenna of the transmit array is active, and the different transmit antennas are switched between for each measurement block according to a fixed pattern. Here, the order in which the individual antennas are activated is generally not the same as the order in which the antennas are spatially arranged in the transmit array. Thus, the relative motion of the radar target leads to characteristic phase shifts between the signals received from the different transmit antennas due to the time offsets at which the signals were transmitted, which are distinct from the phase shifts caused by angle-dependent propagation time differences when the signals are emitted at a certain angle relative to the normal of the transmit array. Again, phase errors due to relative motion can be corrected based on high-resolution Doppler measurements.

[0022] Embodiments of the invention which operate with a combination of code division multiplexing and time division multiplexing are also contemplated.

[0023] Similarly, embodiments in which the code matrix is ​​not square are also contemplated. If the number of columns of the code matrix (code instances) is less than the number of transmit antennas, then under-determined equations are obtained upon decoding, which can be solved with plausible additional assumptions (e.g., regarding the number of simultaneously located radar targets). Conversely, if there are more code instances than transmit antennas, then over-determined equations are obtained. In that case, greater robustness of the ambiguity resolution is obtained, for example, against signal noise or other interference effects.

[0024] In one embodiment, the receive array has a large aperture that is not completely occluded, so that angle estimates based on the receive array are high resolution but ambiguous, while the transmit array has a small aperture that is completely occluded, thus allowing unambiguous angle measurements at lower resolution, however, in other embodiments the transmit array may have a large aperture and the receive array a smaller aperture.

[0025] The embodiments will be described in detail below with reference to the drawings. [Brief description of the drawings]

[0026] [Figure 1] FIG. 1 is a schematic diagram of the analog portion of a MIMO radar system with independent range and velocity determination. [Diagram 2] FIG. 2 is a diagram showing the frequency of an FMCW transmission signal and the modulation pattern of the transmission signal. [Diagram 3] FIG. 1 is a diagram of an antenna array of a radar system. [Figure 4] FIG. 2 is a diagram of the positioning angle range of the radar system. [Diagram 5] 1 is a block diagram of a digital signal evaluation device according to an embodiment of the present invention; [Figure 6] FIG. 11 is a block diagram of a digital signal evaluation device according to another embodiment of the present invention. [Figure 7] FIG. 11 is a block diagram of a digital signal evaluation device according to another embodiment of the present invention. DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0027] An embodiment of a high-speed chirp MIMO radar system will be described as an example of an FMCW-MIMO radar system in which transmission signal encoding is performed by phase modulation, with reference to Fig. 1 to Fig. 4. Fig. 1 shows a schematic simplified configuration of an analog part of the radar system.

[0028] The frequency modulation device 10 controls a high frequency oscillator 12 which generates a train of identical signals in the form of frequency ramps for a number of transmitting antennas 14. In each of a number of transmitting channels, a respective phase modulator 16 preceding an amplifier 18 modulates the phase of the signal according to a respective code 20 generated by a code generator 22. The phase modulated signal is emitted via one of the transmitting antennas 14. The transmitted signal reflected by an object 24 is received by a number of receiving antennas 26, where it is mixed by a mixer 28 with the non-phase modulated signal component from the high frequency oscillator 12 and brought to a low frequency range. Then, A / D conversion is performed by an A / D converter 30 in the usual way.

[0029] FIG. 2 shows diagrammatically the frequency progression of the transmitted signal during a measurement cycle. Below this is shown the pattern of code blocks 32, with which the transmitted signal is phase modulated. The measurement cycle is divided into a number of measurement blocks 32. A "fast chirp" frequency modulation pattern is used, with a train of relatively "fast" frequency ramps 34, so that the evaluation of distance and velocity can be made substantially independently of one another, for example by means of a two-dimensional Fourier transformation. In particular, the Doppler shift in the ramps can be neglected. Each frequency ramp 34 has a duration T R has.

[0030] According to Fig. 2, the codes 20 for each transmit antenna 14 are set in a code sequence M. The code matrix associates each signal for each transmit antenna with a code value A, B, C, ... of said code 20. Each code value defines the phase with which the phase modulator 16 modulates the signal. Each column of the code matrix, also called code instance I, defines a code division multiplexing pattern that assigns a specific code value to each of the transmit antennas, thereby setting a specific phase modulation. The number of code instances corresponds to the number of transmit antennas transmitting simultaneously.

[0031] Within each measurement block 32 the code instance and thus the multiplexing pattern remain unchanged and are switched to the next code instance upon transition to the respective next measurement block, the number of code instances being equal to the number of measurement blocks 32 in the measurement cycle.

[0032] In the sequence of measurement blocks 32, the multiplexing pattern goes through all code instances I=1,...,m of the code matrix M. In the example shown in Fig. 2, the measurement blocks 32 are consecutive with no gaps and are multiplied by a clock rate of 1 / T M A uniform block duration T M 2. The index C#=1...m in Fig. 2 counts the measurement blocks, while the index TX=1...n (n=m) numbers the transmit antennas. The codes 20 of the code matrix are mutually orthogonal (preferably fully orthogonal, or alternatively quasi-orthogonal, i.e. small cross-correlation between codes). Thus, the signals of the individual transmit antennas are coded by codes and the transmit signals are mutually orthogonal in order to allow signal separation in the receive channel.

[0033] In other embodiments, the measurement blocks may be separated by intervals of the same or different lengths.

[0034] 3, the transmitting antennas 14 form a transmitting array 36 and the receiving antennas 26 form a receiving array 38. In the example shown, both arrays are two-dimensional, allowing MIMO angle measurements in both azimuth and elevation.

[0035] In the receive array 38, the receive antennas 26 are equally spaced in the angular resolution direction y, e.g., azimuth. The distance between the individual receive antennas is so large that a large aperture and a correspondingly high angular resolution can be achieved with just a few antennas. However, the Nyquist uniqueness criterion is not met because the antenna-to-antenna distance is now greater than half the wavelength of the radar radiation.

[0036] In Fig. 4, the field of view of the radar sensor is shown to include angles from -θ to +θ with respect to an axis X perpendicular to the plane of the receiving array 38. The result of the angle measurement is unambiguous only if the positioning angle is within a significantly small interval from -θa to +θa. If larger positioning angles cannot be excluded, the measurement is not unambiguous since there are several angles for which the phase relationship between the signals at the receiving antennas is the same.

[0037] In the example shown in FIG. 3, the receiving antennas 26 are also arranged at equal intervals in the elevation angle direction (angle resolution direction z), and since the antenna intervals are large in this direction as well, non-unique undersampling is performed.

[0038] The transmitting antennas 14 of the transmit array 36 are unevenly spaced in azimuth, with the spacing selected to provide unambiguous angle measurements, but with a significantly smaller aperture than that of the receive array 38, resulting in poor angular resolution. Also, in elevation, the transmit array 36 is configured for unambiguous angle measurements with a small aperture.

[0039] 3 further shows a composite array 40 that is obtained when combining each receive antenna 26 with each transmit antenna 14 such that the propagation time differences of the signal from the transmit antenna to the object and from the object to the receive antenna are additive. Ultimately, it is the aperture of this virtual array 40 that determines the resolution of the radar sensor. However, the signal components emanating from the various transmit antennas 14 need to be separated from each other in the received signal so that the receive array can be disambiguated.

[0040] Also, in the example shown in Figure 3, for each y position of the receive antenna, all z positions are also occupied by the receive antenna, so the two angular resolved directions y and z in the receive array 38 are separated from each other. In contrast, the transmit array 36 is an example of a non-separated array where for some y positions (the two right positions in Figure 3), all z positions are not occupied. In general, a separated array makes data evaluation easier, and a non-separated array requires fewer antenna elements. The decision between separated and non-separated arrays can be made differently for the transmit and receive sides depending on their respective requirements.

[0041] Also, an equidistant arrangement of the antenna elements (in azimuth and / or elevation) facilitates the evaluation of the data, since it allows, for example, the use of a Fast Fourier Transform (FFT), whereas a non-equidistant arrangement of the antennas has the advantage that a unique angular range (FIG. 4) can be optimized for a given aperture, as here is the case for the transmitting antenna 14.

[0042] Overall, all combinations of equidistant and non-equidistant or separated and non-separated geometries are contemplated in the radar systems described herein. Similarly, embodiments are possible in which the transmit array is configured for ambiguous high resolution angle measurements while the receive array is configured for unambiguous angle measurements with lower angular resolution.

[0043] Next, an example of an evaluation device for evaluating a received signal obtained using the antenna array shown in FIG. 3 and the multiplexing pattern shown in FIG. 2 will be described with reference to FIG.

[0044] The digital data provided by the A / D converter 30 are sampled over each complete measurement cycle. The total number of (complex) signal values ​​recorded within a measurement cycle is given by the product of the number of receive antennas 26 of the receive array 38, nRX, the number of measurement blocks, nM, the number of repetitions of the frequency ramp 34 within each measurement block, ns, and the number of scan points on each frequency ramp 34, nf. In a processing stage 42, the data sampled over a measurement cycle are subjected to a four-dimensional Fourier transform (4D-FFT). This results in a four-dimensional spectrum with the dimensions "Azimuth 1", "Elevation 1", "Doppler 1" and "Distance". The "Azimuth 1" dimension indicates the distribution of complex amplitudes over the azimuth positioning angle range based on the data of the receive antennas 26 arranged in a row in the azimuth direction. Correspondingly, the "Elevation 1" dimension indicates the distribution over the elevation angle range based on the data of the receive antennas 26 arranged in a row in the elevation direction. The "Doppler 1" dimension is calculated based on the clock rate ns / T of the frequency ramp. M (The reciprocal of the duration of the ramp T R ) shows the Doppler spectrum obtained by a "slow" scan. It should be noted that the results in the "Azimuth 1" and "Elevation 1" dimensions are ambiguous due to the respective undersampling. The results in the "Doppler 1" dimension are unambiguous depending on the selected clock rate, but have only a small resolution due to the relatively short duration of the individual measurement blocks. The "Distance" dimension shows the distance spectrum based on a "fast" scan on each frequency gradient 34. In this dimension too, the results are unambiguous. For each transmitted gradient a separate spectrum is obtained.

[0045] In the 4D spectrum, each radar target located is represented as a peak at a specific coordinate position. When all radar targets are stationary relative to the radar sensor, the power spectra obtained for each successive measurement block are identical to each other. However, if one of the radar targets has a non-vanishing relative velocity, a significant change in the distance of this object can occur over the duration of the measurement cycle. A so-called migration effect then occurs, in which the position of the range coordinate of the peak in the spectrum moves slightly from one measurement block to the next, the extent of which depends on the relative velocity. This effect can be compensated for with a relatively small amount of computation by moving the range coordinate into the range-velocity subspace, so that the migration of the radar targets in the spectrum is reversed. This correction results in a set of spectra in which all peaks occupy the same position.

[0046] These corrected 4D spectra can then be integrated incoherently (addition of the absolute values ​​of the complex amplitudes) without smearing the peaks. This results in an amplitude distribution in a 4D detection space 44. To each point of this 4D space is assigned a specific value of the amplitude sum, in which each located object is represented in the form of a maximum represented at a specific distance, a specific Doppler shift, a specific azimuth angle and a specific elevation angle, although the last two values ​​are ambiguous, so that only one of several hypotheses regarding the azimuth and elevation angles can be associated with the object. Then, in this detection space 44, the 4D coordinates of the detected peaks are searched, each representing a detection result. For each of these points there are nM complex amplitudes that (before the incoherent integration) form a vector with nM components and are further evaluated to improve the measurement of the relative velocity and at the same time resolve any ambiguity remaining in the angle measurement.

[0047] In addition to this, the amplitudes obtained in the different measurement blocks are (coherently) added in a fusion stage 46 for each detected object, resulting in a higher resolution Doppler spectrum and thus a more accurate result for the relative velocity "Doppler 2", since the measurement period is extended to the entire measurement cycle.

[0048] This more accurate value of the relative speed can be used to correct the phase that occurs in the signal vector as a function of the relative speed, thereby restoring the orthogonality of the codes within the code instances so that a correct decoding is possible, resulting in a decoded signal vector whose components each indicate the phase of the signal emanating from one of the transmit antennas 14. Then, based on the components of this vector that belong to the azimuthally arranged transmit antennas 14, an unambiguous (but low-resolution) value for the azimuth angle can be determined, and correspondingly, an unambiguous value for the elevation angle can be determined based on the components that belong to the elevationally arranged transmit antennas.

[0049] Based on these criteria, ambiguity resolution can be performed in a second detection stage 48. For this purpose, the phase-corrected and decoded signal vectors obtained in the fusion stage 46 are summed together, for example in a coherently added three-dimensional spectrum. This spectrum has the dimensions "Doppler2", "Azimuth2", and "Elevation2", and the sharpest (and highest) peaks in this spectrum represent the true unambiguous values ​​for the relative velocity, azimuth and elevation of the object. Similarly, multiple targets can be resolved in this three-dimensional space.

[0050] The results obtained in the first detection stage 44, "Azimuth 1", "Elevation 1", are high resolution but ambiguous, while the results obtained in the second detection stage 48 are low resolution but unambiguous. These results can be fused by selecting, as the final and unambiguous value for the azimuth, the value from the high resolution values ​​of "Azimuth 1" that best matches "Azimuth 2". The elevation values ​​are fused in a similar way.

[0051] When searching the spectrum for "Doppler 2", "Azimuth 2", and "Elevation 2", one can take advantage of the fact that results for "Doppler 1", "Azimuth 1", and "Elevation 1" already exist, so only a relatively small region of the detection space in which the searched values ​​should lie is of interest, and therefore the search can be limited to these regions.

[0052] Since each localization object may have a different relative velocity, the calculations in the fusion stage 46 and the calculations in the second detection stage 48 are carried out separately for each detection result obtained in the detection space 44 .

[0053] A particular advantage of performing both velocity and angle estimation is due to the multidimensionality of the detection space, which equalizes the individual peaks. For example, if two radar targets have approximately the same relative velocity, the peaks are difficult to separate if one considers only the dimension of "relative velocity." In contrast, in a two- or three-dimensional detection space with the dimensions of "relative velocity," "azimuth," and "elevation," the peaks are typically spaced significantly apart such that they are easily separated from one another, allowing even relatively small differences in relative velocity to be detected and determined.

[0054] The structure of the radar system shown in FIG. 1 also allows an alternative mode of operation in which the transmit signals are transmitted in time division multiplexing rather than code division multiplexing. In that case, the code generator 22 drives the individual amplifiers 18 in such a way that only one transmit antenna is active at any given time and switches between the transmit antennas in a specific order. These switches then separate the individual measurement blocks. In this case, no coding of the transmit signals is necessary, since the transmit signals are already separated from one another by being transmitted out of time. Nevertheless, a phase correction of the receive signals is also necessary in time division multiplexing in order to compensate for the time offset between the signals transmitted alternately from the transmit antennas 14. Similar to the phase correction for orthogonality recovery in code division multiplexing, in this case too the phase correction is based on a result that can be determined with high accuracy for the relative speed.

[0055] Likewise, a combined code division multiplexing and time division multiplexing operating mode is also conceivable, in which the transmit antennas 14 are divided into groups each transmitting simultaneously and whose signals are coded with a corresponding smaller code matrix.

[0056] There are also various options for the type of signal evaluation shown in FIG.

[0057] 6 shows an example in which the digital data sampled over a measurement cycle is first subjected in a processing stage 42 to only a two-dimensional Fourier transformation of the dimensions "Doppler 1" and "Distance", so that the first detection stage provides for each detected object a "Distance" value and a low-resolution "Doppler 1" value for the relative velocity.

[0058] In a fusion stage 46, the measurement of the relative velocity is improved based on the signals of all measurement blocks, a phase correction for each relative velocity and a subsequent decoding of the phase-corrected signal vector are performed. In this way, a high-resolution value of the relative velocity "Doppler 2" is obtained, and at the same time a set of signals TX associated with the active transmit antennas 14 is obtained. These signals are then used in an angle estimation stage 52 to perform an angle estimation based on the transmit array 36 and the receive array 38, which results in an unambiguous and high-resolution value of the azimuth angle "Azimuth". In this way, in a second detection stage 48, a two-dimensional spectrum of "Doppler 2" and "Azimuth" is obtained for each detection result of the first stage.

[0059] Another variant of the evaluation method is shown in Fig. 7. In this method, in the processing stage 42, a Fourier transformation is performed in three dimensions in the dimensions "Doppler 1", "distance" and "azimuth angle 1" so that in the detection space 44, an unambiguous distance value, an unambiguous value for the relative velocity "Doppler 1" and an ambiguous value "azimuth angle 1" are obtained. Then, in the fusion stage 46, a phase correction and a decoding are performed for each detection result. Then, in the decoded signal, an angle estimation is performed on the basis of the transmit array 36 in the angle estimation stage 52 so that an unambiguous value for the azimuth angle "azimuth angle 2" is obtained. At this stage, a higher resolution value "Doppler 2" is provided. Then, in the second detection stage 48, a peak of the two-dimensional spectrum in the dimensions "Doppler 2" and "azimuth angle 2" is searched for for each detection result, after which the detection results obtained in the detection stage 44 and the second detection stage 48 are fused in the manner already described.

[0060] Also, in a radar system where the transmit array has a large, unobstructed aperture and the receive array has a smaller aperture, unlike Figure 3, the roles of the transmit and receive arrays can be reversed and the evaluation method described above can be applied in a similar manner. In the first stage, the angle estimates are unambiguous but low-resolution values, and in the second stage, only the neighborhood of these values ​​needs to be investigated to improve accuracy.

Claims

1. a transmitting array (36) having a number of transmitting antennas (14) arranged at a distance from one another in an angularly resolved direction (y, z); a receiving array (38) having a number of receiving antennas (26) arranged at a distance from one another in said angular resolution direction; - Control evaluation device and A MIMO radar system comprising: the antenna spacing at each of the transmit and receive arrays is above the Nyquist limit for unambiguous angle measurement, but the antenna spacing at the combined transmit and receive arrays is below this Nyquist limit; The control evaluation device, - transmitting, during each of a number of repeated measurement cycles, a sequence of transmit signals divided into a number of measurement blocks via said transmit array, a uniform multiplexing pattern being applied within each measurement block, said multiplexing pattern being different for each measurement block; - performing a Doppler and angle estimation for the signals received at each measurement block based on said receive array (38); - performing a Doppler correction of the received signals based on said Doppler estimate; - demodulating said Doppler corrected received signal, improving said Doppler and angle estimates based on signals received in different measurement blocks of a measurement cycle, taking into account the phases of the signals transmitted from the different transmit antennas; A radar system configured for:

2. The control evaluation device, - in each measurement cycle, distance measurements are carried out on the objects to be located, separated into measurement blocks, 2. A radar system according to claim 1, configured for performing a velocity-dependent coordinate transformation of the range data obtained in the different measurement blocks prior to the Doppler correction in order to compensate for the relative motion of the object in the time lapse between the measurement blocks.

3. 3. The radar system of claim 1, wherein the multiplexing pattern includes code division multiplexing for at least some of the transmit antennas (14).

4. 4. A radar system according to claim 1, wherein the multiplexing pattern comprises time division multiplexing for at least some of the transmit antennas (14).

5. 5. A radar system according to claim 1, wherein the transmission signal (20) comprises, within each measurement block, a train of frequency ramps (34) whose slopes are configured for distance measurement according to the FMCW principle.

6. 6. The radar system of claim 1, wherein the transmit array is configured for unambiguous angle measurements in at least one angular dimension and the receive array is configured for high-resolution ambiguous angle measurements.

7. 6. The radar system of claim 1, wherein the receive array is configured for unambiguous angle measurements in at least one angular dimension and the transmit array is configured for high-resolution ambiguous angle measurements.

8. 8. A radar system according to claim 1, wherein the receiving antennas (26) and / or the transmitting antennas (14) are arranged equidistant in the angular resolved directions (y,z).

9. A radar system as claimed in any one of claims 1 to 8, wherein angle estimation is performed based on the receiving array (38) separately from the angle estimation based on the transmitting array (36).

10. 10. The radar system of claim 9, wherein the receiving antennas (26) are equidistantly positioned in the angular resolution directions (y,z) and the angle estimation is performed by a Fast Fourier Transform (FFT) based on the receiving array (38).

Images