Motor control device and motor control method

Abstract

A motor control device comprising: a current control unit that calculates a voltage command with respect to the d axis and the q axis of a motor per each specified calculation cycle; a carrier wave generation unit that generates carrier waves; a carrier wave frequency adjustment unit that adjusts the frequency of the carrier waves; a phase calculation unit that calculates a voltage phase of an inverter based on the rotational position of the motor; a divided phase calculation unit that calculates a divided phase obtained by dividing the voltage phase per each of two or more prescribed division numbers; a three-phase voltage conversion unit that converts the voltage command to a three-phase voltage command on the basis of the divided phase; and a PWM control unit that pulse-width modulates the three-phase voltage command using the carrier waves, and generates a PWM pulse signal for controlling the operation of the inverter.

Description

[Technical Field]

[0001] The present invention relates to an apparatus and method for controlling a motor. [Background technology]

[0002] Conventionally, motor control devices have been known that control the operation of an inverter that converts DC power into AC power using multiple switching elements, and drive an AC motor using the AC power output from the inverter, thereby controlling the motor. Such motor control devices are widely used in, for example, railway vehicles and electric automobiles.

[0003] The power losses generated in a motor control device mainly consist of inverter switching losses and motor iron losses. The motor iron losses can be reduced by increasing the inverter switching frequency. However, increasing the switching frequency usually increases the inverter switching losses accordingly, making it impossible to reduce power losses.

[0004] One known solution to this problem is to use switching elements with excellent characteristics during high-frequency operation, such as semiconductor switching elements made of silicon carbide (SiC), in the inverter, and then increase the switching frequency. This effectively reduces the iron loss of the motor while suppressing the increase in switching loss in the inverter to some extent.

[0005] One example of a known motor control device for increasing the switching frequency is described in Patent Document 1. The motor control device in Patent Document 1 has two arithmetic units, one of which performs calculations to control the motor's current, and the other which monitors the first arithmetic unit for abnormalities and performs calculations to detect the motor's magnetic pole position. This achieves a high-speed, highly responsive motor control device using a controller such as a microcomputer. [Prior art documents] [Patent documents]

[0006] [Patent Document 1] Japanese Patent Publication No. 2003-23800 Summary of the Invention [Problem to be solved by the invention]

[0007] In the motor control device described in Patent Document 1, current control calculation and magnetic pole position calculation are shared between two calculation devices, and these calculation processes are executed synchronously. Therefore, the period of the PWM signal output from the motor control device to the inverter to drive the inverter's switching elements matches the calculation period of the current control calculation, and the output period of the PWM signal cannot be made shorter than the calculation period of the current control calculation. Therefore, it is difficult to increase the switching frequency.

[0008] In view of the above problems, a main object of the present invention is to increase the switching frequency of an inverter. [Means for solving the problem]

[0009] A motor control device according to the present invention is connected to an inverter that converts DC power into three-phase AC power and outputs it to a motor, and controls the operation of the inverter to control the drive of the motor using the inverter.The motor control device includes: a current control unit that calculates voltage commands for the d-axis and q-axis of the motor at each predetermined calculation period; a carrier wave generation unit that generates a carrier wave; a carrier wave frequency adjustment unit that adjusts the frequency of the carrier wave; a phase calculation unit that calculates a voltage phase of the inverter based on the rotational position of the motor; a divided phase calculation unit that calculates divided phases by dividing the voltage phase by a predetermined number of divisions of two or more; a three-phase voltage conversion unit that converts the voltage command into a three-phase voltage command based on the divided phase; and a PWM control unit that pulse-width modulates the three-phase voltage command using the carrier wave to generate a PWM pulse signal for controlling the operation of the inverter. A motor control method according to the present invention is a method for controlling the operation of an inverter that converts DC power into three-phase AC power and outputs it to the motor, thereby controlling the drive of the motor using the inverter, and includes calculating voltage commands for the d-axis and q-axis of the motor at predetermined calculation cycles, adjusting the frequency of a carrier wave, calculating a voltage phase of the inverter based on the rotational position of the motor, dividing the calculation cycle of the voltage command by a predetermined division number to obtain a calculation cycle of a divided phase based on the voltage phase, calculating the divided phase, converting the voltage command into a three-phase voltage command based on the divided phase, and pulse-width modulating the three-phase voltage command using the carrier wave to generate a PWM pulse signal for controlling the operation of the inverter. [Effects of the Invention]

[0010] According to the present invention, the switching frequency of the inverter can be increased. [Brief explanation of the drawings]

[0011] [Figure 1] 1 is a diagram showing the overall configuration of a motor drive system including a motor control device according to an embodiment of the present invention; [Figure 2] 1 is a block diagram showing the functional configuration of a motor control device according to an embodiment of the present invention; [Figure 3] FIG. 3 is a block diagram of a division phase calculation unit according to an embodiment of the present invention. [Figure 4] FIG. 10 is a diagram showing an example of the relationship between a PLL trigger and a division phase. [Figure 5] FIG. 10 is a diagram showing how the division phase changes. [Figure 6] 10 is a comparison diagram between three-phase voltage command values ​​obtained by conventional control when the present invention is not applied and three-phase voltage command values ​​when the present invention is applied; FIG. [Figure 7] FIG. 2 is a diagram showing an example of the hardware configuration of a motor control device. DETAILED DESCRIPTION OF THE INVENTION

[0012] Hereinafter, an embodiment of the present invention will be described in detail with reference to the drawings. In this embodiment, an example of application to a motor drive system mounted on and used in an electrically powered vehicle such as an electric vehicle or a hybrid vehicle will be described.

[0013] Figure 1 is a diagram showing the overall configuration of a motor drive system equipped with a motor control device according to one embodiment of the present invention. In Figure 1, the motor drive system 100 includes a motor control device 1, a permanent magnet synchronous motor (hereinafter simply referred to as "motor") 2, an inverter 3, a rotational position detector 4, and a high-voltage battery 5.

[0014] The motor control device 1 controls the operation of the inverter 3 based on a torque command T* corresponding to a target torque required of the motor 2 by the vehicle, thereby generating a PWM pulse signal for controlling the drive of the motor 2. The generated PWM pulse signal is then output to the inverter 3. Details of the motor control device 1 will be described later.

[0015] The inverter 3 has an inverter circuit 31, a gate drive circuit 32, and a smoothing capacitor 33. The gate drive circuit 32 generates gate drive signals for controlling each switching element of the inverter circuit 31 based on a PWM pulse signal input from the motor control device 1, and outputs the gate drive signals to the inverter circuit 31. The inverter circuit 31 has switching elements corresponding to the upper and lower arms of the U, V, and W phases, respectively. By controlling each of these switching elements in accordance with the gate drive signals input from the gate drive circuit 32, the DC power supplied from the high-voltage battery 5 is converted into AC power and output to the motor 2. The smoothing capacitor 33 smoothes the DC power supplied from the high-voltage battery 5 to the inverter circuit 31.

[0016] Motor 2 is a synchronous motor that is rotationally driven by AC power supplied from inverter 3, and has a stator and a rotor. When AC power input from inverter 3 is applied to armature coils Lu, Lv, and Lw provided in the stator, three-phase AC currents Iu, Iv, and Iw are conducted in motor 2, and armature magnetic flux is generated in each armature coil. Attractive and repulsive forces are generated between the armature magnetic flux of each armature coil and the magnetic flux of the permanent magnets arranged in the rotor, generating torque in the rotor and driving the rotor to rotate.

[0017] A rotational position sensor 8 is attached to the motor 2 to detect the rotational position θr of the rotor. The rotational position detector 4 calculates the rotational position θr from the input signal of the rotational position sensor 8. The calculation result of the rotational position θr by the rotational position detector 4 is input to the motor control device 1, and is used in the phase control of AC power, which is performed by the motor control device 1 generating a PWM pulse signal in accordance with the phase of the induced voltage of the motor 2.

[0018] Here, a resolver composed of an iron core and windings is more suitable for rotational position sensor 8, but a sensor using a magnetic resistance element such as a GMR sensor or a Hall element may also be used. Furthermore, rotational position detector 4 may estimate rotational position θr using three-phase AC currents Iu, Iv, Iw flowing through motor 2 or three-phase AC voltages Vu, Vv, Vw applied to motor 2 from inverter 3, without using an input signal from rotational position sensor 8.

[0019] A current detection unit 7 is disposed between the inverter 3 and the motor 2. The current detection unit 7 detects three-phase AC currents Iu, Iv, and Iw (U-phase AC current Iu, V-phase AC current Iv, and W-phase AC current Iw) passing through the motor 2. The current detection unit 7 is configured using, for example, a Hall current sensor. The detection results of the three-phase AC currents Iu, Iv, and Iw by the current detection unit 7 are input to the motor control device 1 and used to generate PWM pulse signals performed by the motor control device 1. Note that while FIG. 2 shows an example in which the current detection unit 7 is configured with three current detectors, it is also possible to use two current detectors and calculate the AC current of the remaining phase based on the fact that the sum of the three-phase AC currents Iu, Iv, and Iw is zero. Alternatively, the pulsed DC current flowing from the high-voltage battery 5 to the inverter 3 may be detected by a shunt resistor or the like inserted between the smoothing capacitor 33 and the inverter 3, and the three-phase AC currents Iu, Iv, Iw may be calculated based on this DC current and the three-phase AC voltages Vu, Vv, Vw applied from the inverter 3 to the motor 2.

[0020] Next, the motor control device 1 will be described in detail. FIG. 2 is a block diagram showing the functional configuration of the motor control device 1 according to one embodiment of the present invention. In FIG. 2, the motor control device 1 has the following functional blocks: a current command generation unit 10, a speed calculation unit 11, a current conversion unit 12, a current control unit 13, a carrier frequency adjustment unit 14, a carrier generation unit 15, a phase calculation unit 16, a divided phase calculation unit 17, a three-phase voltage conversion unit 18, and a PWM control unit 19. The motor control device 1 is configured, for example, by a microcomputer, and these functional blocks can be realized by executing a predetermined program on the microcomputer. Alternatively, some or all of these functional blocks may be realized using hardware circuits such as logic ICs or FPGAs.

[0021] The current command generation unit 10 calculates the d-axis current command Id* and the q-axis current command Iq* based on the input torque command T* and the voltage Hvdc of the high-voltage battery 5. Here, for example, a preset current command map, formula, or the like is used to determine the d-axis current command Id* and the q-axis current command Iq* according to the torque command T*.

[0022] The speed calculation unit 11 calculates the motor rotation speed ωr, which indicates the rotation speed (number of rotations) of the motor 2, from the change over time in the rotation position θr. The motor rotation speed ωr may be a value expressed as either an angular velocity (rad / s) or a rotation number (rpm). These values ​​may also be converted into each other and used.

[0023] The current conversion unit 12 performs dq conversion on the three-phase AC currents Iu, Iv, and Iw detected by the current detection unit 7 based on the rotational position θr determined by the rotational position detector 4, and calculates the d-axis current value Id and the q-axis current value Iq.

[0024] Based on the deviation between the d-axis current command Id* and q-axis current command Iq* output from the current command generating unit 10 and the d-axis current value Id and q-axis current value Iq output from the current converting unit 12, the current control unit 13 calculates a d-axis voltage command Vd* and a q-axis voltage command Vq* corresponding to the torque command T* so that these values ​​match. Here, for example, using a control method such as PI control, the d-axis voltage command Vd* corresponding to the deviation between the d-axis current command Id* and the d-axis current value Id, and the q-axis voltage command Vq* corresponding to the deviation between the q-axis current command Iq* and the q-axis current value Iq are calculated at every predetermined calculation period Tv.

[0025] The carrier frequency adjuster 14 calculates a carrier frequency fc, which represents the frequency of the carrier wave used to generate the PWM pulse signal, based on the rotational position θr determined by the rotational position detector 4 and the rotational speed ωr determined by the speed calculator 11. For example, the carrier frequency adjuster 14 calculates the carrier frequency fc so that the number of carrier waves per rotation of the motor 2 is a predetermined number of carrier waves Nc and the relationship between the phase of the carrier wave and the rotational position θr is constant.

[0026] Carrier wave generating unit 15 generates a carrier wave signal (triangular wave signal) Sc based on the carrier wave frequency fc calculated by carrier wave frequency adjusting unit .

[0027] Based on the rotational position θr, the phase calculation unit 16 calculates the voltage phase (electrical angle) θe of the inverter 3. For example, based on the rotational position θr, the phase calculation unit 16 calculates the voltage phase θe using the d-axis voltage command Vd* and q-axis voltage command Vq* calculated by the current control unit 13, the rotational speed ωr calculated by the speed calculation unit 11, and the carrier frequency fc calculated by the carrier frequency adjustment unit 14, according to the following equations (1) to (4). θe=θr+φv+φdqv+0.5π ···(1) φv=ωr·1.5Tc ···(2) Tc=1 / fc (3) φdqv=atan(Vq* / Vd*) (4)

[0028] Here, φv represents a calculation delay compensation value for the voltage phase, Tc represents the carrier wave period, and φdqv represents the voltage phase from the d-axis. The calculation delay compensation value φv is a value that compensates for the calculation delay of 1.5 control periods that occurs between when the rotational position detector 4 acquires the rotational position θr and when the motor control device 1 outputs a PWM pulse signal to the inverter 3. In this embodiment, 0.5π is added to the fourth term on the right-hand side of equation (1). This is a calculation to convert the voltage phase calculated by the first to third terms on the right-hand side of equation (1) into a sine wave, since the voltage phase is a cosine wave.

[0029] Here, it is preferable that the calculation of the voltage phase θe by the phase calculation unit 16 is performed in synchronization with the calculation of the d-axis voltage command Vd* and the q-axis voltage command Vq* by the current control unit 13. In this way, the value of the voltage phase θe can be updated in accordance with the timing at which the values ​​of the d-axis voltage command Vd* and the q-axis voltage command Vq* are updated.

[0030] The divided phase calculation unit 17 calculates divided phases θe[n] by dividing the voltage phase θe calculated by the phase calculation unit 16 by a predetermined number of divisions Ne (where Ne is a positive integer equal to or greater than 2) based on the calculation cycle Tv of the d-axis voltage command Vd* and the q-axis voltage command Vq* by the current control unit 13 and the carrier frequency fc. In the divided phases θe[n], n is an integer that continuously changes from 0 to Ne-1, and θe[0]=θe. Details of the divided phase calculation unit 17 will be described later.

[0031] Three-phase voltage conversion unit 18 performs three-phase conversion on the d-axis voltage command Vd* and q-axis voltage command Vq* calculated by current control unit 13 using the divided phase θe[n] calculated by divided phase calculation unit 17, and calculates three-phase voltage commands Vu*, Vv*, Vw* (U-phase voltage command value Vu*, V-phase voltage command value Vv*, and W-phase voltage command value Vw*). In this way, three-phase voltage commands Vu*, Vv*, Vw* according to torque command T* are generated.

[0032] The PWM control unit 19 pulse-width modulates the three-phase voltage commands Vu*, Vv*, and Vw* output from the three-phase voltage conversion unit 18 using the carrier signal Sc output from the carrier wave generation unit 15, thereby generating PWM pulse signals for controlling the operation of the inverter 3. Specifically, the PWM control unit 19 generates pulsed voltages for the U, V, and W phases based on a comparison result between the three-phase voltage commands Vu*, Vv*, and Vw* output from the three-phase voltage conversion unit 18 and the carrier signal Sc output from the carrier wave generation unit 15. Then, based on the generated pulsed voltages, it generates PWM pulse signals for the switching elements of the inverter 3. At this time, it logically inverts the PWM pulse signals Gup, Gvp, and Gwp of the upper arms of each phase to generate PWM pulse signals Gun, Gvn, and Gwn of the lower arms. The PWM pulse signals generated by the PWM control unit 19 are output from the motor control device 1 to the gate drive circuit 32 of the inverter 3 and converted into gate drive signals by the gate drive circuit 32. As a result, each switching element of the inverter circuit 31 is controlled to be turned on / off, and the output voltage of the inverter 3 is adjusted.

[0033] Next, we will explain the operation of divided phase calculation unit 17 in motor control device 1. As described above, divided phase calculation unit 17 calculates divided phases θe[n] by dividing the voltage phase θe of inverter 3 by a predetermined number of divisions Ne. Three-phase voltage conversion unit 18 calculates three-phase voltage commands Vu*, Vv*, and Vw* using these divided phases θe[n], thereby enabling PWM control to be performed based on the three-phase voltage commands Vu*, Vv*, and Vw* corresponding to the divided phases θe[n] in a cycle shorter than the calculation cycle Tv of the d-axis voltage command Vd* and the q-axis voltage command Vq* by current control unit 13.

[0034] Fig. 3 is a block diagram of a divided phase calculation unit 17 according to one embodiment of the present invention. The divided phase calculation unit 17 can calculate a divided phase θe[n] based on a voltage phase θe by using the configuration shown in either the block diagram of Fig. 3(a) or the block diagram of Fig. 3(b).

[0035] In the block diagram of FIG. 3( a ), the divided phase calculation unit 17 has the following functional blocks: a current control period storage unit 171 , a period division unit 172 , and a phase division unit 173 .

[0036] The current control period storage unit 171 stores the value of the calculation period Tv of the d-axis voltage command Vd* and the q-axis voltage command Vq* by the current control unit 13, and outputs the value of this calculation period Tv to the period dividing unit 172.

[0037] The period division unit 172 determines the division number Ne used to calculate the division phase θe[n] based on the calculation period Tv of the d-axis voltage command Vd* and the q-axis voltage command Vq* input from the current control period storage unit 171 and the carrier frequency fc calculated by the carrier frequency adjustment unit 14. Specifically, for example, the carrier period Tc is calculated from the carrier frequency fc using the above-mentioned equation (3), and the division number Ne can be calculated using the following equation (5) based on the ratio Tv / Tc of the calculation period Tv to the carrier period Tc. The right-hand side of equation (5) represents an integer value obtained by rounding down the decimal point of the ratio Tv / Tc. Note that the carrier frequency adjustment unit 14 may adjust the value of the carrier frequency fc so that the value of the ratio Tv / Tc is an integer, allowing the value of the ratio Tv / Tc to be used directly as the division number Ne. Ne=int(Tv / Tc) (5)

[0038] The phase dividing unit 173 calculates the value of the divided phase θe[n], which is updated at intervals shorter than the calculation period Tv, based on the division number Ne calculated by the period dividing unit 172 and the voltage phase θe calculated by the phase calculation unit 16. Specifically, for example, in the case where the phase calculation unit 16 calculates the voltage phase θe at the same calculation period Tv as the d-axis voltage command Vd* and the q-axis voltage command Vq*, if the value of the current voltage phase θe is θe1 and the value of the previous voltage phase θe is θe0, the divided phase θe[n] can be calculated using the following equations (6) and (7). Here, n is an integer that continuously changes from 0 to Ne-1 ​​as described above, and is updated at every carrier wave period Tc. θe[n]=θe1+n·Δθe ···(6) Δθe=(θe1-θe0) / Ne ···(7)

[0039] Note that Δθe obtained by equation (7) represents the interval of the divided phases θe[n] calculated by the phase dividing unit 173. That is, the divided phases θe[n] can be calculated by dividing the amount of change θe1-θe0 in the voltage phase θe during the calculation period Tv by the division number Ne to find the interval Δθe of the divided phases θe[n], and then adding an integer multiple of this interval Δθe to the current voltage phase θe1.

[0040] In the block diagram of FIG. 3( b ), the division phase calculation unit 17 has the functional blocks of a PLL trigger output unit 174 and a PLL calculation unit 175 .

[0041] The PLL trigger output unit 174 generates and outputs a PLL trigger for determining the output timing of the divided phase θe[n] based on the timing at which the d-axis voltage command Vd* and the q-axis voltage command Vq* are output from the current control unit 13 (hereinafter referred to as the "voltage command timing") and the carrier frequency fc calculated by the carrier frequency adjustment unit 14. Specifically, for example, the PLL trigger output unit 174 calculates the carrier period Tc using the above-mentioned equation (3) based on the carrier frequency fc, and generates a pulse signal having a predetermined pulse width for each carrier period Tc starting from the voltage command timing, and outputs it as a PLL trigger. Note that in this case as well, the carrier frequency adjustment unit 14 may adjust the value of the carrier frequency fc so that the value of the ratio Tv / Tc becomes an integer, as described in the block diagram of FIG. 3(a).

[0042] In response to the PLL trigger output from the PLL trigger output unit 174, the PLL calculation unit 175 performs phase calculation based on the voltage phase θe calculated by the phase calculation unit 16, thereby calculating divided phases θe[n] obtained by dividing the value of the voltage phase θe by the division number Ne. Specifically, for example, the PLL calculation unit 175 estimates the continuously changing voltage phase θe' by phase calculation based on the calculation results of the voltage phase θe up to now by the phase calculation unit 16, and outputs the value of the voltage phase θe' at that time as the divided phase θe[n] each time a PLL trigger is output. In this way, the divided phase θe[n] can be calculated.

[0043] FIG. 4 shows an example of the relationship between the PLL trigger and the division phase θe[n] in the block diagram of FIG. 3(b). In FIG. 4, the upper part shows an example of a current control trigger corresponding to the voltage command timing by the current control unit 13. The middle part shows an example of the PLL trigger, PWM timer, and division phase θe[n] when the carrier frequency fc is low, and the lower part shows an example of the PLL trigger, PWM timer, and division phase θe[n] when the carrier frequency fc is high. The PWM timer corresponds to the carrier signal Sc generated by the carrier wave generating unit 15, and its value changes periodically with the carrier frequency fc. The PWM control unit 19 compares the value of this PWM timer with the three-phase voltage commands Vu*, Vv*, and Vw* to generate PWM pulse signals for the switching elements of each phase.

[0044] As shown in Figure 4, the higher the carrier frequency fc, the more PLL triggers are output within the period of the current control trigger (calculation period Tv). In addition, since the division number Ne calculated by the above-mentioned equation (5) increases, the number of division phases θe[n] also increases.

[0045] Fig. 5 shows how the division phase θe[n] changes. Fig. 5(a) shows an example of the division phase θe[n] when the carrier frequency fc is low, which corresponds to the case shown in the middle of Fig. 4. Fig. 5(b) shows an example of the division phase θe[n] when the carrier frequency fc is high, which corresponds to the case shown in the bottom of Fig. 4.

[0046] As shown in Figure 5, the interval between the divided phases θe[n] becomes shorter as the carrier frequency fc increases. In other words, it is possible to output the divided phases θe[n] more precisely, regardless of the calculation cycle of the voltage phase θe. The interval between the divided phases θe[n] can be expressed as the interval Δθe calculated using the above-mentioned equation (7).

[0047] 6 is a comparison diagram between the values ​​of the three-phase voltage commands Vu*, Vv*, and Vw* calculated by conventional control when the present invention is not applied and the values ​​of the three-phase voltage commands Vu*, Vv*, and Vw* when the present invention is applied. Here, in conventional control when the present invention is not applied, the three-phase voltage converter 18 calculates the three-phase voltage commands Vu*, Vv*, and Vw* for each calculation period Tv using the voltage phase θe.

[0048] As shown in Fig. 6, by applying the present invention, it is possible to output three-phase voltage command values ​​more precisely than with conventional control. This makes it possible to increase the switching frequency of the inverter 3. Note that the interval between three-phase voltage command values ​​in the present invention is the interval between divided phases θe[n], which can be expressed as the interval Δθe calculated by the above-mentioned equation (7).

[0049] Next, the hardware configuration of the motor control device 1 will be described below. In the motor control device 1 of this embodiment, as described above, the divided phase calculation unit 17 calculates the divided phase θe[n] (hereinafter referred to as the "divided phase calculation") at a cycle shorter than the calculation cycle Tv of the d-axis voltage command Vd* and the q-axis voltage command Vq* by the current control unit 13. Furthermore, the three-phase voltage conversion unit 18 calculates the three-phase voltage commands Vu*, Vv*, and Vw* (hereinafter referred to as the "current control calculation") at the same cycle as the divided phase calculation. Therefore, while the calculation load of the current control unit 13 remains unchanged compared to conventional control to which the present invention is not applied, there is an increase in the calculation load due to the divided phase calculation performed by the divided phase calculation unit 17 and an increase in the calculation load due to the shortened calculation cycle of the current control calculation performed by the three-phase voltage conversion unit 18, resulting in an overall increase in the calculation load. The motor control device 1 must have a hardware configuration that takes this increase in calculation load into account.

[0050] Fig. 7 is a diagram showing an example of the hardware configuration of the motor control device 1. In the motor control device 1 of this embodiment, by adopting any one of the hardware configurations shown in Fig. 7(a), 7(b), and 7(c), it is possible to realize a hardware configuration that can absorb an increase in the calculation load.

[0051] FIG. 7(a) shows an example of a hardware configuration in which current control calculations and split phase calculations are performed by separate cores within a microcomputer. In FIG. 7(a), the motor control device 1 is configured using a microcomputer having core A and core B. Core A performs calculation processing including current control calculations, while core B performs calculation processing including split phase calculations. Note that the PWM timer processing described above may be performed by either core A or core B, or may be performed by both cores in cooperation. Furthermore, other calculation processing performed by the motor control device 1 may also be performed by either core A or core B.

[0052] FIG. 7(b) shows an example of a hardware configuration in which current control calculations and PWM timer processing are performed by a microcomputer, and split phase calculations are performed by a logic calculation circuit. In FIG. 7(b), the motor control device 1 is configured by combining a microcomputer and a logic calculation circuit. The microcomputer performs calculations including current control calculations and PWM timer processing, and the logic calculation circuit performs calculations including split phase calculations. Note that other calculations performed by the motor control device 1 may be performed by either the microcomputer or the logic calculation circuit.

[0053] FIG. 7(c) shows an example of a hardware configuration in which a microcomputer performs current control calculations and a logic calculation circuit performs division phase and PWM timer processing. In FIG. 7(c), the motor control device 1 is configured by combining a microcomputer and a logic calculation circuit. The microcomputer performs calculations including current control calculations, and the logic calculation circuit performs calculations including division phase calculations and PWM timer processing. Note that other calculations performed in the motor control device 1 may be performed by either the microcomputer or the logic calculation circuit.

[0054] In the motor control device 1, by adopting any of the hardware configurations described above, the calculation unit including the divided phase calculation unit 17 and the calculation unit including the three-phase voltage conversion unit 18 can be configured using different hardware. This makes it possible to distribute the calculation load in the motor control device 1 to different hardware and absorb the increase in calculation load compared to conventional control.

[0055] According to the embodiment of the present invention described above, the following advantageous effects are achieved.

[0056] (1) The motor control device 1 is connected to an inverter 3 that converts DC power into three-phase AC power and outputs it to a motor 2, and controls the operation of the inverter 3 to control the driving of the motor 2 using the inverter 3. The motor control device 1 includes a current control unit 13 that calculates voltage commands Vd* and Vq* for the d-axis and q-axis of the motor 2 at a predetermined calculation cycle, a carrier wave generation unit 15 that generates a carrier wave, a carrier wave frequency adjustment unit 14 that adjusts the frequency fc of the carrier wave, a phase calculation unit 16 that calculates a voltage phase θe of the inverter 3 based on the rotational position θr of the motor 2, a divided phase calculation unit 17 that calculates divided phases θe[n] by dividing the voltage phase θe for each predetermined division number Ne of 2 or more, a three-phase voltage conversion unit 18 that converts the voltage commands Vd* and Vq* into three-phase voltage commands Vu*, Vv*, and Vw* based on the divided phases θe[n], and a PWM control unit 19 that pulse-width modulates the three-phase voltage commands Vu*, Vv*, and Vw* using the carrier wave to generate PWM pulse signals for controlling the operation of the inverter 3. This configuration enables the switching frequency of the inverter 3 to be increased.

[0057] (2) The motor control device 1 includes a first calculation unit including a three-phase voltage conversion unit 18 and a second calculation unit including a split-phase calculation unit 17, and the first calculation unit and the second calculation unit are preferably configured using different hardware. Specifically, for example, as shown in FIG. 7(a), the first calculation unit is preferably core A of a microcomputer and the second calculation unit is preferably core B of the microcomputer. Alternatively, for example, as shown in FIGS. 7(b) and 7(c), the first calculation unit is preferably a microcomputer and the second calculation unit is preferably a logic calculation circuit that performs a predetermined logical calculation. In this way, the calculation load of the motor control device 1 can be distributed to different hardware, and an increase in the calculation load can be absorbed.

[0058] (3) The carrier frequency adjuster 14 may adjust the frequency fc of the carrier so that the ratio Tv / Tc of the calculation period Tv of the voltage commands Vd* and Vq* to the period Tc of the carrier is an integer. In this way, the ratio Tv / Tc can be used as the division number Ne as it is, thereby further reducing the calculation load.

[0059] (4) As shown in FIG. 3(a), the division phase calculation unit 17 may have a period division unit 172 that determines the division number Ne using equation (5) based on the calculation period Tv of the voltage commands Vd* and Vq* and the frequency fc of the carrier wave, and a phase division unit 173 that calculates the interval Δθe of the division phase θe[n] using equation (7) based on the change θe1-θe0 of the voltage phase θe in the calculation period Tv of the voltage commands Vd* and Vq* and the division number Ne determined by the period division unit 172, and calculates the division phase θe[n] using equation (6) by adding an integer multiple of the calculated interval Δθe of the division phase θe[n] to the voltage phase θe. 3(b), the divided phase calculation unit 17 may include a PLL trigger output unit 174 that outputs a PLL trigger signal for each carrier wave period Tc, and a PLL calculation unit 175 that performs PLL calculation based on the PLL trigger signal output from the PLL trigger output unit 174 and updates the voltage phase θe for each period of the PLL trigger signal to calculate the divided phase θe[n]. In this way, the divided phase θe[n] can be calculated accurately with a small calculation load.

[0060] Although the above-described embodiment describes an example of application to a motor drive system mounted on and used in an electrically powered vehicle such as an electric vehicle or a hybrid vehicle, the present invention is not limited to this. The present invention can be applied to a motor control device used in any motor drive system, as long as it is connected to an inverter having a plurality of switching elements and controls the operation of the inverter to control the drive of a motor using the inverter.

[0061] Furthermore, the present invention is not limited to the above-described embodiment, and various modifications are possible within the scope of the invention. [Explanation of symbols]

[0062] 1...motor control device, 2...motor, 3...inverter, 4...rotational position detector, 5...high-voltage battery, 7...current detection unit, 8...rotational position sensor, 10...current command generation unit, 11...speed calculation unit, 12...current conversion unit, 13...current control unit, 14...carrier frequency adjustment unit, 15...carrier generation unit, 16...phase calculation unit, 17...division phase calculation unit, 18...three-phase voltage conversion unit, 19...PWM control unit, 31...inverter circuit, 32...gate drive circuit, 33...smoothing capacitor, 100...motor drive system

Claims

1. A motor control device connected to an inverter that converts DC power into three-phase AC power and outputs the converted power to a motor, and that controls operation of the inverter to control driving of the motor using the inverter, a current control unit that calculates voltage commands for the d-axis and q-axis of the motor at predetermined calculation intervals; a carrier wave generating unit that generates a carrier wave; a carrier frequency adjusting unit that adjusts the frequency of the carrier wave; a phase calculation unit that calculates a voltage phase of the inverter based on a rotational position of the motor; a divided phase calculation unit that calculates divided phases by dividing the voltage phase by a predetermined number of divisions equal to or greater than two; a three-phase voltage converter that converts the voltage command into a three-phase voltage command based on the divided phases; a PWM control unit that pulse-width modulates the three-phase voltage command using the carrier wave to generate a PWM pulse signal for controlling the operation of the inverter.

2. 2. The motor control device according to claim 1, a first calculation unit including the three-phase voltage conversion unit; a second calculation unit including the divided phase calculation unit, A motor control device in which the first calculation unit and the second calculation unit are configured using different hardware.

3. 3. The motor control device according to claim 2, the first calculation unit is a first core of a microcomputer, The second calculation unit is a motor control device that is a second core of the microcomputer.

4. 3. The motor control device according to claim 2, the first calculation unit is a microcomputer, The motor control device wherein the second calculation unit is a logic calculation circuit that performs a predetermined logical calculation.

5. 5. The motor control device according to claim 1, The motor control device, wherein the carrier wave frequency adjusting unit adjusts the frequency of the carrier wave so that the ratio of the calculation period of the voltage command to the period of the carrier wave is an integer.

6. 5. The motor control device according to claim 1, The division phase calculation unit a period division unit that determines the number of divisions based on a calculation cycle of the voltage command and a frequency of the carrier wave; a phase division unit that calculates an interval between the divided phases based on an amount of change in the voltage phase in a calculation cycle of the voltage command and the number of divisions determined by the period division unit, and calculates the divided phases by adding an integer multiple of the calculated interval between the divided phases to the voltage phase.

7. 5. The motor control device according to claim 1, The division phase calculation unit a trigger output unit that outputs a trigger signal for each cycle of the carrier wave; a PLL calculation unit that performs PLL calculation based on the trigger signal output from the trigger output unit, and updates the voltage phase for each cycle of the trigger signal to calculate the divided phase.

8. A method for controlling operation of an inverter that converts DC power into three-phase AC power and outputs the converted power to a motor, thereby controlling driving of the motor using the inverter, comprising: Calculating voltage commands for the d-axis and q-axis of the motor at predetermined calculation intervals; Adjust the carrier frequency, calculating a voltage phase of the inverter based on a rotational position of the motor; a value obtained by dividing the calculation period of the voltage command by a predetermined division number is used as a calculation period of a division phase based on the voltage phase, and the division phase is calculated; converting the voltage command into a three-phase voltage command based on the divided phases; A motor control method for generating a PWM pulse signal for controlling the operation of the inverter by pulse-width modulating the three-phase voltage command using the carrier wave.

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