Systems and methods for shaping beams produced by an antenna array

Abstract

An antenna array system and method for simulating QPSK beamforming includes a bi-phase adjuster associated with each transmit antenna configured to apply a 180 degree phase shift to a transmitted signal, and a controller configured to send instructions to the bi-phase adjuster. The receive antennas are connected to a memory operable to store the received signals and a post-processor including a processing unit operable to apply a 90 degree phase shift to selected received signals stored in the memory and further operable to sum the received signals stored in the memory. An optimized antenna array configuration is disclosed.

Description

[Technical Field]

[0001] This application claims the benefit of priority from U.S. Provisional Patent Application No. 62 / 955,487, filed December 31, 2019, U.S. Provisional Patent Application No. 63 / 037,021, filed June 10, 2020, and U.S. Provisional Patent Application No. 63 / 072,316, filed August 31, 2020, the contents of which are incorporated by reference in their entireties.

[0002] The disclosure herein relates to systems and methods for shaping beams produced by an antenna array. In particular, the disclosure relates to applying post-processing to binary phase-shifted signals to simulate multiple-phase-shift keying (MPSK)-based beamforming, such as quadrature phase-shift keying (QPSK) beamforming. An antenna array design optimized for MIMO radar-based imaging is disclosed. [Background technology]

[0003] The use of radar is becoming more and more widespread with the development of RFIC and signal technology. Radar sensors have the advantage of operating in complete darkness, fog, mist, and rain. Radar is an electronic system with the advantages of low cost, low power consumption, and high accuracy. It can be significantly used in various applications, including space shuttle topographic missions, optical systems, geotechnical mapping, meteorological detection, etc. The working efficiency of radar systems is based on reliable and stable radar signals with wide coverage, high directivity, high gain, and low signal-to-noise ratio.

[0004] The directivity achievable by an antenna depends on its physical size relative to the wavelength at the operating frequency. This is true for both mechanically and electronically steered beams. Electronic beam steering involves aligning the phase of the signal to and from the antenna elements in a given direction. The beam shape of an antenna array depends on the phase shift applied to each antenna element in the array. Typically, each antenna element has an a priori implementation-dependent phase shift associated with the transmission lines and amplifiers along the signal path to the antenna element. If no additional phase shift is applied, the resulting beam typically does not have a well-defined beam shape, and the direction from which the reflected beam is received is difficult to determine.

[0005] A well-known method for achieving highly directional beams is to apply a phase shift along each path to the corresponding antenna elements so that transmissions from different elements combine coherently in a given propagation direction. Nevertheless, applying arbitrary phase shifts introduces implementation complexity and sometimes requires resorting to coarse phase control. An example of coarse phase control is selecting one of two or four possible phases, while finer control may allow the selection of eight or sixteen phase values ​​in each phase control path.

[0006] Directivity to the transmitted beam can be achieved through binary phase shift keying (BPSK)-based beamforming. This can be achieved by applying a 0-degree or 180-degree phase shift to the signal transmitted through the selected antenna. Nevertheless, BPSK beamforming carriers typically suffer from coarse phase quantization and a large difference between the optimal desired phase and the actual phase. BPSK beamforming typically produces significant sidelobes that can waste approximately 60% of the transmitted energy. Sidelobe reduction requires finer control of the phase, for example, every 90 degrees instead of 180 degrees. With a 90-degree granularity of phase control, only 20% of the energy is lost to the sidelobes.

[0007] As an illustration, in the imaging context, a transmitting antenna may be scanned with various code sequences over several time intervals (e.g., by switching between antennas over time, or by coding the antenna with a Hadamard code, or by beamforming toward a specific direction). The directional characteristics can be recreated by recursive beamforming combined with inversion of the encoding matrix. Reflections from moving targets can produce phase rotations across these time intervals in a manner detrimental to imaging. The rationale for creating a good beamformer comes from concentrating energy in different directions in each time interval, reducing the effects of phase rotation. Furthermore, when a transmit sweep across a range of frequencies, such as an up-chirp or down-chirp, is transmitted over a time period, the delay between time intervals is further increased.

[0008] As a result, it can be very difficult to pinpoint the location of a target. [Prior art documents] [Patent documents]

[0009] [Patent Document 1] U.S. Patent No. 7,483,367 [Patent Document 2] U.S. Patent No. 10,020,836(B2) [Patent Document 3] U.S. Patent No. 10,804,954(B2) Summary of the Invention [Problem to be solved by the invention]

[0010] Therefore, there remains a need for an effective radar sensor that can be used to accurately detect objects in the area surrounding a moving vehicle. The invention described herein addresses the above-mentioned need. [Means for solving the problem]

[0011] According to one aspect of the subject matter of the present disclosure, a system for shaping beams produced by an antenna array, for example by reducing sidelobes, is introduced. Various systems may include at least one bi-phase adjuster. The phase adjuster or the like may be configured and operable to selectively apply a 180-degree phase shift to a transmitted signal. A controller may be configured to send instructions to the at least one bi-phase adjuster. The system may also include at least one receive antenna, a memory operable to store received signals, and a post-processor operable to apply a 90-degree phase shift to selected received signals stored in the memory, and further operable to sum the received signals stored in the memory.

[0012] In yet another aspect, a method is taught for simulating quadrature phase-shift keying (QPSK) beamforming in an antenna array, where each antenna of the array is connected to a common transmitter through a biphasic modulator. The method can include determining a complex QPSK steering vector required for each transmitting antenna of the array. The steering vector typically has a real component selected from 0 degrees and 180 degrees and an imaginary component selected from 90 degrees and 270 degrees.

[0013] Thus, the transmitter generates an oscillating signal. During a first time interval, for each transmit antenna having a steering vector associated with a real component of 180 degrees, the bi-phase adjuster applies a 180-degree phase shift to the transmitted signal. During a second time interval, for each transmit antenna having a steering vector associated with an imaginary component of 180 degrees, the bi-phase adjuster applies a 180-degree phase shift to the transmitted signal. A post-processor may be used to apply a 90-degree phase shift to the signal received during the second time interval, and the post-processor may sum the signal received during the first time interval with the 90-degree phase-shifted signal received during the second time interval. Optionally, the transmitter may sweep the oscillating signal over a range of frequencies during each time interval.

[0014] Thus, according to various examples, a controller may provide a system for simulating quadrature phase shift keying (QPSK) beamforming in an antenna array, the system comprising a transmitter and an antenna. The controller may be operable to determine a complex steering vector required for each antenna of the array, the complex steering vector including a real component selected from 0 degrees and 180 degrees and an imaginary component selected from 90 degrees and 270 degrees; to instruct a binary phase adjuster to apply a 180-degree phase shift to a transmitted signal for each antenna having a steering vector associated with a 180-degree real component during a first time interval; and to instruct the binary phase adjuster to apply a 180-degree phase shift to a transmitted signal for each antenna having a steering vector associated with a 180-degree imaginary component during a second time interval; and the post processor is operable to apply the 90-degree phase shift to signals received during the second time interval and to sum signals received during the first time interval with the 90-degree phase shifted signals received during the second time interval.

[0015] Optionally, a dedicated two-phase adjuster is provided for each transmit antenna of the array. Additionally or alternatively, an independently controlled switch connects each transmit antenna to the oscillator.

[0016] If appropriate, a gain control unit may be associated with each transmit antenna, and the controller may be further configured to send instructions to the gain control unit. Thus, for each transmit antenna, the controller may be operable to select a required amplitude AR for the real component of the associated steering vector, instruct the associated gain control unit to apply a first gain GR to the transmitted signal during a first time interval, select a required amplitude A1 for the imaginary component of the associated steering vector, and instruct the associated gain control unit to apply a second gain G1 to the transmitted signal during a second time interval, where the second gain G1 is equal to the product of GR and the ratio of A1 to AR. Variously, each antenna may have a dedicated bi-phase adjuster, or a common bi-phase adjuster may be connected to multiple antennas.

[0017] In yet another aspect, a method is taught for simulating quadrature phase-shift keying (QPSK) beamforming in an antenna array, where each antenna of the array is connected to a common transmitter through a binary phase shifter. The method can include determining a required complex QPSK steering vector for each transmitting antenna of the array. The steering vector typically has a real component selected from 0 and 180 degrees and an imaginary component selected from 90 and 270 degrees.

[0018] Thus, the transmitter generates an oscillating signal. During a first time interval, for each transmit antenna having a steering vector associated with a real component of 180 degrees, the bi-phase adjuster applies a 180-degree phase shift to the transmitted signal. During a second time interval, for each transmit antenna having a steering vector associated with an imaginary component of 180 degrees, the bi-phase adjuster applies a 180-degree phase shift to the transmitted signal. A post-processor may be used to apply a 90-degree phase shift to the signal received during the second time interval, and the post-processor may sum the signal received during the first time interval with the 90-degree phase-shifted signal received during the second time interval. Optionally, the transmitter may sweep the oscillating signal over a range of frequencies during each time interval.

[0019] For a better understanding of the embodiments, and to show how they may be carried into effect, reference will now be made, purely by way of example, to the accompanying drawings in which:

[0020] With specific reference now to the drawings in detail, it is emphasized that the details shown are presented by way of example and for illustrative discussion only of selected embodiments, to provide what is believed to be the most useful and easily understood explanation of principles and conceptual aspects. In this regard, no attempt is made to show structural details in more detail than necessary for a fundamental understanding, and the description taken together with the drawings will make clear to those skilled in the art how various selected embodiments may be practiced. [Brief explanation of the drawings]

[0021] [Figure 1A] FIG. 10 illustrates how steering vectors can be generated by BPSK phase shifting the phase of selected antennas by 0 or 180 degrees. [Figure 1B] FIG. 1 illustrates a possible BPSK mechanism for phase shifting the signal to the antenna by 180 degrees. [Figure 1C]FIG. 10 shows how steering vectors can be generated by QPSK phase shifting the phase of selected antennas by 0, 90, 180, or 270 degrees. [Figure 1D] FIG. 1 illustrates possible quadrature modulation mechanisms for phase shifting the signal to the antenna by 0, 90, 180, or 270 degrees. [Figure 2A] 1 is a block diagram that schematically represents selected elements of a first embodiment of a system for simulating quadrature phase shift keying (QPSK) beamforming. [Figure 2B] 10 is a set of graphs illustrating a possible set of profiles illustrating how the phase of the signal transmitted from each transmit antenna of the first embodiment may vary over time. [Figure 2C] 4 is a flowchart illustrating selected steps in a method for simulating quadrature phase shift keying (QPSK) beamforming in a first embodiment system. [Figure 3A] FIG. 10 is a block diagram that schematically represents selected elements of a second embodiment of a system for simulating quadrature phase shift keying (QPSK) beamforming, with each antenna connected to a gain control unit. [Figure 3B] 10 is a set of graphs illustrating a possible set of profiles illustrating how the phase of the signal transmitted from each transmit antenna of the second embodiment may vary over time. [Figure 3C] 10 is a flowchart illustrating selected steps in a method for simulating quadrature phase shift keying (QPSK) beamforming in a second embodiment system. [Figure 4A] FIG. 10 is a block diagram of a system including a common, shifted dual phase shared by all antennas according to a third embodiment. [Figure 4B] 10 is a set of graphs illustrating a possible set of profiles illustrating how the phase of the signal transmitted from each transmit antenna of the third embodiment may vary over time. [Figure 5]FIG. 1 is a block diagram of a system including clustered transmit antennas connected to a common oscillator via dedicated cluster modulators. [Figure 6A] 10 is a graph showing array power factor for a 32-element array with in-phase excitation. [Figure 6B] 10 is a graph showing array power factor for a 32-element array with random phase excitation. [Figure 6C] 10 is a graph showing the array power factor of a 32-element array with complementary sequence-based excitation. [Figure 7A] A diagram showing a possible transmit antenna array. [Figure 7B] A diagram showing a possible square transmit antenna array. [Figure 7C] A diagram showing a possible two-row transmit antenna array. [Figure 7D] A diagram showing a possible staggered two-row antenna array. [Figure 7E] A diagram showing a possible L-shaped MIMO antenna array. [Figure 7F] A diagram showing a possible pie-shaped MIMO antenna array. [Figure 7G] A diagram showing a possible frame-shaped MIMO antenna array. [Figure 7H] A diagram showing a possible split-frame MIMO antenna array. [Figure 8A] 1A-1C illustrate various topologies for the placement of antennas in a Multiple Input Multiple Output (MIMO) array. [Figure 8B] 1A-1C illustrate various topologies for the placement of antennas in a multiple-input multiple-output (MIMO) array. [Figure 8C] 1A-1C illustrate various topologies for the placement of antennas in a multiple-input multiple-output (MIMO) array. [Figure 8D] 1A and 1B show possible alternative topologies for asymmetric placement of antennas in a multiple-input multiple-output (MIMO) array. [Figure 8E]1A and 1B show possible alternative topologies for asymmetric placement of antennas in a multiple-input multiple-output (MIMO) array. [Figure 8F] 1A and 1B show possible alternative topologies for asymmetric placement of antennas in a multiple-input multiple-output (MIMO) array. [Figure 8G] 1A-1C illustrate possible alternative topologies for asymmetric placement of antennas in a multiple-input multiple-output (MIMO) array, showing possible fields of view for such an asymmetric placement. [Figure 8H] 1A and 1B show possible alternative topologies for asymmetric placement of antennas in a multiple-input multiple-output (MIMO) array. [Figure 9A] FIG. 1 is a schematic diagram showing how a virtual array is constructed with virtual elements at the midpoint between each pair of transmit and receive antennas. [Figure 9B] FIG. 1 is a schematic diagram showing how a virtual array is constructed with virtual elements at the midpoint between each pair of transmit and receive antennas. [Figure 10A] Figure 10 shows shifting the line segments in the frame array to obtain a continuous virtual array excluding the central void. [Figure 10B] Figure 10 shows shifting the line segments in the frame array to obtain a continuous virtual array excluding the central void. [Figure 11A] 10A-10C illustrate how auxiliary subarrays can be used to fill in center voids, according to various embodiments. [Figure 11B] 10A-10C illustrate how auxiliary subarrays can be used to fill in center voids, according to various embodiments. [Figure 12A] FIG. 1 shows an exemplary frame array that can serve as a mother array to guide the construction of synthetic arrays in an array-of-arrays format. [Figure 12B] A diagram showing possible synthetic arrays corresponding to the mother array in Figure 12A. [Figure 13A]Figure 1 shows possible linear arrays that can serve to guide the construction of synthetic arrays in an array-of-arrays fashion. [Figure 13B] A diagram showing possible synthetic arrays corresponding to the mother array in Figure 13A. DETAILED DESCRIPTION OF THE INVENTION

[0022] Aspects of the present disclosure relate to systems and methods for shaping transmit beams produced by a radar array. In particular, systems and methods are described for reducing sidelobes and increasing gain and phase linearity over super-hemispherical radar coverage.

[0023] To reduce sidelobes, the signals transmitted by each antenna of the array can be biphase shifted according to a required time phase shift profile. For example, to simulate multiple phase shift beamforming, such as quadrature phase shift keying (QPSK) beamforming, post-processing methods can be applied to the received reflected signals over multiple time periods. Typically, the receiver and transmitter can be synchronized to produce consistent results during the time interval in which the signals are combined.

[0024] Where necessary, detailed embodiments of the present invention are disclosed herein, however, it will be understood that the disclosed embodiments are merely illustrative of the invention, which may be embodied in various and alternative forms. The figures are not necessarily drawn to scale, and some features may be exaggerated or minimized to show details of particular components. Therefore, the specific structural and functional details disclosed herein should not be construed as limiting, but merely as a representative basis for teaching those skilled in the art to various uses of the present invention.

[0025] In various embodiments of the present disclosure, one or more tasks as described herein may be performed by a data processor, such as a computing platform or distributed computing system for executing a plurality of instructions. Optionally, the data processor includes or has access to volatile memory for storing instructions, data, or the like. Additionally or alternatively, the data processor may have access to non-volatile storage, such as, for example, a magnetic hard disk, flash drive, removable media, or the like, for storing instructions and / or data.

[0026] It is expressly noted that the systems and methods of the present disclosure herein may not be limited in their application to the details of construction and the arrangements of components or methods set forth in the Detailed Description or illustrated in the drawings and examples. The systems and methods of the present disclosure may be capable of other embodiments or of being practiced and carried out in various ways and techniques.

[0027] Alternative methods and materials similar or equivalent to those described herein can be used in the practice or testing of embodiments of the present disclosure. Nevertheless, specific methods and materials are described herein for illustrative purposes only. The materials, methods, and examples are not necessarily intended to be limiting.

[0028] Alternative methods and materials similar or equivalent to those described herein may be used in the practice or testing of embodiments of the present disclosure. Nevertheless, certain methods and materials are described herein for illustrative purposes only. The materials, methods, and examples are not necessarily intended to be limiting. Thus, various embodiments may omit, substitute, or add various procedures or components, as appropriate. For example, methods may be performed in an order different from that described, and various steps may be added, omitted, or combined. Furthermore, aspects and components described with respect to certain embodiments may be combined in various other embodiments.

[0029] Reference is now made to FIGS. 1A and 1B. FIG. 1A illustrates how a steering vector can be generated by BPSK phase shifting. Without artificial phase shifting, an array of antennas can produce a range of phase shifts due to the nature of electronic circuits and the like, in addition to the phase produced by wave propagation in the desired steering direction (called the “array factor”). This range of phases is represented by the circled area in FIG. 1A(i). The phasors shown in the figure do not consistently add up. By selectively adding a 180-degree phase shift to all antennas producing phases within the left side of the circle, these phasors can be partially aligned, as shown in FIG. 1A(ii), thereby radiating energy toward the desired steering direction. Thus, each antenna 1116 in the array may be connected to a signal generation oscillator 1112 via a biphasic adjuster 1114, as shown in FIG. 1B. While the BPSK mechanism can actually generate the steering vector 1110, the resulting beam suffers from significant sidelobes and significant loss.

[0030] By providing additional phase shift options, more efficient steering vectors can be generated. Referring to Figures 1C and 1D, a range of phases such as that shown in Figure 1C(iii) can be converted to a net steering vector 1130, such as that shown in Figure 1C(iv), by selectively shifting each transmit signal by 0, 90, 180, or 270 degrees (QPSK), as needed.

[0031] 1D shows a possible hardware arrangement 1140 for producing such a phase shift in the antennas 1148 of the array. Each antenna 1148 of the array may be connected to a signal generating oscillator 1142 via a phase shift mechanism having two parallel arms: an in-phase arm (Re) and a quadrature arm (Im).

[0032] The in-phase arm (Re) includes a first bi-phase adjuster 1144 that can be selectively activated to add a 180 degree phase shift to the oscillating signal, if desired. Alternatively, by not activating the first bi-phase adjuster, the signal is transferred in phase to the transmit antenna.

[0033] The quadrature arm (Im) includes a second bi-phase adjuster 1146 and a quarter-cycle phase adjuster 1145. The quarter-cycle phase adjuster 1145 is configured to add a 90-degree phase shift to the oscillating signal. Thus, when the second bi-phase adjuster 1146 is not activated, a 90-degree phase shift is applied to the signal transferred to the antenna. Alternatively, when the second bi-phase adjuster is activated to add an additional 180-degree phase shift, a total phase shift of 270 degrees is applied to the signal transferred to the antenna, as needed.

[0034] It will be appreciated that such a hardware quadrature modulation mechanism, such as that shown in Figure 1D, can significantly improve the overall steering vector. Nevertheless, the arrangement requires significantly more hardware elements than the simple two-phase adjuster 1120 of Figure 1B. The addition of a quadrature arm for each antenna, including a quarter-cycle phase adjuster that may need to be located physically close to the antenna itself, places significant hardware limitations on the designer of the antenna array circuit.

[0035] A possible solution for generating improved steering vectors using only two-phase adjuster elements is now described.

[0036] 2A, there is shown selected elements of a first embodiment of a system for simulating quadrature phase shift keying (QPSK) beamforming in an antenna array 1200. The system includes a transmitter 1250, an antenna array 1210, a bi-phase adjuster 1220 associated with each transmit antenna, a controller 1230, a receive antenna 1240, and a post processor 1260.

[0037] Transmitter 1250 is configured and operable to generate an oscillating signal for transmission by antenna array 1210. It should be noted that, if appropriate, transmitter 1250 may be further operable to generate a signal that sweeps through a range of frequencies, or a chirp.

[0038] The antenna array 1210 includes several antennas A1 to An, each operable to transmit a signal generated by an oscillator 1270 with a required phase shift. It will be noted that the superposition of the signals transmitted from all of the antennas in the array results in a general signal beam having a characteristic shape.

[0039] The bi-phase adjuster 1220 associated with each transmit antenna An is configured and operable to selectively apply a 180 degree phase shift to the oscillating signal as needed. Alternatively, if the bi-phase adjuster 1220 is not activated, no phase shift is applied to the oscillating signal. Thus, the signal transmitted by the associated antenna is in phase or out of phase with the oscillating signal produced by the oscillator 1270 as needed.

[0040] The controller 1230 is configured to send activation commands to the bi-phase adjuster 1220 so that only the required antennas transmit phase-shifted signals.

[0041] The receive antenna(s) 1240 are configured to receive return signals reflected from the target.

[0042] The post processor 1260 is operable to manipulate the received signals and includes a memory 1280 element and a processing unit 1290. The memory element 1280 is operable to store the received signals. The processing unit is operable to apply a phase shift to selected received signals stored in the memory 1280 and is further operable to sum the received signals stored in the memory 1280.

[0043] In a particular example, the processing unit may apply a 90 degree phase shift to selected received signals and sum these with other received signals to produce the required output signal.

[0044] Thus, the controller may be operable to determine a complex steering vector C=R+jI required for each antenna of the array. The complex steering vector C includes a binary real component R selected from +1 and −1, and a binary imaginary component I selected from +1 and −1. A value of +1 indicates no phase shift is required, and a component of −1 indicates a phase shift is required. Thus, the real component can represent a required phase shift selected from 0 and 180 degrees, and the imaginary component can represent a required phase shift selected from 90 and 270 degrees, all associated with the combination R=+1, I=+1.

[0045] Referring now to the graph of FIG. 2B, there is shown a possible set of profiles illustrating how the phases S1 to Sn of the signals transmitted from each of the transmit antennas A1 to An in the first embodiment may vary over time.

[0046] Note that the phase shift of each antenna remains fixed for a given time interval Δt. Each antenna A receives a unique profile determined by the required steering vector C at that time. Each complex steering vector C may determine the required phase shift between two consecutive time intervals Δt, Δt+1.

[0047] During a first time interval Δti, the controller instructs the bi-phase adjusters 1220 of antennas A1-An having a steering vector Ci associated with a real component Ri of −1 to apply a 180 degree phase shift to the transmitted signal.

[0048] During a second time interval Δti+1, the controller instructs the antenna's bi-phase adjuster 1220 having a steering vector Ci associated with an imaginary I component of −1 to apply a 180 degree phase shift to the transmitted signal.

[0049] Thus, the post processor 1260 may be operable to store in memory the reflected signals received during the first time interval and the second time interval, and the processor unit may then apply a 90 degree phase shift to the signal received during the second time interval before summing the signal received during the first time interval with the 90 degree phase shifted signal received during the second time interval.

[0050] The resulting output signal from the post processor will have the characteristics of a quadrature-shifted signal.

[0051] Referring now to the flowchart of FIG. 2C, selected steps of a method 1400 for simulating quadrature phase shift keying (QPSK) beamforming in the system of FIG. 2A in which the antennas of the array 1210 are connected to a common transmitter via respective bi-phase adjusters 1220 are shown.

[0052] For each transmit antenna of the array, a required complex QPSK steering vector C is determined 1410, which includes a real component R selected from +1 and −1, and a binary imaginary component I selected from +1 and −1.

[0053] The transmitter generates an oscillating signal that is transmitted to each antenna via a two-phase modulator 1420. Optionally, each transmitted signal may sweep over a range of frequencies during each time interval.

[0054] During the first time interval 1430, for each transmit antenna having a steering vector associated with a real component R of +1, the associated bi-phase adjuster applies a 180 degree phase shift to the transmitted signal 1432, the antenna transmits the signal 1434, and the received signal is stored in a post processor memory 1436.

[0055] During a second time interval 1440, for each transmit antenna having a steering vector associated with an imaginary component I of +1, the associated bi-phase adjuster applies a 180 degree phase shift to the transmitted signal 1442, the antenna transmits the signal 1444, and the received signal is stored in a post processor memory 1446.

[0056] The post processor may then apply 1450 a 90 degree phase shift to the signal received during the second time interval 1440 and sum 1460 the signal received during the first time interval to the 90 degree phase shifted signal received during the second time interval.

[0057] A particular feature of the systems and methods described herein is the linear combination of signals received over several time intervals in order to simulate and benefit from the advantages of an enhanced beamformer in a simulated manner. This feature can be extended in various ways that will be apparent to those skilled in the art and are mentioned here as examples.

[0058] In one extension, if the transmitter already supports beamforming using a particular choice of phases (e.g., 4-phase QPSK, 8-phase 8-PSK, etc.) or gain, then a combination of M codewords (two or more) over M time intervals can be used to generate a larger choice of phases by a factor M (e.g., using four time intervals with BPSK or two time intervals with QPSK to generate simulated 8-PSK).

[0059] The simulated QPSK scheme can alternatively be calculated by taking the desired phasor C for each transmit antenna and calculating X = Sgn(Re{C·e -jφ}), where φ is 0 during the first interval and 90 degrees during the second interval, and then jφIn another extension of the invention, the sequence of "modulation" phases φ can be chosen in different ways over time, for example at different scanned frequencies or frames.

[0060] In another extension of the present invention, the signal received over M intervals may be modulated with arbitrary phasors a1,..a2 that do not necessarily have unit gain (rather than a1=1, a2=j as in the QPSK case as described herein). M The beamforming codewords over these M intervals are a1,..a M These linear combinations, weighted by , are chosen in a way that produces the desired characteristics (such as high peak to sidelobe levels).

[0061] The above-described method for implementing QPSK (quadrature) beamforming by using two phase adjusters and two time intervals is presented for illustrative purposes only. This method can be further generalized to implement any even number 2n of phases over n time intervals. For example, with three time intervals, 6-PSK modulation can be realized.

[0062] A method may be implemented in which the transmitter selectively applies a 180 degree phase shift to a particular antenna during N time intervals according to the following condition: In the nth time interval, if the real value of the steering vector rotated by n*180 / N degrees is negative, then the 180 degree phase shift is applied to the kth antenna. Thus, a 180 degree phase shift is applied to the kth antenna if the following formula is true: Real(C k *e -j*φ[n] )<0 where C k is the k-th component of the steering vector, and φ n =πn / N is the rotation sequence.

[0063] Therefore, if appropriate, in a post-processor, a rotation of φn radians for the nth time interval can be applied before summing the signals received in all time intervals.

[0064] Methods such as those described herein can be extended to incorporate additional measures for the desired beamformer by choosing a set of N phase shift sequences such that the median value of the signal transmitted over N time intervals meets the desired measure. For example, effective attenuation for each unique transmit antenna may be required for gain control for apodization and transmitter gain equalization. This can be achieved without analog gain control by using a unique rotation sequence for each unique transmit antenna, e.g., the steering vector for each antenna may be rotated by, for example, an angle step (1-a)*φn, where the value of a is specifically chosen to suit each transmit antenna.

[0065] The multiple time intervals necessary to apply the described method may also be used for other purposes. In one possible embodiment, multiple time intervals may be used to allow for Doppler processing within each frame, to allow for integration times that may be longer than the channel coherence time, and to obtain information about target velocities. Each spatial transmitter direction to be scanned may include N time intervals, and Doppler post-processing may search for a linear phase shift between the intervals that may correspond to the radial velocity. This may be implemented, for example, using a Fast Fourier Transform (FFT) over the time intervals.

[0066] Note that, if appropriate, each time interval may itself involve sweeping the transmitted signal over multiple frequencies using a stepped frequency continuous wave, chirp, or some other frequency function over time during the time interval. Thus, by varying the beamformer between time intervals as described above, sidelobe levels should typically be reduced by phase quantization at any given rate. However, associated beamforming quantization errors may still produce sidelobes at other rates.

[0067] Another feature of the present method is that the spectral shape of the generated sidelobes can be controlled by selecting a specific order for the time intervals so that the quantization noise that generates the sidelobes is largely confined to high frequencies corresponding to higher radial velocities than expected in the particular application. If desired, the phase rotation φn for the nth time interval (where n can take any integer value from 0 to N-1) can be selected as follows: φ n =π * [(n * (N-1) / 2)mod N] Here, "mod" is the modulo operation that returns the remainder when divided by a given integer, and it is assumed that N is an integer multiple of 4. As mentioned above, a 180 degree rotation is calculated by dividing Real(Ck * ej * The post-processor applies a rotation by φn only if φ[n]<0. With this reordering of the time intervals, most of the sidelobe power resides at the Nyquist frequency of the Doppler.

[0068] In applications where higher level sidelobes at the Nyquist frequency of Doppler must be avoided in constructions similar to those mentioned above, a priori estimation of the exact location of these sidelobes in 4D space can be used to constrain and distinguish between real targets and method artifacts without significantly degrading the radar's dynamic range. One such example is the Doppler component v m 4D voxels suspected of having

number

number

[0069] It will be appreciated that other structures may be used to select the phase rotation sequence, or the order in the sequence of steering vectors, to optimize the spectral shape of the quantization noise, as appropriate.

[0070] In the above structure, the known required steering vector is rotated by φ for the two-phase selection. An alternative approach is to use Real(H({b k}) * exp(j * φ n ))) at the maximum value l, where H({b k}) is the unique phase selection b kis a phasor representing the combination of all transmitting antennas in the desired spatial direction. Such a maximization can be done, for example, by K H can be implemented in various ways, such as by an exhaustive search over all two-phase combinations (with K transmit antennas with the option of H = 1). H can be obtained, for example, by analysis of direct measurements of electromagnetic waves reflected by a reference target located in the desired spatial direction.

[0071] The number of time intervals may be selected to achieve the required beamforming accuracy, for example, in terms of sidelobe level, signal-to-noise ratio (SNR) (possibly using longer integration times with additional intervals), and Doppler estimation resolution. On the other hand, the number of time intervals may be limited by other factors, such as the memory capacity and processing power of the electronic components and the avoidance of blurring within the frame. Therefore, the actual number of time intervals selected may be a compromise between all these considerations.

[0072] Since some spatial directions may be more important than others, in terms of the required SNR and Doppler resolution, it may be preferable to allocate more time intervals to these preferred directions and fewer time intervals to other lower priority directions.

[0073] This scanning method may be used in a variety of applications, such as external car radar sensors used for ADAS (Advanced Driver Assistance Systems) or autonomous driving. It will be appreciated that in such applications, the horizontal angular range of interest (azimuth range) is typically wider than the vertical angular range of interest (altitude range). This is because car radar sensors generally do not need to scan below the road surface. Therefore, it may be preferable to align the transmitter antenna in a vertical linear array so that the side lobes are outside the high preferred altitude range. The receiver antenna may be arranged in a horizontal linear array oriented at right angles.

[0074] Other possible applications may include monitoring enclosed spaces such as rooms, stadiums, goal lines, or the like. Still other applications may involve tracking objects within a target area, possibly using large arrays for body scanning. Still other applications will occur to those skilled in the art.

[0075] Applying an acquisition scheme that relies on beamforming can be particularly efficient in terms of constructing 4D images with limited available processing and memory capacity, or alternatively, significantly increasing the scanned 4D volume and optimizing the SNR. One such realization would be to build an acquisition processing pipeline in which a unique angular slice or solid angle (beam [n]) from [N] beams is illuminated through beamforming while processing the previous beam (n-1). This provides advantages for applications such as those described above, allowing the use of simpler processing units and reducing memory constraints as well as the resulting heat dissipation and product size.

[0076] Such interleaving schemes can be extended to vary the acquisition profile for each beam, allowing coverage of larger and more complex arenas, for example, allocating few beams to short-range, wide-field-of-view, high-resolution, and slow-velocity targets, and allocating other beams to long-range, high-velocity, narrow-field-of-view, and limited-resolution scenarios. Interleaving beams and profiles can be managed at any particular level within a single frame or between each successive frame.

[0077] Reference is now made to the block diagram of FIG. 3A, which schematically illustrates selected elements of a second embodiment of a system in which each antenna is connected to a gain control unit 1550 such that quadrature amplitude modulation (QAM) beamforming can be simulated.

[0078] 2A, a dedicated gain control unit 1530 is associated with each transmit antenna, and the controller is further configured to instruct the gain control unit to amplify the transmitted signal by a required gain determined by the complex steering vector.

[0079] The controller may also be operable to determine a required complex steering vector C=R+jI for each antenna of the array, where the steering vector may still have a continuous real component R selected from the range +1>R>−1 and a continuous imaginary component I selected from the range +1>I>−1.

[0080] Thus, the controller may be further operable to select a required amplitude R for the real component of the associated steering vector and to instruct the associated gain control unit to apply the associated first gain GR to the transmitted signal during a first time interval. Similarly, the controller may be operable to select a required amplitude I for the imaginary component of the associated steering vector and to instruct the associated gain control unit to apply a second gain GI to the transmitted signal during a second time interval, the second gain GI being equal to the product of GR and the absolute ratio of I to R.

[0081] Referring to the set of graphs shown in FIG. 3B, the resulting signal produced by each antenna during each time period may therefore be amplitude modulated and phase modulated.

[0082] Referring now to the flowchart of FIG. 3C, selected steps of a method for simulating quadrature amplitude modulation (QAM) beamforming in the system of FIG. 3A in which the antennas of the array are each connected to a common transmitter via associated binary phase adjusters and gain control units 1530 are shown.

[0083] For each transmit antenna of the array, a required complex QPSK steering vector C is determined 1610, having a real component R selected from the range +1>R>-1 and an imaginary component I selected from the range +1>I>-1.

[0084] The transmitter generates an oscillating signal that is transmitted to each antenna via a two-phase modulator 1620. Optionally, each transmitted signal may sweep over a range of frequencies during each time interval.

[0085] During a first time interval 1630, for each transmit antenna having a steering vector associated with a negative real component R, an associated bi-phase adjuster applies a 180 degree phase shift to the transmitted signal 1632. An associated gain control unit amplifies the signal by a first value GR = |R|G0 1633, the antenna transmits the amplified signal 1634, and the received signal is stored 1636 in a post processor memory.

[0086] During a second time interval 1640, for each transmit antenna having a steering vector associated with a negative imaginary component I, the associated bi-phase adjuster applies a 180 degree phase shift to the transmitted signal 1642. The associated gain control unit amplifies the signal by a second value GI = |I|G0 1643. The antenna then transmits the amplified signal 1644, and the received signal is also stored 1646 in the post processor memory.

[0087] Thus, if the post processor applies 1650 a 90 degree phase shift to the signals received during the second time interval 1640 and sums 1660 these signals with the signals received during the first time interval, the resulting signal can have a virtual phase shift of any value desired.

[0088] It should be further noted that while the system described herein includes a dedicated dual phase adjuster for each antenna, an alternative system without phase adjusters may operate by utilizing additional time intervals, as shown in FIG. 4A.

[0089] The use of such a system can be enabled by activating only those antennas with a real component of +1 during a first time period with no phase shift, activating only those antennas with a real component of −1 during a second time period with a 180 degree phase shift applied at the receiver, activating only those antennas with an imaginary component of +1 during a third time period with no phase shift, and activating only those antennas with an imaginary component of −1 during a fourth time period with a 180 degree phase shift applied at the receiver.

[0090] It should be further noted that if each antenna in the system has an independently controllable connection switch 1740, such as that shown in Figure 4A, it may be possible to apply such a phase shift directly from the oscillator 1770 or during post-processing. Additionally or alternatively, a common bi-phase adjuster may be connected to multiple transmit antennas, if desired.

[0091] An example signal profile produced by such an example system is presented in Figure 4B. The post processor may store in memory the received signals from each of the first time period, the second time period, the third time period, and the fourth time period.

[0092] The four signals can be summed by the receiver after applying 0, 180, 90, and 270 degree phase shifts in the first, second, third, and fourth steps, respectively. By summing all these signals, a simulated QPSK steering vector without a phase adjuster can be realized in the system.

[0093] Further extensions of linear combinations of signals received over multiple time intervals will occur to those skilled in the art.

[0094] Referring now to the block diagram of FIG. 5, each antenna A is connected to a gain control unit 530 which includes an antenna-specific amplifier 532 and a phase adjuster 534. 1~5 , B 1~5 5A and 5B schematically represent selected elements of a multiplexed embodiment of a system 500 in which

[0095] In the multiplexed embodiment system 500, each antenna A 1~5 , B 1~5 It is further noted that the antennas may be grouped into clusters 540A, 540B. Each cluster 540A, 540B may be connected to a phase-locked loop chirp oscillator (PLL) 510 via a dedicated cluster modulator 520A, where each antenna cluster is connected to

[0096] It will be appreciated that such a clustered system 500 may combine frequency modulated continuous wave (FMCW) transmission with frequency division multiplexed (FDM) radar techniques, and it is a feature of such an embodiment that it can provide both beamforming and simultaneous transmission, if desired.

[0097] Note further that phased array theory dictates that the spatial beam pattern is governed by the product of an "element factor" and an "array factor." The "element factor" is usually wide and results from the radiation pattern of a single antenna element that illuminates a significant portion of space. The array factor is

number

[0098] Thus, a phased array system can be designed to maximize the power transmitted in a given direction by choosing a phase that will cause transmission to be consistent in that direction (hence the name "phased array"). In such a case, N elements are used to transmit power in N squared (N instead of N) at the expense of other directions in space. 2 ) can produce a power accumulation factor.

[0099] Note that in various situations it is beneficial to enjoy the increased aggregate power of N transmitters, but to avoid consistent accumulation of power in a given direction. Illustratively, these include situations where it is necessary to limit the effective radiated power so as not to exceed safety thresholds, to avoid interference with other systems elsewhere in the space, or to avoid exceeding regulatory thresholds for permitted EIRP.

[0100] It is therefore useful to design excitation phase combinations that result in a reduction of the maximum EIRP rather than an increase. In some cases, it is desirable for a family of excitation phased arrays to meet this criterion.

[0101] A randomly chosen array may avoid perfectly coherent accumulation, but may also leave some "bright spots", similar to a laser speckle pattern. With many elements N and a randomly chosen phase, the radiation in the strongest direction may be increased by a factor of approximately ln(N) on average. Therefore, if tight control of the maximum EIRP is desired, a more systematic approach is required.

[0102] The theory of constellations with low peaks in their Fourier transforms has been well developed. This is sometimes called the Peak-to-Average Power Ratio (PAPR) problem, since the power in the transformed domain is similar to the power in the original domain. One domain where this theory has been successfully applied is in the field of Orthogonal Frequency Division Multiplexing (OFDM) modulation, where constellations in the frequency domain are converted to the time domain, and it is desirable to avoid large peaks to avoid distortion in power amplifiers.

[0103] It should be further noted that low-PAPR sequences can be used for peak EIRP reduction. U.S. Patent No. 7,483,367 (by N. Chayat et al.), incorporated herein by reference in its entirety, describes the construction of a family of low-PAPR sequences for use as "preambles" in OFDM transmissions. The construction is based on the "complementary sequences" invented by Marcel J. Golay in the 1940s. Complementary sequences are pairs of sequences whose autocorrelation results in a delta function. Consequently, each of the sequences clearly has a PAPR of at most 2 (3 dB). An introduction to the properties of complementary sequences can be found in U.S. Patent No. 7,483,367 and its references, all of which are incorporated herein by reference in their entirety.

[0104] A well-known construction allows the generation of a family of sequences of many lengths. The well-known "doubling construction" is m It allows you to generate an array of A n and B n is a complementary sequence of length n, then A 2n =[A n B n ] and B 2n =[An -B n] is a complementary sequence of length 2n. Thus, if

[0011] and [1 -1] are complementary sequences of length 2, then [1 1 1 -1] and [1 1 -1 1] are complementary sequences of length 4, and [1 1 1 -1 1 1 -1 1] and [1 1 1 -1 -1 -1 1 -1] are complementary sequences of length 8.

[0105] The doubling construction for complementary sequences is closely related to Hadamard matrices and Hadamard codes. For example, the rows of the Hadamard matrix H[8*8] are: 1 1 1 1 1 1 1 1 1 -1 1 -1 1 -1 1 -1 1 1 -1 -1 1 1 -1 -1 1 -1 -1 1 1 -1 -1 1 1 1 1 1 1 1 1 1 1 -1 1 -1 1 -1 1 -1 1 1 -1 -1 1 1 -1 -1 1 -1 -1 1 1 -1 -1 1

[0106] When multiplied by the complementary sequence [1 1 1 -1 1 1 -1 1], in the resulting matrix C[8*8]: 1 1 1 -1 1 1 -1 1 1 -1 1 1 1 -1 -1 -1 1 1 1 -1 -1 -1 1 -1 1 -1 1 1 -1 1 1 1 1 1 -1 1 1 1 1 -1 1 -1 -1 -1 1 -1 1 1 1 1 -1 1 -1 -1 -1 1 1 -1 -1 -1 -1 1 -1 -1

[0107] All rows are complementary sequences, preserving the property of the original Hadamard matrix that the rows are orthogonal to each other. Incidentally, this method of generating a family of complementary sequences relates to the CCK (Complementary Code Modulation) method used in the 802.11b wireless LAN standard.

[0108] Arrays can be further extended into higher dimensions. In particular, Golay doubling constructions have been found to allow the formation of two (or more) dimensional arrays by combining subarrays of dimensions other than the original array. For example, the arrays

[0011] and [1 −1] can be used to form two 2*2 complementary arrays: 1 1 1 -1 and 1 1 -1 1 is a complementary 2*2 array. For example, it is possible to shift the array by any directional vector to form complementary 2D patterns. 1 1 0 0 1 -1 and 1 1 0 0 -1 1 Or alternatively, 1 1 0 0 0 0 0 1 -1 and 1 1 0 0 0 0 0 -1 1

[0109] Similarly, when A and B are complementary column vectors of length N, then [AB] and [A -B] are 2D complementary arrays of size N x 2. Similarly, by stacking row arrays A and B vertically, a complementary array of size 2 x N is obtained.

[0110] It turns out that such complementary arrays can be applied to beamforming by applying phases to the signals transmitted from different elements according to a low PAPR array. The maximum EIRP in any direction can be limited by a small factor over the spatial average. Thus, when 0 / 180 phases are applied according to a Golay complementary array, a peak-to-average EIRP factor of at most 2 is guaranteed.

[0111] In one application, a MIMO radar transmits from multiple transmit antenna elements and receives at multiple receive antenna elements. By measuring the response from all transmit antennas to all receive antennas, the process continues to reconstruct the spatial distribution of scatterers. While it would be possible to transmit from one antenna at a time to measure the response, a more efficient method (in terms of transmit power) would be to transmit from multiple antennas simultaneously. By applying a phase to each transmit element and transmitting multiple phase combinations, the responses from multiple elements can now be resolved by multiplying the set of responses by the inverse of a "beamforming matrix," which consists of multiple beamforming vectors.

[0112] If it is desired that all of the transmissions avoid all strong illumination in a spatial direction, then a family of beamforming signature vectors is needed that are preferably orthogonal to each other and all have the property of low PAPR.

[0113] By using a family of modified Hadamard matrices such as those shown above, it is possible to have a set of beamforming signature sequences that achieve this goal.

[0114] As an illustration, the graph in FIG. 6A shows the array power factor of a 32-element array with in-phase excitation, which can be compared to FIGS. 6B and 6C, which show the array power factors of 32-element arrays with random phase excitation and complementary sequence-based excitation, which have much lower peak values.

[0115] In some cases, this goal can be approximated under practical constraints. One such case is when the transmission line between the transceiver and the antenna element has additional phase shifts that need to be compensated for by a phase adjuster. In such cases, if limited resolution (e.g., 2, 4, or 8 phases) of phase shifts is allowed, the optimal set of phases can be quantized to the nearest feasible value. In such cases, the goal of having a low PAPR is achieved in a non-optimal manner, but still with substantially better performance than a randomly chosen signature sequence. For example, the PAPR under practical constraints can be designed not to exceed a factor of 3 (5 dB) or 4 (6 dB) rather than the theoretical limit of 2 (3 dB) for a complementary sequence.

[0116] In other cases, the constellation can be numerically optimized. For example, in the art of OFDM modulation, techniques are known for "PAPR reduction" by starting with modifying a small subset of phases to substantially reduce the PAPR. The procedure can be iterated several times until the PAPR goal is met.

[0117] Any family of sequences or individual sequences with low PAPR properties can be used for peak EIRP reduction, with complementary sequences being just one specific example. Other examples are CAZAC (constant amplitude zero auto-correlation) sequences, Zadoff-Chu sequences, etc.

[0118] The above properties can be used for one-dimensional, two-dimensional, or low-density array shapes, as long as the low-PAPR property of an array with non-zero values ​​at specified locations is satisfied. An example of particular interest is two parallel columns (or rows) of elements. Such a configuration is encountered in a "frame" MIMO array or a "pie-shaped" MIMO array. In such a case, if A and B are complementary arrays, tuning one column of elements according to array A and the other column of elements according to array B results in a two-dimensional complementary array with low-PAPR property.

[0119] In the case of a sparse array geometry, the excitation phase of the sparse array geometry is designed to achieve low-PAPR properties, and additional sets of excitation phases can be formed by applying additional 2D spatially stepped phase patterns to the excitation. Applying stepped phase patterns is equivalent to angularly shifting the low-PAPR beam shape, thus maintaining its low-PAPR properties. This allows for the generation of any number of low-PAPR patterns as needed for radar spatial image reconstruction.

[0120] Various combinations of transmit and receive arrays may be used, such as those shown in Figures 7A-7H, where Figure 7A shows a possible transmit linear antenna array, Figure 7B shows a possible square transmit antenna array, Figure 7C shows a possible two-row transmit antenna array, Figure 7D shows a possible staggered two-row antenna array, Figure 7E shows a possible L-shaped MIMO antenna array, Figure 7F shows a possible pie-shaped MIMO antenna array, Figure 7G shows a possible frame-shaped MIMO antenna array, and Figure 7H shows a possible split-frame MIMO antenna array.

[0121] 8A-8C, several array topologies include an L-shaped array 800A, a pie-shaped array 800B, and a frame array 800C. Such topologies may limit the achievable system tradeoffs (such as angular resolution, field of view, and signal-to-noise ratio).

[0122] Surprisingly, it was found that more efficient utilization of MIMO and beamforming arrays could be created by combining L-shaped and pie-shaped array topologies using antennas with different directivity values. Such novel array topologies could use a combination of omnidirectional (wide beam) and directional antennas across different edges of the array, allowing for enhanced array performance over selected sectors of an arena, for example.

[0123] 8D, by way of illustration, one possible implementation of a combined array is a first asymmetric array 800D. An asymmetric array can be created by connecting many transmit ports along a first section 810D of the pie-shaped array to a wide-beam antenna 812 and fewer transmit ports along a second section 820D of the pie-shaped array to a directional antenna 822.

[0124] 8G, note that the asymmetric array 800D provides most of the high SNR (signal-to-noise ratio) and high angular resolution on the vertical axis of the L-shaped array. This can produce a wide vertical field of view 842 and a narrow vertical field of view 848, while doubling the horizontal angular resolution over a narrow slice 844 within the vertical field of view. Thus, enhanced horizontal angular resolution is achieved in the central region 846 of the field of view.

[0125] The allocation of ports between directional and omnidirectional antennas can be derived from the overall system requirements (SNR and resolution over different angular sectors).

[0126] In a non-limiting example, one possible way to obtain a balanced response from the two asymmetric branches 810, 820 is approximately

number

number

[0127] Other possible embodiments of asymmetric arrays are shown in Figures 8E and 8F. With specific reference to Figure 8E, in a staggered branch array 800E, the directional antennas may be staggered across the vertical axis. Such an arrangement may extend the field of view of the staggered branch 820E (grating-lobe rejection).

[0128] With specific reference to FIG. 8F, a horizontally extended array 800F of highly directional antennas 820F may be extended across the horizontal axis for further enhancement of horizontal resolution.

[0129] Although only an asymmetric array of transmitters is shown here, it will be appreciated that further embodiments may include similarly modified arrays of receive antennas.

[0130] Furthermore, the gain and directivity of the receive and transmit antennas may be different and adjusted to optimize system performance. For example, in a first asymmetric array, a slightly-directional antenna may be implemented for the receive port, whose directivity is between that of a directional and omnidirectional transmit antenna, and tx,dn <D rx <D tx,d is.

[0131] Possible transmission schemes on this topology could be MIMO transmission, or variously using analog beamforming of the transmit antennas to combine the two transmit branches using time domain multiplexing (TDM), orthogonal coding (e.g., Hadamard encoding), frequency domain multiplexing (FDM, e.g., using different RF frequencies for each branch), or other methods as needed.

[0132] Referring to Figure 8H, multiple chips may be connected to an array as needed to provide a multiplexing cluster, such as those described above with respect to Figure 5. Two vertical arrays of transmitters may each undergo beamforming with FDM used to multiplex the transmitted signals.

[0133] In the context of MIMO radar antenna systems, the concept of a "virtual array" (VA) plays an important role. T Direction from the transmitting antenna n signal to the target, and location r R The signal returning to the receiving antenna at is exp(j*k0*( n *(r T +r R ))), i.e., exp(j*2*k0*(n*(r T +r R ) / 2)). For transmit and receive arrays with multiple antenna elements, the collection of all locations (r T +r R) / 2 (the midpoint between the transmit and receive elements) is represented as the VA of the combined transmit and receive antenna array. This idea is illustrated in FIG. 9A. A linear equispaced transmit antenna array is composed of antennas 901a-901b (represented by black squares), and a linear equispaced receive antenna array is composed of antennas 902a-902b (represented by white squares). The VA of the two linear arrays is a square array of locations, whose corners are 903a-903d. Element 903a of the virtual array is the midpoint along line 904a connecting transmit element 901a with receive element 902a. Similarly, element 903d of the virtual array is the midpoint along line 904d connecting transmit element 901b with receive element 902b, and so on. It is worth mentioning that having a virtual array placed on a uniformly spaced grid facilitates efficient FFT-based MIMO radar signal processing algorithms.

[0134] Referring to Figure 9B, the idea is easily extended to an array with multiple linear segment subarrays, specifically a "frame array" consisting of two parallel linear transmit subarrays 911a and 911b and two linear receive subarrays 912a and 912b. The square virtual array in this case is composed of four subarrays 913a through 913d (each surrounded by dashed lines for clarity). VA subarray 913a is composed of the combination of antennas belonging to transmit subarray 911a and receive subarray 912a. Similarly, VA subarray 913b is the combination of 911b and 912a, 913c is the combination of 911a and 912b, and 913d is the combination of 911b and 912b. Achieving the relative locations of the VA subarrays 913a-913d so as to avoid voids may require placing the transmit subarrays 911a-911b and receive subarrays 912a-912b in locations such that the outermost transmit antennas are very close to the outermost receive antennas. This proximity can be detrimental due to "blinding" of receive antenna elements by strong signals originating from adjacent, strongly coupled transmit antenna elements.

[0135] The problem of strong coupling due to the proximity of TX-RX elements is the topic of a set of solutions presented below. Referring to FIG. 10A, "frame array" antenna element segments 1011a-1011b and 1012a-1012b are shifted so that the distal TX and RX antenna elements are spaced as close as the distance between the other antennas in the array. The resulting VA subarrays 1013a-1013d are shifted accordingly. Subarrays 1013a-1013d are still on a uniform grid, but voids 1014 are created where there are no VA elements. Imperfect VAs cause sidelobes during spatial processing. If these voids are a small portion of the VA array, the sidelobes experienced are low.

[0136] It may be desirable to space the TX and RX antenna elements even further. Referring to FIG. 10B, the "frame array" antenna element segments 1011a-1011b and 1012a-1012b are shifted even further, thus further increasing the spacing of the TX and RX antenna elements at the ends. The square VA subarrays 1013a-1013d are now more contiguous, but the void 1014 is increased to 9 = 3 * 3 missing elements in the center of the VA. This causes a further increase in spatial sidelobes. In the context of low-PAPR beamforming, it is noteworthy that staggering the transmit subarrays also enables the use of 2D complementary arrays, as discussed above.

[0137] To further reduce sidelobes, weighting (windowing) applied during FFT-based processing reduces their impact, so that irregular shapes of the outer boundaries of the VA typically have little effect.

[0138] A solution to the problem of gaps in staggered frame VAs is presented herein. Referring to Figures 11A and 11B, the solution is formed by adding auxiliary transmit and receive subarrays 1121 and 1122. In the illustrative example, these include three transmit antenna elements and three receive antenna elements that contribute to a 3*3 virtual subarray 1123. This virtual subarray fills the gaps between subarrays 1113a-1113d, creating a continuous VA. Naturally, this solution can be scaled to different amounts of stagger to reach the limits of allowable TX-RX antenna spacing. Figures 11A and 11B show two possible arrangements of subarrays 1121 and 1122.

[0139] A further solution to the problem of proximity between TX and RX elements is presented as an example in Figures 12A and 12B. The solution relies on a hierarchical structure of arrays of arrays, such as a nested array or Kroneker. Figure 12A shows a "staggered frame" structure similar to that shown in Figure 10A, but with four antennas per linear segment. The "staggered frame" of Figure 12A acts as a mother array that guides the placement of element subarrays in an array-of-arrays fashion, such as that shown in Figure 12B.

[0140] Figure 12B shows a composite array in which each antenna location in the mother array of Figure 12A is replaced by an element subarray, in this case a linear array of four closely packed antennas. In this example, each TX element, such as 1201, is replaced by an element subarray of four horizontally displaced elements, such as 1221, and each RX element, such as 1202, is replaced by an element subarray of four vertically displaced elements, such as 1222. The resulting virtual array is composed of clusters of elements, each of which replaces an element in the original VA; for example, element 1203 of the original VA becomes cluster 1223, which itself contains 16 VA elements.

[0141] Note that in the above-described example of the original mother array of Figure 12A, the VAs have a void 1214 in their centers, and as a result, the VAs of the corresponding hierarchical composite array also have a void 1224 in their centers. It will be appreciated that by using a mother array without a void in its center, such as those illustrated in Figures 11A and 11B, the resulting composite array may have corresponding contiguous VAs without a void in its center.

[0142] It is further noted that the hierarchical structure nicely facilitates practical modular construction of large arrays. Illustratively, a printed circuit module with 24 antennas and corresponding ASICs can be constructed, and this module can then be used repeatedly to construct larger arrays. The interconnections between modules can follow, for example, the techniques described in U.S. Pat. No. 10,020,836 B2, U.S. Pat. No. 10,804,954 B2, and related applications, which are incorporated herein in their entirety.

[0143] The hierarchical construction technique also facilitates the construction of large arrays for use in short-range (so-called "near-field") MIMO radar imaging systems, such as security scanning systems. As an illustration, consider a repeating six-element one-dimensional "mother array." The repeating arrangement is a TTRTRR, with two transmitters preceding a single receiver, which itself is followed by a single transmitter preceding two receivers. Figure 13A shows such a one-dimensional array and the corresponding one-dimensional VA formed thereby. For each element (e.g., 1303) in the VA, a line indicates the TX (e.g., 1301) and RX (e.g., 1302) elements involved in the formation of the VA element. Once the one-dimensional array is established, the large array is replaced by "tiles" of elements composed of linear segments; for example, transit element 1301 is replaced with a tile of transmit element 1311, and receive element 1302 is replaced with a tile of receive element 1312. By ensuring that a single transmit tile and a single receive tile have perfect VA, the composite array is also guaranteed to have perfect VA.

[0144] It should be further noted that the principles illustrated throughout the above discussion using square arrays are applicable to rectangular arrays having different numbers of elements in different directions in the 2D plane. FIG. 13B illustrates this principle, with the elements having rectangular shapes rather than square shapes. Furthermore, the element subarrays of FIG. 13B illustrate that the number of transmit antennas in an array or subarray need not be the same as the number of receive antennas. This property has applications, for example, whenever it is desirable to reduce the time required to scan a transmitter or transmit beam pattern, or when the cost of the transmit elements is substantially higher than the cost of the receive elements.

[0145] Technical and scientific terms used herein should have the same meaning as commonly understood by those skilled in the art to which this disclosure pertains. Nevertheless, it is expected that many related systems and methods will be developed during the life of the patent that matures from this application. Accordingly, the scope of terms such as computing unit, network, display, memory, server, and the like is intended to deductively include all such new technologies.

[0146] As used herein, the term "about" refers to at least ±10%.

[0147] The terms "comprises," "comprising," "includes," "including," "having," and their cognates mean "including but not limited to" and indicate the inclusion of the listed elements but not the exclusion of other elements as a whole. Such terms encompass the terms "consisting of" and "consisting essentially of."

[0148] The phrase "consisting essentially of" means that the composition or method may include additional components and / or steps, but only if the additional components and / or steps do not materially alter the basic and novel characteristics of the claimed composition or method.

[0149] As used herein, the singular forms "a," "an," and "the" can include plural references unless the context clearly dictates otherwise. For example, the term "a compound" or "at least one compound" can include multiple compounds and mixtures thereof.

[0150] The word "exemplary" is used herein to mean "serving as an example, instance, or illustration." Any embodiment described as "exemplary" is not necessarily to be construed as preferred or advantageous over other embodiments, or to exclude the incorporation of features from other embodiments.

[0151] The word "optionally" is used herein to mean "provided in some embodiments and not provided in others." Any particular embodiment of the present disclosure may include multiple "optional" features, so long as such features do not conflict.

[0152] Whenever a range of numerical values ​​is given herein, it is intended to include any recited number (fractional or integer) within the given range. The phrases "ranging / range between" a first and a second designated number, and "from" a first designated number and "to" a second designated number, are used interchangeably herein and are intended to include the first and second designated numbers and all fractional and integer numbers therebetween. It should therefore be understood that expressions in range format are merely for convenience and brevity and should not be construed as an inflexible limitation on the scope of the present disclosure. Thus, expressions in range format should be considered to have specifically disclosed all possible subranges and individual numerical values ​​within that range. For example, a range such as 1 to 6 should be considered to have specifically disclosed subranges such as 1 to 3, 1 to 4, 1 to 5, 7 to 4, 7 to 6, 3 to 6, etc., as well as individual numbers and non-integer intermediate values ​​within that range, e.g., 1, 7, 3, 4, 5, and 6. This applies regardless of the broadness of the range.

[0153] It is understood that certain features of the present disclosure that are, for clarity, described in the context of separate embodiments, may also be provided in combination in a single embodiment. Conversely, various features of the present disclosure that are, for brevity, described in the context of a single embodiment, may also be provided separately, or in any suitable subcombination, or as suitable in any other described embodiment of the present disclosure. Certain features described in the context of various embodiments should not be considered essential features of those embodiments, unless the embodiment is invalid without those elements.

[0154] While this disclosure has been described in conjunction with specific embodiments thereof, it is evident that many alternatives, modifications, and variations will be apparent to those skilled in the art. Accordingly, it is intended to embrace all such alternatives, modifications, and variations that fall within the spirit and broad scope of the appended claims.

[0155] All publications, patents, and patent applications mentioned herein are herein incorporated by reference in their entirety to the same extent as if each individual publication, patent, or patent application was specifically and individually indicated to be incorporated by reference herein. Furthermore, citation or identification of any reference in this application shall not be construed as an admission that such reference is available as prior art to the present disclosure. To the extent section headings are used, they should not be construed as necessarily limiting.

[0156] The scope of the disclosed subject matter is defined by the appended claims and includes both combinations and subcombinations of the various features described above, as well as variations and modifications thereof, which will occur to those skilled in the art upon reading the foregoing description.

Claims

1. 1. A system for simulating quadrature phase shift keying (QPSK) beamforming in signals transmitted from an antenna array, comprising: a transmitter configured and operable to generate a vibration signal; an antenna array; a bi-phase adjuster associated with each transmit antenna, the bi-phase adjuster configured and operable to selectively apply a 180 degree phase shift to the transmitted signal; a controller configured to send commands to the two-phase regulator; at least one receive antenna; a post processor operable to manipulate the received signals, the post processor comprising: a memory operable to store received signals; and a processing unit operable to apply a 90 degree phase shift to selected received signals stored in said memory and further operable to sum the received signals stored in said memory; Equipped with The controller: determining a complex steering vector required for each antenna of the array, the complex steering vector including a real component selected from 0 degrees and 180 degrees and an imaginary component selected from 90 degrees and 270 degrees; instructing the bi-phase adjuster to apply a 180 degree phase shift to the transmitted signal for each antenna having a steering vector associated with a real component of 180 degrees during a first time interval; commanding the bi-phase adjuster to apply a 180 degree phase shift to the transmitted signal for each antenna having a steering vector associated with an imaginary component of 180 degrees during a second time interval; and The post processor: applying a 90 degree phase shift to the signal received during the second time interval; summing the signal received during the first time interval with a 90 degree phase shifted signal received during the second time interval; and system.

2. further comprising a gain control unit associated with each transmit antenna, the controller being further configured to send instructions to the gain control unit; For each transmit antenna, the controller: selecting a required amplitude AR for the real component of the associated steering vector; instructing the associated gain control unit to apply a first gain GR to the transmitted signal during the first time interval; selecting a required amplitude A1 for the imaginary component of the associated steering vector; instructing the associated gain control unit to apply a second gain Gl to the transmitted signal during the second time interval, the second gain Gl being equal to the product of GR and a ratio of A1 to AR; The system of claim 1 , operable to:

3. The system of claim 1 , wherein each antenna has a dedicated phase adjuster.

4. 10. The system of claim 1, wherein each antenna has a dedicated bi-phase adjuster.

5. The system of claim 1 , wherein the transmitter comprises a plurality of transmit antennas grouped into clusters.

6. The system of claim 5 , wherein the phase adjuster is connected to multiple antennas.

7. The system of claim 6 , wherein the phase adjuster comprises a two-phase adjuster.

8. 2. The system of claim 1, wherein the transmitter comprises a plurality of transmit antennas grouped into clusters and a common phase-locked loop chirp oscillator (PLL), each cluster of transmit antennas connected to the common phase-locked loop chirp oscillator (PLL) through a dedicated cluster modulator.

9. 10. The system of claim 1, wherein the antenna array is selected from at least one of the group consisting of a linear antenna array, a square transmit antenna array, a two-row array, a staggered two-row array, an L-shaped array, a pie-shaped array, and a frame array.

10. The system of claim 1 , wherein the antenna array comprises an asymmetric array.

11. 11. The system of claim 10, wherein the asymmetric array comprises an array of wide beam antennas disposed along a first section and an array of directional antennas disposed along a second section.

12. The system of claim 10 , wherein the asymmetric array comprises a horizontally extending array of highly directional antennas.

13. The system of claim 10 , wherein the asymmetric array comprises a staggered array of highly directional antennas.

14. 1. A method for simulating quadrature phase shift keying (QPSK) beamforming in signals transmitted from an antenna array, wherein each antenna of the array is connected to a common transmitter through a binary phase adjuster, the method comprising: determining, for each transmit antenna of the array, a required complex QPSK steering vector having a real component selected from 0 degrees and 180 degrees and an imaginary component selected from 90 degrees and 270 degrees; a transmitter generating a vibration signal; during a first time interval, for each transmit antenna having a steering vector associated with a real component of 180 degrees, the bi-phase adjuster applying a 180 degree phase shift to the transmitted signal; during a second time interval, for each transmit antenna having a steering vector associated with an imaginary component of 180 degrees, the bi-phase adjuster applying a 180 degree phase shift to the transmitted signal; a post processor applying a 90 degree phase shift to signals received during said second time interval; the post processor summing the signal received during the first time interval with a 90 degree phase shifted signal received during the second time interval; A method comprising:

15. The method of claim 14 , wherein the step of the transmitter generating the vibration signal comprises the transmitter sweeping through a range of frequencies during each time interval.

16. For each transmitting antenna, selecting a desired amplitude AR for the real component of the associated steering vector; applying a first gain GR to the transmitted signal during the first time interval; selecting a desired amplitude A1 for the imaginary component of the associated steering vector; applying a second gain Gl to the transmitted signal during the second time interval, the second gain Gl being equal to the product of GR and the ratio of A1 to AR; 15. The method of claim 14, further comprising:

17. 1. A method for simulating quadrature phase shift keying (QPSK) beamforming in an antenna array, wherein each antenna of the array is selectively connected to a common transmitter oscillator via an independently controllable connection switch, the method comprising: determining, for each transmit antenna of the array, a required complex QPSK steering vector having a real component selected from 0 degrees and 180 degrees and an imaginary component selected from 90 degrees and 270 degrees; a transmitter generating a vibration signal; connecting only transmit antennas having steering vectors associated with a real component of 0 degrees to the common transmitter oscillator during a first time interval, the common transmitter oscillator generating a first time interval transmit signal; connecting only transmit antennas having steering vectors associated with a real component of 180 degrees to the common transmitter oscillator during a second time interval, the common transmitter oscillator generating a second time interval transmit signal; connecting only transmit antennas having steering vectors associated with an imaginary component of 90 degrees to the common transmitter oscillator during a third time interval, the common transmitter oscillator generating a third time interval transmit signal; connecting only transmit antennas having steering vectors associated with an imaginary component of 270 degrees to the common transmitter oscillator during a fourth time interval, the common transmitter oscillator generating a fourth time interval transmit signal; a post processor applying a 0 degree phase shift to signals received during said first time interval; a post processor applying a 180 degree phase shift to signals received during said second time interval; a post processor applying a 90 degree phase shift to signals received during said third time interval; a post processor applying a 270 degree phase shift to signals received during said fourth time interval; the post processor summing the signal received during the first time interval, a 180 degree phase shifted signal received during the second time interval, a 90 degree phase shifted signal received during the third time interval, and a 270 degree phase shifted signal received during the fourth time interval; A method comprising:

Images