DC / DC converter and DC power supply device

Abstract

To provide a DCDC converter using a switched capacitor that can be configured with low inductance even when applied to a voltage source.SOLUTION: A DCDC converter 1 includes a reactor 4, an inverter circuit 20 that is connected to the rear stage of the reactor 4, and including capacitors C1 and C2, and at least one bit inverter 20a, 20b with a bridge circuit, a smoothing capacitor Ca that smooths the output of the inverter circuit 20, a short-circuit switch S4 that switches between a charging mode and a discharging mode, and a control unit 5. The control unit 5 feedback-controls a charging period or a discharging period in each cycle according to the amount of error between voltages Vc1 and Vc2 of the capacitors C1 and C2 of the bit inverters 20a and 20b and target values V1 and V2.SELECTED DRAWING: Figure 1

Description

[Technical Field]

[0001] The present invention relates to a DC-DC converter and a DC power supply device using the same. [Background technology]

[0002] Japanese Patent Publication No. 5028525 (Patent Document 1) proposes a DC-DC converter that performs DC-DC conversion (DCDC conversion) on an input voltage from a current source such as a solar cell. This DC-DC converter has an inverter circuit connected in series to a solar cell. The inverter circuit is configured by connecting the AC sides of first and second single-phase inverters in series. Each single-phase inverter is composed of a semiconductor switch element and a capacitor. A short-circuit switch, a rectifier diode, and a smoothing capacitor are connected downstream of the inverter circuit. The DC-DC converter controls the inverter circuit to superimpose the voltage of each capacitor of the first and second single-phase inverters on the voltage of the solar cell's DC power supply, thereby outputting the desired voltage to the smoothing capacitor. This DC-DC converter performs DC-DC conversion using the charging and discharging of the inverter circuit's capacitor, eliminating the need for a large-capacity reactor.

[0003] Japanese Patent Application Laid-Open No. 2011-114964 (Patent Document 2) describes that in a DCDC with a similar configuration to that of Patent Document 1, multiple control modes are switched depending on the voltage of the DC power supply, and the speed at which the voltage of the smoothing capacitor follows the desired output voltage is gradually adjusted. [Prior art documents] [Patent documents]

[0004] [Patent Document 1] Patent No. 5028525 [Patent Document 2] Japanese Patent Application Laid-Open No. 2011-114964 Summary of the Invention [Problem to be solved by the invention]

[0005] In DC-DC converters, current is generally smoothed using a reactor. Reactors are heavy. Therefore, when weight reduction is required, such as in electric aircraft, DC-DC converters with large-capacity reactors are difficult to adopt. The inventions in Patent Documents 1 and 2 above eliminate the need for a current-smoothing reactor, making it possible to miniaturize the device. These inventions use solar cells as the power source. Because solar cells are current sources, even if the voltage of the inverter's capacitor deviates slightly from the specified value due to an error, no large current flows within the circuit.

[0006] The inventors noticed that when the above-mentioned conventional DC-DC converter is applied to a voltage source, even a slight deviation in the inverter voltage from a predetermined value due to an error generates an electromotive force in the circuit, causing a non-negligible current to flow. As a countermeasure to this problem, a high-inductance reactor is inserted to suppress the current, but this ends up hindering weight reduction.

[0007] Therefore, an object of the present invention is to provide a DC-DC converter using a switched capacitor that can be configured with low inductance even when applied to a voltage source. [Means for solving the problem]

[0008] A DC-DC converter according to an embodiment of the present invention includes: a reactor connected in series with a DC voltage source; an inverter circuit including at least one bit inverter connected downstream of the reactor, the bit inverter having a capacitor and a bridge circuit for switching the connection direction of the capacitor; a smoothing capacitor connected downstream of the inverter circuit via a switchable conductive / non-conductive element for smoothing an output from the inverter circuit; a short-circuit switch for switching between a charge mode in which a voltage of the capacitor of the bit inverter is set to a target value using power from the voltage source and a discharge mode in which the voltage of the voltage source and the voltage of the capacitor are superimposed and output to the smoothing capacitor; and a control unit for periodically repeating operation in the charge mode and the discharge mode by controlling the bit inverter and the short-circuit switch. The control unit feedback-controls a charging period of the capacitor in the charge mode or a discharging period of the capacitor in the discharge mode in each period according to an error amount between the voltage of the capacitor of the bit inverter and its target value. [Effects of the Invention]

[0009] According to the present invention, it is possible to provide a DC-DC converter using a switched capacitor that can be configured with low inductance even when applied to a voltage source. [Brief explanation of the drawings]

[0010] [Figure 1] FIG. 1 is a diagram showing an example of the configuration of a DC-DC converter according to an embodiment of the present invention. [Figure 2] 2 is a diagram for explaining an example of operation of the DC-DC converter 1 shown in FIG. 1; [Figure 3] A table showing an example of setting target values ​​for the DC-DC converter. [Figure 4] FIG. 10 is a diagram showing an example of feedback control by a control unit. [Figure 5] FIG. 10 is a diagram showing another example of feedback control by the control unit. [Figure 6] FIG. 10 is a diagram showing another example of feedback control by the control unit. [Figure 7] FIG. 10 is a diagram showing another example of feedback control by the control unit. [Figure 8] FIG. 10 is a diagram showing another example of feedback control by the control unit. [Figure 9] FIG. 5 is a diagram showing a processing example in which feedback control of a target value is added to the example in FIG. 4; [Figure 10] FIG. 10 is a diagram showing an example of voltage and current waveforms when the feedback control of FIG. 9 is performed. [Figure 11] A diagram to explain the difference in current between the start and end of the charging period [Figure 12] Graph showing the simulation results of the current and voltage waveforms of a DC-DC converter [Figure 13] Example of excessive current during initial charging of a DC-DC converter [Figure 14] FIG. 10 is a diagram showing an example of a limiting pulse generated by a limiting circuit. [Figure 15] FIG. 1 is a diagram showing a configuration example of a DC power supply device according to an embodiment of the present invention; [Figure 16] Graph showing an example of the relationship between the generator's angular frequency and output voltage [Figure 17] Graph showing example settings for boost rates M_DCDC and M_ACDC in response to fluctuations in generator output voltage [Figure 18] Flowchart showing an example of operation of the DC-DC converter boost rate setting circuit [Figure 19] FIG. 10 is a diagram illustrating another example of the operation of the AC / DC converter boost rate setting circuit. [Figure 20] FIG. 10 is a diagram showing another example of the configuration of a DC power supply device according to an embodiment of the present invention. DETAILED DESCRIPTION OF THE INVENTION

[0011] A DC-DC converter according to an embodiment of the present invention includes: a reactor connected in series with a DC voltage source; an inverter circuit including at least one bit inverter connected downstream of the reactor, the bit inverter having a capacitor and a bridge circuit for switching the connection direction of the capacitor; a smoothing capacitor connected downstream of the inverter circuit via a switchable conductive / non-conductive element for smoothing an output from the inverter circuit; a short-circuit switch for switching between a charge mode in which a voltage of the capacitor of the bit inverter is set to a target value using power from the voltage source and a discharge mode in which the voltage of the voltage source and the voltage of the capacitor are superimposed and output to the smoothing capacitor; and a control unit for periodically repeating operation in the charge mode and the discharge mode by controlling the bit inverter and the short-circuit switch. The control unit feedback-controls a charging period of the capacitor in the charge mode or a discharging period of the capacitor in the discharge mode in each period according to an error amount between the voltage of the capacitor of the bit inverter and its target value.

[0012] In the DC-DC converter with the above configuration, in charge mode, the bridge circuit of the bit inverter switches the connection direction of the capacitor relative to the voltage source, thereby charging the capacitor of the bit inverter to a target voltage. In discharge mode, the short-circuit switch is switched, and the voltage charged in the bit inverter capacitor and the voltage of the voltage source are superimposed and output to the smoothing capacitor. The control unit repeats the charge mode and the discharge mode at a predetermined cycle. As a result, the voltage of the voltage source is converted to a desired voltage and output. The control unit controls the charging period of the bit inverter to set the capacitor voltage to a target value. In this case, the control unit feedback-controls the charging period of the capacitor in the charge mode or the discharging period of the capacitor in the discharge mode in each cycle, depending on the error between the voltage of the capacitor of the bit inverter and its target value. This reduces the error between the capacitor voltage and the target value. This reduces the electromotive force caused by the deviation in the capacitor voltage, thereby suppressing the occurrence of overcurrent. As a result, the inductance of the reactor used to prevent overcurrent can be reduced. That is, even when applied to a voltage source, a DC-DC converter using a switched capacitor that can be configured with low inductance can be realized.

[0013] A voltage source is a power source that supplies a substantially constant voltage even when the magnitude of the load fluctuates. The current of a voltage source varies depending on the load connected to it. On the other hand, a current source is a power source that supplies a substantially constant current even when the magnitude of the load fluctuates. The voltage of a current source varies depending on the load connected to it.

[0014] When the voltage of the capacitor is lower than the target value, the control unit may perform feedback control to extend the charging period or shorten the discharging period, thereby enabling efficient execution of feedback control.

[0015] The inverter circuit may include a plurality of the bit inverters. In this case, the target values ​​of the capacitor voltages of the plurality of bit inverters may be different from each other. By combining the different target values ​​of the voltages of the plurality of bit inverters, it is possible to set various ratios of the output voltage to the voltage input from the voltage source.

[0016] A DC-DC converter according to an embodiment of the present invention includes a reactor connected in series with a DC voltage source, an inverter circuit including at least one bit inverter connected downstream of the reactor, the bit inverter including a capacitor and a bridge circuit for switching the direction of the capacitor connection, a smoothing capacitor connected downstream of the inverter circuit via a switchable conductive / non-conductive element for smoothing the output from the bit inverter, a short-circuit switch for switching between a charge mode in which the voltage of the capacitor of the bit inverter is set to a target value using power from the voltage source and a discharge mode in which the voltage of the voltage source and the voltage of the capacitor are superimposed and output to the smoothing capacitor, a control unit for periodically repeating operation in the charge mode and the discharge mode by controlling the bit inverter and the short-circuit switch, and a limiting circuit for limiting the conduction time of the short-circuit switch during a predetermined limiting period. The limiting circuit gradually shortens the limiting time during the limiting period to release the limit.

[0017] The DC-DC converter with the above configuration includes a limiting circuit. The control circuit limits the period during which the current flowing through the reactor increases, thereby mitigating the current increase. This prevents excessive current from flowing through the reactor. As a result, the inductance of the reactor used to prevent overcurrent can be reduced. In other words, a DC-DC converter using a switched capacitor can be realized, which can be configured with low inductance even when applied to a voltage source.

[0018] The limiting circuit may limit the conduction time by allowing the short-circuit switch to conduct for a time shorter than a pulse width controlling a charge period in the charge mode and a discharge period in the discharge mode, thereby providing a limited period in an operation in which the charge mode and the discharge mode are cyclically repeated, thereby mitigating an increase in current during the operation.

[0019] The limit period may be set to at least a part of a transient period until the DC-DC converter reaches a steady state, thereby allowing a period in which a sudden increase in current is expected to be set as the limit period.

[0020] The transient period may include at least one of a period for initially charging the capacitor and a transition period when the target value of the capacitor is changed. During the initial charging period of the capacitor and the transition period when the target value is changed (for example, when the boost ratio is changed), the current increases rapidly. By setting these periods as the limitation periods, the increase in current can be efficiently mitigated.

[0021] The control unit may perform feedback control of the target value of the capacitor so as to reduce a difference in current of the voltage source between a start point and an end point of the longest charging period in the charging mode or a longest discharging period in the discharging mode.

[0022] The target value of the capacitor of a bit inverter is often set to a value under ideal conditions. The loop circuit including the voltage source, reactor, and capacitor of the bit inverter deviates from the ideal conditions due to wiring resistance and other subtle factors. This causes a ripple component to occur in the current flowing from the voltage source to the reactor. In the above configuration, the control unit detects the difference in current of the voltage source at the start and end of the longest charging or discharging period in the charging mode and discharging mode cycles, and can feedback control the target value of the capacitor of the bit inverter to reduce this difference. This reduces the error in the average voltage across the reactor caused by subtle factors. In other words, the ripple component of the current flowing through the reactor can be reduced. As a result, the inductance of the reactor can be lowered.

[0023] The control unit may perform feedback control, for example, to increase the target value when the current of the voltage source is greater at the end of a charging period or a discharging period than at the start, and to decrease the target value when the current of the voltage source is smaller at the end than at the start. This allows for efficient reduction of ripple components. The control unit may also control the bit inverter and the short-circuit switch so that the current of the voltage source is the same at the start and end of the longest charging period in the charging mode or the longest discharging period in the discharging mode. Here, the state in which the current of the voltage source is the same at the start and end includes cases where the currents are exactly the same, as well as cases where there is a negligible error from the perspective of reducing ripple components.

[0024] As an example, the voltage source may be configured with a generator and an AC / DC converter that converts the output voltage of the generator into DC and outputs it.

[0025] The inventors have discovered that by providing a DC-DC converter using a switched capacitor in the downstream of a generator whose output voltage fluctuates, it is possible to configure a DC power supply device with low inductance that can output a DC voltage with little fluctuation. Such a DC power supply device is also included in the embodiments of the present invention.

[0026] A DC power supply device according to an embodiment of the present invention includes an AC / DC converter that converts the output voltage of a generator into DC, and a DC / DC converter connected in a stage subsequent to the AC / DC converter. a reactor connected in series to the AC / DC converter; an inverter circuit including at least one bit inverter connected to a subsequent stage of the reactor, the bit inverter having a capacitor and a bridge circuit that switches the direction of connection of the capacitor; a smoothing capacitor connected to a downstream side of the inverter circuit via an element that switches between conductive and non-conductive states, and smoothing an output from the inverter circuit; a short-circuit switch for switching between a charge mode in which the voltage of the capacitor of the bit inverter is set to a target value using the power of the AC / DC converter and a discharge mode in which the output voltage of the AC / DC converter and the voltage of the capacitor are superimposed and output to the smoothing capacitor; and a control unit that controls the bit inverter and the short-circuit switch to periodically repeat operations in the charge mode and the discharge mode. The control unit changes the target value of the charging mode in accordance with the angular frequency or the output voltage of the generator, thereby changing the boost rate of the DC-DC converter.

[0027] In the DC power supply device with the above configuration, a DC-DC converter using a switched capacitor is connected via a reactor to the downstream of the AC-DC converter that converts the generator's output voltage to DC. The DC-DC converter's control unit changes the boost ratio of the DC-DC converter by changing the target value of the capacitor voltage of the bit inverter and the operating mode of the bit inverter in accordance with the generator's angular frequency or output voltage. This allows the DC-DC converter to boost the voltage in response to fluctuations in the generator's output voltage, even if the AC-DC converter's boost ratio is low. This enables the output of a DC voltage with little fluctuation. Furthermore, because the boost configuration uses a DC-DC converter using a switched capacitor, the reactor can be smaller than in a configuration using, for example, a boost chopper circuit. As a result, a DC power supply device that can output a DC voltage with little fluctuation can be configured with low inductance.

[0028] The control unit may change the target value of the voltage of the capacitor of the bit inverter so that the boost ratio of the DC-DC converter increases when the output voltage or angular frequency of the generator decreases. The control unit can change the target value of the voltage of the capacitor of the bit inverter in the charging mode in accordance with the output voltage or angular frequency of the generator, and change the operating mode of the bit inverter to correspond to this target value, thereby switching the boost ratio in accordance with fluctuations in the output voltage of the generator. The operating mode of the bit inverter is controlled, for example, by the pattern of a control signal that controls the switches of the bridge circuit.

[0029] The generator is, for example, a generator whose output voltage varies according to angular frequency. The generator whose output voltage varies according to angular frequency may be, for example, a permanent magnet synchronous generator. A permanent magnet synchronous generator generates field magnetic flux using permanent magnets. Therefore, the permanent magnet synchronous generator is lightweight and highly efficient. The magnitude of the field magnetic flux is constant in a permanent magnet synchronous generator. Therefore, in a permanent magnet synchronous generator, the electromotive force induced in the generator winding is proportional to the angular frequency determined by the rotation speed. The output voltage of a permanent magnet synchronous generator varies according to the angular frequency. By connecting the above-mentioned DC power supply device to such a permanent magnet synchronous generator, the generator's output voltage can be converted into DC power with a nearly constant voltage. The DC power supply device can be configured with low inductance. Therefore, by combining a lightweight and highly efficient permanent magnet synchronous generator with a DC power supply device, a lightweight and highly efficient DC power generation system can be constructed as a whole.

[0030] The angular frequency of a generator is a value that represents the rotational speed of the generator. The angular frequency of a generator is not limited to angular frequency in the strict sense. For example, the rotational frequency, rotation period, and other values ​​that represent the rotational speed of a generator are also considered to be the angular frequency of a generator.

[0031] The DC power supply device may further include an AC / DC converter control unit that controls the boost rate of the AC / DC converter so that the output voltage of the DC / DC converter approaches a constant value. This controls the boost rate of the AC / DC converter in addition to the boost rate of the DC / DC converter. This allows the DC output voltage to be kept constant by more precisely tracking fluctuations in the generator's output voltage. Furthermore, controlling the boost rates of both the DC / DC converter using a switched capacitor and the A / C converter based on the generator's output voltage or angular frequency makes it easier to configure a DC power supply device with low inductance that can output a DC voltage with minimal fluctuations.

[0032] The AC-DC converter control unit may control the step-up rate of the AC-DC converter in accordance with the step-up rate of the DC-DC converter or the output voltage of the DC-DC converter, thereby efficiently keeping the output voltage of the DC-DC converter close to a constant value.

[0033] The control range of the step-up rate of the AC-DC converter by the AC-DC converter control unit may be set to be smaller than the control range of the step-up rate of the DC-DC converter by the control unit. This prevents the reactor of the AC-DC converter from becoming larger, making it easier to reduce the weight.

[0034] The control unit may feedback-control the charging period of the capacitor in the charging mode or the discharging period of the capacitor in the discharging mode in each cycle according to the error between the capacitor voltage of the bit inverter and its target value. This feedback control reduces the error between the capacitor voltage and the target value. This reduces the electromotive force caused by the deviation in the capacitor voltage, thereby suppressing the occurrence of overcurrent. As a result, the inductance of the reactor can be further reduced.

[0035] The DC-DC converter may include a limiting circuit that limits the conduction time of the short-circuit switch during a predetermined limiting period. The limiting circuit may gradually shorten the limiting time during the limiting period and release the limit. The control circuit can limit the period during which the current flowing through the reactor increases, and mitigate the increase in current. This can prevent excessive current from flowing through the reactor. As a result, the inductance of the reactor can be reduced.

[0036] (Example of DC-DC converter configuration) FIG. 1 is a diagram showing an example of the configuration of a DC-DC converter according to an embodiment of the present invention. The DC-DC converter 1 receives a voltage from a DC voltage source 3, converts it, and outputs a DC voltage. The DC-DC converter 1 converts the voltage using a switched capacitor gradation control operation. The DC-DC converter 1 includes a reactor 4, an inverter circuit 20, a smoothing capacitor Ca, a short-circuit switch S4, and a control unit 5. The reactor 4 is connected between the voltage source 3 and the inverter circuit 20. The smoothing capacitor Ca is connected to the subsequent stage of the inverter circuit 20, i.e., its output terminal. The short-circuit switch S4 is connected between the subsequent stage of the inverter circuit, i.e., its output terminal, and the negative terminal of the voltage source 3.

[0037] The inverter circuit 20 includes multiple bit inverters 20a and 20b. Each bit inverter 20a and 20b includes a bridge circuit and capacitors C1 and C2. The bridge circuit includes switches S1, S2, and S3 that switch the direction of connection of the capacitors C1 and C2 to the voltage source 3. That is, the switches S1, S2, and S3 of the bridge circuit switch the direction of the current flowing through the capacitors C1 and C2 depending on the voltage of the voltage source 3. The switches S1, S2, and S3 are semiconductor switch elements. In the example of FIG. 1, a bridge circuit including the switches is connected to both ends of the capacitors C1 and C2. The capacitors C1 and C2 function as a DC power supply.

[0038] The bridge circuit of the bit inverters 20a and 20b includes switches S1 to S3 and diodes D1 to D3 connected in series. Capacitors C1 and C2 are connected to terminals between the diodes of the switches. In FIG. 1, the bit inverter 20a is a full bridge and the bit inverter 20b is a half bridge. The bit inverter 20b may also be a full bridge. The diodes D1 to D3 may be replaced with semiconductor switch elements.

[0039] In the inverter circuit 20, the AC terminals of multiple bit inverters 20a, 20b are connected in series. That is, the inverter circuit 20 is configured by connecting the AC sides of multiple single-phase inverters in series. The inverter circuit 20 can output a voltage that is the sum of the voltages of the capacitors C1, C2 of the multiple bit inverters 20a, 20b. That is, the inverter circuit 20 can superimpose the sum of the outputs of the single-phase inverters on the output of the voltage source 3.

[0040] A smoothing capacitor Ca is connected to the subsequent stage of the inverter circuit 20, i.e., to the output terminal, via a diode D4. The diode D4 is an example of an element that switches between conductive and non-conductive states, and the diode D4 may be replaced with, for example, a semiconductor switch element. The smoothing capacitor Ca smoothes the output voltage of the inverter circuit 20. In the example of FIG. 1, the cathode side of the diode D4 is connected to the positive electrode of the smoothing capacitor Ca, and the anode is connected to the output terminal of the inverter circuit 20. The negative electrode of the smoothing capacitor Ca is connected to the negative terminal of the voltage source 3. A voltage obtained by superimposing the voltages of the voltage source 3 and the capacitors C1 and C2 is applied to the smoothing capacitor Ca.

[0041] The short-circuit switch S4 switches between conduction and non-conduction between the output side of the bit inverter 20b at the most recent stage of the inverter circuit 20 and the negative side of the voltage source. In the example of FIG. 1, the short-circuit switch S4 is a switch that switches between conduction and non-conduction between one terminal of the capacitor C2 of the bit inverter 20b closest to the load and the negative terminal of the voltage source 3. The short-circuit switch S4 may be, for example, a semiconductor switch element or a mechanical switch. The short-circuit switch S4 switches between the charge mode and the discharge mode of the DC-DC converter 1. When the short-circuit switch S4 is on (conductive), the DC-DC converter 1 is in charge mode, and when it is off (non-conductive), the DC-DC converter 1 is in discharge mode. That is, in the charge mode, the short-circuit switch establishes a connection state in which the capacitor of the bit inverter is charged with power from the voltage source. In the discharge mode, the short-circuit switch establishes a connection state in which the voltages of the capacitor of the bit inverter and the voltage source are superimposed and output.

[0042] The semiconductor switch elements that constitute the switches S1 to S3 and the short-circuit switch S4 may be, for example, an insulated gate bipolar transistor (IGBT) or a metal oxide semiconductor field effect transistor (MOSFET).

[0043] The control unit 5 controls the bit inverters 20a, 20b and the short-circuit switch S4. For example, the control unit 5 controls the on / off of the switches S1 to S3 of the bit inverters 20a, 20b and the short-circuit switch S4. Under the control of the control unit 5, the DC-DC converter 1 periodically repeats the operation of the charge mode and the discharge mode. In the charge mode, the short-circuit switch S4 is turned on, and the switches S1 to S3 are controlled so that the voltages of the capacitors C1 and C2 of the bit inverters 20a, 20b reach their target values ​​using the power of the voltage source 3. In the discharge mode, the short-circuit switch is turned off, and the switches S1 to S3 are controlled so that the voltage of the voltage source 3 and the voltage of the capacitors C1 and C2 are superimposed and output to the smoothing capacitor Ca.

[0044] The control unit 5 feedback-controls the charging period or discharging period of the capacitors C1 and C2 in each cycle according to the error between the voltages Vc1 and Vc2 of the capacitors C1 and C2 of the bit inverters 20a and 20b, respectively, and their target values. The control unit 5 detects the voltages Vc1 and Vc2 of the capacitors C1 and C2. The control unit 5 adjusts the length of the charging period or discharging period of the capacitors C1 and C2 in each cycle based on the error between the detected voltages Vc1 and Vc2 and the target values. That is, the length of the charging period or discharging period is feedback-controlled so that the voltages of the capacitors C1 and C2 approach the target values ​​in each cycle. In this way, the control unit 5 feedback-controls the charging period or discharging period of the capacitors C1 and C2 in each cycle so as to reduce the error between the voltages Vc1 and Vc2 and the target values. In this case, the length of one cycle including the charging mode and the discharging mode remains unchanged. For example, the control unit 5 can increase or decrease the length of the charging period or discharging period of each cycle relative to a predetermined length in accordance with the amount of error between the detected voltages Vc1, Vc2 and the target values.

[0045] The charge period or discharge period of each cycle controlled by the control unit 5 is preset according to the target voltage value of the capacitors C1 and C2. However, in actual operation, the voltages Vc1 and Vc2 of the capacitors C1 and C2 may deviate slightly from the target value. The control unit 5 feeds back the amount of this slight deviation to the charge period or discharge period. This suppresses the electromotive force generated by the deviation of the voltages Vc1 and Vc2 of the capacitors C1 and C2 from their target values. This also suppresses overcurrent caused by the electromotive force. As a result, the inductance L0 of the reactor 4, which suppresses the occurrence of overcurrent, can be reduced. For example, the inductance L0 of the reactor 4 can be set to the same level as that of a wiring (L0≦5 μH). In this case, the reactor 4 may be realized by the wiring connecting the voltage source 3 and the inverter circuit 20.

[0046] (Example of operation) FIG. 2 is a diagram illustrating an example of the operation of the DC-DC converter 1 shown in FIG. 1. In the example of FIG. 2, FIGS. 2(a) and 2(b) show the states in the charging mode, and FIG. 2(c) shows the state in the discharging mode. FIG. 2 shows the state changes in one cycle. The states change in one cycle in the order of (a), (b), and (b). In FIG. 2(a), switches S1 and S2 are off, and switches S3 and S4 are on. In FIG. 2(a), current flows in the direction of the arrow, and capacitor C1 is charged to V1, which is the target value. In FIG. 2(b), switches S1, S2, and S4 are on, and switch S3 is off. Current flows in the direction of the arrow in Figure 2(b), and capacitor C2 is charged to the target value V2. In this example, V2 = V0 + V1. In Figure 2(c), switches S1, S2, and S3 are on, and switch S4 is off. Current flows in the direction of the arrow in Figure 2(c), and inverter circuit 20 outputs voltage VR = V0 + V1 + V2, which is the voltage V0 of voltage source 3 superimposed on the voltages of capacitors C1 and C2. For example, by setting the target values ​​to V1 = V0 and V2 = 2 V0, VR can be set to 4 V0. In this case, the DC-DC converter 1 functions as a quadruple boost circuit.

[0047] By changing the operation of one cycle and the setting of the target value, the DC-DC converter 1 can function as a circuit with various boost rates. Figure 3 is a table showing setting examples of target values ​​V1 and V2 when the DC-DC converter 1 is operated with boost rates of 1.3, 1.5, 2, 3, and 4.

[0048] FIG. 4 is a diagram illustrating an example of feedback control by the control unit 5. FIG. 4 shows an example in which the DC-DC converter 1 operates in quadruple boost mode. The upper part of FIG. 4 shows a diagram illustrating an example of feedback control processing. For example, the upper part of FIG. 4 shows an example of feedback processing when charging capacitor C1 to a target value V1=V0 and an example of feedback processing when charging capacitor C2 to a target value V2=2V0.

[0049] In the example shown in the upper part of FIG. 4, a value obtained by amplifying the difference (error) between the target value V1 and the detected voltage Vc1 of the capacitor C1 is added to the Vc1 reference voltage (Vh / 2), and the resulting value VR1 is provided to the PWM generator as a feedback control value. Also, a value obtained by amplifying the difference (error) between the target value V2 and the detected voltage Vc2 of the capacitor C2 is added to the Vc2 reference voltage ((3 / 4)·Vh), and the resulting value VR2 is provided to the PWM generator as a feedback control value. The PWM generator compares the feedback control values VR1 and VR2 with the sawtooth carrier signal Cr and generates pulse signals G1 and G2 with widths that reflect the feedback. The period of the carrier signal Cr is the same as the period of the repetition of the charge mode and the discharge mode of the DC-DC converter 1. G1 is a pulse signal with a duty ratio corresponding to the charging period of the capacitor C1. G2 is a pulse signal with a duty ratio corresponding to the charging period of the capacitor C2. Here, Vh corresponds to the amplitude of the carrier signal Cr. The Vc1 reference voltage (Vh / 2) and the Vc2 reference voltage ((3 / 4)·Vh) are each determined in advance according to the target values V1 and V2. The PWM generator outputs the pulse signal G1 when the carrier signal Cr is less than or equal to VR1 (Cr≦VR1), outputs the pulse signal G2 when VR1 < Cr≦VR2, and outputs the pulse signal G3 when VR2 < Cr≦Vh.

[0050] The switches S1 to S4 are turned on / off according to the signals G1 to G3. As shown in the middle of FIG. 4, the switches S1 and S2 are off when the pulse signal G1 is at the high level (when G1 is output), and are on when the pulse signal G1 is at the low level (when G1 is not output). The switch S3 is off when G2 is output and on when G2 is not output. The switch S4 is off when G3 is output and on when G3 is not output.

[0051] The output voltages of the bit inverters 20a and 20b due to the operations of the switches S1 to S4 based on the pulse signals G1 to G3 by this PWM control are shown in the lower part of FIG. 4. In the lower part of FIG. 4, the input voltage Vin from the voltage source 3, the output voltage Vbit1 of the bit inverter 20a, the output voltage Vbit2 of the bit inverter 20b, and the output voltage Va of the entire inverter circuit 20 are shown.

[0052] In the example shown in the lower part of FIG. 4, the length of the charging period Tcc1 of the capacitor C1 is adjusted by feedback control based on the voltage Vc1 of the capacitor C1 described above. The length of the charging period Tcc2 of the capacitor C2 is adjusted by feedback control based on the voltage Vc2 of the capacitor C2 described above. For example, if the detected voltage Vc1 is Vc1 < V1 (= V0), the charging period Tcc1 is extended, and if Vc1 > V1, the charging period Tcc1 is adjusted to be shortened. When the charging period Tcc1 is extended, the charging of the capacitor C2 further progresses.

[0053] As shown in FIG. 4, in order to realize feedback control, the control unit 5 can have a subtractor that calculates the difference between the detected voltages Vc1 and Vc2 and the target values, an amplifier that amplifies the difference, an adder that adds the amplified value to a reference value, and a PWM generator that generates a PWM signal using the control value output by the adder. At least a part of the functions of such a control unit 5 may be executed by digital processing using a microcomputer or the like.

[0054] FIG. 5 shows an example in which the DC-DC converter 1 operates in triple boost mode. In the example shown in FIG. 3, the target values ​​V1 and V2 are the same as those in the example shown in FIG. 4, but the on / off conditions of the switches S1 and S2 and the Vc2 and Vc1 reference voltages are different from those in the example shown in FIG. 4. In charge mode, the switches S1 and S2 are turned off and the switch S3 is turned on to charge the capacitor C1 to V1=V0. Then, the switches S1 and S2 are turned off and the switch S3 is turned on to charge the capacitor C2 to V2=2V0. In discharge mode, the switch S1 is turned off and the switches S2 and S3 are turned on to output a voltage Va=3V0, which is the superposition of the voltage V0 of the voltage source 3 and the voltage V2=2V0 of the capacitor C2. In discharge mode, the voltage of the capacitor C1 is not superimposed on the output voltage. As an example, the Vc2 reference value and the Vc1 reference voltage are set so that in one cycle, the charging period of capacitor C1, the charging period of capacitor C2, and the discharging period are approximately equal, i.e., the reference duty ratio of G1, G2, and G3 is 1:1:1.

[0055] Figure 6 shows an example in which the DC-DC converter 1 operates in double boost mode. In the example shown in Figure 6, the DC-DC converter 1 charges the capacitor C1 to a target value V1=V0 during the charging period, and outputs a voltage Va=2V0, which is the voltage V0 of the voltage source 3 superimposed on the voltage V1=V0 of the capacitor C1 during the discharging period. Capacitor C2 is not used. The charging period Tcc1 of the capacitor C1 is adjusted based on the error between the voltage Vc1 of the capacitor C1 and the target value V1.

[0056] FIG. 7 shows an example in which the DC-DC converter 1 operates in 1.5x boost mode. In the example shown in FIG. 7, the target values ​​V1 and V2 are both V0 / 2. In the charge mode, switches S1, S2, and S3 are turned off, and capacitors C1 and C2 are connected in series and charged to a voltage of V0 / 2. The discharge period in the discharge mode includes a discharge period in which switches S1 and S2 are turned on and switch S3 is turned off to output a voltage Va=(3 / 2)V0, which is the voltage V0 of the voltage source 3 superimposed with the voltage V0 / 2 of the capacitor C1; and a discharge period in which switch S1 is turned off and switches S2 and S3 are turned on to output a voltage Va=(3 / 2)V0, which is the voltage V0 of the voltage source 3 superimposed with the voltage V0 / 2 of the capacitor C2.

[0057] In the example of Figure 7, feedback control adjusts the length of the discharge period Tcc2 of capacitor C2 according to the error between the voltage Vc2 of capacitor C2 and the target value V2. In this example, the difference between the target value V2 and the voltage Vc2 of capacitor C2 is amplified, and the amplified value is added to the Vc1 reference voltage ((2 / 3)Vh). This value, VR2, is provided to the PWM generator as the feedback control value. For example, if Vc2 > 2V0, VR2 becomes smaller than the Vc2 reference voltage. As VR2 decreases, G2 decreases, and G3 increases accordingly. In other words, the discharge period Tcc2 of capacitor C2 becomes longer. Thus, if Vc2 > 2V0, the discharge period Tcc2 is extended, and discharge progresses. In contrast, if Vc2 < 2V0, the discharge period Tcc2 is shortened, and the discharge amount decreases.

[0058] FIG. 8 shows an example in which the DC-DC converter 1 operates in 1.33x boost mode. In the example shown in FIG. 8, in charge mode, capacitors C1 and C2 are charged to target values ​​V1 and V2. The target values ​​are V1 = V0 / 3 and V2 = (2 / 3)V0. In discharge mode, capacitor C1 is discharged to a voltage of target value V1 = V0 / 3, and then capacitor C2 is discharged to a voltage of target value V2 = (2 / 3)V0. In the example shown in FIG. 8, feedback control is used to adjust the length of the charging period Tcc1 of capacitor C1 in accordance with the error between the voltage Vc1 of capacitor C1 and the target value V1. Feedback control is also used to adjust the length of the discharging period Tcc3 of capacitor C2 in accordance with the error between the voltage Vc2 of capacitor C2 and the target value V2. Through these feedback controls, the lengths of the charging period Tcc1 and discharging period Tcc3 are adjusted so that the voltages of the capacitors C1 and C2 reach the target values ​​V1=V0 / 3 and V2=(2 / 3)V0, respectively.

[0059] (Example of a process that adjusts the target value based on the current of a voltage source) In the above example, the control unit 5 feeds back the voltages Vc1 and Vc2 of the capacitors C1 and C2 to adjust the length of the charging period or the discharging period so as to reduce the error between the voltages Vc1 and Vc2 and the target values ​​V1 and V2. The control unit 5 can further adjust the target values ​​V1 and V2 based on fluctuations in the current of the voltage source 3.

[0060] Fig. 9 is a diagram showing a processing example when feedback control of a target value is added to the quadruple boost mode shown in Fig. 4. In the example shown in Fig. 9, the processing shown by the dashed line is added to the processing in Fig. 4. In the example shown in Fig. 9, the difference i0 t1 -i0 t2 Here, the difference in the current flowing from the voltage source 3 to the reactor 4 at the start and end points of the longest charging period in each cycle is detected. For example, the difference in the current detected in the past cycle (i0 t1 -i0 t2 ) is used to determine the target value. (i0 t1 -i0t2 ) can use the value recorded by control unit 5 as the currently detected value. Note that current i0 may be the detected value of the current flowing between voltage source 3 and reactor 4, or may be the detected value of the current flowing between reactor 4 and inverter circuit 20.

[0061] The difference between the Vc1 target reference value (V0 in this example) and the above current (i0 t1 -i0 t2 The target value V1 of the voltage of capacitor C1 is obtained by subtracting the value Vdi obtained by PI integration of V0 - Vdi (V0 - Vdi). The target value V1 is obtained by doubling the target value V1 and the target value V2 of the voltage of capacitor C2. The PI integration can be calculated, for example, using the following formula: i0 t1 -i0 t2 =e Vdi=K1e+K2∫edt In the above formula, K1 and K2 are constants. The values ​​of K1 and K2 are selected so that the control system does not oscillate. The target values ​​V1 and V2 are used for feedback control during the charging period or the discharging period, as in the case of Fig. 4. The Vc1 target reference value can be set as, for example, a target value under ideal conditions.

[0062] In this way, the current difference (i0 t1 -i0 t2 ) is subtracted from the target reference value to set the target values ​​V1 and V2, and the charging period or discharging period is feedback controlled to reduce the error between the target values ​​V1 and V2 and the voltages Vc1 and Vc2 of the capacitors C1 and C2. t1 -i0 t2 As a result, the ripple component of the current in the reactor 4 can be reduced.

[0063] 10 is a diagram showing an example of waveforms of AC components of the output voltage of the bit inverters 20a and 20b and the reactor current (current of the voltage source) when the feedback control shown in FIG. 9 is performed. In the example shown in FIG. 10, the difference in reactor current (i0 t1 -i0t2 ) is fed back to the target value for the next cycle. Here, the charging period targeted for detecting the current difference is the longest charging period included in one cycle. In the example of FIG. 10, the longer period Tcc1 of the charging periods Tcc1 and Tcc2 in one cycle is targeted. The discharging period may be targeted instead of the charging period. In this case, the longest discharging period included in one cycle may be targeted.

[0064] Figure 11 shows the difference in current between the start and end of the charging period (i0 t1 -i0 t2 11 is a diagram illustrating the relationship between the voltage Vc1 of the capacitor C1 charged during the charging period and the voltage Vc1 of the capacitor C1 charged during the charging period. As shown in FIG. 11, if the voltage Vc1 of the capacitor C1 charged during the charging period is too low compared to the voltage V0 of the voltage source 3, the current at the end of the charging period will be larger than the current at the start of the charging period (i0 t1 <i0 t2 If the voltage Vc1 of the capacitor C1 charged during the charging period is too high compared to the voltage V0 of the voltage source 3, the current at the end of the charging period will be smaller than the current at the start (i0 t1 >i0 t2 ) When the voltage Vc1 of the capacitor C1 charged during the charging period is appropriate for the voltage V0 of the voltage source 3, the current at the start and end of the charging period will be equal (i0 t1 =i0 t2 ).

[0065] When the current at the end of the charging period is greater than the current at the start of the charging period (i0 t1 <i0 t2 ), and their difference (i0 t1 -i0 t ) the target value of the voltage of the capacitor C1 during the charging period is made larger than the target reference value by an amount corresponding to the current at the end of the charging period (i0 t1 >i0 t2 ), and their difference (i0 t1 -i0 tThe target value of the voltage of capacitor C1 during the charging period is made smaller than the target reference value by an amount corresponding to ( ). This adjusts the target value so that the difference in current between the start and end of the charging period is reduced.

[0066] In the above example, the difference in current during the charging period (i0 t1 -i0 t ) is converted into the difference in the average voltage between the voltage source 3 and the bit inverter 20a during the charging period (the difference in the average voltage across the reactor 4), and the converted value is calculated as the value Vdi to be subtracted from the target reference value in feedback control. t1 -i0 t ), the target reference value is changed by a value Vd according to the current difference (i0 t1 -i0 t In this example, the current of the voltage source can be controlled so that it is the same at the start and end of the longest charging or discharging period in one cycle.

[0067] Figure 12 is a graph showing the simulation results of the waveforms of the current i0 and the voltages Vc1, Vc2, and VR when the DC-DC converter 1 is operated in 4x boost mode. Figure 12 shows the results when feedback control of the target value based on the current difference and feedback control of the charging period based on the error between the target value and the capacitor voltage are performed, as shown in Figure 9. In the example shown in Figure 12, the value of the current i0 of the reactor (voltage source) matches at the start point t1 and the end point t2 of the charging period of capacitor C1. Furthermore, even though the reactor inductance is low (L = 1 μH), the current ripple is suppressed to 3%.

[0068] (Example of a limiting circuit) As shown in FIG. 1, the control unit 5 may include a limiting circuit 6. The limiting circuit 6 limits the conduction time of the short-circuit switch S4 during a preset limiting period. The limiting circuit 6 gradually shortens the limiting time during the limiting period and finally releases the restriction. For example, during the limiting period, the limiting circuit 6 sets an allowed time during which the short-circuit switch S4 is allowed to be conducted, in a cycle shorter than the cycle of the charging mode and the discharging mode. The allowed time is gradually lengthened during the limiting period. The allowed time is set to a time shorter than the pulse width that controls the charging period and the discharging period.

[0069] For example, overcurrent can be suppressed by setting a limiting period during a period when overcurrent is likely to flow, such as the initial charging period. During the initial operation of the DC-DC converter 1, when the voltage V0 of the voltage source 3 rises, switches S1 and S2 are turned on so that the smoothing capacitor Ca is also charged to voltage V0. If the DC-DC converter 1 cycles between charging and discharging modes from this state, the voltages of capacitors C1 and C2 are zero, causing excessive current to flow from the voltage source 3 through the reactor 4, as indicated by the arrows in Figure 13. This can be avoided by increasing the inductance L0 of the reactor 4. However, increasing the inductance L0 is not desirable from the perspective of weight reduction. Therefore, by limiting the conduction of the short-circuit switch S4 using the limiting circuit 6 during the initial charging period, the increase in current can be mitigated and excessive current can be prevented.

[0070] 14 is a diagram showing an example of the limiting pulse LP generated by the limiting circuit 6. The switch S4 is allowed to conduct only while the limiting pulse LP is at high level (H). While the limiting pulse LP is at low level (L), the switch S4 is forcibly made non-conductive. The example shown in FIG. 14 is an example in which the switches S1 to S4 operate in the quadruple boost mode as shown in FIG.

[0071] In the example of FIG. 14, the limiting circuit 6 generates, as the limiting pulse LP, a pulse train that is shorter than the operation cycle of the switches S1 to S4. The limiting period is a predetermined period after the initial charging of the smoothing capacitor is completed (the smoothing capacitor Ca is charged to V0) and the switches S1 to S4 start operating in the charge mode and discharge mode. During the limiting period, the switch S4 is controlled to be conductive only during the period when both the generated limiting pulse LP and the pulse for controlling the switch S4 are at high level (H), and to be non-conductive for the rest of the period. As a result, the period during which a path through which an excessive current flows is formed is limited to the period of the limiting pulse. As a result, even if the inductance of the reactor 4 is low, the generation of an excessive current can be suppressed.

[0072] The width of the limiting pulse LP increases stepwise over time. This gradually charges the voltages Vc1 and Vc2 of the capacitors C1 and C2. Once the capacitors C1 and C2 are charged to a certain extent, excessive current will no longer flow even if the width of the limiting pulse LP increases. In the example of FIG. 14, the limiting pulse LP is generated using a sawtooth-shaped carrier signal Cr2. The limiting pulse LP is output (at high level) while the carrier signal Cr2 is higher than the reference voltage VR4. The reference voltage VR4 increases over time. This increases the time that the switch S4 is allowed to conduct over time.

[0073] The period of the limiting pulse LP can be set shorter than the operation period of the switches S1 to S4. This allows the width of the limiting pulse LP to be set sufficiently shorter than the duration of the on or off of the switches S1 to S4. In the example of Fig. 14, the period of the carrier signal Cr2 is the period of the limiting pulse LP. For example, the period of the carrier signal Cr2 can be set to half or less of the periods in the charge mode and the discharge mode.

[0074] Furthermore, by setting the period of the limiting pulse LP so that the operating period of the switches S1 to S4 is not an integer multiple of the period of the limiting pulse LP, it is possible to distribute the permitted time approximately uniformly over the entire range of the period of the switches S1 to S4. In this case, the ratio of the average time during which the switches S1, S2, S3, and S4 are permitted to conduct is the same as the ratio of the width of the original conduction time of S1, S2, S3, and S4. This makes it easier to obtain the specified charging voltage and output voltage even after the period subject to the limit has passed.

[0075] The period limited by the limiting circuit 6 is set to at least a part of the transition period until the DC-DC converter 1 reaches a steady state. An example of the transition period is the period during which the capacitors C1 and C2 are initially charged. Another example of the transition period is the transition period when the target voltage values ​​V1 and V2 of the capacitors C1 and C2 are changed. For example, the transition period when the charge mode and discharge mode are changed in accordance with a change in the step-up ratio of the DC-DC converter 1 is a transition period. However, the transition period is not limited to these examples. When the charge mode and discharge mode are started in a state in which the voltages of the capacitors C1 and C2 are reset and need to be charged from zero, the limiting period can be set from the start.

[0076] (Example of DC power supply device configuration) Fig. 15 is a diagram showing an example of the configuration of a DC power supply device according to an embodiment of the present invention. The DC power supply device 10 is connected to a generator 11. The output voltage of the generator 11 varies depending on the rotation speed, i.e., angular frequency, of the generator 11. As an example, the generator 11 may be a permanent magnet synchronous generator. The DC power supply device 10 converts the AC output voltage of the generator 11 into DC and outputs DC power with a nearly constant voltage.

[0077] The DC power supply device 10 includes an ACDC converter 7 and a DCDC converter 1. The ACDC converter 7 is connected to a generator 11 and converts the output voltage of the generator 11 into DC. The DCDC converter 1 is connected after the ACDC converter 7. That is, the ACDC converter 7 is connected between the generator 11 and the DCDC converter 1. The DCDC converter 1 can be configured similarly to the DCDC converter 1 shown in FIG. 1. However, in the DCDC converter 1 of FIG. 15, the configuration (processing) of the control unit 5 is different from that of FIG. 1. The ACDC converter 7 may be, for example, a PWM converter.

[0078] The control unit 5 controls the boost ratio of the DC-DC converter 1 by changing the target voltages of the capacitors C1 and C2 of the bit inverters 20a and 20b and the operating mode of the bit inverters. The boost ratio of the DC-DC converter 1 is determined based on the angular frequency ωge or the output voltage Vge of the generator 11. The control unit 5 determines the target voltages of the capacitors C1 and C2 of the bit inverters 20a and 20b so that the boost ratio of the DC-DC converter 1 corresponds to the angular frequency ωge or the output voltage Vge of the generator 11. For example, as shown in FIG. 3, target voltages of the capacitors C1 and C2 for achieving the target boost ratio can be set. The operating mode of the bit inverters 20a and 20b is determined according to the target voltages. That is, the boost ratio can be switched by switching the target voltages and the operating mode of the bit inverters 20a and 20b. The control unit 5 can achieve the boost ratio by operating the bit inverters 20a and 20b in an operating mode in which the capacitors C1 and C2 are charged and discharged at a target voltage set according to the boost ratio.

[0079] In the charge mode, the control unit 5 controls the switching of the bit inverters 20a and 20b to charge the capacitors C1 and C2 to a target value. In the discharge mode, the voltages of the capacitors C1 and C2 are superimposed on the output voltage of the AC / DC converter 7 and output to the smoothing capacitor Ca. The control unit 5 controls the short-circuit switch S4 to switch between the charge mode and the discharge mode.

[0080] The DC power supply device 10 shown in FIG. 15 includes a DC-DC converter step-up rate setting circuit 8 and an AC-DC converter step-up rate setting circuit 9. The DC-DC converter step-up rate setting circuit 8 determines a DC-DC converter step-up rate M_DCDC according to the angular frequency ωge or output voltage Vge of the generator 11. The step-up rate M_DCDC is determined to a value that keeps the output voltage VR of the DC-DC converter 1, i.e., the load DC output voltage VDC2, close to a constant value. The DC-DC converter step-up rate setting circuit 8 can switch the step-up rate M_DCDC, for example, when a change in the angular frequency ωge or output voltage Vge of the generator 11 satisfies a predetermined condition. As an example, the step-up rate M_DCDC may be switched stepwise according to the range of the angular frequency ωge or the output voltage Vge. The step-up rate M_DCDC may be set to one of a plurality of predetermined discrete values, such as 1x, 1.33x, 1.5x, 2x, 3x, and 4x. The control unit 5 sets target values ​​for the voltages of the capacitors C1 and C2 in accordance with the determined boost rate M_DCDC, and operates the bit inverter in an operation mode in accordance with the boost rate M_DCDC.

[0081] The ACDC converter step-up rate setting circuit 9 is an example of an ACDC converter control unit. The ACDC converter step-up rate setting circuit 9 determines the ACDC converter step-up rate M_ACDC. The step-up rate M_ACDC is determined to a value that makes the load DC output voltage VDC2 approach a constant value. The ACDC converter step-up rate setting circuit 9 may determine the ACDC converter step-up rate M_ACDC using, for example, the DCDC converter step-up rate M_DCDC determined by the DCDC converter step-up rate setting circuit 8. Alternatively, the ACDC converter step-up rate setting circuit 9 may determine the ACDC converter step-up rate M_ACDC using the load DC output voltage VDC2.

[0082] The ACDC converter 7 is controlled based on the boost ratio M_ACDC determined by the ACDC converter boost ratio setting circuit 9. The ACDC converter 7 boosts the AC voltage input from the generator 11 at the boost ratio M_ACDC and outputs the boosted DC voltage. As an example, the ACDC converter 7 has a bridge circuit composed of diodes and switches connected in parallel. The boost ratio of the ACDC converter 7 can be controlled by PWM-controlled on / off of the bridge circuit switches. Alternatively, the ACDC converter 7 may be composed of, for example, a combination of a rectifier circuit and a boost circuit. In this case, the boost ratio can be controlled by controlling the switching of the boost open circuit, for example. A control unit that controls the boost ratio of the ACDC converter 7 based on the boost ratio M_ACDC may be included in the ACDC converter 7 or may be provided externally.

[0083] The AC / DC converter 7 may include a reactor for suppressing ripple current. A reactor may be connected between the AC / DC converter 7 and the generator 11. If the range of the step-up ratio of the AC / DC converter 7 becomes large, the reactor must also be made larger. From this perspective, it is preferable that the range of the step-up ratio of the AC / DC converter 7 is small.

[0084] (Example of operation) FIG. 16 is a graph showing an example of the relationship between the angular frequency ωge and the output voltage Vge of the generator 11 shown in FIG. 15. For example, the angular frequency and output power of a permanent magnet synchronous generator have characteristics as shown in FIG. 15. In the example shown in FIG. 15, the angular frequency ωge and the output voltage Vge of the generator 11 have a relationship of Vge=kωge. Furthermore, the angular frequency ωge and the output voltage Vge have upper limits, with the maximum values ​​being ωgemax and Vgemax, respectively. The angular frequency ωge is the angular frequency of rotation of the rotor of the generator 11.

[0085] FIG. 17 is a graph showing an example of setting the boost rate M_DCDC and the boost rate M_ACDC in response to fluctuations in the output voltage Vge of the generator 11 in the DC power supply device 10 shown in FIG. 15. FIG. 17 shows an example in which the output voltage Vge (or angular frequency ωge) of the generator 11 changes from the maximum value Vgemax to approximately Vgemax / 5. FIG. 17 also shows an example in which the target value VDC2ref of the load DC output voltage VDC2=the maximum value Vgemax of the output voltage. That is, this is an example in which k0=VDC2ref / Vgemax=1 (k0 is a constant). In this way, the target value VDC2ref can be determined based on the maximum value Vgemax of the output voltage.

[0086] In the example of Figure 17, when the output voltage Vge of the generator 11 drops from the maximum value Vgemax = target value VDC2ref = 5.00 to 3.76 (= 5 / 1.33), the DC-DC converter boost ratio setting circuit 8 increases the boost ratio M_DCDC of the DC-DC converter from 1 to 1.33. Next, when the output voltage Vge drops to 3.33 (= 5 / 1.5), the boost ratio M_DCDC is increased from 1.33 to 1.5. Furthermore, when the output voltage Vge reaches 2.5 (5 / 2), the boost ratio M_DCDC is doubled. When the output voltage Vge reaches 1.67 (5 / 3), the boost ratio M_DCDC is tripled. When the output voltage Vge reaches 1.25 (5 / 4), the boost ratio M_DCDC is quadrupled.

[0087] 17, the AC / DC converter step-up rate setting circuit 9 changes the step-up rate M_ACDC of the AC / DC converter according to the degree of deviation between the load DC output voltage VDC2 and its target value VDC2ref. Specifically, the step-up rate M_ACDC increases as the deviation between VDC2 and the target value increases, and decreases as the deviation decreases.

[0088] In the example shown in FIG. 17, the boost rate M_ACDC of the AC-DC converter follows the output voltage Vge of the generator 11 more closely than the boost rate M_DCDC of the DC-DC converter. That is, the boost rate M_ACDC is updated more frequently than the boost rate M_DCDC. Furthermore, the boost rate M_DCDC is switched stepwise to one of predetermined discrete values, whereas the boost rate M_ACDC is a value calculated according to the output voltage Vge. This allows for precise response to fluctuations in the output voltage Vge of the generator 11, thereby reducing fluctuations in the load DC output voltage VDC2 of the DC-DC converter 1.

[0089] Furthermore, the range of change in the boost rate M_ACDC of the AC / DC converter is smaller than the range of change in the boost rate M_DCDC of the DC / DC converter, at less than half. By keeping the boost rate of the AC / DC converter low, the increase in ripple current can be suppressed. This means that the AC / DC converter does not need a large reactor to suppress the increase in ripple current.

[0090] FIG. 18 is a flowchart showing an example of the operation of the DC-DC converter boost rate setting circuit 8. The DC-DC converter boost rate setting circuit 8 executes the operation shown in FIG. 18 at a predetermined cycle. This allows, for example, updating of the boost rate M_DCDC as shown in FIG. 17 to be realized. In the example shown in FIG. 18, the angular frequency ωge of the generator is compared with a preset numerical range, and the boost rate M_DCDC is determined depending on which numerical range the angular frequency ωge falls within. The numerical range that is the condition for determining the boost rate M_DCDC is determined using the angular frequency or maximum value of the output voltage (ωgemax or Vgemax) of the generator 11 (or the target value VDC2ref of the load DC output voltage) and the boost rate that can be set by the DC-DC converter 1 (in this example, 1, 1.33, 1.5, 2, 3, or 4). Note that in the example shown in FIG. 17, the angular frequency ωge and its maximum value ωwgemax of the generator 11 are used. Instead of the angular frequency ωge and its maximum value ωgemax, the output voltage Vge and its maximum value Vgemax may be used. Moreover, VDC2ref / (k·k0) may be used instead of ωgemax.

[0091] Next, an example of the operation of the AC / DC converter boost rate setting circuit 9 will be described. As an example, the AC / DC converter boost rate setting circuit 9 can perform calculations using the boost rate M_DCDC of the DC / DC converter, the angular frequency of the generator, or the maximum value of the output voltage (or the target value of the load DC output voltage). As a specific example, the boost rate M_ACDC of the AC / DC converter may be calculated using the following formula: M_ACDC=k0·k·ωgemax / (k·ωgemax·M_DCDC) =k0·Vgemax / (Vgemax·M_DCDC) When M_ACDC is set in this way, the load DC output voltage VDC2 is expressed by the following formula. VDC2=k·ωge·M_ACDC·M_DCDC =k·ωge·{k0·k·ωgemax / (k·ωgemax·M_DCDC)}·M_DCDC =k0·k·ωgemax

[0092] FIG. 19 is a diagram illustrating another example of the operation of the AC / DC converter step-up rate setting circuit 9. In the example shown in FIG. 19, the AC / DC converter step-up rate setting circuit 9 compares the load DC output voltage VDC2 with a target value VDC2ref and amplifies the error using PI integration to generate a command value M_ACDC. The command value M_ACDC is a value that controls the step-up rate of the AC / DC converter. The AC / DC converter 7 operates in accordance with the command value M_ACDC, thereby bringing the load DC output voltage VDC2 closer to the target value VDC2ref. In the example shown in FIG. 19, the AC / DC converter 7 is feedback-controlled using the detected load DC output voltage VDC2 so that VDC2 approaches the target value VDC2ref.

[0093] FIG. 20 is a diagram showing another example of the configuration of a DC power supply device according to an embodiment of the present invention. In FIG. 20, the AC-DC converter has three output levels. In the example shown in FIG. 20, the AC-DC converter has a positive line that outputs a positive voltage relative to GND and a negative line that outputs a negative voltage. A DC-DC converter 1 is connected to each of the positive and negative lines. The configuration of each DC-DC converter 1 can be similar to that of the DC-DC converter 1 shown in FIG. 1 or 15. In this way, by providing a DC-DC converter for each of the positive and negative output lines of the AC-DC converter, the generator-side GND and the load-side GND can be shared. This facilitates insulation design and EMC design.

[0094] 20 does not show the control unit 5, the DC-DC converter step-up rate setting circuit 8, and the AC-DC converter step-up rate setting circuit 9. The configuration for these controls may be shared by the two positive and negative DC-DC converters 1, or may be provided for each of the two positive and negative DC-DC converters.

[0095] 15 and 20, the control unit 5 may or may not have a function of feedback-controlling the charging period or discharging period depending on the error between the voltage of the capacitors C1 and C2 and its target value, similar to the control unit in Fig. 1. By having the control unit 5 perform this feedback control, the inductance of the reactor can be further reduced.

[0096] 1, the control unit 5 of the DC power supply device 10 may have a limiting circuit 6. For example, in the DC power supply device 10, when the control unit 5 changes the boost rate M_DCDC of the DC-DC converter 1, the current increases rapidly during a transition period immediately after the change of the boost rate M_DCDC. This transition period may be set as a limiting period, and the limiting circuit 6 may limit the conduction time of the short-circuit switch S4. In the DC power supply device 10, it is expected that the boost rate will be changed frequently, so the limiting circuit 6 is particularly effective in preventing overcurrent.

[0097] The DC power supply device 10 shown in FIGS. 15 and 20 can be applied to, for example, a DC power generation system for an electric aircraft. In this case, the generator 11 is directly or indirectly connected to the rotating shaft of a turbine engine. For example, the configuration may be such that the rotation of the rotating shaft of the turbine engine is transmitted to the generator 11, causing the generator 11 to rotate and generate power. The rotating shaft of the turbine engine may be connected to the generator 11 via a means for converting the rotation speed, such as a gear. The generator 11 may be, for example, a permanent magnet synchronous generator. The rotation speed of the aircraft's engine varies significantly from takeoff to cruising thereafter. For example, the engine rotation speed during cruising may decrease to 1 / 4 to 1 / 5 of that during takeoff. If the generator 11 is a permanent magnet type, the output power of the generator 11 may also decrease to 1 / 4 to 1 / 5, similar to the engine rotation speed. In such a case, by connecting the DC power supply device 10 of this embodiment to the generator 11, a constant DC voltage can be supplied even if the output power of the generator 11 fluctuates. In addition, the reactor can have low inductance, making the entire system lighter.

[0098] The DC power supply device 10 of this embodiment can be applied to systems other than the power generation system of an electric aircraft. For example, the DC power supply device of this embodiment can be applied to a system that converts AC power, which has a widely fluctuating voltage level, into DC power, which has a constant voltage. For example, the DC power supply device described above can be applied to a power generation system that uses the rotation of a motor in an electric vehicle, which is required to be lightweight.

[0099] (Other variations) The present invention is not limited to the above-described embodiments. For example, the boost ratio that can be realized by the DC-DC converter is not limited to the above-described examples. In addition, in the above-described examples, the inverter circuit 20 is configured to include two single-phase inverters, namely, bit inverters 20a and 20b. However, the inverter circuit 20 may include one single-phase inverter (bit inverter), or three or more single-phase inverters (bit inverters). The bridge circuit of the bit inverter may be a full bridge or a half bridge.

[0100] 1, the cathode of diode D4 is connected to the positive electrode of smoothing capacitor Ca. Diode D4 may be arranged so that its anode is connected to the negative electrode of smoothing capacitor Ca. Diode D4 may be replaced with a semiconductor switch element.

[0101] 1, the inverter circuit 20 is connected to the positive electrode of the smoothing capacitor Ca. The inverter circuit may be connected to the negative electrode side of the smoothing capacitor. Alternatively, the inverter circuit may be connected to both the positive and negative electrodes of the smoothing capacitor Ca.

[0102] Furthermore, the configuration of the feedback control is not limited to the examples shown in FIGS. 4 to 9. For example, the control unit 5 may perform feedback control of the charging period or the discharging period using another detection value in addition to the error between the target value and the detected voltage. The control unit 5 may perform feedback control of the charging period or the discharging period using the difference in current (i0 t1 -i0 t ), other detected values ​​may be used to feedback control the target value. Furthermore, the difference in the capacitor voltage or current that is the basis for feedback control is not limited to the value detected in the previous cycle. Feedback control can be performed using values ​​detected in the same cycle or values ​​detected in multiple cycles.

[0103] In the above example, the control unit 5 has both the function of feedback controlling the voltage of the capacitor and the function of a limiting circuit. The control unit 5 may be configured to have only one of these functions.

[0104] (Application example) The DC-DC converter of this embodiment can be used, for example, in devices that require weight reduction. For example, the DC-DC converter of this embodiment can be suitably applied to a DC-DC converter for an electric aircraft. The adoption of a lightweight DC grid is being considered for electric aircraft. A DC-DC converter is required for the DC grid. Conventional DC-DC converters are not suitable for installation on aircraft because the reactor is heavy. The DC-DC converter of this embodiment can be used as a DC-DC converter for electric aircraft because the reactor has low inductance and can be made lightweight. [Explanation of symbols]

[0105] 1: DC-DC converter, 3: voltage source, 4: reactor, 5: control unit, 6: limiting circuit

Claims

1. A reactor connected in series to a DC voltage source that supplies a constant voltage even when the load size fluctuates; an inverter circuit including a plurality of bit inverters connected to a subsequent stage of the reactor, each bit inverter having a capacitor and a bridge circuit that switches the direction of connection of the capacitor; a smoothing capacitor connected to a downstream side of the inverter circuit via an element that switches between conductive and non-conductive states, and that smoothes an output from the inverter circuit; a short-circuit switch for switching between a charge mode in which the voltage of each of the capacitors of the plurality of bit inverters is set to a target value using the power of the voltage source and a discharge mode in which the voltage of the voltage source and the voltage of the capacitors of the plurality of bit inverters are superimposed and output to the smoothing capacitor; a control unit that controls the plurality of bit inverters and the short-circuit switch to periodically repeat the operation of the charge mode and the discharge mode, The control unit feedback-controls the charging period of each of the capacitors of the plurality of bit inverters in the charging mode in each cycle, or the discharging period of the capacitors in the discharging mode, in accordance with the amount of error between the voltage of each of the capacitors of the plurality of bit inverters and its target value.

2. 2. The DC-DC converter according to claim 1, When the voltage of the capacitor is lower than the target value, the control unit performs feedback control to extend the charging period or shorten the discharging period.

3. 3. The DC-DC converter according to claim 1, The inverter circuit includes a plurality of the bit inverters, and the target values ​​of the capacitor voltages of the plurality of bit inverters are different from one another.

4. A reactor connected in series to a DC voltage source; an inverter circuit including at least one bit inverter connected to a subsequent stage of the reactor, the bit inverter having a capacitor and a bridge circuit that switches the direction of connection of the capacitor; a smoothing capacitor connected to a downstream side of the inverter circuit via an element that switches between conductive and non-conductive states, and that smoothes an output from the inverter circuit; a short-circuit switch for switching between a charge mode in which the voltage of the capacitor of the bit inverter is set to a target value using the power of the voltage source and a discharge mode in which the voltage of the voltage source and the voltage of the capacitor are superimposed and output to the smoothing capacitor; a control unit that controls the bit inverter and the short-circuit switch to periodically repeat the operation of the charge mode and the discharge mode, a limiting circuit for limiting the conduction time of the short-circuit switch during a predetermined limit period; The limiting circuit releases the limitation by gradually shortening the limit time during the limitation period.

5. 5. The DC-DC converter according to claim 4, The limiting circuit limits the conduction time by operating to allow the short-circuit switch to conduct for a time shorter than a pulse width that controls a charge period in the charge mode and a discharge period in the discharge mode.

6. 6. The DC-DC converter according to claim 4, The DC-DC converter, wherein the limited period is set to at least a part of a transition period until the DC-DC converter reaches a steady state.

7. 7. The DC-DC converter according to claim 6, The DC-DC converter, wherein the transient period includes at least one of a period for initially charging the capacitor and a transition period when the target value of the capacitor is changed.

8. The DC-DC converter according to any one of claims 1 to 7, The control unit feedback-controls the target value of the capacitor so as to reduce a difference in current of the voltage source between a start point and an end point of the longest charging period in the charging mode or a longest discharging period in the discharging mode.

9. an AC-DC converter that converts the generator's output voltage into direct current; a DC-DC converter connected to a subsequent stage of the AC-DC converter, The DC-DC converter comprises: a reactor connected in series to the AC-DC converter; an inverter circuit including a plurality of bit inverters connected to a subsequent stage of the reactor, each bit inverter having a capacitor and a bridge circuit that switches the direction of connection of the capacitor; a smoothing capacitor connected to a downstream side of the inverter circuit via an element that switches between conductive and non-conductive states, and that smoothes an output from the inverter circuit; a short-circuit switch for switching between a charge mode in which the voltage of each capacitor of the plurality of bit inverters is set to a target value using the power output from the ACDC converter and a discharge mode in which the output voltage of the ACDC converter and the voltage of the capacitors of the plurality of bit inverters are superimposed and output to the smoothing capacitor; a control unit that controls the plurality of bit inverters and the short-circuit switch to periodically repeat the operation of the charge mode and the discharge mode, The control unit changes the target value of the voltage of each capacitor of the plurality of bit inverters in the charging mode in accordance with the angular frequency or the output voltage of the generator, thereby changing the boost rate of the DC-DC converter.

10. 10. The DC power supply device according to claim 9, The DC power supply device further includes an AC-DC converter control unit that controls a step-up rate of the AC-DC converter so that the output voltage of the DC-DC converter approaches a constant value.

11. 11. The DC power supply device according to claim 9 or 10, the control unit feedback-controls a charging period of the capacitor in the charging mode or a discharging period of the capacitor in the discharging mode in each cycle according to an error between the voltage of the capacitor of the bit inverter and its target value.

12. The DC power supply device according to any one of claims 9 to 11, the DC-DC converter includes a limiting circuit that limits a conduction time of the short-circuit switch during a predetermined limiting period; The limiting circuit releases the restriction by gradually shortening the restriction time during the restriction period.

Images