Laser interferometer

Abstract

To provide a laser interferometer capable of suppressing a decrease in demodulation accuracy even when disturbance such as a temperature change is applied.SOLUTION: A laser interferometer comprises: a laser light source that emits laser light; an optical modulator that includes a vibration element driven by a driving signal and superimposes a modulation signal on the laser light; a photodetector that receives the laser light including a sample signal superimposed due to reflection by an object and the laser light including the modulation signal, and outputs a light receiving signal; a calculation section that performs a calculation on the light receiving signal on the basis of a reference signal; and a signal generation section that outputs the driving signal and the reference signal. The calculation section includes: a preprocessing section that outputs a preprocessing signal including a frequency modulation component on the basis of the reference signal; a demodulation processing section that demodulates the sample signal from the preprocessing signal on the basis of the reference signal; and a correction processing section that outputs a correction signal on the basis of an output signal output in response to driving of the vibration element. The signal generation section corrects the driving signal and the reference signal on the basis of the correction signal.SELECTED DRAWING: Figure 1

Description

[Technical Field]

[0001] This invention relates to a laser interferometer. [Background technology]

[0002] Patent Document 1 discloses a laser vibrometer (laser interferometer) that measures vibration velocity based on scattered laser light that has undergone a Doppler shift after irradiating an object with laser light. In this laser vibrometer, the Doppler signal contained in the scattered laser light is extracted by using optical heterodyne interferometry.

[0003] Furthermore, the laser vibrometer described in Patent Document 1 uses a piezoelectric element or a quartz crystal oscillator, and the frequency is shifted by irradiating these vibrating elements with laser light. By using the laser light containing the modulated signal with the shifted frequency in this way as a reference light, the Doppler signal is demodulated from the scattered laser light. Based on the obtained Doppler signal, the vibration velocity of an object can be measured. [Prior art documents] [Patent Documents]

[0004] [Patent Document 1] Japanese Patent Publication No. 2007-285898 [Overview of the project] [Problems that the invention aims to solve]

[0005] For example, in the case of a quartz crystal oscillator, its mechanical resonant frequency is affected by the ambient temperature and changes accordingly. Therefore, when a quartz crystal oscillator is used to shift the frequency of laser light and generate a modulated signal, the modulated signal changes with temperature changes. This reduces the signal-to-noise ratio (S / N ratio) of the modulated signal, leading to a decrease in the demodulation accuracy of the Doppler signal. For this reason, a challenge is to realize a laser interferometer that can suppress the decrease in demodulation accuracy even when disturbances such as temperature changes are applied. [Means for solving the problem]

[0006] A laser interferometer according to an application example of the present invention is, A laser light source that emits laser light, An optical modulator comprising a vibrating element driven by a drive signal, which superimposes a modulation signal onto the laser light using the vibrating element, A photodetector receives the laser light including the sample signal superimposed by reflection from an object, and the laser light including the modulated signal, and outputs a received signal. A calculation unit that performs calculations on the received light signal based on a reference signal, A signal generation unit that outputs the drive signal and the reference signal, Equipped with, The aforementioned arithmetic unit, A preprocessing unit that performs preprocessing to extract frequency modulation components from the received light signal based on the reference signal and outputs a preprocessed signal including the frequency modulation components, A demodulation processing unit that demodulates the sample signal from the preprocessed signal based on the reference signal, A correction processing unit outputs a correction signal based on the output signal that is output in response to the drive of the vibration element, It has, The signal generation unit is characterized by correcting the drive signal and the reference signal based on the correction signal. [Brief explanation of the drawing]

[0007] [Figure 1] This is a functional block diagram showing a laser interferometer according to the first embodiment. [Figure 2] Figure 1 is a schematic diagram showing the sensor head. [Figure 3] Figure 2 is a perspective view showing a first configuration example of an optical modulator. [Figure 4] This is a plan view showing a part of a second configuration example of an optical modulator. [Figure 5] This is a plan view showing a third configuration example of an optical modulator. [Figure 6]This is a conceptual diagram illustrating the generation of multiple diffracted light beams when incident light Ki is incident on the surface of a vibrating element from a direction perpendicular to the surface. [Figure 7] This is a conceptual diagram illustrating an optical modulator configured such that the angle between the direction of propagation of the incident light Ki and the direction of propagation of the reference light L2 is 180°. [Figure 8] This is a conceptual diagram illustrating an optical modulator configured such that the angle between the direction of propagation of the incident light Ki and the direction of propagation of the reference light L2 is 180°. [Figure 9] This is a conceptual diagram illustrating an optical modulator configured such that the angle between the direction of propagation of the incident light Ki and the direction of propagation of the reference light L2 is 180°. [Figure 10] This diagram provides a detailed view of the arithmetic unit within the functional block diagram shown in Figure 1. [Figure 11] Figure 10 is a flowchart showing an example of how the phase amount setting unit sets the phase amount. [Figure 12] Figure 11 shows an example of waveforms representing the pre-processed signal S(t) before calibration of the laser interferometer based on the flow chart, the phase information demodulated in the demodulation processing unit (phase information before unwrapping), and the phase information unwrapped in the signal output unit (phase information after unwrapping). [Figure 13] Figure 11 shows an example of waveforms representing the pre-processed signal S(t) after calibration of the laser interferometer based on the flow chart, the phase information demodulated in the demodulation processing unit (phase information before unwrapping), and the phase information unwrapped in the signal output unit (phase information after unwrapping). [Figure 14] This diagram provides a detailed view of the correction processing unit, which is part of the functional block diagram shown in Figure 1. [Figure 15] This figure shows an example of a circuit for acquiring the output signal Sm from an optical modulator. [Figure 16] This is a functional block diagram showing a laser interferometer according to the second embodiment. [Figure 17] This diagram provides a detailed view of the arithmetic unit and signal generation unit from the functional block diagram in Figure 16. [Figure 18]This is a schematic diagram showing the optical system related to the first modified example. [Figure 19] This is a schematic diagram showing the optical system related to the second modified example. [Figure 20] This is a schematic diagram showing the optical system related to the third modified example. [Figure 21] This is a schematic diagram showing the optical system related to the fourth modified example. [Modes for carrying out the invention]

[0008] The laser interferometer of the present invention will be described in detail below based on the embodiments shown in the accompanying drawings. 1. First Embodiment First, the laser interferometer according to the first embodiment will be described.

[0009] Figure 1 is a functional block diagram showing a laser interferometer according to the first embodiment. Figure 2 is a schematic configuration diagram showing the sensor head 51 of Figure 1.

[0010] The laser interferometer 1 shown in Figure 1 comprises a sensor head 51 and a main body 59. The sensor head 51 is easily miniaturized and lightweight, and is easily portable and easy to install, so it can be placed, for example, near the object 14 shown in Figure 2, which is the object to be measured by the laser interferometer 1. The main body 59 can be placed away from the sensor head 51 and may be housed, for example, in a rack.

[0011] 1.1. Sensor head section The sensor head unit 51 shown in Figure 1 comprises an optical system 50, a current-voltage converter 531, a signal generation unit 61, and a correction processing unit 62. The main unit 59 shown in Figure 1 comprises a pre-processing unit 53, an orthogonal signal generation unit 57, a demodulation processing unit 55, and a signal output unit 559.

[0012] 1.1.1.Optical system As shown in Figure 2, the optical system 50 includes a laser light source 2, a collimating lens 3, an optical divider 4, a half-wave plate 6, a quarter-wave plate 7, a quarter-wave plate 8, an analyzer 9, a photodetector 10, and a frequency shifter type optical modulator 12.

[0013] The laser light source 2 emits outgoing light L1 (laser light). The photodetector 10 converts the received light into an electrical signal. The optical modulator 12 is equipped with a vibrating element 30, which changes the frequency of the outgoing light L1 and generates a reference light L2 (laser light including the modulation signal) that includes the modulation signal. The outgoing light L1 incident on the object 14 is reflected as object light L3 (laser light including the sample signal) that includes a sample signal, which is a Doppler signal originating from the object 14.

[0014] The optical path connecting the optical divider 4 and the laser light source 2 is designated as optical path 18. The optical path connecting the optical divider 4 and the optical modulator 12 is designated as optical path 20. The optical path connecting the optical divider 4 and the object 14 is designated as optical path 22. The optical path connecting the optical divider 4 and the light-receiving element 10 is designated as optical path 24. In this specification, "optical path" refers to the path through which light travels, set between optical components.

[0015] On optical path 18, the half-wave plate 6 and the collimating lens 3 are arranged in that order from the optical splitter 4 side. On optical path 20, the quarter-wave plate 8 is arranged. On optical path 22, the quarter-wave plate 7 is arranged. On optical path 24, the analyzer 9 is arranged.

[0016] The emitted light L1 from the laser light source 2 travels through the optical path 18 and is split into two by the optical splitter 4. One of the split emitted light L1, the first split light L1a, travels through the optical path 20 and is incident on the optical modulator 12. The other split emitted light L1, the second split light L1b, travels through the optical path 22 and is incident on the object 14. The reference light L2 generated by the frequency modulation in the optical modulator 12 travels through the optical paths 20 and 24 and is incident on the photodetector 10. The object light L3 generated by reflection from the object 14 travels through the optical paths 22 and 24 and is incident on the photodetector 10.

[0017] The following provides a further explanation of each part of the optical system 50. 1.1.1.1. Laser light source Laser light source 2 is a laser light source that emits coherent emitted light L1. Preferably, laser light source 2 uses a light source with a linewidth of MHz or less. Specifically, examples include gas lasers such as He-Ne lasers, DFB-LDs (Distributed Feedback Laser Diodes), FBG-LDs (Fiber Bragg Grating Laser Diodes), VCSELs (Vertical Cavity Surface Emitting Lasers), and semiconductor laser elements such as FP-LDs (Fabry-Perot Laser Diodes).

[0018] The laser light source 2 is preferably a semiconductor laser element. This makes it possible to miniaturize the laser light source 2. As a result, the laser interferometer 1 can be miniaturized. In particular, the sensor head section 51, which houses the optical system 50, can be miniaturized and lightened, which is useful in improving the operability of the laser interferometer 1, such as the degree of freedom in installing the sensor head section 51.

[0019] 1.1.1.2. Collimating Lenses The collimating lens 3 is an optical element placed between the laser light source 2 and the optical divider 4, and an aspherical lens is one example. The collimating lens 3 parallelizes the emitted light L1 from the laser light source 2. However, if the emitted light L1 from the laser light source 2 is sufficiently parallelized, for example, if a gas laser such as a He-Ne laser is used as the laser light source 2, the collimating lens 3 may be omitted.

[0020] On the other hand, when the laser light source 2 is a semiconductor laser element, it is preferable that the laser interferometer 1 includes a collimating lens 3 positioned between the laser light source 2 and the optical divider 4. This allows the emitted light L1 from the semiconductor laser element to be parallelized. As a result, the emitted light L1 becomes collimated light, which suppresses the need to enlarge the various optical components that receive the emitted light L1, and thus allows for miniaturization of the laser interferometer 1.

[0021] The collimated light L1 passes through the half-wave plate 6, where it is converted into linearly polarized light with a P-polarized to S-polarized intensity ratio of, for example, 50:50, and then enters the light divider 4.

[0022] 1.1.1.3.Light splitter The optical splitter 4 is a polarized beam splitter placed between the laser light source 2 and the optical modulator 12, and between the laser light source 2 and the object 14. The optical splitter 4 has the function of transmitting P-polarized light and reflecting S-polarized light. Due to this function, the optical splitter 4 splits the emitted light L1 into a first split beam L1a, which is the light reflected by the optical splitter 4, and a second split beam L1b, which is the light transmitted by the optical splitter 4.

[0023] The first split beam L1a, which is S-polarized and reflected by the optical splitter 4, is converted to circular polarization by the quarter-wave plate 8 and incident on the optical modulator 12. The first split beam L1a incident on the optical modulator 12 is f m It undergoes a frequency shift of [Hz] and is reflected as reference light L2. Therefore, the reference light L2 has a frequency f m The modulated signal is in [Hz]. The reference light L2 is converted to P-polarized light when it passes through the quarter-wave plate 8 again. The P-polarized light of the reference light L2 passes through the light divider 4 and the analyzer 9 and is incident on the photodetector 10.

[0024] The second split beam L1b, which is P-polarized after passing through the light splitter 4, is converted to circular polarization by the quarter-wave plate 7 and incident on the moving object 14. The second split beam L1b incident on the object 14 is f dIt undergoes a Doppler shift of [Hz] and is reflected as object light L3. Therefore, object light L3 has a frequency of f d The sample signal is in [Hz]. The object light L3 is converted to S-polarized light when it passes through the quarter-wave plate 7 again. The S-polarized object light L3 is reflected by the light divider 4, passes through the analyzer 9, and is incident on the photodetector 10.

[0025] As mentioned above, since the emitted light L1 is coherent, the reference light L2 and object light L3 are incident on the photodetector 10 as interfering light.

[0026] Alternatively, a non-polarizing beam splitter may be used instead of a polarizing beam splitter. In this case, the half-wave plate 6, quarter-wave plate 7, and quarter-wave plate 8, etc., become unnecessary, allowing for miniaturization of the laser interferometer 1 by reducing the number of components. Furthermore, other types of optical dividers may be used instead of a beam splitter.

[0027] 1.1.1.4. Analyzer Since S-polarized and P-polarized light, which are orthogonal to each other, are independent of each other, simply superimposing them will not produce beats due to interference. Therefore, the light wave obtained by superimposing S-polarized and P-polarized light is passed through an analyzer 9 tilted at 45° with respect to both the S-polarized and P-polarized light. By using analyzer 9, light with common components can be transmitted, causing interference. As a result, in analyzer 9, the reference light L2 and the object light L3 interfere, and |f m -f d Interfering light with a frequency of |[Hz] is generated.

[0028] 1.1.1.5. Photodetector When interfering light is incident on the photodetector 10, the photodetector 10 outputs a photocurrent (received signal) corresponding to the intensity of the interfering light. By demodulating the sample signal from this received signal using a method described later, the movement of the object 14, i.e., its displacement and velocity, can ultimately be determined. Examples of the photodetector 10 include a photodiode. Note that the light received by the photodetector 10 can be any light containing the sample signal and the modulation signal, and is not limited to the interfering light described above. Furthermore, "demodulating the sample signal from the received signal" in this specification includes demodulating the sample signal from various signals converted from the photocurrent (received signal).

[0029] 1.1.1.6. Optical Modulators Figure 3 is a perspective view showing a first configuration example of the optical modulator 12 in Figure 2.

[0030] 1.1.1.6.1. Overview of the First Configuration Example of an Optical Modulator The frequency shifter type optical modulator 12 has an optical modulation oscillator 120. The optical modulation oscillator 120 shown in Figure 3 comprises a plate-shaped vibrating element 30 and a substrate 31 that supports the vibrating element 30.

[0031] The vibrating element 30 is made of a material that, when an electric potential is applied, repeatedly vibrates in a mode that deforms in a direction along the surface. In this example configuration, the vibrating element 30 is a quartz AT resonator that vibrates in thickness shear along the vibration direction 36 in the high-frequency region of the MHz band. A diffraction grating 34 is formed on the surface of the vibrating element 30. The diffraction grating 34 has grooves 32 that have components intersecting the vibration direction 36, that is, a plurality of linear grooves 32 that extend in a direction intersecting the vibration direction 36.

[0032] The substrate 31 has a front surface 311 and a back surface 312, which are in a front-back relationship with each other. A vibrating element 30 is placed on the front surface 311. The front surface 311 is also provided with a pad 33 for applying an electric potential to the vibrating element 30. On the other hand, the back surface 312 is also provided with a pad 35 for applying an electric potential to the vibrating element 30.

[0033] The size of the substrate 31 is, for example, approximately 0.5 mm to 10.0 mm on its longest side. The thickness of the substrate 31 is, for example, approximately 0.10 mm to 2.0 mm. As an example, the shape of the substrate 31 is a square with sides of 1.6 mm and a thickness of 0.35 mm.

[0034] The size of the vibrating element 30 is, for example, approximately 0.2 mm to 3.0 mm on its longest side. The thickness of the vibrating element 30 is, for example, approximately 0.003 mm to 0.5 mm.

[0035] For example, the shape of the vibrating element 30 is assumed to be a square with sides of 1.0 mm and a thickness of 0.07 mm. In this case, the vibrating element 30 oscillates at a fundamental oscillation frequency of 24 MHz. By changing the thickness of the vibrating element 30 or by considering overtones, it is possible to adjust the oscillation frequency within the range of 1 MHz to 1 GHz.

[0036] In Figure 3, the diffraction grating 34 is formed on the entire surface of the vibrating element 30, but it may be formed only on a portion of the surface.

[0037] The magnitude of the optical modulation by the optical modulator 12 is given by the dot product of the difference wave vector between the wave vector of the emitted light L1 incident on the optical modulator 12 and the wave vector of the reference light L2 emitted from the optical modulator 12, and the vector of the vibration direction 36 of the vibrating element 30. In this configuration example, the vibrating element 30 vibrates with thickness shear, but since this vibration is in-plane vibration, optical modulation cannot be achieved even if light is incident perpendicularly to the surface of the vibrating element 30 alone. Therefore, in this configuration example, by providing a diffraction grating 34 on the vibrating element 30, optical modulation is made possible by the principle described later.

[0038] The diffraction grating 34 shown in Figure 3 is a blazed diffraction grating. A blazed diffraction grating is a diffraction grating whose cross-sectional shape is stepped. The linear grooves 32 of the diffraction grating 34 are provided so that their extension direction is perpendicular to the vibration direction 36.

[0039] When a drive signal Sd is supplied (an AC voltage is applied) from the signal generation unit 61 shown in Figures 1 and 2 to the vibration element 30 shown in Figure 3, the vibration element 30 oscillates. The power required for the oscillation of the vibration element 30 (drive power) is not particularly limited, but is small, ranging from about 0.1 μW to 100 mW. Therefore, the sensor head unit 51 can be easily miniaturized and made lighter.

[0040] Furthermore, conventional optical modulators sometimes require a structure to maintain the temperature of the optical modulator, making it difficult to reduce their volume. In contrast, in this configuration, a structure to maintain the temperature is unnecessary, so the volume of the vibrating element 30 is very small. Therefore, from this perspective as well, miniaturization and power saving of the laser interferometer 1 are easily achieved.

[0041] 1.1.1.6.2. Method for forming diffraction gratings The method for forming the diffraction grating 34 is not particularly limited, but one example is to create a mold using a mechanical engraving method (ruling engine) and then form grooves 32 on the electrodes deposited on the surface of the vibrating element 30 of the quartz AT resonator using a nanoimprint method. The reason for using the electrodes is that, in the case of a quartz AT resonator, high-quality thickness-sliding vibrations can be generated on the electrodes in principle. Note that the grooves 32 are not limited to being formed on the electrodes, but may also be formed on the surface of the non-electrode material. In addition, instead of the nanoimprint method, processing methods such as exposure and etching, electron beam lithography, and focused ion beam (FIB) processing may be used.

[0042] Alternatively, a diffraction grating may be formed on the quartz AT oscillator chip using a resist material, and a mirror film made of a metal film or dielectric multilayer film may be provided thereon. By providing a metal film or mirror film, the reflectivity of the diffraction grating 34 can be increased.

[0043] Furthermore, a resist film may be formed on the chip or wafer of the quartz AT resonator, processed by etching, the resist film may be removed, and then a metal film or mirror film may be formed on the processed surface. In this case, since the resist material is removed, the effects of moisture absorption by the resist material are eliminated, and the chemical stability of the diffraction grating 34 can be improved. In addition, by providing a highly conductive metal film such as Au or Al, it can also be used as an electrode to drive the vibrating element 30.

[0044] The diffraction grating 34 may also be formed using techniques such as anodized alumina (porous alumina).

[0045] 1.1.1.6.3. Other Configuration Examples of Optical Modulators The vibrating element 30 is not limited to a quartz crystal oscillator, but may be, for example, a silicon oscillator, a surface acoustic wave (SAW) device, a ceramic oscillator, or the like.

[0046] Figure 4 is a plan view showing a part of the second configuration example of the optical modulator 12. Figure 5 is a plan view showing the third configuration example of the optical modulator 12.

[0047] The vibrating element 30A shown in Figure 4 is a Si resonator manufactured from a Si substrate using MEMS technology. MEMS (Micro Electro Mechanical Systems) refers to micro-electromechanical systems.

[0048] The vibrating element 30A comprises a first electrode 301 and a second electrode 302 adjacent to each other on the same plane with a gap between them, a diffraction grating mounting section 303 provided on the first electrode 301, and a diffraction grating 34 provided on the diffraction grating mounting section 303. The first electrode 301 and the second electrode 302 vibrate, for example, using electrostatic attraction as a driving force, repeatedly approaching and separating from each other in the left-right direction of Figure 4, that is, along the axis connecting the first electrode 301 and the second electrode 302 shown in Figure 4. This allows in-plane vibration to be imparted to the diffraction grating 34. The oscillation frequency of the Si vibrator is, for example, from 1 kHz to several hundred MHz.

[0049] The vibration element 30B shown in FIG. 5 is a SAW device that utilizes surface waves. SAW (Surface Acoustic Wave) refers to elastic surface waves.

[0050] The vibration element 30B includes a piezoelectric substrate 305, a comb-shaped electrode 306 provided on the piezoelectric substrate 305, a ground electrode 307, a diffraction grating placement portion 303, and a diffraction grating 34. When an alternating voltage is applied to the comb-shaped electrode 306, elastic surface waves are excited due to the inverse piezoelectric effect. Thereby, in-plane vibration can be applied to the diffraction grating 34. The oscillation frequency of the SAW device is, for example, on the order of several 100 MHz to several GHz.

[0051] Regarding the device as described above, by providing the diffraction grating 34, optical modulation can be achieved by the principle described later, similar to the case of a crystal AT oscillator.

[0052] When the vibration element 30 is a crystal oscillator, a highly accurate modulation signal can be generated by utilizing the extremely high Q value of the crystal. The Q value is an index indicating the sharpness of the resonance peak. Also, the crystal oscillator has the characteristic of being less affected by disturbances. Therefore, by using the modulation signal modulated by the optical modulator 12 including the crystal oscillator, the sample signal derived from the object 14 can be acquired with high accuracy.

[0053] 1.1.1.6.4. Optical Modulation by Vibration Element Next, the principle of modulating light using the vibration element 30 will be described.

[0054] FIG. 6 is a conceptual diagram illustrating that when incident light K i is incident from a direction perpendicular to the surface of the vibration element 30, a plurality of diffracted lights are generated.

[0055] When the incident light K i is incident on the diffraction grating 34 that is performing thickness-shear vibration along the vibration direction 36, a plurality of diffracted lights K ns are generated due to the diffraction phenomenon, as shown in FIG. 6. n is the diffracted light Kns The order is such that n = 0, ±1, ±2, ... Note that the diffraction grating 34 shown in Figure 6 is not the blazed diffraction grating shown in Figure 3, but rather an example of a diffraction grating made by repeating concaves and convexities. Also, in Figure 6, the diffracted light K 0s The illustration is omitted.

[0056] In Figure 6, the incident light K i The light is incident on the surface of the vibrating element 30 from a direction perpendicular to it, but this angle of incidence is not particularly limited, and the angle of incidence may be set so that it is incident on the surface of the vibrating element 30 at an angle. When incident at an angle, the diffracted light K ns The direction of movement also changes accordingly.

[0057] Depending on the design of the diffraction grating 34, higher-order light with |n|≧2 may not appear. Therefore, in order to obtain a stable modulation signal, it is desirable to set |n|=1. That is, in the laser interferometer 1 shown in Figure 2, it is preferable to arrange the frequency shifter type optical modulator 12 so that ±1st order diffracted light is used as the reference light L2. This arrangement makes it possible to stabilize the measurement by the laser interferometer 1.

[0058] On the other hand, if higher-order light of |n|≧2 appears from the diffraction grating 34, the optical modulator 12 may be arranged so that any of the diffracted light of order ±2 or higher, rather than ±1st order diffracted light, is used as the reference light L2. This allows the use of higher-order diffracted light, thereby enabling higher frequency and miniaturization of the laser interferometer 1.

[0059] In this embodiment, as an example, the incident light K incident on the optical modulator 12 i The optical modulator 12 is configured such that the angle between the direction of entry of the light and the direction of propagation of the reference light L2 emitted from the optical modulator 12 is 180°. Three examples will be explained below with reference to Figures 7 to 9.

[0060] Figures 7 to 9 show the incident light K, respectively. iThis is a conceptual diagram illustrating an optical modulator 12 configured such that the angle between the direction of propagation of the light source and the direction of propagation of the reference light L2 is 180°.

[0061] The optical modulator 12 shown in Figure 7 is equipped with a mirror 37 in addition to the vibrating element 30. The mirror 37 is diffracted by the light K 1s It is positioned to reflect the light back to the diffraction grating 34. At this time, the diffracted light K directed to the mirror 37 1s The angle between the incident angle and the reflection angle at mirror 37 is 180°. As a result, the diffracted light K emitted from mirror 37 and returned to the diffraction grating 34 1s The incident light K is diffracted again by the diffraction grating 34 and incident into the optical modulator 12. i It will travel in the opposite direction to the direction of travel. Therefore, by adding mirror 37, the incident light K i The condition that the angle between the entry direction of the light and the direction of propagation of the reference light L2 is 180° can be satisfied.

[0062] Furthermore, by passing the light through the mirror 37 in this manner, the reference light L2 generated by the optical modulator 12 undergoes two frequency modulations. Therefore, by using the mirror 37 in combination, higher frequency modulation becomes possible compared to when the vibrating element 30 is used alone.

[0063] In Figure 8, the vibrating element 30 is tilted relative to the arrangement in Figure 6. The tilt angle θ in this case is S This is the incident light K mentioned above. i The setting is such that the angle between the entry direction of the light source and the direction of propagation of the reference light L2 is 180°.

[0064] The diffraction grating 34 shown in Figure 9 has a blaze angle θ. B This is a blazed diffraction grating having the following characteristics: The incident light K propagates at an incident angle β with respect to the normal N of the surface of the vibrating element 30. i When the light is incident on the diffraction grating 34, the blaze angle θ is relative to the normal N. B The reference light L2 will return at the same angle. Therefore, the incident angle β is equal to the blaze angle θ. BBy making it equal to the aforementioned incident light K i The condition that the angle between the entry direction of the light and the propagation direction of the reference light L2 is 180° can be satisfied. In this case, the above condition can be satisfied without using the mirror 37 shown in Figure 7, and without tilting the vibrating element 30 itself as shown in Figure 8, so that the laser interferometer 1 can be further miniaturized and its frequency can be increased. In particular, in the case of a blazed diffraction grating, the arrangement that satisfies the above condition is called the "Littrow arrangement," and it has the advantage of being able to particularly increase the diffraction efficiency of the diffracted light.

[0065] Note that the pitch P in Figure 9 represents the pitch of the blazed diffraction grating, and as an example, the pitch P is assumed to be 1 μm. Also, the blaze angle θ B For example, it is set to 25°. In this case, in order to satisfy the above condition, the incident light K i The angle of incidence β relative to the normal N should also be set to 25°.

[0066] The diffraction grating 34 may be provided only if necessary. For example, if the vibrating element 30 is an element that vibrates out of plane, the efficiency of optical modulation of the emitted light L1 incident on the vibrating element 30 can be increased even without using the diffraction grating 34. In such cases, the diffraction grating 34 may be omitted.

[0067] Furthermore, the vibrating element 30 is not limited to a quartz crystal oscillator, but may also be a silicon oscillator or a ceramic oscillator.

[0068] 1.1.2. Current-Voltage Converters The current-voltage converter 531, also known as a transimpedance amplifier (TIA), converts the photocurrent (received signal) output from the photodetector 10 into a voltage signal and outputs it as a photodetection signal.

[0069] An ADC532, as shown in Figure 1, is positioned between the current-voltage converter 531 and the calculation unit 52. The ADC532 is an analog-to-digital converter that converts an analog signal into a digital signal with a predetermined number of sampling bits. The ADC532 is provided in the sensor head unit 51.

[0070] The optical system 50 may include multiple photodetectors 10. In this case, by providing a differential amplifier circuit between the multiple photodetectors 10 and the current-voltage converter 531, differential amplification processing can be applied to the photocurrent, thereby increasing the signal-to-noise ratio (S / N ratio) of the photodetection signal. The differential amplification processing may also be applied to the voltage signal.

[0071] 1.1.3. Signal Generation Section The signal generation unit 61 outputs a drive signal Sd to the optical modulator 12. The signal generation unit 61 also outputs a reference signal Ss to the calculation unit 52.

[0072] As shown in Figure 1, the signal generation unit 61 includes a voltage-controlled oscillator 612 and an amplifier 614.

[0073] The voltage-controlled oscillator 612 is a VCO (Voltage Controlled Oscillator) and has the function of controlling the frequency of the output periodic signal based on the input voltage signal. As a result, the voltage-controlled oscillator 612 generates a reference signal Ss of the desired frequency and outputs it to the amplifier 614 and the arithmetic unit 52. Note that the voltage-controlled oscillator 612 is not limited to a VCO; any oscillator capable of adjusting the frequency of the output periodic signal is acceptable.

[0074] The amplifier 614 has the function of controlling the amplitude of the output periodic signal based on the input control signal. In this way, the amplifier 614 amplifies the input reference signal Ss, generates a drive signal Sd of the desired amplitude, and outputs it to the optical modulator 12.

[0075] 1.2. Arithmetic section Figure 10 is a diagram that shows the arithmetic unit 52 in detail, as shown in the functional block diagram of Figure 1.

[0076] The calculation unit 52 shown in Figure 1 includes a preprocessing unit 53, an orthogonal signal generation unit 57, a demodulation processing unit 55, a signal output unit 559, and a correction processing unit 62.

[0077] The calculation unit 52 performs demodulation processing to demodulate the sample signal originating from the object 14 from the photodetection signal output from the current-voltage converter 531. The sample signal includes, for example, phase information and frequency information. From the phase information, the displacement of the object 14 can be obtained, and from the frequency information, the velocity of the object 14 can be obtained. By obtaining different physical quantities in this way, the laser interferometer 1 can be given functions as a displacement meter and a velocity meter, thereby enhancing its functionality.

[0078] In the calculation unit 52, the circuit configuration is set according to the modulation processing method. In the laser interferometer 1 according to this embodiment, an optical modulator 12 equipped with a vibrating element 30 is used. Since the vibrating element 30 is an element that undergoes simple harmonic motion, the vibration velocity changes moment by moment within the period. For this reason, the modulation frequency also changes with time, and conventional demodulation circuits cannot be used as is.

[0079] Conventional demodulation circuits refer to circuits that demodulate a sampled signal from a photodetector signal that includes a modulated signal modulated using, for example, an acousto-optic modulator (AOM). In an acousto-optic modulator, the modulation frequency does not change with respect to time unless affected by external disturbances such as temperature changes. Therefore, conventional demodulation circuits can demodulate a sampled signal from a photodetector signal that includes a modulated signal with a constant modulation frequency, but they cannot demodulate a signal that includes a modulated signal modulated by an optical modulator 12 whose modulation frequency changes (periodically) with respect to time.

[0080] As described above, the calculation unit 52 shown in Figure 1 includes a pre-processing unit 53, an orthogonal signal generation unit 57, a demodulation processing unit 55, a signal output unit 559, and a correction processing unit 62. The photodetection signal output from the current-voltage converter 531 is first passed through the pre-processing unit 53 and then led to the demodulation processing unit 55. This pre-processing extracts the frequency modulation component from the photodetection signal, obtaining a signal that can be demodulated by a conventional demodulation circuit. Therefore, the demodulation processing unit 55 demodulates the sample signal originating from the object 14 using a known demodulation method. Furthermore, the orthogonal signal generation unit 57 generates an orthogonal signal, a cosine wave signal cos(θ), based on the reference signal Ss output from the signal generation unit 61 and the pre-processed signal S(t) output from the pre-processing unit 53, as shown in Figure 10. m (t)) and sinusoidal signal sin(θ m (t)) is generated.

[0081] The functions of the arithmetic unit 52 described above are realized by hardware including, for example, a processor, memory, external interface, input unit, display unit, etc. Specifically, this is realized by the processor reading and executing a program stored in memory. These components can communicate with each other via an internal bus.

[0082] Examples of processors include CPUs (Central Processing Units) and DSPs (Digital Signal Processors). Alternatively, instead of these processors executing software, FPGAs (Field-Programmable Gate Arrays) or ASICs (Application-Specific Integrated Circuits) may be used to implement the aforementioned functions.

[0083] Examples of memory include HDD (Hard Disk Drive), SSD (Solid State Drive), EEPROM (Electrically Erasable Programmable Read-Only Memory), ROM (Read-Only Memory), and RAM (Random Access Memory).

[0084] External interfaces include, for example, digital input / output ports such as USB (Universal Serial Bus) and Ethernet® ports.

[0085] Examples of input devices include keyboards, mice, touch panels, and touchpads. Examples of display devices include liquid crystal display panels and organic EL (Electro-Luminescence) display panels.

[0086] 1.2.1. Configuration of the pre-processing unit The pre-processing unit 53 shown in Figure 10 includes a first bandpass filter 534, a second bandpass filter 535, a first delay adjuster 536, a multiplier 538, a third bandpass filter 539, a first AGC unit 540, a second AGC unit 541, and an adder 542. AGC stands for Auto Gain Control.

[0087] The photodetection signal output from the current-voltage converter 531 is split into two signals, a first signal S1 and a second signal S2, at the branching section jp1. In Figure 10, the path of the first signal S1 is designated as the first signal path ps1, and the path of the second signal S2 is designated as the second signal path ps2.

[0088] The first bandpass filter 534, the second bandpass filter 535, and the third bandpass filter 539 are filters that selectively transmit signals in a specific frequency band.

[0089] The first delay adjuster 536 is a circuit that adjusts the delay of a signal by using a memory that temporarily stores the signal. The multiplier 538 is a circuit that generates an output signal proportional to the product of two input signals. The adder 542 is a circuit that generates an output signal proportional to the sum of two input signals.

[0090] Next, the operation of the preprocessor 53 will be explained in accordance with the flow of the first signal S1 and the second signal S2.

[0091] The first signal S1 is passed through a first bandpass filter 534 located on the first signal path ps1, and then its group delay is adjusted by a first delay adjuster 536. The group delay adjusted by the first delay adjuster 536 corresponds to the group delay of the second signal S2 by the second bandpass filter 535, which will be described later. This delay adjustment makes it possible to equalize the delay time between the first bandpass filter 534 through which the first signal S1 passes, and the second bandpass filter 535 and third bandpass filter 539 through which the second signal S2 passes. The first signal S1, having passed through the first delay adjuster 536, is input to the adder 542 via the first AGC unit 540.

[0092] The second signal S2 is passed through a second bandpass filter 535 located on the second signal path ps2, and then input to a multiplier 538. In the multiplier 538, the cosine wave signal cos(θ) output from the orthogonal signal generator 57 is applied to the second signal S2. m (t)) is multiplied. Then the second signal S2 is passed through the third bandpass filter 539, then through the second AGC unit 541, and then input to the adder 542. The adder 542 outputs a signal that is proportional to the sum of the first signal S1 and the second signal S2.

[0093] 1.2.2. Pre-processing Next, the preprocessing in the preprocessing unit 53 will be explained. Preprocessing refers to the process of extracting frequency modulation components from the photodetection signal. In the following explanation, as an example, we consider a system in which the frequency of the modulation signal changes sinusoidally, and the displacement of the object 14 also changes in simple harmonic motion in the optical axis direction. Here, E m , E d φ,

[0094]

number

[0095] In this case, the photodetection signal I output from the current-voltage converter 531 PD Theoretically, it can be expressed by the following equation.

[0096]

number

[0097] Note E m , E d , φ m , φ d ,φ,ω m , ω d ,ω0,a m a d These are as follows:

[0098]

number

[0099] Furthermore, the <> in equation (4) represents the time average. The first and second terms of equation (4) above represent the DC component, and the third term represents the AC component. This AC component is I PD·AC Therefore, I PD·AC The equation is as follows:

[0100]

number

[0101] Here, the ν-th order Bessel functions shown in equations (8) and (9) below are known.

[0102]

number

[0103] Expanding equation (5) above into a series using the Bessel functions in equations (8) and (9) above, it can be transformed into equation (10) below.

[0104]

number

[0105] However, J0(B), J1(B), J2(B), ... are Bessel coefficients, respectively.

[0106] As shown above, theoretically, it is possible to extract a frequency band corresponding to a specific order using a bandpass filter.

[0107] Therefore, the aforementioned preprocessing unit 53 performs preprocessing on the AC component of the photodetection signal according to the following flow, based on this theory.

[0108] First, the AC component of the photodetection signal output from the current-voltage converter 531 is normalized in amplitude by the ADC 532. The signal after passing through the ADC 532 is expressed by the following equation (10-1).

[0109]

number

[0110] The signal after passing through ADC532 is then split into two signals, the first signal S1 and the second signal S2, at branching point jp1. The first signal S1 is passed through the first bandpass filter 534. The first bandpass filter 534 has a center angular frequency of ω mIt is set to this. As a result, the first signal S1 after passing through the first bandpass filter 534 is expressed by the following equation.

[0111]

number

[0112] Meanwhile, the second signal S2 is passed through the second bandpass filter 535. The center angle frequency of the second bandpass filter 535 is set to a different value from the center angle frequency of the first bandpass filter 534. Here, as an example, the center angle frequency of the second bandpass filter 535 is set to 2ω m This setting is used. As a result, the second signal S2 after passing through the second bandpass filter 535 is expressed by the following equation.

[0113]

number

[0114] The second signal S2 after passing through the second bandpass filter 535 is multiplied by the multiplier 538, which receives the cosine wave signal cos(θ) output from the quadrature signal generator 57, which will be described later. m (t)) is multiplied. The second signal S2 after passing through multiplier 538 is expressed by the following equation.

[0115]

number

[0116] In equation (13) above, α is the phase difference, or deviation, when the phase of the reference signal Ss is shifted from its original phase. The original phase is the phase when the pre-processed signal S(t) output from the pre-processing unit 53 is a frequency-modulated signal only, or a signal equivalent to that.

[0117] The second signal S2, after passing through the multiplier 538, is passed through the third bandpass filter 539. The center angle frequency of the third bandpass filter 539 is set to the same value as the center angle frequency of the first bandpass filter 534. Here, as an example, the center angle frequency of the third bandpass filter 539 is set to ω m This setting is used. As a result, the second signal S2 after passing through the third bandpass filter 539 is expressed by the following equation.

[0118]

number

[0119] Subsequently, the first signal S1 represented by the above equation (11) has its phase adjusted by the first delay adjuster 536 and its amplitude adjusted by the first AGC unit 540.

[0120] Furthermore, the amplitude of the second signal S2, represented by equation (14) above, is also adjusted by the second AGC unit 541 so that the amplitude of the second signal S2 matches the amplitude of the first signal S1. The first signal S1 after amplitude adjustment is represented by the following equation (14-1), and the second signal S2 after amplitude adjustment is represented by the following equation (14-2).

[0121]

number

[0122] The first signal S1 and the second signal S2 are then added together by the adder 542. The result of the addition is denoted as the preprocessing signal S(t). The preprocessing signal S(t) is expressed by the following equation (15).

[0123]

number

[0124] As shown in equation (15) above, the preprocessing signal S(t) is expressed by an equation that includes a phase difference α. Therefore, when the phase difference α is an integer multiple of π, equation (15) above can be expressed as equation (15-1) below.

[0125]

number

[0126] In this specification, when the preprocessed signal S(t) is expressed by equation (15-1) above, it is said that the phases are aligned. On the other hand, when the preprocessed signal S(t) is expressed by (15) above and the phase difference α is not an integer multiple of π, it is said that the phases are not aligned.

[0127] When the phases are aligned, the pre-processed signal S(t) can be expressed by equation (15-1) above, and can therefore be said to be a signal consisting only of frequency modulation. In such a pre-processed signal S(t), the frequency modulation component is extracted, so the demodulation accuracy of the sampled signal can be improved in the demodulation processing unit 55. On the other hand, when the phases are not aligned, the pre-processed signal S(t) can be said to be a signal in which frequency modulation and amplitude modulation are superimposed. In such a pre-processed signal S(t), it is difficult to improve the demodulation accuracy of the sampled signal in the demodulation processing unit 55.

[0128] 1.2.3. Configuration of the orthogonal signal generation unit The orthogonal signal generator 57 shown in Figure 10 includes a fourth bandpass filter 571, a Hilbert transform filter 572, a second delay adjuster 573 (reference signal delay), a reference signal phase calculator 574, an absolute value calculator 577, a third lowpass filter 578, a phase amount setting unit 579, an adder 580, a cosine generator 581, and a sine generator 582.

[0129] In this embodiment, based on the phase of the reference signal Ss and the amplitude of the pre-processed signal S(t), the orthogonal signal generation unit 57 generates a cosine wave signal cos(θ) which has a waveform that is orthogonal to each other. m (t)) and sinusoidal signal sin(θ m (t)) is generated. In this specification, the process of generating such orthogonal waveforms is referred to as the "orthogonal waveform generation process".

[0130] Between the signal generation unit 61 and the orthogonal signal generation unit 57, the ADC 533 shown in Figure 1 is positioned. The ADC 533 is an analog-to-digital converter that converts an analog signal to a digital signal with a predetermined number of sampling bits. The fourth bandpass filter 571 is a filter that selectively transmits signals in a specific frequency band.

[0131] The Hilbert transform filter 572 performs a Hilbert transform on the reference signal Ss to obtain signal i. The reference signal Ss output from the voltage-controlled oscillator 612 is cos(ω m It is represented as t). m is the angular frequency of the modulation signal from the optical modulator 12, and t is time. The Hilbert transform process shifts the phase of this reference signal Ss by π / 2.

[0132] The second delay adjuster 573 is a circuit that adjusts the signal delay using a memory that temporarily stores the signal, and it introduces a delay in the reference signal Ss that is equivalent to the delay caused by the Hilbert transform process. This is how the signal r is obtained.

[0133] The reference signal phase calculator 574 calculates the phase of the reference signal Ss from the signal i output from the Hilbert transform filter 572 and the signal r output from the second delay adjuster 573. Specifically, it performs an arctangent operation, i.e., atan(i / r), on the ratio of signal i to signal r.

[0134] The absolute value calculator 577 calculates the absolute value of the pre-processed signal S(t) output from the pre-processing unit 53. The third low-pass filter 578 is a filter that cuts out high-frequency signals from the absolute value of the pre-processed signal S(t) output from the absolute value calculator 577.

[0135] The phase amount setting unit 579 has the function of acquiring the envelope of the signal output from the third low-pass filter 578, the function of acquiring the maximum and minimum values ​​(amplitude of the envelope) of the envelope, and the function of outputting the phase amount a.

[0136] The adder 580 outputs a signal proportional to the sum of the output from the reference signal phase calculator 574 and the output from the phase amount setting unit 579. The cosine generator 581 generates a cosine wave signal cos(θ) based on the signal output from the adder 580. m The sine generator 582 generates a sinusoidal signal sin(θ) based on the signal output from the adder 580. m (t)) is generated.

[0137] 1.2.4. Orthogonal waveform generation process In the quadrature waveform generation process, the reference signal Ss is first input to the fourth bandpass filter 571. The fourth bandpass filter 571 has a center angular frequency of ω m It is set to this. The reference signal Ss output from the fourth bandpass filter 571 is split into two parts; one is input to the Hilbert transform filter 572, and the other is input to the second delay adjuster 573.

[0138] The Hilbert transform filter 572 shifts the phase of the reference signal Ss by π / 2 to generate signal i. The second delay adjuster 573 delays the reference signal Ss to generate signal r. Signals i and r are input to the reference signal phase calculator 574.

[0139] The reference signal phase calculator 574 performs the atan(i / r) operation to obtain the phase of the reference signal Ss. The phase of the reference signal Ss is input to the adder 580.

[0140] The absolute value calculator 577 obtains the absolute value of the preprocessing signal S(t). This allows the negative waveform of the preprocessing signal S(t) to be converted to the positive waveform and combined. The signal from the absolute value calculator 577 is input to the third low-pass filter 578.

[0141] The third low-pass filter 578 cuts out signals in the high-frequency band. This makes it easy and highly accurate to acquire the envelope in the phase amount setting unit 579. The signal from the third low-pass filter 578 is input to the phase amount setting unit 579.

[0142] The phase amount setting unit 579 sets the phase amount a to be added to the calculation result atan(i / r) in the adder 580 based on the signal from the third low-pass filter 578. In other words, the orthogonal signal generation unit 57 adjusts the phase of the reference signal Ss. The setting method will be described later.

[0143] The adder 580 calculates the sum of the output from the reference signal phase calculator 574 and the output from the phase amount setting unit 579. Here, the sum is denoted as β, where β is a + atan(i / r). The cosine generator 581 then generates the cosine wave signal cos(θ m The sine generator 582 generates a sinusoidal signal sin(θ) m (t)) is generated. Cosine wave signal cos(θ m (t)) is output to the multiplier 538 and the demodulation processing unit 55 described later, and the sine wave signal sin(θ) m (t)) is output to the demodulation processing unit 55. m (t) is ω m It is t-β.

[0144] 1.2.5. Phase Amount Setting Method In the phase amount setting unit 579, when the phases are not aligned, the phase amount a added by the adder 580 is set so that the effect of the amplitude modulation described above is minimized. As a result, the cosine generator 581 and the sine generator 582 generate a cosine wave signal cos(θ) that takes the phase amount a into account. m (t)) and sinusoidal signal sin(θ m (t)) is generated. Then this cosine wave signal cos(θ) m (t) is reflected in the preprocessed signal S(t) via the multiplier 538, reducing the effect of amplitude modulation in the preprocessed signal S(t). Ultimately, the phases can be aligned. Also, the cosine wave signal cos(θ) when the phases are aligned m (t)) and sinusoidal signal sin(θ m When (t)) is input to the demodulation processing unit 55, demodulation processing from the pre-processed signal S(t) can be performed with high accuracy.

[0145] Figure 11 is a flowchart illustrating an example of how the phase amount setting unit 579 shown in Figure 10 sets the phase amount. The phase amount setting shown in Figure 11 is preferably performed using a standard sample vibrating at a single frequency as the object 14. This allows the phase amount setting unit 579 to more accurately determine the phase amount a to be added in the adder 580. In the phase amount setting method shown in Figure 11, the amplitude of the preprocessing signal S(t) is repeatedly evaluated while gradually changing the value of the phase amount a. The phase amount a at which the amplitude falls below a predetermined value is then stored in memory as the optimal value. After determining the optimal phase amount a, the phase amount a can be fixed.

[0146] Therefore, the phase quantity setting method shown in Figure 11 is performed using the aforementioned standard sample, for example, before measuring the object 14 with the laser interferometer 1. This allows the laser interferometer 1 to be automatically calibrated. Examples of standard samples include piezoelectric elements and quartz oscillators.

[0147] In step S102 shown in Figure 11, the sign function sgn and phase quantity a are first initialized. Specifically, the value 1 is input to the sign function sgn, and the value a0 is input to the phase quantity a. The value a0 can be any value.

[0148] In step S104, the envelope of the pre-processed signal S(t) is obtained via the absolute value calculator 577 and the third low-pass filter 578. Then, in step S104, the maximum and minimum values ​​of the envelope are obtained, and the difference dS0 between the maximum and minimum values ​​is stored in memory. This difference dS0 corresponds to the amplitude of the pre-processed signal S(t).

[0149] In step S106, the phase amount a is updated in the phase amount setting unit 579 and set as an output value by the equation a + sgn * Δa → a. This equation means that a small amount Δa is added to or subtracted from the current value of the phase amount a based on two possible values ​​of the sign function sgn, namely 1 or -1, to obtain a new phase amount a. The small amount Δa is not particularly limited as long as it is a quantity smaller than the phase amount a. The updated phase amount a is output to the adder 580. Then, the orthogonal signal generation unit 57 changes the phase of the reference signal Ss based on the new phase amount a set by the phase amount setting unit 579, and generates a cosine wave signal cos(θ) based on the updated reference signal Ss. m (t)) and sinusoidal signal sin(θ m (t)) is generated. Then, in the preprocessing unit 53, the cosine wave signal cos(θ) m Based on (t), a new preprocessed signal S(t) is generated.

[0150] In step S108, the difference between the maximum and minimum values ​​of the envelope of the new pre-processed signal S(t) is obtained in the same manner as in step S104 and stored in memory. In step S108, this difference is denoted as dS1.

[0151] In step S110, it is determined whether the difference dS1 is less than or equal to a predetermined value. The predetermined value is, for example, the difference between the maximum and minimum values ​​of the envelope when the pre-processed signal S(t) can be considered as a signal with only frequency modulation. Therefore, if the difference dS1 is less than or equal to the predetermined value, it can be determined that the current phase amount a is optimal. Thus, this flow is terminated. On the other hand, if the difference dS1 is greater than the predetermined value, it can be determined that the current phase amount a is not optimal, and the process proceeds to step S112.

[0152] In step S112, the number of times the judgment in step S110 was performed is obtained. Then, it is determined whether the obtained number is equal to or greater than a predetermined number. The predetermined number is, for example, the actual number of repetitions when the optimal phase amount a can be found when the amplitude of the pre-processed signal S(t) is repeatedly evaluated while gradually changing the value of the phase amount a, as described above. In particular, the maximum value of that actual value can be adopted as the predetermined number. Specific examples of the number of repetitions include the number of times the phase amount a was updated.

[0153] If the number of acquired values ​​exceeds a predetermined number, it can be determined that it is difficult to determine the optimal phase quantity a, and the process proceeds to step S114. In step S114, the phase quantity setting unit 579 issues an error. In this case, the aforementioned display unit should be shown that an error has occurred, prompting the user to take action, such as changing the standard sample. After issuing the error, the flow is terminated. On the other hand, if the number of acquired values ​​is less than a predetermined number, the process proceeds to step S116.

[0154] In step S116, it is determined whether difference dS1 < difference dS0 holds true. If it does, it can be determined that the effect of amplitude modulation has been reduced in the new preprocessed signal S(t). It can also be determined that the value of the sign function sgn initialized in step S102 was appropriate. In this case, the process proceeds to step S118. In step S118, the current value of difference dS1 is input to difference dS0. Then, the process returns to step S106. In the second step S106, the phase quantity a is updated again using the equation a + sgn * Δa → a. The sign function sgn in this equation remains unchanged because it has been determined to be appropriate. Then, the preprocessing unit 53 generates a new preprocessed signal S(t) that reflects the updated phase quantity a.

[0155] In the second step S108, the difference between the maximum and minimum values ​​of the envelope of the new pre-processed signal S(t) is obtained and stored in memory. The difference dS1 obtained in the second step S108 is smaller than the difference dS1 obtained in the first step S108. Therefore, the effect of amplitude modulation on the new pre-processed signal S(t) is further reduced, and it approaches a signal that is frequency-modulated only.

[0156] On the other hand, in step S116, if the condition dS1 < dS0 does not hold, that is, if dS1 ≥ dS0, then it can be determined that no reduction in the amplitude modulation effect was observed in the new pre-processed signal S(t). Furthermore, it can be determined that the value of the sign function sgn initialized in step S102 was inappropriate. In this case, the process proceeds to step S120.

[0157] In step S120, the sign function sgn is inverted from the current value. That is, the value obtained by multiplying the current value by -1 is input to the new sign function sgn. Then, the process proceeds to step S118.

[0158] According to the above flow, unless an error is issued, the phase amount a is updated, the pre-processing signal S(t) is updated, and the difference dS0 is updated repeatedly until the difference dS1 falls below a predetermined value. As a result, the phase amount a is adjusted in the phase amount setting unit 579 until the pre-processing signal S(t) can be considered as a signal with only frequency modulation, and the cosine wave signal cos(θ) is adjusted based on the phase-adjusted reference signal Ss. m (t)) and sinusoidal signal sin(θ m (t)) will be generated.

[0159] The cosine wave signal cos(θ) is generated in this way. mBy inputting (t)) to the preprocessing unit 53, the preprocessed signal S(t) can be brought closer to its original phase. As a result, the laser interferometer 1 can be calibrated. After calibration, the object 14 can be measured using the obtained optimal phase amount a. This improves the accuracy of demodulating the sample signal from the photodetection signal, and enables high-precision measurement of the object 14. The above calibration of the laser interferometer 1 can be performed at any timing and frequency.

[0160] The orthogonal signal generation unit 57 and the phase amount setting method have been described above. However, if the phases of the pre-processed signal S(t) are already aligned and calibration is unnecessary, the orthogonal signal generation unit 57 may be omitted. In that case, the reference signal Ss and a signal obtained by shifting its phase by π / 2 can be used as the orthogonal signal.

[0161] 1.2.6. Configuration of the demodulation processing unit The demodulation processing unit 55 performs demodulation processing to demodulate the sample signal originating from the object 14 from the preprocessed signal S(t). The demodulation processing is not particularly limited, but a known method is quadrature detection. The quadrature detection method is a method of demodulation processing that involves mixing mutually orthogonal signals from an external source with the input signal.

[0162] The demodulation processing unit 55 shown in Figure 10 is a digital circuit comprising a multiplier 551, a multiplier 552, an inverting amplifier 553, a first low-pass filter 555, a second low-pass filter 556, a divider 557, and an inverse tangent 558.

[0163] Multipliers 551 and 552 are circuits that generate an output signal proportional to the product of two input signals. Inverting amplifier 553 is a circuit that generates an output signal with a gain of -1, without changing the amplitude, and with the phase of the input signal inverted. First low-pass filter 555 and second low-pass filter 556 are filters that cut out signals in the high-frequency band, respectively.

[0164] The divider 557 is a circuit that generates an output signal proportional to the quotient of two input signals. The arctangent arithmetic unit 558 is a circuit that outputs the arctangent of the input signals. The output signal from the arctangent arithmetic unit 558 is input to the signal output unit 559.

[0165] The signal output unit 559 takes the phase φ obtained by the arctangent calculator 558 and converts it to information originating from the object 14, including the phase φ d The signal output unit 559 performs phase unwrapping to connect phases when there is a 2π phase gap between two adjacent points. Then, the displacement of the object 14 is calculated from the obtained phase information. This realizes a displacement meter. Furthermore, the velocity of the object 14 can be determined from the displacement. This realizes a speedometer.

[0166] Furthermore, the demodulation processing unit 55 is not limited to a digital circuit, but may also be an analog circuit. The analog circuit may include an F / V converter circuit or a ΔΣ counter circuit.

[0167] Furthermore, the signal output unit 559 described above may be configured to calculate frequency information originating from the object 14. Based on the frequency information, the velocity of the object 14 can be calculated.

[0168] 1.2.7. Demodulation Process In the demodulation process, the pre-processed signal S(t) is first split into two at the branching section jp2. One of the resulting signals is then multiplied by the multiplier 551, which multiplies the sine wave signal sin(θ) output from the sine generator 582. m The (t) is multiplied via the inverting amplifier 553. In other words, the multiplier 551 multiplies the preprocessed signal S(t) by the sinusoidal signal sin(θ). m The signal obtained by inverting the phase of (t) -sin(θ m (t)) is mixed. For the other signal after splitting, the cosine wave signal cos(θ) output from the cosine generator 581 is multiplied in the multiplier 552. m Multiply the preprocessed signal S(t) by the cosine wave signal cos(θ). In other words, the multiplier 552 multiplies the preprocessed signal S(t) by the cosine wave signal cos(θ). m Mix (t)).

[0169] The signal that has passed through the multiplier 551 is then passed through the first low-pass filter 555 and input to the divider 557 as signal y. The signal that has passed through the multiplier 552 is then passed through the second low-pass filter 556 and input to the divider 557 as signal x. In this specification, signals x and y together are referred to as the "mixed signal". The divider 557 performs a division of signal y by signal x, and passes its output y / x through the inverse tangent arithmetic unit 558 to obtain the calculation result atan(y / x). This allows the phase information of the sampled signal to be obtained.

[0170] Subsequently, by inputting the calculation result atan(y / x) to the signal output unit 559, the displacement and velocity of the object 14 are output.

[0171] Figure 12 shows an example of waveforms representing the pre-processed signal S(t) before calibration of the laser interferometer 1 based on the flow shown in Figure 11, the phase information demodulated by the demodulation processing unit 55 (phase information before unwrapping), and the phase information unwrapped by the signal output unit 559 (phase information after unwrapping).

[0172] As shown in Figure 12, the waveform of the pre-processed signal S(t) before calibration shows a large change in amplitude, resulting in a signal where frequency modulation and amplitude modulation are superimposed. Consequently, the amplitude of the envelope of the pre-processed signal S(t) also increases. Therefore, the phase information before unwrapping shown in Figure 12 exhibits irregular waveforms at the locations indicated by the arrows, and the waveform of the phase information after unwrapping is discontinuous. In this case, it is difficult to accurately determine the displacement of the object 14 from the phase information after unwrapping.

[0173] Figure 13 shows an example of waveforms representing the pre-processed signal S(t) after calibration of the laser interferometer 1 based on the flow shown in Figure 11, the phase information demodulated by the demodulation processing unit 55 (phase information before unwrapping), and the phase information unwrapped by the signal output unit 559 (phase information after unwrapping).

[0174] As shown in Figure 13, the waveform of the pre-processed signal S(t) after calibration shows little change in amplitude and is almost entirely frequency-modulated. Therefore, the amplitude of the envelope of the pre-processed signal S(t) is also small. Consequently, the phase information before unwrapping shown in Figure 13 does not have an irregular waveform. Furthermore, the waveform of the phase information after unwrapping shown in Figure 13 is a continuous waveform. In this case, the displacement of the object 14 can be accurately determined from the phase information after unwrapping.

[0175] 1.2.8. Correction Processing Unit As shown in Figure 1, the correction processing unit 62 receives the reference signal Ss output from the signal generation unit 61 and the output signal Sm output in response to the driving of the optical modulator 12. The correction processing unit 62 also outputs a frequency control signal Sf1 (correction signal) to the voltage-controlled oscillator 612. Furthermore, the correction processing unit 62 outputs an amplification factor control signal Sam (correction signal) to the amplifier 614.

[0176] The correction processing unit 62 is implemented, for example, on an FPGA, and is preferably located in the sensor head unit 51. This allows the physical distance between the correction processing unit 62 and the optical modulator 12 to be shortened, thereby suppressing, for example, a decrease in the signal-to-noise ratio of the output signal Sm due to the influence of electromagnetic noise.

[0177] Figure 14 is a diagram that shows in detail the correction processing unit 62, which is part of the functional block diagram shown in Figure 1.

[0178] The output signal Sm from the optical modulator 12 is input to the first offset removal unit 631 shown in Figure 14. The first offset removal unit 631 has the function of removing the DC (direct current) component and extracting the AC (alternating current) component. The output signal Sm that has passed through the first offset removal unit 631 is input to the correction processing unit 62.

[0179] The reference signal Ss from the voltage-controlled oscillator 612 is input to the second offset removal unit 632 shown in Figure 14. The second offset removal unit 632 has the function of removing the DC (direct current) component and extracting the AC (alternating current) component. The reference signal Ss that has passed through the second offset removal unit 632 is input to the correction processing unit 62 and the ADC 533.

[0180] The correction processing unit 62 shown in Figure 14 includes an absolute value calculator 621, a multiplier 622, a multiplier 623, a fourth low-pass filter 624, a fifth low-pass filter 625, an amplitude gain setting unit 626, and a frequency setting unit 627.

[0181] The absolute value calculator 621 calculates the absolute value of the output signal Sm that has passed through the first offset removal unit 631.

[0182] Multipliers 622 and 623 are circuits that output a signal proportional to the product of two input signals. In multiplier 622, both input signals are output signals Sm. Therefore, multiplier 622 outputs a signal proportional to the square of output signal Sm. On the other hand, in multiplier 623, the two input signals are output signal Sm and reference signal Ss. Therefore, multiplier 623 outputs a signal proportional to the product of output signal Sm and reference signal Ss.

[0183] The multipliers 622 and 623 may use elements such as Gilbert cells, or they may be circuits that perform logarithmic transformation on the two input signals using an operational amplifier, perform addition and subtraction, and then perform inverse logarithmic transformation.

[0184] The fourth low-pass filter 624 and the fifth low-pass filter 625 are filters that cut out high-frequency signals from the input signal. The transmission frequency band of the fourth low-pass filter 624 and the fifth low-pass filter 625 should be a band that can remove frequencies of twice the frequency of the drive signal Sd or more, and preferably a band that can remove frequencies above the frequency of the drive signal Sd.

[0185] The signal output from the multiplier 622 and passed through the fourth low-pass filter 624 becomes a signal with a value corresponding to the amplitude of the output signal Sm, as will be described later. The amplitude gain setting unit 626 has the function of determining the amplitude (target amplitude) that should be set for the drive signal Sd based on this signal. The amplitude gain setting unit 626 then controls the gain (amplification factor) to be set for the amplifier 614 of the signal generation unit 61 so that the amplitude of the drive signal Sd becomes the target amplitude. Examples of control logic include feedback control such as PI control and PID control. The amplitude gain setting unit 626 outputs an amplification factor control signal Sam corresponding to the gain to be set to the amplifier 614.

[0186] Amplifier 614 amplifies the amplitude of the drive signal Sd based on the amplification factor control signal Sam. This corrects the amplitude of the drive signal Sd.

[0187] The signal output from the multiplier 623 and input via the fifth low-pass filter 625 is a signal with a value corresponding to the phase difference between the output signal Sm and the reference signal Ss, as will be described later. Here, the phase of the output signal Sm corresponds to the phase of the drive signal Sd. Also, the phase of the drive signal Sd corresponds to the phase of the reference signal Ss. Therefore, the frequency setting unit 627 has the function of determining the frequency (target frequency) to be set for the reference signal Ss. The frequency setting unit 627 then controls the voltage to be set for the voltage-controlled oscillator 612 of the signal generation unit 61 so that the frequency of the reference signal Ss becomes the target frequency. Examples of control logic include feedback control such as PI control and PID control. The frequency setting unit 627 outputs a frequency control signal Sf1 corresponding to the frequency to be set toward the voltage-controlled oscillator 612.

[0188] The voltage-controlled oscillator 612 generates a reference signal Ss with a frequency corresponding to the frequency control signal Sf1. This corrects the frequency of the reference signal Ss. In addition, this also corrects the frequency of the drive signal Sd.

[0189] 1.2.9. Acquisition of output signal from optical modulator Figure 15 shows an example of a circuit for acquiring the output signal Sm from the optical modulator 12.

[0190] The output signal Sm may be a signal obtained by detecting the current flowing through the vibrating element 30 of the optical modulator 12, or a signal obtained by detecting the voltage applied to the vibrating element 30. For example, if the output signal Sm is obtained by detecting the current flowing through the vibrating element 30, the current value flowing through the vibrating element 30 is detected using a current shunt monitor 64, as shown in Figure 15. The current shunt monitor 64 shown in Figure 15 includes a shunt resistor 641 and an operational amplifier 642, and detects the current value flowing through the vibrating element 30 by converting it into a voltage value. This gives the output signal Sm, which is a voltage signal. The obtained output signal Sm is converted into a digital signal by the ADC 643 and output to the first offset removal unit 631.

[0191] In addition to the above method, other methods for detecting the current flowing through the vibrating element 30 include using a Hall element and winding a coil around the current path to detect the electromotive force.

[0192] 1.2.9. Correction Process Next, the correction process in the correction processing unit 62 will be explained. The correction process refers to correcting the drive signal Sd and the reference signal Ss by changing the setting values ​​of the voltage-controlled oscillator 612 and the amplifier 614 based on the correction signal output from the correction processing unit 62.

[0193] If the output signal Sm from the optical modulator 12 is, for example, a voltage signal, the output signal Sm before passing through the first offset removal unit 631 is expressed by the following equation (16).

[0194]

number

[0195] In the above equation (16), V QOM This is the voltage value of the output signal Sm. Also, A mis a coefficient corresponding to the amplitude of the output signal Sm, and α m1 is the phase difference of the output signal Sm with respect to the reference signal Ss, where -π / 2 < α m1 < π / 2 is satisfied. Further, O QOM is the DC component of the output signal Sm.

[0196] Then, the output signal Sm after passing through the first offset removal unit 631 is represented by the following formula (16-1).

[0197]

Equation

[0198] On the other hand, the reference signal Ss before passing through the second offset removal unit 632 is represented by the following formula (18).

[0199]

Equation

[0200] In the above formula (18), V OSC is the voltage value of the reference signal Ss. Also, v OSC is a coefficient corresponding to the amplitude of the reference signal Ss, and O OSC is the DC component of the reference signal Ss.

[0201] Then, the reference signal Ss after passing through the second offset removal unit 632 is represented by the following formula (18-1).

[0202]

Equation

[0203] The output signal Sm after passing through the first offset removal unit 631 is split into two. And one of the output signals Sm, after passing through the absolute value calculator 621 and then being squared by the multiplier 622, is represented by the following formula (16-2).

[0204] [Number]

[0205] After that, by passing through the fourth low-pass filter 624, only the first term on the right side of the above formula (16-2) is extracted. As a result, the output signal Sm after passing through the fourth low-pass filter 624 is represented by the following formula (16-3).

[0206] [Number]

[0207] As represented by the above formula (16-3), the input signal V input to the amplitude gain setting unit 626 QOM 2 becomes a signal without time variation. Therefore, in the amplitude gain setting unit 626, for the output signal Sm represented by the above formula (16-3), the target coefficient A m is used as the control target value for feedback control by substituting the value obtained by substituting it into the above formula (16-3). Then, an amplification rate control signal Sam corresponding to the control target value is output toward the amplifier 614 of the signal generation unit 61. As a result, the amplitude gain in the amplifier 614 can be changed, and the amplitude of the drive signal Sd can be corrected to the target amplitude.

[0208] The other output signal Sm divided into two is multiplied by the reference signal Ss by the multiplier 623. As a result, the signal output from the multiplier 623 is represented by the following formula (17-2).

[0209] [Number]

[0210] After that, by passing through the fifth low-pass filter 625, only the first term on the right side of the above formula (17-2) is extracted. As a result, the output signal Sm after passing through the fifth low-pass filter 625 is represented by the following formula (17-3).

[0211]

Number

[0212] As represented by the above formula (17-3), the input signal V input to the frequency setting unit 627 QOM ·V OSC has a coefficient Am, a coefficient v OSC and a phase difference α m1 in the right side. Among these, the coefficient v OSC is known. On the other hand, the coefficient A m satisfies 0 < A m and is controlled to converge to the target coefficient A m as described above. Therefore, the input signal V QOM ·V OSC also becomes a signal without time change. Thus, in the frequency setting unit 627, for example, the value obtained by substituting the target phase difference α m1 into the above formula (17-3) is used as the control target value for feedback control. Then, the frequency control signal Sf1 corresponding to the control target value is output toward the voltage controlled oscillator 612 of the signal generation unit 61. Thereby, the frequency of the reference signal Ss output from the voltage controlled oscillator 612 can be changed, and the frequency of the reference signal Ss can be corrected to the target frequency. Also, the frequency of the drive signal Sd can be corrected to the target frequency.

[0213] Note that the target phase difference αm1 can be determined based on, for example, the relationship between the phase difference between the drive signal Sd and the output signal Sm in the vibration element 30 vibrating at the mechanical resonance frequency. Specifically, in such a vibration element 30, it is known that the phase of the output signal Sm is delayed by about 90 [deg] with respect to the input drive signal Sd. Also, a phase delay δ [deg] may occur in the process until the output signal Sm is input to the correction processing unit 62. Considering these, the target phase difference αm1 can be, for example, 90 + δ [deg]. The phase delay δ can be obtained by experiments or simulations.

[0214] ed Furthermore, temperature changes and other factors can alter the mechanical resonance frequency and the efficiency with which the vibration element 30 converts input power into vibration. A change in this conversion efficiency will result in a change in the amplitude of vibration of the vibration element 30. Therefore, in the correction process, priority is given to correcting the frequency of the reference signal Ss and the frequency of the drive signal Sd. Then, if necessary, the amplitude of the drive signal Sd is corrected. By performing the correction process in this order, the aforementioned frequency and amplitude can be efficiently controlled to the desired values.

[0215] Furthermore, considering the control in the frequency setting unit 627 described above, it is desirable to converge the control of the signal input to the amplitude gain setting unit 626 faster than the control of the signal input to the frequency setting unit 627. This suppresses instability of the target control value in the frequency setting unit 627, thereby preventing instability in the correction process.

[0216] The amplitude gain setting unit 626 and the frequency setting unit 627 are constructed by combining operational amplifiers or the like to perform feedback control operations, such as PID control. In this case, in order to converge the control of the signal input to the amplitude gain setting unit 626 faster than the control of the signal input to the frequency setting unit 627, the crossover frequency of the open-loop transfer function of the control loop in the operation of the amplitude gain setting unit 626 should be set higher than the crossover frequency of the open-loop transfer function of the control loop in the operation of the frequency setting unit 627.

[0217] By performing the correction process described above, the following effects can be obtained. When the mechanical resonance frequency of the vibrating element 30 changes due to external disturbances such as ambient temperature changes, gravity changes, vibrations, and noise, the frequency and amplitude of the vibration of the vibrating element 30 change, causing a decrease in the signal-to-noise ratio of the modulated signal. This leads to a problem where the demodulation accuracy of the sampled signal decreases.

[0218] On the other hand, by performing the above correction process, even when a disturbance such as a temperature change is applied, the frequency and amplitude of the vibration of the vibration element 30 can be maintained constant respectively. That is, even when a disturbance such as a temperature change is applied, it is possible to correct so that the frequency and amplitude of the vibration of the vibration element 30 do not change. Thereby, a decrease in the S / N ratio of the modulation signal can be suppressed. As a result, even when a disturbance such as a temperature change is applied, the accuracy of the preprocessing and demodulation processing in the arithmetic unit 52 can be improved, and a decrease in the demodulation accuracy of the sample signal can be suppressed.

[0219] Also, even when a disturbance such as a temperature change is applied and the mechanical resonance frequency changes, the frequency of the drive signal Sd can be made to follow, so that the vibration element 30 can be continuously driven in the vicinity of the mechanical resonance frequency of the vibration element 30. Thereby, since the drive efficiency of the vibration element 30 is increased, power consumption reduction of the laser interferometer 1 can be achieved. When the vibration element 30 is driven by, for example, an oscillation circuit, it is difficult to drive in the vicinity of the mechanical resonance frequency of the vibration element 30. As reasons for this, there are various restrictions on the circuit configuration of the oscillation circuit, which is not realistic.

[0220] 1.2.10. Effects of the First Embodiment As described above, the laser interferometer 1 according to the present embodiment includes a laser light source 2, an optical modulator 12, a light receiving element 10, an arithmetic unit 52, and a signal generation unit 61. The laser light source 2 emits emitted light L1 (laser light). The optical modulator 12 includes a vibration element 30 that is driven by a drive signal Sd, and superimposes a modulation signal on the emitted light L1 using the vibration element 30. The light receiving element 10 receives object light L3 (laser light including a sample signal) including a sample signal superimposed by reflection on the object 14 and reference light L2 (laser light including a modulation signal), and outputs a received signal. The arithmetic unit 52 performs arithmetic operations on the received signal based on the reference signal Ss. The signal generation unit 61 outputs a drive signal Sd and a reference signal Ss.

[0221] Furthermore, the calculation unit 52 includes a preprocessing unit 53, a demodulation unit 55, and a correction unit 62. The preprocessing unit 53 performs preprocessing to extract frequency modulation components from the received signal and outputs a preprocessed signal S(t) that includes frequency modulation components. The demodulation unit 55 mixes the preprocessed signal S(t) with an orthogonal signal to obtain signals x and y (mixed signals), and then demodulates the sample signal from signals x and y. The correction unit 62 outputs a correction signal, which is an amplification control signal Sam and a frequency control signal Sf1, based on the output signal Sm that is output in response to the driving of the vibrating element 30. Then, the signal generation unit 61 corrects the drive signal Sd and the reference signal Ss based on the amplification control signal Sam and the frequency control signal Sf1.

[0222] In this configuration, even when disturbances such as temperature changes are applied, the frequency and amplitude of the drive signal Sd can be made to follow the changes in the mechanical resonance frequency and vibration amplitude of the vibrating element 30. This makes it possible to maintain a constant vibration frequency and amplitude of the vibrating element 30. As a result, a decrease in the S / N ratio of the modulated signal can be suppressed, and a decrease in the demodulation accuracy of the sample signal can be suppressed. This makes it possible to realize a laser interferometer 1 that can measure the displacement and velocity of the object 14 with high precision even when disturbances are applied. Furthermore, by using the signal generation unit 61 and the correction processing unit 62, the vibrating element 30 can be driven near its mechanical resonance frequency, thereby reducing the power consumption of the laser interferometer 1.

[0223] Furthermore, it is preferable that the correction processing unit 62 is configured to correct the frequency of the reference signal Ss and the frequency of the drive signal Sd based on the phase difference between the output signal Sm and the reference signal Ss.

[0224] With this configuration, the phase difference between the output signal Sm, which is output in response to the driving of the optical modulator 12, and the reference signal Ss can be fed back to the above frequency. The phase difference between the output signal Sm and the reference signal Ss directly reflects the effect of disturbances on the modulated signal, making it suitable as an input signal for feedback control. Therefore, with this configuration, the effects of disturbances can be corrected in real time, and a laser interferometer 1 with particularly high resistance to disturbances can be realized.

[0225] Furthermore, it is preferable that the correction processing unit 62 is configured to correct the amplitude of the drive signal Sd based on the amplitude of the output signal Sm.

[0226] With this configuration, the amplitude of the output signal Sm, which is output in response to the driving of the optical modulator 12, can be fed back into the amplitude of the driving signal Sd. Since the amplitude of the output signal Sm directly reflects the effect of disturbances on the modulation signal, it is suitable as an input signal used for feedback control. Therefore, with the above configuration, the effects of disturbances can be corrected in real time, and a laser interferometer 1 with particularly high resistance to disturbances can be realized.

[0227] Furthermore, the signal generation unit 61 includes a voltage-controlled oscillator 612 and an amplifier 614. The voltage-controlled oscillator 612 generates a reference signal Ss. The amplifier 614 adjusts the amplitude of the reference signal Ss and outputs it as a drive signal Sd.

[0228] With this configuration, the amplifier 614 generates a drive signal Sd based on the reference signal Ss generated by the voltage-controlled oscillator 612. This allows the signal generation unit 61 to correct the frequency of the reference signal Ss and the frequency of the drive signal Sd based on the frequency control signal Sf1. Furthermore, the amplitude of the drive signal Sd can be individually corrected based on the amplification factor control signal Sam.

[0229] Furthermore, in this embodiment, a voltage-controlled oscillator 612 is used as the oscillator in the signal generation unit 61. This allows the use of a general-purpose voltage-controlled oscillator 612, making it easy to reduce the cost of the signal generation unit 61.

[0230] Furthermore, as mentioned above, the vibrating element 30 is preferably a quartz crystal oscillator. This allows for the generation of a highly accurate modulation signal by utilizing the extremely high Q factor of the quartz crystal. As a result, the sample signal originating from the object 14 can be demodulated with higher accuracy. Examples of quartz crystal oscillators include AT quartz crystal oscillators, SC-cut quartz crystal oscillators, and tuning fork type quartz crystal oscillators.

[0231] Furthermore, in this embodiment, the orthogonal signal generation unit 57 generates the aforementioned orthogonal signal based on the phase of the reference signal Ss and the amplitude of the pre-processed signal S(t). In addition, the orthogonal signal generation unit 57 adjusts the phase of the reference signal Ss based on the amplitude of the pre-processed signal S(t). Then, the adjusted signal is a cosine wave signal cos(θ m By inputting (t)) to the preprocessing unit 53, the preprocessed signal S(t) can be made closer to a signal with only frequency modulation, thereby improving the accuracy of demodulating the sample signal originating from the object 14 from the received light signal.

[0232] With this configuration, even if the phase of the reference signal Ss is shifted from its original phase, it can be calibrated. This makes it possible to realize a laser interferometer 1 that can measure the displacement and velocity of the object 14 with high precision.

[0233] Furthermore, in this embodiment, the orthogonal signal generation unit 57 has a phase amount setting unit 579 that sets the phase of the orthogonal signal from the amplitude of the preprocessing signal S(t). As described above, the phase amount setting unit 579 has the function of setting the phase amount a to be added by the adder 580. Then, the orthogonal signal generation unit 57 adjusts the phase of the reference signal Ss based on the phase amount a and generates the orthogonal signal, i.e., the cosine wave signal cos(θ) m (t)) and sinusoidal signal sin(θ mA quadrature signal (t) is generated. This quadrature signal is mixed with the preprocessed signal S(t) in the demodulation processing unit 55. By appropriately setting the phase amount a, the phases of the preprocessed signal S(t) and the quadrature signal can be aligned. As a result, the demodulation processing unit 55 can perform demodulation processing from the preprocessed signal S(t) with high accuracy.

[0234] In this embodiment, the phase amount setting unit 579 sets the phase of the orthogonal signals such that the difference between the maximum amplitude of the pre-processed signal S(t) and the minimum amplitude of the pre-processed signal S(t) is less than or equal to a predetermined value.

[0235] This allows the phase amount setting unit 579 to efficiently find the optimal phase amount a so that the pre-processed signal S(t) is a signal consisting only of frequency modulation or a signal equivalent thereto.

[0236] Furthermore, in this embodiment, the orthogonal signal generation unit 57 includes a Hilbert transform filter 572, a second delay adjuster 573 (reference signal delay adjuster), and a reference signal phase calculator 574. The Hilbert transform filter 572 performs a Hilbert transform on the reference signal Ss to obtain signal i. The second delay adjuster 573 introduces a delay to the reference signal Ss to obtain signal r. The reference signal phase calculator 574 performs an arctangent operation on the ratio of signal i / signal r to obtain the phase of the reference signal Ss.

[0237] With this configuration, the phase of the reference signal Ss can be obtained instantaneously without sampling the reference signal Ss. Therefore, in the orthogonal signal generation unit 57 having such a phase amount setting unit 579, the phase of the reference signal Ss can be reflected in the orthogonal signal in real time.

[0238] 2. Second Embodiment Next, a laser interferometer according to the second embodiment will be described.

[0239] Figure 16 is a functional block diagram showing the laser interferometer 1a according to the second embodiment. Figure 17 is a detailed diagram showing the calculation unit 52 and the signal generation unit 61a from the functional block diagram of Figure 16.

[0240] The second embodiment will be described below, focusing on the differences from the first embodiment, and similar matters will be omitted from the description. In Figures 16 and 17, components similar to those in the first embodiment are denoted by the same reference numerals.

[0241] This embodiment is the same as the first embodiment, except that the configuration of the signal generation unit 61a and the correction processing unit 62a is different.

[0242] As shown in Figure 16, the signal generation unit 61a includes a numerically controlled oscillator 616, a DAC 615, and an amplifier 614.

[0243] As shown in Figure 17, the correction processing unit 62a includes multipliers 661 and 662, a sixth low-pass filter 663, a seventh low-pass filter 664, an amplitude phase calculation unit 665, a frequency setting unit 627, and an amplitude gain setting unit 626.

[0244] 2.1. Signal Generation Unit As shown in Figure 16, the signal generation unit 61a includes a numerically controlled oscillator 616, a DAC 615, and an amplifier 614.

[0245] The numerically controlled oscillator 616 generates periodic signals such as sine waves and cosine waves by reading address data, which is added at regular clock intervals, from a ROM table that stores the numerical values ​​for one period of a sine wave or cosine wave. This allows the numerically controlled oscillator 616 to generate a reference signal Ss of the desired frequency with high precision and output it to the DAC 615. The DAC 615 is a digital-to-analog converter that generates an analog reference signal Ss based on the input digital reference signal Ss.

[0246] The amplifier 614 amplifies the input reference signal Ss, generates a drive signal Sd of the desired amplitude, and outputs it to the optical modulator 12.

[0247] The numerically controlled oscillator 616 includes a cumulative adder 651, an absolute value calculator 577, a third low-pass filter 578, a phase amount setting unit 579, an adder 652, a first period signal generator 653, and a second period signal generator 654.

[0248] The cumulative adder 651 cumulatively adds the frequency control signal Sf2 output from the frequency setting unit 627 of the correction processing unit 62a. The frequency control signal Sf2, as will be described later, is the amount of phase advance per unit time step corresponding to the frequency to be set in the reference signal Ss. The cumulative adder 651 accumulates and adds these phase advance amounts to calculate the cumulative sum value. The obtained cumulative sum value is output to the first period signal generator 653.

[0249] The first period signal generator 653 includes a ROM (Read Only Memory) that stores the numerical values ​​for one period of sine and cosine waves. The first period signal generator 653 reads out the numerical value at the address corresponding to the cumulative sum. This makes it possible to generate sine and cosine wave signals with frequencies corresponding to the frequency control signal Sf2. The cosine wave signal is output as a reference signal Ss to the DAC 615 of the signal generation unit 61a and the multiplier 661 of the correction processing unit 62a, respectively. The sine wave signal is output as a reference signal Ss' to the multiplier 662 of the correction processing unit 62a.

[0250] As described above, the phase amount setting unit 579 sets the phase amount a to be added to the cumulative sum in the adder 652. The adder 652 calculates the sum of the cumulative sum and the phase amount a. The obtained sum of the cumulative sum and the phase amount a is output to the second period signal generator 654.

[0251] The second period signal generator 654 includes a ROM (Read Only Memory) that stores the numerical values ​​for one period of sine and cosine waves. The second period signal generator 654 reads out the numerical value at the address corresponding to the sum of the cumulative sum and the phase amount a. As a result, a sine wave signal sin(θ) is generated with a phase offset of the phase amount a added at a frequency corresponding to the frequency control signal Sf2. m (t) and the cosine wave signal cos(θ) m (t)) can be generated. Cosine wave signal cos(θ m (t)) is output to the multiplier 538 and the demodulation processing unit 55 described later, and the sine wave signal sin(θ) m (t)) is output to the demodulation processing unit 55.

[0252] The above describes an example configuration of the numerically controlled oscillator 616, but the configuration of the numerically controlled oscillator 616 is not limited to the above.

[0253] 2.2. Correction Processing Unit As shown in Figure 16, the correction processing unit 62a receives the output signal Sm, which is output in response to the driving of the optical modulator 12. The correction processing unit 62a acquires the phase difference between the output signal Sm and the reference signal Ss, and the amplitude of the output signal Sm, by quadrature detection.

[0254] Furthermore, the correction processing unit 62a has the function of outputting a frequency control signal Sf2 (correction signal) to the numerical control oscillator 616, and the function of outputting an amplification factor control signal Sam (correction signal) to the amplifier 614.

[0255] The output signal Sm from the optical modulator 12 is input to the correction processing unit 62a via the ADC 543 shown in Figure 16. The ADC 543 is an analog-to-digital converter. The output signal Sm, converted to a digital signal by the ADC 543, is split into two, as shown in Figure 17. One of the output signals Sm is multiplied by the reference signal Ss in the multiplier 661. The signal output from the multiplier 661 passes through the sixth low-pass filter 663 and is input to the amplitude-phase calculation unit 665 as signal I. The other output signal Sm is multiplied by the reference signal Ss' in the multiplier 662. The signal output from the multiplier 662 passes through the seventh low-pass filter 664 and is input to the amplitude-phase calculation unit 665 as signal Q.

[0256] The transmission frequency bands of the sixth low-pass filter 663 and the seventh low-pass filter 664 are preferably such that they can remove frequencies above the frequency of the drive signal Sd.

[0257] The amplitude-phase calculation unit 665 performs the atan(Q / I) calculation to calculate the phase of the output signal Sm. The amplitude-phase calculation unit 665 outputs the phase difference between the output signal Sm and the reference signal Ss to the frequency setting unit 627. The amplitude-phase calculation unit 665 also performs (I 2 +Q 2 ) 1 / 2 The amplitude of the output signal Sm is calculated by performing the following calculations. The amplitude phase calculation unit 665 outputs the calculated amplitude to the amplitude gain setting unit 626. The amplitude phase calculation unit 665 may, but is not limited to, a hardware demodulation circuit such as CORDIC (COordinate Rotation Digital Computer).

[0258] The frequency setting unit 627 has the function of determining the target frequency of the reference signal Ss. The frequency setting unit 627 then controls the frequency control signal Sf2 so that the frequency of the reference signal Ss becomes the target frequency, and outputs the frequency control signal Sf2 to the numerical control oscillator 616.

[0259] The numerically controlled oscillator 616 generates a reference signal Ss based on the frequency control signal Sf2. This corrects the frequency of the reference signal Ss.

[0260] The amplitude gain setting unit 626 has the function of determining the target amplitude of the drive signal Sd. The amplitude gain setting unit 626 then controls the amplification control signal Sam so that the amplitude of the drive signal Sd becomes the target amplitude, and outputs the amplification control signal Sam to the amplifier 614.

[0261] Amplifier 614 amplifies the amplitude of the drive signal Sd based on the amplification factor control signal Sam. This corrects the amplitude of the drive signal Sd.

[0262] 2.3. Effects of the Second Embodiment As described above, the laser interferometer 1a according to this embodiment comprises a laser light source 2, an optical modulator 12, a photodetector 10, a calculation unit 52, and a signal generation unit 61a. The laser light source 2 emits an outgoing light L1 (laser light). The optical modulator 12 includes a vibrating element 30 driven by a drive signal Sd, and uses the vibrating element 30 to superimpose a modulation signal onto the outgoing light L1. The photodetector 10 receives object light L3 (laser light including the sample signal) including a sample signal superimposed by reflection from the object 14, and reference light L2 (laser light including the modulation signal) including a modulation signal, and outputs a received signal. The calculation unit 52 performs calculations on the received signal based on a reference signal Ss. The signal generation unit 61a outputs the drive signal Sd and the reference signal Ss.

[0263] Furthermore, the calculation unit 52 includes a pre-processing unit 53, a demodulation unit 55, and a correction unit 62a. The pre-processing unit 53 performs pre-processing to extract frequency modulation components from the received signal and outputs a pre-processed signal S(t) that includes frequency modulation components. The demodulation unit 55 mixes the pre-processed signal S(t) with an orthogonal signal to obtain signals x and y (mixed signals), and then demodulates the sample signal from signals x and y. The correction unit 62a outputs a correction signal, which is an amplification control signal Sam and a frequency control signal Sf2, based on the output signal Sm that is output in response to the driving of the vibrating element 30. Then, the signal generation unit 61a corrects the drive signal Sd and the reference signal Ss based on the amplification control signal Sam and the frequency control signal Sf2.

[0264] In this configuration, even when disturbances such as temperature changes are applied, the frequency and amplitude of the drive signal Sd can be made to follow the changes in the mechanical resonance frequency and vibration amplitude of the vibrating element 30. This makes it possible to maintain a constant vibration frequency and amplitude of the vibrating element 30. As a result, a decrease in the S / N ratio of the modulated signal can be suppressed, and a decrease in the demodulation accuracy of the sample signal can be suppressed. This makes it possible to realize a laser interferometer 1a that can measure the displacement and velocity of the object 14 with high precision even when disturbances are applied. Furthermore, by using the signal generation unit 61a and the correction processing unit 62a, the vibrating element 30 can be driven near its mechanical resonance frequency, thereby reducing the power consumption of the laser interferometer 1a.

[0265] Furthermore, in this embodiment, the correction processing unit 62a acquires the phase difference between the output signal Sm and the reference signal Ss, and the amplitude of the output signal Sm, by quadrature detection. With quadrature detection, the phase difference and amplitude can be acquired instantaneously. Therefore, the correction processing can be performed in real time.

[0266] Furthermore, in this embodiment, a numerically controlled oscillator 616 is used as the oscillator in the signal generation unit 61. The numerically controlled oscillator 616 can generate periodic signals based on numerical values ​​read from the ROM table. Therefore, the numerically controlled oscillator 616 can generate high-precision reference signals Ss, Ss', and high-precision cosine wave signals cos(θ) without being affected by noise or the like. m (t)) and sinusoidal signal sin(θ m (t)) can be output. This improves the accuracy of pre-processing and demodulation processing in the calculation unit 52, and improves the demodulation accuracy of the sample signal. As a result, a laser interferometer 1a capable of measuring the displacement and velocity of the object 14 with higher accuracy can be realized.

[0267] 3. Modified Optical Systems Next, the first to fourth modified examples of the optical system 50 will be described.

[0268] Figure 18 is a schematic diagram showing the optical system 50A according to the first modified example. Figure 19 is a schematic diagram showing the optical system 50B according to the second modified example. Figure 20 is a schematic diagram showing the optical system 50C according to the third modified example. Figure 21 is a schematic diagram showing the optical system 50D according to the fourth modified example.

[0269] The following describes the first to fourth modified examples of the optical system 50. The following explanation will focus on the differences from the optical system 50 described above, omitting explanations of similar items. In Figures 18 to 21, the same reference numerals are used for items similar to those in Figure 2. Furthermore, some optical elements are omitted from the illustration in Figures 18 to 21.

[0270] The optical system 50A shown in Figure 18 is the same as the optical system 50 shown in Figure 2, except that the light incident on the photodetector 10, the optical modulator 12, and the object 14 is different. Specifically, in the optical system 50A shown in Figure 18, the emitted light L1 is incident on the photodetector 10 and the optical modulator 12. The optical modulator 12 shown in Figure 18 modulates the emitted light L1 to generate a reference light L2 that includes the modulated signal. This reference light L2 is then incident on the object 14. The object light L3, which includes the sample signal generated by the reflection of the reference light L2 from the object 14, is then incident on the photodetector 10. Therefore, the photodetector 10 shown in Figure 18 receives the object light L3 (laser light including the sample signal and the modulated signal) which includes the sample signal and the modulated signal, as well as the emitted light L1.

[0271] The optical system 50B shown in Figure 19 is the same as the optical system 50A shown in Figure 18, except that the arrangement of the photodetector 10, the optical modulator 12, and the object 14 is different.

[0272] In other words, the optical systems 50A and 50B according to the first and second modified examples include a laser light source 2, an optical modulator 12, and a photodetector 10. The laser light source 2 emits an outgoing light L1. The optical modulator 12 modulates the outgoing light L1 using a vibrating element to generate a reference light L2 including a modulated signal. The photodetector 10 receives object light L3 (laser light including a sample signal and a modulated signal) which includes a sample signal and a modulated signal generated when the reference light L2 is reflected by the object 14, as well as the outgoing light L1, and outputs a received signal.

[0273] The optical system 50C shown in Figure 20 is the same as the optical system 50A shown in Figure 18, except that the arrangement of the optical modulator 12 and the object 14 is different, and the light incident on the photodetector 10, optical modulator 12, and object 14 is different. Specifically, in the optical system 50C shown in Figure 20, the emitted light L1 is incident on the photodetector 10 and the object 14. The emitted light L1 is reflected by the object 14 to generate object light L3. This object light L3 is then incident on the optical modulator 12. The optical modulator 12 shown in Figure 20 modulates the object light L3 to generate reference light L2, which includes the sample signal and the modulated signal. This reference light L2 is incident on the photodetector 10. Therefore, the photodetector 10 shown in Figure 20 receives the reference light L2 (laser light including the sample signal and the modulated signal), which includes the sample signal and the modulated signal, as well as the emitted light L1.

[0274] The optical system 50D shown in Figure 21 is the same as the optical system 50C shown in Figure 20, except that the arrangement of the photodetector 10, the optical modulator 12, and the object 14 is different.

[0275] In other words, the laser interferometers equipped with the optical systems 50C and 50D according to the third and fourth modified examples include a laser light source 2, an optical modulator 12, and a photodetector 10. The laser light source 2 emits an outgoing light L1. The optical modulator 12 includes a vibrating element having a vibration component in a direction intersecting the incident plane of object light L3, which includes a sample signal generated when the outgoing light L1 is reflected by the object 14. The vibrating element modulates the object light L3 and generates a reference light L2 including the modulated signal. The photodetector 10 receives the reference light L2 (laser light including the sample signal and the modulated signal) including the sample signal and the modulated signal, as well as the outgoing light L1, and outputs a received signal.

[0276] The optical systems 50A, 50B, 50C, and 50D described above also have the same functions as the optical system 50 mentioned above.

[0277] Although the laser interferometer of the present invention has been described above based on the illustrated embodiment, the laser interferometer of the present invention is not limited to the above embodiment, and the configuration of each part can be replaced with any configuration having a similar function. In addition, the laser interferometer according to the above embodiment may have other arbitrary components added to it. Furthermore, the laser interferometer of the present invention may include both of the above embodiments. In addition, each functional part of the laser interferometer of the present invention may be divided into multiple elements, or multiple functional parts may be integrated into one.

[0278] The laser interferometer of the present invention can be applied not only to the displacement meters and speedometers mentioned above, but also to, for example, vibration meters, inclinometers, distance meters (measuring instruments), etc. Furthermore, applications of the laser interferometer of the present invention include optical comb interferometry technology that enables distance measurement, 3D imaging, spectroscopy, etc., and optical fiber gyroscopes that realize angular velocity sensors, angular acceleration sensors, etc.

[0279] Furthermore, two or more of the laser light source, optical modulator, and photodetector may be mounted on the same substrate. This facilitates miniaturization and weight reduction of the optical system, as well as improving ease of assembly.

[0280] Furthermore, although each of the above embodiments has a so-called Michelson-type interference optical system, the laser interferometer of the present invention is also applicable to other types of interference optical systems, such as those having a Mach-Zehnder-type interference optical system. [Explanation of symbols]

[0281] 1…Laser interferometer, 1a…Laser interferometer, 2…Laser light source, 3…Collimating lens, 4…Optical divider, 6…Half wave plate, 7…Quarter wave plate, 8…Quarter wave plate, 9…Analyzer, 10…Photodetector, 12…Optical modulator, 14…Object, 18…Optical path, 20…Optical path, 22…Optical path, 24…Optical path, 30…Vibrating element, 30A…Vibrating element, 30B…Vibrating element, 31…Substrate, 32…Groove, 33…Pad, 34…Diffraction grating, 35…Pad, 36…Vibration direction, 37…Mirror, 45…Circuit element, 50…Optical system, 50A…Optical system, 50B…Optical system, 50C…Optical system, 50D…Optics System, 51...Sensor head unit, 52...Calculation unit, 53...Pre-processing unit, 55...Demodulation processing unit, 57...Orthogonal signal generation unit, 59...Main unit, 61...Signal generation unit, 61a...Signal generation unit, 62...Correction processing unit, 62a...Correction processing unit, 64...Current shunt monitor, 120...Optical modulation oscillator, 301...First electrode, 302...Second electrode, 303...Diffraction grating mounting unit, 305...Piezoelectric substrate, 306...Comb-shaped electrode, 307...Ground electrode, 311...Front surface, 312...Back surface, 531...Current-voltage converter, 532...ADC, 533...ADC, 534...First bandpass filter, 535...Second bandpass Filter, 536…First delay adjuster, 538…Multiplier, 539…Third bandpass filter, 540…First AGC section, 541…Second AGC section, 542…Adder, 543…ADC, 551…Multiplier, 552…Multiplier, 553…Inverting amplifier, 555…First low-pass filter, 556…Second low-pass filter, 557…Divider, 558…Inverse tangent calculator, 559…Signal output section, 571…Fourth bandpass filter, 572…Hilbert transform filter, 573…Second delay adjuster, 574…Reference signal phase calculator, 577…Absolute value calculator, 578…Third low 579...Phase amount setting unit, 580...Adder, 581...Cosine generator, 582...Sine generator, 612...Voltage-controlled oscillator, 614...Amplifier, 615...DAC, 616...Numerically controlled oscillator, 621...Absolute value calculator, 622...Multiplier, 623...Multiplier, 624...4th low-pass filter, 625...5th low-pass filter, 626...Amplitude gain setting unit, 627...Frequency setting unit, 631...1st offset removal unit, 632...2nd offset removal unit, 641...Shunt resistor, 642...Operational amplifier, 643...ADC, 651...Cumulative adder, 652...Adder,653...First period signal generator, 654...Second period signal generator, 661...Multiplier, 662...Multiplier, 663...Sixth low-pass filter, 664...Seventh low-pass filter, 665...Amplitude phase calculation unit, K, -2s ...diffracted light, K -1s ...diffracted light, K 0s ...diffracted light, K 1s ...diffracted light, K 2s ...diffracted light, K i …Incoming light, L1…Outgoing light, L1a…First divided light, L1b…Second divided light, L2…Reference light, L3…Object light, N…Normal, P…Pitch, S1…First signal, S2…Second signal, S102…Work process, S104...process, S106...process, S108...process, S110...process, S112...process, S114...process, S116...process, S118...process, S120...process, cos(θ m (t))…cosine wave signal, sin(θ m (t)...Sine wave signal, S(t)...Pre-processed signal, Sa...Amplitude signal, Sam...Amplification control signal, Sd...Drive signal, Sf1...Frequency control signal, Sf2...Frequency control signal, Sm...Output signal, Ss...Reference signal, Ss'...Reference signal, jp1...Branch section, jp2...Branch section, ps1...First signal path, ps2...Second signal path, i...Signal i, r...Signal r, I...Signal I, Q...Signal Q, x...Signal x, y...Signal y, β...Incident angle, θ B ...blaze angle, θ S ...angle of inclination

Claims

1. A laser light source that emits laser light, An optical modulator comprising a vibrating element driven by a drive signal, which superimposes a modulation signal onto the laser light using the vibrating element, A photodetector receives the laser light including the sample signal superimposed by reflection from an object, and the laser light including the modulated signal, and outputs a received signal. A calculation unit that performs calculations on the received light signal based on a reference signal, A signal generation unit that outputs the drive signal and the reference signal, Equipped with, The aforementioned arithmetic unit, A preprocessing unit that performs preprocessing to extract frequency modulation components from the received light signal based on the reference signal and outputs a preprocessed signal including the frequency modulation components, A demodulation processing unit that demodulates the sample signal from the preprocessed signal based on the reference signal, A correction processing unit outputs a frequency control signal based on the output signal that is output in response to the driving of the vibration element, It has, The laser interferometer is characterized in that the signal generation unit corrects the frequency of the drive signal and the frequency of the reference signal based on the frequency control signal.

2. The laser interferometer according to claim 1, wherein the correction processing unit outputs the frequency control signal based on the phase difference between the output signal and the reference signal.

3. The laser interferometer according to claim 2, wherein the correction processing unit acquires the phase difference between the output signal and the reference signal by quadrature detection.

4. The correction processing unit outputs an amplification control signal based on the amplitude of the output signal. The laser interferometer according to any one of claims 1 to 3, wherein the signal generation unit corrects the amplitude of the drive signal based on the amplification control signal.

5. The laser interferometer according to claim 4, wherein the correction processing unit acquires the amplitude of the output signal by quadrature detection.

6. The signal generation unit, An oscillator that generates the aforementioned reference signal, An amplifier that adjusts the amplitude of the reference signal and outputs it as the drive signal, A laser interferometer according to any one of claims 1 to 5.

7. The laser interferometer according to claim 6, wherein the oscillator is a voltage-controlled oscillator that generates the reference signal by voltage control.

8. The laser interferometer according to claim 6, wherein the oscillator is a numerically controlled oscillator that generates the reference signal by numerical control.

9. The laser interferometer according to any one of claims 1 to 8, wherein the vibrating element is a quartz crystal oscillator.

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