Motor control device and power conversion system

Abstract

Provided is a motor control device that is able to reduce loss by controlling a current phase angle on the basis of a change feature amount of inductance.　This motor control device is connected to a power converter that converts power from direct current power into three-phase alternating current power and drives a motor with the three-phase alternating current power, the motor control device controlling the power conversion of the power converter, wherein the motor control device comprises: an inductance change detecting unit that calculates an inductance change feature amount that represents a change in the inductance of the motor in a first phase region that includes a phase in which the absolute value of any of the three-phase alternating currents is the maximum; and a current phase angle control unit that controls the current phase angle on the basis of the inductance change feature amount.

Description

Technical Field

[0001] The present invention relates to a motor control device and a power conversion system.

Background Art

[0002] The motor control device determines a three-phase AC voltage command that satisfies a torque command or the like and reduces losses (copper loss and iron loss) in a three-phase synchronous motor, and controls an inverter based on the three-phase AC voltage command.

[0003] As a technique for suppressing such losses, there is Patent Document 1. In Patent Document 1, in claim 1, "the carrier frequency adjustment unit adjusts the phase difference between the voltage command and the carrier so as to reduce the eddy current loss generated in the magnet of the rotor of the AC motor according to the d-axis current supplied to the AC motor and the rotational speed of the AC motor", in paragraph 0061, "the eddy current loss We is represented by the proportional relationship shown in the following formula (11)", in paragraph 0063, "formula (11) can be represented by replacing it with the proportional relationship shown in the following formula (12)", and in paragraph 0072, "the fixed triangular wave phase determination unit 1633 determines the value of the carrier phase difference Δθcarr based on the d-axis current sum sum calculated by the d-axis current sum calculation unit 1632. Here, the value of the carrier phase difference Δθcarr is determined so that the value of the d-axis current sum sum becomes minimum" is described.

[0004] Furthermore, paragraph 0076 of Patent Document 1 states that "In the voltage phase error calculation unit 163, the carrier phase difference Δθcarr is determined as described above, and the voltage phase error Δθv is calculated. This makes it possible to determine the voltage phase error Δθv such that the sum of d-axis currents sum is minimized according to the d-axis current Id and the motor rotation speed ωr. As a result, the carrier frequency fc can be set by changing the phase difference between the voltage command to the inverter 3 and the carrier wave used for pulse width modulation in order to reduce eddy current losses in the magnets of the rotor of the motor 2. As a result, the rise in magnet temperature Tmag can be suppressed, and irreversible demagnetization can be prevented." [Prior art documents] [Patent Documents]

[0005] [Patent Document 1] Japanese Patent Publication No. 2022-18168 [Overview of the project] [Problems that the invention aims to solve]

[0006] However, iron losses such as eddy current losses depend on the magnetic flux generated in the coil, and this magnetic flux can be reduced by increasing the d-axis current. Therefore, iron losses can sometimes be suppressed when the d-axis current is not minimized. Consequently, there was a challenge in appropriately selecting the d-axis current in order to minimize losses.

[0007] Furthermore, since iron loss also varies with inductance, there was a challenge in that, in order to suppress losses, it was necessary to detect changes in inductance and control the current according to the changes in inductance.

[0008] Therefore, the present invention aims to provide a motor control device and a power conversion system that can reduce losses by controlling the current phase angle based on the characteristic change in inductance. [Means for solving the problem]

[0009] To solve the above problems, the motor control device of the present invention is connected to a power converter that performs power conversion from DC power to three-phase AC power and drives a motor with the three-phase AC power, and controls the power conversion of the power converter, and comprises an inductance change detection unit that calculates an inductance change feature quantity representing the change in the inductance of the motor in a first phase region that includes the phase in which the absolute value of any of the three-phase AC currents is maximum, and a current phase angle control unit that controls the current phase angle based on the inductance change feature quantity.

[0010] Furthermore, the power conversion system of the present invention comprises, for example, the motor control device and the power converter. [Effects of the Invention]

[0011] According to the present invention, it is possible to provide a motor control device and a power conversion system that can reduce losses by controlling the current phase angle based on the characteristic change of inductance. [Brief explanation of the drawing]

[0012] [Figure 1] This is an overall configuration diagram of a motor drive system equipped with a motor control device. [Figure 2] This is a block diagram showing the functional configuration of a motor control device according to an embodiment. [Figure 3] This diagram illustrates the sampling period for the average current. [Figure 4] This diagram illustrates the sampling period for the average current. [Figure 5] This diagram shows an example of the relationship between the current I flowing through a coil and the magnetic flux Φ generated in the coil. [Figure 6] This diagram illustrates the sampling period of the current used to calculate the inductance change feature. [Figure 7] This diagram illustrates the sampling period of the current used to calculate the inductance change feature. [Figure 8]This diagram illustrates the relationship between carrier frequency and current ripple. [Figure 9] This diagram illustrates the relationship between three-phase alternating current and carrier frequency. [Figure 10] This diagram illustrates the relationship between loss and current phase angle. [Modes for carrying out the invention]

[0013] The present invention will now be described in detail with reference to the drawings. However, the present invention is not limited to the embodiments described below. These embodiments are merely illustrative, and the present invention can be implemented in various modified and improved forms based on the knowledge of those skilled in the art. Furthermore, in the drawings used in the following description, common devices and equipment are denoted by the same reference numerals, and descriptions of devices, equipment, and operations already described may be omitted.

[0014] Figure 1 is an overall configuration diagram of a motor drive system equipped with a motor control device. In Figure 1, the motor drive system 100 of this embodiment includes a motor control device 1, a motor 2, an inverter (power converter) 3, a rotational position detector 4, a high-voltage battery 5, a current detection unit 7, and a rotational position sensor 8.

[0015] The motor control device 1 receives the rotational position θr of the motor 2 from the rotational position detector 4. It also receives Iu, Iv, and Iw, representing the three-phase AC currents flowing through the motor 2, from the current detection unit 7, and a torque command T* from a higher-level control device (not shown). Based on this input information, the motor control device 1 generates gate signals to control the drive of the motor 2 and outputs them to the inverter 3. This controls the operation of the inverter 3 and the drive of the motor 2. Further details of the motor control device 1 will be explained later.

[0016] The inverter 3 includes an inverter circuit 31, a PWM signal drive circuit 32, and a smoothing capacitor 33. The PWM signal drive circuit 32 generates a PWM signal for controlling each switching element included in the inverter circuit 31 based on the gate signal input from the motor control device 1, and outputs it to the inverter circuit 31. The inverter circuit 31 has switching elements corresponding to the upper arms and lower arms of the U phase, V phase, and W phase, respectively. By controlling these switching elements according to the PWM signal input from the PWM signal drive circuit 32, the DC power supplied from the high-voltage battery 5 is converted into AC power and output to the motor 2. The smoothing capacitor 33 smoothes the DC power supplied from the high-voltage battery 5 to the inverter circuit 31.

[0017] The high-voltage battery 5 is a DC voltage source of the motor drive system 100, and outputs a power supply voltage Hvdc to the inverter 3. The power supply voltage Hvdc of the high-voltage battery 5 is converted into a variable-voltage, variable-frequency pulsed three-phase AC voltage by the inverter circuit 31 and the PWM signal drive circuit 32 of the inverter 3, and is applied to the motor 2 as the line voltage. Thereby, based on the DC power of the high-voltage battery 5, AC power is supplied from the inverter 3 to the motor 2. Note that the power supply voltage Hvdc of the high-voltage battery 5 varies according to its state of charge.

[0018] The motor 2 is a three-phase motor that is rotationally driven by the AC power supplied from the inverter 3, and has a stator and a rotor. In this embodiment, an example of using a permanent magnet synchronous motor as the motor 2 will be described, but it is not limited thereto. When the AC power input from the inverter 3 is applied to the three-phase coils Lu, Lv, and Lw provided in the stator, three-phase AC currents Iu, Iv, and Iw are conducted in the motor 2, and magnetic fluxes are generated in each coil. By generating attractive and repulsive forces between the magnetic fluxes of each coil and the magnet magnetic flux of the permanent magnet arranged on the rotor, torque is generated on the rotor, and the motor 2 is rotationally driven.

[0019] A rotational position sensor 8 is attached to the motor 2 to detect the rotational position θr of the rotor. The rotational position detector 4 calculates the rotational position θr from the input signal of the rotational position sensor 8. The result of the rotational position θr calculation by the rotational position detector 4 is input to the motor control device 1 and is used in the phase control of AC power, which is performed by the motor control device 1 generating a pulsed gate signal in accordance with the phase of the induced voltage of the motor 2.

[0020] Here, a resolver consisting of an iron core and windings is more preferable for the rotational position sensor 8, but there is no problem in applying magnetoresistive elements such as GMR (Giant Magneto Resistive effect) sensors or sensors using Hall elements. Any sensor that can measure the magnetic pole position of the rotor can be used as the rotational position sensor 8. In addition, the rotational position detector 4 may estimate the rotational position θr using the three-phase AC currents Iu, Iv, Iw flowing through the motor 2 or the three-phase AC voltages Vu, Vv, Vw applied to the motor 2 from the inverter 3, without using the input signal from the rotational position sensor 8.

[0021] A current detection unit 7 is positioned in the current path between the inverter 3 and the motor 2. The current detection unit 7 detects the three-phase AC currents Iu, Iv, and Iw (U-phase AC current Iu, V-phase AC current Iv, and W-phase AC current Iw) that energize the motor 2. For example, a Hall current sensor can be used for the current detection unit 7. The detection results of the three-phase AC currents Iu, Iv, and Iw by the current detection unit 7 are input to the motor control device 1 and used to generate the gate signal performed by the motor control device 1. Although Figure 1 shows an example in which the current detection unit 7 is composed of three current detectors, the AC current of the remaining one phase may be calculated from the fact that the sum of the three-phase AC currents Iu, Iv, and Iw is zero, using only two current detectors. Alternatively, the pulsed DC current flowing from the high-voltage battery 5 to the inverter 3 may be detected by a shunt resistor or the like inserted between the smoothing capacitor 33 and the inverter 3, and the three-phase AC currents Iu, Iv, and Iw may be determined based on this DC current and the three-phase AC voltages Vu, Vv, and Vw applied from the inverter 3 to the motor 2.

[0022] Next, we will describe the details of the motor control device 1. Figure 2 is a block diagram showing the functional configuration of the motor control device 1 according to the embodiment.

[0023] The motor control device 1 has the following functional blocks: a current command generation unit 10, a speed calculation unit 11, a current conversion unit 12, a current control unit 13, a carrier frequency adjustment unit 14, a carrier wave generation unit 15, a phase calculation unit 16, a three-phase voltage conversion unit 18, a gate signal generation unit 19, an inductance change detection unit 21, a current phase angle control unit 22, and a carrier frequency switching unit 23. Here, the current command generation unit 10, the current conversion unit 12, the current control unit 13, and the three-phase voltage conversion unit 18 function as a three-phase AC voltage command generation unit. The configuration of the three-phase AC voltage command generation unit is not limited to this and may include other functions. The motor control device 1 is configured, for example, by a microcomputer, and these functional blocks can be realized by executing a predetermined program in the microcomputer. Alternatively, some or all of these functional blocks may be realized using hardware circuits such as logic ICs or FPGAs (Field Programmable Gate Arrays).

[0024] The speed calculation unit 11 calculates the motor rotation speed ωr, which represents the rotational speed (rotational speed) of the motor 2, from the time change of the rotational position θr of the motor 2. Note that the motor rotation speed ωr may be expressed as either angular velocity (rad / s) or rotational speed (rpm). These values ​​may also be converted to and used interchangeably.

[0025] The carrier frequency adjustment unit 14 determines the carrier frequency fc, which represents the frequency of the carrier wave used to generate the gate signal, based on the rotation speed ωr obtained by the speed calculation unit 11. For example, the carrier frequency fc is determined so that the synchronous PWM carrier number Nc, which represents the number of carrier waves per period of the voltage waveform in synchronous PWM control, is a predetermined integer. Here, the synchronous PWM carrier number Nc can be set as a number that satisfies the condition Nc = 3 × (2 × n - 1) among multiples of 3. In this condition, n represents any natural number, and for example, n=1 (Nc=3), n=2 (Nc=9), n=3 (Nc=15), etc. can be selected. The synchronous PWM carrier number Nc can also be changed according to the rotation speed ωr. The carrier frequency adjustment unit 14 then determines fc based on the rotation speed ωr and the synchronous PWM carrier number Nc.

[0026] Furthermore, the carrier frequency adjustment unit 14 adjusts the carrier frequency fc according to the switching flag FlgFrqSw from the carrier frequency switching unit 23, which will be described later. For example, the carrier frequency adjustment unit 14 uses a coefficient K to set the carrier frequency fc = K·ωr·Nc / 2π, and an example in which K is changed according to the value of the switching flag FlgFrqSw will be described, but it is not limited to this. If the coefficient K when the switching flag FlgFrqSw=1 is made larger than the coefficient K when the switching flag FlgFrqSw=0, it is possible to switch the carrier frequency so that when the switching flag FlgFrqSw=1 it is a high-frequency carrier, and when the switching flag FlgFrqSw=0 it is a low-frequency carrier. Note that the method of switching the carrier frequency fc is not limited to this. In addition, the carrier frequency adjustment unit 14 may take the rotation position θr of the motor 2 as input, and enable carrier frequency switching only when the rotation position θr of the motor 2 is within a specific range. As a result, the carrier frequency switching unit 23 performs calculations only when the rotational position θr of the motor 2 is within a specific range, thereby reducing the processing load on the motor control device 1.

[0027] The carrier wave generation unit 15 generates carrier wave signals Sc for each of the three-phase AC voltage commands Vu*, Vv*, and Vw* based on the carrier frequency fc determined by the carrier wave frequency adjustment unit 14.

[0028] The phase calculation unit 16 calculates the estimated rotational position θe of the motor 2 based on the rotational position θr, the rotational speed ωr, and the carrier frequency fc using the following equations (1) to (3).

[0029] θe = θr + φv ···(1) φv=ωr·1.5Tc ···(2) Tc = 1 / fc ···(3) Here, φv represents the calculation delay compensation value for the voltage phase, and Tc represents the carrier wave period. The calculation delay compensation value φv is a value that compensates for the calculation delay of 1.5 control cycles that occurs between the time the rotation position detector 4 acquires the rotation position θr and the time the motor control device 1 outputs a gate signal to the inverter 3.

[0030] Figures 3 and 4 illustrate the sampling period for the average current value. In Figures 3 and 4, the vertical axis represents current, the horizontal axis represents time, and an example of the current waveform for one phase of a three-phase alternating current is shown. Actual currents experience current ripple and oscillate at a frequency close to the carrier frequency fc. For the sake of simplicity, the frequency of the current ripple is assumed to be equal to the carrier frequency fc below, but is not limited to this. As shown in Figure 3, the current conversion unit 12 acquires the average current value at sampling periods Ts determined based on the carrier frequency fc, for example, Ts = 1 / fc. If the carrier frequency fc is high, the average current value may be acquired at integer multiples of 1 / fc, for example, at Ts = 2 / fc as shown in Figure 4. A longer sampling period reduces the number of times the average value is acquired, thus reducing the processing load on the motor control device 1.

[0031] Figure 5 shows an example of the relationship between the current I flowing through a coil and the magnetic flux Φ generated in the coil. The horizontal axis represents the current I, and the vertical axis represents the magnetic flux Φ. When the current I is small, the relationship between the current I and the magnetic flux Φ is linear, and when the current I is large, the relationship between the current I and the magnetic flux Φ becomes nonlinear. Hereafter, in the curve representing the relationship between the current I and the magnetic flux Φ, the part where the relationship between the current I and the magnetic flux Φ is linear will be called the linear part, and the part where the relationship between the current I and the magnetic flux Φ is nonlinear will be called the nonlinear part.

[0032] Since the inductance L of a coil is defined as Φ = LI, if we calculate the inductance L at point P1 on the linear part and point P2 on the nonlinear part, the inductance L at point P1 is equal to the slope L0 of the line connecting the origin and P1, and the inductance L at point P2 is equal to the slope L2 of the line connecting the origin and P2. Thus, at any point on the linear part, the inductance L takes a constant value L0, but as shown in the inductance L2 in Figure 5, the inductance L at any point on the nonlinear part takes a different value from L0 and changes depending on the point. In other words, when the relationship between current I and magnetic flux Φ is linear, the inductance L takes a constant value, but when the current increases and the relationship between current I and magnetic flux Φ becomes nonlinear, the inductance L changes. This is because magnetic saturation occurs when the current increases. Also, the difference between the inductance L at a point on the nonlinear part and the inductance L0 in the linear part increases as the current increases. To detect such a change in inductance from inductance L0, it is desirable to directly calculate the inductance L from the magnetic flux Φ and current I, but there is a problem in that it is difficult to obtain the magnetic flux Φ. Therefore, in this embodiment, the tangent to the curve shown in Figure 5, i.e., dΦ / dI, is defined as the inductance change feature quantity ΔL (=dΦ / dI), and the change in inductance from inductance L0 is detected using the inductance change feature quantity ΔL.

[0033] The inductance change feature ΔL at point P1 on the linear portion coincides with the inductance L0 in the linear portion. Similarly, the inductance change feature ΔL is L0 at any point on the linear portion. Therefore, the inductance change feature ΔL at any point on the linear portion is constant at ΔL = L0. On the other hand, the inductance change feature ΔL at any point on the nonlinear portion, such as point P2, takes a different value from the inductance change feature ΔL = L0 in the linear portion, and approaches 0 as the current increases. This change characteristic of the inductance change feature ΔL, that is, it is constant at L0 in the linear portion and deviates from the inductance change feature L0 in the nonlinear portion as the current increases, is similar to the change characteristic of inductance L. Therefore, by monitoring the inductance change feature ΔL, changes in inductance can be detected. Furthermore, as will be described later, unlike inductance, the inductance change feature ΔL can be easily calculated without using magnetic flux Φ. Therefore, in this embodiment, the inductance change detection unit 21 calculates the inductance change feature quantity ΔL to detect the change in inductance.

[0034] As shown in Figure 2, the inductance change detection unit 21 receives the three-phase AC currents Iu, Iv, and Iw, and the previous values ​​Vu*_z, Vv*_z, and Vw*_z of the three-phase AC voltage command as input. The inductance change detection unit 21 then calculates the inductance change characteristic quantity ΔL for the phase with the largest absolute value of the current among the three-phase AC currents Iu, Iv, and Iw, using the formula V = ΔL × (dI / dt). Note that V = ΔL × (dI / dt) can be derived by applying the aforementioned dΦ / dI = ΔL to the equation V = dΦ / dt which holds true between the voltage V applied to the coil and the magnetic flux Φ generated in the coil.

[0035] Figures 6 and 7 illustrate the sampling period of the current used to calculate the inductance change feature. In Figures 6 and 7, the vertical axis represents current and the horizontal axis represents time, showing an example of the current waveform for one phase of a three-phase AC current. As shown in Figure 6, the inductance change detection unit 21 calculates dI / dt using the minimum value I_min and the maximum value I_max of the current, which are shown as square plots, from the actual current. Therefore, the inductance change detection unit 21 acquires the current value at the timing when the current ripple reaches its minimum and maximum values. For example, as mentioned above, when the average value of the current (circular plot) is acquired at each sampling period Ts, the inductance change detection unit 21 acquires the minimum value of the current ripple at time (4N+1) × Ts / 4 (N is an integer) and the maximum value of the current ripple at time (4N+3) × Ts / 4. In this case, the time interval between the minimum value I_min and the maximum value I_max of the current ripple acquired within the sampling period Ts of the average value of the current is approximately Ts / 2. Therefore, the inductance change detection unit 21 calculates dI / dt as (I_max-I_min) / (Ts / 2). The inductance change detection unit 21 then substitutes the calculated dI / dt and the previous value of the AC voltage command for the phase in which dI / dt was calculated into V=ΔL×(dI / dt) to calculate the inductance change feature quantity ΔL, and outputs the inductance change feature quantity ΔL to the current phase angle control unit 22.

[0036] In the above, the inductance change feature quantity ΔL was calculated for the phase with the largest absolute value among the currents Iu, Iv, and Iw of each phase, but this is not limited to that. For example, the inductance change feature quantity ΔL may be calculated for each phase regardless of the magnitude of the absolute values ​​of the currents Iu, Iv, and Iw of each phase, and the one inductance change feature quantity ΔL that deviates the most from L0 may be output to the current phase angle control unit 22.

[0037] When the carrier frequency fc is high, the inductance change detection unit 21 may alternately acquire the minimum value I_min and the maximum value I_max of the current ripple every 1 / fc, as shown in Figure 7. In this case, the inductance change detection unit 21 considers the acquired minimum value I_min of the current ripple (filled square plot) as the minimum value of the current ripple for the next period (dotted square plot) and calculates dI / dt.

[0038] As shown in Figure 2, the current phase angle control unit 22 determines the current phase angle θcrr corresponding to the inductance change feature quantity ΔL calculated by the inductance change detection unit 21. Here, the current phase angle θcrr is the leading phase angle from the q-axis of the current vector determined by the three-phase AC currents Iu, Iv, and Iw. It is desirable for the current phase angle control unit 22 to have pre-acquired and stored the current phase angle θcrr that minimizes loss for each inductance change feature quantity ΔL. For example, the current phase angle θcrr that minimizes loss can be pre-acquired for each inductance change feature quantity ΔL through experiments or analysis. Also, if the motor characteristics are clear, the current phase angle θcrr that minimizes loss can be calculated and pre-acquired for each inductance change feature quantity ΔL based on a mathematical formula. The current phase angle control unit 22 refers to the pre-acquired current phase angle θcrr for each inductance change feature quantity ΔL and outputs the current phase angle θcrr corresponding to the input inductance change feature quantity ΔL.

[0039] The carrier frequency switching unit 23 determines whether to switch the carrier frequency based on the current phase angle θcrr and the estimated rotation position θe of the motor 2. Therefore, the direction of the current vector is determined from the estimated rotation position θe of the motor 2 and the current phase angle θcrr, and the phase angles of the U-phase current, V-phase current, and W-phase current are determined. Based on the phase angles of the U-phase current, V-phase current, and W-phase current, the carrier frequency switching unit 23 outputs a switching flag FlgFrqSw as described later. For example, if two types of carrier frequencies, high frequency and low frequency, are prepared, the switching flag FlgFrqSw is set to 1 when the carrier frequency is set to high frequency, and to 0 when the carrier frequency is set to low frequency. Details of the operation of the carrier frequency switching unit 23 will be described later.

[0040] The current command generation unit 10 calculates the d-axis current command Id* and the q-axis current command Iq* based on the input torque command T*, power supply voltage Hvdc, and current phase angle θcrr. Here, for example, a pre-set current command map or a mathematical formula representing the relationship between the d-axis current Id, q-axis current Iq, and motor torque is used to determine the d-axis current command Id* and q-axis current command Iq* according to the torque command T*, power supply voltage Hvdc, and current phase angle θcrr. Alternatively, after determining the d-axis current command Id* and q-axis current command Iq* that satisfy the requirements of the torque command T* and power supply voltage Hvdc, the d-axis current command Id* and q-axis current command Iq* can be corrected using the current phase angle θcrr to determine the final d-axis current command Id* and q-axis current command Iq*.

[0041] The current conversion unit 12 performs a dq conversion on the three-phase AC currents Iu, Iv, and Iw detected by the current detection unit 7, based on the rotational position θr obtained by the rotational position detector 4, and calculates the d-axis current value Id and the q-axis current value Iq.

[0042] The current control unit 13 calculates the d-axis voltage command Vd* and q-axis voltage command Vq* corresponding to the torque command T*, based on the deviation between the d-axis current command Id* and q-axis current command Iq* output from the current command generation unit 10 and the d-axis current value Id and q-axis current value Iq output from the current conversion unit 12, so that these values ​​match. Here, for example, a control method such as PI control is used to determine the d-axis voltage command Vd* corresponding to the deviation between the d-axis current command Id* and the d-axis current value Id, and the q-axis voltage command Vq* corresponding to the deviation between the q-axis current command Iq* and the q-axis current value Iq.

[0043] The three-phase voltage conversion unit 18 calculates and outputs three-phase AC voltage commands Vu*, Vv*, and Vw* (U-phase voltage command Vu*, V-phase voltage command Vv*, and W-phase voltage command Vw*) based on the d-axis current command Id* and q-axis current command Iq* output from the current command generation unit 10 and the estimated rotation position θe calculated by the phase calculation unit 16.

[0044] The gate signal generation unit 19 uses the carrier signal Sc output from the carrier wave generation unit 15 to pulse-width modulate the three-phase AC voltage commands Vu*, Vv*, and Vw* output from the three-phase voltage conversion unit 18, respectively, and generates gate signals to control the operation of the inverter 3. Specifically, based on the comparison result between the three-phase AC voltage commands Vu*, Vv*, and Vw* output from the three-phase voltage conversion unit 18 and the carrier signal Sc output from the carrier wave generation unit 15, it generates pulsed voltages for each of the U, V, and W phases. Then, based on the generated pulsed voltages, it generates pulsed gate signals for the switching elements of each phase of the inverter 3. At this time, the gate signals Gup, Gvp, and Gwp of the upper arm of each phase are logically inverted to generate the gate signals Gun, Gvn, and Gwn of the lower arm. The gate signals generated by the gate signal generation unit 19 are output from the motor control device 1 to the PWM signal drive circuit 32 of the inverter 3, and converted into PWM signals by the PWM signal drive circuit 32. This controls the on / off state of each switching element in the inverter circuit 31, thereby adjusting the output voltage of the inverter 3.

[0045] Figure 8 illustrates the relationship between carrier frequency and current ripple. The vertical axis represents current, and the horizontal axis represents time. It extracts the portion of the current waveform for one phase of a three-phase AC current during a time width T1. The carrier frequency in Figure 8(b) is assumed to be higher than that in Figure 8(a). In Figure 8(a), two periods of current ripple are included within a time width T1. On the other hand, in Figure 8(b), four periods of current ripple are included within the same time width T1. The amount of current ripple included within the same time width T1 is greater in Figure 8(b) with a higher carrier frequency. Therefore, the current difference ΔI and time width ΔT used in the calculation of dI / dt are smaller in Figure 8(b) with a higher carrier frequency. As ΔI and ΔT become smaller, the accuracy of the dI / dt calculation also improves, so dI / dt can be calculated more accurately in Figure 8(b) with a higher carrier frequency. This also improves the accuracy of the inductance change feature ΔL calculated using dI / dt. Therefore, the motor control device according to this embodiment increases the carrier frequency in the phase region where a change in inductance occurs, that is, the phase region that includes the phase in which the absolute value of any of the three-phase AC currents is maximum. This makes it possible to detect changes in inductance with high accuracy.

[0046] Figure 9 illustrates the relationship between three-phase AC current and carrier frequency. Figure 9 shows a current waveform of the three-phase AC current superimposed with a gate signal, where the horizontal axis represents phase. In Figure 9, PR1 represents the first phase region, and PR2 represents the second phase region. The linewidth of the gate signal represents the carrier frequency; the lower the carrier frequency, the wider the linewidth of the gate signal. As shown in Figure 9, the carrier frequency adjustment unit 14 increases the carrier frequency in the first phase region PR1, which includes the phase where the absolute value of any of the three-phase AC currents is maximum. Within one period (2π) of the phase of the three-phase AC current, there are three phases where Iu, Iv, or Iw is maximum, and three phases where Iu, Iv, or Iw is minimum. Therefore, the first phase region PR1 is preset to include six phases where Iu, Iv, or Iw is maximum or minimum. Figure 9 shows an example where the first phase region PR1 is defined as six regions where the linewidth of the gate signal is narrow, i.e., near the six phases where Iu, Iv, or Iw is at its maximum or minimum. However, it is not limited to this example. The carrier frequency switching unit 23 sets the switching flag FlgFrqSw to 1 when the phase of the three-phase AC current calculated based on the current phase angle θcrr and the estimated rotation position θe of the motor 2 is within the first phase region PR1. The carrier frequency adjustment unit 14 then sets the carrier frequency to a higher frequency when the switching flag FlgFrqSw is 1. In this way, by increasing the carrier frequency in the phase region where inductance changes occur, changes in inductance can be detected with high accuracy.

[0047] Furthermore, as shown in Figure 9, the carrier frequency adjustment unit 14 lowers the carrier frequency in the second phase region PR2, which includes a phase where the absolute value of any of the three-phase AC currents is zero. Within one period (2π) of the phase of the three-phase AC current, there are six phases where Iu, Iv, or Iw is zero. Therefore, the second phase region PR2 is pre-set to include these six phases where Iu, Iv, or Iw is zero. Figure 9 shows an example where the second phase region PR2 is set to include the vicinity of the six phases where Iu, Iv, or Iw is zero, i.e., the six regions where the linewidth of the gate signal is thick, but it is not limited to this. The carrier frequency switching unit 23 sets the switching flag FlgFrqSw to 0 when the phase of the three-phase AC current calculated based on the current phase angle θcrr and the estimated rotation position θe of the motor 2 is within the second phase region PR2. The carrier frequency adjustment unit 14 then lowers the carrier frequency when the switching flag FlgFrqSw is 0. In this way, switching losses can be suppressed by lowering the carrier frequency in the phase region where changes in inductance are less likely to occur.

[0048] Figure 10 illustrates the relationship between loss and current phase angle. As shown in Figure 10, the current phase angle at which loss is minimized deviates from the current phase angle at which current is minimized. In this invention, instead of controlling the current phase angle to minimize current, loss can be minimized by calculating the inductance change characteristic quantity from the measured current value and controlling the current phase angle based on the calculated inductance change characteristic quantity. [Explanation of Symbols]

[0049] 1...Motor control device, 2...Motor, 3...Inverter, 4...Rotation position detector, 5...High-voltage battery, 7...Current detection unit, 8...Rotation position sensor, 10...Current command generation unit, 11...Speed ​​calculation unit, 12...Current conversion unit, 13...Current control unit, 14...Carrier frequency adjustment unit, 15...Carrier generation unit, 16...Phase calculation unit, 18...Three-phase voltage conversion unit, 19...Gate signal generation unit, 21...Inductance change detection unit, 22...Current phase angle control unit, 23...Carrier frequency switching unit, 31...Inverter circuit, 32...PWM signal drive circuit, 33...Smoothing capacitor, 34...Voltage detection unit, 100...Motor drive system

Claims

1. A motor control device connected to a power converter that performs power conversion from DC power to three-phase AC power and drives a motor with the three-phase AC power, and which controls the power conversion of the power converter, An inductance change detection unit calculates an inductance change feature quantity representing the change in the inductance of the motor in a first phase region that includes the phase in which the absolute value of any of the three-phase AC currents is maximized. The system includes a current phase angle control unit that controls the current phase angle based on the inductance change characteristic quantity. A motor control device characterized by the following features.

2. A motor control device according to claim 1, The unit further comprises a three-phase AC voltage command generation unit that generates a three-phase AC voltage command, The inductance change detection unit calculates the inductance change characteristic quantity based on the three-phase AC voltage command and the three-phase AC current of the motor. A motor control device characterized by the following features.

3. A motor control device according to claim 2, The inductance change detection unit calculates the inductance change characteristic quantity from the time derivative of the three-phase AC current and the three-phase AC voltage command. A motor control device characterized by the following features.

4. A motor control device according to claim 2, A gate signal generation unit that pulses width modulated the three-phase AC voltage command and generates a gate signal for controlling the power conversion of the power converter, The system includes a carrier frequency adjustment unit that adjusts the frequency of the carrier wave used for pulse width modulation based on the current phase angle and the rotation position of the motor, The carrier frequency adjustment unit raises the carrier frequency in the first phase region to a higher frequency than the carrier frequency in the second phase region, which includes a phase in which the absolute value of any of the three-phase alternating currents becomes zero. A motor control device characterized by the following features.

5. A motor control device according to any one of claims 1 to 4, The power converter comprises A power conversion system characterized by the following features.

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